Std 643-1980

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ANSI/IEEE Std 643-1980

An American National Standard IEEE Guide for Power-Line Carrier Applications

Sponsor

Power System Communications Committee of the IEEE Power Engineering Society Approved March 15, 1979

IEEE Standards Board Approved July 6, 1982

American National Standards Institute

© Copyright 1980 by The Institute of Electrical and Electronics Engineers, Inc 345 East 47th Street, New York, NY 10017, USA No part of this publication may be reproduced in any form, in an electronic retrieval system or otherwise, without the prior written permission of the publisher.

Approved March 15, 1979 IEEE Standards Board Joseph L. Koepfinger, Chair Irvin N. Howell, Jr, Vice Chair Ivan G. Easton, Secretary G. Y. R. Allen William E. Andrus C. N. Berglund Edward Chellotti Edward J. Cohen Warren H. Cook R. O. Duncan Jay Forster

Harold S. Goldberg Richard J. Gowen H. Mark Grove Loering M. Johnson Irving Kolodny W. R. Kruesi Leon Levy J. E. May

Donald T. Michael* R. L. Pritchard F. Rosa Ralph M. Showers J. W. Skooglund W. E. Vannah B. W. Whittington

* Member emeritus

Foreword (This Foreword is not a part of IEEE Std 643-1980, IEEE Guide for Power-Line Carrier Applications.)

In June 1954 the AIEE Committee Report, Guide to Application and Treatment of Channels for Power-Line Carrier, was published in the AIEE Transactions. A revision of the guide has not been prepared since that time. Substantial changes in available equipment and practices have occurred, and preparation of an up-to-date-guide is long overdue. Hopefully this guide will meet that need. The use of power line carrier (PLC) for the purposes of voice communication, protective relaying, supervisory operations, etc, is not unique to North America. While it has not been possible to describe where some differences may occur in application techniques and philosophies, the basic principles remain the same in all parts of the world. These differences, plus the related national and international standards, sometimes give rise to variations in equipment as compared to North American practice. Coordination with the related standards is achieved by maintaining contact with the appropriate organizations, such as IEC/TC33, 38, and 57; CIGRE 34, 35, and 36; CCITT; CCIR; CISPR; and ITU. Both IEC and CIGRE are published PLC guides emphasizing international practices. This guide was prepared by the Carrier Subcommittee of the Power System Communications Committee of the IEEE Power Engineering Society. At the time this guide was approved the members of the subcommittee were: H. I. Dobson, Chair J. W. Hagge, Vice Chair W. P. Bartley* S. J. Bogdanowicz L. H. Brookes T. J. Comerford

* Past Chairman

ii

J. C. Gambale J. Gohari P. R. Hanson D. T. Jones R. L. Linden

J. C. Lynn R. E. Ray C. Stillhard I. A. Whyte

CLAUSE

PAGE

1.

Scope ...................................................................................................................................................................1

2.

References ...........................................................................................................................................................2 2.1 Periodicals and Books ................................................................................................................................ 2 2.2 Standards Publications ............................................................................................................................... 3 2.3 Applicable Document in Preparation ......................................................................................................... 3

3.

Power-Line-Carrier Channels .............................................................................................................................3 3.1 Channel Description................................................................................................................................... 3 3.2 Characteristic Impedance ........................................................................................................................... 4 3.3 Power-Line Noise ...................................................................................................................................... 7 3.4 Channel Bandwidth and Modulation ....................................................................................................... 12 3.5 Channel Losses ........................................................................................................................................ 13 3.6 Modal Analysis ........................................................................................................................................ 21 3.7 Cross-Station Attenuation ........................................................................................................................ 27 3.8 Intrabundle Channels ............................................................................................................................... 29 3.9 Line Coupling Methods ........................................................................................................................... 33 3.10 Insulated Shield Wires ............................................................................................................................. 36

4.

Channel-Performance Evaluation .....................................................................................................................38 4.1 Factors Involved in Channel Performance............................................................................................... 38 4.2 Single-Function Individual-Channel Case ............................................................................................... 42 4.3 Multifunction, Multichannel Case ........................................................................................................... 47

5.

Coupling Components.......................................................................................................................................49 5.1 5.2 5.3 5.4 5.5

6.

Line Traps ................................................................................................................................................ 49 Coupling Capacitors................................................................................................................................. 58 Line Tuners and Bypasses........................................................................................................................ 59 Coaxial Cables and Tuning Leads ........................................................................................................... 69 Hybrids and Separation Filters................................................................................................................. 70

Frequency Selection ..........................................................................................................................................73 6.1 6.2 6.3 6.4 6.5 6.6 6.7

Factors Influencing Selection................................................................................................................... 73 Requirements of New Facilities ............................................................................................................... 74 Existing Frequencies ................................................................................................................................ 76 Frequency Planning.................................................................................................................................. 77 Line Coupling and Tuning ....................................................................................................................... 77 Noise and Line Attenuation ..................................................................................................................... 78 Power Cable Circuits ............................................................................................................................... 78

iii

CLAUSE 7.

Future Trends ....................................................................................................................................................78 7.1 7.2 7.3 7.4

8.

iv

PAGE

Introduction .............................................................................................................................................. 78 Electronic Equipment............................................................................................................................... 78 System Improvements.............................................................................................................................. 79 Applications ............................................................................................................................................. 79

Bibliography......................................................................................................................................................79

An American National Standard IEEE Guide for Power-Line Carrier Applications

1. Scope The purpose of this guide is to provide application information to users of carrier equipment as applied on power transmission lines. Information related to the expanding usage of carriers on distribution lines is not specifically covered. Detailed equipment design information is avoided as this is primarily the concern of equipment manufacturers. The Carrier Subcommittee has endeavored to provide curves and tables to present performance data. Complex formulas have been avoided to free the user from extensive calculations. The text provides British units of measurement followed by approximate metric equivalents in parentheses. It is hoped that this guide will be useful to the initial user of power line carriers as well as to the experienced application engineer. Detailed references have been provided in the guide for those who require more information. Material on power line carrier channel characteristics is presented in Section 3 along with discussions on intrabundle conductor systems and insulated shield wire systems. Section 4 provides procedures for the calculation of channel performance. Data for the calculations are drawn from various sections of the guide. Coupling components are discussed in Section 5. Major coupling components considered are line traps, coupling capacitors, line tuners, coaxial cables, hybrids, and filters. Frequency selection practices are discussed in Section 6, future trends in Section 7, and a bibliography is listed in Section 8. This guide supersedes the now outdated AIEE Committee Report, Guide to the Application and Treatment of Channels for Power-Line Carrier [1].1 There is unavoidable overlap in coverage between this IEEE guide and the PLC guides published by the IEC [2] and CIGRE [3]. However, all three scopes are sufficiently different that each has its own unique value.

1The

numbers in brackets correspond to the references listed in Section 2 of this guide.

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IEEE Std 643-1980

IEEE GUIDE FOR POWER-LINE

2. References 2.1 Periodicals and Books [1] AIEE COMMITTEE REPORT. Guide to Application and Treatment of Channels for Power-Line Carrier, AIEE Transactions on Power Apparatus and Systems, vol 73, Part III, June 1954, pp 417–436. [2] IEC COMMITTEE REPORT. Manual for the Planning of (SSB) Powerline Carrier Systems, in preparation, 1979. [3] CIGRE COMMITTEE REPORT. Guide on Power Line Carrier, in preparation, 1979. [4] Electrical Transmission and Distribution Reference Book, Westinghouse Electric Corporation, 1964. [5] PODSZECK, H. K. Carrier Communications on Power Lines, Berlin-Heidelberg-New York, Springer-Verlag, 1972, 4th ed. [6] JANISCHEWSKYJ, W. Characteristics of RI and TVI Sources, IEEE Tutorial Course Text, 76-CH1163-5-PWR, July 1976. [7] Tatsuo Udo, Mikio Kawal. Fault Generated Impulse Noise Voltage in a Transmission Line, IEEE Transactions on Power Apparatus and Systems, vol PAS-86, June 1967, pp 678–684. [8] LAFOREST, J. J., and WHEPLEY, E. A. Radio Influence Voltage/Radio Influence Field Ratios for 500-kV Lines, AIEE Conference Paper, 1962. [9] Power Line Carrier Application Guide, Lynchburg, VA, General Electric Company. [10] AVILA, C. F., and CORRY, A. F. Underground Transmission in the United States, IEEE Spectrum, vol 7, Mar 1970, pp 42–48. [11] PERZ, M. C. A Method of Analysis of Power Line Carrier Problems on Three-Phase Lines, IEEE Transactions on Power Apparatus and Systems, vol PAS-83, July 1964, pp 686–691. [12] HEDMAN, D. E. Propagation on Overhead Transmission Lines, Parts I and II, IEEE Transactions on Power Apparatus and Systems, vol PAS-84, Mar 1965, pp 200–211. [13] COMBS, E. E. New High-Pass Coupling Network for Power Line Carrier Operation: Design Considerations, IEEE Transactions on Power Apparatus and Systems, vol PAS-86, Dec 1967, pp 1522–1527. [14] BRESTKINA, E.E. Measurement and Calculation of Intrabundle HF Communication Paths, CIGRE Paper 35-03, presented at Paris, France, Aug 21-29, 1974. [15] LAUTENSACH, H., MARTIN, R. E., NOCKER, H., and SCHUMM, E. Intrabundle Carrier Communication Using the Insulated Bundle Conductors of a Power Line, CIGRE Paper 35-04, presented at Paris, France Aug 30-Sept 7, 1978. [16] HASLER, E. F, MARTIN, R. E., and PULLEN, F. D. Communication Systems Using Bundle Conductor Overhead Power Lines, IEEE Transactions on Power Apparatus and Systems, vol PAS-94, Mar/Apr 1975, pp 344– 349. [17] FARMER, G. E. The Use of Insulated Ground Wires on a Transmission Line for Communication Channels, IEEE Transactions on Power Apparatus and Systems, vol 82, Dec 1963, pp 884–891.

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[18] WOOD, T. D, JR. Application and Operating Experience of Carrier Communications over Insulated Static Wire, presented at the 1966 IEEE Winter Power Meeting, New York, NY, Jan 30-Feb 4, Paper 31-CP-66-41. [19] ALSLEBEN, E., FINCKH, B., and LAUTENSACH, H. Telecommunication by Means of Aerial Cables on Power Lines, CIGRE Paper 35-05, presented at Paris, France, Aug 28-Sept 6, 1972.

2.2 Standards Publications [20] ANSI/IEEE C37.90-1978, Relays and Relay Systems Associated with Electric Power Apparatus [21] ANSI C63.2-1979, Specifications for Electromagnetic Noise and Field-Strength Instrumentation, 10 kHz to 1 GHz [22] ANSI C93.1-1972, Requirements for Power Line Coupling Capacitors [23] ANSI C93.2-1978, Requirements for Power Line Coupling Capacitor Voltage Transformers [24] ANSI C93.3-1978, Requirements for Line Traps [25] ANSI/IEEE Std 430-1976, Procedures for Measurement of Radio Noise from Overhead Power Lines

2.3 Applicable Document in Preparation2

3. Power-Line-Carrier Channels 3.1 Channel Description A power-line-carrier (PLC) channel includes the signal path from the transmitting electronic equipment at one terminal, through its tuning equipment, over the power line, through the tuning equipment at the receiving end, and into the electronic equipment at the receiving terminal. In bidirectional applications a similar return path is provided. In planning a channel, consideration must be given to confining carrier signals to a desired path and to excluding unwanted signals from it. These functions can be achieved to a very practical extent through the use of line traps, coupling capacitors, and line tuners to provide high-impedance blocks in the undesired paths and low-impedance paths in the desired directions. A typical PLC channel, utilizing center phase-to-ground coupling, is shown in Fig 1. Note that this is a bidirectional application with both directions of transmission sharing a common path. A transmitter signal from one end (station A) passes through a line tuner which, in conjunction with the coupling capacitor, provides a low-impedance path between the transmitter and the line. The tuner contains an impedance matching transformer, so that the electronic equipment and the line each interface with the proper impedance. The line trap provides a high impedance at the channel frequencies to minimize loss of carrier signal power into the low impedance of the station bus. Similarly the other line trap at station B largely confines the signal power in its intended path to the electronic equipment at that end.

2When

the following document is completed, approved and published, it will become a part of this listing. Requirements for Line Traps, ANSI 93.3

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Figure 1—Diagram of a PLC Channel The electronic equipment at the terminal ends varies significantly, depending on the purpose for which the channel is intended. PLC channels are applied as line relaying or transferred trip channels. They are also used as single-sideband voice, data, supervisory, and control channels. Single-sideband voice channels and other functions are sometimes applied as combined channels. Included in the definition of PLC channels are other arrangements in which power cables, power cables in series with overhead power lines, or insulated ground wires are utilized as transmission paths. Various coupling methods are used for these combinations. Insulated shield wires are not normally recommended for protective relaying applications.

3.2 Characteristic Impedance The characteristic impedance [4], [5] of a transmission line (sometimes called surge impedance) is defined as the ratio between the voltage and the current of the traveling wave on a line of infinite length. This ratio of voltage to its corresponding current at any point in the line is a constant impedance, Z0. It is a function of the per-unit-length series resistance, series inductance, shunt capacitance, and shunt conductance of the line, and is independent of line length. The constant characteristic impedance of a two-wire line can be expressed as +

V Z 0 = ------ = + I

R + jωL --------------------G + jωC

(1)

where R L G C ω

= resistance per unit length, in ohms = inductance per unit length, in henrys = shunt conductance per unit length, in mhos (siemens) = shunt capacitance per unit length, in farads = 2πf, f being frequency in Hz

In practice at PLC frequencies the quantities jωL and jωC are large by comparison with R and G, so that the latter can be neglected, and the expression for characteristic impedance can be reduced to Z0 =

4

L ---C

(2)

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By applying conventional formulas for L and C to Eq 2, D D Z 0 = 276 log ---- × ---- ≥ 20 r r

(3)

is obtained, where D is the distance between conductors and r is their radius in the same units. Eq 3 expresses the characteristic impedance of a line consisting of two aerial wires. For a single aerial conductor at a height h above ground and radius r, the characteristic impedance is 2h Z 0 = 138 log -----r

(4)

Figure 2—Geometric-Mean Radius of Conductor Bundles, GMRc For bundled conductors the geometric mean radius (GMR) is used for r in Eqs 3 and 4. The GMR is defined in Fig 2 for three arrangements. In the case of a three-phase transmission line the calculation of the characteristic impedance is more involved and is further complicated by the use of bundled conductors. If a transmission line is terminated in its characteristic impedance, no energy will be reflected from the termination, and the sending-end behavior is the same as though the line were infinitely long. An impedance network of six impedances, as shown in Fig 3, is required to terminate a three-phase line in its characteristic impedance. Since a transmission line is seldom, if ever, terminated in its characteristic impedance network, the impedance seen by a set of coupling equipment connected to the transmission line, either phase to phase or phase to ground, will be reflected energy on the uncoupled phases affected by reflected energy on the uncoupled phases.

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Figure 3—Terminating Network for a Three-Phase Line Another and more practical value frequently called characteristic impedance is the value of the impedance to which the carrier coupling equipment is matched in order to obtain minimum mismatch and thus achieve maximum power transfer. This value of characteristic impedance is affected by the terminating impedance of the phase(s) not used in the coupling circuit. Measurements indicate that for phase-to-phase coupling, the (practical) characteristic impedance is not very dependent on the terminating impedance of the uncoupled phase. However, much larger differences occur for phase-to-ground coupling, as the terminating impedance of the uncoupled phases varies from an open circuit to a short circuit. As shown in Eq 3, the characteristic impedance is based on the radius of the conductors and the distance between conductors. In general both dimensions increase with higher voltages so that the ratio remains nearly the same. Therefore there is very little difference in the characteristic impedances of lines of various voltages as long as only one conductor is used for each phase. Lower values of characteristic impedance will exist on extra-high-voltage (EHV) transmission lines where bundled conductors are used with an effective radius that is much larger than the radius of a single conductor. Table 1 shows the range of values that can be expected from a wide variety of lines. Table 1—Range of Characteristic Impedances for PLC Circuits on Overhead Lines Transmission Line Conductor (Each Phase)

Characteristic Impedance Phase to Ground (ohms)

Phase to Phase (ohms)

Single wire

350500

650–800

Bundled (2-wire)

250–400

500–600

Bundled (4-wire)

200–350

420–500

The values of the characteristic impedance of power cables vary greatly from those for overhead lines, and there is also a large variation among different types of cables. In general there has not been much information published on power cables, such as the high-frequency characteristic impedance, and it may be required to perform measurements on the actual cable used for a particular circuit. Generally the characteristic impedance of a power cable will be between 10 Ω and 60 Ω.

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3.3 Power-Line Noise 3.3.1 Noise Types Noise voltage within the carrier frequency spectrum, which exists on every power line, must be considered in the design of PLC circuits to ensure that the required degree of reliability is provided. Different types of noise have different influences on various carrier facilities. This section describes the essential characteristics of noise and identifies those types that have the most serious effects on the reliability3 of carrier circuits (see ANSI/IEEE C37.90.1978, [20]. Power-line noise has generally been placed in two categories, called random noise and impulse noise. Random noise has a continuous frequency spectrum and has been compared to white noise, while impulse noise consists of sharp, separate impulses, produced by specific electrical discharges. Power-line noise is predominantly of the impulse type in which the pulse peaks are well above the average level and the space between pulses is occupied by random noise. For PLC applications, in lieu of the distinction between random noise and impulse noise, it is frequently useful to distinguish between noise that occurs as a continuous function of time and noise that occur as short. duration noise bursts which are nonperiodic. 3.3.2 Causes and Effects of Noise Random noise can be caused by thermal agitation in the power-line conductors, certain atmospheric and static discharges, and low-level corona discharges. Small discharges at many different points, although individually impulsive, together add up to random noise. Because of their high repetition frequency, many sources of periodically recurring impulses create noise that is essentially continuous as a function of both time and frequency. Sources of this type of noise include rotating machinery, power rectifiers, arc furnaces, and high-level corona discharges which are frequently produced during heavy rain. Lightning, line faults, circuit breaker operation, arcing of disconnect switches, and many lesser transients produced within a power station are examples of short-duration noise bursts and sharp, well-separated noise impulses which do not recur at periodic intervals. Intelligibility in telephone circuits, error rate in data circuits, dependability in protective relaying circuits, reproduction quality in facsimile circuits, and many other similar characteristics of PLC services depend largely on the signal-tonoise ratio determined by noise, which is present as a continuous function of time. Since these services are expected to operate continuously, design parameters for a required minimum signal-to-noise ratio will be determined by the highest level that this noise may be expected to attain. Bad weather increases the line noise level. Thunderstorms produce discharges which result in line noise as great as 10 times (20 dB higher than) the fair-weather figure. Corona generated on the surface of transmission line conductors during wet weather is sometimes the predominant source of worst case noise, especially on EHV lines. Severe corona noise may be as much as 30 dB above fair-weather levels. Corona is a luminous discharge due to ionization of the air surrounding a conductor when the voltage gradient at any point on its surface exceeds a critical value. Current impulses (or short trains of impulses) resulting from these discharges may be generated during each positive half-cycle of the power-frequency voltage, resulting in a basic burst repetition frequency of 180 Hz for a 60 Hz, three-phase system. Corona noise has an amplitude characteristic that decreases with increasing frequency. Its energy is continuous throughout the spectrum from frequencies below the 3The

definition of reliability as applied to relays and relay systems can also be of benefit in studies of PLC circuits. Reliability is the degree of certainty that a circuit (or system) will perform correctly whenever required and resist misoperation at all times. These two aspects of reliability are defined separately as dependability and security. Dependability in a system is the ability to operate properly when called upon to do so in the presence of any noise which exists at the time. Security is the ability to resist false response due to interference from any source which occurs while the system is in service.

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PLC band to beyond the high-frequency (HF) radio band. Corona noise generally becomes insignificant at frequencies in the mid-VHF region, except during ice and snow conditions. The mechanism for corona generation and the factors that influence it have been described in numerous publications [6]. Various transmission-line parameters affect line noise mainly because they affect corona. Constant and variable factors, such as voltage class, voltage fluctuations, line geometry (including conductor size and bundling arrangements), conductor surface condition, and weather, all affect conductor corona generation and, hence, noise. In comparison to corona noise on conductors, noise produced by corona on insulators, hardware, and station apparatus is usually of secondary concern at PLC frequencies. Similarly, gap-type noise generated in faulty hardware, cracked insulators, or improperly bonded high-voltage apparatus often causes VHF problems such as television interference when it occurs, but produces insignificant disturbance in the PLC band. Nonrecurrent high-level noise bursts are usually not of great concern in the dependability of most carrier circuits. For example, a loud noise burst in a telephone circuit, while bothersome, does not impair the intelligibility of the circuit following the burst. Similarly, isolated hits causing a number of momentary errors in a data circuit do not seriously impair its error rate on a day-in, day-out basis. The most detrimental effects which a high-level noise burst might produce are possible damage to circuit components and loss of security in circuit functions such as direct transferredtrip protective relaying. The noise characteristics on a transmission line during a fault are rather complex [7]. When a fault occurs, there is an initial burst of HF noise which usually decays within 1.5–4 ms. The actual peak value, lasting about 2 µs, may be well over 1000 V in amplitude. This is followed by a period of much lower noise level until the circuit breaker interrupts the fault current, at which time another short burst of HF noise occurs. A diminishing low-level noise exists during deionization of the fault arc which lasts for about 16 ms. Although lightning during adverse weather is sometimes troublesome, the source of noise that has the most detrimental influence on the security of protective relay circuits is the arc that occurs during the operation of a disconnect switch. This arc creates a burst of noise impulses that will last for at least a second. During most of the arcing time the impulse repetition frequency is 720 Hz for a 60 Hz, three-phase system. The frequency spectrum of this noise burst does not have a decreasing amplitude characteristic like that of corona and random noise. Instead it increases gradually in amplitude up to some frequency which is dependent on the electrical length of the switchyard buses and other apparatus connected to the arcing switch. Typically, maximum amplitude occurs at a frequency near 1 MHz. This high-intensity noise contains impulses frequently exceeding 1000V peak across a terminated coaxial cable. 3.3.3 Measuring Noise Noise measurements have been classified according to the responses of various measuring instrument detectors to different noise characteristics. These distinctions are useful because of their different effects on various kinds of carrier receivers. The peak value of power-line noise is the maximum voltage amplitude of recurring impulses. It is these impulses, for example, that affect trigger circuits such as are found in electronic switching devices. The quasi-peak value of noise is a reference level related to peak amplitude and to impulse repetition rate. It is measured by a detector circuit with a fast charging time and a relatively slow discharge time (typically 1 ms and 600 ms, respectively; see ANSI C63.2-1979, [21]. For pulses occurring at high repetition rates the quasi-peak value approaches the peak value. Quasi-peak noise is a measure of the masking effect of noise as a background for speech. The average value of noise is its average voltage over a finite period of time. It is defined as the area under the amplitude-time curve divided by the base length (time period). Average noise affects receiver-detector dc output for telegraphic functions, for continuous wave relaying, or for on-off pulse functions.

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The rms value of noise is the effective voltage of a reference sine wave which would have the same average power level as the noise being measured. RMS noise is of secondary importance in carrier equipment. Peak values of noise may be measured by certain peak-reading voltmeters, including standard noise meters. So-called memory voltmeters can retain the peak reading of the highest amplitude impulse (lasting 1 µs or longer) which occurs during any arbitrary measuring period. Peak values of noise may also be measured by using a cathode ray oscilloscope; however, this usually takes a fairly elaborate setup, including photographic equipment. Quasi-peak values may be measured only by noise meters that have appropriate detector characteristics. Although used extensively for measurements of noise affecting broadcast radio, quasi-peak measurements have not been used to any great extent in carrier applications. Average values of noise may be measured by selective carrier-frequency voltmeters of known bandwidth. Also, average noise can be derived from audio noise measurements made at the output of a single-sideband carrier receiver. Since noise energy is distributed throughout the frequency spectrum, measured values will be a function of the bandwidth of the measuring instrument. As might be expected, changing the bandwidth influences peak and quasipeak readings more than average and rms readings. Measured peak and quasi-peak values would be lowered by approximately 6 dB if the bandwidth were reduced 2:1. The measured values of average and rms noise would be lowered by 3 dB under the same conditions. Measured power-line noise often displays erratic frequency characteristics or severe standing wave patterns. The noise level on a power line is determined by both generation and propagation of noise energy. Propagation phenomena, such as attenuation, reflections, and absorptions, affect noise voltages in the same way as they affect desired carrier signals. The amplitude of power-line noise decreases with increasing frequency. A rule of thumb (very popular in the past) is that the noise amplitude varies inversely as the frequency. This would mean that the noise level drops 6 dB each time that the frequency is doubled. Recent investigations have clarified that the slope of the amplitude-frequency curve is not as steep as this rule would indicate. The curves of Fig 4 portray average noise in a 3 kHz bandwidth for a number of different line voltage classifications and also for both fair and adverse weather conditions. For the user who has no direct measurements of his own circuits, these values may be applied for design purposes.

Figure 4—Transmission-Line Noise Levels in a 3 kHz Bandwidth for Different Line Voltage Ranges. (a) 34.5-161 kV; (b) 230-345 kV; (c) 500 kV; (d) 765 kV. It is Estimated that the Values Given for Adverse Weather are not Exceeded more than 1% of the Time and Those for Fair Weather not more than 25% of the Time.

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3.3.4 Determination of PLC Noise by the RIV/RI Ratio [8] A useful relationship in both carrier work and radio noise theory is the ratio of radio influence voltage (RIV) to radio interference (IRI). RIV is the term generally applied to the radio frequency (RF) line noise potential which exists on the transmission-line conductor. RIV is usually expressed in microvolts (µV) or in decibles (dB) above 1 µV. RI refers to the electromagnetic noise field at ground level in the vicinity of the transmission line. RI is usually expressed in microvolts per meter (µV/m) or in dB above 1 µV/m. The value of this relationship to applied PLC is that if the RIV/RI ratio is known, an unknown power-line noise level can be determined without interrupting in-service carrier facilities by measuring the RI along the line and applying an appropriate conversion factor. Many utilities have established radio noise measuring programs for EHV lines from which RI data at 1 MHz are readily available. The proper method of measuring RI has been adequately described (see ANSI/IEEE Std 430-1976, [25]). The usual procedure is to take enough readings to establish a lateral profile of RI as a function of the horizontal distance from a point directly under the center of the transmission line to a point some 200 ft (60 m) or more from the outer phase conductor. RI measurements must be made using an appropriate instrument. Noise meters meeting the requirements of ANSI C63.2-1979 for the frequencies under consideration have an average bandwidth of about 5 kHz. The actual bandwidth of each individual noise meter and any dependence of the instrument's bandwidth on frequency will be covered in its instruction manual. These meters are capable of measuring peak, quasi-peak, and average values of noise. The average value, commonly called field intensity (FI), is the reading used for deriving RIV in the carrier band. Table 2 lists conversion factors (for a typical single-circuit power line) by which RIV on a 50 Ω impedance base and at a frequency of 150 kHz can be obtained in µV (or in dB above 1 µV) from RI measurements made at either 150 kHz or 1 MHz. For this purpose the preferred RI measuring location is 100 ft (30 m) horizontally from the outer phase of the line. Should the surrounding terrain prevent measurements at 100 ft (30 m) conversion factors for 0 ft and 50 ft (15 m) also appear in the table. However, unless absolutely necessary, measurements at zero reference (directly under the outer phase conductor) are not recommended. In broadcast band radio noise work, for RI comparison purposes, a reference horizontal distance of 50 ft (15 m) from the outer phase of the line is considered standard. Table 2—RI to RIV Conversion Factors

RI Measuring Frequency

Lateral Distance from Outer Conductor

Voltage Multiplier

Correction Factor (dB)

(ft)

(m)

1.0 MHz

100

30

240

47.5

1.0 MHz

50

15

83

38.5

1.0 MHz

0

0

45

33.0

150 kHz

100

30

60

35.5

150 kHz

50

15

21

26.5

150 kHz

0

0

11

21.0

If the RI data are in µV/m the voltage multiplier from Table 2 is used to provide RIV in µV. If the RI data are measurements taken in dB above 1 µV/m, the dB correction factor is added to provide RIV in dB above 1 µV. Table 3 lists the equivalent noise level in dBm for a number of values of RIV in µV. Values between those listed may be interpolated on a logarithmic voltage scale or computed directly. If the RIV level is in dB above 1 µV, it is only necessary to algebraically add −107 dB to that level to obtain dBm.

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Table 3—Conversion from µV across 50 Ω to dBm µv

dBm 100

−67.0

300

−57.5

500

−53.0

1000

−47.0

2000

−41.0

4000

−35.0

6000

−31.5

8000

−29.0

10 000

−27.0

20 000

−21.0

40 000

−15.0

60 000

−11.5

80 000

− 9.0

100 000

− 7.0

In either case the noise level in dBm is the one that would have been measured within the same bandwidth as that of the radio noise meter used to make the initial RI measurements. The equivalent noise level in any other bandwidth is readily obtainable by applying a correction factor (in dB) equal to 10 times the common logarithm of the bandwidth ratio. For example, if the noise meter bandwidth were 5 kHz and power-line noise within a 3 kHz bandwidth is desired, the correction factor would be approximately 2 dB. As an application example, assume that an RI measurement of 40 µV/m at I MHz is made at a distance of 100 ft (30 m) from the outer conductor. From Table 2 the multiplier is 240. Therefore the RIV across a 50 Ω carrier termination at 150 kHz is RIV = 240 × 40 = 9600 µV From Table 3 (by interpolation) this corresponds to a level of approximately –27.5 dBm. If the RI data has been in dB above 1 µV/m (20 log 40 = 32), the decibel correction factor of 47.5 dB from the table would be added: RIV = 32 + 47.5 = 79.5 dB above 1 µV This voltage level across 50 Ω converts to dBm as follows: RIV = 79.5 − 107 = −27.5 dBm If the noise level in a 3 kHz bandwidth is desired, but the noise meter bandwidth is 5 kHz, then the 2 dB correction previously mentioned should be deducted: RIV = (−27.5) − (2) = −29.5 dBm

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3.4 Channel Bandwidth and Modulation 3.4.1 General Bandwidth Requirements While a frequency band from 200 to 6100 Hz permits transmission of speech without loss of fidelity, rials range is seldom employed in practical voice communications systems. With a signal-to-average,noise ratio of 30 dB, word intelligibility (the percentage of a group of nonrelated words that can be recognized and understood after transmission over a communication channel) of greater than 95% can be obtained when a frequency band from 300 to 2200 Hz is used. This bandwidth affords even greater intelligibility when related words, such as those found in normal sentence structures, are used. If greater fidelity is required, such as the requirements of a commercial telephone channel, an expanded bandwidth of 300–3400 Hz can be used. Expansion beyond this bandwidth is normally reserved for highspeed data channels and is not normally found in PLC voice communication systems. While it is obvious that different functions require different bandwidths, the use of non-standard bandwidths on various PLC equipment tends to waste frequencies allocated to PLC because of the guard bands of different widths required to prevent crosstalk between channels. In recent years the PLC industry has tended to standardize the singlesideband (SSB) channel bandwidth to 4 kHz and vary the utilization of this band to fit whatever purpose is required by the user. Thus after subtraction of approximately 300 Hz from each end, a 3400 Hz band is available for data, voice, voice pins, tone relaying, and so on. The guard bands are standard and minimize the amount of spectrum not being used for active communication. Further, there is some tendency toward allocating all services into frequency slots that are multiples of 4 kHz to provide for more orderly growth and easier, frequency planning. Bandwidth requirements for other types of PLC channels, such as telemetry, blocking relaying, transferred trip, supervisory control, and low-speed data transmission, are varied, but generally require less bandwidth than voice. Successful 30 ms transferred-trip operations can be carried out over a channel of less than 250 Hz in bandwidth; however, wider bandwidths are used when greater speeds are required. When guard bands are included, this channel occupies a 500 Hz band. Commercial carrier relaying equipment of this bandwidth is commonly available. Frequencyshift carriers used with supervisory control, teletype, etc, may have even lower bandwidth requirements. 3.4.2 Modulation Types Amplitude modulation (AM) is obtained by varying the amplitude of a constant-frequency carrier in proportion to the amplitude of a modulating waveform, such as a voice or tone signal. The resultant modulated signal contains the original carrier frequency plus frequencies above and below the carrier, called sidebands. The frequencies in these sidebands differ from the carrier frequency by the frequencies in the modulating waveform. These sidebands (upper and lower) each occupy the same bandwidth as the modulating signal and contain the same intelligence. For 100% modulation the power in each sideband is equivalent to one fourth the power contained in the carrier. The total bandwidth required for AM transmission is twice the highest frequency of the modulating wave. Because of its bandwidth requirements and susceptibility to noise interference, AM is not used extensively today in PLC voice channels. The broader the receiver passband, the more energy it will pass, and the greater the effect of noise, except for coherent detection systems which have the same signal-to-noise ratio as SSB systems. AM is still used by some utilities for service voice channels. Keyed carrier (on-off AM) is commonly used for line relaying. Single-sideband modulation (SSB) is AM with one sideband removed. In addition the carrier is usually suppressed at the transmitter and reinserted at the receiver to further reduce unnecessary power transmission, since one sideband contains all the intelligence in the modulating signal. This allows the entire power output of the transmitter to be concentrated in one intelligence-bearing sideband. The SSB bandwidth requirement is one half that required for AM. Hence it has the advantage of providing more efficient spectrum utilization. Additional advantages of SSB over conventional AM channels are a higher signal-to-noise ratio and a reduced susceptibility to corona modulation.4

4Severe corona is believed to cause channel attenuation to vary in cadence with individual cycles of the power frequency transmission-line voltage.

This causes the amplitude of the received carrier signal to vary at a power frequency rate, hence the term corona modulation.

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Frequency modulation (FM) results from varying the frequency of a constant-amplitude carrier in proportion to the amplitude of a modulating waveform and at its rate. An FM wave can contain many sidebands, with equal numbers above and below the carrier. Its maximum amount of frequency shift is known as frequency deviation. The single-tone modulation index is the ratio of frequency deviation to tone frequency. The bandwidth required for FM transmission is dependent on the modulation index. With a modulation index greater than unity, the bandwidth is equal to twice the frequency deviation. As the modulation index decreases below unity, the bandwidth approximates twice the modulating frequency. Like AM, FM is not in extensive use today for voice transmission because of large bandwidth requirements. FM does provide a relatively good signal-to-noise ratio, which increases as the modulation index increases. It finds use in noisy environments. Frequency-shift keying (FSK) is a method by which a continuous carrier is transmitted, with the capability of being shifted to another frequency in order to initiate some function. Two-frequency and three-frequency equipments are available commercially. In two-frequency operation (sometimes called binary FSK) the transmitted carrier can be shifted in frequency to initiate a single function. In three-frequency operation the transmitted carrier is the center frequency and can be shifted to a lower frequency to initiate one function and to a higher frequency to initiate a second function. In other applications the FSK carrier can be continuously keyed in a manner to represent either analog or digital quantities. FSK is particularly adaptable to protective relaying, teletype, telemetering, supervisory control, and data communication channels. Since a carrier is always present in a properly operating channel, a loss of carrier condition can initiate a channel alarm. This results in an improvement over an on-off AM channel.

3.5 Channel Losses 3.5.1 Losses Loss is the reduction in power between two points on a communication channel. It is convenient in practice to express such losses in terms of dB, so that losses in various portions of a channel can be added to obtain the total loss. Loss in dB is defined as follows: P loss = 10 log -----1P2 V loss = 20 log -----1V2 I loss = 20 log ----1 I2

(5A)

(5B) (5C)

where P is power, V is voltage, and I is current, and the subscripts designate points of measurement in the channel. The latter two equations are valid only if the circuit impedances are equal at the points where the two currents (or voltages) are measured. The total attenuation of a PLC communication channel consists of the line attenuation (due to resistance, radiation, induction, etc), shunt attenuation (power flowing into paths other than the desired channel), bypass losses, and coupling losses. Mismatch losses will also be present where the terminal and coupling equipment impedances are not perfectly matched to the line.

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3.5.2 Attenuation in Overhead Lines At carrier frequencies a typical power line operates as a transmission line that is long compared to a wavelength of the carrier frequency. The performance of such a transmission line can be predicted reasonably well by applying modal theory (see 3.6), provided that all the necessary line parameters are known and a suitable computer program is available to the application engineer. For most application problems, however, adequate performance predictions can be made based on attenuation measurements made on typical lines. Line losses are primarily a function of the following parameters: Carrier frequency Type of line construction Line geometry Phase conductor size, material, surface, condition, etc Ground wire size, material, location, etc Method of coupling Type and location of transpositions Weather conditions Ground conductivity Insulator leakage Line losses increase with increasing carrier frequency due to increasing radiation loss, conductor loss, dielectric losses, and coupling to the ground wires, the ground, and the tower structure. These losses can be reduced by using smaller phase-wire spacing-to-height ratios and higher conductivity materials in the phase wires. The line voltage imposes limitations on phase-wire spacing and height, effective phase conductor cross section, and the number and separation of conductors per phase. HV lines exhibit a decreasing loss with increasing voltage. For economic reasons, conductors of more than one material are frequently used on power lines. A typical configuration is a core of steel wires with an outer covering of aluminum wires. The losses in such a conductor depend on the geometry and material of the wires making up the conductor. Since carrier currents are induced in the ground wires, ground-wire conductivity influences overall line losses. Lowloss ground wires are important near the coupling points on the line if single-phase-to-ground coupling is used, since large carrier currents must sometimes flow in them to establish the low-loss propagation mode without suffering substantial coupling loss. The loss in a line depends on the method of coupling used. Single-phase-to-ground (usually the center phase for horizontal structures) coupling is most frequently used. In situations requiring lower losses or higher probabilities of getting a signal through line faults, interphase or three-phase coupling methods are becoming increasingly popular. A single transposition in the middle of a long line can introduce a maximum of 6 dB loss. Usually the losses are not that great, even where several transpositions are involved or the line is short. Weather conditions can introduce significant loss, particularly where hoar frost is formed on the lines. The ratio of line attenuation with frost to fair-weather attenuation ranges from about 4:1 at 50 kHz to 5:1 at 250 kHz. Rain can double the line attenuation, particularly if it increases the conductivity of contaminants, such as heavy fly ash deposits, on the insulator surfaces. Poor ground conductivity can cause increased line losses. This is an important factor where single-phase-to-ground coupling is used and no ground wires (or ground wires of poor conductivity) are used at the coupling point. Poor ground conductivities occur most frequently in dry, sandy soils. A good station ground mat can substantilly reduce the effects of poor ground conductivity on coupling losses, provided it extends along the line for several spans. An estimate of the line losses can be obtained by performing the operations indicated in Fig 5, using additional data from Tables 4–6 [9]. 14

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Figure 5—Method of Determining Attenuation for an Overhead Transmission Line Table 4—Line-Voltage Multipliers Line Voltage (kV)

Fair Weather

Adverse Weather*

34.5

1.46

2.19

69

1.20

1.80

115

1.11

1.66

138

1.00

1.50

230

0.78

0.98

345

0.72

0.90

500

0.54

0.68

765

0.50

0.63

*Under certain severe frost conditions, extreme losses can occur.

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Table 5—Coupling Correction Factors

< 5 mi (8 km)*

Correction (dB) > 50 mi (80 km)*

150 mi (240 km)*

Mode 1 coupling

0

−2

—

Center phase to outer phase

0

0

—

0

1

—

Type of Coupling/ Shield Wires

Center phase to ground: A1 or Cu wire Steel wire †

No shield wire

1

4

—

8

8

—

0

—

13

Outer phase to ground: A1 or Cu wire Steel wire

1

—

15

No shield wire†

19

—

19

*Linear interpolation should be used to determine loss between limits of line length. †Subject to wide variations.

Table 6—Transposition Corrections* Correction (dB) Transposition

100 mi (160 km)†

1

0

6

2–4

0

8

5 or more

0

10

*Apply for 345 k V and higher †Linear interpolation should be used to determine loss between limits of line length.

3.5.3 Attenuation in Power Cables Carrier propagation losses per unit length in a power cable are substantially larger than those experienced with overhead phase wires. The specific loss encountered with a particular cable depends on its construction and the method of coupling. Individual power cable manufacturers may be able to supply attenuation and characteristic impedance information. The two power cable types generally used in underground applications are single-conductor self-contained cables and single- or three-conductor pipe-type cables [10]. It should be noted that experimental work is being done on cryogenic cables, but they are not yet being applied. Single-conductor self-contained cables utilize a hollow oil-filled conductor and an oil-impregnated paper insulation. An extruded metallic shield encases the insulation. The pipe-type cable utilizes a solid conductor with oil-impregnated paper insulation which is wrapped with synthetic tape layers and thin metallic tapes. One or three conductors are housed in a pipe which is filled with high-pressure oil (or gas). In some applications the fluid is pumped through the pipe to provide a more uniform temperature distribution and cooled to remove heat from the cable. Protective skid wires are spiraled around the conductor or conductors to protect them during insertion. The conductivity of these skid wires affects carrier attenuation. Characteristic impedances in power cables vary over a range from about 10 to 60 Ω.

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Representative values of attenuation on a power cable are shown in Fig 6 for a single-phase-to-ground input. Mutual coupling between phases in a three-conductor pipe-type power cable varies with frequency. At lower frequencies, attenuation with interphase coupling is substantially lower than with phase-to-ground coupling.

Figure 6—Phase-to-Ground Attenuation on a 138 kV Pipe-Type Cable 3.5.4 Attenuation at Discontinuities When a relatively short line is terminated in a load that is different from its characteristic impedance, the variation in line input impedance due to reflected energy can cause a transmitter to operate inefficiently. The effects of wide variations in input impedance can be minimized by a proper selection of frequencies provided that the distance to the mismatch and the magnitude of the mismatch termination are known. Any change in the line could lead to an input impedance change, however. In some cases a mismatch can be overcome by adjustment of the line tuner and changing the impedance-transformer ratio. Where a mismatch point is isolated from the input end of a line by substantial line loss, there is little change in input impedance due to reflected energy, and the only additional loss is due to the energy reflected from the mismatch. The loss due to an impedance discontinuity between a line of characteristic impedance Z0 and a load impedance Z1 is given in dB by the formula

Z0 + Z1 mismatch loss = 20 log -----------------2Z 0 Z 1

(6)

Sometimes a tap line is connected to the existing power line between two stations. If the line is short (low loss), its behavior as a stub at all carrier frequencies on the line should be investigated. If the stub is terminated in a high impedance and its length is equal to an odd number of quarter-wave-lengths at the carrier frequency, a low impedance will be reflected across the main line at the junction. For example, if the stub length is one quarter-wavelength at a given carrier frequency and the end of the stub is terminated in an open circuit, a short-circuit impedance will be reflected across the main line. This will result in an essentially infinite loss to the carrier-frequency signal on the line. Similarly, if a short stub is terminated in a low impedance at a carrier frequency and its length is an even number of quarter-wavelengths at that frequency, a low impedance will again be reflected across the junction. If a high loss is present on a line due to a stub, the situation may be eliminated by changing the carrier frequency. However, this cannot always be done, and the most effective method for minimizing a problem with a stub is usually to install a line trap at the input to the stub line.

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3.5.5 Attenuation in a Combined Overhead Line and Cable When an overhead line connects directly to a power cable, a substantial impedance mismatch occurs. This causes signal attenuation due to energy reflection. Typically the overhead line might have a characteristic impedance of 300 Ω (phase to ground), and the power cable might have a characteristic impedance of 20 Ω, resulting in an impedance mismatch ratio of 15:1. This is found to result in a 6.3 dB mismatch loss (Fig 7). As discussed in the previous section, the input impedance to the overhead line may vary substantially from its characteristic impedance if a mismatch is present and the overhead line has low attenuation. Particularly bad situations occur where overhead line lengths are multiples of one quarter-wavelength at the carrier frequency. This can readily be seen for the case of a quarter-wavelength line segment, since it acts as an impedance transformer in which the input impedance Zin is defined by the expression (for a lossless line)

2

Z Z in = -----0R

(7)

where Z0 is the characteristic impedance and R is the load. For Z0 = 300 Ω and R = 20 Ω, Zin = 4500 Ω, which is very high compared to the 300 Ω characteristic impedance. A half-wavelength line segment is equivalent to two quarterwavelength segments back to back, and the input impedance to the overhead line would equal the load resistance of 20 Ω, which is very low compared to 300 Ω. Other multiples of one quarter-wavelength result in either a high- or a lowinput impedance. Fig 7 is a graph of the mismatch loss as a function of the in-phase impedance ratio. Mismatch losses also occur when there is a phase angle associated with the mismatch ratio. For this reason, mismatch losses due to wide variations in the input impedance to the overhead line cannot be directly avoided, even by using frequencies that result in an odd number of one-eighth-wavelengths over the line length, because the lower mismatch ratio assumes a complex nature. However, for these in-between frequencies the resistive component of input impedance is usually within the range of adjustment of the impedance-matching transformer, and the reactive component can be offset by adjustment of the line tuner. Fairly efficient operation within a limited bandwidth is possible. The deleterious effect of reflections on the input impedance to an overhead line is greatly reduced if the line loss exceeds 5 dB.

Figure 7—Mismatch Loss The input impedance to the cable end of a combined system is also sensitive to the impedance mismatch at the junction of the cable and overhead line. Frequencies for which the power cable length corresponds to an odd multiple of an

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eighth-wavelength are to be preferred over those for which the length is any multiple of a quarter-wavelength. As with the overhead line, the effect of reflections can usually be neglected for cables having more than 5 dB attenuation. Although the effect of reflected energy on the input impedances to the overhead line and the power cable can be minimized by careful frequency selection and compromise adjustments of line-tuning equipment at both ends, the present application trend is to provide line trap isolation for the carrier circuit between the overhead line and the power cable. Coupling capacitors, line tuners, and impedance-matching transformers can then be used to provide the proper impedance to both the overhead line and the cable, and mismatch loss as well as frequency selection problems can be eliminated. 3.5.6 Coupling Losses 3.5.6.1 General Coupling losses are the losses that occur between an individual transmitter or receiver and the power line to which it is coupled. They include hybrid and after losses associated with the coupling circuits, tuner losses, coaxial cable losses, coupling capacitor losses, and shunt losses. 3.5.6.2 Carrier-Frequency Separation Equipment Several transmitters and receivers on a common coupling circuit require isolation elements to prevent undue signal absorption, interference, and intermodulation. These signal separation elements (typically hybrid units, trap units, series L/C units, and various filters) cause attenuation of the signal. The magnitude of loss varies widely, depending on the type, characteristics, and combination of the separation elements. Values must be obtained from the manufacturer for specific configurations and equipment. Refer to 5.5 for further discussion. The attenuation caused by these various elements when they are connected between a carrier transmitter (or receiver) and the line tuner is given in Table 7. 3.5.6.3 Coaxial Cable Transmitter/receiver equipment is normally connected by coaxial cable to a line tuner located near or in the base of the coupling capacitor. A representative type of coaxial cable, RG-8/U, has an attenuation that varies with frequency, as shown in Fig 8.

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Table 7—Attenuation of Separation Elements Separation Equipment Hybrid

Attenuation (dB) 3.5

Skewed hybrid Transmitter arm Receiver arm

0.5 12

L/C unit

1–3

Filter (low pass, band pass, or high pass)

1

Figure 8—Attenuation in RG-8/U Coaxial Cable 3.5.6.4 Line Tuner Losses A detailed discussion of various line tuners is presented in 5.3, including curves for determining losses in various classes of tuners. For quick estimates, values to be expected with a 0.003 µF coupling capacitor, a 300 Ω line impedance, and a frequency of 100 kHz are as follows:

Single-tuned resonant coupler

0.5 dB

Double-tuned resonant coupler

1.0 dB

Bandpass coupler

1.0 dB

3.5.6.5 Coupling Capacitor Losses A high dissipation factor in a coupling capacitor can produce an effective series resistance of several ohms at carrier frequencies. This resistance will cause I2R losses to the carrier channel. The impedance level associated with PLC on overhead transmission lines is high enough that this loss is usually only a fraction of 1 dB. It may consequently be either neglected or considered a part of the line tuner loss. On low-impedance power cables, however, series coupling capacitor loss may be several dB and represent a significant portion of the total coupling losses. The capacitance of a normal coupling capacitor is a function of temperature. Changes in capacitance caused by changes in temperature can result in detuning of the resonance between the coupling capacitor and its associated line tuner. The selectivity associated with most coupling circuits on overhead transmission lines is usually broad enough (except at very low frequencies) that detuning caused by temperature changes can be neglected. But PLC channels on 20

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low-impedance power cables may have extremely high selectivity. For these circuits, detuning during cold weather can cause severe increases in loss. One example of a coupling capacitor's dependence on temperature is illustrated in Fig 9. Different capacitors may have different characteristics. Variations in most modem coupling capacitors are not this severe. Manufacturers can provide data on new coupling capacitors at the time of purchase if requested to do so.

Figure 9—Typical Effect of Temperature on a Coupling Capacitor 3.5.6.6 Shunt Losses Shunt losses are the losses due to the transmission of the PLC down undesired paths. These losses normally occur at the points where signals are coupled to the power line and are usually small when a proper line trap is used. For design purposes, a shunt loss of 1 dB at each end of a line is a reasonable estimate. If the value of shunting impedance is known, shunt loss can be determined accurately by the formula

Z+Z shunt loss = 10 log ---------------S- ( dB ) ZS

(8)

where Z Zs

=nominal circuit impedance =shunting impedance

3.6 Modal Analysis 3.6.1 General The development of long EHV transmission lines with transpositions has generated new problems for the carrier communications engineer. In the past, when lines were short, general rules of thumb were applied to carrier applications. If the actual attenuation varied as much as 30–50% from the expected value it was not serious, since there was a large margin of transmitter power and receiver sensitivity. The advent of long EHV lines dictates a more accurate

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approach to PLC analysis, since it is not desirable to increase transmitter power, and the need still exists to predict the PLC channel performance in advance of line construction. Modal analysis is a mathematical tool that provides a more accurate analysis of carrier propagation, much like symmetrical components provide a method for analysis of unbalanced three-phase systems. Modal analysis is not only a mathematical tool, but it can also be readily identified in the physical world. The following discussion of modal analysis will center around the physical aspects. Excellent treatments of the analytical aspects are available [11], [12]. 3.6.2 Description The propagation of electromagnetic energy on a single-conductor line is given by the following relation, provided no reflections exist: V x = V sε

– γx

(8A)

V I x = -----xZ0

(8B)

where Vs Zo x Vx, Ix γ

=sending-end voltage =characteristic impedance =distance from sending end, in miles or kilometers (km) =voltage and current at distance x from sending end =propagation constant

The propagation constant is composed of two quantities: γ = α + jβ

(8C)

where α is an attenuation constant in nepers per mile (or nepers per km) and β is a phase constant in radians per mile (or radians per km). (1 neper = 8.686 dB.) The propagation constant is dependent on the physical properties of the conductor used plus its geometry with respect to ground. It has been observed that if carrier energy is applied to a single conductor of a multiconductor line, then the propagation along the line does not follow the above equation. Instead, the propagation of energy on the line depends on the number of conductors and usually involves all of them. Analysis of a multiconductor line shows that several modes of energy propagation may take place simultaneously. It can also be shown that the number of natural modes of propagation on a multiconductor line is equal to the number of conductors involved in the propagation of energy. The analysis of the multiconductor line is accomplished using a matrix equation: [ V ] = [ V s ]ε

–[ γ m ] x

(9)

where m is an arbitrary integer to identify the modes. The solution of the above matrix equation will yield the voltage and current relationship of the modes. Each mode has its own propagation constant and characteristic impedance, and each mode propagates in a manner that is independent of the other modes. The voltage and current at any location on any one conductor is the vector sum of the individual mode voltages and currents existing on that conductor at the particular distance x from the transmitter.

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The remainder of the discussion will deal with the specific case of a single-circuit three-phase line of fiat construction. This is a common type of construction for EHV circuits as well as many lower voltage transmission lines. A typical example is shown in Fig 10. The assumption is also made that the shield wires are grounded at each tower and are at constant potential along their length. Thus they do not generate any transmission modes. Based on this assumption there are three natural modes of propagation which will be labeled mode 1, mode 2, and mode 3. Each mode has its propagation constant (γ1, γ2, and γ3) and characteristic impedance (Z01, Z02, and Z03. Other transmission-line configurations, such as vertical or triangular ac lines, dc lines, or power cables, will each have its own set of modal parameters. The modal properties described in the following paragraphs are not appropriate for any except the general arrangement shown in Fig 10.

Figure 10—Typical Single-Line Construction Mode 15 is the least attenuated of the three modes and makes long-distance carrier communications possible. It normally has current flowing out the two outer phases and returning via the center phase. Mode 1 attenuation is not only low, but it is also reasonably independent of frequency throughout the PLC range. Mode 2 has its current flowing out on one outside phase and returning on the other outside phase. No mode 2 current exists in the center phase. Mode 2 losses are greater than those of mode 1, and its losses are more dependent on frequency. Mode 3 propagates current almost equally on all three phases and has a ground return. This mode has such a high rate of attenuation that it may be neglected beyond a short distance (approximately 10 mi or 16 km) from the transmitter. The vector relationships of modal currents (or voltages) in the three phase conductors are shown in Fig 11. Current (and voltage) distributions among the phase conductors are given in Table 8. These quantities are normalized to the phase A quantities. The factors p and q depend on the line under study. The factor p can range from about −1.6 to −2.0, and q will have a range of about 1.1 to 1.3.

Figure 11—Modal Vector Relationships 5In some recent literature the lowest loss mode has been identified as mode 3. The definition given here, which identifies mode 1 as the lowest loss

mode, is in agreement with most earlier IEEE publications and with current practice in most parts of the world.

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Table 8—Phase Distribution of Modal Current and Voltage Phase

Mode 1

Mode 2

Mode 3

A

1

1

1

B

p

0

q

C

1

−1

1

As stated before, each mode has an independent propagation constant. Based on several examples of experimental data taken for lines from 345 to 765 kV, the general range for the values of attenuation a and the relative phase velocity (which is dependent upon the phase constant β) are shown in Table 9. Table 9—Atteuation and Relative Phase Velocity for Modes 1, 2, and 3 Quantity

Mode 1

Mode 2

Mode 3

in dB/mi

0.01–0.03

0.09–0.10

—

in dB/km

0.006–0.018

0.0054–0.06

—

in dB/mi

—

—

1.5–3.0

in dB/km

—

—

0.9–1.8

in dB/mi

0.07–0.09

0.4–0.5

—

in dB/km

0.042–0.054

0.24–0.3

—

1.0

0.98–0.995

0.9

Attenuation: at 30 kHz

at 100 kHz

at 300 kHz

Relative phase velocity

The attenuation constant ranges for modes 1 and 2 are given for frequencies of 30 kHz and 300 kHz. The attenuation can be expected to vary in almost a linear fashion between these two frequencies. The phase constant is represented in terms of the relative propagation velocity with respect to the velocity of mode 1, which is nearly the speed of light in free space. These general ranges can be used to calculate line attenuation, keeping in mind that the modal quantities add vectorally to produce the phase quantities. 3.6.3 Coupling to the Power Line When a carrier transmitter is coupled to the power line, it is usually done using single-phase-to-ground or a form of phase-to-phase coupling. All of the generally used methods of coupling generate different portions of mode 1, mode 2, and mode 3 power. Since mode 1 is the least attenuated, then it is desirable to generate as much mode 1 as possible. At the coupling terminal, the phase voltages and currents are set by the coupling configuration, and are known. The mode voltages generated must satisfy these boundary conditions. They can be calculated by solving the set of equations:

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V a = V a1 + V a2 + V a3

(11A)

V b = pV a1 + qV a3

(11B)

V c = V a1 – V a2 + V a3

(11C)

If currents are used, I is substituted for V in the preceding equations. When the magnitudes of Va1, Va2 and Va3 have been calculated the modal power at the transmitter can be found. PT is equal to the total transmitter power less the losses in the coupling equipment: PT = P1 + P2 + P3

(12)

where P1, P2, and P3 are the powers of modes 1, 2, and 3, respectively:

2

2

( 1 + p + 1 )V a1 P 1 = ------------------------------------Z 01

(13A)

2 2V a2

P 2 = -----------Z 02

(13B) 2

2 1 )V a3

(1 + q + P 3 = -------------------------------------Z 03

(13C)

Now that the modal powers are known, the modal coupling efficiency can be calculated:

P η 1 = -----1PT P η 2 = -----2PT P η 3 = -----3PT

(14A)

(14B)

(14C)

where η1, η2, and η3 are the modal coupling efficiencies. It is desirable to make η1 as close to unity as possible and η2 and η3 as close to zero as possible. The loss of transmitter power to modes 2 and 3, in dB, is α 1 = – 10 log η 1

(15)

This loss may or may not be a real loss to the receiver, depending on the line length. Table 10 lists values of α1 as conversion losses for several common coupling methods. The table is based on the assumption that the line is sufficiently long that the received power in modes 2 and 3 is negligible compared to the power in mode 1. Refer to 3.5 for practical guidelines.

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Table 10—Mode 1 Single-End Conversion Losses

Type of Coupling Mode 1

Conversion Loss (dB) 0

Center phase to outer phase (conductors driven electrically out of phase)

1.2

Center phase to ground

1.8

Outer phase to ground

7.0

Double phase to ground (two outer conductors driven electrically in phase)

4.8

The modal coupling efficiency is often expressed in dB and means the same as the modal conversion loss, except that the sign is reversed. For example, center-phase-to-ground coupling, which has a modal conversion loss of 1.8 dB, is said to have a coupling efficiency of -1.8 dB. 3.6.4 Transpositions in the Line On a long transmission line of fiat construction, the 60 Hz impedance between phases is unbalanced. In order to aid in the removal of unbalanced currents, transpositions are placed at strategic points along the line. These transpositions affect not only power frequencies but also carrier frequencies. At carrier frequencies transpositions are assumed to be transparent; that is, they do not reflect any of the incident RF energy. They do, however, act as mode converters. The value of the modal quantities at the input of the transposition will be different from the modal quantities at the output. As an example, even if only mode 1 signals are present at the input of the transposition, all three modes of propagation are present at the output. Most of the output energy will be in modes 1 and 2. However, the mode 1 quantity at the output is 6 dB below its input value. If the generated mode 2 signal is completely attenuated before reaching the receiver or the next transposition, then 6 dB is the loss represented by the transposition. The figure of 6 dB loss is a maximum, and this simple example does not usually hold in practice. This is because quantities other than mode 1 are usually present at the input to the transposition, and the generated mode 2 is not completely attenuated by the time the receiver is reached. The exact procedure for handling transpositions in attenuation calculations is as follows: 1) 2) 3) 4)

Calculate the modal quantities at the input to the transposition, both phase and magnitude. Add the modal quantities vectorially to get phase quantities. Transpose the phase quantities per line transposition. Reconvert phase quantities into modal quantities and continue to the receiving end or next transposition. The conversion to modes is the same method as that described for coupling conversion (Eqs 11).

3.6.5 Receiving Terminal At the receiving terminal the modal components are converted into phase voltages and currents. These voltages and currents then become the received signals, provided the line is properly terminated. Proper termination for all modes of propagation would require a network of six impedances (Fig 3). In most examples of actual practice, each coupled phase is terminated with an impedance to ground. As a result, some reflections will occur, but these usually do not significantly degrade the desired signals.

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3.7 Cross-Station Attenuation 3.7.1 Carrier Runaround Unintentional propagation of carrier-frequency signals through or around a power station (sometimes called carrier runaround) is an undesirable but inevitable phenomenon that takes place wherever PLC is applied. The reduction in magnitude of these signals as they pass through a station is called cross-station attenuation. Because of the importance of minimizing random propagation of interfering signals, a high value of cross-station attenuation is very desirable. It is also desirable for more specific reasons such as the application of carrier-frequency repeaters. 3.7.2 Cross-Coupling Mechanisms 3.7.2.1 General Cross coupling of carrier signals through a station takes place through a network consisting of series-connected line traps and the shunting impedance of the power bus. A substantial amount of carrier energy also passes through untrapped phases of the power line on which the carrier signal is applied. Another path frequently exists by virture of the mutual coupling between parallel lines entering a station. 3.7.2.2 Line Trap Network The passage of signals through this network (Fig 12) will depend on the series impedance of each line trap and the shunting impedance of the station bus. The nature of trap impedance is discussed in 5.1. The bus impedance consists of many individual impedances in parallel. Power transformers, potential transformers, and shunt reactors are all selfresonant at low frequencies and exhibit capacitive impedance characteristics at carrier frequencies. Every other item of power apparatus connected at the bus junction, including bus conductors, bushings, circuit breakers, and current transformers, has a natural capacitance with respect to ground. The impedance of power lines terminating on the bus will vary over a wide range because of standing wave patterns and the nonuniform placement and tuning of line traps. The resultant net bus impedance usually has only capacitive and resistive components. Its magnitude will fluctuate as a function of frequency but will in general be low (100 Ω or less), except in very small stations.

Figure 12—Cross-Coupling Path through Line Trap Network 3.7.2.3 Uncoupled Phases Although a carrier-frequency signal may be coupled to only one phase of a three-phase power line, it is never confined to such a simple path. At distances of 10 mi (16 km) or more from the point of origin, signal strengths of the same order of magnitude may exist on all three phases. Since it is routine practice to trap only the coupled phase, carrier energy on the other phases is free to pass through a receiving station bus, hindered only by the associated impedance discontinuity and a reduction in signal strength caused by the dispersal of energy among various other lines and shunt paths. At a receiving location, energy levels on unused phases are frequently found to be as high as on the coupled phase. The effectiveness of a line trap here in reducing the signal level at stations beyond the receiving station is, at most, only about 3 dB. The line trap, however, is very effective in protecting the receiver itself from other sources of interference. Copyright © 1998 IEEE All Rights Reserved

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3.7.2.4 Parallel Lines Frequently several transmission lines will enter a station over the same or adjacent rights-of-way, with parallels extending over several spans. A considerable amount of carrier energy may be exchanged, particularly where two lines are constructed as a double circuit on common towers. 3.7.3 Cross-Station Attenuation Measurement Cross-station attenuation of a locally originated signal is almost always greater than cross-station attenuation of a signal originated at a distant location. For measuring with a local signal, the procedure is to connect a carrierfrequency signal generator into the line tuner on one line and measure the received signal level at the terminated line tuner on the opposite line. The difference between the inserted signal level and the received level is called near-end cross-station attenuation. To measure far-end cross-station attenuation it is necessary to insert the carrier-frequency signal at the far end of the first line. Both lines under consideration must have properly terminated line tuners at the station being tested. The difference in the received signal level measured on the two respective terminations is far-end cross-station attenuation. Near-end and far-end values are compared in Fig 13 at a substation that does not involve parallel lines.

Figure 13—Example of Near-End and Far-End Cross-Station Attenuation Near-end cross-station attenuation has most significance at transmitter locations and at stations where simple amplifier-type repeaters are applied. It is a measure of the influence that a local transmitter might have on other carrier facilities in the same station. The permissible gain of a repeater may be determined by deducting the desired singingpoint margin from the measured value of near-end cross-station attenuation. Line traps in the coupled phase are normally very effective in providing high values of near-end cross-station attenuation. Far-end cross-station attenuation is more significant at receiver locations and at frequency-frogging6 repeater installations. Frequency-frogging repeater operation generally requires a minimum value of far-end cross-station attenuation of about 20 dB throughout its transmitting and receiving bands for satisfactory operation. Far-end cross-

6Frequency frogging is a technique based on telephone-line carrier practice wherein transmitting and receiving frequencies are exchanged. Identical

frequencies are transmitted in both directions from a repeater and identical frequencies are also received from each direction. It is rarely used in modern PLC systems.

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station attenuation is also a measure of the influence of a received carrier signal on carrier facilities coupled to other lines in the receiver station. 3.7.4 Methods for Increasing Cross-Station Attenuation In most carrier applications the standard use of line traps with each carrier coupling circuit provides adequate rejection to the passage of interfering signals. In applications where the value of cross-station attenuation is not adequate, special measures shall be considered. When the attenuation to a carrier signal's passage through the line trap network exceeds a certain magnitude, crosscoupling through other paths (usually the unused phases) becomes predominant. The exact level where this cross-over takes place is difficult to establish. Special treatment of the coupled phase beyond normal practice without also treating the uncoupled phases is therefore not usually beneficial. Trapping of uncoupled phases on one or more lines can produce substantial improvements. Where it is not considered practical to install traps in series with unused phases, it is sometimes feasible to install coupling capacitors connected between the station bus and ground. The presence of the capacitor lowers the impedance of the bus and increases attenuation to the passage of carrier signals. This bus treatment may be advantageously applied to both the coupled and the uncoupled phases. Lowering the station bus impedance by the addition of substantial shunt capacitance has also been recommended for other aspects of carrier behavior such as the efficiency of wide-band coupling filters [13]. Coupling-capacitor voltage transformers (CCVT) are sometimes connected to aU three phases of a line or a station bus (to supply potential for relaying and other 60 Hz functions). In this case carrier performance is improved at no additional cost. When a coupling capacitor has been installed. on a station bus, it is possible to tune it to ground to provide an efficient short circuit at one frequency. Particularly if applied to all three phases, this technique can provide an exceptionally high magnitude of cross-station attenuation at the one frequency. However, there is a risk that the inductive characteristic of the tuned circuit at frequencies higher than its resonance can produce parallel resonance with the capacitive impedance of the bus. This will permit lower than normal values of cross-station attenuation to exist at the frequency of this antiresonance. In some situations, separate single-frequency phase-to-ground circuits are coupled to each of the three phases of a power line. If a cross-coupling problem occurs on one of these circuits, it is possible to make use of the existing coupling capacitors on the other phases to effect a tuned short circuit to ground. The procedure is to add components converting the line tuners to double-frequency operation, tune the second frequency section to the frequency of the problem circuit, and connect it to ground. If a cross-station attenuation problem exists between two nonparallel lines in a station, a fair degree of improvement may be realized by coupling one carrier circuit center phase to ground and the opposing carrier circuit outer phase to outer phase (mode 2 coupling). The cross-coupled carrier energy will approach a bridge-type null to whatever extent balance exists in both symmetrical circuits. There is no practical method of reducing mutual coupling between parallel transmission lines. Generally the only defense against interference which it can cause is careful selection of frequencies, so that all carrier circuits can operate in a compatible manner. Additional data on parallel lines will be found in 6.2.2.

3.8 Intrabundle Channels 3.8.1 Basic Concept Intrabundle or bundled-phase communication channels can provide a means for sending PLC signals on a single phase of a power line without the need for a ground return path.

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The approach involves the transmission of signals on two or more conductors within a multiconductor phase bundle in which the individual conductors are insulated from each other. Theoretical and experimental work has been carried out on such channels. Experimental intrabundle lines have been operated in the USSR [14] and Germany [15]. Intrabundle channels are in actual operating status in both Norway and Bavaria. 3.8.2 Line Configuration Bundled conductors are normally used on 345 kV and higher voltage lines to reduce problems due to corona discharges. Typically conductive separators are used to maintain separation between conductors in the phase bundle and to equalize the voltage on all conductors in the bundle. In order to utilize a bundled-conductor phase as an intrabundle channel, insulated separators must be used, so that independent carrier currents can circulate in the subconductors in the phase bundle. The unavailability of insulated spacers currently represents a technical problem for the application in intrabundle systems. However, Brestkina [14] report the use of 5000 glass-plastic spacers on their experimental line for a year and a half without failures. At the terminal ends of an intrabundle channel, means must be provided to terminate the channel in its proper characteristic impedance. This could be done by utilizing two line traps and two coupling capacitors at each end in a conventional approach, but a cumbersome mechanical arrangement would result. A more effective coupling is needed. Hasler [16] suggested a quarter-wave coupling scheme for wide bandwidths over an intrabundle channel for frequencies in the range of 540–2140 kHz. (Later studies indicate that for several contributing reasons, 1000 kHz is likely to be the practical upper limit [3].) They further suggested the mounting of two coupling capacitors in a single porcelain, since both capacitors would be subject to the same Line potential, and a cost saving should result. 3.8.3 Experimental Line An experimental, single-circuit, horizontally spaced, 330 kV line, 61.4 km long, with two-wire insulated phase bundles, has been operated in the USSR as previously mentioned. Glass-plastic spacers were installed with a mean separation of 35 m. Other line parameters were reported as follows:

Phase separation

9.5

m

Subconductor separation

40.0

cm

Phase conductor height (at tower)

21.8

m

Ground wire height (at tower)

30.0

m

Ground wire spacing

12.2

m

Subconductor diameter

2.35

cm

Ground wire diameter

9.4

mm

Attenuation and crosstalk measurements were made on the experimental line when deenergized, and noise measurements were made on the energized line. The attenuation and crosstalk measurements were made with the test equipment Coupled through large capacitance values to the line, whereas the noise measuring equipment was coupled through conventional coupling arrangements. These measurements covered the band from 30 through 500 kHz.

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3.8.4 Line Characteristics The calculated attenuation for the experimental intrabundle line described in 3.8.3 is shown in Fig 14. Close correlation between measured and computed values was reported. For comparison, the dashed line represents a typical attenuation curve for a center-phase-to-ground carrier channel. It can be seen that the intrabundle line attenuation is substantially lower.

Figure 14—Intrabundle Channel Attenuation for a 300 kV Line, Dashed Line— Phase-to-Ground Attenuation for Comparison Fig 15 contains the curve of Fig 14 (curve A) plus two additional calculated curves (B and C) for frequencies from 500 to 2000 kHz [16]. The properties of the dual subconductors for each of the curves in Fig 15 are listed in Table 11. The subconductor of curve C has a much larger aluminum cross-sectional area than the subconductor of curve B, and hence its attenuation is substantially lower.

Figure 15—Calculated Intrabundle Channel Attenuation with Different Conductors and Spacings. Conductors A, B, and C are identified in Table 11

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Table 11—Intrabundle Conductor Configurations [14], [16] Fig 15 Parameter Intrabundle conductor spacing, cm Subconductor diameter, cm

Curve A

Curve B

Curve C

40

30

30

2.35

1.96

2.86

Aluminum cross-sectional area, mm2

300

184

486

Steel core cross-sectional area, mm2

28

43

63

36/1

30/7

54/7

Strands: aluminum/steel

Table 12 tabulates calculated multiplying factors [14] to account for additional attenuation in an intrabundle line resulting from the formation of a 2-cm thick ice coating on the subconductors. The attenuation increase is large. Table 12—Multiplying Factors for Intrabundle Channel Attenuation to Account for Icing Frequency (kHz)

Multiplying Factor

50

8.0

100

8.8

200

12.4

300

14.6

400

14.2

500

12.8

Near-end crosstalk was measured on the experimental line between two intrabundle coupled phases and between an intrabundle channel and a phase-to-ground PLC channel on the same line. The mean value of measured attenuation was 60 dB between the two intrabundle channels and 40 dB between the intrabundle channel and the phase-to-ground channel. Far-end crosstalk attenuation is shown in Fig 16. 3.8.5 Recent Experiments Studies conducted in Germany [15] have provided data on the use of a quarter-wavelength coupling filter. In addition, these studies included observations of intrabundle channel performance during line faults. It was determined that clashing of subconductors during a fault caused momentary circuit failure. The circuit was judged unsuitable for protective relaying applications requiring through-fault transmission. A comprehensive discussion and additional performance data may be found in the CIGRE guide [3]. 3.8.6 Application Considerations Although intrabundle channels have been successfully used in some countries, many technical problems still exist, particularly with regard to material availability for large-scale applications in countries where the first application has not yet been made. Experience with line-coupling methods, insulated separator performance, and fault behavior on an intrabundle coupled line is still limited.

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Figure 16—Far-End Crosstalk Attenuation. (a) Between Two Intrabundle Channels; (b) Between an Intrabundle Channel and a Phase-to-Ground PLC Channel. Solid Lines—Calculated; Dashed Lines—Measured. The intrabundle technique has several significant advantages. 1) 2) 3) 4)

Lower fair-weather line losses than conventional carrier channels High crosstalk attenuation between channels Higher density application of channels Less interference with or from radio services in the PLC band.

It is possible that the intrabundle channel will become a useful technique in future applications of PLC, particularly in ice-free areas.

3.9 Line Coupling Methods There are several ways for feeding one or more conductors of a three-phase power line so that PLC signals will propagate down the line. Modal theory (see 3.6) shows that carrier signal currents generally flow in all three phases of a power line as well as in any static wires present. The efficiency of coupling signals to and from the lines depends on the particular coupling arrangement selected. Coupling methods used include the following: 1) 2) 3) 4) 5)

Mode 1 Center phase to ground Center phase to outer phase Outer phase to ground Intercircuit

The first four methods apply to one three-phase power line, but the last method is used to couple signals into two separate power lines.

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Mode 1 coupling is theoretically the lowest loss method for coupling carrier signals to and from the power lines. It has the advantage of providing a very high channel dependability, since it can withstand a single-phase or a two-phase-toground fault near a terminal without the carrier signal being eliminated. It has the disadvantage of high cost since three sets of tuning equipment are required. An arrangement that provides a good approximation for mode 1 coupling is shown in Fig 17. Two isolation transformers (IT) are connected so that the current supplied to the center conductor of the power line is 180° out of phase with the equal currents supplied to the two outer conductors and equal in magnitude to their sum. The modal coupling efficiency for mode 1 coupling is 0 dB (or 100%; see 3.6).

Figure 17—Mode 1 Coupling The simplest and most frquently used coupling method is the center-phase-to-ground scheme shown in Fig 18. It utilizes the minimum amount of equipment and is fairly efficient. The center-phase-to-ground modal coupling efficiency is −1.8 dB at each end of the line. Additional losses can occur, however, if static wires are not used on the power line, and if the soil in the vicinity of a terminal has poor conductivity. Center-phase-to-outer-phase (interphase) coupling, shown in Fig 19, is the next most frequently used method and is becoming increasingly popular. In this arrangement the two currents are equal in magnitude and 180° out of phase. It has the advantage of being somewhat more efficient than center-phase-to-ground coupling, and provides a −1.2 dB modal efficiency per line end. This method provides greater channel dependability than center-phase-to-ground coupling, since a single-phase fault at the substation will not eliminate the carrier signal. Further, its coupling loss is not significantly affected if static wires are not used and if the ground conductivity is poor. Intercircuit coupling, Fig 20, is generally a center-phase-to-center-phase coupling between two separate power lines in close proximity to each other. This arrangement provides two redundant paths for the carrier signal. Any of the line tuning methods discussed in 4.3 can be used with these couplin arrangements. When using any multiconductor coupling method, care should be taken to provide proper phasing.

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Figure 18—Center-Phase-to-Ground Coupling

Figure 19—Adjacent-Phase-to-Phase Coupling

Figure 20—Intercircuit Coupling Copyright © 1998 IEEE All Rights Reserved

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3.10 Insulated Shield Wires 3.10.1 Insulated Shield Wire Chanels The overhead shield wires (ground wires) on a power transmission line can be used as a carrier transmission medium by insulating the conductors from the towers [17], [18]. These conductors (either a single shield wire or a shield-wire pair) are usually isolated from the tower with an insulator having an air-gap breakdown of 15–25 kV. Thus the original purpose of the overhead shield wires, from a lightning breakdown point of view, is not compromised. A benefit resulting from the use of overhead shield wires is the reduction in 60 Hz drainage currents induced in the shield wires. The resulting power savings may be significant for a shield-wire pair where the conductors are properly transposed. Insulating the shield wires causes a slight increase in the zero-sequence impedance of a power-line circuit which can cause a corresponding increase in overvoltages associated with line-to-ground faults. In most systems this increase can be neglected; however, in systems with high ground impedance it may be significant. The cost of the coupling equipment required for an insulated-shield-wire application is less than that required for PLC coupling. PLC applied to phase conductors requires coupling capacitors and wave traps, and the cost of the coupling capacitors increases substantially at higher transmission-line voltages. In contrast, the cost of insulated-shield-wire coupling equipment remains relatively independent of system voltage. The shield-wire channel has an advantage in that it does not require a powerline outage when equipment maintenance is needed, whereas an outage is required for major maintenance of PLC coupling equipment. Major requirements for a shield-wire system include conductors and conductor insulation, transpositions, and terminating equipment. 3.10.2 Conductors and Conductor Insulation The conductors used on a new insulated-shield-wire channel are usually aluminum-jacketed steel cables. Insulation is sometimes added to utilize existing steel wires. The characteristic impedance of a single overhead insulated-shieldwire line is approximately 500 Ω, whereas the balanced pair configuration has a characteristic impedance of about 900 Ω. The shield-wire conductor or conductor pair is ordinarily insulated from ground at each tower. An exception might be where the shield wires are intended to serve as a parallel coupler to induce a carrier signal into the phase conductors. Theoretically such coupling can be effected by a relatively short segment of insulated shield wire at each end of the power line. No experimental data have been published. Insulator requirements are not critical as long as the voltage breakdown is adequate. 3.10.3 Transpositions Whenever two shield wires are used as a pair, a suitable transposition scheme is necessary for the 60 Hz power saving benefit to be realized, to provide balance in the communication circuit, and to make the termination requirements less difficult. The objective in selecting transposition locations is to equalize the induced 60 Hz current in the two wires. This is accomplished by equally dividing the line length over which each wire occupies each position in the line configuration. The maximum distance between transpositions will be determined by how much voltage should be permitted to exist on the shield wires during heavy power-line loading. Depending on the line current and user practice, guidelines limiting this distance to values from about 6 to 30 mi (10 to 50 km) have been used. Induced voltage per unit distance, which is very pertinent to the selection of this value, can be determined by a relatively easy computation [2].

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3.10.4 Shield-Wire Terminating Equipment A suitable protective coupling device must be used at each end of the shield-wire channel to provide a drainage path to ground for the induced 60 Hz influence and lightning currents while coupling the carrier signal to the shield wires. An example of a shield-wire coupler is shown in Fig 21. Other coupler arrangements have been used successfully. For example, some do not require the additional capacitors in series with the coupling capacitors. Many installations have been made using a balanced cable pair instead of coaxial cable for the drop connection. Most of the generally accepted two-wire coupling methods have also been adapted for single-wire operation, except those with iron-core drainage coils.

Figure 21—Insulated Shield-Wire Coupler Drainage coils in two-wire couplers may be either iron-core or air-core inductors. Iron-core drainage coils provide both economic and space advantages. Their performance is adequate where the induced power-frequency currents in the two halves of the drainage inductor are essentially equal, as in well-balanced two-wire circuits. Significant unbalance in drainage currents will cause core saturation. Air-core drainage coils are necessary in single-wire applications or wherever unbalance is severe. 3.10.5 Performance Flashover of the shield wire insulators will cause the carrier frequency attenuation of the circuit to increase, but usually not enough to cause a communication outage. The duration and the amount of the increase depend on the cause of the flashover and the power circuit conditions which accompany it. For example, where lightning causes a flashover which occurs only on the shield wires, the added loss will be small and its duration typically in the order of 1 ms or less. Measurements have indicated an increase in loss ranging from 1 to 8 dB, dependent on frequency, during flashovers near the end of a line. Flashovers which accompany a power-line fault will last until the fault is cleared—typically about 60 ms or longer. The increase in attenuation will also be higher, depending on how many phases are involved in the fault. During wet weather an allowance must be made for approximately a 20% increase in attenuation. Insulated shield wires are more susceptible to increased losses due to frost than phase wires as they do not have significant currentinduced self-heating. Also, complete ice bridges are sometimes formed over the relatively small insulators. Complete communication circuit outages have been experienced during severe icing conditions.

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Transmitter and receiver terminal equipment used on shield-wire channels can be identical to that used on PLC channels. Most of the transmitters used currently have 1 W or greater power output. A frequency range from 8 kHz to 500 kHz is technically feasible for shield wire channels. Because high noise and interference are present at low frequencies on the shield wires, the coupling equipment must be designed to provide a band-pass or high-pass filter characteristic with a cut-off frequency of about 5 kHz. Frequencies below this cutoff are attenuated about 40 dB. Insulated-shield-wire channels are used for voice, supervisory control, alarm, and telemetry functions. Some protective relaying channels have been applied on insulated shield wires, but there is limited application experience. 3.10.6 Other Systems Using Shield Wires Experiments are being actively pursued using composite cables in which the outer portion of the cable is the transmission-line shield wire. In separate systems the core of the composite cable may contain optical fibers or conventional metallic cable conductors. Such experiments with optical fibers are just beginning. However, extensive data and experience have already been obtained with the use of aerial cables on power lines in Europe [19].

4. Channel-Performance Evaluation 4.1 Factors Involved in Channel Performance Two basic principles govern satisfactory channel performance. The first is the ability of the receiver to detect the transmitted signal, and the second is the relative magnitude of the desired signal with respect to the magnitude of interfering noise, called signal-to-noise ratio (SNR). The desired signal incurs losses in transit from transmitter to receiver. Therefore the transmitter must have sufficient power to provide an acceptable signal level to the receiver after having suffered the attenuation cause by the channel elements (see 3.5 and Section 5). Two basic noise types of interest are random (white) noise and impulse noise. Noise has been discussed in 3.3 and will not be covered in this section except for its effects. The signaling function, the type of carrier transmitter/receiver, and the receiver design features will determine whether impulse or random noise is most important. However, as a basis for application, random noise is more predictable and lends itself to making an analysis of carrier performance. Random noise generally exists at a constant power level over the whole line length. However, faulty insulators, faulty hardware, or a localized storm on a long line can cause higher noise levels at specific locations. These will incur the same attenuation as the desired signal in reaching the receiver. Interfering signals from other communication equipment can cause misoperation of carrier receivers. Two examples are: 1) 2)

Beat frequencies caused by intermodulation when adjacent carrier transmitters do not have adequate isolation from each other Alien frequencies which are in the same operating band or very near to the same operating frequency as the affected receiver

Adjacent frequencies generated by transmitters at or near the same location must be chosen with consideration given to the selectivity of the receiving equipment (Section 6). Proper determination of channel selectivity, frequency assignments, isolation obtainable from attenuating equipment, and the required circuit configuration (including line traps) must be made for successful carrier applications.

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Receiver noise tolerance is given in terms of the minimum permissible SNR. This is usually expressed in dB. The SNR is determined as follows: SNR (dB) = signal level (dBm) – noise level (dBm) A graphical representation of signal and noise on a transmission line is shown in Fig 22. It should be noted that both signal and noise are attenuated by the coupling equipment and circuits at the receiving end. Thus the SNR does not change from the line side of the coupling capacitor to the receiver terminals. The frequency dependence of channel attenuation and noise should be considered in carrier applications. Fig 23 shows the general effect of the frequency on signal attenuation and noise levels. Attenuation (per unit of line length) increases with increasing frequency, while noise (independent of distance) decreases with increasing frequency. Although carrier signal loss per unit length may be high at the higher frequencies, the total attenuation for many lines will be acceptable, and a good SNR might still be obtained because noise is lower a the higher frequencies. On very long lines file utilization of lower frequencies may be more desirable or even necessary.

Figure 22—Noise and Signal Levels on EHV Lines

Figure 23—Typical Frequency Characteristics of Fair-Weather Noise and Attenuation on 500 kV Lines

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The carrier equipment characteristics, as well as channel attenuation and noise, affect the overall performance of a carrier channel. The minimum SNR that win give satisfactory operation can vary widely and depends on the equipment characteristics, type of modulation, design features, and signal function. Manufacturers' data and recommendations should be used in planning carrier applications. Typical permissible SNR values in use for modern carrier equipment are given in Table 13. Table 13—Signal-to-Noise Ratios

Modulation

SNR* (dB)

In-Band SNR (dB)

60 baud

FSK

3

20

300 baud

FSK

10

20

1200 baud

FSK

15

20

Line protection

AM (on-off)

15–20

13†

Line protection

FSK

10

13

Slow speed

FSK

0–5

13

Medium speed

FSK

3–7

13

High speed

FSK

5–10

13

SSB

25–30

25–30

FM

25

25

Function Telemetering/data

Relaying

Transferred trip

Voice

*Based on noise in a 3 kHz bandwidth. †SNR should be higher than the receiver sensitivity margin setting.

It should be noted that for some equipment the limiting application condition will be receiver sensitivity, and for other equipment it will be noise. Illustration of the first limitation requires a definition of sensitivity margin setting. It is customary to adjust the receivers on an AM (on-off) carrier relaying channel for a standard sensitivity margin (typically between 12 and 15 dB for most utilities). This means that the sensitivity setting will allow the receiver to operate with a reduction in signal level equal to but not more than this amount. In the illustrative example (Fig 24) a minimum received signal of +10 dBm would satisfy the 20 dB minimum SNR requirement, but a received signal of +15 dBm is required to provide a 15 dB sensitivity margin. Fig 25 illustrates the second situation where noise is the limiting factor. In this example receiver sensitivity is adequate, but the total path attenuation must be limited to that value which will provide an SNR that permits adequate intelligibility in the voice circuit. For single-function equipment, channel-performance evaluation is relatively straightforward, once total path attenuation and noise levels are known. The total power of the transmitter is dedicated to performing a single function, and its output directly represents the available power.

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Figure 24—Operating Range Limitations of an AM (On-Off) Carrier Channel

Figure 25—Operating Range of SNR of a Multifunction Carrier Channel Multifunction, multichannel carrier (usually SSB equipment) requires the predefinition and calculation of a modulation assignment for each of the functions that share in common the transmitter's available power output. These assignments are usually based on the limits of voltage linearity of the transmitting amplifier rather than its thermal power capability. An amplifier capable of an undistorted single-frequency output of 20 W might not be distortion free if it is adjusted to produce two simultaneous signals of 10 W each from the common output. For this reason it is generally customary that multifunction assignments be made on the basis of allocated voltage rather than power. As an example, the rms voltage of a 20 W single-frequency signal on a 50 Ω coaxial cable is 31.6 V. If this voltage were

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divided equally between two signals (15.8 V each), the resultant power would be only 5 W for each signal. Of course some diversity can be assumed when a relatively large number of equal functions are operated through one common amplifier. A maximum permissible simultaneous modulation level of the transmitter (as recommended by its manufacturer) should be observed in dividing and assigning modulating signal levels. The effective power level is then calculated in terms of dBm for each function. This effective power for each function may be examined individually to determine if its SNR is acceptable.

4.2 Single-Function Individual-Channel Case A typical protective relaying PLC channel configuration utilizing single-function equipment for each relaying channel is shown in Fig 26. The AM (on-off) carrier (frequency f1 ) would be used for line protection, and the FSK carrier, one channel in each direction, for transferred-trip relaying. Because impulse noise, bad weather, and line faults can be critical to high-speed relaying, conservative practice usually demands greater operating margins and SNRs than for less essential functions. PLC relaying applications are therefore examined very closely. Total path attenuation is calculated from reference data, and the manufacturer's equipment performance specifications together with noise data are examined to predict overall channel performance. The parameters associated with the PLC configuration in Fig 26 are as follows. The three frequencies f1, f2, and f3 are 100, 140, and 141.5 kHz, respectively. The line tuner at each station is a two-frequency resonant type (see 5.3) operating with a 0.001 µF coupling capacitor and a fixed wide-band line trap which provides 400 Ω blocking impedance ZT at each frequency. The phase-to-ground characteristic impedance Z0 of each 230 kV line terminating in either station is 400 Ω. The bus impedance ZA at station A is assumed to be 600 Ω and ZB at station B is 700 Ω. The power transformer impedance ZTR at station B is assumed to be 1000 Ω at each carrier frequency. For this example it will be assumed that the manufacturer's performance specifications stipulated the following:

Operating Range (dB)

Minimum SNR* (dB)

Am carrier, 10 W

40

20

FSK carrier, 1 W

60

10

Carrier

*Based on 3 kHz bandwidth noise measurement.

A PLC signal attenuation calculation requires that the losses caused by each PLC equipment component (or assembly) and the attenuation caused by each power system element be determined, and that their arithmetic sum be obtained. This sample calculation will show these losses as they are encountered by a signal originating at the transmitter at station A and traveling to the receiver at station B.

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Figure 26—Application of Single-Function Individual Carrier Channels At station A, 200 ft (60 m) of coaxial cable at 100 kHz has a 0.11 dB loss (Fig 8). The two-frequency line tuner loss and the coupling capacitor loss are dependent on the equivalent load impedance ZL1 and the capacitance of the coupling capacitor. The value of ZL1 is calculated from the parallel combination of Z0 and Z1. Z1 in turn consists of ZT in series with the parallel combination of the five lines and equivalent bus impedance. Note that bus impedance is due to capacitance to ground of equipment insulating bushings and bus insulators. The simplifying assumption that all impedances have the same phase angle is made, although this is not strictly correct. Thus, 1 Z 1 = Z T + --------------------------------- = 400 5 ⁄ Z0 + 1 ⁄ ZA

1 + -----------------------------------------( 5 ⁄ 400 ) ( 1 ⁄ 600 ) Z1 = 470 Ω and Z 0Z 1 ( 400 ) ( 470 ) Z L1 = -----------------= ---------------------------- = 216Ω 400 + 470 Z0 + Z1

From Fig 43 at 100 kHz the coupling loss for single-frequency resonant tuning with a 216 Ω load impedance and a 0.001 µF coupling capacitor is approximately 1.1 dB. As stated in 5.3, the attenuation for a two-frequency line tuner is approximately twice that of the single-frequency tuner. Thus the coupling loss is listed as 2.2 dB. Next the shunt loss incurred by the signal in the direction of Z1 is found, using Eq. 8 from 3.5.6.6: Z0 + Z1 400 + 470 Shunt loss = 10 log -----------------= 10 log -----------------------Z1 470 = 2.67 dB

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The line attenuation is found using Fig 5 and Tables 4 and 5. The total line attenuation for fair-weather conditions is as follows: (dB/mi) × (mi) × (line voltage multiplier) + (coupling correction) + (transposition correction) = line attenuation 0.096 × 80 × 0.78 × + 1.0 + 0 = 6.99 dB For adverse-weather conditions the total line attenuation is as follows: 0.096 × 80 × 0.98 + 1.0 + 0 = 8.53 dB The shunt loss at the receiving end is found from the equation Z0 + Z2 shunt loss = 10 log -----------------Z2

where 1 Z 2 = Z T + -------------------------------------------------------- = 500Ω 3 ⁄ Z 0 + 1 ⁄ Z B + 1 ⁄ Z TR

Thus shunt loss is 400 + 500 10 log ------------------------ = 2.55 dB 500

The coupling loss is dependent on ZL2: Z 0Z 2 ( 400 ) ( 500 ) = ---------------------------- = 222Ω Z L2 = -----------------900 Z0 + Z2

From Fig 43 for 100 kHz, 222 Ω load, and a 0.001 µF capacitor, the single-frequency line tuning loss is approximately 1.05 dB. For a two-frequency tuner it is twice as much, or 2.1 dB. Similar to the calculation for station A, the loss in the coaxial cable is 0.11 dB. The results are given in Table 14 for the AM carrier channel. All entries are rounded to the nearest tenth of a dB. A subtotal is calculated for the attenuation from the transmitter at station A to the 230 kV lineside terminal of the coupling capacitor at station B. This is for use in the SNR calculation, since signal and noise are attenuated equally from that point on. It should be noted that even though losses and SNRs are shown to a tenth of a dB in the examples, the accuracy of these estimates is only good to the nearest dB at best. For the FSK carrier losses the same procedures are followed, recognizing that the carrier frequency is now 140 kHz. Note also that the FSK circuit includes a carrier-frequency hybrid (HYB in Fig 26) for which a fixed 3.5 dB loss allowance must be made at each end of the line (see 3.5 and 5.5). The summary of losses for the FSK circuit is given in Table 15.

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Table 14—AM Carrier Channel Losses Loss (dB) Fair Weather

Adverse Weather

Coaxial cable

0.1

0.1

Coupling (2-frequency line tuner)

2.2

2.2

Shunt

2.7

2.7

Channel Component

Line attenuation

7.0

8.5

(12.0)

(13.5 )

Shunt

2.6

2.6

Coupling

2.1

2.1

Subtotal

Coaxial cable Total loss

0.1

0.1

16.8

18.3

Table 15—FSK Carrier Channel Losses Loss (dB) Channel Component

Fair Weather

Adverse Weather

Hybrid

3.5

3.5

Coaxial cable

0.1

0.1

Coupling (2-frequency line tuner

1.5

1.5

Shunt

2.7

2.7

Line attenuation

8.8

10.8

(16.6)

(18.6)

Shunt

2.6

2.6

Coupling

1.5

1.5

Coaxial cable

0.1

0.1

Hybrid

3.5

3.5

24.3

26.3

Subtotal

Total loss

Noise levels can be estimated from Fig 4. The conditions assumed for this example are for fair weather, −35 dBm at 100 kHz and −36 dBm at 140 kHz; and for adverse weather, −18 dBm at 100 kHz and −19 dBm at 140 kHz. Operating ranges and minimum SNRs for the equipment were given at the beginning of this section. This completes the information necessary to calculate the predicted performance of these single-function carrier channels. To illustrate the estimation of channel performance, Figs 27 and 28 show operating margins and SNRs. For the AM channel the total adverse-weather attenuation of 18.3 dB (Table 14) is well within the 40 dB operating range of the carrier equipment and will easily permit the receiver sensitivity to be adjusted for a 15 dB margin setting. The adverse-weather SNR (Fig 27) is SNR = 26.5 − (-18) = 44.5 dB Copyright © 1998 IEEE All Rights Reserved

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Figure 27—AM Carrier Channel Performance. All Received Levels are Referenced to the HV Terminal of the Coupling Capacitor

Figure 28—FSK Carrier Channel Performance This SNR is above the required minimum of 20 dB by a margin of 24.5 dB. This examination shows that satisfactory operation is to be expected.

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For the 140 kHz FSK channel the adverse-weather attenuation is 26.3 dB (Table 15). This is well within the capability of the carrier equipment, which has a maximum range of 60 dB. It should be remembered, however, that line faults can cause additional attenuation, and if tripping through a fault is required, this would leave a much smaller margin. The adverse-weather SNR (Figure 28) is SNR = 11.4 − (−19) = 30.4 dB This SNR is above the acceptable minimum of 10 dB by a margin of 20.4 dB. Thus operating range and SNR parameters indicate a successful application.

4.3 Multifunction, Multichannel Case 4.3.1 Introduction Dual-channel SSB equipment is used to illustrate the SNR criteria which usually limit the application of multifunction carrier equipment. Three values must be determined before the SNR of each applied function can be calculated, that is, effective transmitted power, path attenuation, and line noise. These values are discussed separately in the following paragraphs, and an example is given for calculating a typical SNR for three functions. The configuration of the PLC arrangement for this example is as shown in Fig 29. The parameters involving the power apparatus are similar to those in the single-function example (Fig 26), except that the characteristic impedance of the 345 kV line is about 340 Ω. For the purpose of calculating line attenuation, the line is 100 mi (161 km) long and has steel shield wires. The line tuner is a wide-band (bandpass) type. A carrier frequency of 128 kHz is assumed for the channel to be evaluated. This channel is to provide a voice circuit plus one tone for signaling to support the voice function. In addition, there will be four tones to carry relatively important telemeter readings at a 60 baud rate.

Figure 29—Application of Multifunction Carrier Channels

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4.3.2 Effective Transmitted Power From a modulation plan calculated for all the channels and functions of Fig 29 the following values of effective power for the individual functions in the example were derived from the manufacturer's data:

Voice

+26 dBm

Telemeter tones, each

+20 dBm

Signaling tone

+16 dBm

These values will be used to determine the SNR after each signal is reduced by the path attenuation from the transmitter at station A to the high-voltage terminal of the coupling capacitor at the receiver end (station B). 4.3.3 Path Attenuation The total path attenuation is comprised of shunt losses, coupling losses, and line attenuation. Computations to obtain these values are the same as those described for the single-function carrier application and will not be repeated in detail. For this illustration it is assumed that the combined coupling and shunt losses for the transmitter at station A are 6 dB. The values for the transmitting end only are used in the SNR calculation since receiving-end losses reduce both signal and noise by the same amount and therefore do not affect the SNR. Values of line attenuation during fair weather and adverse weather are 12 and 15 dB, respectively. 4.3.4 Noise Noise values (in a 3 kHz band width) for both fair- and adverse-weather conditions are required to determine if the SNRs are satisfactory. The values estimated for this example (from Fig 4) are -34 dBm and -17 dBm for fair weather and adverse weather, respectively. 4.3.5 Results of Evaluation Using methods similar to those in the single-function example in 4.2, the SNRs of each function may be evaluated for both fair and adverse weather. Pertinent relationships for the voice function are shown in Fig 30. The fair-weather SNR of 42 dB is very good; however, for adverse-weather conditions the SNR of 22 dB is barely acceptable, and methods for improvement should be studied (see 4.3.6). SNR evaluation for the signaling tone and telemeter tones are given here only in terms of adverse-weather performance. The manufacturer's modulation plan allocated a +16 dBm power level to the signaling tone. Considering the adverse-weather loss of 21 dB, the received signal level to be expected is −5 dBm. The adverse-weather noise level is −17 dBm: SNR = −5 − (−17) = 12 dB The required SNR for an FSK signaling function is about the same as for slow-speed (60 baud) telemetering, or 5 dB (Table 13). Since 5 dB is acceptable, the SNR of 12 dB represents a 7 dB margin. The modulation plan allocated +20 dBm effective power level to each telemeter tone. Performing similar calculations, a 16 dB SNR is obtained. According to Table 13, a 5 dB SNR for 60 baud telemetering tones is adequate. Thus this application is acceptable by a margin of 11 dB.

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Figure 30—Single Attenuation and SNR for the Voice Function Chosen for Illustration. Received Signals Are Referenced to the HV Terminal of the Coupling Capacitor 4.3.6 Considerations for Improvement Means for improving the marginal voice SNR include increasing transmitter power, reallocation of effective power outputs for the various functions, and the use of compandors in the voice circuit. An increase in transmitter power is expensive and is usually done only as a last resort. Reallocation of effective power levels for the different functions is sometimes practical. In this example, if the effective output level of the four telemeter tones were each reduced from +20 to +16 dBm, the voice function could be increased to approximately 31 dBm without increasing the peak-voltage loading of the multifunction transmitter. This would raise the voice SNR during adverse weather to about 27 dB. Compandors in a voice circuit usually produce a 10 to 20 dB improvement in SNR. Thus with compandors, at least a 32 dB SNR could be expected. Selection from options of this type is a matter of judgment on the part of the user. In this case it appears that the use of a compandor is likely the most practical solution, since it would provide a fully adequate SNR for the voice function and still retain the entire 11 dB margin for each telemeter tone function.

5. Coupling Components 5.1 Line Traps 5.1.1 Introduction The function of a line trap is to present a high impedance at the carrier frequency or frequencies being used while introducing a negligible impedance at the power frequency, thereby preventing the carrier signal from (1) being dissipated in the station equipment, (2) being gounded in the event of a fault outside the carrier transmission path, or (3) being attenuated by a tap line or a branch of the main transmission path. Line traps prevent out-of-phase reception Copyright © 1998 IEEE All Rights Reserved

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which can result if multiple-path transmission of a carrier signal is permitted to take place. They permit a greater choice of carrier frequencies by minimizing interference from other carrier channels. In effect, the purpose of a line trap is to ensure that the carrier transmission path remains isolated, at carrier frequencies, from the rest of the system. 5.1.2 Application of Line Traps When choosing a line trap, a careful evaluation of the purpose of the carrier channel, the degree of reliability required, the attenuation characteristics of the transmission path, and any future alterations to be made in the carrier channel must be made. Line traps are available with a variety of performance characteristics which can allow the selection of the type most suitable for present and future applications. Line traps may be tuned or untuned. The tuned type of line trap (by far the most prevalent) is essentially a parallel L/ C network with variations in the tuning circuit (or tuning pack as it is normally called), which provides for the choice of single-frequency, double-frequency, or wide-band tuning. Tuning packs are normally designed to operate in a specific band of the carrier-frequency spectrum. These bands are available from manufacturers in standard ranges. The untuned line trap is s simple inductance which establishes carrier transmission path isolation by virtue of the very high inductive reactance of its coil. 5.1.3 Resonant Traps 5.1.3.1 General Resonant traps are available for single-frequency and two-frequency applications. They are designed to block only one or two carrier-frequency signal bands and typically, have less than 0.5 mH inductance. 5.1.3.2 Single-Frequency Traps The single-frequency line trap is the simplest of the tuned traps available. It is tunable to parallel resonance at any frequency within its nominal range. Its blocking band is usually defined as the band of frequencies over which the impedance magnitude is greater than 400 ω. The circuit diagram and impedance characteristics for a single-frequency trap are shown in Figs 31 and 32, respectively. Frequencies fA and fB in Fig 32 define the limits of the 400 Ω bandwidth. The electrical characteristics of the singlefrequency line trap are discussed further in 5.1.5.2. 5.1.3.3 Two-Frequency Traps The two-frequency trap, shown schematically in Fig 33, has a blocking band around two distinct resonant peaks, as shown in Fig 34. 5.1.4 Wide-Band Traps 5.1.4.1 General Wide-band traps are more suitable for multichannel applications because it is very difficult to design resonant traps for more than two frequencies. The wide-band trap is tuned to a specific frequency band, such 90 to 200 kHz, and channels can be placed anywhere in the band. In this type of line trap a tuning device is combined with the inductance of the main trap coil to provide a minimum blocking impedance across the entire band. Two types of wide-band traps are used. These are the fixed wide-band and the adjustable wide-band. The untuned inductor, although not complying with the definition in the preceding paragraph, is also frequently classified as a wideband line trap.

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Figure 31—Single-Frequency Line Trap

Figure 32—Impedance of a Single-Frequency Line Trap

Figure 33—Two-Frequency Line Trap

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Figure 34—Impedance of a Two-Frequency Line Trap 5.1.4.2 Fixed Wide-Band Traps Fixed wide-band line traps are factory constructed for a specific frequency band and cannot be adjusted in the field. The schematic of a fixed wide-band trap is shown in Fig 35. Representative impedance characteristics are shown in Fig 36. 5.1.4.3 Adjustable Wide-Band Traps The adjustable wide-band line trap, shown schematically in Fig 37, has a tuning device which permits positioning the blocking band in various frequency regions. This type of tuning also permits the selection of different values of minimum impedance. The impedance characteristics of the adjustable wide-band trap are essentially the same as those of the fixed wideband trap (Fig 36).

Figure 35—Wide-Band Line Trap

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Figure 36—Impedance of a Typical Wide-Band Line Trap

Figure 37—Adjustable Wide-Band Line Trap 5.1.4.4 Untuned Line Inductors Untuned line inductors are high-inductance line traps (0.5 mH or higher) that do not require a tuning pack. Their application usually involves highpass coupling networks. An advantage of an inductor is that the adjustment and maintenance of a tuning pack are not required. An occasional disadvantage is that under certain switching conditions it is possible that the inductor can series resonate with the capacitance of switchyard apparatus. This is most likely to occur under open circuit-breaker conditions, that is, a deenergized line. Some line inductors are sail-resonant at frequencies within the carrier band. Such inductors generally have a low Q (typically 8–10) and a high series resistive component. Self-resonant inductors are more advantageously applied in band-pass coupling networks.

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5.1.5 Electrical Characteristics of Line Traps 5.1.5.1 General The carrier-frequency characteristics of line traps are considered in this section. Power-frequency and mechanical characteristics are considered in 5.1.6. 5.1.5.2 Single-Frequency Traps The single-frequency line trap provides blocking impedance for a single frequency or for a narrow band of frequencies. The schematic and impedance curves for the single-frequency trap are shown in Figs 31 and 32, respectively. Curves for impedance Z, resistive component R, and reactive component X are shown separately in Fig 32. The lowest resonant frequency, f0 to which a single-frequency trap might be tuned with a given capacitor C1 is determined from 1 f 0 = -----------------------2π L 1 C 1

(17)

Higher frequencies are tuned by adjusting the tap on L1 away from the end of the coil. The trap coil then serves as an autotransformer to reduce the effective value of C1. The inductance L1 is also reduced to some extent but, unless there is an excessive number of turns between the tap and the trap terminal, the reduction of L1 is not significant. Note that the entire trap coil is always included between the main terminals. The impedance at resonance Z0 for the single-frequency trap can be calculated from the equation Z 0 = 2π f 0 L 1 Q

(18)

The factor Q (quality factor) is equal to the inductive reactance divided by the series resistance in the circuit. If it is assumed that the Q remains constant with frequency, then it is apparent that the resonant-frequency impedance of a line trap is proportional to the trap inductance and to the resonant frequency. The Q of a line trap is sometimes intentionally lowered by connecting a resistor in series with the capacitor in the tuning pack. While this admittedly lowers the trap impedance at its resonant frequency, it broadens the resistive component. The purpose of this change is to counteract degradation to the carrier signal which can occur in some switchyards if the bus capacitance resonates with the inductive reactance component of trap impedance just below its resonant frequency. For frequencies near resonance the impedance and its phase angle can be calculated from the universal resonance curve shown in Fig 38. For frequencies far off resonance the magnitude of the impedance can be approximated using the equation 2πf L Z = ---------------1 2 α –1

(19)

where α

= f/f0

Eq 19 is precise for an infinite Q (ideal coil).

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Figure 38—Universal Resonance Curve for a Parallel-Resonant Tuned Circuit The phase angle θ of the impedance can be determined from the equation 1 tan θ = Q  1 – ------ 2  α

(20)

Two or more carrier channels may be used with a single-frequency trap if their frequencies fall within the bandwidth of the trap. A maximum of 12% spacing is recommended as a rule of thumb. However, the actual usable percentage spacing varies with resonant frequency and may be much higher. 5.1.5.3 Two-Frequency Traps The two-frequency trap provides two resonant peaks of impedance, each of which is similar to a single-frequency resonance. In general the impedance at each resonant frequency is approximately half the impedance that could be obtained with a single-frequency trap. Figs 33 and 34 show the schematic and impedance characteristics, respectively, for the two-frequency trap. As indicated in Fig 33, both resonant circuits L1C1 and L2C2 are tuned to the higher of the two frequencies. Resonance at the lower frequency is then established by selection of the proper value of capacitor C3 and setting the second tap on the main coil, L1. It is generally recommended that the two resonant frequencies be separated by 25 kHz or 25% of the higher frequency, whichever is greater. Closer spacing results in distortion in the shape of the impedance curve between peaks and an increasing lack of symmetry in each blocking band. This effect can be observed by comparing Fig 39 with Fig 40. There are no restraints on maximum frequency spacing. The two resonant frequencies are not required to fall within the same nominal tuning range which may apply to its individual tuning packs. 5.1.5.4 Fixed Wide-Band Traps These line traps provide a wide-band impedance characteristic. To obtain the desired characteristic, circuit components are added to the main trap coil to form a terminated half-section filter (Fig 35). Resonant circuits L1C1 and L2C2 constitute the filter elements, and the resistor R provides the termination. The terminating resistance is normally equal to the nominal characteristic impedance Ro of the filter. This resistance also represents the minimum impedance value within the bandwidth of the trap (Fig 36). In some versions of the fixed wide-band line trap, more complex circuitry is used to obtain a flatter (more desirable) impedance characteristic. Some currently available fixed wideband line traps have an impedance level of 400 ω; others have 600 or 800 Ω.

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Figure 39—Impedance of a Two-Frequency Line Trap with Normal Frequency Spacing

Figure 40—Impedance of a Two-Frequency Line Trap with Close Frequency Spacing The equation for bandwidth (BW) of a wide-band line trap is 2

8.89T L 1 f 0 BW = --------------------------- ( kHz ) R

(21)

where f0 T L1 R

= geometric mean frequency (GMF), in kHz = detuning factor (approximately 0.9) = main coil inductance, in mH = terminating resistance, in Ω

The actual band limits f1 and f2 can be determined once the GMF and BW are known:

56

f1 =

2 2 BW  BW --------- + f 0 – -------- 2  2

(22a)

f2 =

2 2 BW  BW --------- + f 0 + -------- 2  2

(22b)

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5.1.5.5 Adjustable Wide-Band Traps An adjustable wide-band line trap consists of an inductance coil with an adjustable tuning pack which can be either field or factory adjusted. Typically the coil inductance ranges from 0.265 to 1.59 mH. The circuit is electrically equivalent to the fixed wide-band trap in that it forms a terminated half-section band-pass filter. The bandwidth and band limits of an adjustable wide-band trap may be computed using the same formulas as for the fixed wide-band trap, Eqs 21 and 22. The detuning factor T may be slightly lower for some settings of the main-coil tap. Minimum impedance levels range from 400 to 1000 Ω. Relationships between bandwidth and impedance level for two values of trap inductance are shown in Fig 41.

Figure 41—Bandwidths Obtainable with Wide-Band Traps of 0.265 mH and 0.53 mH Inductance and Different Impedance Levels 5.1.5.6 Untuned Line Inductors Line inductors are available having inductance ratings from 0.53 to 2.65 mH with a variety of intermediate values. A line inductor has a reactive impedance which varies directly with frequency and is directly proportional to inductance. 5.1.6 Power-Frequency and Mechanical Characteristics of Line Traps The power-frequency and mechanical characteristics which must be met in the design and manufacture of line traps as prescribed by the appropriate American National Standards.7 Table 16 lists continuous current ratings and their related thermal and mechanical ratings.

7More

detailed information will be found in new publications now in preparation. See 2.3 footnote 2

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Table 16—ANSI Current Ratings Continuous Current Rating 1-Second Thermal Fault Current Rating (amperes rms)

Mechanical Current Rating (amperes peak)

400

15 000

38 250

800

20 000

51 000

1200

36 000

91 800

1600

44 000

112 200

2000

63 000

160 650

3000

63 000

160 650

4000

80 000

204 000

Continuous Current Rating (amperes rms)

Line traps can be mounted in either vertical or horizontal position. However, the mechanical aspects can take various forms related to the particular situation. Manufacturers have varying capabilities to meet mounting requirements, and, therefore, each particular situation has to be resolved when it occurs. Seismic problems, abnormal wind conditions, environmental considerations, lack of space, etc, all contribute to the station designer's difficulty in determining how and where a line trap will be mounted.

5.2 Coupling Capacitors 5.2.1 General Efficient propagation of PLC signals for point-to-point communication requires an efficient transfer of carrier energy between the transmitter or receiver and the HV line. Coupling circuits have taken several forms over the years, but present practice utilizes a coupling capacitor as the major component. The requirements and essential characteristics of coupling capacitors have been standardized in ANSI C93.1-1972, [22]. 5.2.2 Construction Conventionally coupling capacitors are made using a paper/liquid dielectric system. Strips of Kraft paper are interleaved with strips of aluminum foil and wound into rolls. These rolls are flattened and stacked in a hollow porcelain insulator with external sheds. The rolls are connected in to provide a large voltage Withstand capability. In order to provide different capacitance values for a specified voltage rating, rolls with different cross-sectional areas are used. (This also requires different porcelain diameters.) When all rolls are in place, the capacitor unit is filled with a suitable fluid and sealed. The combination of the paper and the liquid forms a reliable dielectric system. The capacitor units are mounted on a metal base housing. This base unit usually contains a drainage coil to provide a safe low-impedance path to ground for power-frequency current, but has a high impedance to carrier-frequency signals. Protective gaps limit transient voltages, and a grounding switch is provided to eliminate high potential from the carrier lead during maintenance or repair work. Single capacitor units are available for line-to-line voltages in the range from 34 to 161 kV. Where larger ratings are required, combinations of single units are stacked to provide the necessary rating. For a given type of coupling capacitor the value of capacitance is inversely proportional to the rated voltage. Table 17 lists the approximate range of coupling capacitances available for various voltage classes.

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Table 17—Range of Coupling Capacitor Sizes Voltage Class (kV)

Capacitance Range (µF)

34

0.004–0.010

46

0.004–0.015

69

0.003–0.015

92

0.002–0.020

115

0.0019–0.020

138

0.001–0.016

161

0.0012–0.014

230

0.0009–0.010

287

0.0006–0.007

345

0.0005–0.006

500

0.0014–0.005

765

0.0023–0.005

5.2.3 Application A coupling capacitor is operated with a line tuner to form a resonant circuit or a band-pass or high-pass filter at carrier frequencies. Reference should be made to 5.3 for a discussion of various types of tuners. The bandwidth available for a bandpass tuner/coupling capacitor combination is proportional to the value of the coupling capacitance for a specified GMF. A single coupling capacitor provides for a phase-to-ground feed of the carrier signal on one phase of a power line. Where the application requires the the signal be fed to more than one phase, additional coupling capacitors are required. 5.2.4 Coupling Capacitor Voltage Transformer (CCVT) In some applications an additional function is performed by the coupling capacitor in conjunction with additional circuitry in the base housing. The coupling capacitor is utilized as a power-frequency voltage divider with a tap being brought out of the bottom capacitor unit. This tap is connected to a reactor/transformer combination (also protective gaps and other components) which converts the tap voltage to 115 V output when rated voltage is applied to the capacitor stack. The CCVT provides potential that can be used for line synchronism checks or as inputs for protective relays and metering equipment. The normal carrier-frequency operation of the coupling capacitor is unaffected by the CCVT circuitry. (See ANSI C932-1978 , [23].)

5.3 Line Tuners and Bypasses 5.3.1 Tuner Types Associated with each coupling capacitor, in its function of coupling carrier circuits into the power line or cable, certain auxiliary apparatus, generally known as line-tuning equipment, is required. The line-tuning equipment is connected in series with the HV coupling capacitor to provide for the following: Copyright © 1998 IEEE All Rights Reserved

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1) 2) 3) 4) 5)

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Efficient coupling of carrier transmitters and receivers to the power line or cable Carrier bypasses around power transformers, switches, and other discontinuities in the power line at carrier frequencies Attenuation of undesired signals Protection of personnel and electronic equipment from the high voltage of the power line Impedance matching of carrier equipment to the power line or cable

Line-tuning equipment may be very simple or fairly complex, depending on the number of carrier signals to be passed through the coupling capacitor. The number of signals that can be coupled through a single coupling capacitor depends on the value of the capacitance, the signal frequencies, the signal bandwidths, and the complexity in the line-tuning equipment. The most commonly used line-tuning equipment is available in standard units, having various arrangements of variable inductors and capacitors which can be combined to form series and parallel resonant circuits for one or more channels in the carrier-frequency range. The types of tuners in use are the single-frequency resonant, two-frequency resonant, wide-band band pass, and wide-band high pass. 5.3.2 Resonant Tuners 5.3.2.1 General Resonant line tuners are assemblies containing elements which form series resonance with the reactance of the coupling capacitor at one or more frequencies. 5.3.2.2 Single-Frequency Line Tuners The simplest application of resonant tuning involves the coupling of one frequency between a single phase of the power line and ground (single-phase-to-ground coupling) using one coupling capacitor. An impedance-matching transformer and a line-tuning inductor comprise the basic elements of the tuner, as shown in Fig 42. A protective gap and grounding switch are contained in the tuner, and some also include a drainage coil and a compensating capacitor for operation at higher frequencies. These elements isolate the transmitter and receiver equipment from the 60 Hz power-line voltage, cancel (by series resonance) the inductive and capacitive reactances to obtain a low-resistance path for the carrier signal, and provide matching of the low-impedance transmitter/receiver equipment (and the associated coaxial cable) to the high impedance of the overhead transmission line. For power cable circuits, special low-ratio impedance matching transformers are used. Higher coupling losses are incurred when coupling to cable circuits because of the low ratio of power cable characteristic impedance to the series impedance of the coupling components.

Figure 42—Single-Frequency Line Tuner

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Losses caused by the line tuner and coupling capacitor in combination vary with the frequency, the capacitance of the coupling capacitor, the line-tuner characteristics, and the apparant impedance of the power circuit. A graphical method of determining the losses introduced by a single-frequency line tuner is provided by Fig 43.

Figure 43—Coupling Loss with a Resonant Single-Frequency Line Tuner. Locate Frequency on Scale 1, Move Right to Intersection with Curve for Coupling Capacitance, then Downward to Point Opposite Load Resistance on Scale 2. Scales 1 and 3 Can Be Used to Find Reactance of Couping Capacitors The single-frequency phase-to-ground coupling loss for a 138 kV power cable circuit appears in Fig 44. Quite often more than one frequency must be coupled to a line using the same coupling capacitor in combination vary with the fre-too great, the single-frequency tuner can be operated off resonance for all but one of the frequencies. An off-resonance tuner introduces a reactive component into the transmission path, but in a number of applications to the resultant additional loss can be tolerated. Provided total attenuation is not too great, a narrow-band frequency-shift type channel can operate practically unaffected. It is common practice to run a number of such channels closely spaced through a single-frequency tuner.

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Figure 44—Total Coupling Loss of a 0.005 µF Coupling Capacitor and a Single-Frequency Phase-to-Ground Line Tuner on a 138 kV Power Cable 5.3.2.3 Two-Frequency Line Tuners Where two signals with substantial frequency separation are to be coupled to the line, a two-frequency tuner should be used. The circuit is shown in Fig 45. To permit independent tuning, parallel resonant trap units are inserted in each path and tuned to block the frequency of the opposite path. These elements introduce losses, but if the two frequencies are adequately separated, the loss in each trap unit is low at the series-resonant path frequency.

Figure 45—Two-Frequency Line Tuner. Protector Unit, Compensating Capacitor, and Housing Have Been Omitted for Clarity When using a two-frequency resonant line-tuner to couple two or more frequencies onto a line, adequate frequency spacing must be observed. The recommended spacing is 25% of the higher frequency or 25 kHz, whichever is greater. In general the coupling loss of a two-frequency tuner is about twice that of a single-frequency tuner (other factors being equal). Conversely, the bandwidth of each path is approximately half that of a single-frequency tuner, provided the frequency-spacing rules are observed.

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If three resonant paths are desired, the complexity of the tuner is increased considerably. A three-frequency tuner is shown in Fig 46 where each branch contains antiresonant trap units tuned to the respective resonant frequencies carried by the other two branches of the circuit. Because of this complexity, a two-frequency tuner generally is considered to be the practical upper limit for tach assemblies.

Figure 46—Three-Frequency Line Tuner 5.3.3 Wide-Band Tuners 5.3.3.1 General Multifrequency application problems encountered in the use of resonant tuners prompted the development of wideband coupling networks. The wide-band approach to coupling and tuning uses reactance elements to form band-pass or high-pass filter networks. Generally this type of tuner is essential when it is necessary to pass a broad band of frequencies with low loss over a single phase wire. An example of a broad passband requirement would be the application of a four-channel SSB system to the same phase of a power line. A four-channel system requires a 16 kHz bandwidth for both the send and the receive directions and is usually provided with a suitable guardband in between. This broad passband is normally not obtainable with resonant-type line tuners. 5.3.3.2 Band-Pass Line Tuners A band-pass tuner, together with its associated coupling capacitor, forms a band-pass filter as illustrated in Fig 47(a). The tuner assembly also normally includes an impedance-matching transformer (illustrated) and protective elements (not shown). The bandwidth which a band-pass tuner can provide is proportional to the coupling capacitance, the nominal design value of load impedance, and the square of the GMF.

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Figure 47—Wide-Band Line Tuners. (a) Band Pass. (b) High Pass. Fig 48 presents a graph from which the theoretical bandwidth for a band-pass filter can be determined for a nominal design impedance level of 350 Ω. Compromise settings (that is, the design of element values for an impedance level higher than the actual line impedance) can usually produce reasonably satisfactory performance over much wider bandwidths than those shown. With such settings, the bandwidth between frequency limits which actually produce a 3 dB maximum insertion loss under controlled test conditions is almost twice the bandwidth given by Fig 48. However, care should be exercised in utilizing this 3 dB bandwidth since a lower insertion loss may be required in some applications. Also, impedance fluctuations due to standing waves on the PLC channel can cause fairly severe insertion loss variations in a compromise-set tuner's frequency response. Table 18 presents the insertion loss of a typical band-pass line tuner near the GMF as a function of coupling capacitance, GMF, and line impedance. Band-pass line tuners are available as either fixed wide band or adjustable wide band. The fixed wide band tuner provides a somewhat larger bandwidth than the adjustable wide band tuner at the expense of a somewhat larger nonuniform insertion loss across its passband. Adjustable wide-band tuners can provide a lower and more uniform insertion loss. Detailed information on the specific performance of either type tuner should be obtained from the manufacturer.

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Figure 48—Theoretical Bandwidth of a Band-Pass Tuner (Designed for a Load Impedance of 350 Ω) as a Function of Its GMF. Compromise Settings Can Provide Wider Passbands Table 18—Attenuation of Center Frequecies of a Band-Pass Tuner Coupling Capacitance (µF) 0.01

0.005

0.003

0.002

0.0015

0.001

Geometric-Mean Frequency (kHz) 30 70 200 30 70 200 30 70 200 30 70 200 30 70 200 30 70 200

Loss (dB) with Equivalent Load Impedances Shown 200 Ω 400 Ω 800 Ω 1.2 0.8 0.7 0.8 0.7 0.6 0.7 0.65 0.6 1.6 1.15 0.8 1.05 0.8 0.7 0.8 0.7 0.7 2.3 1.3 0.95 1.4 0.95 0.7 1.0 0.7 0.6 3.2 1.8 1.3 1.6 1.15 0.9 1.2 0.9 0.8 3.9 2.3 1.6 2.2 1.4 1.0 1.4 1.0 0.8 6.0 3.1 2.2 3.2 1.7 1.5 1.9 1.1 1.0

5.3.3.3 High-Pass Line Tuners A high-pass tuner, together with its associated coupling capacitor, forms a high-pass filter as illustrated in Fig 47(b). The complete tuner network includes an impedance-matching transformer and protective elements. High-pass tuners are frequently designed for mounting inside the base of the coupling capacitor. Copyright © 1998 IEEE All Rights Reserved

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This type of tuner will pass a band of frequencies whose low-frequency cutoff limit is determined by the capacitance of its associated coupling capacitor and the nominal design value of load impedance. The high-pass tuner has no theoretical upper frequency limit. In performance this tuner frequently exhibits substantial input impedance variations in the lower part of its passband. Generally the high-pass tuner is recommended where multifrequency wide-hand types of equipment are to be applied over a single phase wire and where higher frequencies can be utilized. For a specified coupling capacitance and impedance level, the low-frequency cutoff for the highpass tuner is not as low as the low-frequency limit available with a band-pass tuner. Fig 49 shows the theoretical cutoff frequencies of a high-pass tuner which would be obtained when the design values of nominal impedance exactly match the load impedance. As with the band-pass tuner, more generous passbands may be obtained by using compromise settings of the element values. Fig 50 shows some typical responses which occur when compromise-set tuners are operated into low impedances. Note that these computed curves represent performance with a constant pure resistance load. Performance with low values of actual line impedance may have severe periodic variations due to standing waves. 5.3.4 Bypasses 5.3.4.1 General PLC systems frequently require the bypassing of signals around a discontinuity, such as an open circuit breaker, a transformer, or lines of different voltages. A bypass may involve the transfer of energy for a single channel, or it may involve the transfer of a wide band of carrier frequencies. Bypassing can be accomplished using line tuning networks between the base leads of coupling capacitors connected to the line at each end of the discontinuity involved. Where the coupling capacitors are separated by less than about 100 ft (30 m), the tuning arrangement can be handled as a short bypass. In this case, less equipment is required than for the long bypass, which is required where greater separations are involved. It is common practice by most users to install line traps on both sides of the discontinuity to be bypassed in order to minimize unnecessary shunt losses.

Figure 49—Theoretical Cutoff Frequencies of a High-Pass Tuner as a Function of Coupling Capacitor Size for Various Load Impedances

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Figure 50—Insertion Loss of a High-Pass Line Tuner with Various Compromise Adjustments. Cutoff Frequencies Are: (a) 50 kHz; (b) 80 kHz; (c) 120 kHz 5.3.4.2 Short Bypasses In applications involving a short bypass, a simplified arrange ment with one line tuner and high-impedance lead-in wire can be used. The high-impedance wire should be kept as short as practical to minimize carrier losses. Typical short bypasses using a single-frequency tuner, a two-frequency tuner, and a band-pass tuner are shown in Fig 51. Two sets of protective elements (not shown) are included, one associated with each coupling capacitor lead. There are no impedance-matching transformers. 5.3.4.3 Long Bypasses The most commonly used long bypass arrangement is shown in Fig 52 in which a normal line tuner is associated with each coupling capacitor and a coaxial cable is used as the link between them. With this arrangement the distance between the coupling capacitors can be as much as several thousand feet, depending upon the losses in the coaxial cable at the frequency involved. If a carrier transmitter or receiver terminal is to be coupled into the power line at the bypass point, a coaxial cable can be run from either tuner back to the equipment. This will allow signals to be coupled to or from the lines. In some installations a separate coaxial cable is run from each tuner to the local terminal, and the direct bypass cable is omitted. In either case a 2:1 impedance mismatch will be present.

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Figure 51—Typical Short Bypasses. (a) Single Frequency. (b) One Version of Two-Frequency. (c) Bandpass.

Figure 52—Typical Long (Normal) Bypass for a Phase-to-Ground PLC Channel

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5.4 Coaxial Cables and Tuning Leads 5.4.1 Introduction Coaxial cables and leads are an integral part of the coupling and tuning portions of a PLC channel. Three specific types of conductors are normally used: insulated single conductor, coaxial cable, and triaxial cable. 5.4.2 Insulated Single Conductors An insulated single conductor is used to connect a coupling capacitor to line-tuning equipment or outdoor transmitting and receiving equipment. It can also be used as the interconnecting lead for short bypasses. Bare conductors and coaxial cables should be avoided for these applications, since one will introduce excessive leakage currents and the other will introduce excessive stray capacitance. Since a single conductor is at a highimpedance point when connected between a coupling capacitor and a line tuner, stray capacitance to ground and leakage currents can affect the coupling circuit performance. The stray capacitance can cause a reduction in bandwidth, and the leakage currents can cause a loss in carrier power. To reduce stray capacitance and leakage currents, either of the following methods may be used: 1)

2)

An insulated single conductor should be run as directly as possible between its required terminations. It should be mounted on insulators and fed through bushings at each end. The conductor insulation should be unbroken between its ends to maintain low leakage. A neoprene-insulated single conductor can be idled in a nonmagnetic flexible metal conduit which is sheathed in a vinyl jacket. The jacket should be isolated from the shield with Teflon washers spaced about 6 in (15 cm) apart. If the conductor has a significant portion of its length outside the shield, it should be mounted on insulators and fed through bushings at its ends as in (1).

A typical insulated carrier lead, 0.48 inches (1.22cm) in diameter, consists of a single AWG No 8, 19-strand conductor having robber insulation and a neoprene outer sheath. 5.4.3 Coaxial Cables This type of cable is normally used for a low-impedance interconnection between a line tuner and a transmitter/ receiver or between line tuners in a long bypass. It is sometimes used between an impedance-matching transformer in a coupling capacitor base and a transmitter/receiver. In these applications the copper braid (shield) which forms the outer conductor of the cable should be adequately grounded at the transmitter/receiver end only (or at only one end of a bypass). If both shield ends are grounded, large sure currents can flow under certain conditions, causing saturation of the impedance-matching transformer and resulting in an inoperative carrier channel. A representative type of coaxial cable, RG-8/U, has an outer diameter of 0.405 in (1.029 cm). Its center conductor is AWG No 12 (7 strands of No 21 copper wire) surrounded by polyethylene insulation. A braided shield made of AWG No 36 copper strands forms the outer concentric conductor. The outer covering is a polyvinyl plastic jacket. RG-8/U coaxial cable has a nominal characteristic impedance of 52 Ω. Its attenuation as a function of frequency is given in 3.5.6.3 (Fig 8). 5.4.4 Triaxial Cables On transmission lines operating at voltages greater than 230 kV, triaxial cable is normally used instead of coaxial cable. This cable provides an additional heavy shield which does not carry signal currents. The outer shield is capable of carrying large induced surge currents under fault conditions and is grounded at both ends. This arrangement Copyright © 1998 IEEE All Rights Reserved

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provides very effective shielding against both magnetic and electrostatic induction so that surges induced in the signal leads are small. Construction of a triaxial cable in effect consists of adding more insulation and another shield surrounding a stranded coaxial cable. For example, the outer plastic jacket of an RG-8/U coaxial cable might be omitted and replaced by a layer of polyethylene insulation and another shield of braided copper and, finally, the outer jacket of polyvinyl chloride. In other versions the outer shield is made of lead and no outer plastic jacket is provided at all. 5.4.5 Insulation Requirements In some cable installations, specifications may call for safe operation under high-temperature conditions. Polyethylene has a maximum service temperature of 80 °C, and, therefore, it must be replaced by other dielectrics where hightemperature operation is required. Chlorosulfonated polyethylene and silicone rubber compounds are examples of materials that have been used in high-temperature cables.

5.5 Hybrids and Separation Filters 5.5.1 Introduction 5.5.1.1 General Increasing use is being made of wide-band coupling methods by which all carrier facilities connected to a given power line can be served by a single coaxial cable. Whenever two or more carrier terminals are connected to such a wide-band path, some form of external isolation is usually required to prevent undesirable interaction such as the following. 5.5.1.2 High-Level Intermodulation Distortions The output selectivity of carrier transmitters is not usually sufficient to prevent intermodulation when two or more transmitters are connected to operate in parallel on adjacent frequencies. 5.5.1.3 Low-Level Intermodulation Distortion Sometimes protective devices such as shunt diodes are connected between a carrier receiver input filter and its line terminals. Because of nonlinearity in the diodes, out-of-band signals can create intermodulation products which cause in-band interference. 5.5.1.4 Bridging Losses Most carrier receivers have sufficiently high input impedance that excessive bridging losses are avoided without special preventive means. However, the output circuitry of carrier transmitters does not usually provide an out-of-band impedance sufficiently high that other frequencies can be used indiscriminately. 5.5.1.5 Transient Influence Receiver input filters are generally not capable of carrying high transient currents such as those produced by disconnect switch arcs. Inductive elements in the filter are frequently saturated by such high currents, and the filter momentarily passes high-level broad-band noise. This can cause overload and saturation of the receiver's electronic circuitry.

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5.5.2 Separation Filters The most often used separation filter circuit is the simple series resonant L/C unit shown in Fig 53. Such units are available as standard items from manufacturers of line-tuning equipment.

Figure 53—Frequency Separation with Series L/C Units The bandwidth coupled into a 50-Ω coaxial cable circuit through a typical series L/C unit is dependent upon the L:C ratio and the resonant frequency at which the unit is tuned. For example, capacitance settings may be provided to permit 3 dB bandwidths from approximately 1 to 8 kHz when tuned to 50 kHz or from 17 to 604 kHz when tuned at 400 kHz. In general, for a given center frequency, a higher L:C ratio provides more selectivity at the expense of a slightly higher insertion loss. The minimum separation between two L/C units should be 25 kHz or 10% of the higher frequency. The range of Q's available with L/C units is from 60 to 280. More specific information on the selectivity characteristics is available from the manufacturers. When the process of frequency separation requires more selectivity or higher out-of-band impedance than simple series L/C units can provide, more elaborate filters may be used. As an example, separation of the whole coupled frequency band into high-, medium-, and low-frequency subbands may be accomplished with a circuit of the form shown in Fig 54.

Figure 54—Frequency Separation with Coordinated Half-Section Filters Special-purpose filters used for the damping of disconnect-switch disturbances and other high-level transients require high out-of-band rejection, particularly to frequencies above the carrier band. Predominant frequencies in arc noises are determined largely by the electrical length of switchyard bus segments connected to the arcing switch. These range generally from around 500 kHz to about 2 MHz.

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Some benefits can be obtained from the use of low-pass filters connected in series with the main coaxial cable. However, the most effective reduction of high-noise influence on a given receiver results from the use of a highly selective band-pass filter in series with the coaxial cable connected to the receiver or receiver assembly. This is particularly helpful for FSK transferred-trip applications. Design values of highly selective constant-K band-pass filters include extremely-low-inductance shunt elements (for example, less than 1.0 µH). The availability of such low-inductance units of satisfactory current rating is limited. The incorporation of step-up impedance transformation into the filter design can provide relief from this problem by changing the require shunt inductance to a higher, more practical value. If the ratio of impedance transformation is selected properly, a half-section filter can be terminated directly into a high-impedance receiver. Such high-impedance circuits should be placed near the receiver. These filters may be constructed using standard modular line tuner components in arrangements which the user can design based on textbook methods. Standard components are well suited for the construction of special filters because the cores in the variable inductors are specifically designed to let the coil carry relatively high currents with a minimum of saturation and detuning. Some manufacturers offer custom-designed filter assemblies and can provide technical assistance or advice concerning what is needed for a given situation. 5.3.3 Hybrids When a carrier transmitter operates at a frequency very close to that of another transmitter or receiver, the selectivity requirements are often so demanding that ordinary filters cannot provide enough isolation to prevent intermodulation and interference. In such situations, a carrier-frequency hybrid can usually provide the required degree of isolation between any two carrier terminals. Hybrids are available in several configurations. The most frequently used types are the balanced resistive, skewed, and balanced reactive hybrids. The hybrid transformer is basically a bridge circuit. In a typical application for a balanced resistive hybrid, two transmitters are fed into separate input terminals of the hybrid such that their combined signals pass from the hybrid to the line. With proper balance in the hybrid, a high degree of isolation is achieved between the two equipment (input) ports being fed by the transmitters. This isolation is called “transhybrid loss” and is typically 20 dB or more throughout a usable band of frequencies. Signals entering the line port of the hybrid split their power equally coming out of the two equipment ports. A loss of slightly more than 3 dB is encountered by signals passing through a balanced hybrid. The circuit for a balanced resistive hybrid is shown in Fig 55.

Figure 55—Balanced Resistive Hybrid Where a transmitter and a receiver are to be coupled to the power line, a skewed hybrid is frequently used. This is an unsymmetrical device which favors its transmitter port. The transmitted signal is attenuated by only about 0.4 dB in passing through a skewed hybrid. Attenuation to the receiver port is usually about 12 dB, but this does not cause a problem since the SNR of the incoming signal is established at the line tuner and does not change in passing through the hybrid. A balanced reactive hybrid is used to interface with the line tuner. It provides a degree of reactive control such that an effective resistive load is presented to the hybrid, even though the input impedance of the line tuner has a reactive component. 72

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In a theoretical sense, balancing a hybrid requires that the balance network have an impedance characteristic which matches the line impedance quite closely at all frequencies within the range of interest. In practice, a corrective network is usually adjusted to cancel the reactive component of line impedance, and the transformer ratio is adjusted to take care of the magnitude. The precision of balance obtainable with an adjustable reactance hybrid (transhybrid loss between the two equipment ports) is sometimes greater than 60 dB at a single frequency. However, this degree of balance is not normally available over a significant band of frequencies and, in addition, it may deteriorate from time to time as line impedance changes with temperature and other variables. An example showing the interconnection of several hybrids is shown in Fig 56. For this example transmitters T1 and T2 lose 6.5–8 dB in reaching the coaxial cable to the line tuner, and transmitter T3 loses 3.5–4 dB. The receivers lose about 15.5 dB in signal strength between the coaxial cable and their inputs.

Figure 56—Hybrid Interconnections

6. Frequency Selection 6.1 Factors Influencing Selection The range of frequencies which most United States users regard as the PLC spectrum is from 30 kHz to 300 kHz. In Canada limited use is made of frequencies from 10 to 490 kHz. Many countries use frequencies as high as 500 kHz. Several factors must be considered in selecting suitable frequencies for a particular application and for utilizing this spectrum efficiently. These factors include the following: 1)

2)

Requirements for the new facilities a) Bandwidth and frequency spacing b) Isolation from sources of interference Existing frequencies a) Interference to other services

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b) Costs of frequency changes Frequency planning Line coupling and tuning Noise and line attenuation Power cable circuits

When considering new facilities for any relatively large carder system, the first and usually most difficult consideration is determining the frequencies of existing facilities already operating in the spectrum. Before making a preliminary frequency study, however, the requirements of the desired new facilities must be established and borne in mind.

6.2 Requirements of New Facilities 6.2.1 Bandwidth and Frequency Spacing Spacing requirements for relaying channels are specified by the minimum frequency separation allowed between the center frequencies of the channels. This applies to both on-off channels and FSK transferred-trip channels. Auxiliary voice channels, sometimes used with relaying channels, require additional frequency spectrum. For SSB voice channels the minimum frequency separation is specified from the band edge of one channel to the nearest band edge of the adjacent channel. Spacing between relaying and SSB channels is specified from the center frequency of the relaying channel to the nearest band edge of the SSB channel. Where minimum frequency separation between two dissimilar channels has not been specified, the larger of the minimum separations individually recommended for the two types of equipment should be used. Table 19 lists representative frequency separation requirements for several equipment types. The required separation is presented for 15 dB of minimum external isolation. 6.2.2 Isolation from Sources of Interference The minimum spacing requirements of 6.2.1 relate generally to isolation between carrier services on the same line. Some guidelines for avoiding interference from sources on other lines and in other stations are discussed in the following paragraphs. A common guideline for assigning relaying frequencies is to use two-line-section separation to provide sufficient isolation for channels on the same frequency. This provides the cross-station attenuation of three buses, which is about 45 dB. The guide line suggesting two-line-section separation is not always appropriate. For example, some utilities apply a three-section separation for direct transferred-trip applications. When substantial doubt exists, isolation measurements should be made. Where two power lines are near each other and parallel for a few miles, there is signal coupling between the lines. Consequently channels operating on the same frequency may interfere with each other. Fig 57 shows calculated results for two representative cases of coupling between power lines. In areas where the carrier density is high, it will sometimes be very difficult to find space in the frequency spectrum for new channels by use of ordinary guidelines and rules. In such situations it may be necessary to resort to some means of providing greater isolation, so that frequencies can be “packed” more efficiently. Various methods can be used to increase isolation from interferring channels or abnormal noise sources. Interference from channels in other power-line sections can be reduced by using line traps to isolate carrier signals in one section from those in other sections. When a high degree of isolation is needed, line traps may be used in all three phases. It may be necessary to use line traps and coupling capacitors to form bandstop filters in each phase.

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High speed

Medium speed

Low speed

2*

Without Voice

*Increase spacing 25% for each 25 kHz above 90 kHz.

SSB Voice

FSK

Wide band with voice

Wide band without voice

Narrow band with voice

Narrow band without voice

AM (on-off) line protection

Equipment Type

8

8

With Voice

Narrow Band

5

8

5

Without Voice

8

8

8

8

With Voice

Wide Band

AM (On-Off) Line Protection

0.5

8

5

8

2

Low Speed

1.5

1.5

8

5

8

2.5

Medium Speed

3

3

3

8

5

8

4.5

High Speed

FSK Unidirectional

4.5

3

1.5

Low Speed

4.5

3

3

Medium Speed

FSK Bidirectional

Table 19—Minimum Frequency Spacing in kHz Between PLC Terminals Coupled to the Same Power-Line Conductor for 15 dB of External Isolation

4.5

4.5

4.5

High Speed

4

4.5

4

4

8

4

8

4

SSB Voice

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Isolation between transmitters or between transmitters and receivers on adjacent channels on the same conductor can be achieved hybrids and filters (see 5.5). Isolation can also be achieved by coupling adjacent channel transmitters on different phase conductors of the same power line. This isolation may be substantial at the near (transmitting) end of the line but almost negligible at the distant (receiving) end. Coupling to different phases can often solve an otherwise difficult line-tuning problem. Carrier transmitters on the same frequency may sometimes be operated on different lines in the same station. The cross-station attenuation between two lines which are equipped with resonant traps will frequently exceed 40 dB at the center frequency.

Figure 57—Coupling Between Parallel Power Lines with Center Phase Separations. (a) 1000 ft (300 m); (b) 200 ft (60 m). SNR Is Based on the Assumption of Approximately Equal Signal Levels on Each Line

6.3 Existing Frequencies 6.3.1 Frequency Survey When new PLC channels are to be installed, a survey should be mede of frequencies already used within at least two line sections of each section in which new channels are to be installed. The survey should also include frequencies in use on any lines that parallel or cross the line section. When several new carrier services are to be added to a large system with complex interconnections, the frequency survey must also consider frequencies used by neighboring utilities. Most PLC users routinely exchange frequency information with other utilities. Users of PLC should be aware that there are many other users of frequencies in the same spectrum, including navigational radio facilities, and care is required to avoid interference with these users. The use of these frequencies for PLC applications in the United States is not licensed by the Federal Communications Commission (FCC). In Canada a form of licensing is granted by the Department of Communications (DOC) permitting use of frequencies to 490 kHz with a number of specific restrictions. The use of PLC frequencies in both countries is permitted on a strict noninterference basis.

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6.3.2 Frequency Changing Costs If it is impossible to find adequate space in the spectrum for new carrier channels, existing frequencies must be rearranged to provide spectrum space. Often it will be found that frequency changing costs are a significant part of the total price of adding new carrier facilities. When more than one frequency rearrangement scheme is being considered to accomplish the same purpose, an economic comparison is necessary. An estimate of frequency changing costs determined by the use may include: 1) 2) 3) 4) 5) 6)

Engineering and drawing changes New parts, such as crystals and filters Labor costs for retuning and testing transmitters and receivers Parts and labor for retuning of line-coupling system Transmission line and communication circuit outage costs Overheads and contingencies

6.4 Frequency Planning 6.4.1 General Considerations Time and effort devoted to long-range planning of carrier system growth can result in higher efficiency in the use of the available spectrum. Achieving a maximum frequency density on a carrier system should be a goal that is given its appropriate share of consideration each time new frequency assignments are made. A plan for approaching this goal should be maintained. To be effective, such a plan must be reevaluated whenever actual growth of the carrier system deviates from what has been anticipated. 6.4.2 Relay Channel Frequency Plans What might be an appropriate plan for one utility might not be appropriate for another. For example, if most lines are relatively short and low-frequency SNR is not a problem, it may be feasible to group relay channel frequencies in the lower part of the spectrum, thus reserving higher frequencies for SSB telephone systems, which usually require more bandwidth. Most utilities find that they can repeat the use of relay channel frequencies at several locations to good advantage (assuming proper isolation is maintained). When several channels are installed on a line section, the minimum frequency separation suggested in Table 19 should be considered. If different types of channels are to be used on the same line, several frequency combinations should be studied to see if a best arrangement exists. 6.4.3 SSB Telephone Channel Frequency Plans Typical SSB telephone channels in North America have a nominal bandwidth of 4 kHz per channel per direction. If possible, channels should be assigned to frequency bends between integer multiples of 4 kHz. Other than the wider spectrum requirements, no special rules are required for SSB telephone channels. The minimum frequency separations suggested in Table 19 should be considered.

6.5 Line Coupling and Tuning The limitations of the commonly used methods of coupling must be kept in mind during the process of frequency assignment. Any frequency grouping assigned on one line must fit into the coupled bands that can be provided on one or more phases by conventional coupling methods. For example, assume that four new channels are to be added and the preliminary frequency survey has shown that the only clear-channel frequencies available are 34, 58, 118 and 198 kHz. It is not feasible to couple these frequencies on a single phase-to-ground or phase-to-phase path. If two phases are available, one solution may be to use two-frequency tuning to couple 34 and 118 kHz on one phase and 58 Copyright © 1998 IEEE All Rights Reserved

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and 198 kHz on the other. The choices which must be considered in the acceptance or rejection of this arrangement would be the retuning costs and the relative difficulty of clearing other frequencies as opposed to the disadvantages of the proposed arrangement. One disadvantage is a higher cost for double-frequency tuning on two phases. A second disadvantage is a higher attenuation and poorer frequency response usually obtained on outer-phase-to-ground circuits as opposed to center-phase-to-ground due to modal conversion losses. Another economic consideration sometimes affecting frequency assignment is the cost of larger coupling capacitors or higher inductance line traps which can provide wider bandwidths.

6.6 Noise and Line Attenuation After the factors in a preliminary study have been considered, the additional items of noise and line attenuation should be considered. Noise on a power line is a decreasing function of frequency, whereas attenuation increases with frequency. The optimum frequency therefore depends on the combined effect of these frequency-dependent factors and may be different on any two lines. On horizontal single-circuit lines up to about 100 mi (160 km) long, the slope of the attenuation versus frequency function is fairly slight for adjacent-phase-to-phase and center-phase-ground coupled circuits. On the other hand, during adverse weather noise will be about 6–10 dB higher at 50 kHz than at 200 kHz. Under this condition, the best SNR for a given transmitter power may occur at frequencies such as 150 or 200 kHz. With longer lines the optimum frequency will be correspondingly lower. It will also be considerably lower on circuits with coupling which does not involve the center phase. For short lines the easiest and most economical means for increasing the SNR or received power level is to increase transmitted power. Where long lines are involved and 10 W or 100 W transmitters are required, a further significant increase in transmitted power is economically and practically undesirable. The compromise between increasing transmitted power and reducing channel loss is an important consideration over a substantial range of line lengths.

6.7 Power Cable Circuits Many factors have to be considered when selecting frequencies for use on power cable circuits. Among these are the type of cable, length of cable, transmitter power, return path (skid wire), and number of frequencies to be used. Because of high attenuation and the difficulty of obtaining adequate bandwidth, only frequencies between 30 and 70 kHz are normally used. Line losses in oil-filled pipe-type cables range from 1.5 to 2.5 dB/mi (about 1–1.5 dB/km) at 30 kHz and from 3 to 5 dB/mi (about 23 dB/km) at 70 kHz. Power cable circuits which exceed about 10 mi (16 km) in length will generally require transmitter equipment with a power output of 100 W.

7. Future Trends 7.1 Introduction The trend of the future in PLC will see improvements in the areas of equipment design, transmission efficiency, and applications. It is anticipated that these improvements will tend to be of an evolutionary nature rather than of a revolutionary one.

7.2 Electronic Equipment In the area of electronic equipment, improved equipment designs utilizing more sophisticated circuits and techniques made possible by integrated circuits can be expected. This will enable the production of smaller, less costly, more reliable, and more versatile equipment.

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In the area of transmission equipment it is expected that transmitters and amplifiers with greater power outputs will become available for use in areas requiring greater signal strength. It is anticipated that some improvement in transmitter frequency stability and reliability will also be attained. In the area of receiving equipment it is anticipated that circuits using more sophisticated techniques will be utilized in separating desire signals from incoming noise, and more effective noise monitoring means will be extensively used. While it is also anticipated that receivers of greater sensitivity will be available, this will not materially increase the effective range of carrier equipment, unless improvements are also made in reducing line noise, since present receivers are usually noise limited and not signal limited. It is anticipated that the overall selectivity of receivers will be improved using new triter techniques. The circuits available in integrated form should enable new types of signal discriminators and demodulators to be utilized. In addition, the availability of large-scale digital integrated circuits will increase the use of logic in the receivers to improve their overall security and dependability as well as to reduce the cost of receiver and transmitter designs.

7.3 System Improvements At the present time, limitations in the range of PLC channels are due to insufficient SNRs at the carrier receivers. To increase the range, increased levels of signal must be coupled to the line. While this can be accomplished with increased transmitter power levels, the improvements along this line are limited, since it would be very costly to increase power significantly above the 100 W level now available. It is anticipated that the increase in signal level will be accomplished with improvements in coupling efficiency, such as greater utilization of mode 1 coupling. A development being considered is the intrabundle channel which utilizes insulated wires in one line phase for the transmission of carrier signals. Another promising possibility is a separately suspended coaxial cable for the exclusive use of carrier communications. Other improvements to be expected are the development and use of more efficient shielded signal cables such as improved coaxial cable, triaxal cable, and video cable pairs.

7.4 Applications In the area of application it is expected that PLCs will see increased use in their present applications. They may make more widespread use of frequencies above 300 kHz. More sophisticated protective relaying systems will utilize PLCs to a greater extent than before. Greater use will be made of the transmission of digitally encoded information for data and control purposes. PLC applications are expanding from HV transmission lines to low-voltage power distribution networks, where it will be used for automated meter reading, revenue billing, selective load shedding, and control of distribution system operations. New areas of application not even thought of at the present time will probably occur. In summary, it can be said that the explosive use of communications affecting all areas of the frequency spectrum will naturally extend to the PLC communications field.

8. Bibliography AIEE COMMITTEE REPORT Report of Methods of Measurements for the Application of Power Line Carrier, AIEE Transactions on Power Apparatus end System, vol 80, Feb 1962, pp 1046–1052. ALSLEBEN, E. Experimental Values of the Characteristics Determining the Behavior of High-Voltage Power Networks at the Frequencies Used by Carrier Current Communication Circuits, CIGRE Paper 340, 1958.

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Applied Protective Relaying, Westinghouse Electric Corporation, 1976. COMERFORD, T. J. Carrier Signal Attenuation Measurements on Pipe Type Cable, presented at the Pennsylvania Electric Association Relay Committee Meeting, Bedford Springs, PA, May 23–24, 1974. EHV Transmission Line Reference Book, New York, Edison Electric Institute, 1968. GROSSER, E. H., and STEEVE, E. J. Isolated Phase Pipe-Type Cable in Aluminum Fiberglass Pipe, IEEE Transactions on Power Apparatus and Systems, vol PAS-92, Mar/Apr 1973, pp 484–493. HAMSHER, D. H., Ed. Communication Systems Engineering Handbook. New York: McGraw-Hill, 1967. HASLER, E. F. The Design and Predicted Performance of Intrabundle Insulation, IEEE Transactions on Power Apparatus and Systems,vol PAS-94, Mer/Apr 1975, pp 338–343. JOHNSON, W. C. Transmission Lines and Networks. New York: McGraw-Hill, 1950. JONES, D. E. Operation of a Power Line Carrier System During Sustained Line Faults, AIEE Transactions on Power Apparatus and Systems, vol 79, Aug 1960, pp 556–560. LaFOREST, J. J., BARETSKY, M., JR, and MacCARTHY, D. D. Radio-Noise Levels of EHV Transmission Lines Based on Project EHV Research, IEEE Transactions on Power Apparatus and Systems, vol PAS-85, Dec 1966, pp 1213–1230. MATICK, R. E. Transmission Lines for Digital and Communication Networks, New York, McGraw-Hill, 1969, pp 268–309. PEEK, F. W. JR. Dielectric Phenomena in High. Voltage Engineering, New York, McGraw-Hill, 1929, 3rd ed. PULLEN, F. D. Signal-Coupling System for Intrabundle Communication on High-Voltage Line, Electronic Letters, vol. 9, Mar 8, 1973. PULLEN, F. D. The Calculated Electromagnetic Fields Surrounding Carrier-Bearing Power Line Conductors, IEEE Transactions on Power Apparatus and System, vol PAS-94, Mar/Apr 1975, pp. 530–538. PULLEN, F. D. Wide Bandwidth Capabilities of Intrabundle Communication on High Voltage Lines, IEEE Transactions on Power Apparatus and Systems, vol PAS-94, Mar/Apr 1975, pp 539–543. ROACH, C. L., and ALLEN, G. Y. R. Protection of Communication Circuits Serving Electric Power Systems, CIGRE Paper 308, 1966. ROBERTSON, A. N. 765-kV Power Line Carrier Communications, Parts I, II, and III, IEEE Transactions on Power Apparatus and Systems, vol PAS-91, Mar/Apr 1972, pp 575–591. Reference Data for Radio Engineers, New York, Howard W. Sams, 6th ed. SAKRISON, D. J. Communication Theory: Transmission of Waveforms and Digital Information, New York, Wiley, 1968. SCHWARTZ, M. Information Transmission, Modulation, and Noise, New York, McGraw-Hill, 1959. WOHLGEMUTH, D. G., GILLIES, D. A., and DIETRICH, R. E. Carrier-Current Transfer-Trip-Relaying Field Testa and Operation Experience, AIEE Transactions on Power Apparatus and Systems, vol 81, Aug 1962, pp 225–235.

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