Trilogy of Magnetics: Design Guide for EMI Filter Design, SMPS & RF Circuits [5 ed.] 9783899291575, 9783899294002

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TRILOGY

OF MAGNETICS

TRILOGY

OF MAGNETICS

DESIGN GUIDE for EMI Filter Design, SMPS & RF Circuits

IMPRINT PUBLISHER Würth Elektronik eiSos GmbH & Co. KG Max-Eyth-Str. 1 · 74638 Waldenburg · Germany Tel. +49 (0) 79 42 945-0 · Fax +49 (0) 79 42 945-400 [email protected] · www.we-online.com R E A L I S AT I O N Jana Haisch AUTHORS Dr. Thomas Brander // Alexander Gerfer // Bernhard Rall // Heinz Zenkner I L L U S T R A T I O N S // MZ Dokuments ∙ 86156 Augsburg ∙ Germany G R A P H I C D E S I G N // DIE NECKARPRINZEN ∙ 74072 Heilbronn ∙ Germany T Y P E S E T T I N G // Ideen im Kopf ∙ 75050 Gemmingen ∙ Germany P D F R E F I N I N G // Zeilenwert® GmbH ∙ 07407 Rudolstadt ∙ Germany P U B L I S H I N G H O U S E // Swiridoff Verlag ∙ 74653 Künzelsau ∙ Germany EDITIONS 1st edition January 2001 2nd extended and revised edition January 2002 3rd extended and revised edition November 2005 4th extended and revised edition April 2009 5th extended and revised edition November 2018 I S B N 978-3-89929-157-5 I S B N ( E - B O O K ) 978-3-89929-400-2

Content BASIC PRINCIPLES

11 – 272

The most important laws and foundations of inductive components, equivalent circuit diagrams and simulation models give the reader a basic knowledge of electronics.

COMPONENTS

273 – 474

This chapter introduces inductive components and their special properties and areas of use. All relevant components are explained, from EMC components and inductors to transformers, RF components, circuit protection components, shielding materials and capacitors.

A P P L I CAT I O N S

475 – 788

In this chapter, the reader will find a comprehensive overview of the principle of filter circuits, circuitry and numerous industrial applications that are explained in detail based on original examples.

APPENDICES A technical dictionary and alphabetical key word index for quick searches complete this book.

This book includes references to different simulation libraries and design software. To access all of these tools, please visit www.we-online.com/toolbox.

789 – 826

Preface

Dear Readers, Inductive components in electronic circuit design – whether for voltage conversion, filtering, or to safeguard the EMC – are anything but trivial and can be a headache for many designers. As a technician or engineer, what do I need to consider when it comes to component selection? How do I find the best component for my application when there are so many versions to choose from? How important are the datasheet parameters? What are the differences between the core materials?

This widespread uncertainty and the largely lacking practical approach in training for selecting and using inductive components in circuit design was the reason we wrote a book. The great success and immense popularity of our application manual „Trilogy of Magnetics“ show that we were on the right track. You are currently holding the fifth completely revised and modernized version in your hands. Thousands of engineers have found support for tens of thousands of designs in this manual. This fills us with pride and gratitude.

The book focuses on the selection of components, circuitry and layout recommendations for a wide array of magnetics components, always keeping in mind an EMC point of view.

We strongly recommend that all readers pay particular attention to Chapter 1 – The Basics. Only if you understand, consistently observe and apply the basics, will you be able to select and use the right inductive components. In addition to these basics, we also address the topic of simulation. The first chapter has been considerably extended in the new edition and we have pooled some of the theory from other sections here for better understandability. This includes filter basics, which has been extended to higher frequencies, transformer equivalent circuits, updated Ethernet and Power-over-Ethernet, basics of switched-mode power supplies including feedback compensation, wireless power transmission and RF basics. In addition, new information on skin, proximity effect and AC losses is included.

We have extended the existing chapters or added new ones to reflect the growing importance of topics such as wireless power transmission, RF, energy harvesting, and the new SiC (silicon carbide) and GaN (gallium nitride) switching devices.

The components section lists all Würth Elektronik magnetic component families clearly in a compact format, always followed by a page with their most important features and characteristics. This allows you to browse quickly the entire range of available components and their general specifications.

At this point I would also like to thank the external authors from our circle of customers and from well-known manufacturers. Their experience and solutions in basic principles and circuit technology are incorporated here and will certainly be of great benefit to you. The trilogy is rounded off by the explanation of important technical terms. Thus, you have a compact and practical reference book for your daily work, but also for teaching, studies and research.

As always, we look forward to receiving your suggestions and hope you enjoy the „Trilogy of Magnetics“.

Sincerely yours

Alexander Gerfer CTO Würth Elektronik eiSos Group

Thank you! Special thanks for their constructive assistance go to: Michael Bairanzade, ON Semiconductor, Toulouse, France Dipl.-Ing. (FH) Markus Braun, Fujitsu Siemens Computers GmbH, Augsburg, Germany Dipl.-Ing. (FH) Hans Dieter Frank, Würth Elektronik iBE GmbH, Thyrnau, Germany Dipl.-Ing. (FH) Ralf Frank, HF-Entwicklung Frank, Salem, Germany Dipl.-Ing. (FH) Andreas Merkle, NewTec GmbH, Pfaffenhofen a. d. Roth, Germany Dr. Andreas Schiff, ICS GmbH, Tettnang, Germany Dipl.-Ing. (FH) Anestis Terzis, DaimlerChrysler, Ulm, Germany Andre Schwarz, Matrix Vision GmbH, Oppenweiler, Germany Ansgar Trotter, Honeywell GmbH, Schönaich, Germany Jörg Meyer, Iris GmbH, Berlin, Germany Matthias Mühle, Philips Medizinsysteme GmbH, Böblingen, Germany Jon Harper, Fairchild Semiconductor GmbH, Fürstenfeldbruck, Germany Power Integration, San Jose, USA Nils Dirks, Dirks Compliance Consulting, Herrsching, Germany Dean Huumala, Wurth Electronics Midcom Inc., Watertown, USA Nigel Smith, Texas Instruments, Freising, Germany Alain Lafuente, Würth Elektronik eiSos, Saint Priest Cedex, France Sébastien Chadal, Würth Elektronik eiSos, Saint Priest Cedex, France Jochen Baier, Würth Elektronik eiSos, Waldenburg, Germany Ralf Stiehler, Würth Elektronik eiSos, Waldenburg, Germany Lorandt Fölkel, Würth Elektronik eiSos, Waldenburg, Germany Rudolf Hauser, Supertex Inc., Gräfelfing, Germany George Slama, Wurth Electronics Midcom Inc., Watertown, USA Stefan Hellwig, Würth Elektronik eiSos, Waldenburg, Germany

Suggestions and criticisms on this book gratefully received by: Würth Elektronik eiSos GmbH & Co. KG Alexander Gerfer Tel.: +49 (0) 79 42 945 - 0 Email: [email protected]

All rights reserved © Würth Elektronik eiSos GmbH & Co. KG, Waldenburg, November 2018 The work and all of its parts are copyrighted. Any use outside the narrow limitations of copyright law without approval of the publishing house is not permitted and is liable to prosecution. This applies in particular to copying, translating, microfilming, other types of processing and to storing and processing in electronic systems. This also applies to the utilization of individual illustrations and text excerpts.

The Authors Bernhard Rall, born in 1928, studied at the Technical University of Hanover after the war and imprisonment. In 1955, Bernhard Rall begun at the Telefunken Research Institute, which later belonged to AEG and Daimler. Mr. Rall left the institute as Chief Engineer in order to carry out his own research on vehicle EMC within the same establishment. His interest lay in the ingenious, practical solution of complex problems. Bernard Rall contributed to this textbook with the description and measurements performed in the chapter “Equivalent circuits and simulation models”, as well through practical examples of power noise suppression using ferrites and in the dimensioning and simulation of a coil filter.

Dr.-Ing. Heinz Zenkner studied electrical engineering with a focus on telecommunication and high frequency engineering and obtained his PhD in this field. He has been a officialy appointed and sworn expert for EMC for many years. In addition to numerous scientific publications, he frequently contributes as an author in many works on EMC. Mr. Zenkner has worked at various Universities, at the German Chamber of Industry and Commerce (IHK) and at numerous seminars. He has been involved with industrial electronics for years now, from the first idea of product through to series production. His special interest lies in Wireless Energy Transfer, to which he has developed and patented his own theoretical and practical concepts.

Alexander Gerfer, born 1965, worked in the field of research and development for precision measuring instruments following his training as a radio and television technician. This was followed by a degree on electrical engineering at the Technical University of Cologne. While studying, Alexander Gerfer published numerous application circuits and construction guidelines from the field of consumer electronics. After his degree, he worked in electronic component distribution and from 1997 on, he was head of R&D department at Würth Elektronik. Today Alexander Gerfer is Managing Director and Chief Technology Officer of Würth Elektronik eiSos GmbH & Co. KG.

Dr. Thomas Brander, born 1970, graduated from the Ruprecht-Karls University of Heidelberg with a degree in physics before gaining his PHD on physical age determination of iron ores. He then worked as a developer for a transformer manufacturer where he designed a wide range of inductors and transformers for high power applications (up to 100 kW). Since 2004, he has been responsible for the product area of transformers at Würth Elektronik and is now Head of Product Management for EMC & Inductive Solutions.

Trilogy of Magnetics

Design Guide for EMI Filter Design, SMPS & RF Circuits

Basic principles

11

I Basic principles Part 1: Basic principles

12

1 1.1 1.1.1 1.1.2 1.1.3 1.2 1.3 1.4 1.5 1.6 1.6.1 1.6.2 1.7 1.7.1 1.7.2 1.8 1.8.1 1.9 1.9.1 1.9.2 1.9.3 1.9.4 1.9.5 1.10 1.11 1.12 1.13 2 2.1 2.2 2.3 2.4 3 3.1 3.2 3.3 3.4 3.4.1 3.5 3.6 4 4.1 4.2 4.3 4.4 4.5 4.6

Basic principles of inductive components . . . . . . . . . . . . . . . . . 14 Ampère’s Law and magnetic field strength H. . . . . . . . . . . . . . . . . . . 16 Straight current conductor. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 Toroidal coil. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 Long solenoid. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 Magnetic induction B . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 Magnetic flux Φ. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 Faraday’s law. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22 Core materials and their losses . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 Permeability µ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 Complex permeability. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32 Comparing core materials. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34 Inductance L . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36 Definition of inductance L. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36 Definition of the A L value. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38 Impedance Z . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39 Self-resonant frequency (SRF). . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41 Resistive losses. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41 Copper losses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42 Skin Effect. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42 Proximity Effect . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44 AC loss calculations. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45 Definition of quality factor Q . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49 Temperature behavior. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50 Rated current. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52 Saturation current. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52 Differentiating EMC ferrite ↔ inductor. . . . . . . . . . . . . . . . . . . . . . . 53 Equivalent circuits and simulation models. . . . . . . . . . . . . . . . . 54 The most important types of equivalent circuits. . . . . . . . . . . . . . . . . 55 EMC ferrite equivalent circuits. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 66 Simulation with LTspice. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74 Design of optimized EMC filters for real operating environments. . . . . 78 Filter Basics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 86 Filter circuits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 86 The principle of filtering, functionality and structure of filters . . . . . . . 87 The low-pass filter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 89 Filter circuitry. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 95 Filter ground reference. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 95 Symmetrical filter/common-mode filter. . . . . . . . . . . . . . . . . . . . . . . 102 Filters for frequencies over 500 MHz . . . . . . . . . . . . . . . . . . . . . . . . 112 Transformer Basics. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 115 Functionality of a transformer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 115 Parasitic Effects. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 119 Transformer: Parasitic parameters and equivalent circuit . . . . . . . . . . 121 Function and application areas of transformers. . . . . . . . . . . . . . . . . 130 Requirements for data and signal transformers. . . . . . . . . . . . . . . . . 130 A transformer’s effect on return loss. . . . . . . . . . . . . . . . . . . . . . . . 131

5 5.1 5.2 5.3 5.4 5.5 5.6 5.7 5.8 5.9 6 6.1 6.2 6.3 6.4 6.5 6.6 6.7 6.7.1 6.8 6.8.1 6.9 6.10 6.11 6.12 6.13 6.14 6.15 6.16 7 7.1 7.2 7.3 7.4 7.5 7.6 7.7 8 8.1 8.1.1 8.1.2 8.1.3 8.1.4 8.1.5 8.1.6 8.1.7 8.2 8.2.1 8.2.2

Ethernet and Power-over-Ethernet Basics . . . . . . . . . . . . . . . . . 148 The history of Ethernet . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 148 The OSI reference model. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 148 What is Ethernet (standard 802.3)? . . . . . . . . . . . . . . . . . . . . . . . . . 150 The different types of encoding for Ethernet . . . . . . . . . . . . . . . . . . . 158 Bob Smith termination . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 161 Power over Ethernet (PoE). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 163 Important safety points to consider. . . . . . . . . . . . . . . . . . . . . . . . . . 164 Infrastructure and signal integrity. . . . . . . . . . . . . . . . . . . . . . . . . . . 164 Power classes, supply voltage. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 167 Switched mode power supply (SMPS). . . . . . . . . . . . . . . . . . . . . 169 Basic circuits. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 169 Buck converter/step-down converter . . . . . . . . . . . . . . . . . . . . . . . . 170 Boost converter/step-up converter. . . . . . . . . . . . . . . . . . . . . . . . . . 180 SEPIC switching controller with low input ripple current. . . . . . . . . . . 184 Input filter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 186 Transformers in switched mode power supplies. . . . . . . . . . . . . . . . . 190 The flyback converter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 190 Quasi-resonant flyback. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 193 The forward converter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 195 Active clamp forward . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 197 The push-pull converter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 199 The half-bridge converter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 201 The full-bridge converter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 202 Isolated soft switching topologies. . . . . . . . . . . . . . . . . . . . . . . . . . . 203 LLC converters. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 203 Current transformers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 205 Gate drive transformers. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 206 An introductrion to frequency compensation. . . . . . . . . . . . . . . . . . . 208 Wireless Power Transfer Basics . . . . . . . . . . . . . . . . . . . . . . . . . 240 Transmission paths for wireless power transfer. . . . . . . . . . . . . . . . . 240 Basics. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 242 Construction and calculation of the resonant circuit. . . . . . . . . . . . . . 243 Coupling and efficiency. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 246 Shielding. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 248 EMC measurements. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 249 The dominating standards. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 250 RF basics. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 252 RF inductor characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 252 Inductance (L) and tolerance (%) . . . . . . . . . . . . . . . . . . . . . . . . . . . 252 Self-Resonant Frequency (SRF) . . . . . . . . . . . . . . . . . . . . . . . . . . . . 253 Quality factor (Q) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 255 DC resistance (R DC). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 256 Rated current (IR) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 256 Size. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 256 Structures. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 257 S-Parameters – Basic principles . . . . . . . . . . . . . . . . . . . . . . . . . . . 258 Basic theoretical principles . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 258 Conceptual design of a matching circuit by means of a Smith chart. . . 263

13

I Basic principles 1 Basic principles of inductive components Magnetism

The basis for understanding inductors is provided by magnetism and a few fundamental electromagnetic field laws, revealing clear and fundamental knowledge of inductors and ferrites. The most important phenomena and laws are from basic physics lessons:

Fig. 1.1: Bar magnet • Every magnet has a north and south pole (The earth is an enormous m ­ agnet!) • If an existing magnet is divided, a new one is created. The magnet created also has a north and south pole. This division can be performed down to the molecular level without losing the magnetic effect. • Every magnet is surrounded by a magnetic field, which is represented by the field line model. • Magnetic field lines are closed loops. They have neither a beginning nor end. • There are magnetizable materials (e.g. iron) and non-magnetizable m ­ aterials (e.g. aluminum). Ferromagnetic ­materials

14

The following analysis concerns a class of magnetizable materials, the ­ferromagnetic materials.

N

S

N

S N

S S

N

N S

N

N

N

N

N

S

N

S

N

N

N

S N

S

S

S

N

S

S

S

N

N

N

S

N S S

N S

N

S

N

S

S

N

N

S

S

N

S unordered

Fig. 1.2: Elementary magnets Every magnetic material is composed of a finite number of the smallest e­ lementary magnets, configured randomly in the unmagnetized state. Therefore the sum of the magnetic effect is zero. These orientate themselves under the influence of an external magnetic field.

N

S

N

S

N

S

N

S

N N N N

S S S S

N

S

N

S

N

S

N

S

N

S

N

S

N

S

N

S

ordered

Fig. 1.3: Saturation If all elementary magnets are oriented in the magnetic field, one speaks of saturation of the material.

Saturation 15

I Basic principles Once the externally applied magnetic field is removed, two effects can o­ ccur: a) The material becomes unmagnetized once again: One speaks of a soft magnetic ­material b) The material remains magnetized: One speaks of a hard magnetic material Ampère’s Law

1.1 Ampère’s Law and magnetic field strength H A magnetic field is created around an electric conductor passing a current. This magnetic force field is a vector quantity perpendicular to the generating current. Field lines represent the magnetic force field. For a current carrying conductor, these form closed concentric circles. Integrating in a counter clockwise direction along a field line, the H and each distance increment (dr) are always in the same direction. A complete circulation provides the magnetic boundary potential:

Fig. 1.4: Magnetic field strength H of a long conductor If several conductor currents pass through the enclosed area, the sum of the currents must be on the right side of the equation (observing their d­ irectional signs):



16



(1.1)

The magnetic field is measured by means of the magnetic field strength H and is defined by the currents generating the field. The field strength of any conductor configuration may be determined from the BiotSavart Law:

Biot-Savart Law

Fig. 1.5: BIOT SAVART’s Law: Describing the field strength outside a straight current conductor Accordingly, the small section (ds) of the conductor carrying current (I) m ­ akes the contribution:





(1.2)

where (a) is the angle between the direction of the line element (ds) and its connection (r) to the point (P) at which the field strength (dH) exists. The field strength (H) results from integrating over the entire length of the conductor.

Magnetic field strength H

The contour integral of H along a closed line is equal to the total c­ urrent through the area across this closed path. The magnetic field strength is given by the total current through the surface enclosed by the magnetic field line and the length of this field line. If it is assumed that the same currents flow in N discrete conductors, as in the case of a coil, the equation simplifies to:

(1.3)



N = number of conductors within the closed path l I = current per conductor The unit of magnetic field strength (H) is A/m.

17

I Basic principles Examples of typical conductor arrangements of the magnetic field strength (practical formula): 1.1.1 Straight current conductor Straight conductor

Fig. 1.6: Magnetic field strength (H) of a straight conductor Toroidal coil

18

1.1.2 Toroidal coil

Fig. 1.7: Magnetic field strength H of a toroidal coil

1.1.3 Long solenoid

Long solenoid

Fig. 1.8: Magnetic field strength H in a solenoid This illustration shows the size of the magnetic field strength (H) inside a long solenoid to be dependent on the current I, the coil length l and the winding turns N. Example: Ferrite sleeve 742 700 9 on a conductor with dc current of I = 10 A Sleeve dimensions: OD = 17.5 mm; ID = 9.5 mm; l = 28.5 mm (→ average diameter = 13.5 mm and ra = 6.75 mm)

Fig. 1.9: Field strength in air and in a ferrule

Field strength

Question: What size has the field strength H1 in air and field strength H2 in ferrule (centered conductor)? Answer: The field strengths H1 in air and H2 in the ferrule are the same. 19

I Basic principles The field strength is given by:



(1.4)

1.2 Magnetic induction B Magnetic flux density

The magnetic induction or magnetic flux density is a physical value with B as symbol. B is the number of magnetic flux lines that pass perpendicularly through a defined area. A potential is induced in a conductor loop if the magnetic field passing through the conductor loop changes with time.

Fig. 1.10: Experimental configuration for magnetic induction Magnetic induction B

The surge in potential over the area of the loop is known as the magnetic induction B. Like the magnetic field strength, the magnetic induction B is a vector quantity. The following relationship applies for the magnetic induction B: (1.5)



The magnetic induction (B) is the quotient of the induced potential surge

20



(1.6)

and the product of the winding turns (N) and the windings area (A) of the induction coil.

The unit of magnetic induction (B) is the Tesla (T) = Vs/m2. The magnetic induction B and the field strength H in air are proportional to one another. The constant of proportionality is the magnetic field constant (µ0), given by experimental measurement.

µ0 = 4p · 10–7 Vs/Am

Magnetic field ­constant

(1.7)

In vacuum and also with sufficient accuracy for air, this leads to:

B = µ0 · H

(1.8)

The magnetic induction (BL) in air for the above example is given by: (1.9)



1.3 Magnetic flux F

Magnetic flux Φ

The magnetic flux (F) is the scalar product of the magnetic induction (flux density) (B) and the area vector (dA).

(1.10)



If (B) passes perpendicular through the area and the field is homogenous:

Φ = B · A

(1.11)

The unit of magnetic flux (F) is the same as that of the voltage surge (Vs) (Voltsecond) or Weber (Wb).

21

I Basic principles Faraday’s law

1.4 Faraday’s law Up until now we have considered static magnetic fields. If the magnetic flux changes with time, a voltage u is induced (Faraday’s law).

(1.12)



u = induced voltage t = time The polarity of the voltage is such that a current is generated in a closed circuit whose induced magnetic field opposes the original magnetic flux, i.e. it tends to reduce the magnetic field (Lenz’s law – Figure 1.11).

Lenz’s law

Fig. 1.11: Representation of Lenz’s law. The imposed magnetic field induces a current in the direction such that its induced magnetic field opposes the imposed field Taking a winding with N turns, Faraday’s law can be expressed in the following form.

(1.13) A l I L AL value

 = cross section of the coil = length of the coil or of the magnetic circuit = current through the coil = inductance of the coil [H(enry) = Vs/A]

So the inductance limits the change in current I once a voltage u is a­ pplied. It can be calculated from the coil data:

22

AL = AL value; mostly in nH/N2



(1.14)

The energy stored in the magnetic field is subject to the following r­ elationships:





(1.15)

The energy stored in the volume V is composed of both the magnetic field strength H and the magnet flux density B. For transformers and inductors with ferromagnetic cores, the flux density is limited by saturation and is constant throughout the magnetic circuit. If an air gap is introduced (material with permeability µ~1), the field strength is highest in this air gap with H = B/µ. It follows that the energy density is highest in the air gap. One also speaks of the energy being stored in the air gap. Comparing magnetic fields with electrical fields, analogies emerge b­ etween certain parameters. These are summarized in Table 1.1: Magnetic field

Electric field

Magnetic flux Φ [Wb]

Electrical current I [A]

Magnetic flux density B [T]

Current density J [A/m2]

Magnetic field strength H [A/m]

Electric field E [V/m]

Permeability µ

Permittivity e [m/W]

Inductance L [H]

Capacitance C [F]

Magnetic energy Emag=1/2 L I2

Electric energy Eelec=1/2 C U2

Analogies

Tab. 1.1: Analogies between magnetic and electric fields

1.5 Core materials and their losses Introducing solids into a magnetic field, their behavior can be classified in three groups: • diamagnetic materials: bismuth, water, nitrogen, copper, silver, gold • paramagnetic materials: magnesium, lithium, tantalum • ferromagnetic materials: iron, nickel, cobalt Diamagnetic and paramagnetic materials have a relative permeability close to one. They are therefore only of limited suitability in the construction of inductive components. Ferromagnetic materials have a r­ elative permeability between 10 and 100 000. In order to understand ferromagnetic core materials, the internal structure of these materials must be examined closer. The atoms in ferromagnetic materials (termed magnetic materials as follows) have a magnetic moment. In the non magnetized state, the ­magnetic moments of the atoms are aligned in all spatial directions, whereby atoms show a preferred direction within limited cells (Weiss domains). The boundaries between these Weiss domains are termed as Bloch walls.

Bloch walls

23

I Basic principles If an external magnetic field is now applied, this attempts to orientate the magnetic moments along the magnetic field direction, whereby the crystal direction attains the preferred direction. This occurs because the Weiss domains with a magnetic moment in the direction of the field grow at the expense of neighboring domains. This is described as shifting of the Bloch walls. This is a reversible process within certain ­limits. If the field strength is increased, the Bloch walls jump from one defect site to the next. This is then no longer reversible. If all domains are aligned, a further increase in the magnetic field rotates the magnetic moments from their crystal direction. Here one speaks of rotating processes. Hysteresis curve (B-H-curve)

This behavior is reflected in the hysteresis curve (also known as the B-H curve, Figure 1.12). Reversible Bloch wall shifts predominate in the ­lower region of the new curve. In the middle region, where the magnetic flux density B rises almost linearly with the field strength H, the irreversible jumps of the Bloch domains (Barkhausen jumps) may be identified. In the saturation range where the rise in the magnetic flux density is very much slower, the rotating processes predominate. Further growth of the Weiss domains is no longer possible.

Fig. 1.12: The B-H curve shows the magnetic flux density as a function of an imposed magnetic field

Remanent flux density Br Coercive field strength HC

24

Reducing the field strength, many of the shifted Bloch walls r­ emain stuck at defect sites. The magnetic flux density falls along an­other curve. Magnetic flux is still present even when the field strength has returned to zero. This refers to remanent flux density Br. In order to reset the flux density to zero, a certain negative field strength has to be applied, the coercive field strength HC.

The hysteresis curve profile is dependent on the material. According to the value of the coercive field strength, “soft magnetic” and “hard mag­netic” materials are distinguished. • soft magnetic materials: HC < 1000 A/m are mainly used in inductive components • hard magnetic materials HC > 10000 A/m are mainly used as permanent magnets and with electromagnets The area within the hysteresis curve corresponds to the core losses per cycle. At higher frequencies there are also eddy current losses.

Soft magnetic ­materials Hard magnetic materials Core losses

A frequently asked question concerns core material losses and the resulting power dissipation through ripple current in the storage choke. We will provide some basic information in this section. Expensive measuring techniques are required to determine core losses. These are generally based on the measurement of parameters defined for toroidal ferrites. In order to obtain relatively accurate results, phase-accurate power amplifiers and multiplying measuring instruments are required for power measurement with a low phase difference. The classical specification of core losses is given by the “Steinmetz formula” named after its inventor:

Steinmetz formula

PCORE = k · f a · Bb(1.16)

B = peak value of induction, Pcore is the average power dissipation per unit volume and f the frequency of the sinusoidal measurement voltage. For ferrites the coefficient “a” is between 1.1~1.9 and the coefficient “b” is in the range 1.6 ~ 3. An iterative approach is required for other materials in order to identify the coefficients. However, in switching controller applications we find rectangular voltages across the storage choke. For a 50% duty cycle, the accuracy of the Steinmetz formula is already reduced, for small or large duty cycles errors of 100% can arise!

25

I Basic principles

Fig. 1.13: Example for the error in the Steinmetz formula with a duty cycle > 50% (f = 100 kHz; MnZn core) The accuracy of the Steinmetz formula is further compromised by: • ignoring the dc premagnetization (→ a different B-H curve is produced!) • ignoring the current harmonics • ignoring the Vµsec product (higher Vµsec product ~ higher losses) • temperature dependence of the core material (many core materials only attain their loss minimum at elevated temperatures!)

26

Fig. 1.14: Power dissipation in the core material as a function of temperature based on the example of a MnZn power ferrite Extended Equations

MSE

Even though there are other equations which separate hysteresis and eddy current loss to overcome the problem of non-sinusoidal waveforms, the empirical Steinmetz equation has proven to be the most useful tool for sinusoidal flux-waveforms, because it provides better accuracy and is quite simple to use. Hence there are extensions to this power loss equation to make it accurate for non-sinusoidal flux waveforms. One such extension is the, “Modified Steinmetz Equation” (MSE) which has been widely utilized.





Pv = K · feqα–1 · Bpkβ · f (1.17)

where

f feq = 2π · (D–D2) 

feq being the equivalent frequency with respect to change in the duty cycle (D) of nonsinusoidal waveforms.

27

I Basic principles GSE

Due to some disadvantages posed by MSE, a later modification brought forth the Generalized Steinmetz Equation (GSE) – as shown here.





Pv = K · feqα–1 · Bpkβ



where Beq is

1 4



T

0

|

dB | dt dt 

(1.18)

Since the GSE & MSE core loss charts are also based on sinusoidal excitation, there exists some limitations. There are also some other equations developed by core manufacturers which works best with their specific cores. The major disadvantages of the Steinmetz and its extended equations are: • Dependent on core manufacturer’s empirical data, (for core loss charts). One must depend on the core manufacturer’s data and the passive component manufacturer does not have any control over the test setup. • Low accuracy with pulsating and triangular waveforms. Because the core loss graphs are created using the data from sinusoidal excitation. • Due to errors occurred in parameter conversion, the extension of Steinmetz models works best only for 50 % duty cycle and limited frequency range. • Only confine to components made with particular materials or by manufacturer. • Due to the complexity of estimating magnetic path length, the estimation of core loss using existing equations for iron powder materials and metal alloys is not only challenging, but the accuracy varies widely. For these reasons, all calculations with the classical Steinmetz formula and its extensions must always be evaluated in the light of these facts. The safest method of assessing whether a design of a storage choke works optimally is efficiency measurement on the switching controller and the measurement of selfheating in operation (taking into account the heat coupling with hotter components such as diodes and switching transistors!). REDEXPERT®

28

Würth Elektronik’s REDEXPERT® design tool has the world’s most accurate core loss model for simulating inductor losses under real world conditions. It includes all types of core materials, core shapes, air gaps and fringing effects and the ac loss effects of wire and winding structure. Based entirely on actual measured data using pulsating square wave voltages applied across the inductor with varying frequency, pulse width and temperature the model will accurately predict the losses in buck, boost and SEPIC converters. The tool is complete with charts, markers, selectable sort criteria and the ability to compare up to 16 products.

Example: Material 1P2400 MnZn-ferrites are mainly used for transformers working in the range between 50 and 500 kHz. At Würth Elektronik the material 1P2400 is used, which is characterized as follows.

MnZn ferrite

Table 1.2 provides an overview of the most important parameters of the material. The permeability µ is around 2400. Initial permeability

µi

25 °C

2400

Saturation flux density

Bs

25 °C

510 mT

100 °C

390 mT

Remanence

Br

25 °C

110 mT

100 °C

60 mT

25 °C

13 A/m

100 °C

6.5 A/m

25 °C

600 kW/m3

60 °C

300 kW/m3

80 °C

260 kW/m3

100 °C

300 kW/m3

120 °C

380 kW/m3

100 kHz

5 x 10–5

Coercive field strength Specific core losses

HC pv

Relative loss factor

tanδ/µi

Curie temperature

TC

215 °C

Resistivity

ρel.

6.5 Ωm

Density

ρmech

4800 kg/m3

Tab. 1.2: Characterization of the material 1P2400 Figure 1.15 shows the specific core losses as a function of magnetic flux density at different frequencies for a sinusoidal input and temperatures of +23 °C and +80 °C. The core losses at +80 °C are lower than at +23 °C. As magnetic components usually work in the range between +60 °C and +100 °C due to self-heating, one can assume the +80 °C curves when calculating losses. The Steinmetz coefficients may be obtained from the curves and by entering into the Steinmetz formula the losses at other frequencies and flux densities may be interpolated.

29

I Basic principles

Fig. 1.15: Specific losses dependent on the magnetic flux density As the magnetic behavior of ferrites is not linear, the respective Steinmetz coefficients only apply within certain ranges. Also, switching magnetics are usually not driven sinusoidally so the calculated core loss values may deviate in reality.

1.6 Permeability µ Permeability

Permeability describes an important effect in ferromagnetic materials. If a ferromagnetic material is placed in a magnetic field, it is observed that the magnetic flux becomes concentrated in this material. Analogous to electric resistance, the

30

ferromagnetic material presents a good conductor for the field lines. Permeability may therefore be described as a magnetic conducting or penetrating property.

Fig. 1.16: Ferromagnetic material in a magnetic field The factor by which the induction (B) changes through the introduction of the material is called the relative permeability (µr).

Relative permeability µr

(1.19)



The equation for relative permeability is extended for the space filled by the material:

B = µo · µr · H

(1.20)

The induction in the core material (BF) in our example on page 19 with the ferrite sleeve 742 700 9 (a constant relative permeability of µr = 800 is assumed) is given by:



(1.21)

The relative permeability of the material is however not constant but s­ trongly ­non-linear. The permeability of a material is essentially dependent on: • The magnetic field strength H (dependent on operating conditions → hysteresis curve) • The frequency f (frequency dependent complex permeability) • The temperature T (→ temperature drift, → Curie temperature) • The material used

31

I Basic principles Typical permeabilities µr

Complex permeability

Typical permeabilities µr: • Iron powder cores, superflux cores   50 …    150 • Manganese-zinc cores 300 … 20000 • Nickel-zinc cores  40 …  1500 1.6.1 Complex permeability

Fig. 1.17: Frequency dependence of permeability and impedance



µ = µI – jµII(1.22)

The introduction of complex permeability allows separation into an ideal (zero loss) inductive component and a frequency dependent resistive component which represents the losses of the core material. This treatment can be applied to all core materials and clearly differentiates between inductors and EMC ferrites. The inductive component is represented by (µI) and the resistive component by (µII). The following applies to transformation on the impedance level:

Z = jwµLo(1.23)

with L0 = inductance of an air coil of the same construction and field distribution, without core material (µr = 1).

32

Series impedance Z

For the series impedance (Z):

Fig. 1.18: Equivalent impedance circuit diagram



Z = RS + jwLS = jwLo (µIS – jµIIS)(1.24)

Multiplying out and dividing into real and imaginary parts provides the f­ollowing relationship: • Loss component • Inductance component

RS = wL0µS|| XLS = wL0µS|

Fig. 1.19: Loss angle (d) Loss angle tan d

For the loss angle (d), tan d is given by:

(1.25)



A large angle (d) means a high core loss; the phase relationship between voltage and current at the inductor is less than 90°. Furthermore: µS| = µi (µi = initial permeability) µS|| = µi · tan d

33

I Basic principles Parallel equivalent circuit

Similarly, inductance and resistance can also be presented as a parallel equivalent circuit; the following relationships apply:

(1.26)



These frequency dependent components can be measured with the aid of an impedance analyzer and represented in an associated graph:

Impedance curve

Fig. 1.20: Impedance curve for SMD ferrite 742 792 034 Observations from the above measurement graph: • The inductance is stable in a certain frequency range, to show strong f­requency dependence above approx. 10 MHz. Above 100 MHz the inductance falls sharply, down to zero at approx. 250 MHz. • The loss component (R) grows continuously with frequency and reaches the same value as the X component at the ferromagnetic resonance frequency. The resistance value rises until the high MHz range and dominates over the impedance (Z). The component shown here – a SMD ferrite – serves the user as a broadband ­absorber or filter component, as a result of its broadband loss resistance (R). 1.6.2 Comparing core materials

Frequency dependent loss components 34

­ ithin a Core materials can only be used effectively in the construction of inductors w limited frequency range, as a result of the frequency dependent loss components. Core losses rise sharply above a typical frequency limit. The core material may then be

used as a filter component (absorber element). This relationship and the limits of core materials are i­llustrated in the following graph: Inductive parts of impedance

XL - impedance reactance

100 % 80 % 60 % 40 % 20 % 0% 10 kHz

100 kHz

1 MHz

10 MHz

100 MHz

Frequency XL (Fe)

XL (MnZn)

XL (NiZn)

R - impedance resistance

Fig. 1.21: Inductive parts of impedance and their frequency ­dependence for various core materials Resistive parts of impedance

100 % 80 % 60 % 40 % 20 % 0% 10 kHz

100 kHz

1 MHz

10 MHz

100 MHz

1 GHz

Frequency R (Fe)

R (MnZn)

R (NiZn)

Fig. 1.22: Resistive parts of impedance and their frequency dependence for various core materials Observations: • Iron powder materials (Fe): May be used as pure inductance up to a­ pprox. 400 kHz; the R loss component dominates thereafter up to a­ pprox. 10 MHz (also beyond depending on the core material). The core is no longer effective in the frequency range above approx. 20 MHz.

Iron powder (Fe)

35

I Basic principles Manganese-zinc (MnZn)

Nickel-zinc (NiZn)

• Manganese-zinc cores are inductive up to frequencies around 20 MHz – 30 MHz, typically with losses rising above 10 MHz. The core material is no longer effective in the frequency range above – approx. 80 MHz. • Nickel-zinc cores are inductive up to frequencies around 60 MHz, above this, the core material shows losses up to frequencies of 1 GHz and more. This quantitative comparison illustrates why nickel-zinc ferrites have b­ ecome predominant in the EMC field. The core material can perform an ­effective filter function in the frequency range of greatest interest.

Inductance L

1.7 Inductance L Not only do magnetic materials possess a magnetic field, every current carrying ­conductor creates a magnetic field itself. Energy can be temporarily stored in the magnetic field. This effect is t­echnically exploited in inductors, consisting of one or more wire windings. There are various types of inductors or coils: • Air coils (without ferrite material) • Choke coils with iron powder core or ferrite core • Toroidal core coil • Rod core coil • SMD types are becoming increasingly important as a result of their small size. Besides wound SMD inductors, multilayer inductors are ­becoming increasingly established. All coils share a special behavior described in more detail by the following definitions. 1.7.1 Definition of inductance L A circuit element which responds to a current change with a counter-­voltage, shows inductive properties. An inductor is a passive component, whose reactance, produces a counter-voltage, the self-induction voltage. The self-induction voltage (Uind) at the terminals of the inductor is dependent on the rate of current change (di/dt) and a constant of proportionality, the inductance (L):



36



(1.27)

The inductance (L) of the coil is dependent on the core material, the geometry of the core and the number of winding turns. The following equation applies generally for calculating an inductance (L):



(1.28)



The unit of inductance (L) is the Henry (H) = Vs/A The inductance of cores with an inserted air gap can be calculated on the basis of the following formula: leff = lmean + (lgap · µr)(1.29)

lmean lgap µ r µ o Aeff leff N

= mean magnetic path length in the core (without air gap) = path length of the air gap(s) = relative permeability = 4p · 10–7 Vs/(Am) = effective cross-sectional area of the coil core* = effective path length in the coil core* = number of winding turns

* more details for the determination of Aeff; leff see appendix → core constants

This formula inserted into the formula for general inductance calculation produces:



(1.30)



This also allows an air gap width to be determined if the required inductance L and the other parameters are known. Here it must be borne in mind that the above formula only applies if μr is large and the air gap length is much smaller than the mean length in the core. In order to include fringing effects and their effect on inductance, McLyman proposes the following form of calculation for fringing effects F:

w L lgap Agap F

 = = = =

Fringing factor

(1.31)

core window length path length of the air gap(s) cross-sectional area of the air gap fringing factor 37

I Basic principles The result is that the inductance LF changes by the calculated value Lgap times the fringing factor F: LF = F · Lgap(1.32)

The positive influence of the air gap is an increased saturation current for the same core size. A disadvantage is that to attain a given L value, the number of turns now has to be increased and so, if no additional winding space is available for larger wire, the dc resistance of the winding also increases. Under no circumstances should the number of turns be reduced to compensate the fringing effect – this additionally increases the the flux density and can lead to premature saturation. The required air gap width for a given inductance L, taking into consideration the fringing factor F, can be calculated to a first approximation as follows:





(1.33)

Practical values: • A 1 mm long piece of conductor has an inductance of approx. 1 nH • Air coils up to 2000 nH • Multilayer inductors 10 nH … 10 µH • Barrel-type inductors 1 µH … 1 mH • Storage and filter chokes 0.1 µH … 10 mH

AL value

1.7.2 Definition of the AL value To save the user from calculating the effective magnetic length (leff) and area (Aeff), the corresponding AL value is given for toroidal cores and sleeves. This represents the effective inductance for one winding turn and must be multiplied by the square of the winding turns (N) to give the actual inductance (L).

38



(1.34)

The (AL) value is the inductance (L), typically given in nH, assuming the winding turns N = 1. Thus, given the AL value, the required number of coil turns can be f­ound without having to take the long route of considering the core’s ­geometric data:



(1.35)



Example: Required inductance 100 µH; the core has an AL value of 250 nH





(1.36)

Result: The core must have 20 turns to generate an inductance of 100 µH.

1.8 Impedance Z

Impedance Z

If an inductor is operated with ac voltage, it is clear that it represents a different resistance than in dc operation. The resistance for an ac voltage applied to the terminals of the coil is c­ alled impedance (Z).

R

Z XL

Z

XL

ϕ R

Fig. 1.23: Relationship between impedance, reactance and resistance





(1.37)

The impedance (Z) is frequency dependent and is composed of the geometric sum of the loss resistance (R) and the reactance (XL) of the ideal coil (L).

39

I Basic principles The reactance XL is defined as follows: XL = 2 · p · f · L

(1.38)

Observation: Impedance rises with increased frequency. This linear relationship continues to infinitely high frequencies for an ideal coil.

Fig. 1.24: Impedance curve for real inductors However, as a consequence of the frequency dependence of permeability, the construction of the coil and parasitic capacitance, the applicability of coils at high frequencies is limited. The impedance declines rapidly from the self-resonant frequency; the inductive nature of the coil disappears.

40

1.8.1 Self-resonant frequency (SRF)

Self-resonant ­frequency

L

C

ACR DCR

Fig. 1.25: Equivalent circuit of the real inductance Every inductor also has capacitive coupling arising from its winding turns and multiple layers. These parasitic capacitances are symbolized by a capacitor (C) in the equivalent circuit. This capacitance in the coil forms a parallel resonant circuit with the inductance.

Windings capacitance

At the self-resonant frequency, the input energy oscillates back and forth between the elements of inductance and capacitance. No more external energy is absorbed (ideal coil). If a coil is operated above its resonance, the capacitance dominates and in fact acts as a capacitor. In practice, coils should be operated well below their resonant frequency.

1.9 Resistive losses No active power (heat loss) is dissipated in the reactance XL due to the 90° phase shift between voltage and current. The total coil losses can be combined into the loss resistance (R), which is connected in series with the ideal inductance (L). This results in the equivalent circuit of real inductance (see Figure 1.25).

Coil loss

As the losses in R are however frequency dependent, the dc resistance (DCR) is also always defined in the data sheet specifications. This is dependent on the wire material used or the construction type of SMD inductors and is found at room temperature by a simple resistance measurement. The value of the dc resistance has a direct influence on the increase in temperature of the coil. Prolonged exceeding of the current rating should therefore be avoided.

41

I Basic principles Frequency dependent losses

The total losses of the coil consist of both the losses in the dc resistance DCR and the following frequency dependent components: • The losses in the core material (magnetic hysteresis loss, eddy-current loss) • Additional losses in the conductor from the skin effect (current displacement at high frequencies) • Magnetic field losses of the neighboring windings (proximity effect) • Radiation losses • Losses from additional magnetic shielding All these loss components can be combined into a loss resistance (R). This loss resistance is primarily responsible for defining the quality of the inductor. Unfortunately, mathematical determination of the loss resistance R is not possible. Therefore inductors are usually measured over the entire frequency range with an impedance analyzer. This measurement provides the individual components XL(f), R(f) and Z(f). The quality factor is defined as a quality characteristic of the inductor. 1.9.1 Copper losses

DC Copper Loss

DC Copper Loss The copper losses for inductive components are composed of direct current losses and eddy current losses. The direct current losses are calculated with Ohm’s law:

PV = R · IRMS2

(1.39)

PV = power loss R = dc resistance IRMS = effective current At higher frequencies, there are also losses due to skin effect and proximity effect. These eddy current losses may be explained directly with Faraday’s law. The current flowing through a conductor generates a magnetic field around this conductor. This magnetic field changes rapidly as a result of the high frequency, so that a voltage is induced in the conductor and in the neighboring conductors. According to Lenz’ law this voltage generates a current that opposes the original current. Thus additional currents are produced in the conductor. In the center they oppose and cancel but on the outside surface they add increasing the current density. Currents are also induced into neighboring conductors. Skin Effect

1.9.2 Skin Effect Considering a single isolated conductor, one speaks of the skin effect. For conductors through which high frequency currents flow, current only flows on the outer skin of the conductor (Figure 1.26). The penetration depth at which the current density has fallen to a value of 1/e (~36.8%) is given by:

42

d=

δ ρ µ 0 µ r f



ρ pµ0µr  f 

(1.40)

= penetration depth = resistivity of material = permeability of free space = relative permeability of material (i.e. copper) = frequency

Current flowing through Magnetic field Eddy currents

Fig. 1.26: Illustrates eddy currents that cause skin effect. Induced currents cancel flowing current in the center and add on the surface. Current density ­distribution

Current

Resulting current distribution

Cross-section Fig. 1.27: Cross section of conductor showing distribution of current density The penetration depth at 50 Hz is 9.38 mm, at 100 kHz it is 0.21 mm.

43

I Basic principles 50 kHz = 0.30 mm

100 kHz = 0.21 mm

500 kHz = 0.09 mm

round wire d = 0.48 mm

10 kHz = 0.66 mm

Fig. 1.28: Current distribution in a conductor at various frequency. Arrow indicates skin depth at given frequency. Proximity Effect

1.9.3 Proximity Effect The proximity effect plays a far greater role in multi-layer windings in both inductors and transformers, whereby neighboring conductors generate fields displaced by current. With inductors the field forces increase with the number of layers whereas in transformers the oppose flowing currents of primary and secondary help to cancel some of the effects. When current flows in the same direction in adjacent wires, the magnetic force between the wires cancels. Areas of low magnetic force have low current therefore the current concentrates on the outer surfaces of the pair as shown below. This is the pattern across a layer of winding in an inductor or transformer. In order to illustrate the principle we use conductors which are much larger in diameter than the skin depth.

Fig. 1.29: Current density in wires with current flowing in the same direction When current flows in the opposite direction in adjacent wires, the magnetic force between the wires is intensified. This means the currents will concentrate on the adjacent surfaces. This is the pattern between the primary and secondary winding of a transformer.

44

Fig. 1.30: Current density in wires with current flow in opposing directions

Fig. 1.31: Current density for a transformer winding. The primary winding is split equally on each side of the secondary winding. Currents concentrate on facing surfaces 1.9.4 AC loss calculations Dowell

Dowell

The possibility to calculate eddy current losses for simple geometries is described by Dowell in 1966. He proposed reducing the problem to one dimension by converting wire windings to equivalent foil windings and normalizing the thickness to the skin depth. The result is Fr which is the ratio of ac/dc resistance. From the chart in Figure 1.33 it is easy to see that ac resistance increases rapidly with increasing wire thickness and increasing number of winding layers.

45

I Basic principles d p d

p

h h h h bw h bwf h

Fig. 1.32: Conversion of litz, round, square or rectangular wire to equivalent foil. First convert round to square, bring together into a foil (bw) and adjust for porosity (bwf). Similarly for rectangular wire. For litz wire use the square root of the number of strands for the number of layers and use the diameter of one strand as the round size.. For foil use the dimensions directly.

2

Fr =

2 N -1 R AC sinh 2A +sin 2A =A + R DC cosh 2A -cos 2A 3

sinh A -sin A cosh A +cos A 

(1.41)

Where

A= d p δ N 46

= wire diamater = wire pitch = skin depth = Number of Layers

π 4

3 4

d δ

d p

(1.42) 

For litz wire substitute N with Nlitz = N  k  where k = number of strands



(1.43)

Equation 1.41 from Kazimierczuk, M. High-Frequency Magnetic Components, 2nd ed. S­ ingapore: John Wiley & Sons, 2014, pp. 306–351.

m

Fr = Rac/Rdc

1000

10 9 8 7 6 5 4 3.5 3 2.5 2 1.5 1 0.5

100

10

1 0

1 ϕ = thickness * F/skin depth

10

Fig. 1.33: Dowell’s curves showing rapid increase in ac resistance factor (Fr) as the wire thickness relative to skin depth (ϕ) and number of winding layers (m) increase The number of layers (m) corresponds to the number of layers in a section where the MMF goes from zero to a maximum or minimum peak or from the peak to zero. A simple MMF diagram is shown in Figure 1.34 for a normal and an interleaved foil winding. In the normal winding the primary has of two layers and the secondary has three. In the interleaved version, there are two primaries of one layer each and the secondary in the middle. The magnetic field strength within a winding rises from the inside to the outside, because evermore turns (ever greater currents) are enclosed by field lines. The magnetic field of the secondary winding is opposite to the primary field. This serves to reduce the magnetic field. The reduction in the magnitude of the H field is plain to see. In the interleaved version the secondary is portioned into two 1.5 layer sections based on the MMF going to zero through the middle turn when using Dowell’s curves. The theory has been further developed by Carsten and others. The mathematical description far exceeds the scope of this book.

MMF diagram

47

I Basic principles

core center leg

It is far more important here to understand the options at hand to limit eddy current losses which are dependent on the magnitude of the magnetic field. This can be achieved, as illustrated, by interleaving the windings. This reduces the absolute value of the magnetic field and therefore also the eddy current losses.

MMF

x primary winding secondary winding

core center leg

Normal non-interleaved

MMF

x

primary winding secondary winding

Interleaved Fig. 1.34: Magnetic field profile in a transformer with different winding configurations 48

Thin, flat conductors, e.g. copper foil, can also be used for winding. The thickness should be of the order of the penetration depth. This should only be used for small numbers of turns, because for higher numbers of turns, the larger number of layers causing higher eddy current losses. A further option for reducing eddy currents is to wind with small insulated wires rather than a large wire. Here care must be taken that the single strand wires connected in parallel to have the same current distribution. High frequency litz wires offers an option, whereby single wires are twisted with one another so that on average each wire strand has the same position in the magnetic field. Care must also be taken with this option that the number of layers is does not become too great. 1.9.5 Definition of quality factor Q

Q factor

The quality factor (Q) is the ratio between the stored energy (XL: reactance) and the losses (R). It is an indicator of how ideal an inductor is. The component of external input energy converted into heat in the loss resistance R does not contribute towards the energy stored in the magnetic field. The larger these losses are, the poorer the inductor acts as a buffer and the lower the Q. The Q factor is defined as follows:





(1.44)

Practical values: • Air coil • Ferrite choke • SMD multiplayer inductors

Q up to 400 Q up to 150 Q up to 60

The quality-frequency graph helps select the best inductor construction for the particular application.

49

I Basic principles

Fig. 1.35: Quality factor-frequency graph Observations: The Q factor rises to a peak value and then declines. Constant small losses in resistance R of the inductor can be assumed up to the peak Q value. Beyond the peak value, significant losses become evident, and the inductance also varies on account of non-linearity of the ferrite material. The operating range with the smallest losses can be defined up to the Q turning point. If the inductor is used at higher frequencies, the losses increase rapidly.

1.10 Temperature behavior Coils having ferrite core materials show a variable inductive behavior with change in the ambient temperature. This is typically significant in ungapped cores, often increasing by a factor of two to three over room temperature values at high temperatures. Usually permeability peaks just before the ferrite reaches its curie temperature. The curie temperature is the point where the ferrite material loses all its permeability – hence the inductance goes to zero. The condition is not permanent. When the ferrite cools down the previous permeability is restored. Generally the higher the initial permeability for the ferrite the lower the curie temperature.

50

8000

initial permeability

6000

4000

2000

0 –50°C

50°C 150°C Temperature

250°C

Fig. 1.36: Initial permeability as a function of temperature for Ferroxcube 3C96 ­material The use of gaps in inductors reduces the variation significantly with larger gaps giving more linear performance over temperature.

Fig. 1.37: Temperature drift of a multilayer inductor

Temperature drift

If high demands are placed on the stability of filter circuits constructed with inductors (e.g. in measurement technology), it is expedient to select a coil with an almost linear temperature curve. This is why air cores are used in RF applications. The inductance change ΔL with respect to the inductance rating L of the coil is lowest in this case. 51

I Basic principles Rated current

1.11 Rated current The rated current, which may be carried by an inductor, is defined as the current that causes the inductor to heat up above ambient temperature from self-heating due to losses (dc and ac). This is different from saturation current and may be a larger value. Both are limiting values with the lower value being the inductor’s useable limit. If the component is operated at its rated current it heats up above the ambient temperature by the temperature stated in the datasheet (∆T = 40 K (typical). It must then be determined if the resulting temperature of the component is suitable for the application. It has to be checked that by operating that the part does not exceed the operating temperature otherwise, a component with a higher rated current capacity needs to be selected.

Saturation current

1.12 Saturation current The saturation current of an inductor is the current at which the initial inductance value has dropped by a percentage specified in the datasheet. For inductors this can range from 10 to 40% and must be carefully checked as the value varies among components and manufacturers. On the material B-H curve this is the point where the curve starts to flatten and permeability starts decreasing. Materials like ferrite have a shape bend or knee, changing very quickly whereas materials like powdered iron have a less pronounced knee. These are sometimes referred to as hard and soft saturation characteristics. Note! Especially for switching controller applications or applications with high capacitive loads or high inrush currents, the peak current flowing through the inductor can be significantly higher at the moment of switching-on then in regular operation. This may lead to total saturation of the component and therefore to potential consequential faults in the electronics. It is advisable to understand and to limit the current or to activate soft-start functions.

52

Fig. 1.38: Inductance-current graph

1.13 Differentiating EMC ferrite ↔ inductor The terminology used by Würth Elektronik clearly distinguishes between i­nductors and EMC ferrites in regard to the quality of the inductor: EMC ferrites are based on Ni-Zn materials. This material has poor quality factors (Q  102, which is a reason for the high capacitance in the equivalent circuit. The parallel resistance Rp falls to between 1/2 and 1/3 of the maximum v­ alue at high frequencies. The higher the initial permeability µi, the lower is the 45° limit frequency fg = Rp/(2 · p · Lp). The application range of this m ­ aterial as an interference power absorber starts above approx. 2 · fg.

NiZn ferrites

Fig.: 1.52: Equivalent circuits on nickel-zinc ferrites Figure 1.52 shows equivalent circuits for Ni-Zn ferrites, which have a lower Cp but are only useful above 20 MHz due to their low permeability. Classical EMC interference sources have a spectrum concentrated well below 20 MHz. The nickel-zinc ferrites, which are otherwise useful up to high frequencies as a result of their low parallel capacitance, have a problem: To become effective down to low frequencies ( ~40 MHz); in the lower frequency range – i.e. where the capacitors are of very low impedance – its attenuation drops quite quickly. Bandwidths of 500 MHz and more can be achieved with this technique without any difficulties. At very high frequencies, the capacitors (even connected in parallel) are of high impedance sooner or later and the filter attenuation drops significantly. The exception here is filtering of power systems as previously mentioned. If the parallel capacitor array of the filter is an optimally configured power system, it can be of very low impedance even into the GHz range. If this parallel capacitor array is now supplemented with a suitable inductor considerable attenuation can be achieved up to 2 GHz and beyond. The use of this filter is a very effective measure for reducing differential mode radiation from power systems, as this radiation typically shows a very pronounced broadband interference level due to the large number of active digital components. But let us return to our “classical” filter, consisting of three 100 nF capacitors parallel and a choke (742 792 093) in series and its insertion loss shown in Figure 1.67 and Figure 1.68. In simplified terms, the insertion loss specifies how much less power is converted in the load as a result of the filter inserted.

83

I Basic principles Insertion loss





(1.66)

Looking at the formula for calculating insertion loss (Equation 1.66), it is immediately apparent that besides the impedance of the filter, also the source and load impedance affect the result (POUT = UOUT2/ZOUT). For measuring instruments (e.g. network analyzers), the convention has become established as the industry standard of using both the source and load impedance as R = 50 W. Logically, this convention is also applied for the simulation of filters. The source and load impedances that EMC filters encounter in their subsequent applications are generally different and nothing like 50 W, but some other (complex) impedance instead. This, however, means that it is only possible to predict the insertion loss of a filter if the actual terminations ZIN(f) and ZOUT(f) are known. For manufacturers of EMC filters for example, this condition can practically never be fulfilled, as the manufacturer cannot know at all in which environment the customer will later use their filter. It is therefore quite common to specify filter attenuation curves for ZIN = ZOUT = 50 W, to at least allow a certain comparability between filters. Unfortunately the nagging doubt remains of not really knowing what kind of attenuation profile the filter will actually produce. In order to give an impression of what has to be expected in the worst possible case, the “approximate worst case method” was defined in CISPR 17. This involves considering the filter in two different termination combinations: 1. ZIN = 0.1 Ω, ZOUT = 100 Ω 2. ZIN = 100 Ω, ZOUT = 0,1 Ω For filters which, for example, only consist of a single series choke, both cases result in the same attenuation profile – the filter is symmetric. As significantly higher attenuation can be achieved by using a parallel impedance (e.g. capacitor), an L structure – as with the SILENT filters – is very common. Nevertheless, there is a dramatic difference between the two CISPR-17 attenuation profiles for this type of filters (Figure 1.70).

84

100/0.1

50 Ω

–CISPR17 100/0.1

Fig. 1.70: Insertion loss of the filter for various terminations Whereas the 0.1/100 curve even promises slightly higher attenuation, the attenuation of the 100/0.1 combination is around 30 dB (!) poorer than the 50 Ω profile. Looking at the filter circuit in Figure 1.64, this is easy to explain: The filter termination (load impedance) of 0.1 Ω is connected in parallel with the capacitors. If the impedance of the capacitors at the observed frequency is greater than the 0.1 Ω of the load, the larger proportion of current flows through the load where it converts POUT accordingly (Equation 1.66). At the other extreme, i.e. with a load impedance of 100 Ω, the capacitors are of comparatively very low impedance so the majority of the interference current flows through the capacitors and not through the load. This relationship can be clearly shown: The choke on its own provides attenuation of around –20 dB at 40 MHz and 190 MHz (Figure 1.69). Taking a look now at Figure 1.70, the attenuation difference between the 50 Ω curve (– – – curve) and the 100/0.1 case (red color), it is apparent that this “mismatching“ at 40 MHz leads to a drop in attenuation of around 48 dB, however only around 36 dB at 190 MHz. As the choke shows the same attenuation at both frequencies, the difference of approx. 12 dB is attributable to the parallel branch. This is very much lower impedance at low frequencies (40 MHz) than at higher frequencies (190 MHz) and therefore makes a greater contribution to the overall attenuation at lower frequencies. If the parallel branch is now rendered largely ineffective by the 0.1 Ω termination connected in parallel, the damage at low frequencies is especially high.

85

I Basic principles This insight allows several specific recommendations for action to be derived: • The attenuation profile from the datasheet should therefore be taken as a first information only! • If the L filter structure described above is used, take care that the capacitor is on the side of the filter where the higher terminal impedance is expected!

T filter

• The “safe bet”: If you don’t wish to think about the terminal impedances at all or measurement/estimation is not possible, the filter is expanded to a T structure using a further choke of the same type. This increases the attenuation once again, but, above all, the filter is tolerant towards low impedance terminations. This approach saves development workload at the expense of higher unit costs and is therefore not recommended for high volume production.

For the sake of completeness, it should be mentioned that, particularly for larger unit volumes, it can be extremely useful to determine the terminations in the actual component group by measurement or simulation in order to then develop both a performance and cost optimized filter with the aid of suitable simulation software.

3 Filter Basics 3.1 Filter circuits The filter is a commonly adopted measure to reduce the noise coupling from A to B. The principle applies in reducing interference emissions as well as in increasing interference immunity. The coupling path A ↔ B can exist on the system, device, component group or even on the component level. Often the useful signal must not be affected, depending on the application, which means that filtering may be complex as required or relatively uncomplicated where the frequencies of noise and useful signal are sufficiently separated. The filtering of an audio output (e.g. headphones) of a PC superimposed with video signals in the 200 MHz range is a simple matter; a video signal with 200 MHz bandwidth, including 100 MHz harmonics from the clock signal ­ eans. cannot be filtered under 200 MHz using conventional m

86

3.2 The principle of filtering, functionality and structure of filters Energy cannot be created or destroyed, it can only be converted into another form of energy – the law of conservation of energy tells us this.

Principle of filters

Electrical noise energy does not disappear through the use of shielding, hinged ferrites or filters. The noise energy should not arise in the first place or where it ­arises should be converted into heat. This conversion or absorption is best achieved at the component or printed circuit board level and in many instances the housing or cable shielding can be significantly simpler to implement and t­herefore more cost effective. Filters can convert the noise energy into heat. Condition: A current must flow leading to a voltage drop. This is easy on paper, but where should the voltage drop happen? Often circuit examples as in Figure 1.71 may be found and there is great astonishment to see that line A rather than line B emits interference after the insertion of a capa­citor.

Fig. 1.71: Example of an insufficient filtering measure using a capacitor What has happened? The capacitor C leads the noise superimposed on line A to ground, so that a circuit is formed through line A to the noise source (e.g. clock ­generator, video controller) via ground, whose magnetic field induces noise energy in the neighboring circuit, in this example line B.

Noise energy

The counterpart of the circuit in Figure 1.71 is given in Figure 1.72.

87

I Basic principles

Fig. 1.72: Example of insufficient filtering measure with a choke

Cable as antenna

There is also great astonishment here when the peripheral cable of port A still emits interference, in spite of the choke. What has happened? From the electrical perspective, a simple cable is a monopole, which radiates an electric field and has a high impedance of typically several kW at the base (e.g. chassis connection). If a choke is connected in series with too high an impedance of several 100 W, the noise current is only slightly changed, it still drops along the cable or more precisely it is radiated by it. A useful, d­ efined filter therefore comprises at least two components, of which at least one must be frequency selective. Possible combinations are shown in Figure 1.73.

Fig. 1.73: Basic filter circuit (low-pass) Source and sink impedances

Low-pass filter

88

Which components should be deployed depends on the source and sink impedances. The impedances on both sides of the circuit must be mis­matched in the noise frequency range, but matched in the useful frequency range. The inputs and outputs of filters as shown in Figure 1.73 are not reversible; their effectiveness can only be guaranteed if they are cor­rectly constructed. The filter in Figure 1.73 is a low-pass filter. Depending on which useful frequencies should be allowed through relative to the noise frequencies, high-pass, band-pass and band-stop filter circuit versions are possible (Figure 1.74).

Fig. 1.74: Various basic filter circuits To assess and classify filters, parameters such as corner frequency, bandwidth, quality factor, steepness, phase shift etc. are used in filter technology. These parameters are of lesser importance for EMC purposes, as these parameters assume a defined input and output, source and sink, impedance. In most cases however, this decisive information is missing, rendering reliable calculation of filter parameters impossible.

3.3 The low-pass filter

Low-pass filter

The low-pass filter is the most commonly used filter circuit in EMC. How­ever, to improve understanding and to evaluate the effectiveness of the filter, some detailed observations may be helpful. The observation should be made on the basis of the two most commonly used low-pass circuit varieties (Figure 1.75).

Fig. 1.75: Low-pass filters of first and second order 89

I Basic principles Laplace transformation

The converted form of the Laplace transformation is used to improve the mathematical representation of filters or quadrupoles in the frequency domain. The Laplace transformation in the complex variable domain results, with



Frequency response

with r = s + jw

(1.67)

the transformed function in the frequency domain, i.e. from a technical v­ iewpoint the circuit’s frequency response. Functions in the complex variable domain can be found using some mathematical rules from b­ asic time domain functions. For example, for

(1.68)



and for

(1.69)

90



Fig. 1.76: Calculation table Both functions can be presented graphically (Figure 1.77).

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I Basic principles

Fig. 1.77: Transfer function Fx(p) against angular frequency As p is to the first power in F1(p), the filter shows an increase in attenuation of 20 dB/ dec (filter 1st order n = 1); the filter with the complex variable function F2(P) has, with a p of second power, an attenuation response of 40 dB/dec (filter 2nd order n = 2). Shifts of the angular frequency axis depend on time constants. At w0 (w1 : F1(P)) there is an attenuation of 3 dB for F1(P) and an attenuation of 6 dB for F2(P). Resonance point

92

In contrast to the RC filter, the LC filter has a resonance point. The resistive losses of the inductor must be taken into account to describe the series resonance. This results in the equivalent circuit in Figure 1.78.

Fig. 1.78: Simplified equivalent low-pass filter circuit taking resistive coil l­osses into account Impedance

The impedance Z(w) is given by



(1.70)



Resonance occurs if the imaginary component is 0, i.e.



(1.71)



It follows that:





(1.72)

Furthermore, at resonance



(1.73)

The amplitude and phase responses of the filter in Figure 1.78 are presented graphically in Figure 1.79.

Amplitude response Phase response

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I Basic principles

Fig. 1.79: Amplitude and phase response of the filter in Figure 1.78 In practice this means that • the choke must have a constant high resistive impedance component over the required filter bandwidth to keep the resonant amplitude as low as possible and its bandwidth as broad as possible. • both the useful and the noise signal frequencies should lie below the r­ esonant frequency of the low-pass filter. • quality factor and loss values of the capacitor generally play a lesser role if the resistive components of the choke are high. • broadband, critical useful signals must lie within the linear phase response of the filter (far below resonance) to avoid distortion. • additional resonance phenomena occur due to each additional parasitic element. • source and sink impedances must also be taken into account with the f­ilter properties. The result is a network with source, filter and sink as shown in Figure 1.80.

94

Source and sink ­impedance (complex)

SMD Ferrite

Ri

LQ

RLQ DCR

LP CP RLS

RP LS

CQ

RS Capacitor

Source

RP Filter

RS CS

C

Sink

Fig. 1.80: Example of a network, taking parasitic impedances of the components into account

3.4 Filter circuitry Passive filters are a network consisting of passive components (resistors, chokes and capacitors) composing a frequency dependant voltage divide.

Frequency dependant voltage dividers

Frequency dependence means a change in the electrical properties with frequency. The most commonly used filter, the LC low-pass, works on the basis that the impedance of the inductor rises with increasing frequency and the impedance of the capacitor falls with increasing frequency. This would solve most EMC problems in theory, were it not for some side-effects, which reduce the filter function, sometimes even negating it altogether.

LC low-pass

3.4.1 Filter ground reference Weaknesses of filter reference grounds One of the most important conditions for useful function of an LC filter is the capacitor’s “ground reference” (Figure 1.81).

Ground

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I Basic principles

Fig. 1.81: Circuit diagram and faulty construction of an LC low-pass filter Parasitic inductance

Every additional impedance in series with the capacitor, whether of para­sitic origin “inside” the capacitor, caused by layout or construction, reduces the effectiveness of the filter. Long connections between the capacitor and ground are additional unwanted series inductances – regardless of whether the inductance comes from the connecting legs of the capacitor, the conductor tracks or bolts on the printed circuit board mounting. Designers and layout specialists are often faced with seemingly almost insoluble problems in this regard, as restrictions such as the space availability within the printed circuit board, number of conductor layers, positioning of ground bolts, bonding of the ground layer to the chassis, are often defined at the outset. If however the conditions for a functional filter are considered from the outset, a useful circuit can be constructed despite this or that compromise. What is wrong with the ground reference construction in Figure 1.81?

Layout

Branch line

Capacitive coupling

• The conductor track between the capacitor and the via (^) is too long. A 1 cm conductor track roughly equates to an inductance of 10 nH. • There is no connection to the chassis in the immediate vicinity of the f­ilter. • The connection between the inductor and the capacitor is a branch line, and the filter output is incorrectly positioned. • The conductor track between capacitor and ground is inductive. • The filter input is coupled with the ground connection of the capacitor, short-circuiting the choke at high frequencies. • Capacitive coupling is generated between the L and C path, which rises with increasing frequency. Figure 1.82 shows a comparison of two layouts. The left layout illustrates the listed weaknesses from Figure 1.81, the right layout is optimized in these regards.

96

Fig. 1.82: Faulty and optimized layouts of an LC low-pass filter The following points favor functionality: • The constriction avoids a noise bypass. • The perpendicular configuration of components lowers the capacitive coupling between the inductor and capacitor. • Both vias and the direct board mounting device facilitate a low impedance ground connection for the capacitor.

Construction (RF)

Vias

In Figure 1.82, a board mounting screw is directly adjacent to the capacitor, which en­ ables a perfect ground reference for the capacitor. Realistically, not every filter capacitor can be “fitted” with a screw to the chassis. There are however numerous realistic alternatives on offer, enabling an effective high-frequency ground. Alternatives to improve the filter reference ground Various options to improve the reference ground are shown as follows. • Combination of several capacitors:

Improved reference ground

Additional vias facilitate low impedance ground routing

Fig. 1.83: Ground island for filter capacitors

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I Basic principles • Reinforcing ground by the use of several layers:

Fig. 1.84: Ground reinforcement from several layers Bonding on the chassis

• Ground layer bonding by means of contact or spring strips on the c­ hassis:

Fig. 1.85: Ground bonding on the chassis

98

Examples of these contact strips are Shielding gasket

Conductive shielding gasket WE-LT

Order Code 302 030 1

A (mm)

B (mm)

Type

 3.0

1.0

A

302 100 2

10.0

2.0

A

302 350 3

35.0

3.0

A

303 643 61

 6.4

3.6

B

Fig. 1.86: Specifications of different conducting textile seals Isolation of the filter reference ground from the signal ground The signal ground is often affected by noise in highly integrated, fast sys­tems. This noise arises as an effect of voltage drop to ground between the switching sources, e.g. a clock chip IC1 and the sink IC2, e.g. a memory chip (Figure 1.87).

Filter reference ground Signal ground

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I Basic principles

Fig. 1.87: Digital circuit with line and ground inductances The ground connections on the clock chip IC1 have a non-negligible inductance. Similarly, the ground layer of the circuit board has an inductance, which can also take on remarkable proportions with an increasing number of vias. The current from IC1 to IC2 through the conductor track and back through the ground layer to IC1 causes a voltage drop UGND:

Ground noise



(1.74)

Ground noise cannot be avoided, but it can be partly reduced. As can be seen from the equation, the essential contributors are: • the signal rise time • the signal current • the lead inductances

Lead inductance

100

EMC compatible circuit design not only involves limiting signal rise times. Moreover, care should be taken that the lead inductances are kept as low as possible. Well designed digital components generally have several GND connections, which may not simply be connected in parallel on the circuit layout. Each GND connection must be separately bonded to the ground s­ urface (Figure 1.88).

Fig. 1.88: Correct and incorrect ground routing layout for a digital chip This measures are however insufficient to reduce the ground noise at the capacitors of a filter group. Isolation of the signal ground from the filter ground is an effective method.

Ground noise

This is achieved in the layout through an isolation channel. Both areas must then of course have a secure ground reference to the chassis via mounting screws of similar HF compatible bonding measures. The galvanic connection between signal and filter ground, sometimes required for circuitry reasons, can be a 0 W resistor or a ferrite choke. The component should be placed where the most quiet reference ground for the signal is expected. An example is given in Figure 1.89.

Isolation channel

Fig. 1.89: Layout example of ground isolation between signal and filter reference ground planes with connection suitable for low speed signals

Ground isolation

101

I Basic principles 3.5 Symmetrical filter/common-mode filter Common mode filters are mainly used in two areas: • in symmetrical signal transmission • in asymmetrical cases if the reference potential is subject to noise. Symmetrical signal transmission The transmission path Common-mode signal transmission

Common-mode signal transmission is used to increase interference immunity of long data and signal transmission lines and, at the same time, to ­reduce the radio interference emissions of signal and data harmonics. The common-mode signal is isolated from the reference potential, i.e. it is ­“ground-free” between sensor and receiver (Figure 1.90).

Fig. 1.90: Symmetrical signal transmission with different conductor types

102

Filter circuitry There are different circuitry techniques to balance signals. The higher the decoupling of the signal is from ground and the better the balance of the signal is, relative to ground, the better the circuit performs. Just as important is the choice of conductor type between transmitter and receiver. Twisted conductor pairs (a, b, c, d in Figure 1.90) are generally con­sidered preferable, as these show a greater interference immunity than u­ ntwisted. An untwisted but shielded cable (Figure 1.90, d), is only suit­able in the low frequency range up to a few kHz. Above this, the symmetry of the wires to the shielding is inadequate in most cases. Transmission p­ aths with unshielded twisted conductors may be earthed on one of the two ends (Figure 1.90, b); this is often necessary for simple (single-ended) transmission end stages for circuitry reasons. The highest interference immunity and the lowest interference emission is expected with twisted conductors in a shielded configuration. A single ended shielded connection ­however, only shields the electric field component, irrespective of whether the signal harmonics arise from the signal to be transmitted or is of external origin. Only if the cable shield is connected at both ends is the expected shield a­ ttenuation achieved. If potential equalization currents (usually via the p­ rotective earthing conductor) flow as a result of the devices’ differing ­reference potentials, the cable shield can be connected to one end via a c­ apacitor (e.g. 1 nF). Most currents in the 50–150 Hz range are reduced by the high reactance of the capacitor. As the capacitor has to have good RF properties, e.g. low inductance, in order to couple the cable shield with low impedance, an SMD type must be used. The capacitor must either have a high breakdown voltage of 250 Veff or a multilayer varistor must be connected in parallel against sporadic bursts (Figure 1.91). The response voltage of the varistor must be lower than the permitted voltage at the capa­citor, but of course greater than the long-term potential difference of the two devices. Experience shows that the potential difference is around 10 Veff (50­–150 Hz). If the voltage is higher, the potential equalization network of the sys­tem should be checked.

Twisted conductor pairs

Shielded cable

Shield attenuation

Varistor

Potential equalization network

For the varistor you can choose our WE-VS 825 41 140. With the varistor breakdown voltage of 24 V and a max. RMS voltage of 14 Veff this varistor suppresses all the incoming burst packages. Furthermore it takes effect as an additional noise filter due to its capacitance of 1.1 nF.

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I Basic principles

Fig. 1.91: Circuitry principles to connect a cable shield with a potential difference between the device grounds

Twisted pair cables

104

The suitable type of cable for symmetrical transmission is twisted and shielded (Figure 1.90, c). The shield is connected on both ends of the respective device grounds (chassis, not pcb!). There are however still factors to be observed: There are some shielded twisted pair cables whose internal conductors are only loosely electrically coupled with one another. This means that the coupling capacitance of each individual conductor to the cable shield is high. The transmission path of the symmetrical 150 W twisted pair conductor becomes ever more like two separated 75 W coaxial conductors (Figure 1.92).

Fig. 1.92: Conversion of twisted pair conductors in two separate coaxial conductors er: Effective dielectric constant between wires The consequence of the inadequate coupling of the two wires is not only a complete asymmetry of both paths and therefore inadequate electro­magnetic compatibility; the consequence is also a serious deterioration in the signal quality. Figure 1.93 shows a recipe for the qualitative assessment of a twisted pair.

105

I Basic principles

Fig. 1.93: Qualitative assessment of the coupling properties of a twisted pair cable Shielding measures

Despite all shielding measures, despite high quality shielding of a twisted pair cable, electromagnetic incompatibilities sometimes still occur in practice, or more precisely, problems in radio interference emission and in interference immunity occur. The causes are often • inadequate layout (asymmetrical, capacitive and inductive coupling with other parts of the circuit) • faulty circuitry at the transmitter/receiver (asymmetrical, incorrect matching) • weak, coupled ground system If nothing can be done about the above points, filters are often the only s­ olution.

Common-mode filter

Common-mode filter (symmetrical filter) circuitry Common mode filters are rarely used apart from in power supplies. On the one hand, this may be because chip manufactures fail to make the necessity clear for some applications for marketing reasons, on the other, that the subject of filters is often only seen as a superfluous cost factor in t­oday’s “fully integrated chip world” and commonmode filters are somewhat more complex than L-C low-pass filters. Common-mode filters may not only reduce the emitted interference from the device, but also from the surroundings into the device. Depending on the filter type, potential isolation, improved matching properties and higher signal quality may be achieved depending on the filter type. Figure 1.94 shows various versions of common-mode filter circuit.

106

Fig. 1.94: Common-mode filter circuit versions On Figure 1.94a: The filter galvanically isolates the transmitter from the receiver by means of the transformer. A further advantage of the transformer is to make the signal symmetrical.

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I Basic principles

Useful signal Series-connected impedance

The transmitter does not need to have a push-pull stage. The capacitors C1 and C2 connected after the transformer filter the asymmetrical noise component, C3 the symmetrical. The capacitors of course also load the useful signal. Their effectiveness with respect to the useful signal, as well as the noise signal, essentially depends on the ­series-connected impedance (Figure 1.95).

Signal load

Fig. 1.95: Signal load properties in consideration of the filter transformation ratio

Transformation ratio

The transformation ratio of the transformer must be considered in regard to the useful signal. This leads to the relationship in Figure 1.95a and 1.95b:





108

(1.75)





(1.76)

Voltage divider relationship

and the voltage divider relationship from 1.95c:

(1.77)



A lower internal resistance and a higher transformation ratio reduce the i­nfluences on the signal, however, the parasitic properties, e.g. coupling c­ apacity of the transformer, are not taken into consideration. The parasitic capacitance – the coupling capacitance generated between the primary and secondary side of the transformer – primarily couples high frequency asymmetrical noise signals in the secondary filter stage. The effectiveness of the first filter stage is significantly reduced, especially with increasing frequency (Figure 1.96).

Parasitic properties RF asymmetrical noise signals

Fig. 1.96: Filter stage, taking parasitic coupling capacity of the transformer into ­consideration The transformation ratio of the transformer falls with increasing frequency as the influence of the coupling capacitance increases. If the internal resistances of the signal and noise sources are low, i.e. small com­pared with Ztot, the noise level at Ztot will be large. To reduce the high frequency noise components the filter must be extended. In Figure 1.94a, the chokes L1 and L2 and the capacitors C4 and C5 have been added. C6 and C7 on the receiver side reduce the any noise coupling from the transmission cable and high frequency noise emitted back from the receiver.

Transformation ratio

RF noise components

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I Basic principles On Figure 1.94b: The relevant differences to the filter from Figure 1.94a are discussed as f­ollows. Galvanic decoupling

Current-compensated choke

Galvanic decoupling takes place here by means of C1 and C2. Push-pull c­ ontrol is necessary for this circuit, whose outputs show the same impedance to ground. To reduce asymmetrical, high frequency noise, also within the signal bandwidth, the current-compensated choke L1 is used. The c­ apacitors C5, C3/C4 and C6/C7 serve to reduce the common-mode noise components; however they also influence the useful signal in combination with the source impedance and the common-mode impedance of the choke L1. If the useful signal is badly affected by harmonics, additional filter stages must be integrated (see Figure 1.94c). On Figure 1.94c:

Galvanic isolation

Only the essential differences to the previous circuit versions will be discussed here. Choke L1 with the associated capacitors C1 and C2 filter the low frequency range as previously described in Figure 1.94b. The high frequency range and high frequency signal components are filtered with L2, L3 and the associated capacitors C3, C4 and C5. It should be observed here ­however that between transmitter and receiver there is no galvanic isolation of any kind. The receiver side filter facilitates an effective reduction in noise signal components towards the receiver, which are, for example, ­externally excited via the cable path. On the output side, from the receiver towards the cable path, L4, L5 and the associated capacitors C6 and C7 ­reduce radio interference. On Figure 1.94d: The chokes L1 and L2 with the associated capacitors C1/C2/C3 reduce high frequency noise. The high frequency noise can often only be inadequately reduced in practice, as

Ground reference

• the capacitors C1 and C2 do not have a sufficient ground reference • the capacitance of the capacitor C3 is cannot selected high enough • the filter is dropped for “marketing” reasons • design preconditions do not allow for an adequate filter

Cable shield

An additional problem is the inadequate bonding of the cable shield with the chassis. The cable shield must be coupled to the chassis with “very” low impedance for high frequency transmission. How much is “very”? If the impedance of a common-mode cable is 75 W and the cable shield is connected to the chassis via an impedance of 0.5 W, there is a voltage drop Usc at the shield-chassis junction of



110



(1.78)

With





(1.79)

there is an effective shielding attenuation of 43.5 dB. 43.5 dB which is not enough for most applications. It should be at least 50 dB, but the impedance between cable shield and chassis often rises far above 0.5 W due to inadequate sheet-metal construction. The result is radiation of radio interference, which can be reduced by fitting a ferrite ring or ferrule. Shielded and unshielded cables must be distinguished when assessing functionality. Fitting a ferrite to a shielded cable introduces a high-loss (resistive) impedance into the noise signal path, which reduces the noise current in the ­effective frequency range. The lower the coupling between the inner conductor and the cable shield, the lower is the effect of the ferrite on the signal carrying conductor, reliable numerical values are however not available here. The ferrite has the effect of a current-compensated choke for unshielded cables ­(Figure 1.97), therefore the useful signal is not affected.

Shielding attenuation

Ferrite ring/ferrule

Noise signal path

Current-compensated choke Unshielded cables

Fig. 1.97: Functional principle of the toroidal ferrite used on unshielded conductors The insertion attenuation and the impedance can be increased by increasing the winding turns, but no more than 3 turns should be wound, as the capacitive coupling between the turns reduces the insertion at­tenuation above the resonant point (Figure 1.98).

Winding turns

111

I Basic principles

Fig. 1.98: Relationship between the number of winding turns, insertion impedance and resonant frequency of a toroidal ferrite

3.6 Filters for frequencies over 500 MHz Filters for frequencies over 500 MHz, including those for use in HF interfaces for antenna ports or for high-speed video, require special attention. The components, such as capacitors and inductors, must be considered separately and the layout must be specifically designed to work at these frequencies. The key considerations in the design of the filters are the frequency range which must not be affected, and the frequency range to be damped (filtered). This makes signal bandwidth an important parameter. A general distinction can be made between: High-speed signals

• High-speed signals, which are delay-critical but can be limited in its bandwidth (e.g. LVDS, HDMI), and • HF signals, with limited bandwidth, where the harmonics and (where necessary) subharmonics are to be filtered (e.g. RF WiFi).

112

Filter layout for frequencies over 500 MHz

Filter layout

What exactly turns a PCB design into one for use with frequencies over 500 MHz? First and foremost: high-speed design. These are designs where the signals have fast transitions that switch so rapidly that the changeover is complete before the signal has passed along the conductive track and reached its destination – the load. In this situation, the signal can be reflected back to the source, meaning that the original signal can be distorted or destroyed. Broadband signals like this, with high-frequency harmonic content, or short rise times, can also radiate out from the source and couple into neighboring conductive tracks, causing problems there. This means that the focus here is not just EMC, but functionality above all. With regard to circuit board design, matching and delay – i.e. length of conductive tracks – must be considered. In the case of HF signals with limited bandwidth, the main focus is on damping the desired signal – i.e. matching. The signal propagation time, or the synchronization of single signal channels to one another (e.g. LVDS, HDMI), is dependant on the lengths of the conductive tracks. If the conductive tracks are too long, or of insufficiently equal lengths, there will be time delays between the signals, which lead to malfunctions. How long is too long? If the switching speed of the transitions increases, the electric current passing through the conductor will behave differently. It will no longer conduct like water down a pipe; instead, most of the current is concentrated more on the surface of the conductor (this is known as the skin effect), and some of the energy moves as an electromagnetic wave. This electromagnetic wave is not conducted by its own conductor, but is carried through the material surrounding the conductor. This slows the movement of the signal, depending on the material. Delay-critical signals:

Signal delay

If there is a mismatch with the load, i.e. at the end of the conductor, a portion of the signal energy is reflected. If the next signal is already on its way from the source, due to the long lines and hence the long signal delay, interference will result between the “old” reflected portion of the signal and the new signal already sent. In summary, if the running time along this line is the same as or longer than the rise time of the signal, the integrity of this signal is doubt. This length is known as the critical length – the design should not feature longer lines than this. A rule of thumb for analysis of circuit board layout says that if the conductive track is longer than a third of the rise time, reflections may occur. For example, if the signal at the source has a rise time of 1 ns, then a conductive track longer than 0.33 ns – this is about 5 cm with an FR4 material – should be expected to create problems for signal integrity.

113

I Basic principles Calculation: The speed at which electrical energy can flow along a conductor is described as propagation and can be defined as follows: VP =



C εR 

(1.80)

where: Vp = propagation speed C = speed of light (299.792458 mm/ns) εR = dielectric constant The dielectric constant, eR for FR4 is 4.2. Therefore the speed of a signal in FR4 can thus be given as: V P FR4 =



299.792458 4.2

= 146.28

mm ns 

(1.81)

Using the “1/3 Rise Time” rule of thumb, the following applies: LL ≥



tR C x 3 εR 

(1.82)

where: LL tR C εR

= length of conductive track (in mm) = signal rise time (in ns) = speed of light (299.792458 mm/ns) = dielectric constant (4.2 for FR4)

If the signal rise time is tR = 1 ns then the length of conductive track above which signal integrity problems can be expected can be calculated for FR4 to be 48.75 mm But if the signal rise time tR = 100 ps, the critical length LL shrinks to 5 mm! Synchronization

Synchronization critical signals: If the signal is delay-critical because of a high data rate and it is divided across multiple channels, such as with LVDS, the signals of individual channels must reach the load at the same time. The signal propagation times of the individual channels may at most have a specific time offset between one another.

114

Continuing with the calculations above, the time offset of the signals can also be determined. The maximum permissible difference in length between two conductive tracks is calculated as follows:

Using the maximum permissible time offset tv (in ns), the maximum difference in length of the conductive tracks, ∆L (in mm), can be calculated. E.g. if tv = 0.1, the maximum difference in length, ∆L, is



ΔL =

V P (FR4) 146.28 mm/ns = = 14.6 mm tV 0.1 ns 

(1.83)

A filter is a frequency dependant voltage divider, which accordingly alters its impedance with the frequency. The connection between the signal source, via the filter, to the load takes place via the conductive track. In the desired frequency range, the transmission system must be adjusted so that the signal can be transmitted over its complete bandwidth while being affected as little as possible. In order to avoid reflections and to ensure the best possible signal transmission in the desired frequency range, the system, consisting of the signal source, the conducting path, filters and signal sink must be adjusted to the impedance. With the transmission system adjusted to the signal frequency no reflections will take place, even if the conductive track lengths exceed the critical length. Matching, as previously mentioned, is essential, i.e. the impedances of sources, lines and sinks must all be equal and the filter must stay “neutral” in the transmission frequency range. The impedance of the conductive track is determined by the dimensions of the routing (width and height of the copper track) and the properties and dimensions of the surrounding materials. Through careful arrangement of the conductive track structure and by calculating its dimensions and properties, a specific desired impedance for the conductive track can be achieved. We would like to point out here that high-performance design software is available for this purpose.

4 Transformers Basics 4.1 Functionality of a transformer A transformer is constructed from at least two windings with the number of turns NP on the primary side and NS on the secondary side. For the sake of simplicity, we consider an ideal transformer with a turns ratio of 1: 1. In the first step, we consider a transformer with open secondary winding NS (­Figure  1.99).

115

I Basic principles UP

IP +

t

+

IP

US

UP -

t

-

US

t

Fig. 1.99: Principle of a transformer with zero load. This ideal trans­former is wound as a bifilar so as to ignore parasitic effects A voltage uP is applied to winding NP. Due to the inductance of the winding, this produces a linearly rising current IP. The following relationship applies.





(1.84)

Winding NS also encloses this magnetic flux. Due to the change of the magnetic flux, a voltage is produced in this.



uS = –NS ·

dΦ dt 

(1.85)

If both equations are solved according to the change of flux and set equal, the following is obtained for the voltage transformation:



uS =

NS u NP P 

A current does not flow in winding NS as the winding is open.

116

(1.86)

If we now connect winding NS to a load resistance RL (Figure 1.100), the voltage in NS produces a current flow due to the load resistance: iS =



uS u N = P · S RL RL NP 

(1.87)

UP

t

IP

US

+ +

UP

US

t

IS

IS

-

t IP

t

Fig. 1.100: The same transformer but with a load The primary current is now composed of the transformed secondary current and the already linearly rising magnetization current.



i P = i S* +

up · dt L 

(1.88)

IS* transformed secondary current on the primary side. As no output can be generated, the transformed output is the same as the primary output put into the system. If the magnetization current is ignored, the following ­applies:



PP = up · iP = uS · iS = uP ·

NS N i → iS = P iP NP S NS 

(1.89)

117

I Basic principles Therefore currents are transformed reversed like voltages. The following also applies: RL =



uS u N u N2 N = P · S = P · S2 = RP · S iS iS NP iP NP NP

  2

(1.90)

Resistances are thus transformed with the squared turns ratio. This also a­ pplies for inductances, capacitances and impedances. The magnetizing current is not transferred to the secondary side. It is required to generate the magnetic field. The aim of the transformer design must therefore be to keep the magnetizing current as small as possible. There are two possibilities here: • Insertion of a highly permeable core to increase the primary inductance. This causes the magnetizing current to rise less steeply and is therefore smaller (Figure 1.101).

IP

without core with core

t Fig. 1.101: Magnetizing current of a transformer with and without a h­ ighly permeable core • Shorter voltage pulses with higher frequency are generated, as the rise in current stops at the end of the voltage pulse and starts again at the original point for the next pulse (Figure 1.102).

Fig. 1.102: Magnetizing current for a transformer at different driving f­requencies

118

4.2 Parasitic Effects In reality, there are still effects which influence the behavior of transformers. The most important are: • the leakage inductance • the coupling capacitance (capacitance between the windings) • the winding capacitance (capacitance within a winding) The Leakage Inductance If two windings are considered, the complete flux is not coupled with the other winding. A part of the field lines closes outside the other winding. This part of the inductance is called leakage inductance. In order to understand how the leakage inductance can be minimized, the parameters which influence it must be known. If a long cylindrical coil (Figure 1.103) is considered, its inductance is produced from:





(1.91)

lw = Length of the coil N = Number of windings A = Area of the coil

Lw

Fig. 1.103: Long solenoid If a second winding is now wound over it (Figure 1.104), the leakage inductance is produced from





(1.92)

119

I Basic principles Lw

Fig. 1.104: Long solenoid with second winding Where A is the area between the two windings. It can be calculated using:





(1.93)

MLT = Mean length turn – length of one turn Hins = Distance between the windings (insulation thickness) H1, H2 = Winding height of the windings 1 and 2 The leakage inductance is thus independent of the core material and of the air gap. For the minimization of the leakage inductance, either the length of the coil must be increased (wide windings) or the distance between the windings must be reduced (e.g. bifilar winding) or the turns reduced. Figure 1.105 shows various more or less optimal winding designs. For existing geometry, the most often used means is a interleaved design (Figure 1.105d), where one winding is wound between the other winding divided into two halves. The winding length is doubled in this way. The winding with the highest number of turns is split. Whether it is the primary or secondary makes no difference.

120

A) Best design: flat coil; small mechanical distance

B) Less favorable: large distance D2

C) As well unfavorable; additional overlap is missing

D1 D3

D2

D) First and second winding are interleaved

D4

D5

For better coupling primary winding can be split. This reduces the effective distance between the windings which reduces the leakage inductance to 25% of regular winding however, at the cost of higher winding complexity.

Fig. 1.105: Different winding structures The Coupling Capacitance The coupling capacitance between the two windings can be imagined as a plate capacitor between the two windings. It follows from this that it can be reduced by increasing the distance, reducing the area or changing the permittivity (dielectric constant) of the material between the them. All result directly in increasing the leakage inductance. The Winding Capacitance The winding capacitance establishes itself turn by turn as these are insulated from each other and at a different potential. It is higher the more layers are required within a winding. It can be reduced by various winding techniques, e.g. Z-winding (the wire is returned to the start side after each layer).

4.3 Transformer: Parasitic parameters and equivalent circuit

Interwinding ­capacitance

Intrawinding ­capacitance

Transformer

Which parasitic parameters are there in the transformer and how can they be measured and subsequently represented in a simulation model? We want to investigate this on the basis of a transformer and investigate using LTspice. “Ideal” transformer models are usually used to make it as easy as possible for the developer and to reduce the computation time in LTspice. Only the inductance values for the primary and secondary are required here, as well as the coupling factor K (here in statement K1 Lp LS set to 1 = ideal).

121

I Basic principles LTspice

K1 Lp Ls 1

in

V1

Lp

Ls

out

R1

AC 1

Fig. 1.106: Ideal transformer in LTspice Coupling factor

The simulation results are far closer to practice if the coupling factor is already taken into consideration:





(1.94)

because transformers have leakage inductances of 2% ~ 8% ( = a K of 0.9899 ~ 0.9592) depending on the construction.

122

We use the following equivalent circuit for further consideration and to determine the parasitic elements: CWW

Rpri

CWpri

LLpri

LLsec

Lpri

Lsec

Rsec

CWsec

Fig. 1.107: Transformer equivalent circuit Cww: winding – winding coupling capacitance Cwpri: primary-side winding capacitance Cwsec: secondary-side winding capacitance LLpri: total leakage inductance (primary + transferred secondary leakage inductance) Rpri: primary dc resistance Rsec: secondary dc resistance Lpri: primary inductance Lsec: secondary inductance The transformer 750 310 136 (see Figure 1.108) from Würth Elektronik serves as an example. The given parameters are: turns ratio → 6.11 : 1, resistance of the windings (1 & 4) and (5 & 8), the primary inductance, Lpri.

123

I Basic principles

Fig. 1.108: Dimensions and circuit of transformer 750 310 136 Measuring the primary and secondary inductance To measure the primary and secondary inductance, the respective winding not being measured must remain open. An inductance of 26.87 μH (100 kHz/100 mV) was measured between pins 5 & 8 and 939 μH (100 kHz/100 mV) between pins 1 & 4.

5

1

Measure across Pin 5 & 8

Pin 1 & 4 open

8

4

Fig. 1.109: Measuring the secondary inductance Turns ratio

The turns ratio n The turns ratio n can be calculated as follows:



124



For LPRI = 939 μH and LSEC = 26.87 uH the calculated turns ratio is 5.91.

(1.95)

Total leakage inductance

Leakage inductance

Primary leakage inductance and transferred secondary inductance can be measured by short-circuiting the secondary winding (pins 5 & 8) and measuring between pins 1 & 4.

Short circuit secondary Pin 5 & 8

5

1 Measure across Pin 1 & 4

8

4

Fig. 1.110: Measuring the total leakage inductance across the primary The result for LLpri is 36.5 μH

Please note: The leakage (also called stray) inductance is in series with the transmission path. The leakage inductance describes that part of the magnetic field, which is not enclosed by the other windings and so does not contribute to the coupling. The leakage inductance results simply from the mechanical arrangement of the windings against each other. Techniques to decrease the leakage inductance usually result in an increase of the coupling capacitance. The total leakage inductance (primary inductance + transferred secondary leakage inductance) is measured by measuring at short circuited secondary winding (Please note: To not distort the measurement result a low impedance short circuit is necessary).

Many applications demand as small a leakage inductance as possible. It can be minimized using various winding techniques. The windings should be as wide as possible. An interleaved construction also helps, as in the case of the proximity effect. However, these techniques increase the coupling capacitance between the primary and secondary sides. DC current winding resistances Rpri and Rsec between pins 5 & 8 and 1 & 4, see Figure 1.109, respectively can be measured with an ohmmeter (e.g. Metra Hit 27I). Rpri: 265 mW and Rsec: 858 mW 125

I Basic principles Coupling capacitance

Coupling capacitance Additional parasitic parameters include the coupling capacitance (capacitance between the primary and secondary sides) and the winding capacitance (capacitance between the turns of a winding). The influence of coupling capacitance on the circuit can be reduced by shielding windings between the primary and secondary sides. However, minimization of the coupling capacitance by winding in several sections or by inserting thick insulation between the primary and secondary side directly causes an increase in leakage inductance. The coupling capacitance can be measured directly. The winding capacitance is measured indirectly via the resonance between the main inductance and the capacitance. An LCR bridge is used to measure from winding to winding, in this case between pins 5 & 1. For measurement reasons both windings should be separately short-circuited so the measurement result is not distorted.

Measure across Pin 5 & 1 Short circuit secondary Pin 5 & 8

5

1

8

4

Short circuit Primary Pin 1 & 4

Fig. 1.111: Measuring the coupling capacitance Cww: 32 pF Winding capacitances

Winding capacitances The winding capacitances can only be determined indirectly from the resonances with the main inductance (Lpri & Lsec). The impedance with the secondary side “open” is measured with an impedance analyzer. The winding capacitance of the primary side is then calculated from the resonant frequency.

Fig. 1.112: Resonant frequency measurement taking the example transformer 126

The resonant frequency is 875 kHz, the measurement resulted in Lpri with 939 μH.



(1.96)



Lpri = main inductance Cwpri = winding capacitance = resonant frequency f Rearranging the formula for Cwpri results in Cwpri 35 pF and for Cwsec 1.2 nF The same approach is also taken on the secondary side. This produces the following simulation equivalent circuit: Cww

V1 AC 1 Rser=50

32pF

Rpri

LLpri

0.858

36.5µH

in

Cwpri

939µH

.ac oct 25 1Hz 50000kHz

out

0.265

Lpri

35pF

Rsec

Lsec

Cwsec

R1

26.87µH

1.2nF

50

K1 Lpri Lsec 1

Fig. 1.113: Simulation equivalent circuit The simulation then produces the following transfer frequency response for the ­transformer: Frequency response Transformer

V(out)/V(in)

-6dB

-12dB -18dB -24dB -30dB -36dB -42dB -48dB -54dB -60dB -66dB -72dB

1Hz

10Hz

100Hz

1KHz

10KHz

100KHz

Fig. 1.114: Transfer frequency response of the transformer

1MHz

10MHz

127

I Basic principles The discrete equivalent circuit can presented in further simplified form, because LTspice offers the option of including the coupling factor, Rpri, Rsec; Cwsec and Cwpri in the components Lpri and Lsec, and of defining the leakage inductance through the K statement. Cww in

V1 AC 1 Rser=50

out

32p Lpri 939µH

Lsec 26.87µH

R1 50

K1 Lpri Lsec 0.9806 .ac oct 25 1Hz 50000kHz

Fig. 1.115: Coupling factor by K-statement In this case: Parallel capacitance corresponds to Cwsec. Series resistance corresponds to Rsec. The coupling factor is calculated from: Coupling factor



(with LPRI: 939 μH; LLpri: 36.5 μH) and then enter in the LTspice text editor as SPICE ­directive.

Fig. 1.116: LTspice text editor

128

(1.97)

Simulated transformer frequency response in LTspice is identical.

V(out)/V(in)

-6dB

-12dB -18dB -24dB -30dB -36dB -42dB -48dB -54dB -60dB -66dB -72dB 100KHz

1MHz

10MHz

Fig. 1.117: Simulated transformer frequency response in LTspice. Upper frequency portion shown. Further calculation formulas for separating the equivalent parameters for the models with mutual inductance Lm:

Rpri

LLpri

LLsec

Rsec

Lm

Fig. 1.118: Simplified transformer equivalent circuit for simulation in SPICE



(1.98)



From this follows for the different inductances:

Lm = k · Lopen(1.99)



LLKG(PRI) = (1 – k) · Lopen(1.100)





(1.101) 129

I Basic principles The values for the resistances are determined by simple measurement with the ohmmeter. This model does not consider core losses, any capacitances or the frequency dependence of resistances due to the skin and proximity effects.

4.4 Function and application areas of transformers As a result of their functionality, transformers can be used for various tasks: • Isolation: Transformers are constructed of several windings. Depend­ing on the ­additional isolation, various potentials can be ­separated or isolated from one another. • Voltage transformation: Voltages are transformed proportionally to the turns ratio. • Current transformation: Currents are transformed inversely to the turns ratio (chapter 1/6.14). • Impedance matching: Impedances are transformed as the square of the turns ratio. This gives rise to various applications for transformers: • Voltage (power) supplies: Here the main functions of the trans­former are voltage transformation and isolation. • Current converters: Here the main function is to convert high ­currents into small measurable currents. • Pulse transformers, e.g. drive transformers for transistors: The main function is isolation; sometimes higher voltages are also required to drive a transistor. • Data transformers: Here the main function is also isolation. In a­ ddition, sometimes different impedances have to be matched or voltages increased.

4.5 Requirements for data and signal transformers Transformers are used on data lines mainly for isolation and impedance matching. The signal should be largely unaffected in this case. From chapter 1/4.1.2 we know that the magnetizing current is not transferred to the secondary side. For this reason, the transformer should have the highest possible main inductance. The signal profiles are usually rectangular pulses, i.e. they include a large number of harmonics. For the transformer, this means that its transformation properties should be as constant as possible up to high frequencies. Taking a look at the transformer equivalent circuit (chapter 1/4.3), it is a­ pparent that the leakage inductances contribute to addition frequency-dependent s­ ignal attenuation. The leakage inductances should therefore be kept as low as possible. Signal transformer bifilar

130

Signal transformers therefore usually deploy ring cores with high permeability. The windings are at least bifilar; wound with twisted wires is better still. Because the power transfer is rather small, DCR is of minor importance.

The direct parameters, such as leakage inductance, interwinding capacitance etc. are usually not specified in the datasheets for signal transformers, but rather the associated parameters, such as insertion loss, return loss, etc. The most important parameters are defined as follows: Insertion loss

• Insertion loss IL: Measure of the losses caused by the transformer



(1.102)



Uo = output voltage Ui = input voltage • Return loss RL: Measure of the energy reflected back from the transformer due to imperfect impedance matching



Return loss

(1.103)



ZS = source impedance ZL = load impedance • Common Mode Rejection: Measure of the suppression of dc inter­ference • Total harmonic distortion: The relationship between the total energy of the harmonics and the energy of the fundamental • Bandwidth: Frequency range in which the insertion loss is lower than 3 dB

Common mode ­rejection Total harmonic distortion Bandwidth

4.6 A transformer’s effect on return loss Return loss Return loss is an expression in decibels (dB) of the power reflected on a transmission line from a mismatched load in relationship to the power of the transmitted incident signal. The reflected signal disrupts the desired signal and if severe enough will cause data transmission errors in data lines or degradation in sound quality on voice circuits.

Return loss

The equation for calculating return loss in terms of characteristic complex line impedance, ZO, and the actual complex load, ZL, is shown below:





(1.104)

131

I Basic principles Expanding the return loss equation to terms of resistance and reactance we have this formula:





(1.105)

Since return loss is a function of line and load impedance, the characteristic impedance of a transformer, inductor or choke will affect the return loss. A simple impedance sweep of a magnetic component reveals that the impedance varies over frequency, hence the return loss varies over frequency. We will discuss further the effects of a transformer on return loss later. Now let’s explore the relationship of return loss to other common reflection terms. Reflection coefficient

Reflection Coefficient While return loss is generally used to denote line reflections in the magnetics industry; a more common term in the electronics industry for reflections is the complex reflection coefficient, gamma, which is symbolized either by the Roman character G or more commonly the equivalent Greek character Γ (gamma). The complex reflection coefficient Γ has a magnitude portion called ρ (rho) and a phase angle portion Φ (Phi). Those of you familiar with the Smith Chart know that the radius of the circle encompassing the Smith Chart is rho equal to one. The reflection coefficient, gamma, is defined as the ratio of the reflected voltage signal in relationship to the incident voltage signal. The equation for gamma is below:





(1.106)

Keep in mind that just as impedance is a complex number, so is gamma and it may be expressed either in polar format with rho and Phi or in rectangular format:

Γ = ρ∠Φ = Γr + jΓi(1.107)

Return loss expressed in terms of gamma is shown in the equation below:

Standing wave ratio

132

RL = –20 Log10 |Γ| dB(1.108)

Standing wave ratio The reflections on a transmission line caused by impedance mismatches reveal themselves in an envelope of the combined incident and reflected wave forms. The standing

wave ratio, SWR, is the ratio of the maximum value of the resulting RF envelope EMAX to the minimum value EMIN.

Fig. 1.119: Standing wave ratio The standing wave ratio expressed in terms of the reflection coefficient is shown below:





(1.109)

Transmission loss

Transmission loss

The last signal reflection expression that we will discuss is the transmission loss. Transmission loss is simply the ratio of power transmitted to the load relative to the incident signal power. Transmission loss in a lossless network expressed in terms of the reflection coefficient is shown below:

TL = –10 Log10 (1–|Γ|) dB(1.110)

Don’t forget that the magnitude of gamma (|Γ|) is rho (ρ) and either form can be found in publications and documents regarding reflections. Related terms Reviewing the complex reflection coefficient formula we can see that the closer matched the load impedance ZL is to the characteristic line impedance ZO the closer to zero the reflection coefficient is. As the mismatch between the two impedances increase the reflection coefficient increases to a maximum magnitude of one.

133

I Basic principles The table below shows how the varying complex reflection coefficient relates to SWR, return loss and transmitted loss. As can be seen a perfect match results in SWR equal to 1 and an infinite return loss. Similarly an open or short at the load will result in return an infinite SWR and 0 dB of return loss. VSWR

lΓl

1.0

0.00

Return loss (dB) ∞

1.05

0.02

32.3

Transmitted loss (dB) 0.00

0.00

1.1

0.05

26.4

0.01

1.2

0.09

20.8

0.04

1.5

0.20

14.0

0.18

2.0

0.33

9.50

0.50

3.0

0.50

6.00

1.25

9.0

0.80

1.94

4.44

50.0 ∞

0.96

0.35

1.00

0.00

11.14 ∞

Tab. 1.3: Relationship between reflection coefficient – standing wave ratio Plotted on a Smith Chart the relationship is even more evident as constant values of all four parameters are graphed on the chart as circles.

Smith chart 134

Fig. 1.120: Smith chart

Maximum power transfer Maximum power transfer is obtained from the source to the load when the source impedance is equal to the complex conjugate of the load impedance. This not only maximizes power but minimizes reflection energy back to the source.

Power matching

RS + XSj = RL – XLj(1.111)

Fig. 1.121: Complex source and load Return loss with matched load

Matching

Let’s take an example of a matched line and load. Let’s say that ZO = 100 W in an ADSL application and that it is terminated with a purely resistive load of 100 W.

Fig. 1.122: Return loss

135

I Basic principles





(1.112)

where: ZO = 100 + 0j W ZL = 100 + 0j W Since the load and source are purely resistive, the return loss will be the same at any frequency. Substituting and calculating shows that RL = ∞. Mismatching

Return loss with unmatched load Let’s take the same example of an ideal transformer, but with a slightly unmatched load. Let’s say that ZO = 100+0j W as before, but now we will calculate return loss at a number of purely resistive load impedances to show how return loss if affected by mismatch. The resistive load is used again so that the return loss will be independent of frequency. RL (W)

Return Loss (dB)

80.0

19.1

95.0

31.8

99.0

46.0

99.9

66.0

100.1

66.0

101.0

46.0

105.3

31.8

125.0

19.1

Tab. 1.4: Return loss at mismatching The results show that return loss is a function of mismatch and without regard to which direction the mismatch is in. If we look at the case of a slightly mismatched line versus load we see that it is independent of frequency if the line and load are purely resistive. Also note that if the match was perfect, the return loss would be infinite.

136

Fig. 1.123: Return loss Return loss with nearly ideal transformer Now let’s take the same example of a matched line and load, but add in a 1 : 1 transformer which is ideal except for having a primary inductance of LPRI = 600 µH. Again we assume the line impedance is a purely resistive 100 W as well as the load impedance. When we had an ideal transformer with both line and load impedances purely resistive, our return loss did not vary over frequency and was the same at any frequency. Now however, the inductance will vary over frequency thereby causing the effective load to vary over frequency. The return loss calculation also becomes more complex due to the load impedance now being complex. Rather than go through all of the complex impedance calculations, here are the steps required to calculate the return loss. Step 1: Using impedance transformation calculations, transform the impedance to the same side of the ideal transformer as the primary inductance. In this case the ideal transformer is a 1 : 1 transformer and the load does not change.

137

I Basic principles

Fig. 1.124: Transformer Return loss Step 2: Combine the XL the current ZL = RL+0j with a resultant ZL’ which is complex.



(1.113)



Fig. 1.125: Return loss with impedance ZL‘ Step 3: Calculate return loss using the resultant load and the original resistive line impedance.

(1.114)

138



Results: Looking at the results over frequency we can see that the inductance at the lower end is mismatched due the inductance shorting out the load. The lower the

primary inductance the more the load will be shunted. Looking at the graphed results we see that return loss due to the primary inductance will behave much like a filter in that it has a knee which will vary with inductance and the slope after the knee is 20 dB per decade. Frequency (kHz)

Return Loss (dB)

0

0.00

1

0.02

10

1.95

100

17.62

1000

37.55

10000

57.55

Tab. 1.5: Return loss with 600 µH Lpri at an ideal transformer

Fig. 1.126: Return loss with 600 µH Lpri Return loss with leakage inductance added

Fig. 1.127: Return loss with leakage inductance Now let’s add leakage inductance of 1 µH to the same transformer under the same load conditions. The effective load is calculated in the same manner with ZL’ the reactance of the primary in parallel with the load impedance after transformation. The ZL’’ is ZL’ with the series leakage inductance reactance added to it.

139

I Basic principles

Fig. 1.128: Return loss with leakage inductance and ZL‘

ZL″ = XLKGj + ZL′(1.115)

Using the same return loss formula we can then calculate the return loss at various frequencies. From the graphed results we see that the high frequency return loss is affected by the leakage inductance. Frequency (kHz)

Return Loss (dB)

0

0.00

1

0.02

10

1.95

100

17.42

1000

27.01

10000

10.43

Tab. 1.6: Return loss with 600 µH Lpri at a leakage inductance of 1 µH

Fig. 1.129: Return loss with 600 µH Lpri and 1 µH leakage inductance

140

For most transformers the primary and leakage inductances will have the greatest effect on return loss, providing that the turns ratio chosen effectively matches the load resistance to the line impedance.

Return loss with a less-than-ideal transformer With the linear transformer model that is typically used in low frequency transformer design applications, we can calculate theoretical return loss based on lumped para­ meter analysis. With the exception of interwinding capacitance, we can reduce the linear transformer model to a load impedance by either combining the elements in parallel or series. Keep in mind that the secondary dc resistance and the ZL have to be transformed by dividing by n2 when brought to the line side of the model.

Fig. 1.130: Return loss of real transformers Interwinding capacitance can not be so simply modeled because it resides on neither the line nor the load side of the model and can not be transformed into the equivalent load. At low frequencies the interwinding capacitance acts as an open across the transformer and typically can be ignored. In fact most modeling programs for transformers ignore interwinding capacitance as leakage inductance and primary inductance are the dominant elements. However in certain designs where interwinding capacitance is fairly large and the operating frequencies are high, it can become a very significant factor. Suffice it to say that if interwinding capacitance needs to be included in the model it would be wise to use a more sophisticated analysis program such as LTspice. Let’s take a look now at the linear transformer model for the theoretical ADSL transformer shown below with a load that is just off of the ideal 25 Ω for a perfect match. We will take this and model the effect of the various elements looking at it parameter by parameter.

Fig. 1.131: Return loss of ADSL transformer 141

I Basic principles Return loss: Effect of DCRs The effect of dc resistance on return loss in the example below highlights two observations. First of all even though the secondary resistance is lower at 1.5 W in comparison to the primary resistance of 3.0 W, the effect on return loss is much greater. The reason for this is that the 1.5 W secondary when reflected to the primary side of the transformer is seen as 6.0 W. Also note that a lower return loss number is only slightly affected by other elements that have significantly better return loss when standing alone. The return loss when due only to the secondary resistance is roughly 30 dB while the return loss due to the primary resistance is roughly 37 dB. When combined, the net effect is a return loss of 27 dB.

Fig. 1.132: Return loss Leakage inductance

Return loss: Effect of leakage inductance and distributed capacitance The effects on return loss by the leakage inductance and the distributive capacitance parameters of a transformer are interesting to compare as well. We see from the example below that the effects due solely to leakage inductance reveal a decaying return loss at the rate of 20 dB per decade. Now taking a look at the distributed capacitance we see that it causes a high-end decay at the same rate with the knee at a higher frequency. The comparison gets interesting when we looked at the combined effect. When combined the net result is an improvement of return loss. Why is this? If you remember in our previous discussion, the return loss is a function of mismatch without regard to which direction the mismatch is in. With this example the mismatch is in opposite

142

directions so the addition of the distributed capacitance effect actually improves the overall return loss. Thinking about this in analytical terms, what is happening in the equivalent circuit? The reflected load is being increased by the reactance due to the leakage inductance thereby causing mismatch. However the reactance of the distributed capacitance is in parallel there by reducing the mismatch back toward the optimal 100 W reflected load.

Fig. 1.133: Return Loss with leakage inductance Return loss: Effect of interwinding capacitance As mentioned earlier, the effect of interwinding capacitance is very difficult to calculate using simple equivalent impedance transformations. The problem is that the interwinding capacitance is shared by both windings and is not clearly on one side of the ideal transformer or the other. The impact to the circuit model is therefore not so straight forward and requires more sophisticated modeling techniques. The example below was modeled with PSPICE rather than with simple calculations.

Interwinding ­capacitance

Typically however, the interwinding capacitance has very little effect on the return loss in comparison to the leakage inductance and can be ignored. A word of warning is in order however since cases where leakage inductance is very low while at the same time the interwinding capacitance is very high, interwinding capacitance can become a factor to reckon with.

143

I Basic principles

Fig. 1.134: Return Loss and interwinding capacitance Core losses

Return loss: Effect of resistive core loss and inductance In this example we compare return loss due to the primary inductance as well as to the resistive core loss assuming a core loss of Rc equals 10 k (W). As can be seen from the return loss due to the combined effect, the resistive core loss has very minimal impact. In very low frequency applications, such as audio, the resistive core loss can be a factor.

144

Fig. 1.135: Return loss and core loss/L-value Return loss: Effect of all parameters Finally, looking at the effects of all parameters combined we can determine which are the significant factors in a typical transformer application. As can be seen from the results below, the leakage inductance and the primary inductance are the driving factors. While the other parasitic parameters do play a role in shaping the return loss response, they play a relatively minor role in a typical transformer design.

145

I Basic principles

Fig. 1.136: Return loss with all parameters A closer look at dominating parameters In closing we take a closer look at the dominating parameters of the transformer. The top graph shows the return loss of various models in comparison to the ideal transformer with the slightly mismatched load. Then the lower graph just zooms in on the non-ideal transformer cases.

Practical tip: These graphs highlight the fact that primary and leakage inductance are the parameters that typically dictate return loss and that there is justification to ignoring interwinding capacitance in most applications.

146

Fig. 1.137: Return loss and influence of the dominating parameters Lprim/Lleak

147

I Basic principles 5 Ethernet and Power-over-Ethernet Basics 5.1 The history of Ethernet Ethernet comes from the word Ether, which represented the “space” where electromagnetic waves propagated according to 18th and 19th century scholars. The Ethernet protocol was first used at the University of Hawaii in the early 1970s, in order to simplify communication between the various campuses on the island. It was based on a radio protocol named Aloha (Hello). At the time, manufacturers had all developed their own communication protocols. It rapidly became clear that it would be impossible to connect these systems without creating an international standard. In 1977, the International Standards Organization (ISO) created a complete set of recommendations on these compatibility requirements to allow connection between open systems, i.e. systems using the same protocols and standards, to exchange data. The initial results obtained with Ethernet in 1980 drove XEROX, Intel and Digital to join forces to develop this network and hence put an end to all the private protocols. In 1983 the IEEE finalized the standard 802.3 for the Ethernet. In 1984, ISO adopted and published an update of its recommendations under the name: Reference OSI (Open Systems Interconnection). This update has become an international standard and serves as a guide. The adventure could now begin …

5.2 The OSI reference model The model is based on a principle of Julius Caesar: Divide and rule. The basic principle describes networks as a group of stacked layers (7), which form the interface between the local application and the data transmission equipment. Companies use this model to ensure their systems communicate correctly with those of different manufacturers.

148

Fig. 1.138: OSI reference Imagine a message sent from sender A to receiver B. The message generated by an application from A, will pass through the successive layers via the interfaces between each layer until it finally reaches the physical media. When the message arrives at its destination, it goes back up through the layers of receiver B via the different interfaces, ending up at the application assigned to process it. Functions of the layers The Physical layer provides the mechanical, electrical and functional means to maintain and deactivate the physical connections used for data transmission. The design of the physical layer can really be considered as part of the electrical engineer’s domain. The physical elements include the media (connector (e.g. RJ45) and cable), encoders, modulators, repeaters, transceivers which includes the hubs used to connect various branches of the network together. The Data Link layer manages the transmission of data frames, acknowledgements, flow control, error detection and recovery providing error free data. The data link elements includes bridges which join networks of different protocols and switches which are intelligent hubs. A layer 2 switch works at this level using forwarding rules based on the MAC address The Network layer sends the data packets to the layers. It therefore manages the destination addresses and routing and regulates transmission. The network elements are routers which use only source and target address to route or filter transmissions and switches operating at level 3 (forwarding based on IP ad-

149

I Basic principles dresses). Internet based phones (VoIP) use this level to achieve higher priority on the network for smooth voice communication. The Transport layer complements the work of previous layers, controlling errors and optimizes the transmission. The transport elements include gateways which are a type of advanced router that is capable of using different protocols. This layer is in a position to read the data from the source protocol and to package it in another target protocol. The Session layer controls sessions, defining the rules for connections between sender and receiver. The Presentation layer formats the data (encryption, compression, etc.) so that the receiver can understand it. The Application layer is the final layer, where the applications actually using the network (e-mail, chat, etc.) interface with the user.

5.3 What is Ethernet (standard 802.3)? Basic principle In an Ethernet network, all the systems are connected to a single transmission line, formed by a cable. There are several physical layouts, depending on the cable chosen and the size of the whole LAN. These include bus, star, token ring, ring, mesh and tree topologies. Communication is achieved with the help of a protocol. CSMA/CD (Carrier Sense Multiple Access with Collision Detect), is a multiple access protocol that tracks the carrier and detects collisions. Any system is allowed to transmit over the line at any time; this is achieved very simply: • Each system checks that the line is free before transmitting. • If two systems transmit simultaneously, there is a collision. • When a collision occurs, the two systems stop sending and wait for a random length of time before transmitting again. Summary of different versions of Ethernet We have seen that standard 802.3 governs the way of transmitting the signal, but it also specifies the type of cable and the connectors. All twisted pair systems use the 8P8C connector commonly referred to as RJ-45. 150

Both 10BASE-T and 100BASE-TX use only two cable pairs, one for transmitting and one for receiving. 1000BASE-T and higher use all four cable pairs operating in full duplex mode (transmitting and receiving), where the switches (or routers) look after the access sequences so collisions no longer arise. Here is a summary of the different variants of the Ethernet protocol. Name

Maximum Length

Cable Required

Cable Bandwidth

Spectral Bandwidth

10BASE-T

100 m

UTP Cat. 3, 2 pairs

16 MHz

10 MHz

100BASE-T

100 m

UTP Cat. 5e, 2 pairs

100 MHz

31.25 MHz

1000BASE-T

100 m

UTP Cat. 5e, 4 pairs

100 MHz

62.5 MHz

2.5GBASE-T

100 m

UTP Cat. 5e, 4 pairs

100 MHz

100 MHz

5GBASE-T

100 m

UTP Cat. 6, 4 pairs

250 MHz

200 MHz

10GBASE-T

100 m

UTP Cat. 6A, 4 pairs

500 MHz

400 MHz

25GBASE-T

30 m

Cat. 8, 4 pairs

1600 MHz

1000 MHz

40GBASE-T

30 m

Cat. 8, 4 pairs

1600 MHz

1600 MHz

Tab. 1.7: Ethernet variants using twisted pair wiring The acronym includes all the information needed to know the characteristics of the variant. The first number is the speed in MHz or GHz (G). ‘BASE’ means broadband and ‘T’ means twisted pair. Protocol Structure

Fig. 1.139: Structure of Ethernet “variants” in relation to OSI model 151

I Basic principles To simplify writing, the Ethernet card is usually abbreviated to NIC (Network Interface Card). A bit like the OSI model, Figure 1.139 shows us how to move up the successive layers of the standard 802.3. Depending on the speed chosen (and therefore the variant), there are obligations and technical constraints to respect in both the software and hardware. The MAC software layer is often integrated in a microcontroller, in which case the physical layer is controlled by an active device named PHY. If the MAC layer is not included in the microcontroller, then the active device is commonly called COMBO (MAC/PHY). Many semiconductor manufacturers have solutions to meet these needs, such as ­Cirrus Logic, Intel, Marvell, SMSC, etc. We will see this later.

WE-LAN

WR-MJ

TXP

J1

TXN

J2

RXP

J3

RXN

J4

WE-RJ45 LAN

Fig. 1.140: Shows the basic diagram of a physical layer. The most commonly used variants are 100BASE-TX and 1000BASE-T in the tertiary sector. In the primary and secondary sectors 1000BASE-Lx or 1000BASE-Sx (both fiber optic) are mainly used. There will also be a shift in the near future towards 10GBASE-T or similar. The different versions are reviewed here in more detail. 10 Mbps Ethernet: 10BASE-T This variant offers a speed of 10 Mbps for serial communications. It uses Manchester encoding (see Figure 1.101) for robustness but it does double the signal frequency to 20 MHz. Although 10BASE-T can be used on a standard telephone cable, the most 152

popular link is two pairs of a UTP cable (Unshielded Twisted Pair) category 3 or higher, terminated at each end with an 8-pin RJ45 connector. It also accepts half or full duplex mode, respectively meaning simultaneous or nonsimultaneous transmission and reception. 10BASE-T is the first protocol including an integrity test determining network status. When turned on, the PMA (Physical Medium Attachment) sends a link request (NLP: Normal Link Pulse) to the NIC at the other end of the line, repeating this every 16 ms if there is no response. The link is established once the two NICs have exchanged a valid NLP. The link between the connector and the active device is often called a LAN transformer but is in fact an assembly of several components. For 10BASE-T it may include: • A transformer for galvanic insulation (which may have a transformation ratio other than 1) • A common mode choke to filter • A low pass filter (Fc (Center frequency) at about 17–18 MHz) • Resistors for distortion The choice of active device performing the “PHY” function determines the elements used to build this transformer assembly. Whether or not the common mode inductor is included with the LAN transformer assembly depends on whether interference is likely to occur on the transmission line. Since Würth Elektronik is a specialist in EMC filtering, it is wise to seek a recommendation for a transformer assembly that includes it.

153

I Basic principles TD+ 1

1:1

16

TX+

14

CTX

15

TX-

10

RX+

CRD 6

11

CRX

RD-

9

RX-

CTD 3 TD-

2

RD+ 7

1:1

8

Fig. 1.141: Diagram of WE-LAN 749020100A To optimize integration, RJ45 connectors exist that include the LAN transformers, common mode chokes, filters, terminations and LEDs built in . There is a wide choice of LAN transformers on the market for 10BASE-T with different structures for the TX and RX. The two key points are knowing the turns ratio the active device needs and whether it needs a low pass filter. The transformer and common mode choke structure is the basis for building LAN transformers for Ethernet variants with faster throughputs. 100 Mbps Ethernet: Fast Ethernet In 1995 three separate standards were created to achieve 100 Mbps. 100BASE-TX and 100BASE-T4 were developed in 1995, and 100BASE-T2 in 1998. Each were defined with different encoding principles and MDI interface connections. • 100BASE-TX uses 4B/5B encoding and can transmit over 2 pairs of a Cat. 5 UTP cable (STP Cat. 1) or over optical fiber. • 100BASE-T4 uses an 8B/6T code over a UTP cable, Cat. 3 or better. • 100BASE-T2 uses a PAM5 encoding over 2 pairs of a UTP cable, Cat. 3 or better. As is often the case when several solutions are developed for the same purpose, not all of them gain widespread recognition.

154

100BASE-X, 100BASE-TX or 100BASE-FX This protocol was designed to accept 2 transmission modes.

Fig. 1.142: Reference model for 100BASE-TX The encoding is identical, only the physical interface differs. Most manufacturers offer both in their IC. The internal structure of a LAN transformer often includes, not only the transformer, but also a common mode choke and possibly a shunt inductor. These components are repeated for each pair making the LAN transformer more like an assembly or module. The key points to bear in mind are as follows: The use of shunt inductors is not determined by the PHY manufacturer but by the characteristics of the network that, depending on the load, may have capacitive ­properties, inductance being designed to offset this effect. If your transformer has a 1:1 turn ratio and a common mode choke, you can install it in any order you want, despite the concerns of some engineers. There are some secondary component differences (see the Bob Smith recommendations) but it works in both cases. See our references 749 020 100A or 749 010 0110A. There is also a very broad range of RJ45s including the LAN transformer with or without the control LEDs (link, collision, etc.) in through hole or SMD version. The characteristics are identical,

155

I Basic principles the only aim being to save space on the PCB. (Würth Elektronik also has a line that includes varistors too). NB: Manufacturers of PHY (according to the Ethernet standard) allow visual checking (via LEDs) of line operation, usually giving the following information: Connection established, Tx, Rx, collisions, etc. It is useful to have this option with an “office” type application, to rapidly check line status, and even more useful with on-board applications, which is why we find these two options with RJ45 LAN transformer. 1000 Mbps Ethernet: Gigabit Ethernet After the 100BASE-T took off, a group of professionals from about 120 companies working in the network, computer and semiconductor fields set up the Gigabit Ethernet Alliance to develop a new standard (Standard 802.3ab in June 1999), with a view to increasing bandwidth (speed) to 1000 Mbps! Why a new speed? Once 100 Mbps became commonplace, network administrators were increasingly faced with saturation problems on some sections of the network. In fact, the cost of a 10/100 Mbps Ethernet card has significantly dropped over time, so network speed has naturally risen. The higher speed on a twisted pair was therefore a logical consequence. Previously this market is predominantly that of corporate networks, but today the falling prices are making it commonplace as well. Please note that the Cat. 5e twisted cables are certified for frequencies up to 100 MHz. The move to 1000 MHz was far from trouble-free, requiring a complete overhaul of the physical layer! To sum up, the standard had to create a level 5 (PAM5) encoding to transmit two bits per symbol at 125 Mbps simultaneously over the 4 pairs of a Cat. 5e UTP cable. Various factors should be considered in the selection of a LAN transformer, such as EMC aspects, layout guidelines, especially for high-speed applications. For this reason Würth Elektronik offers a large selection of discrete and integrated RJ45 transformers with various configurations. 2.5/5 GMbps Ethernet In 2016, IEEE ratified the IEEE 802.3bz standard defining 2.5GBASE-T and 5GBASET, boosting the traditional network speed up to five times using existing cabling. This variant uses the same encoding at 10G but at one quarter or one half the speed, enabling it to operate over Cat. 5e cabling 10 GMbps Ethernet With Gigabit Ethernet becoming more commonplace, the network carriers needed to increase their speed. In 2002 the first 10 Gigabit Ethernet standard was published. 156

Subsequently standards were added to encompassed different media types (fiber, wire, etc.) and in 2009 all the standards were consolidated into IEEE Std 802.3-2003an. This needs a special twisted pair defined as Cat. 6a. 25G, 40G Mbps Ethernet The push for increasing speed continues with 25 and 40 Gigabit defined in IEEE 802.3bq-2016. Using special Cat. 8 cabling, they are limited to 30 m and is currently the limit for twisted pair cabling. Higher speeds and increased distances use fiber optic cables.

D1+

J1

J2

D1-

75 Ω

D2+

J3

J6

D2-

75 Ω

D3+

J4

J5

D3-

75 Ω

D4+

J7

J8

D4VCC

75 Ω

4x0.1u F

1000pF 2kV

GND

Fig. 1.143: Example of RJ45 LAN transformer with integrated filter, 7499111001A

157

I Basic principles 5.4 The different types of encoding for Ethernet A binary signal is a series of 0s or 1s. Much used in an electronic communication, it is nonetheless very foolish and not very efficient to send it as it is over a data line. In fact, there is no way of checking that the message sent has been correctly received (there is no way of checking for corrupted data). To address this problem, many encoding processes of varying complexity check the integrity of a signal, add conditions for start, stop, etc. In this instance, we shall simply give the principle underlying the various codes used in the Ethernet protocol. Manchester or Biphase code Use: Ethernet 10BASE5, 10BASE2, 10BASE-T. Its principle makes it one of the simplest codes. A binary signal of period T, the 1 or 0 bit is represented by a transition of the signal to T/2 instead of a state.

Fig. 1.144: Binary signal of period T The advantage is that it is not very sensitive to transmission errors. On the other hand, it can be sensitive to line interference.

158

nB/mB encoding Use: 4B/5B: Fast Ethernet (100BASE-TX); 8B/10B: Gigabit Ethernet. This is block-by-block encoding: 4 bits of the signal are encoded over 5 bits selected from a transcoding table. Group of 4 bits

4B5B Symbol

0000

11110

0001

01001

0010

10100

0011

10101

0100

01010

0101

01011

0110

01110

0111

01111

1000

10010

1001

10011

1010

10110

1011

10111

1100

11010

1101

11011

1110

11100

1111

11101

Tab. 1.8: Block-by-block encoding This is a transcoding code rather than a transmission code per se, ensuring no more than 2 consecutive 0s are transmitted. The 8B/6T code is based on the same principle. It encodes 8 bits of signal into 6 ternary bits (i.e. tristate bits, or bits that can have one of 3 values: –V, 0, +V) giving a total of 729 symbols. An NRZI or MLT-3 code is often added for the transmission. The advantage is that there are many unused symbols left, which can be assigned to control the transmission, for instance.

159

I Basic principles NRZI encoding (Non Return to Zero Inverted) Use: Fast Ethernet (100BASE-Fx) A signal transition is imposed for each 1 bit and none for the 0.

Fig. 1.145: Signal transition A long sequence of 0s prevents signal transitions (the 4B/5B code imposes no more than 2 consecutive 0s). MLT-3 encoding (Multi Level Transition 3) Use: Fast Ethernet (100BASE-Tx; 100BASE-T4) The 1 bits are successively encoded in 3 states (+1 V, 0 V, –1 V), where the 0 bits retain the previous state. This runs up 31.25 MHz for a symbol rate of 125 Mbps. However the efficiency is only 80% resulting in a throughput of 100 Mbps.

Fig. 1.146: Signal transition Here again a long sequence of 0s can desynchronize the receiver.

160

PAM 5 encoding (Pulse Amplitude Modulation 5) Use: 100BASE-T2; Gigabit Ethernet This works rather like MLT 3, but uses a signal encoded on 5 levels (–1 V, 0.5 V, 0 V, 0.5 V, 1 V) and each symbol also has a different phase. This also runs up 31.25 MHz but encodes two bits per symbol. Over four pairs simultaneously (2 bits x 125 Mbps x 4 pairs) this results in a throughput of 1000 Mbps.

Fig. 1.147: Signal encoding

5.5 Bob Smith termination The Bob Smith termination is frequently used for better protection from EMI and ESD interference. It is based on the pair-to-pair relationships of the twisted pairs within the cable which form transmission lines themselves. Since the cable is basically symmetric, each pair has the same relationship with each other pair. By terminating these pair-to-pair transmission lines with a load matching their characteristic impedance, it effectively dissipates the energy from the standing waves minimizing emissions. Here is an example of one of the several available versions.

161

I Basic principles The unused pairs (4, 5, 7, 8) are linked through a network with 75 Ω resistance to a capacitor of 0.001 μF (2 kV) connected to the frame ground. This provides a “path” to discharge any interference coming from the cable.

Fig. 1.148: Schematics for Bob Smith termination Using a center tapped transformer, the midpoints of the data lines are also connected to the same type of capacitor to allow the discharge of any HF noise. These midpoints (on the chip) are often connected to the VCCA through a capacitor (0.01 μF, 2 kV) connected to the ground (we can also add a ferrite) to minimize noise from the power feed in the twisted pair. In cases where a 2-wire common mode choke is used on the cable side, an autotransformer is used to provide a center tap. This presents a high impedance to the differential signal but a low impedance to the common mode currents because their polarity cancel the magnetic field.

162

Transformer

Common mode choke

Autotransformer

Active PHY

Cat 5 pair 75 R

1000 pF, 2 kV

Fig. 1.149: Bob Smith termination using an autotransformer.

5.6 Power over Ethernet (PoE) Power over Ethernet (also known as PoE) is a technology that powers IP phones, wireless LAN access points, security network cameras and other IP-based devices, in parallel with the data via the existing or in new Ethernet infrastructure. PoE integrates data and power in the same cable and does not affect simultaneous network operation. PoE uses several methods to supply a nominal 48 Vdc via shielded/unshielded twistedpair wiring at predefined power levels called classes. The standardization of PoE technology ensures interoperability between devices from different manufacturers. The association for the standardization of PoE is the Institute of Electrical and Electronic Engineers (IEEE). The two current standards are: IEEE 802.3af-2003 This standard, commonly known as PoE, provides up to 15.4 W of power from each port. After cable losses the powered device can use a maximum of 12.95 W. IEEE 802.3at-2009 This standard, commonly known as PoE+, provides up to 30 W of power from each port . After cable losses the powered device can use a maximum of 25.5 W. A third standard to allow for increased power is in the approval stages. IEEE 802.3bt (not released) This standard, commonly referred to as 4PPoE, provides up to 99.9 W of power from each port. After cable losses the powered device can use a maximum of 74.5 W.

163

I Basic principles 5.7 Important safety points to consider There are numerous challenges in implementing a PoE solution. First and foremost, the solution must meet the relevant communication and safety standards. The design of the Power Sourcing Equipment (PSE port) must take technological aspects, relating to safety, into account. • The PSE port must not damage the existing cable infrastructure or equipment when connected to it. It may not damage devices. Therefore power will only be provided if a PoE-enabled device is detected. • The PoE system must be protected against overload and short-circuits. • Overheating must be avoided under all circumstances to prevent adjacent materials catching fire. • Protection must be provided against PSE ports being connected in parallel and against crossed connections. • In accordance with the Ethernet standard, a galvanic isolation of 1500 Vac is required between the device’s internal electronics and the interface, i.e. the RJ-45 connector. Furthermore, the PoE unit must be isolated from the basic electronics of the switch. This requires a galvanically isolated power supply and galvanically isolated data communication between the switch and the PoE network which is provide by the LAN transformer assembly and power transformer.

5.8 Infrastructure and signal integrity Supplying direct current into an Ethernet network can increase the susceptibility of the system to interference and degrade the Ethernet signal, impairing data communication. Therefore, the supply voltage fed into the system must be free of interference and noise. The supply voltage can, in principle, be supplied in two different ways: Mode A – Endspan (10/100BASE-TX) The power supply is coupled using phantom power via the data pairs: 1/2 and 3/6, as shown in Figure 1.150. Supplying the power via the center tap of the transformer ensures that the bidirectional data flow is maintained, regardless of the status of the module. This mode is used for the installation of endpoint Ethernet switches and the connecting of appropriate 802.3af-ready devices, via structured building cabling.

164

Switch/Hub (PSE)

Terminal device (PD) 1

1

2

2

4

4

5

5

7

7

Data

Power supply for phantom powering

Data

8

8

3

3

6

6

Remote power supply

Data

Data

Fig. 1.150: PoE power supply principle using phantom power via the signal wires Mode B – Midspan (10/100BASE-TX) The power supply is supplied via the spare pairs: 4/5 and 7/8. The wires of each pair are short-circuited. Figure 1.151 shows a schematic representation. This option is useful for supplying additional power between the Ethernet switch and the RJ45 distribution panel in structured building cabling, if midspan devices are installed.

Switch/Hub (PSE)

Terminal device (PD) 1

1

2

2

4

4

5

5

7

7

Data

Power supply for phantom powering

Data

8

8

3

3

6

6

Remote power supply

Data

Data

Fig. 1.151: PoE power supply principle with power supplied via the spare pairs in the cable Mode A is the preferred way of providing PoE for 10BASE-T and 100BASE-TX applications. This frees the spare pairs 4/5 and 7/8 which can be used to supply additional power or for any other purposes. Even on the same cable, one may have two 100BASE-TX communications. Mode A also works on Gigabit Ethernet 1000BASE-T(x) however there are no spare pairs. 165

I Basic principles If the additional power supply is only required selectively within the structured building cabling, as is the case when integrating wireless access points, the use of midspan devices with mode B is recommended. Here, the connection (RJ45 distribution panel) can be supplied with additional power, to which the terminal device is also connected. The additional power is supplied via the spare pairs 4/5 and 7/8. To ensure proper functioning and efficiency of the wiring system with PoE, the wiring components must be considered. The additional dc current through the cable resistance creates heat. The amount of power being supplied, the size of the cable and the number of cables grouped together in bundles must be considered. For low power Cat. 3 or Cat. 5/5e cable is suitable to 100m. For higher power, the larger conductors of Cat. 6 or Cat. 7 cable are more appropriate. Category

Shielding

Wire size Max Speed

Max Bandwidth

Cat. 3 UTP

unshielded

24 AWG

10 Mbps

16 MHz

Cat. 5 UTP

unshielded

24 AWG

10/100 Mbps

100 MHz

Cat. 5e UTP

unshielded

24 AWG

1000 Mbps/1 Gbps

100 MHz

Cat. 6 F/UTP Shielded or unshielded

23 AWG

1000 Mbps/1 Gbps

>250 MHz

Cat. 6a FTP

shielded

23 AWG

10,000 Mbps/10 Gbps 500 MHz

Cat. 7 S/FTP shielded

23 AWG

10,000 Mbps/10 Gbps 600 MHz

Tab. 1.9: Cable categories and characteristics The quality of the RJ45 connectors used, which have to carry electrical power and the data, determines the operational reliability of the network throughout its service life. PoE systems are built up in “star topology”, so that each device is connected to the central hub by a separate channel. Figure 1.152 shows the elements common to all PoE systems, their basic electrical design and their connections.

PSE – Switch/Hub

PD – Device I MAX : 350 mA 3/6 or 4/5

UOUT : 44 - 57 Vdc P MIN : 15.4 W

UIN : 37 - 57 Vdc P MAX : 12.95 W 1/2 or 7/8 L MAX : 100 m R MAX : 20 Ω UDROP : 7 V P LOSS : 2.45 W

Fig. 1.152: Electrical parameters of a PoE connection between hub and terminal device 166

The standard defines some electrical parameters that must be observed in the PoE design. The values shown apply to all power classes: • The operating voltage is normally 48 Vdc, but can vary between 44 and 57 Vdc. It should always be below the maximum SELV requirement of 60 Vdc. • Under the current standards the maximum current supplied by the hub should be in the range between 350 or 600 mA dc to protect the Ethernet cables from overheating due to their parasitic resistance. Under the proposed standard this would increase to 960 mA dc. • The above values result in a required power of at least 15.4 W at the hub output. After cable losses of 2.45 W, caused by parasitic wire resistance of approximately 20 Ω per pair (maximum), the power available at the terminal device is 12.95 W.

5.9 Power classes, supply voltage Figure 1.153 shows a simplified block diagram of the interface of a terminal device. Power can be supplied either by means of the data pairs 1/2 and 3/6, or via the spare pairs 4/5 and 7/8. The data interface is a standard transformer with a center tap on the primary winding. The data passes through the transformer to the interface module.

Signal Interface (Rx/Tx) 1

2

Controller / Application

3

6

R-Sign.

R-Class

Power Switch

Isolation and rectification

4

5 7 8

Fig. 1.153: Block diagram of the PoE components of a terminal device The 48 Vdc supply is taken from the center tap and passes through a PoE interface block, which acts as an intelligent DC-DC converter. The power class is determined using an intelligent power source and a preset terminating resistor. Table 1.10 shows an overview of the electrical data defined according to IEEE 802.3af.

167

I Basic principles Parameter

Min.

Max.

Units

Signature resistor

23.75

26.25

kΩ

Startup time (up to I > 10 mA)

–

300

ms

Power consumption

–

12.95

W

Operating voltage range

36

57

V

Turn-on range (UP_MIN)

30

42

V

Input current (at UP = 36Vdc)

10

350

mA

Peak input current

–

400

mA

Tab. 1.10: Electrical data of the PoE interface of the terminal device (PD) according to IEEE 802.3af The IEEE 802.3af (PoE) standard supports devices with an output power up to 15.4 W per port, the IEEE 802.3at (PoE+) standard up to 25.5 W, and the proposed IEEE 802.3bt (4PPoE) standard up to 74.9 W. Table 1.11 shows the difference in para­ meters. Parameter

Proposed IEEE 802.3.af IEEE 802.3at Proposed Type 1 Type 2 IEEE 802.3bt IEEE 802.3bt Type 4 Type 3

Maximum power from hub (PSE)

15.4 W

34.2 W

2 x 30 W

99.9 W

Available power at the terminal device (PD)

12.95 W

25.5 W

53.5 W

74.9 W

Voltage range at hub (PSE)

44.0–57.0 V

50.0–57.0 V

50.0–57.0

52.0–57.0

Voltage range at the terminal device (PD)

37.0–57.0 V

42,5–57.0 V

37.0–57.0

41.2–57.0

Maximum current

350 mA

600 mA

600 mA

960 mA

Power management

4 classes

1 class

6 classes

2 classes

Supported cable types

Cat. 3 and Cat. 5e

Cat. 5e and above

Cat. 5e and above

Cat. 5e and above

Supported modes

A, B

A, B

A, B, 4-pair

4-pair

Tab. 1.11: Comparison of parameters between standards. Note the values in the proposed standards may change before approval.

168

Once a device has been detected, the hub performs a classification to determine the maximum power that should be provided. The hub supplies 15.5–20.5 Vdc and limits the current to 100 mA for a period of 6 to 75 ms. The terminal device responds by drawing a particular amount of current, representing the power class. The terminal device is assigned to a power class: 0 (standard class) indicates that a full 15.4 watts

should be provided, classes 1–3 indicate equal or lower power levels and class 4 and above are for higher power levels. Terminal devices which do not support classification are assigned to class 0. Caution is required when defining the class, as cable losses can affect the classification. Devices in lossy cable systems typically suffer from a lack of energy resources and can lead to sporadic failures. The current and proposed classifications are listed in Table 1.12. Class

Type

PSE Output Power (W)

PD Input Power (W)

0

802.3af

15.4

0.44 – 12.95

1

4

0.44 – 3.84

1 or 3

1

PD Type

2

7

3.84 – 6.49

1 or 3

3

15.4

6.49 – 12.95

1 or 3

12.95 – 25.5

2 or 3

4

802.3at

30

5

802.3bt (Proposed)

45

40

3

60

53.5

3

6 7

75

62

4

8

99.9

74.9

4

Tab. 1.12: Power classification of current and proposed standards PoE+ can supply 30 W at the hub, but due to line losses, the maximum available power on the terminal side is only 25.5 W. Power is supplied by using only 2 pairs of an Ethernet cable. PoE+ is backward compatible with IEEE 802.3af devices; where these are used, the power is limited.

6 Switched mode power supply (SMPS) 6.1 Basic circuits Switched mode power supplies (switching regulators, switch mode power supplies, or switchers) are used to generate one or more operating voltages from an existing variable input voltage (ac or dc)(Figure 1.154). Switch mode power supplies are generally far more efficient than linear power supplies.

Fig. 1.154: Application example for a switch mode power supply

169

I Basic principles The simplest switching regulators are the step-down converter (buck converter) and the step-up converter (boost converter). These converters can either generate a lower or higher output voltage. The SEPIC (single-ended primary inductance converter) is frequently used if the output voltage lies within the input voltage range, i.e. the voltage must be converted both upwards and downwards. The switching regulators above all have no galvanic isolation between the input and output voltages. Should this be required, a switching regulator topology with a transformer has to be selected. The flyback converter is the simplest option. However, the flyback converter is usually not the topology of choice for higher powers (above around 150 W). In this case, circuit topologies derived from the step-down converter, i.e. forward converters or push-pull converters, are deployed. These excel by virtue of their higher efficiency, although they require considerably more components (e.g. additional storage inductors, more active switches). Table 1.13 provides an overview of switching regulator topologies and the inductive components required in the respective circuits. Switching topology

Inductive components

Step-down (buck)

Power inductor

Step-up (boost)

Power inductor

Step-down/up (Buck-boost)

Power inductor

SEPIC (single ended primary inductance converter)

Coupled inductor or two power inductors

Flyback

(storage) transformer

Forward

Transformer, power inductor

Push-Pull

Transformer, power inductor

Half Bridge

Transformer, power inductor

Full Bridge

Transformer, Power inductor

Tab. 1.13: The most important topologies and inductive components used in the circuits

Step-down converter

6.2 Buck converter/step-down converter Figure 1.155 shows the basic circuit of a step-down converter. The voltage applied at the node S1/D1/L1 (referred to as the switching node SW) alternates between the input voltage and negative forward voltage of the diode.

L1

S1

SW

UIN

C1

170

D1

Fig. 1.155: Step-down converter – basic circuit

UOUT

C2

The dc output voltage corresponds to the mean value of the input voltage UIN and the duty cycle with tON as the time S1 is closed and tOFF the time S1 is open. This results in:



UOUT = UIN ·

t ON = UIN · D t ON + t OFF 

(1.116)

Duty cycle

Where D = duty cycle

D=



tON tON + t OFF

(1.117)

As the duty cycle can only be a value between zero and approaching one (considering internal power losses), the output voltage is always smaller than the input voltage and essentially only depends on the duty cycle. Switch S1 as shown can be an IC with an internal switch or an external switching transistor (bipolar or FET). The diode D1 can be a Schottky diode (higher losses) or a second switching transistor (FET → synchronous rectifier). While the switching transistor S1 is closed, the diode D1 is reverse biased; the current flows through inductor L to the output capacitor C (and to a connected load). The slope of the inductor current rises according to the following function:



SON =

UOUT – UIN L 

(1.118)

Where SON = slope of the current = dIL/dt As soon as the on-time tON is finished, the control circuit opens the switch S1. The voltage at the inductor reverses and thus switches the diode D1 into the forward direction. The energy stored in the inductor’s magnetic field now charges the output capacitor via the diode D1. The slope of the inductor current drops during the off period according to the following function:



SOFF =

UOUT – UD L 

(1.119)

171

I Basic principles Where SOFF = slope of the current In the next switching cycle S1 closes again and the process repeats itself. The average inductor current is equal to the output current of the step-down converter. The maximum inductor current is given by the following function in continuous mode:

I L(PEAK) = IOUT +



UOUT UIN 2· f ·L 

UOUT 1 –

(1.120)

The inductance can be determined by the function: L=



(UIN – UOUT ) · D ΔIL · f 

(1.121)

Where ∆IL is the ripple current and VOUT represents VOUT + VDIODE

Continuous mode

Figure 1.156 shows the associated waveforms for current and voltage when operating in continuous mode.

U IN

S1

I IN (A VG) 0 0

D1

-U IN

L1

0

I OU T 0

U IN -U OU T 0

I OU T

-U OU T

0

U OU T

C1 0

t ON

0

t OFF

Fig. 1.156: Voltages (left) and currents (right) in continuous mode. Shaded a­ reas show ideal equal energy in and out of the inductor L1 and output ­capacitor C1.

172

Discontinuous mode

Discontinuous mode

If the inductor current drops to zero during the time in which S1 is open (off-time), this is d­ escribed as the discontinuous mode (Figure 1.157).

UIN UIN-UOUT 0

IIN(AVG) 0

D1

0 -UOUT -UIN

IOUT(AVG) 0

L1

0 -UOUT

IOUT 0

C1

UOUT 0

0

S1

tON tOFF DT Fig. 1.157: Voltages (left) and currents (right) in discontinuous mode (waveforms are simplified). Shaded areas show ideal equal energy in and out of the inductor L1 and output capacitor C1. The current at which this occurs is given by:



IOUT ≤

U OUT U IN 2·f ·L 

U OUT 1 –

(1.122)

The inductance can be determined by the function:



L=

(UIN – UOUT) · D ΔI L · f 

(1.123)

The duty cycle is no longer constant (at steady state UIN and UOUT) but the output ­voltage is still maintained regulated.

173

I Basic principles Continuous Mode

Discontinuous Mode

Advantages

Output voltage independent from change of load, Good to control, Small peak current, Low radiation

Uses small L values

Disadvantages

Requires larger L values

Output voltage depends on DCR load, High peak currents, More interference

Tab. 1.14: Comparison of operating modes Ringing

Ringing One challenge related to power losses and EMI in discontinuous mode is ringing during the switching-off processes. This is a resonant excitation of the junction capacitance of the diode in parallel to the switching capacitance of the transistor with the inductor. These oscillations can cause interference and they consume additional energy. The ringing frequency can be calculated and used to determine values for a snubber to attenuate the oscillations. See Chapter 4 – Snubber design.

f RING =

1 2

L ( C SWITCH + C DIODE ) 

(1.124)

Typical values for CSWITCH ~ 80 pF and CDIODE = 200 pF to 1000 pF. Snubber

Snubber A detailed look at additional parasitic effects reveals further potential for overshoot and oscillation in the circuit: Interference in the region of 20 MHz to 50 MHz is the result of parasitic inductances in series with the input capacitor, to the switching controller and the diode – in conjunction with the diode capacitance. The parasitic inductances give rise to ~ 0.1 µH and coupled with 500 pF of the diode capacitance, a damped oscillation of 25 MHz is created on the rising edge. It should be stringently ensured that these oscillations are attenuated (snubber). This effect is largely reduced with strategic arrangement of components and layout. 6.2.1 Isolated output from buck converter

174

A low power, isolated output can be added to the buck converter with the addition of a second winding to the inductor. Connected to a separate diode, D2 and capacitor, C3 as illustrated below it will provide a quasi-regulated output. When the switch, S1 turns off the polarity of the voltage across the inductor reverses making the non-dot end of the second winding, N2 positive allowing D2 to conduct. As long as current flows through the inductor, the output voltage will be held constant and therefore the second output will be as well. The capacitor needs to be large enough to maintain the voltage during the switch S1’s on period since no current can flow through D2 during that time.

The second output voltage is proportional to the first by the turns ratio. Remember to account for diode drops.

UO2 =



N2 · UO N1 

(1.125)

D2 UOUT2

C3 N2

S1

L1

UIN

UOUT

N1

C1

D1

C2

Fig. 1.158: A second, isolated output from buck regulator 6.2.2 PolyPhase™ for high output currents

Multi-phase

Today’s processor generations operate with low supply voltages of 1 to 3.3 V and have current inputs of up to 90 A! This is too much for the single phase step-down converters previously introduced – the losses are too great and thus the efficiency too low. Long battery operating times are required, especially for portable devices. The switched currents generate strong electric and magnetic interference fields for single phase switching converters; the danger of pick-up in other stages is very high. The conventional switching converter therefore requires large filter capacitors at its inputs and outputs for interference suppression, which must also have an absolutely small ESR. The storage chokes must be able to buffer the necessary energy – which requires low-loss cores. The costs for a single-phase step-down converter are relatively high for such high currents and are considerably influenced by the noise suppression capacitors and storage chokes.

175

I Basic principles UIN + Module 1 CIN

L1 COUT

UIN Module 2

L2

Fig. 1.159: Schematic circuit diagram of a dual phase step-down converter In multiphase operation, two or more converter stages are arranged in parallel and with its clock frequency, the first module drives the second module – such that they both work in counter-phase. Both modules work from the same input capacitor (CIN), see Figure 1.159. The ripple on the input current and output voltage and ripple current in the output capacitor are only half the size as in a single converter with the same output power or two in-phase parallel operated converters. The interference suppression at the input and output of the switching converter is reduced as a result of doubling of the ripple frequency, which happens at the same time (smaller interference suppression capacitors).

176

Fig. 1.160: Comparison of current curves for single phase and dual phase converters The comparison of current curves of single and dual phase switching converters in Figure 1.160 shows the input ripple current ICIN to fall inversely with the number of phases and also that the ripple frequency is multiplied by the number of phases as it is increased. Theoretically, the input capacitor could therefore be four times smaller than with the single-phase converter.

177

I Basic principles Input ripple current

Fig. 1.161: Normalized input ripple current versus duty factor for 1 … 6 phases Figure 1.161 illustrates this relationship. A duty factor of 0.5 leads to a minimal input current ripple for a dual phase converter!

178

Output ripple current

Fig. 1.162: Normalized output ripple current as a function of the duty factor Duty cycle

Duty cycle

The output ripple current of the dual phase converter also drops to zero if a duty factor of 0.5 is set (Figure 1.162). Therefore one attempts to operate a multiphase converter as close as possible to one of these “critical” points, as here the interference suppression outlay is lowest. The “critical” points of minimal output ripple current and, at the same time, minimal input ripple current of multiphase converters are generally given by:



UOUT i = with i = 1,2, …, N – 1 UIN N 

(1.126)

Where N is the number of phases. The theoretically optimal phase configuration can thus be determined from the voltage input/output voltage ratio.

179

I Basic principles Step up converter

6.3 Boost converter/step-up converter The basic circuit is shown in Figure 1.163. The voltage applied at the node L1/S1/D1 (referred to as the switching node SW) alternates between the output voltage plus the diode forward voltage and the voltage across the RDSON of the transistor S1.

L1

D1

SW

UIN

C1

UOUT

S1

C2

Fig. 1.163: Step-up converter – basic circuit The dc output voltage corresponds to the mean value of the input voltage UIN and with tON as the time in which S1 is closed and tOFF the time S1 is open – results in

UOUT = Duty cycle

U IN U IN = t ON 1 – D 1– t ON+ t OFF 

(1.127)

Where D = duty cycle



D=

t ON t ON + t OFF

(1.128)

As the duty cycle can only be a value between nearly zero and approaching one, the output voltage is always higher than the input voltage and so essentially depends on the duty cycle. The switch S1 as shown can be an IC with an internal switch or an external switching transistor (bipolar or FET) and the diode D1 can be a Schottky diode. While the switching transistor S1 is closed, diode D1 is reverse biased; the current flows through the inductor L1 and builds up the magnetic field. The output capacitor C2 feeds the output side. If switch S1 is now opened, the voltage on the storage inductor reverses and is in series with the input voltage. This recharges the capacitor C2 (reduced by the forward voltage drop of diode D1). The associated curve profiles for both operating modes are shown in Figure 1.164.

180

U OUT +U IN U IN

S1

I IN (AVG) 0

0 0

D1

L1

I OUT

-U OUT -(U OUT +U IN )

0

U IN 0

I OUT

-U OUT

0 I IN -I OUT

U OUT

C1

0

0 t1

-I OUT

t2

Fig. 1.164: Voltages (left) and currents (right) in continuous mode

S1

UIN 0

IIN(AVG) 0

D1

0 -UOUT -(UIN+UOUT)

IOUT(AVG) 0

L1

0

C1

UOUT 0

IIN+IOUT 0

0 -IOUT

tON tOFF DT Fig. 1.165: Voltages (left) and currents (right) in discontinuous mode (waveforms are simplified)

Here it should be considered that the inductor current is directly proportional to the input/output ratio and can very quickly attain a high peak value IPEAK.



IPEAK = I OUT

U OUT U IN 

(1.129)

181

I Basic principles On the other hand, the mean diode current is equal to the output current; the mean value of the current through the switching transistor is given by:



I AVG = I OUT

UOUT – UIN UIN 

(1.130)

There are even higher ratios of inductor to output current in the discontinuous mode. Here it is important to observe that the step-up converter cannot easily be connected to protect against short-circuiting, as the input and output sides are directly coupled via the diode D1.

182

Overview design data:

1)

without considering ESL

Tab. 1.15: Design data

183

I Basic principles 6.4 SEPIC switching controller with low input ripple current SEPIC

The SEPIC switching controller (Single Ended Primary Inductance Converter) has constant output voltage over a wide input voltage range. The input voltage can be smaller, the same or larger than the output voltage. At the same time, the output voltage has short-circuit protection. The increased switching effort was previously considered as disadvantageous because the application circuits usually use two separate choke coils. Due to a sophisticated design of the two choke coils as only one transformer-coupled inductance, a) the component and space requirements can be significantly reduced b) either the input ripple current or the output ripple current can be reduced to zero using a sophisticated design of the choke coils. This enables a very compact design of the circuit and low filter complexity (smaller c­ apacitors) on the input or output side and also makes this type of controller particularly interesting for EMC critical applications. Further benefits of the SEPIC circuit: • No flow of current from the input to the output of the circuit in the switched-off state; this is prevented by the coupling capacitor CC • The output protection diode for battery charging applications is already integrated in the circuit by D1 • The capacitor CC completely absorbs the energy of the leakage inductance; ­therefore no snubber circuit is required SEPIC Circuit Diagram L1

CC

D1 UOUT

UIN

CIN

S1

L2

COU T

Fig. 1.166:  SEPIC basic circuit During start-up, prior to closing the switch S1 the capacitor CC is charged to the potential of the input voltage UIN. The output voltage is zero and no further current flows. The switch S1 now closes (Figure 1.167). Across the coil L1 the input voltage UIN is applied, thus energy is stored in the air gap of the inductor core. The coupling capacitor was previously charged to the input voltage UIN and now feeds the coil L2 which also stores energy in the magnetic field. The diode D1 is in reverse bias so that no energy transfer to the output takes place here. The load is fed from the output capacitor COUT. 184

L1

CC

UIN

UOUT

CIN

I L1 I L2

L1

CC

L2

COUT

D1

UIN

UOUT

CIN

I L1 I L2

L2

COUT

Fig. 1.167: Equivalent circuit for the switching status in the SEPIC controller a) S closed and B) S open If the switch S1 now opens the voltage at the coils reverses. The diode D1 now conducts and feeds the output circuit and charges the capacitor COUT. The current also flows from L1 to recharge the coupling capacitor CC If the end of the second cycle is reached, a new switching cycle starts and the switch S1 closes again. SEPIC Controller with LTC 1871 and Calculations:

Fig. 1.168: Circuit diagram for SEPIC controller with Analog Devices LTC1871 Duty Cycle: The duty cycle for the continuous operating mode is:



D=

Uout + UD Uin + Uout + UD 

(1.131)

185

I Basic principles where UD = diode forward voltage. If the input voltage UIN is close to the output voltage UOUT, the duty cycle will be 50%. Depending on the maximum possible duty cycle of the controller IC, the maximum possible output voltage can be determined as follows: UOUT(MAX) = (UIN + UD ) ·



D MAX 1 – D MAX 

(1.132)

The maximum duty cycle for the LTC1871 is typically 92%. Inductance Value and Inductance Currents: Example: With a ripple current factor of r = 0.3 (e.g. 30%), an output current of 1 A, an output voltage of 12 V, an input voltage of 6–20 V and the switching frequency of 280 kHz, the inductance value in the uncoupled state emerges as: Ripple current with ripple current factor r: ΔI = r · IOUT(MAX)



D MAX 0.68 = 0.3 · 1 · = 0.6375 A (1.133) 1 – D MAX 1 – 0.68 

Inductance required for ripple current at worst case UIN(MIN) and DMAX: U IN(MIN) · D MAX 6 V · 0.68 = = 22.8 µH ΔI · f 0.6375 A · 280 kHz 

L1 = L2 =



(1.134)

Accordingly, the ripple current for a nominal 13 Volts input voltage for example is: DI =



Input filter

Uin · Dmax 13 V · 0.53 = = 1.08 A L·f 22.8 µH · 280 kHz 

6.5 Input filter L

RD +

CIN CD

Output Impedance ZOUT

Input Impedance ZIN

-

Fig. 1.169: Input filter circuit for buck converter

186

Converter

(1.135)

All switching regulators/converters need an input capacitor to stabilize the voltage and provide a low impedance energy reservoir for proper operation. Nevertheless there remains a component of ripple current that cannot be ignored which flows back on the supply line. This leads to additional electromagnetic interference and additional filtering with an inductor is recommended. Two aspects must be taken into account here: All regulators, whether linear or switching, have a negative input resistance. This is easy to understand: The purpose of a regulator is usually to keep its output voltage constant at a given load, so the output power is constant. If, for example, the input voltage increases, the input current must decrease accordingly – this is exactly the definition of a negative input resistance. This arrangement can oscillate if there is a resonance-capable structure in the input circuit where the positive attenuation resistance is not sufficient to compensate the negative input resistance! The L/C ratio is also important. In practice, an electrolytic capacitor is usually required whose ESR usually prevents oscillation. However since the ESR is largely undefined, uncontrollable, temperature and age-dependent, a ceramic or low-induction film capacitor with a series connected defined damping resistance is connected in parallel with the electrolytic capacitor. This must be determined carefully, since the negative input resistance is non-linear and depends among other things on the input voltage and the load! Because of the sharp current pulses that the converter draws, it is quickly considered to connect an upstream LC filter in order to keep them away from the power source. Such a filter must therefore be carefully designed and adapted to the regulator According to Middlebrook the design criteria is to make the input filter cutoff frequency lower than the averaging output filter cutoff frequency and to make sure the input filter output impedance is much lower than the converter’s open-loop input impedance for the converter to remain stable. It’s important to note the input capacitor is considered part of the input filter and not the converter.

ZOUT(filter) VRAMP, the comparator’s output is high, and when VEA < VRAMP its output is low. As VEA changes in accordance with the operating conditions, the duty cycle generated by the comparator changes, correcting the converter’s output.

Pulse-width modulator

Fig. 1.190: Simple modulator

The converter shown is “ideal” in the sense that switching losses are not considered.

2)

211

I Basic principles For a step-down converter, the average voltage output by the PWM is given by: VOUT = D × VIN(1.163)

where D is the duty cycle of the modulator’s output. Since the duty cycle depends upon comparing VEA to VRAMP it can be shown that the modulator’s gain is given by:



(1.164)



(1.165)







(1.166)

In practice, GPWM does not vary with frequency, but does vary with input voltage. This has to be taken into account when designing the compensation circuit because the compensation has to be validated over the entire range of possible input voltages. Fortunately, many modern DC/DC converters feature modulators that implement a function known as voltage feed-forward (see Figure 1.191). Voltage-feed-forward

Fig. 1.191: Modulator with voltage feed-forward 212

The voltage feed-forward function automatically compensates changes in the input voltage, greatly improving the converter’s response to line transients, but also making GPWM independent of VIN. Modulator stages using voltage feed-forward generate a ramp waveform whose amplitude is proportional to the input voltage, i.e. VRAMP = k × VIN(1.167)

The PWM gain is now given by:













(1.168)

(1.169)

(1.170)

For example, TI’s TPS40200 DC/DC controller features a PWM in which VRAMP is equal to exactly one tenth of VIN, resulting in a constant modulator gain equal to 20 dB. Continuous vs. discontinuous conduction The previous discussion made a significant assumption; namely, that the converter was operating in continuous conduction mode (CCM). This means that current is continuously flowing in the inductor. Conversely, the term discontinuous conduction mode (DCM) means the inductor current is zero for some part of the switching cycle.

Continuous mode Discontinuous mode

CCM and DCM are probably easiest understood by considering what happens in a step-down converter, as illustrated in Figure 1.192.

213

I Basic principles

Fig. 1.192: CCM vs. DCM

Ripple current r

In a step-down converter the inductor current rises and falls linearly about the average value, which is equal to the converter’s output current. The amount by which inductor current rises and falls is called the ripple current and is typically 20–30% of the output current. As long as the output current is greater than IRIPPLE/2 the inductor current never falls to zero and the converter operates in CCM. Critical conduction is said to occur when the output current is exactly equal to IRIPPLE/2, in which the inductor current reaches zero for an instant, but immediately begins to rise again. For any value of output current below IRIPPLE/2 a non-synchronous converter will operate in DCM. This happens because the converter’s rectifying diode is only able to conduct in one direction; once the inductor current reaches zero, it stays at this value until the start of the next switching cycle. Synchronous converters, on the other hand, always operate in CCM. This is because the MOSFET used as the rectifier is able to conduct current in both directions. Consequently, once inductor current has reached zero it continues at the same rate of change and goes negative, if necessary (see Figure 1.193).

Fig. 1.193: Synchronous converters always operate in CCM 214

Whether a converter operates in CCM or DCM is important because the two modes exhibit different transfer functions. The compensation scheme must work correctly with whichever mode the converter operates in, and must often work with both CCM and DCM if the converter’s load varies greatly. For example: consider the well-known transfer function of a simple step-down ­converter: VOUT = VIN × D(1.171)

This expression is actually only valid for converters operating in CCM. If the converter operates in DCM, its transfer function is given by:

(1.172)



As can be seen, in DCM the modulator gain changes as the load changes. Output filter

Output filter

In voltage mode control, the buck, boost, flyback and SEPIC topologies all exhibit a second-order double-pole LC filter characteristic. For the buck topology, the filter’s break frequency is given by:



(1.173)



For the boost and flyback topologies, the filter’s break frequency varies with the converter’s duty cycle, and is given by:



(1.174)



Furthermore, if the capacitor exhibits significant ESR (Equivalent Series Resistance), at some frequency its impedance will change from primarily capacitive to resistive in nature. The resulting frequency response will exhibit a zero at a frequency given by:





ESR

(1.175) 215

I Basic principles Since this ESR-zero changes the frequency response of the forward path it must be taken into account when designing the compensation circuit. Figures 1.194 and 1.195 show the gain and phase response of two 100 µF capacitors: one is ideal, and purely capacitive; the other has an ESR of 0.3 Ω.

Fig. 1.194: Capacitor impedance vs. frequency

Fig. 1.195: Capacitor phase vs. frequency 216

Generally speaking, low-ESR capacitors such as ceramic types are desirable because they reduce output voltage ripple (because the inductor ripple current flows through the ESR). In practice, however, low-ESR capacitors are often too expensive or too large to obtain the required output capacitance. As a result, many voltage regulator ICs are designed specifically for use with a capacitors having an ESR in a certain range, and make assumptions about the nature of the filter’s response. Whether the output capacitor exhibits significant ESR or not is important because above the break frequency of the ESR zero the slope of the filter’s roll-off changes from –40 dB per decade to –20 dB per decade (see Figure 1.196). As previously stated, achieving a loop gain slope of –20 dB per decade at crossover is one of the main aims of compensation circuit design, and if crossover occurs at a frequency higher than the ESR zero, the compensation circuit need only exhibit a flat gain response for stable operation (see sections on Type-I, Type-II and Type-III compensation schemes). Furthermore, the phase of an ideal LC filter tends towards –180° at high frequencies whereas the phase of an LC filter with ESR tends towards –90° (see Figure 1.197).

Fig. 1.196: Gain response of an LC filter with and without ESR

217

I Basic principles

Fig. 1.197: Phase response of an LC filter with and without ESR Right half-plane zeros RHP zero

The output filter of the boost and flyback topologies operating in CCM also exhibits a strange mathematical phenomenon known as a right half-plane (RHP) zero. A RHP zero exhibits the same +20 dB per decade gain response as a left half-plane (LHP) zero but its phase response has a 90° lag (a left half-plane zero on the other hand exhibits a 90° phase lead). The combination of this phase lead and its dependence on line and load make RHP zeros very difficult to compensate: in practice, the only reliable way to deal with a RHP zero is to ensure that crossover occurs well below the lowest possible RHP zero frequency. The frequency of the RHP zero in a boost converter is given by:





(1.176)

The frequency of the RHP zero in a flyback converter is given by:

218

where RO is the equivalent load resistance.



(1.177)

Error amplifier The error amplifier compares the regulator’s output with a reference and generates an output signal intended to correct any errors. Since the reference voltage is almost always significantly lower than the regulator’s output, a potential divider is used to adapt the error amplifier to whatever output voltage is required by an application. The basic form of the error amplifier is shown in Figure 1.198.

Error amplifier

Fig. 1.198: Basic error amplifier configuration In practice, Z1 typically includes a resistor between VOUT and the error amplifier’s inverting input. This, together with RBIAS and VREF, sets the regulators dc output voltage. What is not immediately apparent is that RBIAS does not affect the ac response of the above circuit and can therefore be ignored for frequency compensation purposes (RBIAS has been omitted from the following error amplifier circuits). Thus, if the output voltage of a properly compensated circuit is to be changed, it is always recommended that only RBIAS is changed, since this will not affect the circuit’s compensation.

Frequency compensation

Three types of compensation are commonly used in voltage regulator circuits. They are called Type-I, Type-II and Type-III; and which is the best one to use for a particular application depends on the converter topology and the components used in the forward path. Type I compensation

Type I compensation

Type I compensation (see Figure 1.199) is the simplest. Its frequency response has a single pole at the origin and so its gain decreases at a rate of 20 dB per decade, crossing the 0 dB axis





(1.178)

219

I Basic principles

Fig. 1.199: Type-I Compensation Type I correction is only suitable for voltage regulator circuits in which the phase shift in the forward path is very small, which is rarely fulfilled in practice; a purely integrating regulator will therefore always oscillate or cause large deflections. Therefore, there are always very large capacitors in such circuits that make the control circuit extremely slow, which is only acceptable for battery chargers. Use is strongly discouraged, as such behavior is unacceptable in the majority of applications

220

Fig. 1.200: Gain and phase response of Type-I compensation Type II compensation Type II compensation (see Figure 3.197) is the normal, almost exclusively used one. The proportional component is absolutely necessary for damping in order to achieve a stable and fast controller behavior in practice with minimum undershooting/overswinging with load changes. The C2 capacitor is primarily used to bridge the usually slow control amplifier for the high switching frequency of the switching regulator, as it cannot follow it. Depending on the application, several R-C links may be required. This type of controller also has the highest gain at dc, resulting in a high static accuracy of the output voltage. In the middle of the frequency range of interest, the frequency response is flattened before it finally drops to zero as the frequency increases further.

Type II compensation

221

I Basic principles

Fig. 1.201: Type-II compensation

222

Fig. 1.202: Gain and phase response of Type-II compensation

The main parameters of this circuit are given by:







(1.179)



(1.180)





(1.181)

Type-III compensation

Type III compensation

Type-III compensation is even more complicated (see Figure 1.203). Its pole at the origin is followed by two zeros and two poles, which creates two separate regions of constant gain and phase, although in many applications the pairs of poles and zeros occur at the same frequency creating a zigzag response that has no flat areas.

Fig. 1.203: Type-III compensation

223

I Basic principles

Fig. 1.204: Gain and phase response of Type-III compensation

224

The main parameters of this circuit are given by:



(1.182)











(1.183)

(1.184)













(1.185)

(1.186)

(1.187)

Crossover frequency

Crossover frequency

At some stage in the design of the compensation circuit the designer will have to choose a crossover frequency, i.e. the frequency at which the total loop gain is equal crossover to 0 dB. The higher the crossover frequency, the faster the loop responds and the better its transient performance is. However, there are both theoretical and practical reasons why very high crossover frequencies are not used and in practice crossover frequencies somewhere between one tenth and one sixth of the switching frequency are generally used. K factor

K factor

The K factor is often used in the design of compensation circuits. It simplifies a lot of the mathematics by providing pre-calculated values for some of the key relationships. It is very simple, and describes the positions of the pole(s) and zero(s) in the compensation circuit relative to the crossover frequency, as follows:





(1.188)





(1.189) 225

I Basic principles If frequency is plotted logarithmically, fco lies exactly mid-way between fz and fp. This is important because it turns out that the circuit’s phase boost is always at its maximum at this point, a useful characteristic when designing compensation circuits. When use of the K factor to design compensation circuits was first introduced typical switching frequencies were much lower, and consequently it is more common today to have compensation poles and zeros not located symmetrically about the crossover frequency. Tables 1.16 through 1.18 show the phase and gain response of Type-I and Type-II compensation circuits assuming the poles and zeros are not located symmetrically about the crossover frequency but are in fact related to it by two K factors, as follows:



(1.190)





(1.191)



Note: For the Type-III response it is assumed that both poles and both zeros occur at the same frequency.

K2

K1

10

20

30

40

50

60

70

80

90

100

1.2

–225.5°

–222.7°

–221.7°

–221.2°

–221.0°

–220.8°

–220.6°

–220.5°

–220.4°

–220.4°

1.4

–221.2°

–218.4°

–217.4°

–217.0°

–216.7°

–216.5°

–216.4°

–216.3°

–216.2°

–216.1°

1.6

–217.7°

–214.9°

–213.9°

–213.4°

–213.2°

–213.0°

–212.8°

–212.7°

–212.6°

–212.6°

1.8

–214.8°

–211.9°

–211.0°

–210.5°

–210.2°

–210.0°

–209.9°

–209.8°

–209.7°

–209.6°

2

–212.3°

–209.4°

–208.5°

–208.0°

–207.7°

–207.5°

–207.4°

–207.3°

–207.2°

–207.1°

3

–204.1°

–201.3°

–200.3°

–199.9°

–199.6°

–199.4°

–199.3°

–199.2°

–199.1°

–199.0°

4

–199.7°

–196.9°

–195.9°

–195.5°

–195.2°

–195.0°

–194.9°

–194.8°

–194.7°

–194.6°

5

–197.0°

–194.2°

–193.2°

–192.7°

–192.5°

–192.3°

–192.1°

–192.0°

–191.9°

–191.9°

6

–195.2°

–192.3°

–191.4°

–190.9°

–190.6°

–190.4°

–190.3°

–190.2°

–190.1°

–190.0°

7

–193.8°

–191.0°

–190.0°

–189.6°

–189.3°

–189.1°

–188.9°

–188.8°

–188.8°

–188.7°

8

–192.8°

–190.0°

–189.0°

–188.6°

–188.3°

–188.1°

–187.9°

–187.8°

–187.8°

–187.7°

9

–192.1°

–189.2°

–188.2°

–187.8°

–187.5°

–187.3°

–187.2°

–187.1°

–187.0°

–186.9°

10

–191.4°

–188.6°

–187.6°

–187.1°

–186.9°

–186.7°

–186.5°

–186.4°

–186.3°

–186.3°

Tab. 1.16: Phase change through a Type-II compensation circuit

226

K2 1.2

K1 10 –181.0°

20

30

40

50

60

70

80

90

100

–175.3°

–173.4°

–172.5°

–171.9°

–171.5°

–171.2°

–171.0°

–170.9°

–170.8°

1.4

–172.5°

–166.8°

–164.9°

–163.9°

–163.4°

–163.0°

–162.7°

–162.5°

–162.3°

–162.2°

1.6

–165.4°

–159.7°

–157.8°

–156.9°

–156.3°

–155.9°

–155.6°

–155.4°

–155.3°

–155.2°

1.8

–159.5°

–153.8°

–151.9°

–151.0°

–150.4°

–150.0°

–149.7°

–149.5°

–149.4°

–149.3°

2

–154.6°

–148.9°

–146.9°

–146.0°

–145.4°

–145.0°

–144.8°

–144.6°

–144.4°

–144.3°

3

–138.3°

–132.6°

–130.7°

–129.7°

–129.2°

–128.8°

–128.5°

–128.3°

–128.1°

–128.0°

4

–129.5°

–123.8°

–121.9°

–120.9°

–120.4°

–120.0°

–119.7°

–119.5°

–119.3°

–119.2°

5

–124.0°

–118.3°

–116.4°

–115.5°

–114.9°

–114.5°

–114.3°

–114.1°

–113.9°

–113.8°

6

–120.3°

–114.6°

–112.7°

–111.8°

–111.2°

–110.8°

–110.6°

–110.4°

–110.2°

–110.1°

7

–117.7°

–112.0°

–110.1°

–109.1°

–108.6°

–108.2°

–107.9°

–107.7°

–107.5°

–107.4°

8

–115.7°

–110.0°

–108.1°

–107.1°

–106.5°

–106.2°

–105.9°

–105.7°

–105.5°

–105.4°

9

–114.1°

–108.4°

–106.5°

–105.5°

–105.0°

–104.6°

–104.3°

–104.1°

–104.0°

–103.8°

10

–112.8°

–107.1°

–105.2°

–104.3°

–103.7°

–103.3°

–103.1°

–102.9°

–102.7°

–102.6°

Tab. 1.17: Phase change through a Type-III compensation circuit

K1

Type-II

Type-III

10

0 dB

20.0 dB

20

0 dB

26.0 dB

30

0 dB

29.5 dB

40

0 dB

32.0 dB

50

0 dB

34.0 dB

60

0 dB

35.6 dB

227

I Basic principles K1

Type-II

Type-III

70

0 dB

36.9 dB

80

0 dB

38.1 dB

90

0 dB

39.1 dB

100

0 dB

40.0 dB

Tab. 1.18: Gain change between fZ and fCO for Type-II and Type-III compensation circuits These tables are extremely useful because they tell us what the relative positions of the poles and zeros will do to the compensation circuit’s phase response at the crossover frequency. The general procedure for designing a compensation circuit is as follows: Bode plots

• Generate the modulator Bode plots. • Select a suitable crossover frequency using the rule-of-thumb that fCO should be somewhere between fCO/10 and fCO/6. • From the gain Bode plot, determine whether Type-II or Type-III compensation will be used. • From the Bode plots, determine the modulator gain at the crossover frequency. • Place the compensation circuit’s zero(s) approximately one octave below the output filter’s break frequency, and calculate the value of





(1.192)

• Calculate the maximum phase lag through the compensation circuit and, using Table 1.17 or 1.18, calculate the minimum value of K2 achieving this phase lag. Calculate the frequency of the compensation circuit’s pole(s) using

fP = K2 × fCO(1.193)

• Calculate the individual component values in the compensation circuit. The following examples show how this works in practice. Example #1

228

First, the Bode plots of the modulator response must be generated. There are a number of ways to do this, but probably the quickest and easiest way is to use a circuit simulation program, such as LTspice. Figure 1.205 shows the modulator equivalent circuit used to generate the Bode plots shown in Figure 1.206. U1 is an ideal op-amp and is used to simulate the 20 dB gain of the PWM block; R3 simulates C1’s ESR; and R4 simulates a load of 2.5 A at 3.3 V.

Fig. 1.205: Modulator equivalent circuit

Fig. 1.206: Modulator bode plots Next, the break frequencies of the LC filter and its ESR zero are calculated:



(1.194)











(1.195)

(1.196) 229

I Basic principles





(1.197)

It can be seen from the above values that the LC filter’s break frequency and its ESR zero are almost coincident, which means that the filter effectively exhibits a single-pole response, and the simpler Type-II compensation can be used. The rule-of thumb stated previously recommended a crossover frequency between 1/10 and 1/6 of the switching frequency, which in this case means somewhere between 30 kHz and 60 kHz. From the Bode plots, it can be seen that the modulator gain in between these two frequencies ranges from about –5.3 dB to –11.4 dB. The error amplifier therefore has to provide somewhere between 5.3 dB and 11.4 dB of gain during the flat region of its response, which is a reasonable figure. If greater than 20 dB were required, it would probably be better to choose a lower crossover frequency that required less gain from the error amplifier. A crossover frequency of 40 kHz is therefore chosen. At 40 kHz the phase lag through the modulator is 94°, which means that to ensure a minimum phase margin of 45° at this frequency the phase lag through the compensation circuit should not be greater than:



θCOMP = 360º – θPM – θLC(1.198)

θCOMP = 360º – 45º – 94º = 221º

(1.199)

where: θCOMP is the phase lag through the compensation circuit θPM is the minimum desired phase margin θLC is the phase lag through the output filter Using our previous rule-of-thumb, the Type-II compensation zero is placed at 1 kHz, approximately one octave below the output filter’s break frequency. This boosts the phase around the loop by approximately 45° before the output filter’s phase lag starts to have an effect. The K1 parameter can now be calculated as follows:



230



(1.200)

From the K1 = 40 column of Table 1.17 the minimum value of K2 required to achieve an overall phase lag around the loop of less than 221° can be determined. In this case any value for K2 from 1.2 upwards will achieve the required phase response. Choosing

a value of K2 = 2 ensures that the circuit exhibits a phase margin comfortably higher than 45°. The Type II compensation pole is therefore located at a frequency given by:

fP = K2 × fCO = 2 × 40 kHz = 80 kHz

(1.201)

From its Bode plots it can see that at 40 kHz the modulator gain is –7.82 dB. To achieve an overall loop gain of 0 dB at crossover, the error amplifier’s closed-loop gain at this frequency must therefore be 7.82 dB. All the information needed to calculate the values of the compensation components is now available: fCO = 40 kHz fZ = 1 kHz fP = 80 kHz G = 7.82 dB (2.46)

R2 = G × R1(1.202)

R2 = 2.46 × 100 kW = 246 kW(1.203)









(1.204)





(1.205)

(1.206)





(1.207)

The overall response using these values is shown in Figure 1.207. Crossover occurs at 35 kHz (it has moved from the design value of 40 kHz because the standard component values used are not exactly the calculated values), and gain and phase margins are a healthy 23.2 dB and 56.7° respectively. 231

I Basic principles

Fig. 1.207: Gain and phase loop response The above example makes a significant assumption: That the error amplifier’s openloop gain bandwidth is high enough to achieve the desired closed loop transfer characteristic. The error amplifier in the TPS40200 has a minimum open-loop unity gain of 123.5 dB and a single dominant pole roll-off, which means that gain rolls off at 20 dB per decade. The open-loop gain at 40 kHz is therefore given by: A40 kHz = 123.5 dB – 20 × log (40 kHz) = 31.5 dB (37.6)

(1.208)

With this level of open-loop gain, the circuit’s practical performance is close enough to the calculated response that the circuit works acceptably. If the open-loop gain were much lower this might not be the case. Figure 1.208 shows the compensation circuit’s ideal response and its practical response using the error amplifier in the TPS40200.

232

Fig. 1.208: Practical vs. ideal compensation response Example #2 This example uses the circuit shown in the data sheet (Ref: Figure 44) of the TPS40200 (SLU659G-2014). This application, a buck converter with an output of 16 V at 1 A from an input of 18 to 50 V has a switching frequency of 200 kHz and uses output capacitors with negligible ESR, so Type-III compensation is required. Figure 1.209 shows the modulator’s equivalent circuit and Figure 1.210 shows its Bode plots.

Fig. 1.209: Example 2 equivalent circuit

233

I Basic principles

Fig. 1.210: Example 2 modulator response The break frequency of the output filter is given by:



(1.209)







(1.210)

With a switching frequency of 200 kHz, the crossover frequency should be somewhere between the following values:









(1.211)

(1.212)

A preliminary value of 25 kHz is chosen. Positioning the double-pole of the Type-III compensation circuit approximately one octave below the output filter’s break frequency yields a value for fZ of 300 Hz and thus K1 = 83.

234

At 25 kHz the modulator’s phase lag is 180° and therefore the maximum phase lag through the compensation circuit is given by:



θCOMP = 360º – θPM – θLC(1.213)

θCOMP = 360º – 45º – 180º = 135º

(1.214)

From the K1 = 80 column of Table 1.17 it can be seen that a minimum value for K2 of 3 is required to ensure adequate phase margin. At 25 kHz the modulator’s gain is –44.5 dB, which means that the compensation circuit’s gain at 25 kHz must be 44.5 dB. In a Type-III compensation circuit the gain at fZ is set by R2 but increases by 20 dB per decade until the compensation poles are reached. In this case the gain increase between fZ and fCO is given by: GINCREASE = 20 × log (K1) = 20 × log (83) = 38.3 dB

(1.215)

and thus the gain required at fZ is given by: GZ = GCOMP – GINCREASE(1.216)

GZ = 44.5 dB – 38.3 dB = 6.2 dB

(1.217)

The compensation circuit component values can now be calculated using the following parameters: fZ = 300 Hz fCO = 25 kHz fP = 75 kHz GZ = 6.2 dB (2.04) R2 = GZ × R1(1.218)

R2 = 2.04 × 100 kW = 205 kW(1.219) 235

I Basic principles



(1.220)











(1.222)









(1.223)

(1.224)







(1.221)



(1.225)

(1.226)





(1.227)

Figure 1.211 shows the final response of the circuit using the above values. Crossover is 21 kHz. Phase and gain margin are 58.4° and 16.6 dB respectively.

236

Fig. 1.211: Example 2 gain and phase loop response Testing a circuit’s compensation

Control loop test

Even with the best design and simulation tools, there is no getting escaping the fact that at some stage you will have to test your circuit under worst-case conditions, to ensure it meets your application’s requirements. The most complete way to do this is to use a network analyzer, which will provide you with real-world Bode plots, allowing you to see exactly what your circuit is doing. The network analyzer works by injecting a signal into the control loop, sweeping the signal through the frequency range of interest, and measuring the phase and gain introduced by the loop (see Figure 1.212).

Fig. 1.212: Gain and phase measurement using a network analyzer For engineers without access to a network analyzer, or for those just wanting to make a quick sanity check, a lot of useful information can still be obtained by performing tests in the time domain using a transient tester. This device – a typical circuit for one

237

I Basic principles is shown in Figure 1.213 – applies a load step to the voltage regulator under test, and the circuit’s response is observed on an oscilloscope. Transient tester circuit

Fig. 1.213: Simple transient tester circuit In general, the modulation capability of a control amplifier or control loop at high frequencies is very small, there is a high risk that the amplifier will be overdriven and distorted at high frequencies, resulting in incorrect results. With small signals in this frequency range, however, noise and interference are added. Therefore, an oscilloscope must be used to ensure that no overloads occur during the entire measurement. The important thing to look for is the ringing on the output immediately following the application of the load step. The frequency of the ringing indicates the control loop’s crossover frequency, and the number of periods occurring before the ringing dies out indicates the phase margin. Figure 1.214 shows the typical response of a circuit with different phase margins to the same transient condition. As can be seen, about one and a half periods indicates a phase margin of the order of 45°, which is generally considered acceptable. If significantly more ringing than this is observed, it is probably best to double-check the circuit’s performance on a network analyzer before going to mass production.

238

Fig. 1.214: Gain and phase measurement using a transient tester

Fig. 1.215: Transient response vs. phase margin

239

I Basic principles 7 Wireless Power Transfer Basics 7.1 Transmission paths for wireless power transfer There are variety of methods by which energy can be transferred in a contactless manner. The field of wireless power transfer can be divided into several technologies and categories, depending on different factors such as distance from the transmitter source and changes in the electromagnetic field.

Wirelss Power Transfer

Near Field Technology

Far Field Technology

Electromagnetic Induction

Electromagnetic Radiation

Electrodynamic Induction

Electrostatic Induction

Magnetic Induction

Magnetic Resonance

Microwave Power Transmission

Laser Power Beaming

Fig. 1.216: Methods of wireless power transfer Near field technology Near field

Wireless near field power transfer can transmit energy over a distance measuring one or more outer diameters of the transmission coil size or less than one wavelength (λ). Near field energy is a non-radiative power transmission technique but radiation losses may occur. In the normal case, only ohmic losses occur. Near field technology is divided into further classes: • Electromagnetic induction • Electromagnetic resonance • Electrostatic induction Electromagnetic induction

Induction

240

Wireless power transfer by means of electromagnetic induction in the near field is transmitted up to a distance of 1/6 of the wavelength of the transmission frequency. When current flows through a straight conductor the magnetic field strength H and the static magnetic field B are generated around it. When the wire is wound into a coil, the

magnetic field is amplified and it takes the form of a magnetic rod with a north and south pole around the coil. Ampere’s law states that the magnetic flux around a coil is directly proportional to the current flowing through the coil. The magnetic field strength of a coil is defined by the flux. The more turns the coil has, the greater the magnetic field around the coil. It is also possible to remove the electric current and place a permanent magnet inside the coil instead. A current is induced in the coil by mechanical movement of the permanent magnet and thus of the magnetic field. A current is also induced when the coil is moved over the permanent magnet. Thus, voltage and current can be induced by a changing magnetic field – whether the permanent magnetic moves or the coil moves. This process is called electromagnetic induction, which is the basic principle of a transformer. If two coils are not close enough, the entire primary magnetic flux will not flow through the secondary coil, resulting in poor coupling and leakage inductance. Leakage inductance is always present in a coupled coil system because the magnetic coupling of two coils is never ideal. This requires the transmitting and receiving coils to be well aligned with only one TXRX pair at a time. Electromagnetic resonance When the coupling between two galvanically isolated coils is poor, the result is leakage inductance. One of the reasons for poor efficiency between two coils with low coupling is the secondary leakage inductance, which is much greater than the load used on the secondary coil. This leakage inductance requires a large induced voltage on the secondary circuit. A higher current on the primary coil results in a higher induced voltage on the secondary side and thus generates higher losses. Therefore, it is common to reduce secondary leakage inductance by using a capacitance in a resonant circuit.

Electromagnetic resonance

In this case the transmitting and receiving coils can be misaligned. One transmitter can serve several receivers. Electrostatic induction Electrostatic induction is a method for wireless power transfer between the electrodes of a capacitor. A high frequency alternating voltage is generated on the plates of a capacitance that are positioned close to each other. This generates an electric field, resulting in a current displacement, in turn making the electric field the power source. The power transmission is proportional to the distance between the plates.

Electrostatic ­induction

241

I Basic principles 7.2 Basics Faraday’s law of induction describes the basic principle of energy transmission.

Fig. 1.217: Magnetic flux between coils If a current flows through the primary coil (transmitter coil), a magnetic flux Φ is generated. This magnetic flux Φ generates an inducted voltage in the secondary coil (receiver coil) according to the law of induction (see Figure 1.217). Since the transmitter and receiver coils are spatially separated from each other, only part of the magnetic flux Φ penetrates the receiver coil and can be used to supply a load. This magnetic coupling between the two coils is described by the ratio of magnetic fluxes Φ21/Φ1. From the equivalent circuit of an ideal transformer it is possible to derive an equivalent circuit diagram (Figure 1.218) for inductive coupled energy transmission.

R1 IP

Fig. 1.218: Equivalent circuit

242

L leak 1 2 IM

L leak 21

R2

IS M

RL

The coupling factor k is defined as follows: k=



M

(1.228)

L1 · L2 

M is the mutual inductance between the two coils. L1 and L2 are the self-inductances of the coils. The coupling factor depends very much on the offset of the coils in the x, y and z directions. Coupling can be improved by attaching ferromagnetic material to the coil to concentrate the magnetic flux. The ferrite material also acts as shielding against circuits in the vicinity of the transmitter and receiver coils by reducing the induced interference voltage. The coupling factor for an ideal transformer is 1, but with an inductively coupled system maximum couplings of 0.2–0.7 are achieved. The quality of the coils has a direct influence on the coupling and the efficiency of wireless energy transfer. The quality is defined as follows:



Q=

X L ω0 · L = RL R 

(1.229)

The efficiency of an inductively coupled energy transmission can be described with the following formula:



η=

POUT PIN 

(1.230)

PIN is the input power and POUT is the output power at the load RL. In loosely coupled inductive energy transmission systems, higher energy efficiency can be achieved using resonance mode by adding additional capacitances on the transmitter and receiver side.

7.3 Construction and calculation of the resonant circuit The structure of a wireless energy power system is shown pictorially in Figure 1.219.

Receiver Ferrite Receiver Coil Transmitter Coil Transmitter Ferrite

Fig. 1.219: Cross section of transmitter and receiver coils with ferrite 243

I Basic principles Accordingly, the simplified equivalent circuit diagram (Figure 1.220) can be derived from the equivalent circuit diagram (Figure 1.218).

CS CP CD

LP

LS RS

Fig. 1.220: Equivalent circuit of resonant circuits For greater compatibility, the Qi standard allows the receiver coil (Rx coil) to be used with many different transmitter coils (Tx coil). This changes the self-inductance of the secondary coil (Ls). The test setup to measure self-inductance is defined in the WPC (Wireless Power Consortium) specification. A 3.4 mm non-metallic spacer and ferrite shield (Tx ferrite) are designed to simulate components from the transmitter side onto the receiver coil. The Rx coil is placed on the spacer and the inductance (L`s) is measured at 100 kHz at 1V. The inductance (Ls) of the Rx coil without Tx coil is measured under the same measuring conditions. In the simplified equivalent circuit diagram, a parallel capacitance CP = 100 nF is defined on the primary side. The WPC specification defines different Tx coils with different inductance values. The following formulas from the WPC specification are used to calculate the series and parallel resonance capacities CS and CD on the secondary side at specific frequencies:



CS =

CD =



1 (100 kHz · 2π)2 · L’s



(1.231)

1 (1.0 MHz · 2π)2 · (LS –

1 CS 

(1.232)

Where L’s is the measured inductance on the test fixture and Ls is the inductance without the Tx coil. CS, calculated at a frequency of 100 kHz is used to enhance the power transfer efficiency. CD at a frequency of 1.0 MHz is used enable a resonant detection method. The optional switch is to be closed until the first packet transmits. 244

Finally, the quality can be calculated on the receiver side. This must be greater than 77.

Q =



2π · 1.0 MHz · L S Rs 

(1.233)

RS is the dc resistance of the Rx coil. Influence of resistance Litz wire is used to achieve a higher coil quality and thus a higher efficiency of the overall system. This results in a lower ac resistance, which is more frequency-independent and thus reduces the skin and proximity effects. The stranded wire also results in a lower dc resistance.

Influence of ­resistance

10000 1000

Proximity Effect

Impedance ( )

100 10

Skin Effect

1 0.1 0.01 10 kHz

100 kHz

1 MHz Frequency

10 MHz

100 MHz

Q

Fig. 1.221: Impedance measurement of coil over frequency The Q factor depends on the self-inductance of the coil, and thus also on its geometric and material properties, on its dc and ac resistance over the frequency.

245

I Basic principles 100

Impedance ( )

10

1

0.1 10 kHz

100 kHz

1 MHz Frequency

10 MHz

100 MHz

Q

Fig. 1.222: Q factor plot of coil over frequency. The maximum Q factor for the WPC standard should be between 100 and 205 kHz. The coil quality Q can be increased by additional ferromagnetic shielding material. Depending on the material used and the selected application (frequency), the coil quality can be improved. Reducing the copper cross-section increases the ac resistance at the resonance frequency and decreases the quality factor Q of the coil, thus reducing the efficiency of the entire contactless energy transmission path.

7.4 Coupling and efficiency The efficiency is directly proportional to the coil quality Q and to the coupling factor k. The coupling of a system is strongly influenced by the position of the transmitter coil in relation to the receiver coil.

246

Lateral Misalignement

Angular Misalignement

Vertical Misalignement

Fig. 1.223: Type of coil misalignment In a 2D FEMM simulation with an angled receiver coil, the coupling factor is reduced significantly with increasing angle and distance from the transmitter.

Fig. 1.224: A 2D FEMM simulation with an angled receiver coil

247

I Basic principles 1.0 0.9 0.8

Coupling Factor

0.7

0.6 0.5 0.4 0.3 0.2 0.1 0.0 0 mm

2 mm

0°

10°

4 mm 6 mm 8 mm Vertical Displacement 20°

30°

10 mm 40°

Fig. 1.225: Coupling reduction with increasing angle and distance from transmitter.

7.5 Shielding The ferromagnetic shielding material has a great influence on the overall behavior of wireless energy transfer. It serves to increase the self-inductance, improve or deteriorate the coil quality Q, coupling k and efficiency and also provides shielding against other circuit components by induced interference voltages. In addition, the magnetic field strength and the magnetic flux density between the transmitter and receiver coils are concentrated, as can be seen in Figure 1.226.

Fig. 1.226: A 2D FEM image of flux lines between transmitter (bottom) and receiver coil (top) with ferrite shielding for one half of cross section. Center axis at left edge. 248

In the case of vertical, horizontal or oblique displacement of the transmitter and receiver coils, the magnetic flux and magnetic flux density are mainly limited to the transmission system. There is no z-direction extension. From this it can be deduced that the magnetic field in the inductive coupling is limited to the area between the coils by the use of suitable ferrite shielding, as can be seen in Figure 1.226. Additional shielding measures are not necessary.

7.6 EMC measurements Since energy is transmitted in all wireless power applications, compliance with the EMC limits is not trivial. The challenge is that the transmitter and receiver coils behave like a transformer with a very poor coupling factor and a very large air gap. This results in a stray electromagnetic field in the vicinity of the coils. EMC measurements have shown that interference can occur in the broadband spectrum of the fundamental wave up to the frequency range of 80 MHz. The interference voltage levels should be maintained within the limit values with some margin. In general it can be stated that the limit values (e.g. for EN55022 Class B) can represent a hurdle in development that should not be underestimated. Here is an example of a spectrum in interference voltage measurement (9 kHz–30 MHz / Limit Class B):

Fig. 1.227 The H field (dI/dt) can inductively couple interference currents into adjacent line paths. A larger distance or shielding usually helps to prevent this. However the E-field (dV/dt) decouples very easily capacitively to earth. This can be observed when measuring the interference voltage and the interference field strength. These common mode disturbances must be countered in the low (kHz) and higher (MHz) frequency range. 249

I Basic principles Since the E-field is the main cause of EMC problems in wireless power applications, appropriate measures must be taken: • A slotted (smaller eddy current) metal surface (e.g. circuit board with copper) should be arranged under the coils (especially transmitter) and the circuit. This must be connected to the circuit ground or housing via a capacitor (e.g. 1–100 nF/2000 V WE-CSMH). As a result, large parts of the E-field are short-circuited to the source and no longer spread over ground. • Seal off the transmitter and receiver coils and their control with sufficient metal shielding and/or absorber material (WE-FAS/WE-FSFS). • If the leakage currents permit, Y capacitors (2 x 4.7 nF max.) can reduce the interference level over a wide spectrum (WE-CSSA). • To filter common mode interference in the low frequency range (50 kHz–5 MHz), a current-compensated choke from the following series can be used, depending on the operating voltage and current: WE-CMB / WE-CMBNC/WE-UCF/WE-SL/WE-FC • To filter common mode interference in the higher frequency range (5 MHz– 100 MHz), a current-compensated choke from the following series can be used, depending on the operating voltage and current: WE-CMBNiZn/WE-CMBNC/WESL5HC/WE-SCC • Capacitors between +/– L/N from these series can also be connected to differential mode (push-pull), depending on the operating voltage: WE-FTXX / WE-CSGP. • Since very high ac currents flow throughout the entire circuit, depending on the application, a compact and low-inductance PCB layout is decisive for success in EMC. The components of the power stage and the resonant circuit should be placed very close to each other and connected inductively with large copper surfaces (polygons). In general, it is recommended to contact the responsible EMC laboratory at an early stage of development in order to carry out measurements during development. Changes late in development are always associated with a lot of money and great effort.

7.7 The dominating standards The success of these solutions depends on maintaining a standard for transmitters and receivers. The system will only establish itself on the market if it is guaranteed that the device can be operated independently of the manufacturer at any supply station that meets the standard. What are the standard approaches and what is the technology behind them? WPC

Qi

250

Wireless Power Consortium (WPC)

Qi is an open source standard and currently the dominant among mobile devices. • Energy transmission with inductive coupling over short distance (mm range) • Transmitter (Tx) and receiver (Rx) coils are inductively coupled coils. • The magnetic field is concentrated in the narrow area between the transmitter and receiver coils • Each transmitter can only serve one receiver • Various performance classes (5W, 15W, higher in planning up to 2.4kW) • Frequency range 87–205 kHz • Coil shapes: wound on ferrite or printed onto circuit board • The most established solution worldwide Airfuel

Airfuel

AirFuel is a merger of the A4WP (Alliance for Wireless Power) and the PMA (Power Matter Alliance) • The first charging method is magnetic resonance from the previous standard A4WP: A transmitter resonant circuit provides the energy at a resonant frequency. Receivers tuned to the resonance frequency can take over the energy • Greater distance in z-direction (50mm) and exact positioning of the receiver ­unnecessary • One transmitter can serve several receivers simultaneously • Performance class planned for smartphones and tablets, currently up to 22W • Frequency ranges: Energy 6.78 MHz (ISM band), data 2.4 GHz (BTLE) • The second charging method is an inductively coupled solution from the previous standard PMA • The inductively coupled solution uses a different protocol and a different transmission frequency band (details for members only) than the WPC’s Qi solution • No direct compatibility with Qi, dual standard receivers are available Proprietary Systems All standards have their advantages and disadvantages. Approximately 30% of applications in industrial or medical technology are not reliant on being compatible with other manufacturers or other terminal devices. With this fundamental elimination of compatibility, manufacturers from all industries, apart from the consumer and automotive industries, are developing proprietary system solutions. These proprietary solutions do not always require customer-specific passive components for wireless power transfer allowing for standard WE-WPCC transmitter and receiver coils can be used.

251

I Basic principles 8 RF basics 8.1 RF inductor characteristics In order to assess and compare the RF inductors, it is necessary to understand in detail the main characteristics of an inductor for high frequency applications. These characteristics are: the inductance value with its corresponding tolerance, the quality factor, the self-resonant frequency, the dc resistance, the rated current and size. Inductance

8.1.1 Inductance (L) and tolerance (%) In most radio frequency applications, high order filters, tuned circuits or impedance matching applications, it is very important that the inductance curve is as flat as possible at the working frequency point (see please Figure 1.228).

Inductance

100 nH

10 nH

1 nH 1 MHz

10 MHz

100 MHz Frequency

1 GHz

10 GHz

L 744 901 115

Fig. 1.228: Inductance curve of WE-TCI 0402 (PN: 744 901 115) In addition, the inductance should be independent of the current and the temperature at the required frequency. This is the reason why most of the RF Inductors have ceramic or air cores, because they have a very low thermal coefficient, which allows high inductance stability. Both ceramic and air have no magnetic properties and according to the inductance equation of a coil (1), if µr ≈1 the inductance value can only increase with the number of turns. This is the reason why RF inductors with ceramic or air cores only reach inductance values on the range of nH. In case that larger inductance values in the range of µH are needed, ferrite cores (whose permeability µr>>1), as in our series WE-RFI and WE-RFH, are required. 252

L =



µr µo A eff N 2 leff 

(1.234)

L: inductance µr: relative permeability µo: free space permeability (4π · 10–7 Vs/1 Am) Aeff: effective cross-sectional area of the coil core leff: effective path length in the coil core In many RF applications such as filtering, matching or oscillator circuits, as well as inductance stability, it is also very important to have tight inductance tolerances where the real inductance value is as close as possible to the nominal value. In the specifications, both the inductance value and its tolerance are shown at a certain frequency point. 8.1.2 Self-Resonant Frequency (SRF) Below the SRF the inductor behaves as an inductor. Exactly at the SRF the inductor only has real losses (i.e. it behaves as a pure resistor) and beyond the SRF it behaves as a capacitor. In addition, as you can see in the diagram (Figure 1.229), the ­impedance has its maximum value at the SFR. SRF

10 µH

100 kΩ

5 µH

75 kΩ

50 kΩ

Inductor

–5 µH

–10 µH 0 MHz

|Z|

Ls

Capacitor

0 µH

25 kΩ

250 MHz 500 MHz 750 MHz Frequency Ls

0 kΩ 1 GHz

|Z|

Fig. 1.229: Ls (red) and |Z| (black) of an inductor WE-RFH 1008 (PN: 744758256A) As shown in the Figure 1.230, between the wires or internal electrodes of any inductor there is a distributed capacitance. Considering this parasitic capacitance, the equivalent circuit results in the Figure 1.231. 253

I Basic principles

Fig. 1.230: Schematic of a RF inductor. Parallel wires act like electrodes of a ­capacitor generating a distributed capacitance

L

R

C

Fig. 1.231: Equivalent circuit of a RF inductor: L represents the inductance, R the losses of the wire and C the distributed capacitance It is important to mention that there are several additional parasitic effects that increase with frequency. The S-parameters describe accurately the properties of the component as a function of frequency, which account for all parasitic phenomena. Therefore, Würth Elektronik offers the S-parameters of every RF inductor and Modelithics models of most. Modelithics measures the S-parameters of an inductor on different substrate types and thicknesses, creating global models that scale the substratesensitive parasitic effects resulting in very accurate simulations. The relation between the inductance (L), the distributed capacitance (Cp) and the selfresonant frequency (SRF) is shown in the Equation 1.235: SRF =



1 2π LC p 

(1.235)

Thus, the SRF is the frequency point at which the parasitic capacitance forms a parallel resonance with the inductance, or in other words, the frequency point where the capacitance is canceling out the inductance (i.e. both reactances are equal XL = XC).

254

From the previous equation, it is also visible that increasing the inductance and/or the parasitic capacitance lowers the SRF, and vice versa. This is the reason why the larger the inductance value, the lower the SRF.

In most cases, the inductance value must be stable and as close to the desired value as possible. In the diagram, it is visible that for this purpose, the working frequency point should be far away from the SRF. A conservative rule of thumb is to work in a frequency a decade below the SRF. There are exceptions for example, in case that the RF inductor is used as a choke for a certain frequency range, it is convenient that the commented range is close to the SRF of the inductor. In this way, the impedance will be at its maximum. 8.1.3 Quality factor (Q)

Q factor

The Q factor is one of the first things that every RF engineer takes into account to assess the quality of an RF inductor. The Q factor is the ratio (therefore is unit less) between the stored energy (XL: reactance) and the losses (RS). Higher Q factor means less losses and therefore less attenuation of the signal (minimizing the power consumption).

Q=



X L ωL = RS  RS

(1.236)

800

80

7000

700

70

6000

600

60

5000

500

50

4000

400

3000

300

30

2000

200

20

1000

100

20

40

Q

8000

RS (Ω)

XL (Ω)

Since the inductance is roughly constant, from Equation 1.236 it is deducible that the Q factor increases with the frequency. In Figure 1.232, it is visible that for low frequencies the losses raise nearly linearly and therefore the Q factor too. However, for higher frequencies parasitic effects (such as skin effect) appear, increasing the losses faster and consequently the Q factor reaches its maximum and starts to decrease.

10

0 0

100

200

300

400

500

Frequency (MHz)

Fig. 1.232: Q (black), XL (grey) and RS (red) of an inductor WE-RFH 1008 (PN: 744758256A)

255

I Basic principles Depending on the manufacturer, the Q factor is given either as minimum or as typical value for a certain frequency point. In the case of Würth Elektronik, the Q factor is given as the minimum value in order to guarantee our customers a minimum of quality. 8.1.4 DC resistance (RDC) RDC (or DCR) is the resistance of the inductor for dc current. Although for higher frequencies the losses are larger due to effects such as skin effect or proximity effect, the RDC is a good and easy starting point to evaluate the losses of the RF inductor. Thicker wire means lower RDC, but usually means a larger component size. The Q factor and the RDC, are part of the total losses (RS) and inversely proportional meaning smaller RDC results in a larger Q factor. 8.1.5 Rated current (IR) IR indicates the current at which the inductor increases its temperature by a specific value (ΔT) (Figure 1.233). The increase depends on the component series (in our case: ΔT = 15 K, ΔT = 20 K or ΔT = 40 K). In standard RF applications, the current is usually small, and therefore this parameter plays a secondary role. 80

Temperature Rise (K)

70 60 50 40 30 20 10 0 0

10

20

30

40

50

Current (A)

Fig. 1.233: IR curve of an inductor WE-AC HC (PN: 7449152090) The rated current is specified as the maximum dc current (in A or mA) that can pass through the inductor without reaching the maximum rated ambient temperature. 8.1.6 Size In RF circuits size matters! In a market where ever-smaller circuits are required, inevitably engineers place a lot of importance on this parameter. Würth Elektronik offers RF inductors from 0201 up to 1208 (inches): 256

Inches (Metric)

WE-KI

WE-KI HC

1 nH–120 nH

1 nH–51 nH 1.8 nH–390 nH

WE-RFI

WE-RFH

0201 (0603) 0402 (1005) 0603 (1608)

1.6 nH–1 µH

0805 (2012)

2.2 nH–1.8 µH

0.47 µH–10 µH

1008 (2520)

3.3 nH–1 µH

1.2 µH–47 µH

1210 (3225)

22 nH–1 µH

WE-MK

WE-TCI

1 nH–33 nH

1 nH–10 nH

1 nH–270 nH

1 nH–27 nH

1 nH–470 nH 0.47 µH–10 µH

Tab. 1.19: Sizes and inductance ranges of WE RF inductor series

In the case of the air core inductors, the size is given in mm and it depends on the inductance value (i.e. number of turns). All the mentioned characteristics are interrelated. An inductor of size 0402 cannot have as many turns as an inductor of 0805 and therefore the maximal inductance value will be lower. Smaller size means thinner wire, and that results in a larger Rdc and lower Q factor. Therefore, engineers must take into account some trade-offs between size, performance and structure in order to select the right RF inductor for their applications. Once the parameters specified in the datasheet of a RF inductor are understood, the next step is to analyze every series in detail. Thus, the advantages and the particular features of every RF inductor structure and series are pointed. Wire wound inductors Multilayer inductors With core 8.1.7 Structures

Thin film inductors

Air core

Würth Elektronik offers three different kind of RF inductor structures: wire wound (with and without core), multilayer and thin film inductors. Table 1.20. shows an overview of our RF inductor series.

WE-KI/WE-KI HC/WE-RFI/WE-RFH

WE-CAIR/WE-AC HC

Tab. 1.20: RF inductor structures offered by WE

WE-MK

WE-TCI 257

I Basic principles Wire wound: As the name states, this structure is formed by winding a copper wire around a core or just “the air”. Compared with other structures, the wire is thicker and therefore the losses are lower. As we have seen before, low losses mean: low Rdc, high Q factor and high rated current. Moreover, the number of turns possible is fairly high, so a broad inductance range can be achieved with this structure. However, this structure has disadvantages. Due to the wire thickness and their proximity to each other, the capacitive effect between them is considerably high when the number of turns is high. This relatively high parasitic effect results in a lower SRF comparing with other structures. Multilayer: This structure is formed by many ceramic layers with printed conductors stacked on top of each other. The coil is then built connecting the conductors through vias. This fabrication process enables very small sizes and the best inductors in terms of price. On the other hand, due to the narrow size of the conductors, the losses are higher than the wire wound structure. This results in: large Rdc, quite low Q factor and low rated current. Thin film: The thin film technology consists of printing the conductor on a ceramic layer using a photolithographic process. This very accurate and repeatable process, which consequently provides a very tight inductance tolerance. Furthermore, the thin film inductors are extremely thin and very small sizes are also possible. Since the chip surface is that small, the number of windings is quite limited and therefore the inductance range is considerably low in comparison with other structures.

S-Parameters

8.2 S-Parameters – Basic principles 8.2.1 Basic theoretical principles Whereas the approach of measuring voltage and current is useful at low frequencies, problems arise at high frequencies. The threshold, above which the high frequency approach is required, depends on the dimension of the object under investigation compared with the operating wavelength. This is the case as soon as the dimensions of the components and leads are no longer small compared with the wavelength.

Z-Parameters

258

The analysis of a RF network by measuring the Z-Parameters is only of l­imited suitability for the following reasons: • It is very difficult to achieve an open lead for high frequencies. • Active components are unstable if they are operated in a short-circuit or open mode.

• The voltages and currents vary dependent on the lead length. • Falsification of the measurement value due to parasitic capacitances and inductances of the measuring probe. It is favorable to use the S-parameters derived in the following section for characterization in the RF range. The advantage is a simple treatment of the impedance transformation effect of leads by means of a Smith chart. The relationship between the frequency and the wavelength is as follows:

Fig. 1.234: Relationship between frequency and wavelength In high frequency technology a network is viewed with the aid of wave theory. The twoport network in Figure 1.235 is considered by way of illustration

Fig. 1.235: Two-port network with forward and reverse waves It is assumed that there are infinitesimally short leads at the input and output of the two-port network with wave resistance ZL. On the input lead there exist an incident (generator) wave with Uh1 and Ih1 and a reflected wave with Ur1 and Ir1.

259

I Basic principles The following relationships arise for voltage and current at the input:

(1.237)





At the output:

(1.238)





The relationship between the incident and reflected waves is described by the scattering parameters (S-parameters) as follows:



(1.239)



The following relationship results in matrix notation:





(1.240)

The scattering parameters can be determined as follows: The reflection factor provides the relationship between the voltages of the reflected wave to the incident wave on the output side we have adaption (gate 2), that means the load resistor is identical with the wave resistor ZL.

260

Fig. 1.236: Input reflection factor (Port 1) S11

Fig. 1.237: Output reflection factor (Port 2) S22

Fig. 1.238: (Forwards) transmission coefficient S21

Fig. 1.239: (Reverse) transmission coefficient

261

I Basic principles Input reflection

In operation with a load impedance Zl differing from ZL, the input reflection factor can be calculated with

Fig. 1.240: Input reflection factor Matching

Wave resistance matching at the input and output If equiphase voltages arise on the lead, the result is a maximum voltage, whereas counter-phase voltage components lead to a voltage minimum.

Fig. 1.241: Wave resistance matching

262

The relationship between Umax and Umin is determined by mismatching and is termed voltage standing wave ratio (VSWR).





(1.241)

Without reflections, the reflection factor rL is zero and the VSWR is 1. Besides the standing wave ratio, the return loss (unit dB) is a common measure of mismatching. The return loss Ar results from the reflection factor and is expressed as the decibel reciprocal magnitude of the reflection factor:





VSWR

Return loss

(1.242)

If the load impedance Z is matched to the wave resistance ZL no reflections occur and the reflection factor is 0. The case of mismatching and the occurrence of a standing wave are represented in the following figure.

Mismatching

Fig. 1.242: Example of a standing wave 8.2.2 Conceptual design of a matching circuit by means of a Smith chart The approach for matching an antenna to the internal resistance of the generator and the basic theoretical principles of the Smith chart are explained in this section. A Smith chart can be used as an aid for designing the matching network. Smith chart: The basic theoretical principles as well as the special features of the Smith chart are explained as follows.

263

I Basic principles Reflection factor

The complex reflection factor is calculated from the normalized impedance using the following equation, whereby the characteristic impedance of the transmission line is termed Z0.





(1.243)

In the ideal case, the impedance Z (for example, the input impedance of an antenna) is identical with the characteristic impedance Z0. No reflections occur, one speaks of (ideal) matching.

Reflection factor plane

The impedance is a complex number, which can be represented in an infinite twodimensional semi-plane. The real component is entered on the x axis, the imaginary component on the y axis. Only the semi-plane with positive effective resistances (i.e. x > 0) is considered, as negative real parts only arise with active circuits. The Smith chart is generated through iso­gonal conformal mapping from the transformation of the normalized impedance semi-plane onto the reflection factor plane, i.e. onto the inside of the unit circle. Accordingly, the semi-plane with the negative real component would lie outside the unit circle in the Smith chart. For positive real parts of impedances, the magnitude of the reflection factor is always less than 1, for negative real components, the magnitude of the reflection factor is greater than 1. The normalized impedance is therefore mapped onto a limited region resulting in a handy chart. The outer circle of the chart boundary corresponds to the magnitude 1 of the reflection factor. A complete Smith chart may be seen in Figure 1.243. The meaning of the circles shown and the method of working with Smith charts are now explained step-by-step.

264

Smith chart

Fig. 1.243: Smith chart The circles shown represent the coordinate grid in the r plane for the associated ­impedance. For reasons of clarity, only the coordinate lines for the real component Re{Z/Z0} = R = const are shown in Figure 1.244. The reflection factor plane is known as the Re{r},Im{r} plane. The equations of the circles are given without derivation: The center of the circles of the constant real component





(1.244)

Normalization of the real component





(1.245)

265

I Basic principles The center of the circles of the constant real component



(1.246)



is located at



(1.247)



and the radius is given by



Z=0

0

0.2

0.5

1

short

Fig. 1.244: Circles with Imaginary component

(1.248)



2

Z

∞

open

= const

The circles for the constant imaginary component Im{Z/Z0} = I(1.249)

266

are described with the equation.





(1.250)

whereby the center of the circle is located at





(1.251)

and the radius is given by



(1.252)



1j 2j

0.5j

0.2j inductive

capacitive -0.2j

-0.5j

-2j -1j

Fig. 1.245: Circles with

= const

As a short is defined by an impedance of 0, the corresponding point is entered at the point z = 0 as shown in Figure 1.245. The open is defined as an infinite impedance and is also entered. The upper half of the Smith chart with positive imaginary component represents inductive impedances, the lower half capacitive impedances. Figure 1.246 illustrates how the magnitude and phase of the reflection factor can be read off easily by entering the normalized impedance. The magnitude of the reflection factor corresponds to the distance between the entered point and the origin. This measured distance is related to the radius of the chart, which corresponds to a magnitude

Reflection factor 267

I Basic principles of 1. The margined angular scale of the chart specifies the value for the phase of the reflection factor.

Fig. 1.246: Reading off the reflection factor A further application option for the Smith chart is the simple transition from the impedance plane to the admittance plane and vice versa. The transition between the impedance and the admittance is achieved by mirroring around the origin, i.e. at the center of the Smith chart. The scaling on the Smith chart shows that the reflection factor can be read off with good precision. For most applications, this degree of precision is entirely adequate and, for example, is greater than the degree of precision with which cable lengths can be cut to size in practice.

Admittance plane

268

The impedance is changed by parallel and series connexion of concentrated components. The Smith chart offers a simple means of representing these shifts. The impedance plane is preferred for use with series connections, whereas the admittance plane is recommended for parallel connections. The transition is performed by point mirroring around the origin. Conversely, starting with the shift read off the Smith chart, the values of the components can be determined. The formulas specifying the relationship between component value and shift are listed in the following table.

Tab. 1.20: Relationship between component value and shift For example, the introduction of an inductor in series increases the imaginary component of the impedance, whereas the real component remains unchanged. This corresponds to an upward shift along a circle with constant real component in the Smith chart. The parallel connection of a resistor leads to an increase in the real component of the admittance. Accordingly, there must be a shift to the right along a circle of constant imaginary component. The shift directions are illustrated in Figure 1.247.

Fig. 1.247: Shift directions in the Smith chart

269

I Basic principles Lossless line

The impedance transformation from a lossless line can be easily represented by rotation around the chart center. The following equation describes the transformation for a lossless line from the end terminals to the input terminals. This assumes that the Smith chart is normalized to the characteristic impedance of the transformation line.





(1.253)

The angle of rotation





(1.254)

is proportional to the normalized line length





(1.255)

The rotation is clockwise starting from the impedance to be transformed. The directions of rotation are independent of whether one is in the impedance or admittance plane. The scale zero point of the normalized lead length has been set arbitrarily, as only the difference reading is of relevance. Matching

The approach for antenna matching After careful calibration, the input reflection factor or the input impedance is measured at the operating frequency. The value obtained is entered in a Smith chart with the help of which a matching network is designed. The aim of matching is to achieve a shift to the desired impedance (e.g. 50 Ω) with just a few components. A test circuit should then be built and the matching verified by measurement. The values found can be used as starting values and the matching further optimized experimentally.

270

The use of the Smith chart to design the matching network is now described on the basis of an example. The operating frequency is 2.45 GHz. The measured input impedance of the antenna normalized to 50 Ω is 0.5–0.3 j and is entered in the Smith chart. A possible path to the center (50 Ω) starts with the upward shift along the circle with constant real component 0.5. To arrive at the center, a capacitance must be connected in parallel in the Smith chart. Here the dashed circle is drawn through the center (the target point) as shown in Figure 1.249 as a helpful construction. The point of intersection of this dashed circle with the circle of constant real component 0.5 defines the length of the shift. This is given by Dz = 0.5 j – (–0.3 j) = 0.8 j. As the admittance plane is recommended for the parallel connection of a blind element, the obvious transition to the admittance plane is by means of point mirroring around the chart center. The mirror point is at 1 – j. The midpoint is reached by shifting upwards

along the circle with constant real component 1. The length of the shift is given by Dy = 0 – (–1j) = j. The component values are now determined taking the relationships from table 1.20.

(1.256)



Solution of the equations for the component values results in Ls = 2.6 nH and Cp = 1.3 pF. The matching network is shown in Figure 1.248. To verify the result, the input impedance is now calculated with the following equation and 50 W obtained with the values determined.

(1.257)  Finally, reference is made to the fact that there can be several transformation paths. “Natural” circuits are preferable, i.e. circuits such as those in Figure 1.248 in which the capacitor is connected to ground and the inductor is connected in series.

Fig. 1.248: Matching network

Matching network

271

I Basic principles

Fig. 1.249: Impedance transformation in the Smith chart

272

Trilogy of Magnetics

Design Guide for EMI Filter Design, SMPS & RF Circuits

Components

273

II Components Part 2: Components

274

1 2 2.1 2.2 2.2.1 2.3 2.4 2.5 2.5.1 2.5.2 2.6 2.6.1 2.6.2 2.7 2.8 2.9 2.10 2.11 2-12 2.13 2.14 3 3.1 3.2 3.3 3.4 3.5 3.6 3.7 3.8 3.9 3.10 3.11 3.12 3.13 3.14 4 4.1 4.2 4.3 4.4 4.5 4.6 4.7 4.8

Overview of components. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 276 EMC Components. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 286 Various forms of ferrites . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 286 WE-CBF SMD Ferrite. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 287 Design guidelines. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 293 WE-PBF, WE-SUKW High Current Beads . . . . . . . . . . . . . . . . . . . . . . 300 WE-MPSB SMD Multilayer Power Suppression Bead . . . . . . . . . . . . . 303 Through Hole Components . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 304 WE-UKW 6-Hole Ferrite Bead. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 304 WE-MLS Ferrite Bridge. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 307 Snap ferrites . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 309 STAR-TEC, STAR-RING, STAR-CLIP. . . . . . . . . . . . . . . . . . . . . . . . . . 309 STAR-GAP Snap ferrites with a defined air gap . . . . . . . . . . . . . . . . . 317 WE-MI Multilayer Inductor. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 323 WE-FI Radio Interference Suppression Choke . . . . . . . . . . . . . . . . . . 325 Common Mode Chokes. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 329 WE-CNSW Common Mode Chokes for Data and Signal Lines . . . . . . . 330 WE-SL, WE-SLM, WE-SL1, WE-SL2, WE-SL3, WE-SL5. . . . . . . . . . . . 331 WE-LF, WE-CMB, WE-FC Common Mode Chokes for Power Lines . . . . 336 WE-ExB Common Mode Power Line Choke. . . . . . . . . . . . . . . . . . . . 343 WE-LPCC Common Mode Power Line Choke. . . . . . . . . . . . . . . . . . . 344 Power Magnetics – Inductors . . . . . . . . . . . . . . . . . . . . . . . . . . . 345 WE-PMI, WE-GF, WE-GFH – Power Inductors. . . . . . . . . . . . . . . . . . . 345 WE-MAPI Shielded SMD Metal Alloy Power Inductor. . . . . . . . . . . . . . 348 WE-SI Power Inductors. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 349 WE-PD SMD Power Inductors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 357 WE-TPC, WE-HCI, WE-HCC Power Inductors . . . . . . . . . . . . . . . . . . . 368 WE-HCF SMD High Current Inductor. . . . . . . . . . . . . . . . . . . . . . . . . 378 WE-PD2 HV, WE-PD HV, WE-TI HV High Voltage Inductors. . . . . . . . . . 379 WE-PFC Power Factor Correction Choke. . . . . . . . . . . . . . . . . . . . . . 380 WE-EHPI Energy Harvesting Coupled Inductor. . . . . . . . . . . . . . . . . . 382 WE-DD Double Chokes. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 383 WE-DPC SMD Dual Power Choke. . . . . . . . . . . . . . . . . . . . . . . . . . . 387 WE-MCRI SMD Molded Coupled Inductor . . . . . . . . . . . . . . . . . . . . . 388 WE-MTCI SMD Multi-Turn Ratio Coupled Inductor . . . . . . . . . . . . . . . 389 WE-DPC HV, WE-CPIB HV, WE-TDC HV SMD Coupled Inductors. . . . . . 390 Power Magnetics – Transformers . . . . . . . . . . . . . . . . . . . . . . . . 390 WE-FLEX & WE-FLEX + Transformers. . . . . . . . . . . . . . . . . . . . . . . . . 390 WE-FLEX HV Flexible Transformer High Voltage. . . . . . . . . . . . . . . . . 398 WE-PoE Power-over-Ethernet Transformers. . . . . . . . . . . . . . . . . . . . 399 WE-PoEH Power over Ethernet High Power Transformer. . . . . . . . . . . 401 WE-LLCR Resonant Converter Transformer. . . . . . . . . . . . . . . . . . . . 402 WE-UNIT Offline Transformers. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 402 WE-GDT Gate Drive Transformers. . . . . . . . . . . . . . . . . . . . . . . . . . . 404 WE-CST Current Sense Transformers. . . . . . . . . . . . . . . . . . . . . . . . 404

5 5.1 6 6.1 6.2 6.3 6.4 6.5 7 7.1 7.2 7.3 7.4 7.5 7.6 8 8.1 8.2 8.3 8.4 8.5 9 9.1 9.2 9.3 9.4 9.5

Wireless Power Transfer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 407 WE-WPCC, Wireless Power Transfer Coils. . . . . . . . . . . . . . . . . . . . . 407 Signal & Communications. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 408 WE-LAN Transformers. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 408 WE-LAN HPLE – 1000BASE-T – High Performance, Low EMI . . . . . . . 412 WE-RJ45LAN/WE-RJ45 HPLE Transformers integrated with RJ45 connector. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 413 WE-LAN 10G LAN Transformer PoE/PoE+ . . . . . . . . . . . . . . . . . . . . . 417 WE-DSL Telecom Transformers . . . . . . . . . . . . . . . . . . . . . . . . . . . . 418 RF Inductors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 422 RF Inductors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 422 WE-KI, WE-KI HC, WE-FRI, WE-RFH Ceramic Wire Wound Inductors. . . 423 WE-MK Multilayer Ceramic Inductor. . . . . . . . . . . . . . . . . . . . . . . . . 425 WE-TCI Thinfilm Chip Inductors. . . . . . . . . . . . . . . . . . . . . . . . . . . . 427 WE-Cair, WE-AC HC High Current Air Coil. . . . . . . . . . . . . . . . . . . . . 428 WE-AC HC Current Air Coil . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 429 LTCC Components . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 430 LTCC (Low Temperature Co-fired Ceramic) . . . . . . . . . . . . . . . . . . . . 430 WE-LPF Multilayer Chip Low-Pass Filter. . . . . . . . . . . . . . . . . . . . . . 431 WE-BPF Multilayer Chip Band-Pass Filter. . . . . . . . . . . . . . . . . . . . . 432 WE-BAL Multilayer Chip Balun. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 434 WE-MCA Multilayer Chip Antenna. . . . . . . . . . . . . . . . . . . . . . . . . . . 435 ESD and Surge Protection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 436 Basic principles . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 436 Varistors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 438 The ESD suppressor. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 459 TVS diodes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 466 Design layout. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 472

275

II Components 1 Overview of components

EMC Components Ferrites for PCB Assembly WE-TMBS Z @ 100 MHz: IR: RDC: Frequency Range:

10 ~ 2100 Ω 150 ~ 7500 mA 0.09 ~ 2.1 Ω 6 ~ 3000 MHz

WE-CBF Z @ 100 MHz: IR: RDC: Frequency Range:

5 ~ 2700 Ω 50 ~ 6000 mA 0.008 ~ 1.5 Ω 6 ~ 2000 MHz

WE-CBF HF Z @ 1 GHz: IR: RDC: Frequency Range:

244 ~ 2005 Ω 50 ~ 600 mA 0.25 ~ 1.8 Ω 300 ~ 3000 MHz

WE-MPSB Z @ 100 MHz: IR: RDC: Frequency Range:

8 ~ 600 Ω 2100 ~ 10.500 mA 1.0 ~ 80.0 mΩ 1 ~ 3000 MHz

WE-PBF Z @ 100 MHz: IR: RDC: Frequency Range:

42 ~ 98 Ω 6 A 0.6 ~ 0.9 mΩ 6 ~ 2000 MHz

WE-PF IR: RDC: Frequency Range:

WE-CMS Z @ 25 MHz: Z @ 200 MHz: IR: Frequency Range:

276

4.5 ~ 10 A 9 ~ 30 mΩ 1 ~ 100 MHz



20 - 34 Ω 30 ~ 52 Ω 5A 1 ~ 3000 MHz

WE-SUKW Z @ 25 MHz: Z @ 100 MHz: Frequency Range:

272 ~ 425 Ω 416 ~ 580 Ω 0.1 ~ 800 MHz

WE-UKW Z @ 25 MHz: Z @ 100 MHz: Frequency Range:

145 ~ 920 Ω 230 ~ 1240 Ω 0.1 ~ 500 MHz

WE-MLS Z @ 25 MHz: Z @ 100 MHz: Frequency Range:

115 ~ 292 Ω 150 ~ 334 Ω 10 ~ 800 MHz

WE-WAFB Z @ 10 MHz: Z @ 100 MHz: Frequency Range:

28 ~ 65 Ω 70 ~ 130 Ω 1 ~ 1000 MHz

Ferrites for Cable Assembly WE-STAR-BUENO Z @ 25 MHz 1 turn: Z @ 100 MHz 1 turn: Cable diameter: Frequency Range:

100 ~ 180 Ω 150 ~ 250 Ω 2.5 ~ 8.5 mm 1 ~ 1000 MHz

WE-STAR-TEC LFS Z@ 1 MHz 1 turn:  Z@ 10 MHz 1 turn:  Cable diameter:  Frequency Range: 

20 ~ 94 Ω 32 ~ 65 Ω 3.5 ~ 21 mm 300kHz ~ 30 MHz

WE-STAR-TEC Z @ 25 MHz 1 turn: Z @ 100 MHz 1 turn: Cable diameter: Frequency Range:

98 ~ 306 Ω 182 ~ 525 Ω 3.5 ~ 25 mm 1 ~ 1000 MHz

WE-STAR-GAP Z @ 25 MHz 1 turn: Z @ 500 MHz 1 turn: Cable diameter: Frequency Range:

28 ~ 35 Ω 345 ~ 400 Ω 4.5 ~ 12.5 mm 100 ~ 2000 MHz

WE-STAR-RING Z @ 25 MHz 1 turn: Z @ 100 MHz 1 turn: Cable diameter: Frequency Range:

64 ~ 142 Ω 119 ~ 327 Ω 8 ~ 27 mm 1 ~ 1000 MHz

WE-TOF Z @ 25 MHz 1 turn: Z @ 100 MHz 1 turn: Cable diameter: Frequency Range:

WE-STAR-FLAT Z @ 25 MHz 1 turn: Z @ 100 MHz 1 turn: No. of Pins: Frequency Range:

42 ~ 97 Ω 101 ~ 194 Ω 26 ~ 50 1 ~ 1000 MHz

WE-AFB LFS Z @ 25 MHz 1 turn: Z @ 100 MHz 1 turn: Cable diameter: Frequency Range:

46 ~ 300 Ω 70 ~ 451 Ω 3.3 ~ 17.5 mm 0.15 ~ 30 MHz

WE-STAR-CLIP Especially for the fixation of STAR-TEC and STAR-FIX Snap Ferrites

WE-AFB Z@ 1 MHz 1 turn:  Z@ 10 MHz 1 turn:  Cable diameter:  Frequency Range: 

30 ~ 130 Ω 40 ~ 100 Ω 4.55 ~ 12.5 mm 1 ~ 1000 MHz

WE-NCF Z @ 25 MHz 1 turn: Z @ 100 MHz 1 turn: Cable diameter: Frequency Range:

48 ~ 100 Ω 93 ~ 200 Ω ≤ 7.8 ≤ 26.5 mm 1 ~ 1000 MHz

WE-SAFB Z @ 25 MHz 1 turn: Z @ 100 MHz 1 turn: Cable diameter: Frequency Range:

20 ~ 144 Ω 40 ~ 278 Ω 0.55 ~ 4 mm 1 ~ 1000 MHz

WE-SPLITRING Z @ 25 MHz 1 turn: Z @ 100 MHz 1 turn: Cable diameter: Frequency Range:

48 ~ 100 Ω 93 ~ 200 Ω ≤ 7.8 ≤ 26.5 mm 1 ~ 1000 MHz

WE-RIB Z @ 25 MHz 1 turn: Z @ 100 MHz 1 turn: Cable diameter: Frequency Range:

20 ~ 101 Ω 45 ~ 176 Ω 0.8 ~ 3.5 mm 1 ~ 1000 MHz

WE-SFA Z @ 25 MHz 1 turn: Z @ 100 MHz 1 turn: No. of Pins: Frequency Range:

27 ~ 148 Ω 57 ~ 267 Ω 10 ~ 64 1 ~ 1000 MHz

WE-FLAT Z @ 25 MHz 1 turn: Z @ 100 MHz 1 turn: Types: Frequency Range:

1 ~ 90 Ω 42 ~ 166 Ω round, square, edged 1 ~ 1000 MHz



25 ~ 110 Ω 37 ~ 200 Ω 3.0 ~ 33.4 mm 1 ~ 1000 MHz

EMC Filters WE-CLFS IR ILeak RDC

1.5 ~ 20 A 0.173 ~ 0.785 mA 15 ~ 300 mΩ

WE-FLAT Ferrite for Flexible Printed Circuit Boards Z @ 25 MHz 1 turn: 7 ~ 71 Ω Z @ 100 MHz 1 turn: 19 ~ 130 Ω Types: round, square 1 ~ 1000 MHz Frequency Range:

WE-FCAC Easy fixation for flat cores on ribbon cables Max. No. of Poles: 16 ~ 40 1 ~ 1000 MHz Frequency Range:

277

II Components Filter Chokes WE-MI L: IR: RDC: Frequency Range:

WE-SD L: IR: RDC: Frequency Range:

WE-FI L: IR: RDC: Frequency Range:



0.047 ~ 10 µH 3 ~ 300 mA 0.15 ~ 2.1 Ω 1 ~ 100 MHz

2 ~ 10 µH 2.5 ~ 15 A 1.7 ~ 33 mΩ 0.01 ~ 90 MHz

8.2 ~ 860 µH 0.4 ~ 5 A 0.01 ~ 0.45 Ω 0.01 ~ 0.3 MHz

Common Mode Power Line Chokes WE-CNSW Z @ 100 MHz: IR: RDC: Frequency Range: Number of Windings:

45 ~ 10000 Ω 90 ~ 2000 mA 0.05 ~ 8.0 Ω 1 ~ 3000 MHz 2

WE-CNSW HF Z @ 100 MHz: IR: RDC: Frequency Range: Number of Windings:

60 ~ 120 Ω 280 ~ 600 mA 0.22 ~ 0.30 Ω 10 ~ 10000 MHz 2

WE-SLM L: IR: RDC: Frequency Range: Number of Windings:

11 ~ 470 µH 300 ~ 400 mA 0.18 ~ 0.58 Ω 0.01 ~ 600 MHz 2

WE-SL1 L: IR: RDC: Frequency Range: Number of Windings:

10 ~ 330 µH 300 mA 0.16 ~ 0.3 Ω 0.01 ~ 600 MHz 2

WE-SL2 L: IR: RDC: Frequency Range: Number of Windings:

278

10 ~ 20000 µH 0.2 ~ 1.6 A 0.08 ~ 2.6 Ω 0.01 ~ 600 MHz 2

WE-SL3 L: IR: RDC: Frequency Range: Number of Windings:

20 ~ 100 µH 450 ~ 700 mA 0.14 ~ 0.45 Ω 9 ~ 600 MHz 2~3

WE-SL5 L: IR: RDC: Frequency Range: Number of Windings:

120 ~ 4700 µH 350 ~ 2500 mA 0.025 ~ 0.72 Ω 9 ~ 600 MHz 2

WE-SL5 HC L: IR: RDC: Frequency Range: Number of Windings:

5 ~ 30 μH 1.2 ~ 5 A 0.0055 ~ 0.06 Ω 1.4 ~ 300 MHz 2

WE-SL L: IR: RDC: Frequency Range: Number of Windings: 

35 ~ 4700 µH 0.2 ~ 2.7 A 0.035 ~ 0.85 Ω 0.01 ~ 600 MHz 2~4

WE-SCC L: IR: RDC: Frequency Range: Number of Windings:

1 ~ 1000 µH 150~ 4750 mA 0.01~ 4.30 Ω 0.1 ~ 300 MHz 2

WE-UCF L: IR: RDC: Frequency Range: Number of Windings:

0.013 ~ 100 mH 0.15 ~ 10 A 0.0027 ~ 8.5 Ω 0.01 ~ 400 MHz 2~4

WE-CMB L: IR: RDC: Frequency Range: Number of Windings:

0.5 ~ 39 mH 0.3 ~ 35 A 1.7 ~ 3000 mΩ 0.1 ~ 100 MHz 2

WE-CMBNC L: IR: RDC: Frequency Range: Number of Windings:

0.4 ~ 190 mH 0.9 ~ 32 A 1.1 ~ 430 mΩ 0.001 ~ 300 MHz 2

WE-CMB HC L: IR: RDC: Frequency Range:

0.175 ~ 0.7 mH 5 ~ 10 mA 2.7 ~ 13 Ω 0.1 ~ 100 MHz

WE-CMB HV L: IR: RDC: Frequency Range: Number of Windings:

0.7 ~ 4.7 mH 6.8 ~ 21.5 A 3.5 ~ 44 mΩ 0.005 ~ 5 MHz 2

WE-FCL L: IR: RDC: Frequency Range: Number of Windings:

3.9 ~ 100 mH 1.25 ~ 6.0 A 50 ~ 900 Ω 0.01 ~ 100 MHz 2

WE-CMB NiZn L: IR: RDC: Frequency Range: Number of Windings:

14 ~ 110 µH 1.5 ~ 10 A 2.7 ~ 80 mΩ 0.1 ~ 100 MHz 2

WE-LPCC L: IR: RDC: Frequency Range: Number of Windings:

120 - 450 µH 10 ~ 23.5 A 1.4 ~ 9.6 mΩ 0.01 ~ 100 MHz 2

WE-ExB L: IR: RDC: Frequency Range: Number of Windings:

47 ~ 1000 µH 4.5 ~ 15 A 4.6 ~ 42 mΩ 0.1 ~ 200 MHz 2

WE-TPB L: IR: RDC: Frequency Range: Number of Windings:

0.52 ~ 12 mH 6 ~ 24 A 3 ~ 65 mΩ 0.01 ~ 30 MHz 3

WE-CMBH L: IR: RDC: Frequency Range: Number of Windings:

1 ~ 7 mH 3.5 ~ 10 A 12.5 ~ 80 mΩ 0.1 ~ 100 MHz 2

WE-TPB HV L: IR: RDC: Frequency Range: Number of Windings:

0.2 ~ 208 mH 7.2 ~ 46 A 1.6 ~ 85 mΩ 0.005 ~ 5 MHz 3

WE-LF L: IR: RDC: Frequency Range: Number of Windings:

0.4 ~ 50 mH 0.3 ~ 6 A 0.02 ~ 2.6 Ω 0.1 ~ 100 MHz 2

WE-LF SMD L: IR: RDC: Frequency Range: Number of Windings:

0.7 ~ 47 mH 0.3 ~ 5.25 A 0.03 ~ 2.6 Ω 0.1 ~ 100 MHz 2

WE-TFC L: IR: RDC: Frequency Range: Number of Windings:

1.8 ~ 25 mH 0.25 ~ 1.0 A 0.31 ~ 3.60 Ω 0.001 ~ 100 MHz 2

WE-TFCH L: IR: RDC: Frequency Range:

WE-FC L: IR: RDC: Frequency Range: Number of Windings:

1.8 ~ 25 mH 0.25 ~ 1.0 A 0.31 ~ 3.60 Ω 0.01 ~ 100 MHz

ESD and Surge Protection WE-TVS Operating Voltage: Ctyp: Channels: Uni/Bidirectional, Rail-to-Rail

1.2 ~ 20 VDC 0.18 ~ 830 pF 1 ~ 8 (+VDD)

WE-VE/WE-VEA Operating Voltage: 5 ~ 24 VDC Ctyp: 0.2 ~ 120 pF 0201 ~ 0805/0508 ~ 0612 Size:

WE-TVSP Operating Voltage: Imax: VClamp: Size:

5 ~ 100 VDC 2.5 ~ 326.1 A 9.2 ~ 162 V SMAJ, SMBJ, SMCJ, SMDJ

0.82 ~ 33 mH 1.25 ~ 5 A 0.065 ~ 2.5 mΩ 0.001 ~ 100 MHz 2

279

II Components ESD and Surge Protection WE-VS Operating Voltage: Imax: Wmax: Ctyp: Size: WE-VD Operating Voltage: Imax: Wmax: Diameters: Reference for cURus/CQC/VDE

5.5 ~ 56 VDC 10 ~ 200 A 0.02 ~ 1.1 J 70 ~ 3600 pF 0402 ~ 1206

18 ~ 1465 VDC 0.1 ~ 10 kA 0.7 ~ 496 J 5 ~ 20 mm

Power Magnetics Single Coil Power Inductors WE-PMI L: IR: RDC: Frequency Range:

280

0.11 ~ 10 μH 450 ~ 4000 mA 7 ~ 500 mΩ 0.1 ~ 100 MHz

WE-PMMI L: IR: RDC: Frequency Range:

0.025 ~ 1.5 µH 1600 ~ 7000 mA 9 ~ 100 mΩ 0.1 ~ 100 MHz

WE-PMCI L: IR: RDC: Frequency Range:

0.24 ~ 2.2 µH 1200 ~ 3600 mA 23 ~ 195 mΩ 0.05 ~ 6 MHz

WE-GF L: IR: RDC: Frequency Range:

0.1 ~ 1000 µH 30 ~ 450 mA 0.32 ~ 50 Ω 0.1 ~ 100 MHz

WE-GFH L: IR: RDC: Frequency Range:

1.0 ~ 220 µH 75 ~ 1600 A 81 ~ 9126 mΩ 0.1 ~ 100 MHz

WE-LQ L: IR: RDC: Frequency Range:

1 ~ 2200 µH 0.04 ~ 1.8 A 0.08 ~ 63 Ω 10 ~ 1000 MHz

WE-LQS L: IR: RDC: Frequency Range:

0.16 ~ 10000 µH 0.13 ~ 6.85 A 6 ~ 22800 mΩ 0.01 ~ 10 MHz

WE-LQSH L: IR: RDC: Frequency Range:

0.47 ~ 10 μH 0.58 ~ 6.4 A 18 ~ 816 mΩ 0.05 ~ 6 MHz

WE-LQFS L: IR: RDC: Frequency Range:

1.0 ~ 470 μH 0.26 ~ 4.47 A 21 ~ 2803 mΩ 0.1 ~ 10 MHz

WE-MAPI L: IR: RDC: Frequency Range:

0.33 ~ 47 µH 0.39 ~ 9.6 A 7.2 ~ 2300 mΩ 0.05 ~ 6 MHz

WE-TPC L: IR: RDC: Frequency Range:

0.056 ~ 1500 µH 0.08 ~ 8.5 mA 0.0035 ~ 9 Ω 0.1 ~ 10 MHz

WE-SPC L: IR: RDC: Frequency Range:

0.22 ~ 100 µH 0.40 ~ 5.30 A 0.014 ~ 1.133 Ω 0.01 ~ 10 MHz

WE-PD L: IR: RDC: Frequency Range:

0.47 ~ 1500 µH 0.2 ~ 23.5 A 0.003 ~ 9.44 Ω 0.1 ~ 10 MHz

WE-PDF L: IR: RDC: Frequency Range:

0.22 ~ 27 μH 4.3 ~ 19 A 1.95 ~ 42.5 mΩ 0.1 ~ 10 MHz

WE-PD2SR L: IR: RDC: Frequency Range:

1.2 ~ 220 μH 0.67 ~ 4.85 A 8.5 ~ 876 mΩ 0.1 ~ 10 MHz

WE-XHMI L: IR: RDC: Frequency Range:

0.18 ~ 33 μH 4.7 ~ 20.0 A 1.32 ~ 31.0 mΩ 0.05 ~ 5 MHz

WE-PD2 L: IR: RDC: Frequency Range:

0.12 ~ 2200 µH 0.18 ~ 10 A 0.004 ~ 5.3 Ω 0.1 ~ 10 MHz

WE-LHMI L: IR: RDC: Frequency Range:

0.1 ~ 100 µH 1 ~ 32.5 A 0.60 ~ 500 mΩ 0.05 ~ 5 MHz

WE-PD3 L: IR: RDC: Frequency Range:

1 ~ 1000 µH 0.19 ~ 3.9 A 0.027 ~ 3.2 Ω 0.1 ~ 10 MHz

WE-FAMI L: IR: RDC: Frequency Range:

3.0 ~ 22.0 µH 3.7 ~ 14.5 A 3.8 ~ 36 mΩ 0.01 ~ 10 MHz

WE-PD4 L: IR: RDC: Frequency Range:

0.47 ~ 10000 µH 0.07 ~ 18 A 0.002 ~ 39 Ω 0.1 ~ 10 MHz

WE-TI L: IR: RDC: Frequency Range:

1 ~ 68000 µH 0.05 ~ 9 A 0.006 ~ 125 Ω 0.1 ~ 10 MHz

WE-HCI L: IR: RDC: Frequency Range:

0.13 ~ 82 µH 3.5 ~ 41.5 A 0.35 ~ 34 mΩ 0.1 ~ 3 MHz

WE-TIF L: IR: RDC: Frequency Range:

100 ~ 10000 µH 0.21 ~ 2.6 A 90 ~ 10700 mΩ 0.1 ~ 10 MHz

WE-HCC L: IR: RDC: Frequency Range:

0.22 ~ 10 μH 4.4 ~ 27 A 1.5 ~ 41 mΩ 0.1 ~ 3 MHz

WE-TIS L: IR: RDC: Frequency Range:

1.3 ~ 6800 µH 0.05 ~ 8.5 A 0.007 ~ 40 Ω 0.1 ~ 10 MHz

WE-HCF L: IR: RDC: Frequency Range:

0.7 ~ 680 μH 12 ~ 36 A 0.83 ~118.3 mΩ 0.1 ~ 3 MHz

WE-SI L: IR: RDC: Frequency Range:

12 ~ 1619 μH 0.5 ~ 5 A 0.008 ~ 0.7 Ω 0.01 ~ 0.1 MHz

WE-HCFT L: IR: RDC: Frequency Range:

01.5 ~ 65 μH 17.2 ~ 75 A 0.4 ~ 13.13 mΩ 0.1 ~ 3 MHz

WE-PD HV L: IR: RDC: Frequency Range:

220 ~ 3300 μH 0.24 ~ 1.30 A 0.30 ~ 6.5 Ω 0.1 ~ 10 MHz

WE-HCM L: IR: RDC: Frequency Range:

0.072 ~ 0.47 μH 24 ~ 65 A 0.15 ~ 0.37 mΩ 0.1 ~ 3 MHz

281

II Components Single Coil Power Inductors WE-PD2 HV L: IR: RDC: Frequency Range:

WE-TI HV L: IR: RDC: Frequency Range:

330 ~ 2200 μH 0.15 ~ 0.43 A 1.0 ~ 7.2 Ω 0.1 ~ 10 MHz

220 ~ 5600 µH 0.18 ~ 0.9 A 0.6 ~ 12 Ω 0.05 ~ 1 MHz

PFC Chokes WE-PFC L: IR: RDC1: RDC2:

150 ~ 1800 μH 0.3 ~ 3.0 A 78 ~ 1550 mΩ 140 ~ 1200 mΩ

Dual Coil Power Inductors

282

WE-EHPI L: IR: RDC: Frequency Range:

7.5 ~ 75000 µH 0.0035 ~ 2.29 A 1.5 ~ 1.9 mΩ 0.001 ~ 0.3 MHz

WE-TDC L: IR: RDC: Frequency Range:

0.33 ~ 22 µH 0.7 ~ 4.5 A 0.0145 ~ 0.48 Ω 0.01 ~ 10 MHz

WE-DD L: IR: RDC: Frequency Range:

1.3 ~ 470 µH 0.3 ~ 8.6 A 0.011 ~ 1.73 Ω 0.1 ~ 10 MHz

WE-DCT L: IR: RDC: Frequency Range:

0.091 ~ 100 µH 1.1 ~ 14.5 A 3.5 ~ 290 mΩ 0.05 ~ 3 MHz

WE-CFWI L: IR: RDC: Frequency Range:

0.8 ~ 4.4 µH 12.0 ~ 28 A 1.6 ~ 9.6 mΩ 0.1 ~ 3 MHz

WE-DPC L: IR: RDC: Frequency Range:



1 ~ 47 µH 0.9 ~ 4.5 A 25 ~ 350 mΩ 0.1 ~ 10 MHz

WE-MTCI L: IR: RDC: Frequency Range:

10 ~ 297 µH 0.45 ~ 0.95 A 349 ~ 5200 mΩ 0.1 ~ 3 MHz

WE-DPC HV L: IR: RDC: Frequency Range:

1 ~ 47 µH 0.6 ~ 2.9 A 32 ~ 1.200 mΩ 0.1 ~ 10 MHz

WE-CPIB HV L: IR: RDC: Frequency Range:

4.7 ~ 47 μH 0.55 ~ 1.45 A 105 ~ 1.000 mΩ 0.1 ~ 10 MHz

WE-TDC HV L: IR: RDC: Frequency Range:

5.6 ~ 33 μH 0.75 ~ 1.4 A 190 ~ 700 mΩ 0.1 ~ 10 MHz

WE-MCRI L: IR: RDC: Frequency Range:

1 ~ 47 µH 2.3 ~ 17 A 4.5 ~ 240 mΩ 0.1 ~ 3 MHz

Wireless Power Transmission WE-WPCC Wireless Power Transmitter Coil 2.8 ~ 24 µH L: 30 ~ 220 Q: 2.0 ~ 18 A IR: 10 ~ 255 mΩ RDC:

WE-WPCC Wireless Power Array L: 6.4 ~ 12.5 µH µHQ: 100 ~ 145 8.0 ~ 10.0 A IR: 38 ~ 56 mΩ RDC:

WE-WPCC Wireless Power Receiver Coil L: 1.4 ~ 47.0 µH Q: 10 ~ 50 0.40 ~ 5.0 A IR: 0.08 ~ 1200 Ω RDC:

WE-PoEH - PoE and PoE+ powered devices - Flyback or Forward Transformer - d esigned for 12V, 24V or 48V input of Switching Mode Power Supply

WE-FB for LT3573, LT3751, LT3574, LT3575, LT3748

WE-UOST Ui: UO1: IO1:

85 ~ 265 Vac 5 ~ 24 V 0.56 ~ 3.0 A

WE-LLCR Ui: Uo: P:

360 ~ 400 Vdc 12, 24 or 48 Vdc 150, 200 or 250W

WE-UNIT Ui: UO1: IO1:

85 ~ 265 Vac 5 ~ 24 V 0.13 ~ 2.0 A

WE-GDT L: RDC1: RDC2: RDC3:

260 ~ 650 µH 520 ~ 1200 mΩ 150 ~ 600 mΩ 170 ~ 600 mΩ

WE-GDTI L: RDC1: RDC2: RDC3:

735 ~ 1800 µH 1000 ~ 1600 mΩ 600 ~ 1300 mΩ 650 ~ 1300 mΩ

Power Transformers WE-FLEX suitable for all switch mode power supply topologies like: Buck-Converter, Boost-Converter, SEPIC-Converter, Flyback-Converter, Forward-Converter and Push-Pull-Converter

WE-FLEX+ suitable for all switch mode power supply topologies like: Buck-Converter, Boost-Converter, SEPIC-Converter, Flyback-Converter, Forward-Converter and Push-Pull-Converter

WE-FLEX HV suitable for all switch mode power supply topologies like: Buck-Converter, Boost-Converter, SEPIC-Converter, Flyback-Converter, Forward-Converter and Push-Pull-Converter

WE-PoE suitable for Power over Ethernet ICs

WE-CST for Switch Mode Power Supply and AC current detection

WE-PoE+ Compliant with the 30W PoE+ objectives of IEEE802.3at Suitable for PoE+ powered devices

283

II Components Signal & Communications AS-Interface Inductor WE-ASI L: IR: RDC:

3.0 ~ 18.0 mH 0.08 ~ 0.24 A 10.0 ~ 72.0 Ω

WE-LAN 10G Speed: Ports: Temp. Range: PoE:

WE-LAN AQ Speed: Ports: Temp. Range:

WE-RJ45 LAN Speed: Ports: Temp. Range: PoE:

WE-KI L (±2% or ±5%): Q: SRF: IR: Sizes:

1 ~ 390 nH 18 ~ 46 880 ~ 16000 MHz 170 ~ 2300 mA 0402, 0603

10/100/1000 MBit/s 1~4 -40 to +125°C 350 ~1500 mA

WE-RFI L (±5%): Q: SRF: IR: Sizes:

0.47 ~ 47 µH 15 ~ 45 17 ~ 375 MHz 45 ~ 500 mA 0805, 1008

10.000 MBit/s 1 -40 to +85°C 350 ~ 1500 mA

WE-RFH L (±5%): Q: SRF: IR: Sizes:

0.56 ~ 10 µH 15 ~ 45 40 ~ 415 MHz 300 ~ 760 mA 1008

1.000 MBit/s 1~4 -40 to +85°C

10 ~ 10.000 MBit/s 1~4 -40 to +85°C 350 mA ~ 600 mA

WE-EPLE USB-A connector with integrated circuit protection device and EMI noise reduction

WE-TCI L (±0.1nH or 2%): Q: SRF: IR: Sizes:

1 ~ 22 nH 8 ~ 13 2800 ~ 9000 MHz 90 ~ 700 mA 0201, 0402

WE-MK L (±5%): Q: SRF: IR: Sizes:

1 ~ 470 nH 8 ~ 20 250 ~ 17000 MHz 100 ~ 600 mA 0201, 0402, 0603

WE-CAIR L (±5%): Q: SRF: IR: Sizes:

1.65 ~ 120 nH 100 ~ 140 1.1 ~ 12.5 GHz 1.5 ~ 4 A 1322, 1340, 3136, 3168, 4248

WE-AC HC L (±20%): Qtyp: SRFtyp: IR: Sizes:

284

1 ~ 1500 nH 16 ~ 60 200 ~ 12500 MHz 100 ~ 1360 mA 0402, 0603, 0805, 1008

WE-KI HC L (±2%): Q: SRF: IR: Sizes:

LAN Transformers WE-LAN Speed: Ports: Temp. Range: PoE:

RF Inductors

22 ~ 146 nH 163 ~ 280 332 ~ 867 MHz 19 ~ 40 A 1010, 1212

LTCC Components

Automotive Standard Products

WE-LPF Low-Pass Filter Frequency Range: 902 ~ 5875 MHz 0603, 0805 Sizes: Wireless Communication Systems like Bluetooth, WiFi 2.4 & 5.0 GHz, ZigBee … Low insertion loss in passband and high attenuation in stopband

WE-BPF Band-Pass Filter Frequency Range: 2400 ~ 5920 MHz 0805, 1008 Sizes: Wireless Communication Systems like Bluetooth, WiFi 2.4 & 5.0 GHz, ZigBee … Low insertion loss in passband and high attenuation in stopband WE-BAL Balun Frequency Range: 2400 ~ 5875 MHz Sizes: 0603, 0805 Wireless Communication Systems like Bluetooth, WiFi 2.4 & 5.0 GHz, ZigBee … Low loss SMD Balun with balanced impedance of 50-200 ohms WE-MCA Multilayer Chip Antenna Frequency Range: 423 ~ 5875 MHz Wireless Communication Systems like GSM 900, ISM 868/2400, GPS, Bluetooth, WiFi 2.4 & 5.0 GHz, ZigBee …

Automotive Standard Products

WE-TEFA Z @ 25 MHz 1 turn: Z @ 100 MHz 1 turn: Cable diameter: Frequency Range:

22 ~ 83 40 ~ 131 7.5 ~ 15,5 1 ~ 1000 MHz

WE-PDA L: IR: RDC: Frequency Range:

1 ~ 1000 µH 0.23 ~ 9.20 A 8.5 ~ 7200 mΩ 0.1 ~ 10 MHz

WE-PD2SA L: IR: RDC: Frequency Range:

1.2 ~ 220 μH 0.67 ~ 4.85 A 13 ~ 876 mΩ 0.1 ~ 10 MHz

WE-PD2A L: IR: RDC: Frequency Range:

0.33 ~ 470 µH 0.36 ~ 10.8 A 0.004 ~ 1960 Ω 0.1 ~ 10 MHz

WE-MAIA L: IR: RDC: Frequency Range:

0.33 ~ 47.0 µH 0.39 ~ 9.8 A 7.2 ~ 2300 mΩ 0.05 ~ 6 MHz

WE-CHSA L: IR: RDC: Frequency Range:

0.33 ~ 12.0 μH 5.0 ~ 26.0 A 1.34 ~ 28.9 mΩ 0.1 ~ 10 MHz

WE-CBA Z @ 100 MHz: IR: RDC: Frequency Range:

20 ~ 2200 Ω 10 ~ 5000 mA 0.008 ~ 1 Ω 6 ~ 2000 MHz

WE-LHCA L: IR: RDC: Frequency Range:

10.22 ~ 100 μH 1.8 ~ 21.2 A 1 ~ 450 mΩ 0.05 ~ 5 MHz

WE-AEFA Z @ 25 MHz 1 turn: Z @ 100 MHz 1 turn: Cable diameter: Frequency Range:

60 ~ 325 Ω 84 ~ 460 Ω 3.3 ~ 15.0 mm 1 ~ 500 MHz

WE-RCIT L: IR: RDC: Frequency Range:

2.0 ~ 10 μH 2.5 ~ 15 A 1.7 ~ 33 mΩ 0.01 ~ 90 MHz

285

II Components 2 EMC Components 2.1 Various forms of ferrites Ferrite used primarily for EMC comes in the form of beads, cores, rods, sleeves or toroids (see Figure 2.1). EMC ferrite

Fig. 2.1: Various forms of ferrite All other forms have the same technical functionality; they only differ in their application. Ferrite cores or rods are more sensitive to external interference fields and themselves have a larger magnetic stray field. Coils with toroidal cores however, have a lower stray field and are also less sensitive to interference (Figure 2.2). Pot cores have the most effective shielding and also the lowest stray field. As a result of defined internal routing of the magnetic flux, these coils are largely free from external stray fields. The coil with a ferrite core is more sensitive to magnetic field inter­ference than the air coil. It is often necessary to shield coils to spatially ­restrict stray fields.

Fig. 2.2: Field line orientation for a rod and toroidal coil

286

2.2 WE-CBF SMD Ferrites

Fig. 2.3: SMD ferrite WE-CBF SMD ferrites for EMC applications are constructed in multilayer technology and use a nickel-zinc as ferrite material.

SMD ferrites

The special feature of this material composition is that the real (loss) component essentially determines the impedance above 50 MHz. This means it is a filter component, which absorbs, i.e. converts to heat, a broad interference spectrum without being ground referenced. Würth Elektronik distinguishes their ferrites in High Speed and Wide Band ferrites. The permeability of the Wide Band types is µi = 200. High Speed SMD Ferrites have smaller permeabilities. Due to this fact High Speed types do not influence high ­frequent signals up to 1 MHz.

287

II Components 10000

Impedance [Ω]

1000

100

10

HighSpeed: 742 792 716 1 WideBand: 742 792 729 1

1 1

10

100

1000

10000

Frequency [MHz]

Fig. 2.4: HighSpeed vs. Wideband SMD ferrites are available in a wide spectrum of sizes from 0402–1812 with current carrying capability up to 6 A as well as impedances up to 2000 W at 100 MHz. Chip bead

SMD ferrites are also often known as chip beads or chip bead inductors or impeders. It is really an inductor with high losses for frequencies above 10 MHz. To prevent confusion, the term SMD ferrite has shown itself to be favorable.

288

Datasheet specifications: The impedance of the SMD ferrite is clearly described with the following parameters (Figure 2.5):

Fig. 2.5: Impedance curve for the SMD ferrite 742 792 13 Below the ferromagnetic resonant frequency, the impedance of the component is ­essentially determined by the inductive component. In the range between 50 MHz– 100 MHz, the situation is reversed (see Figure 2.5 SMD ferrite 742 792 13); the “R” loss component dominates with increasing frequency and the inductive component tends towards zero. In this example, the inductive component is around 1.5 µH (f = 10 MHz) and the impedance Z attains a value of 600 W at f = 100 MHz.

Resonant frequency

DC resistance (DCR):

DC resistance

The DC resistance results from the internal length and layer thickness of the multilayer meander in the SMD ferrite. This is measured at room temperature. The maximum expected DC resistance is determined from a production lot and this worse case scenario is published as in the data sheet specifications. Values in a range between a few milliohms up to even 1 W (depending on the type) are obtained. Because of their very low DC resistance, SMD ferrites are significantly superior to inductors of the same construction and size and largely avoid problems of voltage drop or potential differences.

289

II Components Current rating: Maximum DC current Self-heating

An additional parameter is the maximum DC current the SMD ferrite can withstand. This is defined for SMD ferrites such that at the current specified, the self-heating in the case of: • signal line ferrites with small rated currents remains < +20 °C • high current SMD ferrites remains < +40 °C Precautions: SMD ferrites should not be operated with a higher current (either peak v­ alue of the AC current or permanent DC current) than the maximum current ratings specified in the datasheet. A higher current would cause overheating and under certain circumstances would reduce the service life of the SMD ferrite. If SMD ferrites are in circuits with surge currents (e.g. switch-on current peak or DC filter circuits with high capacitance values), the WE-MPSB series of multilayer power suppression beads have defined peak current capability. The on line tool REDEXPERT shows the effects of DC bias on impedance and reactance. Take care when using SMD ferrites for high current ratings > 1 A and ambient temperatures of above +85 °C that the current rating beyond +85 °C ambient temperature must be reduced (derating). The characteristic curve from Figure 2.6 (derating curve WE-CBF) shows e.g. that at an ambient temperature of +100 °C, the maximum current rating may then only be 60% of the datasheet value without overstressing the component. The lower curve is intended for very critical applications; this shows that the maximum current may then only be 50% of the datasheet value.

Fig. 2.6: Derating curve WE-CBF

290

SMD ferrite applications: The classic applications for this miniature ferrite are in: • Data line filters • Supply voltage decoupling • Ground decoupling For high frequency applications like HDD and fast bus signals Würth Elektronik offers the SMD ferrites WE-CBF HF. The “High Frequency” SMD ferrites have a modified internal layout which increases effective suppression frequency range. Consequently the impedance at 1 GHz is up to 3 times higher.

10000,00

Impedance [Ω]

1000,00

100,00

742 862 160 WE-CBF HF 742 792 65 WE-CBF

10,00

1,00 1

10

100

1000

10000

Frequency [MHz]

Fig. 2.7: SMD ferrites WE-CBF vs. WE-CBF HF The SMD ferrite needs no ground reference for its filter or absorption function! ­Owing to the “sensitivity” of the ferrite material, even small signal ­currents suffice for the filter effect. The filter effect is raised by: • The specific use of bypass capacitors against ground (selection dependent on the interference spectrum and the damping frequency range → see Chapter III/1 Filter circuits • Low impedance source and drain

Filter effect Bypass capacitor

Source and sink impedances

291

II Components Typical specifications Size 0402: Order Code

Impedance (Ω) @ 100 MHz

DCR (Ω)

Rated ­current max. (mA)

max. Impedance

Type

742 792 731 1

120

0.09

1200

 200 Ω @ 450 MHz

High Current

742 792 729 1

600

0.60

300

 900 Ω @ 250 MHz

Wide Band

742 792 796

1000

1.50

200

1200 Ω @ 200 MHz

Wide Band

Impedance (Ω) @ 100 MHz

DCR (Ω)

Rated ­current max. (mA)

max. Impedance

Type

Size 0603: Order Code

742 792 60

  40

0.15

400

  60 Ω @ 1000 MHz

High Speed

742 792 66

1000

0.60

200

1350 Ω @   140 MHz

Wide Band

742 792 693

2200

0.80

50

2250 Ω @   110 MHz

Wide Band

Order Code

Impedance (Ω) @ 100 MHz

DCR (Ω)

Rated ­current max. (mA)

max. Impedance

Type

742 792 063

  60

0.02

3000

  90 Ω @ 500 MHz

High Current

742 792 040

 600

0.15

 200

 800 Ω @ 200 MHz

High Current

742 792 093

2200

0.60

 200

3000 Ω @   80 MHz

Wide Band

Order Code

Impedance (Ω) @ 100 MHz

DCR (Ω)

Rated ­current max. (mA)

max. Impedance

Type

742 792 15

80

0.03

3000

 160 Ω @ 550 MHz

High Current

Size 0805:

Size 1206:

292

Size 0603 HF: Order Code

742 863 122

Impedance (Ω) 100 MHz

1 GHz

max.

DCR (Ω)

Rated ­current max. (mA)

Type

220

250

500 Ω @ 600 MHz

0.25

600

High Current

742 862 160

600

850

2200 Ω @ 550 MHz

1.50

100

High Speed

742 861 210

1000

1100

1000 Ω @ 450 MHz

1.80

50

Wide Band

Tab. 2.1: Typical specifications of various SMD ferrites 2.2.1 Design guidelines The design engineer should add the appropriate solder pads for chip beads at the prototype stage. This allows preselection of different ferrites ­under laboratory conditions. The following nomogram should be of some help to select the ferrite impedance for the required damping. Würth Elektronik ­offers an extensive set of samples with which the desired filter circuit can be quickly achieved.

Fig. 2.8: Nomogram to determine the required SMD ferrite The insertion attenuation is defined as the logarithmic magnitude of the r­ atio of interference amplitude of the undamped system, to the damped system with impedance ZF. This approach leads to the formula for the insertion loss (Figure 2.9):

Insertion loss

293

II Components

Fig. 2.9: Four pole equivalent circuit to calculate the insertion loss



(2.1)



ZA = ZB = system impedance; ZF = ferrite impedance Nomogram

Experienced data

The nomogram in Figure 2.8 and the above formula can be used in practice to quickly find the required ferrite, which may possibly be further optimized by measurement in an EMC laboratory as the system impedances (ZA and ZB) in the HF range cannot be found by simple calculation. Never­theless, v­ alues drawn from previous experience can be used in circuit ­board design: • Ground surfaces impedances from impedances from • Supply voltage lines impedances from • Video, clock and data lines • Long data lines impedances from

1 W to 2 W 10 W to 20 W 50 W to 90 W 90 W to 150 W and higher

Example calculations: Example 1: SMD ferrite 742 792 66; impedance ZF at f = 100 MHz : 1000 W Type 0603: DCR ≤ 0.6 W; I ≤ 200 mA deployed as a data line filter: system impedance 300 W





According to the above formula, the insertion attenuation at 100 MHz: 8.5 dB

294

(2.2)

Example 2: SMD ferrite 742 792 15; impedance ZF at f = 100 MHz: 80 W Type 1206: DCR ≤ 30 mW; I ≤ 3000 mA deployed as a supply voltage decoupler: system impedance 10 W





(2.3)

According to the above formula, the insertion attenuation at 100 MHz: 14 dB Example 3: Exceeding the limit value curve at 100 MHz by 3 dB; Necessary safety margin to the limit value curve = 5 dB; → required insertion loss = 8 dB System impedance = 50 W the system impedance of 50 W is extracted from the nomogram Figure 2.8: ZF ≈ 180 W → selected: 220 W e.g. size 0603 Würth Elektronik order code 742 792 63: 220 W: 0.3 W DCR; IMAX = 500 mA How much ferrite impedance is required for interference suppression? How do you determine which ferrite is best selected? Step 1: Analyze the EMC measurement graph The measurement graph from the EMC laboratory or spectrum analyzer s­ erves as the basis to determine the required ferrite impedance. This allows the insertion loss to be defined: This is determined from where the limit value on the EMC laboratory measurement graph is exceeded, with an additional a safety reserve (3…6 dB). If the limit value is exceeded at several frequencies, the highest interference level should be used for initial orientation. Step 2: Estimate the interference source/sink impedance The analysis in the EMC test laboratory or with spectrum analyzer/sniffer probes should clarify which PCB track propagates interference. If the cause is known, the impedances ZA and ZB are defined taking into consideration the items under ‘Experienced data’ on the previous page. 295

II Components Step 3: Select ferrite impedance

Supply voltage (suppression)

Example: Interference maximum at 200 MHz, Interference source → supply voltage system → definition ZA = ZB = 10 W Required signal-to-noise ratio = 20 dB → reading off the ferrite impedance at 200 MHz from the nomogram.

Fig. 2.10: Determination of ferrite impedance: Example Supply voltage system Test results and optimization options Developers often use “too much” ferrite impedance in the case of inter­ference and then obtain partly inconsistent measurement results in the r­ epeated EMC test. Results such as “The ferrite does not work!” or “There are new interference frequencies, the ferrite was the wrong selection” are frequently voiced a­ rguments against suppression with EMC ferrites. The cause of this phenomenon is not simply the EMC ferrite with its parasitic effects (see chapter I/2 Equivalent circuits and simulation models), but the interaction of: • the complete circuit board with its parasitic elements (1 mm conductor track ~ 1 nH distributed inductance, 1 via ~ 0.5 nH!) • the filter capacitors or blocking capacitors with their HF properties as a result of their parasitic inductance and ESR (equivalent circuit!) • and of course the placement of the ferrite (close to the interference source!)/­ conductor track layout.

296

It should always be attempted to use as little ferrite impedance as possible to keep the parasitic elements as low as possible. Optimization can be achieved through: 1) Simulation → Simulation model with LTspice 2) Evaluation of the EMC measurement results and use of the nomogram in the ­selection of the EMC ferrite Once the ferrite determined from the nonogram has been deployed, the e­ ffectiveness of the measure must be proven by repeated testing. Only 4 cases can occur here: a) Insertion loss achieved → this proves that the source/sink impedances really do correspond to the ­assumption previously made; the selected EMC ferrite can be designed in. b) Lower insertion loss achieved

Fig. 2.11: Evaluation of measurement result – Supply voltage system is of high impedance in the interference frequency range

297

II Components c) Higher insertion loss attained

Fig. 2.12: Evaluation of measurement result: Supply voltage system is of low ­impedance in the interference frequency range d) No effect The impedances ZA = ZB are of far higher impedance in the interference frequency range than originally assumed – the filter circuit must be expanded in this case to ­appropriately “focus on” the interference current and lead it to ground.

298

Filter circuit topology

L-C-filter

p-filter

T-filter

Fig. 2.13: Suitable filter circuits for various impedances ZA and ZB A further cause for the unaltered interference properties despite the filter measures implemented is often to be found in unidentified coupling of the filter. Example: • Capacitive coupling Very quickly proven with burst tests! If devices fail this test, the cause often lies in underestimated capacitive coupling. The task is then to locate the sensitive areas of the circuit (apply interference to the leads individually) and to specifically “harden” with EMC measures. An overview and a basic sample calculation of the possible impact of coupled burst interference in the device, despite assumed decoupling with isolating elements such as optocouplers etc.

Burst test

299

II Components

Coupling ­capacitances

Optocoupler

1 ~ 5 pF

Solid state relay

5 ~ 10 pF

Electromagnetic relay

10 ~ 100 pF

Switch mode power supply/DC-DC converter

up to 1000 pF

Tab. 2.2: Typical coupling capacitances of galvanic isolating elements What interference current is coupled with a burst test at 2 kV and rise time of 2 ns if the coupling capacitance is 1 pF?





(2.4)

This interference current now directly affects the circuit inputs. Dependent on its impedance, a peak current of this nature is sufficient to derail the e­ lectronics. Discrete purchased DC/DC converter modules in particular are not to be underestimated. Despite their transformer decoupling, these often have enormous coupling capacitances. Additional coupling can be caused by: • Inductive coupling • Impedance coupling (often termed galvanic coupling in the past) For corrective measures, we refer to the subsequent chapters in this book.

2.3 WE-PBF, WE-SUKW High Current Beads High Current Beads have due to their construction a very low DCR. Therefore they can carry a higher current compared to multilayer beads. If higher impedance is requested while having also a high current, Würth Elektronik offers a 2.5 turn PCB ferrite the 5-Hole Ferrite Bead WE-SUKW. The bottom side of the ferrite is plated with 100% tin finish and dipped into solder.

300

WE-SUKW

Fig. 2.14: WE-SUKW 5-hole ferrite bead Order Code

A (mm)

B (mm)

C (mm)

D (mm)

E (mm)

F (mm)

Impedance (Ω)

Material

25 MHz

100 MHz

742 751 1

8,0 max.

5,0±0,25

4,6±0,5

5,5±0,3

2,0 max.

0,5 ref.

272

416

5 W 700

742 751 2

11,0 max.

4,65±0,5

5,0±0,5

8,5±0,5

2,0 max.

0,5 ref.

425

580

5 W 700

2.5 Turns

Tab. 2.3: Typical specifications

742 751 1

742 751 2 1000

Impedance (Ω)

Impedance (Ω)

1000

100

100

10

10 1

10

100

Frequency (MHz)

1000

1

10

100

1000

Frequency (MHz)

B

B

Fig. 2.15: Typical performance of 5-Hole Ferrite Beads WE-SUKW The WE-PBF series is a single flat wire through a ferrite and can handle a rated current of 6 A. For a better conductivity the copper wire is plated with a 5 µm tin layer. The WE-PBF can be used for mains line applications and the parts are released for a RMS current of 5 A. The operating temperature range of –55 °C to +125 °C allows nearly ­every lighting application. For higher reliability Würth Elektronik offers a glued solution as well. This part is especially recommendable if the part is used in vibratory applications.

301

II Components WE-PBF

Fig. 2.16: WE-PBF High current flat wire beads Order Code

Order Code Glued version

A (mm)

B (mm)

C (mm)

D (mm)

E (mm)

25 MHz

100 MHz

742 793 0

742 793 90

4.0±0.2

3.0±0.2

2.55±0.15 1.27±0.3 1.45±0.1

28

742 793 1

742 793 91

8.5±0.3

3.0±0.2

2.55±0.15 1.27±0.3 1.45±0.1

58

742 793 2

742 793 92

7.8±0.3

4.75±0.2

3.0±0.15

65

1.27±0.3 1.45±0.1

Impedance (Ω)

DCR max. (mW)

IN (A)

42

0.6

6

91

0.9

6

98

0.9

6

2.5 Turns

Tab. 2.4: Typical specifications

742 793 0

742 793 1

10

1

742 793 2

1000

1000

100

100

Impedance (Ω)

Impedance (Ω)

Impedance (Ω)

100

10

1 1 IF BW 10 kHz START 1 MHz

10

100 POWER 0 dBm

1000 SWP 134.5 msec STOP 1.8 GHz

1 1

10

IF BW 10 kHz START 1 MHz

100 POWER 0 dBm

1000 SWP 134.5 msec STOP 1.8 GHz

1

10

IF BW 10 kHz START 1 MHz

Z

XL

100 POWER 0 dBm

Frequency (MHz)

Frequency (MHz) Z

10

1000 SWP 134.5 msec STOP 1.8 GHz

Frequency (MHz) R

Z

XL

R

Fig. 2.17: Typical characteristics of Flat Wire High Current SMD Ferrite Beads WE-PBF All high current PCB beads can be soldered with the Würth Elektronik standard solder profile.

302

2.4 WE-MPSB SMD multilayer power suppression bead WE-MPSB

Fig. 2.18: WE-MPSB SMD Order Code

Z@ 100 MHz (Ω)

Zmax (Ω)

TC Zmax

IR 1 (mA)

RDC1 typ (mΩ)

RDC1 max. (mΩ)

742 792 280 8

8

25

1930 MHz

9500

2.5

5

742 792 282 60

26

39

515 MHz

6500

5

8

742 792 286 00

60

99

458 MHz

5100

8.5

15

742 792 281 11

110

135

226 MHz

4100

14.5

20

Z @ 100 MHz: Impedance @ 100 MHz; Zmax: Maximum Impedance; TC Zmax: Maximum Impedance (Test cond.); IR 1: Rated Current; RDC1 typ: DC Resistance; RDC1 max.: DC Resistance

Tab. 2.5: Typical specifications of some WE-MPSB SMD line filters The Multilayer Power Suppression Beads (WE-MPSB) series has been developed based on the requirements of circuits that load the multilayer ferrites with temporary peak currents exceeding the rated current. The maximum pulse load capabilities of the multilayer ferrites have been determined by measurement using an in-house test routine that is different from that of the fuses. Pulses in this context are understood as temporary current peaks with a time limitation less than 8 ms. The appropriate approach in the search for a common standard for the measurement of pulse load capacity for SMD ferrites has been found in the definition of the melting integral for fuses. A pulse of 8 ms duration, according to the standard, is applied to the fuse to “give the current time” to heat the fuse for the determination of the I²t value of the fuses. If the fuse withstands this, the current is increased until the increase results in destruction of the fuse. In doing so, a pause of 10 s between the pulses is required to give the component the necessary time for regeneration (cooling down). Würth Elektronik has developed a test routine based on this fuse standard for the multilayer ferrites. Using the 8 ms pulse duration, current, starting from 1 A and in ever increasing amounts is applied to the multilayer ferrite until destruction. The rectangular pulse shape was selected for all tests as this loads the component with the highest possible energy for the pulse duration although it will only very rarely be applicable in practice. 303

II Components Z vs f

Temp vs Ir

50 45 40 35

Impedance (Ohms)

Temperaturanstieg in Grad K

100

30 25 20

15 10

5 0

1 1

10

100

1000

0

0.5

1

WE-CBF 600Ω (742792040)

WE-MPSB 600Ω (74279220601)

1.5

2

2.5

3

Strom in A

Frequency (MHz) WE-CBF 600Ω (742792040)

WE-MPSB 600Ω (74279220601)

Fig. 2.19: Comparison of the impedance and rated current load capability of WE-CBF and WE-MPSB In comparison with existing multilayer structures, the layer structure has been optimized to have higher current load capability using lower resistances. The WE-MPSB Series is thus ideally suitable for use in circuits with pulse currents. Use the on line tool REDEXPERT to see the effects of DC bias on impedance and reactance.

2.5 T hrough Hole Components 2.5.1 WE-UKW 6-Hole Ferrite Bead “VHF suppression choke” 6-hole ferrite bead VHF choke

The 6-hole ferrite bead is a special type of inductor construction. This component is widely used and known under the name “VHF choke”. Its advantage is in its high current loading capacity (3 A) with high impedance (500–1000 W). A summary of specifications is given in Table 2.2.

WE-UKW

Fig. 2.20: WE-UKW 6-hole ferrite bead

304

Order Code

A B ∅ C ∅ D Impedance (mm) (mm) (mm) (mm) (Ω) 25 MHz

100 MHz

Material

Windings

742 750 4

40

10

6

0.48

702

773

3 W 1200

2.5

742 750 43

38

10

6

0.48

920

961

3 W 1200

3

742 752 43

10

3.5

6

0.50

920

961

3 W 1200

3

Tab. 2.6: Parameters of some 6-hole ferrite beads Separation of the individual windings within the coil by ferrite reduces the capacitive coupling. Competent selection of the ferrite material can ensure that the resistive component of the impedance is high, which keeps the quality factor low so resonances cannot occur. Figures 2.21 and 2.22 show the impedance and phase of the 6-hole ferrite 742 750 4. The electrical parameters of the ferrite are shown in the equivalent circuit of Figure 2.23.

Fig. 2.21: Impedance and phase of the 6-hole ferrite bead 742 750 4 in the ­frequency range from 500 kHz to 30 MHz

305

II Components

Fig. 2.22: Impedance and phase of the 6-hole ferrite bead 742 750 4 in the ­frequency range from 0 MHz to 500 MHz

Fig. 2.23: Equivalent circuit and electrical parameters of the 6-hole ferrite bead 742 750 4 The inductance of 31 µH, measured with a conventional LCR measuring bridge, is really rather surprising. The inductance arises from the 2 ½ turns through the ferrite body. However, as the ferrite material causes the decoupling of the windings already mentioned, a high impedance is achieved over the whole frequency range up to 500 MHz, which has no resonant points. This component is mainly suited to power supply applications.

306

2.5.2 WE-MLS Ferrite Bridge

Ferrite bridge

A type of “choke array” should not go unmentioned WE-MLS

Fig. 2.24: WE-MLS choke array

Order Code

Description

A (mm)

B (mm)

C D E F Impedance (mm) (mm) (mm) (mm) (Ω) 25 MHz

Type

100 MHz

742 730 01

Ferrite bridge 3-lines

7.62

5.08

10

5.8

2.54 2.54

212

264

1

742 730 02

Ferrite bridge 4-lines

10.88

5.49

10

3.19 2.54 2.54

209

249

2

742 730 021 Ferrite bridge 4-lines 11.2 max 11.2 max

5

5.0

2.54 7.62

136

170

2

742 730 022 Ferrite bridge 4-lines 11.2 max 11.2 max

8

2.5

2.54 7.62

208

248

2

742 730 023 Ferrite bridge 4-lines 11.2 max 11.2 max

11

5.0

2.54 7.62

292

334

2

Tab. 2.7: Electrical and dimensional parameters of some ferrite bridges

307

II Components

Fig. 2.25: Possible switching options of the ferrite bridge (layout design) The ferrite bridge has three or four chambers depending on the type. A 0.6 mm ­ utual induc­diameter (22 AWG) wire passes through each chamber. The coupling or m tance Lµ of the chambers with one another is different due to the mechanical layout of the wire windings. The mutual inductance Lµ also falls from chamber to chamber, i.e. Lµ is greater between chamber 1 and chamber 2 than between chamber 1 and chamber 3 (Figure 2.26).

Fig. 2.26: Schematic representation of the ferrite bridge

Common mode choke

308

Comparing the damping characteristics of the ferrite bridge in Figure 2.27 with those of current compensated chokes it is apparent that this component can be used as a common mode choke.

Fig. 2.27: Impedance graph for the ferrite bridge in different configurations Advantages of this common mode choke are high current loading capacity up to 4 A and its damping up to 1000 MHz without resonance. The choke is therefore particularly suited for use in power supply applications and in signal transmission in a signal bandwidth < 5 MHz. The impedance of a chamber is represented by curve B in Figure 2.27, curve A is the signal insertion damping, the differential mode impedance.

Differential mode impedance

2.6 Snap ferrites 2.6.1 STAR-TEC, STAR-RING, STAR-CLIP Split ferrite sleeves or ferrite rings (toroids) are often used in radio interference suppression (Figure 2.28). The safety key technology enables the subsequent assembling of the snap ferrites.

309

II Components STAR-TEC

Fig. 2.28: STAR-TEC beads Order Code

A (mm)

B (mm)

C (mm)

Impedance (Ω) (1 Turn)

Material

(2 Turns)

25 MHz

100 MHz

25 MHz

100 MHz

Cable ∅ mm

742 711 42

32.5

18.8

13.2

 98

182

401

709

3.5–5.0

742 711 11

40.5

23.7

18.2

175

320

770

800

3.5–5.0

742 711 12

40.5

23.7

18.2

176

321

773

806

742 711 31

40.5

24.5

21.0

145

246

607

755

742 711 32

40.5

24.5

21.0

141

241

603

755

7.5–8.5

742 712 21

42.2

33.6

29.5

151

270

641

783

8.5–10.5

742 712 22

42.2

33.5

28.8

145

265

638

779

10.5–12.5

4.5–6.0 4 W 620

6.5–7.5

Tab. 2.8: Typical specifications

742 711 31/742 711 32 1000

100

100

Impedance (Ω)

Impedance (Ω)

742 711 42 1000

10

1

1 1

10

100

Frequency (MHz) B

A

Fig. 2.29: Typical performance

310

10

1000

1

10

100

Frequency (MHz) B

A

1000

STAR-RING

Fig. 2.30: STAR-RING Order Code

A (mm)

B (mm)

C (mm)

Impedance (Ω) (1 Turn)

Material

(2 Turns)

25 MHz

100 MHz

25 MHz

100 MHz

Cable ∅ mm

742 714 3

25.5

19.5

21.0

64

106

291

504

4 W 620

 8.0

742 714 4

34.5

17.0

30.0

70

129

175

490

4 W 620

 11.0

Tab. 2.9: Typical specifications

742 714 4 1000

100

100

Impedance (Ω)

Impedance (Ω)

742 714 3 1000

10

1

10

1 1

10

100

Frequency (MHz) B

A

1000

1

10

100

1000

Frequency (MHz) B

A

Fig. 2.31: Typical performance

311

II Components STAR-CLIP

Fig. 2.32: STAR-CLIP mounting Order Code

suitable for

742 771 1

742 711 11 / 742 711 12 / 742 716 33 / 742 717 33 / 742 727 33

742 771 3

742 711 31 / 742 711 32

Tab. 2.10: Different snap ferrites and new fixation of the STAR-TEC snap ferrites AL value

The AL value, the link between inductance and the number of w ­ inding turns, is often specified in the spectrum of ferrite data. The following applies for the AL value in nH:



N: winding turns

This inductance factor is however only an approximation and is valid • only in the coil‘s lower frequency range (far below the resonant f­requency) • not for coil bodies with intense stray fields (rod cores) • only for nearly fully wound coil bodies

312

(2.5)

AL values of some selected Würth Elektronik toroidal cores/ferrite sleeves Order Code

Core material

Calculated [nH/N2]*)

742 701 13

4W620

930

742 701 0

4W620

1170

742 701 707

7W850

1178

742 701 2

4W620

1100

742 701 4

3W800

1800

742 701 04

4W620

510

742 701 15

3W800

1300

742 701 5

4W620

730

742 700 77

7W380

1743

742 700 752

7W850

1885

742 700 53

3W800

3160

742 700 5

4W620

2000

742 700 9

3W800

2790

742 700 790

7W380

1501

742 700 95

4W1500

3780

Tab. 2.11: Typical characteristics of different toroidal ferrites/axial ferrite beads Explanation: *) The calculation is based on the generally applicable formula for AL ­values for ­toroidal cores. The AL value is to be determined for the ferrite ring 742 701 111. The 51 winding turns around the coil body were wound close together so that the coil body is completely wound. The inductance curve against frequency is shown in Figure 2.33.

313

II Components

Fig. 2.33: Inductance and quality factor as a function of frequency for the f­errite ring 742 701 111 (51 turns) The wound ferrite has an inductance of 1.9 mH at 150 kHz. The AL value is calculated as follows:





(2.6)

The AL value rises to 961 at 1 MHz. A low quality factor of approx. 22 at 150 kHz implies a broad resonant maximum. Figure 2.34 shows the impedance-phase curve as a function of frequency. The capacitive characteristic of the coil predominates above around 2 MHz. The resonance is however corres­pondingly broadband as expected from the low quality value.

314

Fig. 2.34: Inductance-phase curve as a function of frequency for the 51 turn ferrite ring 742 701 111 Snap ferrites are often used to damp cable sheath currents in order to r­ educe cable electromagnetic radiation. The frequency range above ­approx. 30 MHz to 500 MHz are of interest in this regard. Ferrites with a very low i­nductive permeability component may be manufactured given the appropriate material selection, assuring broadband applicability. The real and imaginary impedance parts are shown in Figure 2.35 against ­frequency for the snap ferrite 742 711 1.

Cable sheath currents

Snap ferrite

Fig. 2.35: Imaginary and real parts versus frequency for the snap ferrite 742 711 1 315

II Components The rapidly declining imaginary component above approx. 50 MHz is clearly identifiable in the plot. The rapidly increasing real impedance component prevents resonances and ensures high insertion losses. Impedance, phase, inductance and quality factor are plotted against frequency in Figures 2.36 and 2.37. The anticipated low quality factor and low in­ductance are a ­consequence of the high loss component and the high real permeability component of the ferrite material, as previously mentioned.

Fig. 2.36: Impedance and phase of the snap ferrite 742 711 1

Fig. 2.37: Inductance and quality factor of the snap ferrite 742 711 1 316

Practical tips:

Selecting EMC-ferrites

• The inner diameter of the ferrite should be matched as well as ­possible to the outer diameter of the cable, i.e. the air gap between the cable and core should be as small as possible. • The choice of a long ferrite sleeve means a proportional increase in ­impedance; greater damping is also to be expected. • The greater the volume of a ferrite, the higher the premagnetization or signal related DC and AC currents can be without high impedance loss.

If the snap ferrite surrounds the conductor, the parallel equivalent resistance has the effect of a transformed series resistance, which is high compared with the input impedance, and therefore considerably reduces the interference current. This effect with snap ferrites can be impressively confirmed in the EMC lab and further optimized by measurements.

Snap Ferrites

2.6.2 STAR-GAP – Snap ferrites with a defined air gap The aim in developing the new snap ferrites is to increase the impedance and the suppression effect in the high frequency range and with DC bias. As has already been achieved for transformers, it is intended to reduce the effects of DC bias on ferrite impedance. When the snap ferrite is closed, a defined air gap is produced, which guarantees the desired characteristics at all times in every operating mode. Furthermore, the effect of external forces applied, e.g. cable bending, is almost completely excluded. On the basis of various measurement results, the impedance for frequencies > 100 MHz is illustrated and explained, in which frequency range snap ferrites with a defined air gap are to be recommended. The effect of DC bias on impedance is also described. The accurate measurement method especially developed for cable-mounted ferrites for the observed frequency range is described in the following section.

STAR-GAP

Fig. 2.38: Snap ferrite with defined air gap – STAR-GAP 742 716 33S 317

II Components Method for characterising cable-mounted ferrites

Fig. 2.39: Test circuit

Fig. 2.40: FEKO simulation model of the test circuit Snap ferrites are suitable for decoupling line-bound interference on round cables. The special feature of the ferrite material composition is its high i­mpedance loss component, through which the radiated currents are absorbed, i.e. transformed into heat. A common method on the market for characterising ferrites is therefore the measurement of impedance as a function of frequency. Using this measurement method, a dependence of the measured impedance on the nature of the windings and the length of the connecting wires is particularly apparent at high frequencies (> 100 MHz). The impedance measurement result obtained in this manner not only characterises the ferrite, but also includes the impedance transformation through the lead. An additional effect is that the measurement set-up radiates at high frequencies, because the wire loop, including the connecting wires, is no longer small compared with the wavelength.

318

For this reason, Würth Elektronik thoroughly investigated this matter and developed a measurement technique suitable for characterising ferrites – the reverse transformation method. The measurement circuit shown in Figure 2.39 was reconstructed precisely with field calculation program FEKO, which is based on the method of moments. This allows impedance measurement at the 50 Ω connection point to be simulated. The test circuit simulation model is presented in Figure 2.40. With the aid of the optimisation program OPTFEKO, the impedance of the test object assumed (and to

be measured) in the FEKO calculation is now changed until the impedance measured at the 50 Ω connection point coincides with the numerically calculated value. The transformation effect of the lead is taken into account in this method. Here the lead does not have to show a constant wave resistance, as FEKO can only determine the current distribution in the lead on the basis of its geometrical dimensions and the voltages applied, i.e. without knowledge of the wave resistance. The radiation of the lead at high frequencies is also taken into account as part of this numeric field calculation. In Figure 2.41, the impedance result obtained with the new method is compared with that obtained from the conventional method based on the example of the snap ferrite STAR-GAP 742 716 33S with a defined 0.8 mm air gap. With the conventional method, a lead length of 165 mm was used for measuring the ferrite on an impedance analyzer. The effect of this lead on the measurement result can only be ignored if its length is short compared with the wavelength. The threshold, up to which length a lead can be considered as short, which depends on the required accuracy, is 1/100 to a maximum of 1/10 of the wavelength. Using a 165 mm length wire, the wavelength can therefore be a maximum of tenfold, corresponding to a limit frequency of 180 MHz. From Figure 2.41 it can be seen that the measurement results still coincide at a frequency of 100 MHz, however they deviate significantly beyond this value. The cause is the excess wire length between the ferrite and the impedance analyzer, which together with the surrounding ground surfaces forms an undefined wave resistance for the lead in the conventional method. As a result of the non-negligible length, this effects a transformation of the impedance of the ferrite to be measured, i.e. the impedance measured with the impedance analyzer can deviate significantly from the actual impedance of the ferrite beyond approx. 100 MHz. This means that the conventional method of characterising ferrite properties are suitable up to a frequency of approx. 100 MHz; the reverse transformation method is more accurate above this frequency.

319

II Components

Fig. 2.41: Impedance of the snap ferrite STAR-GAP 742 716 33S with 0.8 mm air gap with simple and double cable feed-through: Comparison of new and conventional measurement methods Impedance Snap ferrites

Comparison of the impedance of a snap ferrite with and without air gap

Fig. 2.42: Magnitude, real and imaginary components of impedance for a double cable feed-through: Ferrite STAR-TEC 742 711 32 without air gap and ferrite STAR-GAP 742 716 33S with a defined 0.8 mm air gap; the ferrite materials and the core dimensions are identical 320

The impedance of the snap ferrite STAR-TEC without air gap 742 711 32 is determined for a double cable feed-through according to the newly developed method described above. This result is compared with the impedance curve of a snap ferrite STAR-GAP with a defined 0.8 mm air gap (742 716 33S) and otherwise identical core dimensions and material composition (Figure 2.42).

STAR-TEC

The following relationships apply for the impedance Z and the complex permeability µr.







(2.7)

(2.8)

Figure 2.42 shows that for frequencies above approx. 225 MHz, both the magnitude and real component of the impedance of the ferrite with defined air gap are significantly higher than the corresponding values for the ferrite without air gap. The maximum value of the impedance magnitude as well as the maximum value of the real component, which absorbs radiated currents, occurs at the resonant frequency of the configuration and is approx 1.8 kΩ for a ferrite with defined air gap; the maximum value of the corresponding curves for the ferrite without air gap is only around 1.1 kΩ. It was possible to increase in the loss component of a factor of 1.6 by introducing a defined air gap. It is also apparent that the effect of the air gap is to shift the attenuation maximum towards higher frequencies. This can be explained with an equivalent circuit. The snap ferrite with cable is represented by a lossy coil with a parallel c­ apacitance (see Figure 2.43). The inductance is reduced by the air gap and the resonant frequency of the parallel resonant circuit is increased. The overall magnetic resistance is increased by the air gap in the magnetic circuit. A further advantage of the air gap is a lower effect of temperature and change in field strength on the permeability.

Fig. 2.43: Equivalent circuit for the snap ferrite with double cable feed-through 321

II Components Further investigations examine the effect of the air gap with the use of just a single cable feed-through, i.e. by fitting a snap ferrite on an extended lead. The impedance curve of the snap ferrite without air gap 742 711 32 is compared with the results of the ferrite with 0.8 mm air gap. Figure 2.44 illustrates that the impedance of the ferrite is indeed higher than the impedance of ferrites with an air gap up to about 340 MHz. However at very high frequencies the opposite is the case. Only with a single cable feed-though does the advantage of the ferrite with air gap, i.e. an increase in impedance in the high frequency range, become apparent.

Selection of snap ferrites: 200–800 MHz:

STAR-GAP with a double cable feed-through

> 800 MHz:

STAR-GAP with a single cable feed-through

Comparison of Figure 2.42 with Figure 2.44 shows the relationship between winding turns and impedance. The impedance increases with increasing winding turns; ­however, the point of maximum impedance is shifted towards lower frequencies.

Fig. 2.44: Magnitude of the impedance of the ferrite STAR-TEC without air gap, as well as of the ferrite STAR-GAP with defined 0.8 mm air gap with a single cable feed-through

322

The influence of DC bias on the impedance of the ferrite The effects of DC bias on the impedance values of the ferrites presented is investigated as follows. This entailed increasing the DC bias in 1 A steps from 0 A to 5 A. For all measurements, saturation effects are observed at comparably low frequencies up to approx. 200 MHz, whereas at higher frequencies the DC bias had a negligibly small effect on impedance (cf. Figure 2.45).

DC bias Saturation

In the measured range from 100 MHz to 200 MHz, a defined air gap leads to the impedance of ferrites with air gaps changing significantly less as a result of the DC bias than the impedance of ferrites without air gaps. The characteristics of a ferrite with a defined air gap are changed less by a DC bias, such that the suppression effect with a DC bias of 5 A does not significantly differ compared with the effect achieved without DC bias. The design with air gap is therefore less susceptible to saturation effects.

Fig. 2.45: The effect of DC bias: Measurement of STAR-TEC 742 711 32 and snap ferrite STAR-GAP 742 716 33S with 0.8 mm air gap and double cable feed-through

2.7 WE-MI Multilayer Inductor Coil within ferrite If the wire windings on the outside of a conventional “coil” are mounted i­nside the core body, the multilayer SMD inductor WE-MI (Figure 2.46) is created.

323

II Components SMD Multilayer inductor WE-MI

Fig. 2.46: SMD-Multilayer inductor WE-MI

Fig. 2.47: Structural schematics of multilayer SMD inductors Size 0603:

324

DCR (Ω)

(MHz) 25

240

0.50

50

10

 85

0.60

25

Test frequency

(µH)  0.1

15

 1.0

35

Inductance

744 796 2 744 796 5

Rated Current

Selfresonant frequency (MHz)

Quality factor Q (min)

Order Code

(mA)

Size 0805: DCR (Ω)

(MHz) 25

140

0.60

200

45

10

41

1.00

30

Quality factor Q (min)

Test frequency

Selfresonant frequency (MHz)

DCR (Ω)

Rated Current

Test frequency

(µH) 0.47

25

4.7

Inductance

Inductance

744 790 32 744 790 6

Rated Current

Selfresonant frequency (MHz)

Quality factor Q (min)

Order Code

(mA)

Size 1206: Order Code

(µH)

(MHz)

(mA)

744 791 3

 0.33

20

25

145

0.5

250

744 791 53

 3.30

45

10

 48

0.7

 50

744 791 8

10.00

50

 2

 26

0.6

 25

Tab. 2.12: Typical characteristics of the SMD-Multilayer inductor WE-MI The WE-MI inductors achieve low DC resistances through their compact structure. The ferrite body magnetically shields the component, significantly reducing external interference and cross-talk. The multilayer inductor may be seen as a “compromise” between the ceramic inductor and SMD ferrite. This component is especially suitable as an inductor in filters and resonant circuits where low interference from external signals is required and in circuits with high packing density.

Practical tips: • Do not operate in the self-resonant range • Observe max. current loading capacity • Low DC resistance, therefore also suitable for low-voltage systems

2.8 WE-FI Radio interference suppression choke Toroidal cores of radio interference suppression chokes are made of compacted iron powder and have a very low stray field. High current loading c­ apacity is achieved through high saturation magnetization. The useable maximum upper frequency range of this component extends from a few MHz to approx. 30 MHz depending on the type.

RF-suppression chokes

Frequency range 325

II Components The impedance-phase curve against frequency of a typical toroidal core choke (744 707 0) is shown in Figures 2.48 and 2.49.

Fig. 2.48: Impedance and phase of the toroidal core choke 744 707 0 (100 µH) against frequency (0 MHz–5 MHz)

Fig. 2.49: Impedance and phase of the toroidal core choke 744 707 0 (100 µH) against frequency (0.5 MHz-30 MHz)

326

It may be seen from Figure 2.49 that the resonant frequency is at 4.1 MHz. Above 4.1 MHz the capacitive character of the choke predominates, at 30 MHz the impedance has fallen to approx. 200 W. The impedance is almost linear (Figure 2.48) up to the resonant frequency of 4.1 MHz.

Resonant frequency

Due to the frequency dependence of complex permeability, calculations are only feasible in limited frequency bandwidths and sufficiently far (in the linear range) below the resonant frequency. Figure 2.50 shows the equivalent circuit of the choke in the 25 kHz to 1 MHz range, Figure 2.51 the equivalent circuit in the 1 MHz to 5 MHz range. The values were found using an impedance analyzer, calculating on the basis of the components’ equivalent circuit. Due to high non-linearity, the equivalent circuit can be simulated over a wide frequency range with just 3 components.

Fig. 2.50: Equivalent circuit with associated measured and simulated ­impedance-phase curves

327

II Components

Fig. 2.51: Equivalent circuit with associated measured and simulated ­impedance-phase curves From the measurement curves impedance and phase an increase in eddy-current losses and a decrease in the complex permeability can be seen. With





(2.9)

the impedance can be calculated as a function of frequency. The broken line in Figures 2.50 and 2.51 are the simulated curves with the values given by the equivalent circuits. It shows that without considering the complex permeability and its frequency dependence, the calculation can be performed with limited accuracy.

328

Practical tip: Here another word on the saturation of the ferrite ring: The effective core cross-sectional area is inversely proportional to the saturation current and proportional to the impedance. This means that wherever possible, the larger core cross-section area should be chosen.

Saturation

2.9 C  ommon Mode Chokes Saturation effects caused by high signal currents, or DC current super­imposed on the signal, reduce the effectiveness of the choke. The use of standard inductors in the signal path adversely impairs the useful signal. Current compensation circumvents these disadvantages. In current compensation, the “useful return current” must be passed through the choke. In this way, the useful current does not contribute to the magnetization of the core.

Current ­compensation

Current compensated choke

Fig. 2.52: Construction and circuit diagram of a current compensated choke Push-pull signals

Common-mode signals

Differential Mode

Common Mode

Symmetrical voltage

Asymmetrical voltage

Tab. 2.13: Common expressions Current-compensated chokes can be manufactured with different ferrite geometries; the best known are ring core and ribbed core. Different core materials enable their use in various frequency ranges. A very well known component, but one not designed as a common mode choke, is the snap ferrite or the split ferrite sleeve.

Snap ferrite 329

II Components The effect of current-compensated chokes on coupled interference is, above all, used for data and signal lines. They are often the only option to avoid the interference suppression component affecting the useful signal.

2.10 Common Mode Chokes for Data and Signal Lines SMD Common Mode Line Filter WE-CNSW Sizes: 0805; 1206 WE-CNSW

Fig. 2.53: WE-CNSW common mode chokes Order Code

Tolerance (%)

DCR (Ω)

Rated Current (mA)

Impedance @ 100 MHz (Ω)

Size

744 231 091

±25

0,3

370

  90

0805

744 232 601

±25

0,8

260

 600

1206

744 232 222

±25

1,2

200

2200

1206

Tab. 2.14: Characteristic data of some common mode WE-CNSW data line filters The WE-CNSW component series is the only data line filter in the Würth Elektronik product range not based on a ring core, either in size 0805 or 1206. Only for this reason is it possible to achieve such a compact common mode component. However, as it is a closed ferrite material system, the stray field remains negligibly small. Typical applications for the WE-CNSW are USB, Firewire and High Speed Data Lines.

330

2.11 WE-SL, WE-SLM, WE-SL1, WE-SL2, WE-SL3, WE-SL5 SMD Common Mode Line Filter WE-SL WE-SL

Fig. 2.54: WE-SL common mode filter Order Code

Inductance (µH)

Tole­ rance (%)

DCR (Ω)

Rated Current (mA)

Impedance (Ω)

744 202

4 x 1000

±35

0.400

 350

5000

744 205

4 x   100

±35

0.100

 700

 900

744 207

2 x    35

±35

0.035

2700

1100

Tab. 2.15: Illustration and specifications of some WE-SL SMD line filters In contrast to WE-CNSW, the complete WE-SL series, i.e. also the newly developed types WE-SL1, WE-SL3 and WE-SL5, include ring cores. As a result, stray fields can almost be excluded. The different geometries and, above all, the very flat package heights of the various types, offer potential solutions for every application.

WE-SL

High current ratings, as used in low-voltage applications, are also available. Despite their compact construction, the WE-SL line filter also offers 4x ­current-compensated versions.

331

II Components SMD Common Mode Line Filter WE-SLM WE-SLM

Fig. 2.55: WE-SLM common mode line filter Order Code

Inductance (µH)

DCR max. (Ω)

Rated Current (mA)

max. Impedance (Ω)

744 242 110

 11

0.18

300

 800

744 242 220

 22

0.23

300

1500

744 242 330

 33

0.27

300

2000

744 242 510

 51

0.32

300

2500

744 242 101

100

0.58

300

4000

Tab. 2.16: Typical specifications of some WE-SLM SMD line filters The WE-SLM data line filter offers high in common mode impedance with a smaller footprint than WE-SL1. A wider working temperature range is achieved through the use of NiZn ferrite. At the same time, the leakage inductance is lower so the signal is less affected. The WE-SLM can therefore also be used at high signal frequencies.

332

SMD Common Mode Line Filter WE-SL1 WE-SL1

Fig. 2.56: WE-SL1 common mode line filter Order Code

Inductance (µH)

Tole­ rance (%)

DCR (Ω)

Rated Current (mA)

Impedance (Ω)

744 212 100

2 x   10

±40

0.24

300

1200

744 212 510

2 x   51

±40

0.16

300

 300

744 212 101

2 x 100

±40

0.22

300

 500

744 212 331

2 x 330

±40

0.30

300

2000

Tab. 2.17: Typical characteristic data of some common mode SMD line filters WE-SL1 The data line filter WE-SL1 above excels by virtue of its low space requirement, both in terms of package height as well as footprint. To nevertheless achieve adequate attenuation values, manganese-zinc was chosen as the basic material.

333

II Components SMD Common Mode Line Filter WE-SL2 WE-SL2

Fig. 2.57: WE-SL2 common mode line filter Order Code

Inductance (µH)

Tole­ rance (%)

DCR (Ω)

Rated Current (mA)

Impedance (Ω)

744 220

2 x 4700

±50

0.75

 500

20000

744 222

2 x 1000

±50

0.31

 800

 6000

744 227*

2 x    51

±30

0.16

1000

 5500

744 226*

2 x    10

±30

0.08

1600

  920

* Also available as sectionally wound types Tab. 2.18: Typical characteristic data of some common mode SMD line filters WE-SL2 Sectional winding

Bifilar winding

334

The WE-SL2 line filter is produced both in a) sectional as well as in b) bifilar winding technology. The separated construction of the sectional winding allows both the attenuation of high frequency symmetrical frequency components, as well as the suppression of asymmetrical interference components. However, if the quality of the useful signal is too greatly affected, the original bifilar winding technology should be chosen.

SMD Common Mode Line Filter WE-SL3 WE-SL3

Fig. 2.58: WE-SL3 common mode line filter Order Code

Inductance (µH)

Tole­rance (%)

DCR (Ω)

Rated Current (mA)

Impedance (Ω)

744 253 200

3 x   20

+50/–30

0.16

500

1250

744 252 220

2 x   22

+50/–30

0.14

700

1600

744 252 510

2 x   51

+50/–30

0.25

500

3300

744 253 101

3 x 100

+50/–30

0.45

450

5000

Tab. 2.19: Typical characteristic data of some common mode SMD line filters WE-SL3 The WE-SL3 date line filter represents an advancement of the WE-SL2 component series. Despite being half the package height, almost the same performance can be attained, at least for low inductance values. Additionally a 3x current-compensated version has been developed, which is mainly used for low voltages.

335

II Components SMD Common Mode Line Filter WE-SL5 WE-SL5

Fig. 2.59: WE-SL5 common mode line filter Order Code

Inductance (µH)

Tole­rance (%)

DCR (Ω)

Rated Current (mA)

Impedance (Ω)

744 272 121

2 x   120

±40

0.025

2500

  460

744 272 471

2 x   470

±40

0.065

1600

 1750

744 272 222

2 x 2200

±40

0.30

 750

 7500

744 272 472

2 x 4700

±40

0.72

 350

13000

Tab. 2.20: Typical characteristic data of some common mode SMD line filters WE-SL5 Manganese-zinc is used exclusively as the basis for the WE-SL5 line filter. This allows the frequency band in the single and double digit megahertz range to be covered.

2.12 Common Mode Chokes for Power Lines SMD Common Mode Power Line Choke WE-LF for mains voltage applications WE-LF

Fig. 2.60: WE-LF common mode choke

336

Order Code

Inductance (mH)

Rated Current (A)

DCR (W)

Housing

744 612 1010

10

0.7

0.55

SV

744 672 1027

27

0.6

0.70

MH

744 632 1033

33

0.8

0.85

LV

Tab. 2.21: Typical specifications of some WE-LF common mode chokes

Horizontal version SH, MH, LH, XH

C

C

Vertical version SV, MV, LV, XV

G A

A

F

F

B G B

D

E

E

D

H Housing

A (mm)

B (mm)

C (mm)

D (mm)

E (mm)

F (mm)

G (mm)

H (mm)

612/SV

18.5

13.5

20.5

15.0

10.0

3.0

0.6

15.0

622/MV

23.5

16.0

25.5

10.0

12.5

3.0

0.6

10.0

632/LV

26.5

18.5

30.5

12.5

15.0

3.0

0.6

12.5

642/XV

32.5

21.5

35.5

12.5

17.5

5.0

0.8

12.5

662/SH

18.0

18.0

13.0

10.0

15.0

3.0

0.6

–

672/MH

23.0

23.0

14.5

12.5

20.0

3.0

0.6

12.5

682/LH

28.5

28.5

17.0

15.0

25.0

3.0

0.6

15.1

692/XH

33.5

33.5

20.0

20.0

30.0

3.5

0.8

20.1

Tab. 2.22: Dimensions of the common mode choke WE-LF The maximum permissible component temperature of the WE-LF component series is +125 °C. If the permissible excess temperature of 55 K is subtracted, an allowable ambient temperature of +70 °C still remains. For even higher temperatures, the rated current must be reduced according to the following formula.





(2.10) 337

II Components WE-LF SMD Common Mode Power Line Choke WE-LF SMD

In its electrical characteristics the WE-LF SMD in size S equals the sizes SH and SV of the THT version WE-LF. In addition to the already existing advantages of the WE-LF family, reflow soldering is possible with the WE-LF SMD. The known excellent ­mechanical stability of this family is as well given for the SMD version.

Fig. 2.61: WE-LF SMD common mode choke Order Code

L (mH)

IN (A)

DCR (W)

744 663 400 07

0.7

4.0

0.03

744 663 200 1

1.0

2.0

0.06

744 663 200 2

2.2

2.0

0.10

744 663 200 3

3.3

1.5

0.15

744 663 100 7

6.8

1.0

0.30

744 663 101 0

10

0.7

0.55

744 663 002 7

27

0.4

1.20

744 663 003 9

39

0.4

1.70

744 663 004 7

47

0.3

2.60

Tab. 2.23: Typical specification of some WE-LF SMD common mode chokes

338

WE-CMB Common Mode Power Line Choke for mains applications WE-CMB Vertical mount

Fig. 2.62: WE-CMB common mode choke Order Code

Inductance (mH)

Rated Current (A)

DCR (W)

Type

744 821 120

20

0.5

1.00

XS

744 822 110

10

1.0

0.36

S

744 823 333

3.3

2.5

0.06

M

744 824 622

2.2

6.0

0.02

L

744 825 1201

1.0

12.0

0.009

XL

744 826 1418

1.8

14.0

0.0079

XXL

Tab. 2.24: Typical specification of some WE-CMB common mode chokes The special construction of the WE-CMB current-compensated choke allows the reduction of undesirable parasitic effects for mains applications. The a­ lmost perfectly selected core/winding relationship allows a very high current for a comparable footprint. However, if conducting components or packaging parts are placed in the immediate vicinity of a WE-CMB, the required safety separation must be ­ensured, as enamelled wires are not considered to be insulated components. In most cases, the use of a WECMB is, however, unproblematic as insulated components, such as capacitors, provide the necessary separation. Of course, the corresponding packaging version of a WE-LF can be chosen.

339

II Components Common Mode Power Line Choke WE-CMB NiZn WE-CMB NiZn

Fig. 2.63: WE-CMB NiZn common mode choke Order Code

Inductance (µH)

DCR (mW)

Rated Current (A)

Type

744 841 414

 14

15

4

XS

744 841 330

 30

26

3

XS

744 841 247

 47

40

2

XS

744 841 210

100

80

1.5

XS

744 842 1016

16

10.0

2.7

S

744 842 932

32

8.5

5.0

S

744 842 742

42

6.5

8.1

S

744 842 565

65

5.0

13.0

S

744 842 311

110

3.0

31.0

S

Tab. 2.25: Typical specification of some WE-CMB common mode chokes The NiZn variants are a specialty within the WE-CMB series and are available in the sizes XS and S. In contrast to comparable mains chokes, nickel-zinc is used here as the base material. This makes it possible to effectively suppress common-mode ­interference components up to the high frequency range. This product variation is also suitable for increasing the interference immunity against incident RF radiation and burst signals. The main frequency component of the interference phenomena mentioned is within the working range of the choke and can therefore be effectively attenuated.

340

Fig. 2.64: Insertion loss (common mode) “WE-CMB NiZn Size XS”

Fig. 2.65: Insertion loss (common mode) “WE-CMB NiZn Size S” WE-CMBNC Common Mode Power Line Choke Nanocrystalline

WE-CMB NC

The WE-CMBNC is a new member of the WE-CMB series. It is a VDE certified series of common mode chokes with highly permeable nanocrystalline core material. The innovative design results in a small size which delivers outstanding broadband attenuation performance, high rated currents and low DC resistance values. Inductance values are stable to 150 °C with it handles high currents and operates over a large frequency range 1 kHz to 300 MHz.

341

II Components WE-FC Current compensated power line choke

Common Mode Power Line Choke WE-FC

Multi-chamber coil body Fig. 2.66: WE-FC common mode choke Order Code

Inductance min. (mH)

Rated Current max. (A)

DCR (W)

Type

744 864 0395

0.82

2.0

0.065

ET

744 864 0398

3.30

1.1

0.250

ET

744 864 0402

10.0

0.6

0.720

ET

744 864 0405

33.0

0.3

2.000

ET

744 864 0406

0.82

2.0

0.065

UT

744 864 0410

2.70

1.2

0.190

UT

744 864 0414

6.80

0.7

0.470

UT

744 864 0418

33.0

0.3

2.500

UT

Tab. 2.26: Typical specification of some common mode power line chokes WE-FC The WE-FC current-compensated choke has roughly twice the leakage inductance r­ elative to comparable toroidal chokes. The effect on symmetrical interference increases without having to use an additional inductor. Parasitic parallel capacitance is reduced as a result of the construction with a multi-chamber coil body. The impedance profile is raised at higher frequencies. At the same time, resilience against burst and surge pulses is improved.

342

2.13 WE-ExB Common Mode Power Line Choke WE-ExB Two core inductor

Fig. 2.67: WE-ExB Common Mode Power Line Choke Order Code

L1 (µH)

IR 1 (A)

RDC1 max. (mΩ)

744844470

2x47

15

2x4.6

744844101

2x100

14

2x6

744844221

2x220

12

2x9

744844471

2x470

9

2x16

744844102

2x1000

4.5

2x42

UR 1 (V (AC))

250

L1: Inductance; IR 1: Rated Current; RDC1 max.: DC Resistance; UR 1: Rated Voltage

Tab. 2.27: Typical specification of some WE-ExB chokes The use of two core materials, MnZn and NiZn offer noise suppression over a very broad frequency range from 100 kHz to 100 MHz. Useful fro combined noise and burst filter applications in power supplies and motors.

40

50

attenuation (dB)

30 25 20 15

744 844 102 744 844 471 744 844 221 744 844 101 744 844 470

45 40 35 attenuation (dB)

744 844 102 744 844 471 744 844 221 744 844 101 744 844 470

35

30 25 20 15

10

10

5

5

0 0.1

1

10

100

0 0.1

frequency (MHz)

Fig. 2.68: Typical performance of some WE-ExB chokes

1

10

100

frequency (MHz)

343

II Components 2.14 WE-LPCC Common Mode Power Line Choke WE-LPCC Low profile

Fig. 2.69: WE-LPCC Common Mode Power Line Choke Order Code

L1 (µH)

IR 1 (A)

RDC1 max. (mΩ)

7448680200

120

23.5

1.4

7448680180

150

20

1.9

7448680140

230

16

3.6

7448680120

330

12

6

7448680100

450

10

9.6

UR 1 (V (AC))

UT 1 (V (AC))

250

1500

L1: Inductance; IR 1: Rated Current; RDC1 max.: DC Resistance; UR 1: Rated Voltage; UT 1: Insulation Test Voltage

Tab. 2.28: Specifications of some WE-LPCC chokes The WE-LPCC common mode power line choke series provides stable leakage inductance for differential mode filtering with high current capabilities in a low profile package.

344

3 Power Magnetics – Inductors 3.1 WE-PMI – Power Multilayer Inductors WE-PMI

Fig. 2.70: Power Multilayer Inductors WE-PMI The miniaturization of SMD components, especially inductors, is a widespread trend in portable devices, as it is especially storage chokes that frequently require the most space. Wired components are out of the question in these orders of magnitude. This is where the newly developed WE-PMI types (744 797 x) apply. In order to allow minimization of the coil volume, the switching controller IC is driven at ever-higher switching frequencies. Switching controllers like the Micrel MIC2285 already work with 4 MHz. The dimensions of the storage chokes required can therefore be reduced by up to 90%. The compact WE-PMIs in 1008 package (2.5 mm x 2.0 mm x 1.0 mm) not only offer high rated currents (up to 2.4 A), but also a lower DCR than comparable WE-MI types (multilayer inductors). Order Code

Inductance ±20% (µH)

DCR max. (mW)

IR (A)

Isat (A)

Type

744 797 871 47

0.47

90

1.9

0.7

Low RDC

744 797 872 10

1.0

120

2.5

0.5

Low RDC

744 798 872 10

1.0

290

1.00

1.2

High Current

744 798 872 22

2.2

400

0.85

1.1

High Current

744 798 872 47

4.7

530

0.75

8.5

High Current

744 798 872 68

6.8

650

0.70

6.0

High Current

744 798 873 10

10

700

0.65

3.5

High Current

Tab. 2.29: Typical characteristics of several Power Multilayer Inductors WE-PMI The saturation current of the WE-PMIs relates to the typical inductance drop of –30% from the zero current inductance. The rated current is defined for the common selfheating of DT = 40 K with respect to the ambient temperature.

345

II Components

Fig. 2.71: Inductance vs. DC current The used NiZn core material allows the use of the WE-PMI series up to 10 MHz. Due to multilayer type the WE-PMI are especially suitable for power supplies in portable devices.

Fig. 2.72: Inductance vs. frequency

346

SMD – Wire Wound Inductor WE-GF/WE-GFH The WE-GF wound SMD inductors are available in SMD packages (Figure 2.73). They essentially differ in their mechanical construction. The WE-GF is completely embedded in plastic and can therefore handle high humidity very effectively. It can therefore be loaded with higher currents at the same inductance in relation to its package volume.

WE-GF

Fig. 2.73: WE-GF Order Code

DCR Rated SelfTest Induc- Tole­ Quality current tance rance factor frequency resonant max. frequency Q (mA) (Ω) (MHz) (MHz) (min) % (µH)

744 764 001   0.10

±20

28

100

700

0.44

450

744 764 01

  1.00

±10

30

7.96

120

0.70

400

744 764 10

 10.00

±10

30

2.52

 36

2.10

150

744 764 20

100.00

±10

20

0.796

 10

11.0

40

Tab. 2.30: Specifications reflect a wide spectrum for the inductor WE-GF In Figure 2.74 the layout of the WE-GF is shown graphically. A wire-wrapping surrounds a ferrite body. The special ferrite mixture faciliates a wide inductance spectrum despite the miniature ferrite core.

Fig. 2.74: Inductor construction of resistor-type WE-GF

347

II Components 3.2 WE-MAPI Shielded SMD Metal Alloy Power Inductor WE-MAPI

Fig. 2.75: WE-MAPI Metal Alloy Power Inductor The WE-MAPI is the world’s smallest metal alloy power inductor with unmatched efficiency. The innovative design features set it apart. The winding is contacted directly with the component’s termination pad without soldering, welding or using a clip. Saving the clip allows the effective coil diameter to be increased, reducing the number of turns needed for the same inductance values. This is directly expressed in reduced DC resistance (RDC) of the winding.

Fig. 2.76: Self-shielding winding and core design for improved EMI behavior of the WE-MAPI series In the application, the start of the coil winding (indicated by a mark) is usually connected with the switching node of the switching controller. Consequently, the spatial expansion of the “hot” switching node is minimized and coupling effects through the outer part of the winding at a “stable” potential are also shielded.

348

The core consists of an innovative metal alloy pressed around the winding. This provides high inductance values in a small package size. At the same time, the special construction of the core has a self-shielding effect. The core material is temperature stable with only a slight drift and “soft” saturation behavior. A protective layer is also

applied around the core rendering the surface resistant against environmental influences. Small size, high current ratings, low ac losses and excellent temperature stability make WE-MAPI inductors well suited for multiphase converters and small, high efficiency power supplies such as used in battery powered devices. Order Code

L1 (µH)

Tol. L

IR 1 (A)

ISAT 1 (A)

RDC1 typ (mΩ)

RDC1 max. (mΩ)

744 383 130 033

0.33

1.9

4.9

65

84

744 383 130 047

0.47

1.7

4.5

77

101

744 383 130 056

0.56

1.65

4

90

113

744 383 130 068

0.68

1.55

3.8

101

126

744 383 130 082

0.82

1.45

3.6

115

144

±30%

744 383 130 10

1

1.4

3.4

127

159

744 383 130 12

1.2

1.3

3.2

140

174

744 383 130 15

1.5

0.95

2.7

189

237

744 383 130 22

2.2

0.85

2.5

337

388

L1: Inductance; Tol. L: Inductance (Tol.); IR 1: Rated Current; ISAT 1: Saturation Current; RDC1 typ: DC Resistance; RDC1 max.: DC Resistance

Tab. 2.31: Specifications for size 1610 WE-MAPI inductors

3.3 WE-SI P  ower Inductors Switched-mode power supplies are becoming ever more widespread. The semiconductor manufacturers have made their contribution, offering a wide range these integrated circuits with simplified circuit design. Care must be taken in the selection of the appropriate storage choke to fully u­ tilize the advantages of switching regulators. The selection of cores and windings of the WE-SI, WE-TPC and WE-PD series of storage chokes are optimized for use in switching converters and DC-DC converters.

Switched-mode power supply

Storage choke

A standard series of storage chokes has been compiled following recommendations from various switching converter IC manufacturers, e.g. Analog Devices, STMicroelectronics, Texas Instruments, Exar, Diodes Inc., MPS, ON Semiconductor, Semtech, Maxim Integrated (see Würth Elektronik catalog). This program is supplemented by special customized solutions.

349

II Components Toroidal core WE-SI WE-SI

Fig. 2.77: Storage choke WE-SI Toroidal core

Toroidal storage chokes are ideal from the EMC perspective: The magnetic field lines mainly pass through the core. The stray field and associated c­ oupling in neighboring conductor tracks or components remain small. In the field of switching converters, storage chokes serve to buffer electrical energy and, at the same time, to smooth the output current.

Energy of magnetic field

The energy stored in the core in this process is:





(2.11)

To enable high energy storage and to minimize the resulting core losses, the toroidal core volume is divided into many electrically isolated regions. The iron powder used in our storage chokes therefore has three-dimensional, uniformly distributed, microscopic air gaps, which reduce eddy-current losses. The disadvantage of reduced permeability is balanced by greater maximum energy storage and lower losses. Furthermore, these cores are extremely well suited for use in applications with high DC premagnetization. Datasheet specification terms Open-circuit ­inductance

Inductance ­measurement

350

Open-circuit inductance L0: If the inductor is operated without DC premagnetization or with only a small AC ­current, the open-circuit inductance L0 results. This value may be measured with sufficiently sensitive inductance measuring equipment for small AC voltages e.g. 0.1–0.5 V and a fixed measuring frequency between 1 kHz and 100 kHz, depending on the inductance v­ alue.

Inductance rating LN:

Rated inductance

In addition to the small AC voltage amplitude, the specified DC current is superimposed and the resulting inductance measured. Current rating IN:

Rated current

The DC current, for which the inductance and wire thickness are specified and whose specifications are optimized. As shown in the graph below, inductance only saturates with a much larger current.

Fig. 2.78: Inductance with DC premagnetization (744 114); IN = 1 A; LN = 100 µH DC resistance DCR:

DC resistance

The windings resistance value is measured with an ohmmeter at an ambient temperature of +25 °C. The test current for resistance measurement is a small DC current, which does not lead to a significant temperature increase in the wire. As values in the milliohm range are measured here, a 4-wire measurement must be made to minimize measurement errors. Magnetic field energy E: The energy, for which the core data and windings of the coil is optimized. This is specified in microjoules.

Energy

The following simple and practical formula can be used for d­ imensioning a storage choke. A brief extract from the extensive core m ­ aterial program and the following table should provide an overview of the choke dimensioning process. Depending on the application, further specifications from the core material data may be necessary.

Dimensioning of toroidal cores

351

II Components Iron core material data The following table shows an overview of the most commonly used materials and their applications. Iron powder cores

Application Temp. coeff. (+ppm/°C)

Material

Color identification

Permeability (µr)

3W5540

red

55

+385

power factor correction; storage choke f > 50 kHz

3W7538

red/white

75

+825

radio interference suppression, 50-Hz-choke, storage choke f < 70 kHz

3W7544

red/blue

75

+650

storage choke f > 70 kHz; 50-Hz-choke, power factor correction

Tab. 2.32: Materials and their applications Operating ­temperature

Operating temperature:

Losses

The operating temperature of the iron powder core may be from –55 °C to +125 °C. Prolonged core operation above +75 °C however results in increased losses.

Insulation voltage

Insulation voltage:

Coating

The protective coating of the toroidal core uniquely identifies the core material by color and serves to protect against environmental effects and provides electrical isolation from the windings. Epoxy resin coatings are used and an insulation dielectric strength of 500 VDC is achieved as standard. Higher i­nsulation voltages can also be offered.

AL value

AL value: ­ inding turns for For every size of core an AL value is specified to simply calculate the w the required choke; the tolerance is ±10%. The standard means of measuring the AL value is at B = 1 mT and f = 10 kHz.

352

Fig. 2.79: Effective permeability with DC premagnetization

Material 3W5540 AL-value (nH/N2)

Material 3W7538 AL-value (nH/N2)

Material 3W7544 AL-value (nH/N2)

Core no.

da (mm)

di (mm)

h (mm)

l (cm)

A (cm2)

V (cm3)

lw (cm)

25.0

37

35

11

11.2

 5.82

 4.04

2.68

0.099

0.266

1.84

24.0

33

33

13

12.7

 7.70

 4.83

3.19

0.112

0.358

2.01

39.5

58

54

18

17.5

 9.40

 6.35

4.23

0.242

1.030

2.77

31.0

46

42

20

20.2

12.60

 6.35

5.14

0.231

1.190

2.80

46.5

71

63

20

20.2

12.60

 9.53

5.14

0.347

1.780

3.44

47.0

70

64

23

22.9

14.00

 9.53

5.78

0.395

2.280

3.64

70.0

93

95

27

26.9

14.50

11.10

6.49

0.659

4.280

4.49

Tab. 2.33: Specifications of iron powder cores da di h l A V lW

= outer diameter = inner diameter = height = effective magnetic length = effective magnetic cross-sectional area = effective magnetic volume = winding wire length for 1 turn 353

II Components Wire table

Wire AWG

Wire Ø (mm)

typ. resistance/length DCR/lW (mΩ/cm)

Permissible current for maximum temperature rise (A) 10 °C

25 °C

40 °C

28

0.3

2.4200

0.64

1.07

1.38

26

0.4

1.3790

0.90

1.52

1.97

24

0.5

0.8790

1.29

2.17

2.81

22

0.6

0.5750

1.83

3.09

4.00

20

0.8

0.3390

2.62

4.41

5.70

19

0.9

0.2640

3.12

5.26

6.81

18

1.0

0.2160

3.72

6.27

8.11

17

1.1

0.1660

4.45

7.50

9.70

16

1.3

0.1320

5.33

8.97

11.60

15

1.4

0.1040

6.35

10.70

13.80

14

1.6

0.0843

7.60

12.80

16.60

13

1.8

0.0664

9.03

15.20

19.70

Tab. 2.34: Wire table Storage choke ­calculation

Storage choke calculation: The following demonstrates how a storage choke can be calculated for a switching converter application: Example: switching converter (step-down controller – storage choke) Requirements: Inductance rating LN = 100 µH Current rating (DC) IN = 1 A Peak current through the inductance Imax = 1.5 A Ripple current = 20% of Imax = 0.3 A (see Chapter III/Applications) Switching frequency f = 52 kHz A maximum AC flux density BAC = 0.05 T is recommended for iron powder cores (to ensure low core losses). Also the inductance should be selected so the ripple current does not exceed 20%–30% of the maximum current.

Core volume

354

Step 1: Choice of the core material and the necessary core volume (V) As the switching frequency is just 50 kHz, we firstly select the material 3W7538 with µr = 75 (see Table 2.31).

 Selected core: Magnetic data:

(2.12)

3W7538, as switching frequency < 70 kHz; core no. 13 da = 12.7 mm; di = 7.7 mm; h = 4.83 mm l = 3.19 cm; A = 0.112 cm2; V = 0.358 cm3 AL value: 33 nH/N2 AL value

Step 2: Required winding turns L in nH AL value in nH/N2



(2.13)



The final number of winding turns must be increased as a result of current dependent permeability. The correction factor for the AL value is determined from the “effective permeability against DC premagnetization” graph (see Figure 2.79).





(2.14)

At H = 1724 A/m on the graph in Figure 2.79 → Effective permeability with DC premagnetization = 80% of the initial p­ ermeability To be certain that the full inductance rating of 100 µH exists with a DC c­ urrent of 1 A, the final number of winding turns is calculated as:





Inductance rating

(2.15)

Step 3: Determination of the DC resistance The wire diameter can be ascertained from the relevant wire tables for the required current of 1 A, e.g. 22 AWG (d = 0.6 mm). This limits the self-heating of the wire to less than +10 °C.

Wire diameter

355

II Components DC resistance

The DC resistance of the windings is given by:



 (2.16)

AC flux density

Step 4: Check for max. AC field flux density



 (2.17)

with: Inductance rating Ripple current Core cross-sectional area Winding turns Peak voltage of the choke Duration of peak voltage

L in H DI in A A in cm2 N Us in V (during “t”) t in s

Step 5: Calculation of core losses Core losses

The losses in the core material may be calculated from the following f­ormula:





with Frequency AC field flux density Core losses

(2.18)

f in Hz B in mT PC in mW/cm3

The constants for the various materials are: Material

a

b

c

d

3W5540

8.0*108

1.7*108

9.0*105

3.1*10–14

3W7538

1.0*109

1.1*108

1.9*106

1.9*10–13

3W7544

1.0*109

1.1*108

2.1*106

6.9*10–14

Tab. 2.35: Material constants

356

For our examples this leads to:



(2.19)



The total core losses of the selected core are:





(2.20)

Copper losses

The losses in the windings equal: PWdg = I2max · DCR = 1.5 A2 · 0.07165 W = 161 mW

(2.21)

The total losses of the storage choke are low at around 370 mW and the choke calculated is well suited for the application.

3.4 WE-PD SMD Power Inductors The WE-PD series of storage chokes uses highly dynamic and low loss NiZn ferrite cores. They are suitable as chokes in switching control applications up to a clock frequency of approx. 10 MHz and offer high current loading capacity and low DC resistances.

WE-PD Storage choke

There are a multitude of construction types available for the different types of application: • Magnetically shielded series WE-PD and WE-PD3 • Unshielded versions series WE-PD2 and WE-PD4

Fig. 2.80: SMD power inductors WE-PD 357

II Components Datasheet specifications Inductance

Inductance L: Different measurement conditions apply for the different construction s­ eries. The inductance is stated at a certain test frequency and measurement voltage (see data sheet).

Rated current

Current rating IN: The current rating of the inductor is specified as the DC current at which inductor exceeds the permitted tolerance limits (DL) or the self generated heating (DT) exceeds a certain limit. The smaller of the currents defined by the two conditions is termed the current rating of the inductor. This is how­ever not the saturation current, which is higher than the current rating. DC resistance DCR:

DC resistance

The windings resistance value is measured with an ohmmeter at an a­ mbient temperature of +20 °C. The test current for resistance measurement is a small DC current, which does not lead to a significant temperature increase in the wire. As values in the milliohm range are measured here, a 4-wire measurement must be made to minimize measurement errors. Operating temperature:

Operating ­temperature Self-heating

The ambient temperature when operating the WE-PD series of storage chokes at full current rating load should generally range from –40 °C to +85 °C. The self-heating of the component must be taken into account at higher ambient temperatures in order that the permissible solder joint temperature is not exceeded or the wire insulation damaged. The wire used can withstand a temperature of up to +150 °C. The ferrite core itself may be used over a far greater temperature range (approx. –50 °C to about +125 °C far away from the Curie temperature of about 220 °C). However, in this case, the tolerance limits of the ­inductor may be exceeded due to the temperature dependence of permeability. As a rule:

Operating temperature = ambient temperature + self-heating < +125 ºC

358

Fig. 2.81: Derating curve WE-PD The above curve assumes that self-heating is permissible up to a maximum temperature sum (component + ambient temperature) of +125 °C. The current must be reduced above an ambient temperature of +85 °C. The curve is intended for critical applications, in which the coil itself should only generate a small amount of self-heating.

Derating

Critical applications

Insulation resistance: The insulation resistance between windings and coil core is more than 100 MW for a test voltage of 500 VDC.

Insulation resistance Test voltage

Typical specifications Size 7332 (WE-PD): DCR (Ω)

DCR max. (Ω)

Rated Current (A)

Saturation Current (A)

Order Code

Inductance (µH)

744 778 10

 10

0.068

0.072

1.83

2.20

744 778 20

100

0.585

0.790

0.62

0.76

Size 1260 (WE-PD): Order Code

Inductance (µH)

DCR (Ω)

DCR max. (Ω)

Rated Current (A)

Saturation Current (A)

744 771 10

 10

0.018

0.025

5.00

5.50

744 771 20

100

0.150

0.160

1.53

1.70 359

II Components Size 1280 (WE-PD): Order Code

Inductance (µH)

DCR (Ω)

DCR max. (Ω)

Rated Current (A)

Saturation Current (A)

744 770 01

1.20

0.005

0.007

12.00

16.60

744 770 10

10

0.019

0.022

6.20

6.60

DCR (Ω)

DCR max. (Ω)

Rated Current (A)

Saturation Current (A)

Size 4532 (WE-PD2): Order Code

Inductance (µH)

744 773 0

1

0.014

0.049

4.00

5.72

744 773 10

10

0.118

0.182

1.45

1.74

Size 5848 (WE-PD2): Order Code

Inductance (µH)

DCR (Ω)

DCR max. (Ω)

Rated Current (A)

Saturation Current (A)

744 774 10

10

0.078

0.100

2.20

2.50

Size 7850 (WE-PD2): Order Code

Inductance (µH)

DCR (Ω)

DCR max. (Ω)

Rated Current (A)

Saturation Current (A)

744 775 10

10

0.044

0.070

2.30

2.95

Size 1054 (WE-PD2): Order Code

Inductance (µH)

DCR (Ω)

DCR max. (Ω)

Rated Current (A)

Saturation Current (A)

744 776 10

10

0.028

0.060

2.98

3.24

Tab. 2.36: Specifications of some WE-PD series SMD storage chokes Saturation current Isat: Saturation current

360

The saturation current is the DC current at which the zero current inductance is reduced by a certain percentage. The percentage of inductance decrease is however not standardized and can be defined differently for each package type. In datasheets – and especially when comparing data from different manufacturers – very close attention must be paid

to the definition point. A printout of the measurement curve “Inductance versus DC pre­magnetization” is better still. Here the user can, check in detail how the inductor behaves in the case of overload or at the moment it switches on. An example for a standardized curve is shown in Figure 2.82.

Fig. 2.82: Standardized curve WE-PD Volt-µsec product As a result of their effective magnetic area Aeff, storage chokes and transformers can only be driven to a maximum value – the Volt-µsec product. The following calculation rule applies for the step-down controller to determine the necessary Vµsec product of storage chokes (Et):





(2.22)

with Et = Vµsec product, Uin(max) the maximum input voltage in Volts, Uout the output v­ oltage of the controller and f the switching frequency in Hz.

361

II Components With increasing switching frequency, the necessary Vµsec product of the storage choke becomes lower; however, with increasing input voltage it becomes higher. This relationship is again illustrated in Figure 2.83 below.

Fig. 2.83: Vµsec product with variable input voltage and different switching frequencies for the step-down controller

Practical tips: For some types in the WE-PD series there is also information on the Vµsec product in Table 2.35. If this information is missing, it can be read off from the measurement curve “Inductance versus DC current premagnetization”. Here you locate the inductance plateau where you read off the associated current (Imax) and residual inductance (Lres) and calculate the Vµsec product as follows: Et (Lx) = Lres · Imax(2.23)

With Lres in µH and Imax in A. Hard saturation

362

For ferrites, the saturation curve shows a very steep decline beyond a certain DC current value (“hard saturation”). For this reason it is recommended to reduce the Vµsec product (Et (Lx)) calculated in this way and therefore to then optimize inductor selection.

Ferrite storage chokes (e.g. WE-PD/ WE-TPC) Et (Lx) = 0.7 · Lrest · Imax > Etop(2.24) Et (Lx) = Vµsec product of the storage choke Etop = required µsec product of the circuit Iron powder core storage chokes, Superflux, WE-PERM etc. have a constant decrease in inductance due to the DC-premagnetization (soft saturation).

Soft saturation

As a rule:

Vµsec (Lx) = Lrest · Imax > Et(2.25)

Order Code

Inductance (µH)

ISAT (A)

Vμsec

744 770 10 744 770 122 744 770 147 744 770 20 744 770 222 744 770 30

10 22 47 100 220 1000

6.6 5 3 2.4 1.49 0.7

59 99 127 216 295 630

744 771 10 744 771 122 744 771 147 744 771 20 744 771 220 744 771 30

10 22 47 100 220 1000

5.6 3.77 2.6 1.7 1.2 0.5

50 75 110 153 238 450

744 777 10 744 777 122 744 777 147 744 777 20 744 777 222 744 777 30

10 22 47 100 220 1000

2.6 1.7 1.1 0.75 0.54 0.25

23 34 47 68 107 225

363

II Components Order Code

Inductance (µH)

ISAT (A)

Vμsec

744 778 10 744 778 122 744 778 147 744 778 20 744 778 222 744 778 30

10 22 47 100 220 1000

2.2 1.4 1 0.76 0.42 0.18

20 28 42 68 83 162

Tab. 2.37: ISAT measurement data and Vμsec data of the series WE-PD WE-PD core material parameters The WE-PD core material parameters are described by the following power loss formula: Steinmetz formula Pcore = 1.264 · 10–10 · f 1.274 · B1.769(2.26)

The corresponding curve is shown in Figure 2.84.

Fig. 2.84: WE-PD core material losses (NiZn power ferrite) (1 Gauss = 10–4 T)

364

Practical tip: In REDEXPERT total ac losses are based on measured data using a hard switching DC-DC converter with real switches and triangular current waveforms. Individual parts are tested over a frequency range of 10 kHz to 10 MHz, and duty cycles from 0.1 to 0.9. Using these numbers the well-known Steinmetz-equivalent parameters can be calculated, but only over narrow ranges. The parameters below are for 100 to 500 kHz at a duty cycle of 0.5. Especially at higher frequencies where the ac wire losses become more dominant, use REDEXPERT which consists of the dc loss, ac core loss and ac wire loss. WE-PD 1210 PCORE = 151446 ∙ f 0.687 ∙ (1.311 ∙〖10–4 ∙ ∆I)2.053 (W) WE-PD 1280 PCORE = 131881 ∙ f 0.724 ∙ (1.647 ∙〖10–4 ∙ ∆I)2.115 (W) WE-PD 1260 PCORE = 21143 ∙ f 0.820 ∙ (2.415 ∙〖10–4 ∙ ∆I)2.054 (W) WE-PD 1245 PCORE = 11522 ∙ f 0.933 ∙ (2.218 ∙〖10–4 ∙ ∆I)2.017 (W) WE-PD 1050 PCORE = 380 ∙ f 1.088 ∙ (5.807 ∙〖10–4 ∙ ∆I)2.057 (W) WE-PD 1030 PCORE = 90 ∙ f 1.433 ∙ (6.192 ∙〖10–4 ∙ ∆I)2.103 (W) WE-PD 7345 PCORE = 896 ∙ f 1.133 ∙ (5.245 ∙〖10–4 ∙ ∆I)2.038 (W) WE-PD 7332 PCORE = 550 ∙ f 1.142 ∙ (6.086 ∙〖10–4 ∙ ∆I)2.029 (W)

With DI = ripple current (peak-peak) in amperes due to the inductance.

(2.27)

(2.28)

(2.29)

(2.30)

(2.31)

(2.32)

(2.33)

(2.34)

365

II Components Power loss and temperature increase in the component Power loss Temperature increase

Now the power loss can be approximately determined, the question arises of the temperature rise of the component in operation. Measurement curves can be generated for the rise in temperature of the components with DC currents. Here the questions are to be resolved: • How was the component measured? – mounted on a PCB with a lot of copper (= cooling element!) or – only the component via a thin and poor heat-conducting connection • After what time was the temperature read off from the component (thermal time constant!) The following approximation formulas can be useful in the design phase; however they do not obviate measurement under real operating conditions. Determination of total power loss in the storage choke:

Copper losses

a) Copper losses:

Core material losses

(2.35)



b) Core material losses from empirical formulas This results in the total power loss (without further losses such as the skin effect etc. …):

REDEXPERT

(2.36)

Würth Elektronik’s online tool REDEXPERT has a highly accurate AC loss model based on measured data from variable frequency and variable duty cycle rectangular pulses. It can easily and quickly compare multiple inductors for losses and temperature rise. It is more accurate than using core loss calculations and dc copper losses because it also contains the winding and core ac losses. www.we-online.com/redexpert

366



Practical tip: Temperature increase in the component (large surface)



Temperature increase

(2.37)



with Ptot in (W) and the surface area A in (mm2)

Practical tip: The following approximation formulas for temperature increase can be specified for the WE-PD and WE-DD series storage chokes: WE-PD Size 6033, WE-DD „XS“:



(2.38)



WE-PD Size 6050:



(2.39)



WE-PD Size 7332, WE-DD „S“:





(2.40)

WE-PD Size 7345, WE-DD „M“:





(2.41)

WE-PD Size 1260, WE-DD „L“:





(2.42)

367

II Components WE-PD Size 1245:



(2.43)



WE-PD Size 1280, WE-DD „XL“:





(2.44)

WE-PD type „XXL“, WE-DD „XXL“:





(2.45)

• Suitable for switching regulator IC’s from Texas Instruments, ­Analog Devices, Maxim Integrated, etc.

3.5 WE-TPC, WE-HCI, WE-HCC P  ower Inductors WE-TPC series SMD storage chokes WE-TPC

Fig. 2.85: SMD Power Inductor WE-TPC The WE-TPC “Tiny Power Choke“ series of storage chokes is usually for applications for which the packing density and the package height is important. The new WE-TPC 2811 (744 028 xxx) is currently the smallest wire-wound inductor Würth Elektronik has produced with dimensions of 2.8 x 2.8 x 1.1 mm. 368

Order Code

L1 (µH)

IR 1 (A)

ISAT 1 (A)

RDC1 typ (mW)

RDC1 max (mW)

744 028 000 056

0.056

4.5

6

11.5

15

744 028 000 15

0.15

3

3.6

18

25

744 028 000 33

0.33

2.8

2.4

27

35

744 028 000 47

0.47

2.5

2

36

42

744 028 000 82

0.82

2

1.6

53

65

744 028 001

1

1.75

1.5

65

85

744 028 002

2.2

1.3

1

125

155

0.85

185

220

744 028 003

3.3

1

744 028 004

4.7

0.85

0.7

265

310

744 028 006

6.8

0.75

0.55

325

400

Tab. 2.38: Typical characteristics of the power inductor WE-TPC size 2811 These chokes are mostly used for switching controllers that have several outputs integrated in one IC, e.g. LTC3544B. This IC has 4 outputs at which different voltages, output currents and switching frequencies are adjustable.

Fig. 2.86: Circuit LTC3544B The core material used for the WE-TPC series is NiZn and is therefore suitable for switching frequencies up to 10 MHz. The saturation current is defined as a –35% inductance drop in relation to the zero current inductance, which is usually typical for inductances in this small package. The WE-TPC series is magnetically shielded and therefore especially suitable for switching controllers in mobile applications.

369

II Components SMD high-current inductor series WE-HCI WE-HCI

Fig. 2.87: SMD High current inductor WE-HCI Laptop computers and motherboards of modern computers are equipped with processors, whose clock frequencies may be 1 GHz or more. Processor manufacturers rely on low supply voltages in order to maintain losses in i­ntegrated circuits within tolerable limits and to attain the required switching speeds. These lie between 1 … 3.3 Volts depending on the processor generation. Components require large currents at the same time. Current inputs of up to 60 A per processor are not a rarity and cannot be ­handled by well known switching regulators. The multiphase ­switching converters fulfill the intelligent power management concept required. Because of the high switching frequency and the output current requirements, one needs only small inductances with high current capability and low losses. This technology is further described in applications. SMD high current inductors WE-HCI

MPP cores CoolMµ® cores

370

The WE-HCI series of SMD high-current inductor is suitable for these applications. The used core materials WE-PERM, WE-PERM2 and WE-Superflux are based upon a special high purity alloy consisting of various kinds of iron powder. These iron powders show significantly lower core losses than conventional iron powder cores. Core losses are nearly as low as those of MPP cores, however, the WE-Superflux material can be operated up to frequencies of over 1 MHz. At the same time, the costs of these cores are lower than those of MPP and CoolMµ® cores.

Core losses WE-HCI

Fig. 2.88: Core power loss of Superflux material WE-HCI Thermal aging

Thermal aging

Thermal aging can lead to the destruction of the organic binder used, especially at high operating temperatures with standard iron powder material. A thermal avalanche effect can consequently occur, which may finally destroy the core material. For standard iron powder materials and those not thermally treated for reasons of their production process, the rule of thumb applies that the maximum temperature of +125 °C measured at the component should not be exceeded for a prolonged period. There is no thermal aging for the material used here and now also available for ring cores – Superflux 200.

Superflux-200

Thanks to the polymer binder specially developed for this purpose, the core material is suitable for temperatures up to +200 °C. The maximum operating temperature can even be raised to +320 °C for special applications.

371

II Components

Fig. 2.89: Core material losses against operating hours (no thermal aging identifiable) WE-HCI core material WE-PERM has again been optimized for core loss compared with the WE-HCI series consisting of the core material WE-Superflux. The core material WE-PERM can be used for switching frequencies above 1 MHz. This series also uses rectangular flat wire, which proves to be a major advantage over conventional round wire designs in terms of AC resistance loss. WE-PERM

Fig. 2.90: Power core loss of the material WE-PERM of the WE-HCI 372

A flat wire (Figure 2.91) inside the choke is used in place of the conventional r­ ound wire. Flat wire windings offer the following advantages:

Flat wire

Fig. 2.91: Flat wire winding • Large wire surface – thus lower high-frequency losses (skin effect) • Low winding capacitance – hence higher self-resonant frequency • Low DC resistance – thus lower self-heating at high prolonged currents • High packing density and therefore smaller component size than c­ omparable chokes with round wire (Figure 2.92) • High operating temperature up to max. +150 °C

Fig. 2.92: Comparison of packing density for equal inductance between flat wire and round wire windings Through the combination of WE-Superflux, WE-PERM and WE-PERM2 low-loss core material and flat wire windings inside the core, a series of SMD high-current inductors is created with the following characteristic data:

SMD high-current inductors 373

II Components DCR ±10% (mΩ)

IR @ 50K (A)

Isat –30% [A]

Ve [cm3]

Ae [cm2]

Le [cm]

Windings (N)

Core Material

1.00

9.00

8.50

13.00

0.12

0.11

1.10

5.50

WE-Superflux

1.55

14.00

6.50

9.00

0.12

0.11

1.10

6.50

WE-Superflux

2.20

1.75

11.50

9.00

13.00

0.12

0.11

1.14

8.50

WE-Superflux

7x7x4

3.30

2.75

18.00

6.50

11.00

0.12

0.11

1.14

10.50

WE-Superflux

7050

7x7x5

8.50

6.80

32.50

4.00

5.20

0.16

0.11

1.42

15.50

WE-Superflux

744 314 101

7050

7x7x5

10.00

7.50

41.60

3.50

4.00

0.16

0.11

1.42

16.50

WE-Superflux

744 323 150

1030

10 x 10 x 3

1.50

1.28

6.60

12.00

18.00

0.27

0.19

1.46

4.50

WE-Superflux

744 323 220

1030

10 x 10 x 3

2.20

1.90

11.38

9.00

10.50

0.27

0.19

1.46

5.50

WE-Superflux

744 355 210 0

1040

10 x 10 x 4

1.00

0.78

3.25

16.00

20.00

0.33

0.19

1.71

4.50

WE-PERM

744 355 215 0

1040

10 x 10 x 4

1.50

1.10

5.10

14.00

17.00

0.33

0.19

1.71

5.50

WE-PERM

744 325 420

1050

10 x 10 x 5

4.20

3.30

6.80

11.00

14.00

0.31

0.18

1.73

8.50

WE-Superflux

744 325 550

1050

10 x 10 x 5

5.50

4.00

11.20

10.00

12.00

0.31

0.18

1.73

9.50

WE-Superflux

744 313 025

1335

13 x 13 x 3,5

0.25

0.22

0.72

24.00

60.00

0.44

0.25

1.74

1.50

WE-Superflux

744 313 068

1335

13 x 13 x 3,5

0.68

0.47

1.80

22.00

40.00

0.44

0.25

1.74

2.50

WE-Superflux

744 355 032 0

1350

13 x 14 x 5

3.20

2.20

5.50

16.00

15.00

0.60

0.32

1.89

6.50

WE-PERM

744 355 048 0

1350

13 x 14 x 5

4.80

3.55

10.50

11.00

13.00

0.60

0.32

1.89

7.50

WE-PERM

744 355 147

1365

13 x 14 x 6,5

0.47

0.41

0.67

30.00

50.00

0.71

0.35

2.04

2.50

WE-PERM

744 355 182

1365

13 x 14 x 6,5

0.82

0.64

0.90

27.00

35.00

0.71

0.35

2.04

3.50

WE-PERM

744 355 668 0

1890

18 x 18 x 9

6.80

5.60

4.10

18.50

27.00

1.06

0.43

2.47

10.50

WE-PERM2

744 355 611 00

1890

18 x 18 x 9

10.00

8.20

6.90

15.00

21.00

1.06

0.43

2.47

13.50

WE-PERM2

Order Code

Type

Size

Lo (µH)

Ln (µH)

744 310 115

7030

7x7x3

1.15

744 310 200

7030

7x7x3

2.00

744 311 220

7040

7x7x4

744 311 330

7040

744 314 850

Tab. 2.39: Specifications for the WE-HCI SMD high-current inductors Currently available sizes of the WE-HCI high current inductors: Size

LN (µH)

IR (A)

Isat (A)

DCR (mΩ)

7030

7.0 x 6.9 x 3.0

0.11 ~ 1.55

6.5 ~ 22.0

9.0 ~ 48.0

0.91 ~ 14.2

7040

7.0 x 6.9 x 3.8

0.18 ~ 3.50

6.0 ~ 21.0

7.0 ~ 32.0

1.10 ~ 19.5

7050

7.0 x 6.9 x 4.8

0.19 ~ 7.50

3.5 ~ 20.0

4.0 ~ 28.0

1.00 ~ 33.0

1030

10.6 x 10.6 x 2.8

0.18 ~ 1.90

9.0 ~ 22.0

15.0 ~ 50.0

0.82 ~ 11.4

1040

10.5 x 10.2 x 4.0

0.13 ~ 3.00

8.0 ~ 25.0

8.0 ~ 60.0

0.58 ~ 14.1

1050

10.5 x 10.2 x 4.7

0.14 ~ 4.10

10.0 ~ 25.0

12.0 ~ 58.0

0.51 ~ 10.3

1335

12.9 x 12.8 x 3.3

0.22 ~ 2.45

12.0 ~ 24.0

14.0 ~ 60.0

0.75 ~ 8.10

1350

13.0 x 12.8 x 4.7

0.17 ~ 7.80

8.5 ~ 29.0

10.0 ~ 60.0

0.5 ~ 14.1

1365

13.2 x 12.8 x 6.2

0.17 ~ 16.0

6.0 ~ 32.0

6.5 ~ 65.0

0.35 ~ 24.7

1890

18.3 x 18.2 x 8.9

0.72 ~ 40.0

6.8 ~ 41.5

7.0 ~ 65.0

0.54 ~ 33.5

Tab. 2.40: Sizes of the WE-HCI high current inductors The values specified in the data sheet are: • Open-circuit inductance L0. tested at 100 MHz with 0.25 VAC • Inductance rating LN at current rating IN and self-heating < +50 °C 374

• Min. inductance at max. current IMAX and self-heating < +100 °C • Min. DC windings resistance DCRMAX at Ta = +25 °C Figure 2.93 shows the typical behavior of WE-HCI high current inductors using the example of the 0.82 µH choke 744 355 182 of component size 13.2 x 12.8 x 6.2 mm.

Fig. 2.93: Inductance curve and self-heating against current (744 355 182) The component has an inductance of 0.65 μH at the specified rated current of 27 A and demonstrates typical self-heating of +50 °C. The inductance is very stable under current load; the limiting factor is the self-heating of the component. Even at a current load of 37 A the inductance does not drop more than 30% from the open-circuit inductance. The self-heating is well over +100 °C, however. Conclusion: The WE-HCI series represents a highly dynamic and robust series of storage chokes, especially suited for use in high-current switching converters and multiphase or polyphase converters. Additional application areas are in high-current interference suppression chokes and as a replacement for rod core chokes. SMD-High Current Cube Inductor WE-HCC The WE-HCC series is available in two different core materials, iron powder and ferrite. The WE-HCC is optimized for power supplies running at high switching frequencies (up to 5 MHz) with high ripple current. Due to the ferrite material the core losses are significantly lower compared to iron powder material which is mainly used in high current inductors designs. On the other hand, if it is required to have an excellent saturation behavior over temperature for filter circuits or for power supplies, the WE-HCC is a good choice as well. With a maximum operating temperature of +125 °C the WE-HCC is able to handle 27 A DC current without heat sink. 375

II Components WE-HCC

Fig. 2.94: WE-HCC high current cube inductor Order Code

A (mm)

B (mm)

C (mm)

D (mm)

E (mm)

F (mm)

L0 ±20% (µH)

LN typ. (µH)

DCR ±10% (mW)

IR (A)

Isat (A)

Size

744 334 003 0

8.4

7.9

7.2

2.3

1.50

2.75

0.30

0.30

1.40

20.5

36.0

8070

744 334 033 0

8.4

7.9

7.2

2.3

1.50

2.75

3.30

0.40

6.50

14.0

8.5

8070

744 333 002 2

10.9

10.0

9.3

3.0

1.60

3.50

0.22

0.22

0.60

21.5

60.0

1090

744 333 100 0

10.9

10.0

9.3

3.0

1.60

3.50

10.0

5.50

20.7

9.0

8.0

1090

744 332 002 2

12.1

11.4

9.5

3.5

2.00

3.50

0.22

0.22

0.53

27.0

60

1210

744 332 100 0

12.1

11.4

9.5

3.5

2.00

3.50

10.0

7.50

14.40

9.0

10

1210

744 331 002 2

12.1

11.4

9.5

3.5

2.15

3.90

0.22

0.21

0.51

27

85

1210

744 331 047 0

12.1

11.4

9.5

3.5

2.15

3.90

4.7

4.50

8.9

11.5

33

1210

Tab. 2.41: Typical characteristics of the high current cube inductors WE-HCC

Due to the design and material a current of 60 A and more can be achieved which is more than enough for the most power supply designs. Especially for multiphase designs, it is getting more important for certain switching topologies (currents sense) to have a tight DCR tolerance.

376

Attenuation 0 –2 –4

Insertion loss (dB)

–6 –8 –10 –12 –14 744 331 047 0 IRMS = 0 A

–16

744 331 047 0 IRMS = 11 A

–18 –20 0

10

20

30

40

50

60

70

80

90 100 110 120 130 140 150 160 170 180 190 200 210 220 230 240 250

f (MHz)

Fig. 2.95: Insertion loss of the WE-HCC 744 331 047 0 iron powder with and without DC load The WE-HCC (iron powder) is designed for extreme high current applications where it is important to have an excellent saturation behavior. High current filter applications like motor units, frequency converters and inverters or Class D amplifiers will benefit of this characteristic due to the temperature-resistant core material. As shown in the graph, the frequency response of the WE-HCC (iron powder) is stable over DC current without any performance decreases. The black curve describes the attenuation over frequency without DC current; the red curve shows the attenuation with 11 A DC load. This series is perfect for high current filter application with a need for high a­ ttenuation in the frequency range from 5 MHz to 100 MHz – even over DC current.

377

II Components 3.6 WE-HCF SMD High Current Inductor WE-HCF

Round wire Litz wire Flat wire Fig. 2.96: WE-HCF SMD High Current Inductors with round, litz or flat wire coils. Order Code

L1 (μH)

LR (μH)

IR 1 (A)

ISAT 1 (A)

RDC 1 (mΩ)

7443630070

0.7

0.7

32

75

0.83

7443630140

1.4

1.39

31.5

60

1.08

7443630220

2.2

2.19

28

52

1.5

7443630310

3.1

3.07

26

45

2.09

7443630420

4.2

4.14

24

38

3.04

7443630550

5.5

5.4

22

33

4

7443630700

7 6.

83

21

30

5.61

7443630860

8.6

8.46

17

25

7.19

7443631000

10

9.8

16

23

7.96

7443631500

15

14.7

14

21

8.7

7443632200

22

21.1

12.5

15

10.65

7443633300

33

19.4

12

11

11.4

7443634700

47

12.45

12

8.5

12.2

L1: Inductance; LR: Rated Inductance; IR 1: Rated Current; ISAT 1: Saturation Current; RDC 1: DC Resistance

Tab. 2.42: Typical specification of some WE-HCF Inductors

378

744 363 330 0

40

744 363 220 0

35

744 363 150 0 744 363 100 0

30

744 363 055 0

25

744 363 007 0

20

80

temperature (ºC)

inductance (µH)

100

744 363 470 0

45

60

40

744 363 470 0 744 363 100 0

15 10

744 363 055 0

20

744 363 022 0

5

744 363 007 0 0

0 0

1

10

100

0

5

10

15

20

25

30

35

40

45

current (A)

current (A)

Fig. 2.97: Typical characteristics of high current WE-HCF inductors The WE-HCF (core material MnZn) is optimized for boost applications running at high switching frequencies with high ripple current. Available with flat, round or litz wire to optimize the DC and AC losses. Furthermore the core losses are significantly lower compared to the iron powder material. The WE-HCF fits perfectly for high switching frequencies up to 5 MHz. The maximum operating temperature is +125 °C and the WE-HCF is able to handle 36 A DC current without any external heat sink. The WE-HCF 2013 is suitable for filters in Class D amplifiers, since efficiency, reliability and the resulting sound quality are influenced in a positive manner.

3.7 WE-PD HV, WE-PD2 HV, WE-TI HV – High Voltage Inductors High voltage inductor WE-PD HV WE-PD2 HV WE-TI HV Size:

L: I R: ISAT: RDC:

WE-PD HV

WE-PD2 HV

WE-TI HV

7.3 x 4.5 mm 10 x 6 mm 12 x 10 mm 0.22 ~ 3.3 mH 0.26 ~ 1.3 A 0.25 ~ 2.0 A 0.3 ~ 5.5 Ω

7.8 x 5 mm 10 x 5.4 mm

8.0 x 9.5 mm

0.56 ~ 2.2 mH 0.15 ~ 0.41 A 0.2 ~ 0.38 A 1.7~ 6.0 Ω

0.22 ~ 2.2 mH 0.32 ~ 0.9 A 0.32 ~ 1.3 A 0.5 ~ 3.9 Ω

Tab. 2.43: Illustration and specification of some WE-PD HV, WE-PD2 HV, WE-TI HV Würth Elektronik offers three families for a total of six series of power inductors in shielded and unshielded surface mount technologies along with unshielded throughhole technology that have been designed specifically to operate safely in the presence of differential voltages up to 400 VDC. Unlike the inductors used for filtering, these energy-storing inductors can be used in offline buck or buck-boost regulators circuits where they endure differential voltages

379

II Components equal to or exceeding the peak input voltage. For universal AC input (85–265 Vac) voltages after rectification, this approaches 400 Vdc. For safe operation this requires a minimum spacing between terminals of 1.6 mm and special consideration of the wire crossovers. The following families of inductors are guaranteed to operate properly with up to 400 VDC according to Würth Elektronik Standard 1516: WE-PD HV, WE-PD2 HV and WE-TI HV. Furthermore, this guarantee remains valid even after three reflow processes, making Würth Elektronik the first manufacturer in the world to provide such a robust guarantee.

3.8 WE-PFC Power Factor Correction Choke WE-PFC Power factor ­correction

Fig. 2.98: The many different packages of WE-PFC inductors WE-PFC Power Factor Correction inductors are available a wide variety of inductances and up to 250 W. Designed to operate over –40 °C to +125 °C with universal input or European input voltages. The wide selection of package styles include EER28, EE20/10/6, RM10, RM12, RM14, EFD25, EFD30 and PQ38/11 making them suitable for most applications. Order Code

L1 (μH)

n

ISAT 1 (A)

RDC1 max (mΩ)

760804110

250

44:5

5.6

150

200

760802112

375

12:1

5.4

360

180

760801020

500

12:1

4

360

180

760801030

650

10,71:1

3.9

480

260

760801130

750

90:8

3.9

720

320

L1: Inductance; n: Turns Ratio; ISAT 1: Saturation Current; RDC1 max: DC Resistance 1; RDC2 max: DC Resistance 2

Tab. 2.44: Typical characteristics of RM10 series of WE-PFC inductors

380

RDC2 max (mΩ)

800 760 801 130 760 801 030 760 801 020 760 802 112 760 804 110

700 600

Inductance (µH)

500 400 300 200 100 0 0

1

2

4

3

5

6

7

8

Current (A)

Fig. 2.99: Typical performance of RM-10 series of WE-PFC inductor Input voltage: 85–265 VAC Power (W)

Order Code

Size

L (μH)

IR (A)

Isat (A)

25

760 800 080

EE20/10/6

1800

0.3

1.0

75

760 801 130

RM10

750

0.9

3.9

75

760 801 131

EFD30

750

0.9

3.3

100

760 802 122

RM12

450

1.2

5.6

125

760 803 200

EER28

150

1.5

12.0

150

760 804 310

RM12

300

2.4

7.4

150

760 806 302

PQ38/11

300

3.0

5.1

200

760 805 410

RM14

225

2.4

12.4

250

760 806 400

RM14

180

3.0

13.2

Input voltage: 195–265 VAC Power (W)

Order Code

Size

L (μH)

IR (A)

Isat (A)

75

760 801 030

RM10

650

0.4

3.9 2.6

75

760 801 031

EFD25

650

0.4

75–100

760 801 020

RM10

500

0.6

4.0

75–100

760 801 021

EFD30

500

0.6

4.2

100–150

760 802 112

RM10

375

0.8

5.4

100–150

760 802 113

EFD30

375

0.8

4.0

150–200

760 804 110

RM10

250

1.1

5.6

150–200

760 804 111

EFD30

250

1.1

4.8

200–250

760 805 210

RM12

200

1.4

8.4

200–250

760 805 211

EFD30

200

1.4

6.0

250

760 806 200

RM12

150

1.4

11.0

250

760 806 201

EFD30

150

1.4

6.8

Tab. 2.45: Application recommendations 381

II Components 3.9 WE-EHPI Energy Harvesting Coupled Inductor WE-EHPI

Fig. 2.100: WE-EHPI Energy harvesting Coupled Inductor The WE-EHPI coupled inductor series is specifically design to work with energy harvesting ICs like the LT3108 in applications where remote sensors are powered from energies in their own environment. Small footprint, low DCR, multiple turns ratios plus welded contacts make for a reliable and optimized inductor. Order Code

L1 (µH)

74488540070

7

74488540120

13

74488540250

25

Tol. L ±20%

L2 (µH)

n

IR 1 (A)

ISAT 1 (A)

RDC1 typ (Ω)

RDC2 typ (Ω)

70000

1 : 100

1.9

1.3

0.085

205

33000

1 : 50

1.7

1

0.09

135

10000

1 : 20

1.5

0.7

0.2

42

L1: Inductance 1; Tol. L: Inductance (Tol.); L2: Inductance 2; n: Turns Ratio; IR 1: Rated Current; ISAT 1: Saturation Current; RDC1 typ: DC Resistance 1; RDC2 typ: DC Resistance 2

Tab. 2.46: Typical characteristics of WE-EPHI inductors

The latest ICs can work in the milliwatt range. By using these new technologies, the power supply can be specifically developed so your batteries can be continuously charged or even replaced. Energy can be harvested from the temperature difference of the environment with the aid of a Peltier element and a boost converter based on the LTC3108 as shown below.

382

Fig. 2.101: Typical application circuit

3.10 WE-DD double chokes The WE-DD series of double chokes with two separate windings is available to expand the standard spectrum of the WE-PD series of storage chokes. WE-DD

Fig. 2.102: WE-DD double chokes in various sizes Size S, M

Size L, XL (744 874 xxx / 744 873 xxx)

383

II Components Size L, XL (744 871 xxx / 744 870 xxx)

Size XXL

Size

A (mm)

B (mm)

C (mm)

D (mm)

E (mm)

S

7.3

4

1.0

2.7

4.0

M

7.3

4.8

1.0

2.7

4.0

L

12.5

6.5

1.5

4.9

7.3

XL

12.5

8.5

1.5

4.9

7.3

XXL

12.5

10.5

1.5

4.9

7.3

Tab. 2.47: Dimensions of WE-DD chokes

7.3 x 7.3 x Height 744 878 xxx 4.0 mm 744 877 xxx 4.8 mm

12 x 12 x Height 744 871 xxx 6.5 mm 744 870 xxx 8.5 mm

12 x 12 x Height 12 x 12 x Height 744 874 xxx 6.5 mm 744 870 9xxx 10.5 mm 744 873 xxx 8.5 mm

1

4

1

2

1

2

1

2

2

3

4

3

4

3

4

3

Fig. 2.103: Pin layout of the double chokes WE-DD Features: • Two separate windings on a common ferrite core • Available in a 1 : 1 winding (standard), but also in other winding ratios (customer-specific) 384

• Bifilar winding for minimal leakage inductance / high coupling factor (k ~ 0.985 ... 0.990) or separate layer winding with increased leakage inductance • 744 874 xxx/744 873 xxx: Low leakage, bifilar winding • 744 871 xxx/744 870 xxx: High leakage, seperate layer winding • Operating voltage up to 80 VDC • Isolation voltage 100 VDC max Size Order Code

7x7

12x12

12x12

low leakage

high leakage

744 878

744 877

744 874

744 873

744 871

744 870

S

M

L

XL

L

XL

XXL

straight

straight

straight

straight

crossed

crossed

crossed

Coilcraft (MSD Serie)

–

(MSD7342)

–

–

MSD1260

MSD1278

–

Pin Layout

–

crossed

–

–

crossed

crossed

–

Coiltronics (DRQ Serie)

(DRQ73)

(DRQ74)

–

–

DRQ125

DRQ127

–

Pin Layout

crossed

crossed

–

–

crossed

crossed

–

Size Pin Layout

744 870 9

Tab. 2.48: Crossreference Applications: • SEPIC switching controllers (functional principle – see Chapter I/6.4) • CUK switching controller (switching controller with negative output voltage) • Switching controllers with second, unregulated output voltage (auxiliary voltage) Operating temperature: The ambient temperature of the WE-DD double choke under full rated current load must be between –40 °C and +85 °C. The self-heating of the component must be taken into account at higher ambient temperatures in order that the permissible solder joint temperature is not exceeded or the wire insulation damaged. The wire used can withstand a temperature of up to +150 °C. The ferrite core itself may be used over a temperature range (of approx. –50 °C to +125 °C (far away from the Curie temperature)). However, in this case, the inductance tolerance limits may be exceeded due to the temperature dependence of permeability.

385

II Components Datasheet definitions information: Electrical characteristics (744 870 220) Characteristics Test ­conditions

Tole­ rance

L 1. L 2

22.0

µH

±20%

DC resistance (per winding)

DCR1.2 typ

0.062

W

typ.

DC resistance (per winding)

DCR1.2 typ

0.080

W

max.

Inductance (per winding)

1 kHz/0.25 V

Value

Rated current (per winding)

DT = 40 K

IN1. IN2

2.45

A

max.

Saturation ­current (per winding)

DL/Lo = –10%

Isat

5.20

A

typ.

SRF

SRF

10.0

MHz

typ.

Rated voltage

UDC

80.0

V

max.

Tab. 2.49: Electrical characteristics of the shielded power transformers WE-DD Rated current

Rated current: In contrast to other double chokes on the market with datasheet specifications unsuitable in practice, we use a very conservative method of determining the rated current. The self-heating for the pair of windings passing maximum current should not sum to more than +40 °C. So the rated current is determined for each winding, which passes current on its own, leading to a temperature increase of up to +20 °C and is specified as IN1 and IN2 respectively. If both windings pass their rated current at the same time, this leads to self-heating totaling +40 °C.

DC resistance

DC resistance: Correspondingly, the specification for the winding resistance of the two individual windings is found from individual measurements. Attention: Please compare the datasheet specifications carefully – often the parallel configuration of the two windings is found in the literature as rated current / DC resistance, which suggests a higher rated current and lower DC resistance. In practice this is of course, not the application for this series of chokes!

Saturation current

386

Saturation current: In the case of the double choke with the same inductance, it is sufficient if just one of the windings carries the saturation current. The second winding is inevitably reduced in its inductance. The value is identical for the same inductance; the saturation current per winding is specified for dissimilar inductance values.

For all other calculations, revert to the equations and information in the section covering WE-PD (such as core losses, estimation of self-heating, etc.).

3.11 WE-DPC SMD Dual Power Choke WE-DPC

Fig. 2.104: WE-DPC SMD Dual Power Choke Order Code

L1 (µH)

Tol. L1

IR 1 (A)

ISAT 1 (A)

RDC1 typ (mΩ)

RDC1 max (mΩ)

7448841010

1

±30%

2.9

5

32

45

7448841015

1.5

±30%

2.7

3.7

48

68

7448841022

2.2

±30%

2.5

3.2

50

72

7448841033

3.3

±30%

1.9

2.6

84

120

7448841047

4.7

±30%

1.6

1.9

102

145

7448841068

6.8

±30%

1.4

1.7

168

240

7448841082

8.2

±30%

1.3

1.6

180

265

7448841100

10

±20%

1.2

1.5

210

300

7448841150

15

±20%

0.9

1.4

325

465

L1: Inductance 1; Tol. L1: Inductance 1 (Tol.); IR 1: Rated Current 1; ISAT 1: Saturation Current; RDC1 typ: DC Resistance 1; RDC1 max: DC Resistance 1

Tab. 2.50: Specification of some WE-DPC power chokes The WE-DPC SMD Dual Power Choke series are miniature inductors with two identical windings in a magnetically shielded package. Ideal for flyback, SEPIC or as a second output on buck converters. They can also be used with series or parallel connection in buck or boost applications.

387

II Components 3.12 WE-MCRI – SMD Molded Coupled Inductor WE-MCRI Molded coupled inductor

Fig. 2.105: WE-MCRI – SMD Molded Coupled Inductor Order Code

L1 (µH)

7448990010

1

7448990015

1.5

7448990022

2.2

7448990033

3.3

7448990047

4.7

7448990068

6.8

7448990082

8.2

7448990100

10

7448990150

15

7448990220

22

7448990330

33

7448990470

47

n

IR 1 (A) 12.5 7.5

1:1

5 4.5 3.4 2.3

ISAT1 (A)

RDC1 typ (mΩ)

fres 1 (MHz)

43.5

4.5

35

34

8.8

29

29.5

10.5

20

28.2

23.5

19

24.2

36

17.5

21.2

46

13

18.5

56

9.5

17

62

9

9.5

78

8.5

8.1

95

6.5

6.8

130

5.5

6

216

4.5

L1: Inductance 1; n: Turns Ratio; IR 1: Rated Current 1; ISAT1: Saturation Current 1; RDC1 typ: DC Resistance 1; fres 1: Self Resonant Frequency

Tab. 2.51: Illustration and specification of some WE-MCRI coupled inductors The WE-MCRI are made from a molded composite core material that provides high saturation current and compact design. The automated winding process has very accurate wire positioning which results in high consistency. The result is lower leakage inductance and higher rated currents than comparable components. Magnetically shielded with a 1:1 turns ratio they can operate from 40 °C to +125 °C. 388

3.13 WE-MTCI SMD Multi-Turn Ratio Coupled Inductor WE-MTCI

Fig. 2.106: WE-MTCI SMD Multi-Turn Ratio Coupled Inductor Order Code

L1 (µH)

L2 (µH)

n

IR 1 (A)

ISAT1 (A)

RDC1 typ (mΩ)

744889015100

10

22.5

1:1.5

0.95

1.5

349

744889020100

10

40

1:2

0.95

1.5

358

744889030100

10

90

1:3

0.95

1.5

363

744889015220

22

49.5

1:1.5

0.6

1

662

744889020220

22

88

1:2

0.6

1

712

744889030220

22

198

1:3

0.6

1

732

744889015330

33

74.25

1:1.5

0.45

0.75

1338

744889020330

33

132

1:2

0.45

0.75

1383

744889030330

33

297

1:3

0.45

0.75

1466

L1: Inductance 1; L2: Inductance 2; n: Turns Ratio; IR 1: Rated Current ; ISAT 1: Saturation Current; RDC1 typ: DC Resistance 1

Tab. 2.52: Typical specification of some WE-MTCI coupled inductors The WE-MCTI is the smallest coupled inductor available with multiple turns ratios. It’s an ideal solution for buck or boost converters that need a second unregulated output at a different voltage. Can also be used in an auto-transformer configuration or even flyback.

389

II Components 3.14 WE-DPC HV, WE-CPIB HV, WE-TDC HV SMD Coupled Inductors WE-DPC WE-CPIB WE-TDC High voltage

Fig. 2.107: WE-DPC HV, WE-CPIB HV, WE-TDC HV SMD Coupled Inductors Order Code

L1 (µH)

L2 (µH)

7448845047

4.7

4.7

7448845100

10

10

7448845220

22

22

n

1:1

IR 1 (A)

IR 2 (A)

ISAT1 (A)

ISAT2 (A)

RDC1 typ (mΩ)

1.45

1.45

2.2

2.2

105

1.2

1.2

1.55

1.55

200

0.8

0.8

1

1

430

7448845330

33

33

0.65

0.65

0.9

0.9

660

7448845470

47

47

0.55

0.55

0.73

0.73

800

L1: Inductance 1; L2: Inductance 2; n: Turns Ratio; IR 1: Rated Current 1; IR 2: Rated Current 2; ISAT1: Saturation Current 1; ISAT2: Saturation Current 2; RDC1 typ: DC Resistance 1

Tab. 2.53: Illustration and specification of some WE-CPIB HV coupled inductors This series of High Voltage (HV) coupled inductors are specially designed to provide up to 2 kVac isolation and functional isolation for a working voltage of 250 Vrms. All are magnetically shielded to improve EMI. They are suitable for a multitude of applications from non isolated buck, boost, SEPIC to isolated flyback to or even as an isolated second output.

4 Power Magnetics – Transformers 4.1 WE-FLEX & WE-FLEX+ Transformers Transformer WE-FLEX

390

The WE-FLEX and WE-FLEX+ series transformers (Figure 2.108) are especially well suited for DC-DC converters in the lower power range. As a result of their winding structure, consisting of six individual windings, each with the same number of turns, and the various air gaps available, the Flex transformers are very flexible in their application. Table 2.54 shows the different sizes, and Table 2.55 the most important electrical parameters.

WE-FLEX

Fig. 2.108: WE-FLEX transformers for switch mode power supplies

Type

A (mm)

B (mm)

C (mm)

D (mm)

E (mm)

ER11/5

12.9

9.2

13.0

0.7

6.2

ER14.5/6

16.3

12.0

16.8

0.7

7.4

EFD15

17.5

16.0

22.1

0.7

8.3

EFD20

21.0

21.0

29.5

0.7

10.8

Tab. 2.54: Mechanical dimensions of the WE-FLEX series of transformers

391

II Components ER 11/5 Part without air gap for buck derived topologies like forward and push-pull converters

Order Code

Lbase (µH)

Rated Currentbase (A)

Voltµsecbase (µVs)

DCRbase (mΩ)

LS base (µH)

749 196 101

198.6

0.55

32.9

344

0.21

Parts with air gap for storage applications like flyback converters and storage chokes

Order Code

Lbase (µH)

Rated Currentbase (A)

Isatbase (A)

DCRbase (mΩ)

LS base (µH)

749 196 111

27.4

0.55

0.22

344

0.21

749 196 121

14.7

0.55

0.54

344

0.21

749 196 131

10.9

0.55

0.73

344

0.21

749 196 141

8.5

0.55

0.96

344

0.21

ER 14.5/6 Part without air gap for buck derived topologies like forward and push-pull converters

Order Code

Lbase (µH)

Rated Currentbase (A)

Voltµsecbase (µVs)

DCRbase (mΩ)

LS base (µH)

749 196 201

140

0.95

48.3

159

0.17

Parts with air gap for storage applications like flyback converters and storage chokes

Order Code

Lbase (µH)

Rated Currentbase (A)

Isatbase (A)

DCRbase (mΩ)

LS base (µH)

749 196 211

21.6

0.95

0.36

159

0.17

749 196 221

11.6

0.95

0.84

159

0.17

749 196 231

8.3

0.95

1.2

159

0.17

749 196 241

6.6

0.95

1.55

159

0.17

EFD 15 Part without air gap for buck derived topologies like forward and push-pull converters

392

Order Code

Lbase (µH)

Rated Currentbase (A)

Voltµsecbase (µVs)

DCRbase (mΩ)

LS base (µH)

749 196 301

153.8

0.97

39.8

140

0.13

Parts with air gap for storage applications like flyback converters and storage chokes

Order Code

Lbase (µH)

Rated Currentbase (A)

Isatbase (A)

DCRbase (mΩ)

LS base (µH)

749 196 311

23.3

0.97

0.33

140

0.13

749 196 321

14.2

0.97

0.63

140

0.13

749 196 331

9.3

0.97

1.09

140

0.13

749 196 341

7.9

0.97

1.33

140

0.13

EFD 20 Parts without air gap for buck derived topologies like forward and push-pull converters

Order Code

Lbase (µH)

Rated Currentbase (A)

Voltµsecbase (µVs)

DCRbase (mΩ)

LS base (µH)

749 196 500

87.1

1.91

65.6

30

0.18

749 196 501

196

1.7

98.4

71.1

0.24

Parts with air gap for storage applications like flyback converters and storage chokes

Order Code

Lbase (µH)

Rated Currentbase (A)

Isatbase (A)

DCRbase (mΩ)

LS base (µH)

749 196 510

9.9

1.91

1.17

30

0.18

749 196 520

5.3

1.91

2.53

30

0.18

749 196 530

4.3

1.91

2.91

30

0.18

749 196 540

3.4

1.91

4.18

30

0.18

749 196 511

22.3

1.7

0.49

71.1

0.24

749 196 521

12.0

1.7

1.73

71.1

0.24

749 196 531

9.7

1.7

2.2

71.1

0.24

749 196 541

7.6

1.7

2.46

71.1

0.24

Tab. 2.55: Electrical data of the WE-FLEX series of transformers

393

C

II Components

E

Sizes

A (mm)

B (mm)

C (mm)

D (mm)

E (mm)

ETD29

37.21

35.56

31.75

4.2

0.8

ETD34

40.39

39.62

31.49

4.2

0.8

ETD39

45.72

44.45

33.78

4.2

0.8

B

D

A

Tab. 2.56: Mechanical dimensions of the WE-FLEX transformers +

ETD29 Parts without air gap for buck derived topologies like forward and push-pull converters

Order Code

Lbase (µH)

IN base (A)

Isatbase (A)

DCRbase (mΩ)

LS base (µH)

749 197 101

284

2.2

0.04

45

0.45

Parts with air gap for storage applications like flyback converters and storage chokes

Order Code

Lbase (µH)

IN base (A)

Isatbase (A)

DCRbase (mΩ)

LS base (µH)

749 197 111

75.1

2.2

0.40

45

0.45

749 197 121

46.3

2.2

0.83

45

0.45

749 197 131

24.3

2.2

1.83

45

0.45

749 197 141

15.0

2.2

3.10

45

0.45

ETD34 Parts without air gap for buck derived topologies like forward and push-pull converters

394

Order Code

Lbase (µH)

IN base (A)

Isatbase (A)

DCRbase (mΩ)

LS base (µH)

749 197 201

374.4

2.5

0.04

51

0.32

Parts with air gap for storage applications like flyback converters and storage chokes

Order Code

Lbase (µH)

IN base (A)

Isatbase (A)

DCRbase (mΩ)

LS base (µH)

749 197 211

113.8

2.5

0.33

51

0.32

749 197 221

69.4

0.67

0.33

51

0.32

749 197 231

36.1

1.58

0.33

51

0.32

749 197 241

22.0

2.91

0.33

51

0.32

ETD39 Parts without air gap for buck derived topologies like forward and push-pull converters

Order Code

Lbase (µH)

IN base (A)

Isatbase (A)

DCRbase (mΩ)

LS base (µH)

749 197 301

326.7

3.2

0.04

34

0.35

Parts with air gap for storage applications like flyback converters and storage chokes

Order Code

Lbase (µH)

IN base (A)

Isatbase (A)

DCRbase (mΩ)

LS base (µH)

749 197 311

128.5

3.2

0.32

34

0.35

749 197 321

77.3

3.2

0.68

34

0.35

749 197 331

39.4

3.2

1.65

34

0.35

749 197 341

23.7

3.2

3.25

34

0.35

Tab. 2.57: Electrical Data for WE-FLEX transformers

Fig. 2.109: Schematic representation of the WE-FLEX transformers By using the appropriate connections of the windings on the circuit b­ oard allows v­ arious inductances and transformers with different turns ratios to be generated ­(Figure  2.110). 395

II Components

Fig. 2.110: The use of a Flex transformer with a turns ratio 1 : 3 with ­layout ­recommendations When calculating the resulting currents and inductances, it must be kept in mind that the six windings are wound on the same core – so they are magnetically coupled. Normally, when connecting two d­ iscrete inductances in series, the inductances are added (Equation 2.46). Connected in parallel, the reciprocal values of the inductances add together (Equation 2.47), i.e. connecting two equal inductances in parallel produces half the inductance.



(2.46)







(2.47)

However because the windings share the same core the results are different. Windings in parallel maintain the same inductance and those in series increase by the square of the number of windings.

396

Fig. 2.111: Parallel and series connection of the WE-FLEX transformers

Lbase inductance of a winding

Lbase

Connecting windings on a common core means adding the number of turns. These are squared in the calculation of the resulting inductance. As the number of turns of the single windings for WE-FLEX is identical, the resulting inductance is proportional to that are on square of the number of windings connected in series (Equation 2.48).



Lseries = Lbase · n2(2.48)

For parallel connection, the number of turns stays the same; only the conductor cross-section changes. So with parallel connection there is no change in inductance (Equation 2.49).



LParallel = Lbase(2.49)

Definition of Lsatbase To determine Isatbase all six windings were connected in parallel and the inductance measured as a function of current. The saturation current determined in this way is the total saturation current of the component. Isatbase is then the measured saturation current divided by six.



Isatbase

(2.50)



Consequently, the total saturation current of the transformer is six t­imes Isatbase. This total saturation current can now be distributed over the current-carrying windings. If, for example, three windings are connected in series, then six times Isatbase can be distributed over these three windings. Hence, each of these windings may carry current of twice Isatbase.





(2.51)

In the case of parallel connection of three windings, these can also carry twice Isatbase. For parallel connection, the individual currents add together to produce the total current, six times Isatbase. The following r­ ules generally apply:

Isat parallel = 6 · Isatbase(2.52) 397

II Components nI number of current-carrying windings Rated current

INbase

In contrast, the rated current, which is a property of the wire diameter, cannot be distributed to other windings. The resulting rated current for series and parallel connections is given by the following formulas:

LNseries = INbase(2.53)



LNParallel = n · INbase(2.54)

The saturation current is not crucial when it comes to dimensioning transformers as forward converters or push-pull converters, but rather the voltage-µs product. The voltage-µs product or V-µs product is proportional to the number of turns, so that the V-µs product from Table 2.57 has to be multiplied by the number of windings connected in series. Recall that Udt = Ldi. The windings of the FLEX transformers are tested against each other with 500 VDC. There is no additional insulation layer between the i­ndividual windings, so they can only be used in the low voltage range (SELV: < 60 VDC). Examples of how the suitable FLEX transformers are selected are d­ escribed in chapter III/8 on the various switching controller ­circuits and supported by the online software tool, REDEXPERT.

4.2 WE-FLEX HV Flexible Transformer High Voltage WE-FLEX HV High voltage

Fig. 2.112: WE-FLEX HV Flexible Transformer High Voltage

398

Order Code

LBASE (µH)

749196348

7.9

749196338

9.3

749196328

14.2

IR BASE (A)

∫ UdtBase (V-μs)

RDC BASE (mΩ)

LS Base (µH)

ISAT BASE (A) 1.41 1.13

0.83

61.3

145

0.13

0.72

749196318

23.3

0.43

749196308

153.8

0.06

LBASE: Inductance Base; IR BASE: Rated Current Base; ∫UdtBase: Voltage-µSecond Base; RDC BASE: DC Resistance Base; LS Base: Leakage Inductance Base; ISAT BASE: Saturation Current Base

Tab. 2.58: Specification of some EFD15 size WE-FLEX HV ­transformers WE-FLEX HV Flexible Transformer High Voltage series is specifically designed to provide up to 1.5 kVac isolation voltage between the individual windings. EFD size parts provide Basic insulation for a working voltage of 250 Vrms when they are connected appropriately. The windings of flexible transformers can be connected in more than 375 possible transformer configurations and 125 choke configurations providing for quick prototyping possibilities.

4.3 WE-PoE Power-over-Ethernet Transformers

Transformer WE-PoE

A DC-DC converter is also required for the power supply in Power over Ethernet. This has to sat­isfy the following tasks: • PoE protocol for detection and power classification • Voltage regulation to the required output voltage • Isolation of 1.5 kVac in accordance with IEC 60950 and IEC 62368-1 Only a DC-DC converter with a transformer meets the isolation requirement. The leading semiconductor manufacturers have developed ICs that implement both the PoE protocol, as well as voltage regulation. Examples of these include LTC4267 (Analog Devices), LM5071 and TPS23750 (Texas Instruments). These ICs were developed for flyback converters with switching frequencies of between 200 und 400 kHz. The WE-PoE series transformers are suitable for these ICs. Table 2.59 shows the most important parameters, such as output power, output voltage, etc. In the case of the transformers for 1.8 V, 3.3 V and 5 V, the output voltages are distributed over two or three separate windings. So up to three loads in the PD can be powered with the same voltage. The 12 V variant has two output taps of 3.3 V and 5 V to be more f­lexible. All transformers in this series have an auxiliary winding with 10–12 V to power the IC.

399

II Components Order Code

Primary inductance (µH)

Leakage inductance (µH)

Output voltage (V)

Output current (A)

749 119 133

400

4

3.3

3 x 0.4

4

ER 11/5

749 119 150

400

4

5

3 x 0.27

4

ER 11/5

749 119 218

210

4.5

1.8

3 x 1.3

7

ER 14.5/6

749 119 233

210

2.5

3.3

0.7/1.4

7

ER 14.5/6

749 119 250

210

2.5

5

3 x 0.47

7

ER 14.5/6 ER 14.5/6

Power

Size

(W)

749 119 291 2

210

2.5

12/5/3.3

0.58

7

749 119 318

120

2.5

1.8

3 x 2.4

13

EFD 15

749 119 333

120

3.5

3.3

3 x 1.35

13

EFD 15

749 119 350

120

2.5

5

3 x 0.9

13

EFD 15

749 119 391 2

120

1.5

12/5/3.3

1.2

13

EFD 15

749 119 933

127

1.3

3.3

2x2

13

EP 13

749 119 933 1

127

2.8

3.3

2x2

13

EP 13

749 119 950

127

1.3

5.0

2 x 1.3

13

EP 13

749 119 950 1

127

2.5

5.0

2 x 1.3

13

EP 13

749 119 911 2

127

1.3

12

2 x 0.55

13

EP 13

749 119 921 2

127

2.3

12

2 x 0.55

13

EP 13

Tab. 2.59: Electrical parameters of the PoE transformers The transformers are tested with 1.5 kVac between the primary and s­ econdary sides and therefore comply with the international standards IEC 60950 and IEC 62368-1. In September 2009 the Standard IEEE802.3at was released. With that the PoE ­Standard IEEE 802.3af was extended for power up to 30 W (25.5 W after cable losses). For those PoE+ objectives the WE-PoE series was enlarged. Order Code

Inductance (μH)

Turns Ratio

Output Voltage (V)

Output ­Current (A)

Auxiliary Voltage (V)

DCR N1 (mΩ)

DCR N2 (mΩ)

DCR N3 (mΩ)

Size

750 310 925

57

8:1:2:2

3.3

9

7

60

8

350

EFD 20

749 119 433

42

11 : 1 : 3.3

3.3

9

11

99

3.2

220

EFD 20

749 119 450

65

7:1:3

5

6

15

116

4

230

EFD 20

749 119 450 1

42

7 : 1 : 2.25

5

6

12

84

5

225

EFD 20

750 310 926

40

5 : 1 : 1.25

5

6

7

50

10

300

EFD 20

749 119 491 2

42

3:1:1

12

2.5

12

61

18

180

EFD 20 EFD 20

750 310 927

112

3 : 1 : 0.5

12

2.5

6

150

50

70

750 310 743

38

6,67 : 1 : 4

3.3

7.5

12

82

2

160

EP13

750 310 744

5 : 1 : 2.5

5

5

82

3

220

–

–

EP13

750 310 742

1 : 2 : 1.1

12

2.1

85

25

155

–

–

EP13

Tab. 2.60: Electrical parameters of the WE-PoE+ transformers

400

4.4 WE-PoEH Power over Ethernet High Power Transformer WE-PoEH

Fig. 2.113: WE-PoEH Power over Ethernet High Power Transformer

Order Code

Version

Ui

UO1 (V)

IO1 (A)

IO2 (A)

L1 (µH)

n

LSmin. (µH)

ISAT 1 (A)

∫ Udt (µVs)

7491195331

Flyback

9 V (DC)–57 V(DC)

3.3

3.2

3.2

21

4:1:1:3

0.3

6.2

–

749119550

Flyback

33 V (DC)–57 V(DC)

5

3

3

48

5.33:1:1:2

0.7

3.6

–

7491195112

Flyback

33 V (DC)–57 V(DC)

12

1.5

1.5

41

2.67:1:1:1

0.5

4.2

–

749119633

Forward

33 V (DC)–57 V(DC)

3.3

8

8

100

6:1:1:3

0.5

1

106.5

749119650

Forward

33 V (DC)–57 V(DC)

5

7

7

100

4:1:1:2

0.5

1

106.5

7491196112

Forward

33 V (DC)–57 V(DC)

12

2.5

2.5

100

1.71:1:1:0­.86

0.5

1

106.5

Ui: Input Voltage; UO1: Output Voltage 1; IO1: Output Current 1; IO2: Output Current 2; L1: Inductance; n: Turns Ratio; LSmin.: Leakage Inductance; ISAT 1: Saturation Current; ∫Udt: Voltage-µSecond

Tab. 2.61: Typical specification of some WE-PoEH transformers WE-PoEH Power over Ethernet High Power Transformer series in the EPQ13 size package handle up to 60 W output power. Available for both flyback and forward topologies with multiple secondaries for the most common voltages.

401

II Components 4.5 WE-LLCR Resonant Converter Transformer WE-LLCR

Fig. 2.114: WE-LLCR Resonant Converter Transformer Order Code

UO (V)

760895431

12

760895441

24

760895451

48

f

L1 (µH)

Tol. L

LSmin. (µH)

Tol. LS

n

ISAT 1 (A)

35 : 2 : 2:3 70–120

600

±10%

100

±10%

35 : 4 : 4:3

2

35 : 8 : 8:3

UO: Output Voltage; fswitch: Switching Frequency; L1: Inductance; Tol. L: Inductance (Tol.); LSmin.: Leakage Inductance; Tol. LS: Leakage Inductance (Tol.); n: Turns Ratio; ISAT 1: Saturation Current

Tab. 2.62: Typical specification of some WE-LLCR transformers WE-LLCR transformers are designed with a defined leakage inductance for use without an additional resonant inductor. Intended to operate after a power factor correction circuit with an input voltage of 360–400 Vdc, output power level from 150–250 W, and switching frequency from 70–120 kHz these transformers provide for a high efficiency power supply.

Offline transformers WE-UNIT

402

4.6 WE-UNIT Offline Transformers The WE-UNIT series transformers (Figure 2.115) are designed for worldwide mains input voltages. The input voltage can span the range from 85 VAC (e.g. Japan) to 265 VAC (e.g. Germany). In contrast to 50 Hz transformers, which have to be switched from 110 V to 230 V, modern switch mode power supplies regulate this. At the same

time optimized IC switching regulators are available and make the requirements for low standby losses possible.

Fig. 2.115: WE-UNIT mains transformers A switched DC voltage of 120 V to 385 V is applied at the transformer. Many IC manufacturers have brought ICs in the low power range onto the market in which MOSFETS are already integrated. Consequently the circuit complexity and the amount of external components is minimized. Order Code

Inductance (mH)

Turns ratio

DCR1 (Ω)

DCR2 (mΩ)

DCR3 (mΩ)

Output power (W)

749 118 105

2.8

18.9 : 1

11

50

–

3

749 118 101 2

2.8

8.1 : 1

11

290

–

3

749 118 102 4

11

1200

–

3

2.8

4:1

749 118 115

2.8

18.9 : 1

8

50

–

3

749 118 111 2

2.8

8.1 : 1

8

290

–

3

749 118 112 4

2.8

4:1

8

1200

–

3

749 118 205

0.9

19 : 1 : 1

4.3

26

26

9

749 118 201 2

0.9

9.5 : 1 : 1

4.3

75

85

9

749 118 202 4

0.9

4.4 : 1

4.3

210

–

9

749 118 215

0.9

19 : 1 : 1

4.3

28

28

9

749 118 211 2

0.9

9.5 : 1 : 1

4.3

94

102

9

749 118 212 4

0.9

4.4 : 1

4.3

155

–

9

Tab. 2.63: Electrical parameters of WE-UNIT series transformers The WE-UNIT series transformers are designed for power of 3 W or 9 W and various output voltages (Table 2.63). The isolation is configured for mains input voltage. A special feature is that the required air gaps and leakage paths are integrated. The 9 W transformers have shield­ing between the primary and secondary side.

403

II Components 4.7 WE-GDT Gate Drive Transformer WE-GDT Gate drive

Fig. 116: WE-GDT – Gate drive transformers RDC 1 RDC 2 (mΩ) (mΩ)

Order Code

L1 (µH)

LSmin. (µH)

760301105

260

1.7

600

760301104

330

1.9

760301103

350

2.3

760301106

370

760301108

460

760301107

650

R

n

∫ Udt (µVs)

CWW 1 (pF)

600

600 mΩ

1:1:1

25.2

9

620

270

270 mΩ

2:1:1

28.6

7

700

170

170 mΩ 2.5:1:1

29.4

7

1.8

520

640

–

1:1

30.2

10

2.3

730

150

–

2.5:1

33.6

9

2.9

1200

600

–

1.5:1

40.8

9

L1: Inductance; LSmin.: Leakage Inductance; RDC 1: DC Resistance 1; RDC 2: DC Resistance 2; RDC 3: DC Resistance 3; n: Turns Ratio; ∫Udt: Voltage-µSecond; CWW 1: Interwinding Capacitance

Tab. 2.64: Typical specification of some WE-GDT transformers The WE-GDT gate drive transformers offered in two sizes, EP5 and EP7 provide up to 2500 Vac isolation, multiple turns ratios and low leakage in a small compact package. Suitable for power or signal applications requiring galvanic isolation.

Current sense ­transformers

404

4.8 WE-CST Current Sense Transformers There are two modes of regulating the output voltage of a switched mode power supply. For voltage regulation (Voltage Mode, VM), the o­ utput voltage is measured directly and compared with a “reference voltage”. For current regulation (Current Mode, CM), the primary current is measured. The output voltage serves as a reference here.

WE-CST

Fig. 2.117: WE-CST current sense transformers To measure the primary current, the voltage can be tapped by a burden resistor. A current sense transformer is often used for higher primary currents to reduce losses and provide isolation. The current is then determined with a burden resistor RT by measuring voltage (Figure 2.118).

NPRI (1 turn)

NSEC (N turns)

Burden resistor

RT

Fig. 2.118: Application example for current sense transformers The voltage measured at RT is given by:





(2.55)

URT = voltage at the burden resistor RT = burden resistor The WE-CST series current converters were developed for primary currents up to 10 A. The mechanical dimensions and the electrical values are given in Table 2.65 and 2.66.

405

II Components Order Code

Inductance min. (µH)

Rated current (A)

DCR N1 max. (mΩ)

DCR N2 max. (Ω)

Turns ratio

High voltage test (VAC)

749 251 020

80

10

6

0.20

1 : 20

500

749 251 030

180

10

6

0.48

1 : 30

500

749 251 040

320

10

6

0.90

1 : 40

500

749 251 050

500

10

6

1.40

1 : 50

500

749 251 060

720

10

6

1.75

1 : 60

500

749 251 070

980

10

6

2.20

1 : 70

500

749 251 100

2000

10

6

5.50

1 : 100

500

749 251 125

3000

10

6

6.50

1 : 125

500

Tab. 2.65: Mechanical dimensions for the WE-CST series current sense transformers

A (mm)

B (mm)

C (mm)

7,7

6,9

5,33

Tab. 2.66:  Mechanical drawing of the WE-CST current sense ­­transformers As described in the functionality of a transformer, a magnetizing current also appears here. This affects the result as a measurement error. The magnetizing current must therefore be considered when selecting a current converter. It can be estimated with the following formula:



406



(2.56)

This is clarified with an example: A current converter with the following properties is sought: Input currents: Ii Frequency: f Burden voltage: URT Accuracy: 10%

= = = =

1 A–5 A 100 kHz 0.1 V at 1 A 0.5 V at 5 A

For a turns ratio of 1:100, Equation 2.55 gives a burden resistance of 10 Ω. The accuracy for the input current of 1 A should be better than 0.1 A, i.e. the magnetizing current carried over to the secondary side must be smaller than 0.001 A. Equation 2.57 calculates the minimum inductance as:





(2.57)

It is apparent from Table 2.65 that 749 251 100 is very well suited for this application.

5 Wireless Power Transfer 5.1 WE-WPCC, Wireless Power Transfer Coils WE-WPCC Wireless Power Transfer

Fig. 2.119: WE-WPCC Wireless power transfer coils 407

II Components Wireless Power Transfer Coils are available in a variety of sizes for applications up to 300 W. All QI standard compliant coils use Litzwire for the highest Q-value and high permeability shielding to concentrate flux and shield sensitive components. Power arrays with multi transmit coils are also available. Dual standard (QI and Airfuel Alliance) receiving coils allow devices to be charged by stations from either standard. Mix and match transmit and receive coils using the online tool REDEXPERT.

60

200

760 308 104 113 760 308 100 141 760 308 101 302

760 308 104 113 760 308 100 141 760 308 101 302

160

40

Q-factor

temperature rise (K)

50

30

120

80

20 40

10 0

0

2

4

6

8

0

10

12

current (A)

10

100

1000

10000

frequency (kHz)

Fig. 2.120: Performance of some WE-WPCC transmitter coils

6 Signal & Communications LAN transformers

6.1 LAN transformers WE-LAN and WE-LAN RJ45 series Ethernet transformers

Ethernet transformers

The WE-LAN series Ethernet transformers (Figure 2.121) are not simple transformers, but rather modules in which, depending on the number of ports for which they are suitable, have at least two transformers and a certain number of current-compensated chokes.

WE-LAN

408

Fig. 2.121: LAN transformers series WE-LAN

As the name denotes, they are designed as transformers for Ethernet networks. Ethernet is the most widespread form of local network (Local Area Network – LAN). Ethernet is operated at different transmission speeds. The requirements for Ethernet are described in the IEEE802.3 standards:

IEEE802.3

• 10 Base-T: transmission rate 10 Mbps > standard IEEE802.3 • 100 Base-T: transmission rate 100 Mbps > standard IEEE802.3u • 1000 Base-T: transmission rate 1000 Mbps > standard IEEE802.3ab • Power over Ethernet: independent of the transmission rate > IEEE802.3af

10 Base-T 100 Base-T 1000 Base-T Power over Ethernet

For these standards, the transmission medium is a copper cable with unshielded, twisted pairs (UTP) of type Cat. 5 or better. The other IEEE802.3 series standards refer to glass fiber cable transmission.

Cat. 5

For 10 Base-T and 100 Base-T, one wire pair is used for the transmission channel (transmit) and one for the reception channel (receive). Two of the four wire pairs in the Cat. 5 cable remain unused. In the case of 1000 Base-T, all four wire pairs are used in both directions. According to the EN60950 standard (safety of information technology equipment), Ethernet is classified as a TNV 1 circuit (telecommunication network voltage and be isolated against a SELV circuit (safe extra low voltage). The test voltage stipulated by the standard must be at least 1.5 kVac with a test duration of 1 min. Similar requirements are listed in IEC 62368-1 for ES1 circuits. As a result of this isolation voltage, it is essential to use a transformer between the network and the Ethernet terminal device. Two transformers are required for 10/100 Base-T networks and four transformers for 1000 Base-T networks. The necessary number of transformers is i­ntegrated in the WE-LAN series modules. As described in the previous section, ring core transformers are used. To avoid further external components, e.g. current-compensated ­chokes (data line filters), these are also integrated in the transformer modules (see Figure 2.122).

Fig. 2.122: Typical circuit diagram for a LAN transformer module

409

II Components The relevant Ethernet standards for 100 Base-T call for transformers with a minimum inductance of 350 µH and DC current premagnetization of 8 mA. This serves to ensure functionality even with small asymmetries of the wire pairs. The turns ratio is defined by the Ethernet controller used. Whereas 10 Base-T often uses turns ratios of 1 : 1.414 or 1 : 2.5 on at least one of the lines, 100 and 1000 Base-T almost exclusively work with a turns r­ atio of 1 : 1. This is because the number of secondary turns is so large – due to the high primary inductance combined with a higher number of turns than with 10 Base-T – that they can no longer be accommodat­ed on a small ring core. Winding with twisted wires is also no longer possible, which can still be tolerated for 10 Base-T, but leads to a de­terioration in transmission properties for 100 Base-T. The minimum insertion loss and the maximum values for return loss, crosstalk and common mode rejection are also defined over the entire frequency range. Table 2.67 provides an overview of the WE-LAN series transformer modules with the most important electrical parameters. 10 Base-T Order Code

749 090 010

Inductance (µH)

Turns ratio

DCR (Ω)

Leakage Inductance (nH)

Interwinding Capacitance (pF)

Rx

Tx

Rx

Tx

1–3

14–16

6–8

9–11

Rx

Tx

Rx

Tx

120

20

1:1

1:2

0.9

0.9

0.9

0.9

400

200

6

9

Ports

1

100 Base-T Return Loss Order Code

Differential to Common Mode Rejection

Induc- Inser1–30 40 50 60–80 tance tion MHz MHz MHz MHz (µH) loss @ 100 Ω @ 100 Ω @ 100 Ω @ 100 Ω (dB) (dB) (dB) (dB) (dB)

30 MHz (dB)

60 MHz (dB)

Crosstalk

60–100 60 MHz MHz (dB) (dB)

100 MHz (dB)

Ports

749 010 011

350

–1.1

–16

–14

–13

–10

–38

–38

–30

–40

–33

1

749 010 012

350

–1.1

–16

–14

–13

–10

–38

–38

–30

–40

–33

1

749 010 013*

350

–1.1

–18

–16

–14

–12

–

–

–30

–

–35

1

749 010 014

350

–1.0

–18

–16

–14

–12

–

–

–30

–45

–35

1

749 010 040

350

–1.0

–18

–14

–13

–12

–37

–37

–25

–40

–33

4

* different pinout as 749 010 014

410

100 Base-T Power over Ethernet Return Loss Order Code

Differential to Common Mode Rejection

1–30 Induc- Inser40 50 60–80 tance tion MHz MHz MHz MHz (µH) loss @ 100 Ω @ 100 Ω @ 100 Ω @ 100 Ω (dB) (dB) (dB) (dB) (dB)

30 MHz (dB)

60 MHz (dB)

Crosstalk

60–100 60 MHz MHz (dB) (dB)

100 MHz (dB)

Ports

749 013 011

350

–1.2

–16

–14

–13

–12

–

–

–35

–

–35

1

749 013 010

350

–1.2

–16

–14

–13

–10

–43

–37

–33

–40

–35

1

749 013 020

350

–1.2

–16

–14

–13

–10

–50

–43

–35

–37

–33

2

749 013 021

350

–1.2

–16

–14

–13

–12

–43

–37

–33

–37

–31

2

749 013 022

350

–1.1

–16

–14

–13

–12

–43

–37

–33

–37

–33

2

749 013 040

350

–1.1

–16

–14

–13

–12

–43

–37

–33

–40

–35

4

1000 Base-T Return Loss Order Code

Differential to Common Mode Rejection

Induc- Inser1–30 40 50 60–80 tance tion MHz MHz MHz MHz (µH) loss @ 100 Ω @ 100 Ω @ 100 Ω @ 100 Ω (dB) (dB) (dB) (dB) (dB)

30 MHz (dB)

60 MHz (dB)

Crosstalk

60–100 60 MHz MHz (dB) (dB)

100 MHz (dB)

Ports

749 020 010

350

–1.0

–18

–16

–12

–10

–43

–37

–33

–40

–35

1

749 020 011

350

–1.0

–18

–16

–12

–10

–43

–37

–33

–40

–35

1

749 020 013

350

–1.0

–18

–15

–12

–10

–42

–

–33

–

–35

1

749 020 100

350

–1.0

–18

–15

–12

–10

–42

–

–33

–

–35

0.5

1000 Base-T Power over Ethernet Return Loss Order Code

Differential to Common Mode Rejection

Induc- Inser1–30 40 50 60–80 tance tion MHz MHz MHz MHz (µH) loss @ 100 Ω @ 100 Ω @ 100 Ω @ 100 Ω (dB) (dB) (dB) (dB) (dB)

30 MHz (dB)

60 MHz (dB)

Crosstalk

60–100 60 MHz MHz (dB) (dB)

100 MHz (dB)

Ports

749 023 010

350

–1.1

–1.0

–2.0

–18

–12

–12

–10

–43

–37

–33

1

749 023 020

350

–1.1

–1.0

–2.0

–18

–12

–12

–10

–41

–37

–33

2

Tab. 2.67: Electrical data of LAN-transformers WE-LAN

411

II Components 6.2 W  E-LAN HPLE – 1000BASE-T High Performance, Low EMI WE-LAN HPLE High performance Low EMI

The High Performance and Low EMI (HPLE) RJ45 ports feature integrated magnetic filters that improve noise attenuation in the range of 100 MHz to 500 MHz. Better noise attenuation means effectively meeting class B compliance FCC15 or CISPR22 tests. The discrete HPLE series can be used for gigabit speeds and PoE applications. The Common Mode Rejection between 1 MHz–100 MHz has to be at least –30 dB to meet the IEEE802.3xx standards, but there is no reference as to how much attenuation (Common Mode Rejection) is needed above 100 MHz. One of the main issues during compliance testing is the noise generated by the internal clock of the PHY chip. The IC controller clock usually runs at 125 MHz and the resulting harmonics may appear up to a range of 875 MHz. See Figure 2.128. The first three harmonics of the125 MHz fundamental frequency are the main source of noise and interference. To solve this issue, Würth Elektronik designed their HPLE RJ45 ports specifically for very high Common Mode Rejection in the frequency range of 100 MHz to 400 MHz. See Figure 2.129. The HPLE series has several different configurations which may or may not be optimal with certain PHY’s. Please check with your vendor for the most compatible HPLE port. Replacing standard RJ45 ports with HPLE ports improves performance significantly. Comparison tests between standard and HPLE ports show that the1st noise harmonic goes down from 45 dBuV to 20 dBuV, the 2nd harmonic from 50 dBuV down to 37 dBuV and the 3rd from 47 dBuV to 39 dBuV.

Fig. 2.123: Schematics of standard and HPLE LAN transformers

412

6.3 W  E-RJ45 LAN/WE-RJ45 HPLE Transformers integrated with RJ45 connector WE-RJ45 LAN and WE-RJ45 HPLE are LAN transformers integrated into a RJ45 connector case. These integrated RJ45 connectors perform the same task as the Ethernet transformers of the WE-LAN series. WE-RJ45 LAN

Fig. 2.124: WE-RJ45 LAN transformer integrated RJ45 connector Inductance Dielectric Rating Turns Ratio

350 µH min

100 kHz, 100 mV, 8 mADC

1500 Vrms

1 Minute

1:1

±2%

Insertion Loss

–1.0 dB max

1 MHz – 100 MHz

Return Loss

–18 dB min.

1 MHz – 30 MHz

–16 dB min.

30 MHz – 45 MHz

–14 dB min.

45 MHz – 60 MHz

–12 dB min.

60 MHz – 80 MHz

Crosstalk

–35 dB min.

1 MHz – 100 MHz

Common Mode Rejection

–35 dB min.

1 MHz – 100 MHz

–25 dB min.

100 MHz – 500 MHz (for HPLE)

Tab. 2.68: Typical electrical characteristics In addition, a “Bob Smith Termination”, consisting of a resistance network of 4x 75 Ω and a capacitor of 1000 pF for HF interference suppression of the unwanted signal, is also integrated. This termination with the resistance and capacitance values mentioned above has emerged as optimum interference suppression combination from a study carried out by Robert W. Smith. High-frequency interference transmitted via the Ethernet cable can be discharged with this termination via a pin or directly with the metal screen connected to ground.

413

II Components

Fig. 2.125: CMC on cable side and Bob Smith Termination WE-RJ45 HPLE The High Performance & Low EMI series of the RJ45 plug connector range particularly provides another common mode choke on the IC side before the Ethernet transformer. WE-RJ45 HPLE

Fig. 2.126: CMC on PHY/Cable side and Bob Smith Termination The High Performance & Low EMI series of the RJ45 plug connectors with integrated wideband filter components for 10/100/1000 Base-T-Ethernet and Power-overEthernet (PoE) applications from Würth Elektronik have been specially developed for interference suppression in the frequency range between 100 MHz and 500 MHz; as a result, the compliance with the limit values is realized compactly in one component.

414

Fig. 2.127: Common Mode Rejection The IEEE802.3xx standards are referred to for the development of Ethernet interfaces. According to these, the Common Mode Rejection must show an attenuation of –35 dB in the range of 1 MHz–100 MHz in order to meet the requirements. What are the requirements for the attenuation over 100 MHz? The interface must conform with the higher level EMC Directives. Figure 2.128 shows the interference spectrum of a standard filtered Ethernet interface. The peaks can be clearly seen; these are produced by the clock frequency of the PHY chip in the range from 125 MHz to 875 MHz.

Fig. 2.128: Typical interference spectrum of an Ethernet interface

415

II Components A reference measurement Figure 2.129 showed the following results: With the High ­Performance & Low EMI RJ45 connector, the interference peaks were reduced as follows: • Fundamental frequency from 45 dBμV to 20 dBμV • Harmonic from 50 dBμV to 37 dBμV • Second harmonic from 47 dBμV to 39 dBμV

Fig. 2.129: Reduction of the interference peak using High Performance The WE-RJ45 LAN and WE-RJ45 HPLE series provide the benefit that RJ45 connector and Ethernet transformer as well as the Bob Smith Termination integrated into one component: this saves space and components on the PCB. Due to the compact connections between connector and transformer, the series also provides better EMC characteristics. 10/100 Base-T Standard Type/HPLE Type

416

Standard Type

HPLE Type

749 901 100 2

749 901 100 13

749 901 112 1

749 901 112 15

749 921 100 2

749 921 100 3

749 921 112 2

749 921 112 3

749 941 100 0

749 941 100 1

749 941 112 1

749 941 112 2

Speed

10/100 Mbs

1000 Base-T Standard Type/HPLE Type Standard Type

HPLE Type

749 911 100 7

749 911 100 5

749 911 161 3

749 911 161 4

749 931 100 0

749 931 100 1

749 931 161 0

749 931 161 1

749 951 100 0

749 951 100 1

749 951 161 0

749 951 161 1

Speed

1000 Mbs

Tab. 2.69: Standard type/HPLE type of the RJ45 LAN connectors

6.4 WE-LAN 10G LAN Transformer PoE/PoE+ WE-LAN 10G 10G LAN with PoE+

Fig. 2.130: WE-LAN 10G LAN Transformer

417

II Components Order Code

L (mm)

Pw (mm)

H (mm)

W (mm)

PoE

Number of Ports

Operating Temperature

749050010U

13.97

15.11

6.6

13.5

non-PoE

0 °C up to +70 °C

749053013

13.97

15.11

6.6

13.5

PoE (up to 350 mA)

–40 °C up to +85 °C

749053011

13.97

15.11

6.6

13.5

PoE (up to 350 mA)

749052012

13.97

15.11

6.6

13.5

PoE (up to 600 mA)

0 °C up to +70 °C

749052051

18.29

16

8.8

14.7

PoE+ (up to 1 A)

–40 °C up to +85 °C

1

0 °C up to +70 °C

+

Auto MDIX

Yes

L: Length; Pw: Pin to Pin (Middle); H: Height; W: Width

Tab. 2.70: Typical specification of some WE-LAN 10G LAN transformers The WE-LAN 10G transformers are single port 10 Gbit/s devices with integrated common mode chokes. They are complaint with IEEE 802.3an, 802.3at, 802.3at and the upcoming 802.3bt capable of up to 1 A of current in PoE applications.

6.5 Telecom transformers DSL transformer

WE-DSL Digital Subscriber Line transformers The WE-DSL series Digital Subscriber Line transformers (Figure 2.131) are specialized line interface transformers that are designed to optimize the performance of DSL chipsets. With each chipset varying in power drive levels, impedance characteristics and signal spectrum the transformers are very much chipset dependant in their design.

WE-DSL

Fig. 2.131: WE-DSL Digital Subscriber Line transformers DSL (Digital Subscriber Line) DSL technology has its origins in the competition between the telecom and cable television industry to offer each other’s services; video and data services. While the hybrid fiber/coax cable network was already a high-bandwidth network, the telecom industry had to look to some other technology to increase the bandwidth of their aging analog network. 418

ADSL (Asymmetrical Digital Subscriber Line)

ADSL

The ADSL technology formed the backbone of the telecom industry’s initial foray into the high speed data and video services market. Asymmetrical in nature, the ADSL technology reserves the bulk of the available bandwidth for down­loading data rather than uploading. This is consistent with the requirements of video services as well as the traffic pattern of the typical i­nternet user. ADSL data rates generally are in the 1 to 16 Mbits/s d­ ownstream and 1 to 2 Mbit/s upstream with POTS service concurrently available on the same line. Various flavours and generations of ADSL are, and have been, promoted such as ADSL2, ADSL2+, ADSL+, RADSL, etc. HDSL (High-Speed Digital Subscriber Line)

HDSL

Following on the heels of ADSL technology was HDSL which addresses the needs of those users requir­ing a symmetrical service with as much bandwidth available for upstream data traffic as downstream. The HDSL services are targeted to service providers, businesses and Small Office Home Office (SOHO) ­customers. Generally speaking the HDSL technology is a replacement for the older T1/E1 technology. Like ADSL, the HDSL technology is ­promoted in a host of variants such as SHDSL, SDSL, HDSL-2, HDSL-4 MDSL, IDSL, g.SHDSL etc. VDSL (Very High Speed Digital Subscriber Line)

VDSL

The latest generation DSL technologies are the VDSL and VDSL2 technologies. VDSL and VDSL2 i­ncreases the maximum available download bit rate to over 100 Mbit/s at short loop lengths. VDSL technologies allow for either symmetric or asymmetric access and support high bandwidth applications such as HDTV in addition to telephone and data services. Transformer Parameters Inductance: Inductance requirements vary widely between not only DSL technologies, but also between chipsets. ADSL and VDSL inductance requirements can be below 100 µH while HDSL inductance specifications are over 3 mH. Depending on the specific requirements of the chipset this inductance can be any value in between and is almost always toleranc­ed to a level of ± 5 to 10 percent. The inductance specification is a­ lmost always specified by the IC manufacturer and is dependant on a host of conflicting requirements including: • The transformer’s need to handle DC current • Whether or not the transformer has to perform a filtering function for concurrent POTS operations • Signal bandwidth • The insertion loss • The return loss characteristics • The impedance characteristics of the chipset 419

II Components DC current handling: Typically only HDSL type technologies require the transformer to operate with a DC current applied through the transformer. This requirement is due to the fact that HDSL type equipment is often called on to provide power to remote terminal devices. Typical DC current requirements for HDSL products are 60 to 100 mA. Turns ratio: Turns ratios, like inductance, can vary greatly from chipset to chipset. While most transformers have a single primary and secondary winding, it is not uncommon for some chipsets to require separate secondary windings for transmit and receive or even a separate auxiliary winding. Factors affecting the turns ratio include: • The architecture of the chipset • The return loss characteristics required • The impedance characteristics of the chipset • Whether or not the signal voltage needs to be stepped up or down Leakage inductance

Leakage inductance: Leakage inductance requirements are almost always held to a maximum value. In rare instances where the chipset is particularly sensitive to variations in impedance, the leakage inductance will be held to a ­tolerance requirement. While this is possible to achieve, it does result in a design that is very susceptible to variations in manufacturing and should be avoided if at all possible. Factors that dictate the leakage inductance specification include: • The insertion loss requirements • Signal bandwidth • The return loss characteristics required • The impedance characteristics of the chipset Interwinding capacitance: Interwinding capacitance may or may not be specified by the IC manufacturers when defining the transformer requirements depending on the transceivers sensitivity to capacitance. However, regardless of whether or not it is specified, interwinding capacitance affects the overall performance and is typically limited by the following requirements: • The insertion loss • Signal bandwidth • The return loss characteristics • The impedance characteristics of the chipset • Longitudinal balance DC resistance:

420

Besides affecting the power transfer efficiency of the transformer, DC resistance also affect to a lesser degree these DSL line transformer r­ equirements:

• The insertion loss requirements • The return loss characteristics • The impedance characteristics of the chipset • Longitudinal balance Total harmonic distortion: General: The total harmonic distortion THD denotes the ratio of the effective value of all harmonics of a signal to the effective value of the total signal.





Total harmonic ­distortion

(2.58)

For sinusoidal signals, the total harmonic distortion is used as a measure of non-linear distortions! The smaller the THD, the more the signal corresponds to the original! Total harmonic distortion is a measure of how much the DSL transformer will distort the desired signal. The harder a transformer is driven, the more the core approaches saturation. As the core approaches saturation the signal begins to distort. While total harmonic distortion is a concern for all DSL transformer, it is not as critical for VDSL transformers where the operation of the core is at higher frequency; hence a lower flux density and further away from saturation. HDSL transformers however utilize the lower end of the frequency bandwidth quite extensively and are often driven quite hard at these frequencies for longer loop lengths. This results in a need for good total harmonic distortion characteristics at a high flux density level. Crosstalk:

Crosstalk

While crosstalk is always a concern for IC manufacturers, it is typically not an issue with the line interface transformers. Almost all line interface transformers for DSL are constructed on some EP style of core which is by virtue of its’ shape, self shielding. Voltage/safety isolation: Voltage/safety isolation is always a consideration as it is the primary purpose of the DSL line interface transformer in the first place. Depending on the safety agency requirements and intended application the isolation as well as creepage and clearance requirements for the transformer will vary. Typically transformers need to supply basic or supplementary isolation for a working voltage of 250 V. In summary each of these parameter affect the others to a greater or lesser degree. The crux of DSL line interface design is balancing all of these parameters to create a solution that meets the customer requirements in a cost effective manner.

421

II Components 7 RF Inductors 7.1 RF Inductors RF inductor

RF inductors are inductors that are used in the high frequency range. These inductors use iron free ceramic cores which do not saturate making them frequency and current independent. Further very important advantages are the linear impedance over a wide frequency range, i.e. XL = 2 p f · L

(2.59)

and a very high quality factor Q (typ. 50–100). The following relationship exists for an inductance L and the winding turns n:

L ~ n2(2.60)

The air coil has a parasitic capacitance dependent on winding diameter, wire thickness as well as layer and winding structure, which mainly exists between neighboring windings (Figure 2.132), but for SMD components also o­ ccurs between both the connection pads.

Fig. 2.132: Schematic of a RF inductor. Parallel wires act like electrodes of a ­capacitor generating a distributed capacitance. The self-resonant frequency, above which the component shows capacitive characteristics, results from the inductance and the parasitic capacitance.

422

7.2 W  E-KI, WE-KI HC, WE-FRI, WE-RFH Ceramic wire wound ­inductors WE-KI, WE-KI HC WE-FRI WE-RFH Ceramic wire wound

Ceramic SMD inductors

Fig. 2.133: Example of ceramic WE-KI SMD inductors Size 0402A: Order Code

744 765 210A

Quality factor Q

Testfrequency

(µH)

(min)

(MHz)

0.10

20

Inductance

Quality factor Q

Inductance

L 150

Q 150

Selfresonantfrequency

DCR

Rated Current

(MHz)

(Ω)

(mA)

1300

2.52

100

Selfresonantfrequency

DCR

Rated Current

(MHz)

(Ω)

(mA)

1400

0.63

400

Selfresonantfrequency

DCR

Rated Current

(MHz)

(Ω)

(mA)

1200

0.43

500

Size 0603A: Order Code

744 761 210A

(µH)

(min)

0.10

35

Inductance

Quality factor Q

Testfrequency L

Q

(MHz) 150

150

Size 0805A: Order Code

744 760 210A

(µH)

(min)

0.10

60

Testfrequency L

Q

(MHz) 150

500

423

II Components Size 1008A: Order Code

Inductance

Quality factor Q

Testfrequency L

Q

(MHz)

Selfresonantfrequency

DCR

Rated Current

(µH)

(min)

(MHz)

(Ω)

(mA)

744 762 210A

0.10

60

100

350

1100

0.18

1000

744 762 310A

1.00

35

25.0

500

310

3.30

120

Tab. 2.71: Electrical parameters of some ceramic SMD inductors The solder pads on the coil former are designed to be very small (Figure 2.134), in order that the inductors achieve a high self-resonant frequency.

Fig. 2.134: Schematic construction of a ceramic inductor Self-resonant frequencies of well above 10 GHz are thus attained. The WE-RFH series is available if higher current carrying capacities are required. The components are wound on ceramic or ferrite cores depending on the inductance value. WE-RFH

Fig. 2.135: Ferrite SMD Inductor WE-RFH

424

The range of wound RF inductors is rounded off with the WE-RFI series. The components wound on ferrite stand out by virtue of their very high inductance values in relation to their package size.

7.3 WE-MK Multilayer ceramic inductor The construction of the SMD inductor (Figure 2.136) WE-MK differs from that of a conventional air coil in that the “coil” is printed on a ceramic substrate. This presents the following advantages:

WE-MK Multilayer ceramic

• Very low inductances achievable • Very low parasitic capacitance and therefore high resonant frequency • Relatively high current carrying capacity of typ. 300 mA WE-MK

Fig. 2.136: WE-MK multilayer ceramic SMD inductor The specifications of various multilayer ceramic SMD inductors are shown in Tables 2.72, 2.73, 2.74. Size 0201: 100

30 744782012 744782056 74478215 74478233

744782012 744782056 25

74478215

Q

Inductance (nH)

74478233 20

10

15

10

5

1

1

10

100

0

1000

Frequency (MHz)

1

10

100

1000

Frequency (MHz)

Fig. 2.137: Size 0201 typical performance Order Code

Inductance (nH)

SelfL/QQuality factor Frequency resonant frequency Q (MHz) (MHz) (min)

DCR

Rated current

(Ω)

(mA)

744 782 015

1.5

17

100/800

>13000

0.18

300

744 782 10

10.0

20

100/800

4000

1.20

250

744 782 33

33.0

17

100/800

1500

2.30

200

Tab. 2.72: Size 0201 typical specifications

425

II Components Size 0402: 1000

50 744784012A 744784082A 744784127A

40

744784215A 100 Q

Inductance (nH)

744784012A 744784082A 744784127A 744784215A

30

20 10

10

1

1

10

100

0

1000

1

Frequency (MHz)

10

100

1000

Frequency (MHz)

Fig. 2.138: Size 0402 typical performance Order Code

Inductance (nH)

SelfL/QQuality factor Frequency resonant frequency Q (MHz) (MHz) (min)

DCR

Rated current

(Ω)

(mA)

744 784 13A

1.5

8

100

>15000

0.13

300

744 784 010A

10.0

8

100

3700

0.45

250

744 784 47A

47.0

8

100

1200

1.30

150

Tab. 2.73: Size 0402 typical specifications Size 0603: 1000

80 744786012 70

74478606 74478613

60

74478625

100 50 Q

Inductance (nH)

744786012 74478606 74478613 74478625

40

30 10 20

10

1

1

10

100

1000

Frequency (MHz)

Fig. 2.139: Size 0603 typical performance

426

0

1

10

100

1000

Frequency (MHz)

Order Code

Inductance (nH)

744 786 02

SelfL/QQuality factor Frequency resonant frequency Q (MHz) (MHz) (min)

2.2

 8

744 786 112

12.0

744 786 110

100.0

DCR

Rated current

(Ω)

(mA) 600

100

12000

0.15

 8

100

3200

0.5

600

12

100

750

2

600

Tab. 2.74: Size 0603 typical specifications The preferred field of application of these components lies in the high-­frequency range, in filter circuits and oscillator circuits.

7.4 WE-TCI Thinfilm Chip Inductors

Thinfilm Chips WE-TCI

Fig. 2.140: WE-TCI Thinfilm Chip Inductors

10

25 744900012 744900039

744900022 744900056

744900012 744900022 744900039

20

Q

Inductance (nH)

744900056

15

10

5

1

10

100

1000

Frequency (MHz)

0

10

100

1000

Frequency (MHz)

Fig. 2.141: Typical characteristics for WE-TCI size 0201 inductors 427

II Components The WE-TCI can be characterized with extreme accuracy which results in very a tight tolerance product. Tolerances of 2% and 1% (upon request) are available. Therefore it’s possible to realize precise application behavior.

Air Coil

7.5 WE-CAIR, WE-AC HC High Current Air Coil

WE-CAIR

Fig. 2.142: WE-CAIR Air Coil Inductor The air coil inductors WE-CAIR are characterized by very high quality factors even in the low frequency range (Q > 100). The self-resonant frequency is very high. The air coil inductors WE-CAIR guarantee a greater rated current compared to other RF inductors. In addition, the air coil inductors WE-CAIR have a high temperature stability.

428

7.6 WE-AC HC High Current Air Coil The WE-AC HC is characterized by very high current and even better quality factor. WE-AC HC

Fig. 2.143: WE-AC HC High Current Air Coil

Order Code

L1 (nH)

7449150023

23

7449150046

46.5

7449150079

79

7449150111

111

7449150146

146

Tol. L

TC L

Qmin.

TC Q

IR 1 (A)

RDC1 max. (mΩ)

fres 1 (MHz)

30

1.2

867

28

1.62

581

23

2.11

422

186

22

2.11

374

163

19

3.33

332

191 223 ±20%

1 MHz

184

100 MHz

L1: Inductance; Tol. L: Inductance (Tol.); TC L: Inductance (Test cond.); Qmin.: Q-Factor; TC Q: Q-Factor (Test cond.); IR 1: Rated Current; RDC1 max.: DC Resistance; fres 1: Self Resonant Frequency

Tab. 2.75: Typical specification of some WE-AC HC inductors

100000

100000 744 915 014 6 744 915 011 1 744 915 007 9 744 915 004 6 744 915 002 3

744 915 014 6 744 915 011 1 744 915 007 9 744 915 004 6 744 915 002 3

10000 inductance (nH)

inductance (nH)

10000

1000

100

10

1000

100

10

1

1 1

10

100

1000

10000

1

frequency (MHz)

10

100

1000

10000

frequency (MHz)

Fig. 2.144: Performance of some WE-AC HC inductors

429

II Components 8 LTCC Components 8.1 LTCC (Low Temperature Co-fired Ceramic)

LTCC technology

The use of LTCC material is gaining importance in applications in the field of communications. This is a multi-layer process used for the production of sophisticated HF components with good performance characteristics. The ceramic-based LTCC technology is an inexpensive substrate technology with which up to 50 layers are stacked on top of each other. The silver or gold conductor tracks are printed on the “green” ceramic tape once the holes for the vias have been laser or mechanically generated. The various layers are then collected together and are pressed in a pressure chamber. Following lamination, the stack is sintered in a furnace, i.e. burned together at around +850 °C in process furnaces. LTCC can be used to produce multi-layer modules in which cavities, Rs, Ls and Cs are integrated. Good thermal conductivity and a low TCE (Temperature Coefficient of Expansion) are achieved. The components are hermetically sealed, i.e. are stable against mechanical and thermal stress. The dielectric properties (er, tan d, the thickness) can be controlled very effectively.

Fig. 2.145: LTCC production process The main advantage of this technology is the high degree of miniaturization and the ever-lower space requirement associated with it. This is illustrated by the following simple example. Rather than discreetly constructing a low-pass filter, there is the o­ ption of integrating the inductances and capacitances, e.g. as size 0603. The ­integration is an important aspect in the field of RF applications. 430

Fig. 2.146: LTCC low-pass filter, size 0603

Fig. 2.147: Discreetly constructed low-pass filter

8.2 WE-LPF Multilayer Chip Low-Pass Filter WE-LPF Low pass filter

Fig. 2.148: WE-LPF Multilayer Chip-Low Pass Filter HF components based on LTCC technology are designed for use in various frequency bands from around 900 MHz to 6 GHz. Application areas for these components are, for example, Wireless LAN, Bluetooth®, HomeRF, PCS, GSM, DECT and PHS. The high precision LTCC technology gives rise to low-loss RF components with reproducible, guaranteed properties and a low space requirement. WE-LPF low-pass filters for diverse applications are produced in the package types 0603 to 0805. The limit frequencies of these compact low-pass filters range from 900 MHz to 5.5 GHz. The insertion loss of the low-pass filter in the admission range is very low. The low-pass filters show comparably high attenuation in the blocked range.

Low-pass filters

431

II Components Order Code

Limit frequency f0 (MHz)

Max. insertion loss IL @ BW (dB)

Min. attenu­ ation @ 2 x f0 (dB)

Min. attenu­ ation @ 3 x f0 (dB)

Max. VSWR @BW

748 112 024

2450

0.5

35

25

1.5

748 111 009

915

0.5

30

30

1.5

Tab. 2.76: Electrical parameters of some low-pass filters WE-LPF

0

attenuation (dB)

–10

–20 –30

–40 –50

0

1

2

3

4

5

6

frequency (GHz) S21 in dB S11 in dB

Fig. 2.149: Typical attenuation curve

8.3 WE-BPF Multilayer Chip Band-Pass Filter WE-BPF Band-pass filter

432

Fig. 2.150: WE-BPF Multilayer Chip Band-Pass Filter

7

8

9

The smallest band-pass filter package type is 0805. A high level of attenuation is achieved over a very wide blocking range. The insertion loss in the admittance range is very low at approx. 2 dB, however somewhat higher than the insertion loss of low-pass filters. A summary of some WE-BPF band-pass filters is given in the following t­able. Order Code

Frequency range (MHz)

Max. insertion loss @BW (dB)

748 351 124

2400– 2500

2.3

42@ 1710– 1990 MHz

30@ 2100 MHz

30@ 4800– 5000 MHz

35@ 7200– 7500 MHz

2

748 351 024

2400– 2500

1.8

30@ 1710– 1785 MHz

25@ 1850– 1910 MHz

25@ 4800– 5000 MHz

20@ 7200– 7500 MHz

2

VSWR

Min. attenuation (dB)

Tab. 2.77: Band-pass filter WE-BPF size 1008 (selection) The example of the band-pass filter for Bluetooth® serves to illustrate where attention should be paid in the selection of components.

Band-pass filters for Bluetooth®

0

attenuation (dB)

–10 –20 –30 –40 –50 –60

0

2

4

6 8 frequency (GHz) S21 in dB S11 in dB

10

12

Fig. 2.151: Frequency response of the band-pass filter To effectively suppress frequencies in the mobile phone range, the attenuation in the range from 900 MHz (GSM900) to 2.1 GHz should be as high as possible. The attenuation in the WLAN/HyperLAN range should also be high. For this reason, specific poles have been placed at these frequencies for the above filter. 433

II Components Baluns

8.4 WE-BAL Multilayer Chip Balun

WE-BAL

Fig. 2.152: WE-BAL Multilayer Chip Balun

0

0

–5 return loss (dB)

insertion loss (dB)

–1

–2

–3

–15

–20

–4

–5

–10

2

2.2

2.4 2.6 frequency (GHz)

2.8

–25

2

2.2

2.4 2.6 frequency (GHz)

2.8

Fig. 2.153: Typical performance curves for WE-BAL

Marchand type

434

Symmetry transformers (baluns) with low insertion loss, as well as low amplitude and phase differences, can be achieved in LTCC technology. The asymmetrical (unbalanced) impedance is 50 Ω, the symmetrical (balanced) impedance can be 50 Ω, 100 Ω or 200 Ω. Two types of baluns are distinguished: LC type and Marchand type. In the case of the latter, no DC bias may be applied. It is recommended to connect a capacitor to ground in front of the pin where the DC voltage is input. Information on external circuitry and layout recommendations may be found in the datasheets.

8.5 WE-MCA Multilayer Chip Antenna

Chip antennas WE-MCA

Fig. 2.154: WE-MCA Multilayer Chip Antenna The WE-MCA chip antennas based on LTCC technology stand out by virtue of their compact constructions and low weight and are suitable for applications in the fields of GPS, Bluetooth, 802.11b+g, UNII and 802.11a as well as for 868–960 MHz, also available as dual & triple band. In relation to their size, the antennas offer a large bandwidth and are easy to match. The input impedance is 50 Ω. The typical gain, depending on the antenna type, is around 1 dBi. The radiation properties of the antennas can be termed plane omnidirectional. Alignment charts are presented in the ­datasheets.

Fig. 2.155: Evaluation board for the antenna 748 891 0245 The electrical parameters of the chip antennas are measured with test boards as shown in Figure 2.155. Many factors, e.g. the board type (dielectric constant) and the layout, affect the electrical properties of a chip antenna. The component values for the matching network shown in the datasheet cannot be assumed for this reason. These values only apply for the test boards used; they may however be used as starting

435

II Components values for determining the matching network in the application. The return loss can be significantly improved by matching the antenna. To achieve the performance shown in the datasheet, it is important to ensure sufficient separation between the antenna and the ground plane. The antenna is preferably placed at a corner point of the board to avoid being completely surrounded by a ground plane. Deviations from the suggested layout instructions can lead to changes in the alignment characteristics and input impedance. In general, care must be taken that the separation between the long side of the antenna and the ground plane is at least 5 mm. The separation between the antenna surface and the housing must be at least 1 mm. The microstrip feed lead can be considered as part of the antenna system. It is recommended to connect the ground plane surrounding the input lead on the edges to the lower ground with vias. This serves to ­minimize the electric field on the edges and hence its influence on the antenna. The efficiency η is the ratio between the radiant power and the input power. The ­efficiencies of the chip antennas derived from numerical calculation are presented in the following table. This includes antennas operating in the frequency range 2400–2500 MHz. Essential differences are the antenna dimensions and the gain. The standing wave ratio is a maximum of 2 for all antennas within the bandwidth. Order Code

Frequency range (MHz)

Max. gain (dBi)

Typ. gain (dBi)

Efficiency (%)

Length x width (mm)

748 891 0245

2400–2500

3

1

78

9.5 x 2.0

748 892 0245

2400–2500

1.3

0

76

7.6 x 3.5

748 893 0245

2400–2500

0.5

–0.5

60

3.2 x 1.6

748 894 0245

2400–2500

2

0.5

75

7.0 x 2.0

Tab. 2.78: Electrical characteristics of some multilayer antennas WE-MCA

9 ESD and Surge Protection 9.1 Basic principles In order to use the correct overvoltage protection element, one needs to know what type of overvoltage and which characteristics are expected. There are three common types of overvoltages. Natural overvoltages

436

Natural overvoltages: Lightning strikes are the cause of most natural overvoltages. Direct and indirect effects are distinguished. Indirect effects include the residual voltage generated by a distant lightning strike in the power grid or when induction in a building from lightning is con-

ducted away to earth through the building’s lightning rod. The common aspect of these overvoltage phenomena is the impossibility to predict the frequency and the intensity of the overvoltage. Industrial overvoltages: This type of overvoltage mainly occurs with switching events, e.g. switching off inductive loads, such as motors or frequency inverters. Compared to a lightning strike, the overvoltage and the associated current surge are usually very much lower, but often recur periodically. An overvoltage pulse of this type is usually not destructive, but its recurrence can have destructive consequences. The IEC 61000-4-5 standard deals with the immunity against industrial overvoltages as well as standardized surge immunity tests. The standardized hybrid pulse consisting of a 1.2/50 μs voltage pulse is induced in an open-circuit and an 8/20 μs current pulse in a closed-circuit. The specification 1.2/50 μs relates to a rise time Tr of 1.2 μs and a decay time to half value Td of 50 μs for the voltage or current pulse (cf. Figure 2.156). The group of components under test must each withstand 5 positive and 5 negative pulses. Surge-Impulse

Fig. 2.156: Shape of the surge impulse, defined in IEC 60060-1 respectively IEC 60469-1 Electrostatic discharges: Everybody knows these. Electrostatic discharges arise through friction between materials with differing dielectric constants. If a charged material meets a conductor the charged material discharges – the electrostatic discharge. Here we will just look at the discharge between humans and materials (HBM – Human Body Model).

Electrostatic ­discharges

One characteristic of an electrostatic discharge (ESD) is a very short current rise time. Humans cause a typical ESD pulse is described in the IEC 61000-4-2 standard (Figure 437

II Components 2.157). The current rise time lies between 0.7 and 1 ns and is therefore around 8000 times faster than the current rise during a surge pulse. ESD Pulse

Fig. 2.157: Shape of current of an ESD impulse of the HBM, defined in IEC 61000-4-2

Varistors

9.2 Varistors Functionality Varistors are components whose resistance depends on the voltage applied to both terminals. The dependence is symmetrical and non-linear. If a ramp voltage is applied to a varistor, its resistance changes very rapidly within a small voltage range and passes from the high resistance (several megohms ) to the low resistance (Ohm range) state. The main properties of varistors are described by the current-voltage curve. The varistor voltage (also known as rated voltage) is understood as the point on the characteristic curve above which the current profile is exponential.

438

Fig. 2.158: Current-voltage curve of an varistor with drawn varistor voltage The base material of today’s metal oxide varistors (MOVs) is zinc oxide (ZnO) in form of very small ZnO grains (10–100 µm). In the past silicon carbide (SiC) was another common base material. Two zinc oxide grains form a micro-varistor with a breakdown voltage of around 3 V (see Figure 2.159). The more of these micro-varistors there are connected in series, the higher the varistor voltage of the complete varistor. The more micro-varistors there are connected in parallel, the higher the current carrying capacity. The addition of further metallic compounds can further influence the behavior of the varistor.

Fig. 2.159: Intergranular layout of a varistor The varistor acts as a pressure relief valve. In normal operation (the varistor in its resting state) a very low leakage current IL flows through the varistor. The varistor has a resistance of several megohms.

Leakage current

439

II Components In the overvoltage situation, the varistor responds and shorts the circuit almost completely. The varistor now has a very small resistance. Nearly all the current is discharged through the varistor.

Fig. 2.160: Different operating stages of a varistor The equivalent circuit of a varistor can be described as follows: Equivalent circuit of a varistor

Fig. 2.161: Equivalent circuit of a varistor

440

LZ: CIG: RIG: RVar: RZ:

conducting inductance (1nH/mm) intergranular capacitance intergranular resistance (several MW) ideal varistor (0 to ∞ W) line resistance (a few mW)

WE-VD disk varistors The core of the disk varistor is made of zinc oxide as described above. In addition, the components are metallized by a silver coating (in some cases copper coating) on both sides for contacting the electrical terminals. Lastly, the insulation and the protective sheathing is applied.

Fig. 2.162: Production stages of disk varistors The following print is on the coating of the WE-VD disk varistor:

Fig. 2.163: Print on WE-VD Disk varistors Disk varistors are available in sizes 5 mm, 7 mm, 10 mm, 14 mm and 20 mm and voltages of 18 VDC (14 VRMS) to 1465 VDC (1000 VRMS). The maximum peak current ranges from 100 A through to 10000 A, the max. energy compatibility from 0.07 J up to 496 J. Besides the Standard series, there is also the High Surge series. Both are mechanically the same, but the High Surge series differs with its higher current loading capacity and a higher energy absorption capacity. A special sample kit (Order Number: 820999) is available for developers appropriate for supply lines of 24 VDC to 400 VRMS. This includes varistors from 26 VDC (20 VRMS) to 895 VDC (680 VRMS). 441

II Components The WE-VD disk varistors are licensed (component-dependent) for VDE, UL and CSA. The coating is flame-retardant and self-extinguishing in accordance with UL 94-V0. Serie

Order Code

VDE File No.

Standard

820x5

40016986

HighSurge

820x4

40016998

UL File No. E244196

Tab. 2.79: Overview of the different safety approvals of WE-VD disk varistors SMD varistors

SMD varistors (WE-VS) The construction of SMD varistors is similar to that of SMD capacitors, however with the difference that zinc oxide is used in place of the dielectric.

WE-VS

Fig. 2.164: Structural layout of SMD varistors In order to ensure outstanding solderability, all WE-VD SMD varistors have a nickel barrier layer. The connection pad is structured as follows:

varistor body

Tin Nickel Interior electrodes

Silver

Fig. 2.165: Metallization of the connection of WE-VS SMD varistors A Standard and a HighSurge series is also available for the SMD varistors. The package types available are 0402, 0603, 0805 and 1206. The maximum current loading capacity ranges from 10 A to 200 A, the maximum energy absorption capacity from 0.02 J to 1.1 J.

Response time 442

The WE-VS SMD varistors vary from disk varistors in their significantly faster response time. Because of the absence of lead wire inductance, WE-VS SMDs have a response time of