Inside switching power supplies

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zyxwvutsrqponmlkjihgfedcbaZYXWVUTSRQPONMLKJIHGFEDCBA no power. Actual voltage drop varlow noise and ripple, and comA DVA NT A GES OF R EGUL A T ED zyxwvutsrqponmlkjihgfedcbaZYXWVUTSRQPONMLKJIHGFEDCBA ies depending on the transistor plete freedom from EMI. It does, .,_switching power supplies are too and the current level, but is usuhowever, waste power in the reggreat to ignore. These versatile ally less than or equal to one volt. ulating transistor, especially at supplies are well known for their (Power FET’s respond better in high load currents. Regulating high efficiency, cool operation, high-current applications.) widely-varying inputs is a probsmall size, and the ability to work Power dissipation still occurs in lem because power dissipation with a wider range of input voltfilter resistor Rl, reducing overall increases as Vu,,,, goes up. ages than their linear countercircuit efficiency Now let’s look at how pulseparts. Once limited to highwidth modulation (PWM) conpower or high-efficiency applicatrols voltage. As shown in Fig. 2, Practical circuits tions, they are now finding their 91 is alternately turned on and To reduce resistive power way into low-power, low-cost conoff by the PWM control circuitry losses, switching regulators use sumer goods. The output is R-C filtered to obL-C, rather than R-C filters, as Because the control elements tain a DC average. If Ql is always shown in Fig. 3. When 91 is on, used in switching regulators are VUNREG is applied to inductor Ll off, the output voltage will be always either fully on or fully off, zero: if it is always on, the output and Dl is reverse-biased. The inthey have low power consumpwill equal the input. The output ductive current supplies the load tion and require little or no heat voltage will be proportional to the and also charges output capacisinking. Small-size high-freduty cycle, which is the ratio of tor C2. quency transformers can be the “on” time to the total period. When 91 turns off, the inducused, and, since regulation effitive current continues, flowing ciency is not too affected by the V OU T = VIN X (T ON ~T OT AL) through Dl. The diode conducts input-to-output voltage differen= ‘/IN x Duty Cycle until the inductor current retial, it’s possible to handle two-toduces to zero, or until Ql is again If 91 were ideal (no voltage drop one input variations, such as turned on, whichever occurs in the on state) it would dissipate 115/230 volt operation. Switching regulators do, however, have disadvantages. A primary drawback is their complexity, and therefore circuit cost. They also exhibit failure modes not seen in simple linear regulators, and can radiate substantial electromagnetic interference (EM11 if not properly designed. Fortunately, a number of IC’s have been developed that not only include most of the complex circuitry, but also overcome common failure modes, which well look at later. Now we’ll concentrate on basics. TI-Jk

Inside SWITCHING POWER SUPPLIES

Switching regulator basics Let’s begin by reviewing a linear (non-switching) regulator as shown in Fig. 1. Op-Amp ICl compares feedback voltage V,, to reference voltage V,,, . If V,, is too high, Ql’s base voltage decreases or, if it’s too low, the base voltage increases, until V, equals V,,, . At equilibrium, Ql’s emitter-collector voltage drop equals Vu,,,, - V=,. The transistor power dissipation, W, equals WUNREG -VREG) X 1. A well-designed linear regulator can provide excellent regulation and transient response,

Learn the basics of switching regulators-the heart of switching power supplies.

HARRY L. TRIETLEY

converter” (Fig. 8) is better suited for high-pow er supplies. When 91 turns on, the unregulated input is applied to the first winding and Dl is reversed-biased. The p rimary current begins to rise and a voltage is induced in the output w inding. Output current flows through D2 and Ll. When Ql turns off, the collapsing field induces reverse-polarity voltages in all three w indings. Since Ql is off and D3 is reversebiased, their w indings carry no current. Current flow s through the middle winding, known as a “ reset” winding, and D2 becomes forward-biased. During that time the inductive current in Ll flows through D3. As long as D2 conducts, the reset winding is connected to the’ inp ut v o ltage. That co nd itio n co ntinues until the current reduces to zero. There are two adv antag es o f that c irc u it: the average primary current is zero, and the w ind ing v o ltages are w e ll- d e f ine d d u ring the o f f portion of the cycle. A smaller c o re c an b e u se d , and hig h flyback voltages are not a problem. To maintain zero average current, the on time must never be longer than the off time, so the duty cycle is limited to 50%. The output voltage, V,,,:,, is VUNREG x N x Duty Cycle. The output and input grounds are tied to gether in Fig. 8 fo r proper feedback voltage. To provide input-to-output isolation it is also necessary to isolate the feedback. We will discuss ways to do that in a future issue. Finally, the push-pull circuit shown in Fig. 9 is similar to a DCto -DC inverter, but with pulsew id th mo d ulatio n ad d ed . That circuit provides the best efficiency in high-power converters. The primary w inding’ s center tap is connected to VU,,&. Transistors Ql and Q2 are under the control of the switching regulator c irc u it. The y are alte rnate ly pulsed on, connecting first one end of the primary and then the o ther to co mmo n. Raising the duty cycle increases the average applied voltage, and therefore the output voltage. Each transistor’s duty cycle is limited to 50%. (we must not have both turned on at once). but since there are tw o, the overall duty cycle can app ro ach 100% . A g ain, iso lated

FIG. 8-IN A FORWARD CONVERTER direct-coupled feedback provides optimum regulation, but no input-output isolation.

FIG. 4-A PUSH-PULL CONVERTER, similar to a DC-DC inverter but with pulse-width modulation, provides best efficiency for high-power supplies.

CLOCK

CLOCK RAMP COMPARATOR

FIG. IO-ADDING A STEERING FLIP-FLOP and a pair of modulator with push-pull output.

feedback is needed if input-tooutput isolation is required. The output voltage, V,,,,is the same as the forward converter V U N REG x N x Duty Cycle.

Controlling the two transistors requires a change in the control circuitry, so let’s examine the IC again. Figure 10 is similar to Fig.

NOR gates

produces a pulse-width

5, w ith output-steering circuitry added. The clock pulses toggle the steering flip-flop. At the start o f each cycle, w hen the co mparator’s output is low, the NOR gate w ho se Q input is low w ill turn on. The other remains off until the start of the next cycle toggles the flip-flop. Figure 10

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CONTINUOUS CURRENT. a

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Q--f+j+ DISCONTINUOUS CURRENT. b

FIG. ll-CONTINUOUS (a) and discontinuous (b) inductor current.

SEC;o"/; RESISTANCE

RESISTANCE

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INDUCTANCE FIG. 12-A TRANSFORMER MODEL showing winding resistances and leakage inductances. Stray capacitances and core losses are not included.

shows the timing wavetorms. An IC of that type is very versatile, and can be used in all the circuits we have examined. Sing le- o u tp u t c o ntro l is im p lemented by simply paralleling Ql and Q2. For forward converters, the 50% duty cycle limitation is easily provided by using only Ql as the drive.

Which one should I use? W e hav e exam ined six circ u its- three w itho u t transf o rmers (buck, bo o st, and buckboost) and three w ith (flyback, fo rw ard , and p ush-p ull). Let’ s take some time now to compare the advantages and drawbacks of those techniques. Transfo rmer-co up led circuits are more flexible in stepping voltages up and down, and can prov id e inp ut-to -o utp ut iso latio n. Negative outputs require only reversal of the rectifier diodes, and multiple secondaries can be used to provide multiple output voltag es. The m ain d raw back o f transfo rmer-co upled circuits is the cos, and the size of the trans-

former itself. w ith a balanced current. A lso , both produce lower output-ripple The cho ice am o ng transforcurrent than the flyback. As a remerless circuits is often simple. sult, smaller transfo rmers and Use the buck circuit (Fig. 3) for filter components may be used. voltage stepdown, where the outInput peak and output ripple curput is lower than the input; the rents are higher in the forw ard boost circuit (Fig. 4) for step-up; converter, because its duty cycle or the buck-boost circuit (Fig. 6) is limited to under 50%. Both are for polarity inversion. A ll three well-suited for use at tens to hunuse the same number of compodreds of w atts, but for highest nents and have similar control p o w er (esp ecially abo v e 1000 requirements. One performance w atts) a p ush-p ull co nv erter difference is w orth noting: the buck co nverter tend s to- hav e should be chosen. lower output ripple because the Discontinuous operation inductor aids in filtering the outFor most efficient operation in put current. any o f the circuits w e’ v e d isWhen designing those circuits c u ssed , the ind u c to r c u rrent you must take into account the should flow continuously: otherpeak voltages and currents in the w ise, rip p le currents w ill intransistors and diodes to ensure crease and regulation may suffer. that tho se co mpo nents o perate That effect is most apparent in w ithin their sp ecified ratings. transfo rmerless circuits. Tho se The buck converter operates with circuits depend on enera’ stored lo w er p eak currents than the during the on cycle being transothers, due to the filtering action ferred to the o utp ut w hen the of the inductor. Peak currents in transistor turns off. If the inducthe transisto r and d io d e equal tance is too low, all of its stored the o utp ut currents, w hile the energy will be transferred to the p eak v o ltages equal the inp ut o u tp u t b ef o re the transisto r voltages. turns back on. In a boost converter, peak tranContinuous operation results sisto r and d io d e currents, I,,,, w hen the p eak-to -p eak rip p le equalzyxwvutsrqponmlkjihgfedcbaZYXWVUTSRQPONMLKJIHGFEDCB current in the ind ucto r is less (OUT __. X WOU T /~I N ). The peak voltage kquals the outthan twice the inductor’s load. or put voltage. D C av erag e current: in o ther In a buck-bo o st sup p ly, the w ords, w hen the inductance is peak current, I,,,, equals large enough that the negative excursio n o f its rip p le nev er louT/l-Duty Cycle. reaches zero. Figure 11-a shoivs The peak voltage equals the sum continuous operation. while Fig. of the input and output voltages. 1 l-b shows discontinuous operaOne drawback of the boost cirtion. For example, in the buck cuit sho uld be mentio ned . Beconverter of Fig. 3. continuous c au s e th e i n p u t i s d i re c tl y o p eratio n means that ind ucto r connected to the output through current is always flo\ving into the the inductor and diode, it is not lo ad . M aintaining co ntinuo us possible to use short-circuit limo p eratio n in a sw itching regiting in the circuit. ulator is usually a simple matter The flyback converter (Fig. 7) of choosing a large enough inretains the advantages (cost and ductor. simplicity) and drawbacks (hi&h Discontinuous operation norpeak currents, high ripple, and mally occurs at low output loads. DC coil current) of a transformerw hen the DC current is so low co up led circuit. It’ s the best that the negative excursion canchoice w hen a simple, low -cost no t be kep t abo v e z ero . Fo rcircuit is needed to regulate up to tunately. d isco ntinuo us o p eratens of w atts. Peak sw itch curtio n is no t d isastro u s. o nly rent, I,,,, of a flyback converter is annoying, if it only happens unIOUT x W x VIN + V OU T )/V I N Fo rw ard and p ush-p ull co nder abnormally light loading. A verters (Figs 8. and 9) are best for d ecrease in regulatio n and increase in ripple are the usual rere g u latin g h ig h e r p o w e r, g w hether iso latio n is need ed o r sult. no t. Bo th require extra w ind On the other hand, if the prob- g ings, inductors and circuitry, but lem o ccurs und er heav y lo ad s ;; bo th p ro v id e the transfo rm er due to poor design (improper in- ‘D 61

ductor selection) the result may be core saturation, excessive current spikes and destruction of components such as the switching transistor.

current of AI, the inductor value can be found by L = Ei2(Al)f

best to start with a little extra inductance, then optimize it experimentally Inductor values Inductors and transformers in the medium to high microhenLet’s finish our discussion by ries are common. looking at inductors and transSwitching regulators operate formers. The design of switchat high frequencies and fast ing- regu lator magnetics is a risetimes, and switching trancomplex subject which we cansients can produce peak voltages higher than the values given earnot cover completely in this article. We will, however, briefly dislier. ‘h-ansformers with switched cuss some of the more important primary currents are the main concepts such as physical size, source of that problem. construction, ratings, and leakA major source of primary-side age inductance. spikes is leakage inductance. Our first consideration is size. Figure 12 shows a transformer The inductance of a choke or model including winding resistra ns former mu s t b e la rge tances and leakage inductances. enough to keep ripple current (Winding capacitances are not shown.) In an ideal transformer within acceptable bounds and to maintain continuous operation. there would be perfect magnetic The core must not saturate at its coupling between the primary highest current. Some of the deand secondary. A voltage spike could not appear across the prisign tradeoffs inclu de size, power, filtering and transient remary unless a proportional spike sponse. Larger inductances and was seen on the secondary If cores provide highest power and there was a load across the secondary, especially when a capacilowest ripple, but with slow recovery from transients. tor is used, spikes would not Cores should be a ferrite mateoccur. rial or powdered iron-laminaIn reality, a small portion of the tions are not suitable for highflux produced by the primary is frequency operation. Toroidal not coupled to the secondary. cores minimize EM1 because they Electrically, that means that a tend to be self-shielding. Air gaps small part of the primary’s inducusually are needed to prevent sattance is not coupled to the securation with unbalanced DC curondary, and vice-versa. Transrents. The gap reduces the core’s former leakage inductance is permeability, requiring larger represented in Fig. 12. Switched structures to achieve the reprimary currents produce spikes quired inductance. When buying in the leakage inductance. an inductor or transformer make Leakage inductance can be minimized, but not completely sure it is rated for the frequencies and DC currents you will be apeliminated, by proper transplying to it. The affect of saturaformer design. The best approach is a bifilar winding, where tion could be the destruction of switching transistors, control the primary and secondary are IC’s or other components in the wound together, their wires intercircuit. mixed in the same coil. That may An approximate inductance not be possible in transformers value can be calculated from requiring high primary-to-secbasic inductor theory. Inductor ondary breakdown voltages. It’s current increases linearly with sometimes necessary to add Zener diodes and/or small capactime when a DC voltage is appliedzyxwvutsrqponmlkjihgfedcbaZYXWV AI=ExTIL itors, across the primary to prowhere AI is the change in current tect the switching transistors in amps, E is the applied voltage and diodes. in volts, T is time in seconds, and In the second and final part of L is inductance in henrys. this article, we’ll look at some more protective and safeguard If your circuit operates at a frequency in hertz equal to lAY, the circuitry provided in switching maximum voltage across the inregulator IC’s. We’ll also examine ductor is E and you want to desome IC families with which you should be familiar. sign for a peak-to-peak ripple R-E It’s

of each cycle, the oscillator pulse , The SG3524/5/0/7 IC IhI OPR LAST EDITION, WE EXAMINEDzyxwvutsrqponmlkjihgfedcbaZYXWVUTSRQPONMLKJIHGFEDCBA &eb-asics of switching regulator power supplies. Now we’ll dive into some real-world applications. Well examine the 3524/ 5/ 7 and 3842 IC families in detail, summarize others, and show some typical circuits. In the process we’ll study how to select components for those circuits and learn m o re abo ut ho w switching regulator IC’s are protected against such problems as startup current surges, undervoltage, and overload. We’ll finish up with some basic troubleshooting hints. Let’s first start off with an explanation of the standard nomenclature used in naming the IC’s we will discuss in this article. The first digit “ 1” indicates full military temperature range of -55 to 150°C “ 2” indicates an industrial temperature range of -25 to 85°C. and “ 3” is a commercial temperature range of 0 to 70°C. Hereafter, we will refer only to the commercial version IC’s, with prefix “ 3.” Keep in mind that all those IC’s are also available in military and industrial versions. A suffix of “ A, $9 “ I$” or “ C” indi_ cates an enhanced version of the IC, which we will discuss in more detail later in this article. Manufacturers may use many different prefixes, some of which include SG-Signetics, SGS-Tho mso n, Motorola, and Linear Technology. CS-Cherry Semiconductor. XR-Exar. CA-GE-RCA. IC-IPS

Figure 1 shows the internal circuit of the switching regulator IC 563524. In that circuit, the oscillator produces both ramp and pulse outputs. Ignoring the current limit (CL) and shutdown circ u its f o r the m o m ent, the comparator’s output goes high when the ramp exceeds the output of the error amplifier. The NOR gates then go low, turning the output transistors off. Each NOR gate can be high only when its three inputs are low. The oscillator output toggles the flip-flop, enabling one gate, and then the other to respond to the comparator. That action gates one transistor on at a time, providing push-pull operation. The selected transistor turns on at the start of each cycle, and turns off as soon as the ramp exceeds the error signal. At the end

momentarily forces both gates low, protecting against the possibility of both transistors being on at the same time. The current-limit amplifier protects against current overloads. Its output is an open-collector type-open-circuit w hen high, pull-down to ground when low. The current-limit amplifier and the shutdown transistor can be used to force the comparator output high, shutting down both transistors. Figure 2 shows the SG3524 in a simple DC-DC converter. The oscillator frequency of about 60 kHz is set by R5 and C2. (The flipflop divides the push-pull output frequency to 30 kHz.1 The current-limit amplifier goes low when its input exceeds 0.2 volts, limiting Rll’s current to 2 amps in case of overload or transformer

Inside SWITCHING POWER SUPPLIES “&‘%?&_ _

.

HARRY L. TRIETLEY

LTSG-Linear ‘Technology LM-National Semiconductor. UC-Unitrode, Motorola, Linear Technology, and Signetics. UD-SGS-Thomson. IP-IPS .

LA S-Lambda.zyxwvutsrqponmlkjihgfedcbaZYXWVUTSRQPONMLKJIHGFEDCBA

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We Tl take an in-depth look at two switching regulator families, with some tions, and guide you through basic troubleshooting techniques.

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saturation. Transistors Ql and Q2 are used for switching transformer current. (The on-chip transistors are rated at only 100 mA.1 Supply pulses produced by the circuit are filtered by C4. The output of the error amplifier is proportional to the difference between the reference input (pin 2) and the feedback (pin 1). If the output increases, the error voltage drops. The ramp then reaches the error voltage more quickly and the transistors turn off sooner, until the output

is reduced back to 5 volts. Since the feedback voltage and ground are directly connected, input-tooutput isolation is not provided. Resistors R6 and R7 limit the current through the internal drive transistors, which are used to switch zyxwvutsrqponmlkjih Ql and Q2. Frequency compensation for closed-loop stability is provided by RlO and C 3. Tra nsistors Ql and Q2 should be high-speed switching power transistors rated at least 5 amps and 60 volts. Shottky or fast-recovery diodes should be

used for Dl and D2. Be&& tl& output is balanced, the transformer core does not need to be gapped, a small ferrite core will do. At high frequencies, the equivalent series resistance (ESR) of filter capacitor C5 is higher than its capacitive impedance. Low series - res is ta nce electrolytics should be used, preferably capacitors designed specifically for switching supplies.

The enhanced SG3524A Figure 3 shows the enhanced version SG3524A, which is pincompatible and interchangeable with the non-A version. The enhanced version adds an undervoltage lockout circuit which disables the regulator until its input rises above 8 volts. That holds current drain to standby levels during turn-on, guarding against problems during startup, surges, and brownouts. A pulsewidth modulator latch is also added, which eliminates multiple pulsing in noisy environments. Set by the comparator and reset by the clock pulse, it can switch only once per comparison cycle. Further protection is provided 4 by thermal protection circuitry (not shown). Performance specifications also are improved-the 5-volt reference is trimmed more closely ( + 1%) and the error amplifiers output can swing up to the 5-volt rail. Let’s look at one more member of this family, and an application. Figure 4 shows the workings of the SG3525A/7A. The 3525A and 3527A differ only in their output logic; the 3525A is low when off, while the 3527A is high when off. (The pinouts of the 3525A/7A do not match those of the 3524 IC series.) Operation is similar to the 3524, but with added features. The oscillator has a sync input, making it easy to lock the frequencies of several supplies, eliminating problems with beat frequencies in multiple-supply boards or systems. The shutdown circuit (also included in the 3524A) and soft-start feature simplify the design of protective circuitry, as will be seen in the next application. The totem pole (push-pull) outputs, rated at maximum 500 mA, provide fast,

charge transistor (pin 7) allows control of the discharge time at the end of each ramp. That provides an ensured stoptime between output pulses so that, even with switching delays, both transistors cannot be on at once. A 47-nanosecond time constant is provided by R16-C2. The 5-volt reference (pin 16) is connected to the noninverting input (pin 2) by current limiting resistor R3, while C9 provides high-frequency bypassing. Negative feedback voltage is divided by Rl-R4, dividing the 6-volt output down to 5 volts. The basic theory of operation is similar to that of Fig. 2; the ramp is compared to the error signal to control the onoff switching of the outputs. On each cycle, the internal flip-flop selects either output A or B. The selected output is switched high at the start of each ramp and reset to low by the latch when the ramp voltage exceeds the error amp’s output. As in Fig. 2, the direct feedb a ck connection means there is no input to output isolation. Compensation for closed-loop stability is provided by R6, R7 and C4. Switching spike currents are limited byR10, Rll, and R12 in the output stages. Components C5 and R17 act as a “snubber,” limiting switching transients from the primary. When input power is first applied, Ql will be off and soft-start capacitor C3 will be discharged. As C3 is charged from the internal 50-i~,A current source, its voltage will rise, gradually increasing the voltage to the pulsewidth modulator (PWM) control’s bottom input. That gradually increases the amount of time per cycle that the output is turned on, providing a “soft” rise of the output voltage, which allows the filter capacitors to charge slowly, reducing startup current surges. If R9’s current exceeds about 3 amps (0.7-volt drop), Ql will turn on, energizing the shutdown circuit which pulls pin 8 low and discharges C3. As the current drops below 3 amps, Ql turns off, C7 discharges, the shutdown input goes low and the soft-start capacitor provides a “soft” recovery for the power supply. Pow er tr a n s for mer Tl is wound on an EE25 ferrite core (0.25-inch center leg). It feeds a

co nv entio nal full-w av e brid ge, p ro v id ing + and - o u tp u ts . Coupled inductor T2, consisting of tw o coils w ound on a cylindrical ferrite core, and the output capacitors filter the output to 50 milliv o lts p eak-to -p eak. Transistors Q2 and Q3 are 50-volt, 5amp, N - c h a n n e l p o w e r MOSFET’s. Fast-recovery diodes must be used in the rectifier due to the high frequency: Dl-D4 are lOO-volt, & diodes w ith 35nanosecond recovery.

Current-mode regulators We now turn to a different class o f sw itching regulato rs-current mode. Although the basic operating theo ry rem ains the sam e (p ulse-w id th mo d ulatio n), current-mode sw itching regulators differ in that the internal ramp is e lim inate d . In its p lac e , the ramp-like increase in the transfo rm er’ s ind uctiv e current is used for control. Figure 6 shows the basics of a current-mo d e co mp arato r. The pulse from an R-C clock sets the flip-flop, producing a high output. FET Ql turns on and transformer current begins to flow. As the inductive current ramps upward, the feedback from currentsensing resisto r R2 increases. Eventually the feedback voltage equals the error amplifiers outp ut, at w hich p o int the co mp arato r resets the flip -flo p , Ql then turns off until the next clock pulse. A s w ith previo us regulato rs, the feedback voltage, VFB, represents the filtered output. If the feedback becomes lower or high52

,

er than the reference voltage, the error signal will increase or decrease accordingly, increasing or decreasing the on time until the proper voltage is restored. Current-mode regulation offers two major advantages; pulseby-pulse current limiting, and feedforward line regu lation. Notice that the circuit in Fig. 6 contains no cu rrent- sensing comparator. Instead, each current pulse ends as soon as it exceeds the level set by the error amplifier. No matter what the cause of overload, whether transformer saturation, an output short, or input overvoltage, the circuit will limit current instantly Pulse-by-pulse limiting also eliminates the need for a separate soft-start circuit.

FEEOFOR1IJ6RD LINE REGULlU%M CLOCK

“IN

1 1 1 1

I I L_ __--...-_c-____*

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FIG. 7-FEEDFORWARD compensation of input variations is achieved when the ramp rate of the transformer’s primary current increases as the input voltage increases.

Feedforward line regulation is illustrated by the waveform shown in Fig. 7. With a fixed load,a the input voltage suddenly increases. On the very next pulse,

the inductive current, I, ramps more quickly due to the increased transformer voltage. Since the feedback and the error I) signal have not changed, the limit is reached more quickly and the pulse width becomes shorter. Changes in line voltage are, therefore, compensated before they have a chance to affect the output.

-

a

UC3842/3/4/5 Figure 8 shows the block diagram of current-mode PWM controller IC UC3842. Compared with the circuit in Fig. 6, the UC3842 adds an undervoltage lockout and an output NOR gate. The undervoltage lockout, with hysteresis, disables the output pulses until Vc, rises above 16 volts. Once started, it will not drop out unless Vc, goes below 10 volts, a feature which prevents constant toggling between “operate” and “lockout.” When disabled, the output (pin 6) goes to a h i g h - i m p e d a n c e s ta te . A “bleeder” resistor should be connected from pin 6 to ground to prevent leakage current from turning the switching FET on. The output NOR gate implements lockout, but also serves another protective fu nction. When the oscillator pulse is high, the NOR output will be low, the OR output high, and pin 6 low. The output cannot go high until the clock goes low. The clock is set up so that timing capacitor Cl charges through Rl, and discharges through the constant current sink. By choosing a larger capacitor and smaller resistor, the charging time (clock low) can be decreased and the discharge time (clock high) increased. That allows you to establish the maximum on time, or duty cycle, which is especially important in circuits where duty cycles higher than 50% can lead to transformer core saturation. The D2-D4-Rl-R2 network between the error amplifier and the current-sensing comparator reduces the error signal so that excessive power is not lost in the current-sensing resistor. The one-volt Zener diode clamps the error signal so the maximum turn-off level will never exceed one volt. UC3843 is similar to the 3842 but has a lower lockout voltage.

Intended for use at lower voltages, it operates at 8.4 volts, and drops out at 7.9 volts. UC3844 and UC3845 (not shown1 have one added feature; a flip-flop which disables the output on alternate clock cycles. That guarantees the duty cycle will always be less than 50% for circuits where that is critical.

An off-line flyback converter Figure 9 shows an SGS-Thomson UC3842 IC in an “off-line” flyback regulator. The circuit provides + 5 volts at 4 amps and ? 12 volts at 300 mA, and can deliver

27 watts. The term “off-line” means that the regulator is on the primary side of the transformer and operates directly “off the line.” The primary advantage of such a circuit is that large amounts of power can be coupled through a small, high-frequency transformer. Line operation requires high-voltage transistors and diodes, and prevents direct coupling between the output and the feedback circuit. The line voltage is rectified and filtered by BRl and Cl. Initial startup current to the IC is pro-

5 < ; 2 53

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vided by Rl. The UC3842.s underwith its low operating voltage two bifilar, parallel IO-turn 30 voltage lockout circuitry prevents and 200-i.& current drain, is ideAWG windings. Now let’s take a startup until the voltage on C2 al for battery and micropower aplook at how an optoisolator can reaches 16 volts. The 50-kHz opplications. Companion microbe used in a switching regulator. erating frequency is set by R6power device 4391 provides C6, with a maximum duty cycle regulated negative outputs from Optocoupled feedback of about 95%. The internal 5-volt positive supplies. LT1070 is the Optocouplers provide a convesupply is filtered by C5 to elimionly IC in the listing housed in a nient way of coupling isolated nate switching spikes. Currentpower IC package. feedback. Figure 10 shows a cirmode feedback is provided by cuit in which the 5-volt secondRlO, while Cl4 and R5 are used ary of a switching regulator is Troubleshooting hints for frequency compensation. controlled. If the output goes When troubleshooting switchOnce the circuit has started, above 5 volts, the inverting input ing regulators, always begin with voltage feedback comes from the the obvious. Check for input decreases below 2.5 volts and the lo-turn control winding. The optocoupler’s LED current depower and output shorts, broken voltage at pin 2 is compared to creases. That decreases the couwires, defective connectors, solthe internal 2.5-volt reference. der bridges, defective solder plers output transistor current, The voltage difference increases increasing V,, until the isolated join ts , b a d cop p er tr a ces , or decreases the duty cycle until output returns to 5 volts. scorched components, and so the voltage at pin 7 equals 13.1 on. It’s surprising how often a volts. Allowing for diode voltage good visual inspection can undrops, that corresponds to a peak cover a problem. voltage of about 14.6 volts on the Make sure you have a data control winding. The control-tosheet, pinouts of the control IC, s econda ry tu rns ra tios a re and a circuit schematic, preferachosen to produce 5- and 12-volt bly with voltages and waveforms. DC outputs. Notice that control There is such a wide variety of is from the control winding’s voltIC’s and operating modes that it’s age, the outputs are only indidifficult to troubleshoot on an inrectly regulated. Power losses due tuitive basis. Figure 11 shows a to currents in the windings, di“generic” block diagram, which odes and inductor will affect the may help you to think through outputs. Five-volt regulation is the circuit function-by-function. FIG. lO-OPTOCOUPLER FEEDBACK alWhen breadboarding tempo10% accurate, while the -+ 12-volt lows precise control of an isolated output. regulator has 5% accuracy rary components, remember that Transistor Ql is a 500-volt, 5switching regulators produce amp power MOSFET The diodes fast, high-current pulses. ConA wide selection of IC’s are fast-recovery diodes. A “snubductor size and lead dress are imOnce a new IC technology is esber” network is formed by D3-C9portant. The input filter capacitablished, the offerings multiply R12 to hold turn-off spikes below tor should be close to the IC, not a as designs advance and the mar91’s breakdown voltage. Snubber foot away. If the main source of ket expands. Switching regD4-C8-Rll slows the turn-off rise power is at a distance, add a sevulators are no exception. Voltage time until 91’s current has had a eral hundred microfarad input mode, current mode, single-endchance to decay, bypass capacitor next to the IC. ed and push-pull IC’s cover a wide ‘h-ansformer design is imporEven though you may undervariety of power levels and usertant: the air gap must be large stand the operation of switching specific applications. enough to prevent core saturaregu la tors , trou b les hooting Table 1 summarizes some of tion but small enough to mainthem can be difficult. The IC and the many IC families available. tain the required inductance. its circuitry perform many funcMost of the devices shown can be (Note that an air gap is not tions, and the failure of one can mu ltiple- s ou rced. The pa rt needed in balanced push-pull circause improper operation of the number prefixes vary from mancuits.) In the Fig. 9 circuit, an rest. For example, failure of the ufacturer to manufacturer, and EC35 ferrite core is used (+&inch feedback circuit may lead to overmany offer additional, propriedia. center leg, Ferroxcu b e voltage, overcurrent, and shuttary devices. EC35-3C8) with a 0.5 mm gap in down by one of the protective It’s not possible to fully dethe center leg. features. Is the circuit dead, unscribe all devices in an abbreviThe primary winding consists stable or out of regulation? That ated table, but the listing should of 45 turns of 26 AWG wire. The alone may often narrow the help direct you to data sheets for 12-volt windings are each 9 turns search to one particular part of IC’s to meet your needs. The 8z of 30 AWG wire, wound together the circuit. pin devices tend to be simpler to g (bifilar). The 5-volt secondary is The following hints may help apply, while the 16-pin and larger E only 4 turns, but instead of using you pinpoint the problem to a IC’S generally offer more compli$ a heavier gauge wire, four bifilar, specific area of the circuit. After 1 cated protective and “housekeepd 4-turn windings of 26 AWG wire the visual inspection, check the ing” features. 6 are used, with their ends conoutput for shorts or overloads The 35241517 and 3842-7 fam5 netted in parallel. The control and check the input source, recilies have been fully covered in 2 (feedback) winding consists ofzyxwvutsrqponmlkjihgfedcbaZYXWVUTSRQPONMLKJIHGFEDCB tifier, filter, and transformer. this article. The 4191-3 family, lid