Silicon-Based High-Sensitivity Broadband Receiver 981970880X, 9789819708802

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Xiaojun Bi

Silicon-Based High-Sensitivity Broadband Receiver

Silicon-Based High-Sensitivity Broadband Receiver

Xiaojun Bi

Silicon-Based High-Sensitivity Broadband Receiver

Xiaojun Bi School of Integrated Circuits Huazhong University of Science and Technology Wuhan, Hubei, China

ISBN 978-981-97-0880-2 ISBN 978-981-97-0881-9 (eBook) https://doi.org/10.1007/978-981-97-0881-9 © The Editor(s) (if applicable) and The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2024 This work is subject to copyright. All rights are solely and exclusively licensed by the Publisher, whether the whole or part of the material is concerned, specifically the rights of translation, reprinting, reuse of illustrations, recitation, broadcasting, reproduction on microfilms or in any other physical way, and transmission or information storage and retrieval, electronic adaptation, computer software, or by similar or dissimilar methodology now known or hereafter developed. The use of general descriptive names, registered names, trademarks, service marks, etc. in this publication does not imply, even in the absence of a specific statement, that such names are exempt from the relevant protective laws and regulations and therefore free for general use. The publisher, the authors, and the editors are safe to assume that the advice and information in this book are believed to be true and accurate at the date of publication. Neither the publisher nor the authors or the editors give a warranty, expressed or implied, with respect to the material contained herein or for any errors or omissions that may have been made. The publisher remains neutral with regard to jurisdictional claims in published maps and institutional affiliations. This Springer imprint is published by the registered company Springer Nature Singapore Pte Ltd. The registered company address is: 152 Beach Road, #21-01/04 Gateway East, Singapore 189721, Singapore Paper in this product is recyclable.

Acknowledgements

Many people deserve my thanks for their help in completing this book. I would like to thank my colleagues and students in the group of microwave and optoelectronics integrated circuits (MOIC) at the Huazhong University of Science and Technology for their support over the years. In particular, Dr. Zilan Cao, Dr. Zhen Gu, Mr. Jian Li, Mr. Ziang Xu, Mr. Caodi Sheng, Mr. Yang Hu, and Miss. Jingyu Xie made helpful contributions and suggestions. I am also grateful to the following people for providing or editing photographs: Mr. Qiang Ma and Miss. Huawei Bu. Generating an approximately 200-page error-free manuscript is next to impossible, but minimizing the number of errors is possible. I am grateful to the following teacher and graduate students for their thorough inspection of every word, figure, and equation: Prof. Xiaohong Wang, Mr. Jinxun Jin and Miss. Jingyi Wang. Because of their reviews, many errors were identified and corrected.

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Contents

1

Background of High-Sensitivity Receiver . . . . . . . . . . . . . . . . . . . . . . . . 1.1 High-Sensitivity Receiver in Wireless Systems . . . . . . . . . . . . . . . . 1.1.1 Radiometer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.1.2 Communication Systems . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.2 High-Sensitivity Optoelectronic Receiver in Wired Systems . . . . . 1.3 Advantages and Limitations of Silicon-Based Technology . . . . . . . 1.4 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

1 1 1 3 5 6 8 8

2

Silicon Device Limitations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.1 Passive Devices . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.1.1 Loss Mechanism of Passives . . . . . . . . . . . . . . . . . . . . . . . . . . 2.1.2 Transmission Line . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.1.3 Inductor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.1.4 Capacitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.1.5 Transformer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.2 Active Devices . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.2.1 Transistor Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.2.2 f T /f max . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.2.3 Noise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.3 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

9 9 10 11 13 16 18 20 20 23 25 27 27

3

High-Sensitivity Radiometer Architecture . . . . . . . . . . . . . . . . . . . . . . . 3.1 Radiometer Principles . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.2 Typical Radiometer Architectures . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.2.1 Total Power Radiometer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.2.2 Dicke Radiometer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.2.3 Null-Balance Radiometer . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.2.4 Pulse Noise Injection Radiometer . . . . . . . . . . . . . . . . . . . . . 3.3 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

29 29 31 31 33 35 37 38 38 vii

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4

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High Sensitivity W-Band Radiometer . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.1 Comparison Between the Conventional Dicke Architecture and the Proposed Radiometer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.2 Proposed SPDT-DA Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.2.1 Operation Principle of the Proposed SPDT-DA . . . . . . . . . . 4.2.2 Design Considerations of the SPDT-DA Switching TLs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.3 Proposed Receiver Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.3.1 LNA Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.3.2 Detector Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.3.3 Receiver Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.4 Experimental Results of the Radiometer . . . . . . . . . . . . . . . . . . . . . . 4.4.1 SPDT-DA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.4.2 LNA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.4.3 Detector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.4.4 Receiver Performance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.4.5 Imaging Testing Experiment . . . . . . . . . . . . . . . . . . . . . . . . . . 4.5 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

41

Interstage Reflectionless Radiometer . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.1 Conventional Radiometer Architectures with Interstage Absorptivity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.1.1 Radiometer Based on Circulator . . . . . . . . . . . . . . . . . . . . . . 5.1.2 Radiometer Based on Attenuator . . . . . . . . . . . . . . . . . . . . . . 5.1.3 Radiometer Based on Quadrature Couplers . . . . . . . . . . . . . 5.2 Radiometer with Reflectionless Matching Network . . . . . . . . . . . . . 5.2.1 Architecture of the Reflectionless Radiometer . . . . . . . . . . . 5.2.2 Structure of the Proposed RLMN . . . . . . . . . . . . . . . . . . . . . . 5.2.3 Transmission Poles Introduced by RLMN . . . . . . . . . . . . . . 5.3 Enhanced Performance of Radiometer with Reflectionless Matching Network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.3.1 Stability Improvement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.3.2 Linearity Enhancement Analysis . . . . . . . . . . . . . . . . . . . . . . 5.3.3 Conversion Gain Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.3.4 Noise Figure Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.4 Design of V-Band Radiometer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.4.1 Low Noise Amplifier with RLMN . . . . . . . . . . . . . . . . . . . . . 5.4.2 Mixer Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.5 Experimental Results of the Proposed Radiometer . . . . . . . . . . . . . . 5.5.1 Reflectionless LNA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.5.2 Radiometer Performance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.6 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

65

42 43 43 45 50 50 51 51 53 53 55 56 56 60 61 62

66 66 66 68 68 68 70 72 75 75 78 82 84 86 86 90 91 91 94 95 96

Contents

6

ix

A Dual-Band Radiometer Utilizing a Distributed Active Hotand Cold-Load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.1 Architecture of the L- and C-Band Radiometer . . . . . . . . . . . . . . . . 6.2 Wideband AHCL Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.2.1 Wideband Distributed Noise-Generation Network . . . . . . . 6.2.2 Working Mechanism of the Active Cold-Load . . . . . . . . . . . 6.2.3 Working Mechanism of the Active Hot-Load . . . . . . . . . . . . 6.2.4 Implementation and Measurement Results . . . . . . . . . . . . . . 6.3 Dual-Band Radiometer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.3.1 Dual-Band Reflectionless SPDT Switch . . . . . . . . . . . . . . . . 6.3.2 Wideband Low-Noise Amplifier . . . . . . . . . . . . . . . . . . . . . . 6.3.3 Dual-Band Reflectionless Bandpass Filter . . . . . . . . . . . . . . 6.3.4 Diplexer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.3.5 L- and C-Band Detectors Design . . . . . . . . . . . . . . . . . . . . . . 6.4 Implementation and Measurement Results . . . . . . . . . . . . . . . . . . . . 6.4.1 Responsivity Measurement . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.4.2 Thermal Sensitivity and NETD Measurements . . . . . . . . . . 6.5 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

97 98 98 99 100 102 103 104 104 106 107 108 110 110 113 113 115 116

7

High Sensitivity Optical Receiver Architecture . . . . . . . . . . . . . . . . . . . 7.1 Noise Analysis of the TIA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7.2 The Basic Noise Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7.3 The Frequency Response Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . 7.4 Influence of Gain Tuning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7.5 Trade-Off in Noise Performance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7.6 The CMOS Implementation of TIA . . . . . . . . . . . . . . . . . . . . . . . . . . 7.7 The BJT Implementation of TIA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7.8 Comparison of the MOS and BJT Implementation . . . . . . . . . . . . . 7.9 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

119 119 120 122 124 125 126 128 130 130 131

8

Ultra-Large Dynamic Range CMOS Transimpedance Amplifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8.2 PrA with the Three Controlled Current Paths . . . . . . . . . . . . . . . . . . 8.2.1 Amplifying Stage of the PrA . . . . . . . . . . . . . . . . . . . . . . . . . 8.2.2 Three Current Bleeding Paths in the PrA . . . . . . . . . . . . . . . 8.2.3 Gain Tuning Mechanisms in the Controlled Current Paths . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8.2.4 Noise Performance of the PrA . . . . . . . . . . . . . . . . . . . . . . . . 8.2.5 Non-linearity of the PrA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8.3 TIA Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8.3.1 AGC Circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8.3.2 Other Wideband Blocks: VGA, PoA and Buffer . . . . . . . . .

133 133 135 135 136 139 140 142 143 144 147

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8.3.3 Other Analog Blocks: DCOC with OPAMP, BGR, LDO . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8.4 Experimental Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8.5 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

147 149 152 153

High Sensitivity and Dynamic-Range 25 GBaud Silicon Receiver . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9.1 Background . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9.2 Current-Controlled DC Adjustment Path . . . . . . . . . . . . . . . . . . . . . . 9.2.1 Structure of the DC Adjustment Path . . . . . . . . . . . . . . . . . . 9.2.2 Sensitivity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9.2.3 Dynamic Range and Linearity . . . . . . . . . . . . . . . . . . . . . . . . 9.2.4 Bandwidth . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9.3 25 GBaud Differential Receiver Design . . . . . . . . . . . . . . . . . . . . . . . 9.3.1 TIA Core Circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9.3.2 Post Amplifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9.3.3 Buffer Stage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9.4 Experimental Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9.5 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

155 155 157 158 160 162 164 165 165 166 169 169 174 174

10 Summary and Conclusions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 177

About the Author

Dr. Xiaojun Bi is a full professor at the School of Integrated Circuits, Huazhong University of Science and Technology, Wuhan, China. He received B.S. and M.S. degrees from Huazhong University of Science and Technology (HUST), Wuhan, China, in 2005 and 2007, respectively, and Ph.D. degree from the National University of Singapore (NUS), Singapore in 2013. From 2007 to 2008, Dr. Bi worked as a research associate in the Institute of Micro-electronics, Chinese Academy of Sciences (IMECAS), Beijing, China. From 2009 to 2013, he was a research scholar with NUS and the Institute of Micro-electronics (IME), Agency for Science, Technology and Research (A*STAR) engaged in silicon-based millimeter-wave ICs for THz imaging and Gb/s wireless communication. From 2013 to 2015, he was a research scientist with IME, A*STAR and worked on high-speed IC design. In December 2014, he joined the School of Optical and Electronic Information/School of Integrated Circuits, Huazhong University of Science and Technology, Wuhan, China, where he has been a Full Professor since November 2020. Dr. Bi’s current research interests include IC design for high-speed communications and mmWave/THz imaging. More specifically, he and his team have developed silicon-based wideband driver amplifier, transimpedance amplifier, and multiplexer for 4×25 GB/ 56 GB/100 GB+ wireline communications, siliconbased W-band and V-band high-sensitivity receivers and hybrid modules for imaging and remote sensing including the world’s first single-pole-double-throw distributed amplifier that achieves the lowest switching xi

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About the Author

loss on silicon at W-band, silicon-based Q-band power amplifier and transmitter for high-speed wireless communications. Dr. Bi served as an Associate Editor for IEEE Access (2019–2022). He is a technical reviewer for the Journal of Solid-State Circuits, IEEE Transactions on Circuits and Systems I, IEEE Transactions on Microwave Theory and Techniques, etc.

Abbreviations and Acronyms

ADC AGC AHCL BPF BW CF DAC DCOC FDFC-OPAMP IF LDO LNA LO LPF MIM NETD NF PA PD PON PrA Q RF RLMN SNR SOC SPDT TIA TL VGA

Analog to digital converter Automatic gain control Active hot and cold load Bandpass filter Bandwidth Center frequency Digital to analog converter DC offset cancellation Fully differential folded cascode operational amplifier Intermediate frequency Low dropout regulator Low noise amplifier Local oscillator Lowpass filter Metal-insulator-metal Noise equivalent temperature difference Noise figure Power amplifier Photodetector Passive optical network Pre-amplifier Quality factor Radio frequency Reflectionless matching network Signal-to-noise ratio System on chip Single-pole double-throw Transimpedance amplifier Transmission line Variable gain amplifier

xiii

Chapter 1

Background of High-Sensitivity Receiver

In recent years, the high frequency circuits based on silicon technology have achieved tremendous development, which promotes the high integration, small size, low cost and high performance of wireless and wire-line communication systems or sensing systems. High-sensitivity receiver as the core component in both transmission systems, directly determines the ability of detecting or transporting signals. The receiver as presented in this book mainly includes the receivers for wireless communication systems, remote sensing systems, and optical communication systems. In addition, low-cost silicon processes that are compatible with digital back-end circuits facilitate more functionality and reconfigurability into the receivers, which further enhances the performance of receivers. The high integration of silicon-based receivers also facilitates implementing receiver arrays which are highly demanded in phased array system, multi-lane optical system, etc.

1.1 High-Sensitivity Receiver in Wireless Systems 1.1.1 Radiometer Microwave receivers, as a kind of passive remote sensing device, have attracted much attention and have been widely applied in passive security scanning, low visibility navigation, stealth target detection, nimble guidance ammunition, and other fields. The detection or imaging resolution accuracy of these mentioned application systems for the monitoring target is directly determined by the receivers’ sensitivity. For example, in relation to the measurement of microwave radiation, radiometers have made significant achievements in atmospheric, ocean and land detection [1], and have important practical significance in retrieving atmospheric temperature and humidity profiles. For example, the MP 3000 mm wave radiometer developed by Radiometrics company of the United States achieves a temperature resolution of © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2024 X. Bi, Silicon-Based High-Sensitivity Broadband Receiver, https://doi.org/10.1007/978-981-97-0881-9_1

1

2

1 Background of High-Sensitivity Receiver

0.1 K (with an integration time of 0.01–2.5 s) [2], which is mainly applied to measure the water vapor and liquid water density distribution in clouds. In the aspect of passive detection imaging, the receiver does not emit electromagnetic waves, so there is no electromagnetic pollution, and it is more suitable for concealment work than active imaging. Meanwhile, the Vela125 [3] passive imaging security instrument developed by Millivision company consists of 64 radiometer channels to form a receiving array. The system center frequency is 94 GHz with a bandwidth of 5 GHz, and the scanning field of view is 26° × 26° with a resolution of 1.5 cm at a distance of 1 m. Any object can be regarded as a radiation source, which emits electromagnetic waves, and the amplitude characteristics of the electromagnetic waves change with the outside temperature. Based on this characteristic, the microwave radiometer can be used to detect different targets by receiving the microwave energy radiated by the monitored targets themselves (Fig. 1.1). However, the microwave signal radiated by the observation target is extremely weak, which is smaller than the noise power of the radiometer system. Therefore, the radiometer is a kind of receiver with high sensitivity and gain. The critical parameters such as sensitivity, linearity and stability of receivers directly determine the detection resolution ability of the radiometer system. Among them, the sensitivity, also known as noise equivalent temperature difference, is one of the most important indexes of the radiometer, which is used to characterize the radiometer ability to distinguish brightness. The sensitivity of the radiometer mainly depends on the noise figure of the receiver, the minimum bandwidth of the system, and the additional temperature variation caused by the fluctuation of the system gain. On the other hand, for different application scenarios of radiometers, the calibration methods are also different due to the particularity of their working environments. For example, the on-orbit two-point calibration method is adopted in the spaceborne millimeter-wave radiometer, which means that the cosmic cold sky background is used as the low temperature reference source and the absorbing black body is used as the high temperature reference source. However, due to the attenuation of millimeter wave in the atmosphere, the microwave radiation of the cold space background cannot be effectively used in ground-based radiometers. Therefore, the wideband cold and hot noise source based on active transistors are usually adopted as the reference source. Radiometer for detection Antenna

LNA

Integrator Vout Filter

Object

Fig. 1.1 Architecture of the radiometer

Detector

1.1 High-Sensitivity Receiver in Wireless Systems

3

Object

...

#n

#2 Antenna

LNA

#1

Mixer Filter

ADC

Lowpass Amplifier Filter Local Oscillator

Computer

Fig. 1.2 Architecture of the synthetic aperture imaging system based high-sensitivity receivers

Furthermore, based on the high-sensitivity multi-channel receiver array, an architecture of synthetic aperture imaging system is introduced as an example in Fig. 1.2, to intuitively demonstrate the application of receivers in the imaging system. As shown in Fig. 1.2, the typical synthetic aperture radiometer imaging system structure mainly consists of antenna array, receiver array, and digital signal processing circuit. The detected signal can be received by the antenna and transmitted to the receiver, where the RF signal can be low-noise amplified, and converted to an IF signal by the mixer. After sampling, the digital signal is formed and transmitted to the digital signal processor. The brightness temperature of the targeted image can be reconstructed by various image processing algorithms, and displayed on the monitor. As the key component of this imaging system, the receiver is mainly used to measure the thermal radiation signal of the target received by the antenna and generate a signal proportional to the difference between the reference load with known temperature and the antenna temperature. It is worth noting that both the receiver and the noise source used for calibration are required to have the same phase in each channel.

1.1.2 Communication Systems Over the past few decades, wireless communication services have achieved considerable development around the world. As the critical component of the whole wireless communication system, RF transmitters and receivers directly determine the signal transmission quality. Due to that the spectrum below 6 GHz has been gradually occupied, millimeter-wave transmitters and receivers with wide spectrum resources have been used in the new generation of 5G/6G communication systems to achieve high transmission rate. The RF transmitter and receiver, located between the digital baseband sampling circuit and the antenna, are mainly used to modulate the baseband signal to the RF carrier for transmission, or to down-convert the received RF signal to the baseband signal. A typical transceiver architecture is shown in Fig. 1.3. In the transmitter,

4

1 Background of High-Sensitivity Receiver

Mixer Digital Circuit

DAC

IF

LNA

PA Antenna

RF

Mixer RF

IF

ADC

Digital Circuit

LO

LO

Transmitter

Receiver

Fig. 1.3 Typical architecture of the wireless communication system based on the transmitter and receiver

the digital signal is first converted to an analog signal through the digital-to-analog converter (DAC). After filtering and amplification, the signal is up-mixed with the local oscillator signal generated by the frequency synthesizer. Then, the output RF signals with high power are realized by the power amplifier. While, at the receiving end, the RF signal from the antenna is received and amplified by the low noise amplifier, and down-mixed with the local oscillator signal. After filtering and amplification, the signal is converted to digital signal by the analog-to-digital converter (ADC). Similar to the radiometer system mentioned above, the sensitivity of the receiver is also a key indicator in the communication system, which directly affects the received signal quality. The receiver sensitivity is used to characterize the ability to receive weak signals. The higher the sensitivity of the receiver, the lower signal power can be received. The minimum signal power detectable at the receiver input can be expressed as: Pin−min (d Bm) = −174(d Bm/H z) + N F(d B) + 10 log10 (BW ) + S N Rmin (d B)

(1.1)

where, NF, SNRmin and BW represent the noise figure, the minimum output signalto-noise ratio, the operational bandwidth of the receiver, respectively. When the bandwidth and the minimum output SNR are determined, then the sensitivity of the system is mainly determined by the noise figure. For the radiometer composed of multistage circuits, the noise figure of cascaded stages can be expressed as: N F = N F1 +

N F2 − 1 N F3 − 1 + + ··· G1 G1G2

(1.2)

N Fn (n = 1−∞) and G n (n = 1−∞) indicate the noise figure and gain of the nth circuit, respectively. Therefore, the noise figure of the system is mainly determined by the first stage. In other words, the first stage LNA in the receiver is usually required to have a low noise figure and a large gain.

1.2 High-Sensitivity Optoelectronic Receiver in Wired Systems

5

1.2 High-Sensitivity Optoelectronic Receiver in Wired Systems Different from the RF receiver in wireless transmission system, the wired fiber communication system employs a high sensitivity optical receiver to process the photocurrent signal. For the high-speed transmission demand of a great deal of data information, such as big data platforms, cloud computing and other scenarios, fiberbased optical communication technology shows obvious advantages in ultra-large bandwidth, low loss and cost, which is very suitable for high-speed signal transmission (Fig. 1.4). For instance, based on 16 nm FinFET CMOS technology, an optical receiver with 100 Gbps PAM4 signal transmission rate has been implemented by Cisco. Similarly, based on 22 nm FinFET CMOS technology, Intel has implemented a 128 Gbps PAM4 optical receiver (Fig. 1.4). Figure 1.5 shows a typical optical communication transmitter and receiver system architecture. The optical transmitter used for modulating the electrical signal to the optical signal is mainly composed of a parallel-series converter, a driver amplifier and a laser. The input parallel data is converted to single-way serial data and then amplified with high output swing in the driver amplifier for driving the laser. After transmission through a long optical fiber with several kilometers, the modulated optical signal can be demodulated by the optical receiver to recover the original electrical signal. The optical receiver is mainly composed of photodiode, transimpedance amplifier (TIA) and serial-parallel converter. Firstly, the input optical signal is converted to a current signal by the photodiode. Then, it is converted to a voltage signal and amplified with introducing low noise in the TIA stage. It’s worth noting that the optical signal in

Fig. 1.4 Typical wired optical communication system

6

1 Background of High-Sensitivity Receiver Fiber

... Parallel-series converter

Photodiode Electrooptic modulator

...

Driver Input data

TIA

Optical Transmitter

Output data

Series-parallel converter

Optical Receiver Laser

Fig. 1.5 Typical architecture of high-speed optical communication systems based on the optical transmitter and optical receiver

the optical fiber is weak, resulting in a very small electrical signal transformed by the optical detector. Therefore, the noise in the optical receiver should be as small as possible to ensure that the sensitivity of the optical receiver is high enough, and the optical fiber communication system can work normally. The performance of the optical receiver can be judged by the indicators of sensitivity, dynamic range, bit rate, etc. The sensitivity of an optical receiver is defined as the minimum incident optical power that can be processed under the required BER conditions. The higher the sensitivity, the weaker the optical signal can be detected by the optical receiver. Similarly, the sensitivity is mainly affected by the noise in the optical receiver. The dynamic range of the optical receiver is defined as the ratio of the maximum input optical power to the minimum detectable optical power. The larger the ratio is, the larger the dynamic range of the optical receiver is, and the smaller the influence caused by the input optical signal variation on the optical receiver is. The sensitivity of optical receiver can be calculated by Eq. (1.3), with the power of 1 mW as a reference. Psen = 10 log10 (P/1 mW) P = 2Qir ms /R

(1.3)

1.3 Advantages and Limitations of Silicon-Based Technology On the other hand, microwave integrated circuits are implemented in III–V technologies such as GaAs, GaN, InP, etc. As shown in Table 1.1, the III–V technologies are shown to be superior to silicon in terms of noise figure, f t , f max and power handling capability. However, driven by the demand for low cost consumer

1.3 Advantages and Limitations of Silicon-Based Technology

7

electronics and highly integrated systems, silicon process is more attractive. The study of silicon-based millimeter-wave integrated circuits has been popular in recent years due to the development of complementary metal oxide semiconductor (CMOS) and bipolar complementary metal oxide semiconductor (BiCMOS) technologies. Benefitting from down scaling, CMOS/BiCMOS transistors became small enough, and consequently fast enough, to operate in the millimeter-wave range and beyond. As a consequence, RF building blocks including power amplifier, low noise amplifier, mixer, oscillator, frequency synthesizer, etc., have been gradually implemented in CMOS and BiCMOS technologies for microwave and millimeter-wave bands. Therefore, it has great significance and application value to study the high sensitivity receiver based on silicon technology. Nonetheless, having faster transistors doesn’t necessarily mean that all obstacles in realizing millimeter-wave integrated circuits in CMOS or BiCMOS technologies have been cleared. Some mechanisms in the CMOS and BiCMOS process may result in a poor performance of millimeter-wave integrated circuits. For instance, conductive losses in metals, the lossy silicon substrate, and radiation caused by metal patterns all may result in a low Q passive connection or component [9]. Many techniques such as new types of transmission line or shielding have to be used to alleviate these problems. In addition, the transistor performance may be degraded directly from the layout dependent f T and f max , due to the parasitic effects that generally exist at such a high frequency [9]. Therefore, this book mainly studies the circuit structure and design methods of novel high sensitivity receivers, utilizing these novel circuit designs to compensate the performance limitations of silicon technology, so as to achieve higher performance receivers. Table 1.1 Comparison between silicon-based and III–V compound semiconductor technologies [4–8] Silicon

InP

GaAs

1.12

1.35

1.42

Electronic mobility 1400 (cm2 /V/s)

5400

8500

Saturated electron 1.0 velocity (107 cm/s)

2.5

2.1

Breakdown field strength (MV/cm)

0.3

0.5

0.4

Dielectric constant

11.7

12.5

12.9

Technology

28 nm CMOS [5]

0.105 um SiGe HBT [6]

100 nm InP HEMT [7]

40 nm GaAs mHEMT [8]

f T / f max (GHz)

349/265(NMOS) 242/184(PMOS)

505/720

300/700

400/600

Integration ability with logical circuit

Yes

Yes

No

No

Process cost

Low

Low

High

High

Bandgap (eV)

8

1 Background of High-Sensitivity Receiver

1.4 Summary In this chapter, the practical application of the high sensitivity receiver in wireless and wired transmission system, and the impact of sensitivity on system performance are mainly introduced. Besides that, the parameters of silicon-based technology and III– V technologies are compared to illustrate the research advantages of silicon-based receiver. In subsequent chapters, several kinds of high sensitivity receivers, which are W-band low-switching-loss radiometer [10], V-band reflectionless receiver [11], Dual-band radiometer with wideband noise source for calibration [12], 2.5 Gb/s transimpedance amplifier in optical receiver [13] and 25 Gb/s high sensitivity optical receiver [9], will be discussed in detail.

References 1. R.R. Ferraro, G. Skofronick-Jackson, Y. Hong, K. Zhang, 4.02—Precipitation, in Comprehensive Remote Sensing, ed. by S. Liang (Elsevier, Oxford, 2018), pp. 4–24 2. T. Hewison, C. Gaffard, Radiometrics MP3000 Microwave Radiometer Performance Assessment (2003) 3. T.D. Williams, N.M. Vaidya, A compact, low-cost, passive MMW security scanner, in Proceedings of the Passive Millimeter-Wave Imaging Technology VIII (Orlando, Fla, USA, 2005), pp. 109–116 4. http://www.ioffe.ru/SVA/NSM/Semicond/index.html 5. M.-T. Yang et al., RF and mixed-signal performances of a low cost 28nm low-power CMOS technology for wireless system-on-chip applications, in 2011 Symposium on VLSI Technology— Digest of Technical Papers (Kyoto, Japan, 2011), pp. 40–41 6. B. Heinemann et al., SiGe HBT with fx/fmax of 505 GHz/720 GHz, in 2016 IEEE International Electron Devices Meeting (IEDM) (San Francisco, CA, USA, 2016), pp. 3.1.1–3.1.4. https:// doi.org/10.1109/IEDM.2016.7838335 7. V. Camarchia, R. Quaglia, A. Piacibello, D.P. Nguyen, H. Wang, A.-V. Pham, A review of technologies and design techniques of millimeter-wave power amplifiers. IEEE Trans. Microw. Theory Tech. Microw. Theory Tech. 68(7), 2957–2983 (2020) 8. OMMIC[Online]. https://gmsystems.com/uploads/3/4/4/4/34441255/foundry___fab__ser vices.pdf 9. X.J. Bi, et al., High sensitivity and dynamic-range 25 Gbaud/s silicon receiver chipset with current-controlled DC adjustment path and cube-shape Ge-on-Si PD. IEEE Trans. Circ. Syst. I, Reg. Pap. 67(11), 3991–4001 (2020) 10. X.J. Bi et al., A low switching-loss W-band radiometer utilizing a single-pole-double-throw distributed amplifier in 0.13 µm SiGe BiCMOS. IEEE Trans. Microw. Theory Tech. Microw. Theory Tech. 64(1), 226–238 (2016) 11. X.J. Bi, et al., An interstage-reflectionless V-band radiometer with capacitor-reused absorptive matching in 0.13 µm SiGe BiCMOS. IEEE Trans. Circ. Syst. I. Reg. Pap. 68(11), 4589–4602 (2021) 12. X.J. Bi et al., An L- and C-band radiometer utilizing distributed active hot and cold loads with 156% fractional bandwidth. IEEE Trans. Microw. Theory Tech. Microw. Theory Tech. 70(3), 1841–1855 (2022) 13. X.J. Bi, et al., Analysis and design of an ultra-large dynamic range CMOS transimpedance amplifier with automatically-controlled multi-current-bleeding paths. IEEE Trans. Circ. Syst. I, Reg. Pap. 66(9), 3266–3278 (2019)

Chapter 2

Silicon Device Limitations

Thanks to the advantages of low cost and good compatibility brought by the silicon technology, it is quite popular to implement microwave and high-speed optoelectronic circuits on silicon. For millimeter wave applications, silicon technology is inferior to III-V semiconductors in many aspects. For example, the carrier mobility of silicon is relatively low, which results in the poor gain and f T /f max of the device. The bandgap of silicon is relatively small, so the breakdown voltage of silicon devices is low, and they perform poorly in high-power applications. In addition, the silicon process makes it difficult to achieve high resistivity or semi-insulating substrates, which results in high losses in passive devices and poor isolation. These limitations are severe and have a substantial impact on high-frequency or high-speed circuits. In this chapter, the device model will be analyzed from two aspects: passive devices and active devices, and the limitations of silicon devices will also be discussed in detail.

2.1 Passive Devices An important factor leading to the wide application of millimeter wave integrated circuits is that the wavelength is greatly reduced, and the size of more passive circuits, including antennas, filters and transmission lines, are significantly reduced, so that they can be integrated on the same substrate. Although the quality factor of on-chip passive devices is usually lower than that of off-chip devices, for applications that can bear the limited performance and favor high integration, engineers still can implement the passives on-chip. For passives implemented on silicon, the loss mechanism, typical passive component structures as well as their models are discussed as follows.

© The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2024 X. Bi, Silicon-Based High-Sensitivity Broadband Receiver, https://doi.org/10.1007/978-981-97-0881-9_2

9

10

2 Silicon Device Limitations

2.1.1 Loss Mechanism of Passives Before discussing passive devices on-chip, we need to understand the energy loss mechanism of passive devices on silicon substrates, which is the basis of device model analysis. Passive devices on silicon substrates mainly suffer from the conducting loss of metal conductors, loss caused by dielectric, loss caused by radiation and insertion loss caused by circuit structure, like matching, coupling coefficient, etc.

2.1.1.1

Conductor Loss

The loss of conductor at low frequency is due to the limited conductivity of different metals. When the current flows through the conductor, it incurs a certain amount of energy loss, which is usually characterized by DC resistance. When the high-frequency signal passes through the metal wire, the energy loss is aggravated due to the skin effect and proximity effect (Fig. 2.1). For individual conductors, according to Maxwell’s equation, when the alternating current flows through the conductor, the induced magnetic field will be excited, and the change in the magnetic field will form the induced electric potential, which will also affect the current distribution. At high frequency, the magnetic field distribution is usually uneven, which will cause the current to move from the conductor center with a strong magnetic field to the conductor surface with a weak magnetic field, so that the current density on the metal surface is greater than the metal center, then the effective conductive area of the metal conductor becomes smaller and the loss increases, which is the so-called skin effect. The strength of the skin effect is usually expressed by the skin depth, which represents the depth from the metal surface when the current flows through the metal conductor. This can be expressed by the following formula:  δ=

1 2 =√ ωμσ π f μσ

(2.1)

where δ represents the skin depth of the metal conductor, σ denotes conductivity, ω denotes the frequency of current and μ denotes permeability. The loss of conductor due to skin effect can be expressed by skin resistance as follows: Rskin =

1 σδ

(2.2)

For multiple adjacent conductors, there is a proximity effect, which means that when the current in the opposite direction flows through two conductors, the magnetic field on the adjacent side of the conductor counteracts each other, and the current concentration on this side will also reduce the effective conductive area of the conductor.

2.1 Passive Devices

11 Skin Effect

Proximity Effect

Alternating Current Magnetic filed caused by an electric current

Displacement current flowing to the substrate Eddy Current

Magnetic field induced by the displacement current flowing to the substrate

Si substrate

Fig. 2.1 Field distribution and loss mechanism of metal wires on silicon substrates

2.1.1.2

Dielectric Loss

The loss of silicon substrate mainly lies in two aspects (Fig. 2.2). One is the electric field coupling mode. When high-frequency current flows through metal conductors, part of the signal is coupled to the substrate through capacitor. The higher the frequency, the smaller the capacitive reactance of the capacitor, and the easier the energy is coupled to the substrate; The other one is the magnetic field coupling method. When alternating current flows through the conductor, it generates an induced magnetic field around the conductor, which causes an induced electric potential. Due to the defect of low silicon resistivity, vortex current is formed in the substrate. For intuitive understanding, the conductor and substrate can be regarded as primary coils and secondary coils (transformer coupling) respectively.

2.1.2 Transmission Line As the frequency increases to the millimeter band, shorter transmission lines play an important role in chip, enabling larger electrical structures such as quarter-wavelength transmission lines to be achieved without consuming too much area. Moreover, the signal line and ground are always together, which makes its reference plane clearly defined and the impedance can be calculated clearly [1].  Z0 =

R + jω0 L G + jω0 C

(2.3)

12

2 Silicon Device Limitations Alternating current flowing through the conductor

Vortex current induced by magnetic field

Conduction current and displacement current flowing to the substrate

Fig. 2.2 Loss mechanisms of silicon-based substrates

γ=

 (R + jω0 L)(G + jω0 C) = α + jβ

(2.4)

The fundamental difference between transmission line and general circuit theory is that its physical size and wavelength can be compared, thus bringing wave behavior. As mentioned earlier, the signal line of a transmission line always appears together with the ground, and it has at least two conductors, so it is often described by a two-conductor wire model that differentiates the transmission line into a very small length and is modeled by a lumped element. Two complex parameters, or four real parameters, need to be introduced. R, L, G, C are the quantities of unit length, which represent series resistance, series inductance, shunt conductance and shunt capacitance of per unit length (Fig. 2.3). The inductance is the total self-inductance of two conductors, the capacitance comes from the proximity of the double conductor, the series resistance R is caused by the limited conductivity of the metal conductor in the silicon process, while the shunt conductivity is introduced by the low resistivity silicon substrate (i.e., the substrate coupling described in the previous section) and when the transmission line is implemented on a semi-insulating substrate such as GaAs, the distributed conductivity is almost zero. d V (z) = −(R + jωL)I (z) dz

(2.5)

d I (z) = −(G + jωC)V (z) dz

(2.6)

2.1 Passive Devices

13

Fig. 2.3 Transmission line model and distributed RLGC lossy equivalent model

i(z,t) + v(z,t) z

∆z i(z,t)

+ v(z,t)

i(z+∆z,t) R∆z

L∆z G∆z

+ C∆z

-

v(z+ z,t) -

∆z

2.1.3 Inductor Inductors have a wide range of applications in microwave circuits, such as RF chokes, suppression of parasitics, impedance matching, etc. Inductors generally come in the form of spiral inductors or short transmission lines. Lumped inductors are smaller in size compared to distributed circuit equivalents and demonstrate higher inductance at the same size, while distributed elements exhibit scalability and improved isolation. The inductance of a short transmission line depends only on the electrical length and the characteristic impedance Z 0 . When the characteristic impedance of the transmission line is defined, an arbitrary inductance can be obtained by varying the length, and the loss mechanism is similar to that of a transmission line. On-chip spiral inductors are made by spiral winding of single or multi-layer metal wires and can be classified as circular inductors, octagonal inductors, and rectangular inductors based on their physical structure, as shown in Fig. 2.4. In general, silicon-based technology supports rectangular and octagonal inductor structures, but fewer technologies support circular inductor structures. The main inductor parameters considered in circuit design are the inductance L and the quality factor Q. The ideal inductor value is fixed and has an infinite quality factor, independent of frequency. Actual inductors are non-ideal and are accompanied by various losses and parasitics. The actual inductor can be considered as a complex two-port network and its input impedance [1] is defined as: Z in = R + j X =

V I∗ V 2P 2[Pl + 2 jω(Wm − We )] = = = 2 2 I |I | |I | |I |2

(2.7)

where Z in is the input impedance, R is the resistance part, X is the reactance part, the port voltage and current are V and I, the angular frequency of the current is ω, and P is the power of a particular port at a specific frequency, W m and W e represent the maximum magnetic field energy and electric field energy, respectively. To derive the equivalent inductance, the 2-port network can be considered as a π-shaped structure, as shown in Fig. 2.5 [1]. The 2-port network parameters are denoted using the Y-parameter. For a passive reciprocal network, Y 12 = Y 21 . If the 2-port network is a symmetrical structure, there

14

2 Silicon Device Limitations

Fig. 2.4 On-chip inductors with different physical structures Fig. 2.5 Equivalent π-network model of two-port network

-Y12

P1

Y11+Y12

[ YY

11 21

Y12 Y22

P2

]

Y22+Y12

is Y 11 = Y 22 . The silicon-based substrate is lossy, and when energy flows from port 1 to port 2, partial energy flows to the substrate and introduces loss, so when one of the ports is grounded (assumed to be port 2), the input admittance of another port is: Yin = (−Y12 ) + (Y11 + Y12 )

(2.8)

The input impedance can be expressed as: 1 1 = = Rse + jωL se Y11 (Y11 + Y12 ) + (−Y12 )

(2.9)

Therefore, the effective inductance of the inductor is expressed as:

L=

Im(Z in ) = ω

  Im 1 Y11 ω

(2.10)

If the substrate is an insulating substrate, energy only transports between the two ports, and it does not reduce in the form of substrate loss. Meanwhile, Y 11 + Y 22 , Y 22 + Y 12 in Fig. 2.6a equal zero, and the input impedance becomes: 1 1 = −Y12 (−Y12 ) + (Y11 + Y12 ) The effective inductance of the insulated substrate is expressed as:

(2.11)

2.1 Passive Devices

15

-Y12

P1

P1 Y11-Y12

Y22+Y12 Y11+Y12

Y11

-Y12

(a) -Y12

P1 Y11+Y12

P2

-Y12

P1

P2

Y22+Y12 Y11+Y12

Y22+Y12

(b) Fig. 2.6 Network transformation under a single-ended drive and b differential drive

   Im −1 Y12 Im(Z in ) = L= ω ω

(2.12)

When the inductor is applied to differential structure, the differential impedance between the two ports in Fig. 2.6b can be expressed in terms of the Y-parameter as:     1 Y11 + 2Y12 + Y22 1 1 // = + Zd = − 2 Y12 Y11 + Y12 Y22 + Y12 Y11 Y22 − Y12

(2.13)

The differential inductance can be expressed as: Ld =

Im(Z d ) ω

(2.14)

Note that if the substrate is insulated, the expression of the differential inductance is the same as the expression of the single-ended inductance of the insulated substrate. Quality factor is another parameter that evaluates the performance of an inductor and is generally expressed as Q. There are many forms of expressions for quality factor, and the defining is: Q = 2π

total ener g y stor ed ener g y consumed in a period

(2.15)

That is, the quality factor is the ratio of the total energy stored in the inductor to the energy it consumes in a period, and the total energy stored is the sum of the electric field energy and the magnetic field energy, and all other defined forms can be derived from Eq. (2.15).

16

2 Silicon Device Limitations

Take the stored energy and consumed energy into the above equation to derive another definition of quality factor: 

W ¯ m − W ¯ e

Im(Y11 ) Wm − We Im(Z in ) = 2w =− Q = 2π = Pl T Pl Re(Z in ) Re(Y11 )

(2.16)

The inductance can be used to define the quality factor through the expression of single-ended input impedance: ωL R

(2.17)

Im(Z in ) Im(Y11 ) ωL = =− R Re(Z in ) Re(Y11 )

(2.18)

Q= Q=

Similarly, the quality factor of the inductor on insulated substrate can be obtained. For differential inductors, the quality factor is calculated as:   Y11 +2Y12 +Y22 Im 2 Im(Z d ) Y11 Y22 −Y12   = Qd = Y +2Y +Y Re(Z d ) Re 11Y Y 12−Y 2 22 11 22

(2.19)

12

The quality factor of differential inductor on insulated substrate can be simplified as:  Im 1 Y12 Im(Z d )  = Qd = Re(Z d ) Re 1 Y12

(2.20)

2.1.4 Capacitor In microwave or millimeter wave circuits, different capacitor forms are used for different applications. The capacitors in matching networks and resonators are usually realized by transmission lines, while decoupling and DC blocking are generally realized by lumped capacitors. The capacitor of the transmission line depends on its electrical length and characteristic impedance Z 0 , which is the same as that of the distributed inductor, and the loss mechanism and transmission line are also similar. The lumped capacitor needs to accurately model its behavior over a wide frequency band, which is convenient for circuit design. For ideal decoupling and DC blocking, the DC impedance of the capacitor is infinite and the impedance at the frequency of interest is zero, so the capacitor requires a high quality factor and a self-resonance frequency (SRF) higher than the frequency of interest. However, even if SRF is lower than the frequency of

2.1 Passive Devices

17

interest (capacitor becomes inductive), it can still be used as a decoupling capacitor and DC block as long as the impedance at the frequency of interest is small enough. If the impedance is not small enough, then accurate modeling is required to introduce parasitic parameters into the circuit for analysis. Interdigital capacitors, also known as MOM (metal-oxide-metal) capacitors, are easily manufactured in modern silicon-based process. The interdigital capacitor is connected to either port of the device by a large number of parallel fingers as shown in Fig. 2.7a. Each finger is stacked with all available metal layers, which reduces resistance and increases quality factor and SRF. Each layer of metal is connected with as many vias as possible, providing additional capacitance from the vias to the vias. However, in the presence of silicon substrate, there is substrate parasitic capacitance, which can be alleviated by eliminating the lowest metal layer, but the quality factor of the capacitor will be reduced. In more advanced silicon processes, additional high-k dielectric layers are provided. They can be used to constitute MIM (metal–insulator-metal) capacitors. The advantage of MIM capacitors is that they have a higher density and may occupy smaller area than interdigital capacitors. Therefore, the size and additional cost of capacitors need to be carefully considered. An equivalent circuit model of a symmetrical DC blocking capacitor is shown in Fig. 2.8. Series R1, C 1, L 1 compose the model of a capacitor, parallel branch C 2 is the capacitor introduced by oxides, C 3 is the capacitor of a silicon substrate, and R2 is produced by a very small conductivity of silicon. When modeling the capacitor, the element values in the equivalent circuit need to be optimized so that the network parameters and measurement results are as close as possible. It is important to note that the value of R1 tends to be only a few hundred m, and it is difficult to extract the corresponding value only through S parameter optimization because the 50  resistor in series with the port overshadows R1. Therefore, the Y and Z parameters are used for optimization simultaneously. The R1 C 1 L 1 branch and the C 2 C 3 R2 branch have smaller and larger impedance, respectively. Correspondingly, they play a dominant role in the Y and Z parameters, respectively.

(a) Fig. 2.7 MOM capacitor structure a top view b section view

(b)

18 Fig. 2.8 Equivalent circuit model of a symmetrical DC blocking capacitor

2 Silicon Device Limitations

R1

C1

L1

C2 C3

C2 R2

R2

C3

2.1.5 Transformer Transformers have many unique applications in RF and millimeter-wave circuits. They can drive balanced circuits such as Gilbert cell mixers for AC coupling and impedance matching. The compact size of the transformer, AC coupling and the ability to provide DC voltage make millimeter-wave circuit design more convenient. However, compared with inductances, the physical structure of transformers on silicon substrates is more complex, usually consisting of two or more metal coils coupled by magnetic fields, and the modeling and design of transformers are more difficult. The common passive transformers on silicon substrates are mainly planar and stacked. The primary and secondary coils of a planar transformer use the same layer of metal and are essentially two ring inductances interlaced with each other, as shown in Fig. 2.9. Stacked transformers are the same, but their primary and secondary coils use two different layers of metal. The advantages of stacked transformers are as follows: Because the design rules of the silicon-based process restrict the spacing of metal wires, the horizontal spacing of planar transformer coils is significantly larger than the vertical spacing of stacked transformer coils, which makes the magnetic field coupling of stacked transformers stronger. In addition, the two coils of the cascade structure are easy to achieve symmetry and occupy less area. Stacked transformers also have their drawbacks. In general, the lower metal layer of the silicon-based technology is thinner, which makes the lower metal coil present higher parasitic resistance. The coupling capacitance between two coils in a stacked structure is also larger than that in a planar structure. Combined with the coupling characteristics of transformers and the loss mechanism of silicon-based process, a general transformer structure can be represented by a lumped equivalent circuit model, as shown in the following Fig. 2.10. It can be seen that the transformer’s equivalent circuit model shown in Fig. 2.10 only has two ports. The other two ports are grounded because the four-port test is more demanding and the de-embedding is more complex. The equivalent circuit is mainly composed of four parts. The first part is the coupling part between coils, which is composed of mutual inductance M 12 and capacitance C 12 . The mutual inductance M 12 represents the magnetic field coupling between coils and C 12 represents the electric field coupling between coils. The second part mainly characterizes the loss

2.1 Passive Devices

19

Fig. 2.9 Planar and stacked transformer structures on silicon substrates C12 P2

P1 M12 L2

L1 Cox1

Cox2 Cs1

R11 RSub1

R2

R1 L11 L21

Cs2

Substrate Effect

R21

CSub1

RSub2 R12

L12 L22

Skin Effect & Proximity Effect

CSub2

R22

Fig. 2.10 Lumped equivalent circuit model of a transformer

and inductor characteristics of the metal wire itself, which is composed of Ri L i Rij L ij (i, j = 1, 2). Ri is the direct current resistance of the coil, L i is the low frequency inductive property of the coil, Rij and L ij represent the skin effect and proximity effect of the metal coil at high frequencies. The third part mainly characterizes the coupling between the metal coil and the silicon-based substrate, parasitic effect and loss characteristics of the substrate, which is composed of C oxi , Rsubi and C subi . C oxi is the capacitance of the oxide layer medium through which part of the signal flowing down the metal coil at high frequencies is coupled to the substrate. Rsubi and C subi represent the loss of conductivity and parasitic capacitance characteristics of the silicon substrate, respectively. The fourth part is the parasitic capacitance of the metal coil itself, expressed in C si .

20

2 Silicon Device Limitations

2.2 Active Devices In the past few decades, the low operating speed of silicon-based technology has been considered as the main problem. However, in recent years, the speed of siliconbased active devices has achieved considerable progress. What is important is that the silicon-based technology has the ability to integrate RF front-end circuits and baseband circuits, without external links. In addition, the size of silicon wafers is larger than that of GaAs and other compounds, which makes the silicon-based technology, especially CMOS, relatively cheaper. Although the performance is slightly insufficient, it is still becoming a dominant technology for operating frequency of millimeter wave and beyond.

2.2.1 Transistor Model Circuit designers usually use compact models to design and verify silicon integrated circuits. A compact model is the interface between technology and circuit design. Circuit designers do not need to learn a process through expensive and timeconsuming experiments, but only need to become familiar with the compact model. Therefore, the compact model should be geometrically scalable and accurate over a wide range of temperature and bias voltages. At present, some practical compact models have been developed for digital, analog and RF applications [2–5]. These models combine physical and empirical methods to develop general equations, and use a large number of them to describe the behavior of devices. Several parameters will be embedded in the equation to customize it for the required process or device. These parameters are determined through the complex parameter extraction process, and finally form the circuit components which are user-friendly to designers. Most of the available parameters are extracted for the purpose of low GHz frequencies, making these compact models less ideal for millimeter wave applications. The main reason is that the parameter extraction is completed at a lower frequency, which causes problems when extrapolating to high frequencies. The mechanisms of some devices are not well captured at low frequencies, so they cannot be accurately modeled, which has a great impact on the performance of devices at high frequencies. At millimeter wave frequencies, some distributed and parasitic effects need to be further considered in order to accurately simulate the operating characteristics. Gate resistance: As the operating frequency of the transistor continues to increase, the effect of gate resistance becomes significant that it cannot be ignored. These effects mainly include the following aspects: (1) the thermal noise, due to the gate resistance, increases the noise figure of the transistor; (2) the gate resistance affects the impedance matching at high frequencies; (3) the gate resistance reduces the transistor maximum oscillation frequency f max . Because of these effects, gate resistance has become a parasitic element that must be considered in high-frequency transistor modeling studies. Figure 2.11 shows a distributed gate resistance model [6]. As

2.2 Active Devices

21

Reltd

Cox S

D Rch

Fig. 2.11 Schematic diagram of distribution characteristics of gate resistance, channel resistance and gate capacitance

shown in the figure, the gate resistance of the model is divided into two parts: gate distributed resistance Rgeltd and channel distributed resistance Rgch . Rg = Rgeltd + Rgch

(2.21)

Among them, Rgeltd is hardly affected by bias conditions and is frequencyindependent, and its value can be obtained by the gate sheet resistance Reltd . Rgch is used to simulate the static channel resistance of the DC channel resistance and the excess diffusion channel resistance caused by the change of the channel charge distribution when it is excited by the AC gate voltage. Source and drain resistance: The source and drain resistances of transistors not only contribute to the DC voltage drop, but also affect the RF performance of the device, so in order to accurately characterize the high frequency of the device, the model needs to involve the source and drain resistances. The source and drain series resistance are mainly composed of the following parts: wiring resistance Rm , contact resistance Rco , via resistance Rvia , sheet resistance Rsh , LDD area sheet resistance Rldd (the sum of accumulation resistance and diffusion resistance). As shown in Fig. 2.12, usually the contact resistance Rco and the LDD area sheet resistance Rldd are the dominant components in the total resistance, they are less affected by the bias conditions [7, 8]. Coupling between source and drain: when the transistor operates at high frequencies, the coupling effect between source and drain has a significant influence on the output characteristics of the transistor. Especially in the radio frequency and Fig. 2.12 Schematic diagram of source resistance and drain resistance

Rin Gate

Rvia Rsalicide Rldd

Source

Substrate

Rco

Drain

22

2 Silicon Device Limitations

Fig. 2.13 Coupling model between source and drain

Gate

Source

Cds

Drain

Rds Substrate

millimeter wave frequency, the coupling between source and drain has dominate impact. This coupling effect can be simulated by placing a resistor and capacitor in parallel between the source and drain, as shown in Fig. 2.13. Substrate resistance: Substrate coupling effects occur in transistors using silicon technology due to the high conductivity of silicon materials and incomplete isolation between the device and the substrate. For MOSFET, there is only capacitive isolation between the device and the substrate, and at high frequencies, the signal passes through the substrate [9], and the high-frequency path of the substrate can have a great impact on the small-signal characteristics. Studies have shown that the substrate resistance has a 20% or more effect on the total output admittance of the MOSFET when the device operates at microwave frequencies or beyond. In addition, the coupling between the MOSFET drain and substrate results in a reduction in RF amplifier gain, efficiency, and f max . Therefore, the effects of substrate parasitic must be considered when studying silicon-based transistor models. Based on different understandings and analysis methods of the coupling characteristics of the silicon substrate, different models can be used to simulate the substrate parasitism. It is generally considered that when the frequency is in the several GHz range, the substrate behaves as a distributed resistance network [10], and the more resistors the network contains, the higher the accuracy and the wider the applicable frequency range. However, the more resistors the substrate network contains, the higher the complexity of the model is and the more complicated the parameter extraction becomes. Therefore, in the process of model research, the corresponding substrate network can be selected according to the accuracy and efficiency requirements. Figure 2.14 shows several commonly used substrate resistor networks. Parasitic capacitance: Parasitic capacitance is an important part of transistor model research and has a significant impact on the device’s RF performance. As shown in Fig. 2.15, the parasitic capacitance, or extrinsic capacitance of MOSFET mainly includes the following parts [11]: overlap capacitances C GSO , C GDO , C GBO between the gate and the heavily doped source region, drain region, and substrate; overlap capacitance C GSOL , C GDOL between the gate and lightly doped source and drain regions; external fringing capacitance C FO between gate and source and drain; internal fringing capacitance C FI between gate and source and drain; source and drain PN junction capacitance C JS , C JD ; substrate capacitance C SUB . The intrinsic capacitance of MOSFET, including gate to source capacitance C GSI , gate to drain capacitance C GDI , gate to substrate capacitance C GBI are also simply shown in Fig. 2.15.

2.2 Active Devices

23 D

S

CDB

CSB RDSB1

RSB

D

S

RDB

RDSB2

CDB

CSB RSB

B

B D

S

RDB

RDSB1

D

S

CSB

CDB

RSB

RDB

CDB

CSB

RSUB

B

B

Fig. 2.14 Several commonly used substrate resistor networks Fig. 2.15 MOSFET parasitic capacitance

Gate CFO CGSO CGSOL

Source

CFI

CGSI

CGDI

CFI

CGDOL CGDO CFO

Drain

CGBI CJS CGBO

Substrate

CJD

2.2.2 fT /fmax Modern semiconductor processes often quote two figures of merit to measure the performance of a transistor or active device, f T and f max . f T is the transition frequency of the transistor. The gain of the transistor at high frequencies is controlled by the capacitive element. In practice, the high frequency capability of the transistor is defined by the short-circuit current gain of the common emitter or common source. When the current gain drops to unity (i.e., 0 dB), this frequency is called the transition frequency. We take MOSFET as an example to illustrate. The following figure shows the common-source configuration and small-signal equivalent circuit of a MOS transistor (Fig. 2.16). The input current ii is:

24

2 Silicon Device Limitations io Cgd

io

Cgs +Cgb

gmvgs

ii ii vgs vgs

Fig. 2.16 Common-source configuration and small-signal equivalent circuit of a MOS transistor

i i = s Cgs + Cgb + Cgd vgs

(2.22)

i o = gm vgs

(2.23)

The output current io is:

Solving the two expressions yields the small-signal current gain as: io gm

= ii s Cgs + Cgb + Cgd

(2.24)

When the gain is unity, then: ω = ωT =

Cgs

gm + Cgb + Cgd

(2.25)

Therefore, the expression for transition frequency: fT =

gm 1 1 ωT = 2π 2π Cgs + Cgb + Cgd

(2.26)

Typical MOS device capacitance C gs is much larger than that of C gb + C gd , so the latter can be ignored in the expression. f max is the maximum oscillation frequency, and its definition comes from the unilateral gain, which was proposed by Mason [12]. If the transistor is modeled as a linear two-port network and its Y-parameter is given, then the Mason unilateral gain is written as: U=

1 |y21 − y12 | 4 g11 g22 − g12 g21

(2.27)

where gij is the real part of yij . In most cases, U is a monotonically decreasing function of frequency. When U drops to unity, the device no longer has the ability to amplify

2.2 Active Devices

25

power. The frequency at this time is called the maximum oscillation frequency, which is a very important figure of merit in the circuit design at the microwave frequency.

2.2.3 Noise In this section, we will discuss the effects of electrical noise in silicon-based integrated circuits, first by describing the types and mechanisms of noise sources, and then by specifying noise characteristics of typical silicon-based active devices. Shot noise is related to DC current and is present in diodes, MOSFETs, and BJTs. Its source can be seen by the PN junction in the diode. In the case of forward bias, the forward current of the diode consists of holes from the p region and electrons from the n region, which can overcome the potential barrier at the junction. The passage of each carrier through the junction can be modeled as a random event, depending on the carrier having sufficient energy and the velocity at directed towards the junction. The external current I appears as a steady current, actually consisting of a large number of random independent current pulses. By observing with a sensitive oscilloscope, it will be found that there are fluctuations in the current, and I D is the average value of the current. The fluctuation of I is called shot noise, and it is expressed by the mean square error of the mean value, which is written as: i 2 = (I − I D )2 1 = lim T →∞ T

T (I − I D )2 dt

(2.28)

0

It can be shown that this mean square value is: i 2 = 2q I D  f

(2.29)

where q is the electronic charge and Δf is the bandwidth. The equation shows that the mean square value of the noise current is proportional to the measurement bandwidth, that is, the noise spectral density is independent of the frequency value, indicating that the shot noise is a kind of white noise. Thermal noise arises from the random thermal motion of electrons and is not affected by the presence or absence of DC, because the electron thermal velocity in a typical conductor is much greater than the electron drift velocity. In resistor R, thermal noise can be represented by a voltage source in series or a current source in parallel, which are equivalent: v 2 = 4kT R f

(2.30)

26

2 Silicon Device Limitations

i 2 = 4kT

1 f R

(2.31)

where k is the Boltzmann constant, and the equation also states that the spectral density of thermal noise is independent of frequency, and it is also a kind of white noise. There are many sources of flicker noise, but the main reason is the pitfalls caused by crystal defects and impurities. These traps may capture or release carriers, resulting in the generation of noisy signals. Like shot noise, flicker noise is also related to DC current, and the mean square value of the noise current is expressed as follows: i2 = K

Ia f fb

(2.32)

where I is the DC current, K is a constant that depends on the particular device, a is a constant, and b is approximately equal to 1. Therefore, the spectral density of flicker noise is inversely proportional to frequency, so it is also called 1/f noise, which plays a dominant role at low frequencies. The small-signal model of a bipolar transistor (BJT) with noise can be represented by following Fig. 2.17 [13]. Diffusion and drift motions of minority carriers across the collector–base junction can be modeled as random processes. According to the previous introduction to shot noise, the collector current consists of a series of current pulses. The collector current I C exhibits shot noise, represented by the current source i c2 between the emitter and collector. Similarly, the base current I B also generates shot noise due to the random process of carrier injection, represented by the current source i b2 , which also incorporates flicker noise. The physical resistance r b that exists in the base will introduce thermal noise, which is represented by a series voltage source vb2 . vb2 = 4kT rb  f

(2.33)

i c2 = 2q Ic  f

(2.34)

i b2 = 2q I B  f + K 1

I Ba f f

(2.35)

The MOSFET small-signal circuit model with a noise source is shown in Fig. 2.18 [14]. The drain current of the MOSFET is controlled by the gate-source voltage, and the channel between the source and drain is resistive. Therefore, thermal noise represented by a source-drain current source i d2 in the equivalent circuit, is introduced, which is also the main noise of the MOSFET. This current source also contains flicker noise, because the MOSFET current is near the silicon surface and the traps will trap and release carriers.

References

27

Fig. 2.17 Small-signal model of a bipolar transistor with noise source Fig. 2.18 Small-signal model of a MOS transistor with noise source

Cgd

G

 i2g

Cgs

vi

gmvi

D ro

 i2d

S

 i d2 = 4kT

 Ia 2 gm  f + K D  f 3 f

(2.36)

where I D is the drain bias current, K is a device-specific constant, and gm is the transconductance at the device operating point.

2.3 Summary In this chapter, we focus on the analysis of passive and active devices for silicon-based technology. The loss mechanism of passive devices is explained from the principle of physical structure, and the performance of specific devices, such as transmission lines, inductors, capacitors, and transformers, is explained from the perspective of circuit principles. For active devices, the high-frequency behavior and noise sources are also briefly explained, important figures of merit are explained, and the model analysis of typical device MOSFET and BJT in silicon-based technology is carried out.

References 1. 2. 3. 4. 5.

D.M. Pozar, Microwave Engineering (Wiley, New York, 2011) The BSIM 4.4 Manual. http://www-device.eecs.berkeley.edu/bsim3 MOS11. http://www.semiconductors.philips.com/Philips Models/mosmodels/model11/ EKV model website. http://legwww.epfl.ch/ekv/ PSP model website. http://www.nxp.com/Philips Models/mos models/psp/

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2 Silicon Device Limitations

6. X.D. Jin et al., An effective gate resistance model for CMOS RF and noise modelling, in IEDM Technical Digest (1998), pp. 961–964 7. J.G. Chern et al., A new method to determine MOSFET channel length. IEEE Electron Dev. Lett. 1(9), 170–173 (1980) 8. K.L. Peng et al., Basic parameter measurement and channel broadening effect in the submicrometer MOSFET. IEEE Electron Dev. Lett. 5(11), 473–475 (1984) 9. R. Gharpurey, R.G. Meyer, Modeling and analysis of substrate coupling in integrated circuits. IEEE J. Solid-State Circuits 31(3), 344–353 (1996) 10. N.K. Verghese, D.J. Allstot, Computer-aided design considerations for mixed-signal coupling in RF integrated circuits. IEEE J. Solid-State Circuits 33(3), 314–323 (1998) 11. Y.H. Chen, C.M. Hu, MOSFET Modeling and BSIM3 User’s Guide (Springer Science & Business Media, 1999) 12. S. Mason, Power gain in feedback amplifiers. IRE Trans. Circuit Theory CT-1(2), 20–25 (1954) 13. A. Van der Ziel, Noise in Solid State Devices and Circuits (Wiley-Interscience, 1986) 14. D.G. Peterson, Noise performance of transistors. IRE Tran. Electron Dev. 9(3), 296–303 (1962)

Chapter 3

High-Sensitivity Radiometer Architecture

Radiation is generated by the interaction of molecules and atoms, and all kinds of materials radiate electromagnetic energy. For a material to absorb and radiate energy at the same rate, the material needs to be brought into thermodynamic balance with its environment. The microwave radiometer was invented according to the radiation characteristics of the object, which was used to obtain the mathematical relationship between the physical characteristics of the observed target and the amount of energy received by the radiometer. In this chapter, firstly, the principles and mechanisms of microwave radiometers are introduced. Then, two typical radiometers, including the total power radiometer and the Dicke radiometer, are illustrated. The architectures are illustrated and the derivations of the sensitivity formulae are given in detail. In addition, two advanced radiometer architectures based on the Dicke radiometer with higher sensitivity are presented.

3.1 Radiometer Principles The radiometer is one of the most basic sensors in microwave remote sensing. The radiometer itself does not emit microwave signals, but only passively receives the random microwave noise from the target and the environment. The microwave noise power emitted by environmental objectives is quite weak, generally in the order of 10−20 –10−9 W/m2 , which is even smaller than the native noise of the radiometer. Therefore, the microwave radiometer can be considered as a highly sensitive receiver used to receive and record weak random microwave noise. The microwave radiometer is generally composed of three parts, namely the antenna system, the high sensitivity receiver system and the data processing system. The key technical indicators of the radiometer mainly include temperature resolution and spatial resolution, and temperature resolution is also known as sensitivity. The spatial resolution depends on the antenna aperture D, the wavelength λ and the © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2024 X. Bi, Silicon-Based High-Sensitivity Broadband Receiver, https://doi.org/10.1007/978-981-97-0881-9_3

29

30

3 High-Sensitivity Radiometer Architecture

observation distance H. The spatial resolution can be written as W = λH/D. The improvement of spatial resolution is limited by the wavelength length and antenna size, but the radiometer provides a high temperature resolution. In some practical applications, it can make up for its relatively low spatial resolution shortcomings. The noise signal received by the radiometer is inversely proportional to the square of the distance, while the signal received by the active microwave sensor is inversely proportional to the fourth power of the distance. Therefore, the dynamic range of the distance changes detected by the radiometer is larger than that of the active microwave sensor. Since Dicke proposed the first practical switch type radiometer in 1946 [1], its development speed has been rapid. The microwave radiometers used nowadays are basically improved on the Dicke radiometer. From the original Dick switch radiometer to the noise injection control null balance feedback radiometer, the dual reference temperature automatic gain control radiometer and the imaging radiometer. From a single-band simple radiometer to a two-band, four-band, and even more multi-band radiometers. Evolution from single-purpose microwave radiometers to contemporary multipurpose radiometers. The accuracy of radiometer detection of targets has also been continuously improved, and the temperature resolution has developed from a few kelvins at the beginning to the current 0.02 K [2]. Microwave radiometers have a wide range of applications in disciplines such as radio astronomy, environmental monitoring, meteorology, marine, hydrology, agriculture and forestry, as well as in military reconnaissance. In practice, microwave radiometers are used to measure the uncorrelated electromagnetic wave energy emitted (or reflected) by the observed target itself at a certain distance. The radiation intensity and wavelength characteristics emitted by the target itself are related to the physical and chemical properties, temperature and surface state of the target, and the relevant information about the target can be inverse-performed from the data received by the microwave radiometer, ultimately obtaining the required radiation characteristics of the target and thus identifying different types of targets. There are media such as atmosphere, stratum or water between the microwave radiometer and the measured target, and the media has a certain influence on the transmission of remote sensing information. That is, the actual signal strength of the target is attenuated due to the insertion of the medium, while it provides its own information. Therefore, these effects must be considered when designing a radiometer system, selecting operating wavelengths, and interpreting remote sensing data. The microwave radiometer identifies a certain property of the target by detecting the size and change of the noise signal emitted by the target itself. Therefore, the influence of background noise emitted by other objects around the target and the propagation medium must be eliminated in data processing. With the development of electronics, microwave, computer, automation technology, other advanced techniques and equipment, microwave radiometer design, development and test research show a broad prospect. Aviation and space microwave radiometer become more and more widespread, microwave radiometer structure

3.2 Typical Radiometer Architectures

31

type constantly updated, and towards high absolute accuracy, high sensitivity, multipurpose, multi-polarization, full-band coverage and improve the direction of space resolution.

3.2 Typical Radiometer Architectures 3.2.1 Total Power Radiometer As a typical radiometer architecture, the circuit structure of the total radiometer is shown in Fig. 3.1, consisting of a receiving antenna, a high-gain, low-noise amplifier (LNA), a bandpass filter with a constant operational bandwidth, a square-law detector with high responsivity and an integrator [3]. The input signal power is amplified linearly by the LNA and detected by a square-law detector to establish a linear relationship between the DC output voltage and the input signal power. The detected output voltage V out can be expressed as shown in Eq. (3.1), where G0 represents the total gain of the system, T sys represents the noise temperature of the system, k is the Boltzmann’s constant, B is the bandwidth, C represents the responsivity of the detector, G represents RF gain, GP represents analog gain, T A represents the antenna temperature, T N represents the noise temperature. Vout = G 0 Tsys = k BC GG p (T A + TN )

(3.1)

Because the receiver itself has internal noise, the effect of internal sound should be considered in the output apart from the response of the noise signal. If the gain of the total power receiver is completely stable, then the total power radiometer has the highest sensitivity. In fact, the serious disadvantage of the total power radiometer is that it is not easy to ensure the stability of the gain in the case of high gain. This is due to the variation in gain, resulting in output fluctuations, which can be confused with real changes in the external noise signal caused by the output change. The system noise temperature (including antenna temperature and noise generated inside the receiver) is amplified, detected and integrated by the receiver to give a voltage V 0 at the output of the radiometer, which is related to the system noise Antenna Low noise amplifier

Filter

Detector

Integrator Vout

Fig. 3.1 Architecture of the total power radiometer

32

3 High-Sensitivity Radiometer Architecture

temperature T sys through the system gain G0 . Assuming that the radiometer system is linearly varying, then a change in T s results in a positive proportional variation in V 0 . Full differentiation of Eq. (3.1) yields the change in V 0 : δVo = G 0 δTsys + Tsys δG 0

(3.2)

The variation of the input noise temperature can be further obtained: δTsys =

δV0 δG 0 − G0 G0

(3.3)

0 is 0, thus only the noise temperature of the For an ideal radiometer system, δG G0 system contributes to δV0 . Therefore, the sensitivity of the system is limited only by the noise of the system. The radiometer output voltage includes the DC component of V 0 , the fluctuating component due to thermal noise and gain variation. The root-mean-square (RMS) error in T sys caused by thermal noise fluctuations is:

Tsys T A + TN = √ Tsys = √ Bτ Bτ

(3.4)

The error caused by the gain fluctuation of the system is: TG = Tsys

δG 0 G0

(3.5)

Since these two errors are considered statistically independent, the sensitivity of the total power radiometer is: T =

1/ 2  2 Tsys + (TG )2

(3.6)

Substituting Eqs. (3.4) and (3.5) into Eq. (3.6), it can be obtained:  T = Tsys

1 + Bτ

 

δG 0 G0

2 1/ 2

1 + = (T A + TN ) Bτ



δG 0 G0

2 1/ 2 (3.7)

1 0 is about 10–2 , and Bτ is 10–4 for a certain bandwidth and Assuming that δG G0 integration time, the total power radiometer has a sensitivity drop of about 100 times compared to the ideal system, which is due to the gain fluctuation.

3.2 Typical Radiometer Architectures

33

Another difficulty in the application of the total power radiometers is the inability to calculate the variation in the internal noise temperature of the receiver, reducing the accuracy of the measurement and thus limiting the use of total power radiometers. The total power radiometer shows an obvious advantage on simplified architecture and ease of implementation in the millimeter waveband. In order to improve sensitivity, the gain stability of the RF amplifier and the IF amplifier should be improved, with controlling the constant supply voltage and the ambient temperature. In addition, the RMS measurement error caused by noise fluctuation can be reduced by increasing the Bτ . However, increasing the bandwidth B is equivalent to reducing the frequency resolution in exchange for an increase in the radiometer temperature resolution. The lower limit of the integration time τ is decided by the response time of the integrator circuit and its upper limit is decided by requirements such as scanning speed.

3.2.2 Dicke Radiometer In practice, the maximum sensitivity of the power radiometer is only an ideal value obtained from theoretical calculations, since the absolute stability of the radiometer with high gain cannot be guaranteed. To address the issue, the architecture of the Dicke radiometer was proposed by R. H. Dicke in 1946, as shown in Fig. 3.2. Compared to the total power radiometer, the Dickey radiometer adds a switch and reference load to the input port and a synchronous detector circuit between the square law detector and the integrator [4]. For the gain fluctuation of the radiometer, according to the spectrum analysis, the low-frequency component plays a major role, so a switch is added to the input port near the antenna, which switches between the antenna and the reference load with known temperature at a certain rate. If the rate of conversion is higher than the highest frequency in the spectrum, meaning that the change in gain is very slow during a conversion cycle, then the effect of the gain fluctuation can be reduced. This switch is called Dicke switch. The Dicke switch is controlled by a square wave signal. Due to the switch, the radiometer receives the noise signal power from the antenna during the first half cycle of the conversion cycle and the noise power from the reference load during the second half cycle. The signals of the two half cycles are detected by the Antenna

SPDT switch

Low noise amplifier

Filter

Detector

Integrator

 Reference Noise Load

Feedback Loop

Fig. 3.2 Architecture of the Dicke radiometer

Vout

34

3 High-Sensitivity Radiometer Architecture

detector, and then rectified and compared with each other. The final output voltage is proportional to the temperature difference between the antenna and the reference load. Assuming that the radiometer output is V o , the antenna temperature is T N , the reference load temperature is T L , the system internal noise temperature is T sys , the RF gain before square-law detector is G, the analog voltage gain after square-law detector is GP , and the switching square wave period is t s . When the radiometer input is connected to the antenna and reference load respectively, the square law detector output voltages are:   Vo1 = C Gk B T A + Tsys

(3.8)

  Vo2 = C Gk B TL + Tsys

(3.9)

The total output voltage of the radiometer V o is: ⎛ Vo =

G⎜ ⎝ t

t

2

t Vo1 dt −

⎞ ⎟ Vo2 dt ⎠

(3.10)

t 2

0

In terms of Eqs. (3.8), (3.9) and (3.10), V o can also be expressed as: Vo =

1 1 C GG P k B(T A − TL ) = G 0 (T A − TL ) 2 2

(3.11)

Because the switching frequency is high enough, G0 is considered constant when V 0 is determined (In fact, G0 is slowly varying and still causes minor changes in V 0 ). Define the input temperature is T in , equal to (T A – T N ). To simplify the analysis, the following definitions are given: T A + Tsys = T0 TL + Tsys = T0

(3.12)

Thus, V o can also be expressed as: Vo =

 1  G 0 T0 − T0 2

(3.13)

Full differentiation of Eq. (3.13) yields the change in V o : δV0 =

 1 1  G 0 δT0 − δT0 + T1 δG 2 2

The sensitivity expression is given by:

(3.14)

3.2 Typical Radiometer Architectures

  2 2   2 1/ 2 2 T A + Tsys + 2 TL + Tsys 2 G T = + (T A − TL ) Bτ G

35

(3.15)

In order to improve the sensitivity of the Dicke radiometer, it is desired that  2 to be zero. One approach is to adjust the value of T L , so that the (T A − TL )2 G G T A = T L . In this case, the radiometer is regarded as being in balance. Based on this method, the null balance radiometer has been developed. Even under unbalanced conditions, the Dick radiometer is superior to the total power radiometer in terms of its reduced gain fluctuation effect. Another way is to keep the G to be zero, namely the gain of the receiver is required to be very stable. According to this theory, the radiometer with automatic gain control (AGC) has been proposed. These Dicke radiometers are designed to reduce the effect of receiver noise fluctuation and gain fluctuation at the same integration time τ . If the temperature of the reference load is close to the antenna, the sensitivity can be approximately expressed as:   2 T A + Tsys T = √ Bτ

(3.16)

The switch modulating wave and phase detection wave of Dicke radiometer are square waves, so the bandwidth of the video amplifier must be wider to ensure that the switch square wave does not distort, at least several times the switching frequency. However, too wide RF bandwidth is not conducive to interference, and the video amplifier is also susceptible to saturation due to noise, therefore, sine waves are also used as modulation or phase detection. With square wave modulation and sine wave phase detection, the bandwidth of the video amplifier can be narrower, usually using a narrow bandpass filter with a switching modulation frequency as the center frequency. After using the above sine wave synchronous demodulation, the sensitivity can be written as:   π T A + Tsys T = √ 2 Bτ

(3.17)

3.2.3 Null-Balance Radiometer As previously mentioned, in order to further eliminate the influence of gain fluctuation on sensitivity, one of the methods is to make the radiometer in a balanced state, which means trying to control T A or T L to meet T A = T L . In balance state, the output of the detector is equal in both halves of the cycle, so that the integrator output is zero [5]. In remote sensing applications, the antenna temperature changes with the measured target, so that there is usually T A = T L , the output of the integrator is not zero. The

36

3 High-Sensitivity Radiometer Architecture

solution is to use the output signal of the integrator to change T A or T L through a feedback loop, making T A and T L equal again, thus achieve automatic balance. Another method of achieving null balance is gain control, which consists of using separate IF amplification systems for T A and T L and controlling their gains so that their outputs after detection are equal. In addition, gain balance can be performed by means of the attenuator. In this way, after the noise load T L passes through the variable attenuator, the original T L is changed, so that T A and T L are re-equal, and the integrator output is zero at this time, thus a new balance is achieved. The control voltage V o of the variable attenuator is usually used as the radiometer output. When there is no loss in the transmission line and connectors, the T L is related to the attenuation factor L of the attenuator as follows:   1 Tn T0 + 1− (3.18) TL = L L where T 0 is the actual temperature of the attenuator, and T n is the effective noise temperature of the noise load. This architecture (Fig. 3.3) has certain limitations in use, because the attenuation factor of the variable attenuator varies from 1 (no attenuation) to a large value range, so the T L can only change from the maximum limit T n to the minimum limit T 0 , which means that the balance can be established only if T A also changes within this range. If T A exceeds this upper and lower limit, the balance will be destroyed. Generally, it is easy to ensure that T n > T 0 . When the detected target brightness temperature is low, that is, when T n < T 0 , the balance cannot be maintained. The solution is to cool T n and the variable attenuator to reduce T 0 , which leads to the equipment selection being complicated, and it is not suitable for airborne or spaceborne radiometers. Antenna

Feedback Loop

SPDT switch

Low noise amplifier

Filter

Detector

Integrator



Vout

Variable attenuator Reference Noise Load

Feedback OPA

Fig. 3.3 Architecture of the null-balance radiometer with feedback to control load temperature

3.2 Typical Radiometer Architectures

37

3.2.4 Pulse Noise Injection Radiometer Figure 3.4 presents the architecture of the pulse noise injection radiometer. The most stable state is when the PIN diode is on and off. The voltage-controlled oscillator (VCO) is stimulated with a feedback control signal, and the output of the voltagecontrolled oscillator drives the pulse generator [6]. The output of the pulse generator  is a series of narrow rectangular pulses, the pulse width is τ p , the period is t R = 1 f p , and f p is the pulse repetition frequency. These narrow rectangular pulses are used to modulate the PIN diode so that it acts as a switch. When there is no pulse, the PIN diode is in the off state, and its attenuation is as large as 60 dB. At this time, Tn is called T off , which is slightly higher than the ambient temperature T 0 . When there is a pulse, the PIN diode is in the on state, and its attenuation is as small as 2 dB. Meanwhile, Tn is very large, which is called T on . In this way, within the pulse repetition period t R ,  Tn

=

Ton 0 ≤ t ≤ τ p To f f τ p ≤ t ≤ t R

(3.19)

When working with the pulse modulation method, the injected noise source only works during the pulsed time. In order to ensure the variation range of the pulse frequency f p and to fit the upper and lower limits of the antenna temperature, the noise temperature T n of the injected noise source should be large enough. The pulse repetition frequency f p should be much higher than the frequency f s of the Dicke switch, so that there can be more pulses during t s /2. The average value of Tn is:   T  n = τ p f p Ton + 1 − τ p f p To f f

(3.20)

where τ p f p is the total time for the diode to be on in 1 s.

Antenna

Coupler

Feedback Loop

SPDT switch

Low noise amplifier

Filter

Detector

Vout

Reference Noise Load VCO

Pulse Generator Noise Source

Integrator

Feedback OPA

Fig. 3.4 Architecture of the pulse noise injection radiometer

38

3 High-Sensitivity Radiometer Architecture

Generally, the pulse width τ p is kept constant. In order to achieve balance, the control signal output from the feedback loop can be used to control f p to achieve the required injected noise value Tn , so f p can be regarded as the output of the radiometer. The frequency counter can be used to measure f p . There is a linear relationship between f p and T A , as shown in the following formula:   C Tco − To f f − (C − 1)T A   fp = τ p Ton − To f f

(3.21)

where C is the coupling coefficient of the directional coupler, T A is the temperature at the output of the antenna and T co is the temperature of the thermal chamber. When the PIN diode is switched off, the attenuation factor is large and the contribution of the noise source can be ignored at this point, then the above equation can be simplified to: fp =

(Tco − T A )(C − 1)   τ p Ton − To f f

(3.22)

Then replacing the antenna with a calibrated noise source of known temperature, since T 0 is known, and by measuring f p at this time, the quantity is multiplied by the (Tco − TA ) term in Eq. (3.22), so that the system is calibrated.

3.3 Summary In this chapter, we introduce the principles of the radiometer, which are composed of the working theory, basic architectural elements, application fields and development prospects of radiometers. Meanwhile, the typical architectures of the radiometer are presented, including the total power radiometer and the Dicke radiometer. In addition, two derived architectures are introduced based on the Dicke radiometer, namely the null-balance radiometer and the pulse noise injection radiometer.

References 1. R.H. Dicke, The measurement of thermal radiation at microwave frequencies. Rev. Sci. Instr. 17, 268–279 (1946) 2. D. Houtz, R. Naderpour, M. Schwank, Portable L-band radiometer (PoLRa): design and characterization. Remote Sens. 12(2780) 3. J.W. May, G.M. Rebeiz, Design and characterization of W-Band SiGe RFICs for passive millimeter-wave imaging. IEEE Trans. Microw. Theory Tech.Microw. Theory Tech. 58(5), 1420–1430 (2010) 4. L. Zhou, C.-C. Wang, Z. Chen, P. Heydari, A W-band CMOS receiver chipset for millimeter-wave radiometer systems. IEEE J. Solid-State Circuits 46(2), 378–391 (2011)

References

39

5. D. Gu, J.E. Jenkins, Noise synthesis technique in time domain for metrology application. IEEE Trans. Instrum. Meas.Instrum. Meas. 68(6), 2288–2294 (2019) 6. J.M. Tarongi, A. Camps, J.A. Pulido, K-band radiometer designed for academic purposes: Intercomparison of performances as total power, dicke or noise injection radiometers, in 2007 IEEE International Geoscience and Remote Sensing Symposium (2007), pp. 2927–2930

Chapter 4

High Sensitivity W-Band Radiometer

To improve the detection accuracy of target objects, millimeter-wave imaging system is utilized for security checking, airport runway obstacle monitoring, remote sensing, etc. The sensitivity of the RF frontend is the key parameter that ultimately limits the image quality and acquisition time. For imaging systems that require a large number of receivers to form an array, the cost of the W-band radiometers fabricated with III–V technologies is extremely high. Meanwhile, silicon technologies with f T above 200 GHz. Noise equivalent temperature difference (NETD) of the silicon-based radiometer in low contrast environments, such as the typically quoted threshold NETD for W-band radiometers, needs to be as low as 0.5 K to obtain high-quality images indoors [1, 2]. In this chapter, to address the issue of radiometer sensitivity reduction caused by high loss of passive single-pole double-throw switch (SPDT), a distributed amplifier SPDT is proposed in our work and applied to construct the W-band Dicke radiometer. Different from the conventional lossy switch, the proposed SPDT-DA structure features a high and flat gain response which significantly decreases the noise of the radiometer system. Based on the 0.13 μm SiGe BiCMOS process, the Dicke radiometer achieves a total noise figure of 8.4 dB, an RF gain of 42 dB and a sensitivity of 0.21 K within 30 ms. Moreover, to effectively demonstrate imaging capability of the proposed single-pole-double-throw distributed amplifier structure, a passive imaging system is established. Based on the radiometer chip, a two-dimensional imaging experiment with an object distance of 0.7 m is successfully carried out.

© The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2024 X. Bi, Silicon-Based High-Sensitivity Broadband Receiver, https://doi.org/10.1007/978-981-97-0881-9_4

41

42

4 High Sensitivity W-Band Radiometer

4.1 Comparison Between the Conventional Dicke Architecture and the Proposed Radiometer NETD, as a measure of the radiometer sensitivity, can be calculated with Eq. (4.1). Assuming that the pre-detection gain of the receiver is high enough to make the noise contributions of the power detector negligible, then NETD is determined by the receiver’s system noise temperature (Tsys ), RF bandwidth (BW ), integration time (τ ) and gain uncertainty (G/G S ) dominated by the flicker noise of the detector [3].  N E T D1 = Tsys

1 + BW τ



G GS

2 1/2 (4.1)

As shown in Fig. 4.1a, for the typical Dicke radiometer architecture [4, 5], the SPDT switch is inserted between the antenna and the low noise amplifier (LNA). The gain uncertainty in Eq. 4.1 can be eliminated by periodically switching between the antenna and a reference load at a rate higher than the gain fluctuation frequency, which is 1/f noise corner frequency. Due to that only the input noise in one half of the time is received by the Dicke radiometer, the NETD can be simplified as: 2Tsys N E T D2 = √ B Mτ

(4.2)

Obviously, the system noise figure Tsys is mainly contributed by the switching loss and the noise of the LNA circuit. Thus, the reduced loss of SPDT and the gain of the LNA is critical to the achievable sensitivity of the radiometer. Equation (4.2) is based on the assumption that the inputs of the Dicke radiometer are balanced. Since the minimum achievable noise figure of LNA is dominated by the fabrication process, the most direct way to increase sensitivity is to reduce the switching loss. Fig. 4.1 Architectures of a the conventional Dicke radiometer and b the proposed Dicke radiometer

RFin DCout

Ref

SPDT

LNA1-2

Detector

(a) RFin DCout

Ref

SPDT-DA

LNA1

(b)

Detector

4.2 Proposed SPDT-DA Design

43

Nowadays, there are two main types of passive millimeter-wave SPDT switches: one is based on quarter wavelength transmission lines; the other one is utilizing travelling-wave switching lines. Limited by the bandwidth of 1/4λ transmission line, the first type SPDT switch is intrinsically narrow band, and the parasitic capacitance introduced by the switch causes imperfect termination which further increases the switching loss. For example, the switch in 65 nm CMOS technology with a minimum insertion loss of 4 dB has been reported, another switching circuit with a small insertion loss of 1.3 dB also has been achieved utilizing the expensive GaAs technology. Meanwhile, for the switching circuit based on the travelling wave structure, transmission lines are used to extend operational bandwidth, leading to increasing insertion loss and circuit size. Besides that, some novel receiver architectures have also been proposed to address the switching loss issue in receivers. For instance, two separate LNAs before the passive switch [6, 7] or two separate switchable amplifying units [8] are employed to reduce the SPDT loss. However, due to process variation between the two branches, the uncorrelated noise and unavoidable imbalance are introduced and may increase the uncertainty of the calibration, making this method less attractive for radiometers. Although the imbalance and switching loss issues exist in [6, 7] can be eliminated by embedding the Dicke switch in a balanced LNA [9, 10], at least three gain blocks are required in this architecture which inevitably occupy larger chip area and increase power consumption. Therefore, it’s still challenging to achieve the silicon-based radiometer with high sensitivity, wide band, low power consumption and compact chip size. To this end, this chapter proposes a single-pole-double-throw distributed amplifier (SPDT-DA) to seamlessly embed the switching function into the low-noise amplifying block, achieving a low switching-loss and a low NETD of the Dicke radiometer, as demonstrated in Fig. 4.1b. By deploying this SPDT-DA as the first block in the receiver, the implemented W-band Dicke radiometer achieves an NF of 8.4 dB at 91 GHz and a switching loss of 0.93 dB at 91 GHz. The typical switching loss is from 0.31 to 1.63 dB over the frequency range of 85–105 GHz. In the following sections, the design methods and mechanisms of the proposed SPDT-DA, total radiometer, and the imaging system prototype are introduced in detail.

4.2 Proposed SPDT-DA Design 4.2.1 Operation Principle of the Proposed SPDT-DA In order to achieve the switching function without introducing additional loss and redundancy, the distributed transmission lines of the SPDT switch are reused as the input baseline network of the distributed amplifier, as shown in Fig. 4.2. Due to that the calibration and signal branches share the same active amplifying units, the balance can be maintained between them. Compared to the resonant LNA [11],

44

4 High Sensitivity W-Band Radiometer TL1

TL2

TL2

TL1

Port1 TL4

TL3 50 Ω

50 Ω

TL3 Ind2

Ind1

Ind1 Port3 TL4

TL3

TL3 TL4

TL2

TL2

TL1

TL1

50 Ω

S31 (dB)

Port2

f2

f1

Frequency (GHz)

Fig. 4.2 Schematic of the proposed single-pole-double-throw distributed amplifier (SPDT-DA)

the distributed amplifier can achieve a similar noise figure, therefore, the turned-on branch is designed as a distributed amplifier to obtain an acceptable noise figure in a wide bandwidth. As shown in Fig. 4.2, the proposed SPDT-DA consists of a two-stage distributed cascode amplifier connecting to two switchable transmission lines (TLs) by sharing the distributed-line. The minimum achievable noise figure (NFmin ) and gain of each stage can be optimized by selection of heterojunction bipolar transistors (HBTs), Q1 and Q2 . The inductors, together with the parasitic capacitances at collectors, introduce S 31 peakings at the frequency where the LC phase delay exactly matches with that caused by corresponding TL sections connected to the base. For instance, the phase matching between TL 1 and Ind 1 introduces an S 31 peaking at f 2 , as shown in Fig. 4.2. Therefore, these peaking frequencies to synthesize the desired passband characteristics in W-band can be controlled by optimizing the inductances of Ind 1 and Ind 2 . The upper switch array, consisting of S w1 to S w3 , is controlled by Vctr 1 while the bottom array is controlled by V ctr2 . Therefore, by applying two out-of-phase and equal-amplitude pulses to V ctr1 and V ctr2 , either the reference resistor or the antenna is connected periodically to the input of the SPDT-DA. Then, the modulation of the input signal is realized and 1/f noise is eliminated. LC type input and output matching networks are more suitable and employed for impedance matching, since the targeted bandwidth covers W-band with a fractional bandwidth of more than 20%. Specifically, the LC type matching networks are designed to achieve a minimum NF at Port 1 and Port 2, and maximum gain at Port 3 respectively.

4.2 Proposed SPDT-DA Design

45

4.2.2 Design Considerations of the SPDT-DA Switching TLs For the proposed SPDT-DA, the input transmission line network is the critical part. As shown in Fig. 4.3, by decomposing the proposed radiometer and the conventional radiometer into essential noisy and lossy blocks, the noise temperature of the proposed radiometer (TSY S_a ) and conventional radiometer (TSY S_b ) can be derived. The detailed derivation formulas are as follows. In Fig. 4.3a, the proposed radiometer is composed of the SPDT-DA, the low noise amplifier with high gain and the detector with high responsibility. By modulating the input signal with the SPDT-DA, the 1/f noise introduced by the detector and the noise from LNA can be eliminated. Furthermore, utilizing the high gain LNA stage to counteract the noise from the detector, the noise of the radiometer will be mainly donated by the noise figure of the first stage SPDT-DA. Compared with the traditional Dicke radiometer architecture shown in Fig. 4.3b, the noise temperature of our proposed radiometer (TSY S_a ) and the typical radiometer (TSY S_b ) can be calculated as follows, respectively. TSY S_a = (N Fmin L Min1 L Din − 1)T0 ((N FMin L mout L Min2 ) − 1)T0 L Din L Min1 + G CC TSY S_b = (N Fmin L Min2 L S P DT S − 1)T0 +

LNA

Switching Block LDout

TD E T L S P DT S L Min2 GLN A

(4.3) (4.4)

Detector

LMout1 LMin2

LMin1

LDin

NFmin Gcc

TDET NFmin,GLNA

(a)

LSPDTS

LMin2 TDET NFmin,GLNA (b)

Fig. 4.3 Detailed diagrams of a the proposed receiver and b conventional Dicke-radiometer (only the turned-on branch in the SPDT-DA is demonstrated)

46

4 High Sensitivity W-Band Radiometer

where NF min indicates the minimum noise figure of the cascode stage, L Min1 /L Min2 and L Mout1 indicate the loss of the input matching network and the output matching network, respectively. L Din and L Dout indicate the loss of the input distributed base line and output distributed collector line, respectively. L SPDTS indicates the loss of the lossy SPDT switch. GCC and GLNA indicate the gain of each cascode stage and the LNA. T 0 indicates the room temperature. T DET indicates the noise temperature of the detector. When GLNA is bigger than 35 dB, GCC is bigger than 8 dB, and the L Din , L Dout , L Min1 , L Mout1 and L Min2 are less than 1 dB, the third term of T SYS_b can be ignored, and (N Fmin L Min1 L Din − 1)T0 >>

(N Fmin L Mout L Min2 − 1)T0 L Din L Min1 G CC

(4.5)

Therefore, the Eq. (4.3) can be rewritten as TSY S_a ≈ (N Fmin L Min2 L Din − 1)T0

(4.6)

Meanwhile, the second term of TSY S_a also be ignored with the help of the huge gain GLNA . Thus, the Eq. (4.4) can be rewritten as TSY S_b ≈ (N Fmin L Min2 L S P DT S − 1)T0

(4.7)

Due to that L Min1 and L Min2 are roughly equivalent, the difference between TSY S_a and TSY S_b is mainly devoted by the insertion loss of the first switching stage, specifically, the L Din in Fig. 4.3a and L SPDT in Fig. 4.3b. As a result, the low loss switching network is the key point to obtain the low noise temperature of the radiometer system. Two main loss mechanisms in this network, including TL loss and insertion loss between ports have been analyzed and designed as follows. With ignoring the switched off branch, the equivalent circuit of the base-switching-TL network is shown in Fig. 4.4, which consists of four TL sections, including two TL 1 sections and two TL 2 sections. C off represents the equivalent capacitance of the switches at the offstate, C in represents the equivalent capacitance looking into the cascode stage. Since the passive loss of TL is proportional to the physical length and related to the equivalent capacitance of TLs to the ground, it can be reduced by optimizing the TL lengths and Z 0 . When designing TLs, a constant phase velocity should also be maintained by satisfying the following equation [11]:  Z0 =

Lb Cb +

Cin lb

(4.8)

where L b and C b are the per-unit-length inductance and capacitance, C in is the input capacitance of the cascode stage and lb is the length of the base-switching-TL section.

4.2 Proposed SPDT-DA Design

47

Fig. 4.4 Circuit model of base-switching-TL network at off-state

B2

B3

R

2

1

1 B1

TL2

TL1

50 Ω

A

Coff

R

3

TL2 Coff

4

TL1

B

Coff

B2

50 Ω

In addition, the signal amplitudes at Port 2 and Port 3 are also affected by their loads, resulting in different signal strengths at different port locations. In the previously distributed amplifier literature, in order to simplify the analysis, this load effect is usually ignored in the theoretical analysis. The conventional gain expression of the distributed amplifier based on a lumped circuit model considering capacitive load can be derived as: Gf =

n 2 gm2 Z b Z c 4

(4.9)

where gm is the trans-conductance of the gain stage, Z b and Z c are the impedances of the base and collector TL, respectively, and n is the number of the gain stages. In fact, resistive elements are always present at the input of the cascode stage. The power flowing into Port 2 and Port 3 is not equal, caused by different loading impedance between ports. Assuming that the power is delivered from Port 1 to Port 2, Port 2 undertakes the passive loss caused by one TL 1 , while Port 3 undertakes the passive loss caused by one TL 1 and two TL 2 . In order to comprehensively quantify the losses of Port2, Port3 and Port4, even-mode and odd-mode equivalent circuit analysis of the base-switch-TL network is performed, as shown in Fig. 4.5a, b. Since TL is a linear network, the actual transmission response can be obtained by summing the responses of the even-mode and odd-mode excitations. For a four-port base-switching-TL network with 50  loads, the S-parameter can be expressed as: ⎡

B1 ⎢ B2 S1 = ⎢ ⎣ B3 B4

B2 B2 B3 B3

B3 B3 B2 B2

⎤ B4 B3 ⎥ ⎥ B2 ⎦ B1

(4.10)

Since the characteristic impedance of Port 2 and Port 3 are not always uniformly 50 , the four-port-parameter based on varied impedance can be generalized as

−1 S2 = Rz (U + S1 )(U − S1 )−1 Rz + U

 × Rz (U + S1 )(U − S1 )−1 Rz − U

(4.11)

48

4 High Sensitivity W-Band Radiometer Te

Te

2

3 Cin

Cin 1/2

1

A

Coff

Γe

TL2

TL2

TL1

1/2 Coff

1/2 Coff

4 1/2

TL1

B

Coff

Γe

(a)

Te

To

2

3 Cin

Cin 1/2 Γo

1

Coff

TL2

TL2

TL1

A

1/2 Coff

4 1/2

TL1

B

Coff

Γo

(b)

Fig. 4.5 Even- and odd- mode analysis of the base-switching-TL network

  where Rz is a diagonal matrix that can be represented as, Rz = diag (50/Z i ) and Z i represents the loading impedance. The calculated and simulated S-parameters of the base-switching-TL network are compared in Fig. 4.6. It should be noted that, the C off and 1/2C off are loads for TL 1 and TL 2 in Fig. 4.5a, respectively, and they are not equal under either the even-mode or odd-mode excitation. Therefore, the widths of TL 1 and TL 2 are different according to different loading capacitors, for maintaining a constant Z 0 . It can be seen that the minimum |S 21 | and maximum |S 31 | at 94 GHz occur simultaneously when the electrical length approaches 45° in Fig. 4.6. Under the circumstances, most of the input power flowing from Port 1 is “driven” into Port 2 and therefore the input power undertakes less passive loss. Meanwhile,√the power dissipated at the load of Port 4 can be minimized. The total passive loss, (|S 21 |2 + |S 31 |2 ), can be minimized to be less than 0.71 dB if TL 2 is around 50 degrees. Therefore, to reduce the reflection loss at Port 1 (S 11 ) and thermal dissipation loss at Port 4 (S 41 ), the real part of load impedance can be optimized to around 33 , as shown in Fig. 4.7. Based on the above analysis, a cascode stage consisting of two HBTs both with an emitter size of 4 × 0.13 μm × 0.84 μm are utilized in the proposed SPDT-DA. The supply voltage for SPDT-DA is 1.5 V and the base voltage of Q1 is around 0.89 V, which leads to a 4.2 mA collector current of each cascode stage. Three inductor values are all around 290 nH. In order to control the peaking frequency, the inductor in the middle is optimized and slightly increased. The achieved Q-factor has a peak value of 12.3 at around 100 GHz. With the electromagnetic simulation software

4.2 Proposed SPDT-DA Design

49

Fig. 4.6 Calculated S-parameters (in MATLAB) and simulated S-parameters of the four-port base-switching-TL network at 94 GHz

Fig. 4.7 Calculated loss (MATLAB) at the four ports at 94 GHz when the load impedance of Port 2 and Port 3 varies

ANSYS HFSS, the optimum TL 2 electrical length is 40° and the corresponding total passive loss is around 0.83 dB. Due to the limited Q-factor of the on-chip inductor at W-band, smaller gain fluctuation is resulted which helps cover the −3 dB bandwidth of 20 GHz. The final dimensions of the main components used in the SPDT-DA are listed in Table 4.1. Table 4.1 Dimensions of the main components in the SPDT-DA

Q1 , Q2

4 × 0.12 μm × 0.84 μm

Sw1_3

8 × 0.13 μm × 0.84 μm

Ind 1_3

290 pH

50

4 High Sensitivity W-Band Radiometer

4.3 Proposed Receiver Design 4.3.1 LNA Design To address the noise from the detector and reduce NETD in the proposed total power radiometer, LNA is required to have a large gain and bandwidth. At the same time, a bandpass filter is required to prevent the radiometer from being saturated by the signal from the interference band. By a considerable number of silicon-based millimeterwave amplifiers have been reported with high gain [12–17] and low noise [18]. As shown in Fig. 4.8, a Q-enhanced cascode stage is employed in LNA design to obtain gain and selectivity improvement simultaneously [17]. In order to minimize noise figure, the L-type LC matching networks are used as the input matching network. For the inter-stage matching networks, simple shuntseries matching networks are employed. The HBTs (Q1 and Q2 ) in the four cascode stages are both constructed by two parallel 4 × 0.12 μm × 0.84 μm transistors. Each cascode stage can obtain the minimum and maximum stable gain (MSG) at a biasing current density of 1.19 and 1.79 mA/μm, respectively. In this receiver, SPDT-DA can provide around 8 dB gain and the receiver’s NF is dominated by the NF of SPDT-DA. Then, the gain requirement of LNA is alleviated. Therefore, compared with [17], the LNA has a lower bias current density at each stage, which improves the dynamic range and stability. An optimum base biasing voltage of 0.86 V is deployed, resulting in a current density of 0.89 mA/μm for each cascode stage. In addition, to reduce the parasitic coupling effects and increase LNA’s stability, customized passive shielding structures, cascode shielding structures and dc biasing tubes [19] are used in the design. The finalized component values and dimensions of LNA are listed in Fig. 4.8. Vcc

Vcc

TL6

TL6

TL1

Q2

TL2

C2

C2 TL3 Q2

TL2

Q1, Q2: 0.12 µm x 0.84 µmx2 TL1: 27.5° TL2: 10° TL3: 115° , 117° and 121° TL4: 54.5°

Fig. 4.8 LNA schematic

Q1

Q1

C2

C1

TL5

TL5

Q1

Q1

TL6

TL6

TL5

TL5

Vcc

Vcc

TL3 Q 2

TL2

C1: 601.3 fF C2, C3, C4: 12.4 fF C5: 17.1 fF

TL3 Q2

C3

4.3 Proposed Receiver Design

51 Vcc

R1 Vb

DCout C1

RFin

R2 DCref Rb2

Rb1

Q1 TL1

C2 RFin

Q1 TL2

Q1, Q2: 0.12 µm x 0.84 µmx2 R1, R2: 3 kΩ C1, C2: 16.7 fF TL1, TL2: 56°

Fig. 4.9 Detector schematic

4.3.2 Detector Design The detector design targets for a low-noise equivalent power (NEP), which can be interpreted as achieving low output noise voltage and high responsivity simultaneously. Since the switch formed by the SPDT-DA can cancel 1/f noise, the β of the HBT can be lower, corresponding to a low biasing current. In addition, the low bias current facilitates the use of a large load resistance, thereby improving the responsiveness of the detector. Therefore, the detector is biased in class B. As shown in Fig. 4.9, the detector consists of a common emitter pair, which is similar as the detector topology proposed in [9, 20]. The HBTs (Q1 and Q2 ) in the differential pair consist of 2 × 0.12 μm × 0.84 μm transistor. The L-type LC input matching networks are used to match the input impedance to 50  at 94 GHz. Q1 senses the amplified W-band signal at the input, while providing a reference dc voltage at the output for subtraction. The DC voltage at the output contributed by the second order component in Taylor series can be extracted by subtracting the output voltage. The load resistance is chosen to be around 3 k. When the base bias voltage is around 0.8 mV and collector voltage is 1.5 V, the collector biasing current of 142 μA can be obtained. The finalized component values and dimensions of the detector layout are also listed in Fig. 4.9.

4.3.3 Receiver Design Based on the proposed active SPDT-DA structure, combined with high gain LNA and high responsibility differential detector, the high sensitivity receiver has been designed further. Simulations are performed for both the proposed SPDT-DA followed by a common LNA and the conventional passive SPDT switches cascaded with the same LNA, respectively, for a fair comparison with the prior arts of the proposed W-band radiometers. The conventional circuit topology utilizing 1/4 λ TLs

52

4 High Sensitivity W-Band Radiometer

and its design values are shown in Fig. 4.10a. The passive switch used for comparison mainly consists of a 1/4 λ transmission line and a pair of HBT switches. Verified by ANSYS HFSS, the simulated minimum loss of the passive SPDT at W-band is around 2 dB. Figure 4.10b compares the simulated NF and gain of the conventional radiometer and the proposed radiometer. Compared to the radiometer with typical passive SPDT, the proposed receiver has significant improvement in NF. The simulation results show that the NF difference between the two receivers is about 1.9 dB at 80 GHz and about 1.1 dB at 94 GHz. It is worth noting that the advantage on NF of the proposed receiver decreases as the frequency rises to 100 GHz and above, which is caused by Fig. 4.10 a Schematic of the conventional receiver using a passive SPDT. b Simulated performance comparison between the receivers using a passive SPDT and this receiver

DCout

SPDT

LNA

Vtr1 TL3 S1

TL1

Port1

Port3 Port2 TL2

S2 TL4 Vtr2

S1, S2: 0.12 µm x 0.84 µmx8 TL1, TL2: 90° @ 94 GHz TL3, TL4: 70° @ 94 GHz

(a)

(b)

4.4 Experimental Results of the Radiometer

53

the operational bandwidth of LC matching network in the SPDT-DA. The matching network also can be tailed for the other bands with similar bandwidth.

4.4 Experimental Results of the Radiometer 4.4.1 SPDT-DA Based on the 0.13 μm SiGe BiCMOS technology with a f T of 240 GHz and a f max of 290 GHz, the proposed SPDT-DA is implemented. The signal line adopts the top metal layer TM2 with a thickness of 3 μm, and the ground plane adopts the bottom metal layer M1 with a thickness of 0.4 μm. The measured transmission line loss is 1.2 dB/mm at 94 GHz. All capacitors used are MIM capacitors with a capacitance density of 1.5 f F/μm2 . Resistors with a sheet resistance of 250 /square and an accuracy of 10% are used for the loading resistors. The chip core area of SPDT-DA is 1.2 × 1.16 mm and the micrograph of the SPDT-DA is demonstrated in Fig. 4.11. The Vector Network Analyzer Anritsu ME7808B is used for measurements below 110 GHz, including an Anritsu 65 GHz VNA and an OML mmWave module. The input power at the power tips of the VNA system is around −31 dBm, which is verified by VDI Erickson Instruments PM4 power meter. Figure 4.12a indicates the simulated and measured -parameters of the proposed SPDT-DA, including the gain and isolation from the input port to the output port for the two different states. When the RF input branch is turned on, the input signal is amplified by 8.3 dB @ 90 GHz, while the reference signal is suppressed by more than 10 dB over the entire frequency band. Conversely, when the switch control signal is in the opposite phase, the reference input is amplified, while the input signal is suppressed. The above off-state forward isolation is typically higher than that of the switched LNA presented in [9], probably because the proposed topology does not rely on two balanced amplification paths to cancel each other out in the off-state. Figure 4.12b plots the input and output Fig. 4.11 Microphotograph of the proposed SPDT-DA

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Fig. 4.12 a On-state and off-state S31, b Reflection coefficients, and c Noise figure of the standalone SPDT-DA

reflection coefficients with a good matching state from 80 to 100 GHz. With the help of the employed cascode stage and passive shielding structure in the proposed SPDT-DA, the measured reverse isolation is increased to 27 dB. On the other hand, in relation to the noise figure, a Y-factor technique [21] is used for noise characterization. Utilizing a 67 GHz spectrum analyzer (R&S FSU) through an external converter, the measuring frequency range is extended to 110 GHz. A commercial isolator and preamplifier are connected between the chip and the converter to protect the noise source and minimize the measurement uncertainty. As shown in Fig. 4.12c, the measured NF of SPDT-DA with a minimum 7.6 dB at around 88 GHz, and 7.6–8.9 dB from 80 to 100 GHz can be observed. In addition, the linearity performance of SPDT-DA is measured by the millimeter-wave signal generator (R&S SMF 100A), multiplier (RPG AFM6-110), W-band attenuators (RPG WTA-110) and millimeter-wave power meter (Erickson Instrument PM4). The measured IP1dB is −6.4 dBm which is enough for the radiometer application.

4.4 Experimental Results of the Radiometer

55

4.4.2 LNA The LNA chip micrograph is demonstrated in Fig. 4.13. In this LNA, a biasing current of 3 mA is supplied for each stage. The Y-factor technique (ENR of the noise source is around 12 dB) is used for the noise characterization and gain measurement. As shown in Fig. 4.14, the implemented amplifier has obtained an in-band gain of 35–40 dB, NF of no more than 7.5 dB from 79 to 97 GHz and power consumption of ~14.3 mW. With the similar linearity measurement setup as the SPDT-DA, the measured OP1dB of this LNA is around −0.5 dBm and the corresponding IP1dB is around − 36.7 dBm at 83 GHz. The core chip area is around 0.68 × 1.12 mm2 .

Fig. 4.13 LNA microphotograph

Fig. 4.14 a Parameters of the standard alone LNA and b Noise figure of the standard alone LNA

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4 High Sensitivity W-Band Radiometer

Fig. 4.15 Responsivity versus current density of the detector

4.4.3 Detector The area of the detector is 0.58 × 0.83 mm2 . Figure 4.15 indicates the measured responsivity versus current density of the detector operating at 90 GHz. The detector achieves a response of over 20 kV/W at a biasing current density around 200 μA/ μm. At the biasing current density, the peak responsivity of the √standalone detector is around 27.2 kV/W and the minimum NEP is around 2.5 pW/ Hz at 91 GHz. NEP is calculated based on the measured output noise without any input and measured responsivity. Figure 4.16 indicates that the standalone detector has a high-Q input matching. As can be seen, the measured input reflection coefficient is below −15 dB from 89.2 to 94.1 GHz. The input matching bandwidth is limited by the high-Q of the L–C input matching. There is usually a trade-off between peak responsivity and bandwidth. As shown in [22], in the differential detector design, the bandwidth can be improved by deploying an on-chip balun in the input matching network.

4.4.4 Receiver Performance The chip micrograph of the whole receiver with SPDT-DA, LNA and the differential detector is illustrated in Fig. 4.17. Individual blocks were characterized by using a focused ion beam (FIB) to sever the connections between adjacent blocks. In order to isolate SPDT-DA, LNA and the detector, a 46 μm wide metal-stacks (M1 –M5 and TM1 –TM2 ) are deployed, from top metal down to P+ in the substrate. The layout of the matching network has been adjusted to compensate around 35 f F capacitance, which is introduced by the testing GSG pads. The whole chip dimension is 1.2 × 2.85 mm2 . With the aforementioned setup based on Y-factor method, the NF and gain of the proposed receiver are measured. As shown in Fig. 4.18, the performances of

4.4 Experimental Results of the Radiometer

57

Fig. 4.16 Measured and simulated reflection coefficient of the standalone detector

Fig. 4.17 Microphotograph of the W-band receiver

SPDT-DA and LNA finally result in a typical system NF from 7.1 to 9.5 dB in band. The NF of the receiver is around 8.4 dB @ 91 GHz. The experimental results show that the gain in W-band is greater than 40 dB, and the simulated gain differs from the actual gain by several dB in the vicinity of the center frequency of 80–100 GHz. One of the gain peaks is about 5 GHz (5%) higher than the expected frequency. This is mainly due to the band broadening of SPDT-DA and LNA. Considering that the SPDT-DA and LNA combined bias pad (PGPPGP) connection is much longer than the standalone block shown in Fig. 4.17, the resistivity of passive connections to V CC was intentionally set larger in HFSS simulation to maintain a larger than desired gain and BW. Since the voltage potential at the collector is higher than the simulated value, the base–collector junction capacitance of the common-base HBT is lower than simulated, leading to gain peak shifting to higher frequency. Output noise measurements were performed by the Agilent 35670A Dynamic Signal Analyzer. For high resolution, 1600 resolution lines were selected when measuring. The average number is set to 20. The supported highest frequency is

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4 High Sensitivity W-Band Radiometer

Fig. 4.18 Noise figure and gain of the SPDT-DA + LNA

51.3 kHz. The required power supply voltage and the base bias voltages in each block are provided by 1.5 V dry cell batteries and low dropout regulators (LDOs). Therefore, the 50 Hz spur, which is usually introduced by the DC voltage source, is eliminated, and the √ average output noise voltage of the receiver is measured to be around 0.45 μV/ Hz for above 100 kHz, as shown in Fig. 4.19. In order to characterize the responsivity of the receiver, as shown in Fig. 4.20, a signal generator, a multiplier, a 40 dB tunable W-band attenuator, a −19 dB coupler, a millimeter-wave power meter and a spectrum analyzer are used. With this setup, the input signal can be adjusted to −70 dBm. The average responsivity is calculated according to the following equation [23, 24]: Fig. 4.19 Output noise of the receiver

4.4 Experimental Results of the Radiometer Signal Generator: R&S SMF100A

Multiplier: Attenuator: RPG RPG AFM6-110 WTA-110

59 Attenuator: RPG WTA-110

-19 dB Coupler

Spectrum Analyzer: R&S FSU

X6 Receiver Chip Power Meter: Erickson Instrument PM4

Fig. 4.20 Testing setup of the responsivity measurement

∞  A = 0 ∞ 0

2 ( f )d f ( f )d f

(4.12)

The NETD after considering a factor of 2 penalty caused by the switching has been accounted. The equivalent switching loss of each receiver is defined as: L SW = N FR X − N FL N A

(4.13)

Therefore, for a conventional Dicke radiometer using a passive SPDT, the L SW is the insertion loss of the passive SPDT. For this radiometer, NF RX and NF LNA are 8.4 and 7.47 dB at 91 GHz, respectively, with a −3 dB bandwidth of 92 to 102 GHz. According to (4.13), the peak switching loss is 2.53 and 2.12 dB at 87 and 99 GHz, respectively, 0.93 dB at 91 GHz, and 0.31–1.63 dB at other frequencies from 85 to 105 GHz. The measured average responsivity is around 166.1 MV/W, as shown in Fig. 4.21. The simulation results are in good agreement with the measured values, with a peaking frequency approximately 5 GHz (5.3%) lower than that in measurements. Due to the wide bandwidth (>20 GHz) of SPDT-DA and LNA, the frequency response shape of the responsivity and NEP are influenced by S 11 of the detector. The unmodeled capacitive parasitic introduced by the input matching network and testing pad of the detector shifts the valley frequency of the detector’s S 11 as well as the peaking frequency of the responsivity and NEP. At the same time, there is a few dB difference between the simulated and measured peaking responsivities and NEP, which could be caused by the unmodeled resistive effects of the passive √connection at detector’s base. The average√NEP across the whole band is 2.7 fW/ Hz and the lowest NEP is around 1.9 fW/ Hz at 90 GHz. Assuming that the Dicke radiometer is balanced, NETD based on the average NEP and 30 ms integration time is 0.21 K. Moreover, the receiver has an RF gain of about 42 dB and a power consumption of 28.5 mW, which is comparable to W-band imagers in GaAs or InP.

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Fig. 4.21 a Responsivity and b NEP versus frequency with input power of 61.5 dBm. c Responsivity and NEP versus input power @ 90 GHz

4.4.5 Imaging Testing Experiment To verify the improvement effect of the proposed radiometer on the imaging system, the measurement system is displayed in Fig. 4.22a, based on a WR-10 horn antenna and a two-dimensional mechanical scanning system. The object fixed on the scanning system is made up by two characters with aluminum which occupies a size of 40 × 75 mm2 . The object is placed in between the transmitting and receiving antenna with a distance of 70 cm. The moving step of the mechanical scanning system is set to 0.25 mm in the X and Y axis, which corresponds to a scanning resolution of 300 × 160 pixels. According to the power level of the W-band source, the output voltage of the detector varied from 0.019 to 1.2 V. Utilizing this imaging system with the high sensitivity radiometer, two pictures with high-definition images can be obtained easily as shown in Fig. 4.22b.

4.5 Conclusion

61 2D mechanical scanning system

W-band Dielectric Multiplier: Horn Lens 1 RPG AFM6-110 Antenna X6 15.7 GHz

Dielectric Dielectric W-band Lens 3 Lens 2 Horn Antenna

Multi-meter

Receiver

Signal Generator: R&S SMF100A

Motors for X and Y Axis

Program controlling the steps of the motor

(a)

(b) Fig. 4.22 a Setup of the imaging experiment. b Photograph of the imaging test and the obtained images (on the right-hand side)

4.5 Conclusion This chapter proposes a single-pole-double-throw distributed amplifier (SPDT-DA) to improve the NF of the Dicke radiometer. The design methods and technologies of SPDT-DA are expounded in detail. Based on 0.13 μm BiCMOS technology, the proposed radiometer has realized an equivalent switching loss of around 0.93 dB and NF of 8.4 dB at 91 GHz. Moreover, the NETD of the proposed radiometer is only 0.21 K with an integration time of 30 ms. The total power consumption of the radiometer is 28.5 mW. This concept can be widely adopted in systems that require integrated front-end switches.

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References 1. C.D. Dietlein, A. Luukanen, F. Meyer, Z. Popovic, E.N. Grossman, Phenomenology of passive broadband terahertz images, in Proceedings of 4th ESA Workshop on Millimetre-Wave Technology and Applications Workshop (Espoo, Finland, 2006) 2. J.J. Lynch, H.P. Moyer, J.H. Schaffner, Y. Royter, M. Sokolich, B. Hughes, Y.J. Yoon, J.N. Schulman, Passive millimeter-wave imaging module with pre-amplified zero-bias detection. IEEE Trans. Microw. Theory Tech.Microw. Theory Tech. 56(7), 1592–1600 (2008) 3. M. Tiuri, Radio astronomy receivers. IEEE Trans. Antennas Propag. AP-12(7), 930–938 (1964) 4. N. Skou, Microwave Radiometer Systems: Design and Analysis (Artech House, Nor-wood, MA, USA, 1989), pp. 35–37 5. R.H. Dicke, The measurement of thermal radiation at microwave fre- quencies. Rev. Sci. Instrum.Instrum. 17, 268–275 (1946) 6. A. Dyskin, S. Wagner, D. Ritter, I. Kallfass, An active 60–90 GHz single pole double throw switch MMIC. J. Infrared Milli. Terahz. Waves 35, 412–417 (2014) 7. I. Gresham, A 24 GHz SiGe DPST switch with 30 dB gain control for multi-channel receiver integrated circuits, in IET Seminar RF Microwave IC Design (2008), pp. 1–7 8. D. Zito, A. Fonte, Dual-input pseudo-switch RF low noise ampli- fier. IEEE Trans. Circuits Syst. II 57(9), 661–665 (2010) 9. L. Gilreath, V. Jain, P. Heydari, Design and analysis of a W-band SiGe direct-detection-based passive imaging receiver. IEEE J. Solid- State Circuits 46(10), 2240–2252 (2011) 10. D.C.W. Lo, H. Wang, B.R. Allen, G.S. Dow, K.W. Chang, M. Biedenbender, R. Lai, S. Chen, D. Yang, Novel monolithic multifunctional balanced switching low-noise amplifiers. IEEE Trans. Microw. Theory Tech.Microw. Theory Tech. 42(12), 2629–2634 (1994) 11. C.S. Aitchison, The intrinsic noise figure of the MESFET distributed amplifier,” in IEEE MTT-S International Microwave Symposium Digest (1985), pp. 460–466 12. A.Y.-K. Chen, Y. Baeyens, Y.-K. Chen, J. Lin, An 83 GHz high- gain SiGe BiCMOS power amplifier using transmission-line current combining technique. IEEE Trans. Microw. Theory Tech.Microw. Theory Tech. 61(4), 1557–1569 (2013) 13. D.L. Lu et al., A 75.5-to-120.5-GHz, high-gain CMOS low-noise amplifier, in IEEE MTT-S International Microwave Symposium Digest (2012), pp. 1–3 14. T. Mury, M. Tiebout, N.B. Buchanan, V.F. Fusco, F. Dielacher, A 76–84 GHz SiGe power amplifier array employing low-loss four-way differential combining transformer. IEEE Trans. Microw. Theory Tech.Microw. Theory Tech. 61(2), 931–938 (2013) 15. A.Y.-K. Chen, Y. Baeyens, Y.-K. Chen, J. Lin, A low-power linear SiGe BiCMOS low-noise amplifier for millimeter-wave active imaging. IEEE Microw. Compon. Lett. 20(2), 103–105 (2010) 16. X.J. Bi, Y.X. Guo, M.K. Je, Analysis and design of gain enhanced cascode stage utilizing a novel passive compensation network. IEEE Trans. Microw. Theory Tech.Microw. Theory Tech. 61(8), 2892–2900 (2013) 17. X.J. Bi, Y.X. Guo, Y.Z. Xiong, M.A. Arasu, M.K. Je, A 19.2 mW, dB gain and high-selectivity 94 GHz LNA in 0.13 m SiGe BiCMOS. IEEE Microw. Wirel. Compon. Lett. 23(5), 261–263 (2013) 18. Y. Yang, S. Cacina, G.M. Rebeiz, A SiGe BiCMOS W-band LNA with 5.1 dB NF at 90 GHz, in Proceedings IEEE Symposium on Compound Semiconductor Integrated Circuit (2013), pp. 1–4 19. X.J. Bi, Y.X. Guo, Y.Z. Xiong, M.A. Arasu, M.S. Zhang, M.K. Je, Passives design for a high performance W-band amplifier, in IEEE MTT-S International Microwave Symposium Digest (2013), pp. 1–3 20. F. Alimenti, S. Leone, G. Tasselli, V. Palazzari, L. Roselli, D. Zito, IF amplifier section in 90 nm CMOS technology for SoC microwave radiometers. IEEE Microw. Wirel. Compon. Lett. 19(11), 770–773 (2009) 21. Noise figure measurement accuracy—The-factor method, Agilent Technologies, Application Note 57–2

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22. E. Dacquay, A. Tomkins, K.H.K. Yau, E. Laskin, P. Chevalier, A. Chantre, B. Sautreuil, S.P. Voinigescu, D-band total power ra- diometer performance optimization in an SiGe HBT technology. IEEE Trans. Microw. Theory Tech.Microw. Theory Tech. 60(3), 813–826 (2012) 23. H.P. Moyer, R.L. Bowen, J.N. Schulman, D.H. Chow, S. Thomas, J.J. Lynch, K.S. Holabird, Sb-heterostructure low noise W-band de-tector diode sensitivity measurements, in IEEE MTT-S International Microwave Symposium Digest (2006), pp. 826–829 24. T.P. Garcia, S.P. Voinigescu, A passive W-band imaging receiver in 65 nm bulk CMOS. IEEE J. Solid-State Circuits 45(10), 1981–1991 (2010)

Chapter 5

Interstage Reflectionless Radiometer

Exploiting the latest generation of standard silicon processes, several millimeterwave radiometers have already shown the feasibility of System-on-Chip (SoC). As a highly sensitive receiver capable of measuring very low levels of microwave radiation, the high-performance millimeter-wave radiometer demands more and more design techniques to improve the temperature resolution, stability, linearity, etc. The heterodyne receiving structure has been widely used in the radiometer design to relieve the design requirements for detectors. The input signal is as weak as −90 dBm and the radiometer usually appears in arrays with large number of units, therefore a considerably high gain with low power consumption is always required. Besides, a high stability and linearity of the receiver are also significant to the success of the radiometer, since a small amount of reflection or unwanted frequency components may violate the stability and sensitivity of the radiometer system. In this chapter, state-of-the-art radiometer architectures with interstage reflection absorptivity are briefly introduced and analyzed at first. Secondly, a novel V-band heterodyne radiometer in SiGe BiCMOS employing a capacitor-reused absorptive inter-stage matching network is firstly proposed to address the existing issues. The proposed radiometer eliminates the out-of-band reflected signals around the interfaces of LNA and mixer, and achieves a gain of 31.9 dB@56 GHz, an enhanced stability in stopband than conventional LNAs.

© The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2024 X. Bi, Silicon-Based High-Sensitivity Broadband Receiver, https://doi.org/10.1007/978-981-97-0881-9_5

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5.1 Conventional Radiometer Architectures with Interstage Absorptivity 5.1.1 Radiometer Based on Circulator The input signal of the radiometer is as weak as –90 dBm, and practically it is usually composed of arrays of many cells, requiring low power consumption and high gain. In addition, the high stability and linearity of the receiver are significant to the success of radiometers, because small reflections or undesirable frequency components can damage the stability and sensitivity of the radiometer system [1]. However, without decreasing the in-band gain or increasing the system complexity, it is still challenging to minimize inter-stage reflection signals in both passband and stopband at millimeter-wave frequency range. In order to reduce the reflected signals between stages of the radiometer system and improve linearity of the system, conventional radiometers or high-sensitivity receivers usually employ an additional circulator between stages, as shown in Fig. 5.1a. By controlling the LO phase of each receiving channel, the conventional heterodyne radiometer can facilitate synthetic imaging [2]. However, between the multi-stage LNA or LNAs cascaded with a mixer, circulators usually are required to enhance isolation between adjacent stages and therefore improve radiometer sensitivity. Limited by the integration capability of silicon process, the circulator based on ferrite [3] can only be used for off-chip cascade, resulting in this type of receiver can only be realized through hybrid integration with very large size and high-power consumption. Furthermore, based on an ideal pre-amplifier circuit, the absorption architecture based on circulator in Fig. 5.1a is simulated and analyzed, as shown in Fig. 5.1b. The parameters of the ideal amplifier are assumed to be an operating bandwidth of 50–60 GHz, a gain of S 21 = 30 dB, an input/output in-band matching of S 11 = S 22 = −15 dB, unmatched stopbands, and a reverse isolation of S 12 = −30 dB. According to the on-chip circulator performance achieved in reference [4], the insertion loss of the circulator can be defined as 7 dB, the operating bandwidth is 30 GHz, and the isolation between ports is 18 dB. As can be seen from Fig. 5.1b, limited by the operating bandwidth of circulator, the absorptive frequency range on output reflection signals only covers 44–65 GHz, with a minimum reflection absorptive level of around 7 dB.

5.1.2 Radiometer Based on Attenuator By introducing additional attenuators, the interstage reflected signals of the radiometer system can also be absorbed, as shown in Fig. 5.2a. For a fully integrated silicon radiometer consisting of a low noise amplifier and a mixer, the high out-of-band reflection coefficient and parasitic reverse coupling may lead to a greater

5.1 Conventional Radiometer Architectures with Interstage Absorptivity

67

Fig. 5.1 The typical absorptive radiometer with circulator. a architecture; b simulation results based on the typical amplifier parameters

(a) S-Parameters (dB)

40 23 dB

20

0 dB

0 -20 -40 -60 -80

S22