Signaling in high-performance memory systems

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Signaling in High-Performance Memory Systems John Poulton Research Professor Department of Computer Science University of North Carolina at Chapel Hill [email protected] http://www.cs.unc.edu/~jp

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What's a Signaling System? 1 Transmitter

2 Transmission Line

3 Receiver +

Data In

− 4 Terminator

Data Out

VRef

5 Clock(s)

1. Transmitter encodes data as voltage/current signal levels onto the line 2. Transmission line delivers the signal to the receiver 3. Receiver compares signal with reference to recover data 4. Terminator removes signals from line, once they're received 5. Clocks tell transmitter when to drive a new signal, receiver when to sample For this discussion, we'll assume • binary (two-level) signaling • direct encoding ( e.g. VHi = "1", V Lo = "0") ISSCC 1999 Signaling in High-Performance Memory Systems

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A Bus has • multiple transmitters and receivers ("drops"), often paired up as transceivers • transceivers share a common transmission line • any one drop may send; any (or all) drops receive at a given time

This is not a bus:

This isn't a bus, either:

...it's a linear network.

...it's a bidirectional link. ISSCC 1999 Signaling in High-Performance Memory Systems

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Outline 5

28

29

35 36

55 56

• Transmission Lines ° Impedance, propagation, reflection ° Impedance discontinuities ° Termination methods ° Loss mechanisms ° Cross-talk, modes, impedance matrices ° Typical properties of PC-board T-lines

• Signaling Methods ° Voltage-mode vs Current-mode ° Single-ended vs Differential ° References ° Clocking

• Bus-based Signaling Systems ° Problems: lowered Z 0, v, f c ° CMOS , voltage-mode signaling ° SSTL & SLDRAM ° Open-drain systems, BTL, GTL, etc ° RAMBUS ° Summary

• Alternative Signaling Approaches ° Future Prospects ° Serial Links

57

• Wrapup, Questions, Discussion ISSCC 1999 Signaling in High-Performance Memory Systems

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Transmission Lines If I,V change at the drive point, B,E change as well; disturbance propagates away from the drive point at the speed of light E

v= Dielectric ( ε,µ ) I

1 c ≈ εµ εr

x

B

IF the dielectric is homogeneous and µ = µ (as usual) 0

V

dx

(Quasi-TEM guided wave propagation) I(x)/2

C=

− − −−

Q V

L=Φ I

...leads to circuit model

Ldx Rdx

I(x) + + + + I(x+dx)

dV R= I

+ + ++

V(x)

Cdx

Gdx

V(x+dx) − − −−

I(x)/2

G=

dI V

dx

L,R,C,G are per unit length

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Characteristic Impedance Semi-infinite line

Impedance is some value Z 0 Still semi-infinite

Peel off dx worth of line; impedance still Z 0 dx Ldx Rdx

So, original line looks like this:

Z0

Z 0 = Rdx + jωLdx +

Cdx

Gdx

Z0

1 jωCdx + Gdx + 1/Z0

(Z 0–Rdx–jωLdx)( jωCdx + Gdx + 1/ Z0) = 1 Z0(G+ jωC)dx – (R+ jωL)dx(G+ jωC)dx –

Z0 =

R+jωL = G+jωC

(R+jωL)dx =0 Z0

L when R=G=0 C

Impedance of a (near-) lossless line is frequency independent and real--just a resistance!

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Propagation Constant Ldx Rdx

I(x)

–(R+jωL)V(x) ∂V(x) = –(R+jωL)I(x) = ∂x Z0

Z0

V(x)

Gdx V(x+dx)

Cdx

∂V(x) 1/2 = –[(G+jωC)(R+jωL)] V(x) ∂x V(x) = V(0) e–Ax ; A = [(G+jωC)(R+jωL)]

1/2

A = (jωRC + jωGL –ω2 LC) 1/2 = jω LC ( 1–jRC+GL ) Taylor exp. ωLC = jω LC ( 1–jRC+GL ) 2ωLC 1/2

V(x)

GZ 0 = jω LC + R + 2 2Z0 Phase change along line, where wave velocity is

v=

V(0)

1 = 1 LC εµ

e –(α R + α D )x 2πv ω

(x) = V(0) e– jω LCx e –(α R + α D)x GZ 0 Attenuation αR = R αD = 2Z 0 2 conductor loss

Re

V(x) V(0) distance x along line

dielectric loss

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Characteristic Impedance of Some Typical Lines h

w

a s c

R DC =

2ρ wh

C = εw s µ s Z0 = ε w

r

s

s

w

h

s

b

ρ ρ + πa2 π(c 2 –b 2) C = 2πε log(b/a) µ log(b/a) Z0 = ε 2π R DC =

r

2ρ πr2 C = πε log(s/r) µ log(s/r) Z0 = ε π R DC =

ρ πr2 C = 2πε log(2s/r) R DC =

Z0 =

µ log(2s/r) ε 2π

R DC =

ρ wh

2πε C ≈ εw s + log(4s/h) Z0 =

εµ C

Good only for homogeneous dielectrics, where L = εµ/C Caveats: 1. Approximations only, useful for 1rst-order estimates; do NOT use these formulas for anything serious! Fire up your field solver... 2. Good only for homogeneous dielectrics; PC-board microstrips are not homogeneous. 3. Since µ = µ0 for many dielectrics,

µ ε =

µ0 377Ω 1 ε0 • ε r = ε r

Main thing to note: Z 0 is always a function of ratios of dimensions! ISSCC 1999 Signaling in High-Performance Memory Systems

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RC vs LC Behavior Ldx Rdx Cdx

In many situations, it's reasonable to set G=0 (dielectrics are pretty good for many signaling systems) What about R? If jωL > R, then mainly an LC line Every line has some frequency f0 = R 2πL below which it's RC and above which it's LC

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RC & LC Approximations

A typical on-chip wire: 0.5µ 0.5µ SiO 2, εr = 3.9

2µ

L = 0.6 nH/mm C = 73 fF/mm Rdc = 120 Ω/mm f0 = 32 GHz

The RC model works for CMOS! (...well, mostly...)

A typical PC-board wire: 5 mil

6 mil 0.7mil

"FR-4", εr = 4.5

L = 0.5 nH/mm C = 104 fF/mm Rdc = 0.008 Ω/mm f0 = 2.5 MHz

6 mil

The LC model is OK for PC board traces (mostly... R = R(f) from skin effect)

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Signal Returns L

R

Impedance of line AND it's return!

C

G

In striplines, return current in planes is concentrated near signal conductor, so return impedance is significant

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Skin Effect Skin depth:

E δ=

ρ πfµ

depth into conductor where E falls to 1/e of surface value

Skin-depth frequency: fs =

4ρ 4ρ = 2 πµs π µ 0 s2 non-magnetic conductors

s

approximation!

frequency at which skindepth equals half the conductor thickness

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Conductor Loss R(f) = R DC

f fs

α C(f) =

f fs

DC

2Z 0

P(1m) ) P(0) V(1m) = 20 log10 ( ) V(0) = 8.686 • α C

Attn (dB/m) = 10 log 10 (

(x) = e–αCL V(0)

RDC must include returns

Some typical signal conductors Conductor

fs

Attn (@ 1GHz)

RG-223/U

22.5 KHz

0.5 dB/m

5-mil, 0.5 Oz PCB trace

56 MHz

6 dB/m

µ x 0.5µ µ 0.5µ on-chip wire 113 GHz

V(6") = 0.90 @ 1GHz V(0) possibly a concern for busbased memory signaling

same as DC!

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Dielectric Loss 1010 Hz (µwave)

1013 Hz (infrared)

1015 Hz (light)

εr dipolar

ionic

electronic

G(ω)

...so, can define tanδ = G ωC GZ 0 2π f C tanδ L/C αD = = 2 2 = π f tanδ LC

freq

αD =

"RF" range:

π εr f tanδδ c

G ≈ k•ω

only for homogeneous dielectrics!

Some typical PC Board Dielectrics εr

Dielectric "FR-4"

$

GETEK $.$ Teflon

αD @1GHz tanδδ 8.86α

4.7 0.035

7.0 dB/m

3.9 0.010

1.8 dB/m

$$$ 2.25 0.001

V(6") = 0.89 @ 1GHz V(0) so, possibly a concern for memory interconnect

0.14 dB/m

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Terminators Real lines (PCB traces, cables) are fairly low-loss • Good news: signals get from here to there with little attenuation • Bad news: if the signals aren't removed, they'll bounce around and interfere with later signals (a.k.a, inter-symbol interference ). The terminator gets rid of the signal once it's received To see what happens at terminations (and other discontinuities), useful to introduce a mythical voltage generator that represents the traveling wave. Terminator

IF = V/Z 0 V IF 2•V

Z0

V

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General Terminations Mythical voltage divides across line & terminator producing terminator current...

VT = 2V•

ZT Z 0+Z T

IF = V/Z 0

V I T = T = 2V Z T Z0+Z T

ZT

V Z0 2•V

Σ currents into output node = 0

Reflection Coefficient (Γ)

I R = I F – I T = V – IT Z0 V(Z T– Z 0 ) = Z 0(Z T + Z 0) I V Z – Z0 Γ = R= R = T I F VF Z T + Z 0

This situation is highly desirable; forward wave is completely absorbed at the receiver.

IF IR

IT ZT

Γ is +

ZT > Z 0

Γ is −

ZT < Z 0

Γ is 0!

ZT = Z 0

VT

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Source Termination Can just as well put the terminator at the transmitter (source, or series termination) Termination Z0

V/2 0

Forward Wave

V/2

Γ = +1

tf

V V/2

Reverse Wave

V/2

V

V

V/2

V/2

V/2

0

0

0 t=0

t=tf t=2t f

t=tf t=2t f

t=0 reflected "1"

VOUT

tf

V

Open circuit

V

t=0

V

t=t f t=2t f reflected "0"

VIN

VOUT t=0

Tx Current 0

What happens when pulse width < t f ??

t=0

t=t f

t=t f

Current only flows when tx opposes reflected wave, so low power!

t=2t f

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Imperfect Terminations: Capacitance Suppose your 'perfect' terminator is shunted by a capacitor (as it usually is!!): ZS Z0

VIN

C

VOUT

Z0 2•V

Z0

Bandwidth is limited by cutoff frequency of Thévenin network, f0 = 1 πZ0 C If the forward wave is a fast-rise edge, the capacitor generates a large reflection: V IN

V OUT t=t f

V

Thévenin Equiv. C

If the transmitter isn't perfectly matched (Z S = Z 0), the reflection bounces off the transmitter and heads toward the receiver again.

t = Z0C/2

t=0

Z0 /2

C

t=2•t f

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Managing Reflections with Capacitive Termination What happens as you increase the risetime?? V (volts)

1 0.8 0.6 0.4 0.2 0

V (volts)

VIN

1 0.8 0.6 0.4 0.2 0

Z0=50Ω, t f=2.5nsec VOut 0

1

V (volts)

2

3

4 5 6 Time (nsec)

7

8

9

1

2

3

4 5 6 Time (nsec)

1 0.8 0.6 0.4 0.2 0

7

8

9

10

trise = 4•(Z 0C/2) 0

1

2

3

4 5 6 Time (nsec)

1 0.8 0.6 0.4 0.2 0

7

8

9

10

trise = 10•(Z 0C/2) 0

1

2

3

4 5 6 Time (nsec)

7

8

9

10pF

V IN

VOut

10

trise = 1•(Z 0C/2) 0

V (volts)

trise = 0.5•(Z 0C/2)

10

As a rule of thumb, if you want reflection to be (1/n)th of signal, risetime should be n•Z 0C/2. ...or, need to terminate the transmitter as well as the receiver, to absorb reflections. Series termination works fine (no terminator at rx), but bandwidth will be halved, i.e., 1/(2πZ0C) instead of 1/( πZ0C)

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Imperfect Terminations: Stubs Suppose the terminator isn't actually at the end of the line?? Reflection,

Transmission,

Γ = -1/3

1−Γ = 2/3

tf = 2 nsec

ZS

tst = 0.25nsec Z0

VIN

VOUT

Z0

VOUT

Volts

VSTUB

Volts

VIN

Volts

...but that's just the first bounce! 1.2 1 0.8 0.6 0.4 0.2 0 1.2 1 0.8 0.6 0.4 0.2 0 1.4 1.2 1 0.8 0.6 0.4 0.2 0

2•V

Z0

Z0

VSTUB

As with cap's, can mitigate the evil effects of stubs with risetime control. 1.2 1 0.8 0.6

tr = 5•t st

0.4 0.2 0

1

2

3

4 5 6 Time (nsec)

7

8

9

10

0 0

1

2

3

4

5

6

7

8

9

10

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A Typical Stub Termination

Z0=75Ω Z0=50Ω tf=40psec tf=40psec

Z0=50Ω tf=1nsec

0.2pF 5nH trace

50Ω

bond wire

pkg

land Stub! ~ 200 psec

Signal at Tx (0.25nsec edge) Internal signal at Rx pad & at terminator

1.2 1 0.8 0.6 0.4 0.2 0 1.6 1.4 1.2 1 0.8 0.6 0.4 0.2 0

2pF pad & rx

Generally, you'd probably like to avoid this type of behavior. • Edge-rate control • Careful board layout • Advanced packaging • On-chip termination

pad term

0

0.5

1

1.5

2

2.5

3

3.5

4

4.5

5

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T-Line Discontinuities Capacitive discontinuity: works pretty much like the capacitively shunted terminator

Inductive discontinuity: the complementary behavior to the capacitive one

Impedance discontinuity: with Z' < Z 0

Z0

Z ' < Z0

Z0

Impedance discontinuity: with Z' > Z 0

Z0

Z ' > Z0

Z0

One thing they have in common: they're all low-pass! ISSCC 1999 Signaling in High-Performance Memory Systems

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Stub Discontinuities Z0

Z0

Z0

Z0

Open Stub with Z ' < Z 0

Z0

Z0

Open Stub with Z ' = 0.5•Z 0

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Some Common Discontinuities

Vias are small capacitive discontinuities (C ~ 0.1pF typically)

Ground-plane slots are inductive discontinuities (also pretty good antennas!). Often created unintentionally in dense fields of vias.

Z < Z0

Multi-drop signals introduced stub discontinuities.

Trace squeezes introduce in-line low-impedance section.

..or..

Connectors introduce impedance discontinuities, often modeled with a combination of lumped elements.

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I1

Field distribution depends on relative directions of currents in neighboring lines...

I1 I2

I2

...and so does impedance!

"Odd-mode" excitation

impedance =

L+ Lm C Cm

Z Odd =

L– Lm C + Cm

Z Lm

L

C

Z0

L

Cm C

distance between lines ISSCC 1999 Signaling in High-Performance Memory Systems

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Crosstalk V fe Z0

Z0

Cm

Vin

Z0 Z0

Vne

Lm

capacitive coupling

inductive coupling

IF lines are homogeneous AND all ports are terminated... equal, when homogeneous

V in Kne•V in V ne V fe t=0

t=tf

t=2•t f

Vne 1 C m L m C = ( + )= m Vin 4 C L 2C V C L K fe = fe = 1 ( m – m ) = 0 Vin 4 C L

K ne =

In inhomogeneous lines, far-end crosstalk gets worse with • faster edges • longer lines ISSCC 1999 Signaling in High-Performance Memory Systems

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solder-mask & air up here FR-4 down here

v c/ εr C Lm ≠ C L

Z0

Microstrips aren't homogeneous; watch out for far-end crosstalk.

Series-terminated back-wave induces far-end crosstalk on neighbor!

Near-end crosstalk can become a far-end problem if transmitters aren't terminated.

Z0

ZT

ZT ≠ Z 0

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Multiconductor Transmission Lines For 2 lines, the Telegrapher's Equations are d V1(x) = – R11 R 12 • I 1(x) –jω L 11 L 12 • I 1(x) dx V2(x) R21 R 22 I 2(x) L 21 L 22 I 2(x) d I 1(x) = – G11 G12 • V1(x) –jω C 11 C 12 • V1(x) dx I 2(x) G21 G22 V2(x) C 21 C 22 V2(x)

These generalize for N coupled lines to d dx d dx

V I

=–R• =–G•

I –jω L • I

where [L], [C], [R], and [G] are NxN matrices

V – jω C • V u-element Field Solver

Geometry LINPAR ($) Ansoft ($$$) Quad Design

HSPICE w-element [L] [C] [R] [G]

($$$)

Circuit simulation with CMOS I/O cells and pkg models Full-board S/I Analysis, 3D solver

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Signaling - Voltage-Mode vs Current-Mode Vdd

Voltage-Mode Signaling "1" and "0" represented by voltages, usually the two power supplies. Low impedance switches required to get line voltages near power supply voltages; these generate the dreaded "simultaneous switching" noise.

VHI ≈ Vdd VLO ≈ GND

Vdd

Current-Mode Signaling "1" and "0" represented by +/− current.

I HI ≈ +I

In well-designed I-mode signaling, I PS never changes. High-impedance current sources isolate the line from the power supplies.

I LO ≈ −I

Signal is converted to a voltage in the receiver's terminator.

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Signaling - References Built-in Reference PFET/NFET split power supply voltage to get reference. Vinv =

+ − + − ref

β P /β N (VDD+VTP ) 1 + β P /β N

Process dependent

Whims of power supply gods

Local Precision Reference External or internal; usually shared among many receivers. Reference fixed, more or less,

but noise across signal return

impedance isn't cancelled...

+ − +

ref line

VTN +

−

Transmitter-generated (Bundled) Reference Hope that noise injected into signal lines is tracked by reference line; unfortunately, Ref Line sees different impedance at far end, since shared by many Rx's.

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Differential Signaling sig sig

Get rid of pesky reference problem by sending a "reference" (actually signal) along with every signal.

−

GOOD: Almost all sources of noise are now common-mode, so can get by with tiny signal voltages, low power BAD: Two wires required, instead of one. Need to keep wires close together, tightly coupled, same length; sometimes an "interesting" engineering problem. Unproved (but widely believed) Lemma: It is always possible to operate a differential signaling system >2x as fast as a single-ended one, all other things being equal.

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Clocking

tfc

Synchronous Every participant gets same frequency and phase.

tfc

phase recovery

magic box & buffer

data

req ack

Mesochronous Every participant gets same frequency, but unknown phase. Requires a way to recover the phase from the data. Coding (e.g. 8b/10b) is often used to make sure there are sufficient data edges. Plesiochronous Every participant gets nearly the same frequency, slowly drifiting phase. Requires a way to detect when the Rx clock has drifted 1/2 cycle from Tx clock. Asynchronous Dispense with clocks altogether, use (e.g.) request/acknowledge 4-phase handshake to ensure correct sequencing of events.

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Clock Forwarding

sig(n) • •

sig(1) • clk

Send a clock along with (a bundle of) data; clock and data "enjoy" the same delay and impedance discontinuties, so skew between data and clock is minimized. Currently very fashionable, and works quite well up to ~ 1GHz. At higher frequencies, gets harder to keep the delays matched sufficiently well to maintain synchronous operation.

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Why Worry About Signaling? So, what's wrong with the way we've been doing things for years?

V out

Reference Inverters make really terrible receivers. Often as much as 500 mV gap between guaranteed HI and LO. V in 0.5 V

Termination Termination (for minimum tx current) needs to be at about Vinv, requiring a second power supply or voltage divider. In 50Ω environment, divider consumes ~50 mW even without a signal! Transmitter FETs biased into linear region, so low resistance; any reflections from impedance discontinuities bounce off the transmitter and head for the receiver. ISSCC 1999 Signaling in High-Performance Memory Systems

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Signaling? ...cont.

Need ± 1 volt of signal, so ± 20 mA of signal current. Vbond = L bond dI = 5nH• 20mA = 1 volt Yikes! dt 0.1nsec ...and several Tx's may have to share a common supply lead... Voltage-mode signaling systems themselves generate most of the noise that they must overcome. V Noise ∝ VSignal , a loosing proposition. Power ~10-20 mW dissipated in the transmitter, and ~70 mW in the terminator. Does it really take nearly 100 mW to send a bit from one chip to another??

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What's a Bus? Vt

Vt

Z0

Z0

It's a transmission line with multiple discontinuities. Each driver sees a line impedance of Z 0/2, ...so, to maintain receiver voltage, need twice the driver power as in a point-to-point signaling system. This is kinda bad, but what about those discontinuities??

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Effects of Multiple Capacitive Point Discontinuities 1 nsec Z0 = 50Ω

1 0.75 0.5 0.25 0

10 nsec Z0 = 50Ω

1 0.75 0.5 0.25 0

8pF 0

2

4

6

8

10 12 14 16 18 20

1 0.75 0.5 0.25 0

1

0

2

4

6

8

1 2

4

6

8

1 2

4

6

8

1

1

1

1

4

6

8

10 12 14 16 18 20

0

2

4

6

8

10 12 14 16 18 20

0

2

4

6

8

10 12 14 16 18 20

0

2

4

6

8

10 12 14 16 18 20

1 0.75 0.5 0.25 0

8

10 12 14 16 18 20

1 0.75 0.5 0.25 0 0

1

2

1 0.75 0.5 0.25 0

9

10 12 14 16 18 20

1 0.75 0.5 0.25 0 0

1

0

1

1 0.75 0.5 0.25 0

7

10 12 14 16 18 20

4 1 0.75 0.5 0.25 0

1

1

1

1

1

1

1

1

1

1

1

3 0

2

4

6

8

1 0.75 0.5 0.25 0

1 0

10 12 14 16 18 20

2

4

6

8

10 12 14 16 18 20

2

1. Reduced bandwidth 2. Increased delay

3. Reduced impedance 4. Overshoot, ringing

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Lumped Model if t rise >> 2•t fs ...

tfs

Z Loaded = L C

CLoad

=

L∆X

L∆x C∆x+C Load Z0

1 + CLoad/C∆X

C∆X + C Load

So, define "bus factor" B f Bf =

1+CLoad/C∆x =

Charateristic impedance Wave velocity Cutoff frequency Time of flight through a bus segment

Cutoff frequency for a lumped-element line is: 1+CLoad/C Seg Z0 Bf v vBus = Line Bf 1 F0 = π t fs Bf Z Bus =

T0 = B f tfs

1 π LC 1 = π L∆x•(C∆x+C LOAD) 1 = = 1 π t fsB f π∆x LC B f

F0 =

Nice discussion in Tom Lee's The Design of CMOS RadioFrequency Integrated Circuits ... a terrific book!

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Allowable Risetime Out

trise = 1•B f•tfs

trise = 2•B f•tfs

trise = 5•B f•tfs

trise = 10•B f•t fs Rule of thumb: t rise ≥ 5•Bf•tfs to reduce overshoot & ringing and...

FMax =

1 = 1 ≈ 0.15• F0 4•t rise 4•5•T0

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Some Examples Assumes Z 0 = 70Ω, 1/v = 7.2 psec/mm Memory

Pkg

CLoad

CSeg

400-mil TSOP

6.5 pF

1.25 pF

2.49 28 Ω 215 psec 1.48 GHz

VSMP

3 pF

0.42 pF

2.85 25 Ω

82 psec 3.88 GHz

4.06 17 Ω

116 psec 2.72 GHz

1.84 38 Ω

132 psec 2.40 GHz

12mm SDRAM

4mm

SLDRAM (SLIO) SLDRAM (LVCMOS)

6.5 pF

10mm RDRAM

CSP

2.4 pF

1.0 pF

Bf

ZBus tdrop-to-drop

F0

Observations: Try to increase F 0 by making t fs smaller, then Z Bus gets smaller too. Try to increase Z Bus by making Z 0 higher, then F 0 decreases.

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Stubs tfs tstub

trise = 0.5•t stub 5•t fs

trise ≥

1+(C Load+C stub)/C∆x

t stub

trise = 5•t stub

Lines with stubs are complex circuits. Accurate modeling and circuit simulation required!

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Voltage-Mode Bus Signaling – PC-100 500 B

Output Characteristics (Intel PC-100 spec) ROut ~ 6.5Ω

400 Ipin (mA)

10nH

SDRAMs: Power pin per driver (x4's & x8's); shared on x16's.

6.5pF

PMOS

300

B BB

J

100

B B

J

B

0 J 0

Typical Application

J

J B

J J

TSOPs on both sides of DIMMs

NMOS

B BJ J B JJ J

200

0.5

SDRAM in Model

1

1.5 2 Vpin (volts)

2.5

3

B 3.5

Simulation of simple model: in

4

Volts

3

10 mm

2

out

1 0 -1 0

out

DIMM Model

5

10

15 20 25 Time (nsec)

30

35

40

Even without termination, Pd ~ 100 mW/pin

No Terminator! ISSCC 1999 Signaling in High-Performance Memory Systems

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Series Stub Terminated Logic (SSTL) Conventional CMOS Signaling

1.5V

0.1 ns, Z 0=50 Ω

25Ω

Pterm = 40 mW Ptx = 60 mW

1.5V 25Ω

Volts

0.25 ns, Z0=50 Ω 6.5pF

3.5 3 2.5 2 1.5 1 0.5 0

1.8 V

0

SSTL-Style Signaling

1.5V

Vtt=1.5V

40Ω

(two term's)

5

10 15 Time (nsec)

20

25

Pterm = 5 mW (2 term's) Ptx = 12.5 mW

40Ω Ω 25Ω

2.5

Volts

2

650 mV

1.5 1 0.5

40Ω

0

3pF Vref = Vtt

0

5

10 15 Time (nsec)

20

25

+ −

better receiver and (slightly) better reference.

Series resistor isolates stub from line, increases line impedance, reduces ringing, reduces power ISSCC 1999

43 of 59

Signaling in High-Performance Memory Systems

SLDRAM (SyncLink DRAM) Vtt

2 strobe(clk)

Command Link

10+2

command

Clk Cntrl

δ

δ

δ

requests

•••

Controller Memory System

16+2

data

Data Link

4

E,O (clks)

• Uses SSTL-style signaling (alternatively, LVCMOS) • Differential clocks, double-edge-triggered FF's • Source-synchronous timing (controller-adjustable δ's) • 200 MHz (800 MByte/sec) operation

Vtt

Perhaps more relevantly: DDR's use SSTL! ISSCC 1999 Signaling in High-Performance Memory Systems

44 of 59

Wired-OR Bus Signaling VH =Vtt

VH

In wired-OR scheme: Driver supplies the "0's" Terminator supplies "1's"

Vref

Vref

+ −

+ −

Enab•Data

Driver has to sink twice the current as in a push-pull driver to get same signal voltage

Enab•Data

Some example signaling systems: assumed

Family

VH

BTL (Backplane Transmission Logic)

GTL (Gunning Transmission Logic)

RSL (Direct RAMBUS)

VL

Vref

Export power dissipation from on-chip driver to terminator resistor

Zbus

IOL

(two term's) Pd(term)

Pd(tx)

Remarks

2.1

1.0

1.55

30Ω

73mA

80mW

current73mW Bipolar mode

1.2

0.4

0.8

30Ω

53mA

42mW

21mW

Voltage Mode

2.5 1.8

1.5 1.0

2.0 1.4

50Ω 50Ω

40mA 32mA

40mW 26mW

80mW 32mW

Current Mode

Note: power is dissipated only when driving LO, so, averaged over 1's and 0's, actual operating power is about one-half of these values. ISSCC 1999 45 of 59

Signaling in High-Performance Memory Systems

Wired-OR Glitch VHI Vtt V LO

Vtt

tbus

Wired-OR scheme with "voltage-mode" drivers requires two full bus transit times to settle the 'glitch'. tbus Left End

VLL

Left of Middle

Right End

Current-mode drivers get it done in one bus transit time, because no reflections off the drivers.

tbus

tbus Left End

VLL

Left of Middle

Right End

ISSCC 1999 Signaling in High-Performance Memory Systems

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Direct RAMBUS (RDRAM) High-End Application

Low-End Application

Devices on motherboard + expansion board

Multiple "RIMM" Modules

RDRAM

RDRAM

•••

RDRAM

Controller 16+2 Data

RAC

8

Cntl,Addr VRef

2

ClkFM

2

ClkTM

ISSCC 1999 47 of 59

Signaling in High-Performance Memory Systems

RDRAM Clocking

Memory System RDRAM

Controller

DLL

DLL

+−

+−

V tt ClkFromMaster + −

Custom clock oscillator with open-drain driver

ClkToMaster

A variation on source-synchronous timing: Clocks travel in same direction as data for both types of transfers

ISSCC 1999 Signaling in High-Performance Memory Systems

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RDRAM Clocking with Expansion Board Matched Expansion Board

ClkFromMaster

Matched

ClkToMaster Matched loop delays Expansion Connector ClkFromMaster

Matched

ClkToMaster Controller

System Board

These connections not strictly necessary, but included to help average out mismatches

ISSCC 1999 49 of 59

Signaling in High-Performance Memory Systems

RDRAM Signaling VHI Controller + −

Single-ended RDRAM Signaling

VLO

tram

tram

Vtt

tram

VRDRAM

Vref

VController

RDRAM

A variation on source termination; works because RDRAM slaves only talk to Controller Double-ended Signaling

Expansion Board V tt Controller + −

tram

tram

V tt

VRDRAM VController

A stub – t stub must be