IEEE MTT-V053-I12 (2005-12) [53, 12 ed.]

303 117 14MB

English Pages 228 Year 2005

Table of contents :
010 - 01550014......Page 1
020 - 01550015......Page 3
II. M ULTILAYER CPS W ITH F INITE D IELECTRIC D IMENSIONS......Page 4
Fig.€3. Cross-sectional view in $\tau$ -plane after the transfor......Page 5
IV. N UMERICAL R ESULTS......Page 6
Fig.€12. Effective dielectric constant of ME CPS and MC CPS as a......Page 7
V. C ONCLUSION......Page 8
V. Fouad Hanna and D. Thebault, Analysis of asymmetrical coplana......Page 9
I. I NTRODUCTION......Page 10
Fig.€2. Plots of tau ( $\tau$ ) with various gamma ( $\gamma$ )......Page 11
Fig.€5. Schematic of PCM/phase-conjugation polarization modulato......Page 12
Fig.€7. Schematic of PSD.......Page 13
Fig.€10. Plotted minimum jammer attenuation level needed to prev......Page 14
Fig.€14. Demonstration of active polarization modulation by comp......Page 15
A. Full-Duplex Measurement......Page 16
B. Radiation Patterns......Page 17
Fig.€25. Monostatic RCS.......Page 18
D. M. Pozar, Microwave Engineering . New York: Wiley, 1998.......Page 19
II. A NTENNAS, A RRAYS, AND P HASE S HIFTERS......Page 20
TABLE II D IMENSIONS OF THE D ESIGNED P HASE S HIFTER......Page 21
III. 2 20-GHz MMIC-B ASED A MPLIFIERS......Page 22
Fig.€9. System block diagram illustrating the test setup for the......Page 23
Fig.€12. Measured beam-steering patterns at: (a) 8, (b) 16, and......Page 24
A. Hirata, T. Morimoto, and Z. Kawasaki, DOA estimation of ultra......Page 25
II. T HEORETICAL A NALYSIS......Page 27
A. Autocorrelation Function......Page 28
A. Frequency-Domain Measurements......Page 29
Fig. 6. PDFs of the noise at the output of amplifier #1. (a) Lin......Page 30
Fig. 10. Output noise spectral density $S _{yn} (f)$ of amplifie......Page 31
B. Convolution Products of the Noise......Page 32
R. Price, A useful theorem for nonlinear devices having Gaussian......Page 33
A. Geens and Y. Rolain, Noise figure measurements on nonlinear d......Page 34
I. I NTRODUCTION......Page 35
III. E XPERIMENTAL R ESULTS......Page 36
IV. T RANSISTOR M ODELING......Page 37
B. Instability Contour......Page 38
C. Analysis of the Self-Oscillating Mixer Regime......Page 39
D. Hysteresis Prediction......Page 40
Fig.€11. Expanded view of the experimental output power spectrum......Page 41
Fig.€13. Amplifier schematic with stabilization network. The sta......Page 42
Fig.€14. Stabilization action of the parallel resistance analyze......Page 43
D. Teeter, A. Platzker, and R. Bourque, A compact network for el......Page 44
E. de Cos, A. Suárez, and S. Sancho, Envelope transient analysis......Page 45
I. I NTRODUCTION......Page 46
B. Example......Page 47
C. Algorithm......Page 48
Fig.€4. Relationship between input load and: (a) throughput and......Page 49
Fig.€6. Relationship between input load and: (a) throughput and......Page 50
2) Throughput and Fairness Index Performances Versus Number of T......Page 51
3) Computational Complexity Comparison: Finally, we examine the......Page 52
M. Ishida et al., Development of gigabit millimeter-wave broad-b......Page 53
A. Simplified Scheme of the Instrument......Page 54
B. Analytic Description......Page 55
Fig.€3. Difference between individual phase measurements and the......Page 56
10 hints for making better network analyzer measurements, Agilen......Page 57
Fig.€1. Conventional wake-up system with a microprocessor.......Page 58
II. S YSTEM O VERVIEW......Page 59
Fig.€5. (a) Prototype. (b) Conversion-loss graph of up/down phas......Page 60
V. M EASUREMENT R ESULTS......Page 61
B. Adaptive Power-Management Performance......Page 62
Fig.€13. Prototype of the proposed system.......Page 63
C. Retrodirective Array Performance......Page 64
J. B. Hacker, J. Bergman, G. Nagy, G. Sullivan, C. Kadow, H.-K.......Page 65
N. Shinohara and H. Matsumoto, Dependence of DC output of a rect......Page 66
I. I NTRODUCTION......Page 67
Fig.€2. Frequency synthesizer in an MB-UWB transceiver.......Page 68
Fig.€3. Frequency tree diagram.......Page 69
III. F REQUENCY P LANNING......Page 70
IV. S YNTHESIZER A RCHITECTURE FOR C URRENT B AND P LAN......Page 71
V. A LTERNATE B AND P LAN AND S YNTHESIZER A RCHITECTURE......Page 72
Fig.€9. Sideband rejection with amplitude and phase error.......Page 73
VI. M ACROMODEL S IMULATION R ESULTS AND P ERFORMANCE A NALYSIS......Page 74
Fig.€12. Macromodel for the evaluation of the impact of the synt......Page 75
TABLE VII D IFFERENT E VALUATION S CENARIOS OF BER D EGRADATION......Page 76
VII. C ONCLUSIONS......Page 77
T. H. Lee, The Design of CMOS Radio-Frequency Integrated Circuit......Page 78
Fig.€2. Block diagram of the proposed BPSK demodulation scheme.......Page 80
II. ILOs......Page 81
1) Phase Locking: In this case, $f_{\rm lock}=f_{r}$ (i.e., $\th......Page 82
4) Unlocking: In this case, $ {4}\vert f_{r}-f_{\rm lock}\vert>f......Page 83
Fig.€12. Schematic representation of the evolution of angle $% \th......Page 84
B. Bit Rate Estimation......Page 85
C. Conversion Dynamics......Page 86
2) Cases $b$ and $c$: In these cases, we have to take into accou......Page 87
IV. C ONCLUSION......Page 88
F. Ramirez, M. E. de Cos, and A. Suárez, Nonlinear analysis tool......Page 89
Fig.€1. Capacitively coupled DAs for higher power handling using......Page 90
B. Capacitively Coupled Common-Source Stage Analysis......Page 91
C. Analysis of Common-Gate Output Impedance in the Capacitively......Page 92
IV. M ATLAB AND ADS I NVESTIGATION......Page 93
V. D ESIGN AND M EASUREMENTS......Page 94
Y. Ayasli, S. W. Miller, R. Mozzi, and L. K. Hanes, Capacitively......Page 95
J. Shohat, I. D. Robertson, and S. J. Nightingale, 10 Gb/s drive......Page 96
II. R EVIEW OF THE D ESIGN OF U NIFORM - AND S TEPPED -I MPEDANC......Page 97
III. D ESIGN OF N EW PCL F ILTERS......Page 98
Fig.€4. Exact circuit transformations. (a) Basic $S$ -plane sub-......Page 99
Fig.€5. (a) $S$ -plane bandpass prototype and (b) its correspond......Page 100
Fig.€6. Development of a seventh-order PCL filter. (a) Synthesiz......Page 101
V. R ESULTS AND D ISCUSSION......Page 102
M. C. Velazquez-Ahumada, J. Martel, and F. Medina, Parallel coup......Page 103
G. I. Zysman and A. K. Johnson, Coupled transmission line networ......Page 104
II. E XPERIMENTAL P ROCEDURE......Page 105
III. S YSTEM A NALYSIS......Page 106
IV. R ESULTS......Page 107
Fig.€6. Results displaying: (a) simulation of sideband behavior......Page 108
V. C ONCLUSION......Page 109
K. J. Williams, Electro-optical broad-band microwave frequency s......Page 110
I. I NTRODUCTION......Page 111
III. F OUR -P OLE P ARALLEL -C OUPLED F ILTER D ESIGN......Page 112
Fig.€5. Measured and simulated performance of the filter. (a) In......Page 113
TABLE€II D ESIGN P ARAMETERS OF THE SIR ON A S UBSTRATE OF D IE......Page 114
M. del Castillo Velázquez-Ahumada, J. Martel, and F. Medina, Par......Page 115
J. S. Hong and M. J. Lancaster, Microstrip Filter for RF/Microwa......Page 116
II. P RINCIPLES AND S IMULATION......Page 117
C. Analyzing Mode Purity......Page 118
B. Verifying Azimuthal Symmetry......Page 119
IV. C ONCLUSION......Page 120
T. H. Chang and C. F. Yu, Polarization controllable ${\rm TE}_{2......Page 121
I. I NTRODUCTION......Page 122
B. Numerical Results and Analysis......Page 123
Fig.€3. Simulated phase delay in the output wave with respect to......Page 124
B. Numerical Results......Page 125
IV. M EASUREMENT R ESULTS AND A NALYSIS......Page 126
D. M. Pozar, Microwave Engineering, 2nd ed. New York: Wiley, 199......Page 127
Fig.€1. Hartley phasing-type SSB modulator. The arrows indicate......Page 129
3) Monitor Circuit: A sample of the modulator's output signal is......Page 130
2) Limited Selectivity of Monitor Circuit: The selectivity of th......Page 131
V. T EST S ETUP......Page 132
1) Complementary Attenuator Control: The output voltage $V_{\rm......Page 133
E. Monitor Detectors......Page 134
1) Variation of LO Frequency: Measurement results obtained from......Page 135
D. Sensitivity to Operating Parameters......Page 136
VIII. C OMPARISON TO P REVIOUSLY P UBLISHED W ORKS......Page 137
3) Transmit Operation: When setting switch SW1 to TX, the SSB mo......Page 138
T. Brauner, R. Vogt, and W. Bächtold, A versatile calibration me......Page 139
II. R ECTANGULAR C AVITY R ESONATOR......Page 140
III. C AVITY F EEDING S TRUCTURE......Page 141
TABLE€I D ESIGN P ARAMETERS OF C AVITY R ESONATORS U SING O PEN......Page 142
IV. T HREE -P OLE F ILTER......Page 143
Fig. 8. (a) External $Q$ factor $({ Q}_{\rm ext})$ evaluated as......Page 144
M. J. Hill, R. W. Ziolkowski, and J. Papapolymerou, Simulated an......Page 145
J.-S. Hong and M. J. Lancaster, Microstrip Filters for RF/Microw......Page 146
A. Small-Signal Model......Page 148
C. Noise-Parameter Relationships......Page 149
Fig.€3. pHEMT noise equivalent-circuit models of CS, CG, and CD......Page 150
TABLE€IV N OISE P ARAMETER R ELATIONSHIPS B ETWEEN CS AND CD C......Page 151
Fig.€7. Comparison of modeled and measured noise parameter for t......Page 152
Fig.€10. Comparison of noise parameters in CS, CG, and CD config......Page 153
A. Cappy, Noise modeling and measurement techniques, IEEE Trans.......Page 154
I. I NTRODUCTION......Page 155
III. W AVEGUIDE C HARACTERISTICS AND T RANSITION D ESIGN......Page 156
B. Measured Results......Page 157
A. Design......Page 158
B. Measured Results......Page 159
G. Pringent, E. Rius, F. L. Pennec, S. L. Maguer, C. Quendo, G.......Page 160
Website......Page 162
240 - 01550037......Page 163
250 - 01550038......Page 164
260 - [email protected] 228

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DECEMBER 2005

VOLUME 53

NUMBER 12

IETMAB

(ISSN 0018-9480)

PAPERS

Quasi-Static Solutions of Multilayer Elliptical, Cylindrical Coplanar Striplines and Multilayer Coplanar Striplines With Finite Dielectric Dimensions—Asymmetrical Case. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . V. Akan and E. Yazgan Mutually Exclusive Data Encoding for Realization of a Full Duplexing Self-Steering Wireless Link Using a Retrodirective Array Transceiver. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . K. M. K. H. Leong and T. Itoh Ultra-Wideband Low-Cost Phased-Array Radars . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . C. T. Rodenbeck, S.-G. Kim, W.-H. Tu, M. R. Coutant, S. Hong, M. Li, and K. Chang Noise Behavior of Microwave Amplifiers Operating Under Nonlinear Conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . L. Escotte, E. Gonneau, C. Chambon, and J. Graffeuil Global Stability Analysis and Stabilization of a Class-E/F Amplifier With a Distributed Active Transformer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . S. Jeon, A. Suárez, and D. B. Rutledge Frequency Channel Blocking Scheme in Mesh-Topology Millimeter-Wave Broad-Band Entrance Networks. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .J. Sangiamwong, K. Tsukamoto, and S. Komaki Effect of Cable Length in Vector Measurements of Very Long Millimeter-Wave Components . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . A. Simonetto, O. D’Arcangelo, and L. Figini Adaptive Power Controllable Retrodirective Array System for Wireless Sensor Server Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . S. Lim, K. M. K. H. Leong, and T. Itoh Frequency Planning and Synthesizer Architectures for Multiband OFDM UWB Radios . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . C. Mishra, A. Valdes-Garcia, F. Bahmani, A. Batra, E. Sánchez-Sinencio, and J. Silva-Martinez BPSK to ASK Signal Conversion Using Injection-Locked Oscillators—Part I: Theory . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . J. M. López-Villegas and J. J. Sieiro Cordoba Investigation of Drain-Line Loss and the Kink Effect in Capacitively Coupled Distributed Amplifiers. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . J. Shohat, I. D. Robertson, and S. J. Nightingale Parallel-Coupled Line Filters With Enhanced Stopband Performances . . . . . . . . . . . . . W. M. Fathelbab and M. B. Steer RF Frequency Shifting via Optically Switched Dual-Channel PZT Fiber Stretchers . . . C. S. McDermitt and F. Bucholtz Design of Microstrip Bandpass Filters With Multiorder Spurious-Mode Suppression . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . C.-F. Chen, T.-Y. Huang, and R.-B. Wu High-Performance Circular -Mode Converter. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .C.-F. Yu and T.-H. Chang

3681 3687 3697 3704 3712 3723 3731 3735 3744 3757 3767 3774 3782 3788 3794

(Contents Continued on Back Cover)

(Contents Continued from Front Cover) Compact U-Shaped Dual Planar EBG Microstrip Low-Pass Filter . . . . . . . . . . . . . . . . . . . . . S. Y. Huang and Y. H. Lee Investigation of a Self-Calibrating SSB Modulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . D. M. Treyer and W. Bächtold Low-Loss LTCC Cavity Filters Using System-on-Package Technology at 60 GHz . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . J.-H. Lee, S. Pinel, J. Papapolymerou, J. Laskar, and M. M. Tentzeris Relationships Between Common Source, Common Gate, and Common Drain FETs . . . . . . . . . . . . J. Gao and G. Boeck Millimeter-Wave Substrate Integrated Waveguides and Filters in Photoimageable Thick-Film Technology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . D. Stephens, P. R. Young, and I. D. Robertson

3799 3806

Information for Authors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

3839

3817 3825 3832

CALLS FOR PAPERS

Special Issue on Applications of Ferroelectrics in Microwave Technology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

3840

2005 INDEX OF MTT TRANSACTIONS .

3840

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IEEE Periodicals Transactions/Journals Department Staff Director: FRAN ZAPPULLA Editorial Director: DAWN MELLEY Production Director: ROBERT SMREK Managing Editor: MONA MITTRA Senior Editor: CHRISTINA M. REZES IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES (ISSN 0018-9480) is published monthly by the Institute of Electrical and Electronics Engineers, Inc. Responsibility for the contents rests upon the authors and not upon the IEEE, the Society/Council, or its members. IEEE Corporate Office: 3 Park Avenue, 17th Floor, New York, NY 10016-5997. IEEE Operations Center: 445 Hoes Lane, P.O. Box 1331, Piscataway, NJ 08855-1331. NJ Telephone: +1 732 981 0060. Price/Publication Information: Individual copies: IEEE Members $20.00 (first copy only), nonmember $69.00 per copy. (Note: Postage and handling charge not included.) Member and nonmember subscription prices available upon request. Available in microfiche and microfilm. Copyright and Reprint Permissions: Abstracting is permitted with credit to the source. Libraries are permitted to photocopy for private use of patrons, provided the per-copy fee indicated in the code at the bottom of the first page is paid through the Copyright Clearance Center, 222 Rosewood Drive, Danvers, MA 01923. For all other copying, reprint, or republication permission, write to Copyrights and Permissions Department, IEEE Publications Administration, 445 Hoes Lane, P.O. Box 1331, Piscataway, NJ 08855-1331. Copyright © 2005 by The Institute of Electrical and Electronics Engineers, Inc. All rights reserved. Periodicals Postage Paid at New York, NY and at additional mailing offices. Postmaster: Send address changes to IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, IEEE, 445 Hoes Lane, P.O. Box 1331, Piscataway, NJ 08855-1331. GST Registration No. 125634188.

Digital Object Identifier 10.1109/TMTT.2005.862199

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Quasi-Static Solutions of Multilayer Elliptical, Cylindrical Coplanar Striplines and Multilayer Coplanar Striplines With Finite Dielectric Dimensions—Asymmetrical Case Volkan Akan, Student Member, IEEE, and Erdem Yazgan, Member, IEEE

Abstract—In this paper, fast, accurate, and simple analytic closed-form expressions are presented in order to calculate the quasi-TEM parameters of multilayer elliptical coplanar stripline (CPS), multilayer cylindrical CPS, and multilayer CPS, with finite dielectric dimensions by using conformal mapping techniques. The obtained computer-aided-design-oriented expressions are quite accurate and easy to apply in designing microwave integrated circuits and antenna applications. The results have also been compared with the ones of another conformal mapping method available in the literature and also confirmed analytically. Index Terms—Computer-aided design (CAD)-oriented formulas, conformal mapping, coplanar stripline (CPS), multilayer cylindrical coplanar stripline (MC CPS), multilayer elliptical coplanar stripline (ME CPS).

I. INTRODUCTION

O

WING TO the developments in microwave integrated circuit (MIC) and monolithic microwave integrated circuit (MMIC) design techniques, coplanar transmission lines have been employed widely. Mixers, amplifiers, switches, and matching circuits such as baluns can be given as examples of the applications of coplanar transmission lines. Recently, elliptical and cylindrical geometries have been extensively analyzed by many authors as in [1]–[5]. The elliptical and cylindrical transmission lines have been studied for applications of slotted lines, transition adapters, and baluns, as mentioned in [6]. However, according to the literature, the quasi-static solutions of the multilayer coplanar stripline (CPS) with finite dielectric dimensions have not been studied. The main purpose of this paper is to survey the effects of finite dielectric dimensions to quasi-TEM parameters of CPS with multilayer dielectrics, as well as to obtain the quasi-TEM parameters of multilayer elliptical coplanar stripline (ME CPS) and multilayer cylindrical coplanar stripline (MC CPS) via the conformal mapping technique (CMT). It is known that in order to increase the mechanical strength and the average power handling of a CPS, another dielectric substrate is mounted below the fragile substrate.

Manuscript received October 6, 2004; revised April 27, 2005. This work was supported by the Hacettepe University Research Foundation under Contract 0302 602 012. The authors are with the Department of Electrical and Electronics Engineering, Hacettepe University, 06800 Beytepe, Ankara, Turkey (e-mail: [email protected]; [email protected]). Digital Object Identifier 10.1109/TMTT.2005.856080

Fig. 1. Cross-sectional view of multilayer asymmetrical CPS with finite dielectric dimensions.

In this study, the thickness of metal strips are assumed as infinite-small thin and the dielectric materials are lossless. Since the structure to be analyzed work in the quasi-TEM mode, a quasi-static approach and CMT can be used. As is known, the quasi-TEM approximation can be used exclusively for lowfrequency range. Nevertheless, it has been reported in [7] and [8] that the quasi-static parameters of coplanar transmission lines are slightly sensitive to changes in frequency up to 40 GHz according to the dispersion characteristics of them. Therefore, the quasi-TEM parameters of CPS can be evaluated using CMT. In addition, analytic closed-form expressions obtained in this paper are appropriate for computer-aided design (CAD) applications. Firstly, a multilayer CPS with finite dielectric dimensions will be analyzed. Afterwards, in Section III, ME CPS and MC CPS are examined, respectively. In Section IV, the results are compared with the method proposed in [1] and analytical confirmation is also presented.

II. MULTILAYER CPS WITH FINITE DIELECTRIC DIMENSIONS The cross-sectional view of a multilayer CPS to be analyzed is seen in Fig. 1. As shown in this figure, an asymmetrical CPS is above the layered two dielectric substrates with dielectric conand , respectively. There is also another dielectric stants superstrate that has dielectric constant on the CPS. The total and the widths of metal strips width of the configuration is and , as seen in Fig. 1. The dielectric materials’ heights are , and for , , and , respectively. are ,

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Fig. 2. Realized illustration for the equivalent dielectric constant " where = 1; 2; 3.

(for h )

The total line capacitance per unit length of the multilayer CPS with finite dielectric dimensions can be expressed as the sum of four capacitances , , , and , as given in [9] (1) Here, is the air capacitance after removing the overall diis the capacitance of the dielectric layer electric material; and equivalent dielectric constant ( with thickness ). is also the capacitance of dielectric material with thickness and having dielectric constant ( ). is the capacitance of dielectric material with thickLastly, and having dielectric constant ( ). After ness giving these definitions, partial capacitances can be calculated. For the calculation of air-capacitance , the below equation must be written similar to the expression given in [2] (2)

Fig. 3. Cross-sectional view in -plane after the transformation using Jacobi elliptic function.

where the height of the Therefore, for the case dielectric is , can be written as follows: (5) For the case where the height of the dielectric is , can be written similarly as follows: (6) and, lastly, for the case is then dielectric is ,

, where the height of the

(7) The air-capacitance

is also given by (8)

where

In order to calculate , , and , Fig. 2 can be realized with defined heights as above. By using the Jacobian elliptic function used as in [1] and [2] and seen as follows:

(9)

(3)

is again the complete elliptic integral of the first kind, is the modulus, and is complementary of and . Thus, the total line capacitance per unit length is written as

is the complete elliptic integral of the first kind; , for , , and , reis the modulus spectively, and where

(10) Moreover, the effective dielectric constant can be written as

for

(11) where for

(12) (4)

Assuming that the dielectric-air borders are magnetic walls, then Fig. 2 is transformed to Fig. 3. Via the Schwarz–Christoffel transformation, partial capacitances can also be calculated.

The characteristic impedance and the guided wavelength are given by (13)

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Fig. 4. Cross-sectional view of ME CPS. Fig. 6. Effective dielectric constant of multilayer asymmetrical CPS as a function of h =(s + w ) for " = 12:9, " = " = 1.

The MC CPS is also transformed into a multilayer CPS, as shown in Fig. 1, by using the following function: (18) The related parameters in Fig. 1 can then be written as (19) and (20) Fig. 5.

Cross-sectional view of MC CPS after transformation of ME CPS.

and (14) The developed formulations written above will give a simple analysis procedure for the determination of the quasi-TEM parameters of ME CPS and MC CPS. These are shown in Section III. III. ME CPS AND MC CPS The cross-sectional view of ME CPS is seen in Fig. 4. The focal distance relations of the confocal elliptic cylinders are (15) Here, and are semimajor and semiminor axes of elliptic cylinders, respectively, where , as shown in Fig. 4. By using the following function, the ME CPS that is seen in Fig. 4 is transformed to MC CPS (as shown in Fig. 5) (16) The radii of the last structure can then be written as (17)

After writing these expressions, the quasi-TEM parameters of the ME CPS and MC CPS are straightforward enough to be obtained by using the CMT. IV. NUMERICAL RESULTS In order to testify that the method used in this study is feasible, initially the finite dielectric dimensions of the multilayer asymmetrical CPS have been increased for , . As seen from Fig. 6, while increasing the dimensions, the effective dielectric constant approaches (see for the case ), as mentioned in [1], [10], and [11]. Also, in Fig. 7, the characteristic impedance of the multilayer asymmetrical CPS is shown as a function of , and it is seen that decreases as increases. After indicating analytical verification, a second dielectric substrate is added below the first one, and a superstrate above the structure leads to an increase in the value of the effective dielectric constant, as seen in Fig. 8, and adversely brings about reducing the value of the characteristic impedance, as shown in Fig. 9. However, for this case, by examining Figs. 8 and 9, it can be shown that when is increased, increases and decreases. The important point is that these changes are slightly sensitive to variation of . Despite the fact that increasing of affects and similar to the preceding case, these changes are more remarkable than the effect of ( ). This can be clearly seen from Figs. 10 and 11. From Figs. 6–11, the effect of enlarging the dielectric dimension is also shown as increasing the value

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Fig. 7. Characteristic impedance of multilayer asymmetrical CPS as a function of h =(s + w ) for " = 12:9, " = " = 1.

Fig. 10. Effective dielectric constant of multilayer asymmetrical CPS as a function of h =(s + w ).

Fig. 8. Effective dielectric constant of multilayer asymmetrical CPS as a h )=(s + w ) . function of (h

Fig. 11. Characteristic impedance of multilayer asymmetrical CPS as a function of h =(s + w ).

Fig. 9. Characteristic impedance of multilayer asymmetrical CPS as a function of (h h )=(s + w ) .

Fig. 12. Effective dielectric constant of ME CPS and MC CPS as a function of 2=(2 + + ) for G = (2 + + ) = 40 and " = 12:9, " = " = 1.

of the effective dielectric constant and decreasing the value of the characteristic impedance.

From Figs. 12–15, some general expected facts can be observed easily for ME CPS and MC CPS. Firstly, as the ratio of

0

0

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Fig. 13. Characteristic impedance of ME CPS and MC CPS as a function of 2=(2 + + ) for G = (2 + + ) = 40 and " = 12:9, " = " = 1.

Fig. 16. Comparisons of the effective dielectric-constant values of ME CPS and MC CPS as a function of 2=(2 + + ) for G = (2 + + ) = 30 by using the method in [1] and the method of this paper.

Fig. 14. Effective dielectric constant of ME CPS and MC CPS as a function of 2=(2 + + ) for G = (2 + + ) = 40 .

Fig. 17. Comparisons of characteristic impedance values of ME CPS and MC CPS as a function of 2=(2 + + ) for G = (2 + + ) = 30 by using the method in [1] and the method of this paper.

increased, decreases and increases. In addition, while asymmetry ratio ( ) decreases, and increases. In Figs. 16 and 17, the comparisons of the results of the values of and are shown by using the method proposed in [1] and via the method presented in this paper. The results are very close to each other. Therefore, this is the second proof of the correctness of this study. V. CONCLUSION

Fig. 15. Characteristic impedance of ME CPS and MC CPS as a function of 2=(2 + + ) for G = (2 + + ) = 40 .

In this study, a multilayer asymmetrical CPS with finite dielectric dimensions has been studied in order to calculate the quasi-TEM parameters, and the derived results have been presented. Step-by-step related conformal transformations have then been realized to analyze the ME CPS and MC CPS. The obtained closed-form expressions are quite easy and accurate to utilize in CAD tools. Furthermore, the effect of altering the physical parameters of multilayer asymmetrical CPS with finite dielectric dimensions, ME CPS and MC CPS can be illustrated

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and interpreted easily as compared to numerical methods. In this paper, three layer dielectric substrates have been inserted for the studied structures. The number of dielectric layers can be increased and then obtained structures can be examined easily by using the proposed method in this study. Finally, it can be emphasized that numerical methods are unwieldy and slow since they consume much more time and memory in computers while calculating the related parameters as compared with the presented method for the given transmission-line structures. REFERENCES [1] Z. W. Du, K. Kong, J. S. Fu, Z. Feng, and B. Gao, “CAD models for asymmetrical, elliptical, cylindrical and elliptical cone coplanar strip lines,” IEEE Trans. Microw. Theory Tech., vol. 48, no. 2, pp. 312–316, Feb. 2000. [2] V. Akan and E. Yazgan, “A simple formulation for quasi-static solutions of elliptical, cylindrical and asymmetrical coplanar strip lines,” Microwave Opt. Technol. Lett., vol. 41, no. 1, pp. 18–21, Apr. 2004. [3] , “Quasistatic TEM characteristics of multilayer elliptical and cylindrical coplanar waveguides,” Microwave Opt. Technol. Lett., vol. 42, no. 4, pp. 317–322, Aug. 2004. [4] C. J. Reddy and M. D. Deshpende, “Analysis of coupled cylindrical striplines filled with multilayered dielectrics,” IEEE Trans. Microw. Theory Tech., vol. 36, no. 9, pp. 1301–1310, Sep. 1988. [5] A. Görür, M. Duyar, and C. Karpuz, “Analytic formulas for calculating the quasi-static parameters of a multilayer cylindrical coplanar strip line,” Microwave Opt. Technol. Lett., vol. 22, no. 6, pp. 432–436, Sep. 1999. [6] L. R. Zeng and Y. Wang, “Accurate solutions of elliptical and cylindrical striplines and microstriplines,” IEEE Trans. Microw. Theory Tech., vol. MTT-34, no. 2, pp. 259–264, Feb. 1986. [7] J. B. Knorr and K. D. Kuchler, “Analysis of coupled slots and coplanar strips on dielectric substrate,” IEEE Trans. Microw. Theory Tech., vol. MTT-23, pp. 541–547, Jul. 1975. [8] S. S. Bedair and I. F. Wolff, “Fast, accurate and simple approximate analytic formulas for calculating the parameters of supported coplanar waveguides for (M)MIC’s,” IEEE Trans. Microw. Theory Tech., vol. 40, no. 1, pp. 41–48, Jan. 1992.

[9] E. Chen and S. Y. Chou, “Characteristics of coplanar transmission lines on multilayer substrates: Modeling and experiments,” IEEE Trans. Microw. Theory Tech., vol. 45, no. 6, pp. 939–945, Jun. 1997. [10] V. Fouad Hanna and D. Thebault, “Analysis of asymmetrical coplanar waveguides,” Int. J. Electron., vol. 50, no. 3, pp. 221–224, 1981. [11] , “Theoretical and experimental investigation of asymmetric coplanar waveguides,” IEEE Trans. Microw. Theory Tech., vol. MTT-32, no. 12, pp. 1649–1651, Dec. 1984.

Volkan Akan (S’05) was born in Konya, Turkey, in 1979. He received the B.S. and M.S. degrees in electronics engineering from Hacettepe University, Ankara, Turkey in 2001 and 2004, respectively, and is currently working toward the Ph.D. degree at Hacettepe University. Since 2001, he has been a Research Assistant with the Department of Electrical and Electronics Engineering, Hacettepe University. His main research areas are microwave planar and nonplanar circuits, CAD-oriented (M)MIC circuit design, and analytical and numerical electromagnetic analysis.

Erdem Yazgan (M’91) received the B.S. and M.S. degrees from the Middle East Technical University, Ankara, Turkey, in 1971 and 1973, respectively, and the Ph.D. degree from Hacettepe University, Ankara, Turkey in 1980, all in electrical engineering. Since 1990, she has been a Professor with the Department of Electrical Engineering, Hacettepe University. In 1989, she was a Visiting Professor with Essex University, Essex, U.K. In 1994, she was with the Electroscience Laboratory, Ohio State University, Columbus. Her research interests include high-frequency propagation, low-altitude radar systems, mobile communications, MICs, reflector and microstrip antennas, Gaussian beam solutions, conformal mapping, and medical electronics.

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Mutually Exclusive Data Encoding for Realization of a Full Duplexing Self-Steering Wireless Link Using a Retrodirective Array Transceiver Kevin M. K. H. Leong, Member, IEEE, and Tatsuo Itoh, Fellow, IEEE

Abstract—A two-way communication link is established between an interrogator and retrodirective antenna array. The link is realized by using standard time-domain modulation for data uplink to the retrodirective array, while outgoing data is coded onto the antenna polarization sense of the signal sent back to the interrogator. Using these noninterfering mutually exclusive modulations schemes, this system architecture is able to resolve the difficulty of dealing with the corruption of the outgoing retrodirected signal by the incoming interrogation signal inherent when using phase-conjugation mixers (PCMs). This paper will discuss the development of a novel PCM also used for polarization modulation and a polarization state detector for demodulation. Analysis of circuit design tolerances and affects of jammers on proper polarization detection is also shown. Demonstration of simultaneous data transfer is presented in simulation and measurement along with the array’s automatic beam-steering ability. Index Terms—Antenna array, beam steering, mixers, mobile communication, phase conjugation, polarization, retrodirective array, transceiver.

I. INTRODUCTION

R

ETRODIRECTIVE antenna arrays [1]–[14] are a unique type of antenna array that is able to automatically detect a received signal’s direction of incidence and respond by retransmitting a signal in response back to the source of signal origination without prior knowledge of the incoming angle. This is done without relying on sophisticated digital signal-processing algorithms. Instead, the array performs this task using either a Van Atta array arrangement [1] or by using phase-conjugation techniques. Being that the retrodirective array is able to perform these target tracking and response functions using purely analog circuits, they have been discussed in the context of implementing high-speed adaptive communication links for mobile users. Thus far, retrodirective arrays have been investigated for use as data transponders [2]. This scenario makes use of the array’s natural ability to respond to an interrogating signal. Stored or collected data can be transmitted from the retrodirective array to a mobile interrogator. A retrodirective system capable of providing semiduplex communication was discussed in [3]. The array was implemented using reconfigurable multifunctional Manuscript received October 19, 2004; revised June 28, 2005. This work was supported by the U.S. Army Research Laboratory and by the U.S. Army Research Office under Grant DAAD19-01-1-0496. The authors are with the Department of Electrical Engineering, University of California at Los Angeles, Los Angeles, CA 90095-1594 USA (e-mail: [email protected]) Digital Object Identifier 10.1109/TMTT.2005.856081

mixers that could be switched from a direct-down-converting mixer for receiving to a phase-conjugation mixer (PCM) for data retransmission. In this method, data cannot be transmitted and received at the same time. Full-duplex communication capability requires that data be received and retransmitted simultaneously. However, since in retrodirective arrays the returned signal is merely a phase-reversed version of the incoming wave, any data transmitted by the interrogator would also be retrodirected. Under these circumstances, full-duplex communication is not straightforward to accomplish. Reference [4] proposes the use of three frequency bands to implement full-duplex operation. A modulated signal along with a single-tone pilot signal is transmitted by the interrogator using two separate frequencies. The retrodirective array is able to receive and demodulate the transmitted data using a conventional receiver, while the received pilot tone is phase conjugated and re-modulated and sent back to the interrogator. In this way, intermixing of received and transmitted data is avoided. In [5], another method of full-duplex communication using a retrodirective array is introduced. In this proposed communication link, the interrogator transmits a binary phase-shift keying (BPSK) modulated signal. The carrier is recovered by the retrodirective array by digitally squaring the BPSK modulated signal at each element using an edge detector. This recovered carrier is re-modulated with a new output message signal, amplified and retransmitted retrodirectively. A similar analog circuit approach was demonstrated in [6]. The carrier signal used for re-modulation and retrodirective transmission was extracted using a diode-limiting circuit. In this paper, a novel array and system architecture for retrodirective full-duplex communication is introduced and demonstrated. The basis of the proposed communication scheme is to use two noninterfering mutually exclusive modulation forms for uplink and downlink to and from the retrodirective array. Specifically, uplink data is modulated using conventional time-domain modulation (e.g., amplitude shift keying (ASK), BPSK), whereas the downlink data sent by the retrodirective array is coded on to the antenna polarization state [14], [15] (e.g., right-handed (RH) circular polarization and left-handed (LH) circular polarization) of the array. The uplink data is received at the retrodirective array using conventional down-conversion mixers, while a polarization state detector can be used to decode the independent message contained in the polarization of the retrodirected signal. Polarization modulation along with the influences of phase conjugation on circular and elliptical polarization is discussed in Section II.

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Section III details the essential link components, including a novel phase-conjugation/polarization modulating mixer and a polarization state detector. Analysis of design tolerances and effects of jammers on proper polarization detection is shown. A system link simulation is detailed in Section IV, while Section V presents experimental verification of the full-duplex link and antenna pattern measurements of the retrodirective array. II. PROPOSED METHOD Full-duplex communication requires that information be sent to and from each party simultaneously. A unique difficulty in implementing this communication scenario exists for retrodirective arrays. This comes from fact that, in most retrodirective array architectures that use PCMs, whatever signal is transmitted to the array is simply retransmitted back to the interrogator. Under these circumstances, one-way communication is possible by interrogating the array with an RF tone signal, and letting the retrodirective array respond with data modulated onto the return signal. However, if the interrogator transmits data to the retrodirective array, difficulties arise when this received signal is to be modulated again with data stored at the retrodirective array. In this case, some type of carrier recovery is needed. As an option for implementing a full-duplex link, the use of mutually exclusive modulation schemes, namely, time domain and antenna polarization modulation, is proposed. Information carried in the time domain does not depend on the polarization of the carrier wave and, likewise, carrier-wave polarization is not influenced by the data it carries in the time domain. Time-domain modulation techniques such as ASK, BPSK, and quadrature phase-shift keying (QPSK) are commonplace [16]. However, antenna polarization is more rarely used. The basic idea for polarization modulation is to encode data as different antenna polarization states. The most general form of polarization is elliptical polarization, where linear and circular polarizations are special cases [17]. Plane-wave polarization is defined by three main factors, which are : 1) axial ratio (AR); 2) rotational sense; and 3) tilt angle ( ). These factors can be determined by knowing the relative power levels and phase differences of the electric field in two orthogonal axes in the transverse plane (e.g., - and -directed for wave traveling in the -direction). The following equations can be used to calculate and are the amplitude of the polarization metrics, where the wave in the - and -direction, respectively, is the relative phase angle between the - and -directed electric-field vectors: (1) (2) (3) (4) Polarization rotational sense is given by the sign of AR; RH polarization is given by a negative AR and a positive value of AR defines LH polarization. Perfect circular polarization is given by and linear polarization is given by .

Fig. 1. Comparisons of plots of AR with various gamma ( ) values as a function of delta ( ) with and without effects adding 180 to delta (modulation).

Fig. 2. Plots of tau ( ) with various gamma ( ) values as a function of delta ( ).

One way of encoding data onto the field polarization is using the rotational sense of the wave as the variable in the data logic states. For example, transmitting an RH polarized wave can represent a digital “1” and transmitting an LH polarized wave can be made to represent a digital “0.” By examining (3) and (4), switching between polarization states can be accomplished by adding or not adding a 180 phase shift to . This can be illustrated by plotting out the AR for all values of (Fig. 1). Furthermore, the effect on can be seen by comparing plots of as a function of , shown in Fig. 2, and resulting from adding a 180 phase shift, shown in Fig. 3. The effect on polarization by applying phase conjugation was also investigated. The phase-conjugation effect is calculated by . It is observed that, under this consimply replacing with dition, the rotational sense of the wave is reversed while maintaining the same absolute value of AR. This is due to the odd symmetry of (3). Due to the even symmetry with respect to of (2), phase conjugation has no influence on tilt angle. Using a combination of the time domain and the discussed form of polarization modulation schemes, a full-duplex communication link can be achieved. The entire communication system

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Fig. 5. Schematic of PCM/phase-conjugation polarization modulator along with additional 90 hybrid used for testing. Fig. 3. Plots of tau ( ) with various gamma ( ) values as a function of delta ( ) with added 180 phase delta shift (modulation).

Fig. 4.

Essential components of proposed communication link.

is comprised of four essential components, as shown in Fig. 4. Data is sent from the interrogator to the retrodirective array using a conventional mixer via a circularly polarized antenna. This signal is received and demodulated by the receiver circuit module of the retrodirective transceiver array. At the same time, the modulated signal is received as an input to the PCMs. The mixer’s phase conjugate the received signal, achieving retrodirective retransmission. Independent modulation of the return signal is accomplished by actively switching the polarization of the return signal. The polarization sense is distinguished by a detector, completing the full-duplex link. Note that no attempt to remove the received data was made. Therefore, the time-domain data is retrodirected back to its source of origination. III. SYSTEM LINK COMPONENTS The proposed communication link system requires the development of two main components. The polarization modulation function is accomplished by developing a novel mixer, which performs as a PCM as well as a polarization modulator. A polarization state detector is also realized for demodulation of data sent from the retrodirective array via polarization modulation. These two circuit elements will be detailed below. A. PCM/Phase-Conjugation Polarization Modulator In any retrodirective array, the core component is the PCM. This specialized mixer is used to “reverse” the incoming wavefront received be the array; retransmitting a beam back to the source of the wave. In this study, the PCM was modified to

also serve as a polarization modulator. The schematic of the mixer is shown in Fig. 5. The mixer consists of a pair of antiparallel Schottky diodes pumped in-phase using a local oscillator (LO) with frequency nearly twice the incoming RF signal. After mixing, the resulting phase conjugated IF signal is outputted at the same ports as the incoming RF. The two branches of the mixer are directly connected to either port of an orthogonally polarized square patch antenna for circular or elliptical polarization reception and transmission. Note that the phase-conjugation behavior of the PCM serves two functions. First, as in a conventional retrodirective array, the array of PCMs reverse the wavefront phase of the incoming wave for retrodirective transmission. More specifically, for the PCM proposed in this paper, it phase conjugates the relative phase received at each arm of each of the dual-polarized antennas in the array. This more local effect reverses the polarization sense, as discussed in Section II. Polarization of the return signal can be controlled by introducing an additional function into the PCM. As discussed in Section II, polarization sense modulation can be accomplished by controlling the relative phase between the IF signals outputted by the two branches of the PCM at each antenna element. Specifically, polarization sense reversal can be accomplished by adding a 180 relative phase shift to the outgoing IF signal. The addition or absence of this extra phase shift allows the polarization of the outgoing wave to be controlled. It should be noted that the retrodirective behavior of the array is not affected at all by the modulation state. This method of polarization modulation can be done in a number of ways. One example is to include a switchable 180 /0 delay line at the LO feeding point. This method would require more circuit board area to implement a delay line. Therefore, polarization modulation is instead accomplished by biasing the antiparallel diode using the symmetric nature of their – curves. By applying positive and negative voltages or ), the direction of the at the data ports of the mixer ( dc current can be controlled and, thus, controlling the relative phase shifts of the IF. For example, if positive bias is applied to the diodes, the top diode of the antiparallel pair will be forward biased, while the bottom diode will be turned off. By applying a negative voltage, the opposite will be true: top diode off and , forward current of each mixer bottom diode on. If branch will have the same direction, leading to 0 phase shift , forward added to the phase-conjugated phase. If

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Fig. 7. Schematic of PSD.

Fig. 6.

Measured mixer conversion loss and isolation.

current of each mixer branch will be antidirectional, leading to an added 180 phase shift to the phase-conjugated phase, reversing the polarization sense. The novel PCM/phase-conjugation polarization modulator was measured using a 90 hybrid coupler to simulate the different signal inputs the mixer would experience while receiving and transmitting circular-polarized (CP) signals. The measureof the 90 ment setup is shown in Fig. 5. When fed from hybrid, the phase relationship for LH CP is realized. Under , the mixer pair will output a signal the condition, 0 , 90 ) and mimicking an LH CP return signal (e.g., the output power will be combined at . The power outputted represents the leakage to the opposite polarization sense, at influencing the polarization purity. If the control voltages were , the output power will combine at , set to representing RH CP retransmission. Conversion loss of better than 9.5 dB is observed over a bandwidth of 150 MHz with 0.2 V. Polarization isolation better than a bias voltage of 18 dB is measured for both RH CP and LH CP inputs. These results are shown in Fig. 6.

Given the parameters of the incoming wave, as well the inherent circuit imbalances present in the PSD, the expected relative power ratio (RPR) can be calculated using simple network relations [18]. Given the parameters of the incoming wave, such as the relative phase shift ( ) between incoming orthogonal field vectors and amplitudes ( , ) of the received wave, as well as the inherent phase ( ) and amplitude ( ) imbalances present in the PSD, the expected RPR can be determined. Referring to Fig. 7, the power at points and can be written as (5) (6) Using superposition, the power at point due to power input at as follows: point is then given by (7) Using the same approach, the power at point due to input and the power at point due power at point is given by and, finally, represents the to input at point is given by power at point due to input at point as follows: (8)

B. Polarization Sense Detector (PSD) The component that is used to demodulate the data encoded onto the polarization sense of the signal returned by the retrodirective array is the PSD. The PSD is able to distinguish if the incoming wave is RH polarized or LH polarized by using the relative phase relationship between the wave’s orthogonal field components. The basic diagram of the PSD is shown in Fig. 7. The inputs to the circuit are connected to an orthogonally polarized square patch antenna. The front-end 90 coupler is used to constructively or destructively sum the powers at one of its two output ports depending on the polarization sense of the signal. By comparing the amplitudes of the received power of the two branches, the polarization sense can be determined. Notice that although both circuit branches receive the same time-domain waveform, the relative amplitude received by each branch reveals the transmitted polarization sense and, thus, the message sent by the retrodirective array. This type of detector is sensitive to the polarization sense of the incoming wave. Therefore, the wave need not necessarily be perfectly RH CP or LH CP; elliptical polarization is also detectable.

(9) (10) The total power at point

and point

is given by (11) (12)

respectively. The difference between power levels detected at point and point or RPR is given by (13) Fig. 8 shows the values of RPR as a function of delta for various values of gamma [refer to (1)], representing different polarization cases. Note that for values of delta from 180 to 0 , RPR is less than 0 dB, meaning that more power will appear at node and delta values greater than 0 power is delivered

LEONG AND ITOH: MUTUALLY EXCLUSIVE DATA ENCODING FOR REALIZATION OF FULL DUPLEXING SELF-STEERING WIRELESS LINK

Fig. 8. Plot of RPR with various gamma values assuming ideal circuit components.

Fig. 9. Plot of RPR with various gamma values assuming nonideal circuit components (gb = 1 dB, pb = 5 ).

to node . This suggests that comparison of relative power difwill determine the poferences between node and node larization sense of the incoming wave. The wave need not be exclusively perfectly CP for correct state detection. However, also notice that the closer to perfect circular polarization, the higher the RPR, suggesting less susceptibility to detection noise. Fig. 9 shows the plot of RPR under the condition that there is dB) and phase ( ) imbalance present in gain ( the PSD. Note the decrease in absolute value of RPR. However, these reductions are tolerable and can be eliminated by using a more ideal component. After passing through the hybrid coupler, the RF signals are amplified and downconverted to an IF frequency. Low-pass filters are used to reject the high-power signal radiated by the transmitter circuit. Since the demodulator operates on the principle of comparing received power levels, gain imbalance and dc offsets become significant. Variable gain op-amp circuits are used to compensate for the channel differences. The dc offset problem becomes considerable when the received power levels are small. In order to improve the accuracy of this technique, a high-precision A/D converter may be used to sample the re-

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Fig. 10. Plotted minimum jammer attenuation level needed to prevent interference. The desired signal has polarization state given by various values of delta and gamma (all cases are LH polarized).

ceived signal. The next step in the demodulating process is to recover the envelope of the signal. This is done using a full-wave rectifier. Finally, the signals from each branch are inputted into a voltage comparator circuit. The function of this circuit is to determine which branch is receiving more power and, thus, detecting the polarization state of the received signal. A general study of how the polarization detection scheme operates in the presence of near-frequency jammer signals was done assuming that both desired and undesired signals were simple sinusoids. Superposition and summation is used to examine the effect of the jammer on the waveform at each branch of the PSD. Note that both amplitude and relative phase is necessary to determine the correct summation waveform. In general, the addition of the jammer results in an AM wave with a near sinusoidal envelope. The frequency of the envelope can be estimated by the frequency difference of the two received signals, while the envelope magnitude and phase can be determined by magnitude and phase of the received signals and gain and phase imbalances of the PSD. Under the condition that LH polarization is the “correct state,” the maximum jammer attenuation level needed for correct detection while exposing the receiver to all possible jammer polarization senses and tilt angles was numerically calculated. Fig. 10 shows the minimum jammer attenuation level needed to ensure correct detection for specific polarization conditions of the desired signal (all cases are LH polarization). From this study, it is observed that for a perfectly CP , gamma ), only 1.6-dB jammer supwave (delta pression is needed for correct state detection. As the desired signal’s polarization becomes closer to linear polarization, the necessary attenuation level increases. It is interesting to note that even a jammer having the same polarization of the desired signal can cause an error in detection. This occurs because the envelope of the signal at both branches are not in-phase so that even if the peak amplitude is much stronger in one branch of the PSD, at some instances in time, the level at this branch may be going through a minimum, while the signal level at the alternate branch goes through a maximum.

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Fig. 11.

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 53, NO. 12, DECEMBER 2005

Simulation schematic of proposed communication link.

IV. SIMULATIONS OF COMMUNICATION LINK A full end-to-end simulation of the proposed system was done using Agilent ADS’s envelope simulation function. A schematic drawing of the simulation setup is shown in Fig. 11. As represented in this figure, the signal flow is from left to right. The circularly polarized transmitter is implemented by using a pre-defined mixer block to upconvert digital baseband data onto a carrier of 5.7 GHz. The modulated RF is then fed into an ideal 90 coupler, which is used to simulate the conditions of a circularly polarized wave. Since the PCMs have a shared input/output port, ideal circulators are used to route the input RF signal into the mixers and re-route the outgoing IF to the polarization detector circuit. Polarization sense modulation is accomplished using the PCM/phase-conjugation polarization modulator. This circuit block uses the “real” mixer circuit, including diode model and microstrip-line characteristics. The and modulation inputs LO was set to 11.6 GHz, while were varied to simulate different modulation conditions. Polarization detection was done by first passing the mixer outputs through an ideal 90 coupler. The envelope of the signals from each of the coupler outputs were extracted using an envelope detector. The magnitude of the two channels was compared in simulation using a “symbolically defined device,” which takes the place of a comparator circuit. In order to verify the polarization modulation concept, the transmitted input signal was modulated with a 10-MHz digital signal with a simple bit sequence of “10” and the communication link was tested using various input scenarios. Fig. 12 shows the envelope of the signals at point and point , respectively, under the condition V. Notice that, for this static case, the voltage level at point is larger than the level at point B, as expected. The 10-MHz envelope of the original transmitted signal can be observed. For comparison, Fig. 13 shows the voltage levels at point and under the conditions, V, V. It can be clearly seen that the polarization state was reversed; the voltage is now larger than the voltage level at point . at point dynamic polarization modulation demonstration was done by at a constant ( V) and driving with setting a 2-MHz digital “10” sequence with a high level of 0.4 V and and low level of 0.4 V. The voltage levels at points are plotted in Fig. 14 for this case. Dynamic polarization switching can be clearly seen. Notice also that the 10-MHz

Fig. 12.

Simulation of polarization demodulation V 1 = V 2 =

Fig. 13.

Simulation of polarization demodulation

0

V

1 = 0:4 V.

V

1 =

00 4 V. :

00 4 V, :

V

2 =

Fig. 14. Demonstration of active polarization modulation by comparing voltage at points A and B with V 1 = 0:4 V and square wave driving V 2 from 0.4 V to 0.4 V at 2 MHz.

0

0

message signal is also present in the waveform. The tracking of the polarization state is observed in Fig. 15, which shows the output of the comparator circuit. Even though the envelope of the original signal is still present, it has no influence on the message carried in polarization.

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Fig. 15. Voltage level at output of comparator, demonstrating polarization detection.

Fig. 17.

Fig. 16.

Schematic of single retrodirective array element.

V. RETRODIRECTIVE TRANSCEIVER ARRAY FABRICATION AND MEASUREMENT Fig. 16 shows the schematic diagram of a single element of the retrodirective transceiver array. Directional couplers are used to route the power received by a set of orthogonally polarized antennas to the PCM, as well as to the receiver section of the circuit. The return IF generated from the PCM passes back out to the antenna and is isolated from the receiver path. Identical control modulation voltages and LO power is fed to each element of the array. A 90 hybrid coupler is placed at the front end of the receiver portion of the element and is followed by a standard receiver chain. In this way, both RH CP and LH CP signals can be received, depending on the choice of output port or ). ( A prototype four-element retrodirective transceiver array and was fabricated on an RT/Duroid substrate with 0.635-mm thickness, and is shown in Fig. 17. Dual-fed square patch antennas for orthogonal polarization reception were integrated with the PCMs and arranged with an array spacing of at 5.82 GHz. LO power was distributed to the PCMs using a Wilkinson power divider ladder. 90 branch-line couplers were also etched onto the array circuit board. An external power combiner and receiver chain was used to complete the receiver module of the array.

Fabricated four-element retrodirective array.

The fabricated polarization state detector uses a dual-fed patch antenna along with a microstrip 90 branch-line coupler. The remainder components include an RF gain block and commercially available mixer for down conversion to baseband. Low-pass filters were used to pass the signal of interest and reject any unwanted signal, especially the signal that is transmitted by the interrogator. Operational amplifier circuits were designed to provide gain and dc offset control. A full-wave rectifier circuit was implemented for envelope detection using a diode bridge circuit. Finally, an off-the-shelf comparator circuit was used as the decision circuit in the PSD. A. Full-Duplex Measurement A data link using the transceiver array was measured by placing the transmitter and polarization state detector at the broadside angle of the array. Modulated data was transmitted to the retrodirective array using a frequency mixer and gain amplifier along with a circularly polarized horn antenna at 5.84 GHz. The 500-kb/s binary sequence received and down converted by the receiver module of the array is shown in Fig. 18. Excellent correlation with the original waveform is observed. Simultaneously, the PCMs were driven with an LO at 11.66 GHz generating and IF of 5.82 GHz. The PCMs were also modulated with digital waveform with a 500-kb/s data rate. The required voltages necessary for polarization modulation were generated by keeping one branch of each mixer pair at a constant bias point of 0.2 V and varying the bias of the other branch from 0.2 to 0.2 V using an arbitrary waveform generator. The PSD was used to demodulate the message signal transmitted by the retrodirective array. The comparator circuit in the detector operates in such a way that when an RH CP is detected, a digital “1” is outputted and a digital “0” represents LH CP.

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Fig. 18. Received data by the retrodirective array receiver module compared with original time-domain data.

Fig. 21.

Bistatic RCS with source at broadside (0 ), V 1 =

V 2.

B. Radiation Patterns

Fig. 19. Comparison of received and transmitted data using polarization modulation with interrogator sending RH CP signal.

Fig. 20. Comparison of received and transmitted data using polarization modulation with interrogator sending LH CP signal.

Fig. 19 shows the demodulated waveform generated by the PSD, while the array was being interrogated with an RH CP wave. In contrast, when interrogated with an LH CP wave, the array response is shown in Fig. 20. By comparing the two figures, the dependence on the interrogator’s polarization of the returned signal’s polarization is observed. Since the PSD was configured the same for both measurements, the received signals appear inverse of each other (i.e., “0” “1” and “1” “0”).

The antenna array’s retrodirective behavior and polarization sensitivity was characterized using two basic methods, bistatic and monostatic radar cross-sectional (RCS) measurements. Bistatic measurements were done by illuminating the array with a single-tone interrogation signal at fixed angles and measuring the array’s response in the array plane of the retrodirective array using a separate receiver antenna. In contrast, for monostatic RCS measurements, the transmitting and receiving antennas are collocated. In both measurements, a single tone at 5.8 GHz was transmitted to the array and the LO frequency was set at 11.61 GHz, generating a return IF at 5.81 GHz. The polarization sensitivity of the array, as well as its ability to control the polarization of the return signal, was confirmed in both bistatic and monostatic RCS pattern measurement. Fig. 21 shows the bistatic RCS pattern measurements with the interrogator located at broadside (0 ) and control voltage set at . In this scenario, it is expected that the retrodirected signal has the same polarization as the interrogator signal due to phase conjugation, as explained in Section II. This behavior was verified by doing two sets of measurements with two different pairs of transmit and receive circularly polarized antennas. First, an RH CP antenna was used as the interrogator, while an LH CP antenna was used to receive the retrodirected signal at various angles (case 1). Next, an LH CP antenna was used as the interrogator, while another LH CP antenna was used as a receiver antenna (case 2). Notice that, as expected, for this input voltage pair, more power is received in case 2. Fig. 22 shows the results of bistatic RCS measurements with a source at broadside, this time . This voltage set reverses with a control voltage set at the polarization sense of the incoming wave. The presented measurements confirm this behavior. The measured results in Figs. 21 and 22 shows a broadside directed main beam when the return is co-polarized with the LH CP receiver and resemble an unshaped pattern when the return is cross-polarized with the receiver. Notice also that the pattern pairs are quite similar. At the main beam angle, the difference between the received RH CP and LH CP is approximately 10 dB. This is consistent with the pattern of a single patch element. Since this experiment could not be done in

LEONG AND ITOH: MUTUALLY EXCLUSIVE DATA ENCODING FOR REALIZATION OF FULL DUPLEXING SELF-STEERING WIRELESS LINK

Fig. 22.

Bistatic RCS with source at broadside (0 ), V 1 =

0V 2 .

Fig. 24. Bistatic RCS with source at

Fig. 25. Fig. 23.

Bistatic RCS with source at

030

,V1 =

V 2.

an anechoic chamber, scattering due to the measurement environment noticeably influences the measured radiation patterns. This is thought to be the reason for ripples present in the radiation patterns along with circuit element imbalances. Figs. 23 and 24 show the same type of measurements with the interrogator now positioned at 30 . Due to the effect of the element pattern, the main beam is slightly squinted to broadside to an angle of 27 in both figures. The same polarization control shown in the broadside measurements are observed. At the main beam angle, there is approximately a 10-dB difference between the received RH CP and LH CP. Furthermore, note that no grating lobes are observed due to the small array spacing. The bistatic RCS pattern can be calculated by taking into account the antenna array directivity along with pattern of a single antenna element. It should be noted that when the array receives the signal from the interrogator, the power is received and processed by each antenna element independently and the re-radiated return signal is focused according to the array dimensions. Furthermore, due to the polarization sensitivity of the array, the signal is actually received independently by each branch of each patch antenna as a linearly polarized wave, then upon re-radiation of the signal, the combination of the two

030

,V1 =

3695

0V 2 .

Monostatic RCS.

branches of each element form a circularly polarized wave. is the The bistatic RCS is given in (14) as follows, where array directivity, is the linearly polarized antenna element directivity, is the circularly polarized antenna element directivity, and is the circuit gain [2], [17]: (14) These calculations are based on measured element patterns and theoretical array directivity. The cross-polarized bistatic RCS is calculated by using the circularly polarized cross-pol pattern element directivity ( ). The calculated results for different interrogator positions and polarizations tested are shown in Figs. 21–24. The monostatic RCS pattern can be calculated in a similar manner with the bistatic RCS [2], [17]. It is given in (15) as follows: (15) Since the array is continuously moving its radiation peak to follow the interrogator, the pattern exhibits no nulls. Fig. 25 shows the measured and calculated monostatic RCS of the array,

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showing a 0- to 5-dB RCS for angles up to 50 . The array’s ability to control the re-radiated polarization was also determined in this measurement. Comparing the co-pol and cross-pol monostatic RCS shows the angular range for which the proposed polarization modulation method is possible. At larger angles away from broadside, the power difference between the two polarization reduces, lessening the distinguishability between the polarization states. The angular range can be maximized by using an antenna element with low cross-pol levels even at wide angles. VI. CONCLUSION A retrodirective full-duplexing link has been demonstrated by using time-domain data modulation for uplink from the interrogator and polarization sense modulation for downlink transmission from retrodirective array to interrogator at data rates up to 500 kb/s. This use of mutually exclusive data modulation techniques allows for the circumvention of the issue of data corruption present when using a simple PCM. Core components needed in this link, including a PCM/phase-conjugation polarization modulator and PSD has been discussed. Application of this technology is most suited for line-of-sight situations since polarization modulation is influenced in a multiple scattering environment. However, most retrodirective array applications are also best in line-of-sight situations as well. Addition of polarization parity checking algorithms may be used to increase the data throughput and reliability. REFERENCES [1] L. G. Van Atta, “Electromagnetic reflector,” U.S. Patent 2 908 002, Oct. 6, 1959. [2] R. Y. Miyamoto, Y. Qian, and T. Itoh, “An active integrated retrodirective transponder for remote information retrieval-on-demand,” IEEE Trans. Microw. Theory Tech., vol. 49, no. 9, pp. 1658–1662, Sep. 2001. [3] , “A reconfigurable active retrodirective/direct conversion receiver array for wireless sensor systems,” in IEEE MTT-S Int. Microwave Symp. Dig., vol. 2, May 2001, pp. 119–1122. [4] P. V. Brennan, “An experimental and theoretical study of self-phased arrays in mobile satellite communications,” IEEE Trans. Antennas Propag., vol. 37, no. 11, pp. 1370–1376, Nov. 1989. [5] L. D. DiDomenico and G. M. Rebeiz, “Digital communications using self-phased arrays,” IEEE Trans. Microw. Theory Tech., vol. 49, no. 4, pp. 677–684, Apr. 2001. [6] K. M. K. H. Leong, Y. Wang, and T. Itoh, “A radar target transceiver using a full duplex capable retrodirective array system,” in IEEE MTT-S Int. Microwave Symp. Dig., vol. 2, Jun. 2003, pp. 1447–1450. [7] C. Y. Pon, “Retrodirective array using the heterodyne technique,” IEEE Trans. Antennas Propag., vol. AP-12, no. 3, pp. 176–180, Mar. 1964. [8] M. I. Skolnik and D. D. King, “Self-phasing array antennas,” IEEE Trans. Antennas Propag., vol. AP-12, no. 2, pp. 142–149, Feb. 1964. [9] C. W. Pobanz and T. Itoh, “A conformal retrodirective array for radar applications using a heterodyne phased scattering element,” in IEEE MTT-S Int. Microwave Symp. Dig., Jun. 1995, pp. 905–908. [10] Y. Chang, H. R. Fetterman, I. Newberg, and S. K. Panaretos, “Microwave phase conjugation using antenna arrays,” IEEE Trans. Microw. Theory Tech., no. 11, pp. 1910–1919, Nov. 1998. [11] S. L. Karode and V. Fusco, “Self-tracking duplex communication link using planar retrodirective antennas,” IEEE Trans. Antennas Propag., no. 6, pp. 993–1000, Jun. 1999. [12] D. M. K. Ah Yo, W. E. Forsyth, and W. A. Shiroma, “A 360 degrees retrodirective self-oscillating mixer array [using HEMTs],” in IEEE MTT-S Int. Microwave Symp. Dig., Jun. 2000, pp. 813–816. [13] T. Brabetz, V. F. Fusco, and S. Karode, “Balanced subharmonic mixers for retrodirective-array applications,” IEEE Trans. Microw. Theory Tech., vol. 49, no. 3, pp. 465–469, Mar. 2001.

[14] K. M. K. H. Leong and T. Itoh, “Full-duplex retrodirective array using mutually-exclusive uplink and downlink modulation schemes,” in IEEE MTT-S Int. Microwave Symp. Dig., vol. 3, Jun. 2004, pp. 1695–1698. [15] M. A. Kossel, R. Kung, H. Benedickter, and W. Bachtold, “An active tagging system using circular-polarization modulation,” IEEE Trans. Microw. Theory Tech., vol. 47, no. 12, pp. 2242–2248, Dec. 1999. [16] S. Haykin, Communication Systems. New York: Wiley, 2001. [17] C. A. Balanis, Antenna Theory: Analysis and Design. New York: Wiley, 1997. [18] D. M. Pozar, Microwave Engineering. New York: Wiley, 1998.

Kevin M. K. H. Leong (S’99–M’04) received the B.S. degree in electrical engineering from the University of Hawaii, Manoa, in 1999, and the M.S. and Ph.D. degrees in electrical engineering from the University of California at Los Angeles (UCLA), in 2001 and 2004, respectively. He is currently a Post-Doctoral Researcher with UCLA. His research interests include planar antennas, millimeter-wave circuits, and mobile communication systems. Dr. Leong was the recipient of the First-Place Best Student Paper Award at the 2001 European Microwave Conference.

Tatsuo Itoh (S’69–M’69–SM’74–F’82) received the Ph.D. degree in electrical engineering from the University of Illinois at Urbana-Champaign, in 1969. From September 1966 to April 1976, he was with the Electrical Engineering Department, University of Illinois at Urbana-Champaign. From April 1976 to August 1977, he was a Senior Research Engineer with the Radio Physics Laboratory, SRI International, Menlo Park, CA. From August 1977 to June 1978, he was an Associate Professor with the University of Kentucky, Lexington. In July 1978, he joined the faculty at The University of Texas at Austin, where he became a Professor of Electrical Engineering in 1981 and Director of the Electrical Engineering Research Laboratory in 1984. During the summer of 1979, he was a Guest Researcher with AEG-Telefunken, Ulm, Germany. In September 1983, he was selected to hold the Hayden Head Centennial Professorship of Engineering at The University of Texas at Austin. In September 1984, he was appointed Associate Chairman for Research and Planning of the Electrical and Computer Engineering Department, The University of Texas at Austin. In January 1991, he joined the University of California at Los Angeles (UCLA) as Professor of Electrical Engineering and Holder of the TRW Endowed Chair in Microwave and Millimeter Wave Electronics. He was an Honorary Visiting Professor with the Nanjing Institute of Technology, Nanjing, China, and with the Japan Defense Academy. In April 1994, he was appointed an Adjunct Research Officer with the Communications Research Laboratory, Ministry of Post and Telecommunication, Japan. He currently holds a Visiting Professorship with The University of Leeds, Leeds, U.K. He has authored or coauthored 350 journal publications, 650 refereed conference presentations, and has written 30 books/book chapters in the area of microwaves, millimeter waves, antennas, and numerical electromagnetics. He has generated 64 Ph.D. students. Dr. Itoh is a member of the Institute of Electronics and Communication Engineers of Japan, and Commissions B and D of USNC/URSI. He served as the editor of the IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES (1983–1985). He serves on the Administrative Committee of the IEEE Microwave Theory and Techniques Society (IEEE MTT-S). He was vice president of the IEEE MTT-S in 1989 and president in 1990. He was the editor-in-chief of IEEE MICROWAVE AND GUIDED WAVE LETTERS (1991–1994). He was elected an Honorary Life Member of the IEEE MTT-S in 1994. He was elected a member of the National Academy of Engineering in 2003. He was the chairman of the USNC/URSI Commission D (1988–1990) and chairman of Commission D of the International URSI (1993–1996). He is chair of the Long Range Planning Committee of the URSI. He serves on advisory boards and committees for numerous organizations. He has been the recipient of numerous awards including the 1998 Shida Award presented by the Japanese Ministry of Post and Telecommunications, the 1998 Japan Microwave Prize, the 2000 IEEE Third Millennium Medal, and the 2000 IEEE MTT-S Distinguished Educator Award.

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Ultra-Wideband Low-Cost Phased-Array Radars Christopher T. Rodenbeck, Member, IEEE, Sang-Gyu Kim, Student Member, IEEE, Wen-Hua Tu, Student Member, IEEE, Matthew R. Coutant, Seungpyo Hong, Student Member, IEEE, Mingyi Li, and Kai Chang, Fellow, IEEE

Abstract—Emerging radar applications require phased arrays that can operate over wide bandwidths to support multiband/multifunction operation. In response to that need, this paper presents a cost-effective implementation for extremely wide-band phased-array radars. Two designs are demonstrated, one operating from 3 to 12 GHz and the other operating from 8 to 20 GHz. These designs incorporate ultra-wideband antipodal tapered slot antennas, a novel cross-polarization suppressed array architecture, piezoelectric true-time-delay phase shifters, and broad-band high-power monolithic amplifiers. The resulting systems provide target detection and beam steering over the complete operating bandwidths. These results exceed the state-of-the-art for phased-array radars in terms of bandwidth and cost and should have direct applications in the development of ultra-wideband and multifunction radar systems. Index Terms—Antenna arrays, monolithic microwave integrated circuits (MMICs), phased arrays, ultra-wideband radar.

I. INTRODUCTION

R

APID progress in telecommunication and radar technology is placing increasing demands on wireless system performance and functionality. Personal communications systems, for example, can now accommodate multiple functions and protocols [e.g., global positioning system (GPS), Universal Mobile Telecommunications System (UMTS), personal communications system (PCS)] within a single RF front end. A parallel process of convergence is beginning to take place within the military arena. Modern naval and aerospace vehicles are required to provide a large and increasing number of sensing functions, leading to a proliferation in the number of onboard phased arrays and radar systems. To reduce the number and collective cost of these systems, current research programs seek to integrate multiple systems and functions operating in different frequency bands into a single ultra-wideband system [1]. Although broad-band phased-array systems are a longstanding topic of interest [2], progress has recently been made using ultra-wideband tapered slot antennas together with true-time delay phase shifters [3]–[5]. In [5], a low-cost fourManuscript received October 19, 2004; revised May 24, 2005. This work was supported in part by the U.S. Army Space Command under Contract DASG60-03-C-0082. C. T. Rodenbeck was with the Department of Electrical Engineering, Texas A&M University, College Station, TX 77843-3128 USA. He is now with Sandia National Laboratories, Albuquerque, NM 87185-0537 USA (e-mail: [email protected]). S.-G. Kim, W.-H. Tu, S. Hong, M. Li, and K. Chang are with the Department of Electrical Engineering, Texas A&M University, College Station, TX 778433128 USA. M. R. Coutant is with TriQuint Semiconductor, Richardson, TX 75080 USA and also with the Department of Electrical Engineering, Texas A&M University, College Station, TX 77843-3128 USA. Digital Object Identifier 10.1109/TMTT.2005.856668

channel 10–21-GHz phased-array transceiver was demonstrated for communications applications. In this paper, extremely wide-band phased-array antennas are demonstrated as an enabling technology for low-cost multiband and multifunction radar systems. Two designs are presented, one operating from 3 to 12 GHz and the other operating from 8 to 20 GHz. Both the 3–12- and 8–20-GHz systems cover frequencies commonly used for military radar and imaging. In addition, the 3–12-GHz system covers the frequencies designated by the U.S. Federal Communications Commission (FCC) for civilian ultra-wideband radar applications such as ground penetration, through-wall imaging, surveillance/security, and medical imaging [6]. Both designs use antipodal tapered-slot antennas (ATSAs) in a novel mirrored array architecture to achieve ultra-wideband performance and low cross-polarization. Beam steering is achieved across the complete operating bandwidths using low-cost piezoelectric transducer (PET) phase shifters. A broad-band monolithic microwave integrated circuit (MMIC) power amplifier (PA) integrated into these designs addresses the traditional challenge of broad-band microwave power generation. The PA operates over the 2–20-GHz decade with more than 29 dBm of output power and 15.8 dB of gain. System tests demonstrating pulse-radar target ranging and phased-array beam steering are performed with excellent results for each array design. This approach is scalable in size and function for application to a variety of systems and has extended bandwidth and reduced cost in comparison with the state-of-the-art. These results are accordingly expected to stimulate further advances in microwave front-end designs for multifunction RF phased arrays and radars. II. ANTENNAS, ARRAYS, AND PHASE SHIFTERS The design of the antenna elements, phase shifters, and antenna arrays are described here. The assembled subsystem is illustrated in the photograph shown in Fig. 1. The key components are a 1 4 -plane array of ATSAs and a multiline PET-based phase shifter. In addition, there is a 1 4 wide-band microstrip power divider. These components are inexpensive, can be easily operated over wide bandwidths, and can be scaled depending on the requirements of the design. The geometry of an ATSA is shown in Fig. 2. The antenna can achieve ultra-wideband performance due to its elegant transition from microstrip line [7], [8]. Two different ATSA element designs are used to cover the respective 8–20- and 3–12-GHz bands. Table I lists the design parameters for both models, identified as “Design A” and “Design B.” Different sizes for the elements are used in Designs A and B in order to provide a satisfactory tradeoff between scan range and element gain for each

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Fig. 3. Measured return loss of the ATSAs.

2 H

Fig. 1. Configuration of a 1 4 -plane array operated by a PET-controlled phase shifter. The 8–20-GHz design is shown.

TABLE II DIMENSIONS OF THE DESIGNED PHASE SHIFTER

Fig. 2. Schematic illustrating an ATSA. TABLE I DIMENSIONS OF THE DESIGNED ATSA

suspended above a series of microstrip lines. The perturber’s triangular shape creates a progressive phase shift that can be varied by using the PET to adjusting the height of the perturber above the microstrip lines. The progressive phase shift between any two neighboring microstrip lines can be calculated as (1)

of the respective bands. The substrates for both designs are selected to provide mechanical rigidity while keeping the effec[9], within to tive thickness, defined as across the operating bandwidth. The microstrip transition at the input is tapered circularly to parallel strips for the antenna feed. The antenna length is chosen according to the empirical guidelines of [9] while also considering the practical stability of the antenna. The flare is tapered exponentially with the opening rate determined using recursive optimization in CST Microwave Studio.1 Fig. 3 shows that the measured return losses of the designed ATSAs are better than 10 dB across the 8–20and 3–12-GHz bands. The measured gain of Design A is 4.9 dBi at 8 GHz, 6.8 dBi at 12 GHz, and 7.8 dBi at 20 GHz. The measured gain of Design B is 4.3 dBi at 3 GHz, 8.2 dBi at 7 GHz, and 13.8 dBi at 12 GHz. A PET-controlled multiline phase shifter is also shown in Fig. 1. A dielectric perturber is attached to a PET actuator and 1CST

Microwave Studio, ver. 4, Darmstadt, Germany, 2003.

where is the progressive length of the perturber above the and are the effective relamicrostrip lines and tive permittivities of the perturbed and unperturbed microstrip lines, respectively [10]. Equation (1) represents a true time delay (TTD) phase shift capable of linear operation over a wide range of frequencies. Two phase shifters are designed to operate over the 8–20and 3–12-GHz bands. Complete design parameters are given in Table II. The maximum progressive phase shifts of 100 at 20 GHz and 80 at 12 GHz are designed to achieve scan angles of 30 and 20 , respectively. Fig. 4 shows the measured -parameters and differential phase shifts for the 8–20-GHz design. The return loss is better than 10 dB for the operating frequency ranges, and the insertion loss is less than 4 dB at 20 GHz with full perturbation. This insertion loss includes the losses due to the connectors, power divider, and line lengths. Fig. 5 shows the measured differential phase shifts for the 3–12-GHz design. Two 1 4 -plane phased arrays of the type shown in Fig. 1 are designed using the ATSA elements and phase shifters discussed above. The arrays are fed using wide-band Chebyshev

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Fig. 6.

Fig. 4. Measured performance of the phase shifter operating from 8 to 20 GHz. (a) S -parameters. (b) Differential phase shift.

Mirrored ATSA array architecture.

array spacing in the 8–20-GHz design is 9.4 mm, corresponding at 8 GHz and at 20 GHz. The spacing in the to at 3 GHz and 3–12-GHz case is 18 mm, meaning at 12 GHz. The measured gain of Design A at broadside scan is 4.8 dBi at 8 GHz, 6.5 dBi at 12 GHz, and 8.9 dBi at 20 GHz. For Design B at broadside scan, the measured array gain is 9.2 dBi at 3 GHz, 12.4 dBi at 7 GHz, and 14.4 dBi at 12 GHz. An important feature of these arrays is that the ATSA elements are combined using a new “mirrored” array architecture [12]. Though able to provide very wide-band performance, ATSA elements and arrays traditionally suffer from high crosspolarization levels due to the placement of the tapered slot flares on different layer of the same substrate. This problem can be eliminated, however, by mirroring the elements in the array, as shown in Fig. 6. Doing so places a null in the cross-polarization pattern in the direction of the co-polarization maximum. This array approach, which is adopted in this paper, can also be applied to two-dimensional arrays. III. 2–20-GHz MMIC-BASED AMPLIFIERS

Fig. 5. Measured differential phase shift of the phase shifter operating from 3 to 12 GHz.

power dividers [11]. The array spacing is chosen to be sufficiently small to prevent grating lobe formation when the beam is steered in the -plane at the highest operating frequency. The

The goal of this research is to demonstrate low-cost ultra-wideband phased arrays for pulse radar applications. Such a demonstration would not be possible, however, without a broad-band PA for the transmitter and low-noise amplifier (LNA) for the receiver. To meet this need, a 2–20-GHz MMIC-based PA and LNA are assembled and incorporated within the system. Fig. 7 illustrates the method of integration. Each MMIC amplifier is solder mounted on a gold/nickel-plated copper–molybdenum carrier. The copper–molybdenum alloy matches the thermal coefficient of expansion for gallium arsenide and provides thermal dissipation for the MMIC; the gold has low resistance to minimize ohmic loss in the ground path; the nickel makes it possible to solder to the carrier. Alumina thin-film networks (TFNs) provide 50- lines at the input and output of the MMIC and dc-bias pads for the bias wires. External bias capacitors are placed between the bias pads and MMIC with care taken to minimize bond-wire lengths in order to ensure low-frequency stability. The capacitors and TFNs are mounted on the carrier using conductive epoxy, and 1-mil gold wire is used for all RF connections between the MMICs and the off-chip components.

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Fig. 7. Assembled PA module including the TGA2509 MMIC, the 50- lines at the input and output, and the bias networks. Fig. 8. Gain and output power of the PA MMIC versus frequency.

LNA MMICs covering the 2–20-GHz decade are currently available from multiple vendors. In this system, two Velocium ALH102C LNA chips are used to build up the receive LNA for the radar. Across the 2–20-GHz band, gain of each device varies from 8 to 12 dB, and noise figure varies from 2 to 4 dB. The power dissipation totals 220 mW for both chips. High-power MMIC amplifiers operating from 2 to 20 GHz have only recently begun to enter the commercial market. This system uses the TriQuint TGA2509 MMIC PA. Developed by a Texas A&M University student, this chip provides over 29 dBm of power with at least 15.8-dB gain across the 2–20-GHz band, performance that surpasses previously published results for amplifiers of this class [13]–[15]. The layout of the chip is included in Fig. 7. The chip size is 7.5 mm . The first stage of this distributed amplifier utilizes cascode cells to enhance the gain while providing an option for automatic gain control. The second stage utilizes common source cells to provide high output power over the entire band. Both stages employ capacitive division on each cell to extend the upper band edge to 20 GHz. Gate bias is provided through the gate termination resistor, while the drain bias is provided through an on-chip low-pass network. The drain voltage is 12 V, and total power consumption is 13.2 W. As illustrated in Fig. 8, small-signal gain exceeds 15.8 dB and saturation power exceeds 29 dBm over the entire 2–20-GHz band. IV. SYSTEM TEST For system-level measurements, the PET-controlled antenna arrays are incorporated with the MMIC-based amplifier modules and tested using the setup shown in Fig. 9. A pulse-modulated synthesized source feeds an HP 8349B wide-band driver amplifier that, in turn, feeds the 2–20-GHz PA module. The PA drives the PET-controlled phased array previously shown in Fig. 1. An identical PET-controlled array is used for the receiver so that both phased arrays can be controlled using the same bias voltage. Alternatively, a single antenna array can be used with the addition of a wide-band duplexer switch. The receiving phased array feeds the 2–20-GHz LNA module, which,

Fig. 9. System block diagram illustrating the test setup for the beam-steerable pulse radar.

in turn, feeds an HP8472B Schottky diode detector and an IF amplifier. The detected waveform is compared with a timing signal to determine target range. A movable 1 1 m aluminum reflector serves as the target. The transmitter projects a pulse-modulated signal onto the target, and the receiver picks up the return signal. The distance between the target and radar is varied, and the radar detects the variation in this distance by measuring the time delay between the received and timing signals. The results presented in this paper use an RF transmit signal that is pulse modulated at a pulse-repetition frequency (PRF) of 20 MHz and at a duty cycle of 10%. All measurements are taken inside a 30-ft indoor range. A plot illustrating a received pulse waveform and timing reference signal is shown in Fig. 10. The measurement shown in this figure uses a transmit signal at 5.8 GHz to range a target located 175 in away from the 3–12-GHz radar system. Ranging tests are conducted at 5.8 GHz for the 3–12-GHz array and at 14 GHz for the 8–20-GHz array. These frequencies are chosen because 5.8 GHz is a popular unlicensed frequency in the 3–12-GHz band and because 14 GHz is in the center of the 8–20-GHz band. As shown in Fig. 11, the measured and expected time delays agree very well. The average error is 0.14 at 5.8 GHz and 0.13 at 14 GHz, which corresponds to an average error of approximately 0.8 in at both frequencies.

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Fig. 10. Detected waveform of the received signal can be compared with the waveform of the timing signal in order to measure target range.

Fig. 11. Comparison between measured and ideal time delays for varying target ranges.

Target position can be determined by varying the beamsteering angle [16]. Figs. 12 and 13 show the steered beam patterns measured from 8 to 20 GHz and from 3 to 12 GHz, respectively. Although beam steering is shown in one direction only, the beam may also be steered in the opposite direction [5]. Each beam is steered approximately 20 away from broadside in the -plane. The maximum external bias voltage to the PET phase shifters is 55–60 V for both frequency ranges. Cross-polarization is less than 17 dB below the co-polarization. This excellent linear polarization is due to the mirrored array approach described in this paper, which reduces cross-polarization levels by more than 20 dB at 20 and 12 GHz. Imperfections in the measured patterns are due to multiple factors: the incomplete progression in the phase shifts, the phase mismatches in the connection points between the phase shifter and the antennas, the imperfect flatness of the individual antenna elements, the mutual coupling between antenna elements, and the reflection of incident energy from the circuit elements. The radiation patterns, nonetheless, can be

Fig. 12.

Measured beam-steering patterns at: (a) 8, (b) 16, and (c) 20 GHz.

further improved by an additional optimization process and by finer fabrication tolerances. Finally, effective isotropic radiated power (EIRP) varies approximately from 41 to 45 dBm over the 3–12-GHz range and from 35 to 38 dBm over the 8–20-GHz range.

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piezoelectric TTD phase shifters, and decade bandwidth 1-W monolithic PAs. Two designs have been presented, one operating from 3 to 12 GHz and the other from 8 to 20 GHz. The technology is inherently scalable to larger arrays and should have wide-ranging impact on ultra-wideband multifunction radar design. ACKNOWLEDGMENT The authors are grateful to T. Ashour, Raytheon Systems Company, Dallas, TX, for arranging the donation of a wire bonding machine to Texas A&M University, College Station, and for critical assistance in assembling the MMIC amplifiers. The authors also wish to thank A. Mitchell, U.S. Army, and R. S. Tahim, RST Scientific Research Inc., Anaheim, CA, for their useful suggestions and encouragement. In addition, the authors wish to thank Dr. J. Carroll, TriQuint Semiconductor, Richardson, TX, for donating the PAs used in this study. The authors also acknowledge Velocium Products, Redondo Beach, CA, for providing the LNAs for a nominal fee. REFERENCES

Fig. 13.

Measured beam steering patterns at: (a) 3, (b) 7, and (c) 12 GHz.

V. CONCLUSION Multiple radar systems can be consolidated within a single multifunction phased-array radar system. This paper has demonstrated an inexpensive approach for producing such a system using cross-polarization suppressed ATSA arrays,

[1] P. F. McManmon, E. A. Watson, and M. T. Eismann, “Suggestions for low cost multifunction sensing,” in Proc. IEEE Aerospace Conf., Aspen, CO, Mar. 1998, pp. 283–306. [2] G. J. Laughlin, E. V. Byron, and T. C. Cheston, “Very wide-band phasedarray antenna,” IEEE Trans. Antennas Propag., vol. AP-20, no. 6, pp. 699–704, Nov. 1972. [3] C. Hemmi, R. T. Dover, F. German, and A. Vespa, “Multifunction wideband array design,” IEEE Trans. Antennas Propag., vol. 47, no. 3, pp. 425–431, Mar. 1999. [4] K. Trott, B. Cummings, R. Cavener, M. Deluca, J. Biondi, and T. Sikina, “Wideband phased array radiator,” in Proc. IEEE Int. Phased Array Systems Technology Symp., Boston, MA, Oct. 2003, pp. 383–386. [5] T. Y. Yun, C. Wang, P. Zepeda, C. T. Rodenbeck, M. R. Coutant, M. Li, and K. Chang, “A 10- to 21-GHz, low-cost, multifrequency, and fullduplex phased-array antenna system,” IEEE Trans. Antennas Propag., vol. 50, no. 5, pp. 641–650, May 2002. [6] “Ultra-wideband transmission systems, first report and order,” FCC, Washington, DC, FCC 02-48, Apr. 2002. [7] E. Gazit, “Improved design of the Vivaldi antenna,” Proc. Inst. Elect. Eng., pt. H, vol. 135, pp. 89–92, Apr. 1988. [8] S. G. Kim and K. Chang, “Ultrawide-band transitions and new microwave components using double-sided parallel-strips lines,” IEEE Trans. Microw. Theory Tech., vol. 52, no. 9, pp. 2148–2152, Sep. 2004. [9] K. S. Yngvesson, “Endfire tapered slot antennas on dielectric substrates,” IEEE Trans. Antennas Propag., vol. AP-33, pp. 1392–1400, Dec. 1985. [10] T. Y. Yun and K. Chang, “Analysis and optimization of a phase shifter controlled by a piezoelectric transducer,” IEEE Trans. Microw. Theory Tech., vol. 50, no. 1, pp. 105–111, Jan. 2002. [11] D. M. Pozar, Microwave Engineering. New York: Addison-Wesley, 1990, pp. 313–318. [12] S. G. Kim and K. Chang, “A low cross-polarized antipodal Vivaldi antenna array for wide-band operation,” in Proc. IEEE Int. AP-S Symp., Monterey, CA, Jun. 2004, pp. 2269–2272. [13] K. I. Jeon, J. H. Lee, S. W. Paek, D. W. Kim, W. S. Lee, C. R. Lim, H. Cha, H. Choi, and K. W. Chung, “A 5 to 27 GHz MMIC power amplifier,” in IEEE MTT-S Int. Microwave Symp. Dig., Boston, MA, Jun. 2000, pp. 541–544. [14] J. J. Komiak, W. Kong, and K. Nichols, “High efficiency wide-band 6 to 18 GHz power amplifier MMIC,” in IEEE MTT-S Int. Microwave Symp. Dig., Seattle, WA, Jun. 2002, pp. 905–907. [15] C. Wang, C. T. Rodenbeck, M. R. Coutant, and K. Chang, “A novel broad-band T/R module for phased array applications in wireless communications,” in IEEE MTT-S Int. Microwave Symp. Dig., Seattle, WA, Jun. 2002, pp. 1325–1328. [16] A. Hirata, T. Morimoto, and Z. Kawasaki, “DOA estimation of ultrawideband EM waves with MUSIC and interferometry,” IEEE Antennas Wireless Propag. Lett., vol. 2, no. 1, pp. 1990–193, 2003.

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Christopher T. Rodenbeck (S’97–M’05) received the B.S. (summa cum laude), M.S., and Ph.D. degrees in electrical engineering from Texas A&M University, College Station, in 1999, 2001, and 2004, respectively. He is currently a Senior Member of the Technical Staff with the Weapons Electronics and Advanced RF Systems Department, Sandia National Laboratories, Albuquerque, NM, where he develops miniaturized microwave circuits and antennas for wireless applications. During the summers of 1998, 1999, and 2000, he was an Intern with the MMIC Design Group, TriQuint Semiconductor, Dallas, TX. He has authored or coauthored 30 papers in the microwave and millimeter-wave fields. Dr. Rodenbeck was the recipient of a Graduate Fellowship from the State of Texas “to advance the state of the art in telecommunications,” two NASA–TSGC Graduate Fellowships, a Texas A&M Graduate Merit Fellowship, and a TxTEC Graduate Scholarship.

Sang-Gyu Kim (S’02) received the B.S.E.E. and M.S.E.E. degrees from Kyungpook National University, Taegu, Korea, in 1991 and 1993, respectively, and the Ph.D. degree from Texas A&M University, College Station, in 2005. From 1993 to 2000, he was a Researcher with SK Telecom, Seoul, Korea. He was with the Central Research Center, Digital Cellular Business Division, and IMT-2000 Development Division, where he was involved in code-division multiple-access (CDMA) cellular system optimization under multipath fading channel and IMT-2000 cellular system development. From 2000 to 2005, he was a Graduate Research Assistant with the Electromagnetics and Microwave Laboratory, Texas A&M University. His research area includes passive and active microwave circuits, wide-band transceivers, and multiple-beam phased-array antenna systems.

Wen-Hua Tu (S’04) received the B.S. degree in communication engineering from National Chiao Tung University, Hsinchu, Taiwan, R.O.C., in 1999, the M.S. degree in communication engineering from National Taiwan University, Taipei, Taiwan, R.O.C., in 2001, and is currently working toward the Ph.D. degree in electrical engineering at Texas A&M University, College Station. Since 2003, he has been a Research Assistant with the Electromagnetics and Microwave Laboratory, Texas A&M University. His research interests include wide-band phased arrays and filters.

Matthew R. Coutant received the B.S. degree in electrical engineering from Oklahoma State University, Stillwater, in 1997, the M.A. degree from Texas A&M University, College Station, in 2000, and is currently working toward the Ph.D. degree at Texas A&M University. He is currently a MMIC Designer with TriQuint Semiconductor, Dallas, TX.

Seungpyo Hong (S’04) received the B.S. and M.S. degrees in electronic engineering from Yonsei University, Seoul, Korea, in 1991 and 1993, respectively, and is currently working toward the Ph.D. degree in electrical engineering at Texas A&M University, College Station. From 1993 to 2000, he was with LG Information and Communications Ltd., Seoul, Korea, where he developed base-station controllers for CDMA cellular systems. His research interests are passive and active microwave circuits and RF system integration.

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Mingyi Li received the B.S. and M.S. degrees in electrical engineering from East China Normal University, Shanghai, China, in 1968 and 1982, respectively. From 1979 to 1982, he was with the East China Normal University, where he was engaged in numerical calculations of electromagnetic (EM) fields and scattering problems. In 1982, he joined the Shanghai Research Institute of Microwave Technology, Shanghai, China, where he was involved in the research and development of millimeter-wave devices and circuits. This activity resulted in -band six-port measurement systems, -band attenuators, phase shifters, -band mismatched calibration standard kits. Since 1987, he has and been a Research Associate with the Department of Electrical Engineering, Texas A&M University, College Station, where he is involved in research on microwave/millimeter-wave circuits and devices, antennas, power combining, numerical analysis, and microwave measurements. He has authored or coauthored over 60 technical papers in the microwave and millimeter-wave fields.

Ka

Ka

Ka

Kai Chang (S’75–M’76–SM’85–F’91) received the B.S.E.E. degree from the National Taiwan University, Taipei, Taiwan, R.O.C., in 1970, the M.S. degree from the State University of New York at Stony Brook, in 1972, and the Ph.D. degree from The University of Michigan at Ann Arbor, in 1976. From 1972 to 1976, he was a Research Assistant with the Microwave Solid-State Circuits Group, Cooley Electronics Laboratory, University of Michigan. From 1976 to 1978, he was with Shared Applications Inc., Ann Arbor, MI, where he was involved with computer simulation of microwave circuits and microwave tubes. From 1978 to 1981, he was with the Electron Dynamics Division, Hughes Aircraft Company, Torrance, CA, where he was involved in the research and development of millimeter-wave solid-state devices and circuits, power combiners, oscillators, and transmitters. From 1981 to 1985, he was with TRW Electronics and Defense, Redondo Beach, CA, as a Section Head, where he developed state-of-the-art millimeter-wave integrated circuits and subsystems including mixers, voltage-controlled oscillators (VCOs), transmitters, amplifiers, modulators, upconverters, switches, multipliers, receivers, and transceivers. In August 1985, he joined the Electrical Engineering Department, Texas A&M University, College Station, as an Associate Professor and became a Professor in 1988. In January 1990, he became E-Systems Endowed Professor of Electrical Engineering. He has authored and coauthored several books, including Microwave Solid-State Circuits and Applications (New York: Wiley, 1994), Microwave Ring Circuits and Antennas (New York: Wiley, 1996; 2nd edition 2004), Integrated Active Antennas and Spatial Power Combining (New York: Wiley, 1996), RF and Microwave Wireless Systems (New York: Wiley, 2000), and RF and Microwave Circuit and Component Design for Wireless Systems (New York: Wiley, 2002). He has served as the Editor of the four-volume Handbook of Microwave and Optical Components (New York: Wiley, 1989 and 1990; 2nd edition 2003). He is the Editor of Microwave and Optical Technology Letters and the Wiley Book Series on “Microwave and Optical Engineering” (over 65 books published). He has authored or coauthored over 450 papers and numerous book chapters in the areas of microwave and millimeter-wave devices, circuits, and antennas. He has graduated over 25 Ph.D. students and over 35 M.S. students. His current interests are microwave and millimeter-wave devices and circuits, microwave integrated circuits, integrated antennas, wide-band and active antennas, phased arrays, microwave power transmission, and microwave optical interactions. Dr. Chang has served as technical committee member and session chair for the IEEE Microwave Theory and Techniques Society (IEEE MTT-S), the IEEE Antennas and Propagation Society (IEEE AP-S), and numerous international conferences. He was the vice general chair for the 2002 IEEE International Symposium on Antennas and Propagation. He was the recipient of the 1984 Special Achievement Award presented by TRW, the 1988 Halliburton Professor Award, the 1989 Distinguished Teaching Award, the 1992 Distinguished Research Award, and the 1996 Texas Engineering Experiment Station (TEES) Fellow Award presented by Texas A&M University.

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Noise Behavior of Microwave Amplifiers Operating Under Nonlinear Conditions Laurent Escotte, Eric Gonneau, Cédric Chambon, and Jacques Graffeuil, Senior Member, IEEE

Abstract—The noise behavior of microwave amplifiers operating under a large-signal condition has been studied in this paper. A Gaussian noise is added to a microwave signal and they are applied at the input of several amplifying devices. Experimental data show a decrease of the output noise spectral density when the power of the microwave signal at the input of the devices increases due to the compression of the amplifiers. A distortion component due to the interaction of the signal and its harmonics with the noise is also demonstrated from a simplified theoretical model. The statistical properties of the signal and the noise have also been investigated in order to verify the Gaussianity of the noise at the output of the nonlinear circuits. We have also observed that the majority of the measured devices show some variations of their additive noise versus the input power level. Index Terms—Microwave amplifiers, noise measurements, nonlinear device, signal and noise, spectral analysis.

I. INTRODUCTION

T

HE mathematical theory of random signals and noise in nonlinear devices has been intensively investigated [1]–[7]. It was applied to the case of square-law detectors, half-wave linear detectors, or amplitude limiters. The effects of noise have been also studied in traveling-wave tube amplifiers [8] and in FM systems [9], [10]. Some of the studies concern only Gaussian noise at the input of the devices [5]–[7], while others rely on the addition of a signal and Gaussian noise [1]–[4]. In the latter, the particular case of a sine wave has also been studied. However, all approaches remain theoretical and suffer from a lack of experimental data. A simple technique to characterize the noise figure (NF) of various silicon-based bipolar transistors has recently been proposed [11] when the devices are operated under nonlinear conditions. The results showed that the NF increases when the power at the input of the transistors increases and that the measured values are strongly correlated to the ones of the measured residual phase noise. The dependence of the NF on the input power of the signal has been also outlined in [12] and a new definition of the NF (noise and distortion figure) has been proposed by other authors [13]. The noise behavior of microwave amplifiers operating under nonlinear conditions must be accurately analyzed in order to evaluate the desensitization Manuscript received February 3, 2005; revised April 29, 2005 and June 6, 2005. L. Escotte, C. Chambon, and J. Graffeuil are with the Laboratoire d’Analyse et d’Architecture des Systèmes, Centre National de la Recherche Scientifique, 31077 Toulouse Cedex, France and also with the Electronic Engineering Department, Paul Sabatier Université, 31077 Toulouse Cedex, France (e-mail: [email protected]). E. Gonneau is with the Laboratoire de Télédétection à Haute Résolution, Paul Sabatier Université, 31029 Toulouse Cedex, France (e-mail: [email protected]). Digital Object Identifier 10.1109/TMTT.2005.856082

Fig. 1. Theoretical model used to calculate the power spectrum at the output of the nonlinear device.

of RF receivers [14], [15]. This also becomes a critical point for each new generation of mobile phones [16] or in the case of multistandards systems. Indeed, some of the noise and signal at the output of the power amplifier can be coupled, through the duplexer, to the low-noise amplifier (LNA) input. The transmit signal, acting as an interferer, further increases the NF of the LNA, and this point needs careful investigations. There is also a special concern for assessing residual phase noise of microwave devices [11], [17] in order to design low-noise oscillators: in this case, the interaction of the input signal with noise is essential. Thus, we propose in this paper to investigate the noise properties of a few different microwave amplifiers operating under nonlinear conditions from both a theoretical and an experimental point-of-view. Section II is then dedicated to the theoretical analysis and some general results are reported. The measurement technique proposed in [11] has been refined and is described in Section III. Moreover, time-domain measurements are reported in order to study the statistical properties of the signal and noise. The variations of the additive noise of the amplifiers versus the input power are analyzed with the help of frequency-domain measurements. Finally, Section IV presents the comparison between experimental data and theoretical results. The variations of the additive noise versus the input power of the amplifiers are then investigated. II. THEORETICAL ANALYSIS The theoretical model of the investigated system is represented in Fig. 1. A white Gaussian noise delivered by a noise source passes through a low-pass filter (LPF) featuring an equivfeaturing a constant alent noise bandwidth . The noise power spectral density (PSD) over the bandwidth is then , which is assumed to be sinusoidal as added to the signal follows: (1) and represent the amplitude and angular frequency of the at the input sine wave, respectively. The power spectrum of of the nonlinear device is given in Fig. 2.

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Furthermore, it can be easily found that, in the case of a sine wave and assuming ergodic process of the signal, when the sum is an odd number. The same behavior can also be found for a Gaussian noise, as previously stated in [2] when the sum is an odd number. and [4], i.e., can then be expressed as The components of Fig. 2. Power spectrum of x(t) at the input of the nonlinear device.

(5) (6)

The frequency of the signal is arbitrarily chosen greater than the noise bandwidth in order to suit the experimental requirements. This can also correspond to the practical case where an interferer or a blocking signal is close, but outside the receiver bandwidth. The model given in Fig. 1 is close to the test set described in Section III-A where both the amplitude of the can be varied. signal and the input noise spectral density at The problem is to determine the power spectrum the output of the device. For that purpose, the nonlinear circuit, supposed to be memoryless, is represented by the following expression:

(7) (8) (9)

(2) where are the coefficients of the polynomial. This type of expression is generally used to describe nonlinearities even if it is well known that it cannot fully describe the complex nonlinear behavior of microwave amplifiers. The nonlinear device is assumed noiseless in a first step. The additive noise contributed by the noise sources located inside the two-port will be taken into account in Section IV. The theoretical analysis is then concentrated on the impact of an external noise entering a nonlinear device in the presence of a large sinusoidal signal. The degree of the polynomial is fixed at 3 in order to reduce the length of the mathematical derivation, which is decomposed in two steps. In the first one, the autocorrelation function is calculated. The output spectrum is then expressed in the second step. A. Autocorrelation Function

(10) where corresponds to the varicorreance of the signal and sponds to the variance of the noise. Concerning the signal, the following equations can be easily obtained: (11) (12) (13) Concerning the noise, and using Price’s theorem [6], the following equations can be derived, which are in agreement with [7]:

of the output signal is The autocorrelation function calculated using the direct method [3]

(14) (15) (16)

(3) is the expected value or the statistical average. Aswhere suming and , , where subscripts 1 and 2 are related to the variables and respectively, (3) can be rewritten as

Equations (11)–(16) are injected into (5)–(10) and the new expressions are then used to calculate the expression of the autocorrelation function of the output signal

(17) (4) The autocorrelation function is then composed of six components, which are derived using the fact that the signal and noise are statistically independent and that the expected values of , , , and are 0.

where the coefficients

and

are as follows: (18)

(19)

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It can be shown that several components can be neglected in and the output noise spectrum. The components are 100 dB lower than the value corresponding to in the noise bandwidth, while the component is found to be even more negligible. and are much lower This is mainly due to the fact that than . It can also be noted that the variance of the noise , which is equal to , is also negligible when comparing its . This simplifies value to the variance of the signal and given by (18) and (19). Therethe expressions of fore, if we concentrate on the noise PSD at the output, can be approximately given as Fig. 3. Output noise spectral density S (f ) showing the different components. The dc component and the spectrum related to the signal are not plotted. f = 3:4 GHz, V = 25 mV, B = 2:3 GHz, and N = kT with T = 290 K, = 33:5, = 70, and = 1450.

(21)

0

B. Output Spectrum The PSD of the output signal is obtained by taking the Fourier transform of the autocorrelation function given by (17)

III. FREQUENCY- AND TIME-DOMAIN EXPERIMENTAL SETUP, CHARACTERISTICS OF INVESTIGATED DEVICES, TIME-DOMAIN RESULTS Firstly, the frequency-domain experimental setup and procedure is described and some essential characteristics of the amplifiers that will be later measured are given. Secondly, the test-set modifications needed for time-domain measurements are presented in order to check the assumption of Gaussianity of the noise at the output of the different devices. A. Frequency-Domain Measurements

(20) represents the Dirac–delta function and the symbol denotes the convolution product. The output spectrum can be decomposed in several components [3], [18]. The first term in the first line in (20) corresponds ). The last term in the first line and to the dc component ( the second line correspond to the interaction of the signal with ). Lines 3 and 4 correspond to the interaction of itself ( the noise with itself ( ). Lines 5–7 in (20) are related ). The to the interaction of the signal with the noise ( different convolution products are given in the Appendix . Fig. 3 represents the different components of the output noise ) calculated spectral density ( from (20) and from the different expressions given in the Appendix . The amplitude of the signal is fixed at 25 mV corre). The value sponding to an input power of 22 dBm ( of is set to 3.4 GHz and is greater than the noise bandwidth , which is equal to 2.3 GHz. The input noise power density is , assuming that corresponds to the stangiven by dard temperature (290 K) and is the Boltzmann’s constant. are those of amplifier #1 described in SecThe coefficients tion III. All the components appearing in Fig. 3 are normalized . It can be seen that the interaction of the with respect to signal with noise impacts in many different bandwidths and that the noise level rises as the input power increases. The different , , and components are overlapping between the frequencies and corresponding to 1.1 and 2.3 GHz, respectively.

The experimental setup used to characterize the different amplifiers in the frequency domain is reported in Fig. 4. Some differences from the one previously described in [11] have been introduced. A spectrum analyzer is now used to measure the different harmonics level at the output of the amplifier. This allows to eval) and uate the output power at 1-dB compression gain ( to calculate the total harmonic distortion (THD) at the output of the device-under-test (DUT). The maximum power delivered by the RF synthesizer at is adjusted for each amplifier in order to obtain the same nonlinear condition fixed at 3-dB compression gain. In this case, the THD calculated up to the third harmonic is close to the one calculated using five harmonics since the deviation is less than 0.7% for all the investigated amplifiers. The LPFs were characterized using a network analyzer in -parameter order to determine their transfer function. The of the LPFs exhibits a minimum value ( 60 dB) at a frequency of 3.4 GHz, which has been chosen for . The power gain is then modeled (up to ) using the Butterworth function, and the equivalent noise bandwidth is calculated using the conventional equation [19]. The value of is equal to 2.3 GHz. The LPF located after the noise source is used in order to match the experimental test set to the theoretical model reported in Fig. 1. The same experiment has also been carried out using a narrow-band filter centered at the same frequency where the noise is measured: results are essentially unchanged and will not be addressed further. The LPF located in the receiver (represented in the dashed box in Fig. 4) is used to attenuate the RF signal at , ensuring a linear behavior of the noise receiver. The noise is measured

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Fig. 4. Experimental setup used to determine the noise PSD at the output of the amplifier operating under nonlinear condition.

at a lower frequency (2 GHz), which is compatible with the different frequency bands of the various microwave elements used in the setup. This measurement principle is identical to the one described in [11] and is also used to characterize the hot small-signal -parameters of microwave power transistors [20]. The DUT is located between two isolators in order to reduce mismatch uncertainties when measuring the noise powers [21]. The noise and RF signal are combined at the input of the DUT with the help of a 3-dB hybrid junction. The losses of the two-port, including the LPF, the 3-dB coupler, and the input isolator, are measured at 2 GHz for appropriate correction in the NF meter. The same thing is also performed for the two-port located between the RF synthesizer and the isolator output at GHz in order to precisely know the power at the input of the DUT. The noise receiver is calibrated when the RF signal is off and without the DUT. The NF meter measures the noise power at an intermediate frequency of 22 MHz [11] when the noise source is alternatively off and on and the corrected noise are then displayed. The power spectral densities relative to characteristics of the measured amplifiers at 2 GHz are reported in Table I. The gain and NF are measured under small-signal conditions (without RF signal). All these devices are commercially available amplifiers and detailed information about their internal configurations is not available.

TABLE I CHARACTERISTICS OF THE MEASURED AMPLIFIERS AT 2 GHz

Fig. 5. PDFs. (a) Sinusoidal signal. (b) Sinusoidal and Gaussian noise. Solid lines correspond to theoretical expressions.

B. Time-Domain Measurements Time-domain measurements are performed with the help of a 6-GHz bandwidth digital oscilloscope that has been substituted for the NF meter and spectrum analyzer of Fig. 4 at the DUT output. The statistical properties of the signal and the noise at the output of the DUT can then be investigated. Two basic experiments are first performed. The first of these consists of measuring the signal delivered by the RF synthesizer and of calculating its probability density function (PDF). In the second experiment, we determine the PDF of the signal plus the noise combined within a 3-dB hybrid junction. The noise source is on and is amplified using a broad-band LNA (10 kHz–1 GHz) in order to be larger than the noise level of the oscilloscope. The results are reported in Fig. 5(a) and (b). 800 000 points are used in the statistical analysis and the width of the classes in the different histograms is limited by the sensibility of the oscilloscope. The PDF of the sinusoidal signal reported in Fig. 5(a) is in agreement with the expected result [19] reported as follows: (22)

Fig. 6. PDFs of the noise at the output of amplifier #1. (a) Linear conditions. (b) Nonlinear conditions.

The maximum value ( ) of the voltage is approximately 10 mV and the calculated value of the density function at 0 V . The PDF of the signal plus the noise corresponds to is plotted in Fig. 5(b). The theoretical plot corresponds to the convolution of the PDF of the signal with the PDF of the Gaussian noise [22]. The amplitude of the signal is 20 mV and the standard deviation of the noise is 3 mV. Here, again, a good agreement is observed between theoretical and measured data. In a second step, we aim at investigating the behavior of the noise at the output of a microwave amplifier. Two cascaded LPFs located after the DUT are used to ensure a better attenuation of the tone at and at the different harmonics. They are needed in order to estimate the PDF of the output noise only. If these filters were not present, the PDF would be modified by the tone and this would lead to a PDF similar to the one of Fig. 5(b). The results corresponding to amplifier #1 are reported in Fig. 6(a) and (b) when the RF signal is off and on, respectively. In the latter,

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Fig. 7. Output noise spectral density S (f ) of amplifier #1 versus input power P at f = 2 GHz. Symbols: measured data. Solid lines: T is constant. Dashed lines: T is constant.

Fig. 9. Output noise spectral density S (f ) of amplifier #3 versus input power P at f = 2 GHz. Symbols: measured data. Solid lines: T is constant. Dashed lines: T is constant.

Fig. 8. Output noise spectral density S (f ) of amplifier #2 versus input power P at f = 2 GHz. Symbols: measured data. Solid lines: T is constant. Dashed lines: T is constant.

Fig. 10. Output noise spectral density S (f ) of amplifier #4 versus input power P at f = 2 GHz. Symbols: measured data. Solid lines: T is constant. Dashed lines: T is constant.

the input power is 10 dBm corresponding to 3-dB compression gain. The density function of the noise at the output of the amplifier without any RF signal [see Fig. 6(a)] closely fits the normal law, which validates the assumption of a Gaussian noise in the mathematical model of Section II. This result is also confirmed when calculating the higher order moments ( and ) of the distribution in order to check the Gaussianity of the noise. The PDF of the noise at the output of the amplifier operating under a nonlinear condition is reported in Fig. 6(b) and is again compared to the normal law. A good agreement is still observed and a deviation less than 7% is obtained between the theoretical value and the measured one of the fourth-order moment . It can, therefore, be stated that the statistical properties of a noise passing through a nonlinear device remain unchanged. This statement is experimentally verified as long as an efficient filtering of the tone pumping the nonlinear device is provided.

is on) at dBm, corresponding to the case of 3-dB compression gain for device #1. It can then be concluded that the behavior of the signal and the noise are very different, as previously stated in [23]. This observation is valid for devices #1–#3. Amplifier #4 exhibits different microwave performances than the other amplifiers (small gain and high NF), but the harmonic distortion is very low. , reported in The variations of the noise PSD versus Fig. 10, are small compared to the other amplifiers and a deviation of approximately 2.5 dB is observed when the noise source is on. This indicates that the noise behavior of the devices operating under nonlinear conditions is closely correlated to the harmonics level. The experimental results are compared to the theoretical model. The noise PSD is given from (21) where and , given in the Appendix, the values of is greater than . The are calculated in the case where is then obtained assuming a single-sided expression of band spectrum for a better comparison with experimental data as follows:

IV. RESULTS AND DISCUSSION The experimental results are reported in Figs. 7–10 for the different amplifiers. The noise power spectral densities relative are plotted versus the power at the input of the DUT to ( ). The measurements are performed when the noise source is off and on, corresponding to a noise temperature at the input of the devices equal to 297 K ( ) and 2625 K ( ), respectively. All the amplifiers exhibit the same behavior. The noise increases. For example, we observe in PSD decreases when Fig. 7, a deviation of 7 dB between the small-signal value of the noise PSD and the value measured (when the noise source

(23) This equation indicates that the noise PSD at the output of the nonlinear device is the sum of three components. The first one is the consequence of the amplification of the input noise. It decreases as the input power increases ( ) due to gain

ESCOTTE et al.: NOISE BEHAVIOR OF MICROWAVE AMPLIFIERS OPERATING UNDER NONLINEAR CONDITIONS

compression. The second component is related to the interaction, or the mixing, of the noise with the fundamental frequency of the signal. The last component corresponds to the interaction of the noise with harmonic 2 of the signal. This last component becomes higher than the other ones when the device is strongly nonlinear. A fraction of the measured output noise is also contributed by the noise generated inside the device itself, which we call additive noise. Its PSD ( ) is added to (23) in order to give the final expression of the output noise PSD, which is compared to experimental data (24) where

is given by (25)

and when the noise The values of correspond to source is off and on, respectively. The main problem is to determine the value of the additive noise. A simple solution consists ( can be written as , of determining the value of is the noise temperature at the output of the ampliwhere fier) without any RF signal (or under small-signal condition) and . This assumption of considering this value as independent of has been used by other authors [13], [23], but it is not realistic since the noise is distributed differently along the different active devices from the input to the output of the amplifier. However, two limiting cases can be evaluated. The first one as a constant (the noise is located at consists of assuming the output of the amplifier). It corresponds to the solid lines in Figs. 7–10. In the other case (dashed lines), the noise source is located at the input of the amplifier (input noise temperature supposed to be constant) and the additive noise is given by . Two sets of coefficients ( constant and constant) can then be extracted for each amplifier. is deterand mined under small-signal condition and the coefficients are adjusted using a least square function in order to fit the measured values of when the noise source is on. In that case, the additive noise ( ) is lower than the noise at the output ). This of the device due to the amplified noise source ( is particularly verified for amplifier #2 featuring a low NF. For this example reported in Fig. 8, the additive noise is negligible (noise source on) and only one set of coefficients is found. A seems to be a good approximation for constant value of this amplifier, but it is not the case for the other devices. As reported in Fig. 9, amplifier #3 exhibits the opposite behavior seems to be a valuable assumption since a constant value of in order to fit the variations of versus when the noise source is on and off. Amplifiers #1 and #4 show a dependence of or on the input power . This is also certainly the higher than 13 dBm. Our expericase for amplifier #2 for mental results show that the noise behavior and particularly the additive noise of the microwave amplifiers operating under nonlinear conditions vary from one device to another. The observed

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or versus are certainly due to the disvariations of tributed nature of the noise sources and the nonlinearities inside the amplifiers. V. CONCLUSION The impact of input noise on small-signal microwave amplifiers operating under nonlinear conditions has been presented in this paper. It is the first time, to our knowledge, that experimental results have been reported on this topic. We have compared the measured values of the output noise PSD with those given by a model based on the mathematical analysis of a sinusoidal signal and a Gaussian noise simultaneously passing through a nonlinear device. A polynomial model of the nonlinear amplifier transfer function has been demonstrated to be efficient in order to predict the interaction of the noise with the signal and its harmonics. The statistical properties of the noise have also been studied showing that it remains Gaussian at the output of the amplifiers driven into large-signal condition. It has further been found that the additive noise depends on the input power. This behavior could be attributed to the distributed nature of both the noise sources and nonlinearities among the different devices of the multistages amplifiers. This last point could be verified in a future study by measuring, for example, cascaded amplifiers with small gain or by using “homemade” amplifiers where nonlinear models of active devices are available. APPENDIX A. Convolution Products of the Signal Assuming a sinusoidal signal, the different PSDs are reported as follows:

B. Convolution Products of the Noise The convolution between following relation:

and

is given by the

where is a temporary variable needed for performing the integration. Assuming that has the shape given in Fig. 2, the following equations can be derived: for for other frequencies.

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The component is given by the following equations:

is noted

and

:

For

for and for for

for for for

for other frequencies. is noted and The component the following two cases are considered to derive its expression. : For

for other frequencies. for and for

C. Convolution Products of the Signal by the Noise and is noted The convolution product between and the following two cases must to be considered for calculating its expression. : For

for and for for other frequencies.

for and for

For

:

for other frequencies. For

: for

for and for

and for for for other frequencies. The component is noted and the following three cases are considered to derive its expression. : For for and for for and for for for other frequencies. :

For

for and for for and for for for other frequencies.

for and for

for for other frequencies. ACKNOWLEDGMENT The authors thank Dr. N. Nolhier and Dr. E. Tournier, both of the Laboratoire d’Analyse et d’Architecture des Systèmes, Centre National de la Recherche Scientifique (LAAS–CNRS), Toulouse, France, for their help during amplifier measurements and mathematical derivation, respectively. REFERENCES [1] S. O. Rice, “Mathematical analysis of random noise,” Bell Syst. Tech. J., vol. 24, pp. 46–156, 1945. [2] D. Middleton, “Some general results in the theory of noise through nonlinear devices,” Quart. Appl. Math., vol. 5, pp. 445–498, 1948. [3] W. B. Davenport, Jr. and W. L. Root, An Introduction to the Theory of Random Signals and Noise. New York: IEEE Press, 1987, pp. 250–276. [4] H. B. Shutterly, “General results in the mathematical theory of random signals and noise in nonlinear devices,” IEEE Trans. Inf. Theory, vol. IT-9, no. 4, pp. 74–84, Apr. 1963. [5] R. F. Baum, “The correlation function of smoothly limited Gaussian noise,” IRE Trans. Inf. Theory, vol. IT-3, no. 9, pp. 193–197, Sep. 1957. [6] R. Price, “A useful theorem for nonlinear devices having Gaussian inputs,” IRE Trans. Inf. Theory, vol. IT-4, no. 6, pp. 69–72, Jun. 1958.

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[7] R. F. Baum, “The correlation function of Gaussian noise passed through nonlinear devices,” IEEE Trans. Inf. Theory, vol. IT-15, no. 7, pp. 448–456, Jul. 1969. [8] O. Shimbo, “Effects of intermodulation, AM–PM conversion, and additive noise in multicarrier TWT systems,” Proc. IEEE, vol. 59, no. 2, pp. 230–238, Feb. 1971. [9] M. L. Liou, “Noise in an FM system due to an imperfect linear transducer,” Bell Syst. Tech. J., pp. 1537–1561, Nov. 1966. [10] T. G. Cross, “Intermodulation noise in FM systems due to transmission deviations and AM/PM conversion,” Bell Syst. Tech. J., pp. 1749–1773, Dec. 1966. [11] G. Cibiel, L. Escotte, and O. Llopis, “A study of the correlation between high-frequency noise and phase noise in low-noise silicon-based transistors,” IEEE Trans. Microw. Theory Tech., vol. 52, no. 1, pp. 183–189, Jan. 2004. [12] S. L. Delage and J. Obregon, “Comments on noise source modeling for cyclostationary noise analysis in large-signal device operation,” IEEE Trans. Electron Devices, vol. 50, no. 10, p. 2183, Oct. 2003. [13] P. M. Lavrador, N. Borges de Carvalho, and J. C. Pedro, “Evaluation of signal-to-noise and distortion ratio degradation in nonlinear systems,” IEEE Trans. Microw. Theory Tech., vol. 52, no. 3, pp. 813–822, Mar. 2004. [14] R. G. Meyer and A. K. Wong, “Blocking and desensitization in RF amplifiers,” IEEE J. Solid-State Circuits, vol. 30, no. 8, pp. 944–946, Aug. 1995. [15] A. Schneider and O. Werther, “Nonlinear analysis of noise in currentsteering variable gain amplifiers,” IEEE J. Solid-State Circuits, vol. 39, no. 2, pp. 290–296, Feb. 2004. [16] T. Gee, “Suppressing errors in W-CDMA mobile devices,” Commun. Syst., vol. 7, no. 3, Mar. 2001. [Online]. [17] M. Rudolph and P. Heymann, “The influence of microwave two-port noise on residual phase noise in GaAs HBTs,” in Proc. 34th Eur. Microwave Conf., Amsterdam, The Netherlands, 2004, pp. 945–948. [18] N. M. Blachman, “The signal x signal, noise x noise, and signal x noise output of a nonlinearity,” IEEE Trans. Inf. Theory, vol. IT-14, no. 1, pp. 21–27, Jan. 1968. [19] A. Ambrozy, Electronic Noise. New York: McGraw-Hill, 1982. [20] T. Gasseling, D. Barataud, S. Mons, J. M. Nébus, J. P. Villotte, J. J. Obregon, and R. Quéré, “Hot small-signal S -parameter measurements of power transistors operating under large-signal conditions in a load–pull environment for the study of nonlinear parametric interactions,” IEEE Trans. Microw. Theory Tech., vol. 52, no. 3, pp. 805–812, Mar. 2004. [21] “Noise figure measurement accuracy: The Y -factor method,” Agilent Technol., Palo Alto, CA, Applicat. Note 57-2, 1976. [22] L. J. Bain and M. Engelhardt, Introduction to Probability and Mathematical Statistics. Boston, MA: PWS-Kent, 1992. [23] A. Geens and Y. Rolain, “Noise figure measurements on nonlinear devices,” IEEE Trans. Instrum. Meas., vol. 50, no. 4, pp. 971–975, Aug. 2001.

Laurent Escotte was born in Nouméa, France, in 1962. He received the Ph.D. degree in optic and microwave communications from the University of Limoges, Limoges, France, in 1988. Since 1989, he has been an Assistant Professor of electronic engineering with the Paul Sabatier Université, Toulouse, France. At the same time, he joined the Laboratoire d’Analyse et d’Architecture des Systèmes du Centre National de la Recherche Scientifique (LAAS–CNRS), Toulouse, France. Since 1999, he has been a Professor of electronic engineering with the Paul Sabatier Université. His current research interests include noise characterization and modeling of active devices and circuits in the microwave and millimeter-wave frequency range. He has authored or coauthored over 50 technical papers and one book.

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Eric Gonneau was born in Saint Pierre, Réunion Island, France, in 1965. He received the Ph.D. degree in signal processing from the Toulouse National Polytechnic Institute (INPT), Toulouse, France, in 1993. Since 1994, he has been an Assistant Professor of electronic engineering with the Paul Sabatier Université, Toulouse, France. Until 2004, he was with the Laboratoire d’Acoustique de Métrologie et d’Instrumentation, where he specialized in sources localization using array processing and active noise reduction on multiple -input–output systems. Since 2005, he has been with the Laboratoire de Télédetection à Haute Résolution, where his current research interests are noise fluctuations, signal and multiresolution image processing. He has authored or coauthored over 30 technical papers.

Cédric Chambon was born in Limoges, France, on August 28, 1980. He received the M.S. degree in electronics from the University of Limoges, Limoges, France, in 2003, and is currently working toward the Ph.D. degree in electronics at the Laboratoire d’Analyse et d’Architecture des Systèmes du Centre National de la Recherche Scientifique (LAAS–CNRS), Toulouse, France. His main field of interest is in the study of noise in microwave devices and circuits, more particularly, noise modeling and noise behavior in microwave amplifiers operating under nonlinear conditions, including the development of specific high-frequency noise measurement techniques.

Jacques Graffeuil (SM’90) was born in Agen, France. He received the Ingénieur degree from the Institut National des Sciences Appliquées, Toulouse, France, in 1969, and the thèse d’Etat degree in electronic engineering from Paul Sabatier Université, Toulouse, France, in 1977. Since 1970, he has been an Assistant Professor with Paul Sabatier Université. At that time, he joined the Laboratoire d’Analyse et d’Architecture des Systèmes, Centre National de la Recherche Scientifique (LAAS–CNRS), Toulouse, France, where he was engaged in research on noise in III–V semiconductor devices. He is currently a Professor of electronic engineering with Paul Sabatier Université and a Senior Scientist with the Microwave Devices and Integrated Circuits Group, LAAS–CNRS. His research initially dealt with Gunn effect devices and then with electrical properties of gallium–arsenide Schottky-barrier field-effect transistors (FETs). His current research involves noise and nonlinear properties of III–V FETs, HBTs and microwave silicon devices, silicon or III–V monolithic-microwave integrated-circuit (MMIC) design and microwave silicon microelectromechanical systems (MEMS). He has authored or coauthored over 150 technical papers and three books. He holds several patents.

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Global Stability Analysis and Stabilization of a class-E/F Amplifier With a Distributed Active Transformer Sanggeun Jeon, Student Member, IEEE, Almudena Suárez, Senior Member, IEEE, and David B. Rutledge, Fellow, IEEE

Abstract—Power amplifiers (PAs) often exhibit instabilities giving rise to frequency divisions or spurious oscillations. The prediction of these instabilities requires a large-signal stability analysis of the circuit. In this paper, oscillations, hysteresis, and chaotic solutions, experimentally encountered in a high-efficiency class-E Fodd PA with four transistors combined using a distributed active transformer, are studied through the use of stability and bifurcation analysis tools. The tools have enabled an in-depth comprehension of the different phenomena, which have been observed in simulation with good agreement with experimental results. The study of the mechanism generating the instability has led to a simplified equivalent circuit from which the optimum stabilization network has been determined. The network enables a global stabilization of the circuit for all the expected operating values of the amplifier bias voltage and input power. This has been achieved with negligible degradation of the amplifier performance in terms of drain efficiency and output power. The stable behavior obtained in simulation has been experimentally confirmed. Index Terms—Bifurcation, chaos, distributed active transformer (DAT), hysteresis, stability analysis, stabilization, switching amplifiers.

I. INTRODUCTION

S

WITCHING amplifiers such as classes D–F have been widely used to achieve high output power with high efficiency [1]. What distinguishes the different classes is the tuning technique of harmonics, the tolerance of transistor output capacitance, and the operating frequency limit. However, all of them are potentially able to achieve 100% drain efficiency by preventing the voltage and current waveforms from overlapping each other at a drain terminal. Since the transistor in switching amplifiers is operated as a switch instead of a current source to achieve high drain efficiency, the input-drive power should be large enough to make the transistor saturated for a certain time of period depending on a duty cycle. A class-E/F amplifier has been proposed by Kee et al. [2] to combine the advantages of both class-E and class-F operations. By selective tuning of harmonic components, the class-E/F amplifier can have the benefits of class F , which are low peak voltage and low rms current, using a simple Manuscript received February 22, 2005; revised May 5, 2005. This work was supported by the Lee Center for Advanced Networking and by the Jet Propulsion Laboratory. S. Jeon and D. B. Rutledge are with the Department of Electrical Engineering, California Institute of Technology, Pasadena, CA 91125 USA (e-mail: sjeon@ caltech.edu; [email protected]). A. Suárez is with the Communications Engineering Department, University of Cantabria, 39005 Santander, Spain (e-mail: [email protected]). Digital Object Identifier 10.1109/TMTT.2005.856083

push–pull configuration. The class E/F is also suitable at higher frequencies than the class E because it can tolerate a larger transistor output capacitance. During the measurements of switching amplifiers, several different behaviors can be observed. Below a certain level of input power, the transistor is completely turned off and only leakage power from the input-drive source, passing through the feed-forward capacitance of the transistor, is obtained at the output. For an intermediate input power range, the output power and drain efficiency of the amplifiers increase rapidly. However, it is also not unusual to observe spurious oscillations, sub-harmonic oscillations, or even chaos when the amplifiers are not completely stable [3]–[5]. As the input drive increases further to a high power level, the amplifiers show a typical switching amplifier operation with high drain efficiency. In switching amplifiers, the gain increase versus the input power for intermediate drive level, besides the negative resistance from nonlinear capacitances, may give rise to oscillations. These oscillations, observed from a certain level of input power, cannot be detected through a small-signal stability analysis of the circuit such as the one based on the -factor and the stability circles. Instead, a large-signal stability analysis must be performed [6]–[8]. The qualitative changes in the observed spectrum, when varying the input power, are the result of bifurcations, or qualitative stability variations [9], taking place in the circuit. In previous studies, large-signal analyses of the mechanisms leading single-ended or power-combining power amplifiers (PAs) to unstable behavior have been carried out [6]–[8], [10], [11]. As an example, the frequency division by two, commonly observed in power-combining amplifiers, has been related to odd-mode instabilities, favored by the symmetries of circuit topology. The analyzed amplifiers were operated in either classes A or AB. No similar study has ever been attempted in the case of switching amplifiers. To devise a proper stabilization procedure for these amplifiers, an understanding of the oscillation mechanism is necessary. This study will be carried out here through the use of accurate stability and bifurcation analysis tools, applied in combination with harmonic balance (HB). In particular, a PA at 29 MHz exhibiting chaos and hysteresis will be analyzed in detail. The circuit contains two push–pull pairs with four vertical double-diffused MOSFETs (VDMOS) mode. The objective is to that are operated in class-E F characterize the instability versus the bias voltage and input power. The instability contour, in terms of these two parameters, will be obtained through a bifurcation-detection technique.

0018-9480/$20.00 © 2005 IEEE

JEON et al.: GLOBAL STABILITY ANALYSIS AND STABILIZATION OF CLASS-E/F AMPLIFIER WITH DAT

Fig. 1. Schematic of the class-E=F PA with sketches of the voltage waveforms at the main circuit nodes. Two push–pull pairs are formed by M –M and M –M , although the transistors in each pair are in different transistor packages. The AG with an ideal bandpass filter inside the dashed box is not a part of the amplifier, but will be connected to one of the drain terminals for the stability analysis in Section V.

A suitable stabilization network will then be designed in order to globally suppress the instability. This paper is organized as follows. In Section II, the class-E F PA operation is briefly summarized. In Section III, the measurement results, showing unstable behavior, are presented. In Section IV, the transistor-modeling efforts, in order to obtain reliable simulation results, are discussed. In Section V, the application of the stability and bifurcation analysis techniques is presented. Finally, Sections VI and VII, respectively, deal with the stabilization procedure and the measurement results. II. OPERATION OF CLASS-E F PA WITH A DISTRIBUTED ACTIVE TRANSFORMER (DAT) A class-E F PA has been designed and implemented with discrete components at 29 MHz. Four VDMOS transistors are combined using a distributed active transformer (DAT) [12], [13]. Fig. 1 shows a schematic of the amplifier with sketches of voltage waveforms at the gate and drain terminals. By the symmetry and double differential drive of the amplifier, each and – ) can be considered to be transistor pair ( – mode independently. A capacitor operated in the class-E F and the magnetization inductance of the output transformer build a parallel resonance at the operating frequency, and present appropriate tuning impedances to the transistors at the fundaoperation. Since mental and odd harmonics for the class-E F each transistor pair is driven in push–pull, as shown in Fig. 1, a virtual ground develops at the center of the pair so that the impedance seen by a transistor is near zero at all odd harmonics factor of the parallel resonant tank is high enough. if the Therefore, the drain voltage waveforms become half-sinusoidal and 180 out-of-phase between each pair of transistors. The secondary circuits of the two output transformers are connected in series, which forms a DAT. The DAT provides each transistor pair with a 2 : 1 output impedance transformation, as well as output power combining. The DAT is made of two pieces of thick copper tape, stacked together, with a cross section of

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Fig. 2. Variation of the gain and drain efficiency of the class-E=F PA versus the saturated output power. The output power is varied by changing the drain bias voltage.

4.8 mm 1.3 mm each. The copper tape provides an inductance required for the output resonant tank with a high- factor of 600, which minimizes the ohmic loss under a high current flowing through them. It should be noted that the center of the primary circuit of the DAT is an ideal point for the dc feed owing to the virtual ground formed at the fundamental and odd harmonics. III. EXPERIMENTAL RESULTS The amplifier was experimentally characterized using a Yaesu FT-840 transceiver, a Bird oil-filled 30-dB 2-kW attenuator as a load, a Bird 4421 power meter, and an Agilent E4407B spectrum analyzer. Fig. 2 shows the measured gain and drain efficiency as a function of output power when the amplifier is driven by sufficiently large input power. The variation of the measured output spectrum at a drain bias V with different input-drive power is presented in Fig. 3. As shown in Fig. 3(a), for low input power, only a leakage signal is obtained at the amplifier output. When the input power reaches W, the output spectrum turns into the one in Fig. 3(b). The continuity of this spectrum suggests chaotic behavior. Two peaks can be seen on each side of the fundamental line at the input frequency of 29 MHz. The distance from each peak to this fundamental line is approximately 4 MHz. As the input power is further increased, this kind of spectrum continues to be observed until the input power reaches W. From this value on, the spectrum becomes the proper one, shown in Fig. 3(c). The amplifier behaves in the expected switching mode with high drain efficiency (Fig. 2). Note that even harmonics are much more attenuated than odd harmonics because of the push–pull operation of the amplifier. The reverse sense of input power variation has also been considered and we have found a hysteresis phenomenon. If the input power is reduced, the chaotic spectrum is observed until W is reached and a mixer-like specthe value trum is obtained. Thus, chaos is observed for the input power W at which it had originated when below the value increasing the power. As we decrease the power further from W, the amplifier behaves in a self-oscillating mixer regime. The input signal at 29 MHz ( ) mixes with a self-oscillation at approximately 4 MHz ( ) and gives the output power

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Fig. 4. Quasi-periodic output power spectrum observed near the bifurcation boundary when the input power is decreased. The circuit behaves in a selfoscillating mixer regime at the input-drive frequency f = 29 MHz and the 4 MHz. oscillation frequency f

tinues to be reduced, the oscillation vanishes at the input power W. To summarize, as the input power increases, the amplifier W (jump to undergoes bifurcations at the values W (extinction of oscillachaotic solution) and tion). When the input power decreases, the amplifier undergoes W (extinction of chaotic bifurcations at the values W (extinction of oscillation). solution) and IV. TRANSISTOR MODELING

Fig. 3. Variation of the measured output power spectrum when increasing the input power. (a) P = 4 W, showing leakage power at the input-drive frequency. (b) P = 10 W, showing a chaotic spectrum. (c) P = 16:5 W, showing the proper spectrum in switching-mode operation.

spectrum of Fig. 4. It is interesting to note that the intermodulation products with even orders at the oscillation frequency ( : positive integer) are stronger than those with odd . This is attributed to the orders at this frequency common-mode oscillation in each push–pull pair, which will be discussed in more detail in Section V-C. If the input power con-

As stated in Section I, one of the objectives of this study is to fully understand the different instability phenomena, involving self-oscillation, chaos, and hysteresis, which have been observed in the measurements. With this aim, stability and bifurcation analysis tools will be applied to the PA in combination with HB. In order for the simulation tools to be successful, accurate models for the different linear and nonlinear elements will be necessary. Thus, special effort has been devoted to the transistor modeling. The active device employed in the amplifier is the ARF473 VDMOS from Advanced Power Technology Inc., Bend, OR [14]. It is a matched pair of power transistors with a maximum drain voltage of 500 V and rms drain current of 10 A for each transistor. The transistor is modeled primarily as a voltage-controlled current source with two nonlinear capacitances [15], [16]. One . This is modeled as a is the drain-to-source capacitance reverse-biased diode in which the parameters of the junction capacitance are fitted in order to match the measured capacitance as a function of the drain bias voltage. The other is the feedback capacitance between the gate and drain. The values of the feedback capacitance are extracted from the data sheet of the transistor, and a junction capacitance model is also used to fit the values. The gate-to-source capacitance is assumed to be constant as a first-order approximation [16]. The parasitic resistances and inductances at both gate and drain are also incorporated in the model as linear elements.

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V. STABILITY ANALYSIS As shown in Section III, different instability phenomena have been observed in the experimental characterization of the amplifier including self-oscillation, chaos, and class-E F hysteresis. Thus, the stability analysis of the amplifier will be a demanding one, involving different kinds of tools. The entire analysis procedure is presented as follows. A. Local Stability Analysis The initial objective is to analyze the stability of the amplifier solution for several values of drain bias voltage and input power at which unstable behavior had been experimentally observed. The stability analysis is based on the linearization of the amplifier circuit about its large-signal steady-state regime at the calculated with HB and 15 harmonic input-drive frequency components. To obtain this linearization, a small-signal current generator is connected to a particular circuit node. Due to the complex topology of the amplifier circuit, different observation nodes must be considered. The generator operates at a frequency , nonrationally related to . The purpose of the generator is to enable the determination of the total impedance function at the frequency at the observation node [17]. This is obtained by taking the ratio of the node voltage to the injected current using the conversion-matrix approach [18]. In this way, a single-input single-output transfer function is calculated to which pole-zero identification will be later applied. V and Initially, the operation conditions of W, for which instability had experimentally been observed, were considered. By sweeping and representing the real and , a critimaginary parts of the admittance function ical resonance at 5.6 MHz was found at the drain terminal of any of the four transistors. Negative conductance and a zero crossing of the susceptance with positive slope were obtained, which are the startup conditions for an oscillation at that frequency [19]. For a more rigorous stability analysis, pole-zero identification [18]. Since all circuit nodes share the same was applied to characteristic equation [20], the pole values are independent of the particular location of the current generator. However, exact pole-zero cancellations may occur at some current-generator locations. Thus, the need for the initial consideration of different observation nodes arises. Applying this technique, a pair of complex-conjugate poles are found on the right-hand side of V the complex plane for the considered conditions of and W, confirming the unstable behavior (Fig. 5). Now the variation of the input power will be considered. When increasing the input power from a very small value, the critical poles evolve, as shown in Fig. 5. The amplifier solution is initially stable with the poles located on the left-hand side of the complex plane. When increasing the input power, the critW. From this ical poles cross the imaginary axis at power value, the amplifier periodic solution becomes unstable. , a Hopf bifurcation [21] is obtained (the additional At sub-index means lower boundary). This Hopf bifurcation gives rise to the onset of an oscillation at the frequency 4.8 MHz, which is determined by the imaginary part of the poles. As the power continues to increase, the poles move further to the right, turn, and cross the imaginary axis again to the left-hand side at

= Fig. 5. Evolution of the critical poles with increasing input power for V 72 V. For simplicity, only poles in the upper half of the complex plane have been represented. The input power has been increased from 5 to 15 W by 1-W step. = 6:1 W The poles cross to the right-hand side of the complex plane for P = 13:5 W. and return to the left-hand side for P

the input power value W (the sub-index denotes upper boundary). At this power value, the oscillation vanishes. This corresponds to an inverse Hopf bifurcation occurring . at The above stability analysis provides the input-power range for which the amplifier periodic solution is unstable and, thus, , an oscillation is generated giving rise unobservable. At , the oscillation is to a self-oscillating mixer regime. At extinguished and the amplifier recovers stability. Note that the stability analysis of the amplifier periodic solution does not enable by itself the prediction of the experimentally observed hysteresis phenomenon. Actually, in the experiment, chaotic and mixer-like spectra had been found for the input power below as well, which is not explained by the previous analysis. Hysteresis is associated with Hopf bifurcations of the subcritical type [21]. The determination of the bifurcation type requires higher order derivatives of the circuit equations about the bifurcation point [22], which is beyond the scope of this paper. A different technique will be used in this study, which will be demonstrated in Section V-D. B. Instability Contour The amplifier circuits generally have one or more parameters, susceptible to be varied in the different applications. The designer will be interested in knowing the parameter ranges, which give rise to unstable operation of the amplifier. In the case of this and the amplifier, the parameters are the drain bias voltage . Thus, the objective will be the determination input power , ) values with unstable behavior. The set of the set of ( will be delimited by the Hopf-bifurcation locus, containing the points at which the oscillation is generated or extinguished, depending on the variation sense of the parameters [21]. To obtain this locus, the continuity of local bifurcations is taken into account, according to which the oscillation amplitude tends to zero at the Hopf bifurcation. Unlike previous study [7], the small-signal current generator, introduced in Section V-A, will be used here to obtain the input-admittance function at the observation node. This generator operates at a frequency

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Fig. 6. Instability contour (solid line). The Hopf-bifurcation locus delimits and P values for which the amplifier periodic solution is unstable. the V Experimental points have been superimposed. Squares indicate the upper border. In the lower border, triangles indicate the onset of instability for increasing input power, whereas stars indicate recovering of stable behavior for decreasing power. The dashed line shows the input power variation in the pole-zero identification of Fig. 5. Note that both analyses show good consistency in bifurcation points.

, nonrationally related with . The admittance is calculated as the ratio between the delivered current and node voltage by means of the conversion-matrix approach. At the bifurcation , both the real and imaginary parts point, occurring for of the input admittance vanish. This oscillation condition is fulfilled for oscillation amplitude tending to zero, as expected at the Hopf-bifurcation point. Thus, the Hopf-bifurcation locus, which and , delimits the unstable behavior region in terms of is obtained by solving the following system: (1) with being the oscillation frequency. The above system (1) is solved through error minimization or optimization, using HB with 15 harmonic components and the conversion-matrix apS and proach. The goals are Real S. There exist three unknowns, i.e., , , and , in . the two equations given by the real and imaginary parts of This gives a curve in the plane defined by the bias voltage and the input power . Note that the oscillation frequency must be included in the calculation, as the frequency is autonomous and, thus, varies along the locus. The application of the above technique to the class-E F PA has provided the instability contour of Fig. 6. The dashed V considline shows the input power variation at ered in the pole-zero identification of Fig. 5. The consistency in the bifurcation points resulting from both analyses should be noted. Through several applications of the pole-zero identification technique, the unstable region is confirmed to be inside the locus. The locus exhibits three points of infinite slope. To pass through these points, we switch the sweep parameter between and . Thus, the entire contour has been traced, enabling accuracy in the determination of the unstable operation region. In Fig. 6, experimental points have been superimposed. In the lower border, two different sets of experimental points are represented. The triangles correspond to the points at which the amplifier becomes unstable for increasing input power. A

chaotic regime is immediately obtained at most of the represented points, as shown in the spectra of Fig. 3. The stars correspond to the oscillation extinction for decreasing input power. The two sets of points show the hysteresis phenomenon discussed in Section III. On the other hand, in the upper border, no hysteresis has been experimentally obtained and only one set of measured points has been represented by squares. This set of points shows good agreement with the upper section of the simulated locus. The instability contour provides the set of points at which the amplifier periodic solution becomes unstable, i.e., at which a pair of complex-conjugate poles cross the imaginary axis to the right-hand side of the complex plane. Thus, in the lower border, the contour must agree with the set of experimental points providing the instability threshold for increasing input power. As can be seen in Fig. 6, the obtained locus enables a good prediction of this set of values represented by triangles. The prediction of the hysteresis interval demands a different procedure to be presented in Section V-D. C. Analysis of the Self-Oscillating Mixer Regime For understanding of the oscillation mechanism, a steadystate analysis of the circuit in its undesired self-oscillating mixer regime has been carried out. The oscillating solution will exist inside the instability contour of Fig. 6. Due to the hysteresis phenomenon, it may also exist for input-power values below the lower border of the instability contour. In order to obtain the oscillating solution in HB, a two-tone analysis must be carried out. One of the fundamentals is the . The other fundamental is the oscilinput-drive frequency lation frequency . By default, HB will converge to the amplifier periodic solution, with zeroes at all spectral lines containing . In order to avoid this, an auxiliary generator (AG) is introduced into the circuit [21] for simulation purposes only. When choosing a voltage AG, this generator is connected in parallel at a circuit node. We will use the drain node, as in Section V-A. The AG operates at the oscillation frequency, i.e., , and must be an open circuit at all other frequencies. Thus, an ideal bandpass filter is used in series with the AG (Fig. 1). Furthermore, the AG must not perturb the circuit steady-state solution. This is ensured by imposing a zero value , to its current-to-voltage relationship and are the current and voltage of the AG, rewhere and , the amplitude and the spectively. For given of the AG are calculated in order to fulfill the frequency condition . Even though the amplifier contains four transistors, only one AG, connected at one of the drain terminals, is necessary to determine the oscillating steady state. To investigate the nature of the oscillation, the phase at each drain terminal of the four transistors has been analyzed at different harmonic frequencies (see Table I). At the oscillation frequency , the two transistors in the same pair are in-phase, whereas the two pairs are 180 out-of-phase. However, at the input-drive frequency , the original phase-shift relationships are maintained, i.e., 180 phase shift between the two transistors in the same pair and 180 phase shift between the two pairs as well. Other phase relationships exist at intermodulation products of the two frequencies. The oscillation can be understood

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PHASE

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TABLE I FREQUENCIES

OF SIGNALS WITH DIFFERENT TERMINAL OF THE FOUR TRANSISTORS (

V V

AT EACH DRAIN ARE DEFINED IN FIG.

1)

Fig. 8. Schematic of the push–pull amplifier showing the virtual-ground and virtual-open planes at the oscillation frequency.

V

V

Fig. 7. Comparison of two simulated drain voltage waveforms and in the same push–pull pair. The slowly varying envelopes at the oscillation frequency appear to be in-phase, whereas the fast-varying carriers are out-of-phase.

as the result of the negative resistance exhibited by the transistor under relatively strong pumping signal and the resonant circuit formed with the equivalent capacitance and inductance seen from the drain terminals. This equivalent circuit will be discussed in detail in Section VI. The two drain voltage waveforms in the same push–pull pair, and , are compared in Fig. 7. It can be seen that the i.e., are in-phase, whereas the fastslowly-varying envelopes at show a 180 phase shift from each other. varying carriers at Since the amplifier is operated in push–pull, the in-phase drain voltage waveforms at the oscillation frequency will ideally be cancelled, presenting no power in the output spectrum. This explains why the intermodulation products of the form ( : positive integer) are much more attenuated , as can be seen in Fig. 4. than those having the form In a practical amplifier, however, no perfect cancellation can occur due to the imperfect symmetry. Taking the above phase relationships into account, different virtual-ground and virtual-open planes can be considered , in the circuit topology. At the input-drive frequency virtual-ground planes exist between any of two adjacent transistors. At the oscillation frequency , two virtual-ground planes are located between the two push–pull pairs, as shown in Fig. 8. In addition, two virtual-open planes develop at the symmetry planes of the pairs. Considering these virtual-ground and virtual-open plane concepts, a simplified equivalent circuit at the oscillation frequency will be obtained in Section VI, which will be useful in efficiently finding the stabilization network.

Fig. 9. Simulation of the undesired self-oscillating mixer regime of the PA. Variation of the oscillation amplitude at the drain terminal is represented with the input-drive power. The points at which the different bifurcations occur are and indicated. denotes a Hopf bifurcation and denotes a turning point. are Hopf bifurcations from the amplifier periodic regime. is a Hopf bifurcation from the self-oscillating mixer regime. and indicate jumps , which is analyzed in of the solution. Chaotic solutions are observed from Section V-E.

H

H

T

J H

J

H

H

D. Hysteresis Prediction To study the hysteresis phenomenon, the evolution of the self-oscillating mixer solution versus the input power will be analyzed. As discussed in Section V-C, an AG is connected to is any drain terminal and the equation in combination with HB. A two-fundamental solved versus and , the latter playing the role of frequency basis at the oscillation frequency , must be considered in the HB simulation. The resulting variation of the oscillation amplitude at , is represented in Fig. 9. the drain terminal, agreeing with V has been assumed. As A constant bias voltage can be seen, the curve exhibits an infinite-slope point or turning point . To pass through this point, the sweep parameter has in the neighborbeen switched to the oscillation amplitude and hood of the turning point, calculating the input power the oscillation frequency for each value. The turning point is responsible for the hysteresis phenomenon. Actually, when the input power is increased, the transition from the stable amplifier behavior to the self-oscillating mixer occurring regime ( in Fig. 9) is due to a Hopf bifurcation

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in the amplifier solution. When the input power is decreased, the transition back to the stable amplifier behavior ( ) is due to the turning point in the self-oscillating mixer solution. Note that the simulated hysteresis interval, in terms of the input power, is in good correspondence with the experimental one shown in Fig. 6. On the other hand, no hysteresis is obtained in the upper , which also agrees with the input-power range, delimited by measurement results. The hysteresis phenomenon is well predicted by Fig. 9. However, in the measurement, an abrupt transition from stable amplifier behavior to the chaotic regime occurred for most values, when the input power was increased. The study of this chaotic solution will require additional tools, to be presented in Section V-E. E. Envelope-Transient Analysis of the Oscillating Solution The envelope-transient enables an efficient analysis of the regimes in which two different time scales may be distinguished. In this technique, the circuit variables are expressed in a Fourier series with time-varying coefficients and a differential-equation system is obtained in these coefficients [23], [24]. The technique is efficiently applied to forced circuits. However, when used for the simulation of an oscillating regime, like that of the unstable amplifier, it generally converges to the coexisting nonoscillating solution in a similar manner to HB. To avoid this, the oscillation must be properly initialized [25]. This can be done through the connection of an AG to the circuit at the initial envelope time . The amplitude and frequency of the AG are determined through a previous HB simulation. Since the AG is used for the initialization of the solution only, it must be disconnected from the . After this disconnection, the circuit will circuit for time evolve according to its own dynamics. The AG disconnection from the circuit can be carried out with the aid of a time-varying resistor in series with the AG [25] changing from zero to a very high value (ideally infinite). The objective will be to analyze the circuit along the entire solution curve of Fig. 9, corresponding to a self-oscillating mixer regime. Thus, the variables are represented here in and as fundamentals, i.e., a Fourier series with . At each point, the AG and frequency , resulting from the HB amplitude analysis in Fig. 9, are used for initialization purposes. From the in Fig. 9, the magnitude of the harmonic components point becomes time-varying [see Fig. 10(a)]. It oscillates at a few hundred kilohertz, the actual oscillation frequency depending on the input power. Thus, there is a second oscillation, in addition to the previous oscillation at approximately 4 MHz ( ). Together with the input-drive frequency, this gives rise to a three-fundamental regime. The simulated spectrum is shown in Fig. 10(b). To verify this qualitatively, an expanded view of the experimental spectrum near the turning point in Fig. 9 is shown in Fig. 11. This confirms the existence of the second oscillation at approximately 500 kHz, in agreement with the envelope-transient simulation, which has enabled the efficient detection of the second oscillation. As has been shown, there are two autonomously generated fundamentals involved in the circuit solution in addition to the input-drive frequency. According to the Ruelle–Takens theorem

Fig. 10. Envelope-transient simulation of the amplifier. (a) Time-domain evolution of the magnitude of the f harmonic component of the drain voltage when a two-fundamental basis at f and f is considered. (b) Spectrum of the harmonic component showing the presence of two oscillation frequencies.

Fig. 11. Expanded view of the experimental output power spectrum about = 72 V and P = 5:15 W. It shows the the input-drive frequency. V coexistence of two oscillation frequencies at approximately 4 MHz and 500 kHz. This is in agreement with the envelope-transient simulation of Fig. 10.

[9], this type of solution is likely to give rise to chaos, which would explain the chaotic spectrum that was observed in the experiment [see Fig. 3(b)]. Actually, chaotic envelope variations

JEON et al.: GLOBAL STABILITY ANALYSIS AND STABILIZATION OF CLASS-E/F AMPLIFIER WITH DAT

have also been obtained in simulation for some values of input power. However, it should be noted that there is limited accuracy in the representation of these solutions using the two-tone basis and . The interval in which the self-oscillating mixer solution at and is unstable has been indicated with stars in Fig. 9. At , the self-oscillating mixer solution becomes unthe point stable due to the generation of the second oscillation frequency. It is a Hopf bifurcation from the self-oscillating mixer regime. Thus, starting from very low input power, the amplifier periodic solution is initially stable, and as the input power is further increased, it suddenly becomes chaotic at the Hopf-bifurcation . This is because the input power for the bifurcation point is larger than the input power for the bifurcation so the to the chaotic regime (see in Fig. 9). solution jumps from The chaotic regime persists until the input power reaches the bifrom which the amplifier periodic solution furcation point becomes stable. When decreasing the input power, the second oscillation vanand the self-oscillating mixer regime (at and ishes at ) becomes stable for a very short input-power interval. At the turning point , the system jumps to the stable amplifier periodic solution ( in Fig. 9). In conclusion, the bifurcation diagram of Fig. 9 gives a satisfactory explanation of the experimental observations of Fig. 3. All the different phenomena observed in the measurements are associated with the occurrence of particular types of bifurcations.

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Fig. 12. Simplified equivalent circuit of the PA at the oscillation frequency after considering the virtual-open and virtual-ground planes.

Fig. 13. Amplifier schematic with stabilization network. The stabilization network consists of a stabilization resistor R , a second harmonic trap, and . a dc-blocking capacitor C

VI. STABILIZATION TECHNIQUE After understanding the different phenomena observed in the measurements, the objective will be the stabilization of the amplifier. For this purpose, a stabilization network will have to be added to the circuit. In order to efficiently obtain the optimum network, a simplified equivalent circuit will be derived here taking into account the virtual-ground and virtual-open planes at the oscillation frequency, identified in Section V-C. Fig. 12 shows the equivalent circuit that corresponds to a quarter section of the amplifier at the oscillation frequency. It is a parallel resonance oscillator composed of the transistor exhibiting negative resistance and the equivalent capacitance, inductance, and load resistance seen from the drain terminal. The connected between two transistor pairs output capacitance is doubled due to the virtual-ground developed at the center of in the the capacitance, and the magnetization inductance output transformer is divided by two due to the virtual-open at the center of the transformer. Note that the RF choke inductance is also a critical element included in the equivalent circuit. The resonance frequency is 5.3 MHz, which is quite close to the oscillation frequency obtained both in measurement and simulation. This confirms the validity of the proposed equivalent circuit. From the schematic of Fig. 12, a simple means to stabilize the amplifier is the addition of a resistor at node N. The value of this resistance must be small enough to avoid the oscillation for and . all the possible operation conditions in terms of It also must not affect the normal operation of the amplifier.

The node N corresponds to the center point of the primary circuit in the output transformers. The push–pull operation introduces a virtual ground at the node for the operating frequency and odd harmonics. Hence, the addition of the resistance at the node N will have little effect over these frequencies. However, the resistor will impose a finite impedance to even harmonics instead of an open circuit, which would be the right teroperation. Thus, a second harmination for the class-E F will be connected in series with the resistor to monic trap at reduce the effect. This will provide an open circuit at the second harmonic frequency. The effect of higher even harmonics on the operation, expected to be small, will be analyzed through simulation. The schematic of the amplifier with the stabilization network is shown in Fig. 13. Once the topology and location of the stabilization network have been determined, the next step will be the calculation of stabilization resistance value, in order to ensure stable ampliand . An fier operation for all the expected values of efficient technique will be applied for this purpose. The technique is based on the plot of the small-signal input admittance . This is calculated at the drain terminal using the small-amplitude current source and the conversion-matrix approach (see with Section V-A). Comparing the frequency variation of pole-zero identification results, it has been possible to associate the instability with the existence of negative conductance and will resonance at the oscillation frequency. Thus, the plot of allow a fast verification of these oscillation conditions.

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Three different values of the stabilization resistance have been considered, i.e., 100, 50, and 15 . For each value, two nested sweeps are carried out in the two amplifier parameters and . For each ( , ) point, an HB calculation is performed, together with a sweep in the current-source frequency , using a conversion matrix. This yields the input seen by the current source. The imaginary admittance part of is then plotted versus the real part. The resulting plots for the original amplifier and amplifier with three indicated resistance values are shown in Fig. 14. Each admittance , ). The same curve corresponds to a pair of values ( frequency-sweep range has been considered for each curve. In the representation, this range has been limited to 3–5.5 MHz for the sake of clarity. For global stability, no crossing of the real axis with negative conductance and positive increase of the susceptance must be obtained [19]. As expected, larger stability ranges are achieved as the stabilization resistance is reduced. For , the amplifier becomes stable for all the operation values of and . This has been rigorously verified by extending the range of the frequency sweep and applying pole-zero identification. Through bifurcation detection, it is possible to directly calculate the stabilization resistance for given and values. To achieve this, the stabilization resistance and the oscillation frequency will be determined in order to fulfill . For each and , the resulting resistance is the maximum value allowed for stable behavior. The resistance value is actually a bifurcation value: the amplifier is unstable for , whereas it is stable for . , a In order to globally determine the variation of sweep of is performed for several values covering the expected operation ranges. For each , the equation is solved to calculate versus , which is shown in Fig. 15. As can be observed, decreases with the bias voltage. On the other hand, as approaches values for which the amplifier periodic solution is stable, this resistance tends to infinity. From Fig. 15, a resistance value smaller than 17 is required for global stabilization of the amplifier. The results are consistent with those obtained from the admittance plots of Fig. 14. In view of the results of Figs. 14 and 15, the resistance value has been chosen for the corrected design of the amplifier. As the final step, the influence of this resistance on the amplifier drain efficiency and output power has been analyzed. This is shown in Fig. 16, where the drain efficiency and output power is traced versus the resistance value. As can be seen, the stabilization resistance has only a small influence. This is due to the fact that the connection point is a virtual ground at the fundamental and odd harmonics, and the second-harmonic trap has been used to maintain the connection point as an open circuit at that frequency. The higher even harmonics turned out to have negligible influence. VII. MEASUREMENTS OF THE STABILIZED AMPLIFIER

Fig. 14. Stabilization action of the parallel resistance analyzed by means of admittance plots. Three resistance values have been considered. (a) No stabilization resistor. (b) R = 100 . (c) R = 50 . = 15 . In (a)–(c), the oscillation condition is satisfied for a certain (d) R parameter range and, thus, global stability is not achieved.

The amplifier has been modified for globally stable behavior by introducing the stabilization network developed in Section VI. A stabilization resistor of 15 in series with the

second harmonic trap and a dc-blocking capacitor is simply connected to the center point of each output transformer. Power

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the oscillation frequency and the feasibility of realizing each component. A mica capacitor and an air-core inductor with 14-AWG copper wire are used for the second harmonic trap. This amplifier has shown global stability over the entire range and , as predicted in the simof operating conditions of ulation. We never observed any oscillation or chaotic regimes. The output power spectrum was similar to Fig. 3(c) whenever the input drive was sufficient to turn on the transistors. The drain efficiency of the amplifier has been measured, which is shown in Fig. 17. As can be seen, the only significant degradation is 1.3% for 240-W output power. VIII. CONCLUSION Fig. 15. Variation of the maximum value of stabilization resistance R versus the input-drive power P obtained through bifurcation analysis. Three different drain bias voltages have been considered.

In this study, the unstable behavior of a class-E/F PA, experimentally exhibiting self-oscillation, chaos, and hysteresis, has been studied. Stability and bifurcation analysis tools have been applied to give satisfactory explanation to the different phenomena observed in the experiment, which are associated with particular kinds of bifurcations. An in-depth analysis of the oscillation mechanism has also been carried out. From this study, a simplified equivalent circuit at the oscillation frequency has been derived, enabling an efficient determination of the topology and location of the required stabilization network. An admittance plot has allowed a quick test for global stabilization. With the inclusion of the stabilization network, the PA behaves in a globally stable manner with minimum degradation in drain efficiency and output power. ACKNOWLEDGMENT

Fig. 16. Variation of the amplifier drain efficiency and output power versus the value of the stabilization resistance. The drain bias voltage is assumed as V = 72 V.

The authors would like to thank S. Weinreb, Jet Propulsion Laboratory, Pasadena, CA, K. Potter and F. Wang, both of the California Institute of Technology, Pasadena, and J.-M. Collantes, University of the Basque Country, Bilbao, Spain, for their advice and discussions. REFERENCES

Fig. 17. Measured drain efficiency versus the output power. Solid line: stabilized PA. Dashed line: original PA. Compared to the original PA, the drain efficiency is degraded by less than 0.4% for all output power levels, except for 240 W, which shows 1.3% degradation.

resistors with a 35-W rating are used in order to handle the simulated current of 0.7 A at even harmonics of the input-drive frequency. The factor of the second harmonic trap is carefully chosen considering a tradeoff between low impedance at

[1] H. L. Krauss, C. W. Bostian, and F. H. Raab, Solid State Radio Engineering. New York: Wiley, 1980. [2] S. D. Kee, I. Aoki, A. Hajimiri, and D. B. Rutledge, “The class-E/F family of ZVS switching amplifiers,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 6, pp. 1677–1690, Jun. 2003. [3] E. Lau, K.-W. Chiu, J. Qin, J. Davis, K. Potter, and D. Rutledge, “High-efficiency class-E power amplifiers—Part 1,” QST, pp. 39–42, May 1997. [4] M. Albulet, RF Power Amplifiers. New York: Noble, 2001. [5] J. Vidkjaer, “Instabilities in RF-power amplifiers caused by a self-oscillation in the transistor bias network,” IEEE J. Solid-State Circuits, vol. SSC-11, no. 10, pp. 703–712, Oct. 1976. [6] A. Anakabe, J. M. Collantes, J. Portilla, J. Jugo, S. Mons, A. Mallet, and L. Lapierre, “Analysis of odd-mode parametric oscillations in HBT multi-stage power amplifiers,” in Eur. Microwave Week—11th GaAs Symp., Munich, Germany, Oct. 2003, pp. 533–536. [7] A. Collado, F. Ramírez, and A. Suárez, “Analysis and stabilization tools for microwave amplifiers,” in IEEE MTT-S Int. Microwave Symp. Dig., Fort Worth, TX, Jun. 2004, pp. 945–948. [8] S. Mons, J.-C. Nallatamby, R. Queré, P. Savary, and J. Obregon, “A unified approach for the linear and nonlinear stability analysis of microwave circuits using commercially available tools,” IEEE Trans. Microw. Theory Tech., vol. 47, no. 12, pp. 2403–2409, Dec. 1999. [9] J. M. T. Thompson and H. B. Stewart, Nonlinear Dynamics and Chaos. New York: Wiley, 1986. [10] D. Teeter, A. Platzker, and R. Bourque, “A compact network for eliminating parametric oscillations in high power MMIC amplifiers,” in IEEE MTT-S Int. Microwave Symp. Dig., Anaheim, CA, Jun. 1999, pp. 967–970.

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[11] M. Mochizuki, M. Nakayama, Y. Tarui, Y. Itoh, S. Tsuji, and T. Takagi, “Nonlinear analysis of f =2 loop oscillation of high power amplifiers,” in IEEE MTT-S Int. Microwave Symp. Dig., Orlando, FL, May 1995, pp. 709–712. [12] I. Aoki, S. D. Kee, D. B. Rutledge, and A. Hajimiri, “Distributed active transformer—A new power-combining and impedance-transformation technique,” IEEE Trans. Microw. Theory Tech., vol. 50, no. 1, pp. 316–331, Jan. 2002. amplifier [13] S. Jeon and D. B. Rutledge, “A 2.7-kW, 29-MHz class-E=F with a distributed active transformer,” presented at the IEEE MTT-S Int. Microwave Symp., Long Beach, CA, Jun. 2005. [14] Adv. Power Technol. Inc., Bend, OR, VDMOS Data Sheet for ARF473, 2003. [Online]. Available: http://www.advancedpower.com. [15] “Spice model for TMOS power MOSFETs,” Motorola Semiconduct. Products Inc., Phoenix, AZ, Applicat. Note AN1043, 1989. [16] Y.-S. Kim and J. G. Fossum, “Physical DMOST modeling for highvoltage IC CAD,” IEEE Trans. Electron Devices, vol. 37, no. 3, pp. 797–803, Mar. 1990. [17] V. Iglesias, A. Suárez, and J. L. García, “New technique for the determination through commercial software of the stable-operation parameter ranges in nonlinear microwave circuits,” IEEE Microw. Guided Wave Lett., vol. 8, no. 12, pp. 424–426, Dec. 1998. [18] J. Jugo, J. Portilla, A. Anakabe, A. Suárez, and J. M. Collantes, “Closedloop stability analysis of microwave amplifiers,” Electron. Lett., vol. 37, pp. 226–228, Feb. 2001. [19] K. Kurokawa, “Some basic characteristics of broad-band negative resistance oscillator circuits,” Bell Syst. Tech. J., vol. 48, pp. 1937–1955, Jul.–Aug. 1969. [20] J. Jugo, A. Anakabe, and J. M. Collantes, “Control design in the harmonic domain for microwave and RF circuits,” Proc. Inst. Elect. Eng., Contr. Theory Applicat., vol. 150, no. 2, pp. 127–131, Mar. 2003. [21] A. Suárez and R. Queré, Global Stability Analysis of Microwave Circuits. Boston, MA: Artech House, 2003. [22] J. Guckenheimer and P. Holmes, Nonlinear Oscillations, Dynamical Systems and Bifurcations of Vector Fields, 3rd ed. Berlin, Germany: Springer-Verlag, 1990. [23] E. Ngoya and R. Larcheveque, “Envelope transient analysis: A new method for the transient and steady state analysis of microwave communication circuits and systems,” in IEEE MTT-S Int. Microwave Symp. Dig., San Francisco, CA, Jun. 1996, pp. 1365–1368. [24] J. C. Pedro and N. B. Carvalho, “Simulation of RF circuits driven by modulated signals without bandwidth constraints,” in IEEE MTT-S Int. Microwave Symp. Dig., Seattle, WA, Jun. 2002, pp. 2173–2176. [25] E. de Cos, A. Suárez, and S. Sancho, “Envelope transient analysis of self-oscillating mixers,” IEEE Trans. Microw. Theory Tech., vol. 52, no. 4, pp. 1090–1100, Apr. 2004.

Sanggeun Jeon (S’05) received the B.S. and M.S. degrees in electrical engineering from Seoul National University, Seoul, Korea, in 1997 and 1999, respectively, the M.S. degree in electrical engineering from the California Institute of Technology, Pasadena, in 2004, and is currently working toward the Ph.D. degree at the California Institute of Technology. From 1999 to 2002, he was a Full-Time Instructor of electronics engineering with the Korea Air Force Academy, Cheongwon, Korea. His research interests include high-efficiency PAs and nonlinear stability analysis. Mr. Jeon was the recipient of the Third Place Award in the Student Paper Competition at the 2005 IEEE Microwave Theory and Techniques Society (IEEE MTT-S) International Microwave Symposium (IMS).

Almudena Suárez (M’96–SM’01) was born in Santander, Spain. She received the Electronic Physics and Ph.D. degrees from the University of Cantabria, Santander, Spain, in 1987 and 1992, respectively, and the Ph.D. degree in electronics from the University of Limoges, Limoges, France, in 1993. In 1987, she joined the Electronics Department, University of Cantabria, where she was involved with nonlinear simulation. From May 1990 to December 1992, she was on leave with the Institute de Recherche en Communications Optiques et Microondes (IRCOM), University of Limoges. Since 1993, she has been an Associate Professor (permanent since June 1995) with the Communications Engineering Department, University of Cantabria. She coauthored Stability Analysis of Microwave Circuits (Norwood, MA: Artech House, 2003). Her areas of interest include the nonlinear design of microwave circuits, especially the nonlinear stability and phase-noise analysis and the investigation of chaotic regimes.

David B. Rutledge (S’77–M’77–SM’89–F’93) received the B.A. degree in mathematics from Williams College, Williamstown, MA, the M.A. degree in electrical sciences from Cambridge University, Cambridge, U.K., and the Ph.D. degree in electrical engineering from the University of California at Berkeley. He is currently the Tomiyasu Professor of Electrical Engineering with the California Institute of Technology, Pasadena. He is Director of the California Institute of Technology’s Lee Center for Advanced Networking. He authored the electronics textbook The Electronics of Radio (Cambridge, U.K.: Cambridge Univ. Press, 1999) and coauthored the microwave computer-aided-design software package Puff, which has sold 30 000 copies. His research has been in integrated-circuit antennas, active quasi-optics, computer-aided design, and high-efficiency PAs. Prof. Rutledge was the recipient of the Microwave Prize, the Distinguished Educator Award of the IEEE Microwave Theory and Techniques Society (IEEE MTT-S), the Teaching Award of the Associated Students of the California Institute of Technology, the Doug DeMaw Award of the American Radio Relay League (ARRL), and the Third Millennium Award of the IEEE. He was the editor-in-chief of the IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, and a Distinguished Lecturer of the IEEE Antennas and Propagation Society (IEEE AP-S).

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Frequency Channel Blocking Scheme in Mesh-Topology Millimeter-Wave Broad-Band Entrance Networks Jaturong Sangiamwong, Student Member, IEEE, Katsutoshi Tsukamoto, Member, IEEE, and Shozo Komaki, Senior Member, IEEE

Abstract—This paper proposes a novel frequency channel blocking (FCB) scheme in mesh-topology millimeter-wave broadband entrance networks. In the FCB scheme, any frequency channel is determined to be blocked according to not only the network throughput, but also the newly defined fairness index. As the results, the FCB scheme yields better throughput and fairness performances with a little increase in computational complexity compared to the conventional scheme. Index Terms—Fairness index, frequency channel blocking (FCB), millimeter-wave (MMW) broad-band entrance networks.

I. INTRODUCTION

B

ROAD-BAND fixed wireless access (FWA) systems offer an alternative to cabled access networks, such as asynchronous digital subscriber line (ADSL), cable television (CATV), and fiber to the home (FTTH) because they have the capacity to address broad geographic areas without the costly infrastructure development required in deploying cable links to individual sites, and then may lead to more ubiquitous broad-band access with a low investment [1]. The broad-band FWA industry has matured to a point at which it now has the IEEE 802.16 WirelessMAN [2] standard for second-generation wireless metropolitan area networks, which has recently been updated to IEEE 802.16-2004. Its purpose is to facilitate the optimal use of bandwidth from 10 to 66 GHz, as well as interoperability among devices from different vendors. In addition, IEEE 802.16a extends the air interface support to lower frequencies in the 2–11-GHz band, including both licensed and unlicensed spectra. The progress of the standard has been studied by the keen interest of the wireless broad-band industry to capture the emerging worldwide interoperability for the microwave access (WiMax) market. The WiMax Forum, formed in 2003, promotes the commercialization of IEEE 802.16 and the European Telecommunications Standard Institute (ETSI)’s High-Performance Radio MAN (HiperMAN). At present in Japan, FWA services are mainly provided in 22-, 26-, and 38-GHz bands according to the Association of

Manuscript received March 1, 2005; revised June 14, 2005. This work was supported in part by the Japan Society of the Promotion of Science under Grants-in-Aid for Scientific Research (B) 14350202. The authors are with the Department of Communications Engineering, Graduate School of Engineering, Osaka University, Suita-shi 565-0871, Japan (e-mail: [email protected]; [email protected]. osaka-u.ac.jp). Digital Object Identifier 10.1109/TMTT.2005.856084

Radio Industries and Businesses (ARIB) standards of STD-T58 and STD-T59 [3]. In addition, FWA systems need a clear lineof-sight (LOS) between stations. However, it may be difficult to obtain LOS when the distance between stations is far. One promising way to solve the above problem is the multihop meshtopology network using directional antennas, which results in an increase in coverage area and capacity [4]. The broad-band FWA systems with a mesh configuration were proposed in [5] and [6]. Additionally, according to the tremendous increase of highspeed demand, new frequency bands should be allocated for FWA systems. As discussed during World Radio Conference 2000 (WRC 2000), several new frequency bands, e.g., 32, 52, and 55 GHz, have been reserved internationally for the highdensity fixed service (HDFS) including FWA. A new system exploiting the 32-GHz band has been developed by the BroadBand Millimeter-Wave (MMW) Wireless Access Group, Research and Development Committee Yokosuka Research Park (YRP), Japan [7], [8]. This system has a hierarchical network structure of access networks constructed with point-to-multipoint (P-MP) links for customer premises equipments (CPEs) and a higher level meshtopology entrance network constructed with gigabits per second point-to-point (P-P) links. The entrance network is used to relay traffic from several base stations (BSs) providing P-MP access links, to a center station (CS) connecting to backbone (BB) networks via P-P wireless links. The concept of the entrance network is similarly investigated in the radio access network (RAN) for the digital enhanced cordless telecommunications (DECT) [9] and the fourth-generation (4G) mobile communication [10]. In mesh-topology MMW entrance networks, the interference problem is one of the most challenging issues. We proposed the dynamic resource assignment (DRA) scheme performing the radio path allocation and the sub-optimum frequency channel assignment in [11]. Nevertheless, the throughput degradation problem occurs when traffic load becomes heavy. This is because the heavier traffic load leads more reuse of any channel among different radio links and, thus, interfere with one another stronger. Therefore, to combat the above problem, we enhance the DRA scheme by using the frequency channel blocking (FCB) scheme proposed in [12]. The proposed FCB scheme blocks the use of the frequency channel at any radio link in order to suppress interference level in other links. That is, the proposed FCB scheme sacrifices the throughput of any BS to improve the total network

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throughput performance. However, this may lead to the unfairness problem, which is one of the most important issues in multihop mesh-topology networks [13]. Therefore, in the proposed FCB scheme, any frequency channel is determined to be blocked whether or not based on not only the network throughput, but also the newly defined fairness index. The remainder of this paper is organized as follows. Section II is devoted to define throughput and fairness index considered in this paper and describe the proposed FCB scheme. Section III shows performance evaluations under various weather conditions and then discusses the effectiveness of the proposed FCB scheme. Finally, conclusions are given in Section IV. II. FCB Let us first define network throughput and fairness index as follows. Throughput is the successful received traffic rate in the network (bits per second). If the number of error bits of the received packet, whose length is 1500 B, is no more than 2 bits (maximum tolerable error bits in packet), it will be successfully received. In addition, the impact of retransmission when packet is unsuccessfully received is not considered. is newly defined based on the definition Fairness index shown in [13], and can be written as

(1)

(2)

is the transmitted power, is the antenna gain, is the propagation path loss of radio link th whose dis(in kilometers), is the wavelength of the carrier tance is frequency channel th, and are, respectively, the atmosphere absorption factor and specific rain attenuation in decibels per kilometer, is the rain rate in millimeters per hour, and and are power-law parameters, which depend on frequency, raindrop size, rain temperature, and polarization [14]. The thermal noise power at the receiver is given by

where

(7) where is the Boltzmann constant and is the temperature is the channel bandwidth, and is the noise in kelvin, figure. is the interference signal power from Assuming that radio link th using frequency channel th to radio link th, it can be written as (8) where is the directivity of the transmitting antenna of radio link th whose angle difference toward the receiving anand, similarly, is the ditenna of radio link th is rectivity of the receiving antenna of radio link th whose angle difference toward the transmitting antenna of radio link th is . is the distance between the transmitting antenna of radio link th and the receiving antenna of radio link th, and is the propagation path loss of the interference signal. as the set of radio links using frequency Let us define channel th. The received carrier-to-noise-plus-interference ratio (CNIR) of radio link th using frequency channel th can be obtained as

where

is the number of BSs in network. , , and , respectively, denote the throughput, input load, and normalized throughput of the th BS. Note that the summation of the throughput of all BSs is the network throughput, i.e., (3) Moreover, from (1), we can observe that the fairness index is bounded from 0 to 1. A higher fairness index indicates better fairness between BSs. In the case of perfect fair, in which each BS has the same value of normalized throughput, the fairness index becomes 1. On the other hand, in the case of perfect unfair, in which only one BS has nonzero throughput, the fairness index , which is 0 in the limit as tends to . becomes A. Theoretical Expression is the received carrier power of radio link Assuming that th using frequency channel th, it can be written as (4) (5) (6)

(9) When we forbid the use of frequency channel th at radio link th, which belongs to , the CNIR of radio link th using frequency channel th will become (10) Note that the blocking of frequency channel th at radio link th improves the CNIR of channel th in not only the radio link th, but also the other radio links that belong to . B. Example The example of the FCB is simply clarified as illustrated in Fig. 1, let us consider the four frequency channels network. There are three BSs transmitting traffic to CS in the mesh-topology entrance network. Note that the traffic slot , where and transmitted through network is numbered by , respectively, denote the number of traffic source BSs and the number of used frequency channels. Without consideration of the FCB, it is assumed that frequency channels are assigned as illustrated in Fig. 1(a), and the share of frequency channels between radio links BS2-CS and

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Fig. 2.

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FCB algorithm.

traffic (throughput) in the network will be increased. Moreover, the fairness index of this case becomes 0.641, which overcomes the case without consideration of the FCB. Fig. 1(c) and (d) shows the second FCB executed at radio link BS2-CS and radio link BS3-CS, respectively. Both cases yield the same total network throughput, but a different network fairness index. The second FCB executed at radio link BS2-CS leads the network fairness index of 0.667. On the other hand, the second FCB executed at radio link BS3-CS leads the network fairness index of 0.923. Therefore, we can perceive that the FCB should be executed at each radio link in a manner of round sequential to obtain a high fairness index. C. Algorithm

Fig. 1. Example of FCB. (a) Without consideration of the FCB. (b) First FCB executed at radio link BS2-CS. (c) Second FCB executed at radio link BS2-CS. (d) Second FCB executed at radio link BS3-CS.

BS3-CS yields strong interference to each other and then traffic will be lost in both radio links. On the other hand, the reuse of frequency channels between other radio link pairs yields quite weak interference. By using (1) and (2), the fairness index of the case without consideration of the FCB becomes 0.533. In contrast, with consideration of the FCB, the first FCB is assumed to be executed at radio link BS2-CS, as illustrated in used to transmit traffic Fig. 1(b). The frequency channel in this radio link is blocked. As a result, the traffic transmitted in radio link BS3-CS through the frequency channel will not be interfered with. Thus, the overall successful received

The proposed FCB is performed as detailed in the flowchart shown in Fig. 2. Let us define parameters used in this algorithm as follows: number of BSs in network; number of FCB cancellations. At the beginning, the initial value of network throughput and fairness index, when the FCB scheme is not performed, are calculated. The FCB first tries to forbid one dummy frequency channel. Note that, at each time in blocking one dummy frequency channel, the frequency channel assignment will be redone. After that, the total throughput and the fairness index of the network are calculated and, thus, respectively, set as and . Since both the network throughput and fairness index are significant, in this paper, we propose a novel criterion determining whether the blocking is permitted or not by comparing the product of network throughput multiplied by the fairness

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TABLE I PARAMETERS USED IN CALCULATIONS

Fig. 3. Analysis model under: (a) fine weather condition, (b) rainfall condition I, and (c) rainfall condition II.

index before and after the blocking has been done. If the product after the blocking has been done is more than that before the blocking is done, the blocking will be permitted. That is, the latest blocking will be accepted if the following equation: (11) is satisfied, and will be cancelled otherwise. can be If the FCB is accepted, the new value of and and , respectively. Moreover, the number written as of FCB cancellations ( ) is set to be zero. On the other hand, is added by one. if the FCB is cancelled, is still not satisfied, the algoIf the condition of rithm tries to additionally forbid one more frequency channel at another radio link. In other words, in this proposed FCB algorithm, the algorithm tries to forbid the frequency channel until there is no one blocking acceptance for all BSs, i.e., the algorithm will be terminated when the blocking is cancelled for times running. Finally, plural frequency channels could be forbidden. III. PERFORMANCE EVALUATIONS A. Analysis Model The analysis model is illustrated in Fig. 3. Eight BSs and one CS each, arranged into the 3 3 square mesh topology, are established connections with P-P links using parabolic antennas as the wireless entrance network under the LOS and adaptive white Gaussian noise (AWGN) environments. This square mesh topology is considered in order to evaluate the performance in the severe interference condition. Assume the network in Fig. 3(a) is under the fine weather condition, the network in Fig. 3(b) is under rainfall condition I, of which rain falls at the upper right corner of network, and the network in Fig. 3(c) is under rainfall condition II, of which rain falls at the center of the network.

Fig. 4. Relationship between input load and: (a) throughput and (b) fairness index under fine weather condition.

The rain rate in the rain zone is assumed to be the heavy flat rate of 45 mm/h. This causes the rain attenuation of 10 dB/km by using the calculation from the power-law relationship of an ITU-R Recommendation [14], as shown in (6), where and for the 32-GHz band [15]. Table I lists parameters used in the calculations. The adaptive modulation (quadrature phase-shift keying (QPSK), 16 quadrature amplitude modulation (QAM), 64 QAM, and 256 QAM) is also performed based on the carrier-to-noise ratio (CNR) achieving the bit error rate (BER) of 10 with a margin of 1 dB. In addition, calculations are performed in the following two systems. • System I is designed for a small-size network, which is constructed with 1-km-long P-P links using parabolic anis 28 dBi and front-to-back ratio tennas whose gain

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Fig. 5. Relationship between input load and: (a) throughput and (b) fairness index under rainfall condition I.

Fig. 6. Relationship between input load and: (a) throughput and (b) fairness index under rainfall condition II.

is 15 dB. Maximum transmitted power is set to be 28.6 dBm, i.e., 4.8 dBm/MHz. For example, transmitted power per channel is set to be 23.8 dBm when channel is 20 MHz. bandwidth • System II is designed for a larger network, which is constructed with 3-km-long P-P links using parabolic is 42 dBi and is 30 dB. Maxantennas whose imum transmitted power is set to be 30.4 dBm, i.e., 3 dBm/MHz, i.e., transmitted power per channel is set is 20 MHz. to be 25.6 dBm when

Fig. 4(a) shows the throughput performance in the case of fine weather condition. In system II, it is obvious that even if the FCB scheme is not applied, a quite good throughput can be obtained, which is degraded slightly when compared to that of the idealized optimum. Meanwhile, the use of the FCB scheme gives very little improvement of throughput over the conventional scheme without FCB. In contrast, in system I, we can observe that the throughput of the case without FCB first increases and reaches a peak at an input load of 9 Gb/s and then starts to decrease slightly if the input load increases further. On the other hand, the FCB yields throughput improvement approximately 2 Gb/s over the conventional scheme without FCB when the input load is close to 12 Gb/s. Fig. 4(b) shows the fairness index performance in the case of fine weather condition. We can see that the fairness index of both the conventional and proposed schemes using FCB are almost the same, which is close to 1 in system II. In contrast, in system I, it is obvious that the scheme without FCB remains with the fairness index close to 1 when the input load is less than 9 Gb/s, and yields the deteriorated fairness index when the input load exceeds 9 Gb/s. On the other hand, the fairness index of the case using the FCB scheme is close to 1 even if input load becomes more than 9 Gb/s, which is almost the same as that of the idealized optimum. From Fig. 5(a), in the case of the rainfall condition I, it is clear that the results are similar to those under the fine weather condition shown in Fig. 4(a). The FCB improves the throughput only in

B. Results Throughput and fairness index performances of conventional scheme investigated in [11] are compared to those of the case using the proposed FCB scheme. In addition, performances of case idealized optimum, searching the frequency channel assignment pattern that gives the best throughput performance, are also compared. The calculations of the case idealized optimum are performed in system II, but use different parabolic antennas and are 42 dBi and whose beamwidth is close to 0 , and , respectively. 1) Throughput and Fairness Index Performances Versus Input Load: Let us first evaluate performances. In the case of uniform traffic and number of total frequency channels, is 36. The results under fine weather condition, rainfall condition I, and rainfall condition II are shown in Figs. 4–6, respectively.

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Fig. 7. Relationship between number of total frequency channels (a) throughput and (b) fairness index under fine weather condition.

N

and:

system I. In addition, we also observed that the throughput performance of this rainfall condition I is slightly degraded when compared to that of the fine weather condition. Fig. 5(b) shows the fairness index performance of the case rainfall condition I. In system II, the fairness index of both cases with and without FCB are almost the same, which is close to 1. In contrast, in system I, the fairness index performance degradation of the case conventional scheme using no FCB when input load exceeds 9 Gb/s in the case of rainfall condition I is more noticeable compared to the case of the fine weather condition. On the other hand, the fairness index of the scheme using FCB is close to 1 even if input load becomes more than 9 Gb/s. Fig. 6(a) shows the throughput performance in the case of rainfall condition II. In system I, the scheme using FCB yields the throughput improvement of approximately 3 Gb/s compared to the conventional scheme without FCB when the input load is close to 12 Gb/s. That is, the case when FCB is applied more outperforms the case when FCB is not applied in this rainfall condition II compared to the other weather conditions. In contrast, in system II, the throughput performance of this rainfall condition II is almost the same as that of the other weather conditions. From Fig. 6(b), in the case of rainfall condition II, it is obvious that the fairness index of all schemes are almost the same, which is close to 1 in system II. In contrast, in system I, the fairness index performance degradation of the case without FCB when input load exceeds 9 Gb/s in the case of rainfall condition II is more noticeable compared to the case of rainfall condition I. In

Fig. 8. Relationship between under rainfall condition I.

N

and: (a) throughput and (b) fairness index

addition, the fairness index of the case using FCB decreases a little if the input load exceeds 9 Gb/s. Moreover, from Figs. 4–6, it is clear that the throughput and fairness index improvements by the FCB scheme in the case of the fine weather condition, rainfall condition I, and rainfall condition II are, respectively, more evident. 2) Throughput and Fairness Index Performances Versus Number of Total Frequency Channels: Next we evaluate performances versus the number of total frequency channels in the case of uniform traffic in system I. Note that each link channels for transmitting traffic. In possibly uses maximum fact, the number of channels used in each link depends on traffic. The results under fine weather condition, rainfall condition I, and rainfall condition II are shown in Figs. 7–9, respectively. From Figs. 7(a), 8(a), and 9(a), it is clear that when the FCB is not applied, the throughput performance is dependent on and the weather condition. In the case of the fine weather conof 12 and 24 yields the degraded dition shown in Fig. 7(a), throughput performance since interference becomes larger as decreases. Meanwhile, in the case of rainfall conditions I and II, shown in Figs. 8(a) and 9(a), respectively, of 36 yields the best throughput performance compared to the other cases of (12, 24, and 48) when the input load is more than 9 Gb/s. This is because, in our analysis models, traffic load was disis 48 by the radio path allocation protributed too well when posed in [11]. It thus yielded much inverse-direction-propagated wave, which causes strong interference. Finally, we obtained results opposite to those we have intended, especially in both rain

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Fig. 9. Relationship between under rainfall condition II.

N

and: (a) throughput and (b) fairness index

conditions. In contrast, when using the FCB scheme, since the effect of interference can be well mitigated, the throughput performance is almost the same in all cases of the number of frequency channels and the weather condition. From Figs. 7(b), 8(b), and 9(b), it is obvious that, in the conventional scheme without FCB, the fairness index performance of 12 compared to the other is much degraded in the case of (24, 36, and 48) in all weather conditions. On the cases of other hand, when the FCB is applied, the fairness index perforand weather condimance is almost the same in all cases of tion, which is close to 1. Next, performance evaluations are done in the case of nonuniform traffic, of which input load of each BS is generated randomly from 1 to 1.25 Gb/s. Note that the total input load of network varies from 8 to 10 Gb/s, and its average becomes 9 Gb/s. Fig. 10(a) and (b) shows, respectively, the throughput performance and fairness index performance versus the number of in system I. When the FCB is not total frequency channels applied, we can observe that both the throughput and fairness . In addition, index performances are improved as increasing the case using no FCB yields the best throughput and the best fairness index performances in the case of fine weather condition, and yields the worst throughput and the worst fairness index performances in the case of rainfall condition II. In contrast, when using the FCB scheme, the throughput per. In addition, the formance is almost the same in all cases of throughput performance is almost the same in the cases of fine weather and rainfall condition I, and is a little better than that

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N

Fig. 10. Relationship between and: (a) throughput and (b) fairness index under nonuniform traffic condition, of which input load varies from 8 to 10 Gb/s. TABLE II COMPUTATIONAL COMPLEXITY NORMALIZED BY THAT OF CASE USING CONVENTIONAL SCHEME WITHOUT FCB

in the case of rainfall condition II. Moreover, we can see that the fairness index performance is almost the same in all cases of the number of frequency channels and the weather condition, which is close to 1. 3) Computational Complexity Comparison: Finally, we examine the computational complexity in the processing of frequency channel assignment of the conventional scheme, the proposed scheme using FCB, and the case of idealized optimum, which are in order of , , and , denotes the number of frequency chanrespectively, where denotes nels, i.e., traffic slots, required for all radio links, denotes the the number of total frequency channels, and number of BSs in the network. The computational complexity of the case using FCB and the case of idealized optimum, for example, are determined in the of 36 and input load of 10.25 Gb/s, and then normalcase of ized by the computational complexity of the case using a conventional scheme without FCB. The results are shown in Table II. It is clear that the addition of computational complexity by using FCB is very small compared to the case of an idealized optimum.

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IV. CONCLUSION This paper has proposed the FCB scheme, performed based not only on the network throughput, but also the newly defined fairness index for MMW entrance networks. As shown by the results in system I, when the input load is close to 12 Gb/s, the FCB scheme yields the network throughput improvement of approximately 2–3 Gb/s over the conventional scheme without FCB, and obtains a fairness index of nearly 1. Additionally, performances are dependent on the number of total frequency chanand the weather condition when using the conventional nels scheme. The use of the FCB scheme yields almost the same . Moreover, the use of the FCB performance in all case of scheme leads to a small increase in computational complexity over the conventional scheme. Our proposed scheme is simply applicable to MMW entrance networks by installing the algorithm only in the CS of networks. Unfortunately, the efficiency of the proposed scheme was just shown by simulation results. The effect in a real system will be shown in future studies.

[12]

, “Frequency channel blocking for MMW entrance networks,” IEICE Electron. Express (Japan), vol. 2, no. 1, pp. 19–24, Jan. 2005. [13] D. M. Chiu and R. Jain, “Analysis of the increase and decrease algorithms for congestion avoidance in computer networks,” J. Comput. Networks ISDN Syst., vol. 17, pp. 1–14, Jun. 1989. [14] “Specific attenuation model for rain for use in prediction methods,”, ITU-R Recommendation P.838, 1992. [15] M. Ishida et al., “Development of gigabit millimeter-wave broad-band wireless access system (II)—(4) Estimation for probability distribution of rain attenuation at 32 GHz band,” in Proc. 6th Topical Millimeter Waves Symp., Feb. 2004, pp. 217–220.

Jaturong Sangiamwong (S’03) was born in Chachoengsao, Thailand, on June 1, 1980. He received the B.E. degree in electrical engineering from Chulalongkorn University, Bangkok, Thailand, in 2000, and the M.E. and Ph.D. degrees in communications engineering from Osaka University, Osaka, Japan, in 2002 and 2005 respectively. His research interests include broad-band wireless communication systems.

REFERENCES [1] W. Webb, “Broadband fixed wireless access as a key component of the future integrated communications environment,” IEEE Commun. Mag., vol. 39, no. 9, pp. 115–121, Sep. 2001. [2] IEEE Standard for Local and Metropolitan Area Networks—Part 16: Air Interface for Fixed Broad-Band Wireless Access Systems, IEEE Standard 802.16-2001, 2002. [3] “Fixed wireless access systems development in ARIB,” ARIB, Study Rep., Jun. 1999. [4] P. Whitehead, “Mesh networks: A new architecture for broad-band wireless access systems,” in Proc. IEEE RAWCON, May 2000, pp. 43–46. [5] Y. Kishi, K. Tabata, S. Konishi, and S. Nomoto, “A proposal of multihop mesh network architecture featuring adaptive network control for broadband fixed wireless access systems,” Electron. Commun. Jpn. 1, vol. 87, no. 6, pp. 22–33, Sep. 2004. [6] D. Uchida, M. Sugita, I. Toyoda, and T. Atsugi, “Mesh-type broad-band fixed wireless access system,” NTT Tech. Rev., vol. 2, no. 1, pp. 44–54, Jan. 2004. [7] H. Ogawa, “Millimeter-wave indoor/outdoor wireless access systems and their technologies,” in Proc. IEEE RAWCON, Aug. 2003, pp. 5–8. [8] K. Tsukamoto et al., “Development of gigabit millimeter-wave broadband wireless access system (II)—(1) System review,” in Proc. 6th Topical Millimeter Waves Symp., Feb. 2004, pp. 42–45. [9] M. Celidonio and D. D. Zenobio, “A wide-band two-layer radio access network using DECT technology in the uplink,” IEEE Commun. Mag., vol. 37, no. 10, pp. 76–81, Oct. 1999. [10] T. Otsu, Y. Aburakawa, and Y. Yamao, “Multi-hop wireless link system for new generation mobile radio access networks,” IEICE Trans. Commun. (Japan), vol. E85-B, no. 8, pp. 1542–1551, Aug. 2002. [11] J. Sangiamwong, K. Tsukamoto, and S. Komaki, “Dynamic resource assignment scheme in mesh-topology millimeter-wave broad-band entrance networks,” IEICE Trans. Fundamentals (Japan), vol. E87-A, no. 10, pp. 2668–2675, Oct. 2004.

Katsutoshi Tsukamoto (M’87) was born in Shiga, Japan, on October 7, 1959. He received the B.E., M.E., and Ph.D. degrees in communications engineering from Osaka University, Osaka, Japan, in 1982, 1984, and 1995, respectively. He is currently an Associate Professor with the Department of Communications Engineering, Osaka University, where he is engaged in research on radio and optical communication systems. Dr. Tsukamoto is a member of the Institute of Television Engineers of Japan (ITE). He was the recipient of the 1996 Paper Award presented by the Institute of Electrical, Information and Communication Engineers (IEICE), Japan.

Shozo Komaki (SM’00) was born in Osaka, Japan, in 1947. He received the B.E., M.E., and Ph.D. degrees in electrical communication engineering from Osaka University, Osaka, Japan, in 1970, 1972, and 1983, respectively. In 1972, he joined the NTT Radio Communication Laboratories, where he was engaged in repeater development for a 20-GHz digital radio system and 16and 256-QAM systems. In 1990, he joined the Faculty of Engineering, Osaka University, where he became engaged in research on radio and optical communication systems. He is currently a Professor with Osaka University. Dr. Komaki is a member of the Institute of Electrical, Information and Communication Engineers (IEICE), Japan and the Institute of Television Engineers of Japan (ITE). He was the recipient of the 1977 Paper Award and the 1994 Achievement Award presented by the IEICE.

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Effect of Cable Length in Vector Measurements of Very Long Millimeter-Wave Components Alessandro Simonetto, Ocleto D’Arcangelo, and Lorenzo Figini

Abstract—In millimeter-wave applications, it is sometimes required to measure waveguide components that are several hundred wavelengths long. Depending on the vector network analyzer calibration technique used, the distance between the generator and receiver can be much longer than the one used during calibration when the device-under-test is inserted. If the frequency is swept or drifting, a systematic error is known to appear. A simple way of correcting the result combining two measurements was found. Index Terms—Cables, delay effects, measurement errors, millimeter-wave measurements.

I. INTRODUCTION HE vector network analyzer (VNA) owned by the authors (AB Millimetre VNA1) uses a patented scheme [1], [2] to make millimeter-wave vector measurements with a pair of low-frequency oscillators, whose frequency difference is phase locked to a reference using multipliers and harmonic mixers for millimeter-wave generation and detection. The calculations shown here are made for this VNA, but the results apply to any measuring setup using two oscillators with a heterodyne detection, including VNAs operated in swept frequency mode. Widely used VNAs using millimeter-wave extension kits can also be prone to this effect. When very large components are measured, as the waveguides for the low-frequency instrument of the Planck satellite,2 [3], now under test at the Istituto di Fisica del Plasma (IFP), Milan, Italy [4], [5], the path difference between the deviceunder-test (DUT) and calibration standards can be very large unless a path-preserving calibration technique is used, such as the one shown in [6]. This leads to measurement errors if the frequency stability of the master oscillator is inadequate or the VNA operates in swept frequency mode, as described in [7]. The solutions suggested in [7], reducing the sweep rate or compensating the delay with an additional cable, may sometimes not be applicable with ease. This paper models the error and shows an alternative way to correct the data combining two measurements with cables of different length.

T

Manuscript received May 6, 2005; revised June 9, 2005. The authors are with the Istituto di Fisica del Plasma, Consiglio Nazionale delle Ricerche, 20125 Milan, Italy (e-mail: simonetto@ ifp.cnr.it; [email protected]; [email protected]). Digital Object Identifier 10.1109/TMTT.2005.856085 1AB Millimetre, Paris, France. [Online]. Available: http://www.abmillimetre.com 2[Online]. Available: http://astro.estec.esa.nl/Planck/

Fig. 1. Idealized scheme of the measurement setup. A single reflectometer configuration is used because it increases the effective length of the DUT, enhancing the effect under consideration. X is the master oscillator, O is the slave oscillator, 3 is a times 3 multiplier, bandpass filter is denoted as BPF, and the vector receiver is denoted as VR. , , and are the delays experienced along the three paths.

2

II. THEORY A. Simplified Scheme of the Instrument Fig. 1 shows the model scheme analyzed here. This figure depicts a single reflectometer configuration, with a one-port DUT, but the model is not limited to that. As a simplification, we assume that only a response correction is applied, i.e., the transmission between ports is derived from the ratio of a measurement with the DUT and one with an ideal short (or a thru connection for transmission measurements in a double reflectometer configuration). Unless one uses a calibration technique that keeps the transmission-line length between transmitter and receiver nearly constant at all times, any other method will be equivalent to the simple response correction in this idealized model. The signal generated at the transmitter ( in Fig. 1) is delayed by time along the transmission line to the multiplier (if any) and further along the waveguide to the reference plane of the transmitting port, and possibly from the reference plane of the receiver port to the mixer (the two planes are coincident in the one port configuration shown). The signal generated at the local oscilto the mixer. That lator (LO) ( ) is similarly delayed by time originated at the oscillator is further delayed in transit along the DUT by time . The IF output of the mixer goes through a bandpass filter (BPF in Fig. 1) and then to the vector receiver. The mixer can also be a harmonic one, and a multiplier or comb generator can be placed along the transmission line from the transmitter to the DUT. A times 3 multiplier is shown in Fig. 1. The model can account for that with minor modifications to the algebra. The VNA we consider operates in a stepped-frequency mode, but the generator can take tens or hundreds of milliseconds to

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reach the set frequency so there is a residual frequency drift. The same would apply, of course, for a VNA operated in a sweptfrequency mode. Waiting for the generator to stabilize would solve the problem. However, measurements of long components require narrow frequency steps over a wide bandwidth to achieve sufficient resolution and avoid aliasing when Fourier transforming to perform time-domain analysis. For the waveguides we need to test, this would cause the measurements to last longer than the time stability of the VNA calibration.

We assume that the bandpass filters’ transfer function can be approximated around its central frequency as

(8) where is an effective halfwidth. The validity of this approximation is essential for the subsequent steps. This is granted if the bandpass filter is sufficiently narrow. The measured quantity is the Fourier transform of (4) times

B. Analytic Description Among the many possible conventions on Fourier transforms, we used the definition (1)

(9) that can be written explicitly as a sum of integrals of the type (10)

(2) with . The requirement that the DUT transfer function sents a real component dictates that therefore, it can be written as

where

and

are defined as

repre;

(11) (12)

(3)

The only nonnegligible terms are

where and are even functions of frequency. represents the time delay along the DUT, and is a function of frequency for dispersive devices. The mixer output can be written as the product of LO and RF signals (4) is the LO waveform, is the generator wavewhere form, and, as usual, denotes convolution. The base frequency is switched in steps, and the vector receiver output is averaged over the time interval. By assumption, a small frequency drift is present. Expanding the argument of the oscillators’ sinewave in the neighborhood of instant and retaining only first-order terms in time, one gets (5) hence, the two oscillators’ spectra can be written, neglecting the constant phase shift as (6)

(13) with

, the measurement frequency, defined as (14)

Equation (13) is proportional to the real part of . The vector receiver processes it to separate amplitude and phase information. One can describe that by keeping only the “positive as follows: frequency” part of (13) and discarding

(15) Applying calibration in the form of response correction , as given by (15), to a measuremeans taking the ratio of ment taken in identical conditions with (hence, ). This results in

(7) By assumption, , where is the center of the since passband of the filter. Moreover, the frequency difference between the oscillators is locked to a common reference. We assume that it is constant in time and drop the subscript. The measured quantity is the IF output of the mixer, as given by (4), convoluted with the impulse functions of the bandpass filter and the vector receiver.

(16)

This gives the correct result only under the assumption that is very small (short device) or that the delay is compensated as suggested in [7] so that (17)

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almost exactly. This can be impossible to achieve with cables if the DUT is dispersive, or simply the correct length of low-loss cable may not be available. C. Correction of Data If the measurement is repeated twice with cables of different, but arbitrary length, one can extract the correct information from the combination of measurements even if condition (17) is not met. Of course, the result is more precise if the measurements are as close as possible to the ideal condition. Assuming that (18) Fig. 2. Measured magnitude of the S scattering parameter when the DUT is terminated with a short. The thin curves are labeled with the difference in cable length. Between 1 and 3 m, the increment in cable length is 1 m per curve. The thick curve (almost overlapped to the 3-m one) shows corrected data, computed using (22).

one can rewrite (16) as

0

0

(19) and one can easily see that

(20) If the DUT is dispersive, the (and all the ’s) are functions of frequency. The model predicts that the phase measurement is not affected by the systematic error, at least to first order, therefore, it can be used to measure the group velocity in the DUT, and also its dispersion if a model fitting function (e.g., ideal waveguide transmission) is available. can be estimated easily if the group Knowing the above, velocity in the cables is known with the required approximation, which is normally true. III. MEASUREMENTS The concept was tested using the scheme shown in Fig. 1. The VNA is configured as a one-port reflectometer, and the DUT (a WR28 waveguide 1.78-m long, made with several pieces) is terminated with a short to measure its insertion loss. The quantiand can be varied using cables of different length ties between the VNA (containing the oscillators) and the multiplier and mixer. Measurements were taken with most combinations of the available cables. The quantity defined in (18) can be written as (21) where and are, respectively, the path lengths in waveguide along the directional coupler (1.09 m, including all components along the path from the multiplier to the mixer port) and and along the DUT (twice the physical length), and are the group velocities in the waveguide and cables. The quanis defined as the length of the cable from generator to tity

Fig. 3. Difference between individual phase measurements and their average (excluding the 5-m record). Curves are labeled with the difference in cable length. The phase shift is mostly frequency independent.

0

multiplier ( ) minus the length of the cable from LO to mixer ( ). from 1 to 5 m. The measurements were made varying scattering paFig. 2 shows the measured amplitude of the ) as a function of frerameter (i.e., the square of the DUT quency. As one can see, there is a significant and monotonic variation with the cable path difference. All measurements were made using a calibration with fixed short, variable short, and variable load. The calibration was repeated for each measure, 5 m has very ment. The measurement with the largest large noise-like (but reproducible) features, which suggests that it is close to the acceptable limit. Fig. 3 shows, in the same cases, the difference between the measured phase and its average over the measurements (excluding the 5-m measurement from the , nearly frequency average). There is indeed a trend with independent, but it is fairly small and can be neglected at first order. The DUT length estimated by fitting the experimental phase-versus-frequency curves has no significant dependence on the cable path difference so it can be used to estimate and correct the data with a pair of measurements using (20). If more than two measurements are available, one can also take the

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D. F. Williams, National Institute of Standards and Technology (NIST), Boulder, CO, for his useful comments. REFERENCES

Fig. 4. Comparison between amplitude measurements corrected using the best fit over the set of data [using (22)] and a worst case obtained combining a pair of measurements [using (20)]. Full correction is achieved even if the 5-m initial data were not very good. All the other 14 combinations of the six measurements are of equivalent or better quality.

0

[1] P. Goy, “Antenna vector characterization in the mm- and submm-wave regions,” Microwave J., p. 98, Jun. 1994. [2] P. Goy, M. Gross, and J. M. Raimond, Proc. 15th Int. Infrared and Millimeter Waves Conf., Orlando, FL, 1990, pp. 172–174. [3] N. Mandolesi et al., “Low frequency instrument of Planck,” in Proc. SPIE IR Space Telescopes Instrum., vol. 4850, 2003, pp. 722–729. [4] A. Simonetto et al., “Millimeterwave tests on passive components of Planck–LFI,” in Proc. Joint 29th Int. Infrared and Millimeter Waves/12th Int. Terahertz Electronics Conf., Karlsruhe, Germany, 2004, pp. 435–436. [5] A. Simonetto et al., “Millimeterwave techniques for fusion plasmas and for experimental cosmology,” in Proc. AIP Plasmas in the Laboratory and in the Universe—New Insights and New Challenges Conf., vol. 703, Como, Italy, 2003, pp. 455–458. [6] I. Rolfes and B. Schiek, “LRR—A self-calibration technique for the calibration of vector network analyzers,” IEEE Trans. Instrum. Meas., vol. 52, no. 2, pp. 316–319, Apr. 2003. [7] “10 hints for making better network analyzer measurements,” Agilent Technol., Palo Alto, CA, Applicat. Note 1291-1B, 2001.

logarithm of (19) and fit the measurements with the resulting relation, which is linear in , as follows:

(22) The result of this last procedure (again excluding data with m) is shown by the thick curve in Fig. 2. It nearly m curve because, in that case, overlaps with the is very close to 0.5 over the whole frequency range so condition (17) is met. Fig. 4 shows the same (corrected) data together with the results obtained using a pair of measurements combined using (20). All possible combinations of the available were computed, and one of the worst cases, including the low-quality 5-m data, are shown in this figure. As one can see, was completely removed. the large dependence on IV. CONCLUSIONS While all these calculations were made in the case of the AB Millimetre VNA, the formal approach also applies to sweptfrequency VNAs, and our results show a simple way to recover the measurements’ accuracy should the suggestions of [7] be impracticable. The method we found allows to correct the error caused by swept frequency combined with the unequal path difference between calibration and measurement setups. ACKNOWLEDGMENT The authors are grateful to the anonymous reviewers of this paper’s manuscript and Associate Editor of this TRANSACTIONS,

Alessandro Simonetto received the Physics degree from the Universita’ degli Studi di Milano, Milan, Italy, in 1984. Since then, he has been involved with electron cyclotron emission or electron cyclotron resonance heating experiments in tokamaks with the Istituto di Fisica del Plasma, Consiglio Nazionale delle Ricerche (CNR), Milan, Italy. His current research interests include precision millimeter-wave measurements.

Ocleto D’Arcangelo received the Physics degree from the Universita’ degli Studi di Milano, Milan, Italy, in 2000. He is currently with the Istituto di Fisica del Plasma, Consiglio Nazionale delle Ricerche (CNR), Milan, Italy. His research is related to the characterization of microwave passive components for cosmic microwave background (CMB) observation.

Lorenzo Figini was born in Milan, Italy, in 1980. He received the Physics degree from the Universita’ degli Studi di Milano, Milan, Italy, in 2004. He is currently with the Istituto di Fisica del Plasma, Consiglio Nazionale delle Ricerche (CNR), Milan, Italy, where he is involved in the characterization of passive components of Planck–LFI radiometers.

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Adaptive Power Controllable Retrodirective Array System for Wireless Sensor Server Applications Sungjoon Lim, Student Member, IEEE, Kevin M. K. H. Leong, Member, IEEE, and Tatsuo Itoh, Fellow, IEEE

Abstract—An adaptive power controllable retrodirective array system is presented. It is able to conserve battery power in an idle mode and wake up only when it needs to operate, extending the array system’s lifetime. One application of this technology is for use as wireless sensor servers, which act as a relay point between wireless sensors and remote data collectors. The proposed retrodirective array is fabricated and tested at 5.8 GHz and uses an integrated rectenna and an analog switch, which controls a battery power source. When an RF signal is received by the antenna array, it is split between a rectenna and receiver (RX), where most power is sent to a rectenna. The collected dc voltage wakes up the system by activating a switch connected to a battery and the RX. When there is no interrogation, the switch turns off. Furthermore, the second and third harmonic rejection characteristic of a circular sector antenna is introduced so that it makes the system simpler by eliminating a low-pass filter in the rectenna. For the phase-conjugation retrodirective array, second subharmonic mixers are used by employing antiparallel diode pairs, which enables avoiding expensive high-frequency oscillators. It is experimentally demonstrated that the retrodirective array system with the proposed power management can retransmit the received signal toward the source when the received power is greater than 8.5 dBm. Application of the retrodirective array system as a multifunctional RX array is also investigated. Index Terms—Circular sector antenna, harmonic rejection, portable device, power management, rectenna, retrodirective array, wake-up, wireless sensor.

I. INTRODUCTION

R

ECENTLY, use of wireless sensor networks are widely expanding toward military security, outdoor monitoring, and space-exploring applications [1]–[3]. Such sensors are used for data collection or surveillance. The wireless sensor network typically requires autonomous operation in monitoring and data processing. For outdoor applications, a remote sensor usually needs to be placed in inaccessible places such as on mountain top, jungle, or a deep-space environments. For these conditions, data retrieval from the sensors themselves becomes a challenge. One technology that has been proposed to operate in conjunction with such sensors are retrodirective antenna arrays [4] to be used as a data collection device. Retrodirective array systems are able to respond to an incoming signal by returning a signal back toward the source without a priori knowledge of its location [4]–[9]. This self-beam-steering capability is useful in several mobile applications such as RF identifications (RFIDs) Manuscript received May 9, 2005; revised June 26, 2005. This work was supported by Sony MICRO. The authors are with the Microwave Electronics Laboratory, Department of Electrical Engineering, University of California at Los Angeles, Los Angeles, CA 90095 USA (e-mail: [email protected]). Digital Object Identifier 10.1109/TMTT.2005.856086

Fig. 1.

Conventional wake-up system with a microprocessor.

and autonomous systems. In the application scenario proposed in [4] where the retrodirective array acts as a wireless sensor server , the array is used to locally collect data from various sensors and then it is remotely interrogated to extract this collected data making use of the array’s automatic beam-steering function. Furthermore, a wireless sensor server is typically required to provide multifunctional features such as data collection and processing. Currently, one issue related to RF sensors is the lifetime of the portable device. It is desirable to prolong the battery life for energy-limited environments [1]–[3]. A long operating lifetime of a battery is a critical feature not only in a wireless sensor network area, but also every mobile wireless transceiver [10]. There has been much effort to increase battery life time from a material level [11] to a circuit level [12]. Currently, lithium–ion batteries are commonly used and electrochemical engineers try to find better battery materials. For integrated-circuit (IC) designers, a drop-out voltage is a key parameter to design a circuit since power consumption in CMOS is pro, where is a clock frequency, is a portional to is a supplied voltage. It is also desirable capacitance, and for a portable device to have a prolonged battery life given a size-limited battery and circuit components. Thus, efficient power-management schemes have recently become important in a system level [13]. The well-known solution for efficient power management is a wake-up system [14]–[16]. It can save power by activating the circuit or system only when it needs to operate while it is turned off at other times. A typical wake-up system is shown in Fig. 1. It periodically wakes up from a sleeping mode by using an RC oscillator as a clock generator. Energy can be efficiently saved by properly manipulating the duty period [15]. However, it is not effective when a system is rarely operating since it still consumes power. An alternative approach to overcome this problem is to wake up from a standby mode in which the system consumes lower power than one in active mode. However, these implementations require a complicated microprocessor, as shown in Fig. 1. Recently, Gu and Stankovic reported the novel wake-up system at 433 MHz with a radio-triggered circuit in which a rectified radio signal triggers the sensor network [17].

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Fig. 2. Example of a retrodirective system as a wireless sensor with a block diagram of the proposed concept.

Thus, the network can eliminate the wake-up period and it can be woken up when it needs to perform. This paper introduces the development of a retrodirective array with an integrated adaptive power-management scheme. This adaptation will allow the retrodirective array used for data collection and retrieval to be more power efficient. The proposed system is able to adaptively control system power consumption and, therefore, have prolonged battery life [18]. A similar wake-up system concept used in RF sensors is adopted for use in the retrodirective array. The basic concept can be explained by the block diagram shown in Fig. 2. The system consists of a rectenna, retrodirective array components, an analog switch, and a battery. The rectenna converts RF power into dc power [19] and the converted dc voltage is used to activate the system by way of a switch. In this system, only when an RF signal is received, the system is connected to the battery. Thus, it can avoid wasting power while the array is not active. By adopting this power-management scheme into a retrodirective array system, we can enhance the performance of the system in terms of energy efficiency, which is especially important for mobile portable applications. Integration of the two technologies is beneficial to both the dc voltage extraction and the retrodirective function. Since the retrodirective system is implemented in an array configuration, the amount of collected dc power can be scaled with the number of elements if a rectenna is embedded in each one. Additionally, the retrodirective system only operates when it receives an interrogation signal, and at all other times it is turned off and simply waiting to be interrogated. This makes a retrodirective array a good candidate for this adaptive power-management scheme. Furthermore, the retrodirective array may also be operated as a receiving array, as proposed in [4]. This multifunctionality allows it to operate as a data collection and storage unit for local sensors, as well as a relay to remote interrogators. Therefore, the operation of the rectenna power-management scheme is investigated under communication scenarios. This paper is organized as follows. The principle of a proposed system is explained in Section II. The implementation of the retrodirective array is described in Section III. In Section IV, the rectenna and trigger schematics are presented and discussed. The performance of the system is demonstrated in Section V. Finally, conclusions are discussed in Section VI.

Fig. 3. Proposed system schematic.

II. SYSTEM OVERVIEW Retrodirective arrays only operate while being interrogated by a remote source. In addition, rectennas are used to extract dc power from a received RF power beam. These two conditions make integration of these two technologies ideal. A schematic of this proposed retrodirective/power extraction system is shown in Fig. 3. A single circular sector antenna is employed to feed both the phase conjugators used for retrodirective transmission and the rectenna. A CMOS switch used for array activation is positioned between a battery and bias circuitries of active devices. In this paper, only bias networks of low-noise amplifiers (LNAs) are controlled; however, bias of the local oscillator (LO) can be further controlled. The entire system does not rely on the extracted dc power from the rectenna for operation. The recovered power is rather used as a trigger signal to activate a switch to the onboard battery supply. This reduces the amount of power that must be sent by the interrogator and, therefore, increases range of interrogation. This will be quantified in Section IV. The operating process is explained in the following. The system is initially disconnected from the battery so that it does not consume power. When a carrier signal is received, it is divided to the retrodirective system and a rectenna with a coupled-line coupler. The converted dc power generated by the rectenna activates a switch. When the activating voltage is greater than a threshold voltage of the switch, the switch becomes closed and dc power is supplied to active devices (active mode). If the signal is turned off or the RF signal strength is lower than a certain power level, the rectified voltage becomes lower than a threshold voltage so that the system is returned back to an initial condition (idle mode). The dual use of the receiver (RX) antenna for power delivery to the phase conjugator, as well as the rectenna, allows for close layout integration of the two systems. Both systems benefit from an increased number of antenna elements. The retrodirective

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Fig. 4. Principle of the phase-conjugating retrodirective array. In the proposed = 5:79 GHz, f = 5:81 GHz, and f = 2:9 GHz. system, f

array will have a more directive beam, while the rectennas will be able to collect a larger amount of power assuming that an efficient power-combining scheme is used. In dc power combining, the received RF power is first converted to dc power by individual rectennas and then the rectified dc powers from each rectenna are combined. Alternatively, in RF power combining, received RF power is first combined by power combiners and then rectified by a single rectenna. DC power combining is basically preferable over RF power combining since RF signals incur transmission-line loss. However, this is under the condition that enough RF power can be collected to drive the rectifier diode for efficient rectification. III. RETRODIRECTIVE ARRAY USING PHASE CONJUGATION The phase-conjugating retrodirective array is implemented by way of heterodyne mixing. The principle is illustrated in Fig. 4. In this case, a two-step mixing process is used for phase conjugation. This is different from phase-conjugation mixers, which use an LO frequency that is twice the RF [8]. In this case, phase conjugation is done by first down-converting the incoming RF with an LO frequency slightly higher than the RF. At this point, the lower sideband of this mixing process is actually the phase-conjugated form of the RF. In order to transmit only the phase-conjugated signal, the down-converter is followed by a low-pass filter (LPF), which can filter out undesired signals including the incoming RF signal and LO leakage, and a second mixer is used to up-convert the phase-conjugated signal back to the RF band. This is summarized by the following set of equations:

(1)

Fig. 5. (a) Prototype. (b) Conversion-loss graph of up/down phase conjugators.

In the real implementation of the phase-conjugation mixers, subharmonic mixers [8], [9] are used. Thanks to antiparallel diode pairs (APDPs) used in the mixer, a subharmonic LO frequency at half of the RF frequency is needed for proper down-conversion and up-conversion. Since this approach does not need a high-frequency LO source, it provides a simpler and more economical solution at microwave and millimeter-wave frequencies. An additional benefit for this phase-conjugation scheme is that it is easily adaptable to act as a multifunctional system. Specifically because the first mixer is used for RF down-conversion to baseband, the exact same hardware can be used as a normal RX array. Furthermore, the up-conversion mixer can be used as part of a conventional transmitter (TX). In the proposed system, we choose 5.79 GHz as the receiving RF frequency and 2.9 GHz as the subharmonic LO frequency. The down-converted IF (10 MHz is again up-converted to 5.81 GHz) and undesired signals are then eliminated by an LPF. The prototype of up/down converters is shown in Fig. 5(a). Agilent beam lead Schottky diode pairs (HSCH-5531) are used. The conversion loss of phase-conjugating mixers are plotted in Fig. 5(b). Since the mixers are using second harmonics, the measured conversion loss is over 19.2 dB. The up-converters and down-converters are optimized with an LO power of 5 dBm. In this system, four pairs of phase conjugators are implemented for a retrodirective array. The conjugated phases from the four elements are transmitted and a beam direction is determined by an array factor given by (3)

(2) where

.

where

and

is the distance between elements.

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fore, a critical concern of rectenna design is a conversion efficiency, which is defined by

(4)

Fig. 6.

Layout of the proposed power-management schematic.

In order to compensate for propagation loss and conversion losses of up/down-converters, LNAs are positioned before down-converters and after up-converters. These LNAs are consuming power even if the retrodirective array system is not interrogated. Therefore, a power-management scheme is required for the efficient battery operation. IV. APPLICATION OF ADAPTIVE POWER MANAGEMENT USING TRIGGERING SCHEME The proposed power-triggering schematic is shown in Fig. 6. This power-management circuit simply consists of a rectenna, which is used to extract dc power from incoming RF power and a switch. First, received RF power is divided to a power-management circuit and an RX system by a coupled-line coupler. Since the rectenna in the power-management circuit requires much more driving power than an RX system, 97% of input power is sent to a rectenna through a 15-dB coupler and the RX uses only 3%. It should be noted that this power ratio is kept constant, while a switch is closed during interrogation. The performance of the integrated rectenna used in this system is crucial for overall system performance. A conventional rectenna consists of an antenna, a first LPF, a rectifying diode, and a second LPF. The first LPF suppresses higher harmonic frequencies while passing a fundamental frequency ( ). The second LPF is used for passing the rectified dc signal while suppressing spurious harmonics generated by a diode. The antenna type is chosen depending on application, such as a dual-frequency operation [20], broad-band operation [21], or circular polarization [22]. Since the rectenna will be integrated with the retrodirective array system, it should be planar and compact while keeping high efficiency. A circular sector antenna satisfies these primary requirements. First of all, it has a planar configuration with a moderate gain, even though it has a drawback of a narrow bandwidth. In addition, it can simplify the overall rectenna by eliminating the first LPF. It was reported that the second and third harmonics are suppressed in a circular sector antenna with 240 of a circular sector angle and 30 of a feeding angle from the edge [23]. The rectenna with a circular sector antenna was recently designed by our group and 77.8% of the maximum conversion efficiency was observed at 2.4 GHz [19]. Most applications of rectennas are related to portable RFIDs [24], [25] or in the biomedical implants [26] for a battery-free operation, there-

where is a rectified dc power [ ], is an incident RF is a load resistance [ ], and is a output power [ ], dc voltage across a load resistance [ ]. However, it is difficult to obtain enough power to operate a complex system so that applications are limited to low-power systems and must be delivered power from a short distance and requires high power transmission. Therefore, an efficient battery management scheme is utilized rather than a battery-free operation. In this power management, a rectified dc power activates a battery. Since this triggering mechanism is dc voltage dependent , it is more feasible and a dc voltage can be controlled by for mobile wireless communication systems. To maximize dc-voltage collection, a series pair of rectifier diodes are used as a full-wave rectifier. Since it produces twice the output voltage, it is also known as a voltage doubler, where output voltage of two diodes are added in series [27]. Thus, the overall voltage sensitivity is increased. A detector diode typically has a high RF impedance so that it is difficult to obtain impedance matching. However, in a voltage doubler, the impedances of series diodes are added in parallel and the overall impedance becomes lower. It enables easier reactive matching of the rectifier. Since the rectifier basically acts a trigger, having low threshold voltage is more of a concern than power conversion efficiency. The dc output voltage can be increased with in. However, the voltage saturation point must also creasing be considered in the rectifier circuit design. When the output voltage is larger than the threshold voltage, a dc battery and bias circuitries of all active devices are active. In this application, an analog switch is appropriate. It is crucial to have almost zero quiescent power and good off isolation. A fast switching speed is also required. In Section II, dc and RF power-combining techniques are compared. DC power combining is chosen in this paper by assuming that an RX RF power is enough to activate the switch. The next issue is whether dc power obtained from each rectenna is combined in series or parallel. It might also be a combination of series and parallel. The method to interconnect dc power in array was previously studied in [28]. In order to add up a current, a parallel connection is preferable. For our application, we are more interested in a voltage than a current, thus the obtained dc voltage is summed up by combining in series rather than in parallel, as shown in Fig. 7. As the number of elements in array is increased, such as in a two-dimensional (2-D) configuration, output voltage is also increased. It makes the retrodirective array a good application of the proposed power-management idea. V. MEASUREMENT RESULTS Here, the measurement results of a single element rectenna and the effect of power combining used in the integrated retrodirective array will be presented. Furthermore, the performance

LIM et al.: ADAPTIVE POWER CONTROLLABLE RETRODIRECTIVE ARRAY SYSTEM

Fig. 7.

DC power-combining configuration in series.

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Fig. 9.

Measured: (a) E - and (b) H -field patterns of a circular sector antenna.

Fig. 10.

Simulated comparison of dc series combining and single rectenna.

Fig. 8. Simulated and measured return loss of a circular sector antenna.

of the entire system is measured. The antenna radiation patterns confirming the retrodirective nature of the array and the system wake-up operation are presented. A. Rectenna Performance For the rectenna design, a circular sector antenna is first simulated by Agilent ADS Momentum and then fabricated on RT/Duroid 6010LM with a dielectric constant of 10.2 and a thickness of 50 mil. Since receiving and transmitting frequencies need to be slightly different, the antenna with 5.79 and 5.81 GHz of resonant frequencies are designed for a TX and an RX, respectively. The simulated and measured -parameters are shown in Fig. 8. The second and third harmonic rejection capability of the circular sector antenna should be noted. At the fundamental is 21.1 dB. On the other hand, frequency of 5.79 GHz, exhibits 3.9 and 7.3 dB at 11.58 and 17.37 GHz, respectively. Since the substrate used is suitable only up to -band, the difference from a simulation and measurement at high frequencies is caused from the loss of the substrate. A 10-dB bandwidth is also observed to be 1.5% (90 MHz). In addition, a far-field radiation pattern is measured at 5.79 GHz. Fig. 9 exhibits the normalized - and -field patterns, which are omnidirectional. Cross-polarizations are 17.33 and 11.67 dB lower than co-polarizations of - and -planes, respectively. A 7.5-dBi maximum gain is observed at broadside. Next, a rectifier circuit is designed and combined with the circular sector antenna. An Agilent Schottky detector diode

HSMS-2860 is used as a rectifier diode because this package (HSMS-286C) provides two diodes connected in series. Therefore, space can be saved and parasitic effects can be minimized. In the design of the LPF for dc passing, it is considered to suppress up to fundamental and second harmonic frequencies. It is previously suggested that the dc voltage from each element is connected in series. A single rectifier circuit is first simulated by ADS and then combined in series as the configuration of Fig. 7. Since the output voltage is also dependent on the load resistance until it is saturated, a proper resistor value should be chosen. The relationship of output voltage versus the load is chosen to be 1 M , resistance is plotted in Fig. 10. where it starts saturating for a four-rectenna array. The output M and voltage from four elements is 1.01 V with dBm, while the output voltage from a single element is 0.36 V with the same condition. It is less than four times because the output from dc connection is a little saturated. The measured output voltage versus a power density will be discussed in Section V-B. B. Adaptive Power-Management Performance For the proposed adaptive power-management technique, the sensitivity should be first considered. It is crucial to know the threshold voltage to activate a switch at the given supplied voltage of the dc battery, which is required to be as low as possible. Additionally, it should be noted how much power is consumed in the off mode. Based on the selection guide from these two parameters, Fairchild FSA-1156 is employed as a dc-to-dc switch. This analog switch is a single-pole/single-throw (SPST)

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TABLE I KEY PARAMETERS OF FSA-1156 SPST ANALOG SWITCH

Fig. 12. Measured output voltage comparison of the rectenna and the rectenna with a coupler.

Fig. 11. Simulated return losses of the coupler with the measured data of the circular sector antenna.

switch, which has one possible connection. The key parameters provided by the manufacturer are indicated in Table I. For 3.5-V supply voltage, which is the bias condition of the LNAs, it requires around 0.6 V to switch on. The quiescent current is almost negligible. In addition, since the switch is implemented in submicrometer CMOS technology, it can turn on and off quickly. The adaptive power-management circuit is first constructed with a single rectenna. As shown in Fig. 6, it consists of a circular sector antenna, a 15-dB coupler, a rectifier, a switch, and an LNA. Due to the coupler, the harmonic tuning effect of the antenna is altered, therefore, the output voltage becomes less. However, it can be optimized with the length of the transmission line between the circular sector antenna and rectifier. Fig. 11 shows a simulated result after optimization. A 15-dB coupled-line coupler is simulated with the measured return loss of the circular sector antenna as a load (Fig. 8). The return losses of the coupler with the antenna is 19.2, 5.15, and 12.2 dB at 5.79, 11.58, and 17.37 GHz, respectively. Since the second and third harmonics cannot be simultaneously controlled, the optimization mainly concerns the second harmonic frequency of 11.58 GHz. Fig. 12 shows the measured results of the power-management circuit with a single antenna. The received power can be calculated from the Friis transmission equation given by (5) where is the received power, is the transmitted power, and is the propagation distance between a TX antenna and an RX antenna. and represent gains of TX and RX antennas,

Fig. 13.

Prototype of the proposed system.

respectively. In the current setup at 5.8 GHz, the gain of the TX horn antenna is 11 dBi from the data sheet and the gain of the RX circular sector antenna is 7.5 dBi from the measurement. The TX antenna transmits a 5.8-GHz carrier signal 58.42 cm away from the RX antenna. The power density is obtained from

(6) . where the effective area As expected, it is observed that the rectified output becomes lower after being combined with a coupler since a coupler degrades the harmonic rejection capability. In order to achieve a threshold voltage to activate a switch, 18.53 W cm is required in the adaptive power-management circuit. It can be improved by increasing the number of elements. Finally, the adaptive power-management scheme is integrated with the retrodirective array system on the same substrate. The prototype of the system is shown in Fig. 13. The retrodirective array consists of four elements. The bias voltage at the LNA is

LIM et al.: ADAPTIVE POWER CONTROLLABLE RETRODIRECTIVE ARRAY SYSTEM

Fig. 14.

Measured biased voltage at LNAs in the proposed system of Fig. 13.

Fig. 15.

RCS measurement setup.

Fig. 16.

Monostatic RCS measurement.

measured by changing the transmitted power and is plotted in Fig. 14. It is observed that the proposed system turns on when the received power level is more than 8.5 dBm (11.5 W cm in a power density), which is calculated from (5). It should be noted that the rectenna’s load resistance is affected by the impedance of the LNA’s bias networks. As the number of array elements increases, the system becomes more sensitive to input RF signal.

Fig. 17. Bistatic RCS measurement at source position of: (a) and (c) 33 .

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050

, (b) 0 ,

C. Retrodirective Array Performance The retrodirective system consists of LNAs and phase-conjugating mixers. GaAs monolithic-microwave integrated-circuit (MMIC) amplifiers (HP MGA-86576) with 1.6-dB noise figure . When (NF) are used. It has 12 dB of gain and 6.75 dBm of an LNA is biased at 3.5 V, 95 mA of current is consumed.

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an HP 438A power meter is used. The output voltage of the modulated signal is much increased compared with a single tone, as shown in Fig. 18. It can also be increased with a higher . In the proposed system, the data-rate limitation is mainly due to the bandwidth of the circular sector antenna. VI. CONCLUSION

Fig. 18. Measured output voltage in a single tone and 20-Mb/s BPSK = 1 and 2 V. modulated signals

V

For the subharmonic mixer, Agilent beam lead Schottky diode pairs (HSCH-5531) are used. A total LO power of 6.9 dBm is supplied through Wilkinson power dividers, feeding all array mixers. In order to demonstrate retrodirectivity, monostatic and bistatic radar cross section (RCS) measurements are performed in the far-field condition. The measurement setup is exhibited in Fig. 15. In order to provide enough power to trigger the system, a Hughes traveling-wave tube (TWT-1177H) power amplifier was used as an interrogator. Standard gain horn antennas are used as the TX and RX antennas. They have 11 dBi of gain at 5.8 GHz. For the monostatic RCS, interrogation TX and RX antennas are rotating together while keeping 1.5-m distance from the device-under-test (DUT) where the proposed system is positioned. The retransmitted signal from the DUT is measured with the RX antenna. The measured monostatic pattern is shown in Fig. 16. For the bistatic RCS, when a interrogation TX antenna is 1.5 m away from the DUT and positioned at 50 , 0 , and 33 , the retransmitted signal from the DUT is measured with an RX antenna. In Fig. 17, the measured bistatic RCS results are compared with the theoretical results calculated from an array factor. They are in good agreement. These results show the retrodirective system can operate properly with rectennas to control a power supply. D. System Performance in BPSK Data Transmission Link The phase-conjugation mixer pairs used in the proposed array can also be used as simple down-conversion mixers, as explained in Section III. Therefore, the array can be used as a standard RX front-end, as well as a retrodirective array. This RX function can be used to receive data from sensors or from a remote interrogator sending commands to the array. In this scenario the interrogator must transmit enough power to activate the array, as well as transmit data. Therefore, the effect of rectified voltage obtained by a rectenna from a modulated signal was measured. The adaptive power-management circuit with a single rectenna is used for this measurement. An RF source signal is modulated by a 20-Mb/s binary phase-shift keying (BPSK) with 5.79 GHz of a carrier signal. In order to measure the total transmitted power of the modulated signal,

A novel adaptive power-management scheme is applied to a four-element retrodirective array system at 5.8 GHz. When the RF signal is detected, the system is activated from a sleeping mode by switching on with a dc voltage converted through a rectenna. Without any interrogation, the system is turned off. Since a quiescent current of the switch used is typically 0.1 A, the power consumption is negligible in an idle mode. In order to increase a sensitivity of a power-management circuit, a voltage doubler and a dc combining technique are introduced. For the digital wireless communications, the sensitivity is further enhanced with modulated signals. It is also feasible to produce higher voltage as the number of elements in the antenna array is increased. Therefore, the proposed retrodirective array system can have substantially increased battery life for wireless sensor server applications. ACKNOWLEDGMENT The authors would like to acknowledge the helpful discussion with Dr. C. Kim, Samsung Advanced Institute of Technology, Yongin, Korea, and Dr. J.-Y. Park, Advanced Power Technology, Santa Clara, CA. REFERENCES [1] S. Ci, H. Sharif, and K. Nuli, “Study of an adaptive frame size predictor to enhance energy conservation in wireless sensor networks,” IEEE J. Select. Areas Commun., vol. 23, no. 2, pp. 283–292, Feb. 2005. [2] T.-H. Lin, W. J. Kaiser, and G. J. Pottie, “Integrated low-power communication system design for wireless sensor networks,” IEEE Commun. Mag., vol. 42, no. 12, pp. 142–150, Dec. 2004. [3] S. D. Muruganathan, D. C. F. Ma, R. I. Bhasin, and A. O. Fapojuwo, “A centralized energy-efficient routing protocol for wireless sensor networks,” IEEE Commun. Mag., vol. 43, no. 3, pp. S8–S13, Mar. 2005. [4] R. Y. Miyamoto, K. M. K. H. Leong, S.-S. Jeon, Y. Wang, Y. Qian, and T. Itoh, “Digital wireless sensor server using an adaptive smart antenna/retrodirective array,” IEEE Trans. Veh. Technol., vol. 52, no. 5, pp. 1181–1188, Sep. 2003. [5] L. C. Van Atta, “Electromagnetic reflector,” U.S. Patent 2 908 002, Oct. 1959. [6] C. Y. Pon, “Retrodirective array using the heterodyne technique,” IEEE Trans. Antennas Propag., vol. AP-12, no. 3, pp. 176–180, Mar. 1964. [7] Y. Chang, H. R. Fetterman, I. L. Newberg, and S. K. Panaretos, “Microwave phase conjugation using antenna arrays,” IEEE Trans. Microw. Theory Tech., vol. 46, no. 11, pp. 1910–1919, Nov. 1998. [8] T. Brabetz, V. F. Fusco, and S. Karode, “Balanced subharmonic mixers for retrodirective-array applications,” IEEE Trans. Microw. Theory Tech., vol. 49, no. 3, pp. 465–469, Mar. 2001. [9] J.-Y. Park, K. M. K. H. Leong, G. R. Kim, J. I. Choi, and T. Itoh, “Active retrodirective array with subharmonic phase conjugation mixers,” presented at the Asia–Pacific Microwave Conf., Nov. 2003. [10] C. Smaltz, “Extending battery life of portable devices,” in WESCON Conf. Rec., Sep. 1993, pp. 384–387. [11] Y. Idota, T. Kubota, A. Matsufuji, Y. Maekawa, and T. Miyasaka, “Tinbased amorphous oxide: A high-capacity lithium–ion-storage material,” Science, vol. 276, pp. 1395–1397, May 1997. [12] J. B. Hacker, J. Bergman, G. Nagy, G. Sullivan, C. Kadow, H.-K. Lin, A. C. Gossard, M. Rodwell, and B. Brar, “An ultra-low power InAs/Alsb HEMT -band low-noise amplifier,” IEEE Microw. Wireless Compon. Lett., vol. 14, no. 4, pp. 156–158, Apr. 2004.

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[13] J. M. Daga, C. Papaix, M. Merandat, S. Ricard, G. Medulla, J. Guichaoua, and D. Auvergne, “Design techniques for EEPROM’s embedded in portable systems on chips,” IEEE Des. Test. Comput., vol. 20, no. 1, pp. 68–75, Jan.–Feb. 2003. [14] M. M. Nordman, W. E. Kozlowski, and O. Vahamaki, “Synchronizing low-cost energy aware sensors in a short-range wireless cell,” in Proc. Euromicro Digital Systems and Design Symp., Sep. 2001, pp. 438–445. [15] M. Olivieri, A. Trifiletti, and A. De Gloria, “A low-power microcontroller with on-chip self-tuning digital clock-generatior for variable-load applications,” in IEEE Int. Computer Design Conf., Oct. 1999, pp. 476–481. [16] H. Chen and W. Yen, “A low power and fast wake up circuit,” in IEEE Int. Solid-State Circuits Conf., vol. 2, May 2004, pp. II-293–296. [17] L. Gu and J. A. Stankovic, “Radio-triggered wake-up capability for sensor networks,” in IEEE Real-Time and Embedded Technology and Applications Symp., May 2004, pp. 27–36. [18] S. Lim, K. M. K. H. Leong, and T. Itoh, “Adaptive power controllable retrodirective array system for portable battery-operated applications,” presented at the IEEE MTT-S Int. Microwave Symp., Jun. 2005. [19] J.-Y. Park, S.-M. Han, and T. Itoh, “A rectenna design with harmonic-rejecting circular-sector antenna,” IEEE Antennas Wireless Propag. Lett., vol. 3, pp. 52–54, 2004. [20] Y.-H. Suh and K. Chang, “A high-efficiency dual-frequency rectenna for 2.45- and 5.8-GHz wireless power transmission,” IEEE Trans. Microw. Theory Tech., vol. 50, no. 7, pp. 1784–1789, Jul. 2002. [21] J. A. Hagerty, F. B. Helmbrecht, W. H. McCalpin, R. Zane, and Z. B. Popovic, “Recycling ambient microwave energy with broad-band rectenna arrays,” IEEE Trans. Microw. Theory Tech., vol. 52, no. 3, pp. 1014–1024, Mar. 2004. [22] R. H. Rasshofer, M. O. Thieme, and E. M. Biebl, “Circularly polarized millimeter-wave rectenna on silicon substrate,” IEEE Trans. Microw. Theory Tech., vol. 46, no. 3, pp. 715–718, Mar. 1998. [23] W. F. Richards, J. D. Ou, and S. A. Long, “A theoretical and experimental investigation of annular, annular sector, and circular sector microstrip antennas,” IEEE Trans. Antennas Propag., vol. AP-32, no. 8, pp. 864–867, Aug. 1984. [24] P. Foster and R. Burberry, “Antenna problems in RFID systems,” in Proc. IEE RFID Technology Colloq., vol. 3, Oct. 1999, pp. 1–5. [25] S.-M. Han, J.-Y. Park, and T. Itoh, “Dual-fed circular sector antenna system for a rectenna and a RF receiver,” presented at the Eur. Microwave Conf., Oct. 2004. [26] H. Matsuki, Y. Yamakata, and N. Chubachi, “Transcutaneous DC–DC converter for totally implantable artificial heart using synchronous rectifier,” IEEE Trans. Magn., vol. 32, no. 9, pp. 5518–5120, Sep. 1996. [27] E. Angelo, Electronic Circuits. New York: McGraw-Hill, 1958. [28] N. Shinohara and H. Matsumoto, “Dependence of DC output of a rectenna array on the method of interconnection of its array elements,” Elect. Eng. Jpn., vol. 125, no. 1, pp. 9–17, 1998.

Sungjoon Lim (S’02) received the B.S. degree in electronic engineering from Yonsei University, Seoul, Korea, in 2002, the M.S. degree in electrical engineering from the University of California at Los Angeles (UCLA), in 2004, and is currently working toward the Ph.D. degree in electrical engineering at UCLA. Since 2002, he has been a Graduate Student Researcher with the Microwave Electronics Laboratory, UCLA. In 2005, he was an Intern with MOTIA Inc. His current research interests include microwave and millimeter-wave systems for wireless communications and microwave applications on metamaterials.

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Kevin M. K. H. Leong (S’99–M’04) received the B.S. degree in electrical engineering from the University of Hawaii at Manoa, in 1999, and the M.S. and Ph.D. degree in electrical engineering from the University of California at Los Angeles (UCLA), in 2001 and 2004, respectively. He is currently a Post-Doctoral Researcher with UCLA. His research interests include planar antennas, millimeter-wave circuits, and mobile communication systems. Dr. Leong was the recipient of the First-Place Best Student Paper Award at the 2001 European Microwave Conference.

Tatsuo Itoh (S’69–M’69–SM’74–F’82) received the Ph.D. degree in electrical engineering from the University of Illinois at Urbana-Champaign, in 1969. From September 1966 to April 1976, he was with the Electrical Engineering Department, University of Illinois at Urbana-Champaign. From April 1976 to August 1977, he was a Senior Research Engineer with the Radio Physics Laboratory, SRI International, Menlo Park, CA. From August 1977 to June 1978, he was an Associate Professor with the University of Kentucky, Lexington. In July 1978, he joined the faculty at The University of Texas at Austin, where he became a Professor of Electrical Engineering in 1981 and Director of the Electrical Engineering Research Laboratory in 1984. During the summer of 1979, he was a Guest Researcher with AEG-Telefunken, Ulm, Germany. In September 1983, he was selected to hold the Hayden Head Centennial Professorship of Engineering at The University of Texas at Austin. In September 1984, he was appointed Associate Chairman for Research and Planning of the Electrical and Computer Engineering Department, The University of Texas at Austin. In January 1991, he joined the University of California at Los Angeles (UCLA) as Professor of Electrical Engineering and Holder of the TRW Endowed Chair in Microwave and Millimeter Wave Electronics. He was an Honorary Visiting Professor with the Nanjing Institute of Technology, Nanjing, China, and with the Japan Defense Academy. In April 1994, he was appointed an Adjunct Research Officer with the Communications Research Laboratory, Ministry of Post and Telecommunication, Japan. He currently holds a Visiting Professorship with The University of Leeds, Leeds, U.K. He has authored or coauthored 350 journal publications, 650 refereed conference presentations, and has written 30 books/book chapters in the area of microwaves, millimeter waves, antennas, and numerical electromagnetics. He has generated 64 Ph.D. students. Dr. Itoh is a member of the Institute of Electronics and Communication Engineers of Japan, and Commissions B and D of USNC/URSI. He served as the editor of the IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES (1983–1985). He serves on the Administrative Committee of the IEEE Microwave Theory and Techniques Society (IEEE MTT-S). He was vice president of the IEEE MTT-S in 1989 and president in 1990. He was the editor-in-chief of IEEE MICROWAVE AND GUIDED WAVE LETTERS (1991–1994). He was elected an Honorary Life Member of the IEEE MTT-S in 1994. He was elected a member of the National Academy of Engineering in 2003. He was the chairman of the USNC/URSI Commission D (1988–1990) and chairman of Commission D of the International URSI (1993–1996). He is chair of the Long Range Planning Committee of the URSI. He serves on advisory boards and committees for numerous organizations. He has been the recipient of numerous awards including the 1998 Shida Award presented by the Japanese Ministry of Post and Telecommunications, the 1998 Japan Microwave Prize, the 2000 IEEE Third Millennium Medal, and the 2000 IEEE MTT-S Distinguished Educator Award.

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Frequency Planning and Synthesizer Architectures for Multiband OFDM UWB Radios Chinmaya Mishra, Student Member, IEEE, Alberto Valdes-Garcia, Student Member, IEEE, Faramarz Bahmani, Student Member, IEEE, Anuj Batra, Member, IEEE, Edgar Sánchez-Sinencio, Fellow, IEEE, and Jose Silva-Martinez, Senior Member, IEEE

Abstract—This work presents an analysis on frequency planning and synthesis for multiband (MB) orthogonal frequency-division multiplexing (OFDM) ultra-wideband (UWB) radios operating in the range of 3.1–10.6 GHz. The most important specifications for the frequency synthesizer in an MB-OFDM UWB transceiver are provided. A synthesizer architecture for an existing frequency plan is introduced along with a discussion on its performance and implementation. An alternative frequency plan and its corresponding synthesizer architecture are also proposed. It is shown how this modified frequency plan leads to a significant simplification in the synthesizer realization. The feasible performance of both synthesizer architectures is evaluated through macromodel simulations using realistic models for the building blocks. Finally, system-level simulation results showing the impact of synthesizer spurs on the bit error rate performance of an MB-OFDM UWB receiver in the presence of interferers are provided. The presented results and discussion provide valuable insight for the implementation of a 3.1–10.6-GHz UWB synthesizer. Index Terms—Frequency-band plan, frequency synthesis, multiband (MB) orthogonal frequency-division multiplexing (OFDM), ultra-wideband (UWB).

I. INTRODUCTION

D

UE TO its high channel capacity, an ultra-wideband (UWB) system is an attractive solution for the implementation of very high data rate ( 100 Mb/s) short-range wireless networks. Among the different options for the efficient use of the available UWB spectrum in personal computer and consumer electronic applications, the MB-OFDM approach has received strong support from several industrial organizations [1]. According to the regulations from the Federal Communications Commission (FCC), Washington, DC, UWB devices for communication applications can operate in the 3.1–10.6-GHz frequency band while employing at least 500 MHz of bandwidth (measured at 10-dB points) with a power spectral density (PSD) of less than 41.25 dBm/MHz [2]. In the multiband (MB) orthogonal frequency-division multiplexing (OFDM) proposal [3] the 7500-MHz UWB spectrum is divided into 14 bands of 528 MHz each. The bands are grouped into five band groups, as shown in the upper section of Fig. 1. Only Manuscript received March 14, 2005; revised July 10, 2005. C. Mishra, A. Valdes-Garcia, F. Bahmani, E. Sánchez-Sinencio, and J. SilvaMartinez are with the Analog and Mixed-Signal Center, Electrical Engineering Department, Texas A&M University, College Station, TX 77843 USA (e-mail: [email protected]; [email protected]). A. Batra is with the Digital Signal Processing Solutions Research and Development Center, Texas Instruments Incorporated, Dallas, TX 75243 USA. Digital Object Identifier 10.1109/TMTT.2005.856087

the first band group, corresponding to the lower part of the spectrum (3.1–4.8 GHz), is considered as mandatory by the current standard proposal. The remaining band groups have been defined and left as optional to enable a structured and progressive expansion of the system capabilities. Current efforts from semiconductor companies for the implementation of UWB devices focus on the first band group to achieve a faster time-to-market and affordable power consumption with current CMOS [4] and BiCMOS [5] technologies. The realization of UWB radios for operation in the entire 3.1–10.6-GHz range is an open research area, which leads to various design challenges at both the system and circuit levels. Fig. 2 illustrates the role of a UWB frequency synthesizer in an MB-OFDM direct conversion transceiver. As in other wireless systems, the frequency synthesizer has the crucial function of generating the local oscillator (LO) signal that drives the down-converter in the receiver path and the up-converter in the transmitter. There are at least two demanding requirements that make a frequency synthesizer for an MB-OFDM UWB radio significantly different from the widely explored synthesizers for narrow-band wireless systems, which are: 1) the range of frequencies to be generated spans several gigahertz and 2) the time to switch between different band frequencies within a band group should be less than 9.47 nS [3]. This requirement prevents the use of a standard phase-locked loop (PLL)-based synthesizer as a solution for this application [6]. Some possible ways of performing the frequency generation in a UWB device are discussed in [6]. One approach, which has been employed in [7], is to have multiple frequency sources. The synthesizer presented in [7] generates 4-, 5-, 6-, and 7-GHz tones using a single-output frequency synthesizer for each frequency. Reference [4] follows a similar approach by having three fixed-modulus PLLs; one for each of the three frequencies in band group 1. Reference [8] uses two PLLs (to generate 3960 and 528 MHz) and a single-sideband (SSB) mixer to span the frequencies in band group 1. In [9], two PLLs are again used, but this time in a different fashion to generate seven bands from 3 to 8 GHz. Reference [10] uses two frequency sources (implying two PLLs) and external inputs that are not generated within the system. It generates eight tones from 3.25 to 6.75 GHz with a spacing of 500 MHz between each tone. A more practical strategy, however, is to generate one frequency with a PLL and indirectly generate the other frequencies from auxiliary signals generated in parallel [6]. This technique is used in [1], [11], and [12]. From here onwards, the term “auxiliary signals” or “auxiliary tones”

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Fig. 1. Current band plan from the MB-OFDM proposal (top) and proposed alternate band plan (bottom).

Fig. 2. Frequency synthesizer in an MB-UWB transceiver.

would refer to signals that are generated within the division loop of the PLL. Reference [11] presents a synthesizer for band group 1 based on a divide-by-7.5 structure (which uses standard dividers and SSB mixers); both the reference tone (3960 MHz)

and a 528-MHz tone for up/down conversions are generated using a single PLL. The synthesizer used in [12] is based on a 16-GHz quadrature voltage-controlled oscillator (VCO), eight divide-by-2 structures, two SSB mixers, and two multiplexers to generate seven bands from 3.1 to 8.2 GHz. The diverse characteristics of the UWB synthesizers reported thus far is a clear sign of the challenge involved in the search for an optimum solution. Moreover, none of these current architectures span the entire UWB spectrum licensed by the FCC and considered by the MB-OFDM proposal. Significant research must first be performed at the system level to develop an efficient synthesizer solution (in terms of performance and power consumption) for the requirements of a completely integrated MB-OFDM UWB transceiver. This paper addresses the problem of frequency synthesis for a 3.1–10.6-GHz MB-OFDM radio in the aspects of choice of the band plan, specifications and architecture for the synthesizer, and impact of the expected nonidealities in an integrated implementation. Section II outlines the different specifications to be met by the LO signal. Section III discusses the logic governing the frequency planning for an MB-OFDM UWB radio with a subband spacing of 528 MHz. A frequency-synthesizer architecture for the current band plan is proposed and discussed in Section IV. Section V introduces the possibility of modifying the frequency band plan to simplify the architecture of the synthesizer. Section VI presents simulation results from a macromodel of the proposed synthesizer architectures including the most important nonidealities expected in a hardware implementation. Finally, conclusions are provided in Section VII.

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II. SYNTHESIZER SPECIFICATIONS

TABLE I SUMMARY OF SYNTHESIZER SPECIFICATIONS

In addition to the frequency switching speed, the LO signal must comply with other requirements to ensure proper operation of the MB-OFDM UWB radio. The specifications outlined here assume the OFDM parameters and bit error rate (BER) requirements described in [3] for a 480-Mb/s data transmission and an additive white Gaussian noise (AWGN) channel. A quadrature phase-shift keying (QPSK) constellation is considered for the individual sub-carriers. For a packet error rate of 8% with a 1024-byte packet, the target BER when using a coding rate is 10 , which corresponds to an un-coded BER of approximately 10 . A. Phase Noise The phase noise from the LO in an OFDM receiver has two different effects on the received symbols. It introduces a phase rotation of the same magnitude in all of the sub-carriers and creates inter-carrier interference (ICI) [13]. The first undesired effect is eliminated by introducing pilot carriers with a known phase in addition to the information carriers. On the other hand, phase noise produces ICI in a similar way as adjacent-channel interference in narrow-band systems. Assuming that the data symbols on the different sub-carriers are independent, the ICI may be treated as Gaussian noise. The PSD of a locked PLL can be modeled by a Lorenzian spectrum described by (1) where is the 3-dB bandwidth of the PSD, which has a normalized total power of 0 dB. The degradation ( in decibels) in the signal-to-noise ratio (SNR) of the received sub-carriers due to the phase noise of the LO in an OFDM system can be approximated as [14] (2) where is the OFDM symbol length in seconds (without the cyclic extension), defines the Lorenzian spectrum described is the desired SNR for the received symabove, and bols (in a linear scale, not in decibels). For this system, MHz and the for the target coded BER of 10 is 5.89 (7.7 dB). For dB and the mentioned parameters, can be computed with (2) and is 7.7 kHz. The corresponding Lorenzian spectrum has a power of 86.5 dBc/Hz @ 1 MHz. This phase-noise specification is to be met by the LO signal at the input of the down-conversion mixer, as shown in Fig. 2. As it will be explained in Section II-B, unlike most frequency synthesizers for narrow-band systems, a UWB synthesizer architecture involves a source PLL followed by a series of up and down conversions that will affect the phase noise of the source oscillation. For this reason, the phase-noise specification provided above does not directly correspond to the phase noise at the output of the employed source PLL. General guidelines for the analysis of phase noise in component cascades are provided in [15]. For this application, the most relevant components for phase noise degradation are the mixers employed in the frequency translation operations across the synthesizer architecture. For a given

Fig. 3.

Frequency tree diagram.

offset frequency , the phase noise at the output of a mixer can be estimated as the rms sum of the individual input noise contributions. Hence, given phase-noise relative power densiand (in decibels relative to carrier per ties hertz) at the input of each port of the mixer, the output phase noise can be expressed as (3) Even though in this case the two signals are indirectly derived from the same reference, their noise can be assumed in general to be uncorrelated since the delay from the PLL to each input of a given mixer would be significantly different. The size of an integrated implementation would be small in comparison to the wavelengths involved, but the frequency dividers and the poles in the signal path introduce a delay. As will be evident later, there is at least one frequency divider between the inputs of each mixer. The gain or loss of the mixer amplifies or attenuates all of the frequency components around the frequency of operation by the same amount and, hence, does not affect the phase noise. Moreover, due to the relatively large amplitude (tens of millivolts) of the signals within the synthesizer, the contribution of the thermal noise of the mixers to the phase noise is negligible. For a given UWB synthesizer architecture, the path with the largest number of frequency translations can be analyzed with the use of (3) to find the phase-noise specification for the source PLL.

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Fig. 4.

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Synthesizer architecture (I).

B. In-Phase ( ) and Quadrature ( ) Matching In an OFDM system, the amplitude and phase imbalance between the and channels transform the received time-domain vector into a corrupted vector , which consists of a scaled version of the original vector combined with a term proportional to its complex conjugate . This transformation can be written as [16]

TABLE II SYNTHESIS OF FREQUENCIES FOR CURRENT BAND PLAN

(4) where and are complex constants, which depend on the imbalance. This alteration on the received symamount of bols can have a significant impact on the system performance. The effect of a phase mismatch in the quadrature LO signal on the BER versus SNR performance of the receiver was evaluated considering the system characteristics outlined at the beginning of this section and using a model built in SystemView [17]. Simulation results for uncoded data over an AWGN channel showed that the degradation in the sensitivity is 0.6 dB for 5 of mismatch. This degradation can be reduced with the use of coding and compensation techniques [16]. C. Spurious Content As in other communication systems, the most harmful spurious components of an LO signal are those at an offset equal to multiples of the frequency spacing between adjacent bands (in this case, 528 MHz) since they directly down-convert the transmission of a peer device on top of the signal of interest, as shown in Fig. 2. Using the system-level model described above, it was found that, in order to have a negligible degradation in the sensitivity ( 0.1 dB); the carrier-to-interferer ratio (CIR) at baseband should be at least 24 dB. If two interferers with same power level are present, each of them must be at least 27 dB below the signal of interest. In other words, to tolerate the presence of other UWB transmissions that arrive with comparable power at the antenna of the receiver, the synthesizer spurs that appear at frequencies corresponding to other bands must have an aggregate power of less than 24 dBc. A summary of the synthesizer specifications is given in Table I.

III. FREQUENCY PLANNING The frequency-band plan introduced in [3] is shown in the upper section of Fig. 1. Each band in any band group is 528 MHz away from its adjacent band. Each band’s center frequency is given by MHz

(5)

(band number). where The main objective of frequency planning is to maximize the number of usable bands in the available spectrum while keeping the architecture of the frequency synthesizer simple, compact, and power efficient. As mentioned in Section I, generating each band frequency using a PLL is impractical due to the very fast switching time requirement. The MB-OFDM standard proposal [3] considers that when two UWB devices communicate, they do so using the three (or two) adjacent frequencies of a band group. This implies that the synthesizer needs to hop very fast only between the frequencies of a particular band group. A relatively simple solution for the synthesis of these frequencies is to generate a reference tone (as shown in Fig. 1) for each band

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Fig. 5.

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 53, NO. 12, DECEMBER 2005

Synthesizer architecture (II).

group and the adjacent frequencies through an up or down conversion by 528 MHz. A reference tone in a band group is that tone from which the required adjacent frequencies are derived. From the above discussion, it is clear that, for the generation of any band frequency in any band group, the 528-MHz tone always needs to be available apart from the reference frequency of that band group. A very practical approach involves a PLL-based architecture where the output frequency of the PLL is fixed and the reference tones in the different band groups and the 528-MHz tone are generated (either directly or indirectly) from the auxiliary frequencies (frequencies generated in the process of deriving the PLL reference frequency from the VCO output). The auxiliary frequencies in a PLL will depend on the division ratio and the dividers used in its implementation. In order to have maximum possible auxiliary frequencies that could be derived from a fixed VCO frequency, the division ratio should be implemented with small divisors such as 2 and 3. With the assumption that a divide-by-2 and a divide-by-3 serve as the basic cells in the division loop of a PLL, a frequency tree diagram can be generated as depicted in Fig. 3. This diagram shows the different possible VCO frequencies that can result in a 528-MHz tone by successive division by 2, 3, or both. The tree also shows the different auxiliary frequencies generated in the PLL during the process of generation of the 528-MHz tone. In this way, separate synthesis of 528 MHz is avoided. The reference frequency of the PLL could be further derived from 528 MHz. Fig. 3 provides various choices for the VCO frequency. In order to reduce the number of components and simplify the architecture, the VCO frequency should be chosen such that most of the auxiliary frequencies are the same as the reference tones. Based on Fig. 3, a band plan and a set of auxiliary frequencies can be defined to obtain an efficient synthesizer architecture. A different, but not less important factor to consider in the choice of the frequencies to be used by the MB-OFDM UWB radio is the overlap between the Unlicensed National Information Infrastructure (U-NII) band from 5.15 to 5.825 GHz and the UWB spectrum. While the maximum output power of a UWB transmitter can reach 10 dBm when using 1584 MHz of bandwidth (three bands of 528 MHz), the devices operating in the mentioned U-NII band can have a transmit power of 16 dBm or higher. The interference from wireless local area network

TABLE III SYNTHESIS OF FREQUENCIES FOR ALTERNATE BAND PLAN

(WLAN) radios using the IEEE 802.11a standard are of particular concern due to their widespread use. In [3], it is estimated that an attenuation of 30 dB in the 5.15–5.825-GHz spectrum is required from a front-end filter to tolerate the presence of a 802.11a transmitter at a distance of 0.2 m. Due to the nature of their target applications, MB-OFDM and 802.11a radios will coexist in most environments preventing the effective use of a band group that overlaps with the U-NII band. For these reasons, the synthesizer architectures described in the following sections do not consider a band group in the range of 5.15–5.825 GHz. IV. SYNTHESIZER ARCHITECTURE FOR CURRENT BAND PLAN The frequency tree diagram in Fig. 3 is useful to define the architecture for the frequency synthesizer; each VCO frequency results in different auxiliary frequencies and choices for the architecture. Based on this analysis, an efficient synthesizer architecture for the existing band plan is presented here. The architecture presented in [18] for the generation of seven frequencies (between 3.432–7.920 GHz while avoiding U-NII bands) is based on a PLL that generates a tone at 6336 MHz, and is considered as a starting point for the discussion. Choosing the VCO frequency as 6336 MHz and following the path

MISHRA et al.: FREQUENCY PLANNING AND SYNTHESIZER ARCHITECTURES FOR MB-OFDM UWB RADIOS

Fig. 6.

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Synthesizer architecture (III).

enclosed by the dotted lines (Fig. 3), a possible architecture for the current band plan can be defined as shown in Fig. 4. In contrast to the architecture in [18], this architecture generates 11 frequencies. The shaded italicized frequencies in Fig. 4 correspond to the reference tones for the current band plan shown in Fig. 1. It is important to note that the switching time between bands within a given band group depends only on the switching of the final multiplexer. This feature is shared with both of the other architectures presented below. In this architecture option, the employed mixers need to be SSB and broad-band since they cannot be optimized for a single input or output frequency. In addition, intermediate filtering stages are required to maintain the spectral purity of the signals, which undergo a series of up/down conversions for the generation of a particular frequency. As shown in Fig. 4, one option would be to have bandpass filters at the output of such mixers, either dedicated or capable of being tuned over a wide range of frequencies. This would involve a significant amount of passives, which would increase the required area. Due to the high frequency and wide-band nature of the components involved, the power consumption of this synthesizer implementation may also become a major portion of the entire transceiver power. Hence, to obtain a suitable performance from this solution would be at the cost of significant area and power. A strategy to reduce the power and complexity in the frequency synthesizer is to identify auxiliary frequencies that can be used to generate most of the reference tones with few or no frequency translation operations. From the frequency tree diagram, it can be found that following the path enclosed within the dashed shaded line two of the frequencies, i.e., 6336 and 3168 MHz, are equally spaced (792 MHz) from their reference tones, as shown in Fig. 1. Therefore, having these frequencies at hand, one stage of mixing could be avoided in the generation of the reference tones. Based on these auxiliary frequencies, Table II shows the proposed synthesis of the reference tones for the current band plan. A compact frequency-synthesizer architecture is proposed based on the frequency synthesis described in Table II and is shown in Fig. 5. From Table II, it can be seen that the architecture (I) was modified such that all the reference tone generations involve a final up conversion by a 792-MHz tone, term in the frequency synthesis column for all which is the frequencies. It is important to mention that the reference tone

in band group 5 has changed from 9768 MHz in architecture (I) to 10 296 MHz in architecture (II). A significant reduction in power and area would be expected due to the reduced number of mixers with multiple frequency output. However, this architecture still needs a broad-band SSB up converter for the generation of all the reference tones (up conversion with 792 MHz). Harmonics can be curtailed by low-pass filtering at different stages, but suppressing the unwanted sidebands demands additional filtering (bandpass or band notch) for the different IFs generated in the synthesizer. In the above architecture, this would imply a wide tuning-range bandpass (or notch) filter to cater to the wide range of IFs generated (especially after the up conversion with 792 MHz) apart from the dedicated filtering wherever required (see Fig. 5). One possibility is to have dedicated SSB mixer blocks and filtering for generation of each reference tones, but that would be at the expense of higher power consumption. It must be mentioned here that the last two mixers used to generate the bands adjacent to the reference frequency (up/down conversion by 528 MHz) also have a multiple frequency input and output and would have to be broad-band. However, this structure with two mixers and one multiplexer at the end of the frequency synthesizer is common to all of the architectures presented in this work. Since filtering at the final stage would demand a broad-band tunable filter spanning several gigahertz, it is not practical and is, hence, not employed at the output of the last mixers in any of the architectures. Hence, the aim would be to have the reference frequency as spectrally pure as possible before the final up/down conversion. Therefore, an important consideration is to minimize the number of up/down-conversion operations in the generation of any reference frequency to reduce the spurs within the UWB spectrum. The above discussion highlights some of the most important considerations for the design of a frequency synthesizer in an UWB system. V. ALTERNATE BAND PLAN AND SYNTHESIZER ARCHITECTURE From the frequency tree diagram in Fig. 3, it can be noted that different sets of auxiliary frequencies can be generated in the PLL. In order to further reduce the number of multiple frequency output SSB mixers and avoid reconfigurable filtering

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Fig. 7. SystemView setup for the macromodel.

TABLE IV DSB MIXER SPECIFICATIONS

Fig. 8. SSB mixer block with phase and amplitude error. For an ideal SSB mixer, A and .

1 =0

=

=0

schemes, a branch in the frequency tree can be selected such that most of the reference tones are directly generated in the divider chain (path from the selected VCO frequency to the 528-MHz tone). Looking carefully, it can be found that by moving the first three bands in band group 1 by 264 MHz to the higher side of the frequency spectrum and moving band groups 3–5 by 264 MHz to the lower side of the spectrum (as shown with gray arrows in Fig. 1), two of the reference tones (8448 and 4224 MHz) are generated in the divider chain of the PLL, which completely eliminates the need of any multiple frequency output mixer for the generation of any reference frequency. The corresponding set of auxiliary frequencies for the modified band plan is enclosed with a solid line in the frequency tree of Fig. 3. It is important to mention that this proposed modification in the band plan overlaps with the radio astronomy bands in Japan; however, it does not introduce any overlap with the U-NII band in the U.S. Based on the frequency-generation table (Table III), a modified architecture [synthesizer architecture (III)] is proposed, as shown in Fig. 6. This architecture employs dedicated SSB mixers since each of them generates only one frequency. The most significant advantage of this architecture is that dedicated filtering can be employed at every stage wherever required to obtain a clean spectrum, thereby eliminating the need of

Fig. 9.

Sideband rejection with amplitude and phase error.

reconfigurable filtering schemes. The generation of two reference tones within the divider chain also helps in reducing the complexity. As it will be shown in Section VI, the spurs in this architecture are diminished because of the reduced number of up/down conversions involved in the generation of the reference frequencies. In general, for an MB-OFDM UWB system, a frequency-synthesizer architecture, which minimizes the number of up/down conversions would be preferred.

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TABLE V SPURS ASSOCIATED WITH EACH BAND FREQUENCY FOR SYNTHESIZER ARCHITECTURE (II) WITH NONIDEALITIES

VI. MACROMODEL SIMULATION RESULTS AND PERFORMANCE ANALYSIS In order to obtain further insight on the performance of the proposed architectures (II and III), a macromodel was built in SystemView [17] for each of them. The models consist of divide-by-2 or divide-by-3 blocks, SSB mixer blocks composed of active mixers, low-pass filters, and bandpass filters at intermediate stages. Fig. 7 shows a block diagram of the schematic in SystemView for architecture II generating the 3960- and 4488-MHz frequencies. Since all frequencies are not available at the same time, a block diagram for the generation of all frequencies is not shown. The results presented here do not include any multiplexing. Hence, coupling and switching issues have not been considered here. The SystemView model, as shown in Fig. 7, consists of a sinusoidal source that models the oscillator. A divide-by- token or for of the communications library is used with the divide-by-2 or divide-by-3 implementation. The input to this token could be a sine or square wave, whereas the output is always a rectangular wave. The output of a divide–by-2 circuit has significant harmonic content, which results in multiple spurious tones after subsequent mixing in the later stages of the synthesizer. This is also an issue in an integrated-circuit implementation. For this reason, a first-order low-pass filter is employed at the output of each divider in the macromodel to partially filter out the harmonics. To provide additional suppression for unwanted tones (harmonics, intermodulation products, leakage, and sidebands), dedicated second-order bandpass filters are placed at the output of the SSB mixer blocks. The aim is to have as clean a signal as possible until the final up/down conversion with 528 MHz. The filters used in the macromodel are from the linear systems/filters operator group and they are of continuous time analog type. The bandpass filters used in the macromodels have a quality factor ( ) of 5, which is a realistic assumption for an implementation in current deep-submicrometer CMOS technologies. The SSB mixers are built with the widely used configuration of two double-sideband (DSB) active mixers, as shown in Fig. 8 [19]. The active mixers used were from the RF/analog library.

The specifications used for each of the DSB active mixers (in the SSB mixer) that are shown in Table IV are close to typical values provided in [20] and [21]. Whether the upper or lower sideband is rejected depends on the placement of the phase shifts or the polarity of the summing block. Since a divide-by-2 circuit can result in both and signals, the divider outputs can directly form the inputs to the SSB mixer. However, in the macromodel, the quadrature signals of the divider were generated by adding a time-delay token of the delay operator group. Finally, an analysis sink was used to capture each output frequency. Ideally, to obtain perfect rejection of one of the sidebands, the signals should be in perfect quadrature and there should be no gain mismatch in the signal paths [19]. Fig. 8 shows the SSB mixer block with the nonidealities expected from an actual circuit implementation. The sideband rejection ratio (SBRR) in an SSB mixer with a proportional amplitude error between the two and phase errors and in each of DSB mixer outputs the input signals is given by (see the Appendix )

(6) A plot showing the sideband rejection versus amplitude and total phase error ( ) is shown in Fig. 9. For a particular case of (5%) and (corresponding to a total phase error of 10 ), the sideband rejection is 20.86 dB. Simulations are performed for architectures II and III first with the component models described above, but assuming no phase or amplitude mismatch in the SSB mixer blocks or any shift in frequencies in the intermediate filters. In this scenario, for the generation of all the required tones in both architectures, the level of each spur is at least 26 dB below the desired frequency. This could be tolerated according to the specifications outlined in Section II. From an analysis of these macromodel simulation results, it is observed that the most significant spurs are due to the finite LO leakage to the IF port since, under no amplitude or phase mismatch, the image rejection of the SSB mixer is very high. The isolation between these ports can be improved by proper circuit and layout design techniques. It must

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TABLE VI SPURS ASSOCIATED WITH EACH BAND FREQUENCY FOR SYNTHESIZER ARCHITECTURE (III) WITH NONIDEALITIES

Fig. 10. Output spectrum of the synthesizer architecture (II) for the generation of the 8184-MHz tone.

be mentioned here that the LO leakage in the macromodel is implemented by a feed-forward path from the LO port adding at the output via a gain stage (with attenuation) and not through the LO leakage parameter of the model. To avoid any influence, the parameter in the model was set to a very low value. Next, simulations are performed for a worst case scenario with several nonidealities incorporated in the macromodels. These include – phase mismatch (5 ) in all paths, amplitude mismatch (5%) between the two signal paths in the SSB mixer blocks (see Fig. 8), and a frequency deviation of 10% in the center frequency of the bandpass filters. These are the most important nonidealities expected from an integrated implementation. Even though circuit implementations of frequency dividers are known to yield accurate quadrature outputs, signal routing effects such as crosstalk, loading, mismatch of parasitic components, etc. become relevant at gigahertz frequencies. For this reason, the effect of amplitude and phase mismatch for the signals across the synthesizer must be taken into account. The amplitude and phase mismatches are introduced in the macromodel by changing the gain factor of the gain block and changing the value in the delay token, respectively. It is also important to mention that the deviation considered for the center frequency

Fig. 11. Output spectrum of the synthesizer architecture (III) for the generation of the 9504-MHz tone.

Fig. 12. Macromodel for the evaluation of the impact of the synthesizer spurs on the BER of the UWB receiver.

in each of the bandpass filters is in a way that they enhance a spur while attenuating the fundamental tone. For example, the bandpass filter centered at 3960 MHz (Fig. 7) is shifted by 10%

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TABLE VII DIFFERENT EVALUATION SCENARIOS OF BER DEGRADATION DUE TO THE INTERFERENCE FROM PEER DEVICES

to lower frequencies, i.e., toward that of the alternate sideband frequency. Tables V and VI show the spurious tones produced during the synthesis of each of the 11 frequencies for architectures (II) and (III), respectively, in the described worst case conditions. Since in architecture (III) the 8448-MHz tone is the oscillator output and the 4224-MHz tone is generated by a divide-by-2, these tones do not create any spurious tones in the spectrum of interest and, hence, no spurs are shown for them in Table VI. It must be stressed here that because of the intermediate bandpass filters used in both the architectures, not many spurs appear in the generation of the reference tones. Figs. 10 and 11 show the output spectrum of synthesizer architecture (II) and (III), respectively, for the generation of one particular frequency. The spectrum is normalized with respect to the power of the frequency tone of interest. Fig. 10 shows the generation of the 8184-MHz tone by architecture (II). The most prominent spurious tones are at 9240, 10 296, 8712, and 7392 MHz. Likewise, Fig. 11 shows the generation of 9504-MHz tone by architecture (III), the most significant spurious tones being 10 560, 10 032, and 7920 MHz. The spurious tones generated by the synthesizer that are outside the UWB spectrum can cause interference to other communication systems and also down-convert their emissions; thereby corrupting the received signal. The first effect must be suppressed by proper antenna design and off-chip filtering. On the other hand, if the down-converted interference from other non-UWB devices is narrow-band, it can be tolerated to a certain degree by the inherent interference rejection capabilities of the OFDM with the coded QPSK constellation modulation format employed [18]. From Tables V and VI, it can be noted that both architectures produce spurs at 5808 and 5280 MHz, which fall in the U-NII band. Architecture (II) produces a tone at 5544 MHz, which overlaps with the band used by the HIPERLAN standard. The tones at 2376 and 2640 MHz are close, but not at the populated 2.4-GHz industrial–scientific–medical (ISM) band. It is important to note that neither of the architectures produces any spur in the range of 800 MHz to 2 GHz where the mobile phone [global system for mobile communication (GSM), digital enhanced cordless telecommunications (DECT)] and global positioning system (GPS) standards are located. Moreover, the spurs gener-

ated by both architectures comply with the FCC spectral mask requirements for UWB emissions. As shown in Fig. 2, the adverse effect of unwanted tones at frequencies within the UWB spectrum is that they down-convert the signals from peer UWB devices transmitting at the frequency of the spur, corrupting the signal from the band of interest. The impact of the synthesizer spurs on the BER of a direct-conversion receiver in the presence of other MB-OFDM UWB interferers is analyzed through a system-level model in SystemView. A conceptual description of the employed model is depicted in Fig. 12. The simulations consider the OFDM parameters described in [3] for a 480-Mb/s transmission, uncoded QPSK constellation, and AWGN channel. Under these conditions, the target BER is 10 . As shown with gray blocks in Fig. 12, in the SystemView model, each down-converted interferer is implemented with an independent random bit stream and an OFDM modulator with quadrature amplitude modulation (QAM) constellation. Before the addition to the signal of interest, each interferer is scaled by a factor , which represents the carrier-to-interference ratio at baseband. For example, if the interferer at frequency is received with a power 6 dB higher than the signal of interest and the synthesizer spur at frequency has a power of 26 dBc with respect to the tone of interest, then corresponds to 20 dB. For simplicity, only one interferer is included in Fig. 12. However, the actual simulation setup assumes that there is a UWB peer device transmitting in each of the ten bands different from the one of interest. In this pessimistic scenario, any spur from the synthesizer within the UWB spectrum down-converts an undesired peer transmission. For each architecture, a simulation is performed for the reception of each of the two bands for which the synthesizer shows the largest amount of spurs. The considered simulation scenarios are summarized in Table VII. The simulation results are shown in Figs. 13 and 14. Two different cases are evaluated for each received band; when each of the interferers has the same power as the signal of interest and when it has 6-dB higher power. In both figures, curve “a” represents the performance of the ideal receiver with no spurs in the LO, which is also equivalent to not having any interferer. Fig. 13 depicts the receiver performance in the worst spur scenarios for architecture (II), which correspond to the re-

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BER degradation due to interference from other UWB devices. Nevertheless, the obtained results for a pessimistic scenario with uncoded data remark upon the importance of frequency-planning and architectural design that yields the smallest amount of spurs for each generated frequency. VII. CONCLUSIONS

Fig. 13. BER degradation in the presence of peer interferences due to spurs in the LO from synthesizer architecture (II).

The most important considerations for the selection of a band plan and the design of frequency-synthesizer architecture for an MB-OFDM radio operating in the range of 3.1–10.6 GHz have been investigated. The relationship between the choice of band frequencies and synthesizer complexity has been analyzed. Based on this study, a frequency plan has been proposed to simplify the synthesizer implementation, reduce its power consumption, and potentially improve its spurious performance. Macromodels for the two proposed architectures (corresponding to an existent band plan and the proposed one) are built considering several nonidealities from an integrated implementation and without assuming any nonconventional circuit technique. The simulation results provide significant insight for the implementation of a UWB frequency synthesizer. A detailed summary of the expected spurious components for each generated frequency has been presented and the degradation in the BER performance of the receiver due to the synthesizer spurs in the presence of UWB interferers has been evaluated. It has been seen that both architectures can tolerate certain level of nonidealities expected from a hardware implementation without compromising the receiver performance in the presence of UWB interferers with comparable power. The analysis and simulation results indicate that a proper optimization of the synthesizer architecture can result in a frequency synthesis performance suitable for the operation of an MB-OFDM radio within the UWB spectrum allocated by the FCC. APPENDIX From Fig. 8, the SBRR can be derived as follows. The output of each mixer is given by (A.1)

Fig. 14. BER degradation in the presence of peer interferences due to spurs in the LO from synthesizer architecture (III).

(A.2) ception of the band at 10 296 MHz and the one at 8184 MHz. Fig. 14 shows the receiver performance in the worst spur scenarios for architecture (III), which are the reception of the band at 6336 MHz and the band at 8976 MHz (equivalent in terms of spurs to the reception of the 7920-MHz band). From the BER plots, it is important to observe that when the interferers have the same power as the signal of interest, the degradation in the performance is not significant ( 1 dB in SNR), and the amount of spurs does not seem to make a relevant difference, i.e., the degradation is dominated by the strongest spur. However, when the power of the interferers grows, the degradation in the performance is apparently stronger for the reception cases with a larger number of spurs. It is important to mention that the bit interleaving and forward error correction techniques employed in a complete MB-OFDM radio [3] are expected to reduce the

is a scaled version of given by

with the amplitude error and is

(A.3) Adding

and

at the output, we have

(A.4)

MISHRA et al.: FREQUENCY PLANNING AND SYNTHESIZER ARCHITECTURES FOR MB-OFDM UWB RADIOS

The fundamental tone and the sideband are given by

(A.5)

(A.6) SBRR is a ratio of the power level of the desired (fundamental) signal to that of the sideband and, hence, only the power of the signals at the fundamental and the sideband is required as follows:

(A.7)

(A.8) Hence, the SBRR is given by

(A.9)

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[5] J. Bergervoet, K. Harish, G. van der Weide, D. Leenaerts, R. van de Beek, H. Waite, Y. Zhang, S. Aggarwal, C. Razzell, and R. Roovers, “An interference robust receive chain for UWB radio in SiGe BiCMOS,” in IEEE Int. Solid-State Circuits Conf. Tech. Dig., Feb. 2005, pp. 200–201. [6] R. Harjani, J. Harvey, and R. Sainati, “Analog/RF physical layer issues for UWB systems,” in Proc. 17th Int. VLSI Design Conf., Jan. 2004, pp. 941–948. [7] A. Medi and W. Namgoong, “A fully integrated multi-output CMOS frequency synthesizer for channelized receivers,” in Proc. IEEE Int. Systems-on-Chip Conf., Sep. 2003, pp. 75–78. [8] D. Leenaerts, R. van de Beek, G. van der Weide, J. Bergervoet, K. S. Harish, H. Waite, Y. Zhang, C. Razzell, and R. Roovers, “A SiGe BiCMOS 1 ns fast hopping frequency synthesizer for UWB radio,” in IEEE Int. Solid-State Circuits Conf. Tech. Dig., Feb. 2005, pp. 202–203. [9] J. Lee and D. Chiu, “A 7-band 3–8 GHz frequency synthesizer with 1 ns band-switching time in 0.18 m CMOS technology,” in IEEE Int. Solid-State Circuits Conf. Tech. Dig., Feb. 2005, pp. 204–205. [10] C. Sandner and A. Wiesbauer, “A 3 GHz to 7 GHz fast-hopping frequency synthesizer for UWB,” in Joint Int. Ultra Wideband Systems Workshop/Ultrawideband Systems and Technologies Conf., May 2004, pp. 405–409. [11] C. Lin and C. Wang, “A regenerative semi-dynamic frequency divider for mode-1 MB-OFDM UWB hopping carrier generation,” in IEEE Int. Solid-State Circuits Conf. Tech. Dig., Feb. 2005, pp. 206–207. [12] A. Ismail and A. Abidi, “A 3.1 to 8.2 GHz direct conversion receiver for MB-OFDM UWB communications,” in IEEE Int. Solid-State Circuits Conf. Tech. Dig., Feb. 2005, pp. 208–209. [13] A. G. Armada, “Understanding the effects of phase noise in orthogonal frequency division multiplexing (OFDM),” IEEE Trans. Broadcast., vol. 47, no. 2, pp. 153–159, Jun. 2001. [14] T. Pollet, M. V. Bladel, and M. Moeneclaey, “BER sensitivity of OFDM systems to carrier frequency offset and Wiener phase noise,” IEEE Trans. Commun., vol. 43, no. 2, pp. 191–193, Feb./Mar./Apr. 1995. [15] K. V. Puglia, “Phase noise analysis of component cascades,” IEEE Micro, vol. 3, no. 4, pp. 71–75, Dec. 2002. [16] J. Tubbax, B. Come, L. Van der Perre, S. Donnay, M. Engels, H. De Man, and M. Moonen, “Compensation of IQ imbalance and phase noise in OFDM systems,” IEEE Trans. Wireless Commun., vol. 4, no. 3, pp. 872–877, May 2005. [17] SystemView: Elanix Advanced Dynamic System Analysis for Microsoft Windows, Elanix Inc., Westlake Village, CA, 2001. [18] A. Batra et al., “Multi-band OFDM physical layer proposal,” IEEE, Piscataway, NJ, IEEE 802.15-03/267r1-TG3a, Jul. 2003. [19] J. Rogers and C. Plett, Radio Frequency Integrated Circuit Design. Norwood, MA: Artech House, 2003. [20] B. Razavi, RF Microelectronics. Upper Saddle River, NJ: PrenticeHall, 1998. [21] T. H. Lee, The Design of CMOS Radio-Frequency Integrated Circuits. Cambridge, U.K.: Cambridge Univ. Press, 1998.

ACKNOWLEDGMENT The authors would like to thank J. Balakrishnan, N. Belk, and A. F. Mondragon-Torres, all of the Digital Signal Processing Solutions Research (DSPS) and Development Center, Texas Instruments Incorporated, Dallas, TX, for helpful discussions. REFERENCES [1] A. Batra, J. Balakrishnan, G. R. Aiello, J. R. Foerster, and A. Dabak, “Design of a multiband OFDM system for realistic UWB channel environments,” IEEE Trans. Microw. Theory Tech., vol. 52, no. 9, pp. 2123–2138, Sep. 2004. [2] “First report and order, revision of part 15 of the Commission’s rules regarding ultra-wideband transmission systems,” FCC, Washington, DC, ET Docket 98-153, Feb. 14, 2002. [3] A. Batra et al., “Multi-band OFDM physical layer proposal for IEEE 802.15 Task Group 3a,” IEEE, Piscataway, NJ, IEEE P802.15-03/268r3TG3a, Mar. 2004. [4] B. Razavi, T. Aytur, F.-R. Yang, R.-H. Yan, H.-C. Kang, C.-C. Hsu, and C.-C. Lee, “A 0.13 m CMOS UWB transceiver,” in IEEE Int. SolidState Circuits Conf. Tech. Dig., Feb. 2005, pp. 216–217.

Chinmaya Mishra (S’03) received the B.E. (Hons.) degree in electrical and electronics engineering from Birla Institute of Technology and Science, Pilani, India in 2002, the M.S. degree in electrical engineering from Texas A&M University, College Station, in 2004, and is currently working toward the Ph.D. degree at the Analog and Mixed Signal Center (AMSC), Texas A&M University. In Spring 2002, he was with the DSP Design Group, Texas Instruments India Pvt. Ltd., Bangalore, India, where he worked on formal verification of built-in self-test controllers for memories. During the summer of 2005, he was with WiQuest Communications Inc., Allen, TX, where he was responsible for the design of a frequency synthesizer for a UWB radio. His research interests include RF and microwave circuit design in silicon for UWB applications and low-voltage low-power analog circuit design.

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Alberto Valdes-Garcia (S’00) was born in 1978 and grew up in San Mateo Atenco, Mexico. He received the B.S. degree in electronic systems engineering (highest honors) from the Monterrey Institute of Technology (ITESM), Campus Toluca, Mexico, in 1999, and is currently working toward the Ph.D. degree in electrical engineering at the Analog and Mixed-Signal Center (AMSC), Texas A&M University, College Station. In 2000, he was a Design Engineer with the Broadband Communications Sector, Motorola. From 2001 to 2004, he was a Semiconductor Research Corporation (SRC) Research Assistant at the AMSC working on the development of analog and RF built-in testing techniques. In the summer of 2002, he was with the Read Channel Design Group, Agere Systems, where he investigated wide tuning range gigahertz LC VCOs for mass storage applications. During the summer of 2004, he was with the Mixed-Signal Communications IC Design Group, IBM T. J. Watson Research Center, where worked on the design and analysis of millimeter-wave SiGe power amplifiers. His current research involves system-level and RF circuit design for UWB communications. Mr. Valdes-Garcia has been the recipient of a scholarship from the Mexican National Council for Science and Technology (CONACYT) since the fall of 2000. In 2005, he was the recipient of the Doctoral Thesis Award from the IEEE Test Technology Technical Council (TTTC).

Faramarz Bahmani (S’01) received the M.Sc. (Hons.) degree in electronics engineering from the Tehran University, Tehran, Iran, in 2000, and is currently working toward the Ph.D. degree at the Texas A&M University, College Station. He interned as a Circuit Design Engineer at Alvand Technology Inc., Santa Clara, CA, during the summer of 2005. His research interests include high-speed analog and mixed-signal circuits, highly linear continuous-time filters, and PLL-based frequency synthesizers.

Anuj Batra (M’00) received the B.S. degree (with distinction) in electrical engineering from Cornell University, Ithaca, NY, in 1992, the M.S. degree in electrical engineering from Stanford University, Stanford, CA, in 1993, and the Ph.D. degree in electrical engineering from the Georgia Institute of Technology, Atlanta, in 2000. In 1992, he was with Raytheon E-Systems, Falls Church, VA, where he designed algorithms for a software-defined radio based on the Advanced Mobile Phone Service (AMPS) Standard. In 2000, he joined the Digital Signal Processing Solutions (DSPS) Research and Development Center, Texas Instruments Incorporated (TI), Dallas, TX. Since 2002, he has helped begin an internal UWB development effort within TI and co-developed the time-frequency interleaved OFDM (TFI-OFDM) proposal, which served as the foundation for the MultiBand OFDM proposal. This proposal defines a wireless UWB-based PHY for high-speed communications (up to 480 Mb/s). In addition, he serves as the PHY Technical Chair for the WiMedia/MBOA, a partnership of over 119 of the companies in the CE, PC, home entertainment, semiconductor, and digital imaging segments. In 2004, he was named one of the world’s 100 Top Young Innovators by Technology Review, MIT’s Magazine of Innovation. He is currently a Member, Group Technical Staff with TI. His research interests are in the areas of wireless communications, in particular, the design of high-speed wireless networks, multiuser detection theory, and coexistence between unlicensed wireless devices. Dr. Batra is a member of Eta Kappa Nu and Tau Beta Pi. Since joining TI, he has also been involved in standardization activities for MBOA Special Interest Group (SIG), IEEE 802.15.3a, IEEE 802.11g, IEEE 802.15.2, and Bluetooth SIG.

Edgar Sánchez-Sinencio (F’92) was born in Mexico City, Mexico. He received the communications and electronic engineering degree (Professional degree) from the National Polytechnic Institute of Mexico, Mexico City, Mexico, in 1966, the M.S.E.E. degree from Stanford University, Stanford, CA, in 1970, and the Ph.D. degree from the University of Illinois at Urbana–Champaign, in 1973. In 1974, he held an industrial Post-Doctoral position with the Central Research Laboratories, Nippon Electric Company Ltd., Kawasaki, Japan. From 1976 to 1983, he was the Head of the Department of Electronics at the Instituto Nacional de Astrofísica, Optica y Electrónica (INAOE), Puebla, Mexico. He was a Visiting Professor with the Department of Electrical Engineering at Texas A&M University, College Station, during the academic years of 1979–1980 and 1983–1984. He is currently the Texas Instruments Incorporated (TI), Dallas, TX, J. Kilby Chair Professor and Director of the Analog and Mixed-Signal Center at Texas A&M University. He coauthored Switched Capacitor Circuits (New York: Van Nostrand-Reinhold, 1984) and coedited Low Voltage/Low-Power Integrated Circuits and Systems (Piscataway, NJ: IEEE Press, 1999). His current interests are in the area of RF communication circuits and analog and mixed-mode circuit design. Dr. Sánchez-Sinencio was the general chairman of the 1983 26th Midwest Symposium on Circuits and Systems (CAS). He was an associate editor for IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMS (1985–1987), and an associate editor for the IEEE TRANSACTIONS ON NEURAL NETWORKS. He is the former editor-in-chief of the IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMS—II: ANALOG AND DIGITAL SIGNAL PROCESSING. He is a former IEEE CAS vice president–publications. He was the IEEE CAS Society representative to the Solid-State Circuits Society (2000–2002). He was a member of the IEEE Solid-State Circuits Society Fellow Award Committee (2002–2004). He is currently a member of the IEEE CAS Society Board of Governors. In November 1995, he was awarded an Honoris Causa Doctorate by the National Institute for Astrophysics, Optics and Electronics, Puebla, Mexico, which was the first honorary degree awarded for microelectronic circuit design contributions. He was corecipient of the 1995 Guillemin-Cauer for his work on cellular networks. He was also the corecipient of the 1997 Darlington Award for his work on high-frequency filters He was the recipient of the IEEE CAS Society Golden Jubilee Medal in 1999.

Jose Silva-Martinez (SM’98) was born in Tecamachalco, Puebla, Mexico. He received the B.S. degree in electronics from the Universidad Autónoma de Puebla, Puebla, Mexico, in 1979, the M.Sc. degree from the Instituto Nacional de Astrofísica Optica y Electrónica (INAOE), Puebla, Mexico, in 1981, and the Ph.D. degree from the Katholieke Univesiteit Leuven, Leuven, Belgium, in 1992. From 1981 to 1983, he was with the Electrical Engineering Department, INAOE, where he was involved with switched-capacitor circuit design. In 1983, he joined the Department of Electrical Engineering, Universidad Autónoma de Puebla, where he remained until 1993. He was a co-founder of the graduate program on opto-electronics in 1992. From 1985 to 1986, he was a Visiting Scholar in the Electrical Engineering Department, Texas A&M University, College Station. In 1993, he rejoined the Electronics Department, INAOE, and from May 1995 to December 1998, was the Head of the Electronics Department. He was a co-founder of the Ph.D. program on electronics in 1993. He is currently with the Department of Electrical Engineering (Analog and Mixed Signal Center) Texas A&M University, where he is an Associate Professor. His current field of research is in the design and fabrication of integrated circuits for communication and biomedical application. Dr. Silva-Martinez has served as IEEE Circuits and Systems (CAS) Society vice president Region-9 (1997–1998) and as associate editor for the IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMS—II: ANALOG AND DIGITAL SIGNAL PROCESSING (1997-1998 and 2002-2004) and for the IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMS—I: FUNDAMENTAL THEORY AND APPLICATIONS (since 2004). He was the main organizer of the 1998 and 1999 International IEEE-CAS Tour in region 9, and chairman of the International Workshop on Mixed-Mode Integrated Circuit Design and Applications (1997–1999). He is the inaugural holder of the Texas Instruments Incorporated (TI) Professorship-I in Analog Engineering, Texas A&M University. He was a corecipient of the 1990 European Solid-State Circuits Conference Best Paper Award.

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BPSK to ASK Signal Conversion Using Injection-Locked Oscillators—Part I: Theory José María López-Villegas, Senior Member, IEEE, and Javier Jose Sieiro Cordoba, Member, IEEE

Abstract—This paper presents a new method and circuit for the conversion of binary phase-shift keying (BPSK) signals into amplitude shift keying signals. The basic principles of the conversion method are the superharmonic injection and locking of oscillator circuits, and interference phenomena. The first one is used to synchronize the oscillators, while the second is used to generate an amplitude interference pattern that reproduces the original phase modulation. When combined with an envelope detector, the proposed converter circuit allows the coherent demodulation of BPSK signals without need of any explicit carrier recovery system. The time response of the converter circuit to phase changes of the input signal, as well as the conversion limits, are discussed in detail. Index Terms—Amplitude shift keying (ASK), bifurcation, converters, injection-locked oscillator (ILO), phase shift keyings (PSKs).

I. INTRODUCTION

T

HE DIGITAL phase shift keying (PSK) of a sinusoidal signal is one of the most efficient modulation techniques, both in terms of noise immunity and required bandwidth. Coherent demodulation is the preferred procedure to demodulate PSK signals, especially when optimum error performance is of particular importance [1]. Coherent demodulation requires the availability of a local carrier having the same frequency and phase than the received modulated carrier. However, frequency and/or phase deviations degrade the detection process and, consequently, the system performance. Therefore, local carrier synchronization is a critical issue in most digital communication systems. Carrier recovery is accomplished by using synchronization loops [2]–[5]. The most widely used are the squaring loop and Costas loop, shown in Fig. 1(a) and (b), respectively. However, synchronization time is usually large, leading to loss of data at the beginning of a communication or malfunctioning in burst mode transmissions. Noncoherent demodulation of PSK signals can overcome this problem, however, noise immunity is worst and the bit period has to be known [1]. This study proposes an alternative method for the demodulation of binary phase-shift keying (BPSK) signals, which is based on the use of a coherent BPSK to amplitude shift keying (ASK) converter [6], as depicted in Fig. 2. A simple envelope detector cascaded to the converter acts as the final stage of the demodulation system. The operation of the BPSK to ASK converter

Manuscript received January 31, 2005; revised July 8, 2005. This work was supported by the Spanish Science and Technology Commission under Project TIC2001-2947-C02-01. The authors are with the RF Group, Department of Electronics, University of Barcelona, E-08028 Barcelona, Spain (e-mail: [email protected]). Digital Object Identifier 10.1109/TMTT.2005.859875

Fig. 1. Block diagram of typical BPSK demodulation schemes using carrier recovery systems. (a) Squaring loop. (b) Costas loop.

Fig. 2.

Block diagram of the proposed BPSK demodulation scheme.

relies on two main principles: the superharmonic injection of oscillators and interference phenomena. The former is used to lock the oscillators (in frequency and phase) with the incoming signal, whereas the later is used to generate an amplitude interference pattern that reproduces the original phase modulation. Section II is devoted to the analysis of injection-locked oscillators (ILOs), whereas Section III covers the conversion mechanism, dynamics, and limitations of the coherent BPSK to ASK converter.

0018-9480/$20.00 © 2005 IEEE

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In a first approximation, the applied voltage is the addition and the fundamental component of the of the injected signal oscillation signal , which can be expressed as follows:

(3)

Fig. 3.

Circuit schematic of a second harmonic ILO.

II. ILOs Injection is a usual way to synchronize an oscillator with an incident signal. The injection-locking mechanism, first analyzed by Van der Pol in 1927 [7], has been investigated by several auharthors [8]–[11]. When an injected signal is close to the monic of the free running frequency of the oscillator, the ensemble is known as superharmonic or th harmonic ILO. In the locked state, superharmonic ILOs act as frequency dividers, with the dividing factor being the nearest harmonic order of the ILO to the injected signal. For that reason, these components are also known as injection-locked frequency dividers (ILFDs) [12], [13]. The output of a superharmonic ILO or ILFD could be in any of possible phase states ( being the harmonic order). This is due to the phase uncertainty introduced by the process of frequency division. For example, in the case of a second harmonic ILO, the output frequency is half the frequency of the injected signal and phase uncertainty is equal to . In this study, the interest is focused in a particular implementation of a second harmonic ILO, of which a schematic is shown in Fig. 3. The circuit is a cross-pair oscillator, whose resonant tank consists of an inverter transformer and a pair of varactor diodes. The input port of the injected signal is the center tap of the transformer. Ideally, under common-mode excitation, the transformer acts as a short circuit, therefore, the injected signal is found without distortion at the terminals of the varactors. Due to the nonlinear behavior of the stored charge dependence up , on the applied voltage, the injected signal, at frequency . As mixes with the oscillator signal at frequency close to a consequence, a new current component appears, which modifies the characteristics of the resonant tank. In order to quantify this process, let us consider a second-order approximation for the stored charge versus applied voltage at the varactors

is the voltage amplitude of the fundamental compowhere is the voltage amplitude of the injected nent of the oscillator, is the frequency in the locked state, and and signal, are the corresponding phases. Note that the time dependence of takes into account the evolution of the oscillator frequency from the free-running state to the locked state (i.e., at , , with being the oscillator’s free-running frequency). at any time , i.e., Provided that , by substituting in (2) and after some calculation to evaluate the mixing terms, one can obtain the following expression for the current at the fundamental frequency passing through the varactors:

(4) where angle

is given by (5)

The first term in (4) is the displacement current related to the capacitance of the varactors. The second term is an in-phase current, which can be positive (i.e., dissipative) or negative (i.e., regenerative) depending on the value of angle . This current is responsible for the oscillation rise in parametric analog frequency dividers [12]. According to (4), the capacitance of the varactors in the is given by locked state (6) changes depending on the amplitude of the injected signal and angle . Therefore, it implies a change in the osciland, through (5), also lator frequency, which is reflected in . The differential equation governing the dynamics back in of this process can be obtained through the instantaneous oscillation frequency (7)

(1) is a constant charge depending on the bias conditions, where is the small-signal capacitance at the bias point, is the applied voltage, and is a second-order coefficient evaluated at the bias point of the varactors. According to (1), the current passing through the varactors is given by (2)

where is half the inductance of the inverter transformer working in differential mode. Taking into account that , , and (i.e., ), one finally obtains (8) Equation (8) explains the dynamic of the locking process through the evolution of over time. It establishes the steady-

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A quasi-linear behavior is observed for small values of the varactors bias shift. Note that the maximum and minimum and , respectively, which, achievable values of are according to (5), leads maximum and minimum output phase of and . B. Locking Range and Locking Sensitivity The locking range of the second harmonic ILO can easily be function must be obtained from (9). The argument of the kept from 1 to 1, thus, (11) Fig. 4. Graphical representation of the stability analysis of (8) in the steady state.

A similar expression has been previously reported by other authors [12]. It should be noted that, in contrast with the fundamental locking analysis [8], [9] the second harmonic injection-locking leads to a locking range not dependent on the resonator quality factor. This is due to the fact that injection at the second harmonic does not force the oscillator to oscillate out of its natural or resonant frequency, but actually it changes this frequency. Equation (11) also establishes a relationship between the freand the minimum injected power requency shift quired for locking. According to (11), one can expect (12)

Fig. 5. Steady-state solution of (8) versus varactors bias shift.

state condition, locking range, locking sensitivity, and dynamic behavior. A. Steady-State Solution In the steady state, the value of tions of (8) are two fixed values of angle

is 0. Thus, the solu-

That is to say, the locking sensitivity varies as the relative frequency shift to the square. C. Dynamic Behavior To analyze the dynamic behavior of the locking process, we need to solve (8). Depending on the boundary conditions, we can consider several cases as follows. 1) Phase Locking: In this case, (i.e., ) and (8) reduces to

(9) (13) and Its solution can be written as (10) As is schematically depicted in Fig. 4, any perturbation around will be automatically compensated. Consequently, is will be stable. On the contrary, any perturbation around amplified, indicating that it is an unstable value. According to (9), the second harmonic ILO shown in Fig. 3 can be used as a continuous phase modulator [14]. In the locked state, the output frequency of the second harmonic ILO is always half the injected frequency. A change in the varactors bias conditions then produces a phase shift instead of a frequency has shift. This fact can be observed in Fig. 5, where angle been plotted against the varactor’s bias shift. To evaluate , , and , the depletion capacitance formula of a p–n junction has been used. Moreover, the zero of the bias shift scale corresponds (i.e., ). to the bias value at which

(14) where denotes the initial condition of the phase for time is given by

The

(15) Fig. 6 shows the transient responses of for different values of as a function of the normalized time . In this particular case, regardless of the injected power, the phase locking always takes place. According to (14), (15), and Fig. 6, the input power only determines the speed at which equilibrium conditions are reached, i.e., the higher the power, the faster the response. Consequently, provided the injection time is long

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Fig. 6. Time-dependent solution of (8) in the case of phase locking.

Fig. 8. Time-dependent solution of (8) in the locking threshold.

power is required to assure locking. This situation will be analyzed in the next case. 3) Locking Threshold: This case corresponds to the condi, i.e., . Here, (8) can tion be expressed as follows: (19)

Fig. 7. Time-dependent solution of (8) in the case of phase and frequency locking.

enough, any signal at frequency (for instance noise) can lock the oscillator and change its output phase. 2) Phase and Frequency Locking: This case corresponds and to the double condition . The solution of (8) is now given by

The minus sign applies when and the plus sign applies when . Under these conditions, the solution of (19) is given by

(20)

where

(21)

(16) where (17) and

(18)

(i.e., ), The upper sign applies when (i.e., ). and the lower sign when for Fig. 7 shows an example of the transient response of as a function of normalized time . In different values of this example, angle . It is clearly observed in Fig. 7 that angle acts as an attractor of the dynamic phase trajectoacts as a scatterer. As in previous case 1), ries while angle the injected power determines the speed at which the equilibrium conditions are reached. However, now a minimum injected

Once again, in these equations, the upper sign applies when (i.e., ), and the lower sign applies when (i.e., ). for different values of as a The transient response of is shown in Fig. 8. In this case, function of normalized time is equal to . At the locking threshold, the injected power is the minimum required to assure locking. Under these conditions, the ILO is very unstable. Consequently, any perturbation at the input (for instance, noise) can unlock the oscillator. . 4) Unlocking: In this case, Consequently, at any time , and locking does not occurs, i.e., the injected power is not big enough to assure the locking of the oscillator. Under these conditions, the solution of (8) is given by

(22) where (23)

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Fig. 9. Time-dependent solution of (8) in the unlocked state. The continuous line corresponds to = 2 and the dashed line corresponds to = 1:25. Fig. 11. Schematic representation of the voltage waveforms evolution at varactors leads when the phase of the injected signal changes in . The dashed line corresponds to the fundamental voltage waveform just at the time when the input phase changes. This waveform can evolve in two ways to again reach the steady-state conditions: (a) by decreasing the waveform phase in =2 or (b) by increasing the waveform phase in the same amount.

Fig. 10. Schematic representation of voltage waveforms at varactors leads. The thick line corresponds to the fundamental component and the thin line corresponds to the injected signal. (a) and (b) identify the two possible states due to the phase uncertainty. Marker squares have been inserted to highlight that the same phase relationship between the fundamental component and injected signal is observed in both cases.

and

(24)

The minus sign applies when and the plus sign ap. plies when as a function of the norFig. 9 depicts the dynamics of malized time for two values of parameter . The continuous and the dashed line corresponds to line corresponds to . For comparative purposes, both curves have the same shows a periinitial conditions . It can be observed that odic behavior of which the period depends on the parameter. approaches 1, the period increases and becomes infinity As . This limit situation corresponds to the previous for analysis case of the locking threshold. III. BPSK TO ASK CONVERSION A. Conversion Mechanism Let us consider a second harmonic ILO injected by a reference signal. Frequency and power of the injected signal are such as to assure both frequency and phase locking of the oscillator. According to (5), there are two possible values of the output phase that verify the locking conditions, which differ in . This is schematically depicted in Fig. 10.

Fig. 12. Schematic representation of the evolution of angle when the input phase changes in . Label (a) identifies the case when the steady-state value > 0 and (b) the case < 0.

Now let us assume that the phase of the injected signal changes in . The locking phase conditions are no longer satisfied, thus, the output phase of the ILO changes to again reach the steady state. As is shown in Fig. 11, the change in the output or indistinctly. phase could be either According to our previous analysis of the ILO’s dynamics, when the phase of the input signal changes in , the angle changes from it steady-state value to . Depending on whether the value of is positive (i.e., ) or negative ), the transient of back to the steady-state (i.e., or a decrease conditions will consist in an increase , respectively. This situation is shown in Fig. 12. Taking into account (5), an increment leads to an increment . Hence, the of the output phase of the oscillator response of a phase change of the BPSK input signal is a change when of the phase of the oscillator output or when . Let us now consider the circuit shown in Fig. 13. A BPSK signal of frequency is injected to both ILOs, e.g., using a power splitter. The total injected power is assumed to be enough to assure frequency and phase locking of both oscillators. The are biased so that the free-running frequency varactors of . The varactors of are also biased so that . Finally, the outputs of both and are combined together using a power combiner.

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Fig. 13.

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 53, NO. 12, DECEMBER 2005

Circuit block diagram of the proposed BPSK to ASK converter.

As starting point for our analysis of the BPSK to ASK conversion, and without loss of generality, we can assume that the is the solution of (5) with output phase of the first oscillator (i.e., the steady-state value of for the first oscil, i.e., lator) and

Fig. 14. Phasor diagram of the BPSK to ASK converter circuit. Label (a) corresponds to an arbitrary initial condition for which the superposition of the outputs O and O give a maximum and (b)–(d) correspond to successive changes in of the input phase.

(25) Analogously, (26) where is the output phase and is the steady-state value of for . Under these conditions, the outputs of both and can be expressed as oscillators

(27) This situation is described in the phasor diagram of Fig. 14(a). At the output of the power combiner, they will mainly constructively interfere leading to maximum output amplitude (28) If the phase of the BPSK input signal changes in phase of phase of changes in (27), the outputs are given by

changes in , the output , while the output . Similarly to

(29) However, both outputs will now mainly destructively interfere, leading to minimum amplitude at the output of the power combiner (30)

Fig. 15. Output waveform of the converter circuit of Fig. 13 in response to successive changes in of the input phase. Labels (a)–(d) correspond to the equivalent phasor states in Fig. 14.

as shown in Fig. 14(b). Further changes in of the phase of the BPSK input signal will cause consecutive switches between minimum and maximum output amplitudes following the sequence given in Fig. 14(c), (d), (a), and (b) and so on. Hence, the resulting interference pattern reproduces the phase changes of the injected BPSK signal. In others words, the circuit diagram of Fig. 13 effectively down converts the BPSK input signal at into an ASK output signal at frequency . frequency and delimit a converIt is noteworthy that frequencies sion channel. Only locking frequencies inside the range correspond to BPSK input signals, which are properly converted into ASK signals. Locking frequencies outside this range will cause phase changes at the output of both oscilla( and ) or ( tors of and ) when the input phase changes in . Consequently, no interference pattern will be generated. B. Bit Rate Estimation Fig. 15 shows an example of the output waveform of the converter circuit of Fig. 13 as a function of the normalized time . The time-domain output signals of both oscillators have been

LÓPEZ-VILLEGAS AND SIEIRO CORDOBA: BPSK TO ASK SIGNAL CONVERSION USING ILOs—PART I

Fig. 17.

Fig. 16. Continuous closed lines are the contours of constant normalized transient time. Dashed lines are the contours of constant normalized conversion channel. Both set of curves are represented in the ( ; ) plane.

Normalized bit rate as a function of the normalized channel width.

in (31), the following expression is obtained for the maximum conversion channel width:

0

computed using an arbitrary carrier frequency and the analytical expressions of the phase behavior (14) to (24). This figure shows the resulting interference pattern in response to consecutive phase changes of the BPSK input signal. Here, input phase changes take place every ten normalized time units. After each input phase change, a transient response of the output can be observed lasting approximately five normalized time units. For a better understanding, letter labels in this figure correspond to the different steady states of Fig. 14. It is noteworthy that the conversion dynamics of the new converter circuit is significantly different than that of the classical demodulators based on synchronization loops. In the new approach, the BPSK to ASK converter does not lock to recover the carrier and then demodulates, but relocks for every single bit (i.e., phase change of the input signal). Consequently, the new demodulation scheme will be very useful in burst mode communication where the classical approach fails due to the usually large locking time. The transient response of the converter circuit will be one of the limiting factors of the maximum achievable bit rate. Only can be properly downBPSK signals having a bit rate converted into ASK signals, with being the total transient only time. Using normalized time units, the transient time and . Fig. 16 shows the depends on the steady-state values contours (continuous lines) on the projection of constant plane . Transient times have been evaluated as the required time to achieve 95% of the total output phase change after a sudden input phase change of . In Fig. 16, there are dashed curves corresponding to constant values of . Taking into account that , it can be seen that these curves also correspond to . In constant values of the conversion channel width fact, according to (9), one can obtain (31) For a given injected power, the maximum value of is achieved for . Substituting this condition

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(32) By combining (31) and (32), one finally obtains the following expression for the normalized channel width: (33) According to data in Fig. 16, for a given conversion channel width, the minimum transient response is observed when . Under such condition, the locking frequency is cenand . Moreover, the absolute minimum of tered between the transient time is found to be approximately 3.6 normalized . time units for Fig. 17 shows the normalized bit rate as a function of the normalized channel width. It is noteworthy that a maximum is achieved for a converbit rate of approximately sion channel width equal to the 55%–60% of the maximum . Equivalently, for a given conversion channel width value , the maximum achievable bit rate will be approxiprovided the converter is operating under mately optimal conditions. C. Conversion Dynamics The above discussion of the conversion dynamics has been done considering a steep phase change of the BPSK input signal. However, this phase change usually takes place during a certain transition time . Among the multiple possible trajectories going from one symbol to the other of a BPSK signal, we have consider in detail three simple cases, which are shown in Fig. 18. The first one, labeled , corresponds to a constant decrease of the injected amplitude, a phase change of when the amplitude reaches zero, and a constant amplitude increase up to the initial value. This trajectory is the result of the multiplication of a carrier signal with a quasi-step function from 1 to 1 or vice versa having a certain rise or fall time, respectively. The second trajectory, labeled , corresponds to a continuous phase increase in during the transition time, keeping constant the signal amplitude. Finally, the third trajectory corresponds to a constant signal amplitude during the same time. and a continuous phase decrease in We next analyze these three cases through the evaluation of in Fig. 13, i.e., the one with the condition .

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Fig. 18. Simple trajectories in the constellation diagram between both symbols of a BPSK signal.

1) Case : The study of the dynamic behavior of the converter in response to a phase change of the BPSK signals must be carried out considering three different stages. First, the amplitude of the injected signal decreases from its initial value to zero during a normalized time . Second, the phase of the injected signal suddenly changes in and the amplitude increases from zero to the initial value , also during a time . Finally, the converter system evolves until an equilibrium condition is reached once again. Assuming a linear decrease of the injected amplitude, the dynamics of the first stage is governed by the following differential equation:

(34) and where is the fall time. The domain of integration is the initial condition is . For the second stage, the differential equation to be solved is given by

(35) and the initial conditions being the integration domain . Finally, the differential equation for the third stage is given by (8), which can be rewritten as (36) with an integration domain and the initial condition . Once the solutions for the three stages are known, the output phase change of the converter can be evaluated. According to (5), one obtains

1 ()

Fig. 19. Plot of output phase 8 t as a function of the normalized time for different values of the transit time . The insert shows the BPSK waveform corresponding to the trajectory labeled “a” in Fig. 18.

Equations (34) and (35) have been solved numerically, whereas the solution of (36) is given by (16)–(18). The same , procedure can be applied for solving the output phase of the one with . From the symmetry of the problem, similar expressions will be found. in the case Fig. 19 shows the global solution of and for several values of the normalized transit time . The inset in this figure illustrates the generation procedure of the input BPSK signal. It should be noted that shows a bifurcation behavior the output phase change is smaller depending on the normalized transit time . If than the bifurcation time (in the example shown in Fig. 19, ), then the output phase shift is , as (i.e., ). On the contrary, expected when if the transit time is injected, amplitude increases above the locking threshold. On the contrary, for a long time interval of the oscillators in the free-running state, a wrong final output phase will be reached. Further increase of the transit time reveals the existence of additional bifurcation times, which delimits zones of right and wrong behavior of the converter. Accordingly, the first bifurcation time must be understood as the maximum acceptable fall and/or rise time of the trapezoidal waveform used to generate the BPSK signal. Bifurcation phenomena in harmonic-injected dividers have been analyzed by other authors [15]. However, the analysis has been carried out taking into account only the injected power and not the input phase change dynamics. 2) Cases and : In these cases, we have to take into account two different stages to analyze the dynamics of the converter in response to a phase change of the BPSK signals. First, the phase of the injected signal increases or decreases linearly in during a certain transit time (factor 2 is included for comparative purposes with the previous case). Second, the converter system evolves until an equilibrium condition is again reached. According to (5)–(7), the dynamics of the first stage is governed by the differential equations (38)

(37)

that can be rewritten as (39)

LÓPEZ-VILLEGAS AND SIEIRO CORDOBA: BPSK TO ASK SIGNAL CONVERSION USING ILOs—PART I

Fig. 20. Plot of output phase (t) as a function of the normalized time for different values of the transit time . The insert shows the BPSK waveform corresponding to the trajectory labeled “b” in Fig. 18.

with being by

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Fig. 21. Plot of output phase as a function of the normalized time for different values of the transit time . The insert shows the BPSK waveform corresponding to the trajectory labeled “c” in Fig. 18.

an equivalent injection frequency, which is given

(40) , and the initial conThe integration domain of (39) is and (this last value can be ardition is , the solution bitrarily assigned). Depending on the value of will be one of those described in Section II-C. According to the previous discussion, the transit between symbols of the BPSK signal has to be interpreted just as a to . change in the injection frequency from For the second stage, the equation to solve is given by (8), which, in this case, can be expressed as (41) with the integration domain being and the initial con. dition being Finally, the converter’s output phase change is given by

(42) The plus sign applies when the phase on the input signal increases from 0 to , while the minus sign applies when the phase . decreases from 0 to for and several values Figs. 20 and 21 depict of the normalized transit time , assuming a linear varifrom 0 to , respectively. In both cases, the inset ation of illustrates the generation procedure of the input BPSK signal. In the first case, shown in Fig. 20, the locking frequency increases during the transit time, whereas in by an amount equal to the second case in Fig. 21, the locking frequency decreases by the same amount. It is important to note that the bifurcation behavior is only observed in the second case. To understand these results, we have to take into account that trajectories and are equivalent when the transit time is very short. In that case, the injected oscillators only react to the final value of phase , which is the same in both cases. This can be observed in both Figs. 20 and 21 as an output phase change in

Fig. 22. Normalized bifurcation time as a function of angle for trajectories labeled “a” and “c” in Fig. 18.

, as expected when (i.e., ). On the contrary, if the transit time is long enough, the injected oscillators follow the input phase changes and then the final value of the output phase will depend on the input phase trajectory. In case , and no bifurcation appears. However, in case , and the bifurcation behavior is observed. Opposite because (i.e., results would be obtained for ) is considered. In any case, the bifurcation time must be considered the maximum acceptable transit time between symbols of the BPSK input signal. To conclude, Fig. 22 shows the normalized bifurcation time as a function of angle for trajectories and . Note the important differences in behavior related to the different nature of the bifurcation phenomena. BPSK signals generated using a schema in accordance with trajectory will be better demodulated using strong injected oscillators (i.e., small values of ). On the contrary, BPSK signals generated according to trajectories or will be better demodulated using weak injected oscillators. IV. CONCLUSION A new method and circuit to convert BPSK signals into ASK signals based on the use of second harmonic ILOs has been presented. First, the second harmonic ILOs have been analyzed in detail. Their dynamics in response to phase changes of the injected signal has been studied exhaustively. Hence, an analytical expression, describing the oscillator response to phase changes of

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the BPSK input signal, has been obtained for all possible locking conditions. Second, the conversion mechanism, based on frequency and phase synchronization and interference phenomena, has been studied in detail. As a result, the existence of a conversion channel, controlled by external bias, has been pointed out. Moreover, the out-of-channel rejection of the converter has been discussed in detail. In addition, the dynamics of the conversion mechanism has also been extensively analyzed. The maximum achievable modulation bit rate has been studied as a function of the locking conditions of the converter. From this study, the optimum operating conditions of the converter have been derived, and a relationship between the conversion channel width and the maximum bit rate has been established. Finally, the limitations of the conversion process related to the characteristics of the BPSK signal (i.e., nonnull transit time from one symbol to another) have been considered. Three relevant cases have been studied exhaustively and, as a result, the existence of a bifurcation behavior of the converter response has been evidenced. From this, maximum acceptable transit times between symbols of the input BPSK signal have been obtained. REFERENCES [1] B. Sklar, Digital Communication Fundamentals and Applications. Upper Saddle River, NJ: Prentice-Hall, 1988. [2] J. P. Costas, “Synchronous communications,” Proc. IRE, vol. 47, pp. 2058–2068, 1959. [3] G. L. Do and K. Feher, “An ultra-fast carrier recovery versus traditional synchronizers,” IEEE Trans. Broadcast., vol. 42, no. 1, pp. 42–49, Mar. 1996. [4] S. Mirabbasi, S. Gazor, and K. Martin, “A wide-band carrier-recovery system for multilevel QAM signals,” in Proc. IEEE Int. Circuits Systems Symp., 2000, pp. IV-661–IV-664. [5] H. Sari and S. Moridi, “New phase and frequency detectors for carrier recovery in PSK and QAM systems,” IEEE Trans. Commun., vol. 36, no. 9, pp. 1035–1043, Sep. 1988. [6] J. M. Lopez-Villegas, J. G. Macias, J. Cabanillas, J. J. Sieiro, J. A. Osorio, J. Samitier, J. Bausells, J. Montserrat, and E. Cabruja, “BPSK to ASK converter for RF digital communications,” in IEEE Radio Frequency Integrated Circuits Symp. Dig., 2003, pp. 643–646. [7] B. Van der Pol, “Forced oscillations in a circuit with nonlinear resistance,” Phil. Mag., vol. 3, pp. 65–80, Jan. 1927. [8] R. Adler, “A study of locking phenomena in oscillators,” Proc. IRE, vol. 34, no. 6, pp. 351–357, Jun. 1946. [9] , “A study of locking phenomena in oscillators,” Proc. IEEE, vol. 61, no. 10, pp. 1380–1385, Oct. 1973. [10] M. T. Jezewski, “An approach to the analysis of injection-locked oscillators,” IEEE Trans. Circuits Syst., vol. CS-21, no. 5, pp. 395–401, May 1974.

[11] E. F. Calandra and A. M. Sommariva, “Stability analysis of injectionlocked oscillators in their fundamental mode of operation,” IEEE Trans. Microw. Theory Tech., vol. MTT-29, no. 11, pp. 1137–1143, Nov. 1981. [12] H. R. Rateg and T. H. Lee, “Superharmonic injection-locked frequency dividers,” IEEE J. Solid-State Circuits, vol. 34, no. 6, pp. 813–821, Jun. 1999. [13] S. Verma, H. R. Rateg, and T. H. Lee, “A unified model for injectionlocked frequency dividers,” IEEE J. Solid-State Circuits, vol. 38, no. 6, pp. 1015–1027, Jun. 2003. [14] J. M. Lopez-Villegas, J. G. Macias, J. A. Osorio, J. Cabanillas, J. J. Sieiro, J. Samitier, and N. Vidal, “Continuous phase shift of sinusoidal signals using injection locked oscillators,” IEEE Microw. Wireless Compon. Lett., vol. 15, no. 5, pp. 312–314, May 2005. [15] F. Ramirez, M. E. de Cos, and A. Suárez, “Nonlinear analysis tools for the optimized design of harmonic-injection dividers,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 6, pp. 1752–1762, Jun. 2003.

José María López-Villegas (M’93–SM’03) was born in Barcelona, Spain, in 1962. He received the Physics degree and Ph.D. degree in physics from the University of Barcelona, Barcelona, Spain, in 1985 and 1990, respectively. From 1985 to 1987, he was Visiting Research Fellow with the Laboratoire d’Electronic Philips (LEP), Paris, France, where he was involved in the electrical characterization of III–V compound semiconductor devices. In 1990, he joined the Electronic Department, University of Barcelona, initially as an Assistance Professor and then as a permanent Professor Titular. He is currently the Head of the RF Group, Electronics Department, University of Barcelona. His research and development activities are focused on the design optimization and test of RF systems and circuits performed using silicon technologies. Within this field, he is particularly interested in the modeling and optimization of integrated inductors and transformers for RFICs applications and in the development of new homodyne transceiver architectures.

Javier Jose Sieiro Cordoba (M’04) was born in Barcelona, Spain, in 1971. He received the Physics degree, M.S. degree in electronic engineering, and Ph.D. degree in physics from the University of Barcelona, Barcelona, Spain, in 1995, 1999, and 2001, respectively. From 2002 to 2003, he was a member of the Electronic Components Technology and Materials (ECTM) Group, Delft University of Technology, Delft, The Netherlands, where he was involved with the modeling of passive components and the design of RFIC circuits. Since 2003, he has been with the Department of Electronics, University of Barcelona, where here is currently an Associate Professor. His research interest are the modeling of passive components and the design of homodyne transceiver architectures.

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Investigation of Drain-Line Loss and the S 22 Kink Effect in Capacitively Coupled Distributed Amplifiers Josef Shohat, Ian D. Robertson, Senior Member, IEEE, and Steve J. Nightingale, Fellow, IEEE

Abstract—This paper investigates the practical limits of the capacitively coupled distributed amplifier (DA) in terms of commonsource output impedance. It is shown that the output impedance of the common-source device is considerably affected by the input coupling circuit. The 22 kink effect is more pronounced in the case of the capacitively coupled circuit. The effect on drain-line loss is very marked, and becomes the practical limitation of the technique. The effect is clearly illustrated by practical measurements on a 45-MHz–20-GHz GaAs monolithic-microwave integrated-circuit amplifier. The kink effect is also shown to be relevant to the output impedance and stability of the common-gate stage in the cascode DA topology. Index Terms—Distributed amplifiers (DAs), distributed parameter circuits, high electron-mobility transistor (HEMT), monolithic-microwave integrated-circuit (MMIC) amplifiers.

I. INTRODUCTION

T

HE ORIGINAL analysis of the field-effect transistor (FET) distributed amplifier (DA) was based on the unilatand neglected losses ( eral transistor model and ). Later, the discrepancies between the analysis and practical performance led researchers to gradually introduce was analyzed because these effects into their analysis. First, it was easy to see its effect on the low-frequency response, while for the other parasitic components of the complete transistor model, the design used computer-aided design (CAD) tools and optimization [1]. Further developments [2]–[4] focused on the effect of losses on the DA gain and bandwidth (BW). These have led to better agreement between the analysis predictions and the measured DA performance. In addition, they have given a deeper understanding of the design tradeoffs in terms of optimum number of stages, gate periphery, and gain-BW performance. Achieving high power with the DA is extremely challenging. High-power DA design is different to reactively matched design because the drain load is fixed to approximately 25 by the 50 to both the constant- ladder (where the device sees left- and right-hand sides). The current swing is the main limitation and, to get more power, the device gate periphery must be increased, but this reduces the DA BW because of the in. Hence, the capacitively coupled DA [5], shown in creased Manuscript received April 7, 2005; revised July 13, 2005. This work was supported by the ERA Foundation Ltd. J. Shohat and I. D. Robertson are with the Institute of Microwaves and Photonics, School of Electronic and Electrical Engineering, The University of Leeds, Leeds LS2 9JT, U.K. (e-mail: [email protected]). S. J. Nightingale is with ERA Technology Ltd, Leatherhead, Surrey KT22 7SA, U.K. Digital Object Identifier 10.1109/TMTT.2005.859873

Fig. 1. Capacitively coupled DAs for higher power handling using: (a) common-source and (b) cascode configuration.

Fig. 1, is a very important technique. By increasing the gate periphery, but adding a coupling capacitor in series with the gate, the BW and gain can be kept constant in the ideal case while the power handling increases. However, there are many tradeoffs in this technique relating to the gate and drain losses and the scaling of device parameters [6]. On the gate line, the increased gatewidth reduces the input resistance, thus, gate-line decreases with inloss is lower. However, on the drain line, creased gatewidth, thus, the drain-line loss increases. This paper further investigates the practical limits of using capacitive couinto the analysis. pling by introducing the effect of Schindler et al. [7] reported degradation of gain in the common-source capacitively coupled DA at high frequency and showed that it was beneficial to capacitively couple the last is only part of the device onto the drain line. However, drain-line loss problem because feedback in the device means that the output impedance cannot simply be represented by in parallel with . Feedback in the FET is the origin of the “kink” effect [8]. In addition, some publications on common-source DA circuits (without capacitive coupling) tried to express the output capacitance of transistors as a parallel capacitance with the Miller effect on [9]. structure of It is shown in this paper that the kink effect is even more pronounced in the case of the capacitively coupled common-source DA. The effect on the drain-line loss then becomes the practical limitation of the technique. The kink effect is also shown

0018-9480/$20.00 © 2005 IEEE

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Fig. 2. Output impedance model in standard common-source configuration, excluding R and C .

to be relevant to the stability of the common-gate stage in the cascode DA topology. There is little published information on common-gate and cascode circuit stability. However, the output of the common-gate circuit produces negative resistance, requiring many cascode DA designs to use series resistive feedback on the gate of the common-gate transistor or an inductance between the common-source and common-gate transistor [10], [11]. Careful design and layout of the bias network and grounding can also improve the stability [12]. It is shown that another feature of the kink effect in the common-source device is that the common-gate output stability is improved in the capacitively coupled cascode DA. II. ANALYSIS

Fig. 3. Illustration of the feedback effect. Solid lines: S 22 of FET with 50-

source connected for 100-, 200-, and 600-m devices (up to 30 GHz), showing kink effect. Dashed line: S 22 of 600-m FET when connected in the DA (plotted up to 5 GHz).

This is a very low value that does not degrade the DA performance. Similarly, the real part of the output impedance can be approximated to

A. Simple Common-Source Stage Analysis

(4)

In all the following analyses, the change of input capacitance due to the Miller effect will be neglected and is assumed to be zero. For the standard DA topology (i.e., no capacitive coupling), the analysis also assumes that the input capacitance of the transistor is absorbed perfectly into the constant- artificial transmission line. Since the transistor gate sees the transmission-line impedance both to the left- and right-hand sides, the impedance seen is a constant- pi-section impedance divided by two. Thus, referring to the simplified model in Fig. 2, the inside load impedance at the gate terminal, including the the transistor, is given by

(1) The transistor output impedance, excluding thus,

and

is,

This resistance is very low, but it is isolated from the drain line by the small series capacitor defined in (3). From this analysis of the standard DA case, we can conis absorbed into the constant- netclude that because work, giving a small resistive impedance at the gate terminal, the kink effect is not a major issue. Fig. 3 shows the typical of a FET with a 50- source connected for 100-, 200-, and 600- m Filtronic devices (up to 30 GHz) showing the kink effect, which is most notable for large periphery devices since and create a large feedback current [8]. Also plotted (in the of the 600- m FET when its gate is condashed line) is the nected into the DA constant- network. This is only plotted up to 5 GHz, which is the cutoff frequency. The two markers ( ) are located at 4 GHz and clearly show that the kink effect disapis absorbed and the gate pears in the standard DA because line presents a low impedance (25 ) to the gate terminal. B. Capacitively Coupled Common-Source Stage Analysis

(2) Over most of the frequency range (where ), this can be approximated as a series capacitor with a value of (3)

Fig. 4 shows a simplified schematic diagram of the capacis the gate periphery scaling factor, itively coupled circuit: e.g., means the device gate width is doubled and the seto maintain the same ries coupling capacitor is set equal to gain and BW. The series resistor is required for biasing and the shunt one provides for zero-pole cancellation, giving a constant voltage division ratio at all frequencies. To find the output impedance of the transistor, not including and , we assume that the gate line presents the 50constant- pi-section artificial line impedance both to the left

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KINK EFFECT IN CAPACITIVELY COUPLED DAs

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Fig. 4. Output impedance modeling of the capacitively coupled circuit with biasing resistors.

and right, again giving approximately 25 over most of the DA operating frequency range, but this is now presented to the complete coupling circuit/device combination. The total load , is now at the gate terminal, including the device’s own given by (5)

Fig. 5. Output impedance modeling of the common-gate device. (a) The transistor input and output terminals. (b) The transistor intrinsic model. (c) The simplified model for studying output impedance and stability.

Since the zero and pole are inside the DA operating band, at high frequency, the output impedance transforms from a series equivalent circuit to a parallel one, which is given by (11)

The output impedance is thus given by

(6)

The parallel resistor is the same as the series one while the new parallel capacitor is (12)

Over most of the frequency range (where ), this can be approximated to (7), shown at the bottom of this page. Solving the numerator zeros gives one very high-frequency zero outside the DA frequency band and one very low-frequency time constant. The denominator gives zero close to the a pole after this low-frequency zero, but inside the DA operating band. At low frequency, the output impedance can thus be approximated as

(8) which gives a series capacitor (9) and a resistor of (10)

The high series capacitance (9) means the series resistance is strongly coupled to the drain line, causing losses in the capacitively coupled DA in comparison with the simple commonsource DA case (3). C. Analysis of Common-Gate Output Impedance in the Capacitively Coupled Cascode DA The common-gate device in the cascode stage is located between the common-source device and the drain artificial transmission line. The common-gate device, shown in Fig. 5, , the current feedback, can easily become unstable since is positive. To obtain the simplified model [see Fig. 5(c)], the input impedance (common-source output impedance) and are combined in . Similarly, and are combined in . However, is ignored since it does not contribute to the negative output impedance when the gate terminal is grounded. The output impedance from the simplified model is (13)

(7)

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TABLE I EXTRACTED PARAMETER VALUES FOR FUJITSU FSU01LG

Since has resistive and capacitive components, a capaciwill interact with to form negative resistance at the tive last term in (13). In the standard DA, the negative part of (13) is

Fig. 6. Feedback contribution to output impedance for the Fujitsu FSU01LG. (a) Standard DA. (b) Capacitively coupled circuit.

(14) While in the capacitively coupled DA case, it is

(15) where denotes impedances in parallel. (Note that, in this equation, the FET scaling factor, , in and cancel each other). is We can infer that the capacitively coupled design has a smaller magnitude and is not capacitive. more stable if It can easily be seen in (15) that the magnitude requirement is satisfied since the factor of appears in the denominator of the first two terms of , and the third term is a parallel resistive component, obtained from (10), which must reduce the magnitude of . For the second condition, note that the factor does and the not affect the phase of the two left-hand-side terms of third term is resistive, thus, adding this in parallel will shift the phase closer to the real axis. Thus, the capacitively coupled case is more stable in the cascode configuration and the high coupling factor increases the output transmission-line loss. Note that would have reduced the magnitude without changing the phase feedback is neglected. when the common-source

Fig. 7. Feedback output impedance of a standard common-source circuit with logarithmic frequency scale showing high impedance.

Fig. 8. Feedback output impedance of a capacitively coupled common-source circuit ( = 1000 ) with logarithmic frequency scale showing high series capacitance inside the DA operating band for large scaling factor ( ).

R

N

III. PRACTICAL EXAMPLE A Fujitsu FSU01LG transistor model was used to verify the above analysis. Table I presents the values of the transistor pi model components. These were extracted from the data sheet -parameters. Using the equations above, the feedback component of the output impedance is first evaluated for the standard DA case. The calculation gives a series capacitor (3) of 0.11 pF with 11series resistor (4). With these values, the real part is negligible is very low, thus, and this capacitance added in parallel with the circuit will have high performance operating up to the DA operates up to 8 GHz). cutoff frequency (a DA having this The capacitive-coupling circuit was chosen to have and . The DA circuit has an operating BW of MHz and 8 GHz, while the two numerator zeros are at GHz. The second zero is well outside the DA operating frequency band and can be ignored. The denominator pole GHz. The series-to-parallel crossover frequency of is at and the output impedance (feedback contribution without ) is MHz. A comparison of the two cases is shown

in Fig. 6. It is clear that the large 5-pF capacitor (9) will strongly couple the 320- resistor (10) to the drain line inside the operating band, degrading performance. In the cascode, this coupled resistor will give a more stable common-gate transistor and will lower the output impedance. IV. MATLAB AND ADS INVESTIGATION and MATLAB was used with the equations (i.e., omitting ) to graphically show how the resistance and capacitance vary with frequency, with bias resistance , and the gatewidth scaling factor as parameters for the Fujitsu transistor model. The simulations are shown in Figs. 7–10. Fig. 7, for the standard DA, shows the low series capacitance, which isolates the series resistance, preventing loss. It agrees with the analysis and Fig. 6(a). From the results in Figs. 8 and 9 for the capacitively coupled DA, it can be deduced that a lower and improve the performance of the DA by reducing a lower the series capacitor, and this agrees with the theory and the approximation. However, even though Fig. 9 shows

SHOHAT et al.: INVESTIGATION OF DRAIN-LINE LOSS AND

KINK EFFECT IN CAPACITIVELY COUPLED DAs

Fig. 9. Feedback output impedance of a capacitively coupled common-source = 2) with logarithmic frequency scale showing high series circuit ( capacitance inside the DA operating for large bias resistance ( ).

N

R

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Fig. 11. Schematic for simulating drain-line loss of the six-section commonsource DA.

Fig. 12. Drain-line factors ( ).

N

S 21 response of a six-section DA for various FET scaling

Fig. 10. Feedback output impedance of a capacitively coupled commonsource circuit ( = 2 and = 1000 ) with logarithmic frequency scale showing the transformation from series impedance to parallel.

N

R

that a lower value reduces the output series capacitance value will add to gate-line and reduces the loss, too low an loss. Fig. 10 shows how the feedback output impedance in the capacitively coupled DA varies with frequency: the change from series to parallel equivalent circuit is clearly shown. It agrees with the analysis on the capacitively coupled DA output impedance and Fig. 6(b). ADS was used to simulate the output impedance for the above circuit example. The schematic diagram in Fig. 11 shows the drain line of a DA with six stages. For phase equalization, is chosen so that the sum of , and , which includes the parallel feedback capacitance (12), is equal to the input capacirepresents (2) or (6) for the standard DA tance. between the and capacitively coupled case, respectively. two ends of Fig. 11, representing the drain-line loss, is plotted in Fig. 12. The DA performance degradation effect and the first are clearly evident as increases. This high transzero at reduces the number of mission-line loss with the increased sections that can be used practically and limits the gain–BW product [2]. To look at the stability, simulations were performed on ADS of the output impedance of standard cascode and capacitively coupled cascode cells used in a DA, as shown in Fig. 13, without

Fig. 13. Model setup used to compare the stability of: (a) the standard cascode = 1000 ). DA and (b) the capacitively coupled DA case ( = 2 and

N

R

the drain artificial transmission-line inductance. In this simulation, the common-gate device was chosen to have the same gate width as the common-source one. The magnitude and phase of the output impedance are compared in Fig. 14 and it is clear that the capacitively coupled DA is more stable. Note that the delay ps (extracted from the maximum frequency of the of available -parameter at 4 GHz) was omitted in the simulation, but adding it later does change the phase response a little for both circuits. V. DESIGN AND MEASUREMENTS The practical design used the 0.5- m pseudomorphic high electron-mobility transistor (pHEMT) technology of Filtronic Compound Semiconductors Limited, Newton Aycliffe, U.K., in common-source topology. The DA was designed using six transistors each of 2 75 m width. The gate line was slightly

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Fig. 14. Comparison of the magnitude and phase of the output impedance of the cascode cell (with common-gate C ) between the standard DA and capacitively coupled DA case (N = 2 and R = 1000 ).

Fig. 17. Measured low-frequency S 21 gain response of the six-section DA showing the drain-line loss above 0.5 GHz due to the feedback effect.

VI. CONCLUSION

Fig. 15.

Chip (3.75 mm

2 1 mm).

kink effect This paper has analyzed the origin of the and has related it to the design of the capacitively coupled high power DA for the first time. It has been shown that the drain-line loss is increased significantly, not simply because of the scaling and , but rather because of the internal feedback effect of effect, which is exaggerated by the capacitive-coupling network at the input. This reduces the DA gain BW product. Based on this, the practical limits and accurate performance prediction for the capacitively coupled DA have been investigated analytically. An unexpected benefit of the feedback is the phase change of the cascode output impedance that improves the circuit stability in the cascode topology and may reduce the need for stabilizing circuits. ACKNOWLEDGMENT

Fig. 16.

Measured S -parameters of the six-section capacitively coupled DA.

tapered for improved [13]. After basic design of the DA as the (using Microwave Office), the design used scaling factor for the capacitively coupled DA. A photograph of the chip is shown in Fig. 15 and the measured performance is shown in Fig. 16. To verify the theory presented here, the low-frequency response is shown in Fig. 17: at these frequencies, the transmission-line imaginary component does not affect the response so that the effect of the output feedback impedance can be seen. From this result, the degradation caused by the zero/pole in the output impedance can be identified with a roll off starting at 500 MHz. The gain at low frequency can be flattened with suitable bias circuit design, but the drain-line loss is still a limiting factor at higher frequencies.

The authors would like to thank Dr. J. Mayock, Filtronic Compound Semiconductor Limited, Newton Aycliffe, U.K., for his help in the design and layout stages of the work. For the chip assembly and measurements, the authors would like to thank Dr. P. Steenson and R. Clarke, both of The University of Leeds, Leeds, U.K., for their help. REFERENCES [1] K. B. Niclas, W. T. Wilser, T. R. Kritzer, and R. R. Pereira, “On theory and performance of solid-state microwave distributed amplifiers,” IEEE Trans. Microw. Theory Tech., vol. MTT-31, no. 6, pp. 447–456, Jun. 1983. [2] J. B. Beyer, S. N. Prasad, R. C. Becker, J. E. Nordman, and G. K. Hohenwarter, “MESFET distributed amplifier design guidelines,” IEEE Trans. Microw. Theory Tech., vol. MTT-32, no. 3, pp. 268–275, Mar. 1984. [3] R. C. Becker and J. B. Beyer, “On gain-bandwidth product for distributed amplifiers,” IEEE Trans. Microw. Theory Tech., vol. MTT-34, no. 6, pp. 736–738, Jun. 1986. [4] M. Ross and R. G. Harrison, “Optimization of distributed monolithic GaAs amplifiers using an analytical/graphical technique,” in IEEE MTT-S Int. Microwave Symp. Dig., Jun. 1988, pp. 379–382. [5] Y. Ayasli, S. W. Miller, R. Mozzi, and L. K. Hanes, “Capacitively coupled traveling-wave power amplifier,” IEEE Trans. Microw. Theory Tech., vol. MTT-32, no. 12, pp. 1704–1709, Dec. 1984.

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KINK EFFECT IN CAPACITIVELY COUPLED DAs

[6] B. Agarwal, A. Schmitz, J. Brown, M. Matloubian, M. Case, M. Le, M. Lui, and M. Rodwell, “112-GHz, 157 GHz and 180-GHz InP HEMT traveling-wave amplifiers,” IEEE Trans. Microw. Theory Tech., vol. 46, no. 12, pp. 2553–2559, Dec. 1998. [7] M. J. Schindler, J. P. Wendler, M. P. Zaitlin, M. E. Miller, and J. R. Dormail, “A K=Ka-band distributed power amplifier with capacitive drain coupling,” IEEE Trans. Microw. Theory Tech., vol. 36, no. 12, pp. 1902–1907, Dec. 1988. [8] S. S. Lu, T. W. Chen, and H. C. Chen, “The origin of the kink phenomenon of transistor scattering parameter S 22,” IEEE Trans. Microw. Theory Tech., vol. 49, no. 2, pp. 333–340, Feb. 2001. [9] C. Paoloni and S. D’Agostino, “An approach to distributed amplifier based on a design-oriented FET model,” IEEE Trans. Microw. Theory Tech., vol. 43, no. 2, pp. 272–277, Feb. 1995. [10] H. Shigematsu, M. Sato, T. Suzuki, T. Takahashi, K. Imanishi, N. Hara, H. Ohnishi, and Y. Watanabe, “A 49-GHz preamplifier with a transimpedance gain of 52 dB using InP HEMT’s,” IEEE J. Solid-State Circuits, vol. 36, no. 9, pp. 1309–1313, Sep. 2001. [11] H. Shigematsu, M. Sato, T. Hirose, and Y. Watanabe, “A 54-GHz distributed amplifier with 6-Vpp output for 40-Gb/s LiNbO modulator driver,” IEEE J. Solid-State Circuits, vol. 37, no. 9, pp. 1100–1105, Sep. 2002. [12] M. M. Oda, “A stable GaAs 6–20 GHz high gain and power TWA,” in IEEE MTT-S Int. Microwave Symp. Dig., Jul. 1991, pp. 437–440. [13] J. Shohat, I. D. Robertson, and S. J. Nightingale, “10 Gb/s driver amplifier using a tapered gate line for improved input matching,” IEEE Trans. Microw. Theory Tech., to be published.

Josef Shohat received the B.Eng. degree from Brunel University, West London, U.K., in 2000, and is currently working toward the Ph.D. degree at The University of Leeds, Leeds, U.K. From 2000 to 2001, he was with 3G.Com, Natanya, Israel, where he was involved with analog and RF circuits for 3G network systems. From 2001 to 2002, he was with Tiaris, Herzliya, Israel, where he was involved with RF circuits for wide-band cable modems. His research interests include circuit analysis, wide-band and high-efficiency amplifier design, and amplifier stability. Mr. Shohat was the recipient of the Institution of Electrical Engineers (IEE) Prize for academic achievement

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Ian D. Robertson (M’96–SM’05) received the B.Sc. (Eng.) and Ph.D. degrees from King’s College London, London, U.K., in 1984 and 1990, respectively. From 1984 to 1986, he was with the Monolithic Microwave Integrated Circuit (MMIC) Research Group, Plessey Research (Caswell) Ltd. Since then, he has held academic posts with King’s College London and the University of Surrey. In June 2004, he became the Centenary Chair in Microwave and Millimeter-Wave Circuits with The University of Leeds. He is currently the Honorary Editor of the IEE Proceedings—Microwaves, Antennas & Propagation. He edited MMIC Design (London, U.K.: IEE, 1995) and coedited RFIC & MMIC Design and Technology (IEE: London, U.K., 2001, 2nd ed.). He has authored or coauthored over 330 papers in the areas of microwave integrated circuit (MIC) and MMIC design. Dr. Robertson has organized numerous colloquia, workshops, and short courses for both the Institution of Electrical Engineers (IEE), U.K., and the IEEE.

Steve J. Nightingale (M’82–SM’96–F’02) received the DipEE degree in 1974, and the Ph.D. degree from the University of Kent, Kent, U.K., in 1980. He was with Philips Research Laboratories, Redhill, U.K., and Hamburg, Germany, and with General Electric, Syracuse, NY. In 1986, he joined Thorn EMI Electronics, Hayes, Middlesex, U.K., initially as a Technology Manager and then as Manager of the Radiation Department. In 1996, he joined ERA Technology Ltd., Leatherhead, Surrey, U.K., to develop a business designing, developing, and manufacturing ultra-wideband amplifiers and components for optical communications applications. He is currently Chief Consultant with Electronic Systems, ERA Technology Ltd. He has been a Visiting Professor with The University of Leeds. He is currently a Royal Academy of Engineering Visiting Professor with the University of Surrey, Surrey, U.K. He has authored or coauthored numerous publications in the microwave field and contributed to four books. He has filed six patents in the areas of GaAs MMIC and hybrid circuit design applied to millimeter-wave radiometers, radars, communication systems, and opto-electronics. He has given over 20 specialized invited lectures to universities and industries in Europe, the U.S., and China. Dr. Nightingale was director of the European Microwave Association (EuMA) and chairman of the Institution of Electrical Engineers (IEE), U.K., Electronics and Communications Division. He was chairman of European Microwave Week (EuMW) and the European Microwave Conference (EuMC), London, U.K., in September 2001.

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Parallel-Coupled Line Filters With Enhanced Stopband Performances Wael M. Fathelbab, Senior Member, IEEE, and Michael B. Steer, Fellow, IEEE

Abstract—A new class of parallel-coupled line filters with broad stopband response is introduced. The design is based on the synthesis of bandpass prototypes with pre-defined upper stopband characteristics. The new filters have uniform- and steppedimpedance resonators, some of which are loaded by open-circuited stubs at their open-circuited ends. A seventh-order filter implementation is presented with a fundamental passband centered at 1 GHz. The measured wide-band transmission characteristic of the filter demonstrated a broad upper stopband and was in agreement with simulations. The performance of the new filter is also compared with the characteristic of a conventionally designed filter to highlight the advantages of the proposed design method. Index Terms—Capacitively loaded resonators, high-pass/bandpass filter prototypes, parallel-coupled line (PCL) filters, spurious stopband rejection, uniform/stepped-impedance resonators.

I. INTRODUCTION

P

ARALLEL-COUPLED line (PCL) filters [1] do not require ground connections and, thus, are easily fabricated in planar form. They have traditionally been used to realize passband bandwidth ratios ranging from as small as 1.02 : 1 to as large as 3 : 1 [2]. Another advantage of PCL filters is that their resonators can be folded into hairpin form, making them suitable for applications with space restrictions [18], [19]. However, when this class of filters is realized on an inhomogeneous media such as microstrip or coplanar waveguide, it suffers from poor upper stopband performance and typically has spurious passbands centered at harmonics of the fundamental passband center frequency, . Considerable efforts to improve the stopband performance of microstrip PCL filters have been directed at suppressing . This was done the harmonic spurious passband located at by compensating for the difference between the even- and odd-mode phase velocities of the inhomogeneous media in a variety of ways. In one approach, the coupled lines were loaded by lumped capacitors [3] or meandered [4]. Other approaches introduced ground-plane apertures underneath the coupled lines [5]–[7] or utilized suspended substrates with dielectric overlay [8], [9]. Other attempts to suppress the undesired spurious passbands introduced geometrical perturbations to the uniform-impedance resonators in the coupling regions [10]–[14], loaded them by split-ring resonators [15], or alternatively utilized stepped-impedance resonators [16]. More recently, a Manuscript received June 9, 2005. This work was supported by the U.S. Army Research Office as a Multi-disciplinary University Research Initiative on Multifunctional Adaptive Radio Radar and Sensors under Grant DAAD19-01-1-0496. The authors are with the Department of Electrical and Computer Engineering, North Carolina State University, Raleigh, NC 27695-7911 USA (e-mail: [email protected]). Digital Object Identifier 10.1109/TMTT.2005.859871

PCL filter with uniform-impedance resonators loaded by opencircuited stubs at all its open-circuited ends demonstrated suppression of the second- and third-ordered harmonic passbands [17]. However, the design approach presented requires generalization. This paper presents a circuit-oriented design approach for PCL filters based on classical network synthesis techniques. The advantages of this approach are primarily design insight and exploitation of all parameters of a transfer function leading to optimum network topologies. The new PCL filters are derived from synthesized bandpass prototypes with controlled upper stopband responses. This is done by systematic application of appropriate circuit transformations on the initial bandpass prototype resulting in a class of PCL filters comprising uniformand stepped-impedance resonators some of which are loaded by open-circuited stubs at their open-circuited ends. This paper begins with an outline in Section II of the traditional design procedure for PCL filters using uniform- and stepped-impedance resonators. The new design method is then presented in Section III. In Section IV, a bandpass filter prototype is synthesized and appropriately transformed to a realizable PCL filter. Finally, in Section V, the measured characteristic of the new filter is contrasted to the measured performance of a PCL filter utilizing optimized stepped-impedance resonators. II. REVIEW OF THE DESIGN OF UNIFORM- AND STEPPED-IMPEDANCE PCL FILTERS The traditional design of PCL filters is based on the -plane being the high-pass prototype depicted in Fig. 1(a) with Richards variable defined as (1) In (1), and are the real and complex frequency variables, respectively, and is the center frequency of the fundamental passband in the -plane. This prototype can be synthesized in the -plane by exact methods [22], [25] and then transformed to the -plane using (1). The rectangular blocks in the -plane prototype of Fig. 1(a) are known as unit elements (U.E.s) [25] and become transmission lines in the -plane, whereas the capacitors represent open-circuited stubs in the -plane. The distributed elements in the -plane are all quarter-wave-length long at . Now each building block in the -plane prototype comprises a pair of series capacitors separated by a U.E., and this can be replaced by a PCL section using the equivalence shown in Fig. 1(b). The critical property of the transition from the -plane to the corresponding -plane is that the -plane response of

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Fig. 2. (a) Type 1 stepped-impedance resonator from [16]. (b) Pair of coupled stepped-impedance resonators.

Fig. 1. (a) S -plane high-pass prototype. (b) Subsection equivalence.

the prototype becomes a periodic bandpass function with spurious passbands centered at odd multiples of . This is an intrinsic property of the prototype of Fig. 1(a). An additional complexity arises when this filter is realized on an inhomogeneous transmission media due to the inequality between the even- and odd-mode phase velocities. This effect can be investigated by matrix (see the Appendix) of each PCL utilizing the full section in a filter for a known even- to odd-mode velocity ratio. It can be shown that the effect of the inhomogeneous media creates additional spurious passbands centered approximately at even [3]–[16]. multiples of To some extent it is possible to alter the positions of the spurious passbands of conventional PCL filters by using stepped-impedance resonators. An excellent account of this technique presenting useful design equations can be found in [16]. Fig. 2(a) shows a Type 1 stepped-impedance resonator [16]. The design approach using Type 1 reswhere onators is briefly explained as follows. The uniform resonators of each PCL section in a conventional filter [see Fig. 1(b)] are staggered by a defined amount. This effect is consequently compensated for by setting the characteristic impedances of the uncoupled line sections of the resonators to a fixed high value . Consequently, the length of each uncoupled line in the filter is then optimized until the return loss of the fundamental passband is restored. A pair of stepped-impedance resonators is shown in Fig. 2(b). Intuitively one could conclude that a change in the stopband performance of a filter will depend on . Now the amount of stagger and the impedance ratio the effect of the inhomogeneous media can be investigated as discussed above. This process could be iterated on until

a suitable stopband performance is reached. In general, this technique offers limited stopband improvement and is best suited to applications where a small shift in the spurious passbands would block harmonics of the input signal from passing through the filter. Thus, the spurious passbands of a filter are not eliminated, but in effect, the stopband characteristic is altered to suit a particular specification [16]. There is also a Type 2 resulting in filters stepped-impedance resonator with with slightly different stopband behavior to those utilizing Type 1 resonators [16]. In the following, we investigate how the design of PCL filters based on -plane bandpass prototypes would lead to better stopband performances. III. DESIGN OF NEW PCL FILTERS The prototype proposed here is shown in Fig. 3(a). It is classified as an -plane bandpass prototype with the Richards variable redefined as (2) In (2), and are as defined previously, but now is the center frequency of the first upper stopband in the -plane. Similar to the -plane high-pass prototype discussed in Section II, this prototype can be synthesized in the -plane by exact methods [23]–[25] and then transformed to the -plane using (2). In the -plane, the U.E.s, capacitors, and inductors of Fig. 3(a) become transmission lines and open- and short-circuited stubs, respectively, and are all a quarter-wave-length long at . It has been shown in [26] that the -plane response of an -plane bandpass prototype is periodic with a fundamental passband centered at and spurious passbands centered at and

(3)

Thus, synthesis based on an -plane bandpass prototype has great flexibility in controlling the location of the first pair of spurious passbands in the -plane. This is possible by selecting a large commensurate frequency to shift the upper spurious

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Fig. 3. (a) S -plane bandpass prototype. (b) Real frequency response for one commensurate frequency . (c) Response after doubling .

f

f

passbands further away from the fundamental passband. Essentially this is the design rationale of the new PCL filters, which closely resembles the design of conventional bandpass filters such as combline [20], [21], [24], [25]. A typical -plane response of the proposed prototype is depicted in Fig. 3(b) where it is seen that the series -plane LC sections (i.e., -plane short- and open-circuited stubs) contribute to multiple transmission zeros at where . By doubling the commensurate frequency (i.e., moving from to ) leads to the response shown in Fig. 3(c) where the spurious responses are shifted leaving behind a broad stopband region. Now we focus our attention on the circuit structure of Fig. 3(a). Through a number of circuit transformations we convert this into a form that can be implemented using distributed

S

Fig. 4. Exact circuit transformations. (a) Basic -plane sub-circuit. (b) Application of relevant Kuroda transformations. (c) Splitting of the 1 : transformer. (d) Distribution of the series capacitor. (e) -plane subsection comprising a PCL loaded by an open-circuited stub at the left open-circuited end.

f

N

elements. Consider the -plane sub-circuit shown in Fig. 4(a) comprising a U.E. separating a pair of series capacitors fol-

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lowed by a series inductor. It is feasible to transform the series LC section across the U.E. by using relevant Kuroda transformations [25], [26]. This leads to the sub-circuit of Fig. 4(b), whose element values are related to those of Fig. 4(a) through the following relations:

(4) Now the series capacitor and the U.E. of characteristic form a single PCL section if a transimpedance transformer. The former is extracted from the existing value of the turns ratio of this transformer is (5)

leaving a

transformer of turns ratio (6)

transformer together This step leads to Fig. 4(c). The with the series capacitor and U.E. is the -plane model of an open-circuited coupled-line section in homogenous media [25]. and just Thus, let us for a while ignore the shunt capacitor investigate the sub-circuit of Fig. 4(d). If a relevant Kuroda , then transformation is applied to spread the series capacitor the following relations:

(7) hold for the circuits of Fig. 4(d). Hence, the modal impedances of the PCL section loaded by an open-circuited stub (of characteristic impedance ) at the left open-circuited end are then evaluated [26] using (7) to give

(8) This concludes the transformation of the subsection of Fig. 4(a) into that of Fig. 4(e). Fig. 5(a) shows an -plane bandpass prototype where the sub-circuits enclosed in the dashed boxes can be systematically transformed as just described. The transformation results in the -plane filter shown in Fig. 5(b). This is only one among many that could be transformed using the new method. In practical terms, it might be desirable not to transform all the inductors across the U.E.s to form PCLs loaded by open-circuited stubs. This is typically due to extreme element values that might lead to realization difficulties on a printed circuit board (PCB).

Fig. 5. (a) S -plane bandpass prototype and (b) its corresponding f -plane filter comprising PCLs loaded by open-circuited stubs at the left open-circuited ends. (c) An S -plane bandpass prototype; and (d) its corresponding f -plane filter comprising PCLs loaded by open-circuited stubs at either the left- or the right-hand-side open-circuited ends. The resulting filters are derived after application of the circuit transformation scheme described in Fig. 4 on the sub-circuits enclosed in the dashed boxes of the S -plane prototypes.

To a great extent these constraints can be resolved dependent on how the initial -plane bandpass prototype is transformed. For this reason, let us apply the transformation steps to the subsections (enclosed in the dashed boxes) of the -plane prototype shown in Fig. 5(c). This prototype is identical to that of Fig. 5(a), except that some of the inductors are now divided into pairs. Upon transformation, the -plane filter illustrated in Fig. 5(d) results. As seen from Fig. 5(d), the filter inevitably has some series short-circuited stubs in its main signal path. We propose approximating these stubs by sections of high-impedance transmission lines as follows. A section of transmission line of short electrical length at and of high characteristic impedance has an input impedance approximately equal to (9) where is the quarter-wave-length frequency of the transmission line. On the other hand, a series short-circuited stub of and has an input characteristic impedance is resonant at impedance at the center of the fundamental passband equal to (10) The above equations are approximately equal for passband bandwidths up to an octave (i.e., 2 : 1 bandwidth ratio) and,

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hence, can be simultaneously solved for the resonant frequency of the transmission line to give (11)

Subsequently the physical length of the transmission line can be evaluated for a given substrate specification. In general, the higher the line impedance that can be realized, the better the approximation. However, over half of the total number of -plane inductors in a prototype is transformed into open-circuited stubs and, thus, the shape of the -plane transfer function of a prototype in the stopband region is maintained after the above approximation. This completes the filter design approach. IV. EXPERIMENTAL PCL FILTER Design of a filter is best illustrated using an example following the common approach in filter papers. The outline specification is to obtain a bandpass filter at 1 GHz with a broad stopband response. A seventh-order approximation function was first developed in the -plane [23]–[25]. In the -plane, the response has a fundamental passband center frequency of of 2.75 GHz, which 1 GHz and a commensurate frequency according to (3), leads to the existence of a first pair of spurious passbands centered at 4.5 and 6.5 GHz. The return loss of the fundamental passband was 16 dB and its bandwidth ratio was 1.22 : 1 (i.e., 20%) leading to lower and upper band-edge frequencies of 0.9 and 1.1 GHz, respectively. Following standard synthesis, a physically symmetrical -plane network results and, thus, only half of it is shown in Fig. 6(a). We shall now examine the inter-resonator couplings after transforming all the inductors in the prototype into open-circuited stubs to yield a PCL filter similar in topology to that of Fig. 5(b). The objective here is twofold: firstly, to numerically demonstrate the circuit transformation process discussed in Section III, but secondly, to see how the element values of the resulting filter are distributed. From the first box [of the prototype of Fig. 6(a)], we have the following parameters:

Fig. 6. Development of a seventh-order PCL filter. (a) Synthesized S -plane prototype in a 50- system. (b) After splitting some of the inner inductors into pairs. (c) Topology of the f -plane filter depicting inter-resonator couplings. (d) Filter’s electrical layout. (The X X is the plane of symmetry.)

0

which, according to (6), leaves a ratio

transformer of turns

(12) (15) corresponding to a subsection such as that of Fig. 4(a). Now, according to the relations of (4), a set of element values are evaluated to give

This takes us to a sub-circuit of similar topology to that of Fig. 4(c). The next step is to evaluate the element values of the sub-circuit of Fig. 4(d) using (7) to give (16)

(13)

Now, according to (8), the modal impedances of the first pair of PCLs loaded by an open-circuited stub of characterat the left—hand-side istic impedance open-circuited end are calculated as

This transforms the subsection to that of Fig. 4(b). A transformer is then extracted from the transformer using (5). The value of the turns ratio of this transformer is

(17)

(14)

This concludes the transformation of the first subsection to a topology similar to that of Fig. 4(e). In a similar fashion, the

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sub-circuits enclosed in the second and third boxes are also transformed leading to the following element values:

(18) and

Fig. 7. PCB layout of the implemented filter.

(19) It must be appreciated that each transformed subsection has a transformer associated with it. Each of these transformers is eliminated by scaling down the modal impedances of the PCL sections together with the characteristic impedances of the stubs in the filter. This step leads to a symmetrical -plane filter compatible with that of Fig. 5(b) having the following element values:

the impedance matrix representing this circuit is, therefore, necessary at this point. This step changed the system impedance of the filter from 50 to 50/2.5 and imposes the placement of an impedance inverter of value 31.62 at each port of the filter. Each inverter is then approximated by a transmission line and the scaled series short-circuited stubs [near the filter ports, see Fig. 6(c)] of characteristic impedance (364.926/2.5) are subsequently transformed into open-circuited stubs using a Kuroda transformation. For a seventh-order filter, there is only a pair of series shortcircuited stubs left in the main line of the filter that must be approximated by high-impedance transmission lines. From Fig. 6(c), the scaled characteristic impedance of one of these short-circuited stubs is (737.934/2.5) . The highest characof a transmission that is realizable on teristic impedance our board is 137 corresponding to a 7-mil-wide (0.177 mm) track. Thus, using (11), the quarter-wave-length frequency of the line is evaluated as follows: GHz

(20) It is desired to construct the filter on an FR4 board with a substrate thickness of 62 mil (1.57 mm), relative dielectric constant of 4.7, and loss tangent of 0.016. Practical constraints imposed by the PCB manufacturer are that any track width and spacing between tracks must not exceed a minimum of 7 mil (0.177 mm). From (20), the coupling coefficients of the PCLs were calculated and the LinCal sub-routine of Agilent’s Advanced Design System (ADS)1 was utilized to obtain a rough estimate of physical dimensions. Unfortunately, the resulting dimensions were unrealizable on this type of board and, also as seen from (20), some of the impedances of the stubs are low. Therefore, transformation of the initial prototype to the proposed network topology of Fig. 5(d) was done. The steps involved are shown in Figs. 6(b) and (c). It can now be seen from the circuit of Fig. 6(c) that the modal impedances are actually high and would still be unrealizable on our PCB. Scaling down 1ADS,

ver. 2003A, Agilent Technol., Palo Alto, CA.

(21)

At this stage, tuning of some of the element values of the filter is required to obtain a flat return loss over the passband. This leads to the electrical layout of the filter, as shown in Fig. 6(d). Examination of the inter-resonator couplings and stub impedances show that they can now be easily realized on our PCB. The electrical layout of filter was then converted into physical dimensions using ADS and the filter constructed. The overall length of the filter is 9449 mil (240 mm), as shown in Fig. 7. V. RESULTS AND DISCUSSION The measured in-band performance of the constructed filter is shown in Fig. 8. The center frequency of the passband is offset by 65 MHz (6.5%), which could be compensated for by shortening the physical lengths of the transmission-line resonators. The midband insertion loss was 5.1 dB and the return loss was greater than 10 dB, as shown in Fig. 8(a). The simulated and measured wide-band scattering parameters of the filter from dc up to 10 GHz are illustrated in Fig. 8(b). This filter was designed in Section IV to theoretically have its first pair of spurious passbands centered at 4.5 and 6.5 GHz should it be realized on a homogenous media. However, the simulated (dark) plot of Fig. 8(b) predicts that the first two spurious passbands are approximately centered at 5.25 and 7.25 GHz instead of 4.5 and 6.5 GHz. This difference is associated with the

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Fig. 9. Measured and simulated wide-band frequency responses of a conventional fifth-order PCL filter utilizing optimized (Type 1) steppedimpedance resonators [16].

VI. CONCLUSION

Fig. 8. Measured and simulated performances of the implemented PCL filter of Fig. 7. (a) In-band frequency response. (b) Wide-band frequency response.

inequality between the even- and odd-mode phase velocities of the microstrip media and also partially due to approximating the inner pair of short-circuited stubs in the -plane filter by high-impedance transmission lines. In summary, the measured transmission performance of the filter followed the simulation and achieved an attenuation level exceeding 39.9 dB (i.e., an effective rejection level of approximately 35 dB) up to 10 GHz. As seen from Fig. 8(b), the stopband of the filter has a tendency to show a spurious passband at the region where the theoretical spurious passbands would have existed. This is around 6 GHz, as predicted by the simulation. At this stage, the performance of the new PCL filter is compared with the characteristic of a conventional filter designed for the same fractional bandwidth, but of fifth order. The order of the filter was lowered to reduce the insertion loss level of its fundamental passband. The conventional filter has optimized stepped-impedance resonators [according to Fig. 2(a)] with the following parameters: mil

mm

(22)

Its measured characteristic is shown in Fig. 9. As expected, a shift in the positions of the spurious passbands is observed, which complies with the argument presented in Section II and [16]. The measured midband insertion loss of the filter was 3.6 dB. Comparison of the stopband performances of the new and conventional filters highlights the improvements gained from the new design methodology.

A circuit-oriented approach for the design of a new class of PCL filters has been presented in this paper. The PCL filters are driven from synthesized -plane bandpass prototypes after appropriate circuit transformations. As a result, the filters have uniform- and stepped-impedance resonators, some of which are loaded by open-circuited stubs at their open-circuited ends. In comparison with the performances of conventional filters utilizing uniform- or stepped-impedance resonators, the new filters have broader stopband characteristics. This has been illustrated experimentally by the implementation of practical filters. It is believed that the new class of PCL filters will find wide usage in advanced RF/microwave front-end transceivers. APPENDIX The matrix of a pair of uniform open-circuited coupled lines in an inhomogeneous media [27] is

(23) and are the modal impedances and where the even- and odd-mode phase lengths.

and

are

REFERENCES [1] S. B. Cohn, “Parallel-coupled transmission-line resonator filters,” IRE Trans. Microw. Theory Tech., vol. MTT-6, no. 4, pp. 223–231, Apr. 1958. [2] B. J. Minnis, “Printed circuit coupled-line filters for bandwidths up to and greater than an octave,” IEEE Trans. Microw. Theory Tech., vol. MTT-29, no. 3, pp. 215–222, Mar. 1981. [3] I. Bahl, “Capacitively compensated high performance parallel coupled microstrip filters,” in IEEE MTT-S Int. Microwave Symp. Dig., Jun. 1989, pp. 679–682. [4] S. M. Wang, C. H. Chi, M. Y. Hsieh, and C. Y. Chang, “Miniaturized spurious passband suppression microstrip filter using meandered parallel coupled lines,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 2, pp. 747–753, Feb. 2005. [5] M. C. Velazquez-Ahumada, J. Martel, and F. Medina, “Parallel coupled microstrip filters with ground-plane aperture for spurious band suppression and enhanced coupling,” IEEE Trans. Microw. Theory Tech., vol. 52, no. 3, pp. 1082–1086, Mar. 2004.

FATHELBAB AND STEER: PCL FILTERS WITH ENHANCED STOPBAND PERFORMANCES

[6] [7]

[8]

[9]

[10]

[11]

[12] [13] [14] [15]

[16] [17] [18] [19] [20] [21] [22] [23]

, “Parallel coupled microstrip filters with floating ground-plane conductor for spurious-band suppression,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 5, pp. 1823–1828, May 2005. L. Zhu, H. Bu, and K. Wu, “Broadband and compact multi-pole microstrip bandpass filters using ground plane aperture technique,” Proc. Inst. Elect. Eng.–Microwave Antennas Propag., vol. 149, pp. 71–77, Feb. 2002. J. T. Kuo, M. Jiang, and H. J. Chang, “Design of parallel-coupled microstrip filters with suppression of spurious resonances using substrate suspension,” IEEE Trans. Microw. Theory Tech., vol. 52, no. 1, pp. 83–89, Jan. 2004. J. T. Kuo and M. Jiang, “Enhanced microstrip filter design with a uniform dielectric overlay for suppressing the second harmonic response,” IEEE Microw. Wireless Compon. Lett., vol. 14, no. 9, pp. 419–421, Sep. 2004. T. Lopetegi, M. A. G. Laso, J. Hernandez, M. Bacaicoa, D. Benito, M. J. Garde, M. Sorolla, and M. Guglielmi, “New microstrip ‘wiggly line’ filters with spurious passband suppression,” IEEE Trans. Microw. Theory Tech., vol. 49, no. 9, pp. 1593–1598, Sep. 2001. T. Lopetegi, M. A. G. Laso, F. Falcone, F. Martin, J. Bonache, J. Garcia, L. Perz-Ceivas, M. Sorolla, and M. Guglielmi, “Microstrip ‘wiggly line’ filters with multispurious rejection,” IEEE Microw. Wireless Compon. Lett., vol. 14, no. 11, pp. 531–533, Nov. 2004. B. S. Kim, J. W. Lee, and M. S. Song, “An implementation of harmonic-suppression microstrip filters with periodic grooves,” IEEE Microw. Wireless Compon. Lett., vol. 14, no. 9, pp. 413–415, Sep. 2004. J. T. Kuo, W. H. Hsu, and W. T. Huang, “Parallel coupled microstrip filters with suppression of harmonic response,” IEEE Microw. Wireless Compon. Lett., vol. 12, no. 10, pp. 383–385, Oct. 2002. , “Tapped wiggly-coupled technique applied to microstrip bandpass filters for multi-octave spurious suppression,” Electron. Lett., vol. 40, pp. 46–47, Jan. 2004. J. Garcia-Garcia, F. Martin, F. Falcone, J. Bonache, I. Gil, T. Lopetegi, M. Laso, M. Sorolla, and R. Marques, “Spurious passband suppression in microstrip coupled line band pass filters by means of split ring resonators,” IEEE Microw. Wireless Compon. Lett., vol. 14, no. 9, pp. 416–418, Sep. 2004. M. Makimoto and S. Yamashita, “Bandpass filters using parallel coupled striplines stepped impedance resonators,” IEEE Trans. Microw. Theory Tech., vol. MTT-28, no. 12, pp. 1413–1417, Dec. 1980. P. Cheong, S. W. Fok, and K. W. Tam, “Miniaturized parallel coupledline bandpass filter with spurious-response suppression,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 5, pp. 1810–1816, May 2005. E. G. Cristal and S. Frankel, “Hairpin-line and hybrid hairpin-line/halfwave parallel-coupled-line filters,” IEEE Trans. Microw. Theory Tech., vol. MTT-20, no. 11, pp. 719–728, Nov. 1972. U. H. Gysel, “New theory and design for hairpin-line filters,” IEEE Trans. Microw. Theory Tech., vol. MTT-22, no. 5, pp. 523–531, May 1974. I. Hunter, Theory and Design of Microwave Filters. London, U.K.: IEE Press, 2001. G. Matthaei, L. Young, and E. M. T. Jones, Microwave Filters, Impedance-Matching Networks, and Coupling Structures. Norwood, MA: Artech House, 1980. M. Horton and R. Wenzel, “General theory and design of optimum quarter-wave TEM filters,” IEEE Trans. Microw. Theory Tech., vol. MTT-13, no. 5, pp. 316–327, May 1965. H. J. Orchard and G. C. Temes, “Filter design using transformed variable,” IEEE Trans. Circuit Theory, vol. CT-15, no. 12, pp. 385–408, Dec. 1968.

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[24] R. J. Wenzel, “Synthesis of combline and capacitively loaded interdigital bandpass filters of arbitrary bandwidth,” IEEE Trans. Microw. Theory Tech., vol. MTT-19, no. 8, pp. 678–686, Aug. 1971. [25] J. A. G. Malherbe, Microwave Transmission Line Filters. Norwood, MA: Artech House, 1979. [26] B. J. Minnis, Designing Microwave Circuits by Exact Synthesis. Norwood, MA: Artech House, 1996. [27] G. I. Zysman and A. K. Johnson, “Coupled transmission line networks in an inhomogeneous dielectric medium,” IEEE Trans. Microw. Theory Tech., vol. MTT-17, no. 10, pp. 753–759, Oct. 1969.

Wael M. Fathelbab (M’03–SM’05) received the Bachelor of Engineering (B.Eng.) and Doctor of Philosophy (Ph.D.) degrees from the University of Bradford, Bradford, U.K., in 1995, and 1999 respectively. From 1999 to 2001, he was an RF Engineer with Filtronic Comtek (U.K.) Ltd., where he was involved in the design and development of filters and multiplexers for various cellular base-station applications. He was subsequently involved with the design of novel RF front-end transceivers for the U.K. market with the Mobile Handset Division, NEC Technologies (U.K.) Ltd. He is currently a Research Associate with the Department of Electrical and Computer Engineering, North Carolina State University, Raleigh. His research interests include network filter theory, synthesis of passive and tunable microwave devices, and the design of broad-band matching circuits.

Michael B. Steer (S’76–M’82–SM’90–F’99) received the B.E. and Ph.D. degrees in electrical engineering from the University of Queensland, Brisbane, Australia, in 1976 and 1983, respectively. He is currently a Professor with the Department of Electrical and Computer Engineering, North Carolina State University, Raleigh. In 1999 and 2000, he was a Professor with the School of Electronic and Electrical Engineering, The University of Leeds, where he held the Chair in microwave and millimeter-wave electronics. He was also Director of the Institute of Microwaves and Photonics, The University of Leeds. He has authored approximately 300 publications on topics related to RF, microwave and millimeter-wave systems, high-speed digital design, and RF and microwave design methodology and circuit simulation. He coauthored Foundations of Interconnect and Microstrip Design (New York: Wiley, 2000). Prof. Steer is active in the IEEE Microwave Theory and Techniques Society (IEEE MTT-S). In 1997, he was secretary of the IEEE MTT-S. From 1998 to 2000, he was an elected member of its Administrative Committee. He is the Editor-in-Chief of the IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES (2003–2006). He was a 1987 Presidential Young Investigator (USA). In 1994 and 1996, he was the recipient of the Bronze Medallion presented by the Army Research Office for “Outstanding Scientific Accomplishment.” He was also the recipient of the 2003 Alcoa Foundation Distinguished Research Award presented by North Carolina State University.

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RF Frequency Shifting via Optically Switched Dual-Channel PZT Fiber Stretchers Christopher S. McDermitt and Frank Bucholtz, Member, IEEE

Abstract—We demonstrate a new technique for RF serrodyne frequency shifting with ultrahigh sideband suppression. This is accomplished using two parallel optically switched piezo fiber stretchers driven with two out-of-phase serrodyne modulations. We achieved sideband suppression of nearly 50 dB, an improvement of almost 30 dB compared to the standard fiber stretching approach. The frequency shifter is easily integrated into photonic and RF systems, is functional in certain broad-band applications, and allows for tunable translations. Index Terms—Microwave frequency translation, optical signal processing, piezo (PZT) fiber stretchers, serrodyne modulation. Fig. 1.

Important characteristics of serrodyne modulation.

I. INTRODUCTION

R

F PHASE shifters play an important role in modern electronic systems. The fundamental goal of most frequency shifters is to maximize the suppression of sidebands. As shown in Fig. 1, serrodyning—applying a sawtooth phase modulation—is a conventionally used method for frequency shifting. Optical frequency shifters with high sideband suppression include serrodyne modulation of waveguides [1] and switched channel moving reflectors [2]. However, RF frequency shifts require 10 larger ramps in delay time and are produced by physically changing the fiber length—and, therefore, the time delay—with a sawtooth driven piezoelectric actuator [3]. Unfortunately, a perfect linear sawtooth with infinitely fast flyback is impossible to achieve with existing techniques due to mechanical and electrical restraints inherent in the fiber and piezoelectric ceramics. The finite flyback time of the sawtooth ramp is the dominant factor in degrading serrodyne performance and limits this approach to sideband suppressions of just over 20 dB. The piezo (PZT) stretcher, being a mechanical device, has an upper frequency limit of a few hundred hertz at best. However, an optimal linear frequency shift of only 10–100 Hz is useful and important in applications such as phased-array radars. In this paper, we demonstrate a new technique for RF frequency shifting with ultrahigh sideband suppression. This is accomplished by incorporating optically switched dual-channel PZT fiber stretchers that are simultaneously driven with two out-of-phase serrodyne modulations. This is similar to the moving reflectors system for optical frequency shifting [2], except that the PZT fiber stretchers are capable of producing desirably large time delays without the issue of power fluctuations. Accordingly, RF frequency shifts, as opposed to the optical shifts in [4]–[6], are obtainable. Manuscript received May 19, 2005; revised July 5, 2005. C. S. McDermitt is with Sachs Freeman Associates Inc., Largo, MD 20774 USA (e-mail: [email protected]). F. Bucholtz is with the Optical Sciences Division, Code 5652, Naval Research Laboratory, Washington, DC 20375 USA. Digital Object Identifier 10.1109/TMTT.2005.859870

Fig. 2. Block diagram for: (a) single channel and (b) switched dual-channel serrodyne modulation of optical fiber showing distributed feedback laser (DFB), Mach–Zehnder modulator (MZM), high-speed switch, piezo (PZT), photodiode (PD), and electrical spectrum analyzer (ESA).

II. EXPERIMENTALPROCEDURE The standard fiber-stretching technique for serrodyne modulation is depicted in Fig. 2(a). As previously mentioned, typical results produced from this type of system are limited to approximately 20–30 dB of sideband suppression. This is due to the flyback restraints inherent in the fiber and piezo ceramic. Fig. 2(b) shows an architecture that effectively eliminates the issue of flyback. The system consists of a distributed feedback laser followed by a Mach–Zehnder modulator (MZM) biased at quadrature and a high-speed optical switch. Each output of the switch contains a serrodyne modulated piezoelectric fiber stretcher. The stretchers comprised 120 m of 80- m fiber wound in multiple turnaround PZT actuators (Polytec PI, P-845.60). The two channels are then recombined with an optical coupler and the output is detected by a photodiode. It is critical to match

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is the optical power level, is the optical (angular) where frequency, and is a factor relating power to electric-field amplitude. The optical power is given by (2) RF and dc voltages applied to the MZM produces net phase modulation (3)

Fig. 3. Switch theory for avoiding flyback with the black solid sawtooth corresponding to the phase modulation seen at the photodiode.

not only the optical path lengths of each channel, but also to match the RF power levels in order to prevent undesirable amplitude modulations. As opposed to a dual-output MZM, an optical switch was used as the switching mechanism due to extinction benefits. Synchronized switching could also be used to provide maximum extinction between channels by replacing the 2 1 coupler with another switch. The advantage of this system is achieved by switching back and forth between two out-of-phase serrodyne modulations to avoid the finite electromechanical flyback of each individual channel. Accordingly, the linear ramp occurs in the electrical domain and the nearly zero flyback time (300 ns) occurs in the optical domain via optical switching. This is illustrated in Fig. 3 with the black solid sawtooth corresponding to the actual phase modulation seen at the photodiode. There are several key parameters displayed in Fig. 3. Both in phase within sawtooth modulations must ramp from 0 to their respective transistor transistor logic (TTL) “on” states. This effectively requires a phase ramp greater than 0 to since only a portion of each ramp is utilized. For the actuators used here, this required a voltage ramp of approximately 42 V pk–pk. The ramps can have either negative or positive slope depending on the sign of the desired frequency shift. The dc offset voltages must also be adjusted so that the zero degree phase shift of one ramp is aligned with the corresponding zero degree shift of the other ramp. Furthermore, “flyback time” is not necessarily the appropriate term for the switched dual-channel technique as the transfer from one phase ramp to the other is an optical switching time. This “transfer time” introduces a minor problem of its own because during the 300-ns transition there is no power in either channel. The photodetector sees a minor phase spike when the signal vanishes temporarily.

is the (static) bias phase shift, is the where is the amplitude of the applied phase modulation amplitude, is the peak voltage required to produce RF voltage, and peak phase modulation. . Hence, For a balanced MZM biased at quadrature, , and the electric field at one of the MZM outputs is

(4) After passage through the PZT stretcher, the phase of each component of the electric field incident on the photodetector is shifted by the periodic serrodyne phase modulation . Therefore, the field incident on the photodetector is

(5) Assume the exponential containing the serrodyne modulation can be expanded in a Fourier series (6) where

and

(7)

III. SYSTEM ANALYSIS To analyze the system, we first calculate the response for a single PZT stretcher with no switching. In Fig. 2(a), the electric field at the output of the laser can be written (1)

Here, is the period of the serrodyne modulation. The rms RF power in the th sideband of the RF fundamental frequency is given by (8)

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time delay (transfer time) between the rising edge of the switch function and the onset of the linear ramp. Hence,

and (12) Fig. 4. Scheme for switched dual-channel linear phase ramps of amplitude 2q and finite optical transfer time (F ).

where is the responsivity (amperes per watt) of the photodeis the Bessel function of the first kind of order . tector and In ideal serrodyne modulation, all the power is transferred from the fundamental to one of the sidebands, effectively producing a frequency shift. For a perfectly linear serrodyne ramp, power in the unwanted sidebands is exactly zero when: 1) the total phase excursion of and the ramp is an exact integer multiple of , 2) the flyback time is zero. In this case, (7) reduces to

(9)

As before, we perform a Fourier expansion of the function and obtain a rather complicated expression for the (see the Appendix) However, for expansion coefficients , we obtain for the dual-switched configuration,

where, now,

(13)

and is the RF for which proaches zero,

where

. In the limit as

ap-

(14) (10) IV. RESULTS

However, for a practical system with finite flyback time, it is straightforward to show that

(11) Hence, the serrodyne performance is degraded by the ratio of the flyback time to the modulation period . For example, ratio sideband suppression of 20, 40, and 60 dB requires an of 0.1, 0.01, and 0.001, respectively. At 10 Hz, the 300-ns optical switching speed allows for theoretical sideband suppression of 110 dB. The newly developed system practically eliminates the issue of flyback. The theory of serrodyne modulation is further supported in [7]–[9]. To calculate the RF response for the dual-switched scheme in Fig. 2(b), we use the following approach (see Fig. 4). Consider and define two serrodyne the time interval and as shown. Here, denotes a modulation functions

Here, we compare the results of the conventional fiberstretching frequency translator to the newly developed switched dual-channel technique. In addition, we briefly consider the possibility of using the new system for broad-band and frequency-hopping applications. The results for a single-channel 20-Hz frequency shifter are shown in Fig. 5(a). The serrodyne frequency was 10 Hz and pk–pk phase ramp was used with the foresight that it would be needed for the switched dual-channel method. The sideband suppressions obtained were approximately 20 dB at 18 GHz. This agrees with earlier results [3]. Utilizing the switched dual-channel technique produced sideband suppression of nearly 50 dB, as seen in Fig. 5(b). It is important to note that, here, the frequency shift is twice the “usual” serrodyne shift due to the fact that each phase ramp must go in a shorter time frame. The geometry in Fig. 3 from 0 to phase ramp must be used in order for each shows that a 0 to phase ramp within the appropriate channel to complete a 0 to TTL cycle.

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Fig. 5. Frequency shifting results for: (a) single-channel 10-Hz serrodyne modulation with 4 pk–pk phase ramp and (b) switched dual-channel 20-Hz serrodyne modulation with 2 pk–pk phase ramp.

Fig. 6. Results displaying: (a) simulation of sideband behavior for various frequency shifts that occur at their respective RF input frequencies where sideband suppression equals 10 log[(S ) 2=(S ) 2] and (b) compilation of S compared to S versus RF input frequency with the system adjusted and left at optimal performance for 18 GHz.

With the serrodyne parameters optimized for maximum performance at 18 GHz, the broad-band behavior of sideband suppression was analyzed for the switched dual-channel technique. Ceteris paribus, the RF input frequency, was varied from 6 to 40 GHz. The results are presented in Fig. 6(a) and (b) where the curves are a compilation of the sideband with maximum power for a given input frequency compared to the power left at the carrier frequency (i.e., the 9-, 18-, 27-, and 36-GHz curves correspond to frequency shifts of 10, 20, 30, and 40 Hz, respectively). The Appendix and Fig. 7 provide a detailed discussion of the power in each sideband as a function of RF input frequency. The behavior of the sideband suppression is simply due to the fact that the amount of phase shift for a given physical stretch varies linearly with frequency. Accordingly, the system achieves a peak performance at every half-integer multiple of since those frequencies the pre-optimized carrier frequency correspond to integer multiples of peak-to-peak phase ramps. Fig. 7 clearly depicts the concept that sidebands experience peak power exclusively once at a corresponding half-integer multiple of 18 GHz, while all other sidebands are suppressed at that frequency. Optimizing the system at a lower carrier frequency would decrease the spacing of the peak performance intervals

and create a broad-band system that is functional at a wide range RF input frequencies. Consequently, the input signal may contain a band of carefully chosen frequencies that will each be shifted by a different multiple of the serrodyne modulation frequency. Broad-band frequency shifting techniques are discussed in [10] and [11]. Larger frequency shifts are obtainable by simultaneously increasing the serrodyne frequency and TTL switching frequency. Effectively, this changes the slope of the phase ramp and each within the appropriate TTL channel still rises from zero to state. This fact makes it theoretically possible to rapidly change frequency shifts. Due to the mechanical sensitivity of the current system, minor tuning of the sawtooth voltages was required to obtain the data shown in Fig. 8. Displayed are 20- and 40-Hz frequency shifts that were produced by 10- and 20-Hz switched dual-channel serrodyne modulations. For single channel fiber stretching, the flyback becomes a larger percentage of the period as the serrodyne frequency increases. A unique feature of the switched dual channel technique is that the flyback time is an optical switching time that is not affected as the serrodyne frequency is increased. With regard to larger frequency shifts, the current system is limited by the drive frequency of the voltage

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V. CONCLUSION We have demonstrated a highly efficient RF frequencyshifting technique that provides 50 dB of sideband suppression. This represents an improvement of almost 30 dB compared to previous results with single channel fiber stretching. The calculated sideband suppression for the flyback to period ratio was 110 dB, demonstrating that the flyback discontinuity is no longer the limiting factor of the system. However, the transfer time introduced by the optical switch now plays a role since the photodetector temporarily sees no power during this event. We believe further improvements are possible by reducing transfer time and by eliminating the small amount of TTL jitter that was noticed. Using variable attenuators in each channel to provide power level matching and reduce the effects from channel-to-channel amplitude modulations, a cleaner RF source, and mechanical designs to reduce sensitivity are additional possibilities for improved results. For the demonstrated system, the sideband suppression varies with RF frequency. This is advantageous in broad-band applications since the sideband suppression is maximized at ever half-integer multiple of the carrier frequency. Furthermore, even at nonoptimal carrier frequencies, reasonable sideband suppressions are obtainable. This is particularly true at higher frequencies where the worst case sideband suppression is 20 dB. As evident in Fig. 6(b), these troughs continue to increase with frequency. Accordingly, it is possible to use this technique to design an RF frequency shifter that is operational over a wide range of input frequencies. Larger frequency shifts are obtainable by increasing the serrodyne frequency or by implementing additional channels. Fig. 7. System initially optimized for 18 GHz. (a) Theoretical sideband power as a function of the RF input frequency with F =T = 0:003 where power equals 10 log[(S ) 2]. (b) Measured sideband power as a function of the RF input frequency.

APPENDIX The following discussion is intended to provide a better understanding of the sideband suppression as a function of RF carrier frequency (Figs. 5–7). The dependence of on is calculated as follows. Given an applied voltage , the change in of the fiber in the stretcher is proportional to , i.e., length kV, where the constant depends on the PZT material used in the stretcher, as well as the mechanical details of the stretcher assembly, length of fiber, etc. At RF frequency , this length change corresponds to a phase shift

(15)

Fig. 8. Frequency hopping: various frequency shifts using switched dualchannel technique. The shifts were accomplished by simultaneously changing the serrodyne and TTL switching frequencies.

amplifiers. Additional channels could also be added in parallel to provide larger shifts.

where is the speed of light, is the core refractive index, is the group velocity in the fiber, and . Suppose now that the sawtooth peak-to-peak voltage and flyback time are fixed such that for one particular RF frequency , . As the RF frequency varies, the value of then (for fixed ) is given by

(16)

McDERMITT AND BUCHOLTZ: RF FREQUENCY SHIFTING VIA OPTICALLY SWITCHED DUAL-CHANNEL PZT FIBER STRETCHERS

Performing a Fourier expansion of the function , we obtain for the imaginary and real parts of the Fourier coefficients and

where period of sawtooth; dead time (hence, on flyback);

fraction of period spent

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[4] E. Voges, O. Ostwald, B. Schiek, and A. Neyer, “Optical phase and amplitude measurements by single sideband homodyne detection,” IEEE J. Quantum Electron., vol. QE-18, no. 1, pp. 124–129, Jan. 1982. [5] S. Ozharar, F. J. Quinlan, S. Gee, and P. J. Delfyett, “A novel TDM method for serrodyne modulation with high sideband suppression for arbitrary waveform generation,” presented at the Lasers and Electro-Optics Conf., Baltimore, MD, 2005. [6] R. A. Soref, “Voltage-controlled optical/RF phase shifter,” IEEE J. Lightw. Technol., vol. LT-3, no. 5, Oct. 1985. [7] R. C. Cumming, “The serrodyne frequency translator,” Proc. IRE, vol. 45, pp. 175–186, Feb. 1957. , “Frequency translation by modulation of transit-time devices,” [8] Stanford Univ., Stanford, CA, vol. 39, Aug. 1955. [9] G. Klein and L. Dubrowsky, “The DIGILATOR, a new broad-band microwave frequency translator,” IEEE Trans. Microw. Theory Tech., vol. MTT-15, no. 3, pp. 172–179, Mar. 1967. [10] S. T. Winnall, A. C. Lindsay, and G. A. Knight, “A wide-band microwave photonic phase and frequency shifter,” IEEE Trans. Microw. Theory Tech., vol. 45, no. 6, pp. 1003–1006, Jun. 1997. [11] K. J. Williams, “Electro-optical broad-band microwave frequency shifter,” U.S. Patent 6 043 926, Mar. 28, 2000.

; ; ; ; . Assuming an ratio of 0.003, the theoretical power in each individual sideband is shown in Fig. 7(a), while the measured power is presented in Fig. 7(b). ACKNOWLEDGMENT The authors wish to thank K. J. Williams and J. L. Dexter, both of the Optical Science Division, Naval Research Laboratory (NRL), Washington, DC, for their support and initial work on fiber-optic frequency shifters. Additional thanks go to M. S. Rogge and V. J. Urick, both of the NRL, for valuable insight. REFERENCES [1] L. M. Johnson and C. H. Cox, “Serrodyne optical frequency translation with high sideband suppression,” IEEE J. Lightw. Technol., vol. 6, no. 1, pp. 109–112, Jan. 1988. [2] R. C. Gutierrez, “Doppler phase shifting using dual, switched phase shifting devices,” U.S. Patent 6 426 828, Jul. 30, 2002. [3] J. L. Dexter and C. J. Murphy, “Comparison of fiber-optic and a p-i-n diode RF phase shifter,” Naval Res. Lab., Washington, DC, NRL/FR/5715-95-9759, 1995.

Christopher S. McDermitt was born in Harrisburg, PA, on November 29, 1977. He received the B.S. degree in physics from Bloomsburg University, Bloomsburg, PA, in 2000, and is currently working toward the M.S. degree at George Mason University, Fairfax, VA. He is currently with Sachs Freeman Associates Inc., Largo, MD, as a Research Physicist with the Optical Sciences Division, Naval Research Laboratory, Washington, DC. His research interests include photonics signal processing for phased array radars, fiber-optic interferometric systems, and enhancement of microwave photonic links.

Frank Bucholtz (M’81) was born in Detroit, MI, on April 4, 1953. He received the B.S. degree in physics and mathematics from Wayne State University, Detroit, MI, in 1975, and the M.S. and Ph.D. degrees in physics from Brown University, Providence, RI, in 1977 and 1981, respectively. From 1981 to 1983, he was a National Research Council (NRC) Post-Doctoral Research Associate with the Naval Research Laboratory, where he conducted research in the area of ferrimagnetic devices for microwave signal processing. He is currently a member of the Optical Sciences Division, Naval Research Laboratory, Washington, DC. His research interests include fiber-optic sensors, hyperspectral imaging, and analog microwave photonics.

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Design of Microstrip Bandpass Filters With Multiorder Spurious-Mode Suppression Chi-Feng Chen, Ting-Yi Huang, and Ruey-Beei Wu, Senior Member, IEEE

Abstract—This paper proposes a bandpass filter design method for suppressing spurious responses in the stopband by choosing the constitutive resonators with the same fundamental frequency, but staggered higher order resonant frequencies. The design concept is demonstrated by a four-pole parallel-coupled Chebyshev bandpass filter and a compact four-pole cross-coupled elliptic-type bandpass filter. Each filter is composed of four different stepped-impedance resonators (SIRs) for which a general design guideline has been provided in order to have the same fundamental frequency and different spurious frequencies by proper adjusting the impedance and length ratios of the SIR. Being based on knowledge of the coupling coefficients and following the traditional design procedure, the resultant filter structures are simple and easy to synthesize. The measured results are in good agreement with the simulated predictions, showing that better than 30-dB rejection levels in the stopband up to 5 4 0 and 8 2 0 are achieved by the Chebyshev and quasi-elliptic filters, respectively. Index Terms—Coupling coefficient, elliptic-type bandpass filter, microstrip filter, stepped-impedance resonator (SIR).

I. INTRODUCTION

I

N MICROWAVE communication systems, the bandpass filter is an essential component, which is usually used in both receivers and transmitters. Thus, the quality of bandpass filters is extremely important. Planar filters are currently a popular structure because they can be fabricated using printed circuit technology and are suitable for commercial applications due to their small size and lower fabrication cost. In the past, the parallel-coupled microstrip bandpass filter was one of the most popular filters in communication systems due to its advantages of ease in manufacture, ease of synthesis method, low cost, and high practicality [1]. With the advent of widespread applications of mobile communication systems, the development of bandpass filters has emphasized compact size and high performance. Thus, the elliptic or quasi-elliptic function bandpass filters realized by cross-coupling were introduced in order to reduce the circuit size and improve the selectivity of bandpass filters [2]–[5]. The cross-coupled bandpass filters have attracted much attention because they exhibit a single pair of transmission zeros in the stopband to reject possible interference, which can be caused by the multipath effect.

Manuscript received March 13, 2005; revised July 6, 2005. This work was supported in part by the National Science Council under Grant NSC 93-2752-E002-003-PAE. The authors are with the Department of Electrical Engineering and Graduate Institute of Communication Engineering, National Taiwan University, Taipei, Taiwan, R.O.C. Digital Object Identifier 10.1109/TMTT.2005.859869

In order to enhance the circuit performance, bandpass filters must also have wide stopband responses. For traditional microstrip half-wavelength resonator bandpass filters, due to their unequal even- and odd-mode phase velocity, the first spurious passband is centered at twice the midband frequency, the second spurious passband at triple the midband frequency, and so on. Poor out-of-band response may happen at each higher order resonant frequency of the resonators. Several methods have been proposed to resolve this problem. One popular solution is to impose additional band reject filters to reject the spurious frequencies with a larger circuit size and insertion loss as tradeoff. In addition, the quarter-wavelength resonator filters can also be used to reject the twice midband frequency, but its main drawback is the requirement in vias for microstrip realization [6], [7]. Another way is to eliminate the difference in propagation speeds of the even and odd modes in coupled lines, e.g., using corrugated coupled microstrip lines [8] and introducing apertures in the ground plane along each coupled section [9]. The suppression of spurious frequencies can also be achieved by introducing over-coupling to the end stages and increasing the image impedance of the filter [10]. The wiggly-line filter [11] can also reject the harmonic passband of the filter by using a continuous perturbation of the width of the coupled lines following a sinusoidal law. Nevertheless, these structures mentioned above only can improve the rejection char. In addition, the stepped-impedance acteristics of the filter at resonators (SIRs) [12], split-ring resonators [13], and wigglyline resonators [14] are also effective for suppressing the spurious passbands. Although many studies have been done on suppressing the harmonic response, as mentioned above, most of these techniques need a very complicated procedure for filter design. In this paper, we propose a simple design method, which can avoid the presence of spurious modes without resorting to extra filter or stub elements. The concept is demonstrated by two filter designs. One is the microstrip parallel-coupled bandpass filter and the other is the cross-coupled bandpass filter, where each filter is composed of SIRs. The major design idea is to irregularly distribute the higher order spurious frequencies of these SIRs over the stopband without overlap, thus excellent stopband rejection levels can be obtained, which would be better than that using the similar concept in [15]. The theory and guidelines for selecting the geometric parameters of SIRs are proposed in Section II. The design procedures for the two filters are given in Sections III and IV. The simulation results are compared with the measured data. Finally, brief conclusions are drawn in Section V.

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CHEN et al.: DESIGN OF MICROSTRIP BANDPASS FILTERS WITH MULTIORDER SPURIOUS-MODE SUPPRESSION

Fig. 1.

Structure of the SIR. (a) K = Z

=Z < 1. (b) K = Z =Z > 1. Fig. 2. Resonant electric length versus stepped percentage impedance ratio K = 0:25; 0:5; 1; 2; 4 as a parameter.

II. THEORY Most filters consist of several resonators, which have the same resonant frequency at the desired band. However, it is recognized that the higher order resonant modes of these resonators might result in spurious response in the stopband. To suppress the spurious response, the constitutive resonators are chosen to have the same fundamental frequency, but staggered higher order resonant frequencies. Since all higher order modes of the resonators have different resonant frequencies, they cannot build up to cause significant spurious response. A simple kind of resonator that can have some fundamental frequency, but different higher order resonant frequencies is the SIR. Fig. 1 shows a typical structure of a half-wavelength SIR. The SIR was originally presented not only to control the spurious responses, but also to reduce the resonator size [16], [17]. The resonance conditions can be described by the equation or where

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(1)

is the impedance ratio of the SIR defined as (2)

Herein, the fundamental resonance and the spurious resonances can be adjusted by choosing a suitable impedance ratio for the SIR. As a first-order approximation, assume that the propagation constants of the transmission-line sections are only weakly dependent on the width so that the electric angles in (1) are proportional to the corresponding physical lengths. Define the stepped percentage as the portion of the transmission line 2 to the whole transmission line, i.e.,

with

There are various solutions for , which are dependent on the choice of and . Fig. 2 shows the relation between the resoand nant electric length versus with as a parameter. It can be found that in the nonstepped case of or, equivalently, or . This means that the th resonant mode occurs when the total length equals times the half waveto the length. The ratio of the first spurious frequency is 2. In cases where , this fundamental frequency ratio becomes higher than 2, and it is smaller than 2 if . The deviation from 2 is more significant for larger . In order to obtain the dimensions of each resonator, the design procedures are as follows. • To decide the locations of the higher order spurious frequencies of each SIR. Herein, the higher order spurious frequencies must be located irregularly over the stopband in order to obtain an overall satisfactory stopband performance. • To decide the and values. The SIRs can be designed with fixed fundamental frequency, but various higher order spurious frequencies by choosing diffrom Fig. 2 and, thereby, the ferent combinations of and dimensions of each SIR can be determined. In an actual design, the two parameters should be slightly tuned to consider the full-wave properties such as the equivalent capacitances of the step discontinuities, the dependence of the propagation constants on the width of the transmission lines, and so on. Also, and , at for compactness, it is usually chosen which minimizes the total length.

(3) III. FOUR-POLE PARALLEL-COUPLED FILTER DESIGN

is the total electric angle of the SIR. where Substituting (3) into (1) yields

or (4)

To demonstrate the proposed concept, without loss of generality, consider the design of a parallel-coupled bandpass filter. The Chebyshev filter is composed of four different SIRs, as illustrated in Fig. 3. By properly adjusting the impedance and length ratios of the SIRs, the values of the fundamental frequency and spurious frequencies can then be tuned simultane-

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Fig. 3.

Layout of the four-pole parallel-coupled bandpass filter.

TABLE I DESIGN PARAMETERS OF THE SIR ON A SUBSTRATE OF DIELECTRIC CONSTANT 3.38 AND THICKNESS 0.508 mm

Fig. 4. Fabricated filter.

ously. In our design process, the four SIRs were designed with the parameters listed in Table I. As can be seen, each SIR has been chosen the same fundamental frequency and different spurious frequencies and, thereby, the spurious passbands of the filter can be rejected with the combination of different SIRs structure. In the above design, the major design effort is focused on scattering off the first spurious frequencies of the constitutive SIRs. Nonetheless, it turns out from Table I that all the other higher spurious modes happen to distribute themselves irregularly over the stopband. Hence, they cause no significant concern in the spurious response in even higher frequency bands. The four-pole parallel-coupled Chebyshev bandpass filter with a 0.1-dB ripple level was designed with the given specifications. The center frequency of the filter is 1.5 GHz, and the fractional bandwidth is 5.6%. In order to determine the physical dimensions of the filter, the coupling coefficients and the external quality factor can be found as

(5) A full-wave simulator IE3D has been used to design the filter and extract the above parameters. The coupling coefficient is and evaluated from two dominant resonant frequencies given as (6) where represents the coupling coefficient between resonators and . The proposed filters are designed to be fabricated using copper metallization on a Rogers RO4003 substrate with a

Fig. 5. Measured and simulated performance of the filter. (a) Insertion and return loss. (b) Wide-band response.

relative dielectric constant of 3.38, a thickness of 0.508 mm, and a loss tangent of 0.0027. Fig. 4 presents a photograph of the circuit. The size of the filter is 119.3 mm 28.5 mm, i.e., by 0.23 , where is the guided approximately 0.97 wavelength on the substrate at the center frequency. The fabricated filter was measured with an Agilent E8364B network analyzer. The measured and simulated results of the filter are illustrated in Fig. 5(a) and (b). The measured passband return loss is below 16 dB and the passband insertion loss is approximately 2.9 dB. The insertion loss would be attributed

CHEN et al.: DESIGN OF MICROSTRIP BANDPASS FILTERS WITH MULTIORDER SPURIOUS-MODE SUPPRESSION

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TABLE II DESIGN PARAMETERS OF THE SIR ON A SUBSTRATE OF DIELECTRIC CONSTANT 3.38 AND THICKNESS 0.508 mm

Fig. 6. (a) Layout of the four-pole elliptic-type bandpass filter. (b) Typical coupling structure.

mainly to the conductor loss. The measured results are in good agreement with the simulated predictions. Fig. 5(b) shows the wide-band response of this filter from 0.5 to 8.5 GHz. With this new structure, it is significant that the stopband rejection of the filter is better than 30 dB from 1.59 to 8.14 GHz, which shows that the filter has a very wide stopband rejection of up to 5.4 . It is interesting to compare the current result with those by other multispurious rejection filters described in [12] and [13], showing stopband rejection of up to approximately 8.2 and 4 , respectively, at which the second passband appears. On the contrary, the second passband will not show up by our design since it is very unlikely that the spurious modes of all the four resonators coincide at the same frequency. In addition to the ease of the design, this current method can obtain a quite wide stopband response with few numbers of resonators.

factor

. The coupling matrix [ ] and the external quality are found to be

(7) (8) The filter design is based on the knowledge of the coupling coefficients of the three basic coupling structures, which are the electric, magnetic, and mixed couplings, as described in [4]. In addition, the 0 feeding structure is introduced to realize the input and output ports of the filter [5]. A filter with 0 feeding structure could create two extra transmission zeros on opposite sides of the passband. Thus, the selectivity and out-of-band rejection of the filter can be significantly increased. For the th spurious frequency, the coupling coefficient can also be evaluated from [18] as follows:

IV. FOUR-POLE CROSS-COUPLED FILTER DESIGN The four-pole quasi-elliptic bandpass filter is also designed to demonstrate this concept. The configuration of the compact cross-coupled bandpass filter is shown in Fig. 6(a), which is composed of four different SIRs. The four SIRs were designed with the parameters listed in Table II. Each SIR has also been chosen with the same fundamental frequency and different spurious frequencies. Furthermore, each SIR is folded to achieve cross-coupling and to be compact. Fig. 6(b) shows the typical coupling structure of the four-pole cross-coupled bandpass filter, where each node represents a resonator. The solid lines and dotted line represent the direct coupling routes and the cross-coupling route, respectively. The multipath effect is introduced to exhibit a single pair of transmission zeros near the passband at finite frequency. The cross-coupled bandpass filter was designed with the given specifications. The center frequency of the filter is 1.51 GHz, and the fractional bandwidth is 4.5%. The lumped circuit element values of the low-pass prototype filter are found , , , , and to be

(9) and represent the resonance frequencies of reswhere onators and , respectively. This formulation is used to extract the coupling coefficient of any two asynchronously tuned coupled resonators. It should be noted that if this formulation is applied for synchronously tuned coupled resonators, then it simplifies to (6). Therefore, the calculated results show that the cou, , , and pling coefficients of are about one order less than that described in (7) for all the – ). This shows that there are very spurious modes ( weak couplings between either the direct coupling routes or the cross-coupling routes. It is anticipated that the spurious passbands can be rejected to a low level. The proposed filters are fabricated on the same Rogers RO4003 substrate as mentioned in Section III. Fig. 7 presents a photograph of the filter. The size of the filter is 0.22 39.2 mm 27.2 mm, i.e., only approximately 0.32 , where is the guided wavelength on the substrate at the center frequency.

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Fig. 7.

IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 53, NO. 12, DECEMBER 2005

Fabricated filter.

the extra transmission zeros is at 1.28 GHz and the other is at 1.83 GHz. The measured passband return loss is below 17 dB and the passband insertion loss is approximately 2.7 dB. The insertion loss would be attributed mainly to the conductor loss. The measured results are in good agreement with the simulated predictions. Fig. 8(b) shows the wide-band response of this filter from 0.5 to 13 GHz. As can be seen, the filter has a rejection level lower than 30 dB for the spurious passbands. The stopband rejection of the filter is better than 30 dB from 1.57 to 12.36 GHz. This implies that the filter exhibits an excellent upper stopband rejec. As a result, the proposed compact elliptiction of up to type filter not only has high selectivity, but also has a wide upper stopband. In addition, it is found that the attenuation level in the stopband depends on the order of SIRs. The larger the order of SIRs, the better the rejection level in the stopband with a larger circuit size and insertion loss as tradeoff. V. CONCLUSION This paper has presented a simple and efficient design method for microstrip bandpass filters with multiorder spurious-mode suppression. The method has been demonstrated by applying it to a parallel-coupled bandpass filter and a cross-coupled bandpass filter. With a combination of different SIR structures, the spurious passbands of the filter can be rejected by properly adjusting the spurious frequencies of each SIR to different values. Both theoretical and experimental results are presented to verify this concept. The measured results are in good agreement with the simulated predictions, showing that better than 30-dB rejection levels in the stopband up to 5.4 and 8.2 are achieved by the Chebyshev and quasi-elliptic filters, respectively. As a result, this method is quite useful for applications in mobile communication systems when good stopband response and high selectivity are required. REFERENCES

Fig. 8. Measured and simulated performance of the filter. (a) Insertion and return loss. (b) Wide-band response.

The measured and simulated results of the filter are illustrated in Fig. 8(a) and (b). It can be expected that there are a single pair of transmission zeros near the passband due to the cross-coupling effect. The effect of the two transmission zeros at 1.44 and 1.58 GHz is observed. It can be also clearly observed that there are two extra transmission zeros on opposite sides of the passband due to the zeroth-degree feeding structure. One of

[1] D. M. Pozar, Microwave Engineering, 2nd ed. New York: Wiley, 1998, ch. 8. [2] J. S. Hong and M. J. Lancaster, “Couplings of microstrip square openloop resonators for cross-coupled planar microwave filters,” IEEE Trans. Microw. Theory Tech., vol. 44, no. 11, pp. 2099–2109, Nov. 1996. , “Cross-coupled microstrip hairpin-resonator filters,” IEEE Trans. [3] Microw. Theory Tech., vol. 46, no. 1, pp. 118–122, Jan. 1998. [4] , “Design of highly selective microstrip bandpass filters with a single pair of attenuation poles at finite frequencies,” IEEE Trans. Microw. Theory Tech., vol. 48, no. 7, pp. 1098–1107, Jul. 2000. [5] C. M. Tsai, S. Y. Lee, and C. C. Tsai, “Performance of a planar filter using a 0 feed structure,” IEEE Trans. Microw. Theory Tech., vol. 50, no. 10, pp. 2362–2367, Oct. 2002. [6] C. C. Chen, Y. R. Chen, and C. Y. Chang, “Miniaturized microstrip crosscoupled filters using quarter-wave or quasi-quarter-wave resonators,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 1, pp. 120–131, Jan. 2003. [7] C. H. Wang, Y. S. Lin, and C. H. Chen, “Novel inductance-incorporated microstrip coupled-line bandpass filters with two attenuation poles,” in IEEE MTT-S Int. Microwave Symp. Dig., 2004, pp. 1979–1982. [8] J. T. Kuo, W. H. Hsu, and W. T. Huang, “Parallel coupled microstrip filters with suppression of harmonic response,” IEEE Microw. Wireless Compon. Lett., vol. 12, no. 10, pp. 383–385, Oct. 2002. [9] M. del Castillo Velázquez-Ahumada, J. Martel, and F. Medina, “Parallel coupled microstrip filters with ground-plane aperture for spurious band suppression and enhanced coupling,” IEEE Trans. Microw. Theory Tech., vol. 52, no. 3, pp. 1082–1086, Mar. 2004.

CHEN et al.: DESIGN OF MICROSTRIP BANDPASS FILTERS WITH MULTIORDER SPURIOUS-MODE SUPPRESSION

[10] J. T. Kuo, S. P. Chen, and M. Jiang, “Parallel-coupled microstrip filters with over-coupled end stages for suppression of spurious responses,” IEEE Microw. Wireless Compon. Lett., vol. 13, no. 10, pp. 440–442, Oct. 2003. [11] T. Lopetegi, M. A. G. Laso, J. Hernández, M. Bacaicoa, D. Benito, M. J. Garde, M. Sorolla, and M. Guglielmi, “New microstrip wiggly-line filters with spurious passband suppression,” IEEE Trans. Microw. Theory Tech., vol. 49, no. 9, pp. 1593–1598, Sep. 2001. [12] J. T. Kuo and E. Shih, “Microstrip stepped impedance resonator bandpass filter with an extended optimal rejection bandwidth,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 5, pp. 1554–1559, May 2003. [13] J. G. García, F. Martín, F. Falcone, J. Bonache, I. Gil, T. Lopetegi, M. A. G. Laso, M. Sorolla, and R. Marqués, “Spurious passband suppression in microstrip coupled line band pass filters by means of split ring resonators,” IEEE Microw. Wireless Compon. Lett., vol. 14, no. 9, pp. 416–418, Sep. 2004. [14] T. Lopetegi, M. A. G. Laso, F. Falcone, F. Martin, J. Bonache, J. Garcia, L. Perez-Cuevas, M. Sorolla, and M. Guglielmi, “Microstrip ‘wigglyline’ bandpass filters with multispurious rejection,” IEEE Microw. Wireless Compon. Lett., vol. 14, no. 11, pp. 531–533, Nov. 2004. [15] S. Denis, C. Person, S. Toutain, S. Vigneron, and B. Theron, “Improvement of global performances of bandpass filters using nonconventional stepped impedance resonators,” in Proc. 28th Eur. Microwave Conf. Dig., vol. 2, 1998, pp. 323–328. [16] M. Makimoto and S. Yamashita, “Bandpass filters using parallel coupled stripline stepped impedance resonators,” IEEE Trans. Microw. Theory Tech., vol. 28, no. 12, pp. 1413–1417, Dec. 1980. [17] M. Sagawa, M. Makimoto, and S. Yamashita, “Geometrical structures and fundamental characteristics of microwave stepped-impedance resonators,” IEEE Trans. Microw. Theory Tech., vol. MTT-45, no. 7, pp. 1078–1085, Jul. 1997. [18] J. S. Hong and M. J. Lancaster, Microstrip Filter for RF/Microwave Applications. New York: Wiley, 2001, ch. 8.

Chi-Feng Chen was born in PingTung, Taiwan, R.O.C., on September 3, 1979. He received the B.S. degree in physics from the Chung Yuan Christian University, Taoyuan, Taiwan, R.O.C., in 2001, the M.S. degree in electrophysics from the National Chiao Tung University, Hsinchu, Taiwan, R.O.C., in 2003, and is currently working toward the Ph.D. degree in communication engineering at National Taiwan University, Taipei, Taiwan, R.O.C. His research interests include the design of microwave filters and associated RF modules for microwave and millimeter-wave applications.

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Ting-Yi Huang was born in Hualien, Taiwan, R.O.C., on November 12, 1977. He received the B.S. degree in electrical engineering and M.S. degree in communication engineering from National Taiwan University, Taipei, Taiwan, R.O.C., in 2000 and 2002, respectively, and is currently working toward the Ph.D. degree in communication engineering at National Taiwan University. His research interests include computational electromagnetics, the design of microwave filters, transitions, and associated RF modules for microwave and millimeter-wave applications.

Ruey-Beei Wu (M’91–SM’97) received the B.S.E.E. and Ph.D. degrees from National Taiwan University, Taipei, Taiwan, R.O.C., in 1979 and 1985, respectively. In 1982, he joined the faculty of the Department of Electrical Engineering, National Taiwan University, where he is currently a Professor. He is also with the Graduate Institute of Communications Engineering, National Taiwan University, which was established in 1997. From March 1986 to February 1987, he was a Visiting Scholar with IBM, East Fishkill, NY. From August 1994 to July 1995, he was with the Electrical Engineering Department, University of California at Los Angeles. He was also appointed Director of the National Center for High-Performance Computing (1998–2000) and has served as Director of Planning and Evaluation Division since November 2002, both under the National Science Council. His areas of interest include computational electromagnetics, transmission line and waveguide discontinuities, microwave and millimeter-wave planar circuits, and interconnection modeling for computer packaging.

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High-Performance Circular TE01-Mode Converter Ching-Fang Yu and Tsun-Hsu Chang, Member, IEEE

Abstract—This study presents the design and cold testing of a -band TE01 -mode converter. A wave is efficiently converted from the TE10 rectangular waveguide mode into the TE01 circular waveguide mode. This converter comprises a power-dividing section and a mode-converting section. The field pattern and the working principle of each section are analyzed and discussed. A prototype was built and tested. Back-to-back transmission measurements exhibit excellent agreement to the results of computer simulations. The measured optimum transmissions are 97% with a 1-dB bandwidth of 5.8 GHz centered at 34.0 GHz. The angle-independent transmissions manifest high mode purity and the field pattern is directly demonstrated on a temperature-sensitive liquidcrystal sheet. In addition to exhibiting a high conversion efficiency, high mode purity, and broad bandwidth, this converter is also easy to construct and is structurally simple. Index Terms—Converter, coupler, gyrotron, mode purity, wave launcher.

I. INTRODUCTION

T

HE circular waveguide mode, featuring azimuthally symmetric electric field and low wall loss, has drawn much attention in relation to a variety of applications. These include electron-cyclotron maser-based devices, gyrotrons [1]–[13], plasma processing systems [14]–[16], and the next linear colliders and antennas [17], [18]. Gyrotrons require highperformance converters for many years. It launches a wave into an interaction circuit or couples the wave out for further applications. The performance of the mode converter directly affects the stability and tunability of the gyrotrons. mode can be classified Means of generating a circular into two categories according to the coupling methods involved. One is inline coupling [8], [11], [12], [14]–[23] and the other is sidewall coupling [2]–[5], [6], [7], [9], [10], [13], [24]. The former, using a deformed waveguide structure, gradually converts a wave into the desired mode. The transition length is typically long and multiple modes can be excited during the converting process. Among them, the Tantawi converter [6], [17], [19], Marie transducer [21], [22], serpentine/corrugated converter [8], [11], [20], and sector converter [23] are typical. In addition, Guo’s converter [12], taking advantage of the wave nature, efficiently converts the waves between circular and circular modes. On the other hand, the latter generally uses a smooth waveguide with coupling holes on the

Manuscript received March 16, 2005; revised July 27, 2005. This work was supported by the National Science Council of Taiwan under Contract NSC-932112-M-007-019. The authors are with the Department of Physics, National Tsing Hua University, Hsinchu, Taiwan, R.O.C. (e-mail: [email protected]; [email protected]). Digital Object Identifier 10.1109/TMTT.2005.859866

Fig. 1. Schematic diagram of the TE -mode converter. The converter consists of two sections—the power-dividing section and mode-converting section.

sidewall. The typical coupling method is the multihole directional coupler, which is based on the field vector superposition theorem [3], [7]. Recently, coaxial couplers have been generally adopted because of their high converting efficiency [2], [4], [5], [10], [13], [24]. However, the existence of the unwanted modes during the transition could interact with an electron beam, resulting in a serious mode-competition problem for gyrotron applications. Thus, shortening the converting length and enhancing the mode purity help to eliminate the complicated mode-competition problem. This study presents a sidewall coupling mode converter, which takes advantage of the wave nature to excite a pure mode. This converter allows the electron beam to pass through it and to interact with the wave and, thus, it is especially suitable for gyrotron amplifier/oscillator applications. The rest of this paper is organized as follows. Section II shows the principle of operation and numerical results, and discusses the field forming processes and mode purity analysis. Section III describes fabrication and experimental results. Back-to-back measurements are present and field profiles are displayed. Finally, Section IV presents conclusions. II. PRINCIPLES AND SIMULATION Fig. 1 schematically depicts the circular -mode converter, which is a two-port device. A wave injected into port 1 mode) is split into four signals of (rectangular waveguide equal amplitude with special orientations in the power-dividing section. These four signals are then led into a circular waveguide mode, in the mode-converting section to form a circular which will then emerge from port 2. Numerical simulation is conducted using a full-wave solver—High Frequency Structure Simulator (HFSS).1 1HFSS, Ansoft Corporation, http://www.ansoft.com/

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-MODE CONVERTER

Fig. 2. HFSS simulation results. (a) Distribution of the electric-field strength of the power-dividing section viewed at the middle cross section of the rectangular waveguide. (b) Frequency response of the reflection at port 1; ports 1a–d are assumed to be matched to eliminate the effect of multiple reflections.

A. Power-Dividing Section The input power is first divided into two signals of equal amplitude through a Y-shaped power divider. Each signal is then further divided through another Y-shaped power divider. Therefore, the cascaded Y-shaped power dividers quarter the input power. Fig. 2(a) shows the cross section of the electric-field distribution and the electric-field direction. The field strength is expressed as a grayscale image and the length of arrows. At the ends of the four output ports (ports 1a–d), the field strength are the same, but the electric-field orientations vary by 90 , indicating that all of the field strengths are the same, but the directions are counterclockwise, as viewed from the top. Such a power division is also discussed in [25] and [26]. Fig. 2(b) shows the reflection coefficient of the input port. Ports 1a–d are well terminated to prevent the complications caused by multiple reflections. Only the reflection at port 1 is examined. Optimizing the geometry of the Y-shaped splitters minimizes the reflection. The calculated reflection coefficient is better than 30 dB over the entire frequency range. Since the reflection of the power-dividing section is exceptional small, the mode-converting section determines the bandwidth of the coupler, as will be shown in Section II-B. B. Mode-Converting Section Fig. 3(a) presents the outline of the mode-converting section and the cross section of the electric-field distribution. The field strength and direction are expressed in grayscale and by the directions of the arrows. The four signals of equal amplitude, but various orientations are injected into intermediary ports 2a–d.

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Fig. 3. (a) Cross-sectional view of the electric-field distribution with HFSS. (b) The reflection of port 2 with ports 2a–d matched.

Sidewall coupling separates each by 90 along the circumferwaveguide mode. A microwave ence, forming the circular short (waveguide cutoff) is placed at the opposite end to ensure that the wave propagates in the desired direction. The resultant wave emerges from port 2. circular The microwave short is made of a circular waveguide of radius 4.2 mm with cutoff frequency of 43.6 GHz. In gyrotron applications, this hole functions as the beam tunnel. Its diameter and position are optimized for the beam spot size and the wave transmission. In addition, a tapered geometry is used at the joints between the rectangular and circular waveguide, effectively minimizing the reflection. The principle concern in this section is to minimize the reflection at port 2, for which minimization can be achieved by optimizing the position of the waveguide cutoff and the joint geometry. Fig. 3(b) shows the optimized reflection in port 2, with ports 2a–d matched. The center frequency is 34 GHz and the 10-dB bandwidth is around 5.6 GHz. C. Analyzing Mode Purity The next step is to put these two sections together. Fig. 4(a) shows the simulated electric-field strength of the converter on the surface and at the ports. This figure presents the modeconverting process. Given a radius of 6.0 mm, the cutoff frequencies for the first six modes are 14.7, 19.1, 24.3, 30.5, , , , , , and 30.5, and 33.4 GHz for , respectively. Therefore, when the desired mode is being excited, the concentration of the other five modes shall be kept as low as possible. The sidewall couplings prevent the

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Fig. 4. (a) Electric-field strengths on the surface of the converter simulated with HFSS. (b) Calculated transmissions and reflection associated with a single converter.

excitations of TM waves because of the orientation of the electric field. Additionally, the quad-feed structure is unfavorable to the , , and modes. It is instead suitable for a fourfold or a circular symmetric field pattern. In the range mode can be formed. of operating frequencies, only the Therefore, high mode purity is expected. Fig. 4(b) shows the transmission losses of the first six modes and the reflection loss at port 1. A rectangular waveguide mode injected into port 1 was converted into six circular waveguide modes at port 2. The converting efficiency of a specific mode is defined as the output power of this mode at port 2 divided by the input power at port 1. The converting efficiency of the desired mode is very high and those of the other five modes are extremely low (below 0.01%). Close to the center frequency, the converting efficiency of the desired mode is approximately 98.5%. The remaining 1.5% is mainly because of reflection and ohmic loss. The concentrations of the spurious modes are basically less than 40 dB. Numerical calculations have shown an excellent performance. We are now in a position to verify these predictions. III. FABRICATION AND EXPERIMENTAL MEASUREMENT Fig. 5(a) and (b) displays the design drawings and finished parts of the converter. The converter comprises two pieces—a slotted plate and a plane cover. It is made of oxygen-free high-conductivity (OFHC) copper. The slotted plate is machined using a computer numerically controlled (CNC) lathe with a tolerance of 0.01 mm. Plates are aligned with pins and

Fig. 5. (a) Design drawings of the converter including a slotted plane (left) and plane cover (right). (b) The finished parts, which are made of OFHC copper.

fastened with screws. In the cold test, two converters are joined back to back, which allows a direct measurement to be made using a two-port vector network analyzer (VNA), Agilent 8510C, Agilent Technologies, Palo Alto, CA. A. Back-to-Back Measurement Fig. 6(a) shows the experimental setup. Two identical converters are joined back-to-back through a uniform middle section of 1.0 cm. A well-calibrated two-port VNA is employed. The measured results are highly consistent with the HFSS simulation data, as shown in Fig. 6(b). The bandwidths associated with the 1- and 3-dB transmission losses are 5.8 and 7.0 GHz, respectively. The measured converting efficiency of a single converter is approximately 98.5%. The ohmic loss and reflection account for the other is 1.5%. Measurements are made using a rotatable setup to further explore the field symmetry and to examine the existence of competing modes. B. Verifying Azimuthal Symmetry Fig. 7(a) schematically depicts the experimental setup. The angle between the two identical converters can be adjusted. Three angles are used in this measurement. They are 0 , 45 , and 90 . Fig. 7(b) shows the measured results of transmission using these three setups. The transmission is almost independent of the angle for a broad bandwidth, which rules out the existence and circular polarization of the spurious modes. Thus, the excellent agreement between back-to-back measurements and the simulation was demonstrated. The next step is to display the mode in the circular waveguide assofield profile of the ciated with a single converter.

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-MODE CONVERTER

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Fig. 8. Schematic diagram of the experimental setup for directly measuring the field distribution pattern. The power, generated by a TWT and controlled by a sweeper/synchronizer, is injected into the mode converter. A sheet of temperature-sensitive LCD is placed in front of the circular horn, where the grayscale image shows the time-averaged field distribution.

C. Visualizing Field Profile

Fig. 6. (a) Two identical converters joined back to back. (b) Measured and simulated results of the back-to-back transmission.

The following setup is used and measurements are made to visualize the field pattern directly. Fig. 8 schematically depicts the experimental setup and results. The microwave power is provided by the traveling-wave tube (TWT) amplifier (Hughes 1077H, Hughes, Germantown, MD) driven by a synchronizer (Agilent Technologies 8357a). An input power of 0.5 W, monitored using a calibrated circuit and power meter, is injected into the converter. A slightly tapered conical horn is connected at the end of the converter to enlarge the size of the field pattern for visual inspection. A temperature-sensitive liquid crystal display (LCD) sheet, displaying a full-color spectrum when the temperature changes from 25 C to 30 C, is placed in front of the horn. The color spectrum on the LCD is directly correlated with the microwave energy pattern. Fig. 8 displays the circular and azimuthal symmetric field pattern as evidence of the purity of the mode. circular IV. CONCLUSION A -band circular -mode converter has been developed, fabricated, and tested. This converter features high back-to-back converting efficiency (97%), high mode purity , and broad (99.99%), compact converting section bandwidth (17% at a 1-dB transmission). Such a converter is suitable for a variety of applications, especially the gyrotrons. Although the presence of the electron beam might slightly change the performance of the converter, a -band gyrotron backward-wave oscillator (gyro-BWO) is being developed based on this converter. ACKNOWLEDGMENT

Fig. 7. (a) Drawing of the two converters jointed back to back and rotated at an angle of . (b) The measured transmission results for three angles.

The authors would like to thank Dr. L. R. Barnett, University of California at Davis, Prof. K. R. Chu, National Tsing Hua University, Hsinchu, Taiwan, R.O.C., and Prof. Y. S. Yeh, South Taiwan University of Technology, Tainan, Taiwan, R.O.C., for many helpful discussions. The technical support of C. Lee, Ansoft, Taiwan, R.O.C., is greatly appreciated.

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REFERENCES [1] K. R. Chu, “The electron cyclotron maser,” Rev. Mod. Phys., vol. 76, no. 2, pp. 489–540, Apr. 2004. [2] M. Blank, B. G. Danly, and B. Levush, “Experimental demonstration of a -band (94 GHz) gyrotwystron amplifier,” IEEE Trans. Plasma Sci., vol. 27, no. 2, pp. 405–411, Apr. 1999. [3] K. C. Leou, D. B. McDermott, A. J. Balkcum, and N. C. Luhmann, Jr., “Stable high-power TE gyro-TWT amplifiers,” IEEE Trans. Plasma Sci., vol. 22, no. 5, pp. 585–592, Oct. 1994. [4] M. Garven, J. P. Calame, B. G. Danly, K. T. Nguyen, B. Levush, F. N. Wood, and D. E. Pershing, “A gyrotron-traveling wave tube amplifier experiment with a ceramic loaded interaction region,” IEEE Trans. Plasma Sci., vol. 30, no. 3, pp. 885–893, Jun. 2002. [5] D. B. McDermott, H. H. Song, Y. Hirata, A. T. Lin, L. R. Barnett, T. H. Chang, H. L. Hsu, P. S. Marandos, J. S. Lee, K. R. Chu, and N. C. Luhmann, Jr., “Design of a -band TE mode gyrotron traveling-wave amplifier with high power and broad-band capabilities,” IEEE Trans. Plasma Sci., vol. 30, no. 3, pp. 894–902, Jun. 2002. [6] J. P. Tate, H. Guo, M. Naiman, L. Chen, and V. L. Granatstein, “Experimental proof-of-principle results on a mode-selective input coupler for gyrotron applications,” IEEE Trans. Microw. Theory Tech., vol. 42, no. 8, pp. 1910–1917, Aug. 1994. [7] W. Wang, Y. Gong, G. Yu, L. Yue, and J. Sun, “Mode discriminator based on mode-selective coupling,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 1, pp. 55–63, Jan. 2003. [8] W. Lawson, M. R. Arjona, B. P. Hogan, and R. L. Ives, “The design of serpentine-mode converters for high-power microwave applications,” IEEE Trans. Microw. Theory Tech., vol. 48, no. 5, pp. 809–814, May 2000. [9] T. A. Spencer, C. E. Davis, K. J. Hendricks, F. J. Agee, and R. M. Gilgenbach, “Results from gyrotron backward wave oscillator experiments utilizing a high-current high-voltage annular electron beam,” IEEE Trans. Plasma Sci., vol. 24, no. 3, pp. 630–635, Jun. 1996. [10] M. Blank, B. G. Danly, B. Levush, P. E. Latham, and D. E. Pershing, “Experimental demonstration of a -band gyroklystron amplifier,” Phys. Rev. Lett., vol. 79, no. 22, pp. 4485–4488, Dec. 1997. [11] M. J. Buckley and R. J. Vernon, “Compact quasi-periodic and aperiodic TE mode converters in overmoded circular waveguides for use with gyrotrons,” IEEE Trans. Microw. Theory Tech., vol. 38, no. 6, pp. 712–721, Jun. 1990. [12] H. Guo, S. H. Chen, V. L. Granatstein, J. Rodgers, G. S. Nusinovish, M. T. Walter, J. Zhao, and W. Chen, “Operation of a high performance, harmonic-multiplying, inverted gyrotwystron,” IEEE Trans. Plasma Sci., vol. 26, no. 3, pp. 451–460, Jun. 1998. [13] B. Levush, M. Blank, J. Calame, B. Danly, K. Nguyen, D. Pershing, S. Cooke, P. Latham, J. Petillo, and T. Antonsen, Jr., “Modeling and design of millimeter wave gyroklystrons,” Phys. Plasmas, vol. 6, no. 5, pp. 2233–2240, May 1999. [14] R. L. Kinder and M. J. Kushner, “TE excitation of electron cyclotron resonance plasma source,” IEEE Trans. Plasma Sci., vol. 27, no. 1, pp. 64–65, Feb. 1999. [15] Y. Kato, H. Furuki, T. Asaji, and S. Ishii, “Production of multicharged ions in a 2.45 GHz electron cyclotron resonance source directly excited in a circular TE mode cavity resonator,” Rev. Sci. Instrum., vol. 75, no. 5, pp. 1470–1472, 2004. [16] R. Hidaka, N. Hirotsu, N. Tanaka, and Y. Kawai, “Generation of electron cyclotron resonance plasmas using a circular TE mode microwave,” J. Appl. Phys., vol. 72, no. 9, pp. 4461–4462, Nov. 1992. [17] S. G. Tantawi, C. Nantista, N. Kroll, Z. Li, R. Miller, R. Ruth, and P. Wilson, “Multimoded RF delay line distribution system for the next linear collider,” Phy. Rev. ST Accel. Beams, vol. 5, 032001, 2002.

W

W

W

[18] R. D. Wengenroth, “A mode transducing antenna,” IEEE Trans. Microw. Theory Tech., vol. MTT-26, no. 5, pp. 332–334, May 1978. [19] I. Spassovsky, E. S. Gouveia, S. G. Tantawi, B. P. Hogan, W. Lawson, and V. L. Granatstein, “Design and cold testing of a compact TE to TE mode converter,” IEEE Trans. Plasma Sci., vol. 30, no. 3, pp. 787–793, Jun. 2002. [20] M. J. Buckley, D. A. Stein, and R. J. Vernon, “A single-period TE - -TE mode converter in a highly overmoded circular waveguide,” IEEE Trans. Microw. Theory Tech., vol. 39, no. 8, pp. 1301–1306, Aug. 1991. [21] G. R. P. Marie, “Mode transforming waveguide transition,” U.S. Patent 2 859 412, Nov. 4, 1958. [22] W. A. Huting and K. J. Webb, “Comparison of mode-matching and differential equation techniques in the analysis of waveguide transitions,” IEEE Trans. Microw. Theory Tech., vol. 39, no. 2, pp. 280–286, Feb. 1991. [23] F. Sporleder and H. G. Unger, Waveguide Tapers Transitions and Couplers. New York: Peregrinus, 1979, ch. 7. [24] A. H. McCurdy and J. J. Choi, Design and Analysis of a Coaxial Coupler for a 35-GHz Gyroklystron Amplifier, vol. 47, no. 2, pp. 164–175, Feb. 1999. [25] R. J. Barker, N. C. Luhmann, Jr., J. H. Booske, and G. S. Nusinovich, Modern Microwave and Millimeter-Wave Power Electronics. Piscataway, NJ: IEEE Press, 2005, ch. 11. [26] T. H. Chang and C. F. Yu, “Polarization controllable TE mode converter,” Rev. Sci. Instrum., vol. 76, 074703, 2005.

Ching-Fang Yu was born in Taipei, Taiwan, R.O.C., in 1978. He received the B.S. degree in physics from National Cheng Kung University, Taiwan, R.O.C., in 2000, the M.S. degree in physics from National Tsing Hua University, Taiwan, R.O.C., in 2002, and is currently working toward the Ph.D. degree in physics at National Tsing Hua University. His current research includes the design, construction, and characterization of high-order-mode converters and their applications to gyro-BWOs.

Tsun-Hsu Chang (M’99) received the B.S. degree from National Central University, Taoyuan, Taiwan, R.O.C., in 1991, and the Ph.D. degree from National Tsing Hua University, Hsinchu, Taiwan, R.O.C., in 1999. Upon graduation, he was a Post-Doctoral Researcher involved with the development of highpower millimeter-wave sources (gyrotrons) for two years. From 2001 to 2003, he was a Technical Manager with the Silicon Integrated System Corporation, where he was responsible for analyzing high-speed signals and power integrity. He is currently an Assistant Professor with the Department of Physics, National Tsing Hua University. His current research focuses on the nonlinear dynamics and nonstationary behavior of the electron cyclotron maser and characterizes microwave/nanoparticles interaction.

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Compact U-Shaped Dual Planar EBG Microstrip Low-Pass Filter Shao Ying Huang, Student Member, IEEE, and Yee Hui Lee, Member, IEEE

Abstract—In this paper, a novel high-performance compact dual planar electromagnetic bandgap (DP-EBG) microstrip low-pass filter with a U-shaped geometry is proposed. By employing the unique DP-EBG configuration and the U-shaped geometry of the microstrip line (MLIN), the proposed structure achieves a wide stopband with high attenuation and a high selectivity within a small circuit area. Its passband ripple level is low due to the U-shaped geometry and the electromagnetic bandgap (EBG) structure with square patches inserted at the bends of the MLIN. The Chebyshev tapering technique is used to taper components of the proposed structure in order to eliminate ripples caused by the EBG periodicity. The structure was fabricated and the measured results are in good agreement with the simulated results. This novel design demonstrates superior low-pass filtering functionality and can easily be applied to monolithic circuits. Index Terms—Electromagnetic bandgap (EBG), low-pass filters, planar passive filters, tapering techniques.

I. INTRODUCTION

T

HE electromagnetic bandgap (EBG) structure has been a term widely accepted today to call the artificial periodic structure that prohibits the propagation of electromagnetic waves in certain frequency bands at microwave or millimeterwave frequencies. These periodic structures were originally proposed at optical frequencies [1], [2] and are known as a photonic bandgap (PBG) structure or photonic crystal (PC). Analogous to crystals where periodic arrays of atoms produce bandgaps in which the propagation of photon is prohibited, an artificial periodic structure is comprised of periodic macroscopic cells. These periodic structures are scalable over a wide frequency range in the electromagnetic spectrum. Due to their scalability, research has progressed into the range of microwave and millimeter wave [3]. The unique feature of EBG structures is the existence of the bandgap where electromagnetic waves are not allowed to propagate. They have been widely applied to the substrate of microwave circuits such as patch antennas and power amplifiers to improve their performance [4], [5]. The introduction of the planar EBG structure [6], [7] where two-dimensional (2-D) periodic elements are introduced in the ground plane or the microstrip line (MLIN), simplifies the fabrication process of EBG structures while maintaining a similar control on the wave propagation to that of an electromagnetic crystal where three-dimensional (3-D) periodic elements are arranged in a host medium

Manuscript received March 21, 2005; revised July 15, 2005. The authors are with the School of Electrical and Electronic Engineering, Nanyang Technological University, Singapore 639798, Singapore (e-mail: [email protected]; [email protected]). Digital Object Identifier 10.1109/TMTT.2005.859865

(complex and costly fabrication process is required). The only tradeoff of planar EBG structures is that they are not able to control the propagation of waves in the entire 3-D space. Nevertheless, planar EBG structures have attracted much attention because of their prominent stopband characteristic, their ease of fabrication, and their compatibility with monolithic circuits. With the superior filtering functionality associated with the bandgap, planar EBG structures are employed in the design of high-performance microstrip low-pass filters. In [8], a microstrip low-pass filter with a periodic 2-D EBG pattern consisting of 9 3 circles etched in the ground plane was proposed. It is able to achieve a broad rejection band of up to 10 GHz. With the uniplanar compact photonic bandgap (UC-PBG) structure [9] etched in the ground plane of a conventional low-pass filter [10], the performance of the low-pass filter can be improved in terms of attenuation in the stopband and spurious suppression. In a traditional straight one-dimensional (1-D) EBG microstrip structure [7], a good stopband performance is usually obtained by increasing the number or the size of EBG cells, thus resulting in an increase in the circuit area. These EBG structures have cells in a single plane. They are a compromise between good filtering performance and compact physical size. The problem above was well addressed by introducing multiple bends in the MLIN giving rise to an EBG filter structure with an excellent rejection band in a relatively small physical size [11]. The undesirable ripples caused by the meander MLIN are significantly removed by applying several techniques such as to tune the length of the MLIN in a bent unitary period and to modify the length of resonators in the transmission line [12]. In [13], [14], a dual planar EBG (DP-EBG) configuration was proposed where EBG cells are in both the MLIN and the ground plane to obtain good filtering functionality with a small number of EBG cells (small circuit area). In [15], the width of the MLIN in a DP-EBG structure is reduced and tapered in order to obtain excellent stopband and passband performance in a small physical size. In this paper, a novel high-performance compact DP-EBG microstrip low-pass filter structure with a U-shaped MLIN geometry is proposed and implemented. With the unique U-shaped geometry of the MLIN and the DP-EBG configuration [14] with a reduction in the width of the MLIN [15], as well as the adopted Chebyshev distribution [16], the proposed filter structure demonstrates a high selectivity, a low ripple level in the passband, and a rejection of below 20 dB in the range from 3.4 to above 11 GHz within a small circuit area. As compared to previous research on EBG microstrip low-pass filters, the proposed structure achieves better performance with a significant reduction in physical size.

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Fig. 1. Schematic of: (a) a straight EBG microstrip structure, (b) a U-shaped MLIN, and (c) a U-shaped EBG microstrip structure.

In Section II, the U-shaped MLIN geometry is studied and the performance of the U-shaped EBG microstrip structure with square patches inserted in the MLIN is analyzed. The design of the compact U-shaped DP-EBG microstrip low-pass filter structure is proposed in Section III. The proposed structure was fabricated and tested. The measured results are presented and analyzed in Section IV. II. STUDY OF U-SHAPED GEOMETRY

not a conventional stepped-impedance microstrip low-pass filter [17] since it is designed to satisfy the Bragg reflection condition. equals half of the peThe edge of the inserted square patch riod , which is equivalent to an optimal filling factor of 0.25 [6]. Compared to the straight EBG microstrip structure, the U-shaped EBG microstrip structure in Fig. 1(c) is more compact in physical size and may provide additional flexibility in circuit layout designs. The performance of the structure in Fig. 1(c) is affected by the resonator created in the MLIN that is presented in its corresponding U-shaped MLIN [see Fig. 1(b)] because the resonator causes resonance in the transmission of the U-shaped EBG structure. The resonator in an MLIN with is the result of the multiple bends the number of bends reflections between two consecutive or nearby bends (with one bend in between). The resonant frequency is determined by the physical length of the resonator. Base on a study on the meander MLIN [12], the relation between the length of the resonator and the resonant frequency can be expressed using (3) as follows: (3) where is the physical length of the resonator, is an integer, is the guided wavelength corresponding to the resonant and is expressed as the following: frequency . (4) Hence,

A. U-Shaped EBG Microstrip Structure

(5)

Fig. 1(a) shows the schematic of a straight EBG microstrip structure with four square patches inserted in the MLIN at a period of and Fig. 1(b) shows that of a U-shaped MLIN where the distance between the two parallel sections is also . Fig. 1(c) shows the schematic of the U-shaped EBG microstrip structure used in the proposed compact U-shaped DP-EBG microstrip low-pass filter structure. It is a combination of the structures in Fig. 1(a) and (b) with the 90 bending taking place at the center point of the second and the third patch from the leftto right-hand side. The EBG structures in Fig. 1(a) and (c) are single planar (SP) structures since their EBG cells are arranged only in the MLIN. The straight EBG microstrip structure in Fig. 1(a) satisfies the Bragg reflection condition [7], which is expressed by the following equation: (1) where is the wavenumber in the substrate material and is where is the the period of the structure. Since guided wavelength corresponding to the center frequency of the is decided by stopband , (2) where is the speed of light in free space and is the effective dielectric constant. The straight EBG microstrip structure is

As can be seen in Fig. 1(b), the U-shaped MLIN contains two bends. Therefore, it has only one resonator with a length of , as shown by the shadowed area. According to (5), with set to 1, its resonant frequency is the same as the center frequency of the stopband in the straight EBG structure [see Fig. 1(a)] as approximated by (2). As compared to the meander line in [12], the U-shaped MLIN has less bends. As a result, U-shaped structures may have smaller power loss caused by the radiation and scattering that take place at the bend. In the U-shaped MLIN, there is only one resonator and its length is as short as one EBG period. Due to the absence of a resonator with a length longer than , no resonance is introduced at frequencies lower than . B. Numerical Results and Analysis The transmission characteristics of an EBG microstrip structure with multiple bends are determined by the responses of the two individual structures; the corresponding straight EBG structure and the MLIN with a corresponding geometry. In order to study the proposed U-shaped EBG microstrip structure, the three structures in Fig. 1 are simulated using the method-of-moment (MoM)-based software Zeland IE3D. Taconic, with a diof 2.43 and a thickness of 0.76 mm, electric constant is used as the substrate. The center frequency of the stopband is set to 7.5 GHz, resulting in a period of the structure of 13.78 mm. The edge of the square patch is set to 6.89 mm,

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Fig. 3. Simulated phase delay in the output wave with respect to the input wave in a bent unitary period and in a straight unitary period.

Fig. 2.

Simulated S -parameters of structures in Fig. 1. (a) S . (b) S .

which equals half of the period. The width of the MLIN is 2.32 mm corresponding to a characteristic impedance of 50 at - and -pa7.5 GHz. Fig. 2(a) and (b) shows the simulated rameter, respectively. As can be seen in Fig. 2(a), the U-shaped MLIN shows slight attenuation in the higher frequency range of 6 GHz and above. The small resonance in the transmission is due to the resonator where reflections take place between the bends. With a physical length of 13.78 mm, the resonator causes slight resonance centered at approximately 7.5 GHz, which is the same as that predicted using (5). The straight EBG microstrip structure shows a prominent stopband centered at approximately 7 GHz, which indicates that it satisfies the Bragg reflection condition. The U-shaped EBG microstrip structure (a combination of the two above-mentioned structures) obtains a wider stopband with higher attenuation and an increase in selectivity, yet without compromising the ripple level in the lower passband. The low passband ripple level of the U-shaped EBG structure is -parameter in indicated by the low sidelobe level of its Fig. 2(b). The results reveal that the U-shaped MLIN geometry is able to enhance the filtering functionality of the EBG structure with square patches inserted in the MLIN. The enhancement is ob-

tained without producing any additional ripple. This is mainly because in the proposed EBG structure with a U-shaped MLIN geometry, the absence of the resonator with a length that is avoids resonance at longer than one period of the structure frequencies lower than the center frequency of the stopband. In an EBG microstrip structure with bends, the difference between the phase delay of a bent unitary period and that of a straight one causes an increase in the passband ripple level. As was reported in [12], for the meander EBG microstrip structure with circles etched in the ground plane below a bend, the bending of a unitary period introduces a reduction in the phase delay, thus increasing the passband ripple level. A study on the bent section of the proposed U-shaped EBG structure in Fig. 1(c) is performed. Fig. 3 shows the phase delay of the bent unitary period (shadowed area) and that of the straight unitary period (meshed area). As indicated in Fig. 3, in the U-shaped EBG structure with a square patch inserted in the MLIN at a bend [see Fig. 1(c)], the phase delay of the bent unitary period is the same as that of the straight unitary period over a wide frequency range from dc to 6.76 GHz. Since the bending of the unitary period does not introduce any difference in phase delay, the proposed U-shaped EBG microstrip structure is able to maintain a low passband ripple level with enhanced stopband performance. Additional power loss caused by the radiation and scattering that take place at the bend is another important issue in an EBG microstrip structure with bends. In the meander EBG microstrip structure in [11] and [12], the radiation and scattering at the bend with an etched circle below introduce a significant amount of additional power loss, thus degrading the performance of the filter structure. The radiation and scattering in a meander microstrip structure can be estimated in terms of percentage using Radiation + Scattering,

(6)

The power loss of the U-shaped EBG microstrip structure in Fig. 1(c) is studied. Fig. 4 shows the simulated radiation and scattering level of the U-shaped EBG structure [see Fig. 1(c)] and that of the straight EBG structure [see Fig. 1(a)] for comparison. As can be seen in Fig. 4, the power loss of the U-shaped

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Fig. 4. Simulated radiation and scattering levels for the U-shaped EBG microstrip structure and the straight EBG structure.

EBG structure is similar to that shown by the straight one over a frequency range dc–9.42 GHz. This is mainly attributed to the small number of bends in the U-shaped structure. The similar radiation and scattering level is also due to the square patch inserted at the bend of the MLIN that eliminates additional power loss introduced at the bend. In single planar EBG (SP-EBG) structures, the proposed U-shaped EBG microstrip structure with patches periodically inserted in the MLIN is a compact structure with good filtering functionality.

Fig. 5. Schematic of the proposed high-performance compact U-shaped DP-EBG microstrip low-pass filter structure. (a) 3-D view. (b) Top view 1. (c) Top view 2.

TABLE I NORMALIZED COEFFICIENTS OF CHEBYSHEV ARRAYS

III. COMPACT U-SHAPED DP-EBG MICROSTRIP LOW-PASS FILTER A. Design By applying the U-shaped MLIN geometry analyzed above, a compact U-shaped DP-EBG microstrip structure is proposed. Fig. 5(a) shows the 3-D schematic of the proposed structure. A U-shaped EBG microstrip structure with square patches periodically inserted in the MLIN is on top of a perturbed ground plane with etched circles creating a dual planar configuration [14]. As indicated in Fig. 5(a), the substrate material has a dielectric constant of and a thickness of . Fig. 5(b) shows the relative location of the patches and the circles and Fig. 5(c) shows their dimensions. The width of the MLIN is set corresponding to a characteristics impedance of 50 and the square patches are inserted at a period of according to the Bragg reflection condition for a center frequency of . Circles are etched exactly below the MLIN with their centers allocated at the center point of two , which constructs anconsecutive square patches other SP-EBG structure [14] with a period of . The etched circles are for enhancing the contrast of the equivalent reactance in the EBG cell, thus improving the stopband performance of the structure. For the same purpose, as was reported in [15], the width of the MLIN above the etched circle is reduced so as to further increase the bandwidth and attenuation of the stopband, (where ). resulting in To eliminate ripples caused by the periodicity of the design, the Chebyshev distribution [16] is adopted to taper the area of

the etched circle and the inserted square patch. It results in the and and the corresponding length corresponding radius and . An optimal filling factor of 0.25 of the edge [6] is used to determine the radius of the central circle . The is set to avoid any edge length of the two central squares overlap between the circle in the ground plane and the square patch in the MLIN in order to obtain a sharp reactance contrast in an EBG cell for ensuring good filtering performance of the structure. The effect of the reduction in the width of has been studied and it was found that the MLIN is also tagood performance can be obtained when pered [15]. A three- and a four-element Chebyshev array with a major-to-minor ratio of 25 dB are used for the proposed design and the normalized coefficients are tabulated and shown in Table I. B. Numerical Results The proposed U-shaped DP-EBG microstrip structure is sim, mm) is used as the ulated. Taconic ( is set to substrate and the center frequency of the stopband 7.5 GHz. Table II shows the numerical values of the parameters set. Fig. 6 shows a comparison between the simulated -parameters of the proposed U-shaped DP-EBG structure and those of the U-shaped SP-EBG microstrip structure [see Fig. 1(c)].

HUANG AND LEE: COMPACT U-SHAPED DP-EBG MICROSTRIP LOW-PASS FILTER

TABLE II PARAMETERS OF THE PROPOSED LOW-PASS FILTER

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TABLE III SIMULATED PERFORMANCE AND CIRCUIT AREA EBG MICROSTRIP STRUCTURES

OF

U-SHAPED

Fig. 7. Fabricated compact U-shaped DP-EBG microstrip low-pass filter structure: MLIN (left), ground plane (right).

U-shaped DP-EBG structure exhibits a much lower sidelobe level than that of the U-shaped SP-EBG structure, which clearly indicates the smooth transmission in the passband of the novel dual planar design. The improvement of the stopband performance and the increase in selectivity of the proposed DP-EBG microstrip structure can mainly be attributed to the dual planar configuration where circles are etched in the ground plane. It can also be attributed to the reduction in the width of the MLIN above the etched circle, which increases the variation of equivalent reactance in an EBG cell. The significant elimination of the passband ripple in this dual planar structure is due to the tapering of all three components of the structure: the etched circles in the ground plane, the inserted square patches in the MLIN, and the (reduction in) width of the MLIN. Within a small circuit area, this proposed U-shaped DP-EBG microstrip structure shows excellent low-pass filtering functionality with superior passband and stopband characteristics. As compared to previous work [8], [10], the proposed design achieves significant improvement in the selectivity and a great reduction in both the physical size and passband insertion loss. Fig. 6. Simulated S -parameters for the proposed U-shaped DP-EBG microstrip low-pass filter structure and the U-shaped SP-EBG microstrip structure. (a) S . (b) S .

As can be seen from Fig. 6(a), the U-shaped DP-EBG structure shows a selectivity of approximately 22.50 dB/GHz, a passband ripple level of 0.20 dB, and an excellent stopband performance. There is a good stopband attenuation of more than 20 dB for frequencies of 3.4 GHz and above. Table III shows the performance and circuit area of the U-shaped DP-EBG structure and those of the U-shaped SP-EBG structure. As can be seen in Table III, within the same circuit area as that of the U-shaped SP-EBG structure as shown in Fig. 1(c), the proposed U-shaped DP-EBG structure achieves a significant increase in the bandwidth and attenuation of the stopband, a higher selectivity, and a -pamuch smaller passband ripple level. Fig. 6(b) shows the rameters of the two structures. As can be seen in Fig. 6(b), the

IV. MEASUREMENT RESULTS AND ANALYSIS The proposed compact U-shaped DP-EBG microstrip low-pass filter structure was fabricated and tested. Fig. 7 shows the MLIN and ground plane of the fabricated structure and Fig. 8 shows its measured and simulated -parameters. As shown in Fig. 7, the physical size of the proposed structure is small. EBG cells are allocated in both the MLIN (the inserted square patches) and ground plane (the etched circles). The input and output port are on the same side of the circuit, which may offer a greater degree of flexibility in the layout of microwave circuits designs. As can be seen in Fig. 8, the measured results show good agreement with the simulated results. The stopband of the measured result is not as smooth as that of the simulated result due to the additional connectors used for the testing of the structure and the nonideal fabrication and material. However, as shown

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in a compact physical size. The passband ripple level of the novel U-shaped DP-EBG structure is very low, which can be attributed to the unique MLIN geometry, as well as the tapering technique (Chebyshev distribution) by means of which the passband ripples caused by the periodicity in the structure are effectively eliminated. The proposed filter structure is easy to fabricate and shows compatibility with monolithic-microwave integrated-circuit (MMIC) technology. The novel design of this structure is able to achieve superior passband and stopband characteristics in a small circuit area. The structure was fabricated and tested for its performance. The good performance of the proposed structure is verified from the measured results, as they are in good agreement with the simulated results.

REFERENCES

Fig. 8. Simulated and measured S -parameters of the proposed U-shaped DP-EBG microstrip low-pass filter structure. (a) S . (b) S .

in Fig. 8(a), the fabricated structure still demonstrates a wide stopband with an attenuation of over 20 dB at the frequencies above 3.4 GHz. The passband ripple level shown in the measured result is as low as 0.33 dB, which is indicated by the low -parameter in Fig. 8(b). A sidelobe level of the measured good selectivity of 22.50 dB/GHz can also be observed for the measured result. The proposed structure is compact in size and is able to achieve excellent passband and stopband performance. This filter structure can easily be applied to circuit applications requiring low-pass filtering functionality. The unique U-shaped MLIN geometry provides additional flexibility for the allocation of components in circuit designs, thus potentially improving the compactness of circuits. V. CONCLUSION In this paper, the design and implementation of a novel high-performance compact U-shaped DP-EBG microstrip low-pass filter structure has been presented. Due to the U-shaped geometry of the MLIN and the dual planar arrangement of EBG structures, this proposed EBG filter structure obtains excellent stopband performance and a high selectivity

[1] E. Yablonovitch, “Inhibited spontaneous emission in solid-state physics and electronics,” Phys. Rev. Lett., vol. 58, no. 20, pp. 2059–2062, May 1987. [2] S. John, “Strong localization of photons in certain disordered dielectric superlattices,” Phys. Rev. Lett., vol. 58, no. 23, pp. 2486–2489, Jun. 1987. [3] E. R. Brown, C. D. Parker, and E. Yablonovitch, “Radiation properties of a planar antenna on a photonic-crystal substrate,” J. Opt. Soc. Amer. B, Opt. Phys., vol. 10, no. 2, pp. 404–407, Feb. 1993. [4] R. Gonzalo, P. De Maagt, and M. Sorolla, “Enhanced patch-antenna performance by suppressing surface waves using photonic-bandgap substrates,” IEEE Trans. Microw. Theory Tech., vol. 47, no. 11, pp. 2131–2138, Nov. 1999. [5] V. Radisic, Y. Qian, and T. Itoh, “Broad-band power amplifier using dielectric photonic bandgap structure,” IEEE Microw. Guided Wave Lett., vol. 8, no. 1, pp. 13–14, Jan. 1998. [6] V. Radisic, Y. Qian, R. Coccioli, and T. Itoh, “Novel 2-D photonic bandgap structure for microstrip lines,” IEEE Microw. Guided Wave Lett., vol. 8, no. 2, pp. 69–71, Feb. 1998. [7] F. Falcone, T. Lopetegi, and M. Sorolla, “1-D and 2-D photonic bandgap microstrip structures,” Microwave Opt. Technol. Lett., vol. 22, no. 6, pp. 411–412, Sep. 1999. [8] M. Bozzetti, A. D’Orazio, M. De Sario, V. Petruzzelli, F. Prudenzano, and F. Renna, “Tapered photonic bandgap microstrip lowpass filters: Design and realization,” Proc. Inst. Elect. Eng.–Microwaves, Antennas, Propag., vol. 150, no. 6, pp. 459–462, Dec. 2003. [9] F. Yang, K. Ma, Y. Qian, and T. Itoh, “A uniplanar compact photonicbandgap (UC-PBG) structure and its applications for microwave circuit,” IEEE Trans. Microw. Theory Tech., vol. 47, no. 8, pp. 1509–1514, Aug. 1999. [10] F. Yang, Y. Qian, and T. Itoh, “A novel uniplanar compact PBG structure for filter and mixer applications,” in IEEE MTT-S Int. Microwave Symp. Dig., vol. 3, Jun. 1999, pp. 912–922. [11] F. Falcone, T. Lopetegi, M. Irisarri, M. A. G. Laso, M. J. Erro, and M. Sorolla, “Compact photonic bandgap microstrip structures,” Microwave Opt. Technol. Lett., vol. 23, no. 4, pp. 233–236, Nov. 1999. [12] T. Lopetegi, M. A. G. Laso, M. Irisarri, M. J. Erro, F. Falcone, and M. Sorolla, “Optimization of compact photonic bandgap microstrip structures,” Microwave Opt. Technol. Lett., vol. 26, no. 4, pp. 211–216, Aug. 2000. [13] T. Akalin, M. A. G. Laso, E. Delos, T. Lopetegi, O. Vanbesien, M. Sorolla, and D. Lippens, “High performance double-sided microstrip PBG filter,” Microwave Opt. Technol. Lett., vol. 35, no. 2, pp. 90–93, Oct. 2002. [14] S. Y. Huang and Y. H. Lee, “Tapered dual-plane compact electromagnetic bandgap microstrip filter structure,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 9, pp. 2656–2664, Sep. 2005. [15] , “A tapered small-size EBG microstrip bandstop filter design with triple EBG structures,” Microwave Opt. Technol. Lett., vol. 46, no. 2, pp. 154–158, Jul. 2005. [16] N. C. Karmakar and M. N. Mollah, “Investigations into nonuniform photonic-bandgap microstripline low-pass filters,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 2, pp. 564–572, Feb. 2003. [17] D. M. Pozar, Microwave Engineering, 2nd ed. New York: Wiley, 1998, pp. 470–473.

HUANG AND LEE: COMPACT U-SHAPED DP-EBG MICROSTRIP LOW-PASS FILTER

Shao Ying Huang (S’05) received the B.Eng. degree from Nanyang Technological University, Singapore, in 2002, and is currently working toward the M.Eng. degree in electrical and electronic engineering at Nanyang Technological University. Her research interest includes the design, development and modeling of EBG structures and their applications.

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Yee Hui Lee (M’95) received the B.Eng. and M.Eng. degrees from the Nanyang Technological University, Singapore, in 1996 and 1998, respectively, and the Ph.D. degree from the University of York, York, U.K., in 2002. Since July 2002, she has been an Assistant Professor with the School of Electrical and Electronic Engineering, Nanyang Technological University, Singapore. Her interest is in evolutionary techniques, computational electromagnetics, and antenna design.

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Investigation of a Self-Calibrating SSB Modulator Daniel Marco Treyer, Member, IEEE, and Werner Bächtold, Fellow, IEEE

Abstract—A method to adaptively calibrate a Hartley phasingtype single-sideband (SSB) modulator for obtaining high carrier and image suppressions is presented. The Hartley SSB modulator is extended by control elements to compensate for circuit nonidealities, a monitor circuit to measure the levels of the undesired residual carrier and image tones, and a digital controller. A direct search algorithm adjusts the control elements to maximize modulator performance. The effectiveness of the technique has been studied on an experimental SSB modulator. Carrier and image suppression ratios of 60 dB have been achieved over a carrier range from 1.40 to 2.35 GHz, a bandwidth of 51%. Index Terms—Calibration, carrier suppression, frequency shifters, image suppression, microwave phase shifters, single-sideband (SSB) mixers, SSB modulators.

I. INTRODUCTION

H

ARTLEY and Weaver phasing-type single-sideband (SSB) modulators are frequently applied in transmitters, frequency synthesizers, and measurement systems. Especially for millimeter-wave transmitters, where the ratio of local oscillator (LO) frequency to the intermediate frequency (IF) can be very high and filtering of spurious signals accordingly difficult, the SSB modulator is a very attractive candidate. Another important application for SSB modulators are frequency shifters. A Hartley phasing-type SSB modulator can be regarded as a vector modulator with the IF inputs fed in quadrature. For perfect image suppression, the transmission frequency responses of the in-phase (I) and quadrature (Q) paths must be identical, and the LO and IF power splitters must provide exact 90 phase shifts. Furthermore, to achieve low carrier leakage, the individual mixers must be perfectly balanced. In practice, it is very difficult to maintain all these requirements to a degree that produces acceptable modulator performance. At millimeter-wave frequencies, the situation becomes even worse due to fabrication tolerances on hybrid microwave integrated circuits (MICs), or inevitable couplings between signal lines on monolithic microwave integrated circuits (MMICs). Current SSB modulator designs rely on well-matched I and Q mixers and complex 90 phase-shift circuits to achieve good performance [1]. Factory calibration by amplitude and phase tuning, as suggested in [2] and [3], has the disadvantage that temperature and frequency changes cannot be compensated for Manuscript received May 4, 2005. D. M. Treyer is with the Laboratory for Electromagnetic Fields and Microwave Electronics, Department of Information Technology and Electrical Engineering, Swiss Federal Institute of Technology, CH-8092 Zurich, Switzerland (e-mail: [email protected]). W. Bächtold, retired, was with the Laboratory for Electromagnetic Fields and Microwave Electronics, Department of Information Technology and Electrical Engineering, Swiss Federal Institute of Technology, CH-8092 Zurich, Switzerland. Digital Object Identifier 10.1109/TMTT.2005.859876

Fig. 1. Hartley phasing-type SSB modulator. The arrows indicate signal phasors.

later on during operation. Methods such as automatic frequency tuning of polyphase filters [4] or the Havens’ amplitude and phase corrector circuit [5] only improve the performance of the 90 phase shifters. Techniques for automatic adjustment of quadrature modulators proposed by Faulkner et al. [6] and Huang and Caron [7] are able to correct amplitude and phase errors, as well as LO leakage, but require a digital signal processor (DSP) and special training signals for calibration. The approach presented in this paper employs uncritical analog radio frequency (RF) signal processing, as introduced in [8] and [9], instead of a DSP. The phasing-type SSB modulator is extended by variable attenuators and phase shifters to correct for amplitude and phase imbalances, and by variable direct current (dc) sources at the mixer IF ports to eliminate carrier leakage. An additional third mixer produces two monitor signals, which are related to the levels of the undesired carrier and image tones at the RF output. A controller automatically adjusts the control elements to maximize carrier and image suppressions. The calibration process neither requires a spectrum analyzer, nor any special training signals. Thus, it can work while the modulator is in normal operation, continuously compensating for temperature and frequency changes. This paper proceeds in Section II with a short review of the Hartley SSB modulator. Section III describes the self-calibrating SSB modulator topology investigated, and analyzes the achievable carrier and image suppressions. The calibration algorithm is outlined in Section IV. A practical test system is illustrated in Section V. Circuits of the variable attenuator, variable phase shifter, and monitor mixer are introduced and illustrated with measured performance data. In Section VI, test results of the realized self-calibrating SSB modulator are presented. Sensitivities of carrier and image suppressions on

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Fig. 2. Extended Hartley SSB modulator topology for self-calibration. (a) Block diagram of the modulator. Quadrature errors are corrected by the variable attenuators and phase shifters. Carrier leakage is canceled by injecting appropriate dc offsets into the mixer IF ports. The monitor circuit detects the levels of the undesired image and carrier components at the RF output. (b) Block diagram of the monitor circuit.

operating parameters and control element settings are discussed. Section VII lists possible circuit improvements, and Section VIII gives a comparison to previously published works. Section IX describes a potential application.

by 40 dB relative to the desired sideband, the amplitude and phase accuracies must be better than 1.4% ( 0.12 dB) and 0.8 , respectively. III. SELF-CALIBRATING SSB MODULATOR

II. HARTLEY SSB MODULATOR

A. Extension of the Hartley SSB Modulator

The principle of the Hartley phasing-type SSB modulator is shown in Fig. 1. At each mixer, the phase relations of the LO, IF, RF upper sideband (USB), and RF lower sideband (LSB) signals are equal to the corresponding frequency relations, i.e., the phase of the USB output is the sum of the LO and IF phases, whereas the phase of the LSB output is the difference between the LO and IF phases. Assuming the LO and IF phases in the I path are 0 , the I path USB and LSB phases are also 0 . If the LO and IF phases in the Q path are 90 , the Q path USB phase is 180 (sum) and the LSB phase is 0 (difference). Given that the I and Q IF signals have equal amplitudes, the I and Q USB and LSB components also have equal amplitudes. Thus, when summing the mixer output signals in the RF combiner, the USB components cancel, while the LSB components add up. Production of the USB and cancellation of the LSB would require to alter the phase relations accordingly. This can be done by interchanging the outputs of the IF or LO 90 splitter, or by replacing the RF 0 combiner by a 180 combiner. To achieve complete suppression of the undesired sideband (image), the LO and IF signals in the I and Q paths must be in perfect quadrature, and the frequency responses of the I and Q mixers and RF paths must be identical. In practice, the image suppression is limited by amplitude and phase imbalances. Deas the relative gain mismatch, and as the total noting phase mismatch (deviation from 90 in radians), the image suppression ratio is

(1) This equation is equal to that of the image-rejection ratio (IRR) for an image-rejection mixer (IRM) [10]. To suppress the image

A block diagram of the extended topology is given in Fig. 2(a). The original Hartley SSB modulator consists of the shaded components only. The purpose of the added components is described below. 1) Carrier or LO Cancellation: The SSB modulator is operated as a vector modulator by injecting two independent dc and into the mixer IF ports through bias tees. signals At the RF output, this results in a linear combination of the I and Q LO signals. If this vector is made equal in magnitude as the inherent carrier leakage, but opposite in phase, the LO tone at the output will be completely canceled. 2) Image Suppression: The amplitude and phase imbalances are corrected by electronically variable attenuators and phase shifters at the RF and LO ports of the I and Q mixers, respectively. Using the same elements in both the I and Q paths ensures that their nominal frequency responses are matched. Complementary control of the same elements produces equal or phase in both paths, changes of attenuation but with opposite signs. 3) Monitor Circuit: A sample of the modulator’s output signal is amplified and fed to a diode where it mixes with itself. The desired output tone acts as the pumping signal. From the resulting mixing products, the frequency components at once and twice the IF frequency are separated and fed to level detectors. See Fig. 2(b) for a block diagram, and Fig. 3 for the (M1) results corresponding spectra. The component at from mixing of the desired signal with the carrier leakage, and from mixing of the carrier leakage with the image tone. Since the carrier and image tones are substantially weaker than the desired signal, the magnitude of M1 is mainly proportional to (M2) the level of the carrier leakage. The component at originates from mixing of the desired signal with the image

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In the worst case, the maximum IF dc power that must be applied to either one of the mixers is dB

(2)

where dBm is the LO level at the mixers, dB is the mixer LO-to-RF isolation, dB is the mixer conversion loss, and 6 dB is a factor to account for the modulator topology. The required accuracy of IF dc power is dB

Fig. 3. Input and output spectra of the monitor mixer in the case of USB suppression.

tone. Hence, the magnitude of M2 is proportional to the level of the residual image tone. 4) Controller: The detectors in the monitor circuit provide the levels of the monitor signals M1 (proportional to the carrier leakage) and M2 (proportional to the image tone) to a microcontroller. Using a direct search algorithm, the controller iteratively deduces the required settings of the variable attenuators, phase shifters, and IF dc-offset sources to minimize the carrier and image tones at the RF output. B. Location of the Control Elements In principle the variable attenuators and phase shifters can be placed anywhere in the signal path from IF to RF. However, the nonideal behavior of practical circuits must be considered. 1) Variable Attenuator: Due to their resistive nature, absorptive attenuators are very broad-band devices. Nevertheless, it has been chosen to put the variable attenuators not into the IF paths, but into the RF paths because signal level and relative signal bandwidth are lowest there, reducing nonlinear and dispersion effects. 2) Variable Phase Shifter: Most phase shifters have significant phase dispersion and, therefore, are not suited for signals with large instantaneous bandwidths. Thus, the phase shifter was placed in the LO paths, where the instantaneous bandwidth is zero. The AM rejection property of balanced mixers weakens the effect of the phase shifter’s parasitic attenuation variation. C. Limitations The maximum amplitude and phase imbalances that can be compensated depend on the attenuation and phase-shift ranges of the control elements. The maximum attainable carrier and image suppressions depend on the resolution of the digital-to-analog converters (DACs) that produce the control voltages for the control elements, linearity of the mixers, sensitivity of the monitor circuit, and resolution of the analog-to-digital converters (ADCs) that sample the monitor signals. Crosstalk among circuit blocks may exist, but if it is time varying it is permitted to change only slowly. An IF signal must be present and its bandwidth must not exceed one octave. D. Accuracy of Control Elements For the mixer IF dc-offset sources, range and resolution are dictated by the LO-to-RF isolation of the mixers and the maximum permissible LO leakage at the SSB modulator RF output.

(3)

where dBm is the IF signal level at the mixers, [dB] is the desired carrier suppression ratio, and 3 dB is a factor to account for the modulator topology. IF dc power can be converted to current through the following relation: A

mW

(4)

The parameters of the SSB modulator discussed later in SecdBm, dB, dB, tion V are dBm, and dB. The required IF dc-offset current ranges become 502 A, and the accuracy is 0.36 A. Using bit

(5)

the DACs for the mixer IF dc-offset sources must, therefore, have a resolution of at least 11 bits. For the variable attenuators and phase shifters, adjustment range and step size depend on the expected maximum amplitude and phase mismatches, respectively, and on the required image suppression. For the SSB modulator investigated here, imbalances of 1.5 dB and 20 have been estimated. To obtain an image suppression of 60 dB, amplitude and phase errors must not exceed 0.012 dB and 0.08 , according to (1). The corresponding DACs must, therefore, have resolutions of at least 7 and 8 bits, respectively. IV. CALIBRATION ALGORITHM The operation of the calibration algorithm is outlined in Fig. 4. The controller alternately attempts to minimize the levels of the residual carrier and image tones. If no improvement is possible anymore, it observes the carrier and image suppressions. When the LO frequency is changed or drift is detected, the carrier and image minimizations are resumed. Iterative calibration is necessary because of the following two effects. 1) Attenuator and Phase Shifter Affect LO Leakage: While adjusting the variable attenuators and phase shifters to minimize the image, LO suppression will degrade. This is because LO leakage cancellation occurs at the RF combiner, which is located after the attenuators and phase shifters. Measured sensitivities will be given in Section VI-C. 2) Limited Selectivity of Monitor Circuit: The selectivity of the filter separating the M1 and M2 signals is limited, as will be shown in Section V-E. Thus, if the LO leakage is much stronger

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TABLE I MICROWAVE COMPONENTS USED FOR THE SSB MODULATOR TEST SETUP

Fig. 4.

Self-calibration algorithm.

points and creates new points in between them. The search interval narrows at each step. Convergence is detected if the control parameter values or the responses of a certain number of contiguous points fall within specified error bounds. If the search gets stuck at the lower or upper limit of the search interval, or when resuming optimization after other parameters have been adjusted, the search interval width is doubled. Fig. 5. Optimization using a direct-search method on a five-point simplex. At each step, the optimum point and its two neighboring points are kept, and two new points are created in between to have five points again.

V. TEST SETUP

than the image or vice versa, the monitor circuit will give erroneous readings for the weaker signal. In the test setup that will be described in Section V, it was observed that: 1) for the uncalibrated SSB modulator, the LO leakage dominates over the image tone and 2) that temperature drift affects LO leakage far more than image suppression. For these reasons, the algorithm always tries to minimize the LO leakage before the image. The carrier is minimized by alternately varying the I and Q mixer IF dc offsets, until the M1 signal level measured by the monitor circuit cannot be further reduced. Likewise, the image is minimized by alternately varying the variable attenuator and phase-shifter settings, until the M2 signal of the monitor circuit is as low as possible. The actual optimizations of the LO and image suppressions by adjusting the I and Q mixer IF dc offsets and the variable attenuators and phase shifters are based on a direct-search algorithm working on one-dimensional simplexes. The principle is illustrated in Fig. 5. When optimizing the LO suppression, or , and the monthe control parameter used is either . When optimizing the image supitor response observed is or , and the pression, the control parameter is either . response is The algorithm first places a number of points across a starting search interval of the control parameter to be optimized, and measures the response at each point. It then identifies the best

A prototype self-calibrating SSB modulator according to the block diagram in Fig. 2 has been built to study the concept. The test setup nominally works at an LO of 1.8 GHz and an IF of 40 MHz. Table I lists the commercial microwave components employed. Additional components used in the test setup that are not shown in the block diagram for simplicity are: 1) 3-dB attenuators at the mixer LO and RF ports to enhance return losses; 2) a low-pass filter at the IF input to reject harmonics of the IF source; 3) high-pass filters after the RF combiner to prevent IF leakage from entering the monitor detectors; 4) low-pass filters before the RF coupler to suppress the LO second harmonic; and 5) dc amplifiers at the monitor detector outputs. The controller is a PIC16F876 from Microchip Technology Inc., Chandler, AZ. The control signals for the attenuators, phase shifters, and IF dc-offset sources are produced by 12-bit DACs. The monitor signals are digitized by 10-bit ADCs. Modulator performance is sensitive to interference picked up by the control voltage lines and IF dc-offset lines, e.g., 50-Hz stray fields from power supply transformers. To avoid performance degradation, the control DACs have been mounted in the same shielded packages as the corresponding attenuators, phase shifters, and current sources. The suppression of harmonics in the IF input signal must be of the same order as the desired image and LO suppression ratios. To obtain independent compensation of amplitude and phase imbalances, the parasitic phase variation of the attenuator and

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Fig. 6. Electronically variable attenuator using p-i-n diodes in a pi configuration.

Fig. 8. Electronically variable phase shifter using varactor tuned resonant shunt loads.

Fig. 7. Measured transmission frequency response of the electronically variable attenuator. The control voltage V is varied from 1.2 to 12 V. The effect of the feed lines (linear phase term) has been subtracted from the phase response.

Fig. 9. Measured transmission frequency response of the electronically variable phase shifter. The control voltage V is set to 0, 1, 2, 3, 4, 5, 6, 8, and 12 V. The effect of the feed lines (linear phase term) has been subtracted from the phase response.

the parasitic amplitude variation of the phase shifter were minimized by the selection of appropriate circuit topologies. Circuit details and measurement data of the custom made variable attenuator, variable phase shifter, and monitor mixer will be discussed below. A. Variable Attenuator Broad-band pi-type resistive attenuator pads using four p-i-n diodes are employed, as shown in Fig. 6. The measured transmission characteristics are shown in Fig. 7 for different control voltage values. The measured return loss is higher than 15 dB for frequencies from 0.5 to 2.6 GHz. The attenuation control characteristic is highly nonlinear and saturates for control voltages above 10 V. In the SSB modulator, only small amplitude corrections are necessary, hence, the control voltages applied will be in the range from 7.5 to 10 V, corresponding to an attenuation range from 2.5 to 4.0 dB, and a parasitic phase shift of 1 dB.

Fig. 10. Measured (a) transmission magnitude ratio of the attenuators and (b) transmission phase difference of the phase shifters for DAC values from 2048 to 2047 in steps of 512. Complementary control of same elements is applied.

0

phase shift range is 20 and the parasitic attenuation variation is less than 0.01 dB . Phase dispersion becomes severe below 1.6 GHz and above 2.0 GHz.

B. Variable Phase Shifter Loaded-line phase shifters consisting of two resonant shunt loads spaced by a quarter-wave transmission line, as depicted in Fig. 8, are used. Quarter-wave spacing of two identical shunt loads partially cancels reflections, improving port return losses. The measured transmission characteristics are shown in Fig. 9 for different control voltage values. The measured return loss is higher than 12 dB for frequencies from 1.2 to 2.2 GHz. The phase shift control characteristic is nonlinear. At 1.8 GHz, the

C. Control Voltages and IF dc-Offset Sources 1) Complementary Attenuator Control: The output voltage of a single DAC is converted to a control voltage for the attenuator in the I path that runs from 7.5 to 10 V as the DAC input word is changed from 2048 to 2047 (12 bits), and a complementary control voltage for the attenuator in the Q path that runs from 10 to 7.5 V. The net effect on the transmission magnitude ratio of the RF paths is shown in Fig. 10(a).

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Fig. 11. Monitor mixer circuit with output amplifier. Port M denotes the monitor IF signal.

2) Complementary Phase-Shifter Control: The output voltage of a single DAC is converted to a control voltage for the phase shifter in the I path that runs from 1 to 10 V as the DAC input word is changed from 2048 to 2047, and a complementary control voltage for the attenuator in the Q path that runs from 10 to 1 V. The net effect on the transmission phase difference of the LO paths is shown in Fig. 10(b). 3) IF dc-Offset Sources: Independent 12-bit DACs are used to generate the IF dc offsets for the I and Q mixers. The current range of the IF dc-offset sources is from 480 to 480 A, and the sensitivity to the smallest change of the DAC word (LSB) is A LSB. The control characteristics of the individual attenuators and phase shifters are nonlinear. However, by careful selection of the nominal control voltage values (at DAC word 0), the net effect of the control elements on the I/Q system has been made to relate almost linearly to the corresponding DAC words, as can be seen from the nearly equally spaced curves in Fig. 10. At 1800 MHz, the sensitivity of the transmission magnitude ratio mdB LSB, and that of the of the attenuators is transmission magnitude phase difference of the phase shifters is m LSB.

Fig. 12. Measured characteristics of the monitor mixer. The RF test signal consists of the two components RF1, which pumps the mixer, and RF2, which is the signal to be downconverted to IF. In the large-signal graph, the desired IF is the tone at 40 MHz, while the second-order intermodulation product is at 80 MHz.

D. Monitor Mixer The circuit diagram of the monitor mixer with output amplifier is given in Fig. 11. The mixer consists of a single Schottky diode embedded in a filter and impedance matching network. The short-circuited quarter-wave stub on the cathode side provides the IF return path. The LC network on the anode side acts as an RF return, IF bandpass filter, and impedance transformer. Measured characteristics of the constructed monitor mixer are illustrated in Fig. 12. At an RF pump signal of 1800 MHz at 0 dBm, and an RF mixing signal of 1760 MHz, the monitor mixer together with the monitor IF amplifier has a measured conversion gain of 16.5 dB, a noise figure of roughly 5 dB, and an input return loss of better than 10 dB. When the level of the RF pump signal is varied from 5 to 5 dBm, the relative conversion factor change is 3 to 1 dB. The measured 3-dB RF bandwidth is from 0.9 to 3.1 GHz, and the 3-dB IF bandwidth is from 20 to 150 MHz. The input referred second order intercept point is at 27 dBm. Note that the 2 RF–2 LO product, which is second order provided the RF level is constant, can degrade the measurement of the image suppression because it falls onto the M2 signal.

Fig. 13.

Monitor detector circuit with diplexer.

To prevent disturbance of the monitor detectors by the very strong RF desired signal, the isolation of the monitor detectors from the monitor mixer RF port must be of the same order as the carrier and image suppression ratios aspired. The monitor mixer RF-to-IF isolation and the upper stopband attenuation of the detector bandpass filters must be adequately high. E. Monitor Detectors The purpose of the monitor detector circuit is to separate the M1 and M2 components in the output signal of the monitor mixer, and to measure their levels (see Fig. 3). The circuit diagram is shown in Fig. 13. The LC diplexer filters are tuned for 40 and 80 MHz, respectively. The two filters consist of single LC tanks with

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Fig. 14. Measured characteristics of the monitor detector circuit with diplexer. (a) Amplitude response. (b) Frequency response.

tapped input elements for impedance matching. If the inductor is tapped, the resulting filter has a lower passband edge slope of 10 dB/decade and an upper passband edge slope of 20 dB/decade, and vice versa if the capacitor is tapped. Thus, good selectivity is obtained when using inductive tapping for the 40-MHz filter and capacitive tapping for the 80-MHz filter. Further, the interaction of the two filters introduces a transmission zero in each filter, located near the center frequency of the other filter. The detectors must have good sensitivity and high dynamic range. This is achieved by the tuned input circuits and by using zero-bias Schottky detectors. The detector output voltage is compressed by a simple diode limiter, saving ADC resolution while still maintaining monotonicity and high dynamic range. Absolute level accuracy and linearity are not important. The junction resistance of the diodes increases as the applied power decreases. This leads to a longer detector response time for weak monitor IF signals, which must be kept in mind when implementing the calibration algorithm. The measured characteristics of the monitor detectors are shown in Fig. 14. The limiter has no effect for signals below 20 dBm. At 0-dBm input power, the detectors alone would produce almost 2 V at the output. The limiter compresses the output voltage by a factor of approximately four, thus saving two bits of ADC resolution. Both detectors have a low-level sensitivity of 0.4 mV W at their center frequency, and a noise bandwidth of roughly 5 MHz. The detector output voltages are amplified to match the input range of the ADCs. To maintain high monitor sensitivity, negative input offset voltages at these dc amplifiers must be avoided. VI. TEST RESULTS The tests discussed below are done with the SSB modulator configured to produce the USB and cancel the LSB. However, nearly equal performance is achieved when configuring the mixer for LSB generation and USB cancellation by reversing the outputs of the IF 90 coupler. Nominal operating parameters MHz, dBm, MHz, and are dBm. A. Convergence of the Self-Calibration Algorithm In the uncalibrated state, i.e., when the attenuators and phase shifters are set to equal values in the I and Q paths and the IF dc

Fig. 15. RF output spectrum before and after calibration. Before calibration, carrier and image suppressions are 20 and 30 dBc, respectively. Calibration improves them by 47 and 42 dB, respectively. The interference around 1845 MHz originates from a DCS1800 system.

0

0

Fig. 16. Exemplary evolution of residual carrier and image levels during calibration of the SSB modulator. Each adjustment step takes approximately 1 ms.

offsets are zero, the levels of the carrier and image tones in the RF output signal are 20 and 30 dBc, respectively (relative to the desired tone). After the automatic calibration is enabled, the carrier and image tones gradually decrease until they remain constant below 60 dBc. The calibration takes approximately 100 ms. Fig. 15 shows the RF output spectra before and after calibration. Fig. 16 depicts the convergence of the carrier and image levels as the algorithm proceeds. Minimizations of the carrier and image tones are not independent. The origin of the intermodulation product at 1880 MHz will be discussed in Section VI-E. The convergence of CRR and IRR exhibit long periods where no improvement is achieved, seen as parallel horizontal segments in Fig. 16. The algorithm could be accelerated by an improved convergence detection method to switch faster to the next control parameter. B. Operating Bandwidths The LO and image suppressions achievable have been measured over a broad range of LO and IF frequencies. At each frequency setting, the modulator has been recalibrated. 1) Variation of LO Frequency: Measurement results obtained from an LO frequency sweep are shown in Fig. 17. Even though the test setup was not optimized for broad-band

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Fig. 18. Measured monitor detector readings when sweeping the attenuator DAC setting. Phase shifter and mixer IF dc offsets are at optimum values. The thin curve in the zoomed graph on the right-hand side is a parabola fit to M2.

TABLE II MEASURED TOLERANCE INTERVALS OF DAC SETTINGS SPECIFIED LO AND IMAGE SUPPRESSIONS

TO

ACHIEVE

Fig. 17. SSB modulator output tones versus LO frequency. The modulator has been recalibrated at each LO frequency.

operation, the LO and image suppressions are at least 60 dB over an LO range from 1400 to 2350 MHz, corresponding to a relative bandwidth of 51%. Over this LO range, the total variations of the attenuator, phase shifter, and IF dc-offset source DAC values are 1250, 350, and 300, respectively. These translate to control parameter variations of 1.1 dB, 2.9 , and 70 A, respectively. It was found that when reducing the LO frequency below 1400 MHz, the monitor circuit is deceived by the second harmonic components 2 LO and 2 USB in the RF spectrum. A mixing product from these tones falls onto the M2 signal, thereby corrupting the measurement of residual image level. Extension of the lower LO frequency limit could be achieved by using an RF low-pass filter having a lower cutoff frequency, and a steeper passband to stopband transition. However, the first measure compromises the upper LO frequency limit, while the second asks for a complex RF filter. Another motivation for good second harmonic suppression is to avoid the 2 LO–USB intermodulation, which falls onto the LSB frequency. The 2 LO components generated by the mixers were found to be much stronger than their fundamental LO leakage. The upper usable LO frequency is limited by the RF low-pass filter, mixers, and phase shifters, which all have considerable insertion losses above 2.2 GHz. 2) Variation of IF Frequency: The calibration achieves CRR and IRR values of more than 50 dBc for IF frequencies from 30 to 50 MHz. The IF range is mainly limited by the selectivity of the monitor detectors.

C. Sensitivity to Control Element Settings The sensitivities of LO and image suppressions on the control element settings are important to know when choosing the ADC and DAC resolutions and when designing the calibration algorithm. Before calibration, the LO and image suppression ratios

may not have distinct optimum points with respect to the modulator control parameters. After calibration, changing any parameter will noticeably affect carrier and image suppressions, as illustrated in Fig. 18, for the attenuator DAC setting. A measure for these sensitivities is the width of the intervals within which the control parameters must remain to achieve specified carrier and image suppressions. Such tolerance intervals have been measured and are tabulated in Table II. It can be concluded that for suppression ratios of 60 dBc, the DAC resolutions of the mixer IF dc-offset sources are just sufficient (12 bit). However, if a tolerance interval of 1 LSB is considered adequate, the DAC resolutions of the variable attenuators and phase shifters may be reduced by up to 4 bits, yielding 8-bit resolutions. These findings are consistent with the accuracy predictions made in Section III-D. If the desired suppression ratios are reduced by 10 dB, the required DAC accuracies decrease by a factor of 3. The tolerance intervals DAC can be converted to current, attenuation, and phase shift by multiplication with the corresponding sensitivi, and given in Section V-C. ties , The graphs of the monitor detector ADC readings of M1 and M2 with respect to the attenuator DAC setting have a parabolic shape near the optimum, as seen in Fig. 18. This can be exploited by the calibration algorithm. The optimum attenuator setting lies at the apex of a parabola that is fit to measured points of the characteristic. This approach also works M2 level versus for optimally adjusting the phase shifter and the IF dc-offset sources. It greatly helps to circumvent convergence problems arising from detector and quantization noise. D. Sensitivity to Operating Parameters Here, we look at the effects of varying the modulator operating parameters after an initial calibration has been performed.

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fails because the signal that pumps the monitor mixer becomes too weak, actually reducing the monitor sensitivity. E. Intermodulation Products

Fig. 19. Effects of operating parameters on LO and image suppressions. Thick lines: the modulator has been calibrated at an LO signal of 1800 MHz and 15.5 dBm, and an IF signal of 40 MHz and 20 dBm. It is not recalibrated while sweeping the operating parameters. Thin lines: the modulator is recalibrated continuously during the sweep.

0

Fig. 20. Sweep of IF signal level. Left: With foregoing calibration at P = 20 dBm. Right: With continuous calibration. The spectrum analyzer noise floor is at 88 dBm.

0

0

Figs. 19 and 20 shown the measured LO and image suppressions when varying the LO frequency, LO level, IF frequency, temperature, and IF level. 1) Without Recalibration: The measurement results show that LO frequency, LO level, and temperature all greatly influence the LO suppression. The IF frequency only affects the image suppression. The instantaneous IF range for an IRR of better than 40 dB spans from 38.0 to 42.5 MHz. The IF level has no effect on the LO and image suppressions, as long as it remains below 10 dBm. 2) With Continuous Calibration: When leaving the calibration active while the operating parameters change, LO and image suppressions of better than 60 dB are maintained over wide operating parameter ranges. The major limitations come from the bandwidth and sensitivity of the monitor circuit. For IF frequencies below 32 or above 47 MHz, the bandwidth limits of the monitor detectors inhibit detection of residual LO and image components. For IF levels below 30 dBm, calibration

When the SSB modulator is configured for USB generation, an undesired intermodulation tone occurs at the frequency . In the following, this component will be designated UIM for the USB intermodulation product. Measurements indicated that the UIM grows by 2 dB per decibel increase of IF level, and that it is particularly strong when the modulator is not calibrated. It has been found that the UIM predominantly is the 2 USB–LO intermodulation product generated in the variable attenuators and RF amplifier. Calibration will minimize the residual LO and, hence, indirectly reduce the UIM, as can be seen from the output spectra shown in Fig. 15, and also from Fig. 20. However, minimizing the LO leakage only reduces the UIM originating from the RF amplifier, but not that originating from the variable attenuators, as cancellation of the LO leakage occurs after the attenuators. When setting up the modulator for producing the LSB, the corresponding tone is at the frequency (LIM). VII. IMPROVEMENTS The test results described above were obtained without optimizing the SSB modulator to meet any specific performance goals. However, practical applications may benefit from the following modifications. 1) IF Bandwidth: Maintaining good image suppression over a wider instantaneous IF bandwidth could be achieved by using a more precise IF quadrature network. A polyphase filter would be a viable candidate. The frequency responses of the monitor detectors must be matched to the IF range. 2) LO Range: The lower end of the LO frequency range could be extended by optimizing the RF low-pass filter, as described in Section VI-B.1. 3) DAC Resolutions: For the modulator investigated, the DAC full-scale ranges of all control voltages can be decreased to match the actual maximum amplitude and phase imbalances, and mixer LO leakages. Also, the step size for amplitude and phase adjustments can be increased (see Section VI-C). 4) Variable Attenuator: Although the attenuator’s parasitic phase shift and delay are small at low attenuation settings, they may still cause significant degradation of image suppression when using the modulator as an upconverter for modulated IF signals. An attenuator with reduced parasitic phase variation, as described in [12] or [13], could give improvement. 5) LO and Image Suppressions: If higher LO and image suppressions were desired, the linearity of the variable attenuators and RF amplifiers would have to be improved to limit the UIM or LIM intermodulation products. Also, more precise adjustment of the control parameters and higher monitor sensitivity would be necessary. VIII. COMPARISON TO PREVIOUSLY PUBLISHED WORKS A comparison of published Hartley phasing-type SSB modulators is given in Table III. It lists reported modulator performances, as well as the methods that were employed to increase

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TABLE III PUBLISHED HARTLEY PHASING-TYPE SSB MODULATORS AND THEIR PERFORMANCE

the carrier and image suppressions. In the IF bandwidth column, the entry “spot” means that the modulator has been tested at a single IF frequency only. For [6], the IRR was calculated from measured quadrature errors. Pros and cons of the investigated self-calibrating Hartley SSB modulator include the following: automatic calibration of LO and image suppression; continuously tracks frequency and temperature changes; no special training signal needed; no spectrum analyzer needed; use of analog signal processing instead of a DSP; calibration takes time, and must be updated continuously; modulation bandwidth must not exceed one octave; modulation signal (IF) is needed during the calibration.

IX. APPLICATION A possible application for a self-calibrating SSB modulator in a smart antenna system is shown in Fig. 21. For an array of low-IF receivers that employ tunable IRMs, an SSB modulator generates pilot tones to facilitate tuning of image rejections and calibration of downconversion factors. Generating the pilot tones by an SSB modulator has the advantage that they can be derived from the same LO as is used for the receive mixers. The SSB modulator can serve the following three purposes. 1) Tuning of Image Rejections: Switch SW1 is set to CAL, and the SSB modulator is configured to produce a tone at the downconverter image frequency. Each IRM is then tuned for best image rejection by adjusting its amplitude and phase balance to minimize its IF output signal level. An example of an electronically tunable IRM has been reported in [14]. 2) Calibration of Downconversion Factors: Switch SW1 is set to CAL, and the SSB modulator is configured to produce a tone at the desired receive frequency. By first injecting the calibration signal from the left-hand side across the calibration signal line (switch SW2 set to position A), and in a second step from the right-hand side (switch SW2 set to position B), the complex downconversion factors of all receive channels can be found using the technique described in [15].

Fig. 21. Tuning and calibration of IRMs in a receiver array. Pilot tones are generated by a self-calibrating SSB modulator, which can also serve as a transmit mixer. The required controller is not shown.

3) Transmit Operation: When setting switch SW1 to TX, the SSB modulator can serve as a low-IF upconverter for signal transmission. Full-duplex operation is possible if the transmit and receive antennas are sufficiently isolated (to prevent selfblocking), and if the transmit and receive radio frequencies are unequal. The latter can be achieved by either using unequal transmit and receive IFs or by employing different sidebands. The pilot tones are distributed to the receiver inputs via a dedicated transmission-line network, and directional couplers or capacitive taps. An alternative pilot tone distribution method would employ the parasitic coupling between transmit and receive antennas. However, this would only allow calibration of the image rejections, but not the downconversion factors. If the mixer circuit used for the receive IRMs is reciprocal with respect to the RF and IF ports, and has some LO suppression when used as an upconverter, the same circuit type could be employed for both the SSB modulator and downconverters. The use of optimally tuned IRMs allows to omit the otherwise indispensable receiver RF image filters and, hence, enables the use of a very low IF, resulting in a considerable reduction of circuit complexity. With the calibration technique described, the

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achievable image rejections are limited by the image suppression of the SSB modulator, e.g., to at least 60 dB when using the modulator investigated in this paper. X. CONCLUSION A technique to automatically calibrate the carrier and image suppressions in a Hartley phasing-type SSB modulator has been presented and demonstrated. The technique uses uncritical analog RF signal-processing blocks and a low-cost controller to automatically compensate for circuit quadrature errors and carrier leakage. A practical self-calibrating SSB modulator that operates at an RF around 1.8 GHz and an IF around 40 MHz has been constructed to investigate the effectiveness of the principle. Unprecedented carrier and image suppression ratios in excess of 60 dB have been achieved reproducibly over wide ranges of LO and IF frequencies and temperature. The technique allows automatic calibration during normal operation of the SSB modulator, thereby continuously tracking any operating parameter changes. Investigations have been made for single-tone IF signals only. The technique is expected to also work for modulated IF signals, provided that the bandwidths of the frequency-selective detectors in the monitor circuit are wide enough. The components that enable calibration of the SSB modulator significantly contribute to the overall circuit complexity and power consumption. However, this is not of concern in systems where high suppression of spurious signals is mandatory such as test equipment or high-power transmitters. Possible applications of the self-calibrating SSB modulator include: 1) frequency shifter for homodyne measurement systems; 2) upconverter in communication transmitters; and 3) pilot tone generator for calibration purposes. The technique is highly suited for monolithic integration. It also enables the use of SSB modulators in low-IF single-conversion millimeter-wave transmitters. Using a similar calibration approach in a Weaver phasing-type SSB modulator may also be considered, as well as adaptation of the concept to optical SSB generation. REFERENCES [1] J. D. K. West, “The application of the asymmetric polyphase filter in an SSB transceiver,” in Proc. South African Communications and Signal Processing Symp., Pretoria, South Africa, Aug. 1991, pp. 85–92. [2] A. Bóveda, F. Ortigoso, and J. I. Alonso, “A 0.7–3 GHz GaAs QPSK/QAM direct modulator,” IEEE J. Solid-State Circuits, vol. 28, no. 12, pp. 1340–1349, Dec. 1993. [3] W. Philibert and R. Verbiest, “A subharmonically pumped I/Q vector -band satellite communication,” in IEEE Radio modulator MMIC for Frequency Integrated Circuits Symp. Dig., Boston, MA, Jun. 2000, pp. 183–186. [4] B. Shi, W. Shan, and P. Andreani, “A 57-dB image band rejection CMOS Gm-C polyphase filter with automatic frequency tuning for Bluetooth,” in Proc. IEEE Int. Circuits and Systems Symp., vol. V, Scottsdale, AZ, May 2002, pp. 169–172.

Ka

[5] I. A. Koullias, J. H. Havens, I. G. Post, and P. E. Bronner, “A 900 MHz transceiver chip set for dual-mode cellular radio mobile terminals,” in IEEE Int. Solid-State Circuits Conf. Tech. Dig., San Francisco, CA, Feb. 1993, pp. 24–26. [6] M. Faulkner, T. Mattson, and W. Yates, “Automatic adjustment of quadrature modulators,” Electron. Lett., vol. 27, no. 3, pp. 214–216, Jan. 31, 1991. [7] X. Huang and M. Caron, “Gain/phase imbalance and dc offset compensation in quadrature modulators,” in Proc. IEEE Int. Circuits and Systems Symp., vol. IV, Scottsdale, AZ, May 2002, pp. 811–814. [8] W. Schiller, “High linearity upconverter with ultra low distortion over a broad frequency band for digital radio links,” in Proc. 16th Eur. Microwave Conf., Dublin, Ireland, Sep. 1986, pp. 157–163. [9] D. M. Treyer and W. Bächtold, “A self-calibration technique for SSB modulators,” in Proc. 34th Eur. Microwave Conf., Amsterdam, The Netherlands, Oct. 2004, pp. 1045–1048. [10] B. Razavi, RF Microelectronics. Upper Saddle River, NJ: PrenticeHall, 1998. [11] B. Schönrock and R. Knöchel, “ -band homodyne network analyzer using a broad-band computer-controlled SSB-modulator,” in Proc. 20th Eur. Microwave Conf., Budapest, Hungary, Sep. 1990, pp. 1467–1472. [12] O. Stoukatch, “The diode controllable attenuators,” in Proc. 2nd IEEE Int. Microwave and Millimeter-Wave Technology Conf., Beijing, China, Sep. 2000, pp. 40–43. [13] S. Walker, “A low phase shift attenuator,” IEEE Trans. Microw. Theory Tech., vol. 42, no. 2, pp. 182–185, Feb. 1994. [14] D. M. Treyer and W. Bächtold, “A tunable -band image-rejection mixer,” in Proc. Asia–Pacific Microwave Conf., New Delhi, India, Dec. 2004. [CD ROM]. [15] T. Brauner, R. Vogt, and W. Bächtold, “A versatile calibration method for small active antenna arrays,” in Proc. 33rd Eur. Microwave Conf., Munich, Germany, Oct. 2003, pp. 797–800.

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Daniel Marco Treyer (S’99–M’04) was born in Aarau, Switzerland, in 1971. He received the El.-Ing. HTL degree from the Höhere Technische Lehranstalt, Brugg–Windisch, Switzerland, in 1994, the M.S.E.E. degree from the University of Illinois at Urbana-Champaign (UIUC), in 1998, and is currently working toward the Ph.D. degree at the Swiss Federal Institute of Technology (ETH), Zurich, Switzerland. During his M.S. thesis, he was involved with the construction and validation of an Fe Boltzmann temperature lidar. Since 1998, he has been with the Laboratory for Electromagnetic Fields and Microwave Electronics, ETH, where his research is focused on microwave and millimeter-wave circuit design.

Werner Bächtold (M’71–SM’99–F’00) was born on October 1, 1939. He received the Diploma and Ph.D. degrees in electrical engineering from the Swiss Federal Institute of Technology (ETH), Zurich, Switzerland, in 1964 and 1968, respectively. From 1969 to 1987, he was with the IBM Zurich Research Laboratory, where he was involved with device and circuit design and analysis of GaAs MESFETs, design of logic and memory circuits of Josephson junctions, and semiconductor lasers for digital communication. He had several assignments with the IBM T. J. Watson Research Center, Yorktown Heights, NY. Since December 1987, he has been Professor of electrical engineering with ETH. He headed the Microwave Electronics Group, Laboratory for Electromagnetic Fields and Microwave Electronics, and was engaged in the design and characterization of MMICs, design and technology of InP-HEMT devices and circuits, and microwave photonics. He retired at the end of March 2005.

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Low-Loss LTCC Cavity Filters Using System-on-Package Technology at 60 GHz Jong-Hoon Lee, Student Member, IEEE, Stephane Pinel, Member, IEEE, John Papapolymerou, Senior Member, IEEE, Joy Laskar, Fellow, IEEE, and Manos M. Tentzeris, Senior Member, IEEE

Abstract—In this paper, three-dimensional (3-D) integrated cavity resonators and filters consisting of via walls are demonstrated as a system-on-package compact solution for RF front-end modules at 60 GHz using low-temperature cofired ceramic 4 open stub has (LTCC) technology. Slot excitation with a been applied and evaluated in terms of experimental performance and fabrication accuracy and simplicity. The strongly coupled cavity resonator provides an insertion loss 0.84 dB, a return loss 20.6 dB over the passband ( 0.89 GHz), and a 3-dB bandwidth of approximately 1.5% ( 0.89 GHz), as well as a simple fabrication of the feeding structure (since it does not require to drill vias to implement the feeding structure). The design has been utilized to develop a 3-D low-loss three-pole bandpass filter for 60-GHz wireless local area network narrow-band ( 1 GHz) applications. This is the first demonstration entirely authenticated by measurement data for 60-GHz 3-D LTCC cavity filters. This filter exhibits an insertion loss of 2.14 dB at the center frequency of 58.7 GHz, a rejection 16.4 dB over the passband, and a 3-dB bandwidth approximately 1.38% ( 0.9 GHz). Index Terms—Bandpass filter (BPF), cavity filters, cavity resonators, low-temperature cofired ceramic (LTCC), millimeter wave, system-on-package (SOP), three-dimensional (3-D) integration.

I. INTRODUCTION

T

HE RAPID growth of wireless local area and personal communication networks, as well as sensor applications, has led to a dramatic increase of interest in the regimes of RF/microwave/millimeter-wave systems [1]. Such emerging applications with data rates in excess of 100 Mb/s require real-estate efficiency, low-cost manufacturing, and excellent performance achieved by a high level of integration of embedded functions using low-cost and high-performance materials. The multilayer low-temperature cofired ceramic (LTCC) system-on-package (SOP) [2] has emerged as a candidate to provide an efficient integration of the RF passives due to its mature multilayer fabrication capability and its relatively low cost [2]. On-package integrated cavity filters using LTCC multilayer technology are a very attractive option for three-dimensional (3-D) RF front-end modules up to the millimeter-wave frequency range because of their relatively

Manuscript received April 29, 2005; revised July 26, 2005. This work was supported in part by the Asahi Glass Company, by the National Science Foundation under CAREER Award ECS-9984761 and Grant ECS-0313951, by the Georgia Electronic Design Center, and by the Georgia Institute of Technology Packaging Research. The authors are with the School of Electrical and Computer Engineering, Georgia Institute of Technology, Atlanta, GA 30332 USA (e-mail: [email protected]). Digital Object Identifier 10.1109/TMTT.2005.859864

high quality ( ) factor compared to stripline/microstrip [3], [4] or lumped-element-type filters [5]. Numerous publications [6]–[15] have dealt with 3-D low-loss and high- resonators/bandpass filters (BPFs) utilizing electromagnetic bandgap (EBG) substrates and micromachined technology. In particular, the recent development of Duroid-based EBG cavity resonators and filters [6], [7] have demonstrated advantages of implementing EBG instead of fully conducting metals [8] as sidewalls in terms of reconfigurability and inexpensive mass production. The idea of using rows of stacked vias (over 1000) as sidewalls has been applied to produce high cavity resonators at -band [9] and low-loss quasi-planar twopole filters based on capacitive loaded cavities at -band [10] using low-temperature LTCC technology. Also, narrow-band two-pole filters, which can be embedded inside packaging, have been implemented in LTCC by employing capacitive loading techniques at -band [11]. For -band applications, planar alumina waveguide filters with coplanar waveguide (CPW) I/O ports have exhibited low insertion loss ( 3 dB) and good stopband rejection, but occupy large real estate because of the twodimensional (2-D) arrangement of the resonators [12], [13]. In this paper, we present the first extensive report, entirely validated by measured data on low-loss 60-GHz ( -band) 3-D multilayer cavity resonators and three-pole BPFs utilizing via fences as sidewalls, enabling a complete passive solution for 3-D compact low-cost wireless RF front-end modules open stub is emin LTCC. The slot excitation with a ployed for 60-GHz cavity resonators and evaluated in terms of -parameters, bandwidth, and fabrication simplicity based on measurement data. After ensuring the performance of the slot excitation technique at -band (55–65 GHz), a compact and low-loss 3-D geometry of three-pole BPFs has been implemented by importing three identical cavity resonators coupled together with moderate external and inter-resonator couplings for 60-GHz wireless local area network (WLAN) narrow-band ( 1 GHz) applications. II. RECTANGULAR CAVITY RESONATOR The proposed cavity resonators are based on the theory of rectangular cavity resonators [16] and all designs are optimized with the aid of a finite-element method (FEM) -based full-wave simulator (HFSS). The cavity resonator is built utilizing conducting planes as horizontal walls and via fences as sidewalls, as shown in Fig. 1. The size ( in Fig. 1) and spacing ( in Fig. 1) of via posts are properly chosen to prevent electromagnetic field leakage and

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the dielectric loss from the filling substrates, and the leakage loss through the via walls, respectively. Since the gap between the via at the highest frequency of interest as menposts is less than tioned, the leakage (radiation) loss can be negligible as mentioned above and the individual quantity of two other factors can be obtained from [18] (3) where is the wavenumber in the resonator , is the surface resistance of the cavity ground planes , is the wave impedance of the LTCC resonator filling, are the length, width, and height of the cavity resonator, respectively, and Fig. 1. Cavity resonator utilizing via fences as sidewalls.

(4)

to achieve stopband characteristic at the desired resonant fremode is obquency [6]. The resonant frequency of the tained by [16] (1) is the resonant frequency, is the speed of light, where the dielectric constant, is the length of the cavity, is the width of the cavity, and is the height of the cavity. Using (1), the initial dimensions of the cavity with perfectly conducting walls are determined for a resonant frequency of 60 GHz for dominant mode by simply indexing , , the and optimized with a full-wave electromagnetic simulator ( mm, mm, mm). The design parameters of the feeding structures are then slightly modified to achieve the best performance in terms of low insertion loss and accurate resonant frequency. In Section III, we discuss the open stub. slot excitation with a To decrease the metal loss and enhance the factor, the vertical conducting walls are replaced by a lattice of via posts. In our case, we used Cassivi’s expressions [17] to get the preliminary design values, and then the final dimensions of the cavity are fine tuned with the HFSS simulator. The spacing ( ) between the via posts of the sidewalls is limited to less than half at the highest frequency of interest guided wavelength so that the radiation losses becomes negligible [6]. It has also been proven that smaller via sizes result in an overall size reduction of the cavity [6]. In our case, we used the minimum size m in Fig. 1) allowed by LTCC of vias ( via diameter design rules. The spacing between the vias has also been set to the minimum via pitch (390 m). In the case of low external coupling, the unloaded ( ) is controlled by three loss mechanisms and is defined by [18] (2) , , and take into account the conductor where loss from the horizontal plates (the metal loss of the horizontal plates dominates, especially for a thin substrate such as 0.3 mm),

is the loss tangent of the LTCC substrate. where All fabricated filters were measured using the Agilent 8510C Network Analyzer and Cascade Microtech probe station with 250- m-pitch air coplanar probes. III. CAVITY FEEDING STRUCTURE Fig. 2(a)–(c) shows the top view, a 3-D overview, and the side view of the microstrip-fed cavity resonator, respectively, using open stub slot excitation technique. Microstrip lines are a utilized to excite the resonator through coupling slots etched in the top metal layer (metal 2) of the cavity, as shown in Fig. 2(c). In order to maximize the magnetic coupling by maximizing magnetic currents, a short is placed at the center of each slot by open stubs. The excitation terminating the feedlines with using an open stub contributes to fabrication simplicity with no need to drill via-holes to short the end of feedlines. It also avoids the loss and inductance effects generated by shorting vias close to the slot, which could be serious in the millimeter-wave frequency range. The accurate design of the external coupling slots is a key issue to achieve low-loss cavity resonators. The external coupling factor is directly related to the input resistance and reactance that can be controlled by the position and size of the coupling aperture [19]. To determine the dimensions of the slots for the optimum response, the coupling slots are initially located at a quarter of the cavity length [SP in Fig. 2(a)] to maximize the coupling [6], and then the slot width [SW in Fig. 2(a)] is in Fig. 2(a)]. varied with the constant slot length [ The dimensions of the coupling slots have been determined to 0.538 0.21 mm . The position of the slots is adjusted to obtain the desired insertion loss, resonant frequency, and input impedances. The optimized results of resonant frequency, insermm tion loss, and bandwidth are obtained with [see Fig. 2(a)]. The solid conducting sidewalls are replaced with rows of vias. The spacing between via rows is set up to the minimum via pitch (390 m). In this process, major filter characteristics such as resonant frequency and insertion loss are not only affected by the via pitch/diameter, but also by the number of rows of vias. The resonator characteristics are investigated with the increase

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Fig. 3. Magnitude of electric field distribution on horizontal plane inside the cavity using slot excitation with an open stub at the resonant frequency (= 59:2 GHz). TABLE I DESIGN PARAMETERS OF CAVITY RESONATORS USING OPEN STUBS

Fig. 2. LTCC cavity resonator employing slot excitation with a open stub. (a) Top view of feeding structure. (b) 3-D overview. (c) Side view of the proposed resonator.

of the number of via rows (1 3). Double and triple rows exhibit almost the same characteristics such as an insertion loss of 1.14 dB, while a single row exhibits 1 dB higher insertion loss due to a higher leakage. Fig. 3 shows the electric field distributions inside the cavity surrounded by rows of vias. It is clearly observed that two rows of vias are sufficient to block the field leakage through the vias. All the final design parameters are summarized in Table I. Although it has been verified that double and triple rows exhibit almost the same performance in simulations, three rows of via posts were used to ensure a high level of leakage block with respect to both the simulation error and the fabrication accuracy. Once the slot size/position and cavity size is determined, the length of the open-circuited stub is optimized to maximize the magnetic coupling. The length [OSL in Fig. 2(a)] of open-cir. The fringing fields genercuited stub is initially set up to ated by open-end discontinuity can be modeled by an equivalent length of transmission line [22], which is determined to be ap. Therefore, the optimum length of the stub proximately due to the effect of fringing field. is approximately The coupling coefficient between the cavity resonator and feeding microstrip line can be analytically evaluated using (5). due to the aperture coupling of microstrip feeding is First,

obtained by utilizing Wheeler’s equivalent energy concept [20] and parallel-plate waveguide model of microstrip line [21] and then the coupling coefficient is given as [15] (5) where is the coupling coefficient of external coupling and is the unloaded from (2). The proposed cavity resonator was fabricated in an LTCC 044 SiO B O glass by the Asahi Glass Company, Kanagawa, of the substrate is 5.4 and Japan. The relative permittivity is 0.0015 at 35 GHz. The dielectric layer its loss tangent thickness per layer is 100 m, and the metal thickness is 9 m. The resistivity of metal (silver trace) is determined to be 2.7 10 m. The top view of the microstrip feedlines and CPW probe pads utilized to excite the embedded cavity resonator is shown in Fig. 4. The overall size is 3.8 mm 3.2 mm 0.3 mm (including the CPW measurement pads). The measured insertion and reflection loss of the fabricated cavity are compared with the simulated results in Fig. 5(a). In the measurement, the capacitance and inductance effects of the probing pads are deembedded by use of Wincal software so that effects, such as those due to the CPW loading, become negligible. The cavity exhibits an insertion loss 0.84 dB, a return loss 20.6 dB over the passband, and a 3-dB bandwidth of approximately 1.5% at a center frequency of 59.2 GHz. The simulation results for the insertion loss ( 0.78 dB) and the return loss ( 22 dB) correlate well with the experiments, but exhibit a slightly increased bandwidth of 2.3% at a center fre-

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due to the fabrication accuracy of the slot design that has been optimized for the original resonant frequencies and not for the shifted frequencies. can be calculated using the following [6]: The (6) (7) Fig. 4. Fabricated input/output microstrip feedlines with an open stub and CPW probe pads utilized to excite the embedded cavity resonator.

dB

(8)

is the loaded , and is the 3-dB bandwidth. The where weak external coupling allows for the apt verification of of the cavity resonator since approaches with the weak external coupling, as described in (6). The weak coupling also . abates sensitivity of the measurement on the amplitude of Using the above definitions, a weakly coupled cavity resonator 20 dB) has been separately investigated in HFSS and ( of 367 at 59.8 GHz compared to the theoretical exhibits a of 372 at 60 GHz from (2)–(4). IV. THREE-POLE FILTER

Fig. 5. Comparison between measured and simulated S -parameters (S 11 and S 21) of 0.3-mm-high cavity resonator using slot excitation with an open stub. (a) Simulation with = 5:4 versus measurement. (b) Simulation with = 5:5 versus measurement.

quency of 59.8 GHz. The center frequency downshift can be attributed to the dielectric constant variation at these high frequencies. The preliminary HFSS simulation presumed that the averaged relative permittivity would be increased to 5.5 across 55–65 GHz. An expanded plot of a comparison of the simulaand the measurement is shown in Fig. 5(b). tion with The coincidence between the center frequencies is observed in Fig. 5(b). The narrower bandwidth in measurements might be

We have designed and fabricated three-pole filters using via walls for 60-GHz WLAN narrow-band ( 1 GHz) applications that consist of three coupled cavity resonators [i.e., Cavity1, Cavity2, Cavity3 in Fig. 6(b)]. The 3-D overview and side view are illustrated in Fig. 6(a) and (b), respectively. The three-pole BPF based on a Chebyshev low-pass prototype filter is developed for a center frequency of 60 GHz, 3-dB insertion loss, 0.1-dB in-band ripple, and 1.67% fractional bandwidth. in To meet design specifications, the cavity height [ Fig. 2(b)] was increased to 0.5 mm (five substrate layers) to and, consequently, to obtain narrower achieve a higher bandwidth. The cavity resonator with 0.5-mm height has been fabricated in LTCC and measured. The comparison between the simulation and the measurement is shown in Fig. 7. An insertion loss of 1.24 dB at the center frequency of 59.2 GHz and a narrow bandwidth of 1.35% ( 0.8 GHz) has been meayields 426 and it is very close to the sured. The theoretical of 424 from a weakly coupled cavity in HFSS. simulated After verifying the experimental performance of a single cavity resonator, the external coupling and inter-resonator coupling are considered for the three-pole filter design. can be defined from the following specifications Firstly, [22]: (9) where are the element values of the low-pass prototype, is the resonant frequency, and is the bandwidth of the filter. The input and output were calculated to be 61.89. The position and size of the external slots [see Fig. 6(a)] are the main . The slots have been poparameters to achieve the desired ) and their length sitioned at a quarter of the cavity length ( has been fixed to [ in Fig. 6(a)]. [shown in Fig. 8(a)] has then been (using full-wave simulations) evaluin Fig. 6(a)] ated as a function of the external slot width [

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Fig. 6. LTCC three-pole cavity BPF employing slot excitation with an open stub. (a) 3-D overview. (b) Side view of the proposed filter.

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Fig. 8. (a) External factor ( ) evaluated as a function of external slot ) as a function width (SW ). (b) Inter-resonator coupling coefficient ( of internal slot width (SW ).

S

S

Fig. 7. Comparison between measured and simulated -parameters ( 11 and 21) of 0.5-mm-high cavity resonator using slot excitation with an open stub.

S

based on the following relationship [22]:

, the size of internal slots [see Fig. 6(a)] is optimized using full-wave simulations to find the two characteristic frequencies ) that are the frequencies of the peaks in the transmis( , sion response of the coupled structure when an electric wall or magnetic wall, respectively, is inserted in the symmetrical plane can then be determined by measuring the amount [22]. that the two characteristic frequencies deviate from the resonant and the characterfrequency. The relationship between ) is defined as follows [22]: istic frequencies ( , (12)

(10) where is the frequency difference between the phase response of . Secondly, the inter-resonator coupling coefficients between the vertically adjacent resonators is determined by [22] (11) where filter.

k

or because of the symmetrical nature of the was calculated to be 0.0153. To extract the desired

Based on the above theory, the physical dimensions of internal slots were determined by using a simple graphical approach displaying two distinct peaks of character frequencies . Fig. 8(b) shows the graphical relationship befor a fixed and internal slot width [ in Fig. 6(a)] varitween in Fig. 6(a)]. ation with the fixed slot length [ was determined to be 0.261 mm corresponding to the refrom Fig. 8(b). After determining the quired initial dimensions of the external/internal slots, the other design parameters such as the open stub length [OSL in Fig. 6(b)] and in Fig. 6(a)] using via the cavity length and width [ and

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TABLE II DESIGN PARAMETERS OF THREE-POLE CAVITY FILTER USING AN OPEN STUB

with the aid of WinCal3.0 software. The filter exhibits an insertion loss 2.14 dB, which is slightly higher than the simulated value of 2.08 dB, and a return loss 16.39 dB compared to a simulated value 18.37 dB over the passband shown in Fig. 9(a) and (b), respectively. In Fig. 9(a), the measurement shows a slightly increased 3-dB fractional bandwidth of approximately 1.53%( 0.9 GHz) at a center frequency 58.7 GHz. The simulated results give a 3-dB bandwidth of 1.47% ( 0.88 GHz) at a center frequency of 60 GHz. The center frequency downshift can be attributed to the fabrication accuracy such as slot positioning affected by the alignment between layers, layer thickness tolerance, and higher dielectric constant at this high frequency range (55–65 GHz) than 5.4 that is the relative permittivity at 35 GHz, as mentioned in Section III. The overall response of the measurement is in excellent agreement with the simulation, except a frequency shift of 1.3 GHz ( 2%). This three-pole filter can be used in the development of three-pole duplexers for millimeter-wave wireless systems. V. CONCLUSION In this study, 3-D integrated cavity resonators and three-pole filters composed of via walls have been successfully demonstrated with excellent performance using LTCC technology at open has been studied 60 GHz. The slot excitation with a and the resulting weak-coupling cavity demonstrated excellent performances in terms of a low insertion loss ( 0.84 dB) at 59.2 GHz and a 3-dB bandwidth of approximately 1.5% ( 0.89 GHz), as well as fabrication simplicity. A low-loss fully integrated three-pole BPF employing a slot excitation with an open stub has been implemented for the first time for 60-GHz WLAN narrow-band ( 1 GHz) applications. It exhibited an insertion loss of 2.14 dB at a center frequency of 58.7 GHz, a return loss 16.39 dB over the passband, and a 3-dB fractional bandwidth of approximately 1.38% ( 0.9 GHz). To the best of the authors’ knowledge, this is the lowest loss reported for a LTCC 3-D integrated narrow-band filter at 60 GHz. The presented structures can be easily integrated within a 3-D LTCC 60-GHz front-end module and can also be used in the development of multipole 60-GHz duplexers for a SOP RF front end. REFERENCES

S

S

Fig. 9. Comparison between measured and simulated: (a) 21 and (b) 11 of three-pole cavity BPF using slot excitation with an open stub.

walls are determined under the design guidelines described in Sections II and III. The initial dimensions of the external/internal slot widths are set up as optimal variables and fine tuned to achieve the desired frequency response using HFSS simulators. The summary of all design parameters for the three-pole filter is given in Table II. Fig. 9(a) and (b) shows the comparison between the simulated and measured -parameters of the BPF. In the measurements, the parasitic effects from the I/O open pads were deembedded

[1] H. H. Meinel, “Commercial applications of millimeter waves: History, present status, and future trends,” IEEE Trans. Microw. Theory Tech., vol. 43, no. 7, pp. 1639–1653, Jul. 1995. [2] K. Lim, S. Pinel, M. F. Davis, A. Sutono, C.-H. Lee, D. Heo, A. Obatoynbo, J. Laskar, E. M. Tentzeris, and R. Tummala, “RF-system-onpackage (SOP) for wireless communications,” IEEE Micro, vol. 3, no. 1, pp. 88–99, Mar. 2002. [3] C. H. Lee, A. Sutono, S. Han, K. Lim, S. Pinel, J. Laskar, and E. M. -band transmitter module,” Tentzeris, “A compact LTCC-based IEEE Trans. Adv. Packag., vol. 25, no. 3, pp. 374–384, Aug. 2002. -band narrow bandpass [4] B. G. Choi, M. G. Stubbs, and C. S. Park, “A filter using LTCC technology,” IEEE Microw. Wireless Compon. Lett., vol. 13, no. 9, pp. 388–389, Sep. 2003. [5] V. Piatnitsa, E. Jakku, and S. Leppaevuori, “Design of a 2-pole LTCC filters for wireless communications,” IEEE Trans. Wireless Commun., vol. 3, no. 2, pp. 379–381, Mar. 2004. [6] M. J. Hill, R. W. Ziolkowski, and J. Papapolymerou, “Simulated and measured results from a Duroid-based planar MBG cavity resonator filter,” IEEE Microw. Wireless Compon. Lett., vol. 10, no. 12, pp. 528–530, Dec. 2000.

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[7] H.-J. Hsu, M. J. Hill, J. Papapolymerou, and R. W. Ziolkowski, “A planar -band electromagnetics bandgap (EBG) 3-pole filter,” IEEE Microw. Wireless Compon. Lett., vol. 12, no. 7, pp. 255–257, Jul. 2002. [8] C. A. Tavernier, R. M. Henderson, and J. Papapolymerou, “A reducedsize silicon micromachined high- resonator at 5.7 GHz,” IEEE Trans. Microw. Theory Tech., vol. 50, no. 10, pp. 2305–2314, Oct. 2002. [9] A. El-Tager, J. Bray, and L. Roy, “High- LTCC resonators For millimeter wave applications,” in IEEE MTT-S Int. Microwave Symp. Dig., Philadelphia, PA., Jun. 2003, pp. 2257–2260. [10] P. Ferrand, D. Baillargeat, S. Verdeyme, J. Puech, M. Lahti, and T. Jaakola, “LTCC reduced-size bandpass filters based on capacitively loaded cavities for band application,” in IEEE MTT-S Int. Microwave Symp. Dig., Long Beach, CA., Jun. 2005, pp. 1789–1792. [11] X. Gong, W. J. Chappell, and L. P. B. Katehi, “Multifunctional substrates for high-frequency applications,” Microw. Wireless Compon. Lett., vol. 13, no. 10, pp. 428–430, Oct. 2003. [12] M. Ito, K. Maruhashi, K. Ikuina, T. Hashiguchi, S. Iwanaga, and K. Ohata, “60-GHz-band dielectric waveguide filters with cross-coupling for flip-chip modules,” in IEEE MTT-S Int. Microwave Symp. Dig., Seattle, WA., Jun. 2002, pp. 1789–1792. [13] S. T. Choi, K. S. Yang, K. Tokuda, and Y. H. Kim, “A -band planar narrow bandpass filter using a new type integrated waveguide transition,” IEEE Microw. Wireless Compon. Lett., vol. 14, no. 12, pp. 545–547, Dec. 2004. [14] M. Chatras, P. Blondy, D. Cros, O. Vendier, C. Drevon, and J. L. Cazaux, “Narrow band micro-machined band pass filter and a surface-mountable topology,” in Proc. 33rd Eur. Microwave Conf., Munich, Germany, Oct. 2003, pp. 813–815. [15] T. Euler and J. Papapolymerou, “Silicon micromachined EBG resonator and two-pole filter with improved performance characteristics,” IEEE Microw. Wireless Compon. Lett., vol. 13, no. 9, pp. 373–375, Sep. 2003. [16] R. E. Collin, Foundations for Microwave Engineering. New York: McGraw-Hill, 1992. [17] Y. Cassivi and K. Wu, “Low cost microwave oscillator using substrate integrated waveguide cavity,” IEEE Microw. Wireless Compon. Lett., vol. 13, no. 2, pp. 48–50, Feb. 2003. [18] D. M. Pozar, Microwave Engineering, 2 ed. New York: Wiley, 1998. [19] D. M. Pozar and D. H. Schaubert, Microstrip Antennas. Piscataway, NJ: IEEE Press, 1995. [20] H. Wheeler, “Coupling holes between resonant cavities or waveguides evaluation in terms of volume ratios,” IEEE Trans. Microw. Theory Tech., vol. MTT-12, no. 3, pp. 231–244, Mar. 1964. [21] W. H. Leighton and A. G. Milnes, “Junction reactance and dimensional tolerance effects on -band 3 dB directional couplers,” IEEE Trans. Microw. Theory Tech., vol. MTT-19, no. 10, pp. 818–824, Oct. 1971. [22] J.-S. Hong and M. J. Lancaster, Microstrip Filters for RF/Microwave Applications. New York: Wiley, 2001.

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Jong-Hoon Lee (S’98) received the B.S. degree in electrical engineering from the Pennsylvania State University, University Park, in 2001, the M.S. degree from the Georgia Institute of Technology, Atlanta, in 2004, and is currently working toward the Ph.D. degree in electrical and computer engineering at the Georgia Institute of Technology. He is a member of the Georgia Tech ATHENA Research Group, National Science Foundation (NSF) Packaging Research Center, and the Georgia Electronic Design Center, Atlanta. He has authored or coauthored over 20 papers in referred journals and conference proceedings. His research interests are packaging technology for microwave/millimeter-wave systems and digital signal processing (DSP)-based predictor to improve the computational efficiency of the simulation. He is currently involved in the development of SOP module for millimeter-wave wireless systems using LTCC ceramic and LCP organic technologies.

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Stéphane Pinel (M’05) received the B.S. degree from Paul Sabatier University, Toulouse, France, in 1997, and the Ph.D. degree in microelectronics and microsystems (with highest honors) from the Laboratoire d’Analyze et d’Architecture des Systemes, Centre National de la Recherche Scientifique, Toulouse, France, in 2000. For three years, he has been involved with a UltraThin Chip Stacking (UTCS) European Project. He is currently a Research Engineer with the Microwaves Applications Group, Georgia Institute of Technology. He has authored or coauthored over 80 journal and proceeding papers, two book chapters, and numerous invited talks. He holds four patents/invention disclosures. His research interests include advanced 3-D integration and packaging technologies, RF and millimeter-waves embedded passives design using organic and ceramic material, RF microelectromechanical systems (MEMS) and micromachining techniques, SOP for RF front-end modules, and system-on-insulator (SOI) RF circuit design. Dr. Pinel has participated and organized numerous workshops. He was the recipient of the First Prize Award presented at the 1998 Society of Electronic and Electro-technique (SEE), the Second Prize Award presented at 1999 International Microelectronics and Packaging Society (IMAPS), and the Best Paper Award presented at the 2002 International Conference on Microwave and Millimeter-Wave Technology, Beijing, China.

John Papapolymerou (S’90–M’99–SM’04) received the B.S.E.E. degree from the National Technical University of Athens, Athens, Greece, in 1993, and the M.S.E.E. and Ph.D. degrees from The University of Michigan at Ann Arbor, in 1994 and 1999, respectively. From 1999 to 2001, he was a faculty member with the Department of Electrical and Computer Engineering, University of Arizona, Tucson. During the summers of 2000 and 2003, he was a Visiting Professor with The University of Limoges, Limoges, France. From 2001 to 2005, he was an Assistant Professor with the School of Electrical and Computer Engineering, Georgia Institute of Technology, Atlanta, where he is currently an Associate Professor. He has authored or coauthored over 120 publications in peer-reviewed journals and conferences. His research interests include the implementation of micromachining techniques and MEMS devices in microwave, millimeter-wave, and terahertz circuits and the development of both passive and active planar circuits on semiconductor (Si/SiGe, GaAs) and organic substrates (LCP, LTCC) for system-on-a-chip (SOC)/ SOP RF front ends. Dr. Papapolymerou currently serves as the vice-chair for Commission D of the U.S. National Committee of URSI and as an associate editor for the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION. During 2004, he was the chair of the IEEE Microwave Theory and Techniques (MTT)/Antennas and Propagation (AP) Atlanta Chapter. He was the recipient of the 2004 Army Research Office (ARO) Young Investigator Award, the 2002 National Science Foundation (NSF) CAREER award, the Best Paper Award presented at the 3rd IEEE International Conference on Microwave and Millimeter-Wave Technology (ICMMT2002), Beijing, China, and the 1997 Outstanding Graduate Student Instructional Assistant Award presented by the American Society for Engineering Education (ASEE), The University of Michigan Chapter. His student was also the recipient of the Best Student Paper Award presented at the 2004 IEEE Topical Meeting on Silicon Monolithic Integrated Circuits in RF Systems, Atlanta, GA.

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Joy Laskar (S’84–M’85–SM’02–F’05) received the B.S. degree (highest honors) in computer engineering with math/physics minors from Clemson University, Clemson, SC, in 1985, and the M.S. and Ph.D. degrees in electrical engineering from the University of Illinois at Urbana-Champaign, in 1989 and 1991, respectively. Prior to joining the Georgia Institute of Technology, Atlanta, in 1995, he held faculty positions with the University of Illinois at Urbana-Champaign and the University of Hawaii. At the Georgia Institute of Technology, he holds the Joseph M. Pettit Professorship of Electronics and is currently the Chair for the Electronic Design and Applications Technical Interest Group, the Director of Georgia’s Electronic Design Center, and the System Research Leader for the National Science Foundation (NSF) Packaging Research Center. With the Georgia Institute of Technology, he heads a research group with a focus on integration of high-frequency electronics with opto-electronics and integration of mixed technologies for next-generation wireless and opto-electronic systems. In July 2001, he became the Joseph M. Pettit Professor of Electronics with the School of Electrical and Computer Engineering, Georgia Institute of Technology. He has authored or coauthored over 210 papers. He has ten patents pending. His research has focused on high-frequency integrated-circuit (IC) design and their integration. His research has produced numerous patents and transfer of technology to industry. Most recently, his research has resulted in the formation of two companies. In 1998, he cofounded the advanced WLAN IC company RF Solutions, which is now part of Anadigics. In 2001, he cofounded the next-generation interconnect company Quellan Inc., Atlanta, GA, which develops collaborative signal-processing solutions for enterprise applications. Dr. Laskar has presented numerous invited talks. For the 2004–2006 term, he has been appointed an IEEE Distinguished Microwave Lecturer for his Recent Advances in High Performance Communication Modules and Circuits seminar. He was a recipient of the 1995 Army Research Office’s Young Investigator Award, 1996 recipient of the National Science Foundation (NSF) CAREER Award, 1997 NSF Packaging Research Center Faculty of the Year, 1998 NSF Packaging Research Center Educator of the Year, 1999 corecipient of the IEEE Rappaport Award (Best IEEE Electron Devices Society journal paper), the faculty advisor for the 2000 IEEE Microwave Theory and Techniques Society (IEEE MTT-S) International Microwave Symposium (IMS) Best Student Paper Award, 2001 Georgia Institute of Technology Faculty Graduate Student Mentor of the Year, a 2002 IBM Faculty Award, 2003 Clemson University College of Engineering Outstanding Young Alumni Award, and 2003 Outstanding Young Engineer of the IEEE MTT-S.

Manos M. Tentzeris (SM’03) received the Diploma degree in electrical and computer engineering from the National Technical University of Athens, Athens, Greece, in 1992, and the M.S. and Ph.D. degrees in electrical engineering and computer science from The University of Michigan at Ann Arbor, in 1993 and 1998, respectively. He is currently an Associate Professor with the School of Electrical and Computer Engineering, Georgia Institute of Technology, Atlanta. During the summer of 2002, he was a Visiting Professor with the Technical University of Munich, Munich, Germany. He has authored or coauthored over 200 papers in refereed journals and conference proceedings and eight book chapters. He has helped develop academic programs in highly integrated packaging for RF and wireless applications, microwave MEMS, SOP-integrated antennas, RF ID’s 3-D integration, and adaptive numerical electromagnetics (finite difference time domain (FDTD), multiresolution algorithms). He is the Georgia Tech National Science Foundation (NSF)-Packaging Research Center Associate Director for RF Research and the RF Alliance Leader. He is also the Leader of the Novel Integration Techniques Sub-Thrust of the Broadband Hardware Access Thrust of the Georgia Electronic Design Center (GEDC) of the State of Georgia. Dr. Tentzeris is member of the Technical Chamber of Greece. He was the 1999 Technical Program co-chair of the 54th ARFTG Conference, Atlanta, GA. He is the vice-chair of the RF Technical Committee (TC16) of the IEEE Components, Packaging, and Manufacturing Technology (CPMT) Society. He was the recipient of the 2006 IEEE Microwave Theory and Technique Society (IEEE MTT–S) Outstanding Young Engineer Award, the 2004 IEEE TRANSACTIONS ON ADVANCED PACKAGING Commendable Paper Award, the 2003 IEEE CPMT Outstanding Young Engineer Award, the 2002 International Conference on Microwave and Millimeter-Wave Technology Best Paper Award (Beijing, China), the 2002 Georgia Tech-Electrical and Computer Engineering (ECE) Outstanding Junior Faculty Award, the 2001 ACES Conference Best Paper Award, the 2000 NSF CAREER Award, and the 1997 Best Paper Award, International Hybrid Microelectronics and Packaging Society.

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Relationships Between Common Source, Common Gate, and Common Drain FETs Jianjun Gao and Georg Boeck, Senior Member, IEEE

Abstract—This paper comprehensively analyzes the relationship between common source (CS), common gate (CG), and common drain (CD) field-effect transistors (FETs). The signal and noise parameters of the CG and CD configuration can be obtained directly by using a simple set of formulas from CS signal and noise parameters. All the relationships provide a bi-directional bridge for the transformation between CS, CG, and CD FETs. This technique is based on the combination of an equivalent-circuit model and conventional two-port network signal/noise correlation matrix technique. The derived relationships have universal validity, but they 40 m gatewidth (number of gate finhave been verified at 2 gers unit gatewidth) double-heterojunction -doped AlGaAs/InGaAs/GaAs pseudomorphic high electron-mobility transistor with 0.25- m gate length. Good agreement has been obtained between calculated and measured results. Index Terms—Common drain (CD), common gate (CG), common source (CS), high electron-mobility transistor (HEMT), MESFET, noise parameters.

I. INTRODUCTION

F

IELD-EFFECT transistors (FETs) and pseudomorphic high electron-mobility transistors (pHEMTs) are widely used in the design of active monolithic microwave and millimeter-wave integrated circuits (MMICs) such as low-noise amplifiers (LNAs), mixers, oscillators, and other RF subsystem components. When used in amplifiers, FETs and high electron-mobility transistors (HEMTs) are almost exclusively operated in a common source (CS) configuration. However, a configuration in the CS mode usually produces more noise than its common gate (CG) counterpart. This is primarily due to the pure matching [1], [2] between optimum and real source impedance with respect to noise. The situation is much more relaxed in CG configuration because the input impedance is and not nearly pure capacitive as in CS. Therefore, a CG configuration has the advantage of being able to match the input to 50 and to the optimum (noise) source impedance at the same time, also especially in broad-band applications. As a consequence, it is much easier to achieve noise values close in a CG than in a CS configuration. In general, this to leads to lower noise values in CG than in CS configuration, values of both configurations are close together, although which we will show in this paper. Hence, a CG configuration is generally more suitable for optical and microwave broad-band Manuscript received April 27, 2005; revised August 8, 2005. J. Gao is with the Institute of RF and Opto-Electronic Integrated Circuits, Radio Engineering Department, Southeast University, Nanjing 210096, China (e-mail: [email protected]). G. Boeck is with the Microwave Engineering Department, Technische Universität Berlin, 10587 Berlin, Germany ([email protected]). Digital Object Identifier 10.1109/TMTT.2005.859863

communication applications. The common drain (CD) configuration has been widely used in the design of the low thermal-resistance oscillator and low-distortion variable-gain amplifier [3], [4]. The complete characterization and modeling of the transistor (CS, CG, and CD) in terms of noise and scattering parameters is necessary for computer-aided design (CAD) of microwave circuits. The -parameter and noise parameter for each configuration can be obtained by measuring test patterns of CS, CG, and CD configurations. However, this method requires two special test structures (CG and CD) for each device size on the wafer, and the nonuniformity across the wafer has to be ignored. A complex mathematical method for determining the noise parameters of active devices based on two- and three-port relationships is proposed by Grosch and Carpenter [5]. This method is complicated and time consuming due to the full characterization of the three-port network, which has to be first obtained from a two-port network. In this paper, we propose a simple, but efficient transformation technique for microwave FET devices. This technique is based on the combination of an equivalent-circuit model and conventional two-port network signal/noise correlation matrix technique. The signal and noise parameters of the CG and CD configuration can be obtained directly by using a simple set of formulas from CS signal and noise parameters. All the relationships provide a bi-directional bridge for the transformation between CS, CG, and CD FETs, respectively. The structure of this paper is as follows. Section II gives the derivation of analytical expressions for the signal and noise parameters based on an accurate equivalent-circuit model and conventional two-port network signal/noise correlation matrix technique. A comparison between the new expressions and experimental data measured for AlGaAs/InGaAs/GaAs pHEMTs is presented in Section III. A conclusion is given in Section IV. II. THEORETICAL ANALYSIS A. Small-Signal Model CS, CG, and CD configurations for the pHEMT devices considered in our studies are shown in Fig. 1(a)–(c), respectively. The intrinsic part with the parasitic series elements is shown in Fig. 2. , , and represent the input, output, and feedback pad capacitances, , , and represent the inductances of the gate, drain, and source feeding lines, respectively, and are the source and drain resistances, and is the , , and are gate–source, distributed gate resistance.

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Similar to (1)–(4), the corresponding and CD configurations are as follows:

-parameters for CG

(9)

Fig. 1. CS, CG, and CD configurations of pHEMT.

(10) (11) (12) (13) (14) (15) Fig. 2.

pHEMT small-signal equivalent-circuit model.

(16) gate–drain, and drain–source capacitances, respectively. is the channel resistance, is the transconductance, is the drain conductance, and is the time delay associated with transconductance. Since the pad capacitances are the same for the three configurations, all the transformation formulas in this paper will not include the effect of pad capacitances.

Comparing the -parameters of the CS, CG, and CD configurations, we can get the -parameter relationships, as shown in Tables I and II. The - and -parameter relationships between the CS, CG, and CD configurations are also given in Tables I and II by using the matrix conversion technique. C. Noise-Parameter Relationships

B. Signal Parameter Relationships The -parameter expressions for the CS configuration can be expressed as follows:

(1) (2)

Fig. 3 shows the noise equivalent circuits of CS, CG, and CD configurations for pHEMT devices. and are the two noise sources at the input of the noiseless pHEMT device for and are for the CG the CS configuration in Fig. 3(a), configuration, as shown in Fig. 3(b), and and are for the CD configuration, as shown in Fig. 3(c). The self- and cross-power spectral densities of the two-port -matrix form can be expressed as noise sources in the

(3) (17) (4) where , , , and trinsic part of the CS device

are the

-parameters of the in-

(5)

The chain noise correlation matrix is easier to obtain from the noise measurement because there is a direct relation between the measured noise parameters ( is the minimum is noise resistance, is optimum source noise figure, is optimum source susceptance, respecconductance, and tively). The chain noise correlation matrix can be expressed in terms of four noise parameters [6]

(6) (7) (8) with

.

(18)

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TABLE I

Z -, A-, AND S -PARAMETER RELATIONSHIPS BETWEEN CS AND CG CONFIGURATIONS

TABLE II

Z -, A-, AND S -PARAMETER RELATIONSHIPS BETWEEN CS AND CD CONFIGURATIONS

Using transformation techniques for noise sources, the relationships between the noise sources in the noise equivalent-circuit models of the CS, CG, and CD configurations [7] can be expressed as follows: (19)

Fig. 3. pHEMT noise equivalent-circuit models of CS, CG, and CD configurations.

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TABLE III NOISE PARAMETER RELATIONSHIPS BETWEEN CS AND CG CONFIGURATIONS

TABLE IV NOISE PARAMETER RELATIONSHIPS BETWEEN CS AND CD CONFIGURATIONS

(20) (21) (22)

The relationship of the noise parameters between the CS, CG, and CD configurations are given in Tables III and IV by using the noise source transformation matrix technique [6]. (valid for GHz), the For the condition noise-parameter relationships between CS, CG, and CD configurations for an intrinsic FET device can be simplified by

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TABLE V EXTRINSIC AND INTRINSIC pHEMT PARAMETERS

Fig. 5. Comparison of modeled and predicted S -parameters for the pHEMT in CG configuration, bias condition: V = 0:0 V, V = 2:0 V, frequency range 1–26 GHz, —: calculated data from equivalent-circuit model, : predicted data from measured noise parameters in CS configuration.

Fig. 6. Comparison of modeled and predicted S -parameters for the pHEMT in CD configuration, bias condition: V = 0:0 V, V = 2:0 V, frequency range : predicted 1–26 GHz, —: calculated data from equivalent-circuit model, data from measured noise parameters in CS configuration. Fig. 4. Comparison of modeled and measured S -parameters for the pHEMT in CS configuration, bias condition: V = 0:0 V, V = 2:0 V, frequency range : measured 1–26 GHz, —: calculated data from equivalent-circuit model, data.

neglecting the influence of the parasitics (23)

(24) (25) (26) (27) (28) From (23)–(28), it can be found that the noise parameters of the CG and CD configuration for an intrinsic FET device can be predicted directly by using a CS configuration. The expressions show that, up to moderate frequencies, the noise parameters of the three different configurations become close to each other if the parasitic elements can be neglected. III. EXPERIMENTAL VERIFICATION In order to verify the equations derived in Section II for the signal and noise parameters, 2 40 m AlGaAs/InGaAs/GaAs pHEMTs with 0.25- m gate length have been characterized [8].

Fig. 7. Comparison of modeled and measured noise parameter for the pHEMT in CS configuration, bias condition: V = 0:0 V, V = 2:0 V, frequency range 1–26 GHz, —: calculated data from equivalent-circuit model, : measured data.

The -parameter measurements for model extraction and verification were made by using an Agilent 8510C network analyzer. DC bias was supplied by an Agilent 4156A. Microwave noise-parameter measurements are carried out on wafer over the frequency range of 1–26 GHz using an ATN microwave noise measurement system NP5.

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Fig. 8. Comparison of modeled and predicted noise parameters for the pHEMT in CG configuration, bias condition: V = 0:0 V, V = 2:0 V, frequency range 1–26 GHz, —: calculated data from equivalent-circuit model, : predicted data from noise measurement of CS configuration.

Fig. 9. Comparison of modeled and predicted noise parameters for the pHEMT in CD configuration, bias condition: V = 0:0 V, V = 2:0 V, frequency range 1–26 GHz, —: calculated data from equivalent-circuit model, : predicted data from noise measurement of CS configuration.

The extracted values of the bias-independent extrinsic small-signal elements are summarized in Table V. Once the values of the parasitic elements are known, all bias-dependent elements can be easily determined by using a direct extraction , , , technique. The corresponding intrinsic parameters , , , and and noise model parameters , , and [9] are also summarized in Table V for a constant drain–source V and gate–drain voltage V, voltage respectively.

Fig. 10. Comparison of noise parameters in CS, CG, and CD configuration for the intrinsic pHEMT device, bias condition: V = 0:0 V, V = 2:0 V.

The measured -parameter and noise parameter without pad capacitances are obtained after deembedding the pad capacitances. Fig. 4 compares the measured and modeled -parameters for a 2 40 m AlGaAs/InGaAs/GaAs pHEMT in the frequency range of 1–26 GHz. An excellent agreement over the whole frequency range is obtained for the CS configuration. To illustrate the efficiency of the -parameter transformation formulas, we compare the measured and predicted results for CG and CD configurations in Figs. 5 and 6. Good agreement can be observed between calculated data from the equivalent-circuit model and data predicted by the transformation formulas.

GAO AND BOECK: RELATIONSHIPS BETWEEN CS, CG, AND CD FETs

Fig. 7 shows the measured and computed noise parameters versus frequency for the pHEMT in the CS configuration at V and V. To illustrate the efficiency of the transformation formulas for noise parameters, we compare the predicted and modeled results for pHEMTs in CG and CD configurations in Figs. 8 and 9. The predicted data for CG and CD configurations are obtained from the measured data of the CS configuration by using the formulas in Tables III and IV. Good agreement is obtained between modeled and predicted data, which verifies the supposed approach for the noise parameters. In Fig. 10, a comparison between CS, CG, and CD configuration noise parameters for the intrinsic pHEMT device are given. At low frequencies, all noise parameters are more or less iden. As can also be taken from tical with one exception, i.e., (23) and (24), in the CS and CG configuration differ from each other by a frequency-independent value. With increasing frequency, the noise-parameter values of the different configurations disperse more and more. While the minimum noise figand optimum source conductances show a weak ures and values disperse more strongly with indivergence, creasing frequency. IV. CONCLUSION In this paper, we have proposed a set of new analytical expressions for the relationship between CS, CG, and CD microwave FETs including MESFETs and HEMTs. All the relationships provide a bi-directional bridge for the transformation between CS, CG, and CD FETs. This technique is based on the combination of an equivalent-circuit model and conventional two-port network signal-noise correlation matrix technique. The validity of the new approach is proved by comparison with a measured -parameter and noise parameter up to 26 GHz. A good agreement has been obtained for AlGaAs/InGaAs/GaAs pHEMTs. REFERENCES [1] K. W. Kobayaski, D. C. Streit, D. K. Umemoto, T. R. Block, and A. K. Oki, “A monolithic HEMT–HBT direct-coupled amplifier with active input matching,” IEEE Microw. Guided Wave Lett., vol. 6, no. 1, pp. 55–57, Jan. 1996. [2] K. B. Niclas, “Active matching with common-gate MESFET’s,” IEEE Trans. Microw. Theory Tech., vol. MTT-33, no. 6, pp. 492–499, Jun. 1985. [3] R. L. Camisa and F. N. Sechi, “Common-drain flip-chip GaAs FET oscillators,” IEEE Trans. Microw. Theory Tech., vol. MTT-27, no. 5, pp. 391–394, May 1979. [4] K. Nishikawa and T. Tokumitsu, “An MMIC low-distortion variable-gain amplifier using active feedback,” IEEE Trans. Microw. Theory Tech., vol. 43, no. 12, pp. 2812–2816, Dec. 1995.

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[5] T. O. Grosch and L. A. Carpenter, “Two-port to three-port noise-wave transformation for CAD applications,” IEEE Trans. Microw. Theory Tech., vol. 41, no. 9, pp. 1543–1548, Sep. 1993. [6] H. Hillbrand and P. Russer, “An efficient method for computer-aided noise analysis of linear amplifier networks,” IEEE Trans. Circuits Syst., vol. CAS-23, no. 4, pp. 235–238, Apr. 1976. [7] J. B. Hagen, “Noise parameter transformations for three-terminal amplifiers,” IEEE Trans. Microw. Theory Tech., vol. 38, no. 3, pp. 319–321, Mar. 1990. [8] J. Gao, C. L. Law, H. Wang, S. Aditya, and G. Boeck, “A new method for PHEMT noise parameter determination based on 50- noise measurement system,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 10, pp. 2079–2089, Oct. 2003. [9] A. Cappy, “Noise modeling and measurement techniques,” IEEE Trans. Microwave Theory Tech., vol. 36, no. 1, pp. 1–10, Jan. 1988.

Jianjun Gao was born in Hebei province, China, in 1968. He received the B.Eng. and Ph.D. degrees from Tsinghua University, Tsinghua, China, in 1991 and 1999, respectively, and the M.Eng. degree from the Hebei Semiconductor Research Institute, Hebei, China, in 1994. From 1999 to 2001, he was a Post-Doctoral Research Fellow with the Microelectronics Research and Development Center, Chinese Academy of Sciences, where he developed a pHEMT optical modulator driver. In 2001, he joined the School of Electrical and Electronic Engineering, Nanyang Technological University (NTU), Singapore, as a Research Fellow involved with semiconductor device modeling and on-wafer measurement. In 2003, he joined the Institute for High-Frequency and Semiconductor System Technologies, Berlin University of Technology, Berlin, Germany, as a Research Associate involved with InP HBT modeling and circuit design for high-speed optical communication. Since 2004, he has been a Full Professor with the Institute of RF and Opto-Electronic Integrated Circuits, Southeast University, Nanjing, China. His main areas of research are characterization, modeling, and on-wafer measurement of microwave semiconductor devices, opto-electronics devices, and high-speed integrated circuits for optical communication.

Georg Boeck (M’93–SM’00) was born in Wertingen, Germany, in 1951. He received the Dipl.-Ing. degree in electrical engineering and Doctoral degree from the Technische Universität Berlin, Berlin, Germany, in 1977 and 1984, respectively. In 1984, he joined the Siemens Research Laboratories, Munich, Germany, where his research areas were fiber optics and GaAs electronics. From 1988 to 1991, he was a Full Professor of electronic devices and circuits with the Fachhochschule Regensburg, Regensburg, Germany. Since 1991, he has been a Full Professor of microwave engineering with the Technische Universität Berlin, Berlin, Germany. His main areas of research are analysis, characterization, modeling and design of planar passive structures, microwave semiconductor devices and integrated circuits (microwave integrated circuits (MICs) and MMICs) up to the millimeter-wave range.

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Millimeter-Wave Substrate Integrated Waveguides and Filters in Photoimageable Thick-Film Technology Daniel Stephens, Paul R. Young, Senior Member, IEEE, and Ian D. Robertson, Senior Member, IEEE

Abstract—This paper presents the design and fabrication of substrate-integrated waveguides and filters for use at millimeter-wave frequencies. The components described are fabricated using photoimageable thick-film materials. Measurements on - and -band waveguide-to-microstrip transitions are presented. Losses resulting from the reduced-height nature of the waveguides are extracted from thru-relect-line calibration standards illustrating the effect of current losses in the broadwalls of the waveguides. The design of fourth-order 0.01-dB ripple Chebyshev resonant cavity filters operating at -, -, -, and -band is presented. The measured results of all components are in excellent agreement with simulated predictions. Index Terms—Ceramics, filters, millimeter waves, substrate integrated waveguides (SIWs), thick-film technology.

I. INTRODUCTION

I

N RECENT years, much interest has developed in systems operating in the millimeter-wave regime. This interest is largely based on the unique spectral characteristics of the atmosphere, affecting the propagation of electromagnetic waves above 10 GHz [1]. The presence of absorption lines and transmission windows within the atmosphere has given way to the development of systems used in a wide variety of applications. These include remote sensing of the H absorption lines (occurring at 22 and 183 GHz) for the monitoring of climate change, passive millimeter-wave imaging (at 94 GHz), capable of all-weather operation, and also the development of wireless communication networks (at 60 GHz), which take advantage of the increased atmospheric attenuation (10–15 dB km ) of the O line to provide for frequency reuse. A practical solution for millimeter-wave transceiver modules required for these applications consists of a system-in-package (SiP) approach with Si, GaAs, and InP devices integrated on a single substrate with embedded passive components. However, the use of conventional planar transmission lines within a SiP results in significant attenuation when operating above 100 GHz. Considerable research has thus been reported on methods of fabricating rectangular waveguides (RWGs) using photolithographical methods. RWGs have excellent power-handling capabilities, are well understood, and are immune from radiation loss and crosstalk. In addition, the attenuation and dispersion characteristics of RWGs can be accurately modeled mathematically or with

Manuscript received May 3, 2005. D. Stephens is with the Microwave Systems Research Group, University of Surrey, Guildford, Surrey GU2 7XH, U.K. P. R. Young is with the Department of Electronics, University of Kent, Canterbury, Kent CT2 7NT, U.K. (e-mail: [email protected]). I. D Robertson is with the Institute of Microwaves and Photonics, The University of Leeds, Leeds LS2 9JT, U.K. (e-mail: [email protected]). Digital Object Identifier 10.1109/TMTT.2005.859862

the aid of three-dimensional electromagnetic simulators. Moreover, a vast array of waveguide components and designs are available to the microwave engineer, including high- filters, compact antennas, power combiners, and broad-band couplers. However, before the advantages of this medium can be drawn upon, the manufacture of these waveguides needs to be accomplished with sufficient accuracy so as to allow for operation at millimeter-wave frequencies. At very high frequencies, micromachining has been successfully used to fabricate hollow waveguides [2], [3]. For ease of integration with other components, the substrate integrated waveguide (SIW) has attracted much interest. Here, the waveguide is dielectric filled and embedded into a substrate, which can also be used for microstrip and CPW lines, lumped elements, and as a chip carrier for integrated circuits. Hence, the SIW is a very promising approach for realizing a millimeter-wave multichip module [4]. The methods that have been used to realize SIWs include multilayer GaAs monolithic microwave integrated circuit (MMIC) [5], low-temperature co-fired ceramic (LTCC) [6], standard microwave integrated circuit (MIC) [7], and thick-film [8] technology. It has been shown that a wide range of components can be integrated including filters [9]–[13], antennas [14], [15], phase shifters [16], and circulators [17]. In LTCC technology, the continuous sidewalls are usually replaced with a periodic row of metallized via-holes, which is more compatible with the processing of the tapes. The propagation and leakage characteristics of a guide with this structure have been studied [18]. The size of the via-holes is quite large in LTCC and this limits the technique currently to the lower end of the millimeter-wave spectrum. The minimum via-to-via spacing requirement results in increased radiation losses at higher millimeter-wave frequencies. Multilayer GaAs MMIC fabrication allows solid sidewalls to be formed. However, the waveguides in [5] suffered from very high losses due to fabrication limitations on the waveguide height. The multilayer thick-film approach to fabrication of SIWs has the advantage that continuous metal sidewalls can be fabricated and low-loss dielectric pastes are available, including materials from Heraeus [19], Dupont [20], and Hibridas [21]. The Hibridas photoimageable technology has previously been shown to be suitable for realizing SIWs up to at least 100 GHz with excellent dimensional tolerances and low dielectric loss. Transitions, filters, and antennas have previously been reported using this technology [8], [12], [15]. With thick-film fabrication, the challenge is that highly three-dimensional structures are difficult to print with high resolution and, thus, waveguide height is limited. This paper, therefore, describes the fabrication process and its limitations, the propagation characteristics

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Fig. 2. Cross section of a thick-film waveguide. TABLE I THICK-FILM WAVEGUIDE WIDTHS FOR STANDARD BANDS

Fig. 1. Photoimageable thick-film process.

of the reduced-height waveguides, and demonstrates the level of performance that can be achieved by presenting results for SIW bandpass filters operating up to 180 GHz. II. FABRICATION OF THICK-FILM WAVEGUIDES Traditionally, thick-film technology has been limited to applications below 20 GHz due to the manufacturing tolerances of the process. Recently, however, the introduction of photoimageable thick-film materials has allowed the technology to produce patterns of much higher resolution, supporting the fabrication of low-loss circuits in the millimeter-wave regime. In this paper, the Hibridas HC4700 silver conductor and HD1000 diphotoimageable pastes are used to fabricate electric millimeter-wave SIWs and resonant cavity filters. The fabrication process is summarized in Fig. 1. First, a uniform metal layer is printed on an alumina substrate [see Fig. 1(i)]. This forms the lower broadwall of the waveguide and the ground plane for the microstrip transition. A dielectric layer is then printed [see Fig. 1(ii)] and photoimaged [see Fig. 1(iii)], forming the trenches required for the waveguide sidewalls and vias for the microstrip–coplanar waveguide (CPW) probe transition. Each dielectric print has a thickness of 10 m (fired), and repeated prints are used to achieve the required guide height. Finally, a conductor layer is printed and photoimaged to form the upper broadwall of the waveguides and the microstrip feed lines [see Fig. 1(iv)]. After printing and imaging, each layer is dried and fired prior to the processing of subsequent layers. Fig. 2 shows a cross section of the fabricated guide structure with the upper and lower broadwalls, and trench vias used to form the waveguide sidewalls. Each trench has a width of

Fig. 3. Fabricated calibration.

W -band waveguide–microstrip transition set for TRL

100 m. The use of trench vias as waveguide sidewalls provides complete isolation and allows for operation at higher millimeter-wave frequencies. The apparent irregularity of the sidewalls is mostly attributed to the polishing performed after the slicing of the substrate. III. WAVEGUIDE CHARACTERISTICS AND TRANSITION DESIGN The Hibridas dielectric paste has a nominal relative permittivity of 8. Table I shows the required waveguide width calculated for the standard -, -, -, and -bands. To facilitate probing of the waveguide, a simple microstrip-to-waveguide transition like that proposed by Deslandes and Wu [7] is used. Use of a coplanar transition has previously been investigated [8]. However, full-wave modeling performed on both structures indicate the microstrip transition to be better suited, as it ensures minimal disruption to the field propagating between the two media. A two-section Chebyshev transformer is employed to alleviate the impedance mismatch between the waveguide and microstrip feed line. Simulations indicate this gives an improvement of 14 dB in return-loss performance compared to a direct transition from microstrip to SIW. Fig. 3 shows a microphotograph of a set of -band back-to-back transitions designed to allow thru-reflect line (TRL) calibration.

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is the dielectric constant of the where is the frequency, is the loss tangent guide, is the free-space wavenumber, cutoff frequency. The conof the dielectric, and is the ductive losses of RWG can be calculated using the ideal modal currents in the waveguide walls and the surface resistivity . By considering the total power flow, the conductive attenuation constant is obtained [22] as follows: (2)

where and are the waveguide width and height, respectively. The total attenuation is simply the sum of the dielectric and . conductive losses and, for thick-film waveguides, , Since then the second term in (2) is much less than the first. Therefore, the conduction losses are approximately inversely proportional to as follows: (3)

Fig. 4. Waveguide back-to-back transition measurements. (a) (b) W -band.

V -band.

To first measure the characteristics of the back-to-back transitions, a broad-band line-reflect-reflect-match (LRRM) calibration using a Cascade Microtech impedance standard substrate (ISS) and Wincal was performed. As shown in Fig. 4, the measurements are in good agreement with simulated predictions, and show dominant mode propagation beginning at 39 and 59 GHz for the - and -band waveguides. An insertion loss of 1 dB is achieved for the -band back-to-back transition and 1.5 dB for the -band one, which have 6.64- and 4.35-mm waveguide lengths, respectively. Return-loss measurements indicate optimal matching at midband for both guides, as was intended with the design of the Chebyshev matching transformer. The bandwidth of the transition is found to be limited by the differing impedance-frequency characteristics of microstrip and waveguide transmission lines. IV. WAVEGUIDE LOSSES A. Loss Mechanisms Losses in a RWG are comprised of dielectric and conductor losses. The dielectric losses of RWG can be calculated by allowing the dielectric constant to become complex. This gives an attenuation constant [22] (1)

For a waveguide of conventional height, the loss tangent of the dielectric filling has a significant effect on the overall losses. is independent of , we find that if the height However, since of the guide is small, as is the case in the current implementation, then the conductive losses dominate. In addition, at millimeter wavelengths, conductor losses increase significantly due to the effect of surface roughness. Conductor surface roughness is a characteristic engineered into thick-film materials to promote adhesion between layers. The processed conductor layer was found to have an rms surface roughness of 0.27 m. Another contributing factor to conductor losses is found in currents flowing down the sidewalls of the waveguide. As the waveguide is built using a multilayer technique, misalignment of successive layers can result in an increase in current losses. However, due , losses in the to the reduced height of the waveguides upper and lower broadwalls tend to dominate and current losses in the sidewalls do not overtly affect the loss characteristic of the waveguides. B. Measured Results To precisely measure the propagation characteristics of the fabricated waveguides, the probe–microstrip and the microstrip–SIW transitions are calibrated out. This is achieved by means of the multiline TRL calibration technique [23]. The calibration standards fabricated include a thru, reflect, and three at the upper, line lengths, each of which correspond to lower, and middle frequencies of the respective guide bands. The chosen line lengths improve the accuracy of the algorithms used in the MultiCal software, as the calibration technique is to an extent frequency dependant. Measurements were taken on waveguides of height 30 and 60 m for both - and -bands. The measurements in Fig. 5 illustrate the significant reduction in attenuation achieved by printing more layers to increase the waveguide height: In going from a 30- m height to 60 m, the

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Fig. 5. Measured attenuation (in decibels per millimeter) for waveguides.

V - and W -band

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Fig. 7. Relative permittivity extracted from the measured relative phase constant versus frequency.

Fig. 8.

H -plane offset filter with equivalent circuit. TABLE II DESIGN AND FABRICATED DIMENSIONS FOR THE 94-GHz FILTER

Fig. 6.

Measured and calculated relative phase constant.

average loss improves from 0.32 to 0.1 dB mm in the -band and from 0.44 to 0.2 dB mm in the -band. By deducting these measured waveguide losses from the back-to-back transition data in Section III, a single microstrip–SIW transition is found to have a loss of only 0.27 dB in the -band and 0.49 dB in the -band. for both - and -band The relative phase constant waveguides is extracted from the calibrated TRL data and is in excellent agreement with theoretical predictions, as shown in Fig. 6. For the calculated values, a relative permittivity of 7.87 was used, based on previous results. The graph demonstrates that the dispersion characteristics of the photoimageable thick-film SIWs can be predicted accurately, even at millimeter wavelengths while using standard waveguide equations. Fig. 7 shows the value of extracted from the measured relative phase constant by manipulating the standard waveguide expression to give (4) The estimated uncertainty in this data is approximately 0.2 based on 10- m uncertainty in the length and width for measurements.

V. WAVEGUIDE FILTERS A. Design The filters presented in this paper are based on an inverter-coupled filter design [24], [25]. Series resonators coupled by appropriate shunt susceptances are used to realize a bandpass-filter characteristic. The series resonators are readily formed using sections of waveguide transmission lines, and -plane offsets are used to realize required shunt susceptances. The -plane offset is chosen due to its ease of manufacture using the thick-film process. The offset can be modeled as at the middle of a transmission line a shunt susceptance of length (Fig. 8). The additional length introduced by the offset is compensated for in the adjacent resonant cavity sections. To synthesize a Chebyshev filter response, the dimensions of the displaced waveguide junctions are calculated using the technique of Hunter [26]. The calculated values obtained for each displaced junction is further characterized using full-wave analysis by modeling the structure in HFSS. The complete

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Fig. 9. Microphotograph of 94-GHz filter with RWG-to-microstrip transitions (total length: 4.34 mm).

Fig. 10.

Fig. 12.

Measured and modeled 150-GHz filter response.

Fig. 13.

Measured and modeled 180-GHz filter response.

Measured and modeled 60-GHz filter response.

TABLE III FILTER PERFORMANCE

B. Measured Results Fig. 11.

Measured and modeled 94-GHz filter response.

structure is subsequently simulated to ensure the filter response is in accordance with the design specification. Fourth-order Chebyshev filters of 0.01-dB ripple, centred at 60, 94, 150, and 180 GHz have been designed and tested. Each filter has been realized using -, -, -, and -band waveguides designed using the procedure described in Section III with a waveguide height of 60 m. Table II shows the dimensional values for the aperture width and length of the cavities for the 94-GHz filter. Fig. 9 shows a microphotograph of the 94-GHz filter. As can be seen from Table II, differences between the designed and fabricated dimensions are minimal. The process, therefore, allows highly accurate designs to be manufactured, which is essential for filter structures operating in the millimeter-wave band.

Measurements performed on the filters used the TRL calibration and deembedding technique mentioned earlier. Removing the losses introduced by the probes and microstrip transitions gives an accurate indication of the passband insertion loss of the fabricated filters. As shown in Figs. 10–13, all the filter measurements are in good agreement with simulated predictions. The upper stopband response degradation observed in the 180-GHz filter is an artifact of the calibration procedure, and not a characteristic of the filter response. Calibration at 180 GHz was extremely challenging with the probe overtravel (skate) distance being comparable to the physical difference in length of the calibration lines. A slight deterioration in filter performance is believed to be due to the effect of dimensional tolerances, which are more significant due to the smaller operating wavelength. The insertion loss and filter bandwidth of the filters are summarized in Table III. The measured response of the filters show performance at least equal to the results obtained in [27]–[29],

STEPHENS et al.: MILLIMETER-WAVE SIWs AND FILTERS IN PHOTOIMAGEABLE THICK-FILM TECHNOLOGY

all of which utilized fabrication procedures far more complex than the one adopted here.

VI. CONCLUSION This paper has demonstrated the potential of thick-film processing for the fabrication of hybrid circuits based on planar microstrip and miniaturized waveguide components. The loss and dispersion characteristics of the fabricated waveguide structures operating up to 110 GHz have been presented. Resonant cavity filters operating at 60, 94, 150, and 180 GHz have been designed and fabricated. The measured responses of the filter characteristics are in good agreement with predicted values, proving that the precise dimensional requirements needed for millimeter-wave circuit design can be achieved with photoimageable thick-film materials. The straightforward fabrication technique, low-loss, and highly predictable performance of the components make this an attractive technology for economical millimeter-wave system development.

ACKNOWLEDGMENT The authors would like to acknowledge the use of MultiCal, which was provided by the National Institute of Standards and Technology (NIST), Boulder, CO.

REFERENCES [1] L. Yujiri, M. Schucri, and P. Moffs, “Passive millimeter-wave imaging,” IEEE Micro, pp. 39–50, Sep. 2003. [2] J. W. Digby, C. E. McIntosh, G. M. Parkhurst, J. W. Hadjiloucas, J. M. Chamberlain, R. D. Pollard, R. E. Miles, D. P. Steenson, N. J. Cronin, and S. R. Davies, “Fabrication and characterization of micromachined rectangular waveguide components for use at millimeter-wave and terahertz frequencies,” IEEE Trans. Microw. Theory Tech., vol. 48, no. 8, pp. 1293–1302, Aug. 2000. [3] W. R. McGrath, C. Walker, M. Yap, and Y. Tai, “Silicon micromachined waveguides for millimeter-wave and submillimeter-wave frequencies,” IEEE Microw. Guided Wave Lett., vol. 3, no. 3, pp. 61–63, Mar. 1993. [4] S. Lucyszyn, “Terahertz multi-chip module (T-MCM) technology for the 21st Century?,” in IEE Multi-Chip Modules RFICs Colloq., London, U.K., May 1998, pp. 6/1–6/8. [5] S. Lucyszyn, D. Budimir, Q. H. Wang, and I. D. Robertson, “Design of compact monolithic dielectric-filled metal-pipe rectangular waveguides for millimeter-wave applications,” Proc. Inst. Elect. Eng., pt. H, vol. 143, no. 5, pp. 451–453, Oct. 1996. [6] H. Uchimura, T. Takenoshita, and M. Fujii, “Development of a ‘laminated’ waveguide,” IEEE Trans. Microw. Theory Tech., vol. 46, no. 12, pp. 2437–2443, Dec. 1998. [7] D. Deslandes and K. Wu, “Integrated microstrip and rectangular waveguide in planar form,” IEEE Microw. Wireless Compon. Lett., vol. 11, no. 2, pp. 68–70, Feb. 2001. [8] M. S. Aftanasar, P. R. Young, I. D. Robertson, J. Minalgiene, and S. Lucyszyn, “Photoimageable thick-film millimeter-wave metal-pipe rectangular waveguides,” Electron. Lett., vol. 37, no. 18, pp. 1122–1123, Aug. 2001. [9] Y. Rong, A. Zaki, J. Gipprich, M. Hageman, and D. Stevens, “LTCC wide-band ridge-waveguide bandpass filters,” IEEE Trans. Microw. Theory Tech., vol. 47, no. 9, pp. 1836–1840, Sep. 1999. [10] D. Deslandes and K. Wu, “Single-substrate integration technique of planar circuits and waveguide filters,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 2, pp. 593–596, Feb. 2003. [11] Z. C. Hao, W. Hong, X. P. Chen, J. X. Chen, K. Wu, and T. J. Cui, “Multilayered substrate integrated waveguide (MSIW) elliptic filter,” IEEE Microw. Wireless Compon. Lett., vol. 15, no. 2, pp. 95–97, Feb. 2005.

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[12] M. S. Aftanasar, P. R. Young, and I. D. Robertson, “Rectangular waveguide filters using photoimageable thick-film processing,” in Proc. 32nd Eur. Microwave Conf., Milan, Italy, Sep. 2002. [CD ROM]. [13] D. Stephens, P. R. Young, and I. D. Robertson, “Design and Characterization of 180 GHz filters in photoimageable thick film technology,” IEEE MTT-S Int. Microwave Symp. Dig., Jun. 2005. [CD ROM]. [14] L. Yan, W. Hong, G. Hua, J. Chen, K. Wu, and T. J. Cui, “Simulation and experiment on SIW slot array antennas,” IEEE Microw. Wireless Compon. Lett., vol. 14, no. 9, pp. 446–448, Sep. 2004. [15] D. Stephens, P. R. Young, and I. D. Robertson, “ -band substrate integrated waveguide slot antenna,” Electron. Lett., vol. 41, no. 4, pp. 165–167, Feb. 17, 2005. [16] W. Che, E. K. N. Yung, and K. Wu, “Millimeter-wave ferrite phase shifter in substrate integrated waveguide (SIW),” in IEEE AP-S Int. Symp. Dig., vol. 4, Jun. 2003, pp. 887–890. [17] W. D’Orazio, K. Wu, and J. Helszajn, “A substrate integrated waveguide degree-2 circulator,” IEEE Microw. Wireless Compon. Lett., vol. 14, no. 5, pp. 207–209, May 2004. [18] F. Xu and K. Wu, “Guided-wave and leakage characteristics of substrate integrated waveguide,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 1, pp. 66–73, Jan. 2005. [19] P. Barnwell and J. Wood, “A novel thick film on ceramic MCM technology offering MCM-D performance,” in IEEE 6th Int. Multichip Modules Conf., Apr. 1997, pp. 48–52. [20] B. Dziurdzia and S. Nowak, “FODEL(R) photoimageable technology in microwave applications,” in IEEE 12th Int. Microwaves and Radar Conf., vol. 2, May 1998, pp. 445–450. [21] S. Muckett and J. Minalgene, “Advances in superfine line thick film materials & processing,” in Microelectronics and Packaging Int. Symp., 2002, pp. 20–22. [22] S. Ramo, J. R. Whinnery, and T. van Duzer, Fields and Waves in Communications Electronics, 3rd ed. New York: Wiley, 1994. [23] R. Marks, “A multiline method of network analyzer calibration,” IEEE Trans. Microw. Theory Tech., vol. 39, no. 7, pp. 1205–1215, Jul. 1991. [24] G. Matthaei, L. Young, and E. M. T. Jones, Microwave Filters, Impedance Matching Networks and Coupling Structures. Norwood, MA: Artech House, 1980. [25] I. C. Hunter, Theory and Design of Microwave Filters, ser. Electromagnetic Wave. London, U.K.: IEE Press, 2001. [26] J. D. Hunter, “The displaced rectangular waveguide junction and its use as adjustable reference reflection,” IEEE Trans. Microw. Theory Tech., vol. MTT-32, no. 4, pp. 387–394, Apr. 1984. [27] S. V. Robertson, L. P. B. Katehi, and G. M. Rebeiz, “Micromachined -band filters,” IEEE Trans. Microw. Theory Tech., vol. 44, no. 4, pp. 598–606, Apr. 1996. [28] E. Rius, G. Prigent, H. Happy, G. Dambrine, S. Boret, and A. Cappy, “Wide- and narrow-band bandpass coplanar filters in the -frequency band,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 3, pp. 784–791, Mar. 2003. [29] G. Pringent, E. Rius, F. L. Pennec, S. L. Maguer, C. Quendo, G. Six, and H. Happy, “Design of narrow-band DBR planar filters in Si-BCB technology for millimeter-wave applications,” IEEE Trans. Microw. Theory Tech., vol. 52, no. 3, pp. 1045–1051, Mar. 2004.

W

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Daniel Stephens received the B.Eng. degree in electrical and electronic engineering from Heriot-Watt University, Edinburgh, U.K., in 2000, and the Ph.D. degree from the Advanced Technology Institute (ATI), University of Surrey, Surrey U.K, in 2005. His doctoral research focused on the development of low-cost millimeter-wave transmission lines and passive components for use in emerging technologies within the field. This concept was successfully demonstrated by the development of waveguides, filters and antennas all operating at millimeter-wave frequencies. His current research interests include the development of low-loss filters and the accurate modeling of transmission lines.

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IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 53, NO. 12, DECEMBER 2005

Paul R. Young (M’00–SM’05) received the B.Eng. and Ph.D. degrees in electronic engineering from the University of Kent, Canterbury, Kent, U.K., in 1994 and 1998, respectively. From 1997 to 1999, he was a Research Scientist with the National Physical Laboratory, Middlesex, U.K., where he developed standards for RF and microwave impedance measurement. He then spent a year as a Research Fellow with the University of Surrey before becoming a Lecturer with the University of Kent in 2001. He has authored or coauthored over 40 papers in journals and conference proceedings. His research interests include guided-wave structures, electromagnetic bandgaps, and microwave measurement techniques.

Ian D. Robertson (M’96–SM’05) received the B.Sc. (Eng.) and Ph.D. degrees from King’s College London, London, U.K., in 1984 and 1990, respectively. From 1984 to 1986, he was with the Monolithic Microwave Integrated Circuit (MMIC) Research Group, Plessey Research (Caswell) Ltd. Since then, he has held academic posts with King’s College London and the University of Surrey. In June 2004, he became the Centenary Chair in Microwave and Millimeter-Wave Circuits with The University of Leeds, Leeds, U.K. He is currently the Honorary Editor of the IEE Proceedings—Microwaves, Antennas & Propagation. He edited MMIC Design (London, U.K.: IEE, 1995) and coedited RFIC & MMIC Design and Technology (IEE: London, U.K., 2001, 2nd ed.). He has authored or coauthored over 330 papers in the areas of MIC and MMIC design. Dr. Robertson has organized numerous colloquia, workshops, and short courses for both the Institution of Electrical Engineers (IEE), U.K., and the IEEE.

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Digital Object Identifier 10.1109/TMTT.2005.859922

Digital Object Identifier 10.1109/TMTT.2005.862201

2005 Index IEEE Transactions on Microwave Theory and Techniques Vol. 53 This index covers all technical items — papers, correspondence, reviews, etc. — that appeared in this periodical during 2005, and items from previous years that were commented upon or corrected in 2005. Departments and other items may also be covered if they have been judged to have archival value. The Author Index contains the primary entry for each item, listed under the first author's name. The primary entry includes the coauthors’ names, the title of the paper or other item, and its location, specified by the publication abbreviation, year, month, and inclusive pagination. The Subject Index contains entries describing the item under all appropriate subject headings, plus the first author’s name, the publication abbreviation, month, and year, and inclusive pages. Subject cross-references are included to assist in finding items of interest. Note that the item title is found only under the primary entry in the Author Index.

AUTHOR INDEX

A Aaen, P.H., J.A. Pla, and C.A. Balanis. On the development of CAD techniques suitable for the design of high power RF transistors; T-MTT Oct 05 3067-3074 Abubakar, A., see Semenov, S.Y., T-MTT Jul 05 2284-2294 Achar, R., see Nakhla, N.M., T-MTT Nov 05 3520-3530 Adahl, A., see Lepine, F., T-MTT Jun 05 2007-2012 Adamiecki, A., see Sinsky, J.H., T-MTT Jan 05 152-160 Adams, G.G., see Johnson, J., T-MTT Nov 05 3615-3620 Adler, E.D., M.C. Calcatera, J.-F. Luy, W.D. Palmer, and D.S. Purdy. Guest editorial [special issue intro. multifunctional RF systems]; T-MTT Mar 05 1005-1008 Ahmed, T., E. Gad, and M.C.E. Yagoub. An adjoint-based approach to computing time-domain sensitivity of multiport systems described by reduced-order models; T-MTT Nov 05 3538-3547 Ahn, D., see Younkyu Chung, T-MTT Feb 05 739-746 Ahn, D., see Jong-Sik Lim, T-MTT Aug 05 2539-2545 Aikio, J.P., and T. Rahkonen. Detailed distortion analysis technique based on simulated large-signal voltage and current spectra; T-MTT Oct 05 30573066 Aja, B., E. Artal, L. de la Fuente, J.P. Pascual, A. Mediavilla, N. Roddis, D. Kettle, W.F. Winder, L.Pi. Cara, and P. de Paco. Very low-noise differential radiometer at 30 GHz for the PLANCK LFI; T-MTT Jun 05 2050-2062 Akan, V., and E. Yazgan. Quasi-static solutions of multilayer elliptical, cylindrical coplanar striplines and multilayer coplanar striplines with finite dielectric dimensions - Asymmetrical case; T-MTT Dec 05 3681-3686 Aksoy, S., M. Antyufeyeva, E. Basaran, A.A. Ergin, and O.A. Tretyakov. Time-domain cavity oscillations supported by a temporally dispersive dielectric; T-MTT Aug 05 2465-2471 Ala-Laurinaho, J., see Koskinen, T., T-MTT Sep 05 2999-3006 Al-Attar, T., and T.H. Lee. Monolithic integrated millimeter-wave IMPATT transmitter in standard CMOS technology; T-MTT Nov 05 3557-3561 Albasha, L., see Clifton, J.C., T-MTT Jun 05 2251-2258 Aldana, R., see Ioakeimidi, K., T-MTT Jan 05 336-342 Alexandrou, S., see Jingjing Zhang, T-MTT Nov 05 3281-3287 Allen, P.E., see Srirattana, N., T-MTT Mar 05 852-860 Allshouse, G.E., see Mazumder, S., T-MTT Mar 05 1065-1071 Allstot, D.J., see Sher Jiun Fang, T-MTT Feb 05 478-487 Alonso, J.I., see Sanchez-Renedo, M., T-MTT Jan 05 191-199 Alonso, J.I., see Gomez-Garcia, R., T-MTT Oct 05 3221-3229 Alter, J.J., see Tavik, G.C., T-MTT Mar 05 1009-1020 Amari, S., see Ofli, E., T-MTT Mar 05 843-851 Amari, S., and U. Rosenberg. New in-line dual- and triple-mode cavity filters with nonresonating nodes; T-MTT Apr 05 1272-1279 Amari, S., and G. Macchiarella. Synthesis of inline filters with arbitrarily placed attenuation poles by using nonresonating nodes; T-MTT Oct 05 3075-3081

Amari, S., and U. Rosenberg. Characteristics of cross (bypass) coupling through higher/lower order modes and their applications in elliptic filter design; T-MTT Oct 05 3135-3141 Amat, E., see Garcia-Garcia, J., T-MTT Jun 05 1997-2006 Amor-Martin, D., see Gomez-Garcia, R., T-MTT Oct 05 3221-3229 Analui, B., J.F. Buckwalter, and A. Hajimiri. Data-dependent jitter in serial communications; T-MTT Nov 05 3388-3397 Anderson, J.P., see Shapiro, M.A., T-MTT Aug 05 2610-2615 Ando, Y., see Inoue, T., T-MTT Jan 05 74-80 Andreani, P., see Strandberg, R., T-MTT Feb 05 660-669 Andrews, J.M., see Qingqing Liang, T-MTT May 05 1745-1755 Andrievski, V.F., see Malyshev, S.A., T-MTT Feb 05 439-443 Anh Do Manh, see Choon Beng Sia, T-MTT Sep 05 3035-3044 Anh-Vu Pham, see Chen, A.C., T-MTT Nov 05 3648-3655 Aniel, F., see Sirbu, M., T-MTT Sep 05 2991-2998 Antonino-Daviu, E., see Valero-Nogueira, A., T-MTT Mar 05 868-873 Antyufeyeva, M., see Aksoy, S., T-MTT Aug 05 2465-2471 Aoki, I., see Seungwoo Kim, T-MTT Jan 05 380-388 Aoki, Y., and K. Honjo. Q-factor definition and evaluation for spiral inductors fabricated using wafer-level CSP technology; T-MTT Oct 05 3178-3184 Aparin, V., and L.E. Larson. Modified derivative superposition method for linearizing FET low-noise amplifiers; T-MTT Feb 05 571-581 Arai, K.I., see Suzuki, E., T-MTT Feb 05 696-701 Arakawa, S., see Suzuki, E., T-MTT Feb 05 696-701 Araneo, R., S. Barmada, S. Celozzi, and M. Raugi. Two-port equivalent of PCB discontinuities in the wavelet domain; T-MTT Mar 05 907-918 Arndt, F. Comments on "A shorted waveguide-stub coupling mechanism for narrow-band multimode coupled resonator filters"; T-MTT Jan 05 414-415 Arndt, F., see Catina, V., T-MTT Nov 05 3562-3567 Artal, E., see Aja, B., T-MTT Jun 05 2050-2062 Aryanfar, F., and K. Sarabandi. Characterization of semilumped CPW elements for Millimeter-wave filter design; T-MTT Apr 05 1288-1293 Asbeck, P.M., see O'Sullivan, T., T-MTT Jan 05 106-114 Asbeck, P.M., see Tsai-Pi Hung, T-MTT Jan 05 144-151 Asbeck, P.M., see Junxiong Deng, T-MTT Feb 05 529-537 Asbeck, P.M., see Dongjiang Qiao, T-MTT Mar 05 1089-1095 Asbeck, P.M., see Feipeng Wang, T-MTT Apr 05 1244-1255 Au Kong Jin, see Grzegorczyk, T.M., T-MTT Apr 05 1443-1450 Au Kong Jin, see Yan Li, T-MTT Apr 05 1522-1526 Au Kong Jin, see Grzegorczyk, T.M., T-MTT Sep 05 2956-2967 Aykut Dengi, see Feng Ling, T-MTT Jan 05 264-273 B Baccarelli, P., P. Burghignoli, F. Frezza, A. Galli, P. Lampariello, G. Lovat, and S. Paulotto. Effects of leaky-wave propagation in metamaterial grounded slabs excited by a dipole source; T-MTT Jan 05 32-44 Baccarelli, P., P. Burghignoli, F. Frezza, A. Galli, P. Lampariello, G. Lovat, and S. Paulotto. Fundamental modal properties of surface waves on metamaterial grounded slabs; T-MTT Apr 05 1431-1442 Bachtold, W., see Treyer, D.M., T-MTT Dec 05 3806-3816 Baek Sang-Hyun, see Il-Joo Cho, T-MTT Jul 05 2450-2457 Baena, J.D., J. Bonache, F. Martin, R.M. Sillero, F. Falcone, T. Lopetegi, M.A.G. Laso, J. Garcia-Garcia, I. Gil, M.F. Portillo, and M. Sorolla. Equivalent-circuit models for split-ring resonators and complementary split-ring resonators coupled to planar transmission lines; T-MTT Apr 05 1451-1461 Baena, J.D., see Garcia-Garcia, J., T-MTT Jun 05 1997-2006 Baginski, M.E., D.L. Faircloth, and M.D. Deshpande. Comparison of two optimization techniques for the estimation of complex permittivities of multilayered structures using waveguide measurements; T-MTT Oct 05 3251-3259 Bahmani, F., see Mishra, C., T-MTT Dec 05 3744-3756 Baillargeat, D., see Valois, R., T-MTT Jun 05 2026-2032 Bairavasubramanian, R., see Pinel, S., T-MTT May 05 1707-1712 Bairavasubramanian, R., see Il Kwon Kim, T-MTT Sep 05 2943-2948 Bajaj, V.S., see Woskov, P.P., T-MTT Jun 05 1863-1869

IEEE T-MTT 2005 INDEX — 2 Bajon, D., see Wane, S., T-MTT Jan 05 200-214 Baker, A., see Semouchkina, E., T-MTT Feb 05 644-652 Bakr, M.H., see Bandler, J.W., T-MTT Sep 05 2801-2811 Balalem, A., see Menzel, W., T-MTT Oct 05 3230-3237 Balanis, C.A., see Aaen, P.H., T-MTT Oct 05 3067-3074 Ball, J.A.R., see Wells, C.G., T-MTT Oct 05 3169-3177 Bandler, J.W., A.S. Mohamed, and M.H. Bakr. TLM-based modeling and design exploiting space mapping; T-MTT Sep 05 2801-2811 Ban-Leong Ooi, see Boon Tiong Tan, T-MTT Jan 05 343-348 Ban-Leong Ooi Compact EBG in-phase hybrid-ring equal power divider; TMTT Jul 05 2329-2334 Baralis, M., see Virone, G., T-MTT Mar 05 888-894 Barmada, S., see Araneo, R., T-MTT Mar 05 907-918 Barnes, F.S., see Jaeheung Kim, T-MTT Aug 05 2622-2627 Basaran, E., see Aksoy, S., T-MTT Aug 05 2465-2471 Basaran, U., N. Wieser, G. Feiler, and M. Berroth. Small-signal and highfrequency noise modeling of SiGe HBTs; T-MTT Mar 05 919-928 Batra, A., see Mishra, C., T-MTT Dec 05 3744-3756 Battat, J., R. Blundell, T.R. Hunter, R. Kimberk, P.S. Leiker, and C.-Y.E. Tong. Gain stabilization of a submillimeter SIS heterodyne receiver; TMTT Jan 05 389-395 Baudrand, H., see Wane, S., T-MTT Jan 05 200-214 Baumann, D., C. Fumeaux, and R. Vahldieck. Field-based scattering-matrix extraction scheme for the FVTD method exploiting a flux-splitting algorithm; T-MTT Nov 05 3595-3605 Bayard, B., see Vincent, D., T-MTT Apr 05 1174-1180 Bayba, A., see Darwish, A.M., T-MTT Sep 05 3052-3053 Bayba, A.J., see Darwish, A.M., T-MTT Jan 05 306-313 Beach, M.A., see Carey-Smith, B.E., T-MTT Feb 05 777-785 Beasley, C., see Popovic, D., T-MTT May 05 1713-1722 Beasley, C., see Popovic, D., T-MTT Sep 05 3053 Beerkens, R., see Dickson, T.O., T-MTT Jan 05 123-133 Behrmann, G.P., see Zhaolin Lu, T-MTT Apr 05 1362-1368 Belaid, M., and Ke Wu. Frequency multiplier using waveguide-based spatial power-combining architecture; T-MTT Apr 05 1124-1129 Belisle, C., see Qi, G., T-MTT Oct 05 3090-3097 Bellaouar, A., see Sher Jiun Fang, T-MTT Feb 05 478-487 Beng Hwee Ong, see Choon Beng Sia, T-MTT Sep 05 3035-3044 Beng Sia Choon, see Choon Beng Sia, T-MTT Sep 05 3035-3044 Benson, T.M., see Sewell, P., T-MTT Jun 05 1919-1928 Berdel, K., J.G. Rivas, P.H. Bolivar, P. de Maagt, and H. Kurz. Temperature dependence of the permittivity and loss tangent of high-permittivity materials at terahertz frequencies; T-MTT Apr 05 1266-1271 Bergveld, H.J., K.M.M. van Kaam, D.M.W. Leenaerts, K.J.P. Philips, A.W.P. Vaassen, and G. Wetkzer. A low-power highly digitized receiver for 2.4-GHz-band GFSK applications; T-MTT Feb 05 453-461 Bernard, S., see Puyal, V., T-MTT Apr 05 1338-1344 Bernardi, P., see Pisa, S., T-MTT Apr 05 1256-1265 Berroth, M., see Basaran, U., T-MTT Mar 05 919-928 Beukema, T., see Floyd, B.A., T-MTT Apr 05 1181-1188 Beyene, W.T., J. Feng, N. Cheng, and Xingchao Yuan. Performance analysis and model-to-hardware correlation of multigigahertz parallel bus with transmit pre-emphasis equalization; T-MTT Nov 05 3568-3577 Bharathan, K., see Lawson, W., T-MTT Jan 05 372-379 Bhattacharya, S., R. Senguttuvan, and A. Chatterjee. Production test technique for measuring BER of ultra-wideband (UWB) devices; T-MTT Nov 05 3474-3481 Bien, F., see Hur, Y., T-MTT Jan 05 246-255 Bien, F., see Moonkyun Maeng, T-MTT Nov 05 3509-3519 Bifeng Rong, see Spirito, M., T-MTT Jul 05 2340-2347 Bing-Jye Kuo, see Ming-Dou Ker, T-MTT Feb 05 582-589 Bing-Jye Kuo, see Ming-Dou Ker, T-MTT Sep 05 2672-2681 Binn Soo Chee, see Fusco, V., T-MTT Feb 05 730-738 Biondi, J.P., see Mazumder, S., T-MTT Mar 05 1065-1071 Birafane, A., and A.B. Kouki. Phase-only predistortion for LINC amplifiers with Chireix-outphasing combiners; T-MTT Jun 05 2240-2250 Biscontini, B., see Lorenz, P., T-MTT Nov 05 3631-3637 Biswas, A., see Srivastava, K.V., T-MTT Jun 05 1960-1967 Blasche, G., see Weimin Zhou, T-MTT Mar 05 929-933 Blayac, S., see Puyal, V., T-MTT Apr 05 1338-1344 Block, T.R., see Yeong-Chang Chou, T-MTT Nov 05 3398-3406 Blondy, P., see Pothier, A., T-MTT Jan 05 354-360 Blundell, R., see Battat, J., T-MTT Jan 05 389-395 Boeck, G., see Jianjun Gao, T-MTT Jan 05 330-335 Boeck, G., see Jianjun Gao, T-MTT Jan 05 417 + Check author entry for coauthors

Boeck, G., see Zinal, S., T-MTT Jun 05 1870-1874 Boeck, G., see Sayed, A., T-MTT Jul 05 2441-2449 Boeck, G., see Gao, J., T-MTT Dec 05 3825-3831 Bogdashov, A., G. Denisov, D. Lukovnikov, Y. Rodin, and J. Hirshfield. Kaband resonant ring for testing components for a high-gradient linear accelerator; T-MTT Oct 05 3152-3155 Boix, R.R., see Leon, G., T-MTT May 05 1739-1744 Bolivar, P.H., see Berdel, K., T-MTT Apr 05 1266-1271 Bolton, P.R., see Ioakeimidi, K., T-MTT Jan 05 336-342 Bonache, J., see Baena, J.D., T-MTT Apr 05 1451-1461 Bonache, J., see Garcia-Garcia, J., T-MTT Jun 05 1997-2006 Bonkee Kim, see Nam, I., T-MTT May 05 1662-1671 Boon Tiong Tan, Jong Jen Yu, S.T. Chew, M.-S. Leong, and Ban-Leong Ooi. A miniaturized dual-mode ring bandpass filter with a new perturbation; TMTT Jan 05 343-348 Booske, J.H., see Popovic, D., T-MTT May 05 1713-1722 Booske, J.H., see Popovic, D., T-MTT Sep 05 3053 Boret, S., see Dickson, T.O., T-MTT Jan 05 123-133 Boria-Esbert, V.E., see Taroncher, M., T-MTT Jun 05 2153-2163 Bornemann, J., see Rambabu, K., T-MTT May 05 1787-1791 Borras, S.C., see Ros, J.V.M., T-MTT Apr 05 1130-1142 Bosisio, R.G., see Xinyu Xu, T-MTT Jul 05 2267-2273 Bosisio, R.G., see Tatu, S.O., T-MTT Sep 05 2768-2776 Boumaiza, S., see Hammi, O., T-MTT May 05 1643-1652 Boumaiza, S., see Helaoui, M., T-MTT Jul 05 2355-2361 Boumaiza, S., see Sirois, J., T-MTT Sep 05 2875-2883 Boumaiza, S., see Taijun Liu, T-MTT Nov 05 3578-3587 Bourgeois, P.-Y., and V. Giordano. Simple model for the mode-splitting effect in whispering-gallery-mode resonators; T-MTT Oct 05 3185-3190 Bouttement, Y., see Tiemeijer, L.F., T-MTT Feb 05 723-729 Boyd, C.R., Jr. Impedance matching considerations for ferrite Faraday rotators; T-MTT Jul 05 2371-2374 Boyraz, O., see Han, Y., T-MTT Apr 05 1404-1408 Brace, C.L., P.F. Laeseke, D.W. van der Weide, and F.T. Lee, Jr. Microwave ablation with a triaxial antenna: results in ex vivo bovine liver; T-MTT Jan 05 215-220 Brandolini, M., P. Rossi, D. Manstretta, and F. Svelto. Toward multistandard mobile terminals - fully integrated receivers requirements and architectures; T-MTT Mar 05 1026-1038 Brandt, J., see Catina, V., T-MTT Nov 05 3562-3567 Brassard, G., see Sirois, J., T-MTT Sep 05 2875-2883 Braun, S., and P. Russer. A low-noise multiresolution high-dynamic ultrabroad-band time-domain EMI measurement system; T-MTT Nov 05 33543363 Brazil, T.J., see Wren, M., T-MTT May 05 1723-1731 Brennan, C., see Dautbegovic, E., T-MTT Feb 05 548-555 Bressan, M., see Mira, F., T-MTT Apr 05 1294-1303 Brewer, F., see Shigematsu, H., T-MTT Feb 05 472-477 Briso-Rodriguez, C., see Sanchez-Renedo, M., T-MTT Jan 05 191-199 Buchanan, N., see Fusco, V., T-MTT Feb 05 730-738 Bucholtz, F., see McDermitt, C.S., T-MTT Dec 05 3782-3787 Buck, T., see Schoebel, J., T-MTT Jun 05 1968-1975 Buck, T., see Morschbach, M., T-MTT Jun 05 2013-2018 Buckwalter, J.F., see Analui, B., T-MTT Nov 05 3388-3397 Bulyshev, A.E., see Semenov, S.Y., T-MTT Jul 05 2284-2294 Bumman Kim, see Seungwoo Kim, T-MTT Jan 05 380-388 Bumman Kim, see Jangheon Kim, T-MTT May 05 1802-1809 Bumman Kim, see Joongjin Nam, T-MTT Aug 05 2639-2644 Bunnjaweht, S., see Chongcheawchamnan, M., T-MTT Jul 05 2458-2462 Bunz, B., see Ghose, A., T-MTT Jun 05 2082-2087 Burghartz, J.N., see Spirito, M., T-MTT Jul 05 2340-2347 Burghignoli, P., see Baccarelli, P., T-MTT Jan 05 32-44 Burghignoli, P., see Baccarelli, P., T-MTT Apr 05 1431-1442 Button, T.W., see Holmes, J.E., T-MTT Jan 05 322-329 Byeong-Ha Park, see Young-Jin Kim, T-MTT Feb 05 606-613 Byoungjoong Kang, Jae-Hyoung Park, Jeiwon Cho, Kihyun Kwon, Sungkyu Lim, Jeonghoon Yoon, Changyul Cheon, Yong-Kweon Kim, and Youngwoo Kwon. Novel low-cost planar probes with broadside apertures for nondestructive dielectric measurement of biological materials at microwave frequencies; T-MTT Jan 05 134-143 Bystrom, M., see Li, Y., T-MTT Oct 05 3121-3129 C Cabedo-Fabres, M., see Valero-Nogueira, A., T-MTT Mar 05 868-873

IEEE T-MTT 2005 INDEX — 3 Caihua Chen, see Zhaolin Lu, T-MTT Apr 05 1362-1368 Calcatera, M.C., see Adler, E.D., T-MTT Mar 05 1005-1008 Calduch, M.T., see Ros, J.V.M., T-MTT Apr 05 1130-1142 Caloz, C., see Sungjoon Lim, T-MTT Jan 05 161-173 Caloz, C., see Horii, Y., T-MTT Apr 05 1527-1534 Camacho-Penalosa, C., see Esteban, J., T-MTT Apr 05 1506-1514 Cameron, R.J., Ming Yu, and Ying Wang. Direct-coupled microwave filters with single and dual stopbands; T-MTT Nov 05 3288-3297 Camiade, M., see Nallatamby, J.-C., T-MTT May 05 1601-1612 Cam Nguyen, see Jeongwoo Han, T-MTT Jun 05 1875-1882 Cangellaris, A.C., see Okhmatovski, V.I., T-MTT May 05 1829 Cao Meng, see Meng Cao, T-MTT Aug 05 2572-2579 Cara, L.Pi., see Aja, B., T-MTT Jun 05 2050-2062 Carchon, G.J., see Schoebel, J., T-MTT Jun 05 1968-1975 Carey-Smith, B.E., P.A. Warr, M.A. Beach, and T. Nesimoglu. Wide tuningrange planar filters using lumped-distributed coupled resonators; T-MTT Feb 05 777-785 Carvalho, N.B., see Pedro, J.C., T-MTT Jan 05 45-54 Carvalho, N.B., see Martins, J.P., T-MTT Jun 05 1982-1989 Cassivi, Y., see Duochuan Li, T-MTT Aug 05 2546-2551 Castejon, P.V., see Pereira, F.D.Q., T-MTT Jan 05 94-105 Catala-Civera, J.M., see Plaza-Gonzalez, P., T-MTT May 05 1699-1706 Catherinot, A., see Pothier, A., T-MTT Jan 05 354-360 Catina, V., F. Arndt, and J. Brandt. Hybrid surface integral-equation/modematching method for the analysis of dielectric loaded waveguide filters of arbitrary shape; T-MTT Nov 05 3562-3567 Cavagnaro, M., see Pisa, S., T-MTT Apr 05 1256-1265 Cazaux, J.-L., see Melle, S., T-MTT Nov 05 3482-3488 Ce-Jun Wei, J.M. Gering, and Y.A. Tkachenko. Enhanced high-current VBIC model; T-MTT Apr 05 1235-1243 Celozzi, S., see Araneo, R., T-MTT Mar 05 907-918 Cendes, Z., see Din-Kow Sun, T-MTT Mar 05 984-992 Ceylan, N., J.-E. Mueller, and R. Weigel. Optimization of EDGE terminal power amplifiers using memoryless digital predistortion; T-MTT Feb 05 515-522 Cha Choong-Yul, see Trung-Kien Nguyen, T-MTT Feb 05 538-547 Cha Choong-Yul, see Choong-Yul Cha, T-MTT Mar 05 881-887 Cha Jeonghyeon, see Jangheon Kim, T-MTT May 05 1802-1809 Chaker, M., see Ouaddari, M., T-MTT Apr 05 1390-1397 Chaki, S., T. Ishida, T. Mizukoshi, H. Yumoto, Y. Sasaki, M. Komaru, and Y. Matsuda. A short stub-matching 77-GHz-band driver amplifier with an attenuator compensating temperature dependence of a gain; T-MTT Jun 05 2073-2081 Chakrabarty, S., see Sharma, S.B., T-MTT Aug 05 2604-2609 Chakraborty, A., see Sanabria, C., T-MTT Feb 05 762-769 Chaloupka, H., see Pepe, G., T-MTT Jan 05 22-31 Chambon, C., see Escotte, L., T-MTT Dec 05 3704-3711 Champeaux, C., see Pothier, A., T-MTT Jan 05 354-360 Chan Chi Hou, see Kam Man Shum, T-MTT Mar 05 895-900 Chandramouli, S., see Hur, Y., T-MTT Jan 05 246-255 Chandramouli, S., see Moonkyun Maeng, T-MTT Nov 05 3509-3519 Chang, K., see Rodenbeck, C.T., T-MTT Dec 05 3697-3703 Chang, S.-F.R., Wen-Lin Chen, Shuen-Chien Chang, Chi-Kang Tu, ChangLin Wei, Chih-Hung Chien, Cheng-Hua Tsai, J. Chen, and A. Chen. A dual-band RF transceiver for multistandard WLAN applications; T-MTT Mar 05 1048-1055 Chang, T.-H., see Yu, C.-F., T-MTT Dec 05 3794-3798 Chang-Auck Choi, see Jaewoo Lee, T-MTT Nov 05 3335-3344 Chang Chi-Yang, see Shih-Ming Wang, T-MTT Feb 05 747-753 Chang Chi-Yang, see Ching-Ku Liao, T-MTT Jul 05 2302-2308 Chang Chun-Fu, see Chun-Fu Chang, T-MTT Jul 05 2383-2388 Chang Han Je, see Jaewoo Lee, T-MTT Nov 05 3335-3344 Chang-Ho Lee, see Mukhopadhyay, R., T-MTT Jan 05 81-93 Chang Ik Soo, see Hyeong Tae Jeong, T-MTT Aug 05 2587-2593 Chang-Lin Wei, see Chang, S.-F.R., T-MTT Mar 05 1048-1055 Chang Shuen-Chien, see Chang, S.-F.R., T-MTT Mar 05 1048-1055 Chang-Soon Choi, Hyo-Soon Kang, Woo-Young Choi, Dae-Hyun Kim, and Kwang-Seok Seo. Phototransistors based on InP HEMTs and their applications to millimeter-wave radio-on-fiber systems; T-MTT Jan 05 256-263 Changyul Cheon, see Byoungjoong Kang, T-MTT Jan 05 134-143 Changyul Cheon, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 Chao-Hsiung Tseng, see Chi-Feng Chen, T-MTT Sep 05 2688-2692 Chao-Hsiung Tseng, and Tah-Hsiung Chu. An effective usage of vector network analyzer for microwave imaging; T-MTT Sep 05 2884-2891 + Check author entry for coauthors

Chao-Huang Wu, see Yo-Shen Lin, T-MTT Jul 05 2324-2328 Chappell, W.J., see Xun Gong, T-MTT Nov 05 3638-3647 Chase, D.R., L.-Y. Chen, and R.A. York. Modeling the capacitive nonlinearity in thin-film BST varactors; T-MTT Oct 05 3215-3220 Chatterjee, A., see Bhattacharya, S., T-MTT Nov 05 3474-3481 Chattopadhyay, G., see Maestrini, A., T-MTT Sep 05 2835-2843 Chee Binn Soo, see Fusco, V., T-MTT Feb 05 730-738 Chek Pin Yang, see Holmes, J.E., T-MTT Jan 05 322-329 Chen, A., see Chang, S.-F.R., T-MTT Mar 05 1048-1055 Chen, A.C., Anh-Vu Pham, and R.E. Leoni, III. Development of low-loss broad-band planar baluns using multilayered organic thin films; T-MTT Nov 05 3648-3655 Chen, C.-F., T.-Y. Huang, and R.-B. Wu. Design of microstrip bandpass filters with multiorder spurious-mode suppression; T-MTT Dec 05 37883793 Chen, J., see Chang, S.-F.R., T-MTT Mar 05 1048-1055 Chen, K.J., see Leung, L.L.W., T-MTT Aug 05 2472-2480 Chen, L.-Y., see Chase, D.R., T-MTT Oct 05 3215-3220 Chen, X.M., see Fan, X.C., T-MTT Oct 05 3130-3134 Chen Caihua, see Zhaolin Lu, T-MTT Apr 05 1362-1368 Chen Chi-Feng, see Chi-Feng Chen, T-MTT Sep 05 2688-2692 Chen Chun Hsiung, see Yo-Shen Lin, T-MTT Jul 05 2324-2328 Chen Chun Hsiung, see Shau-Gang Mao, T-MTT Nov 05 3460-3466 Chen Fanglu, see Li Yang, T-MTT Jan 05 183-190 Cheng, N., see Beyene, W.T., T-MTT Nov 05 3568-3577 Cheng-Hua Tsai, see Chang, S.-F.R., T-MTT Mar 05 1048-1055 Chen Hongsheng, see Yan Li, T-MTT Apr 05 1522-1526 Chen Ji, see Shumin Wang, T-MTT Jun 05 1913-1918 Chen Ji Xin, see Yu Lin Zhang, T-MTT Apr 05 1280-1287 Chen Ji-Xin, see Zhang-Cheng Hao, T-MTT Sep 05 2968-2977 Chen Kangsheng, see Yan Li, T-MTT Apr 05 1522-1526 Chen Kun-Ming, see Ming-Hsiang Cho, T-MTT Sep 05 2926-2934 Chen Min, see Youngcheol Park, T-MTT Jan 05 115-122 Chen Shiou-Li, see Shau-Gang Mao, T-MTT Apr 05 1515-1521 Chen Sin-Ting, see Tzong-Lin Wu, T-MTT Sep 05 2935-2942 Chen Tsung-Wen, see Chi-Feng Chen, T-MTT Sep 05 2688-2692 Chen Tung-Sheng, see Chih-Yuan Lee, T-MTT Feb 05 523-528 Chen-Wei Huang, see Shau-Gang Mao, T-MTT Apr 05 1515-1521 Chen Wen-Lin, see Chang, S.-F.R., T-MTT Mar 05 1048-1055 Chen Xiao-Ping, see Zhang-Cheng Hao, T-MTT Sep 05 2968-2977 Chen Xudong, see Grzegorczyk, T.M., T-MTT Apr 05 1443-1450 Chen Xudong, see Grzegorczyk, T.M., T-MTT Sep 05 2956-2967 Chen Yi-jan Emery, see Yi-jan Emery Chen, T-MTT May 05 1672-1681 Chen Zhizhang, see Shuiping Luo, T-MTT Mar 05 969-976 Cheon Changyul, see Byoungjoong Kang, T-MTT Jan 05 134-143 Cheon Changyul, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 Cheong Pedro, see Pedro Cheong, T-MTT May 05 1810-1816 Chew, S.T., see Boon Tiong Tan, T-MTT Jan 05 343-348 Chia-Sung Chiu, see Ming-Hsiang Cho, T-MTT Sep 05 2926-2934 Chi Chun-Hsiang, see Shih-Ming Wang, T-MTT Feb 05 747-753 Chien Chih-Hung, see Chang, S.-F.R., T-MTT Mar 05 1048-1055 Chien-Chung Wang, see Tzong-Lin Wu, T-MTT Sep 05 2935-2942 Chi-Feng Chen, Ting-Yi Huang, Chao-Hsiung Tseng, Ruey-Beei Wu, and Tsung-Wen Chen. A miniaturized multilayer quasi-elliptic bandpass filter with aperture-coupled microstrip resonators; T-MTT Sep 05 2688-2692 Chih-Hung Chien, see Chang, S.-F.R., T-MTT Mar 05 1048-1055 Chih-Ming Tsai, see Hong-Ming Lee, T-MTT Sep 05 2812-2818 Chih-Ming Tsai, Hong-Ming Lee, and Chin-Chuan Tsai. Planar filter design with fully controllable second passband; T-MTT Nov 05 3429-3439 Chi Hou Chan, see Kam Man Shum, T-MTT Mar 05 895-900 Chi-Hsueh Wang, see Yo-Shen Lin, T-MTT Jul 05 2324-2328 Chih-Yuan Lee, Tung-Sheng Chen, J.D.-S. Deng, and Chin-Hsing Kao. A simple systematic spiral inductor design with perfected Q improvement for CMOS RFIC application; T-MTT Feb 05 523-528 Chi-Kang Tu, see Chang, S.-F.R., T-MTT Mar 05 1048-1055 Chin, K.-S., and J.-T. Kuo. Insertion loss function synthesis of maximally flat parallel-coupled line bandpass filters; T-MTT Oct 05 3161-3168 Chin-Chuan Tsai, see Chih-Ming Tsai, T-MTT Nov 05 3429-3439 Ching-Feng Lee, and Song Tsuen Peng. Systematic analysis of the offsetPLL output spur spectrum; T-MTT Sep 05 3024-3034 Ching-Ku Liao, and Chi-Yang Chang. Design of microstrip quadruplet filters with source-load coupling; T-MTT Jul 05 2302-2308 Chin-Hsing Kao, see Chih-Yuan Lee, T-MTT Feb 05 523-528 Chin-Shen Lin, see Ming-Da Tsai, T-MTT Feb 05 496-505 Chiu, Y.J., see JaeHyuk Shin, T-MTT Feb 05 636-643

IEEE T-MTT 2005 INDEX — 4 Chiu Chia-Sung, see Ming-Hsiang Cho, T-MTT Sep 05 2926-2934 Chiu Hung-Wei, see Hung-Wei Chiu, T-MTT Mar 05 813-824 Chi-Yang Chang, see Shih-Ming Wang, T-MTT Feb 05 747-753 Chi-Yang Chang, see Ching-Ku Liao, T-MTT Jul 05 2302-2308 Chizh, A.L., see Malyshev, S.A., T-MTT Feb 05 439-443 Cho Choongeol, see Choongeol Cho, T-MTT Apr 05 1197-1202 Cho Choon Sik, see Jaeheung Kim, T-MTT Aug 05 2622-2627 Choi, D.K., see Hakala, I., T-MTT Jun 05 2129-2138 Choi Chang-Auck, see Jaewoo Lee, T-MTT Nov 05 3335-3344 Choi Chang-Soon, see Chang-Soon Choi, T-MTT Jan 05 256-263 Choi Hyun-Sik, see Seung-Yup Lee, T-MTT Feb 05 786-793 Choi Jonghoon, see Jonghoon Choi, T-MTT Nov 05 3407-3414 Choi Jung Han, see Jung Han Choi, T-MTT Jun 05 2033-2042 Cho Il-Joo, see Il-Joo Cho, T-MTT Jul 05 2450-2457 Choi Savio, see Daneshmand, M., T-MTT Jan 05 12-21 Choi Woo-Young, see Chang-Soon Choi, T-MTT Jan 05 256-263 Cho Jeiwon, see Byoungjoong Kang, T-MTT Jan 05 134-143 Cho Jeiwon, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 Cho Je-Kwang, see Young-Jin Kim, T-MTT Feb 05 606-613 Cho Kyoung-Joon, see Kyoung-Joon Cho, T-MTT Jan 05 292-300 Cho Ming-Hsiang, see Ming-Hsiang Cho, T-MTT Sep 05 2926-2934 Chongcheawchamnan, M., S. Patisang, S. Srisathit, R. Phromloungsri, and S. Bunnjaweht. Analysis and design of a three-section transmission-line transformer; T-MTT Jul 05 2458-2462 Choon Beng Sia, Beng Hwee Ong, Kiat Seng Yeo, Jian-Guo Ma, and Manh Anh Do. Accurate and scalable RF interconnect model for silicon-based RFIC applications; T-MTT Sep 05 3035-3044 Choongeol Cho, W.R. Eisenstadt, B. Stengel, and E. Ferrer. IIP3 estimation from the gain compression curve; T-MTT Apr 05 1197-1202 Choong-Yul Cha, see Trung-Kien Nguyen, T-MTT Feb 05 538-547 Choong-Yul Cha, and Sang-Gug Lee. A complementary Colpitts oscillator in CMOS technology; T-MTT Mar 05 881-887 Choon Sik Cho, see Jaeheung Kim, T-MTT Aug 05 2622-2627 Chopra, S., see Sehgal, A., T-MTT Sep 05 2682-2687 Cho Sungjoon, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 Chou Yeong-Chang, see Yeong-Chang Chou, T-MTT Nov 05 3398-3406 Chrisostomidis, C.E., and S. Lucyszyn. On the theory of chained-function filters; T-MTT Oct 05 3142-3151 Christopoulos, C., see Sewell, P., T-MTT Jun 05 1919-1928 Chuah, H.T., see Kung, F., T-MTT Apr 05 1189-1196 Chua Lye Heng, see Peng Wang, T-MTT Jan 05 349-353 Chueh Yu-Zhi, see Shau-Gang Mao, T-MTT Nov 05 3460-3466 Chul Dong Kim, see Hyeong Tae Jeong, T-MTT Aug 05 2587-2593 Chul-Soo Kim, see Jong-Sik Lim, T-MTT Aug 05 2539-2545 Chun, C., see Hur, Y., T-MTT Jan 05 246-255 Chun-Cheng Yeh, see Jen-Tsai Kuo, T-MTT Apr 05 1331-1337 Chun-Fu Chang, and Shyh-Jong Chung. Bandpass filter of serial configuration with two finite transmission zeros using LTCC technology; T-MTT Jul 05 2383-2388 Chung Shyh-Jong, see Chun-Fu Chang, T-MTT Jul 05 2383-2388 Chung Younkyu, see Younkyu Chung, T-MTT Feb 05 739-746 Chun-Hsiang Chi, see Shih-Ming Wang, T-MTT Feb 05 747-753 Chun-Hsien Lien, see Ming-Da Tsai, T-MTT Feb 05 496-505 Chun Hsiung Chen, see Yo-Shen Lin, T-MTT Jul 05 2324-2328 Chun Hsiung Chen, see Shau-Gang Mao, T-MTT Nov 05 3460-3466 Chun Young-Hoon, see Young-Hoon Chun, T-MTT Feb 05 687-695 Chu Tah-Hsiung, see Chao-Hsiung Tseng, T-MTT Sep 05 2884-2891 Circa, R., see Rykaczewski, P., T-MTT Mar 05 1056-1064 Clement, T.S., see Williams, D.F., T-MTT Apr 05 1384-1389 Clendenin, J.E., see Ioakeimidi, K., T-MTT Jan 05 336-342 Clifton, J.C., L. Albasha, A. Lawrenson, and A.M. Eaton. Novel multimode J-pHEMT front-end architecture with power-control scheme for maximum efficiency; T-MTT Jun 05 2251-2258 Cognata, A., see Root, D.E., T-MTT Nov 05 3656-3664 Cogollos, S., see Taroncher, M., T-MTT Jun 05 2153-2163 Collado, A., and A. Suarez. Application of bifurcation control to practical circuit design; T-MTT Sep 05 2777-2788 Conciauro, G., see Mira, F., T-MTT Apr 05 1294-1303 Condon, M., see Dautbegovic, E., T-MTT Feb 05 548-555 Copani, T., see Girlando, G., T-MTT Mar 05 952-959 Cormos, D., R. Loison, and R. Gillard. Fast optimization and sensitivity analysis of nonintuitive planar structures; T-MTT Jun 05 2019-2025 Corral, J.L., see Vidal, B., T-MTT Aug 05 2600-2603

+ Check author entry for coauthors

Coskun, A.H., A. Mutlu, and S. Demir. A multitone model of complex enveloped signals and its application in feedforward circuit analysis; TMTT Jun 05 2171-2178 Courcelle, L., see Kerherve, E., T-MTT Jun 05 2145-2152 Coutant, M.R., see Rodenbeck, C.T., T-MTT Dec 05 3697-3703 Cressler, J.D., see Mukhopadhyay, R., T-MTT Jan 05 81-93 Cressler, J.D., see Yi-jan Emery Chen, T-MTT May 05 1672-1681 Cressler, J.D., see Qingqing Liang, T-MTT May 05 1745-1755 Crnkovich, J.G., Jr., see Tavik, G.C., T-MTT Mar 05 1009-1020 Cros, D., see Pothier, A., T-MTT Jan 05 354-360 Cros, D., see Krupka, J., T-MTT Feb 05 702-712 Crowe, T.W., see Zhiyang Liu, T-MTT Sep 05 2949-2955 Curras-Francos, M.C. Table-based nonlinear HEMT model extracted from time-domain large-signal measurements; T-MTT May 05 1593-1600 D Dae-Hyun Kim, see Chang-Soon Choi, T-MTT Jan 05 256-263 Dagli, N., see JaeHyuk Shin, T-MTT Feb 05 636-643 Dahlstrom, M., see Paidi, V.K., T-MTT Feb 05 598-605 Dai Wenliang, see Wenliang Dai, T-MTT Jul 05 2416-2423 Dale, C., see Hadjem, A., T-MTT Jan 05 4-11 Dalmia, S., see Mukherjee, S., T-MTT Jun 05 2196-2210 Dambrine, G., see Six, G., T-MTT Jan 05 301-305 Damm, C., see Mueller, S., T-MTT Jun 05 1937-1945 Daneshmand, M., R.R. Mansour, P. Mousavi, Savio Choi, B. Yassini, A. Zybura, and Ming Yu. Integrated interconnect networks for RF switch matrix applications; T-MTT Jan 05 12-21 Daneshmand, M., and R.R. Mansour. Multiport MEMS-based waveguide and coaxial switches; T-MTT Nov 05 3531-3537 D'Arcangelo, O., see Simonetto, A., T-MTT Dec 05 3731-3734 Darques, M., see Saib, A., T-MTT Jun 05 2043-2049 Darwish, A.M., A.J. Bayba, and H.A. Hung. Accurate determination of thermal resistance of FETs; T-MTT Jan 05 306-313 Darwish, A.M., A. Bayba, and H.A. Hung. Authors' reply [to comments on 'Thermal resistance calculation of AlGaN-GaN devices']; T-MTT Sep 05 3052-3053 Daschner, F., see Schimmer, O., T-MTT Jun 05 2107-2113 Dautbegovic, E., M. Condon, and C. Brennan. An efficient nonlinear circuit simulation technique; T-MTT Feb 05 548-555 Davis, L.E. Comments on "Microwave phase shifter utilizing nonreciprocal wave propagation"; T-MTT Jan 05 414 Dawidczyk, J., see Malyshev, S.A., T-MTT Feb 05 439-443 De Conto, D., see Melle, S., T-MTT Nov 05 3482-3488 de Graaf, J.W., see Tavik, G.C., T-MTT Mar 05 1009-1020 Deguchi, H., see Ohira, M., T-MTT Nov 05 3320-3326 DeJean, G., see Jong-Hoon Lee, T-MTT Jun 05 2220-2229 de Kort, R., see Tiemeijer, L.F., T-MTT Sep 05 2917-2925 Delabie, C.J.P., see Murphy, O.H., T-MTT Jun 05 2063-2072 de la Fuente, L., see Aja, B., T-MTT Jun 05 2050-2062 de la Rubia, V., and J. Zapata. An efficient method for determining TE and TM modes in closed waveguides made up of N cylindrical conductors; TMTT Feb 05 670-678 Deleniv, A.N. Full-wave analysis of coupled strip-slot guiding structures; TMTT Jun 05 1904-1912 Deleniv, A.N., and S. Gevorgian. Open resonator technique for measuring multilayered dielectric plates; T-MTT Sep 05 2908-2916 Delprat, S., see Ouaddari, M., T-MTT Apr 05 1390-1397 de Maagt, P., see Berdel, K., T-MTT Apr 05 1266-1271 Demir, S., see Coskun, A.H., T-MTT Jun 05 2171-2178 Deng, J.D.-S., see Chih-Yuan Lee, T-MTT Feb 05 523-528 Dengi Aykut, see Feng Ling, T-MTT Jan 05 264-273 Deng Junxiong, see Junxiong Deng, T-MTT Feb 05 529-537 Denidni, T.A., see Tatu, S.O., T-MTT Sep 05 2768-2776 Denisov, G., see Bogdashov, A., T-MTT Oct 05 3152-3155 de Paco, P., see Aja, B., T-MTT Jun 05 2050-2062 De Paola, F.M., see Spirito, M., T-MTT Jul 05 2340-2347 Deshpande, M.D., see Baginski, M.E., T-MTT Oct 05 3251-3259 Deslandes, D., and Ke Wu. Analysis and design of current probe transition from grounded coplanar to substrate integrated rectangular waveguides; TMTT Aug 05 2487-2494 De Vita, G., and G. Iannaccone. Design criteria for the RF section of UHF and microwave passive RFID transponders; T-MTT Sep 05 2978-2990 de Vreede, L.C.N., see Spirito, M., T-MTT Jul 05 2340-2347

IEEE T-MTT 2005 INDEX — 5 De Zutter, D., and L. Knockaert. Skin effect modeling based on a differential surface admittance operator; T-MTT Aug 05 2526-2538 Di Alessio, F.L., A. D'Orazio, and R. Lavolpe. Even-harmonic C-band modulator; T-MTT Apr 05 1203-1210 Diaz-Morcillo, A., see Requena-Perez, M.E., T-MTT Jun 05 2114-2120 Dickson, T.O., M.-A. LaCroix, S. Boret, D. Gloria, R. Beerkens, and S.P. Voinigescu. 30-100-GHz inductors and transformers for millimeter-wave (Bi)CMOS integrated circuits; T-MTT Jan 05 123-133 Ding Runtao, see Lei Zhang, T-MTT Sep 05 2752-2767 Din-Kow Sun, L. Vardapetyan, and Z. Cendes. Two-dimensional curlconforming singular elements for FEM solutions of dielectric waveguiding structures; T-MTT Mar 05 984-992 Djordjevic, A.R., and A.G. Zajic. Low-reflection bandpass filters with a flat group delay; T-MTT Apr 05 1164-1167 Do Manh Anh, see Choon Beng Sia, T-MTT Sep 05 3035-3044 Domenech-Asensi, G., J. Hinojosa, J. Martinez-Alajarin, and J. GarrigosGuerrero. Empirical model generation techniques for planar microwave components using electromagnetic linear regression models; T-MTT Nov 05 3305-3311 Donelli, M., and A. Massa. Computational approach based on a particle swarm optimizer for microwave imaging of two-dimensional dielectric scatterers; T-MTT May 05 1761-1776 Dong, X., W.-Y. Yin, and Y.-B. Gan. Perfectly matched layer implementation using bilinear transform for microwave device applications; T-MTT Oct 05 3098-3105 Donghoon Oh, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 Dongjiang Qiao, R. Molfino, S.M. Lardizabal, B. Pillans, P.M. Asbeck, and G. Jerinic. An intelligently controlled RF power amplifier with a reconfigurable MEMS-varactor tuner; T-MTT Mar 05 1089-1095 Dong Kim Chul, see Hyeong Tae Jeong, T-MTT Aug 05 2587-2593 D'Orazio, A., see Di Alessio, F.L., T-MTT Apr 05 1203-1210 Dounavis, A., see Nakhla, N.M., T-MTT Nov 05 3520-3530 Dreher, A., see Truong Vu Bang Giang, T-MTT Jan 05 404-409 Driad, R., see Schneider, K., T-MTT Nov 05 3378-3387 Dubuc, D., see Melle, S., T-MTT Nov 05 3482-3488 Duchamp, J.-M., see Pistono, E., T-MTT Aug 05 2506-2514 Duelk, M., see Sinsky, J.H., T-MTT Jan 05 152-160 Du Huilian, see So, P.P.M., T-MTT Apr 05 1496-1505 Du-Hyun Ko, see Kyung-Whan Yeom, T-MTT Jul 05 2435-2440 Duk-Jae Woo, and Taek-Kyung Lee. Suppression of harmonics in Wilkinson power divider using dual-band rejection by asymmetric DGS; T-MTT Jun 05 2139-2144 Dunsmore, J.A., see Williams, D.F., T-MTT Jan 05 314-321 Duochuan Li, Ping Yang, and Ke Wu. An order-reduced volume-integral equation approach for analysis of NRD-guide and H-guide millimeterwave circuits; T-MTT Mar 05 799-812 Duochuan Li, Y. Cassivi, Ping Yang, and Ke Wu. Analysis and design of bridged NRD-guide coupler for millimeter-wave applications; T-MTT Aug 05 2546-2551 Duochuan Li, and Ke Wu. A generalized surface-volume integral-equation (SVIE) approach for analysis of hybrid planar/NRD-guide integrated circuits; T-MTT Sep 05 2732-2742 Durand, J.-P., see Mazumder, S., T-MTT Mar 05 1065-1071 Duvillaret, L., see Pistono, E., T-MTT Aug 05 2506-2514 E Eaton, A.M., see Clifton, J.C., T-MTT Jun 05 2251-2258 Eccleston, K.W. Compact dual-fed distributed power amplifier; T-MTT Mar 05 825-831 Egorov, V.N., V.L. Masalov, Y.A. Nefyodov, A.F. Shevchun, M.R. Trunin, V.E. Zhitomirsky, and M. McLean. Dielectric constant, loss tangent, and surface resistance of PCB materials at K-band frequencies; T-MTT Feb 05 627-635 Eisenstadt, W.R., see Choongeol Cho, T-MTT Apr 05 1197-1202 Eleftheriades, G.V., and O.F. Siddiqui. Negative refraction and focusing in hyperbolic transmission-line periodic grids; T-MTT Jan 05 396-403 Eleftheriades, G.V., see Kokkinos, T., T-MTT Apr 05 1488-1495 Ellinger, F. 26.5-30-GHz resistive mixer in 90-nm VLSI SOI CMOS technology with high linearity for WLAN; T-MTT Aug 05 2559-2565 El-Tager, A.M., and L. Roy. Study of cylindrical multilayered ceramic resonators with rectangular air cavity for low-phase noise K/Ka-band oscillators; T-MTT Jun 05 2211-2219 Emery Chen Yi-jan, see Yi-jan Emery Chen, T-MTT May 05 1672-1681 Eng, D., see Yeong-Chang Chou, T-MTT Nov 05 3398-3406 + Check author entry for coauthors

Engargiola, G., and A. Navarrini. K-band orthomode transducer with waveguide ports and balanced coaxial probes; T-MTT May 05 1792-1801 Engheta Nader, see Nader Engheta, T-MTT Apr 05 1535-1556 Entesari, K., and G.M. Rebeiz. A differential 4-bit 6.5-10-GHz RF MEMS tunable filter; T-MTT Mar 05 1103-1110 Entesari, K., and G.M. Rebeiz. A 12-18-GHz three-pole RF MEMS tunable filter; T-MTT Aug 05 2566-2571 Ergin, A.A., see Aksoy, S., T-MTT Aug 05 2465-2471 Eric Rius, see Six, G., T-MTT Jan 05 301-305 Esbert, V.E.B., see Ros, J.V.M., T-MTT Apr 05 1130-1142 Esbert, V.E.B., see Mira, F., T-MTT Apr 05 1294-1303 Escotte, L., and J. Graffeuil. Comments on "Microwave noise modeling for InP-InGaAs HBTs"; T-MTT Jan 05 415-416 Escotte, L., E. Gonneau, C. Chambon, and J. Graffeuil. Noise behavior of microwave amplifiers operating under nonlinear conditions; T-MTT Dec 05 3704-3711 Eshrah, I.A., A.A. Kishk, A.B. Yakovlev, and A.W. Glisson. Rectangular waveguide with dielectric-filled corrugations supporting backward waves; T-MTT Nov 05 3298-3304 Esteban, J., C. Camacho-Penalosa, J.E. Page, T.M. Martin-Guerrero, and E. Marquez-Segura. Simulation of negative permittivity and negative permeability by means of evanescent waveguide Modes-theory and experiment; T-MTT Apr 05 1506-1514 Esteban, M., see Lawson, W., T-MTT Jan 05 372-379 Euisik Yoon, see Il-Joo Cho, T-MTT Jul 05 2450-2457 Euisik Yoon, see Sangsoo Ko, T-MTT Sep 05 2789-2800 Eun Kim Ji, see Hyeong Tae Jeong, T-MTT Aug 05 2587-2593 Evins, J.B., see Tavik, G.C., T-MTT Mar 05 1009-1020 Exarchos, M.-N., see Papaioannou, G., T-MTT Nov 05 3467-3473 F Faircloth, D.L., see Baginski, M.E., T-MTT Oct 05 3251-3259 Fai Wong Man, see Hadjem, A., T-MTT Jan 05 4-11 Falcone, F., see Baena, J.D., T-MTT Apr 05 1451-1461 Falcone, F., see Garcia-Garcia, J., T-MTT Jun 05 1997-2006 Fan, X.C., X.M. Chen, and X.Q. Liu. Complex-permittivity measurement on high-Q materials via combined numerical approaches; T-MTT Oct 05 3130-3134 Fanglu Chen, see Li Yang, T-MTT Jan 05 183-190 Fang Sher Jiun, see Sher Jiun Fang, T-MTT Feb 05 478-487 Fan Mingyan, see Li Yang, T-MTT Jan 05 183-190 Farina, M. Comments on "On deembedding of port discontinuities in fullwave CAD models of multiport circuits" and related comments; T-MTT May 05 1829 Farina, M., A. Morini, and T. Rozzi. On the derivation of coupled-line models from EM simulators and application to MoM analysis; T-MTT Nov 05 3272-3280 Fathelbab, W., see Nath, J., T-MTT Sep 05 2707-2712 Fathelbab, W.M., and M.B. Steer. A reconfigurable bandpass filter for RF/microwave multifunctional systems; T-MTT Mar 05 1111-1116 Fathelbab, W.M., and M.B. Steer. Four-port microwave networks with intrinsic broad-band suppression of common-mode signals; T-MTT May 05 1569-1575 Fathelbab, W.M., and M.B. Steer. New classes of miniaturized planar Marchand baluns; T-MTT Apr 05 1211-1220 Fathelbab, W.M., and M.B. Steer. Parallel-coupled line filters with enhanced stopband performances; T-MTT Dec 05 3774-3781 Fear, E.C., see Sill, J.M., T-MTT Nov 05 3312-3319 Feiler, G., see Basaran, U., T-MTT Mar 05 919-928 Feipeng Wang, A.H. Yang, D.F. Kimball, L.E. Larson, and P.M. Asbeck. Design of wide-bandwidth envelope-tracking power amplifiers for OFDM applications; T-MTT Apr 05 1244-1255 Feng, J., see Beyene, W.T., T-MTT Nov 05 3568-3577 Feng Ling, V.I. Okhmatovski, W. Harris, S. McCracken, and Aykut Dengi. Large-scale broad-band parasitic extraction for fast layout verification of 3-D RF and mixed-signal on-chip structures; T-MTT Jan 05 264-273 Feng Xu, and Ke Wu. Guided-wave and leakage characteristics of substrate integrated waveguide; T-MTT Jan 05 66-73 Feng Zhenghe, see Li Yang, T-MTT Jan 05 183-190 Feng Zhiming, see Guofu Niu, T-MTT Feb 05 506-514 Feresidis, A.P., see Yunchuan Guo, T-MTT Apr 05 1462-1468 Fernandes, C.A., see Silveirinha, M.G., T-MTT Apr 05 1418-1430 Ferrari, P., see Pistono, E., T-MTT Aug 05 2506-2514 Ferrer, E., see Choongeol Cho, T-MTT Apr 05 1197-1202

IEEE T-MTT 2005 INDEX — 6 Ferrero, A., see Teppati, V., T-MTT Nov 05 3665-3671 Fickenscher, T., and A. Schwolen. Experimental verification of nonuniform plasma layer model for quartz-silicon image guide phase shifters; T-MTT Jul 05 2375-2382 Figini, L., see Simonetto, A., T-MTT Dec 05 3731-3734 Filicori, F., see Raffo, A., T-MTT Nov 05 3449-3459 Floyd, B.A., S.K. Reynolds, T. Zwick, L. Khuon, T. Beukema, and U.R. Pfeiffer. WCDMA direct-conversion receiver front-end comparison in RFCMOS and SiGe BiCMOS; T-MTT Apr 05 1181-1188 Focardi, P., W.R. McGrath, and A. Neto. Design guidelines for terahertz mixers and detectors; T-MTT May 05 1653-1661 Fok Si-Weng, see Pedro Cheong, T-MTT May 05 1810-1816 Franzon, P.D., see Nath, J., T-MTT Sep 05 2707-2712 Frederick Huang, see Yi Wang, T-MTT Jul 05 2348-2354 Freeman, G., see Yi-jan Emery Chen, T-MTT May 05 1672-1681 Freire, M.J., see Leon, G., T-MTT May 05 1739-1744 Frezza, F., see Baccarelli, P., T-MTT Jan 05 32-44 Frezza, F., see Baccarelli, P., T-MTT Apr 05 1431-1442 Frommberger, M., C. Schmutz, M. Tewes, J. McCord, W. Hartung, R. Losehand, and E. Quandt. Integration of crossed anisotropy magnetic core into toroidal thin-film inductors; T-MTT Jun 05 2096-2100 Frye, R.C., see Youngcheol Park, T-MTT Jan 05 115-122 Fujise, M., see Zhenhai Shao, T-MTT Jul 05 2261-2266 Fumeaux, C., see Baumann, D., T-MTT Nov 05 3595-3605 Fung, A., see Paidi, V.K., T-MTT Feb 05 598-605 Fusco, V., Chee Binn Soo, and N. Buchanan. Analysis and characterization of PLL-based retrodirective array; T-MTT Feb 05 730-738 G Gad, E., see Ahmed, T., T-MTT Nov 05 3538-3547 Galli, A., see Baccarelli, P., T-MTT Jan 05 32-44 Galli, A., see Baccarelli, P., T-MTT Apr 05 1431-1442 Galwas, B.A., see Malyshev, S.A., T-MTT Feb 05 439-443 Gamand, P., see Wane, S., T-MTT Jan 05 200-214 Gan, Y.-B., see Dong, X., T-MTT Oct 05 3098-3105 Gao, J., and G. Boeck. Relationships between common source, common gate, and common drain FETs; T-MTT Dec 05 3825-3831 Gao, S., and P. Gardner. Integrated antenna/power combiner for LINC radio transmitters; T-MTT Mar 05 1083-1088 Gao Huai, see Haitao Zhang, T-MTT Nov 05 3606-3614 Gao Jianjun, see Jianjun Gao, T-MTT Jan 05 330-335 Gao Jianjun, see Jianjun Gao, T-MTT Jan 05 417 Garcia, J.P., see Pereira, F.D.Q., T-MTT Jan 05 94-105 Garcia-Garcia, J., see Baena, J.D., T-MTT Apr 05 1451-1461 Garcia-Garcia, J., F. Martin, F. Falcone, J. Bonache, J.D. Baena, I. Gil, E. Amat, T. Lopetegi, M.A.G. Laso, J.A.M. Iturmendi, M. Sorolla, and R. Marques. Microwave filters with improved stopband based on subwavelength resonators; T-MTT Jun 05 1997-2006 Gard, K.G., L.E. Larson, and M.B. Steer. The impact of RF front-end characteristics on the spectral regrowth of communications signals; T-MTT Jun 05 2179-2186 Gardner, P., see Gao, S., T-MTT Mar 05 1083-1088 Garrigos-Guerrero, J., see Domenech-Asensi, G., T-MTT Nov 05 3305-3311 Gebara, E., see Hur, Y., T-MTT Jan 05 246-255 Gebara, E., see Raghavan, A., T-MTT Nov 05 3498-3508 Gebara, E., see Moonkyun Maeng, T-MTT Nov 05 3509-3519 Gerhard, W., and R.H. Knoechel. LINC digital component separator for single and multicarrier W-CDMA signals; T-MTT Jan 05 274-282 Gering, J.M., see Ce-Jun Wei, T-MTT Apr 05 1235-1243 Gevorgian, S., see Kuylenstierna, D., T-MTT Jun 05 2164-2170 Gevorgian, S., see Deleniv, A.N., T-MTT Sep 05 2908-2916 Geyer, R.G., see Xun Gong, T-MTT Nov 05 3638-3647 Ghali, H.A., and T.A. Moselhy. Broad-band and circularly polarized spacefilling-based slot antennas; T-MTT Jun 05 1946-1950 Ghannouchi, F.M., see Hammi, O., T-MTT May 05 1643-1652 Ghannouchi, F.M., see Helaoui, M., T-MTT Jul 05 2355-2361 Ghannouchi, F.M., see Sirois, J., T-MTT Sep 05 2875-2883 Ghannouchi, F.M., see Taijun Liu, T-MTT Nov 05 3578-3587 Gharaibeh, K.M., and M.B. Steer. Modeling distortion in multichannel communication systems; T-MTT May 05 1682-1692 Gharavi, L., see Hakala, I., T-MTT Jun 05 2129-2138 Ghazel, A., see Helaoui, M., T-MTT Jul 05 2355-2361

+ Check author entry for coauthors

Ghose, A., B. Bunz, J. Weide, and G. Kompa. Extraction of nonlinear parameters of dispersive avalanche photodiode using pulsed RF measurement and quasi-DC optical excitation; T-MTT Jun 05 2082-2087 Ghosh, D., see Nath, J., T-MTT Sep 05 2707-2712 Giang, Truong Vu Bang, see Truong Vu Bang Giang, T-MTT Jan 05 404409 Gil, I., see Baena, J.D., T-MTT Apr 05 1451-1461 Gil, I., see Garcia-Garcia, J., T-MTT Jun 05 1997-2006 Gil, J., see Taroncher, M., T-MTT Jun 05 2153-2163 Gil, J.M. CAD-oriented analysis of cylindrical and spherical dielectric resonators in cavities and MIC environments by means of finite elements; T-MTT Sep 05 2866-2874 Gill, J.J., see Maestrini, A., T-MTT Sep 05 2835-2843 Gillard, R., see Cormos, D., T-MTT Jun 05 2019-2025 Gimeno, B., see Taroncher, M., T-MTT Jun 05 2153-2163 Giordano, V., see Bourgeois, P.-Y., T-MTT Oct 05 3185-3190 Girlando, G., S.A. Smerzi, T. Copani, and G. Palmisano. A monolithic 12GHz heterodyne receiver for DVB-S applications in silicon bipolar technology; T-MTT Mar 05 952-959 Glisson, A.W., see Eshrah, I.A., T-MTT Nov 05 3298-3304 Gloria, D., see Dickson, T.O., T-MTT Jan 05 123-133 Godin, J., see Konczykowska, A., T-MTT Apr 05 1228-1234 Godin, J., see Puyal, V., T-MTT Apr 05 1338-1344 Goldwasser, S.M., see Li, Y., T-MTT Oct 05 3121-3129 Gomez-Garcia, R., see Sanchez-Renedo, M., T-MTT Jan 05 191-199 Gomez-Garcia, R., J.I. Alonso, and D. Amor-Martin. Using the branch-line directional coupler in the design of microwave bandpass filters; T-MTT Oct 05 3221-3229 Gong Xun, see Xun Gong, T-MTT Nov 05 3638-3647 Gonneau, E., see Escotte, L., T-MTT Dec 05 3704-3711 Gonzalez, H.E., see Ros, J.V.M., T-MTT Apr 05 1130-1142 Gook-Ju Ihm, see Trung-Kien Nguyen, T-MTT Feb 05 538-547 Gortz, F.-J., see Pepe, G., T-MTT Jan 05 22-31 Gosalia, K., see Konanur, A.S., T-MTT Jun 05 1837-1844 Goshi, D.S., K.M.K.H. Leong, and T. Itoh. A secure high-speed retrodirective communication link; T-MTT Nov 05 3548-3556 Goto, S., see Ohta, A., T-MTT Jun 05 2121-2128 Gourdon, C., see Nallatamby, J.-C., T-MTT May 05 1601-1612 Goussetis, G., see Yunchuan Guo, T-MTT Apr 05 1462-1468 Gradinaru, S., see Ioakeimidi, K., T-MTT Jan 05 336-342 Graffeuil, J., see Escotte, L., T-MTT Jan 05 415-416 Graffeuil, J., see Escotte, L., T-MTT Dec 05 3704-3711 Grenier, K., see Melle, S., T-MTT Nov 05 3482-3488 Griffin, R.G., see Woskov, P.P., T-MTT Jun 05 1863-1869 Griffith, Z., see Paidi, V.K., T-MTT Feb 05 598-605 Grzegorczyk, T.M., M. Nikku, Xudong Chen, B.-I. Wu, and Jin Au Kong. Refraction laws for anisotropic media and their application to left-handed metamaterials; T-MTT Apr 05 1443-1450 Grzegorczyk, T.M., see Yan Li, T-MTT Apr 05 1522-1526 Grzegorczyk, T.M., C.D. Moss, Jie Lu, Xudong Chen, J. Pacheco, Jr., and Jin Au Kong. Properties of left-handed metamaterials: transmission, backward phase, negative refraction, and focusing; T-MTT Sep 05 29562967 Guangtsai Lei, see Meisong Tong, T-MTT Jul 05 2362-2370 Guann-Pyng Li, see Haitao Zhang, T-MTT Nov 05 3606-3614 Guan Xiang, see Hashemi, H., T-MTT Feb 05 614-626 Gudem, P.S., see Junxiong Deng, T-MTT Feb 05 529-537 Guilin Sun, and C.W. Trueman. Optimized finite-difference time-domain methods based on the (2,4) stencil; T-MTT Mar 05 832-842 Guizhen Zheng, see Pothier, A., T-MTT Jan 05 354-360 Gulck, A., see Schimmer, O., T-MTT Jun 05 2107-2113 Gunnarsson, S.E., see Kuylenstierna, D., T-MTT Aug 05 2616-2621 Guoan Wang, see Papaioannou, G., T-MTT Nov 05 3467-3473 Guofu Niu, Jin Tang, Zhiming Feng, A.J. Joseph, and D.L. Harame. Scaling and technological limitations of 1/f noise and oscillator phase noise in SiGe HBTs; T-MTT Feb 05 506-514 Guofu Niu, see Qingqing Liang, T-MTT May 05 1745-1755 Guo-Wei Huang, see Ming-Hsiang Cho, T-MTT Sep 05 2926-2934 Guoyong Zhang, F. Huang, and M.J. Lancaster. Superconducting spiral filters with quasi-elliptic characteristic for radio astronomy; T-MTT Mar 05 947-951 Guo Yunchuan, see Yunchuan Guo, T-MTT Apr 05 1462-1468 Gupta, M., see Sehgal, A., T-MTT Sep 05 2682-2687

IEEE T-MTT 2005 INDEX — 7 Gupta, R.C., and S.P. Singh. Analysis of the SAR distributions in threelayered bio-media in direct contact with a water-loaded modified box-horn applicator; T-MTT Sep 05 2665-2671 Gupta, R.S., see Sehgal, A., T-MTT Sep 05 2682-2687 H Habicht, W., II, see Tavik, G.C., T-MTT Mar 05 1009-1020 Hadjem, A., D. Lautru, C. Dale, Man Fai Wong, V.F. Hanna, and J. Wiart. Study of specific absorption rate (SAR) induced in two child head models and in adult heads using mobile phones; T-MTT Jan 05 4-11 Hagewood, S.M., see Tavik, G.C., T-MTT Mar 05 1009-1020 Hagness, S.C., see Popovic, D., T-MTT May 05 1713-1722 Hagness, S.C., see Popovic, D., T-MTT Sep 05 3053 Haitao Zhang, Huai Gao, and Guann-Pyng Li. Broad-band power amplifier with a novel tunable output matching network; T-MTT Nov 05 3606-3614 Hai-Ying Yao, Wei Xu, Le-Wei Li, Qun Wu, and Tat-Soon Yeo. Propagation property analysis of metamaterial constructed by conductive SRRs and wires using the MGS-based algorithm; T-MTT Apr 05 1469-1476 Haiyong Xu, see Zhiyang Liu, T-MTT Sep 05 2949-2955 Hajimiri, A., see Hashemi, H., T-MTT Feb 05 614-626 Hajimiri, A., see Analui, B., T-MTT Nov 05 3388-3397 Hakala, I., D.K. Choi, L. Gharavi, N. Kajakine, J. Koskela, and R. Kaunisto. A 2.14-GHz Chireix outphasing transmitter; T-MTT Jun 05 2129-2138 Hakli, J., see Koskinen, T., T-MTT Sep 05 2999-3006 Hale, P.D., see Williams, D.F., T-MTT Apr 05 1384-1389 Halloran, J.W., see Xun Gong, T-MTT Nov 05 3638-3647 Halonen, K.A.I., see Tiiliharju, E., T-MTT Feb 05 679-686 Halonen, K.A.I., see Varonen, M., T-MTT Apr 05 1322-1330 Ha Man-Lyun, see Jong-Min Yook, T-MTT Jun 05 2230-2234 Hammi, O., S. Boumaiza, M. Jaidane-Saidane, and F.M. Ghannouchi. Digital subband filtering predistorter architecture for wireless transmitters; T-MTT May 05 1643-1652 Han, Y., O. Boyraz, and B. Jalali. Ultrawide-band photonic time-stretch a/D converter employing phase diversity; T-MTT Apr 05 1404-1408 Han Choi Jung, see Jung Han Choi, T-MTT Jun 05 2033-2042 Hancock, T.M., see Juo-Jung Hung, T-MTT Feb 05 754-761 Hancock, T.M., and G.M. Rebeiz. A 12-GHz SiGe phase shifter with integrated LNA; T-MTT Mar 05 977-983 Hancock, T.M., and G.M. Rebeiz. Design and analysis of a 70-ps SiGe differential RF switch; T-MTT Jul 05 2403-2410 Han Je Chang, see Jaewoo Lee, T-MTT Nov 05 3335-3344 Han Jeongwoo, see Jeongwoo Han, T-MTT Jun 05 1875-1882 Hanna, V.F., see Hadjem, A., T-MTT Jan 05 4-11 Han She Wing, see Xun Gong, T-MTT Nov 05 3638-3647 Hao Zhang-Cheng, see Zhang-Cheng Hao, T-MTT Sep 05 2968-2977 Happy, H., see Six, G., T-MTT Jan 05 301-305 Harame, D.L., see Guofu Niu, T-MTT Feb 05 506-514 Harris, J.S., Jr., see Ioakeimidi, K., T-MTT Jan 05 336-342 Harris, W., see Feng Ling, T-MTT Jan 05 264-273 Harrison, R.G., see Pistono, E., T-MTT Aug 05 2506-2514 Hartnagel, H.L., see Vicente, C., T-MTT Aug 05 2515-2525 Hartnett, J.G., see Krupka, J., T-MTT Feb 05 702-712 Hartung, W., see Frommberger, M., T-MTT Jun 05 2096-2100 Hashemi, H., Xiang Guan, A. Komijani, and A. Hajimiri. A 24-GHz SiGe phased-array receiver-LO phase-shifting approach; T-MTT Feb 05 614626 Hashimoto, O., see Ohno, T., T-MTT Jun 05 2088-2095 Hashimoto, O., see Pokharel, R.K., T-MTT Sep 05 2726-2731 Hataya, K., see Inoue, T., T-MTT Jan 05 74-80 Hau-Yiu Tsui, and J. Lau. An on-chip vertical solenoid inductor design for multigigahertz CMOS RFIC; T-MTT Jun 05 1883-1890 Havens, R.J., see Tiemeijer, L.F., T-MTT Feb 05 723-729 Havens, R.J., see Tiemeijer, L.F., T-MTT Sep 05 2917-2925 Heikman, S., see Sanabria, C., T-MTT Feb 05 762-769 Heinrich, H., see Nikitin, P.V., T-MTT Sep 05 2721-2725 Helaoui, M., S. Boumaiza, A. Ghazel, and F.M. Ghannouchi. On the RF/DSP design for efficiency of OFDM transmitters; T-MTT Jul 05 2355-2361 Helaoui, M., see Sirois, J., T-MTT Sep 05 2875-2883 Helszajn, J., and J. Sharp. Verification of first circulation conditions of turnstile waveguide circulators using a finite-element solver; T-MTT Jul 05 2309-2316 Heng Chua Lye, see Peng Wang, T-MTT Jan 05 349-353 Hentschel, T. The six-port as a communications receiver; T-MTT Mar 05 1039-1047 + Check author entry for coauthors

Heo, D., see Srirattana, N., T-MTT Mar 05 852-860 Herczfeld, P.R., see Li, Y., T-MTT Oct 05 3121-3129 Hermann, C., M. Tiebout, and H. Klar. A 0.6-V 1.6-mW transformer-based 2.5-GHz downconversion mixer with +5.4-dB gain and -2.8-dBm IIP3 in 0.13-ȝm CMOS; T-MTT Feb 05 488-495 Hernandez, E., see Legarda, J., T-MTT Aug 05 2481-2486 Herranz-Herruzo, J.I., see Valero-Nogueira, A., T-MTT Mar 05 868-873 Hettak, K., G.A. Morin, and M.G. Stubbs. The integration of thin-film microstrip and coplanar technologies for reduced-size MMICs; T-MTT Jan 05 283-291 Hettak, K., G.A. Morin, and M.G. Stubbs. Compact MMIC CPW and asymmetric CPS branch-line couplers and Wilkinson dividers using shunt and series stub loading; T-MTT May 05 1624-1635 Hidalgo-Carpintero, I., see Montejo-Garai, J.R., T-MTT May 05 1636-1642 Hieng Tiong Su, see Yi Wang, T-MTT Jul 05 2348-2354 Hilterbrick, C.L., see Tavik, G.C., T-MTT Mar 05 1009-1020 Hinojosa, J., see Domenech-Asensi, G., T-MTT Nov 05 3305-3311 Hirose, T., see Shigematsu, H., T-MTT Feb 05 472-477 Hirshfield, J., see Bogdashov, A., T-MTT Oct 05 3152-3155 Hoefer, W.J.R., see So, P.P.M., T-MTT Apr 05 1496-1505 Hogan, B.P., see Lawson, W., T-MTT Jan 05 372-379 Holmes, J.E., Chek Pin Yang, P.A. Smith, and T.W. Button. Dielectric helical resonators; T-MTT Jan 05 322-329 Hong, S., see Rodenbeck, C.T., T-MTT Dec 05 3697-3703 Hong Jia-Sheng, see Jia-Sheng Hong, T-MTT Jun 05 1976-1981 Hong Jun Tang, see Yu Lin Zhang, T-MTT Apr 05 1280-1287 Hong-Ming Lee, and Chih-Ming Tsai. Improved coupled-microstrip filter design using effective even-mode and odd-mode characteristic impedances; T-MTT Sep 05 2812-2818 Hong-Ming Lee, see Chih-Ming Tsai, T-MTT Nov 05 3429-3439 Hong Seung-Ho, see Seung-Yup Lee, T-MTT Feb 05 786-793 Hongsheng Chen, see Yan Li, T-MTT Apr 05 1522-1526 Hong Songcheol, see Sangsoo Ko, T-MTT Sep 05 2789-2800 Hongtao Xu, see Sanabria, C., T-MTT Feb 05 762-769 Hong Wang, see Jianjun Gao, T-MTT Jan 05 330-335 Hong Wang, see Jianjun Gao, T-MTT Jan 05 417 Hong Wang, Rong Zeng, and Xiuping Li. An experimental study of carrier heating on channel noise in deep-submicrometer NMOSFETs via body bias; T-MTT Feb 05 564-570 Hong Wei, see Yu Lin Zhang, T-MTT Apr 05 1280-1287 Hong Wei, see Zhang-Cheng Hao, T-MTT Sep 05 2968-2977 Hong-Xing Zheng, and Kwok Wa Leung. An efficient method to reduce the numerical dispersion in the ADI-FDTD; T-MTT Jul 05 2295-2301 Honjo, K., see Aoki, Y., T-MTT Oct 05 3178-3184 Honma, N., see Seki, T., T-MTT Jun 05 2101-2106 Hoppenjans, E.E., see Xun Gong, T-MTT Nov 05 3638-3647 Horii, Y., C. Caloz, and T. Itoh. Super-compact multilayered left-handed transmission line and diplexer application; T-MTT Apr 05 1527-1534 Hornstein, M.K., see Woskov, P.P., T-MTT Jun 05 1863-1869 Hou Chan Chi, see Kam Man Shum, T-MTT Mar 05 895-900 Houng-Jay Yang, see Hsien-Shun Wu, T-MTT Sep 05 2713-2720 How, H., and C. Vittoria. Authors' reply [to Comments on "Microwave phase shifter utilizing nonreciprocal wave propagation"]; T-MTT Jan 05 414 Hrin, G.P., see Tavik, G.C., T-MTT Mar 05 1009-1020 Hsiang, T.Y., see Jingjing Zhang, T-MTT Nov 05 3281-3287 Hsiao Yuan-Wen, see Ming-Dou Ker, T-MTT Sep 05 2672-2681 Hsieh Ming-Yu, see Shih-Ming Wang, T-MTT Feb 05 747-753 Hsien-Shun Wu, Houng-Jay Yang, C.-J. Peng, and C.-K.C. Tzuang. Miniaturized microwave passive filter incorporating multilayer synthetic quasi-TEM transmission line; T-MTT Sep 05 2713-2720 Hsiung Chen Chun, see Yo-Shen Lin, T-MTT Jul 05 2324-2328 Hsiung Chen Chun, see Shau-Gang Mao, T-MTT Nov 05 3460-3466 Hsu Tsun-Lai, see Ming-Hsiang Cho, T-MTT Sep 05 2926-2934 Hu, R., and Tzu-Hsien Sang. On-wafer noise-parameter measurement using wide-band frequency-variation method; T-MTT Jul 05 2398-2402 Hua-Chou Tseng, see Ming-Hsiang Cho, T-MTT Sep 05 2926-2934 Huai Gao, see Haitao Zhang, T-MTT Nov 05 3606-3614 Huang, F., see Guoyong Zhang, T-MTT Mar 05 947-951 Huang, F. Superconducting spiral wide bandpass filters with wide upper stopband; T-MTT Jul 05 2335-2339 Huang, S.Y., and Y.H. Lee. Compact U-shaped dual planar EBG microstrip low-pass filter; T-MTT Dec 05 3799-3805 Huang, T.-W., see Wu, P.-S., T-MTT Oct 05 3106-3114 Huang, T.-Y., see Chen, C.-F., T-MTT Dec 05 3788-3793 Huang Chen-Wei, see Shau-Gang Mao, T-MTT Apr 05 1515-1521

IEEE T-MTT 2005 INDEX — 8 Huang Frederick, see Yi Wang, T-MTT Jul 05 2348-2354 Huangfu Jiangtao, see Yan Li, T-MTT Apr 05 1522-1526 Huang Guo-Wei, see Ming-Hsiang Cho, T-MTT Sep 05 2926-2934 Huang Shao Ying, see Shao Ying Huang, T-MTT Sep 05 2656-2664 Huang Ting-Yi, see Shih-Hao Lee, T-MTT Aug 05 2552-2558 Huang Ting-Yi, see Chi-Feng Chen, T-MTT Sep 05 2688-2692 Hubert, S., see Williams, D.F., T-MTT Jan 05 314-321 Hudec, P., M. Polivka, and P. Pechac. Microwave system for the detection and localization of mobile phones in large buildings; T-MTT Jun 05 22352239 Huei Wang, see Ming-Da Tsai, T-MTT Feb 05 496-505 Huei Wang, see Ming-Fong Lei, T-MTT Mar 05 861-867 Hughes, B., see Konanur, A.S., T-MTT Jun 05 1837-1844 Hui Lee Yee, see Shao Ying Huang, T-MTT Sep 05 2656-2664 Huilian Du, see So, P.P.M., T-MTT Apr 05 1496-1505 Hui Liew Yew, see Yew Hui Liew, T-MTT Aug 05 2633-2638 Hung, H.A., see Darwish, A.M., T-MTT Jan 05 306-313 Hung, H.A., see Darwish, A.M., T-MTT Sep 05 3052-3053 Hung Juo-Jung, see Juo-Jung Hung, T-MTT Feb 05 754-761 Hung Tsai-Pi, see Tsai-Pi Hung, T-MTT Jan 05 144-151 Hung-Wei Chiu, Shey-Shi Lu, and Yo-Sheng Lin. A 2.17-dB NF 5-GHzband monolithic CMOS LNA with 10-mW DC power consumption; TMTT Mar 05 813-824 Hunter, T.R., see Battat, J., T-MTT Jan 05 389-395 Hur, Y., Moonkyun Maeng, C. Chun, F. Bien, Hyoungsoo Kim, S. Chandramouli, E. Gebara, and J. Laskar. Equalization and near-end crosstalk (NEXT) noise cancellation for 20-Gb/s 4-PAM backplane serial I/O interconnections; T-MTT Jan 05 246-255 Hur, Y., see Moonkyun Maeng, T-MTT Nov 05 3509-3519 Huynen, I., see Saib, A., T-MTT Jun 05 2043-2049 Huynen, I., see Torrese, G., T-MTT Oct 05 3238-3243 Hwang In-Chul, see Young-Jin Kim, T-MTT Feb 05 606-613 Hwang Jiunn-Nan, see Jiunn-Nan Hwang, T-MTT Feb 05 653-659 Hwee Ong Beng, see Choon Beng Sia, T-MTT Sep 05 3035-3044 Hyeong Tae Jeong, Ji Eun Kim, Ik Soo Chang, and Chul Dong Kim. Tunable impedance transformer using a transmission line with variable characteristic impedance; T-MTT Aug 05 2587-2593 Hyo-Soon Kang, see Chang-Soon Choi, T-MTT Jan 05 256-263 Hyoungsoo Kim, see Hur, Y., T-MTT Jan 05 246-255 Hyoungsoo Kim, see Moonkyun Maeng, T-MTT Nov 05 3509-3519 Hyun-Sik Choi, see Seung-Yup Lee, T-MTT Feb 05 786-793 I Iannaccone, G., see De Vita, G., T-MTT Sep 05 2978-2990 Idler, W., see Konczykowska, A., T-MTT Apr 05 1228-1234 Ihm Gook-Ju, see Trung-Kien Nguyen, T-MTT Feb 05 538-547 Ik Soo Chang, see Hyeong Tae Jeong, T-MTT Aug 05 2587-2593 Ildu Kim, see Jangheon Kim, T-MTT May 05 1802-1809 Ilic, A.Z., see Ilic, M.M., T-MTT Apr 05 1377-1383 Ilic, M.M., A.Z. Ilic, and B.M. Notaros. Efficient large-domain 2-D FEM solution of arbitrary waveguides using p-refinement on generalized quadrilaterals; T-MTT Apr 05 1377-1383 Il-Joo Cho, Taeksang Song, Sang-Hyun Baek, and Euisik Yoon. A lowvoltage and low-power RF MEMS series and shunt switches actuated by combination of electromagnetic and electrostatic forces; T-MTT Jul 05 2450-2457 Il Kwon Kim, N. Kingsley, M. Morton, R. Bairavasubramanian, J. Papapolymerou, M.M. Tentzeris, and Jong-Gwan Yook. Fractal-shaped microstrip coupled-line bandpass filters for suppression of second harmonic; T-MTT Sep 05 2943-2948 In-Chul Hwang, see Young-Jin Kim, T-MTT Feb 05 606-613 Inoue, A., see Ohta, A., T-MTT Jun 05 2121-2128 Inoue, T., Y. Ando, H. Miyamoto, T. Nakayama, Y. Okamoto, K. Hataya, and M. Kuzuhara. 30-GHz-band over 5-W power performance of shortchannel AlGaN/GaN heterojunction FETs; T-MTT Jan 05 74-80 Ioakeimidi, K., R.F. Leheny, S. Gradinaru, P.R. Bolton, R. Aldana, Kai Ma, J.E. Clendenin, J.S. Harris, Jr., and R.F.W. Pease. Photoelectronic analogto-digital conversion: sampling and quantizing at 100 Gs/s; T-MTT Jan 05 336-342 Iommi, R., G. Macchiarella, A. Meazza, and M. Pagani. Study of an active predistorter suitable for MMIC implementation; T-MTT Mar 05 874-880 Isaksson, M., D. Wisell, and D. Ronnow. Wide-band dynamic modeling of power amplifiers using radial-basis function neural networks; T-MTT Nov 05 3422-3428 + Check author entry for coauthors

Ishida, T., see Chaki, S., T-MTT Jun 05 2073-2081 Ishikawa, T., see Ohta, A., T-MTT Jun 05 2121-2128 Itoh, T., see Sungjoon Lim, T-MTT Jan 05 161-173 Itoh, T., see Younkyu Chung, T-MTT Feb 05 739-746 Itoh, T., and A.A. Oliner. Guest editorial [special issue intro. on metamaterial structures, phenomena, and applications]; T-MTT Apr 05 1413-1417 Itoh, T., see Horii, Y., T-MTT Apr 05 1527-1534 Itoh, T., see Goshi, D.S., T-MTT Nov 05 3548-3556 Itoh, T., see Leong, K.M.K.H., T-MTT Dec 05 3687-3696 Itoh, T., see Lim, S., T-MTT Dec 05 3735-3743 Iturmendi, J.A.M., see Garcia-Garcia, J., T-MTT Jun 05 1997-2006 Iwamoto, M., see Tsai-Pi Hung, T-MTT Jan 05 144-151 J Jaakola, T., see Valois, R., T-MTT Jun 05 2026-2032 Jackson, C. Guest editorial [special issue intro. on the2005 International Microwave Symposium: 2005 Symposium Issue]; T-MTT Nov 05 3264 Jaeheung Kim, Choon Sik Cho, and F.S. Barnes. Dielectric slab Rotman lens for microwave/millimeter-wave applications; T-MTT Aug 05 2622-2627 Jae-Hyoung Park, see Byoungjoong Kang, T-MTT Jan 05 134-143 JaeHyuk Shin, C. Ozturk, S.R. Sakamoto, Y.J. Chiu, and N. Dagli. Novel Trail electrodes for substrate removed low-voltage high-speed GaAs/AlGaAs electrooptic modulators; T-MTT Feb 05 636-643 Jae-Ryong Lee, see Young-Hoon Chun, T-MTT Feb 05 687-695 Jaewoo Lee, Chang Han Je, Sungweon Kang, and Chang-Auck Choi. A lowloss single-pole six-throw switch based on compact RF MEMS switches; T-MTT Nov 05 3335-3344 Jaidane-Saidane, M., see Hammi, O., T-MTT May 05 1643-1652 Jakoby, R., see Mueller, S., T-MTT Jun 05 1937-1945 Jalali, B., see Han, Y., T-MTT Apr 05 1404-1408 Jangheon Kim, Jeonghyeon Cha, Ildu Kim, and Bumman Kim. Optimum operation of asymmetrical-cells-based linear Doherty power Amplifiersuneven power drive and power matching; T-MTT May 05 1802-1809 Jansman, A.B.M., see Tiemeijer, L.F., T-MTT Feb 05 723-729 Jarndal, A., and G. Kompa. A new small-signal modeling approach applied to GaN devices; T-MTT Nov 05 3440-3448 Jarry, P., see Kerherve, E., T-MTT Jun 05 2145-2152 Javadi, H.S., see Maestrini, A., T-MTT Sep 05 2835-2843 Je Chang Han, see Jaewoo Lee, T-MTT Nov 05 3335-3344 Jeiwon Cho, see Byoungjoong Kang, T-MTT Jan 05 134-143 Jeiwon Cho, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 Je-Kwang Cho, see Young-Jin Kim, T-MTT Feb 05 606-613 Jeng, S.-K., see Lai, M.-I., T-MTT Oct 05 3244-3250 Jen-Tsai Kuo, Tsung-Hsun Yeh, and Chun-Cheng Yeh. Design of microstrip bandpass filters with a dual-passband response; T-MTT Apr 05 1331-1337 Jen Yu Jong, see Boon Tiong Tan, T-MTT Jan 05 343-348 Jeon, S., A. Suarez, and D.B. Rutledge. Global stability analysis and stabilization of a class-E/F amplifier with a distributed active transformer; T-MTT Dec 05 3712-3722 Jeong-Geun Kim, see Sangsoo Ko, T-MTT Sep 05 2789-2800 Jeonghoon Yoon, see Byoungjoong Kang, T-MTT Jan 05 134-143 Jeonghoon Yoon, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 Jeonghyeon Cha, see Jangheon Kim, T-MTT May 05 1802-1809 Jeong Hyeong Tae, see Hyeong Tae Jeong, T-MTT Aug 05 2587-2593 Jeong Jinho, see Jinho Jeong, T-MTT Jun 05 1891-1898 Jeong Jinseong, see Younkyu Chung, T-MTT Feb 05 739-746 Jeongwoo Han, and Cam Nguyen. Coupled-slotline-hybrid sampling mixer integrated with step-recovery-diode pulse generator for UWB applications; T-MTT Jun 05 1875-1882 Jeong Yong-Chae, see Jong-Sik Lim, T-MTT Aug 05 2539-2545 Jeong Yoon-Ha, see Seung-Yup Lee, T-MTT Feb 05 786-793 Jerinic, G., see Dongjiang Qiao, T-MTT Mar 05 1089-1095 Jia Lin, see Jianjun Gao, T-MTT Jan 05 330-335 Jiangtao Huangfu, see Yan Li, T-MTT Apr 05 1522-1526 Jian-Guo Ma, see Choon Beng Sia, T-MTT Sep 05 3035-3044 Jianjun Gao, Xiuping Li, Lin Jia, Hong Wang, and G. Boeck. Direct extraction of InP HBT noise parameters based on noise-figure measurement system; T-MTT Jan 05 330-335 Jianjun Gao, Xiuping Li, Hong Wang, and G. Boeck. Authors' reply [to Comments on "Microwave noise modeling for InP-InGaAs HBTs"]; TMTT Jan 05 417 Jianjun Xu, see Lei Zhang, T-MTT Sep 05 2752-2767 Jian-Ming Jin, see Yilmaz, A.E., T-MTT Sep 05 2851-2865 Jian-Ming Jin, see Zheng Lou, T-MTT Sep 05 3014-3023

IEEE T-MTT 2005 INDEX — 9 Jia-Sheng Hong, E.P. McErlean, and B.M. Karyamapudi. A high-temperature superconducting filter for future mobile telecommunication systems; TMTT Jun 05 1976-1981 Ji Chen, see Shumin Wang, T-MTT Jun 05 1913-1918 Jie Lu, see Grzegorczyk, T.M., T-MTT Sep 05 2956-2967 Ji Eun Kim, see Hyeong Tae Jeong, T-MTT Aug 05 2587-2593 Jin Au Kong, see Grzegorczyk, T.M., T-MTT Apr 05 1443-1450 Jin Au Kong, see Yan Li, T-MTT Apr 05 1522-1526 Jin Au Kong, see Grzegorczyk, T.M., T-MTT Sep 05 2956-2967 Jingjing Zhang, S. Alexandrou, and T.Y. Hsiang. Attenuation characteristics of coplanar waveguides at subterahertz frequencies; T-MTT Nov 05 32813287 Jingzhao She, see Li Yang, T-MTT Jan 05 183-190 Jinho Jeong, and Youngwoo Kwon. V-band high-order harmonic injectionlocked frequency-divider MMICs with wide bandwidth and low-power dissipation; T-MTT Jun 05 1891-1898 Jin-Ho Shin, see Joongjin Nam, T-MTT Aug 05 2639-2644 Jin Jian-Ming, see Yilmaz, A.E., T-MTT Sep 05 2851-2865 Jin Jian-Ming, see Zheng Lou, T-MTT Sep 05 3014-3023 Jin-Koo Rhee, see Young-Hoon Chun, T-MTT Feb 05 687-695 Jinseong Jeong, see Younkyu Chung, T-MTT Feb 05 739-746 Jin Tang, see Guofu Niu, T-MTT Feb 05 506-514 Jin Zhenrong, see Yi-jan Emery Chen, T-MTT May 05 1672-1681 Jiun Fang Sher, see Sher Jiun Fang, T-MTT Feb 05 478-487 Jiunn-Nan Hwang A compact 2-D FDFD method for modeling microstrip structures with nonuniform grids and perfectly matched layer; T-MTT Feb 05 653-659 Ji Xin Chen, see Yu Lin Zhang, T-MTT Apr 05 1280-1287 Ji-Xin Chen, see Zhang-Cheng Hao, T-MTT Sep 05 2968-2977 Joe, J., see Yew Hui Liew, T-MTT Aug 05 2633-2638 Johansen, T.K., J. Vidkjaer, and V. Krozer. Analysis and design of wideband SiGe HBT active mixers; T-MTT Jul 05 2389-2397 Johnson, J., G.G. Adams, and N.E. McGruer. Determination of intermodulation distortion in a contact-type MEMS microswitch; T-MTT Nov 05 3615-3620 Jong-Gwan Yook, see Jung-Min Kim, T-MTT Sep 05 2693-2699 Jong-Gwan Yook, see Il Kwon Kim, T-MTT Sep 05 2943-2948 Jong-Heon Kim, see Kyoung-Joon Cho, T-MTT Jan 05 292-300 Jonghoon Choi, and A. Mortazawi. Design of push-push and triple-push oscillators for reducing 1/f noise upconversion; T-MTT Nov 05 3407-3414 Jong-Hoon Lee, G. DeJean, S. Sarkar, S. Pinel, Kyutae Lim, J. Papapolymerou, J. Laskar, and M.M. Tentzeris. Highly integrated millimeter-wave passive components using 3-D LTCC system-on-package (SOP) technology; T-MTT Jun 05 2220-2229 Jong Jen Yu, see Boon Tiong Tan, T-MTT Jan 05 343-348 Jong-Min Yook, Ju-Hyun Ko, Man-Lyun Ha, and Young-Se Kwon. Highquality solenoid inductor using dielectric film for multichip modules; TMTT Jun 05 2230-2234 Jong-Sik Lim, Chul-Soo Kim, D. Ahn, Yong-Chae Jeong, and Sangwook Nam. Design of low-pass filters using defected ground structure; T-MTT Aug 05 2539-2545 Jongsoo Lee, see Mukhopadhyay, R., T-MTT Jan 05 81-93 Jongsoo Lee, see Yi-jan Emery Chen, T-MTT May 05 1672-1681 Jongwoo Lee, see Seungwoo Kim, T-MTT Jan 05 380-388 Joongjin Nam, Jin-Ho Shin, and Bumman Kim. A handset power amplifier with high efficiency at a low level using load-modulation technique; TMTT Aug 05 2639-2644 Jorge, F., see Konczykowska, A., T-MTT Apr 05 1228-1234 Jorge, F., see Puyal, V., T-MTT Apr 05 1338-1344 Joseph, A., see Mukhopadhyay, R., T-MTT Jan 05 81-93 Joseph, A.J., see Guofu Niu, T-MTT Feb 05 506-514 Jourdain, A., see Schoebel, J., T-MTT Jun 05 1968-1975 Juh-Tzeng Lue, see Yan-Shian Yeh, T-MTT May 05 1756-1760 Ju-Hyun Ko, see Jong-Min Yook, T-MTT Jun 05 2230-2234 Junfa Mao, see Wenliang Dai, T-MTT Jul 05 2416-2423 Jung, H., see Psiaki, M.L., T-MTT Oct 05 3082-3089 Jung Han Choi, G.R. Olbrich, and P. Russer. An Si Schottky diode demultiplexer circuit for high bit-rate optical receivers; T-MTT Jun 05 2033-2042 Jung-Min Kim, Woo-Tae Kim, and Jong-Gwan Yook. Resonance-suppressed magnetic field probe for EM field-mapping system; T-MTT Sep 05 26932699 Jung-Mu Kim, Donghoon Oh, Jeonghoon Yoon, Sungjoon Cho, Namgon Kim, Jeiwon Cho, Youngwoo Kwon, Changyul Cheon, and Yong-Kweon

+ Check author entry for coauthors

Kim. In vitro and in vivo measurement for biological applications using micromachined probe; T-MTT Nov 05 3415-3421 Jun Tang Hong, see Yu Lin Zhang, T-MTT Apr 05 1280-1287 Junxiong Deng, P.S. Gudem, L.E. Larson, and P.M. Asbeck. A high averageefficiency SiGe HBT power amplifier for WCDMA handset applications; T-MTT Feb 05 529-537 Juo-Jung Hung, T.M. Hancock, and G.M. Rebeiz. High-power highefficiency SiGe Ku- and Ka-band balanced frequency doublers; T-MTT Feb 05 754-761 K Kai Ma, see Ioakeimidi, K., T-MTT Jan 05 336-342 Kajakine, N., see Hakala, I., T-MTT Jun 05 2129-2138 Kajfez, D. Deembedding of lossy Foster networks; T-MTT Oct 05 3199-3205 Kamei, T., see Utsumi, Y., T-MTT Nov 05 3345-3353 Kam Man Shum, Ting Ting Mo, Quan Xue, and Chi Hou Chan. A compact bandpass filter with two tuning transmission zeros using a CMRC resonator; T-MTT Mar 05 895-900 Kam-Weng Tam, see Pedro Cheong, T-MTT May 05 1810-1816 Kan, Q., see Yeong-Chang Chou, T-MTT Nov 05 3398-3406 Kangaslahti, P., see Varonen, M., T-MTT Apr 05 1322-1330 Kang Byoungjoong, see Byoungjoong Kang, T-MTT Jan 05 134-143 Kang Hyo-Soon, see Chang-Soon Choi, T-MTT Jan 05 256-263 Kangsheng Chen, see Yan Li, T-MTT Apr 05 1522-1526 Kang Sungweon, see Jaewoo Lee, T-MTT Nov 05 3335-3344 Kao Chin-Hsing, see Chih-Yuan Lee, T-MTT Feb 05 523-528 Kaper, V.S., R.M. Thompson, T.R. Prunty, and J.R. Shealy. Signal generation, control, and frequency conversion AlGaN/GaN HEMT MMICs; T-MTT Jan 05 55-65 Karanasiou, I.S., N.K. Uzunoglu, and C.C. Papageorgiou. Authors' reply [to comments on "toward functional noninvasive imaging of excitable tissues inside the human body using focused microwave radiometry"]; T-MTT May 05 1831-1832 Karkkainen, M., see Varonen, M., T-MTT Apr 05 1322-1330 Karkkainen, M.K. Efficient excitation of microstrip lines by a virtual transmission line in FDTD; T-MTT Jun 05 1899-1903 Karyamapudi, B.M., see Jia-Sheng Hong, T-MTT Jun 05 1976-1981 Kasper, E., see Morschbach, M., T-MTT Jun 05 2013-2018 Katehi, L.P.B., see Lu, Y., T-MTT Nov 05 3672-3678 Kaunisto, R., see Hakala, I., T-MTT Jun 05 2129-2138 Kee, S.D., see Seungwoo Kim, T-MTT Jan 05 380-388 Kenney, J.S., see Youngcheol Park, T-MTT Jan 05 115-122 Kenney, J.S., see Wangmyong Woo, T-MTT Jan 05 229-237 Kerherve, E., C.P. Moreira, P. Jarry, and L. Courcelle. 40-Gb/s wide-band MMIC pHEMT modulator driver amplifiers designed with the real frequency technique; T-MTT Jun 05 2145-2152 Ker Ming-Dou, see Ming-Dou Ker, T-MTT Feb 05 582-589 Ker Ming-Dou, see Ming-Dou Ker, T-MTT Sep 05 2672-2681 Kettle, D., see Aja, B., T-MTT Jun 05 2050-2062 Ke Wu, see Feng Xu, T-MTT Jan 05 66-73 Ke Wu, see Duochuan Li, T-MTT Mar 05 799-812 Ke Wu, see Belaid, M., T-MTT Apr 05 1124-1129 Ke Wu, see Yu Lin Zhang, T-MTT Apr 05 1280-1287 Ke Wu, see Ouaddari, M., T-MTT Apr 05 1390-1397 Ke Wu, see Xinyu Xu, T-MTT Jul 05 2267-2273 Ke Wu, see Deslandes, D., T-MTT Aug 05 2487-2494 Ke Wu, see Duochuan Li, T-MTT Aug 05 2546-2551 Ke Wu, see Duochuan Li, T-MTT Sep 05 2732-2742 Ke Wu, see Tatu, S.O., T-MTT Sep 05 2768-2776 Ke Wu, see Zhang-Cheng Hao, T-MTT Sep 05 2968-2977 Ke-Ying Su, and J.-T. Kuo. Application of two-dimensional nonuniform fast Fourier transform (2-D NUFFT) technique to analysis of shielded microstrip circuits; T-MTT Mar 05 993-999 Khuon, L., see Floyd, B.A., T-MTT Apr 05 1181-1188 Kiat Seng Yeo, see Choon Beng Sia, T-MTT Sep 05 3035-3044 Kihyun Kwon, see Byoungjoong Kang, T-MTT Jan 05 134-143 Kim, H., I.-J. Yoon, and Y.J. Yoon. A novel fully integrated transmitter frontend with high power-added efficiency; T-MTT Oct 05 3206-3214 Kim, S.-G., see Rodenbeck, C.T., T-MTT Dec 05 3697-3703 Kimball, D.F., see Feipeng Wang, T-MTT Apr 05 1244-1255 Kimberk, R., see Battat, J., T-MTT Jan 05 389-395 Kim Bonkee, see Nam, I., T-MTT May 05 1662-1671 Kim Bumman, see Seungwoo Kim, T-MTT Jan 05 380-388 Kim Bumman, see Jangheon Kim, T-MTT May 05 1802-1809

IEEE T-MTT 2005 INDEX — 10 Kim Bumman, see Joongjin Nam, T-MTT Aug 05 2639-2644 Kim Chul Dong, see Hyeong Tae Jeong, T-MTT Aug 05 2587-2593 Kim Chul-Soo, see Jong-Sik Lim, T-MTT Aug 05 2539-2545 Kim Dae-Hyun, see Chang-Soon Choi, T-MTT Jan 05 256-263 Kim Hyoungsoo, see Hur, Y., T-MTT Jan 05 246-255 Kim Hyoungsoo, see Moonkyun Maeng, T-MTT Nov 05 3509-3519 Kim Ildu, see Jangheon Kim, T-MTT May 05 1802-1809 Kim Il Kwon, see Il Kwon Kim, T-MTT Sep 05 2943-2948 Kim Jaeheung, see Jaeheung Kim, T-MTT Aug 05 2622-2627 Kim Jangheon, see Jangheon Kim, T-MTT May 05 1802-1809 Kim Jeong-Geun, see Sangsoo Ko, T-MTT Sep 05 2789-2800 Kim Ji Eun, see Hyeong Tae Jeong, T-MTT Aug 05 2587-2593 Kim Jong-Heon, see Kyoung-Joon Cho, T-MTT Jan 05 292-300 Kim Jung-Min, see Jung-Min Kim, T-MTT Sep 05 2693-2699 Kim Jung-Mu, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 Kim Namgon, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 Kim Seungwoo, see Seungwoo Kim, T-MTT Jan 05 380-388 Kim Woo-Tae, see Jung-Min Kim, T-MTT Sep 05 2693-2699 Kim Yong-Kweon, see Byoungjoong Kang, T-MTT Jan 05 134-143 Kim Yong-Kweon, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 Kim Young-Jin, see Young-Jin Kim, T-MTT Feb 05 606-613 Kingon, A.I., see Nath, J., T-MTT Sep 05 2707-2712 Kingsley, N., see Il Kwon Kim, T-MTT Sep 05 2943-2948 King-Yuen Wong, and Wai-Yip Tam. Analysis of the frequency response of SAW filters using finite-difference time-domain method; T-MTT Nov 05 3364-3370 Kintner, P.M., see Psiaki, M.L., T-MTT Oct 05 3082-3089 Kishk, A.A., see Eshrah, I.A., T-MTT Nov 05 3298-3304 Kitazawa, T., see Yamamoto, H., T-MTT Jun 05 2187-2195 Klar, H., see Hermann, C., T-MTT Feb 05 488-495 Klein, N., see Panaitov, G.I., T-MTT Nov 05 3371-3377 Knochel, R.H., see Schimmer, O., T-MTT Jun 05 2107-2113 Knockaert, L., see De Zutter, D., T-MTT Aug 05 2526-2538 Knoechel, R.H., see Gerhard, W., T-MTT Jan 05 274-282 Ko Du-Hyun, see Kyung-Whan Yeom, T-MTT Jul 05 2435-2440 Ko Ju-Hyun, see Jong-Min Yook, T-MTT Jun 05 2230-2234 Kokkinos, T., C.D. Sarris, and G.V. Eleftheriades. Periodic finite-difference time-domain analysis of loaded transmission-line negative-refractiveindex metamaterials; T-MTT Apr 05 1488-1495 Komaki, S., see Sangiamwong, J., T-MTT Dec 05 3723-3730 Komaru, M., see Chaki, S., T-MTT Jun 05 2073-2081 Komijani, A., see Hashemi, H., T-MTT Feb 05 614-626 Kompa, G., see Ghose, A., T-MTT Jun 05 2082-2087 Kompa, G., see Jarndal, A., T-MTT Nov 05 3440-3448 Konanur, A.S., K. Gosalia, S.H. Krishnamurthy, B. Hughes, and G. Lazzi. Increasing wireless channel capacity through MIMO systems employing co-located antennas; T-MTT Jun 05 1837-1844 Konczykowska, A., F. Jorge, W. Idler, and J. Godin. High-sensitivity InP/InGaAs DHBT decision circuit-design and application in optical and system experiments at 40-43 Gbit/s; T-MTT Apr 05 1228-1234 Konczykowska, A., see Puyal, V., T-MTT Apr 05 1338-1344 Kong Jin Au, see Grzegorczyk, T.M., T-MTT Apr 05 1443-1450 Kong Jin Au, see Yan Li, T-MTT Apr 05 1522-1526 Kong Jin Au, see Grzegorczyk, T.M., T-MTT Sep 05 2956-2967 Kornegay, K.T., see Kucharski, D., T-MTT Feb 05 590-597 Ko Sangsoo, see Sangsoo Ko, T-MTT Sep 05 2789-2800 Koskela, J., see Hakala, I., T-MTT Jun 05 2129-2138 Koskinen, T., J. Ala-Laurinaho, J. Saily, A. Lonnqvist, J. Hakli, J. Mallat, J. Tuovinen, and A.V. Raisanen. Experimental study on a hologram-based compact antenna test range at 650 GHz; T-MTT Sep 05 2999-3006 Kosmas, P., and C.M. Rappaport. Time reversal with the FDTD method for microwave breast cancer detection; T-MTT Jul 05 2317-2323 Kouki, A.B., see Birafane, A., T-MTT Jun 05 2240-2250 Ko Won, see Won Ko, T-MTT Jan 05 361-371 Kozyrev, A.B., and D.W. van der Weide. Nonlinear wave propagation phenomena in left-handed transmission-line media; T-MTT Jan 05 238245 Krishnamurthy, S.H., see Konanur, A.S., T-MTT Jun 05 1837-1844 Krowne, C.M. Electromagnetic distributions demonstrating asymmetry using a spectral-domain dyadic Green's function for ferrite microstrip guidedwave structures; T-MTT Apr 05 1345-1361 Krozer, V., see Johansen, T.K., T-MTT Jul 05 2389-2397 Krupka, J., M.E. Tobar, J.G. Hartnett, D. Cros, and J.-M. Le Floch. Extremely high-Q factor dielectric resonators for millimeter-wave applications; T-MTT Feb 05 702-712 + Check author entry for coauthors

Kucharski, D., and K.T. Kornegay. Jitter considerations in the design of a 10Gb/s automatic gain control amplifier; T-MTT Feb 05 590-597 Kuester, E.F., see Spowart, M.P., T-MTT Mar 05 938-946 Kuester, E.F., see Se-Ho You, T-MTT Sep 05 2826-2834 Kuhn, W.B., M.M. Mojarradi, and A. Moussessian. A resonant switch for LNA protection in watt-level CMOS transceivers; T-MTT Sep 05 28192825 Kulas, L., and M. Mrozowski. Low-reflection subgridding; T-MTT May 05 1587-1592 Kung, F., and H.T. Chuah. A study on the stability of bipolar-junctiontransistor formulation in finite-difference time-domain framework; T-MTT Apr 05 1189-1196 Kun-Ming Chen, see Ming-Hsiang Cho, T-MTT Sep 05 2926-2934 Kuo, J.-T., see Ke-Ying Su, T-MTT Mar 05 993-999 Kuo, J.-T., see Chin, K.-S., T-MTT Oct 05 3161-3168 Kuo Bing-Jye, see Ming-Dou Ker, T-MTT Feb 05 582-589 Kuo Bing-Jye, see Ming-Dou Ker, T-MTT Sep 05 2672-2681 Kuo Jen-Tsai, see Jen-Tsai Kuo, T-MTT Apr 05 1331-1337 Kuo Wei-Min Lance, see Yi-jan Emery Chen, T-MTT May 05 1672-1681 Kurz, H., see Berdel, K., T-MTT Apr 05 1266-1271 Kusunoki, S., see Mizusawa, N., T-MTT Nov 05 3327-3334 Kuylenstierna, D., A. Vorobiev, P. Linner, and S. Gevorgian. Ultrawide-band tunable true-time delay lines using ferroelectric varactors; T-MTT Jun 05 2164-2170 Kuylenstierna, D., S.E. Gunnarsson, and H. Zirath. Lumped-element quadrature power splitters using mixed right/left-handed transmission lines; T-MTT Aug 05 2616-2621 Kuzuhara, M., see Inoue, T., T-MTT Jan 05 74-80 Kwang-Seok Seo, see Chang-Soon Choi, T-MTT Jan 05 256-263 Kwok Wa Leung, see Hong-Xing Zheng, T-MTT Jul 05 2295-2301 Kwon Kihyun, see Byoungjoong Kang, T-MTT Jan 05 134-143 Kwon Kim Il, see Il Kwon Kim, T-MTT Sep 05 2943-2948 Kwon Young-Se, see Jong-Min Yook, T-MTT Jun 05 2230-2234 Kwon Youngwoo, see Byoungjoong Kang, T-MTT Jan 05 134-143 Kwon Youngwoo, see Won Ko, T-MTT Jan 05 361-371 Kwon Youngwoo, see Jinho Jeong, T-MTT Jun 05 1891-1898 Kwon Youngwoo, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 Kwyro Lee, see Nam, I., T-MTT May 05 1662-1671 Kyoung-Joon Cho, Jong-Heon Kim, and S.P. Stapleton. A highly efficient Doherty feedforward linear power amplifier for W-CDMA base-station applications; T-MTT Jan 05 292-300 Kyungho Lee, see Seungwoo Kim, T-MTT Jan 05 380-388 Kyung-Suc Nah, see Young-Jin Kim, T-MTT Feb 05 606-613 Kyung-Whan Yeom, and Du-Hyun Ko. A novel 60-GHz monolithic star mixer using gate-drain-connected pHEMT diodes; T-MTT Jul 05 24352440 Kyutae Lim, see Jong-Hoon Lee, T-MTT Jun 05 2220-2229 L LaCroix, M.-A., see Dickson, T.O., T-MTT Jan 05 123-133 Laeseke, P.F., see Brace, C.L., T-MTT Jan 05 215-220 Lahti, M., see Valois, R., T-MTT Jun 05 2026-2032 Lai, M.-I., and S.-K. Jeng. A microstrip three-port and four-channel multiplexer for WLAN and UWB coexistence; T-MTT Oct 05 3244-3250 Lai, R., see Yeong-Chang Chou, T-MTT Nov 05 3398-3406 Lam, S.F., see Nikitin, P.V., T-MTT Sep 05 2721-2725 Lampariello, P., see Baccarelli, P., T-MTT Jan 05 32-44 Lampariello, P., see Baccarelli, P., T-MTT Apr 05 1431-1442 Lanagan, M., see Semouchkina, E., T-MTT Feb 05 644-652 Lanagan, M., see Semouchkina, E.A., T-MTT Apr 05 1477-1487 Lancaster, M.J., see Guoyong Zhang, T-MTT Mar 05 947-951 Lancaster, M.J., see Yi Wang, T-MTT Jul 05 2348-2354 Lance Kuo Wei-Min, see Yi-jan Emery Chen, T-MTT May 05 1672-1681 Lara-Rojo, F., see Rayas-Sanchez, J.E., T-MTT Mar 05 960-968 Lardizabal, S.M., see Dongjiang Qiao, T-MTT Mar 05 1089-1095 Larson, L.E., see Junxiong Deng, T-MTT Feb 05 529-537 Larson, L.E., see Aparin, V., T-MTT Feb 05 571-581 Larson, L.E., see Feipeng Wang, T-MTT Apr 05 1244-1255 Larson, L.E., see Gard, K.G., T-MTT Jun 05 2179-2186 Laskar, J., see Mukhopadhyay, R., T-MTT Jan 05 81-93 Laskar, J., see Hur, Y., T-MTT Jan 05 246-255 Laskar, J., see Srirattana, N., T-MTT Mar 05 852-860 Laskar, J., see Yi-jan Emery Chen, T-MTT May 05 1672-1681 Laskar, J., see Pinel, S., T-MTT May 05 1707-1712

IEEE T-MTT 2005 INDEX — 11 Laskar, J., see Jong-Hoon Lee, T-MTT Jun 05 2220-2229 Laskar, J., see Raghavan, A., T-MTT Nov 05 3498-3508 Laskar, J., see Moonkyun Maeng, T-MTT Nov 05 3509-3519 Laskar, J., see Lee, J.-H., T-MTT Dec 05 3817-3824 Laso, M.A.G., see Baena, J.D., T-MTT Apr 05 1451-1461 Laso, M.A.G., see Garcia-Garcia, J., T-MTT Jun 05 1997-2006 Lau, J., see Hau-Yiu Tsui, T-MTT Jun 05 1883-1890 Lautru, D., see Hadjem, A., T-MTT Jan 05 4-11 Lavolpe, R., see Di Alessio, F.L., T-MTT Apr 05 1203-1210 Lawrenson, A., see Clifton, J.C., T-MTT Jun 05 2251-2258 Lawson, W., M. Esteban, H. Raghunathan, B.P. Hogan, and K. Bharathan. Bandwidth studies of TE0n-TE0(n+1) ripple-wall mode converters in circular waveguide; T-MTT Jan 05 372-379 Lazebnik, M., see Popovic, D., T-MTT May 05 1713-1722 Lazebnik, M., see Popovic, D., T-MTT Sep 05 3053 Lazzi, G., see Konanur, A.S., T-MTT Jun 05 1837-1844 Lee, F.T., Jr., see Brace, C.L., T-MTT Jan 05 215-220 Lee, J.-H., S. Pinel, J. Papapolymerou, J. Laskar, and M.M. Tentzeris. Lowloss LTCC cavity filters using system-on-package technology at 60 GHz; T-MTT Dec 05 3817-3824 Lee, T.H., see Al-Attar, T., T-MTT Nov 05 3557-3561 Lee, Y.H., see Huang, S.Y., T-MTT Dec 05 3799-3805 Lee Chang-Ho, see Mukhopadhyay, R., T-MTT Jan 05 81-93 Lee Chih-Yuan, see Chih-Yuan Lee, T-MTT Feb 05 523-528 Lee Ching-Feng, see Ching-Feng Lee, T-MTT Sep 05 3024-3034 Lee Hong-Ming, see Hong-Ming Lee, T-MTT Sep 05 2812-2818 Lee Hong-Ming, see Chih-Ming Tsai, T-MTT Nov 05 3429-3439 Lee Jae-Ryong, see Young-Hoon Chun, T-MTT Feb 05 687-695 Lee Jaewoo, see Jaewoo Lee, T-MTT Nov 05 3335-3344 Lee Jong-Hoon, see Jong-Hoon Lee, T-MTT Jun 05 2220-2229 Lee Jongsoo, see Mukhopadhyay, R., T-MTT Jan 05 81-93 Lee Jongsoo, see Yi-jan Emery Chen, T-MTT May 05 1672-1681 Lee Jongwoo, see Seungwoo Kim, T-MTT Jan 05 380-388 Lee Kwyro, see Nam, I., T-MTT May 05 1662-1671 Lee Kyungho, see Seungwoo Kim, T-MTT Jan 05 380-388 Leenaerts, D.M.W., see Bergveld, H.J., T-MTT Feb 05 453-461 Lee Sang-Gug, see Trung-Kien Nguyen, T-MTT Feb 05 538-547 Lee Sang-Gug, see Choong-Yul Cha, T-MTT Mar 05 881-887 Lee See Taur, see Sher Jiun Fang, T-MTT Feb 05 478-487 Lee Seung-Yup, see Seung-Yup Lee, T-MTT Feb 05 786-793 Lee Shih-Hao, see Shih-Hao Lee, T-MTT Aug 05 2552-2558 Lee Sungjae, see Sungjae Lee, T-MTT Apr 05 1314-1321 Lee Taek-Kyung, see Duk-Jae Woo, T-MTT Jun 05 2139-2144 Lee Yee Hui, see Shao Ying Huang, T-MTT Sep 05 2656-2664 Lee Yong-Sub, see Seung-Yup Lee, T-MTT Feb 05 786-793 Le Floch, J.-M., see Krupka, J., T-MTT Feb 05 702-712 Legarda, J., J. Presa, E. Hernandez, H. Solar, J. Mendizabal, and J.A. Penaranda. An adaptive feedforward amplifier under "Maximum Output" control method for UMTS downlink transmitters; T-MTT Aug 05 24812486 Leheny, R.F., see Ioakeimidi, K., T-MTT Jan 05 336-342 Lei Guangtsai, see Meisong Tong, T-MTT Jul 05 2362-2370 Leiker, P.S., see Battat, J., T-MTT Jan 05 389-395 Lei Ming-Fong, see Ming-Fong Lei, T-MTT Mar 05 861-867 Lei Zhang, Jianjun Xu, M.C.E. Yagoub, Runtao Ding, and Qi-Jun Zhang. Efficient analytical formulation and sensitivity analysis of neuro-space mapping for nonlinear microwave device modeling; T-MTT Sep 05 27522767 Lei Zhu, see Sheng Sun, T-MTT May 05 1817-1822 Lei Zhu, see Sheng Sun, T-MTT Apr 05 1221-1227 Lei Zhu, see Sheng Sun, T-MTT Sep 05 2844-2850 Leon, G., M.J. Freire, R.R. Boix, and F. Medina. Experimental validation of analysis software for tunable microstrip filters on magnetized ferrites; TMTT May 05 1739-1744 Leong, K.M.K.H., see Goshi, D.S., T-MTT Nov 05 3548-3556 Leong, K.M.K.H., and T. Itoh. Mutually exclusive data encoding for realization of a full duplexing self-steering wireless link using a retrodirective array transceiver; T-MTT Dec 05 3687-3696 Leong, K.M.K.H., see Lim, S., T-MTT Dec 05 3735-3743 Leong, M.-S., see Boon Tiong Tan, T-MTT Jan 05 343-348 Leoni, R.E., III, see Chen, A.C., T-MTT Nov 05 3648-3655 Lepaul, S.B.P., see Sirbu, M., T-MTT Sep 05 2991-2998 Lepine, F., A. Adahl, and H. Zirath. L-band LDMOS power amplifiers based on an inverse class-F architecture; T-MTT Jun 05 2007-2012 Lessin, S.A., see Tavik, G.C., T-MTT Mar 05 1009-1020 + Check author entry for coauthors

Leung, D.L., see Yeong-Chang Chou, T-MTT Nov 05 3398-3406 Leung, L.L.W., and K.J. Chen. Microwave characterization and modeling of high aspect ratio through-wafer interconnect vias in silicon substrates; TMTT Aug 05 2472-2480 Leung Kwok Wa, see Hong-Xing Zheng, T-MTT Jul 05 2295-2301 Le-Wei Li, see Hai-Ying Yao, T-MTT Apr 05 1469-1476 Li, G.P., see Ma, Y., T-MTT Jan 05 221-228 Li, M., see Rodenbeck, C.T., T-MTT Dec 05 3697-3703 Li, Y., M. Bystrom, D. Yoo, S.M. Goldwasser, and P.R. Herczfeld. Coherent optical vector modulation for fiber radio using electrooptic microchip lasers; T-MTT Oct 05 3121-3129 Liang Qingqing, see Qingqing Liang, T-MTT May 05 1745-1755 Liao Ching-Ku, see Ching-Ku Liao, T-MTT Jul 05 2302-2308 Li Duochuan, see Duochuan Li, T-MTT Mar 05 799-812 Li Duochuan, see Duochuan Li, T-MTT Aug 05 2546-2551 Li Duochuan, see Duochuan Li, T-MTT Sep 05 2732-2742 Lien Chun-Hsien, see Ming-Da Tsai, T-MTT Feb 05 496-505 Liew Yew Hui, see Yew Hui Liew, T-MTT Aug 05 2633-2638 Li Guann-Pyng, see Haitao Zhang, T-MTT Nov 05 3606-3614 Li Le-Wei, see Hai-Ying Yao, T-MTT Apr 05 1469-1476 Lim, S., K.M.K.H. Leong, and T. Itoh. Adaptive power controllable retrodirective array system for wireless sensor server applications; T-MTT Dec 05 3735-3743 Limaye, K.U., see Santra, M., T-MTT Feb 05 718-722 Lim Jong-Sik, see Jong-Sik Lim, T-MTT Aug 05 2539-2545 Lim Kyutae, see Jong-Hoon Lee, T-MTT Jun 05 2220-2229 Lim Sungjoon, see Sungjoon Lim, T-MTT Jan 05 161-173 Lim Sungkyu, see Byoungjoong Kang, T-MTT Jan 05 134-143 Lin Chin-Shen, see Ming-Da Tsai, T-MTT Feb 05 496-505 Ling Feng, see Feng Ling, T-MTT Jan 05 264-273 Lin Jia, see Jianjun Gao, T-MTT Jan 05 330-335 Lin-Kun Wu, see Ming-Hsiang Cho, T-MTT Sep 05 2926-2934 Linner, P., see Kuylenstierna, D., T-MTT Jun 05 2164-2170 Lin Yen-Hui, see Tzong-Lin Wu, T-MTT Sep 05 2935-2942 Lin Yo-Shen, see Yo-Shen Lin, T-MTT Jul 05 2324-2328 Lin Yo-Sheng, see Hung-Wei Chiu, T-MTT Mar 05 813-824 Lin Zhang Yu, see Yu Lin Zhang, T-MTT Apr 05 1280-1287 Liu, X.Q., see Fan, X.C., T-MTT Oct 05 3130-3134 Liu Taijun, see Taijun Liu, T-MTT Nov 05 3578-3587 Liu Zhiyang, see Zhiyang Liu, T-MTT Sep 05 2949-2955 Lixin Ran, see Yan Li, T-MTT Apr 05 1522-1526 Li Xiuping, see Jianjun Gao, T-MTT Jan 05 330-335 Li Xiuping, see Jianjun Gao, T-MTT Jan 05 417 Li Xiuping, see Hong Wang, T-MTT Feb 05 564-570 Li Yan, see Yan Li, T-MTT Apr 05 1522-1526 Li Yang, Mingyan Fan, Fanglu Chen, Jingzhao She, and Zhenghe Feng. A novel compact electromagnetic-bandgap (EBG) structure and its applications for microwave circuits; T-MTT Jan 05 183-190 Li Zhengfan, see Wenliang Dai, T-MTT Jul 05 2416-2423 Loison, R., see Cormos, D., T-MTT Jun 05 2019-2025 Long, J.R., see Tasic, A., T-MTT Feb 05 556-563 Lonnqvist, A., see Koskinen, T., T-MTT Sep 05 2999-3006 Lopetegi, T., see Baena, J.D., T-MTT Apr 05 1451-1461 Lopetegi, T., see Garcia-Garcia, J., T-MTT Jun 05 1997-2006 Lopez-Villegas, J.M., and J.J. Sieiro Cordoba. BPSK to ASK signal conversion using injection-locked oscillators - Part I: Theory; T-MTT Dec 05 3757-3766 Lopresto, V., see Pisa, S., T-MTT Apr 05 1256-1265 Lorenz, P., J.V. Vital, B. Biscontini, and P. Russer. TLM-G-a grid-enabled time-domain transmission-line-matrix system for the analysis of complex electromagnetic structures; T-MTT Nov 05 3631-3637 Losehand, R., see Frommberger, M., T-MTT Jun 05 2096-2100 Lou Zheng, see Zheng Lou, T-MTT Sep 05 3014-3023 Lovat, G., see Baccarelli, P., T-MTT Jan 05 32-44 Lovat, G., see Baccarelli, P., T-MTT Apr 05 1431-1442 Lovisolo, G.A., see Pisa, S., T-MTT Apr 05 1256-1265 Lu, Y., L.P.B. Katehi, and D. Peroulis. High-power MEMS varactors and impedance tuners for millimeter-wave applications; T-MTT Nov 05 36723678 Lucyszyn, S. Investigation of Wang's model for room-temperature conduction losses in normal metals at terahertz frequencies; T-MTT Apr 05 1398-1403 Lucyszyn, S., see Chrisostomidis, C.E., T-MTT Oct 05 3142-3151 Ludwig, M., see Schneider, K., T-MTT Nov 05 3378-3387 Lue Juh-Tzeng, see Yan-Shian Yeh, T-MTT May 05 1756-1760 Lu Jie, see Grzegorczyk, T.M., T-MTT Sep 05 2956-2967

IEEE T-MTT 2005 INDEX — 12 Lukovnikov, D., see Bogdashov, A., T-MTT Oct 05 3152-3155 Luo, W.Q., S.Y. Tan, and B.T. Tan. Effects of the ground on power-line communications; T-MTT Oct 05 3191-3198 Luo Shuiping, see Shuiping Luo, T-MTT Mar 05 969-976 Lu Shey-Shi, see Hung-Wei Chiu, T-MTT Mar 05 813-824 Luthi, T., A. Murk, and A. Magun. Design and simulations of a nulling interferometer using a single Martin-Puplett interferometer for doublesideband operation; T-MTT Apr 05 1168-1173 Luthi, T., and C. Matzler. Stereoscopic passive millimeter-wave imaging and ranging; T-MTT Aug 05 2594-2599 Luy, J.-F., see Adler, E.D., T-MTT Mar 05 1005-1008 Lu Zhaolin, see Zhaolin Lu, T-MTT Apr 05 1362-1368 Lye Heng Chua, see Peng Wang, T-MTT Jan 05 349-353 M Ma, Y., and G.P. Li. ESD protection design considerations for InGaP/GaAs HBT RF power amplifiers; T-MTT Jan 05 221-228 Maas, S.A., see Pedro, J.C., T-MTT Apr 05 1150-1163 Macchiarella, G., see Iommi, R., T-MTT Mar 05 874-880 Macchiarella, G., see Tamiazzo, S., T-MTT May 05 1693-1698 Macchiarella, G., see Amari, S., T-MTT Oct 05 3075-3081 Macchiarella, G., and S. Tamiazzo. Design techniques for dual-passband filters; T-MTT Nov 05 3265-3271 Madhow, U., see Munkyo Seo, T-MTT Mar 05 1072-1082 Maeng Moonkyun, see Hur, Y., T-MTT Jan 05 246-255 Maeng Moonkyun, see Moonkyun Maeng, T-MTT Nov 05 3509-3519 Maestrini, A., J.S. Ward, J.J. Gill, H.S. Javadi, E. Schlecht, C. TriponCanseliet, G. Chattopadhyay, and I. Mehdi. A 540-640-GHz highefficiency four-anode frequency tripler; T-MTT Sep 05 2835-2843 Magun, A., see Luthi, T., T-MTT Apr 05 1168-1173 Ma Jian-Guo, see Choon Beng Sia, T-MTT Sep 05 3035-3044 Ma Kai, see Ioakeimidi, K., T-MTT Jan 05 336-342 Makon, R.E., see Schneider, K., T-MTT Nov 05 3378-3387 Maksimovic, D., see Narisi Wang, T-MTT Mar 05 1096-1102 Mallat, J., see Koskinen, T., T-MTT Sep 05 2999-3006 Malyshev, S.A., B.A. Galwas, A.L. Chizh, J. Dawidczyk, and V.F. Andrievski. Frequency conversion of optical signals in p-i-n photodiodes; T-MTT Feb 05 439-443 Man Fai Wong, see Hadjem, A., T-MTT Jan 05 4-11 Mangla, T., see Sehgal, A., T-MTT Sep 05 2682-2687 Manh Anh Do, see Choon Beng Sia, T-MTT Sep 05 3035-3044 Man-Lyun Ha, see Jong-Min Yook, T-MTT Jun 05 2230-2234 Man Shum Kam, see Kam Man Shum, T-MTT Mar 05 895-900 Mansour, R., see Salehi, H., T-MTT Nov 05 3489-3497 Mansour, R.R., see Daneshmand, M., T-MTT Jan 05 12-21 Mansour, R.R., see Daneshmand, M., T-MTT Nov 05 3531-3537 Manstretta, D., see Brandolini, M., T-MTT Mar 05 1026-1038 Mao Junfa, see Wenliang Dai, T-MTT Jul 05 2416-2423 Mao Shau-Gang, see Shau-Gang Mao, T-MTT Apr 05 1515-1521 Mao Shau-Gang, see Shau-Gang Mao, T-MTT Nov 05 3460-3466 Maria, J.-P., see Nath, J., T-MTT Sep 05 2707-2712 Marini, S., see Taroncher, M., T-MTT Jun 05 2153-2163 Marques, R., see Garcia-Garcia, J., T-MTT Jun 05 1997-2006 Marquez-Segura, E., see Esteban, J., T-MTT Apr 05 1506-1514 Martel, J., see Velazquez-Ahumada, Md.C., T-MTT May 05 1823-1828 Marti, J., see Vidal, B., T-MTT Aug 05 2600-2603 Martin, C.B., see Ros, J.V.M., T-MTT Apr 05 1130-1142 Martin, F., see Baena, J.D., T-MTT Apr 05 1451-1461 Martin, F., see Garcia-Garcia, J., T-MTT Jun 05 1997-2006 Martinez, B.G., see Ros, J.V.M., T-MTT Apr 05 1130-1142 Martinez, B.G., see Mira, F., T-MTT Apr 05 1294-1303 Martinez, R., see Nikitin, P.V., T-MTT Sep 05 2721-2725 Martinez-Alajarin, J., see Domenech-Asensi, G., T-MTT Nov 05 3305-3311 Martinez-Guerrero, E., see Rayas-Sanchez, J.E., T-MTT Mar 05 960-968 Martin-Guerrero, T.M., see Esteban, J., T-MTT Apr 05 1506-1514 Martins, J.P., and N.B. Carvalho. Multitone phase and amplitude measurement for nonlinear device characterization; T-MTT Jun 05 19821989 Masalov, V.L., see Egorov, V.N., T-MTT Feb 05 627-635 Massa, A., see Donelli, M., T-MTT May 05 1761-1776 Massaro, A., see Pierantoni, L., T-MTT Jun 05 1856-1862 Massler, H., see Schneider, K., T-MTT Nov 05 3378-3387 Matoglu, E., see Mukherjee, S., T-MTT Nov 05 3621-3630 Matsuda, Y., see Chaki, S., T-MTT Jun 05 2073-2081 + Check author entry for coauthors

Matsuda, Y., see Ohta, A., T-MTT Jun 05 2121-2128 Matzler, C., see Luthi, T., T-MTT Aug 05 2594-2599 Mazumder, S., J.-P. Durand, S.L. Meyer, W.D. Weaver, J.V. Traverse, C.A. Rynas, G.E. Allshouse, J.E. Toland, Jr., and J.P. Biondi. High-band digital preprocessor (HBDP) for the AMRFC test-bed; T-MTT Mar 05 1065-1071 McCarthy, K.G., see Murphy, O.H., T-MTT Jun 05 2063-2072 McCartney, L., see Popovic, D., T-MTT May 05 1713-1722 McCartney, L., see Popovic, D., T-MTT Sep 05 3053 McCord, J., see Frommberger, M., T-MTT Jun 05 2096-2100 McCracken, S., see Feng Ling, T-MTT Jan 05 264-273 McDermitt, C.S., and F. Bucholtz. RF frequency shifting via optically switched dual-channel PZT fiber stretchers; T-MTT Dec 05 3782-3787 McErlean, E.P., see Jia-Sheng Hong, T-MTT Jun 05 1976-1981 McGrath, W.R., see Focardi, P., T-MTT May 05 1653-1661 McGruer, N.E., see Johnson, J., T-MTT Nov 05 3615-3620 McKinley, M.D., see Verspecht, J., T-MTT Apr 05 1369-1376 McLean, M., see Egorov, V.N., T-MTT Feb 05 627-635 Meazza, A., see Iommi, R., T-MTT Mar 05 874-880 Mediavilla, A., see Aja, B., T-MTT Jun 05 2050-2062 Medina, F., see Rodriguez-Berral, R., T-MTT May 05 1613-1623 Medina, F., see Leon, G., T-MTT May 05 1739-1744 Medina, F., see Velazquez-Ahumada, Md.C., T-MTT May 05 1823-1828 Mehdi, I., see Maestrini, A., T-MTT Sep 05 2835-2843 Meisong Tong, G. Pan, and Guangtsai Lei. Full-wave analysis of coupled lossy transmission lines using multiwavelet-based method of moments; TMTT Jul 05 2362-2370 Melcon, A.A., see Pereira, F.D.Q., T-MTT Jan 05 94-105 Melle, S., D. De Conto, D. Dubuc, K. Grenier, O. Vendier, J.-L. Muraro, J.-L. Cazaux, and R. Plana. Reliability modeling of capacitive RF MEMS; TMTT Nov 05 3482-3488 Melville, R., see Youngcheol Park, T-MTT Jan 05 115-122 Mendizabal, J., see Legarda, J., T-MTT Aug 05 2481-2486 Meng Cao, and R. Pietig. Ferrite coupled-line circulator with reduced length; T-MTT Aug 05 2572-2579 Menzel, W., and A. Balalem. Quasi-lumped suspended stripline filters and diplexers; T-MTT Oct 05 3230-3237 Mesa, F., see Rodriguez-Berral, R., T-MTT May 05 1613-1623 Metzger, A.G., see Tsai-Pi Hung, T-MTT Jan 05 144-151 Meyer, P., and W. Steyn. Authors' reply [to Comments on "A shorted waveguide-stub coupling mechanism for narrow-band multimode coupled resonator filters"]; T-MTT Jan 05 415 Meyer, S.L., see Mazumder, S., T-MTT Mar 05 1065-1071 Michielssen, E., see Yilmaz, A.E., T-MTT Sep 05 2851-2865 Midkiff, J.C., see Zhiyang Liu, T-MTT Sep 05 2949-2955 Miller, M.D., see Wangmyong Woo, T-MTT Jan 05 229-237 Milosavljevic, Z.D. Design of generalized Chebyshev filters with asymmetrically located transmission zeros; T-MTT Jul 05 2411-2415 Min Chen, see Youngcheol Park, T-MTT Jan 05 115-122 Ming-Da Tsai, Chin-Shen Lin, Chun-Hsien Lien, and Huei Wang. Broadband MMICs based on modified loss-compensation method using 0.35ȝm SiGe BiCMOS technology; T-MTT Feb 05 496-505 Ming-Dou Ker, and Bing-Jye Kuo. Decreasing-size distributed ESD protection scheme for broad-band RF circuits; T-MTT Feb 05 582-589 Ming-Dou Ker, Yuan-Wen Hsiao, and Bing-Jye Kuo. ESD protection design for 1- to 10-GHz distributed amplifier in CMOS technology; T-MTT Sep 05 2672-2681 Ming-Fong Lei, and Huei Wang. An analysis of miniaturized dual-mode bandpass filter structure using shunt-capacitance perturbation; T-MTT Mar 05 861-867 Ming-Hsiang Cho, Guo-Wei Huang, Lin-Kun Wu, Chia-Sung Chiu, YuehHua Wang, Kun-Ming Chen, Hua-Chou Tseng, and Tsun-Lai Hsu. A shield-based three-port de-embedding method for microwave on-wafer characterization of deep-submicrometer silicon MOSFETs; T-MTT Sep 05 2926-2934 Mingyan Fan, see Li Yang, T-MTT Jan 05 183-190 Ming Yu, see Daneshmand, M., T-MTT Jan 05 12-21 Ming Yu, see Cameron, R.J., T-MTT Nov 05 3288-3297 Ming-Yu Hsieh, see Shih-Ming Wang, T-MTT Feb 05 747-753 Min-Sou Wu, see Shau-Gang Mao, T-MTT Nov 05 3460-3466 Mira, F., M. Bressan, G. Conciauro, B.G. Martinez, and V.E.B. Esbert. Fast S-domain modeling of rectangular waveguides with radially symmetric metal insets; T-MTT Apr 05 1294-1303 Mirshekar-Syahkal, D., see Peng Wang, T-MTT Jan 05 349-353

IEEE T-MTT 2005 INDEX — 13 Mishra, C., A. Valdes-Garcia, F. Bahmani, A. Batra, E. Sanchez-Sinencio, and J. Silva-Martinez. Frequency planning and synthesizer architectures for multiband OFDM UWB radios; T-MTT Dec 05 3744-3756 Mishra, U.K., see Sanabria, C., T-MTT Feb 05 762-769 Mishra, V.V., see Srivastava, K.V., T-MTT Jun 05 1960-1967 Mittra, R., see Semouchkina, E., T-MTT Feb 05 644-652 Miyagawa, H., see Yamamoto, H., T-MTT Jun 05 2187-2195 Miyamoto, H., see Inoue, T., T-MTT Jan 05 74-80 Mizukoshi, T., see Chaki, S., T-MTT Jun 05 2073-2081 Mizusawa, N., and S. Kusunoki. Third- and fifth-order baseband component injection for linearization of the power amplifier in a cellular phone; TMTT Nov 05 3327-3334 Modelski, J.W. Guest editorial [special section into. on selected papers from the 15th International Conference on Microwaves, Radar, and Wireless Communications, MIKON]; T-MTT Feb 05 425-426 Modelski, J.W., see Yashchyshyn, Y., T-MTT Feb 05 427-438 Mohamed, A.S., see Bandler, J.W., T-MTT Sep 05 2801-2811 Mojarradi, M.M., see Kuhn, W.B., T-MTT Sep 05 2819-2825 Moldovan, E., see Tatu, S.O., T-MTT Sep 05 2768-2776 Molfino, R., see Dongjiang Qiao, T-MTT Mar 05 1089-1095 Montejo-Garai, J.R., J.A. Ruiz-Cruz, J.M. Rebollar, M.J. Padilla-Cruz, A. Onoro-Navarro, and I. Hidalgo-Carpintero. Synthesis and design of in-line N-order filters with N real transmission zeros by means of extracted poles implemented in low-cost rectangular H-plane waveguide; T-MTT May 05 1636-1642 Montejo-Garai, J.R., J.A. Ruiz-Cruz, and J.M. Rebollar. Full-wave design of H-plane contiguous manifold output multiplexers using the fictitious reactive load concept; T-MTT Aug 05 2628-2632 Monzo-Cabrera, J., see Plaza-Gonzalez, P., T-MTT May 05 1699-1706 Monzo-Cabrera, J., see Requena-Perez, M.E., T-MTT Jun 05 2114-2120 Moonkyun Maeng, see Hur, Y., T-MTT Jan 05 246-255 Moonkyun Maeng, F. Bien, Y. Hur, Hyoungsoo Kim, S. Chandramouli, E. Gebara, and J. Laskar. 0.18-ȝm CMOS equalization techniques for 10Gb/s fiber optical communication links; T-MTT Nov 05 3509-3519 Moreira, C.P., see Kerherve, E., T-MTT Jun 05 2145-2152 Morgan, J.M., see Williams, D.F., T-MTT Apr 05 1384-1389 Morin, G.A., see Hettak, K., T-MTT Jan 05 283-291 Morin, G.A., see Hettak, K., T-MTT May 05 1624-1635 Morini, A., see Farina, M., T-MTT Nov 05 3272-3280 Moritake, H., see Utsumi, Y., T-MTT Nov 05 3345-3353 Morschbach, M., A. Muller, C. Schollhorn, M. Oehme, T. Buck, and E. Kasper. Integrated silicon Schottky mixer diodes with cutoff frequencies above 1 THz; T-MTT Jun 05 2013-2018 Morsey, J.D., see Okhmatovski, V.I., T-MTT May 05 1829 Mortazawi, A., see Jonghoon Choi, T-MTT Nov 05 3407-3414 Mortazawi, A., see Schulwitz, L., T-MTT Nov 05 3588-3594 Morton, M., see Il Kwon Kim, T-MTT Sep 05 2943-2948 Moselhy, T.A., see Ghali, H.A., T-MTT Jun 05 1946-1950 Moss, C.D., see Grzegorczyk, T.M., T-MTT Sep 05 2956-2967 Mo Ting Ting, see Kam Man Shum, T-MTT Mar 05 895-900 Mousavi, P., see Daneshmand, M., T-MTT Jan 05 12-21 Moussessian, A., see Kuhn, W.B., T-MTT Sep 05 2819-2825 Mrozowski, M., see Kulas, L., T-MTT May 05 1587-1592 Mueller, J.-E., see Ceylan, N., T-MTT Feb 05 515-522 Mueller, S., A. Penirschke, C. Damm, P. Scheele, M. Wittek, C. Weil, and R. Jakoby. Broad-band microwave characterization of liquid crystals using a temperature-controlled coaxial transmission line; T-MTT Jun 05 19371945 Mukherjee, S., B. Mutnury, S. Dalmia, and M. Swaminathan. Layout-level synthesis of RF inductors and filters in LCP substrates for Wi-Fi applications; T-MTT Jun 05 2196-2210 Mukherjee, S., M. Swaminathan, and E. Matoglu. Statistical analysis and diagnosis methodology for RF circuits in LCP substrates; T-MTT Nov 05 3621-3630 Mukhopadhyay, R., Yunseo Park, P. Sen, N. Srirattana, Jongsoo Lee, Chang-Ho Lee, S. Nuttinck, A. Joseph, J.D. Cressler, and J. Laskar. Reconfigurable RFICs in Si-based technologies for a compact intelligent RF front-end; T-MTT Jan 05 81-93 Muller, A., see Morschbach, M., T-MTT Jun 05 2013-2018 Munkyo Seo, see Paidi, V.K., T-MTT Feb 05 598-605 Munkyo Seo, M.J.W. Rodwell, and U. Madhow. Comprehensive digital correction of mismatch errors for a 400-msamples/s 80-dB SFDR timeinterleaved analog-to-digital converter; T-MTT Mar 05 1072-1082 Murakowski, J., see Schuetz, C.A., T-MTT May 05 1732-1738 Muraro, J.-L., see Melle, S., T-MTT Nov 05 3482-3488 + Check author entry for coauthors

Murk, A., see Luthi, T., T-MTT Apr 05 1168-1173 Murphy, A.C., see Murphy, O.H., T-MTT Jun 05 2063-2072 Murphy, O.H., K.G. McCarthy, C.J.P. Delabie, A.C. Murphy, and P.J. Murphy. Design of multiple-metal stacked inductors incorporating an extended physical model; T-MTT Jun 05 2063-2072 Murphy, P.J., see Murphy, O.H., T-MTT Jun 05 2063-2072 Mutlu, A., see Coskun, A.H., T-MTT Jun 05 2171-2178 Mutnury, B., see Mukherjee, S., T-MTT Jun 05 2196-2210 N Nader Engheta, and R.W. Ziolkowski. A positive future for double-negative metamaterials; T-MTT Apr 05 1535-1556 Nah Kyung-Suc, see Young-Jin Kim, T-MTT Feb 05 606-613 Nakamura, K., see Suzuki, E., T-MTT Feb 05 696-701 Nakayama, T., see Inoue, T., T-MTT Jan 05 74-80 Nakhla, M.S., see Nakhla, N.M., T-MTT Nov 05 3520-3530 Nakhla, N.M., A. Dounavis, M.S. Nakhla, and R. Achar. Delay-extractionbased sensitivity analysis of multiconductor transmission lines with nonlinear terminations; T-MTT Nov 05 3520-3530 Nallatamby, J.-C., M. Prigent, and J. Obregon. On the role of the additive and converted noise in the generation of phase noise in nonlinear oscillators; T-MTT Mar 05 901-906 Nallatamby, J.-C., M. Prigent, M. Camiade, A. Sion, C. Gourdon, and J.J. Obregon. An advanced low-frequency noise model of GaInP-GaAs HBT for accurate prediction of phase noise in oscillators; T-MTT May 05 16011612 Nam, I., Bonkee Kim, and Kwyro Lee. CMOS RF amplifier and mixer circuits utilizing complementary Characteristics of parallel combined NMOS and PMOS devices; T-MTT May 05 1662-1671 Namgon Kim, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 Nam-Jin Oh, see Trung-Kien Nguyen, T-MTT Feb 05 538-547 Nam Joongjin, see Joongjin Nam, T-MTT Aug 05 2639-2644 Nam Sangwook, see Jong-Sik Lim, T-MTT Aug 05 2539-2545 Nanver, L.K., see Spirito, M., T-MTT Jul 05 2340-2347 Narisi Wang, Xinli Peng, V. Yousefzadeh, D. Maksimovic, S. Pajic, and Z. Popovic. Linearity of X-band class-E power amplifiers in EER operation; T-MTT Mar 05 1096-1102 Narisi Wang, see Pajic, S., T-MTT Sep 05 2899-2907 Nath, J., D. Ghosh, J.-P. Maria, A.I. Kingon, W. Fathelbab, P.D. Franzon, and M.B. Steer. An electronically tunable microstrip bandpass filter using thin-film Barium-Strontium-Titanate (BST) varactors; T-MTT Sep 05 2707-2712 Navarrini, A., see Engargiola, G., T-MTT May 05 1792-1801 Ndagijimana, F., see Williams, D.F., T-MTT Jan 05 314-321 Nefyodov, Y.A., see Egorov, V.N., T-MTT Feb 05 627-635 Nesimoglu, T., see Carey-Smith, B.E., T-MTT Feb 05 777-785 Neto, A., see Focardi, P., T-MTT May 05 1653-1661 Nguyen Cam, see Jeongwoo Han, T-MTT Jun 05 1875-1882 Nguyen Trung-Kien, see Trung-Kien Nguyen, T-MTT Feb 05 538-547 Nightingale, S.J., see Shohat, J., T-MTT Oct 05 3115-3120 Nightingale, S.J., see Shohat, J., T-MTT Dec 05 3767-3773 Nikitin, P.V., K.V.S. Rao, S.F. Lam, V. Pillai, R. Martinez, and H. Heinrich. Power reflection coefficient analysis for complex impedances in RFID tag design; T-MTT Sep 05 2721-2725 Nikku, M., see Grzegorczyk, T.M., T-MTT Apr 05 1443-1450 Nikolova, N.K., see Rickard, Y.S., T-MTT Jul 05 2274-2283 Nishikawa, K., see Seki, T., T-MTT Jun 05 2101-2106 Nishikawa, T., see Yamamoto, H., T-MTT Jun 05 2187-2195 Niu Guofu, see Guofu Niu, T-MTT Feb 05 506-514 Niu Guofu, see Qingqing Liang, T-MTT May 05 1745-1755 Noren, B., see O'Sullivan, T., T-MTT Jan 05 106-114 Norgren, M. Chebyshev collocation and Newton-type optimization methods for the inverse problem on nonuniform transmission lines; T-MTT May 05 1561-1568 Notaros, B.M., see Ilic, M.M., T-MTT Apr 05 1377-1383 Nouet, P., see Puyal, V., T-MTT Apr 05 1338-1344 Noyel, G., see Vincent, D., T-MTT Apr 05 1174-1180 Nuttinck, S., see Mukhopadhyay, R., T-MTT Jan 05 81-93 O Obregon, J., see Nallatamby, J.-C., T-MTT Mar 05 901-906 Obregon, J.J., see Nallatamby, J.-C., T-MTT May 05 1601-1612 Oehme, M., see Morschbach, M., T-MTT Jun 05 2013-2018

IEEE T-MTT 2005 INDEX — 14 Ofli, E., R. Vahldieck, and S. Amari. Novel E-plane filters and diplexers with elliptic response for millimeter-wave applications; T-MTT Mar 05 843851 Oh Donghoon, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 Ohira, M., H. Deguchi, M. Tsuji, and H. Shigesawa. Novel waveguide filters with multiple attenuation poles using dual-behavior resonance of frequency-selective surfaces; T-MTT Nov 05 3320-3326 Oh Nam-Jin, see Trung-Kien Nguyen, T-MTT Feb 05 538-547 Ohno, T., K. Wada, and O. Hashimoto. Design methodologies of planar duplexers and triplexers by manipulating attenuation poles; T-MTT Jun 05 2088-2095 Ohta, A., A. Inoue, S. Goto, K. Ueda, T. Ishikawa, and Y. Matsuda. Intermodulation distortion analysis of class-F and inverse class-F HBT amplifiers; T-MTT Jun 05 2121-2128 Oh Yong-Hun, see Trung-Kien Nguyen, T-MTT Feb 05 538-547 Okamoto, Y., see Inoue, T., T-MTT Jan 05 74-80 Okhmatovski, V.I., see Feng Ling, T-MTT Jan 05 264-273 Okhmatovski, V.I., J.D. Morsey, and A.C. Cangellaris. Authors' reply [to comments on ""Deembedding of port discontinuities in full-wave CAD models of multiport circuits"]; T-MTT May 05 1829 Okhmatovski, V.I., see Rautio, J.C., T-MTT Sep 05 2892-2898 Oki, A., see Yeong-Chang Chou, T-MTT Nov 05 3398-3406 Okoniewski, M., see Popovic, D., T-MTT May 05 1713-1722 Okoniewski, M., see Popovic, D., T-MTT Sep 05 3053 Olbrich, G.R., see Jung Han Choi, T-MTT Jun 05 2033-2042 Oliner, A.A., see Itoh, T., T-MTT Apr 05 1413-1417 Olivieri, A., see Virone, G., T-MTT Mar 05 888-894 Olivieri, M., G. Scotti, P. Tommasino, and A. Trifiletti. Necessary and sufficient conditions for the stability of microwave amplifiers with variable termination impedances; T-MTT Aug 05 2580-2586 Ong Beng Hwee, see Choon Beng Sia, T-MTT Sep 05 3035-3044 Onoro-Navarro, A., see Montejo-Garai, J.R., T-MTT May 05 1636-1642 Ooi Ban-Leong, see Boon Tiong Tan, T-MTT Jan 05 343-348 Ooi Ban-Leong, see Ban-Leong Ooi, T-MTT Jul 05 2329-2334 Orlianges, J.-C., see Pothier, A., T-MTT Jan 05 354-360 Orta, R., see Virone, G., T-MTT Mar 05 888-894 O'Sullivan, T., R.A. York, B. Noren, and P.M. Asbeck. Adaptive duplexer implemented using single-path and multipath feedforward techniques with BST phase shifters; T-MTT Jan 05 106-114 Ota, H., see Suzuki, E., T-MTT Feb 05 696-701 Ott, R., see Panaitov, G.I., T-MTT Nov 05 3371-3377 Ouaddari, M., S. Delprat, F. Vidal, M. Chaker, and Ke Wu. Microwave characterization of ferroelectric thin-film materials; T-MTT Apr 05 13901397 Ozturk, C., see JaeHyuk Shin, T-MTT Feb 05 636-643 P Pacheco, J., Jr., see Grzegorczyk, T.M., T-MTT Sep 05 2956-2967 Pacheco, P.S., see Ros, J.V.M., T-MTT Apr 05 1130-1142 Padilla-Cruz, M.J., see Montejo-Garai, J.R., T-MTT May 05 1636-1642 Pagani, M., see Iommi, R., T-MTT Mar 05 874-880 Pagani, M., see Raffo, A., T-MTT Nov 05 3449-3459 Page, J.E., see Esteban, J., T-MTT Apr 05 1506-1514 Paidi, V.K., Z. Griffith, Yun Wei, M. Dahlstrom, M. Urteaga, N. Parthasarathy, Munkyo Seo, L. Samoska, A. Fung, and M.J.W. Rodwell. G-band (140-220 GHz) and W-band (75-110 GHz) InP DHBT medium power amplifiers; T-MTT Feb 05 598-605 Pajic, S., see Narisi Wang, T-MTT Mar 05 1096-1102 Pajic, S., Narisi Wang, P.M. Watson, T.K. Quach, and Z. Popovic. X-band two-stage high-efficiency switched-mode power amplifiers; T-MTT Sep 05 2899-2907 Palacios, T., see Sanabria, C., T-MTT Feb 05 762-769 Palmer, W.D., see Adler, E.D., T-MTT Mar 05 1005-1008 Palmisano, G., see Girlando, G., T-MTT Mar 05 952-959 Palomba, F., see Raffo, A., T-MTT Nov 05 3449-3459 Pan, G., see Meisong Tong, T-MTT Jul 05 2362-2370 Panaitov, G.I., R. Ott, and N. Klein. Dielectric resonator with discrete electromechanical frequency tuning; T-MTT Nov 05 3371-3377 Papageorgiou, C.C., see Karanasiou, I.S., T-MTT May 05 1831-1832 Papaioannou, G., M.-N. Exarchos, V. Theonas, Guoan Wang, and J. Papapolymerou. Temperature study of the dielectric polarization effects of capacitive RF MEMS switches; T-MTT Nov 05 3467-3473 Papapolymerou, J., see Pothier, A., T-MTT Jan 05 354-360 Papapolymerou, J., see Ponchak, G.E., T-MTT Feb 05 713-717 + Check author entry for coauthors

Papapolymerou, J., see Pinel, S., T-MTT May 05 1707-1712 Papapolymerou, J., see Jong-Hoon Lee, T-MTT Jun 05 2220-2229 Papapolymerou, J., see Il Kwon Kim, T-MTT Sep 05 2943-2948 Papapolymerou, J., see Papaioannou, G., T-MTT Nov 05 3467-3473 Papapolymerou, J., see Lee, J.-H., T-MTT Dec 05 3817-3824 Paquet, S., see Qi, G., T-MTT Oct 05 3090-3097 Park Byeong-Ha, see Young-Jin Kim, T-MTT Feb 05 606-613 Parker, A.E., and J.G. Rathmell. Broad-band characterization of FET selfheating; T-MTT Jul 05 2424-2429 Parkhomenko, V.N., see Young-Jin Kim, T-MTT Feb 05 606-613 Park Jae-Hyoung, see Byoungjoong Kang, T-MTT Jan 05 134-143 Park Youngcheol, see Youngcheol Park, T-MTT Jan 05 115-122 Park Yunseo, see Mukhopadhyay, R., T-MTT Jan 05 81-93 Parssinen, A., see Sivonen, P., T-MTT Apr 05 1304-1313 Parthasarathy, N., see Paidi, V.K., T-MTT Feb 05 598-605 Pasalic, D., and R. Vahldieck. A hybrid drift-diffusion-TLM analysis of traveling-wave photodetectors; T-MTT Sep 05 2700-2706 Pascual, J.P., see Aja, B., T-MTT Jun 05 2050-2062 Patisang, S., see Chongcheawchamnan, M., T-MTT Jul 05 2458-2462 Paulotto, S., see Baccarelli, P., T-MTT Jan 05 32-44 Paulotto, S., see Baccarelli, P., T-MTT Apr 05 1431-1442 Pearson, L.W., see Xing Wang, T-MTT Jan 05 410-413 Pease, R.F.W., see Ioakeimidi, K., T-MTT Jan 05 336-342 Pechac, P., see Hudec, P., T-MTT Jun 05 2235-2239 Pedreno-Molina, J.L., see Requena-Perez, M.E., T-MTT Jun 05 2114-2120 Pedro, J.C., and N.B. Carvalho. Designing multisine excitations for nonlinear model testing; T-MTT Jan 05 45-54 Pedro, J.C., and S.A. Maas. A comparative overview of microwave and wireless power-amplifier behavioral modeling approaches; T-MTT Apr 05 1150-1163 Pedro Cheong, Si-Weng Fok, and Kam-Weng Tam. Miniaturized parallel coupled-line bandpass filter with spurious-response suppression; T-MTT May 05 1810-1816 Penaranda, J.A., see Legarda, J., T-MTT Aug 05 2481-2486 Peng, C.-J., see Hsien-Shun Wu, T-MTT Sep 05 2713-2720 Peng Song Tsuen, see Ching-Feng Lee, T-MTT Sep 05 3024-3034 Peng-Un Su A 0.25-ȝm CMOS OPLL transmitter IC for GSM and DCS applications; T-MTT Feb 05 462-471 Peng Wang, Lye Heng Chua, and D. Mirshekar-Syahkal. Accurate characterization of low-Q microwave resonator using critical-points method; T-MTT Jan 05 349-353 Peng Xinli, see Narisi Wang, T-MTT Mar 05 1096-1102 Penirschke, A., see Mueller, S., T-MTT Jun 05 1937-1945 Pepe, G., F.-J. Gortz, and H. Chaloupka. Sequential tuning of microwave filters using adaptive models and parameter extraction; T-MTT Jan 05 2231 Pereira, F.D.Q., P.V. Castejon, D.C. Rebenaque, J.P. Garcia, and A.A. Melcon. Numerical evaluation of the Green's functions for cylindrical enclosures by a new spatial images method; T-MTT Jan 05 94-105 Peroulis, D., see Lu, Y., T-MTT Nov 05 3672-3678 Peverini, O.A., see Virone, G., T-MTT Mar 05 888-894 Pfeiffer, U.R., see Zwick, T., T-MTT Mar 05 934-937 Pfeiffer, U.R., see Floyd, B.A., T-MTT Apr 05 1181-1188 Pfeiffer, U.R., and C. Schuster. A recursive un-termination method for nondestructive in situ S-parameter measurement of hermetically encapsulated packages; T-MTT Jun 05 1845-1855 Pham Anh-Vu, see Chen, A.C., T-MTT Nov 05 3648-3655 Philips, K.J.P., see Bergveld, H.J., T-MTT Feb 05 453-461 Phromloungsri, R., see Chongcheawchamnan, M., T-MTT Jul 05 2458-2462 Pienkowski, D., see Rykaczewski, P., T-MTT Mar 05 1056-1064 Pierantoni, L., A. Massaro, and T. Rozzi. Accurate modeling of TE/TM propagation and losses of integrated optical polarizer; T-MTT Jun 05 1856-1862 Pietig, R., see Meng Cao, T-MTT Aug 05 2572-2579 Pillai, V., see Nikitin, P.V., T-MTT Sep 05 2721-2725 Pillans, B., see Dongjiang Qiao, T-MTT Mar 05 1089-1095 Pinel, S., R. Bairavasubramanian, J. Laskar, and J. Papapolymerou. Compact planar and vialess composite low-pass filters using folded steppedimpedance resonator on liquid-Crystal-polymer substrate; T-MTT May 05 1707-1712 Pinel, S., see Jong-Hoon Lee, T-MTT Jun 05 2220-2229 Pinel, S., see Lee, J.-H., T-MTT Dec 05 3817-3824 Ping Yang, see Duochuan Li, T-MTT Mar 05 799-812 Ping Yang, see Duochuan Li, T-MTT Aug 05 2546-2551 Pin Yang Chek, see Holmes, J.E., T-MTT Jan 05 322-329

IEEE T-MTT 2005 INDEX — 15 Piotrowski, J.K., see Schimmer, O., T-MTT Jun 05 2107-2113 Piraux, L., see Saib, A., T-MTT Jun 05 2043-2049 Pisa, S., M. Cavagnaro, V. Lopresto, E. Piuzzi, G.A. Lovisolo, and P. Bernardi. A procedure to develop realistic numerical models of cellular phones for an accurate evaluation of SAR distribution in the human head; T-MTT Apr 05 1256-1265 Pistono, E., P. Ferrari, L. Duvillaret, J.-M. Duchamp, and R.G. Harrison. Hybrid narrow-band tunable bandpass filter based on varactor loaded electromagnetic-bandgap coplanar waveguides; T-MTT Aug 05 2506-2514 Piuzzi, E., see Pisa, S., T-MTT Apr 05 1256-1265 Pla, J.A., see Aaen, P.H., T-MTT Oct 05 3067-3074 Plana, R., see Melle, S., T-MTT Nov 05 3482-3488 Plaza-Gonzalez, P., J. Monzo-Cabrera, J.M. Catala-Civera, and D. SanchezHernandez. Effect of mode-stirrer configurations on dielectric heating performance in multimode microwave applicators; T-MTT May 05 16991706 Pokharel, R.K., M. Toyota, and O. Hashimoto. Analysis on effectiveness of wave absorbers to improve DSRC electromagnetic environment on express highway; T-MTT Sep 05 2726-2731 Polivka, M., see Hudec, P., T-MTT Jun 05 2235-2239 Ponchak, G.E., J. Papapolymerou, and M.M. Tentzeris. Excitation of coupled slotline mode in finite-ground CPW with unequal ground-plane widths; TMTT Feb 05 713-717 Popovic, D., L. McCartney, C. Beasley, M. Lazebnik, M. Okoniewski, S.C. Hagness, and J.H. Booske. Precision open-ended coaxial probes for in vivo and ex vivo dielectric spectroscopy of biological tissues at microwave frequencies; T-MTT May 05 1713-1722 Popovic, D., L. McCartney, C. Beasley, M. Lazebnik, M. Okoniewski, S.C. Hagness, and J.H. Booske. Corrections on "Precision open-ended coaxial probes for In Vivo and Ex Vivo dielectric spectroscopy of biological tissues at microwave frequencies" [May 05 1713-1722]; T-MTT Sep 05 3053 Popovic, Z., see Narisi Wang, T-MTT Mar 05 1096-1102 Popovic, Z., see Pajic, S., T-MTT Sep 05 2899-2907 Portillo, M.F., see Baena, J.D., T-MTT Apr 05 1451-1461 Posukh, V.G., see Semenov, S.Y., T-MTT Jul 05 2284-2294 Pothier, A., J.-C. Orlianges, Guizhen Zheng, C. Champeaux, A. Catherinot, D. Cros, P. Blondy, and J. Papapolymerou. Low-loss 2-bit tunable bandpass filters using MEMS DC contact switches; T-MTT Jan 05 354360 Powell, S.P., see Psiaki, M.L., T-MTT Oct 05 3082-3089 Prather, D.W., see Schuetz, C.A., T-MTT May 05 1732-1738 Prather, D.W., see Zhaolin Lu, T-MTT Apr 05 1362-1368 Presa, J., see Legarda, J., T-MTT Aug 05 2481-2486 Prigent, G., see Six, G., T-MTT Jan 05 301-305 Prigent, M., see Nallatamby, J.-C., T-MTT Mar 05 901-906 Prigent, M., see Nallatamby, J.-C., T-MTT May 05 1601-1612 Prunty, T.R., see Kaper, V.S., T-MTT Jan 05 55-65 Psiaki, M.L., S.P. Powell, H. Jung, and P.M. Kintner. Design and practical implementation of multifrequency RF front ends using direct RF sampling; T-MTT Oct 05 3082-3089 Purdy, D.S., see Adler, E.D., T-MTT Mar 05 1005-1008 Puyal, V., A. Konczykowska, P. Nouet, S. Bernard, S. Blayac, F. Jorge, M. Riet, and J. Godin. DC-100-GHz frequency doublers in InP DHBT technology; T-MTT Apr 05 1338-1344 Q Qi, G., J. Yao, J. Seregelyi, S. Paquet, and C. Belisle. Generation and distribution of a wide-band continuously tunable millimeter-wave signal with an optical external modulation technique; T-MTT Oct 05 3090-3097 Qiao Dongjiang, see Dongjiang Qiao, T-MTT Mar 05 1089-1095 Qi-Jun Zhang, see Lei Zhang, T-MTT Sep 05 2752-2767 Qingqing Liang, J.M. Andrews, J.D. Cressler, and Guofu Niu. Systematic linearity analysis of RFICs using a two-port lumped-nonlinear-source model; T-MTT May 05 1745-1755 Quach, T. Guest editorial [special section intro. on the 2004 IEEE Radio Frequency Integrated Circuits (RFIC) Symposium]; T-MTT Feb 05 451452 Quach, T.K., see Pajic, S., T-MTT Sep 05 2899-2907 Quandt, E., see Frommberger, M., T-MTT Jun 05 2096-2100 Quan Xue, see Kam Man Shum, T-MTT Mar 05 895-900 Quay, R., see Schneider, K., T-MTT Nov 05 3378-3387 Qun Wu, see Hai-Ying Yao, T-MTT Apr 05 1469-1476

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R Raffo, A., A. Santarelli, P.A. Traverso, G. Vannini, F. Palomba, F. Scappaviva, M. Pagani, and F. Filicori. Accurate pHEMT nonlinear modeling in the presence of low-frequency dispersive effects; T-MTT Nov 05 3449-3459 Raghavan, A., see Srirattana, N., T-MTT Mar 05 852-860 Raghavan, A., E. Gebara, E.M. Tentzeris, and J. Laskar. Analysis and design of an interference canceller for collocated radios; T-MTT Nov 05 34983508 Raghunathan, H., see Lawson, W., T-MTT Jan 05 372-379 Rahkonen, T., see Aikio, J.P., T-MTT Oct 05 3057-3066 Raisanen, A.V., see Koskinen, T., T-MTT Sep 05 2999-3006 Rambabu, K., and J. Bornemann. Simplified analysis technique for the initial design of LTCC filters with all-capacitive coupling; T-MTT May 05 17871791 Ramirez, F., see Suarez, A., T-MTT Sep 05 2743-2751 Randall, C.A., see Semouchkina, E.A., T-MTT Apr 05 1477-1487 Ran Lixin, see Yan Li, T-MTT Apr 05 1522-1526 Rao, K.V.S., see Nikitin, P.V., T-MTT Sep 05 2721-2725 Rappaport, C.M., see Kosmas, P., T-MTT Jul 05 2317-2323 Rathmell, J.G., see Parker, A.E., T-MTT Jul 05 2424-2429 Raugi, M., see Araneo, R., T-MTT Mar 05 907-918 Rauscher, C. Design of dielectric-filled cavity filters with ultrawide stopband Characteristics; T-MTT May 05 1777-1786 Rautio, J.C. Deembedding the effect of a local ground plane in electromagnetic analysis; T-MTT Feb 05 770-776 Rautio, J.C., and V.I. Okhmatovski. Unification of double-delay and SOC electromagnetic deembedding; T-MTT Sep 05 2892-2898 Rayas-Sanchez, J.E., F. Lara-Rojo, and E. Martinez-Guerrero. A linear inverse space-mapping (LISM) algorithm to design linear and nonlinear RF and microwave circuits; T-MTT Mar 05 960-968 Rebeiz, G.M., see Juo-Jung Hung, T-MTT Feb 05 754-761 Rebeiz, G.M., see Hancock, T.M., T-MTT Mar 05 977-983 Rebeiz, G.M., see Entesari, K., T-MTT Mar 05 1103-1110 Rebeiz, G.M., see Hancock, T.M., T-MTT Jul 05 2403-2410 Rebeiz, G.M., see Entesari, K., T-MTT Aug 05 2566-2571 Rebenaque, D.C., see Pereira, F.D.Q., T-MTT Jan 05 94-105 Rebollar, J.M., see Ruiz-Cruz, J.A., T-MTT Jan 05 174-182 Rebollar, J.M., see Montejo-Garai, J.R., T-MTT May 05 1636-1642 Rebollar, J.M., see Montejo-Garai, J.R., T-MTT Aug 05 2628-2632 Reimann, M., see Schoebel, J., T-MTT Jun 05 1968-1975 Rejaei, B., see Spirito, M., T-MTT Jul 05 2340-2347 Remley, K.A., see Williams, D.F., T-MTT Jan 05 314-321 Remley, K.A., see Verspecht, J., T-MTT Apr 05 1369-1376 Requena-Perez, M.E., J.L. Pedreno-Molina, J. Monzo-Cabrera, and A. DiazMorcillo. Multimode cavity efficiency optimization by optimum load location-experimental approach; T-MTT Jun 05 2114-2120 Reynolds, S.K., see Floyd, B.A., T-MTT Apr 05 1181-1188 Reznik, A.N. Comments on "Toward functional noninvasive imaging of excitable tissues inside the human body using focused microwave radiometry"; T-MTT May 05 1829-1831 Rhee Jin-Koo, see Young-Hoon Chun, T-MTT Feb 05 687-695 Rickard, Y.S., and N.K. Nikolova. Off-grid perfect boundary conditions for the FDTD method; T-MTT Jul 05 2274-2283 Riet, M., see Puyal, V., T-MTT Apr 05 1338-1344 Riska, J., see Varonen, M., T-MTT Apr 05 1322-1330 Rius Eric, see Six, G., T-MTT Jan 05 301-305 Rivas, J.G., see Berdel, K., T-MTT Apr 05 1266-1271 Robertson, I.D., see Shohat, J., T-MTT Oct 05 3115-3120 Robertson, I.D., see Shohat, J., T-MTT Dec 05 3767-3773 Robertson, I.D., see Stephens, D., T-MTT Dec 05 3832-3838 Roddis, N., see Aja, B., T-MTT Jun 05 2050-2062 Rodenbeck, C.T., S.-G. Kim, W.-H. Tu, M.R. Coutant, S. Hong, M. Li, and K. Chang. Ultra-wideband low-cost phased-array radars; T-MTT Dec 05 3697-3703 Rodin, Y., see Bogdashov, A., T-MTT Oct 05 3152-3155 Rodriguez-Berral, R., F. Mesa, and F. Medina. Appropriate formulation of the Characteristic equation for open nonreciprocal Layered waveguides with different upper and lower half-spaces; T-MTT May 05 1613-1623 Rodwell, M., see Shigematsu, H., T-MTT Feb 05 472-477 Rodwell, M.J.W., see Paidi, V.K., T-MTT Feb 05 598-605 Rodwell, M.J.W., see Munkyo Seo, T-MTT Mar 05 1072-1082

IEEE T-MTT 2005 INDEX — 16 Roelvink, J., and A.G. Williamson. Reactance of hollow, solid, and hemispherical-cap cylindrical posts in rectangular waveguide; T-MTT Oct 05 3156-3160 Rogers, S.D. Electromagnetic-bandgap layers for broad-band suppression of TEM modes in power planes; T-MTT Aug 05 2495-2505 Rolfes, I., and B. Schiek. Multiport method for the measurement of the scattering parameters of N-ports; T-MTT Jun 05 1990-1996 Rong Bifeng, see Spirito, M., T-MTT Jul 05 2340-2347 Rong Zeng, see Hong Wang, T-MTT Feb 05 564-570 Ronnow, D., see Isaksson, M., T-MTT Nov 05 3422-3428 Root, D.E., J. Verspecht, D. Sharrit, J. Wood, and A. Cognata. Broad-band poly-harmonic distortion (PHD) behavioral models from fast automated simulations and large-signal vectorial network measurements; T-MTT Nov 05 3656-3664 Ros, J.V.M., P.S. Pacheco, H.E. Gonzalez, V.E.B. Esbert, C.B. Martin, M.T. Calduch, S.C. Borras, and B.G. Martinez. Fast automated design of waveguide filters using aggressive space mapping with a new segmentation strategy and a hybrid optimization algorithm; T-MTT Apr 05 1130-1142 Rosenberg, U., see Amari, S., T-MTT Apr 05 1272-1279 Rosenberg, U., see Amari, S., T-MTT Oct 05 3135-3141 Rossi, P., see Brandolini, M., T-MTT Mar 05 1026-1038 Rouiller, T., see Vincent, D., T-MTT Apr 05 1174-1180 Roy, L., see El-Tager, A.M., T-MTT Jun 05 2211-2219 Rozzi, T., see Pierantoni, L., T-MTT Jun 05 1856-1862 Rozzi, T., see Farina, M., T-MTT Nov 05 3272-3280 Rudiakova, A.N. BJT class-F power amplifier near transition frequency; TMTT Sep 05 3045-3050 Ruey-Beei Wu, see Shih-Hao Lee, T-MTT Aug 05 2552-2558 Ruey-Beei Wu, see Chi-Feng Chen, T-MTT Sep 05 2688-2692 Ruiz-Cruz, J.A., M.A.E. Sabbagh, K.A. Zaki, J.M. Rebollar, and Yunchi Zhang. Canonical ridge waveguide filters in LTCC or metallic resonators; T-MTT Jan 05 174-182 Ruiz-Cruz, J.A., see Montejo-Garai, J.R., T-MTT May 05 1636-1642 Ruiz-Cruz, J.A., see Montejo-Garai, J.R., T-MTT Aug 05 2628-2632 Runtao Ding, see Lei Zhang, T-MTT Sep 05 2752-2767 Russchenberg, H.W.J., see Yanovsky, F.J., T-MTT Feb 05 444-450 Russer, P. Editorial [special issue intro. on 34th European Microwave Conference]; T-MTT Jun 05 1935-1936 Russer, P., see Jung Han Choi, T-MTT Jun 05 2033-2042 Russer, P., see Braun, S., T-MTT Nov 05 3354-3363 Russer, P., see Lorenz, P., T-MTT Nov 05 3631-3637 Rutledge, D.B., see Seungwoo Kim, T-MTT Jan 05 380-388 Rutledge, D.B., see Jeon, S., T-MTT Dec 05 3712-3722 Rykaczewski, P., D. Pienkowski, R. Circa, and B. Steinke. Signal path optimization in software-defined radio systems; T-MTT Mar 05 1056-1064 Rynas, C.A., see Mazumder, S., T-MTT Mar 05 1065-1071 S Saavedra, C.E., and You Zheng. Ring-hybrid microwave voltage-variable attenuator using HFET transistors; T-MTT Jul 05 2430-2434 Sabbagh, M.A.E., see Ruiz-Cruz, J.A., T-MTT Jan 05 174-182 Saib, A., M. Darques, L. Piraux, D. Vanhoenacker-Janvier, and I. Huynen. An unbiased integrated microstrip circulator based on magnetic nanowired substrate; T-MTT Jun 05 2043-2049 Saily, J., see Koskinen, T., T-MTT Sep 05 2999-3006 Saito, K., see Utsumi, Y., T-MTT Nov 05 3345-3353 Sakamoto, S.R., see JaeHyuk Shin, T-MTT Feb 05 636-643 Salehi, H., and R. Mansour. Analysis, modeling, and applications of coaxial waveguide-based left-handed transmission lines; T-MTT Nov 05 34893497 Samoska, L., see Paidi, V.K., T-MTT Feb 05 598-605 Sanabria, C., Hongtao Xu, T. Palacios, A. Chakraborty, S. Heikman, U.K. Mishra, and R.A. York. Influence of epitaxial structure in the noise figure of AlGaN/GaN HEMTs; T-MTT Feb 05 762-769 Sanchez-Hernandez, D., see Plaza-Gonzalez, P., T-MTT May 05 1699-1706 Sanchez-Renedo, M., R. Gomez-Garcia, J.I. Alonso, and C. Briso-Rodriguez. Tunable combline filter with continuous control of center frequency and bandwidth; T-MTT Jan 05 191-199 Sanchez-Sinencio, E., see Mishra, C., T-MTT Dec 05 3744-3756 Sang-Gug Lee, see Trung-Kien Nguyen, T-MTT Feb 05 538-547 Sang-Gug Lee, see Choong-Yul Cha, T-MTT Mar 05 881-887 Sang-Hyun Baek, see Il-Joo Cho, T-MTT Jul 05 2450-2457

+ Check author entry for coauthors

Sangiamwong, J., K. Tsukamoto, and S. Komaki. Frequency channel blocking scheme in mesh-topology millimeter-wave broad band entrance networks; T-MTT Dec 05 3723-3730 Sangsoo Ko, Jeong-Geun Kim, Taeksang Song, Euisik Yoon, and Songcheol Hong. K- and Q-bands CMOS frequency sources with X-band quadrature VCO; T-MTT Sep 05 2789-2800 Sang Tzu-Hsien, see Hu, R., T-MTT Jul 05 2398-2402 Sang-Won Yun, see Young-Hoon Chun, T-MTT Feb 05 687-695 Sangwook Nam, see Jong-Sik Lim, T-MTT Aug 05 2539-2545 Santarelli, A., see Raffo, A., T-MTT Nov 05 3449-3459 Santra, M., and K.U. Limaye. Estimation of complex permittivity of arbitrary shape and size dielectric samples using cavity measurement technique at microwave frequencies; T-MTT Feb 05 718-722 Sarabandi, K., see Aryanfar, F., T-MTT Apr 05 1288-1293 Sarkar, S., see Jong-Hoon Lee, T-MTT Jun 05 2220-2229 Sarris, C.D., see Kokkinos, T., T-MTT Apr 05 1488-1495 Sasaki, Y., see Chaki, S., T-MTT Jun 05 2073-2081 Sato, R., see Suzuki, E., T-MTT Feb 05 696-701 Savio Choi, see Daneshmand, M., T-MTT Jan 05 12-21 Sayed, A., and G. Boeck. Two-stage ultrawide-band 5-W power amplifier using SiC MESFET; T-MTT Jul 05 2441-2449 Scappaviva, F., see Raffo, A., T-MTT Nov 05 3449-3459 Scheele, P., see Mueller, S., T-MTT Jun 05 1937-1945 Schiek, B., see Rolfes, I., T-MTT Jun 05 1990-1996 Schimmer, O., A. Gulck, F. Daschner, J.K. Piotrowski, and R.H. Knochel. Noncontacting determination of moisture content in bulk materials using sub-nanosecond UWB pulses; T-MTT Jun 05 2107-2113 Schlecht, E., see Maestrini, A., T-MTT Sep 05 2835-2843 Schlechtweg, M., see Schneider, K., T-MTT Nov 05 3378-3387 Schmutz, C., see Frommberger, M., T-MTT Jun 05 2096-2100 Schneider, G.J., see Schuetz, C.A., T-MTT May 05 1732-1738 Schneider, K., R. Driad, R.E. Makon, H. Massler, M. Ludwig, R. Quay, M. Schlechtweg, and G. Weimann. Comparison of InP/InGaAs DHBT distributed amplifiers as modulator drivers for 80-Gbit/s operation; T-MTT Nov 05 3378-3387 Schneider, M., see Schoebel, J., T-MTT Jun 05 1968-1975 Schoebel, J., T. Buck, M. Reimann, M. Ulm, M. Schneider, A. Jourdain, G.J. Carchon, and H.A.C. Tilmans. Design considerations and technology assessment of phased-array antenna systems with RF MEMS for automotive radar applications; T-MTT Jun 05 1968-1975 Schollhorn, C., see Morschbach, M., T-MTT Jun 05 2013-2018 Scholten, A.J., see Tiemeijer, L.F., T-MTT Sep 05 2917-2925 Schreurs, D., see Verspecht, J., T-MTT Apr 05 1369-1376 Schuetz, C.A., J. Murakowski, G.J. Schneider, and D.W. Prather. Radiometric Millimeter-wave detection via optical upconversion and carrier suppression; T-MTT May 05 1732-1738 Schuetz, C.A., see Zhaolin Lu, T-MTT Apr 05 1362-1368 Schulwitz, L., and A. Mortazawi. A compact dual-polarized multibeam phased-array architecture for millimeter-wave radar; T-MTT Nov 05 35883594 Schuster, C., see Pfeiffer, U.R., T-MTT Jun 05 1845-1855 Schwolen, A., see Fickenscher, T., T-MTT Jul 05 2375-2382 Scott, J.B. Investigation of a method to improve VNA calibration in planar dispersive media through adding an asymmetrical reciprocal device; TMTT Sep 05 3007-3013 Scotti, G., see Olivieri, M., T-MTT Aug 05 2580-2586 See Taur Lee, see Sher Jiun Fang, T-MTT Feb 05 478-487 Sehgal, A., T. Mangla, S. Chopra, M. Gupta, and R.S. Gupta. Sub-threshold analysis and drain current modeling of polysilicon thin-film transistor using Green's function approach; T-MTT Sep 05 2682-2687 Se-Ho You, and E.F. Kuester. Guaranteed passive direct lumped-element modeling of transmission lines; T-MTT Sep 05 2826-2834 Seki, T., N. Honma, K. Nishikawa, and K. Tsunekawa. Millimeter-wave highefficiency multilayer parasitic microstrip antenna array on teflon substrate; T-MTT Jun 05 2101-2106 Semenov, S.Y., A.E. Bulyshev, A. Abubakar, V.G. Posukh, Y.E. Sizov, A.E. Souvorov, P.M. van den Berg, and T.C. Williams. Microwavetomographic imaging of the high dielectric-contrast objects using different image-reconstruction approaches; T-MTT Jul 05 2284-2294 Semouchkin, G.B., see Semouchkina, E., T-MTT Feb 05 644-652 Semouchkin, G.B., see Semouchkina, E.A., T-MTT Apr 05 1477-1487 Semouchkina, E., A. Baker, G.B. Semouchkin, M. Lanagan, and R. Mittra. New approaches for designing microstrip filters utilizing mixed dielectrics; T-MTT Feb 05 644-652

IEEE T-MTT 2005 INDEX — 17 Semouchkina, E.A., G.B. Semouchkin, M. Lanagan, and C.A. Randall. FDTD study of resonance Processes in metamaterials; T-MTT Apr 05 1477-1487 Sen, P., see Mukhopadhyay, R., T-MTT Jan 05 81-93 Senguttuvan, R., see Bhattacharya, S., T-MTT Nov 05 3474-3481 Seng Yeo Kiat, see Choon Beng Sia, T-MTT Sep 05 3035-3044 Seo Kwang-Seok, see Chang-Soon Choi, T-MTT Jan 05 256-263 Seo Munkyo, see Paidi, V.K., T-MTT Feb 05 598-605 Seo Munkyo, see Munkyo Seo, T-MTT Mar 05 1072-1082 Serdijn, W.A., see Tasic, A., T-MTT Feb 05 556-563 Seregelyi, J., see Qi, G., T-MTT Oct 05 3090-3097 Seung-Ho Hong, see Seung-Yup Lee, T-MTT Feb 05 786-793 Seungwoo Kim, Kyungho Lee, Jongwoo Lee, Bumman Kim, S.D. Kee, I. Aoki, and D.B. Rutledge. An optimized design of distributed active transformer; T-MTT Jan 05 380-388 Seung-Yup Lee, Yong-Sub Lee, Seung-Ho Hong, Hyun-Sik Choi, and YoonHa Jeong. An adaptive predistortion RF power amplifier with a spectrum monitor for multicarrier WCDMA applications; T-MTT Feb 05 786-793 Sewell, P., T.M. Benson, C. Christopoulos, D.W.P. Thomas, A. Vukovic, and J.G. Wykes. Transmission-line modeling (TLM) based upon unstructured tetrahedral meshes; T-MTT Jun 05 1919-1928 Shao Ying Huang, and Yee Hui Lee. Tapered dual-plane compact electromagnetic bandgap microstrip filter structures; T-MTT Sep 05 26562664 Shao Zhenhai, see Zhenhai Shao, T-MTT Jul 05 2261-2266 Shapiro, M.A., J.P. Anderson, and R.J. Temkin. Synthesis of gyrotron phasecorrecting mirrors using irradiance moments; T-MTT Aug 05 2610-2615 Sharma, A., see Yeong-Chang Chou, T-MTT Nov 05 3398-3406 Sharma, S.B., V.K. Singh, and S. Chakrabarty. Multifrequency waveguide orthomode transducer; T-MTT Aug 05 2604-2609 Sharp, J., see Helszajn, J., T-MTT Jul 05 2309-2316 Sharrit, D., see Root, D.E., T-MTT Nov 05 3656-3664 Shau-Gang Mao, Shiou-Li Chen, and Chen-Wei Huang. Effective electromagnetic parameters of novel distributed left-handed microstrip lines; T-MTT Apr 05 1515-1521 Shau-Gang Mao, Min-Sou Wu, Yu-Zhi Chueh, and Chun Hsiung Chen. Modeling of symmetric composite right/left-handed coplanar waveguides with applications to compact bandpass filters; T-MTT Nov 05 3460-3466 Shealy, J.R., see Kaper, V.S., T-MTT Jan 05 55-65 She Jingzhao, see Li Yang, T-MTT Jan 05 183-190 Sheng Sun, and Lei Zhu. Periodically nonuniform coupled microstrip-line filters with harmonic suppression using transmission zero reallocation; TMTT May 05 1817-1822 Sheng Sun, and Lei Zhu. Guided-wave characteristics of periodically nonuniform coupled microstrip lines-even and odd modes; T-MTT Apr 05 1221-1227 Sheng Sun, and Lei Zhu. Stopband-enhanced and size-miniaturized low-pass filters using high-impedance property of offset finite-ground microstrip line; T-MTT Sep 05 2844-2850 Sher Jiun Fang, A. Bellaouar, See Taur Lee, and D.J. Allstot. An imagerejection down-converter for low-IF receivers; T-MTT Feb 05 478-487 Shevchun, A.F., see Egorov, V.N., T-MTT Feb 05 627-635 She Wing Han, see Xun Gong, T-MTT Nov 05 3638-3647 Shey-Shi Lu, see Hung-Wei Chiu, T-MTT Mar 05 813-824 Shigematsu, H., T. Hirose, F. Brewer, and M. Rodwell. Millimeter-wave CMOS circuit design; T-MTT Feb 05 472-477 Shigesawa, H., see Ohira, M., T-MTT Nov 05 3320-3326 Shih-Hao Lee, Ting-Yi Huang, and Ruey-Beei Wu. Fast waveguide eigenanalysis by wide-band finite-element model-order reduction; T-MTT Aug 05 2552-2558 Shih-Ming Wang, Chun-Hsiang Chi, Ming-Yu Hsieh, and Chi-Yang Chang. Miniaturized spurious passband suppression microstrip filter using meandered parallel coupled lines; T-MTT Feb 05 747-753 Shin JaeHyuk, see JaeHyuk Shin, T-MTT Feb 05 636-643 Shin Jin-Ho, see Joongjin Nam, T-MTT Aug 05 2639-2644 Shiou-Li Chen, see Shau-Gang Mao, T-MTT Apr 05 1515-1521 Shi Shouyuan, see Zhaolin Lu, T-MTT Apr 05 1362-1368 Shohat, J., I.D. Robertson, and S.J. Nightingale. 10-Gb/s driver amplifier using a tapered gate line for improved input matching; T-MTT Oct 05 3115-3120 Shohat, J., I.D. Robertson, and S.J. Nightingale. Investigation of drain-line loss and the S22 kink effect in capacitively coupled distributed amplifiers; T-MTT Dec 05 3767-3773 Shouyuan Shi, see Zhaolin Lu, T-MTT Apr 05 1362-1368 Shuen-Chien Chang, see Chang, S.-F.R., T-MTT Mar 05 1048-1055 + Check author entry for coauthors

Shuiping Luo, and Zhizhang Chen. Iterative methods for extracting causal time-domain parameters; T-MTT Mar 05 969-976 Shumin Wang, and Ji Chen. Pre-iterative ADI-FDTD method for conductive medium; T-MTT Jun 05 1913-1918 Shum Kam Man, see Kam Man Shum, T-MTT Mar 05 895-900 Shyh-Jong Chung, see Chun-Fu Chang, T-MTT Jul 05 2383-2388 Sia Choon Beng, see Choon Beng Sia, T-MTT Sep 05 3035-3044 Siddiqui, O.F., see Eleftheriades, G.V., T-MTT Jan 05 396-403 Sieiro Cordoba, J.J., see Lopez-Villegas, J.M., T-MTT Dec 05 3757-3766 Sik Cho Choon, see Jaeheung Kim, T-MTT Aug 05 2622-2627 Sill, J.M., and E.C. Fear. Tissue sensing adaptive radar for breast cancer detection - experimental investigation of simple tumor models; T-MTT Nov 05 3312-3319 Sillero, R.M., see Baena, J.D., T-MTT Apr 05 1451-1461 Silva-Martinez, J., see Mishra, C., T-MTT Dec 05 3744-3756 Silveirinha, M.G., and C.A. Fernandes. Homogenization of 3-D-connected and nonconnected wire metamaterials; T-MTT Apr 05 1418-1430 Simonetto, A., O. D'Arcangelo, and L. Figini. Effect of cable length in vector measurements of very long millimeter wave components; T-MTT Dec 05 3731-3734 Simovsky, C., see Vincent, D., T-MTT Apr 05 1174-1180 Singh, S.P., see Gupta, R.C., T-MTT Sep 05 2665-2671 Singh, V.K., see Sharma, S.B., T-MTT Aug 05 2604-2609 Sinsky, J.H., M. Duelk, and A. Adamiecki. High-speed electrical backplane transmission using duobinary signaling; T-MTT Jan 05 152-160 Sin-Ting Chen, see Tzong-Lin Wu, T-MTT Sep 05 2935-2942 Sion, A., see Nallatamby, J.-C., T-MTT May 05 1601-1612 Sirbu, M., S.B.P. Lepaul, and F. Aniel. Coupling 3-D Maxwell's and Boltzmann's equations for analyzing a terahertz photoconductive switch; T-MTT Sep 05 2991-2998 Sirois, J., S. Boumaiza, M. Helaoui, G. Brassard, and F.M. Ghannouchi. A robust modeling and design approach for dynamically loaded and digitally linearized Doherty amplifiers; T-MTT Sep 05 2875-2883 Sivonen, P., and A. Parssinen. Analysis and optimization of packaged inductively degenerated common-source low-noise amplifiers with ESD protection; T-MTT Apr 05 1304-1313 Si-Weng Fok, see Pedro Cheong, T-MTT May 05 1810-1816 Six, G., G. Prigent, Eric Rius, G. Dambrine, and H. Happy. Fabrication and characterization of low-loss TFMS on silicon substrate up to 220 GHz; TMTT Jan 05 301-305 Sizov, Y.E., see Semenov, S.Y., T-MTT Jul 05 2284-2294 Smerzi, S.A., see Girlando, G., T-MTT Mar 05 952-959 Smith, P.A., see Holmes, J.E., T-MTT Jan 05 322-329 So, P.P.M., Huilian Du, and W.J.R. Hoefer. Modeling of metamaterials with negative refractive index using 2-D shunt and 3-D SCN TLM networks; TMTT Apr 05 1496-1505 Solar, H., see Legarda, J., T-MTT Aug 05 2481-2486 Songcheol Hong, see Sangsoo Ko, T-MTT Sep 05 2789-2800 Song Taeksang, see Il-Joo Cho, T-MTT Jul 05 2450-2457 Song Taeksang, see Sangsoo Ko, T-MTT Sep 05 2789-2800 Song Tsuen Peng, see Ching-Feng Lee, T-MTT Sep 05 3024-3034 Son Young-Suk, see Young-Jin Kim, T-MTT Feb 05 606-613 Soo Chang Ik, see Hyeong Tae Jeong, T-MTT Aug 05 2587-2593 Soo Chee Binn, see Fusco, V., T-MTT Feb 05 730-738 Sorolla, M., see Baena, J.D., T-MTT Apr 05 1451-1461 Sorolla, M., see Garcia-Garcia, J., T-MTT Jun 05 1997-2006 Souvorov, A.E., see Semenov, S.Y., T-MTT Jul 05 2284-2294 Spirito, M., F.M. De Paola, L.K. Nanver, E. Valletta, Bifeng Rong, B. Rejaei, L.C.N. de Vreede, and J.N. Burghartz. Surface-passivated high-resistivity silicon as a true microwave substrate; T-MTT Jul 05 2340-2347 Spowart, M.P., and E.F. Kuester. An orthogonality-based deembedding technique for microstrip networks; T-MTT Mar 05 938-946 Srirattana, N., see Mukhopadhyay, R., T-MTT Jan 05 81-93 Srirattana, N., A. Raghavan, D. Heo, P.E. Allen, and J. Laskar. Analysis and design of a high-efficiency multistage Doherty power amplifier for wireless communications; T-MTT Mar 05 852-860 Srisathit, S., see Chongcheawchamnan, M., T-MTT Jul 05 2458-2462 Srivastava, K.V., V.V. Mishra, and A. Biswas. A modified ring dielectric resonator with improved mode separation and its tunability characteristics in MIC environment; T-MTT Jun 05 1960-1967 Stapleton, S.P., see Kyoung-Joon Cho, T-MTT Jan 05 292-300 Steer, M. Editorial [special section intro. on 2004 IEEE MTT-S International Microwave Symposium]; T-MTT Jan 05 3 Steer, M.B., see Fathelbab, W.M., T-MTT Mar 05 1111-1116 Steer, M.B., see Fathelbab, W.M., T-MTT May 05 1569-1575

IEEE T-MTT 2005 INDEX — 18 Steer, M.B., see Gharaibeh, K.M., T-MTT May 05 1682-1692 Steer, M.B., see Fathelbab, W.M., T-MTT Apr 05 1211-1220 Steer, M.B., see Gard, K.G., T-MTT Jun 05 2179-2186 Steer, M.B. Special section on the Asia-Pacific Microwave Conference; TMTT Sep 05 2649 Steer, M.B., see Nath, J., T-MTT Sep 05 2707-2712 Steer, M.B., see Fathelbab, W.M., T-MTT Dec 05 3774-3781 Steinke, B., see Rykaczewski, P., T-MTT Mar 05 1056-1064 Stengel, B., see Choongeol Cho, T-MTT Apr 05 1197-1202 Stephens, D., P.R. Young, and I.D. Robertson. Millimeter-wave substrate integrated waveguides and filters in photoimageable thick-film technology; T-MTT Dec 05 3832-3838 Steyn, W., see Meyer, P., T-MTT Jan 05 415 Strandberg, R., P. Andreani, and L. Sundstrom. Spectrum emission considerations for baseband-modeled CALLUM architectures; T-MTT Feb 05 660-669 Stubbs, M.G., see Hettak, K., T-MTT Jan 05 283-291 Stubbs, M.G., see Hettak, K., T-MTT May 05 1624-1635 Suarez, A., and F. Ramirez. Analysis of stabilization circuits for phase-noise reduction in microwave oscillators; T-MTT Sep 05 2743-2751 Suarez, A., see Collado, A., T-MTT Sep 05 2777-2788 Suarez, A., see Jeon, S., T-MTT Dec 05 3712-3722 Su Hieng Tiong, see Yi Wang, T-MTT Jul 05 2348-2354 Su Ke-Ying, see Ke-Ying Su, T-MTT Mar 05 993-999 Sun, M., and Y.P. Zhang. Performance of inter-chip RF-interconnect using CPW, capacitive coupler, and UWB transceiver; T-MTT Sep 05 26502655 Sun Din-Kow, see Din-Kow Sun, T-MTT Mar 05 984-992 Sundstrom, L., see Strandberg, R., T-MTT Feb 05 660-669 Sungjae Lee, and K.J. Webb. The influence of transistor nonlinearities on noise properties; T-MTT Apr 05 1314-1321 Sungjoon Cho, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 Sungjoon Lim, C. Caloz, and T. Itoh. Metamaterial-based electronically controlled transmission-line structure as a novel leaky-wave antenna with tunable radiation angle and beamwidth; T-MTT Jan 05 161-173 Sungkyu Lim, see Byoungjoong Kang, T-MTT Jan 05 134-143 Sun Guilin, see Guilin Sun, T-MTT Mar 05 832-842 Sungweon Kang, see Jaewoo Lee, T-MTT Nov 05 3335-3344 Sun Sheng, see Sheng Sun, T-MTT May 05 1817-1822 Sun Sheng, see Sheng Sun, T-MTT Apr 05 1221-1227 Sun Sheng, see Sheng Sun, T-MTT Sep 05 2844-2850 Su Peng-Un, see Peng-Un Su, T-MTT Feb 05 462-471 Suzuki, E., S. Arakawa, H. Ota, K.I. Arai, R. Sato, and K. Nakamura. EO probe for simultaneous electric and magnetic near-field measurements using LiNbO3 with inverted domain; T-MTT Feb 05 696-701 Svelto, F., see Brandolini, M., T-MTT Mar 05 1026-1038 Swaminathan, M., see Mukherjee, S., T-MTT Jun 05 2196-2210 Swaminathan, M., see Mukherjee, S., T-MTT Nov 05 3621-3630 T Tae Jeong Hyeong, see Hyeong Tae Jeong, T-MTT Aug 05 2587-2593 Taek-Kyung Lee, see Duk-Jae Woo, T-MTT Jun 05 2139-2144 Taeksang Song, see Il-Joo Cho, T-MTT Jul 05 2450-2457 Taeksang Song, see Sangsoo Ko, T-MTT Sep 05 2789-2800 Tah-Hsiung Chu, see Chao-Hsiung Tseng, T-MTT Sep 05 2884-2891 Taijun Liu, S. Boumaiza, and F.M. Ghannouchi. Deembedding static nonlinearities and accurately identifying and modeling memory effects in wide-band RF transmitters; T-MTT Nov 05 3578-3587 Tamiazzo, S., and G. Macchiarella. An analytical technique for the synthesis of cascaded N-tuplets cross-coupled resonators microwave filters using matrix rotations; T-MTT May 05 1693-1698 Tamiazzo, S., see Macchiarella, G., T-MTT Nov 05 3265-3271 Tam Kam-Weng, see Pedro Cheong, T-MTT May 05 1810-1816 Tam Wai-Yip, see King-Yuen Wong, T-MTT Nov 05 3364-3370 Tan, B.T., see Luo, W.Q., T-MTT Oct 05 3191-3198 Tan, S.Y., see Luo, W.Q., T-MTT Oct 05 3191-3198 Tan Boon Tiong, see Boon Tiong Tan, T-MTT Jan 05 343-348 Tang Hong Jun, see Yu Lin Zhang, T-MTT Apr 05 1280-1287 Tang Jin, see Guofu Niu, T-MTT Feb 05 506-514 Taroncher, M., A. Vidal, V.E. Boria-Esbert, S. Marini, S. Cogollos, J. Gil, and B. Gimeno. CAD of complex passive devices composed of arbitrarily shaped waveguides using Nystrom and BI-RME methods; T-MTT Jun 05 2153-2163 Tascone, R., see Virone, G., T-MTT Mar 05 888-894 + Check author entry for coauthors

Tasic, A., W.A. Serdijn, and J.R. Long. Design of multistandard adaptive voltage-controlled oscillators; T-MTT Feb 05 556-563 Tat-Soon Yeo, see Hai-Ying Yao, T-MTT Apr 05 1469-1476 Tatu, S.O., E. Moldovan, Ke Wu, R.G. Bosisio, and T.A. Denidni. Ka-band analog front-end for software-defined direct conversion receiver; T-MTT Sep 05 2768-2776 Taur Lee See, see Sher Jiun Fang, T-MTT Feb 05 478-487 Tavik, G.C., C.L. Hilterbrick, J.B. Evins, J.J. Alter, J.G. Crnkovich, Jr., J.W. de Graaf, W. Habicht, II, G.P. Hrin, S.A. Lessin, D.C. Wu, and S.M. Hagewood. The advanced multifunction RF concept; T-MTT Mar 05 1009-1020 Taylor, K.M., and D.W. van der Weide. Ultra-sensitive detection of protein thermal unfolding and refolding using near-zone microwaves; T-MTT May 05 1576-1586 Temkin, R.J., see Woskov, P.P., T-MTT Jun 05 1863-1869 Temkin, R.J., see Shapiro, M.A., T-MTT Aug 05 2610-2615 Tentzeris, E.M., see Raghavan, A., T-MTT Nov 05 3498-3508 Tentzeris, M.M., see Ponchak, G.E., T-MTT Feb 05 713-717 Tentzeris, M.M., see Jong-Hoon Lee, T-MTT Jun 05 2220-2229 Tentzeris, M.M., see Il Kwon Kim, T-MTT Sep 05 2943-2948 Tentzeris, M.M., see Lee, J.-H., T-MTT Dec 05 3817-3824 Teppati, V., and A. Ferrero. On-wafer calibration algorithm for partially leaky multiport vector network analyzers; T-MTT Nov 05 3665-3671 Tewes, M., see Frommberger, M., T-MTT Jun 05 2096-2100 Theonas, V., see Papaioannou, G., T-MTT Nov 05 3467-3473 Thiel, M., see Truong Vu Bang Giang, T-MTT Jan 05 404-409 Thomas, D.W.P., see Sewell, P., T-MTT Jun 05 1919-1928 Thompson, R.M., see Kaper, V.S., T-MTT Jan 05 55-65 Tiebout, M., see Hermann, C., T-MTT Feb 05 488-495 Tiemeijer, L.F., R.J. Havens, A.B.M. Jansman, and Y. Bouttement. Comparison of the "pad-open-short" and "open-short-load" deembedding techniques for accurate on-wafer RF characterization of high-quality passives; T-MTT Feb 05 723-729 Tiemeijer, L.F., R.J. Havens, R. de Kort, and A.J. Scholten. Improved Yfactor method for wide-band on-wafer noise-parameter measurements; TMTT Sep 05 2917-2925 Tiiliharju, E., and K.A.I. Halonen. An active differential broad-band phase splitter for quadrature-modulator applications; T-MTT Feb 05 679-686 Tilmans, H.A.C., see Schoebel, J., T-MTT Jun 05 1968-1975 Ting-Kuang Wang, see Tzong-Lin Wu, T-MTT Sep 05 2935-2942 Ting Mo Ting, see Kam Man Shum, T-MTT Mar 05 895-900 Ting Ting Mo, see Kam Man Shum, T-MTT Mar 05 895-900 Ting-Yi Huang, see Shih-Hao Lee, T-MTT Aug 05 2552-2558 Ting-Yi Huang, see Chi-Feng Chen, T-MTT Sep 05 2688-2692 Tiong Su Hieng, see Yi Wang, T-MTT Jul 05 2348-2354 Tiong Tan Boon, see Boon Tiong Tan, T-MTT Jan 05 343-348 Tkachenko, Y.A., see Ce-Jun Wei, T-MTT Apr 05 1235-1243 Tobar, M.E., see Krupka, J., T-MTT Feb 05 702-712 Toland, J.E., Jr., see Mazumder, S., T-MTT Mar 05 1065-1071 Tommasino, P., see Olivieri, M., T-MTT Aug 05 2580-2586 Tong, C.-Y.E., see Battat, J., T-MTT Jan 05 389-395 Tong, K.Y., and C. Tsui. A physical analytical model of multilayer on-chip inductors; T-MTT Apr 05 1143-1149 Tong Meisong, see Meisong Tong, T-MTT Jul 05 2362-2370 Torrese, G., I. Huynen, and A. Vander Vorst. An analytical small-signal model for submicrometer nn-i-nn traveling-wave photodetectors; T-MTT Oct 05 3238-3243 Toyota, M., see Pokharel, R.K., T-MTT Sep 05 2726-2731 Traverse, J.V., see Mazumder, S., T-MTT Mar 05 1065-1071 Traverso, P.A., see Raffo, A., T-MTT Nov 05 3449-3459 Tretiakov, Y.V., see Yi-jan Emery Chen, T-MTT May 05 1672-1681 Tretyakov, O.A., see Aksoy, S., T-MTT Aug 05 2465-2471 Treyer, D.M., and W. Bachtold. Investigation of a self-calibrating SSB modulator; T-MTT Dec 05 3806-3816 Trifiletti, A., see Olivieri, M., T-MTT Aug 05 2580-2586 Tripon-Canseliet, C., see Maestrini, A., T-MTT Sep 05 2835-2843 Trueman, C.W., see Guilin Sun, T-MTT Mar 05 832-842 Trung-Kien Nguyen, Nam-Jin Oh, Choong-Yul Cha, Yong-Hun Oh, GookJu Ihm, and Sang-Gug Lee. Image-rejection CMOS low-noise amplifier design optimization techniques; T-MTT Feb 05 538-547 Trunin, M.R., see Egorov, V.N., T-MTT Feb 05 627-635 Truong Vu Bang Giang, M. Thiel, and A. Dreher. A unified approach to the analysis of radial waveguides, dielectric resonators, and microstrip antennas on spherical multilayer structures; T-MTT Jan 05 404-409 Tsai Cheng-Hua, see Chang, S.-F.R., T-MTT Mar 05 1048-1055

IEEE T-MTT 2005 INDEX — 19 Tsai Chih-Ming, see Hong-Ming Lee, T-MTT Sep 05 2812-2818 Tsai Chih-Ming, see Chih-Ming Tsai, T-MTT Nov 05 3429-3439 Tsai Chin-Chuan, see Chih-Ming Tsai, T-MTT Nov 05 3429-3439 Tsai Ming-Da, see Ming-Da Tsai, T-MTT Feb 05 496-505 Tsai-Pi Hung, A.G. Metzger, P.J. Zampardi, M. Iwamoto, and P.M. Asbeck. Design of high-efficiency current-mode class-D amplifiers for wireless handsets; T-MTT Jan 05 144-151 Tseng Chao-Hsiung, see Chi-Feng Chen, T-MTT Sep 05 2688-2692 Tseng Chao-Hsiung, see Chao-Hsiung Tseng, T-MTT Sep 05 2884-2891 Tseng Hua-Chou, see Ming-Hsiang Cho, T-MTT Sep 05 2926-2934 Tsuen Peng Song, see Ching-Feng Lee, T-MTT Sep 05 3024-3034 Tsui, C., see Tong, K.Y., T-MTT Apr 05 1143-1149 Tsui Hau-Yiu, see Hau-Yiu Tsui, T-MTT Jun 05 1883-1890 Tsuji, M., see Ohira, M., T-MTT Nov 05 3320-3326 Tsukamoto, K., see Sangiamwong, J., T-MTT Dec 05 3723-3730 Tsunekawa, K., see Seki, T., T-MTT Jun 05 2101-2106 Tsung-Hsun Yeh, see Jen-Tsai Kuo, T-MTT Apr 05 1331-1337 Tsung-Wen Chen, see Chi-Feng Chen, T-MTT Sep 05 2688-2692 Tsun-Lai Hsu, see Ming-Hsiang Cho, T-MTT Sep 05 2926-2934 Tu, W.-H., see Rodenbeck, C.T., T-MTT Dec 05 3697-3703 Tu Chi-Kang, see Chang, S.-F.R., T-MTT Mar 05 1048-1055 Tung-Sheng Chen, see Chih-Yuan Lee, T-MTT Feb 05 523-528 Tuovinen, J., see Koskinen, T., T-MTT Sep 05 2999-3006 Tzong-Lin Wu, Yen-Hui Lin, Ting-Kuang Wang, Chien-Chung Wang, and Sin-Ting Chen. Electromagnetic bandgap power/ground planes for wideband suppression of ground bounce noise and radiated emission in high-speed circuits; T-MTT Sep 05 2935-2942 Tzuang, C.-K.C., see Hsien-Shun Wu, T-MTT Sep 05 2713-2720 Tzu-Hsien Sang, see Hu, R., T-MTT Jul 05 2398-2402 U Ueda, K., see Ohta, A., T-MTT Jun 05 2121-2128 Ulm, M., see Schoebel, J., T-MTT Jun 05 1968-1975 Unal, C.M.H., see Yanovsky, F.J., T-MTT Feb 05 444-450 Urteaga, M., see Paidi, V.K., T-MTT Feb 05 598-605 Utsumi, Y., T. Kamei, K. Saito, and H. Moritake. Increasing the speed of microstrip-line-type polymer-dispersed liquid-crystal loaded variable phase shifter; T-MTT Nov 05 3345-3353 Uzunoglu, N.K., see Karanasiou, I.S., T-MTT May 05 1831-1832 V Vaassen, A.W.P., see Bergveld, H.J., T-MTT Feb 05 453-461 Vahldieck, R., see Ofli, E., T-MTT Mar 05 843-851 Vahldieck, R., see Pasalic, D., T-MTT Sep 05 2700-2706 Vahldieck, R., see Baumann, D., T-MTT Nov 05 3595-3605 Valdes-Garcia, A., see Mishra, C., T-MTT Dec 05 3744-3756 Valero-Nogueira, A., J.I. Herranz-Herruzo, E. Antonino-Daviu, and M. Cabedo-Fabres. Evaluation of the input impedance of a top-loaded monopole in a parallel-plate waveguide by the MoM/Green's function method; T-MTT Mar 05 868-873 Valletta, E., see Spirito, M., T-MTT Jul 05 2340-2347 Valois, R., D. Baillargeat, S. Verdeyme, M. Lahti, and T. Jaakola. High performances of shielded LTCC vertical transitions from DC up to 50 GHz; T-MTT Jun 05 2026-2032 van den Berg, P.M., see Semenov, S.Y., T-MTT Jul 05 2284-2294 Vander Vorst, A., see Torrese, G., T-MTT Oct 05 3238-3243 van der Weide, D.W., see Brace, C.L., T-MTT Jan 05 215-220 van der Weide, D.W., see Kozyrev, A.B., T-MTT Jan 05 238-245 van der Weide, D.W., see Taylor, K.M., T-MTT May 05 1576-1586 Vanhoenacker-Janvier, D., see Saib, A., T-MTT Jun 05 2043-2049 van Kaam, K.M.M., see Bergveld, H.J., T-MTT Feb 05 453-461 Vannini, G., see Raffo, A., T-MTT Nov 05 3449-3459 Vardapetyan, L., see Din-Kow Sun, T-MTT Mar 05 984-992 Vardaxoglou, J.C., see Yunchuan Guo, T-MTT Apr 05 1462-1468 Varonen, M., M. Karkkainen, J. Riska, P. Kangaslahti, and K.A.I. Halonen. Resistive HEMT mixers for 60-GHz broad-band telecommunication; TMTT Apr 05 1322-1330 Velazquez-Ahumada, Md.C., J. Martel, and F. Medina. Parallel coupled microstrip filters with floating ground-plane conductor for spurious-band suppression; T-MTT May 05 1823-1828 Vendier, O., see Melle, S., T-MTT Nov 05 3482-3488 Verdeyme, S., see Valois, R., T-MTT Jun 05 2026-2032

+ Check author entry for coauthors

Verspecht, J., D.F. Williams, D. Schreurs, K.A. Remley, and M.D. McKinley. Linearization of large-signal scattering functions; T-MTT Apr 05 1369-1376 Verspecht, J., see Root, D.E., T-MTT Nov 05 3656-3664 Vicente, C., and H.L. Hartnagel. Passive-intermodulation analysis between rough rectangular waveguide flanges; T-MTT Aug 05 2515-2525 Vidal, A., see Taroncher, M., T-MTT Jun 05 2153-2163 Vidal, B., J.L. Corral, and J. Marti. Statistical analysis of WDM photonic microwave filters with random errors; T-MTT Aug 05 2600-2603 Vidal, F., see Ouaddari, M., T-MTT Apr 05 1390-1397 Vidkjaer, J., see Johansen, T.K., T-MTT Jul 05 2389-2397 Vincent, D., T. Rouiller, C. Simovsky, B. Bayard, and G. Noyel. A new broad-band method for magnetic thin-film characterization in the microwave range; T-MTT Apr 05 1174-1180 Virone, G., R. Tascone, M. Baralis, O.A. Peverini, A. Olivieri, and R. Orta. A novel design tool for waveguide polarizers; T-MTT Mar 05 888-894 Vital, J.V., see Lorenz, P., T-MTT Nov 05 3631-3637 Vittoria, C., see How, H., T-MTT Jan 05 414 Voinigescu, S.P., see Dickson, T.O., T-MTT Jan 05 123-133 Vorobiev, A., see Kuylenstierna, D., T-MTT Jun 05 2164-2170 Vukovic, A., see Sewell, P., T-MTT Jun 05 1919-1928 W Wada, K., see Ohno, T., T-MTT Jun 05 2088-2095 Wai-Yip Tam, see King-Yuen Wong, T-MTT Nov 05 3364-3370 Wakino, K., see Yamamoto, H., T-MTT Jun 05 2187-2195 Wa Leung Kwok, see Hong-Xing Zheng, T-MTT Jul 05 2295-2301 Wane, S., D. Bajon, H. Baudrand, and P. Gamand. A new full-wave hybrid differential-integral approach for the investigation of multilayer structures including nonuniformly doped diffusions; T-MTT Jan 05 200-214 Wang, C.-H., see Wu, P.-S., T-MTT Oct 05 3106-3114 Wang, H., see Wu, P.-S., T-MTT Oct 05 3106-3114 Wang Chien-Chung, see Tzong-Lin Wu, T-MTT Sep 05 2935-2942 Wang Chi-Hsueh, see Yo-Shen Lin, T-MTT Jul 05 2324-2328 Wang Feipeng, see Feipeng Wang, T-MTT Apr 05 1244-1255 Wang Guoan, see Papaioannou, G., T-MTT Nov 05 3467-3473 Wang Hong, see Jianjun Gao, T-MTT Jan 05 330-335 Wang Hong, see Jianjun Gao, T-MTT Jan 05 417 Wang Hong, see Hong Wang, T-MTT Feb 05 564-570 Wang Huei, see Ming-Da Tsai, T-MTT Feb 05 496-505 Wang Huei, see Ming-Fong Lei, T-MTT Mar 05 861-867 Wangmyong Woo, M.D. Miller, and J.S. Kenney. A hybrid digital/RF envelope predistortion linearization system for power amplifiers; T-MTT Jan 05 229-237 Wang Narisi, see Narisi Wang, T-MTT Mar 05 1096-1102 Wang Narisi, see Pajic, S., T-MTT Sep 05 2899-2907 Wang Peng, see Peng Wang, T-MTT Jan 05 349-353 Wang Shih-Ming, see Shih-Ming Wang, T-MTT Feb 05 747-753 Wang Shumin, see Shumin Wang, T-MTT Jun 05 1913-1918 Wang Ting-Kuang, see Tzong-Lin Wu, T-MTT Sep 05 2935-2942 Wang Wenxiang, see Wenxiang Wang, T-MTT May 05 1833 Wang Xing, see Xing Wang, T-MTT Jan 05 410-413 Wang Yi, see Yi Wang, T-MTT Jul 05 2348-2354 Wang Ying, see Cameron, R.J., T-MTT Nov 05 3288-3297 Wang Yuanxun, see Younkyu Chung, T-MTT Feb 05 739-746 Wang Yueh-Hua, see Ming-Hsiang Cho, T-MTT Sep 05 2926-2934 Ward, J.S., see Maestrini, A., T-MTT Sep 05 2835-2843 Warr, P.A., see Carey-Smith, B.E., T-MTT Feb 05 777-785 Watson, P.M., see Pajic, S., T-MTT Sep 05 2899-2907 Weaver, W.D., see Mazumder, S., T-MTT Mar 05 1065-1071 Webb, K.J., see Sungjae Lee, T-MTT Apr 05 1314-1321 Wehling, J.H. Multifunction millimeter-wave systems for armored vehicle application; T-MTT Mar 05 1021-1025 Wei Ce-Jun, see Ce-Jun Wei, T-MTT Apr 05 1235-1243 Wei Chang-Lin, see Chang, S.-F.R., T-MTT Mar 05 1048-1055 Weide, J., see Ghose, A., T-MTT Jun 05 2082-2087 Weigel, R., see Ceylan, N., T-MTT Feb 05 515-522 Wei Hong, see Yu Lin Zhang, T-MTT Apr 05 1280-1287 Wei Hong, see Zhang-Cheng Hao, T-MTT Sep 05 2968-2977 Weikle, R.M., II, see Zhiyang Liu, T-MTT Sep 05 2949-2955 Weil, C., see Mueller, S., T-MTT Jun 05 1937-1945 Weimann, G., see Schneider, K., T-MTT Nov 05 3378-3387 Wei-Min Lance Kuo, see Yi-jan Emery Chen, T-MTT May 05 1672-1681

IEEE T-MTT 2005 INDEX — 20 Weimin Zhou, and G. Blasche. Injection-locked dual opto-electronic oscillator with ultra-low phase noise and ultra-low spurious level; T-MTT Mar 05 929-933 Wei Xu, see Hai-Ying Yao, T-MTT Apr 05 1469-1476 Wei Yun, see Paidi, V.K., T-MTT Feb 05 598-605 Wells, C.G., and J.A.R. Ball. Mode-matching analysis of a shielded rectangular dielectric-rod waveguide; T-MTT Oct 05 3169-3177 Wenliang Dai, Zhengfan Li, and Junfa Mao. Parameter extraction for on-chip interconnects by double-image Green's function method combined with hierarchical algorithm; T-MTT Jul 05 2416-2423 Wen-Lin Chen, see Chang, S.-F.R., T-MTT Mar 05 1048-1055 Wenxiang Wang Corrections on "Mode discriminator based on modeselective coupling" [Jan 03 55-63]; T-MTT May 05 1833 Wen-Yan Yin Comments on "Thermal resistance calculation of AlGaN-GaN devices"; T-MTT Sep 05 3051-3052 Wetkzer, G., see Bergveld, H.J., T-MTT Feb 05 453-461 Wiart, J., see Hadjem, A., T-MTT Jan 05 4-11 Wieser, N., see Basaran, U., T-MTT Mar 05 919-928 Williams, D.F., F. Ndagijimana, K.A. Remley, J.A. Dunsmore, and S. Hubert. Scattering-parameter models and representations for microwave mixers; T-MTT Jan 05 314-321 Williams, D.F., see Verspecht, J., T-MTT Apr 05 1369-1376 Williams, D.F., P.D. Hale, T.S. Clement, and J.M. Morgan. Calibrated 200GHz waveform measurement; T-MTT Apr 05 1384-1389 Williams, T.C., see Semenov, S.Y., T-MTT Jul 05 2284-2294 Williamson, A.G., see Roelvink, J., T-MTT Oct 05 3156-3160 Winder, W.F., see Aja, B., T-MTT Jun 05 2050-2062 Wing, Z.N., see Xun Gong, T-MTT Nov 05 3638-3647 Wing Han She, see Xun Gong, T-MTT Nov 05 3638-3647 Wisell, D., see Isaksson, M., T-MTT Nov 05 3422-3428 Wittek, M., see Mueller, S., T-MTT Jun 05 1937-1945 Wong King-Yuen, see King-Yuen Wong, T-MTT Nov 05 3364-3370 Wong Man Fai, see Hadjem, A., T-MTT Jan 05 4-11 Won Ko, and Youngwoo Kwon. Improved noise analysis of distributed preamplifier with cascode FET cells; T-MTT Jan 05 361-371 Wood, J., see Root, D.E., T-MTT Nov 05 3656-3664 Woo Duk-Jae, see Duk-Jae Woo, T-MTT Jun 05 2139-2144 Woo-Tae Kim, see Jung-Min Kim, T-MTT Sep 05 2693-2699 Woo Wangmyong, see Wangmyong Woo, T-MTT Jan 05 229-237 Woo-Young Choi, see Chang-Soon Choi, T-MTT Jan 05 256-263 Woskov, P.P., V.S. Bajaj, M.K. Hornstein, R.J. Temkin, and R.G. Griffin. Corrugated waveguide and directional coupler for CW 250-GHz gyrotron DNP experiments; T-MTT Jun 05 1863-1869 Wren, M., and T.J. Brazil. Experimental class-F power amplifier design using computationally efficient and accurate large-signal pHEMT model; TMTT May 05 1723-1731 Wu, B.-I., see Grzegorczyk, T.M., T-MTT Apr 05 1443-1450 Wu, D.C., see Tavik, G.C., T-MTT Mar 05 1009-1020 Wu, P.-S., C.-H. Wang, T.-W. Huang, and H. Wang. Compact and broadband millimeter-wave monolithic transformer balanced mixers; T-MTT Oct 05 3106-3114 Wu, R.-B., see Chen, C.-F., T-MTT Dec 05 3788-3793 Wu Chao-Huang, see Yo-Shen Lin, T-MTT Jul 05 2324-2328 Wu Hsien-Shun, see Hsien-Shun Wu, T-MTT Sep 05 2713-2720 Wu Ke, see Feng Xu, T-MTT Jan 05 66-73 Wu Ke, see Duochuan Li, T-MTT Mar 05 799-812 Wu Ke, see Belaid, M., T-MTT Apr 05 1124-1129 Wu Ke, see Yu Lin Zhang, T-MTT Apr 05 1280-1287 Wu Ke, see Ouaddari, M., T-MTT Apr 05 1390-1397 Wu Ke, see Xinyu Xu, T-MTT Jul 05 2267-2273 Wu Ke, see Deslandes, D., T-MTT Aug 05 2487-2494 Wu Ke, see Duochuan Li, T-MTT Aug 05 2546-2551 Wu Ke, see Duochuan Li, T-MTT Sep 05 2732-2742 Wu Ke, see Tatu, S.O., T-MTT Sep 05 2768-2776 Wu Ke, see Zhang-Cheng Hao, T-MTT Sep 05 2968-2977 Wu Lin-Kun, see Ming-Hsiang Cho, T-MTT Sep 05 2926-2934 Wu Min-Sou, see Shau-Gang Mao, T-MTT Nov 05 3460-3466 Wu Qun, see Hai-Ying Yao, T-MTT Apr 05 1469-1476 Wu Ruey-Beei, see Shih-Hao Lee, T-MTT Aug 05 2552-2558 Wu Ruey-Beei, see Chi-Feng Chen, T-MTT Sep 05 2688-2692 Wu Tzong-Lin, see Tzong-Lin Wu, T-MTT Sep 05 2935-2942 Wykes, J.G., see Sewell, P., T-MTT Jun 05 1919-1928

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X Xiang Guan, see Hashemi, H., T-MTT Feb 05 614-626 Xianmin Zhang, see Yan Li, T-MTT Apr 05 1522-1526 Xiao-Ping Chen, see Zhang-Cheng Hao, T-MTT Sep 05 2968-2977 Xin Chen Ji, see Yu Lin Zhang, T-MTT Apr 05 1280-1287 Xingchao Yuan, see Beyene, W.T., T-MTT Nov 05 3568-3577 Xing Wang, and L.W. Pearson. Design of coupled-oscillator arrays without a posteriori tuning; T-MTT Jan 05 410-413 Xinli Peng, see Narisi Wang, T-MTT Mar 05 1096-1102 Xinyu Xu, R.G. Bosisio, and Ke Wu. A new six-port junction based on substrate integrated waveguide technology; T-MTT Jul 05 2267-2273 Xiuping Li, see Jianjun Gao, T-MTT Jan 05 330-335 Xiuping Li, see Jianjun Gao, T-MTT Jan 05 417 Xiuping Li, see Hong Wang, T-MTT Feb 05 564-570 Xudong Chen, see Grzegorczyk, T.M., T-MTT Apr 05 1443-1450 Xudong Chen, see Grzegorczyk, T.M., T-MTT Sep 05 2956-2967 Xue Quan, see Kam Man Shum, T-MTT Mar 05 895-900 Xu Feng, see Feng Xu, T-MTT Jan 05 66-73 Xu Haiyong, see Zhiyang Liu, T-MTT Sep 05 2949-2955 Xu Hongtao, see Sanabria, C., T-MTT Feb 05 762-769 Xu Jianjun, see Lei Zhang, T-MTT Sep 05 2752-2767 Xun Gong, Wing Han She, E.E. Hoppenjans, Z.N. Wing, R.G. Geyer, J.W. Halloran, and W.J. Chappell. Tailored and anisotropic dielectric constants through porosity in ceramic components; T-MTT Nov 05 3638-3647 Xu Wei, see Hai-Ying Yao, T-MTT Apr 05 1469-1476 Xu Xinyu, see Xinyu Xu, T-MTT Jul 05 2267-2273 Y Yagoub, M.C.E., see Lei Zhang, T-MTT Sep 05 2752-2767 Yagoub, M.C.E., see Ahmed, T., T-MTT Nov 05 3538-3547 Yakovlev, A.B., see Eshrah, I.A., T-MTT Nov 05 3298-3304 Yamamoto, H., H. Miyagawa, T. Nishikawa, K. Wakino, and T. Kitazawa. Full-wave analysis for propagation characteristics of cylindrical coplanar waveguides with finite thickness of conductor; T-MTT Jun 05 2187-2195 Yang, A.H., see Feipeng Wang, T-MTT Apr 05 1244-1255 Yang Chek Pin, see Holmes, J.E., T-MTT Jan 05 322-329 Yang Houng-Jay, see Hsien-Shun Wu, T-MTT Sep 05 2713-2720 Yang Li, see Li Yang, T-MTT Jan 05 183-190 Yang Ping, see Duochuan Li, T-MTT Mar 05 799-812 Yang Ping, see Duochuan Li, T-MTT Aug 05 2546-2551 Yan Li, Lixin Ran, Hongsheng Chen, Jiangtao Huangfu, Xianmin Zhang, Kangsheng Chen, T.M. Grzegorczyk, and Jin Au Kong. Experimental realization of a one-dimensional LHM-RHM resonator; T-MTT Apr 05 1522-1526 Yanovsky, F.J., H.W.J. Russchenberg, and C.M.H. Unal. Retrieval of information about turbulence in rain by using Doppler-polarimetric Radar; T-MTT Feb 05 444-450 Yan-Shian Yeh, Juh-Tzeng Lue, and Zhi-Ren Zheng. Measurement of the dielectric constants of metallic nanoparticles embedded in a paraffin rod at microwave frequencies; T-MTT May 05 1756-1760 Yao, J., see Qi, G., T-MTT Oct 05 3090-3097 Yao Hai-Ying, see Hai-Ying Yao, T-MTT Apr 05 1469-1476 Yashchyshyn, Y., and J.W. Modelski. Rigorous analysis and investigations of the scan antennas on a ferroelectric substrate; T-MTT Feb 05 427-438 Yassini, B., see Daneshmand, M., T-MTT Jan 05 12-21 Yazgan, E., see Akan, V., T-MTT Dec 05 3681-3686 Yee Hui Lee, see Shao Ying Huang, T-MTT Sep 05 2656-2664 Yeh Chun-Cheng, see Jen-Tsai Kuo, T-MTT Apr 05 1331-1337 Yeh Tsung-Hsun, see Jen-Tsai Kuo, T-MTT Apr 05 1331-1337 Yeh Yan-Shian, see Yan-Shian Yeh, T-MTT May 05 1756-1760 Yen-Hui Lin, see Tzong-Lin Wu, T-MTT Sep 05 2935-2942 Yeo Kiat Seng, see Choon Beng Sia, T-MTT Sep 05 3035-3044 Yeom Kyung-Whan, see Kyung-Whan Yeom, T-MTT Jul 05 2435-2440 Yeong-Chang Chou, R. Lai, T.R. Block, A. Sharma, Q. Kan, D.L. Leung, D. Eng, and A. Oki. The effect of RF-driven gate current on DC/RF performance in GaAs pHEMT MMIC power amplifiers; T-MTT Nov 05 3398-3406 Yeo Tat-Soon, see Hai-Ying Yao, T-MTT Apr 05 1469-1476 Yew Hui Liew, and J. Joe. Large-signal diode modeling-an alternative parameter-extraction technique; T-MTT Aug 05 2633-2638 Yi-jan Emery Chen, Wei-Min Lance Kuo, Zhenrong Jin, Jongsoo Lee, Y.V. Tretiakov, J.D. Cressler, J. Laskar, and G. Freeman. A low-power ka-band

IEEE T-MTT 2005 INDEX — 21 Voltage-controlled oscillator implemented in 200-GHz SiGe HBT technology; T-MTT May 05 1672-1681 Yilmaz, A.E., Jian-Ming Jin, and E. Michielssen. A parallel FFT accelerated transient field-circuit simulator; T-MTT Sep 05 2851-2865 Yin, W.-Y., see Dong, X., T-MTT Oct 05 3098-3105 Ying Huang Shao, see Shao Ying Huang, T-MTT Sep 05 2656-2664 Ying Wang, see Cameron, R.J., T-MTT Nov 05 3288-3297 Yin Wen-Yan, see Wen-Yan Yin, T-MTT Sep 05 3051-3052 Yi Wang, Hieng Tiong Su, Frederick Huang, and M.J. Lancaster. Wide-band superconducting coplanar delay lines; T-MTT Jul 05 2348-2354 Yong-Chae Jeong, see Jong-Sik Lim, T-MTT Aug 05 2539-2545 Yong-Hun Oh, see Trung-Kien Nguyen, T-MTT Feb 05 538-547 Yong-Kweon Kim, see Byoungjoong Kang, T-MTT Jan 05 134-143 Yong-Kweon Kim, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 Yong-Sub Lee, see Seung-Yup Lee, T-MTT Feb 05 786-793 Yoo, D., see Li, Y., T-MTT Oct 05 3121-3129 Yook Jong-Gwan, see Jung-Min Kim, T-MTT Sep 05 2693-2699 Yook Jong-Gwan, see Il Kwon Kim, T-MTT Sep 05 2943-2948 Yook Jong-Min, see Jong-Min Yook, T-MTT Jun 05 2230-2234 Yoon, I.-J., see Kim, H., T-MTT Oct 05 3206-3214 Yoon, Y.J., see Kim, H., T-MTT Oct 05 3206-3214 Yoon Euisik, see Il-Joo Cho, T-MTT Jul 05 2450-2457 Yoon Euisik, see Sangsoo Ko, T-MTT Sep 05 2789-2800 Yoon-Ha Jeong, see Seung-Yup Lee, T-MTT Feb 05 786-793 Yoon Jeonghoon, see Byoungjoong Kang, T-MTT Jan 05 134-143 Yoon Jeonghoon, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 York, R.A., see O'Sullivan, T., T-MTT Jan 05 106-114 York, R.A., see Sanabria, C., T-MTT Feb 05 762-769 York, R.A., see Chase, D.R., T-MTT Oct 05 3215-3220 Yo-Sheng Lin, see Hung-Wei Chiu, T-MTT Mar 05 813-824 Yo-Shen Lin, Chi-Hsueh Wang, Chao-Huang Wu, and Chun Hsiung Chen. Novel compact parallel-coupled microstrip bandpass filters with lumpedelement K-inverters; T-MTT Jul 05 2324-2328 Young, P.R., see Stephens, D., T-MTT Dec 05 3832-3838 Youngcheol Park, R. Melville, R.C. Frye, Min Chen, and J.S. Kenney. Dualband transmitters using digitally predistorted frequency multipliers for reconfigurable radios; T-MTT Jan 05 115-122 Young-Hoon Chun, Jae-Ryong Lee, Sang-Won Yun, and Jin-Koo Rhee. Design of an RF low-noise bandpass filter using active capacitance circuit; T-MTT Feb 05 687-695 Young-Jin Kim, Young-Suk Son, V.N. Parkhomenko, In-Chul Hwang, JeKwang Cho, Kyung-Suc Nah, and Byeong-Ha Park. A GSM/EGSM/DCS/PCS direct conversion receiver with integrated synthesizer; T-MTT Feb 05 606-613 Young-Se Kwon, see Jong-Min Yook, T-MTT Jun 05 2230-2234 Young-Suk Son, see Young-Jin Kim, T-MTT Feb 05 606-613 Youngwoo Kwon, see Byoungjoong Kang, T-MTT Jan 05 134-143 Youngwoo Kwon, see Won Ko, T-MTT Jan 05 361-371 Youngwoo Kwon, see Jinho Jeong, T-MTT Jun 05 1891-1898 Youngwoo Kwon, see Jung-Mu Kim, T-MTT Nov 05 3415-3421 Young Yun A novel microstrip-line structure employing a periodically perforated ground metal and its application to highly miniaturized and low-impedance passive components fabricated on GaAs MMIC; T-MTT Jun 05 1951-1959 Younkyu Chung, Jinseong Jeong, Yuanxun Wang, D. Ahn, and T. Itoh. Power level-dependent dual-operating mode LDMOS power amplifier for CDMA wireless base-station applications; T-MTT Feb 05 739-746 Yousefzadeh, V., see Narisi Wang, T-MTT Mar 05 1096-1102 You Se-Ho, see Se-Ho You, T-MTT Sep 05 2826-2834 You Zheng, see Saavedra, C.E., T-MTT Jul 05 2430-2434 Yu, C.-F., and T.-H. Chang. High-performance circular TE01-mode converter; T-MTT Dec 05 3794-3798 Yuan-Wen Hsiao, see Ming-Dou Ker, T-MTT Sep 05 2672-2681 Yuan Xingchao, see Beyene, W.T., T-MTT Nov 05 3568-3577 Yuanxun Wang, see Younkyu Chung, T-MTT Feb 05 739-746 Yueh-Hua Wang, see Ming-Hsiang Cho, T-MTT Sep 05 2926-2934 Yu Jong Jen, see Boon Tiong Tan, T-MTT Jan 05 343-348 Yu Lin Zhang, Wei Hong, Ke Wu, Ji Xin Chen, and Hong Jun Tang. Novel substrate integrated waveguide cavity filter with defected ground structure; T-MTT Apr 05 1280-1287 Yu Ming, see Daneshmand, M., T-MTT Jan 05 12-21 Yu Ming, see Cameron, R.J., T-MTT Nov 05 3288-3297 Yumoto, H., see Chaki, S., T-MTT Jun 05 2073-2081 Yunchi Zhang, see Ruiz-Cruz, J.A., T-MTT Jan 05 174-182

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Yunchuan Guo, G. Goussetis, A.P. Feresidis, and J.C. Vardaxoglou. Efficient modeling of novel uniplanar left-handed metamaterials; T-MTT Apr 05 1462-1468 Yun Sang-Won, see Young-Hoon Chun, T-MTT Feb 05 687-695 Yunseo Park, see Mukhopadhyay, R., T-MTT Jan 05 81-93 Yun Wei, see Paidi, V.K., T-MTT Feb 05 598-605 Yun Young, see Young Yun, T-MTT Jun 05 1951-1959 Yu-Zhi Chueh, see Shau-Gang Mao, T-MTT Nov 05 3460-3466 Z Zajic, A.G., see Djordjevic, A.R., T-MTT Apr 05 1164-1167 Zaki, K.A., see Ruiz-Cruz, J.A., T-MTT Jan 05 174-182 Zampardi, P.J., see Tsai-Pi Hung, T-MTT Jan 05 144-151 Zapata, J., see de la Rubia, V., T-MTT Feb 05 670-678 Zeng Rong, see Hong Wang, T-MTT Feb 05 564-570 Zhang, Y.P., see Sun, M., T-MTT Sep 05 2650-2655 Zhang-Cheng Hao, Wei Hong, Ji-Xin Chen, Xiao-Ping Chen, and Ke Wu. Compact super-wide bandpass substrate integrated waveguide (SIW) filters; T-MTT Sep 05 2968-2977 Zhang Guoyong, see Guoyong Zhang, T-MTT Mar 05 947-951 Zhang Haitao, see Haitao Zhang, T-MTT Nov 05 3606-3614 Zhang Jingjing, see Jingjing Zhang, T-MTT Nov 05 3281-3287 Zhang Lei, see Lei Zhang, T-MTT Sep 05 2752-2767 Zhang Qi-Jun, see Lei Zhang, T-MTT Sep 05 2752-2767 Zhang Xianmin, see Yan Li, T-MTT Apr 05 1522-1526 Zhang Yu Lin, see Yu Lin Zhang, T-MTT Apr 05 1280-1287 Zhang Yunchi, see Ruiz-Cruz, J.A., T-MTT Jan 05 174-182 Zhaolin Lu, C.A. Schuetz, Shouyuan Shi, Caihua Chen, G.P. Behrmann, and D.W. Prather. Experimental demonstration of self-collimation in lowindex-contrast photonic crystals in the millimeter-wave regime; T-MTT Apr 05 1362-1368 Zhengfan Li, see Wenliang Dai, T-MTT Jul 05 2416-2423 Zheng Guizhen, see Pothier, A., T-MTT Jan 05 354-360 Zhenghe Feng, see Li Yang, T-MTT Jan 05 183-190 Zheng Hong-Xing, see Hong-Xing Zheng, T-MTT Jul 05 2295-2301 Zheng Lou, and Jian-Ming Jin. An accurate waveguide port boundary condition for the time-domain finite-element method; T-MTT Sep 05 3014-3023 Zheng You, see Saavedra, C.E., T-MTT Jul 05 2430-2434 Zheng Zhi-Ren, see Yan-Shian Yeh, T-MTT May 05 1756-1760 Zhenhai Shao, and M. Fujise. An improved FDTD formulation for general linear lumped microwave circuits based on matrix theory; T-MTT Jul 05 2261-2266 Zhenrong Jin, see Yi-jan Emery Chen, T-MTT May 05 1672-1681 Zhiming Feng, see Guofu Niu, T-MTT Feb 05 506-514 Zhi-Ren Zheng, see Yan-Shian Yeh, T-MTT May 05 1756-1760 Zhitomirsky, V.E., see Egorov, V.N., T-MTT Feb 05 627-635 Zhiyang Liu, J.C. Midkiff, Haiyong Xu, T.W. Crowe, and R.M. Weikle, II. Broad-band 180° phase shifters using integrated submillimeter-wave Schottky diodes; T-MTT Sep 05 2949-2955 Zhizhang Chen, see Shuiping Luo, T-MTT Mar 05 969-976 Zhou Weimin, see Weimin Zhou, T-MTT Mar 05 929-933 Zhu Lei, see Sheng Sun, T-MTT May 05 1817-1822 Zhu Lei, see Sheng Sun, T-MTT Apr 05 1221-1227 Zhu Lei, see Sheng Sun, T-MTT Sep 05 2844-2850 Zinal, S., and G. Boeck. Complex permittivity measurements using TE11p modes in circular cylindrical cavities; T-MTT Jun 05 1870-1874 Ziolkowski, R.W., see Nader Engheta, T-MTT Apr 05 1535-1556 Zirath, H., see Lepine, F., T-MTT Jun 05 2007-2012 Zirath, H., see Kuylenstierna, D., T-MTT Aug 05 2616-2621 Zwick, T., and U.R. Pfeiffer. Pure-mode network analyzer concept for onwafer measurements of differential circuits at millimeter-wave frequencies; T-MTT Mar 05 934-937 Zwick, T., see Floyd, B.A., T-MTT Apr 05 1181-1188 Zybura, A., see Daneshmand, M., T-MTT Jan 05 12-21

SUBJECT INDEX

A Absorbing media lossy Foster networks. Kajfez, D., T-MTT Oct 05 3199-3205

IEEE T-MTT 2005 INDEX — 22 Accelerator RF systems Ka-band resonant ring for testing components for a high-gradient linear accelerator. Bogdashov, A., + , T-MTT Oct 05 3152-3155 Acoustic devices; cf. Surface acoustic wave devices Acoustic filters; cf. Surface acoustic wave filters Active arrays advanced multifunction RF concept. Tavik, G.C., + , T-MTT Mar 05 10091020 mm-wave high-effic. multilayer parasitic microstrip antenna array, teflon substr. Seki, T., + , T-MTT Jun 05 2101-2106 Active filters 20-Gb/s 4-PAM backplane serial I/O interconnections, equalization and NEXT noise cancellation. Hur, Y., + , T-MTT Jan 05 246-255 image-rejection CMOS LNA design optim. techs. Trung-Kien Nguyen, + , T-MTT Feb 05 538-547 Active networks; cf. Active filters; Impedance converters Adaptive antenna arrays adaptive power controllable retrodirective array system for wireless sensor server applications. Lim, S., + , T-MTT Dec 05 3735-3743 Adaptive control Maximum Output control method for UMTS downlink transmitters, adaptive feedforward amp. Legarda, J., + , T-MTT Aug 05 2481-2486 predistortion RF power amp., spectrum monitor for multicarrier WCDMA appls. Seung-Yup Lee, + , T-MTT Feb 05 786-793 Adaptive filters microwave filters, adaptive models and param. extr., seq. tuning. Pepe, G., + , T-MTT Jan 05 22-31 Adaptive radar breast cancer detect., expt. investig. of simple tumor models, tissue sens. adaptive radar. Sill, J.M., + , T-MTT Nov 05 3312-3319 Adaptive systems duplexer implemented, single-path/multipath feedforward techs., BST phase shifters. O'Sullivan, T., + , T-MTT Jan 05 106-114 hybrid digital/RF envelope predistortion linearization syst. for power amps. Wangmyong Woo, + , T-MTT Jan 05 229-237 Adaptive systems; cf. Adaptive filters Admittance microstrip nets., orthogonality-based deembedding tech. Spowart, M.P., + , T-MTT Mar 05 938-946 on-chip interconnects by double-image Green's fn. method combined, hierarchical algm., param. extr. Wenliang Dai, + , T-MTT Jul 05 24162423 Alkali metal compounds; cf. Lithium compounds Alkaline earth compounds; cf. Barium compounds; Magnesium compounds Alkaline earth metals; cf. Barium Alloys; cf. Boron alloys; Germanium alloys; Silicon alloys Aluminum compounds AlGaN-GaN devices, thermal resist. calc. Wen-Yan Yin, T-MTT Sep 05 3051-3052 AlGaN-GaN devices'), 'Thermal resist. calc. Darwish, A.M., + , T-MTT Sep 05 3052-3053 epitaxial struct., noise fig. of AlGaN/GaN HEMTs. Sanabria, C., + , TMTT Feb 05 762-769 mm-wave appls., extremely high-Q factor dielec. resonators. Krupka, J., + , T-MTT Feb 05 702-712 photoelectronic ADC. Ioakeimidi, K., + , T-MTT Jan 05 336-342 short-channel AlGaN/GaN heterojunction FETs, 30-GHz-band, 5-W power perform. Inoue, T., + , T-MTT Jan 05 74-80 sig. gener., control, freq. conversion AlGaN/GaN HEMT MMICs. Kaper, V.S., + , T-MTT Jan 05 55-65 substr. removed LV high-speed GaAs/AlGaAs electrooptic modulators, Trail electrodes. JaeHyuk Shin, + , T-MTT Feb 05 636-643 Amplifiers broad-band power amp., tunable output matching net. Haitao Zhang, + , T-MTT Nov 05 3606-3614 compact dual-polarized multibeam phased-array archit. for mm-wave radar. Schulwitz, L., + , T-MTT Nov 05 3588-3594 deembedding static nonlinearities and accurately identifying and modeling memory effects, wide-band RF transmitters. Taijun Liu, + , T-MTT Nov 05 3578-3587 mm-wave CMOS cct. design. Shigematsu, H., + , T-MTT Feb 05 472-477 robust modeling and design approach for dynamically loaded and digitally linearized Doherty amps. Sirois, J., + , T-MTT Sep 05 2875-2883 serial communs., data-depend. jitter. Analui, B., + , T-MTT Nov 05 33883397 + Check author entry for coauthors

Amplifiers; cf. Differential amplifiers; Distributed amplifiers; Power amplifiers; Traveling wave amplifiers Amplitude modulation baseband-modeled CALLUM archits., spectrum emission considerations. Strandberg, R., + , T-MTT Feb 05 660-669 hybrid digital/RF envelope predistortion linearization syst. for power amps. Wangmyong Woo, + , T-MTT Jan 05 229-237 X-band class-E power amps., EER operation, linearity. Narisi Wang, + , T-MTT Mar 05 1096-1102 Amplitude modulation; cf. Amplitude shift keying; Quadrature amplitude modulation Amplitude shift keying BPSK to ASK signal conversion using injection-locked oscillators theory, part I. Lopez-Villegas, J.M., + , T-MTT Dec 05 3757-3766 Analog circuits; cf. Analog integrated circuits; Analog processing circuits Analog-digital conversion 2.4-GHz-band GFSK appls., low-power highly digitized receiver. Bergveld, H.J., + , T-MTT Feb 05 453-461 AMRFC test-bed, high-band digital preprocessor (HBDP). Mazumder, S., + , T-MTT Mar 05 1065-1071 mismatch errors for 400-msamples/s 80-dB SFDR time-interleaved ADC, comprehensive digital correction. Munkyo Seo, + , T-MTT Mar 05 10721082 multifrequency RF front ends using direct RF sampling. Psiaki, M.L., + , T-MTT Oct 05 3082-3089 photoelectronic ADC. Ioakeimidi, K., + , T-MTT Jan 05 336-342 ultrawide-band photonic time-stretch a/D converter employing phase diversity. Han, Y., + , T-MTT Apr 05 1404-1408 Analog-digital conversion; cf. Sigma-delta modulation Analog integrated circuits 12-GHz SiGe phase shifter, integr. LNA. Hancock, T.M., + , T-MTT Mar 05 977-983 Analog integrated circuits; cf. BiCMOS analog integrated circuits; Bipolar analog integrated circuits Analog multipliers modified loss-compensation method, 0.35-ȝm SiGe BiCMOS technol., broad-band MMICs. Ming-Da Tsai, + , T-MTT Feb 05 496-505 Analog processing circuits software-defined direct conversion receiver, ka-band analog front-end. Tatu, S.O., + , T-MTT Sep 05 2768-2776 Analog processing circuits; cf. Analog multipliers Angle modulation; cf. Phase modulation Antenna accessories compact EM-bandgap (EBG) struct. and appls. for microwave ccts. Li Yang, + , T-MTT Jan 05 183-190 Antenna arrays distrib. left-handed microstrip lines, effective EM params. Shau-Gang Mao, + , T-MTT Apr 05 1515-1521 mutually exclusive data encoding for realization of a full duplexing selfsteering wireless link using a retrodirective array transceiver. Leong, K.M.K.H., + , T-MTT Dec 05 3687-3696 ultra-wideband low-cost phased-array radars. Rodenbeck, C.T., + , T-MTT Dec 05 3697-3703 Antenna arrays; cf. Adaptive antenna arrays; Microwave antenna arrays; Millimeter wave antenna arrays Antenna measurements hologram-based CATR, 650 GHz, expt. study. Koskinen, T., + , T-MTT Sep 05 2999-3006 metamaterial-based electronically controlled transm.-line struct., leakywave antenna, tunable radiation angle and beamwidth. Sungjoon Lim, + , T-MTT Jan 05 161-173 Antenna radiation patterns LINC radio transmitters, integr. antenna/power combiner. Gao, S., + , TMTT Mar 05 1083-1088 metamaterial-based electronically controlled transm.-line struct., leakywave antenna, tunable radiation angle and beamwidth. Sungjoon Lim, + , T-MTT Jan 05 161-173 scan antennas, ferroelec. substr., rigorous anal. and investigs. Yashchyshyn, Y., + , T-MTT Feb 05 427-438 surface waves, metamaterial grounded slabs, fund. modal props. Baccarelli, P., + , T-MTT Apr 05 1431-1442 Antennas complex impedances, RFID tag design, power refl. coeff. anal. Nikitin, P.V., + , T-MTT Sep 05 2721-2725

IEEE T-MTT 2005 INDEX — 23 double-neg. metamaterials, pos. future. Nader Engheta, + , T-MTT Apr 05 1535-1556 Antennas; cf. Antenna accessories; Antenna arrays; Antenna radiation patterns; Dipole antennas; Directive antennas; Leaky wave antennas; Loop antennas; Microwave antennas; Millimeter wave antennas; Mobile antennas; Monopole antennas; Radar antennas Application specific integrated circuits; cf. Mixed analog-digital integrated circuits Approximation methods nonlin. model testing, designing multisine excit. Pedro, J.C., + , T-MTT Jan 05 45-54 Arrays; cf. Antenna arrays; Phased array radar Array signal processing advanced multifunction RF concept. Tavik, G.C., + , T-MTT Mar 05 10091020 AMRFC test-bed, high-band digital preprocessor (HBDP). Mazumder, S., + , T-MTT Mar 05 1065-1071 Atmospheric precipitation; cf. Rain Attenuators ring-hybrid microwave voltage-variable attenuator, HFET transistors. Saavedra, C.E., + , T-MTT Jul 05 2430-2434 short stub-matching 77-GHz-band driver amp., attenuator compensating temp. depend. of gain. Chaki, S., + , T-MTT Jun 05 2073-2081 Automated highways effectiveness of wave absorbers, improve DSRC EM environ., express highway. Pokharel, R.K., + , T-MTT Sep 05 2726-2731 Avalanche diodes; cf. Avalanche photodiodes; IMPATT diodes Avalanche photodiodes nonlin. params. of dispers. APD, pulsed RF meas. and quasiDC opt. excit., extr. Ghose, A., + , T-MTT Jun 05 2082-2087 B Baluns 60-GHz broad-band telecomm., resistive HEMT mixers. Varonen, M., + , T-MTT Apr 05 1322-1330 act. differential broad-band phase splitter for quadrature-modulator appls. Tiiliharju, E., + , T-MTT Feb 05 679-686 low-loss broad-band planar baluns, multilayered organic thin films. Chen, A.C., + , T-MTT Nov 05 3648-3655 miniaturized planar Marchand baluns, classes. Fathelbab, W.M., + , TMTT Apr 05 1211-1220 surface-passivated high-resist. Si, true microwave substr. Spirito, M., + , T-MTT Jul 05 2340-2347 Band-pass filters branch-line directional coupler in the design of microwave bandpass filters. Gomez-Garcia, R., + , T-MTT Oct 05 3221-3229 compact bandpass filter, 2 tuning transm. zeros, CMRC resonator. Kam Man Shum, + , T-MTT Mar 05 895-900 compact parallel-coupled microstrip bandpass filters, lumped-element Kinverters. Yo-Shen Lin, + , T-MTT Jul 05 2324-2328 compact super-wide bandpass substr. integr. waveguide (SIW) filters. Zhang-Cheng Hao, + , T-MTT Sep 05 2968-2977 coupled strip-slot guiding structs., full-wave anal. Deleniv, A.N., T-MTT Jun 05 1904-1912 cross coupling through higher/lower order modes and their applications in elliptic filter design. Amari, S., + , T-MTT Oct 05 3135-3141 dielec.-filled cavity filters, ultrawide stopband Characteristics, design. Rauscher, C., T-MTT May 05 1777-1786 differential 4-bit 6.5-10-GHz RF MEMS tunable filter. Entesari, K., + , TMTT Mar 05 1103-1110 direct-coupled microwave filters, single and dual stopbands. Cameron, R.J., + , T-MTT Nov 05 3288-3297 dual-passband filters, design techs. Macchiarella, G., + , T-MTT Nov 05 3265-3271 electronically tunable microstrip bandpass filter, thin-film BariumStrontium-Titanate (BST) varactors. Nath, J., + , T-MTT Sep 05 27072712 EM field-mapping syst., reson.-suppressed mag. field probe. Jung-Min Kim, + , T-MTT Sep 05 2693-2699 future mobile telecomm. systs., high-temp. supercond. filter. Jia-Sheng Hong, + , T-MTT Jun 05 1976-1981 gen. Chebyshev filters, asymmetrically located transm. zeros, design. Milosavljevic, Z.D., T-MTT Jul 05 2411-2415

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highly integr. mm-wave pass. components, 3D LTCC syst.-on-package (SOP) technol. Jong-Hoon Lee, + , T-MTT Jun 05 2220-2229 hybrid narrow-band tunable bandpass filter, varactor loaded EM-bandgap CPW. Pistono, E., + , T-MTT Aug 05 2506-2514 init. design of LTCC filters, all-capacitive coupling, simplified anal. tech. Rambabu, K., + , T-MTT May 05 1787-1791 in-line dual- and triple-mode cavity filters, nonresonating nodes. Amari, S., + , T-MTT Apr 05 1272-1279 in-line N-order filters, N real transm. zeros by of extracted poles implemented, low-cost rect. H-plane waveguide, synthesis and design. Montejo-Garai, J.R., + , T-MTT May 05 1636-1642 insertion loss function synthesis of maximally flat parallel-coupled line bandpass filters. Chin, K.-S., + , T-MTT Oct 05 3161-3168 low-loss 2-bit tunable bandpass filters, MEMS DC contact switches. Pothier, A., + , T-MTT Jan 05 354-360 low-loss LTCC cavity filters using system-on-package technology at 60 GHz. Lee, J.-H., + , T-MTT Dec 05 3817-3824 low-refl. bandpass filters, flat group delay. Djordjevic, A.R., + , T-MTT Apr 05 1164-1167 LTCC, metallic resonators, canonical ridge waveguide filters. Ruiz-Cruz, J.A., + , T-MTT Jan 05 174-182 microstrip bandpass filters, dual-passband response, design. Jen-Tsai Kuo, + , T-MTT Apr 05 1331-1337 microstrip bandpass filters with multiorder spurious-mode suppression. Chen, C.-F., + , T-MTT Dec 05 3788-3793 microwave filters, improved stopband, sub-wavel. resonators. GarciaGarcia, J., + , T-MTT Jun 05 1997-2006 miniaturized dual-mode bandpass filter struct., shunt-capacitance perturb. Ming-Fong Lei, + , T-MTT Mar 05 861-867 miniaturized dual-mode ring bandpass filter, perturb. Boon Tiong Tan, + , T-MTT Jan 05 343-348 miniaturized multilayer quasiellipt. bandpass filter, aperture-coupled microstrip resonators. Chi-Feng Chen, + , T-MTT Sep 05 2688-2692 miniaturized parallel coupled-line bandpass filter, spurious-response suppression. Pedro Cheong, + , T-MTT May 05 1810-1816 miniaturized planar Marchand baluns, classes. Fathelbab, W.M., + , TMTT Apr 05 1211-1220 miniaturized spurious passband suppression microstrip filter, meandered parallel coupled lines. Shih-Ming Wang, + , T-MTT Feb 05 747-753 parallel-coupled line filters with enhanced stopband performances. Fathelbab, W.M., + , T-MTT Dec 05 3774-3781 periodically nonuniform coupled microstrip-line filters, harmonic suppression, transm. zero reallocation. Sheng Sun, + , T-MTT May 05 1817-1822 planar duplexers/triplexers by manipulating atten. poles, design methodologies. Ohno, T., + , T-MTT Jun 05 2088-2095 planar filter design, fully controllable second passband. Chih-Ming Tsai, + , T-MTT Nov 05 3429-3439 quasi-lumped suspended stripline filters and diplexers. Menzel, W., + , TMTT Oct 05 3230-3237 RF inductors and filters, LCP substrs. for Wi-Fi appls., layout-level synthesis. Mukherjee, S., + , T-MTT Jun 05 2196-2210 RF low-noise bandpass filter, act. capacitance cct., design. Young-Hoon Chun, + , T-MTT Feb 05 687-695 RF/microwave multifunctional systs., reconfigurable bandpass filter. Fathelbab, W.M., + , T-MTT Mar 05 1111-1116 semilumped CPW elements for Millimeter-wave filter design, charactn. Aryanfar, F., + , T-MTT Apr 05 1288-1293 substr. integr. waveguide cavity filter, defected ground struct. Yu Lin Zhang, + , T-MTT Apr 05 1280-1287 supercond. spiral filters, quasiellipt. charact. for radio astron. Guoyong Zhang, + , T-MTT Mar 05 947-951 supercond. spiral wide bandpass filters, wide upper stopband. Huang, F., T-MTT Jul 05 2335-2339 suppression of second harmonic, fractal-shaped microstrip coupled-line bandpass filters. Il Kwon Kim, + , T-MTT Sep 05 2943-2948 symmetric composite right/left-handed CPW, appls., compact bandpass filters, modeling. Shau-Gang Mao, + , T-MTT Nov 05 3460-3466 tapered dual-plane compact EM bandgap microstrip filter structs. Shao Ying Huang, + , T-MTT Sep 05 2656-2664 waveguide filters, multiple atten. poles, dual-behavior reson. of freq.selective surfaces. Ohira, M., + , T-MTT Nov 05 3320-3326 wide tuning-range planar filters, lumped-distrib. coupled resonators. Carey-Smith, B.E., + , T-MTT Feb 05 777-785

IEEE T-MTT 2005 INDEX — 24 Bandstop filters direct-coupled microwave filters, single and dual stopbands. Cameron, R.J., + , T-MTT Nov 05 3288-3297 dual-passband filters, design techs. Macchiarella, G., + , T-MTT Nov 05 3265-3271 gen. Chebyshev filters, asymmetrically located transm. zeros, design. Milosavljevic, Z.D., T-MTT Jul 05 2411-2415 parallel-coupled line filters with enhanced stopband performances. Fathelbab, W.M., + , T-MTT Dec 05 3774-3781 tapered dual-plane compact EM bandgap microstrip filter structs. Shao Ying Huang, + , T-MTT Sep 05 2656-2664 waveguide filters, multiple atten. poles, dual-behavior reson. of freq.selective surfaces. Ohira, M., + , T-MTT Nov 05 3320-3326 Band-stop filters; cf. Notch filters Barium electronically tunable microstrip bandpass filter, thin-film BariumStrontium-Titanate (BST) varactors. Nath, J., + , T-MTT Sep 05 27072712 Barium compounds adaptive duplexer implemented, single-path/multipath feedforward techs., BST phase shifters. O'Sullivan, T., + , T-MTT Jan 05 106-114 ferroelec. thin-film materials, microwave charactn. Ouaddari, M., + , TMTT Apr 05 1390-1397 scan antennas, ferroelec. substr., rigorous anal. and investigs. Yashchyshyn, Y., + , T-MTT Feb 05 427-438 ultrawide-band tunable true-time delay lines, ferroelec. varactors. Kuylenstierna, D., + , T-MTT Jun 05 2164-2170 Beam steering mutually exclusive data encoding for realization of a full duplexing selfsteering wireless link using a retrodirective array transceiver. Leong, K.M.K.H., + , T-MTT Dec 05 3687-3696 Bessel functions determining TE and TM modes, closed waveguides made up of N cylindrical conductors, efficient method. de la Rubia, V., + , T-MTT Feb 05 670-678 BiCMOS analog integrated circuits compact intell. RF front-end, reconfigurable RFICs, Si-based technols. Mukhopadhyay, R., + , T-MTT Jan 05 81-93 multiple-metal stacked inductors incorporating, extended phys. model, design. Murphy, O.H., + , T-MTT Jun 05 2063-2072 BiCMOS integrated circuits compact intell. RF front-end, reconfigurable RFICs, Si-based technols. Mukhopadhyay, R., + , T-MTT Jan 05 81-93 design of 10-Gb/s AGC amp., jitter considerations. Kucharski, D., + , TMTT Feb 05 590-597 full-wave hybrid differential-integral approach for investig. of multilayer structs. incl. nonuniformly doped diffusions. Wane, S., + , T-MTT Jan 05 200-214 GSM/EGSM/DCS/PCS direct conversion receiver, integr. synthesizer. Young-Jin Kim, + , T-MTT Feb 05 606-613 mm-wave (Bi)CMOS IC, 30-100-GHz inductors and transformers. Dickson, T.O., + , T-MTT Jan 05 123-133 modified loss-compensation method, 0.35-ȝm SiGe BiCMOS technol., broad-band MMICs. Ming-Da Tsai, + , T-MTT Feb 05 496-505 RF-CMOS and SiGe BiCMOS, WCDMA direct-conversion receiver front-end comp. Floyd, B.A., + , T-MTT Apr 05 1181-1188 SiGe HBTs, small-sig. and HF noise modeling. Basaran, U., + , T-MTT Mar 05 919-928 BiCMOS integrated circuits; cf. BiCMOS analog integrated circuits Bifurcation BPSK to ASK signal conversion using injection-locked oscillators theory, part I. Lopez-Villegas, J.M., + , T-MTT Dec 05 3757-3766 control, practical cct. design. Collado, A., + , T-MTT Sep 05 2777-2788 global stability analysis and stabilization of a class-E/F amplifier with a distributed active transformer. Jeon, S., + , T-MTT Dec 05 3712-3722 stabil. ccts. for phase-noise reduction, microwave oscillators. Suarez, A., + , T-MTT Sep 05 2743-2751 Bilinear systems perfectly matched layer implementation using bilinear transforms. Dong, X., + , T-MTT Oct 05 3098-3105 BIMOS integrated circuits; cf. BiCMOS integrated circuits Biological effects of electromagnetic radiation develop realistic num. models of cellular phones for accurate eval. of SAR distrib., human head, procedure. Pisa, S., + , T-MTT Apr 05 1256-1265

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protein thermal unfolding and refolding, near-zone microwaves, ultrasensitive detect. Taylor, K.M., + , T-MTT May 05 1576-1586 SAR distribs., 3-layered bio-media, direct contact, water-loaded modified box-horn applicator. Gupta, R.C., + , T-MTT Sep 05 2665-2671 Biological effects of radiation microwave ablation, triaxial antenna. Brace, C.L., + , T-MTT Jan 05 215220 SAR induced, 2 child head models and adult heads, mobile phones. Hadjem, A., + , T-MTT Jan 05 4-11 Biological organs; cf. Brain; Liver Biological system modeling develop realistic num. models of cellular phones for accurate eval. of SAR distrib., human head, procedure. Pisa, S., + , T-MTT Apr 05 1256-1265 Biological tissues corrections on "Precision open-ended coaxial probes for In Vivo and Ex Vivo dielectric spectroscopy of biological tissues on microwave frequencies" (May 05 1713-1722). Popovic, D., + , T-MTT Sep 05 3053 Biological tissues; cf. Skin Biomedical equipment; cf. Biomedical transducers Biomedical image processing high dielec.-contrast objs., different image-reconstruction approaches, microwave-tomographic imaging. Semenov, S.Y., + , T-MTT Jul 05 2284-2294 Biomedical imaging breast cancer detect., expt. investig. of simple tumor models, tissue sens. adaptive radar. Sill, J.M., + , T-MTT Nov 05 3312-3319 excitable tissues inside human body, focused microwave radiometry, functional noninvasive imaging. Reznik, A.N., T-MTT May 05 18291831 excitable tissues inside human body, focused microwave radiometry ), functional noninvasive imaging. Karanasiou, I.S., + , T-MTT May 05 1831-1832 time reversal, FDTD method for microwave breast cancer detect. Kosmas, P., + , T-MTT Jul 05 2317-2323 Biomedical measurements biol. appls., micromachined probe, in vivo meas. Jung-Mu Kim, + , TMTT Nov 05 3415-3421 low-cost planar probes, broadside apertures for nondestructive dielec. meas. of biol. materials, microwave freqs. Byoungjoong Kang, + , TMTT Jan 05 134-143 Biomedical signal processing in vivo and ex dielec. spectrosc. of biol. tissues, microwave freqs., precision open-ended coaxial probes. Popovic, D., + , T-MTT May 05 1713-1722 Biomedical transducers breast cancer detect., expt. investig. of simple tumor models, tissue sens. adaptive radar. Sill, J.M., + , T-MTT Nov 05 3312-3319 Biothermics; cf. Hyperthermia Bipolar analog integrated circuits InGaP/GaAs HBT RF power amps., ESD protection design considerations. Ma, Y., + , T-MTT Jan 05 221-228 Bipolar integrated circuits; cf. Bipolar analog integrated circuits Bipolar transistor circuits BJT class-F power amp. near transit. freq. Rudiakova, A.N., T-MTT Sep 05 3045-3050 Bipolar transistors stabil. of BJT formulation, FDTD framework. Kung, F., + , T-MTT Apr 05 1189-1196 Bipolar transistors; cf. Heterojunction bipolar transistors; Microwave bipolar transistors; Millimeter wave bipolar transistors; Power bipolar transistors Bolometers terahertz mixers and detectors, design guidelines. Focardi, P., + , T-MTT May 05 1653-1661 Boltzmann equation coupling 3D Maxwell's and Boltzmann's eqns. for analyzing, terahertz photoconductive switch. Sirbu, M., + , T-MTT Sep 05 2991-2998 Bonding processes CAD techniques suitable for the design of high-power RF transistors. Aaen, P.H., + , T-MTT Oct 05 3067-3074 Boron alloys crossed anisotropy mag. core, toroidal thin-film inductors, integrat. Frommberger, M., + , T-MTT Jun 05 2096-2100

IEEE T-MTT 2005 INDEX — 25 Boundary element methods accurate waveguide port boundary condition for time-domain FEM. Zheng Lou, + , T-MTT Sep 05 3014-3023 Boundary integral equations rect. waveguides, radially symmetric metal insets, fast S-domain modeling. Mira, F., + , T-MTT Apr 05 1294-1303 Boundary value problems FDTD method, off-grid perfect boundary conds. Rickard, Y.S., + , T-MTT Jul 05 2274-2283 Bragg gratings wide-band continuously tunable millimeter-wave signal with an optical external modulation technique. Qi, G., + , T-MTT Oct 05 3090-3097 Brain excitable tissues inside human body, focused microwave radiometry, functional noninvasive imaging. Reznik, A.N., T-MTT May 05 18291831 excitable tissues inside human body, focused microwave radiometry ), functional noninvasive imaging. Karanasiou, I.S., + , T-MTT May 05 1831-1832 Brain modeling SAR induced, 2 child head models and adult heads, mobile phones. Hadjem, A., + , T-MTT Jan 05 4-11 Broadband amplifiers 2-stage ultrawide-band 5-W power amp., SiC MESFET. Sayed, A., + , TMTT Jul 05 2441-2449 40-Gb/s wide-band MMIC pHEMT modulator driver amps. designed, real freq. tech. Kerherve, E., + , T-MTT Jun 05 2145-2152 design of 10-Gb/s AGC amp., jitter considerations. Kucharski, D., + , TMTT Feb 05 590-597 fast layout verification of 3D RF and mixed-sig. on-chip structs., largescale broad-band parasitic extr. Feng Ling, + , T-MTT Jan 05 264-273 high av.-effic. SiGe HBT power amp. for WCDMA handset appls. Junxiong Deng, + , T-MTT Feb 05 529-537 on-wafer noise-param. meas., wide-band freq.-var. method. Hu, R., + , TMTT Jul 05 2398-2402 power level-depend. dual-operating mode LDMOS power amp. for CDMA wireless base-station appls. Younkyu Chung, + , T-MTT Feb 05 739-746 RF-CMOS and SiGe BiCMOS, WCDMA direct-conversion receiver front-end comp. Floyd, B.A., + , T-MTT Apr 05 1181-1188 wireless transmitters, digital subband filtering predistorter archit. Hammi, O., + , T-MTT May 05 1643-1652 Broadband communication 60-GHz broad-band telecomm., resistive HEMT mixers. Varonen, M., + , T-MTT Apr 05 1322-1330 act. differential broad-band phase splitter for quadrature-modulator appls. Tiiliharju, E., + , T-MTT Feb 05 679-686 deembedding static nonlinearities and accurately identifying and modeling memory effects, wide-band RF transmitters. Taijun Liu, + , T-MTT Nov 05 3578-3587 highly efficient Doherty feedforward lin. power amp. for W-CDMA basestation appls. Kyoung-Joon Cho, + , T-MTT Jan 05 292-300 measuring BER of UWB devices, prod. test tech. Bhattacharya, S., + , TMTT Nov 05 3474-3481 Broadband networks frequency channel blocking scheme in mesh-topology millimeter-wave broad band entrance networks. Sangiamwong, J., + , T-MTT Dec 05 3723-3730 Butterworth filters miniaturized dual-mode ring bandpass filter, perturb. Boon Tiong Tan, + , T-MTT Jan 05 343-348 C CAD CAD techniques suitable for the design of high-power RF transistors. Aaen, P.H., + , T-MTT Oct 05 3067-3074 quasi-static solutions of multilayer elliptical, cylindrical coplanar striplines and multilayer coplanar striplines with finite dielectric dimensions Asymmetrical case. Akan, V., + , T-MTT Dec 05 3681-3686 Calibration calibrated 200-GHz waveform meas. Williams, D.F., + , T-MTT Apr 05 1384-1389 coupled-line models from EM simulators and appl., MoM anal., derivation. Farina, M., + , T-MTT Nov 05 3272-3280 + Check author entry for coauthors

double-delay and SOC EM deembedding, unification. Rautio, J.C., + , TMTT Sep 05 2892-2898 effect of cable length in vector measurements of very long millimeter wave components. Simonetto, A., + , T-MTT Dec 05 3731-3734 GSM/EGSM/DCS/PCS direct conversion receiver, integr. synthesizer. Young-Jin Kim, + , T-MTT Feb 05 606-613 method, improve VNA calib., planar dispers. media, adding, asymmetrical reciprocal device. Scott, J.B., T-MTT Sep 05 3007-3013 mismatch errors for 400-msamples/s 80-dB SFDR time-interleaved ADC, comprehensive digital correction. Munkyo Seo, + , T-MTT Mar 05 10721082 partially leaky multiport vector net. analyzers, on-wafer calib. algm. Teppati, V., + , T-MTT Nov 05 3665-3671 self-calibrating SSB modulator. Treyer, D.M., + , T-MTT Dec 05 38063816 Capacitance miniaturized dual-mode bandpass filter struct., shunt-capacitance perturb. Ming-Fong Lei, + , T-MTT Mar 05 861-867 on-chip interconnects by double-image Green's fn. method combined, hierarchical algm., param. extr. Wenliang Dai, + , T-MTT Jul 05 24162423 Capacitance measurement substr. removed LV high-speed GaAs/AlGaAs electrooptic modulators, Trail electrodes. JaeHyuk Shin, + , T-MTT Feb 05 636-643 Capacitors differential 4-bit 6.5-10-GHz RF MEMS tunable filter. Entesari, K., + , TMTT Mar 05 1103-1110 RF low-noise bandpass filter, act. capacitance cct., design. Young-Hoon Chun, + , T-MTT Feb 05 687-695 Capacitors; cf. Varactors Carbon; cf. Diamond Carrier mobility; cf. Electron mobility Cascade circuits 2-stage ultrawide-band 5-W power amp., SiC MESFET. Sayed, A., + , TMTT Jul 05 2441-2449 act. differential broad-band phase splitter for quadrature-modulator appls. Tiiliharju, E., + , T-MTT Feb 05 679-686 distrib. preamplifier, cascode FET cells, improved noise anal. Won Ko, + , T-MTT Jan 05 361-371 stopband-enhanced and size-miniaturized low-pass filters, high-impedance property of offset finite-ground microstrip line. Sheng Sun, + , T-MTT Sep 05 2844-2850 Cathodes; cf. Photocathodes Cavity resonator filters low-loss LTCC cavity filters using system-on-package technology at 60 GHz. Lee, J.-H., + , T-MTT Dec 05 3817-3824 Cavity resonators complex permitt. of arbitrary shape and size dielec. samples, cavity meas. tech., microwave freqs., estim. Santra, M., + , T-MTT Feb 05 718-722 cylindrical/spherical dielec. resonators, cavities and MIC environments by of finite elements, CAD-oriented anal. Gil, J.M., T-MTT Sep 05 28662874 dielec. helical resonators. Holmes, J.E., + , T-MTT Jan 05 322-329 low-loss LTCC cavity filters using system-on-package technology at 60 GHz. Lee, J.-H., + , T-MTT Dec 05 3817-3824 reson. Processes, metamaterials, FDTD study. Semouchkina, E.A., + , TMTT Apr 05 1477-1487 tailored and anisotropic dielec. consts., porosity, ceramic components. Xun Gong, + , T-MTT Nov 05 3638-3647 Cavity resonators; cf. Cavity resonator filters Ceramics designing microstrip filters utilizing mixed dielectrics, approaches. Semouchkina, E., + , T-MTT Feb 05 644-652 dielec. helical resonators. Holmes, J.E., + , T-MTT Jan 05 322-329 LTCC, metallic resonators, canonical ridge waveguide filters. Ruiz-Cruz, J.A., + , T-MTT Jan 05 174-182 miniaturized multilayer quasiellipt. bandpass filter, aperture-coupled microstrip resonators. Chi-Feng Chen, + , T-MTT Sep 05 2688-2692 permitt. and loss tangent of high-permitt. materials, terahertz freqs., temp. depend. Berdel, K., + , T-MTT Apr 05 1266-1271 tailored and anisotropic dielec. consts., porosity, ceramic components. Xun Gong, + , T-MTT Nov 05 3638-3647 Chaos global stability analysis and stabilization of a class-E/F amplifier with a distributed active transformer. Jeon, S., + , T-MTT Dec 05 3712-3722

IEEE T-MTT 2005 INDEX — 26 Charge carrier lifetime TW photodetectors, hybrid drift-diffusion-TLM anal. Pasalic, D., + , TMTT Sep 05 2700-2706 Charge carrier mobility coupling 3D Maxwell's and Boltzmann's eqns. for analyzing, terahertz photoconductive switch. Sirbu, M., + , T-MTT Sep 05 2991-2998 Charge carrier processes RF-driven gate current, DC/RF perform., GaAs pHEMT MMIC power amps., effect. Yeong-Chang Chou, + , T-MTT Nov 05 3398-3406 Chebyshev approximation collocation and Newton-type optim. methods for inverse problem, nonuniform transm. lines. Norgren, M., T-MTT May 05 1561-1568 Chebyshev filters differential 4-bit 6.5-10-GHz RF MEMS tunable filter. Entesari, K., + , TMTT Mar 05 1103-1110 dual-passband filters, design techs. Macchiarella, G., + , T-MTT Nov 05 3265-3271 gen. Chebyshev filters, asymmetrically located transm. zeros, design. Milosavljevic, Z.D., T-MTT Jul 05 2411-2415 microstrip bandpass filters with multiorder spurious-mode suppression. Chen, C.-F., + , T-MTT Dec 05 3788-3793 supercond. spiral filters, quasiellipt. charact. for radio astron. Guoyong Zhang, + , T-MTT Mar 05 947-951 Chemical variables measurement; cf. Moisture measurement Circuit analysis anal. of dielec. loaded waveguide filters of arbitrary shape, hybrid surface integral-eqn./mode-matching method. Catina, V., + , T-MTT Nov 05 3562-3567 broad-band poly-harmonic distortion (PHD) behavioral models from fast automated simul. and large-sig. vectorial net. meas. Root, D.E., + , TMTT Nov 05 3656-3664 double-delay and SOC EM deembedding, unification. Rautio, J.C., + , TMTT Sep 05 2892-2898 linearizing FET low-noise amps., modified derivative superposition method. Aparin, V., + , T-MTT Feb 05 571-581 microwave mixers, scatt.-param. models and representations. Williams, D.F., + , T-MTT Jan 05 314-321 RF ccts., LCP substrs., stat. anal. and diagnosis methodology. Mukherjee, S., + , T-MTT Nov 05 3621-3630 stabil. ccts. for phase-noise reduction, microwave oscillators. Suarez, A., + , T-MTT Sep 05 2743-2751 Circuit analysis computing; cf. Circuit simulation Circuit layout; cf. Integrated circuit layout; Printed circuit layout Circuit noise additive and converted noise, gener. of phase noise, nonlin. oscillators, role. Nallatamby, J.-C., + , T-MTT Mar 05 901-906 on-wafer noise-param. meas., wide-band freq.-var. method. Hu, R., + , TMTT Jul 05 2398-2402 RF low-noise bandpass filter, act. capacitance cct., design. Young-Hoon Chun, + , T-MTT Feb 05 687-695 Circuit noise; cf. Integrated circuit noise Circuit optimization asymmetrical-cells-based lin. Doherty power Amplifiers-uneven power drive and power matching, optimum operation. Jangheon Kim, + , TMTT May 05 1802-1809 image-rejection CMOS LNA design optim. techs. Trung-Kien Nguyen, + , T-MTT Feb 05 538-547 intelligently controlled RF power amp., reconfigurable MEMS-varactor tuner. Dongjiang Qiao, + , T-MTT Mar 05 1089-1095 lin. inverse space-mapping (LISM) algm., design lin./nonlin. RF and microwave ccts. Rayas-Sanchez, J.E., + , T-MTT Mar 05 960-968 LTCC, metallic resonators, canonical ridge waveguide filters. Ruiz-Cruz, J.A., + , T-MTT Jan 05 174-182 microstrip-line-type PDLC loaded variable phase shifter, increasing speed. Utsumi, Y., + , T-MTT Nov 05 3345-3353 packaged inductively degenerated common-source low-noise amps., ESD protection, anal. and optim. Sivonen, P., + , T-MTT Apr 05 1304-1313 RF inductors and filters, LCP substrs. for Wi-Fi appls., layout-level synthesis. Mukherjee, S., + , T-MTT Jun 05 2196-2210 robust modeling and design approach for dynamically loaded and digitally linearized Doherty amps. Sirois, J., + , T-MTT Sep 05 2875-2883 small-sig. modeling approach applied, GaN devices. Jarndal, A., + , TMTT Nov 05 3440-3448 Circuit reliability; cf. Integrated circuit reliability

+ Check author entry for coauthors

Circuits baseband-modeled CALLUM archits., spectrum emission considerations. Strandberg, R., + , T-MTT Feb 05 660-669 Circuit simulation microwave filters, adaptive models and param. extr., seq. tuning. Pepe, G., + , T-MTT Jan 05 22-31 Circuit stability stabil. ccts. for phase-noise reduction, microwave oscillators. Suarez, A., + , T-MTT Sep 05 2743-2751 stabil. of microwave amps., variable termination impedances, necessary and sufficient conds. Olivieri, M., + , T-MTT Aug 05 2580-2586 Circuit synthesis 2-stage ultrawide-band 5-W power amp., SiC MESFET. Sayed, A., + , TMTT Jul 05 2441-2449 dual-passband filters, design techs. Macchiarella, G., + , T-MTT Nov 05 3265-3271 high-effic. current-mode class-D amps. for wireless handsets, design. TsaiPi Hung, + , T-MTT Jan 05 144-151 in-line N-order filters, N real transm. zeros by of extracted poles implemented, low-cost rect. H-plane waveguide, synthesis and design. Montejo-Garai, J.R., + , T-MTT May 05 1636-1642 miniaturized planar Marchand baluns, classes. Fathelbab, W.M., + , TMTT Apr 05 1211-1220 RF/microwave multifunctional systs., reconfigurable bandpass filter. Fathelbab, W.M., + , T-MTT Mar 05 1111-1116 synthesis of cascaded N-tuplets cross-coupled resonators microwave filters, matrix rotations, anal. tech. Tamiazzo, S., + , T-MTT May 05 1693-1698 Circuit testing; cf. Integrated circuit testing Circuit topology 40-Gb/s wide-band MMIC pHEMT modulator driver amps. designed, real freq. tech. Kerherve, E., + , T-MTT Jun 05 2145-2152 dual-passband filters, design techs. Macchiarella, G., + , T-MTT Nov 05 3265-3271 g-band (140-220 GHz) and W-band (75-110 GHz) InP DHBT medium power amps. Paidi, V.K., + , T-MTT Feb 05 598-605 Circuit tuning 3rd.- and fifth-order baseband component injection for linearization of power amp., cellular phone. Mizusawa, N., + , T-MTT Nov 05 33273334 broad-band power amp., tunable output matching net. Haitao Zhang, + , T-MTT Nov 05 3606-3614 compact intell. RF front-end, reconfigurable RFICs, Si-based technols. Mukhopadhyay, R., + , T-MTT Jan 05 81-93 differential 4-bit 6.5-10-GHz RF MEMS tunable filter. Entesari, K., + , TMTT Mar 05 1103-1110 expt. class-F power amp. design, computationally efficient and accurate large-sig. pHEMT model. Wren, M., + , T-MTT May 05 1723-1731 high-power MEMS varactors and impedance tuners for mm-wave appls. Lu, Y., + , T-MTT Nov 05 3672-3678 intelligently controlled RF power amp., reconfigurable MEMS-varactor tuner. Dongjiang Qiao, + , T-MTT Mar 05 1089-1095 metamaterial-based electronically controlled transm.-line struct., leakywave antenna, tunable radiation angle and beamwidth. Sungjoon Lim, + , T-MTT Jan 05 161-173 microwave filters, adaptive models and param. extr., seq. tuning. Pepe, G., + , T-MTT Jan 05 22-31 multistandard adaptive voltage-controlled oscillators, design. Tasic, A., + , T-MTT Feb 05 556-563 RF/microwave multifunctional systs., reconfigurable bandpass filter. Fathelbab, W.M., + , T-MTT Mar 05 1111-1116 tunable combline filter, continuous control of center freq. and bandwidth. Sanchez-Renedo, M., + , T-MTT Jan 05 191-199 Circular waveguides anal. of radial waveguides, dielec. resonators, microstrip antennas, spherical multilayer structs., unified approach. Truong Vu Bang Giang, + , T-MTT Jan 05 404-409 determining TE and TM modes, closed waveguides made up of N cylindrical conductors, efficient method. de la Rubia, V., + , T-MTT Feb 05 670-678 Green's fns. for cylindrical enclosures by spatial images method, num. eval. Pereira, F.D.Q., + , T-MTT Jan 05 94-105 high-performance circular TE01-mode converter. Yu, C.-F., + , T-MTT Dec 05 3794-3798

IEEE T-MTT 2005 INDEX — 27 k-band orthomode transducer, waveguide ports and balanced coaxial probes. Engargiola, G., + , T-MTT May 05 1792-1801 multifrequency waveguide orthomode transducer. Sharma, S.B., + , TMTT Aug 05 2604-2609 propag. characts. of cylindrical CPW, finite thickness of conductor, fullwave anal. Yamamoto, H., + , T-MTT Jun 05 2187-2195 TE0n-TE0(n+1) ripple-wall mode converters, circ. waveguide, bandwidth studies. Lawson, W., + , T-MTT Jan 05 372-379 waveguide polarizers, design tool. Virone, G., + , T-MTT Mar 05 888-894 Circulators 1st. circ. conds. of turnstile waveguide circulators, finite-element solver, verification. Helszajn, J., + , T-MTT Jul 05 2309-2316 Circulators; cf. Ferrite circulators; Millimeter wave circulators CMOS analog integrated circuits 0.6-V 1.6-mW transformer-based 2.5-GHz downconversion mixer, +5.4dB gain and -2.8-dBm IIP3, 0.13-ȝm CMOS. Hermann, C., + , T-MTT Feb 05 488-495 linearizing FET low-noise amps., modified derivative superposition method. Aparin, V., + , T-MTT Feb 05 571-581 mm-wave CMOS cct. design. Shigematsu, H., + , T-MTT Feb 05 472-477 on-chip vert. solenoid inductor design for multigigahertz CMOS RFIC. Hau-Yiu Tsui, + , T-MTT Jun 05 1883-1890 CMOS integrated circuits 0.25-ȝm CMOS OPLL transmitter IC for GSM and DCS appls. Peng-Un Su, T-MTT Feb 05 462-471 10-Gb/s fiber opt. commun. links, 0.18-ȝm CMOS equalization techs. Moonkyun Maeng, + , T-MTT Nov 05 3509-3519 1-, 10-GHz distrib. amp., CMOS technol., ESD protection design. MingDou Ker, + , T-MTT Sep 05 2672-2681 20-Gb/s 4-PAM backplane serial I/O interconnections, equalization and NEXT noise cancellation. Hur, Y., + , T-MTT Jan 05 246-255 2.17-dB NF 5-GHz-band monolithic CMOS LNA, 10-mW DC power consumption. Hung-Wei Chiu, + , T-MTT Mar 05 813-824 2.4-GHz-band GFSK appls., low-power highly digitized receiver. Bergveld, H.J., + , T-MTT Feb 05 453-461 90-nm VLSI SOI CMOS technol., high linearity for WLAN, 26.5-30-GHz resistive mixer. Ellinger, F., T-MTT Aug 05 2559-2565 broad-band RF ccts., decreasing-size distrib. ESD protection scheme. Ming-Dou Ker, + , T-MTT Feb 05 582-589 carrier heating, channel noise, deep-submicrometer NMOSFETs via body bias, expt. study. Hong Wang, + , T-MTT Feb 05 564-570 compact intell. RF front-end, reconfigurable RFICs, Si-based technols. Mukhopadhyay, R., + , T-MTT Jan 05 81-93 image-rejection CMOS LNA design optim. techs. Trung-Kien Nguyen, + , T-MTT Feb 05 538-547 interf. canceller for collocated radios, anal. and design. Raghavan, A., + , T-MTT Nov 05 3498-3508 k- and Q-bands CMOS freq. sources, X-band quadrature VCO. Sangsoo Ko, + , T-MTT Sep 05 2789-2800 lin. inverse space-mapping (LISM) algm., design lin./nonlin. RF and microwave ccts. Rayas-Sanchez, J.E., + , T-MTT Mar 05 960-968 LNA protection, watt-level CMOS transceivers, reson. switch. Kuhn, W.B., + , T-MTT Sep 05 2819-2825 low-IF receivers, image-rejection down-converter. Sher Jiun Fang, + , TMTT Feb 05 478-487 microwave on-wafer charactn. of deep-submicrometer Si MOSFETs, shield-based 3-port de-embedding method. Ming-Hsiang Cho, + , TMTT Sep 05 2926-2934 multilayer on-chip inductors, phys. anal. model. Tong, K.Y., + , T-MTT Apr 05 1143-1149 multistandard mobile terminals, fully integr. receivers requirements and archits. Brandolini, M., + , T-MTT Mar 05 1026-1038 on-chip interconnects by double-image Green's fn. method combined, hierarchical algm., param. extr. Wenliang Dai, + , T-MTT Jul 05 24162423 packaged inductively degenerated common-source low-noise amps., ESD protection, anal. and optim. Sivonen, P., + , T-MTT Apr 05 1304-1313 RF amp. and mixer ccts. utilizing complementary Characteristics of parallel combined NMOS and PMOS devices. Nam, I., + , T-MTT May 05 1662-1671 RF-CMOS and SiGe BiCMOS, WCDMA direct-conversion receiver front-end comp. Floyd, B.A., + , T-MTT Apr 05 1181-1188 serial communs., data-depend. jitter. Analui, B., + , T-MTT Nov 05 33883397

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simple systematic spiral inductor design, perfected Q improv. for CMOS RFIC appl. Chih-Yuan Lee, + , T-MTT Feb 05 523-528 std. CMOS technol., monolithic integr. mm-wave IMPATT transmitter. Al-Attar, T., + , T-MTT Nov 05 3557-3561 technol., complementary Colpitts oscillator. Choong-Yul Cha, + , T-MTT Mar 05 881-887 CMOS integrated circuits; cf. CMOS analog integrated circuits Coaxial transmission lines k-band orthomode transducer, waveguide ports and balanced coaxial probes. Engargiola, G., + , T-MTT May 05 1792-1801 liq. crysts., temp.-controlled coaxial transm. line, broad-band microwave charactn. Mueller, S., + , T-MTT Jun 05 1937-1945 waveguide-based left-handed transm. lines, anal., modeling, appls. Salehi, H., + , T-MTT Nov 05 3489-3497 Coaxial waveguides input impedance of top-loaded monopole, parallel-plate waveguide by MoM/Green's fn. method. Valero-Nogueira, A., + , T-MTT Mar 05 868873 waveguide-based left-handed transm. lines, anal., modeling, appls. Salehi, H., + , T-MTT Nov 05 3489-3497 Cobalt alloys crossed anisotropy mag. core, toroidal thin-film inductors, integrat. Frommberger, M., + , T-MTT Jun 05 2096-2100 Code division multiaccess 2.14-GHz Chireix outphasing transmitter. Hakala, I., + , T-MTT Jun 05 2129-2138 3rd.- and fifth-order baseband component injection for linearization of power amp., cellular phone. Mizusawa, N., + , T-MTT Nov 05 33273334 adaptive duplexer implemented, single-path/multipath feedforward techs., BST phase shifters. O'Sullivan, T., + , T-MTT Jan 05 106-114 adaptive predistortion RF power amp., spectrum monitor for multicarrier WCDMA appls. Seung-Yup Lee, + , T-MTT Feb 05 786-793 asymmetrical-cells-based lin. Doherty power Amplifiers-uneven power drive and power matching, optimum operation. Jangheon Kim, + , TMTT May 05 1802-1809 baseband-modeled CALLUM archits., spectrum emission considerations. Strandberg, R., + , T-MTT Feb 05 660-669 class-F and inverse class-F HBT amps., IMD anal. Ohta, A., + , T-MTT Jun 05 2121-2128 deembedding static nonlinearities and accurately identifying and modeling memory effects, wide-band RF transmitters. Taijun Liu, + , T-MTT Nov 05 3578-3587 dual-band transmitters, digitally predistorted freq. multipliers for reconfigurable radios. Youngcheol Park, + , T-MTT Jan 05 115-122 high av.-effic. SiGe HBT power amp. for WCDMA handset appls. Junxiong Deng, + , T-MTT Feb 05 529-537 high-effic. multistage Doherty power amp. for wireless communs., anal. and design. Srirattana, N., + , T-MTT Mar 05 852-860 highly efficient Doherty feedforward lin. power amp. for W-CDMA basestation appls. Kyoung-Joon Cho, + , T-MTT Jan 05 292-300 hybrid digital/RF envelope predistortion linearization syst. for power amps. Wangmyong Woo, + , T-MTT Jan 05 229-237 LINC amps., Chireix-outphasing combiners, phase-only predistortion. Birafane, A., + , T-MTT Jun 05 2240-2250 linearizing FET low-noise amps., modified derivative superposition method. Aparin, V., + , T-MTT Feb 05 571-581 low-IF receivers, image-rejection down-converter. Sher Jiun Fang, + , TMTT Feb 05 478-487 multichannel commun. systs., modeling distortion. Gharaibeh, K.M., + , T-MTT May 05 1682-1692 power level-depend. dual-operating mode LDMOS power amp. for CDMA wireless base-station appls. Younkyu Chung, + , T-MTT Feb 05 739-746 RF-CMOS and SiGe BiCMOS, WCDMA direct-conversion receiver front-end comp. Floyd, B.A., + , T-MTT Apr 05 1181-1188 RF front-end characts., spectral regrowth of communs. sigs., impact. Gard, K.G., + , T-MTT Jun 05 2179-2186 single and multicarrier W-CDMA sigs., LINC digital component separator. Gerhard, W., + , T-MTT Jan 05 274-282 Coils; cf. Solenoids Communication channels single and multicarrier W-CDMA sigs., LINC digital component separator. Gerhard, W., + , T-MTT Jan 05 274-282

IEEE T-MTT 2005 INDEX — 28 Communication equipment serial communs., data-depend. jitter. Analui, B., + , T-MTT Nov 05 33883397 Communication system signaling high-speed elec. backplane transm., duobinary signaling. Sinsky, J.H., + , T-MTT Jan 05 152-160 Communication terminals 6-port, communs. receiver. Hentschel, T., T-MTT Mar 05 1039-1047 multistandard mobile terminals, fully integr. receivers requirements and archits. Brandolini, M., + , T-MTT Mar 05 1026-1038 Compensation planar microwave components, EM lin. regression models, empirical model gener. techs. Domenech-Asensi, G., + , T-MTT Nov 05 3305-3311 Computer applications; cf. CAD Computerized control; cf. Intelligent control Computer networks; cf. Local area networks Conductivity IMD, contact-type MEMS microswitch, determ. Johnson, J., + , T-MTT Nov 05 3615-3620 Conductors differential surface admittance operator, skin effect modeling. De Zutter, D., + , T-MTT Aug 05 2526-2538 FDTD method, off-grid perfect boundary conds. Rickard, Y.S., + , T-MTT Jul 05 2274-2283 parallel coupled microstrip filters, floating ground-plane conductor for spurious-band suppression. Velazquez-Ahumada, Md.C., + , T-MTT May 05 1823-1828 propag. characts. of cylindrical CPW, finite thickness of conductor, fullwave anal. Yamamoto, H., + , T-MTT Jun 05 2187-2195 Conformal mapping quasi-static solutions of multilayer elliptical, cylindrical coplanar striplines and multilayer coplanar striplines with finite dielectric dimensions Asymmetrical case. Akan, V., + , T-MTT Dec 05 3681-3686 Connectors inter-chip RF-interconnect, CPW, capacitive coupler, UWB transceiver, perform. Sun, M., + , T-MTT Sep 05 2650-2655 Contact resistance rough rect. waveguide flanges, pass.-intermodulation anal. Vicente, C., + , T-MTT Aug 05 2515-2525 Control of specific variables; cf. Frequency control Control systems; cf. Reduced order systems Control theory; cf. Adaptive control; Compensation Convergence; cf. Convergence of numerical methods Convergence of numerical methods FEM solns. of dielec. waveguiding structs., 2D curl-conforming sing. elements. Din-Kow Sun, + , T-MTT Mar 05 984-992 Green's fns. for cylindrical enclosures by spatial images method, num. eval. Pereira, F.D.Q., + , T-MTT Jan 05 94-105 Convergence of numerical methods; cf. Numerical stability Converters DVB-S appls., Si bipolar technol., monolithic 12-GHz heterodyne receiver. Girlando, G., + , T-MTT Mar 05 952-959 Converters; cf. Driver circuits; Impedance converters Coplanar waveguides 12-18-GHz 3-pole RF MEMS tunable filter. Entesari, K., + , T-MTT Aug 05 2566-2571 60-GHz broad-band telecomm., resistive HEMT mixers. Varonen, M., + , T-MTT Apr 05 1322-1330 calibrated 200-GHz waveform meas. Williams, D.F., + , T-MTT Apr 05 1384-1389 compact bandpass filter, 2 tuning transm. zeros, CMRC resonator. Kam Man Shum, + , T-MTT Mar 05 895-900 compact MMIC CPW and asymmetric CPS branch-line couplers and Wilkinson dividers, shunt and series stub loading. Hettak, K., + , T-MTT May 05 1624-1635 compact super-wide bandpass substr. integr. waveguide (SIW) filters. Zhang-Cheng Hao, + , T-MTT Sep 05 2968-2977 coupled slotline mode, finite-ground CPW, unequal ground-plane widths, excit. Ponchak, G.E., + , T-MTT Feb 05 713-717 current probe transit. from grounded coplanar, substr. integr. rect. waveguides, anal. and design. Deslandes, D., + , T-MTT Aug 05 24872494 ferroelec. thin-film materials, microwave charactn. Ouaddari, M., + , TMTT Apr 05 1390-1397

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hybrid narrow-band tunable bandpass filter, varactor loaded EM-bandgap CPW. Pistono, E., + , T-MTT Aug 05 2506-2514 inter-chip RF-interconnect, CPW, capacitive coupler, UWB transceiver, perform. Sun, M., + , T-MTT Sep 05 2650-2655 microwave filters, improved stopband, sub-wavel. resonators. GarciaGarcia, J., + , T-MTT Jun 05 1997-2006 planar duplexers/triplexers by manipulating atten. poles, design methodologies. Ohno, T., + , T-MTT Jun 05 2088-2095 propag. characts. of cylindrical CPW, finite thickness of conductor, fullwave anal. Yamamoto, H., + , T-MTT Jun 05 2187-2195 RF switch matrix appls., integr. interconnect nets. Daneshmand, M., + , TMTT Jan 05 12-21 semilumped CPW elements for Millimeter-wave filter design, charactn. Aryanfar, F., + , T-MTT Apr 05 1288-1293 shielded LTCC vert. transits. from DC up, 50 GHz, high performs. Valois, R., + , T-MTT Jun 05 2026-2032 Si Schottky diode DEMUX cct. for high bit-rate opt. receivers. Jung Han Choi, + , T-MTT Jun 05 2033-2042 substr. integr. waveguide cavity filter, defected ground struct. Yu Lin Zhang, + , T-MTT Apr 05 1280-1287 subterahertz freqs., atten. characts. Jingjing Zhang, + , T-MTT Nov 05 3281-3287 surface-passivated high-resist. Si, true microwave substr. Spirito, M., + , T-MTT Jul 05 2340-2347 symmetric composite right/left-handed CPW, appls., compact bandpass filters, modeling. Shau-Gang Mao, + , T-MTT Nov 05 3460-3466 terahertz mixers and detectors, design guidelines. Focardi, P., + , T-MTT May 05 1653-1661 thin-film microstrip and coplanar technols. for reduced-size MMICs, integrat. Hettak, K., + , T-MTT Jan 05 283-291 ultrawide-band tunable true-time delay lines, ferroelec. varactors. Kuylenstierna, D., + , T-MTT Jun 05 2164-2170 wide-band finite-element model-order reduction, fast waveguide eigenanalysis. Shih-Hao Lee, + , T-MTT Aug 05 2552-2558 wide-band supercond. coplanar delay lines. Yi Wang, + , T-MTT Jul 05 2348-2354 Copper high aspect ratio through-wafer interconnect vias, Si substrs., microwave charactn. and modeling. Leung, L.L.W., + , T-MTT Aug 05 2472-2480 Cores; cf. Magnetic cores Correlation InP HBT noise params., noise-fig. meas. syst., direct extr. Jianjun Gao, + , T-MTT Jan 05 330-335 nonlin. model testing, designing multisine excit. Pedro, J.C., + , T-MTT Jan 05 45-54 RF front-end characts., spectral regrowth of communs. sigs., impact. Gard, K.G., + , T-MTT Jun 05 2179-2186 software-defined radio systs., sig. path optim. Rykaczewski, P., + , T-MTT Mar 05 1056-1064 Corundum; cf. Sapphire Coupled circuits insertion loss function synthesis of maximally flat parallel-coupled line bandpass filters. Chin, K.-S., + , T-MTT Oct 05 3161-3168 Coupling circuits compact parallel-coupled microstrip bandpass filters, lumped-element Kinverters. Yo-Shen Lin, + , T-MTT Jul 05 2324-2328 highly integr. mm-wave pass. components, 3D LTCC syst.-on-package (SOP) technol. Jong-Hoon Lee, + , T-MTT Jun 05 2220-2229 microstrip quadruplet filters, source-load coupling, design. Ching-Ku Liao, + , T-MTT Jul 05 2302-2308 miniaturized parallel coupled-line bandpass filter, spurious-response suppression. Pedro Cheong, + , T-MTT May 05 1810-1816 parallel coupled microstrip filters, floating ground-plane conductor for spurious-band suppression. Velazquez-Ahumada, Md.C., + , T-MTT May 05 1823-1828 periodically nonuniform coupled microstrip-line filters, harmonic suppression, transm. zero reallocation. Sheng Sun, + , T-MTT May 05 1817-1822 port discontinuities, full-wave CAD models of multiport ccts., deembedding. Farina, M., T-MTT May 05 1829 port discontinuities, full-wave CAD models of multiport ccts. ), deembedding. Okhmatovski, V.I., + , T-MTT May 05 1829 Crosstalk 20-Gb/s 4-PAM backplane serial I/O interconnections, equalization and NEXT noise cancellation. Hur, Y., + , T-MTT Jan 05 246-255

IEEE T-MTT 2005 INDEX — 29 Cryogenic electronics init. design of LTCC filters, all-capacitive coupling, simplified anal. tech. Rambabu, K., + , T-MTT May 05 1787-1791 Cryogenics very low-noise differential radiometer, 30 GHz for PLANCK LFI. Aja, B., + , T-MTT Jun 05 2050-2062 Cryogenics; cf. Cryogenic electronics Crystal filters millimeter-wave substrate integrated waveguides and filters in photoimageable thick-film technology. Stephens, D., + , T-MTT Dec 05 3832-3838 Crystal filters; cf. Surface acoustic wave filters Current density 2D nonuniform FFT (2-D NUFFT) tech., anal. of shielded microstrip ccts. Ke-Ying Su, + , T-MTT Mar 05 993-999 distrib. preamplifier, cascode FET cells, improved noise anal. Won Ko, + , T-MTT Jan 05 361-371 Current distribution differential surface admittance operator, skin effect modeling. De Zutter, D., + , T-MTT Aug 05 2526-2538 microstrip nets., orthogonality-based deembedding tech. Spowart, M.P., + , T-MTT Mar 05 938-946 Curve fitting low-pass filters, defected ground struct., design. Jong-Sik Lim, + , T-MTT Aug 05 2539-2545 Cyclotron masers high-performance circular TE01-mode converter. Yu, C.-F., + , T-MTT Dec 05 3794-3798 D Data buses 20-Gb/s 4-PAM backplane serial I/O interconnections, equalization and NEXT noise cancellation. Hur, Y., + , T-MTT Jan 05 246-255 multigigahertz parallel bus, transmit preemphasis equalization, perform. anal. and model-to-hardware correl. Beyene, W.T., + , T-MTT Nov 05 3568-3577 Data communication high-speed elec. backplane transm., duobinary signaling. Sinsky, J.H., + , T-MTT Jan 05 152-160 InP HEMTs and their appls., mm-wave radio-on-fiber systs., phototransistors. Chang-Soon Choi, + , T-MTT Jan 05 256-263 multimode J-pHEMT front-end archit., power-control scheme for max. effic. Clifton, J.C., + , T-MTT Jun 05 2251-2258 serial communs., data-depend. jitter. Analui, B., + , T-MTT Nov 05 33883397 Data communication equipment; cf. Modems Data conversion; cf. Analog-digital conversion Data handling; cf. Table lookup Data security secure high-speed retrodirective commun. link. Goshi, D.S., + , T-MTT Nov 05 3548-3556 Delay circuits; cf. Delay lines Delay effects multiconductor transm. lines, nonlin. terminations, delay-extr.-based sensitivity anal. Nakhla, N.M., + , T-MTT Nov 05 3520-3530 short-channel AlGaN/GaN heterojunction FETs, 30-GHz-band, 5-W power perform. Inoue, T., + , T-MTT Jan 05 74-80 Delay lines super-compact multilayered left-handed transm. line and diplexer appl. Horii, Y., + , T-MTT Apr 05 1527-1534 ultrawide-band tunable true-time delay lines, ferroelec. varactors. Kuylenstierna, D., + , T-MTT Jun 05 2164-2170 wide-band supercond. coplanar delay lines. Yi Wang, + , T-MTT Jul 05 2348-2354 Delta modulation; cf. Sigma-delta modulation Demodulation 2.4-GHz-band GFSK appls., low-power highly digitized receiver. Bergveld, H.J., + , T-MTT Feb 05 453-461 Demodulators; cf. Modems Demultiplexing Si Schottky diode DEMUX cct. for high bit-rate opt. receivers. Jung Han Choi, + , T-MTT Jun 05 2033-2042

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Design automation complex pass. devices composed of arbitrarily shaped waveguides, Nystrom and BI-RME methods, CAD. Taroncher, M., + , T-MTT Jun 05 2153-2163 cylindrical/spherical dielec. resonators, cavities and MIC environments by of finite elements, CAD-oriented anal. Gil, J.M., T-MTT Sep 05 28662874 planar microwave components, EM lin. regression models, empirical model gener. techs. Domenech-Asensi, G., + , T-MTT Nov 05 3305-3311 port discontinuities, full-wave CAD models of multiport ccts., deembedding. Farina, M., T-MTT May 05 1829 port discontinuities, full-wave CAD models of multiport ccts. ), deembedding. Okhmatovski, V.I., + , T-MTT May 05 1829 TLM-based modeling and design exploiting space mapping. Bandler, J.W., + , T-MTT Sep 05 2801-2811 Design methodology complex impedances, RFID tag design, power refl. coeff. anal. Nikitin, P.V., + , T-MTT Sep 05 2721-2725 Detectors detect. and localization of mobile phones, large buildings, microwave syst. Hudec, P., + , T-MTT Jun 05 2235-2239 Diamond freq. response of SAW filters, FDTD method. King-Yuen Wong, + , TMTT Nov 05 3364-3370 Dielectric devices capacitive nonlinearity in thin-film BST varactors. Chase, D.R., + , TMTT Oct 05 3215-3220 Dielectric devices; cf. Capacitors; Dielectric resonators; Ferroelectric devices; Piezoelectric devices Dielectric films high-quality solenoid inductor, dielec. film for multichip modules. JongMin Yook, + , T-MTT Jun 05 2230-2234 low-loss broad-band planar baluns, multilayered organic thin films. Chen, A.C., + , T-MTT Nov 05 3648-3655 low-loss TFMS, Si substr. up, 220 GHz, fab. and charactn. Six, G., + , TMTT Jan 05 301-305 Dielectric loaded waveguides rect. waveguide, dielec.-filled corrugations supporting backward waves. Eshrah, I.A., + , T-MTT Nov 05 3298-3304 Dielectric losses complex permitt. of arbitrary shape and size dielec. samples, cavity meas. tech., microwave freqs., estim. Santra, M., + , T-MTT Feb 05 718-722 const., loss tangent, surface resist. of PCB materials, K-band freqs. Egorov, V.N., + , T-MTT Feb 05 627-635 measuring multilayered dielec. plates, open resonator tech. Deleniv, A.N., + , T-MTT Sep 05 2908-2916 Dielectric materials complex permitt. meas., TE11p modes, circ. cylindrical cavities. Zinal, S., + , T-MTT Jun 05 1870-1874 complex permitt. of arbitrary shape and size dielec. samples, cavity meas. tech., microwave freqs., estim. Santra, M., + , T-MTT Feb 05 718-722 const., loss tangent, surface resist. of PCB materials, K-band freqs. Egorov, V.N., + , T-MTT Feb 05 627-635 designing microstrip filters utilizing mixed dielectrics, approaches. Semouchkina, E., + , T-MTT Feb 05 644-652 dielec.-filled cavity filters, ultrawide stopband Characteristics, design. Rauscher, C., T-MTT May 05 1777-1786 helical resonators. Holmes, J.E., + , T-MTT Jan 05 322-329 high-quality solenoid inductor, dielec. film for multichip modules. JongMin Yook, + , T-MTT Jun 05 2230-2234 low-loss broad-band planar baluns, multilayered organic thin films. Chen, A.C., + , T-MTT Nov 05 3648-3655 measuring multilayered dielec. plates, open resonator tech. Deleniv, A.N., + , T-MTT Sep 05 2908-2916 metamaterials, neg. refr. index, 2D shunt and 3D SCN TLM nets., modeling. So, P.P.M., + , T-MTT Apr 05 1496-1505 microwave/mm-wave appls., dielec. slab Rotman lens. Jaeheung Kim, + , T-MTT Aug 05 2622-2627 mode-matching analysis of a shielded rectangular dielectric-rod waveguide. Wells, C.G., + , T-MTT Oct 05 3169-3177 mode-stirrer configurations, dielec. heating perform., multimode microwave applicators, effect. Plaza-Gonzalez, P., + , T-MTT May 05 1699-1706 permitt. and loss tangent of high-permitt. materials, terahertz freqs., temp. depend. Berdel, K., + , T-MTT Apr 05 1266-1271

IEEE T-MTT 2005 INDEX — 30 rect. waveguide, dielec.-filled corrugations supporting backward waves. Eshrah, I.A., + , T-MTT Nov 05 3298-3304 reson. Processes, metamaterials, FDTD study. Semouchkina, E.A., + , TMTT Apr 05 1477-1487 temporally dispers. dielec., time-domain cavity oscills. supported. Aksoy, S., + , T-MTT Aug 05 2465-2471 uniplanar left-handed metamaterials, efficient modeling. Yunchuan Guo, + , T-MTT Apr 05 1462-1468 Dielectric materials; cf. Ferroelectric materials Dielectric measurement; cf. Permittivity measurement Dielectric measurements ferroelec. thin-film materials, microwave charactn. Ouaddari, M., + , TMTT Apr 05 1390-1397 low-cost planar probes, broadside apertures for nondestructive dielec. meas. of biol. materials, microwave freqs. Byoungjoong Kang, + , TMTT Jan 05 134-143 Dielectric properties; cf. Capacitance; Dielectric losses; Permittivity Dielectric resonator oscillators cylindrical multilayered ceramic resonators, rect. air cavity for low-phase noise K/Ka-band oscillators. El-Tager, A.M., + , T-MTT Jun 05 22112219 Dielectric resonators const., loss tangent, surface resist. of PCB materials, K-band freqs. Egorov, V.N., + , T-MTT Feb 05 627-635 consts. of metallic nanoparticles embedded, paraffin rod, microwave freqs., meas. Yan-Shian Yeh, + , T-MTT May 05 1756-1760 cylindrical/spherical dielec. resonators, cavities and MIC environments by of finite elements, CAD-oriented anal. Gil, J.M., T-MTT Sep 05 28662874 helical resonators. Holmes, J.E., + , T-MTT Jan 05 322-329 measuring multilayered dielec. plates, open resonator tech. Deleniv, A.N., + , T-MTT Sep 05 2908-2916 mm-wave appls., extremely high-Q factor dielec. resonators. Krupka, J., + , T-MTT Feb 05 702-712 modified ring dielec. resonator, improved mode separation and tunability characts., MIC environ. Srivastava, K.V., + , T-MTT Jun 05 1960-1967 resonator, discrete electromechanical freq. tuning. Panaitov, G.I., + , TMTT Nov 05 3371-3377 Dielectric resonators; cf. Dielectric resonator oscillators Dielectric waveguides FEM solns. of dielec. waveguiding structs., 2D curl-conforming sing. elements. Din-Kow Sun, + , T-MTT Mar 05 984-992 nonuniform plasma layer model for quartz-Si image guide phase shifters, expt. verification. Fickenscher, T., + , T-MTT Jul 05 2375-2382 Dielectric waveguides; cf. Nonradiative dielectric waveguides; Optical waveguides Differential amplifiers deembedding static nonlinearities and accurately identifying and modeling memory effects, wide-band RF transmitters. Taijun Liu, + , T-MTT Nov 05 3578-3587 Diffraction gratings; cf. Bragg gratings Digital circuits extracting causal time-domain params., iter. methods. Shuiping Luo, + , TMTT Mar 05 969-976 wideband suppression of ground bounce noise and radiated emission, high-speed ccts., EM bandgap power/ground planes. Tzong-Lin Wu, + , T-MTT Sep 05 2935-2942 Digital circuits; cf. Digital filters; Digital integrated circuits Digital control hybrid digital/RF envelope predistortion linearization syst. for power amps. Wangmyong Woo, + , T-MTT Jan 05 229-237 Digital filters differential 4-bit 6.5-10-GHz RF MEMS tunable filter. Entesari, K., + , TMTT Mar 05 1103-1110 Digital integrated circuits 12-GHz SiGe phase shifter, integr. LNA. Hancock, T.M., + , T-MTT Mar 05 977-983 high-speed elec. backplane transm., duobinary signaling. Sinsky, J.H., + , T-MTT Jan 05 152-160 Digital signal processors effic. of OFDM transmitters, RF/DSP design. Helaoui, M., + , T-MTT Jul 05 2355-2361 software-defined direct conversion receiver, ka-band analog front-end. Tatu, S.O., + , T-MTT Sep 05 2768-2776

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Diodes compact and broad-band millimeter-wave monolithic transformer balanced mixers. Wu, P.-S., + , T-MTT Oct 05 3106-3114 Dipole antennas leaky-wave propag., metamaterial grounded slabs excited by dipole source, effects. Baccarelli, P., + , T-MTT Jan 05 32-44 Dipole arrays increasing wireless channel capacity, MIMO systs. employing colocated antennas. Konanur, A.S., + , T-MTT Jun 05 1837-1844 Directional couplers bridged NRD-guide coupler for mm-wave appls., anal. and design. Duochuan Li, + , T-MTT Aug 05 2546-2551 Directional couplers; cf. Millimeter wave directional couplers Direction of arrival estimation detect. and localization of mobile phones, large buildings, microwave syst. Hudec, P., + , T-MTT Jun 05 2235-2239 secure high-speed retrodirective commun. link. Goshi, D.S., + , T-MTT Nov 05 3548-3556 Directive antennas PLL-based retrodirective array, anal. and charactn. Fusco, V., + , T-MTT Feb 05 730-738 Directive antennas; cf. Horn antennas; Lens antennas Discrete Fourier transforms 2D nonuniform FFT (2-D NUFFT) tech., anal. of shielded microstrip ccts. Ke-Ying Su, + , T-MTT Mar 05 993-999 layout verification of 3D RF and mixed-sig. on-chip structs., large-scale broad-band parasitic extr. Feng Ling, + , T-MTT Jan 05 264-273 parallel FFT accelerated transient field-cct. simulator. Yilmaz, A.E., + , TMTT Sep 05 2851-2865 software-defined radio systs., sig. path optim. Rykaczewski, P., + , T-MTT Mar 05 1056-1064 Discrete transforms; cf. Discrete Fourier transforms Discriminators corrections to "Mode discriminator based on mode-selective coupling" (Jan 03 55-63). Wenxiang Wang, T-MTT May 05 1833 Distance measurement stereoscopic pass. mm-wave imaging/ranging. Luthi, T., + , T-MTT Aug 05 2594-2599 Distortion measurement distortion analysis technique based on simulated large-signal voltage and current spectra. Aikio, J.P., + , T-MTT Oct 05 3057-3066 theory of chained-function filters. Chrisostomidis, C.E., + , T-MTT Oct 05 3142-3151 Distributed amplifiers 1-, 10-GHz distrib. amp., CMOS technol., ESD protection design. MingDou Ker, + , T-MTT Sep 05 2672-2681 compact dual-fed distrib. power amp. Eccleston, K.W., T-MTT Mar 05 825-831 drain-line loss and the S22 kink effect in capacitively coupled distributed amplifiers. Shohat, J., + , T-MTT Dec 05 3767-3773 InP/InGaAs DHBT distrib. amps., modulator drivers for 80-Gbit/s operation, comp. Schneider, K., + , T-MTT Nov 05 3378-3387 modified loss-compensation method, 0.35-ȝm SiGe BiCMOS technol., broad-band MMICs. Ming-Da Tsai, + , T-MTT Feb 05 496-505 preamplifier, cascode FET cells, improved noise anal. Won Ko, + , T-MTT Jan 05 361-371 Distributed feedback lasers RF frequency shifting via optically switched dual-channel PZT fiber stretchers. McDermitt, C.S., + , T-MTT Dec 05 3782-3787 Distributed feedback oscillators stabil. ccts. for phase-noise reduction, microwave oscillators. Suarez, A., + , T-MTT Sep 05 2743-2751 Distributed parameter circuits wide tuning-range planar filters, lumped-distrib. coupled resonators. Carey-Smith, B.E., + , T-MTT Feb 05 777-785 Distributed parameter filters; cf. Strip line filters Distributed parameter networks drain-line loss and the S22 kink effect in capacitively coupled distributed amplifiers. Shohat, J., + , T-MTT Dec 05 3767-3773 Distributed parameter networks; cf. Distributed amplifiers Diversity methods ultrawide-band photonic time-stretch a/D converter employing phase diversity. Han, Y., + , T-MTT Apr 05 1404-1408

IEEE T-MTT 2005 INDEX — 31 Doppler effect PLL-based retrodirective array, anal. and charactn. Fusco, V., + , T-MTT Feb 05 730-738 Doppler measurement; cf. Doppler radar Doppler radar 15th International Conference on Microwaves, Radar, and Wireless Communications, MIKON (special section). T-MTT Feb 05 425-450 15th International Conference on Microwaves, Radar, and Wireless Communications, MIKON (special section intro.). Modelski, J.W., TMTT Feb 05 425-426 inform., turbulence, rain by Doppler-polarimetric Radar, retrieval. Yanovsky, F.J., + , T-MTT Feb 05 444-450 Dosimetry SAR induced, 2 child head models and adult heads, mobile phones. Hadjem, A., + , T-MTT Jan 05 4-11 Driver circuits 40-Gb/s wide-band MMIC pHEMT modulator driver amps. designed, real freq. tech. Kerherve, E., + , T-MTT Jun 05 2145-2152 low-cost planar probes, broadside apertures for nondestructive dielec. meas. of biol. materials, microwave freqs. Byoungjoong Kang, + , TMTT Jan 05 134-143 E Eigenvalues and eigenfunctions coupled strip-slot guiding structs., full-wave anal. Deleniv, A.N., T-MTT Jun 05 1904-1912 differential surface admittance operator, skin effect modeling. De Zutter, D., + , T-MTT Aug 05 2526-2538 rect. waveguides, radially symmetric metal insets, fast S-domain modeling. Mira, F., + , T-MTT Apr 05 1294-1303 substr. integr. waveguide cavity filter, defected ground struct. Yu Lin Zhang, + , T-MTT Apr 05 1280-1287 transm. lines, guaranteed pass. direct lumped-element modeling. Se-Ho You, + , T-MTT Sep 05 2826-2834 wide-band finite-element model-order reduction, fast waveguide eigenanalysis. Shih-Hao Lee, + , T-MTT Aug 05 2552-2558 Electric actuators; cf. Microactuators Electrical contacts; cf. Ohmic contacts Electric current; cf. Current density; Current distribution Electric field measurement simultaneous elec. and mag. near-field meas., LiNbO3, inverted domain, EO probe. Suzuki, E., + , T-MTT Feb 05 696-701 Electric fields mode-stirrer configurations, dielec. heating perform., multimode microwave applicators, effect. Plaza-Gonzalez, P., + , T-MTT May 05 1699-1706 Electric resistance; cf. Contact resistance Electric variables control; cf. Gain control; Phase control; Power control; Voltage control Electric variables measurement; cf. Capacitance measurement; Electric field measurement; Frequency measurement; Phase measurement Electrodes freq. response of SAW filters, FDTD method. King-Yuen Wong, + , TMTT Nov 05 3364-3370 substr. removed LV high-speed GaAs/AlGaAs electrooptic modulators, Trail electrodes. JaeHyuk Shin, + , T-MTT Feb 05 636-643 Electromagnetic compatibility low-noise multiresolution high-dyn. ultra-broad-band time-domain EMI meas. syst. Braun, S., + , T-MTT Nov 05 3354-3363 Electromagnetic fields anal. of complex EM structs., TLM-G, grid-enabled time-domain TLM syst. Lorenz, P., + , T-MTT Nov 05 3631-3637 conductive medium, preiter. ADI-FDTD method. Shumin Wang, + , TMTT Jun 05 1913-1918 coupled-line models from EM simulators and appl., MoM anal., derivation. Farina, M., + , T-MTT Nov 05 3272-3280 dielec.-filled cavity filters, ultrawide stopband Characteristics, design. Rauscher, C., T-MTT May 05 1777-1786 double-delay and SOC EM deembedding, unification. Rautio, J.C., + , TMTT Sep 05 2892-2898 field-mapping syst., reson.-suppressed mag. field probe. Jung-Min Kim, + , T-MTT Sep 05 2693-2699 FVTD method exploiting, flux-splitting algm., field-based scatt.-matrix extr. scheme. Baumann, D., + , T-MTT Nov 05 3595-3605 + Check author entry for coauthors

leaky-wave propag., metamaterial grounded slabs excited by dipole source, effects. Baccarelli, P., + , T-MTT Jan 05 32-44 simultaneous elec. and mag. near-field meas., LiNbO3, inverted domain, EO probe. Suzuki, E., + , T-MTT Feb 05 696-701 Electromagnetic heating ablation, triaxial antenna. Brace, C.L., + , T-MTT Jan 05 215-220 FET self-heating, broad-band charactn. Parker, A.E., + , T-MTT Jul 05 2424-2429 mode-stirrer configurations, dielec. heating perform., multimode microwave applicators, effect. Plaza-Gonzalez, P., + , T-MTT May 05 1699-1706 Electromagnetic induction; cf. Inductance Electromagnetic interference low-noise multiresolution high-dyn. ultra-broad-band time-domain EMI meas. syst. Braun, S., + , T-MTT Nov 05 3354-3363 wideband suppression of ground bounce noise and radiated emission, high-speed ccts., EM bandgap power/ground planes. Tzong-Lin Wu, + , T-MTT Sep 05 2935-2942 Electromagnetic propagation 3D-connected/nonconnected wire metamaterials, homogenization. Silveirinha, M.G., + , T-MTT Apr 05 1418-1430 leaky-wave propag., metamaterial grounded slabs excited by dipole source, effects. Baccarelli, P., + , T-MTT Jan 05 32-44 low-refl. subgridding. Kulas, L., + , T-MTT May 05 1587-1592 metamaterial constructed by conductive SRRs and wires, MGS-based algm., propag. property anal. Hai-Ying Yao, + , T-MTT Apr 05 14691476 reson. Processes, metamaterials, FDTD study. Semouchkina, E.A., + , TMTT Apr 05 1477-1487 super-compact multilayered left-handed transm. line and diplexer appl. Horii, Y., + , T-MTT Apr 05 1527-1534 Electromagnetic propagation in absorbing media CPW, subterahertz freqs., atten. characts. Jingjing Zhang, + , T-MTT Nov 05 3281-3287 effectiveness of wave absorbers, improve DSRC EM environ., express highway. Pokharel, R.K., + , T-MTT Sep 05 2726-2731 SAR distribs., 3-layered bio-media, direct contact, water-loaded modified box-horn applicator. Gupta, R.C., + , T-MTT Sep 05 2665-2671 SAR induced, 2 child head models and adult heads, mobile phones. Hadjem, A., + , T-MTT Jan 05 4-11 Electromagnetic propagation in dispersive media 3D-connected/nonconnected wire metamaterials, homogenization. Silveirinha, M.G., + , T-MTT Apr 05 1418-1430 anisotropic media and their appl., left-handed metamaterials, refr. laws. Grzegorczyk, T.M., + , T-MTT Apr 05 1443-1450 hyperb. transm.-line periodic grids, neg. refr. and focusing. Eleftheriades, G.V., + , T-MTT Jan 05 396-403 loaded transm.-line neg.-refr.-index metamaterials, periodic FDTD anal. Kokkinos, T., + , T-MTT Apr 05 1488-1495 surface waves, metamaterial grounded slabs, fund. modal props. Baccarelli, P., + , T-MTT Apr 05 1431-1442 uniplanar left-handed metamaterials, efficient modeling. Yunchuan Guo, + , T-MTT Apr 05 1462-1468 Electromagnetic propagation in plasma media neg. permitt. and neg. permeab. by of evanescent waveguide Modestheory and expt., simul. Esteban, J., + , T-MTT Apr 05 1506-1514 Electromagnetic radiative interference interf. canceller for collocated radios, anal. and design. Raghavan, A., + , T-MTT Nov 05 3498-3508 RF front-end characts., spectral regrowth of communs. sigs., impact. Gard, K.G., + , T-MTT Jun 05 2179-2186 secure high-speed retrodirective commun. link. Goshi, D.S., + , T-MTT Nov 05 3548-3556 Electromagnetic refraction anisotropic media and their appl., left-handed metamaterials, refr. laws. Grzegorczyk, T.M., + , T-MTT Apr 05 1443-1450 left-handed metamaterials, props. Grzegorczyk, T.M., + , T-MTT Sep 05 2956-2967 Electromagnetic scattering anal. software for tunable microstrip filters, magnetized ferrites, expt. validation. Leon, G., + , T-MTT May 05 1739-1744 metamaterials, neg. refr. index, 2D shunt and 3D SCN TLM nets., modeling. So, P.P.M., + , T-MTT Apr 05 1496-1505 particle swarm optimizer for microwave imaging of 2D dielec. scatterers, comput. approach. Donelli, M., + , T-MTT May 05 1761-1776

IEEE T-MTT 2005 INDEX — 32 rect. waveguide, dielec.-filled corrugations supporting backward waves. Eshrah, I.A., + , T-MTT Nov 05 3298-3304 surface waves, metamaterial grounded slabs, fund. modal props. Baccarelli, P., + , T-MTT Apr 05 1431-1442 Electromagnetic shielding simple systematic spiral inductor design, perfected Q improv. for CMOS RFIC appl. Chih-Yuan Lee, + , T-MTT Feb 05 523-528 Electromagnetic surface waves waves, metamaterial grounded slabs, fund. modal props. Baccarelli, P., + , T-MTT Apr 05 1431-1442 Electromagnetism; cf. Maxwell equations Electromechanical filters; cf. Crystal filters Electron beams gyrotron phase-correcting mirrors, irradiance moments, synthesis. Shapiro, M.A., + , T-MTT Aug 05 2610-2615 Electron device noise; cf. Semiconductor device noise Electron device testing; cf. Semiconductor device testing Electronic engineering; cf. Cryogenic electronics Electronic engineering computing; cf. SPICE Electronic equipment testing measuring BER of UWB devices, prod. test tech. Bhattacharya, S., + , TMTT Nov 05 3474-3481 Electronic warfare advanced multifunction RF concept. Tavik, G.C., + , T-MTT Mar 05 10091020 Electron microscopy parallel coupled microstrip filters, floating ground-plane conductor for spurious-band suppression. Velazquez-Ahumada, Md.C., + , T-MTT May 05 1823-1828 Electron mobility 10-Gb/s driver amplifier using a tapered gate line for improved input matching. Shohat, J., + , T-MTT Oct 05 3115-3120 Electron wave tubes; cf. Traveling wave tubes Electro-optical devices; cf. Electro-optical modulation Electro-optical modulation coherent optical vector modulation for fiber radio using electrooptic microchip lasers. Li, Y., + , T-MTT Oct 05 3121-3129 Electrooptic devices simultaneous elec. and mag. near-field meas., LiNbO3, inverted domain, EO probe. Suzuki, E., + , T-MTT Feb 05 696-701 Electrooptic effects calibrated 200-GHz waveform meas. Williams, D.F., + , T-MTT Apr 05 1384-1389 Electrooptic materials/devices CPW, subterahertz freqs., atten. characts. Jingjing Zhang, + , T-MTT Nov 05 3281-3287 Electrooptic modulation substr. removed LV high-speed GaAs/AlGaAs electrooptic modulators, Trail electrodes. JaeHyuk Shin, + , T-MTT Feb 05 636-643 Electrostatic devices capacitive RF MEMS, reliab. modeling. Melle, S., + , T-MTT Nov 05 3482-3488 Electrostatic devices; cf. Capacitors Electrostatic discharges 1-, 10-GHz distrib. amp., CMOS technol., ESD protection design. MingDou Ker, + , T-MTT Sep 05 2672-2681 broad-band RF ccts., decreasing-size distrib. ESD protection scheme. Ming-Dou Ker, + , T-MTT Feb 05 582-589 InGaP/GaAs HBT RF power amps., ESD protection design considerations. Ma, Y., + , T-MTT Jan 05 221-228 packaged inductively degenerated common-source low-noise amps., ESD protection, anal. and optim. Sivonen, P., + , T-MTT Apr 05 1304-1313 Electrostatics; cf. Electric fields Elliptic filters cross coupling through higher/lower order modes and their applications in elliptic filter design. Amari, S., + , T-MTT Oct 05 3135-3141 dual-passband filters, design techs. Macchiarella, G., + , T-MTT Nov 05 3265-3271 E-plane filters and diplexers, ellipt. response for mm-wave appls. Ofli, E., + , T-MTT Mar 05 843-851 future mobile telecomm. systs., high-temp. supercond. filter. Jia-Sheng Hong, + , T-MTT Jun 05 1976-1981 inline filters with arbitrarily placed attenuation poles. Amari, S., + , TMTT Oct 05 3075-3081

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LTCC, metallic resonators, canonical ridge waveguide filters. Ruiz-Cruz, J.A., + , T-MTT Jan 05 174-182 supercond. spiral filters, quasiellipt. charact. for radio astron. Guoyong Zhang, + , T-MTT Mar 05 947-951 synthesis of cascaded N-tuplets cross-coupled resonators microwave filters, matrix rotations, anal. tech. Tamiazzo, S., + , T-MTT May 05 1693-1698 waveguide filters, multiple atten. poles, dual-behavior reson. of freq.selective surfaces. Ohira, M., + , T-MTT Nov 05 3320-3326 Emitter coupled logic 70-ps SiGe differential RF switch, design and anal. Hancock, T.M., + , TMTT Jul 05 2403-2410 Epitaxial layers; cf. Semiconductor epitaxial layers Equalizers 10-Gb/s fiber opt. commun. links, 0.18-ȝm CMOS equalization techs. Moonkyun Maeng, + , T-MTT Nov 05 3509-3519 20-Gb/s 4-PAM backplane serial I/O interconnections, equalization and NEXT noise cancellation. Hur, Y., + , T-MTT Jan 05 246-255 multigigahertz parallel bus, transmit preemphasis equalization, perform. anal. and model-to-hardware correl. Beyene, W.T., + , T-MTT Nov 05 3568-3577 Si Schottky diode DEMUX cct. for high bit-rate opt. receivers. Jung Han Choi, + , T-MTT Jun 05 2033-2042 Equations; cf. Integral equations; Nonlinear equations Equiripple filters; cf. Chebyshev filters; Elliptic filters Equivalent circuits anal. of radial waveguides, dielec. resonators, microstrip antennas, spherical multilayer structs., unified approach. Truong Vu Bang Giang, + , T-MTT Jan 05 404-409 broad-band suppression of TEM modes, power planes, EM-bandgap layers. Rogers, S.D., T-MTT Aug 05 2495-2505 calibrated 200-GHz waveform meas. Williams, D.F., + , T-MTT Apr 05 1384-1389 compact parallel-coupled microstrip bandpass filters, lumped-element Kinverters. Yo-Shen Lin, + , T-MTT Jul 05 2324-2328 dielec.-filled cavity filters, ultrawide stopband Characteristics, design. Rauscher, C., T-MTT May 05 1777-1786 efficient anal. formulation and sensitivity anal. of neuro-space mapping for nonlin. microwave device modeling. Lei Zhang, + , T-MTT Sep 05 2752-2767 high aspect ratio through-wafer interconnect vias, Si substrs., microwave charactn. and modeling. Leung, L.L.W., + , T-MTT Aug 05 2472-2480 high av.-effic. SiGe HBT power amp. for WCDMA handset appls. Junxiong Deng, + , T-MTT Feb 05 529-537 InP-InGaAs HBTs, microwave noise modeling. Escotte, L., + , T-MTT Jan 05 415-416 InP-InGaAs HBTs ), microwave noise modeling. Jianjun Gao, + , T-MTT Jan 05 417 large-sig. diode modeling, alternative param.-extr. tech. Yew Hui Liew, + , T-MTT Aug 05 2633-2638 low-Q microwave resonator, crit.-points method, accurate charactn. Peng Wang, + , T-MTT Jan 05 349-353 multiconductor transm. lines, nonlin. terminations, delay-extr.-based sensitivity anal. Nakhla, N.M., + , T-MTT Nov 05 3520-3530 PCB discontinuities, wavelet domain, 2-port equiv. Araneo, R., + , T-MTT Mar 05 907-918 periodically nonuniform coupled microstrip-line filters, harmonic suppression, transm. zero reallocation. Sheng Sun, + , T-MTT May 05 1817-1822 port discontinuities, full-wave CAD models of multiport ccts., deembedding. Farina, M., T-MTT May 05 1829 port discontinuities, full-wave CAD models of multiport ccts. ), deembedding. Okhmatovski, V.I., + , T-MTT May 05 1829 rect. waveguide, dielec.-filled corrugations supporting backward waves. Eshrah, I.A., + , T-MTT Nov 05 3298-3304 small-sig. modeling approach applied, GaN devices. Jarndal, A., + , TMTT Nov 05 3440-3448 split-ring resonators and complementary split-ring resonators coupled, planar transm. lines, equiv.-cct. models. Baena, J.D., + , T-MTT Apr 05 1451-1461 stopband-enhanced and size-miniaturized low-pass filters, high-impedance property of offset finite-ground microstrip line. Sheng Sun, + , T-MTT Sep 05 2844-2850 super-compact multilayered left-handed transm. line and diplexer appl. Horii, Y., + , T-MTT Apr 05 1527-1534

IEEE T-MTT 2005 INDEX — 33 transistor nonlinearities, noise props. Sungjae Lee, + , T-MTT Apr 05 1314-1321 transm. lines, guaranteed pass. direct lumped-element modeling. Se-Ho You, + , T-MTT Sep 05 2826-2834 waveguide filters, multiple atten. poles, dual-behavior reson. of freq.selective surfaces. Ohira, M., + , T-MTT Nov 05 3320-3326 Error analysis high-speed elec. backplane transm., duobinary signaling. Sinsky, J.H., + , T-MTT Jan 05 152-160 inter-chip RF-interconnect, CPW, capacitive coupler, UWB transceiver, perform. Sun, M., + , T-MTT Sep 05 2650-2655 measuring BER of UWB devices, prod. test tech. Bhattacharya, S., + , TMTT Nov 05 3474-3481 on-wafer meas. of differential ccts., mm-wave freqs., pure-mode NWA concept. Zwick, T., + , T-MTT Mar 05 934-937 partially leaky multiport vector net. analyzers, on-wafer calib. algm. Teppati, V., + , T-MTT Nov 05 3665-3671 planar microwave components, EM lin. regression models, empirical model gener. techs. Domenech-Asensi, G., + , T-MTT Nov 05 3305-3311 Error correction meas. of scatt. params. of N-ports, multiport method. Rolfes, I., + , T-MTT Jun 05 1990-1996 mismatch errors for 400-msamples/s 80-dB SFDR time-interleaved ADC, comprehensive digital correction. Munkyo Seo, + , T-MTT Mar 05 10721082 Errors; cf. Measurement errors Estimation theory; cf. Recursive estimation Etching; cf. Sputter etching Evolutionary computation; cf. Genetic algorithms

F Fading; cf. Fading channels Fading channels ultrawide-band photonic time-stretch a/D converter employing phase diversity. Han, Y., + , T-MTT Apr 05 1404-1408 Faraday effect ferrite Faraday rotators, impedance matching considerations. Boyd, C.R., Jr., T-MTT Jul 05 2371-2374 Fast Fourier transforms; cf. Discrete Fourier transforms Fault diagnosis microwave filters, adaptive models and param. extr., seq. tuning. Pepe, G., + , T-MTT Jan 05 22-31 RF ccts., LCP substrs., stat. anal. and diagnosis methodology. Mukherjee, S., + , T-MTT Nov 05 3621-3630 FDTD methods (2,4) stencil, optimized FDTD methods. Guilin Sun, + , T-MTT Mar 05 832-842 accurate waveguide port boundary condition for time-domain FEM. Zheng Lou, + , T-MTT Sep 05 3014-3023 conductive medium, preiter. ADI-FDTD method. Shumin Wang, + , TMTT Jun 05 1913-1918 coupled slotline mode, finite-ground CPW, unequal ground-plane widths, excit. Ponchak, G.E., + , T-MTT Feb 05 713-717 coupling 3D Maxwell's and Boltzmann's eqns. for analyzing, terahertz photoconductive switch. Sirbu, M., + , T-MTT Sep 05 2991-2998 designing microstrip filters utilizing mixed dielectrics, approaches. Semouchkina, E., + , T-MTT Feb 05 644-652 develop realistic num. models of cellular phones for accurate eval. of SAR distrib., human head, procedure. Pisa, S., + , T-MTT Apr 05 1256-1265 extracting causal time-domain params., iter. methods. Shuiping Luo, + , TMTT Mar 05 969-976 freq. response of SAW filters, FDTD method. King-Yuen Wong, + , TMTT Nov 05 3364-3370 general lin. lumped microwave ccts., matrix theory, improved FDTD formulation. Zhenhai Shao, + , T-MTT Jul 05 2261-2266 left-handed metamaterials, props. Grzegorczyk, T.M., + , T-MTT Sep 05 2956-2967 loaded transm.-line neg.-refr.-index metamaterials, periodic FDTD anal. Kokkinos, T., + , T-MTT Apr 05 1488-1495 low-refl. subgridding. Kulas, L., + , T-MTT May 05 1587-1592 method, off-grid perfect boundary conds. Rickard, Y.S., + , T-MTT Jul 05 2274-2283

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microstrip lines by virtual transm. line, FDTD, efficient excit. Karkkainen, M.K., T-MTT Jun 05 1899-1903 PCB discontinuities, wavelet domain, 2-port equiv. Araneo, R., + , T-MTT Mar 05 907-918 reduce num. dispers., ADI-FDTD, efficient method. Hong-Xing Zheng, + , T-MTT Jul 05 2295-2301 reson. Processes, metamaterials, FDTD study. Semouchkina, E.A., + , TMTT Apr 05 1477-1487 SAR induced, 2 child head models and adult heads, mobile phones. Hadjem, A., + , T-MTT Jan 05 4-11 self-collimation, low-index-contrast photonic crysts., mm-wave regime, expt. demons. Zhaolin Lu, + , T-MTT Apr 05 1362-1368 stabil. of BJT formulation, FDTD framework. Kung, F., + , T-MTT Apr 05 1189-1196 super-compact multilayered left-handed transm. line and diplexer appl. Horii, Y., + , T-MTT Apr 05 1527-1534 time reversal, FDTD method for microwave breast cancer detect. Kosmas, P., + , T-MTT Jul 05 2317-2323 Feedback circuits baseband-modeled CALLUM archits., spectrum emission considerations. Strandberg, R., + , T-MTT Feb 05 660-669 bifurcation control, practical cct. design. Collado, A., + , T-MTT Sep 05 2777-2788 Feedforward systems complex enveloped sigs. and appl., feedforward cct. anal., multitone model. Coskun, A.H., + , T-MTT Jun 05 2171-2178 Ferrimagnetic materials; cf. Ferrites; Garnets Ferrite circulators coupled-line circulator, reduced length. Meng Cao, + , T-MTT Aug 05 2572-2579 Ferrite devices; cf. Ferrite circulators; Ferrite phase shifters; Ferrite waveguides Ferrite phase shifters microwave phase shifter utilizing nonreciprocal wave propag. Davis, L.E., T-MTT Jan 05 414 microwave phase shifter utilizing nonreciprocal wave propag. ). How, H., + , T-MTT Jan 05 414 Ferrites anal. software for tunable microstrip filters, magnetized ferrites, expt. validation. Leon, G., + , T-MTT May 05 1739-1744 ferrite Faraday rotators, impedance matching considerations. Boyd, C.R., Jr., T-MTT Jul 05 2371-2374 mag. thin-film charactn., microwave range, broad-band method. Vincent, D., + , T-MTT Apr 05 1174-1180 Ferrite waveguides EM distribs. demonstrating asymmetry, spectral-domain dyadic Green's fn. for ferrite microstrip guided-wave structs. Krowne, C.M., T-MTT Apr 05 1345-1361 Ferroelectric devices ultrawide-band tunable true-time delay lines, ferroelec. varactors. Kuylenstierna, D., + , T-MTT Jun 05 2164-2170 Ferroelectric films measuring multilayered dielec. plates, open resonator tech. Deleniv, A.N., + , T-MTT Sep 05 2908-2916 thin-film materials, microwave charactn. Ouaddari, M., + , T-MTT Apr 05 1390-1397 Ferroelectric materials scan antennas, ferroelec. substr., rigorous anal. and investigs. Yashchyshyn, Y., + , T-MTT Feb 05 427-438 FET circuits linearizing FET low-noise amps., modified derivative superposition method. Aparin, V., + , T-MTT Feb 05 571-581 FETs AlGaN-GaN devices, thermal resist. calc. Wen-Yan Yin, T-MTT Sep 05 3051-3052 AlGaN-GaN devices'), 'Thermal resist. calc. Darwish, A.M., + , T-MTT Sep 05 3052-3053 distrib. preamplifier, cascode FET cells, improved noise anal. Won Ko, + , T-MTT Jan 05 361-371 RF low-noise bandpass filter, act. capacitance cct., design. Young-Hoon Chun, + , T-MTT Feb 05 687-695 thermal resist. of FETs, accurate determ. Darwish, A.M., + , T-MTT Jan 05 306-313 Fibers; cf. Optical fibers Field effect devices; cf. Field effect transistors

IEEE T-MTT 2005 INDEX — 34 Field effect transistor circuits; cf. MESFET circuits Field effect transistors relationships between common source, common gate, and common drain FETs. Gao, J., + , T-MTT Dec 05 3825-3831 Field effect transistors; cf. High electron mobility transistors Field programmable gate arrays hybrid digital/RF envelope predistortion linearization syst. for power amps. Wangmyong Woo, + , T-MTT Jan 05 229-237 single and multicarrier W-CDMA sigs., LINC digital component separator. Gerhard, W., + , T-MTT Jan 05 274-282 Field strength measurement; cf. Electric field measurement; Magnetic field measurement Films; cf. Thin films Filtering wireless transmitters, digital subband filtering predistorter archit. Hammi, O., + , T-MTT May 05 1643-1652 Filters H-plane contiguous manifold output multiplexers, fictitious reactive load concept, full-wave design. Montejo-Garai, J.R., + , T-MTT Aug 05 2628-2632 inline filters with arbitrarily placed attenuation poles. Amari, S., + , TMTT Oct 05 3075-3081 quasi-lumped suspended stripline filters and diplexers. Menzel, W., + , TMTT Oct 05 3230-3237 reactance of hollow, solid, and hemispherical-cap cylindrical posts in rectangular waveguide. Roelvink, J., + , T-MTT Oct 05 3156-3160 Filters; cf. Active filters; Adaptive filters; Band-pass filters; Band-stop filters; Butterworth filters; Digital filters; High-pass filters; Ladder filters; Lowpass filters; Nonlinear filters; Notch filters; Passive filters; Resonator filters; Waveguide filters Finite difference methods modeling microstrip structs., nonuniform grids and perfectly matched layer, compact 2D FDFD method. Jiunn-Nan Hwang, T-MTT Feb 05 653-659 rect. waveguide, dielec.-filled corrugations supporting backward waves. Eshrah, I.A., + , T-MTT Nov 05 3298-3304 substr. integr. waveguide cavity filter, defected ground struct. Yu Lin Zhang, + , T-MTT Apr 05 1280-1287 substr. integr. waveguide, guided-wave and leakage characts. Feng Xu, + , T-MTT Jan 05 66-73 Finite difference methods; cf. Finite difference time-domain analysis Finite difference time-domain analysis perfectly matched layer implementation using bilinear transforms. Dong, X., + , T-MTT Oct 05 3098-3105 Finite element analysis; cf. Mesh generation Finite element methods 1st. circ. conds. of turnstile waveguide circulators, finite-element solver, verification. Helszajn, J., + , T-MTT Jul 05 2309-2316 accurate waveguide port boundary condition for time-domain FEM. Zheng Lou, + , T-MTT Sep 05 3014-3023 arbitrary waveguides, p-refinement, gen. quadrilaterals, efficient largedomain 2D FEM soln. Ilic, M.M., + , T-MTT Apr 05 1377-1383 complex permitt. of arbitrary shape and size dielec. samples, cavity meas. tech., microwave freqs., estim. Santra, M., + , T-MTT Feb 05 718-722 cylindrical/spherical dielec. resonators, cavities and MIC environments by of finite elements, CAD-oriented anal. Gil, J.M., T-MTT Sep 05 28662874 FEM solns. of dielec. waveguiding structs., 2D curl-conforming sing. elements. Din-Kow Sun, + , T-MTT Mar 05 984-992 IMD, contact-type MEMS microswitch, determ. Johnson, J., + , T-MTT Nov 05 3615-3620 substr. integr. waveguide, guided-wave and leakage characts. Feng Xu, + , T-MTT Jan 05 66-73 substr. removed LV high-speed GaAs/AlGaAs electrooptic modulators, Trail electrodes. JaeHyuk Shin, + , T-MTT Feb 05 636-643 super-compact multilayered left-handed transm. line and diplexer appl. Horii, Y., + , T-MTT Apr 05 1527-1534 transm. lines, guaranteed pass. direct lumped-element modeling. Se-Ho You, + , T-MTT Sep 05 2826-2834 wide-band finite-element model-order reduction, fast waveguide eigenanalysis. Shih-Hao Lee, + , T-MTT Aug 05 2552-2558 Finite element time-domain analysis complex-permittivity measurement on high-Q materials via combined numerical approaches. Fan, X.C., + , T-MTT Oct 05 3130-3134

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FIR digital filters high-speed elec. backplane transm., duobinary signaling. Sinsky, J.H., + , T-MTT Jan 05 152-160 mismatch errors for 400-msamples/s 80-dB SFDR time-interleaved ADC, comprehensive digital correction. Munkyo Seo, + , T-MTT Mar 05 10721082 Flip-chip devices nondestructive in situ S-param. meas. of hermetically encapsulated packages, recursive un-termination method. Pfeiffer, U.R., + , T-MTT Jun 05 1845-1855 Flip-flops high-sensitivity InP/InGaAs DHBT decision cct.-design and appl., opt. and syst. expts., 40-43 Gbit/s. Konczykowska, A., + , T-MTT Apr 05 1228-1234 Fourier transforms CPW, subterahertz freqs., atten. characts. Jingjing Zhang, + , T-MTT Nov 05 3281-3287 extracting causal time-domain params., iter. methods. Shuiping Luo, + , TMTT Mar 05 969-976 Fractals suppression of second harmonic, fractal-shaped microstrip coupled-line bandpass filters. Il Kwon Kim, + , T-MTT Sep 05 2943-2948 Frequency SAW filters, FDTD method. King-Yuen Wong, + , T-MTT Nov 05 33643370 Frequency allocation corrections on "Precision open-ended coaxial probes for In Vivo and Ex Vivo dielectric spectroscopy of biological tissues on microwave frequencies" (May 05 1713-1722). Popovic, D., + , T-MTT Sep 05 3053 Frequency control tunable combline filter, continuous control of center freq. and bandwidth. Sanchez-Renedo, M., + , T-MTT Jan 05 191-199 Frequency conversion 20-Gb/s 4-PAM backplane serial I/O interconnections, equalization and NEXT noise cancellation. Hur, Y., + , T-MTT Jan 05 246-255 540-640-GHz high-effic. 4-anode freq. tripler. Maestrini, A., + , T-MTT Sep 05 2835-2843 bifurcation control, practical cct. design. Collado, A., + , T-MTT Sep 05 2777-2788 dual-band transmitters, digitally predistorted freq. multipliers for reconfigurable radios. Youngcheol Park, + , T-MTT Jan 05 115-122 high-power high-effic. SiGe Ku- and Ka-band balanced freq. doublers. Juo-Jung Hung, + , T-MTT Feb 05 754-761 InP DHBT technol., DC-100-GHz freq. doublers. Puyal, V., + , T-MTT Apr 05 1338-1344 k- and Q-bands CMOS freq. sources, X-band quadrature VCO. Sangsoo Ko, + , T-MTT Sep 05 2789-2800 left-handed transm.-line media, nonlin. wave propag. phenom. Kozyrev, A.B., + , T-MTT Jan 05 238-245 low-IF receivers, image-rejection down-converter. Sher Jiun Fang, + , TMTT Feb 05 478-487 multiplier, waveguide-based spatial power-combining archit. Belaid, M., + , T-MTT Apr 05 1124-1129 v-band high-order harmonic injection-locked freq.-divider MMICs, wide bandwidth and low-power dissipation. Jinho Jeong, + , T-MTT Jun 05 1891-1898 Frequency converters; cf. Microwave frequency converters Frequency division multiplexing; cf. OFDM modulation Frequency domain analysis broad-band poly-harmonic distortion (PHD) behavioral models from fast automated simul. and large-sig. vectorial net. meas. Root, D.E., + , TMTT Nov 05 3656-3664 Chebyshev collocation and Newton-type optim. methods for inverse problem, nonuniform transm. lines. Norgren, M., T-MTT May 05 15611568 comput. time-domain sensitivity of multiport systs. described by reducedorder models, adjoint-based approach. Ahmed, T., + , T-MTT Nov 05 3538-3547 CPW, subterahertz freqs., atten. characts. Jingjing Zhang, + , T-MTT Nov 05 3281-3287 cross coupling through higher/lower order modes and their applications in elliptic filter design. Amari, S., + , T-MTT Oct 05 3135-3141 extracting causal time-domain params., iter. methods. Shuiping Luo, + , TMTT Mar 05 969-976

IEEE T-MTT 2005 INDEX — 35 FET self-heating, broad-band charactn. Parker, A.E., + , T-MTT Jul 05 2424-2429 lin. inverse space-mapping (LISM) algm., design lin./nonlin. RF and microwave ccts. Rayas-Sanchez, J.E., + , T-MTT Mar 05 960-968 low-refl. subgridding. Kulas, L., + , T-MTT May 05 1587-1592 modeling microstrip structs., nonuniform grids and perfectly matched layer, compact 2D FDFD method. Jiunn-Nan Hwang, T-MTT Feb 05 653-659 RFICs, 2-port lumped-nonlin.-source model, systematic linearity anal. Qingqing Liang, + , T-MTT May 05 1745-1755 substr. integr. waveguide cavity filter, defected ground struct. Yu Lin Zhang, + , T-MTT Apr 05 1280-1287 Frequency measurement 2004 IEEE Radio Frequency Integrated Circuits Symposium, RFIC (special section). T-MTT Feb 05 451-626 2004 IEEE Radio Frequency Integrated Circuits Symposium, RFIC (special section intro.). Quach, T., T-MTT Feb 05 451-452 Frequency modulation; cf. Frequency shift keying Frequency response high-speed elec. backplane transm., duobinary signaling. Sinsky, J.H., + , T-MTT Jan 05 152-160 improved coupled-microstrip filter design, effective even-mode/odd-mode charact. impedances. Hong-Ming Lee, + , T-MTT Sep 05 2812-2818 miniaturized microwave pass. filter incorporating multilayer synthetic quasiTEM transm. line. Hsien-Shun Wu, + , T-MTT Sep 05 2713-2720 miniaturized spurious passband suppression microstrip filter, meandered parallel coupled lines. Shih-Ming Wang, + , T-MTT Feb 05 747-753 nonuniform plasma layer model for quartz-Si image guide phase shifters, expt. verification. Fickenscher, T., + , T-MTT Jul 05 2375-2382 waveguide polarizers, design tool. Virone, G., + , T-MTT Mar 05 888-894 wide-band SiGe HBT act. mixers, anal. and design. Johansen, T.K., + , TMTT Jul 05 2389-2397 Frequency selective surfaces waveguide filters, multiple atten. poles, dual-behavior reson. of freq.selective surfaces. Ohira, M., + , T-MTT Nov 05 3320-3326 Frequency shift keying 2.4-GHz-band GFSK appls., low-power highly digitized receiver. Bergveld, H.J., + , T-MTT Feb 05 453-461 Frequency synthesizers 1/f noise and oscillator phase noise, SiGe HBTs, scaling and technol. limitations. Guofu Niu, + , T-MTT Feb 05 506-514 frequency planning and synthesizer architectures for multiband OFDM UWB radios. Mishra, C., + , T-MTT Dec 05 3744-3756 multistandard WLAN appls., dual-band RF transceiver. Chang, S.-F.R., + , T-MTT Mar 05 1048-1055 single and multicarrier W-CDMA sigs., LINC digital component separator. Gerhard, W., + , T-MTT Jan 05 274-282 Functional analysis; cf. Harmonic analysis Function approximation; cf. Chebyshev approximation Functions; cf. Bessel functions; Transfer functions; Transforms G Gain control design of 10-Gb/s AGC amp., jitter considerations. Kucharski, D., + , TMTT Feb 05 590-597 submillimeter SIS heterodyne receiver, gain stabil. Battat, J., + , T-MTT Jan 05 389-395 Galerkin method anal. of NRD-guide/H-guide mm-wave ccts., order-reduced vol.-integral eqn. approach. Duochuan Li, + , T-MTT Mar 05 799-812 arbitrary waveguides, p-refinement, gen. quadrilaterals, efficient largedomain 2D FEM soln. Ilic, M.M., + , T-MTT Apr 05 1377-1383 Gallium compounds 40-Gb/s wide-band MMIC pHEMT modulator driver amps. designed, real freq. tech. Kerherve, E., + , T-MTT Jun 05 2145-2152 60-GHz monolithic star mixer, gate-drain-connected pHEMT diodes. Kyung-Whan Yeom, + , T-MTT Jul 05 2435-2440 act. predistorter suitable for MMIC implement. Iommi, R., + , T-MTT Mar 05 874-880 AlGaN-GaN devices, thermal resist. calc. Wen-Yan Yin, T-MTT Sep 05 3051-3052 AlGaN-GaN devices'), 'Thermal resist. calc. Darwish, A.M., + , T-MTT Sep 05 3052-3053

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distrib. act. transformer, optimized design. Seungwoo Kim, + , T-MTT Jan 05 380-388 effic. of OFDM transmitters, RF/DSP design. Helaoui, M., + , T-MTT Jul 05 2355-2361 epitaxial struct., noise fig. of AlGaN/GaN HEMTs. Sanabria, C., + , TMTT Feb 05 762-769 GaInP-GaAs HBT for accurate predict. of phase noise, oscillators, advanced LF noise model. Nallatamby, J.-C., + , T-MTT May 05 16011612 high-effic. current-mode class-D amps. for wireless handsets, design. TsaiPi Hung, + , T-MTT Jan 05 144-151 hybrid digital/RF envelope predistortion linearization syst. for power amps. Wangmyong Woo, + , T-MTT Jan 05 229-237 InGaP/GaAs HBT RF power amps., ESD protection design considerations. Ma, Y., + , T-MTT Jan 05 221-228 InP HBT noise params., noise-fig. meas. syst., direct extr. Jianjun Gao, + , T-MTT Jan 05 330-335 InP-InGaAs HBTs, microwave noise modeling. Escotte, L., + , T-MTT Jan 05 415-416 InP-InGaAs HBTs ), microwave noise modeling. Jianjun Gao, + , T-MTT Jan 05 417 microstrip-line struct. employing, periodically perforated ground metal and appl., highly miniaturized and low-impedance pass. components fabricated. Young Yun, T-MTT Jun 05 1951-1959 opt. sigs., p-i-n photodiodes, freq. conversion. Malyshev, S.A., + , T-MTT Feb 05 439-443 photoelectronic ADC. Ioakeimidi, K., + , T-MTT Jan 05 336-342 short-channel AlGaN/GaN heterojunction FETs, 30-GHz-band, 5-W power perform. Inoue, T., + , T-MTT Jan 05 74-80 sig. gener., control, freq. conversion AlGaN/GaN HEMT MMICs. Kaper, V.S., + , T-MTT Jan 05 55-65 substr. removed LV high-speed GaAs/AlGaAs electrooptic modulators, Trail electrodes. JaeHyuk Shin, + , T-MTT Feb 05 636-643 thermal resist. of FETs, accurate determ. Darwish, A.M., + , T-MTT Jan 05 306-313 transistor nonlinearities, noise props. Sungjae Lee, + , T-MTT Apr 05 1314-1321 TW photodetectors, hybrid drift-diffusion-TLM anal. Pasalic, D., + , TMTT Sep 05 2700-2706 v-band high-order harmonic injection-locked freq.-divider MMICs, wide bandwidth and low-power dissipation. Jinho Jeong, + , T-MTT Jun 05 1891-1898 wide-band on-wafer noise-param. meas., improved Y-factor method. Tiemeijer, L.F., + , T-MTT Sep 05 2917-2925 Gallium materials/devices deembedding static nonlinearities and accurately identifying and modeling memory effects, wide-band RF transmitters. Taijun Liu, + , T-MTT Nov 05 3578-3587 InP/InGaAs DHBT distrib. amps., modulator drivers for 80-Gbit/s operation, comp. Schneider, K., + , T-MTT Nov 05 3378-3387 presence of LF dispers. effects, accurate pHEMT nonlin. modeling. Raffo, A., + , T-MTT Nov 05 3449-3459 RF-driven gate current, DC/RF perform., GaAs pHEMT MMIC power amps., effect. Yeong-Chang Chou, + , T-MTT Nov 05 3398-3406 small-sig. modeling approach applied, GaN devices. Jarndal, A., + , TMTT Nov 05 3440-3448 Garnets mm-wave appls., extremely high-Q factor dielec. resonators. Krupka, J., + , T-MTT Feb 05 702-712 Gaussian channels 2.4-GHz-band GFSK appls., low-power highly digitized receiver. Bergveld, H.J., + , T-MTT Feb 05 453-461 Gaussian processes; cf. Gaussian channels Genetic algorithms comparison of two optimiz. techniques for the est. of complex permittivities of multilayered struc. using waveguide meas. Baginski, M.E., + , T-MTT Oct 05 3251-3259 intelligently controlled RF power amp., reconfigurable MEMS-varactor tuner. Dongjiang Qiao, + , T-MTT Mar 05 1089-1095 nonintuitive planar structs., fast optim. and sensitivity anal. Cormos, D., + , T-MTT Jun 05 2019-2025 waveguide filters, multiple atten. poles, dual-behavior reson. of freq.selective surfaces. Ohira, M., + , T-MTT Nov 05 3320-3326 Geometrical optics; cf. Ray tracing

IEEE T-MTT 2005 INDEX — 36 Germanium alloys 12-GHz SiGe phase shifter, integr. LNA. Hancock, T.M., + , T-MTT Mar 05 977-983 1/f noise and oscillator phase noise, SiGe HBTs, scaling and technol. limitations. Guofu Niu, + , T-MTT Feb 05 506-514 24-GHz SiGe phased-array receiver-LO phase-shifting approach. Hashemi, H., + , T-MTT Feb 05 614-626 70-ps SiGe differential RF switch, design and anal. Hancock, T.M., + , TMTT Jul 05 2403-2410 act. differential broad-band phase splitter for quadrature-modulator appls. Tiiliharju, E., + , T-MTT Feb 05 679-686 compact intell. RF front-end, reconfigurable RFICs, Si-based technols. Mukhopadhyay, R., + , T-MTT Jan 05 81-93 design of 10-Gb/s AGC amp., jitter considerations. Kucharski, D., + , TMTT Feb 05 590-597 high av.-effic. SiGe HBT power amp. for WCDMA handset appls. Junxiong Deng, + , T-MTT Feb 05 529-537 high-power high-effic. SiGe Ku- and Ka-band balanced freq. doublers. Juo-Jung Hung, + , T-MTT Feb 05 754-761 low-power ka-band Voltage-controlled oscillator implemented, 200-GHz SiGe HBT technol. Yi-jan Emery Chen, + , T-MTT May 05 1672-1681 mm-wave (Bi)CMOS IC, 30-100-GHz inductors and transformers. Dickson, T.O., + , T-MTT Jan 05 123-133 modified loss-compensation method, 0.35-ȝm SiGe BiCMOS technol., broad-band MMICs. Ming-Da Tsai, + , T-MTT Feb 05 496-505 RF-CMOS and SiGe BiCMOS, WCDMA direct-conversion receiver front-end comp. Floyd, B.A., + , T-MTT Apr 05 1181-1188 RFICs, 2-port lumped-nonlin.-source model, systematic linearity anal. Qingqing Liang, + , T-MTT May 05 1745-1755 SiGe HBTs, small-sig. and HF noise modeling. Basaran, U., + , T-MTT Mar 05 919-928 wide-band SiGe HBT act. mixers, anal. and design. Johansen, T.K., + , TMTT Jul 05 2389-2397 Global positioning system multifrequency RF front ends using direct RF sampling. Psiaki, M.L., + , T-MTT Oct 05 3082-3089 Granular materials; cf. Powders Green’s function method 2D nonuniform FFT (2-D NUFFT) tech., anal. of shielded microstrip ccts. Ke-Ying Su, + , T-MTT Mar 05 993-999 anal. of NRD-guide/H-guide mm-wave ccts., order-reduced vol.-integral eqn. approach. Duochuan Li, + , T-MTT Mar 05 799-812 anal. of radial waveguides, dielec. resonators, microstrip antennas, spherical multilayer structs., unified approach. Truong Vu Bang Giang, + , T-MTT Jan 05 404-409 EM distribs. demonstrating asymmetry, spectral-domain dyadic Green's fn. for ferrite microstrip guided-wave structs. Krowne, C.M., T-MTT Apr 05 1345-1361 fns. for cylindrical enclosures by spatial images method, num. eval. Pereira, F.D.Q., + , T-MTT Jan 05 94-105 gen. surface-vol. integral-eqn. (SVIE) approach for anal. of hybrid planar/NRD-guide IC. Duochuan Li, + , T-MTT Sep 05 2732-2742 input impedance of top-loaded monopole, parallel-plate waveguide by MoM/Green's fn. method. Valero-Nogueira, A., + , T-MTT Mar 05 868873 left-handed metamaterials, props. Grzegorczyk, T.M., + , T-MTT Sep 05 2956-2967 on-chip interconnects by double-image Green's fn. method combined, hierarchical algm., param. extr. Wenliang Dai, + , T-MTT Jul 05 24162423 sub-threshold anal. and drain current modeling of polysilicon TFT, Green's fn. approach. Sehgal, A., + , T-MTT Sep 05 2682-2687 TE/TM propag. and losses of integr. opt. polarizer, accurate modeling. Pierantoni, L., + , T-MTT Jun 05 1856-1862 Ground support systems ground on power-line communications. Luo, W.Q., + , T-MTT Oct 05 3191-3198 Gyrotrons CW 250-GHz gyrotron DNP expts., corrugated waveguide and directional coupler. Woskov, P.P., + , T-MTT Jun 05 1863-1869 gyrotron phase-correcting mirrors, irradiance moments, synthesis. Shapiro, M.A., + , T-MTT Aug 05 2610-2615 high-performance circular TE01-mode converter. Yu, C.-F., + , T-MTT Dec 05 3794-3798 + Check author entry for coauthors

TE0n-TE0(n+1) ripple-wall mode converters, circ. waveguide, bandwidth studies. Lawson, W., + , T-MTT Jan 05 372-379 H Harmonic analysis BJT class-F power amp. near transit. freq. Rudiakova, A.N., T-MTT Sep 05 3045-3050 fully integrated transmitter front-end with high power-added efficiency. Kim, H., + , T-MTT Oct 05 3206-3214 Harmonic distortion broad-band poly-harmonic distortion (PHD) behavioral models from fast automated simul. and large-sig. vectorial net. meas. Root, D.E., + , TMTT Nov 05 3656-3664 coupled-slotline-hybrid sampling mixer integr., step-recovery-diode pulse generator for UWB appls. Jeongwoo Han, + , T-MTT Jun 05 1875-1882 Helical waveguides dielec. helical resonators. Holmes, J.E., + , T-MTT Jan 05 322-329 Helmholtz equations determining TE and TM modes, closed waveguides made up of N cylindrical conductors, efficient method. de la Rubia, V., + , T-MTT Feb 05 670-678 HEMT integrate circuits 10-Gb/s driver amplifier using a tapered gate line for improved input matching. Shohat, J., + , T-MTT Oct 05 3115-3120 Heterojunction bipolar transistors 1/f noise and oscillator phase noise, SiGe HBTs, scaling and technol. limitations. Guofu Niu, + , T-MTT Feb 05 506-514 broad-band power amp., tunable output matching net. Haitao Zhang, + , T-MTT Nov 05 3606-3614 class-F and inverse class-F HBT amps., IMD anal. Ohta, A., + , T-MTT Jun 05 2121-2128 efficient anal. formulation and sensitivity anal. of neuro-space mapping for nonlin. microwave device modeling. Lei Zhang, + , T-MTT Sep 05 2752-2767 effic. of OFDM transmitters, RF/DSP design. Helaoui, M., + , T-MTT Jul 05 2355-2361 enhanced high-current VBIC model. Ce-Jun Wei, + , T-MTT Apr 05 12351243 GaInP-GaAs HBT for accurate predict. of phase noise, oscillators, advanced LF noise model. Nallatamby, J.-C., + , T-MTT May 05 16011612 g-band (140-220 GHz) and W-band (75-110 GHz) InP DHBT medium power amps. Paidi, V.K., + , T-MTT Feb 05 598-605 high av.-effic. SiGe HBT power amp. for WCDMA handset appls. Junxiong Deng, + , T-MTT Feb 05 529-537 high-effic. current-mode class-D amps. for wireless handsets, design. TsaiPi Hung, + , T-MTT Jan 05 144-151 high-sensitivity InP/InGaAs DHBT decision cct.-design and appl., opt. and syst. expts., 40-43 Gbit/s. Konczykowska, A., + , T-MTT Apr 05 1228-1234 InGaP/GaAs HBT RF power amps., ESD protection design considerations. Ma, Y., + , T-MTT Jan 05 221-228 InP DHBT technol., DC-100-GHz freq. doublers. Puyal, V., + , T-MTT Apr 05 1338-1344 InP HBT noise params., noise-fig. meas. syst., direct extr. Jianjun Gao, + , T-MTT Jan 05 330-335 InP/InGaAs DHBT distrib. amps., modulator drivers for 80-Gbit/s operation, comp. Schneider, K., + , T-MTT Nov 05 3378-3387 InP-InGaAs HBTs, microwave noise modeling. Escotte, L., + , T-MTT Jan 05 415-416 InP-InGaAs HBTs ), microwave noise modeling. Jianjun Gao, + , T-MTT Jan 05 417 low-power ka-band Voltage-controlled oscillator implemented, 200-GHz SiGe HBT technol. Yi-jan Emery Chen, + , T-MTT May 05 1672-1681 modified loss-compensation method, 0.35-ȝm SiGe BiCMOS technol., broad-band MMICs. Ming-Da Tsai, + , T-MTT Feb 05 496-505 RFICs, 2-port lumped-nonlin.-source model, systematic linearity anal. Qingqing Liang, + , T-MTT May 05 1745-1755 SiGe HBTs, small-sig. and HF noise modeling. Basaran, U., + , T-MTT Mar 05 919-928 wide-band SiGe HBT act. mixers, anal. and design. Johansen, T.K., + , TMTT Jul 05 2389-2397

IEEE T-MTT 2005 INDEX — 37 x-band 2-stage high-effic. switched-mode power amps. Pajic, S., + , TMTT Sep 05 2899-2907 HF transformers 0.6-V 1.6-mW transformer-based 2.5-GHz downconversion mixer, +5.4dB gain and -2.8-dBm IIP3, 0.13-ȝm CMOS. Hermann, C., + , T-MTT Feb 05 488-495 High electron mobility transistors drain-line loss and the S22 kink effect in capacitively coupled distributed amplifiers. Shohat, J., + , T-MTT Dec 05 3767-3773 relationships between common source, common gate, and common drain FETs. Gao, J., + , T-MTT Dec 05 3825-3831 Higher order statistics nonlin. model testing, designing multisine excit. Pedro, J.C., + , T-MTT Jan 05 45-54 High-frequency effects; cf. Skin effect High-pass filters 20-Gb/s 4-PAM backplane serial I/O interconnections, equalization and NEXT noise cancellation. Hur, Y., + , T-MTT Jan 05 246-255 miniaturized planar Marchand baluns, classes. Fathelbab, W.M., + , TMTT Apr 05 1211-1220 parallel-coupled line filters with enhanced stopband performances. Fathelbab, W.M., + , T-MTT Dec 05 3774-3781 semilumped CPW elements for Millimeter-wave filter design, charactn. Aryanfar, F., + , T-MTT Apr 05 1288-1293 High-temperature superconductors future mobile telecomm. systs., high-temp. supercond. filter. Jia-Sheng Hong, + , T-MTT Jun 05 1976-1981 wide-band supercond. coplanar delay lines. Yi Wang, + , T-MTT Jul 05 2348-2354 Horn antennas SAR distribs., 3-layered bio-media, direct contact, water-loaded modified box-horn applicator. Gupta, R.C., + , T-MTT Sep 05 2665-2671 Hot carriers carrier heating, channel noise, deep-submicrometer NMOSFETs via body bias, expt. study. Hong Wang, + , T-MTT Feb 05 564-570 Hydrologic measurements inform., turbulence, rain by Doppler-polarimetric Radar, retrieval. Yanovsky, F.J., + , T-MTT Feb 05 444-450 Hyperthermia SAR distribs., 3-layered bio-media, direct contact, water-loaded modified box-horn applicator. Gupta, R.C., + , T-MTT Sep 05 2665-2671 Hysteresis bifurcation control, practical cct. design. Collado, A., + , T-MTT Sep 05 2777-2788 global stability analysis and stabilization of a class-E/F amplifier with a distributed active transformer. Jeon, S., + , T-MTT Dec 05 3712-3722 I Identification; cf. Parameter estimation IEEE standards effic. of OFDM transmitters, RF/DSP design. Helaoui, M., + , T-MTT Jul 05 2355-2361 frequency channel blocking scheme in mesh-topology millimeter-wave broad band entrance networks. Sangiamwong, J., + , T-MTT Dec 05 3723-3730 IF amplifiers image-rejection CMOS LNA design optim. techs. Trung-Kien Nguyen, + , T-MTT Feb 05 538-547 submillimeter SIS heterodyne receiver, gain stabil. Battat, J., + , T-MTT Jan 05 389-395 Image denoising self-calibrating SSB modulator. Treyer, D.M., + , T-MTT Dec 05 38063816 Image processing; cf. Image denoising; Image reconstruction Image reconstruction high dielec.-contrast objs., different image-reconstruction approaches, microwave-tomographic imaging. Semenov, S.Y., + , T-MTT Jul 05 2284-2294 vector NWA for microwave imaging, effective usage. Chao-Hsiung Tseng, + , T-MTT Sep 05 2884-2891 Imaging; cf. Biomedical imaging; Microwave imaging; Millimeter wave imaging

+ Check author entry for coauthors

Immittance converters broad-band 180° phase shifters, integr. submillimeter-wave Schottky diodes. Zhiyang Liu, + , T-MTT Sep 05 2949-2955 distrib. act. transformer, optimized design. Seungwoo Kim, + , T-MTT Jan 05 380-388 tunable impedance transformer, transm. line, variable charact. impedance. Hyeong Tae Jeong, + , T-MTT Aug 05 2587-2593 IMPATT diodes std. CMOS technol., monolithic integr. mm-wave IMPATT transmitter. Al-Attar, T., + , T-MTT Nov 05 3557-3561 Impedance coupled strip-slot guiding structs., full-wave anal. Deleniv, A.N., T-MTT Jun 05 1904-1912 high-power MEMS varactors and impedance tuners for mm-wave appls. Lu, Y., + , T-MTT Nov 05 3672-3678 input impedance of top-loaded monopole, parallel-plate waveguide by MoM/Green's fn. method. Valero-Nogueira, A., + , T-MTT Mar 05 868873 microstrip-line struct. employing, periodically perforated ground metal and appl., highly miniaturized and low-impedance pass. components fabricated. Young Yun, T-MTT Jun 05 1951-1959 propag. characts. of cylindrical CPW, finite thickness of conductor, fullwave anal. Yamamoto, H., + , T-MTT Jun 05 2187-2195 Impedance converters global stability analysis and stabilization of a class-E/F amplifier with a distributed active transformer. Jeon, S., + , T-MTT Dec 05 3712-3722 Impedance matching 12-GHz SiGe phase shifter, integr. LNA. Hancock, T.M., + , T-MTT Mar 05 977-983 2.17-dB NF 5-GHz-band monolithic CMOS LNA, 10-mW DC power consumption. Hung-Wei Chiu, + , T-MTT Mar 05 813-824 3-sect. transm.-line transformer, anal. and design. Chongcheawchamnan, M., + , T-MTT Jul 05 2458-2462 broad-band RF ccts., decreasing-size distrib. ESD protection scheme. Ming-Dou Ker, + , T-MTT Feb 05 582-589 complex impedances, RFID tag design, power refl. coeff. anal. Nikitin, P.V., + , T-MTT Sep 05 2721-2725 ferrite coupled-line circulator, reduced length. Meng Cao, + , T-MTT Aug 05 2572-2579 ferrite Faraday rotators, impedance matching considerations. Boyd, C.R., Jr., T-MTT Jul 05 2371-2374 LINC radio transmitters, integr. antenna/power combiner. Gao, S., + , TMTT Mar 05 1083-1088 metamaterial-based electronically controlled transm.-line struct., leakywave antenna, tunable radiation angle and beamwidth. Sungjoon Lim, + , T-MTT Jan 05 161-173 packaged inductively degenerated common-source low-noise amps., ESD protection, anal. and optim. Sivonen, P., + , T-MTT Apr 05 1304-1313 std. CMOS technol., monolithic integr. mm-wave IMPATT transmitter. Al-Attar, T., + , T-MTT Nov 05 3557-3561 tunable impedance transformer, transm. line, variable charact. impedance. Hyeong Tae Jeong, + , T-MTT Aug 05 2587-2593 unbiased integr. microstrip circulator, mag. nanowired substr. Saib, A., + , T-MTT Jun 05 2043-2049 Indium compounds effic. of OFDM transmitters, RF/DSP design. Helaoui, M., + , T-MTT Jul 05 2355-2361 GaInP-GaAs HBT for accurate predict. of phase noise, oscillators, advanced LF noise model. Nallatamby, J.-C., + , T-MTT May 05 16011612 g-band (140-220 GHz) and W-band (75-110 GHz) InP DHBT medium power amps. Paidi, V.K., + , T-MTT Feb 05 598-605 InGaP/GaAs HBT RF power amps., ESD protection design considerations. Ma, Y., + , T-MTT Jan 05 221-228 InP DHBT technol., DC-100-GHz freq. doublers. Puyal, V., + , T-MTT Apr 05 1338-1344 InP HBT noise params., noise-fig. meas. syst., direct extr. Jianjun Gao, + , T-MTT Jan 05 330-335 InP HEMTs and their appls., mm-wave radio-on-fiber systs., phototransistors. Chang-Soon Choi, + , T-MTT Jan 05 256-263 InP/InGaAs DHBT distrib. amps., modulator drivers for 80-Gbit/s operation, comp. Schneider, K., + , T-MTT Nov 05 3378-3387 InP-InGaAs HBTs, microwave noise modeling. Escotte, L., + , T-MTT Jan 05 415-416

IEEE T-MTT 2005 INDEX — 38 InP-InGaAs HBTs ), microwave noise modeling. Jianjun Gao, + , T-MTT Jan 05 417 opt. sigs., p-i-n photodiodes, freq. conversion. Malyshev, S.A., + , T-MTT Feb 05 439-443 Inductance high av.-effic. SiGe HBT power amp. for WCDMA handset appls. Junxiong Deng, + , T-MTT Feb 05 529-537 rect. waveguide, dielec.-filled corrugations supporting backward waves. Eshrah, I.A., + , T-MTT Nov 05 3298-3304 Inductors high-effic. current-mode class-D amps. for wireless handsets, design. TsaiPi Hung, + , T-MTT Jan 05 144-151 high-power high-effic. SiGe Ku- and Ka-band balanced freq. doublers. Juo-Jung Hung, + , T-MTT Feb 05 754-761 mm-wave (Bi)CMOS IC, 30-100-GHz inductors and transformers. Dickson, T.O., + , T-MTT Jan 05 123-133 multilayer on-chip inductors, phys. anal. model. Tong, K.Y., + , T-MTT Apr 05 1143-1149 multiple-metal stacked inductors incorporating, extended phys. model, design. Murphy, O.H., + , T-MTT Jun 05 2063-2072 on-chip vert. solenoid inductor design for multigigahertz CMOS RFIC. Hau-Yiu Tsui, + , T-MTT Jun 05 1883-1890 pad-open-short and open-short-load deembedding techs. for accurate onwafer RF charactn. of high-quality passives, comp. Tiemeijer, L.F., + , T-MTT Feb 05 723-729 Q-factor def. and eval. for spiral inductors fabricated using wafer-level CSP technology. Aoki, Y., + , T-MTT Oct 05 3178-3184 RF inductors and filters, LCP substrs. for Wi-Fi appls., layout-level synthesis. Mukherjee, S., + , T-MTT Jun 05 2196-2210 RF low-noise bandpass filter, act. capacitance cct., design. Young-Hoon Chun, + , T-MTT Feb 05 687-695 simple systematic spiral inductor design, perfected Q improv. for CMOS RFIC appl. Chih-Yuan Lee, + , T-MTT Feb 05 523-528 Information rates increasing wireless channel capacity, MIMO systs. employing colocated antennas. Konanur, A.S., + , T-MTT Jun 05 1837-1844 Information theory measuring BER of UWB devices, prod. test tech. Bhattacharya, S., + , TMTT Nov 05 3474-3481 Injection locked oscillators BPSK to ASK signal conversion using injection-locked oscillators theory, part I. Lopez-Villegas, J.M., + , T-MTT Dec 05 3757-3766 dual opto-electron. oscillator, ultra-low phase noise and ultra-low spurious level. Weimin Zhou, + , T-MTT Mar 05 929-933 v-band high-order harmonic injection-locked freq.-divider MMICs, wide bandwidth and low-power dissipation. Jinho Jeong, + , T-MTT Jun 05 1891-1898 Inorganic compounds; cf. Aluminum compounds; Gallium compounds; Indium compounds; Silicon compounds Instruments; cf. Portable instruments Integral equations anal. of dielec. loaded waveguide filters of arbitrary shape, hybrid surface integral-eqn./mode-matching method. Catina, V., + , T-MTT Nov 05 3562-3567 anal. of NRD-guide/H-guide mm-wave ccts., order-reduced vol.-integral eqn. approach. Duochuan Li, + , T-MTT Mar 05 799-812 bridged NRD-guide coupler for mm-wave appls., anal. and design. Duochuan Li, + , T-MTT Aug 05 2546-2551 complex pass. devices composed of arbitrarily shaped waveguides, Nystrom and BI-RME methods, CAD. Taroncher, M., + , T-MTT Jun 05 2153-2163 coupled lossy transm. lines, multiwavelet-based MoM, full-wave anal. Meisong Tong, + , T-MTT Jul 05 2362-2370 gen. surface-vol. integral-eqn. (SVIE) approach for anal. of hybrid planar/NRD-guide IC. Duochuan Li, + , T-MTT Sep 05 2732-2742 Green's fns. for cylindrical enclosures by spatial images method, num. eval. Pereira, F.D.Q., + , T-MTT Jan 05 94-105 TE/TM propag. and losses of integr. opt. polarizer, accurate modeling. Pierantoni, L., + , T-MTT Jun 05 1856-1862 Integral equations; cf. Boundary integral equations; Volterra equations Integrated circuit design 1-, 10-GHz distrib. amp., CMOS technol., ESD protection design. MingDou Ker, + , T-MTT Sep 05 2672-2681 12-GHz SiGe phase shifter, integr. LNA. Hancock, T.M., + , T-MTT Mar 05 977-983 + Check author entry for coauthors

2.17-dB NF 5-GHz-band monolithic CMOS LNA, 10-mW DC power consumption. Hung-Wei Chiu, + , T-MTT Mar 05 813-824 carrier heating, channel noise, deep-submicrometer NMOSFETs via body bias, expt. study. Hong Wang, + , T-MTT Feb 05 564-570 CMOS RF amp. and mixer ccts. utilizing complementary Characteristics of parallel combined NMOS and PMOS devices. Nam, I., + , T-MTT May 05 1662-1671 DVB-S appls., Si bipolar technol., monolithic 12-GHz heterodyne receiver. Girlando, G., + , T-MTT Mar 05 952-959 efficient nonlin. cct. simul. tech. Dautbegovic, E., + , T-MTT Feb 05 548555 fast layout verification of 3D RF and mixed-sig. on-chip structs., largescale broad-band parasitic extr. Feng Ling, + , T-MTT Jan 05 264-273 high-sensitivity InP/InGaAs DHBT decision cct.-design and appl., opt. and syst. expts., 40-43 Gbit/s. Konczykowska, A., + , T-MTT Apr 05 1228-1234 image-rejection CMOS LNA design optim. techs. Trung-Kien Nguyen, + , T-MTT Feb 05 538-547 InGaP/GaAs HBT RF power amps., ESD protection design considerations. Ma, Y., + , T-MTT Jan 05 221-228 interf. canceller for collocated radios, anal. and design. Raghavan, A., + , T-MTT Nov 05 3498-3508 linearizing FET low-noise amps., modified derivative superposition method. Aparin, V., + , T-MTT Feb 05 571-581 lin. inverse space-mapping (LISM) algm., design lin./nonlin. RF and microwave ccts. Rayas-Sanchez, J.E., + , T-MTT Mar 05 960-968 mm-wave (Bi)CMOS IC, 30-100-GHz inductors and transformers. Dickson, T.O., + , T-MTT Jan 05 123-133 mm-wave CMOS cct. design. Shigematsu, H., + , T-MTT Feb 05 472-477 packaged inductively degenerated common-source low-noise amps., ESD protection, anal. and optim. Sivonen, P., + , T-MTT Apr 05 1304-1313 sig. gener., control, freq. conversion AlGaN/GaN HEMT MMICs. Kaper, V.S., + , T-MTT Jan 05 55-65 simple systematic spiral inductor design, perfected Q improv. for CMOS RFIC appl. Chih-Yuan Lee, + , T-MTT Feb 05 523-528 wideband suppression of ground bounce noise and radiated emission, high-speed ccts., EM bandgap power/ground planes. Tzong-Lin Wu, + , T-MTT Sep 05 2935-2942 Integrated circuit design; cf. Integrated circuit layout Integrated circuit interconnections 20-Gb/s 4-PAM backplane serial I/O interconnections, equalization and NEXT noise cancellation. Hur, Y., + , T-MTT Jan 05 246-255 accurate and scalable RF interconnect model for Si-based RFIC appls. Choon Beng Sia, + , T-MTT Sep 05 3035-3044 fast layout verification of 3D RF and mixed-sig. on-chip structs., largescale broad-band parasitic extr. Feng Ling, + , T-MTT Jan 05 264-273 inter-chip RF-interconnect, CPW, capacitive coupler, UWB transceiver, perform. Sun, M., + , T-MTT Sep 05 2650-2655 low-loss TFMS, Si substr. up, 220 GHz, fab. and charactn. Six, G., + , TMTT Jan 05 301-305 on-chip interconnects by double-image Green's fn. method combined, hierarchical algm., param. extr. Wenliang Dai, + , T-MTT Jul 05 24162423 transm. lines, guaranteed pass. direct lumped-element modeling. Se-Ho You, + , T-MTT Sep 05 2826-2834 Integrated circuit layout InGaP/GaAs HBT RF power amps., ESD protection design considerations. Ma, Y., + , T-MTT Jan 05 221-228 pad-open-short and open-short-load deembedding techs. for accurate onwafer RF charactn. of high-quality passives, comp. Tiemeijer, L.F., + , T-MTT Feb 05 723-729 RF inductors and filters, LCP substrs. for Wi-Fi appls., layout-level synthesis. Mukherjee, S., + , T-MTT Jun 05 2196-2210 Integrated circuit measurements 12-GHz SiGe phase shifter, integr. LNA. Hancock, T.M., + , T-MTT Mar 05 977-983 on-wafer meas. of differential ccts., mm-wave freqs., pure-mode NWA concept. Zwick, T., + , T-MTT Mar 05 934-937 Integrated circuit metallization high aspect ratio through-wafer interconnect vias, Si substrs., microwave charactn. and modeling. Leung, L.L.W., + , T-MTT Aug 05 2472-2480 RF switch matrix appls., integr. interconnect nets. Daneshmand, M., + , TMTT Jan 05 12-21 thin-film microstrip and coplanar technols. for reduced-size MMICs, integrat. Hettak, K., + , T-MTT Jan 05 283-291

IEEE T-MTT 2005 INDEX — 39 Integrated circuit modeling accurate and scalable RF interconnect model for Si-based RFIC appls. Choon Beng Sia, + , T-MTT Sep 05 3035-3044 CMOS technol., complementary Colpitts oscillator. Choong-Yul Cha, + , T-MTT Mar 05 881-887 distrib. preamplifier, cascode FET cells, improved noise anal. Won Ko, + , T-MTT Jan 05 361-371 efficient nonlin. cct. simul. tech. Dautbegovic, E., + , T-MTT Feb 05 548555 full-wave hybrid differential-integral approach for investig. of multilayer structs. incl. nonuniformly doped diffusions. Wane, S., + , T-MTT Jan 05 200-214 gen. surface-vol. integral-eqn. (SVIE) approach for anal. of hybrid planar/NRD-guide IC. Duochuan Li, + , T-MTT Sep 05 2732-2742 inter-chip RF-interconnect, CPW, capacitive coupler, UWB transceiver, perform. Sun, M., + , T-MTT Sep 05 2650-2655 multiple-metal stacked inductors incorporating, extended phys. model, design. Murphy, O.H., + , T-MTT Jun 05 2063-2072 planar microwave components, EM lin. regression models, empirical model gener. techs. Domenech-Asensi, G., + , T-MTT Nov 05 3305-3311 Integrated circuit noise 2.17-dB NF 5-GHz-band monolithic CMOS LNA, 10-mW DC power consumption. Hung-Wei Chiu, + , T-MTT Mar 05 813-824 carrier heating, channel noise, deep-submicrometer NMOSFETs via body bias, expt. study. Hong Wang, + , T-MTT Feb 05 564-570 CMOS technol., complementary Colpitts oscillator. Choong-Yul Cha, + , T-MTT Mar 05 881-887 design of 10-Gb/s AGC amp., jitter considerations. Kucharski, D., + , TMTT Feb 05 590-597 distrib. preamplifier, cascode FET cells, improved noise anal. Won Ko, + , T-MTT Jan 05 361-371 DVB-S appls., Si bipolar technol., monolithic 12-GHz heterodyne receiver. Girlando, G., + , T-MTT Mar 05 952-959 image-rejection CMOS LNA design optim. techs. Trung-Kien Nguyen, + , T-MTT Feb 05 538-547 inter-chip RF-interconnect, CPW, capacitive coupler, UWB transceiver, perform. Sun, M., + , T-MTT Sep 05 2650-2655 linearizing FET low-noise amps., modified derivative superposition method. Aparin, V., + , T-MTT Feb 05 571-581 wideband suppression of ground bounce noise and radiated emission, high-speed ccts., EM bandgap power/ground planes. Tzong-Lin Wu, + , T-MTT Sep 05 2935-2942 Integrated circuit packaging miniaturized microwave pass. filter incorporating multilayer synthetic quasiTEM transm. line. Hsien-Shun Wu, + , T-MTT Sep 05 2713-2720 nondestructive in situ S-param. meas. of hermetically encapsulated packages, recursive un-termination method. Pfeiffer, U.R., + , T-MTT Jun 05 1845-1855 Integrated circuit reliability 1-, 10-GHz distrib. amp., CMOS technol., ESD protection design. MingDou Ker, + , T-MTT Sep 05 2672-2681 thermal resist. of FETs, accurate determ. Darwish, A.M., + , T-MTT Jan 05 306-313 Integrated circuits 2004 IEEE Radio Frequency Integrated Circuits Symposium, RFIC (special section). T-MTT Feb 05 451-626 2004 IEEE Radio Frequency Integrated Circuits Symposium, RFIC (special section intro.). Quach, T., T-MTT Feb 05 451-452 Integrated circuits; cf. Analog integrated circuits; Digital integrated circuits; Power integrated circuits; Thin film circuits Integrated circuit technology; cf. Integrated circuit interconnections; Integrated circuit metallization; Integrated circuit packaging; Isolation technology Integrated circuit testing gain compress. curve, IIP3 estim. Choongeol Cho, + , T-MTT Apr 05 1197-1202 Integrated optics TE/TM propag. and losses of integr. opt. polarizer, accurate modeling. Pierantoni, L., + , T-MTT Jun 05 1856-1862 Integration (mathematics) anal. of NRD-guide/H-guide mm-wave ccts., order-reduced vol.-integral eqn. approach. Duochuan Li, + , T-MTT Mar 05 799-812

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Integrodifferential equations full-wave hybrid differential-integral approach for investig. of multilayer structs. incl. nonuniformly doped diffusions. Wane, S., + , T-MTT Jan 05 200-214 Integro-differential equations; cf. Boltzmann equation Intelligent control intelligently controlled RF power amp., reconfigurable MEMS-varactor tuner. Dongjiang Qiao, + , T-MTT Mar 05 1089-1095 Interchannel interference adaptive predistortion RF power amp., spectrum monitor for multicarrier WCDMA appls. Seung-Yup Lee, + , T-MTT Feb 05 786-793 Interconnections; cf. Integrated circuit interconnections Interference (signal); cf. Crosstalk; Electromagnetic interference; Intersymbol interference Interference suppression 20-Gb/s 4-PAM backplane serial I/O interconnections, equalization and NEXT noise cancellation. Hur, Y., + , T-MTT Jan 05 246-255 4-port microwave nets., intrinsic broad-band suppression of commonmode sigs. Fathelbab, W.M., + , T-MTT May 05 1569-1575 adaptive duplexer implemented, single-path/multipath feedforward techs., BST phase shifters. O'Sullivan, T., + , T-MTT Jan 05 106-114 canceller for collocated radios, anal. and design. Raghavan, A., + , T-MTT Nov 05 3498-3508 EM field-mapping syst., reson.-suppressed mag. field probe. Jung-Min Kim, + , T-MTT Sep 05 2693-2699 measuring BER of UWB devices, prod. test tech. Bhattacharya, S., + , TMTT Nov 05 3474-3481 miniaturized parallel coupled-line bandpass filter, spurious-response suppression. Pedro Cheong, + , T-MTT May 05 1810-1816 parallel coupled microstrip filters, floating ground-plane conductor for spurious-band suppression. Velazquez-Ahumada, Md.C., + , T-MTT May 05 1823-1828 secure high-speed retrodirective commun. link. Goshi, D.S., + , T-MTT Nov 05 3548-3556 wideband suppression of ground bounce noise and radiated emission, high-speed ccts., EM bandgap power/ground planes. Tzong-Lin Wu, + , T-MTT Sep 05 2935-2942 Intermodulation; cf. Intermodulation distortion Intermodulation distortion 3rd.- and fifth-order baseband component injection for linearization of power amp., cellular phone. Mizusawa, N., + , T-MTT Nov 05 33273334 asymmetrical-cells-based lin. Doherty power Amplifiers-uneven power drive and power matching, optimum operation. Jangheon Kim, + , TMTT May 05 1802-1809 class-F and inverse class-F HBT amps., IMD anal. Ohta, A., + , T-MTT Jun 05 2121-2128 contact-type MEMS microswitch, determ. Johnson, J., + , T-MTT Nov 05 3615-3620 high av.-effic. SiGe HBT power amp. for WCDMA handset appls. Junxiong Deng, + , T-MTT Feb 05 529-537 linearizing FET low-noise amps., modified derivative superposition method. Aparin, V., + , T-MTT Feb 05 571-581 metamaterial-based electronically controlled transm.-line struct., leakywave antenna, tunable radiation angle and beamwidth. Sungjoon Lim, + , T-MTT Jan 05 161-173 multichannel commun. systs., modeling distortion. Gharaibeh, K.M., + , T-MTT May 05 1682-1692 nonlin. device charactn., multitone phase and amplit. meas. Martins, J.P., + , T-MTT Jun 05 1982-1989 offset-PLL output spur spectrum, systematic anal. Ching-Feng Lee, + , TMTT Sep 05 3024-3034 presence of LF dispers. effects, accurate pHEMT nonlin. modeling. Raffo, A., + , T-MTT Nov 05 3449-3459 RF front-end characts., spectral regrowth of communs. sigs., impact. Gard, K.G., + , T-MTT Jun 05 2179-2186 tunable combline filter, continuous control of center freq. and bandwidth. Sanchez-Renedo, M., + , T-MTT Jan 05 191-199 wireless transmitters, digital subband filtering predistorter archit. Hammi, O., + , T-MTT May 05 1643-1652 X-band class-E power amps., EER operation, linearity. Narisi Wang, + , T-MTT Mar 05 1096-1102 Interpolation low-refl. subgridding. Kulas, L., + , T-MTT May 05 1587-1592

IEEE T-MTT 2005 INDEX — 40 single and multicarrier W-CDMA sigs., LINC digital component separator. Gerhard, W., + , T-MTT Jan 05 274-282 TLM-based modeling and design exploiting space mapping. Bandler, J.W., + , T-MTT Sep 05 2801-2811 uniplanar left-handed metamaterials, efficient modeling. Yunchuan Guo, + , T-MTT Apr 05 1462-1468 Intersymbol interference multigigahertz parallel bus, transmit preemphasis equalization, perform. anal. and model-to-hardware correl. Beyene, W.T., + , T-MTT Nov 05 3568-3577 Inverse problems Chebyshev collocation and Newton-type optim. methods for inverse problem, nonuniform transm. lines. Norgren, M., T-MTT May 05 15611568 in vivo and ex dielec. spectrosc. of biol. tissues, microwave freqs., precision open-ended coaxial probes. Popovic, D., + , T-MTT May 05 1713-1722 particle swarm optimizer for microwave imaging of 2D dielec. scatterers, comput. approach. Donelli, M., + , T-MTT May 05 1761-1776 Ion implantation simple systematic spiral inductor design, perfected Q improv. for CMOS RFIC appl. Chih-Yuan Lee, + , T-MTT Feb 05 523-528 Iron alloys crossed anisotropy mag. core, toroidal thin-film inductors, integrat. Frommberger, M., + , T-MTT Jun 05 2096-2100 Iron compounds; cf. Ferrites Isolation technology full-wave hybrid differential-integral approach for investig. of multilayer structs. incl. nonuniformly doped diffusions. Wane, S., + , T-MTT Jan 05 200-214 Iterative methods conductive medium, preiter. ADI-FDTD method. Shumin Wang, + , TMTT Jun 05 1913-1918 extracting causal time-domain params., iter. methods. Shuiping Luo, + , TMTT Mar 05 969-976 general lin. lumped microwave ccts., matrix theory, improved FDTD formulation. Zhenhai Shao, + , T-MTT Jul 05 2261-2266 Iterative methods; cf. Newton method; Newton-Raphson method

J Jacobian matrices Chebyshev collocation and Newton-type optim. methods for inverse problem, nonuniform transm. lines. Norgren, M., T-MTT May 05 15611568 JFETs short-channel AlGaN/GaN heterojunction FETs, 30-GHz-band, 5-W power perform. Inoue, T., + , T-MTT Jan 05 74-80 Jitter design of 10-Gb/s AGC amp., jitter considerations. Kucharski, D., + , TMTT Feb 05 590-597 multigigahertz parallel bus, transmit preemphasis equalization, perform. anal. and model-to-hardware correl. Beyene, W.T., + , T-MTT Nov 05 3568-3577 photoelectronic ADC. Ioakeimidi, K., + , T-MTT Jan 05 336-342 K Knowledge based systems; cf. Intelligent control L Ladder circuits modified loss-compensation method, 0.35-ȝm SiGe BiCMOS technol., broad-band MMICs. Ming-Da Tsai, + , T-MTT Feb 05 496-505 Ladder filters 20-Gb/s 4-PAM backplane serial I/O interconnections, equalization and NEXT noise cancellation. Hur, Y., + , T-MTT Jan 05 246-255 Ladder networks; cf. Ladder filters Land mobile radio cellular systems 3rd.- and fifth-order baseband component injection for linearization of power amp., cellular phone. Mizusawa, N., + , T-MTT Nov 05 33273334 detect. and localization of mobile phones, large buildings, microwave syst. Hudec, P., + , T-MTT Jun 05 2235-2239 + Check author entry for coauthors

develop realistic num. models of cellular phones for accurate eval. of SAR distrib., human head, procedure. Pisa, S., + , T-MTT Apr 05 1256-1265 EDGE terminal power amps., memoryless digital predistortion, optim. Ceylan, N., + , T-MTT Feb 05 515-522 GSM/EGSM/DCS/PCS direct conversion receiver, integr. synthesizer. Young-Jin Kim, + , T-MTT Feb 05 606-613 linearizing FET low-noise amps., modified derivative superposition method. Aparin, V., + , T-MTT Feb 05 571-581 multimode J-pHEMT front-end archit., power-control scheme for max. effic. Clifton, J.C., + , T-MTT Jun 05 2251-2258 multistandard mobile terminals, fully integr. receivers requirements and archits. Brandolini, M., + , T-MTT Mar 05 1026-1038 RF front-end characts., spectral regrowth of communs. sigs., impact. Gard, K.G., + , T-MTT Jun 05 2179-2186 Laplace equations thermal resist. of FETs, accurate determ. Darwish, A.M., + , T-MTT Jan 05 306-313 Lasers coherent optical vector modulation for fiber radio using electrooptic microchip lasers. Li, Y., + , T-MTT Oct 05 3121-3129 Lasers; cf. Distributed feedback lasers Layout RF ccts., LCP substrs., stat. anal. and diagnosis methodology. Mukherjee, S., + , T-MTT Nov 05 3621-3630 waveguide filters, multiple atten. poles, dual-behavior reson. of freq.selective surfaces. Ohira, M., + , T-MTT Nov 05 3320-3326 Layout of circuit boards multigigahertz parallel bus, transmit preemphasis equalization, perform. anal. and model-to-hardware correl. Beyene, W.T., + , T-MTT Nov 05 3568-3577 Lead bonding nondestructive in situ S-param. meas. of hermetically encapsulated packages, recursive un-termination method. Pfeiffer, U.R., + , T-MTT Jun 05 1845-1855 Leaky wave antennas metamaterial-based electronically controlled transm.-line struct., leakywave antenna, tunable radiation angle and beamwidth. Sungjoon Lim, + , T-MTT Jan 05 161-173 Least squares methods mismatch errors for 400-msamples/s 80-dB SFDR time-interleaved ADC, comprehensive digital correction. Munkyo Seo, + , T-MTT Mar 05 10721082 Lens antennas terahertz mixers and detectors, design guidelines. Focardi, P., + , T-MTT May 05 1653-1661 Lenses left-handed transm.-line media, nonlin. wave propag. phenom. Kozyrev, A.B., + , T-MTT Jan 05 238-245 microwave/mm-wave appls., dielec. slab Rotman lens. Jaeheung Kim, + , T-MTT Aug 05 2622-2627 Life estimation capacitive RF MEMS, reliab. modeling. Melle, S., + , T-MTT Nov 05 3482-3488 Linear algebra; cf. Eigenvalues and eigenfunctions; Vectors Linear approximation 3rd.- and fifth-order baseband component injection for linearization of power amp., cellular phone. Mizusawa, N., + , T-MTT Nov 05 33273334 dual-band transmitters, digitally predistorted freq. multipliers for reconfigurable radios. Youngcheol Park, + , T-MTT Jan 05 115-122 EDGE terminal power amps., memoryless digital predistortion, optim. Ceylan, N., + , T-MTT Feb 05 515-522 hybrid digital/RF envelope predistortion linearization syst. for power amps. Wangmyong Woo, + , T-MTT Jan 05 229-237 large-sig. scatt. fns., linearization. Verspecht, J., + , T-MTT Apr 05 13691376 LINC amps., Chireix-outphasing combiners, phase-only predistortion. Birafane, A., + , T-MTT Jun 05 2240-2250 Maximum Output control method for UMTS downlink transmitters, adaptive feedforward amp. Legarda, J., + , T-MTT Aug 05 2481-2486 Linear circuits compact parallel-coupled microstrip bandpass filters, lumped-element Kinverters. Yo-Shen Lin, + , T-MTT Jul 05 2324-2328

IEEE T-MTT 2005 INDEX — 41 compact planar and vialess composite low-pass filters, folded steppedimpedance resonator, liq.-Crystal-polymer substr. Pinel, S., + , T-MTT May 05 1707-1712 comput. time-domain sensitivity of multiport systs. described by reducedorder models, adjoint-based approach. Ahmed, T., + , T-MTT Nov 05 3538-3547 general lin. lumped microwave ccts., matrix theory, improved FDTD formulation. Zhenhai Shao, + , T-MTT Jul 05 2261-2266 inverse space-mapping (LISM) algm., design lin./nonlin. RF and microwave ccts. Rayas-Sanchez, J.E., + , T-MTT Mar 05 960-968 low-refl. bandpass filters, flat group delay. Djordjevic, A.R., + , T-MTT Apr 05 1164-1167 lumped-element quadrature power splitters, mixed right/left-handed transm. lines. Kuylenstierna, D., + , T-MTT Aug 05 2616-2621 PCB discontinuities, wavelet domain, 2-port equiv. Araneo, R., + , T-MTT Mar 05 907-918 RFICs, 2-port lumped-nonlin.-source model, systematic linearity anal. Qingqing Liang, + , T-MTT May 05 1745-1755 transm. lines, guaranteed pass. direct lumped-element modeling. Se-Ho You, + , T-MTT Sep 05 2826-2834 wide tuning-range planar filters, lumped-distrib. coupled resonators. Carey-Smith, B.E., + , T-MTT Feb 05 777-785 Liquid crystal materials/devices microstrip-line-type PDLC loaded variable phase shifter, increasing speed. Utsumi, Y., + , T-MTT Nov 05 3345-3353 Liquid crystals compact planar and vialess composite low-pass filters, folded steppedimpedance resonator, liq.-Crystal-polymer substr. Pinel, S., + , T-MTT May 05 1707-1712 microstrip-line-type PDLC loaded variable phase shifter, increasing speed. Utsumi, Y., + , T-MTT Nov 05 3345-3353 RF ccts., LCP substrs., stat. anal. and diagnosis methodology. Mukherjee, S., + , T-MTT Nov 05 3621-3630 temp.-controlled coaxial transm. line, broad-band microwave charactn. Mueller, S., + , T-MTT Jun 05 1937-1945 Liquids; cf. Liquid crystals Lithium compounds simultaneous elec. and mag. near-field meas., LiNbO3, inverted domain, EO probe. Suzuki, E., + , T-MTT Feb 05 696-701 Lithography; cf. Photolithography Liver microwave ablation, triaxial antenna. Brace, C.L., + , T-MTT Jan 05 215220 Local area networks effectiveness of wave absorbers, improve DSRC EM environ., express highway. Pokharel, R.K., + , T-MTT Sep 05 2726-2731 interf. canceller for collocated radios, anal. and design. Raghavan, A., + , T-MTT Nov 05 3498-3508 Local area networks; cf. Wireless LAN Loop antennas increasing wireless channel capacity, MIMO systs. employing colocated antennas. Konanur, A.S., + , T-MTT Jun 05 1837-1844 simultaneous elec. and mag. near-field meas., LiNbO3, inverted domain, EO probe. Suzuki, E., + , T-MTT Feb 05 696-701 Losses; cf. Dielectric losses Loss measurement Wang's model for RT cond. losses, normal metals, terahertz freqs. Lucyszyn, S., T-MTT Apr 05 1398-1403 Low-pass filters compact planar and vialess composite low-pass filters, folded steppedimpedance resonator, liq.-Crystal-polymer substr. Pinel, S., + , T-MTT May 05 1707-1712 compact U-shaped dual planar EBG microstrip low-pass filter. Huang, S.Y., + , T-MTT Dec 05 3799-3805 defected ground struct., design. Jong-Sik Lim, + , T-MTT Aug 05 25392545 gen. Chebyshev filters, asymmetrically located transm. zeros, design. Milosavljevic, Z.D., T-MTT Jul 05 2411-2415 stopband-enhanced and size-miniaturized low-pass filters, high-impedance property of offset finite-ground microstrip line. Sheng Sun, + , T-MTT Sep 05 2844-2850 Luminescence; cf. Photoluminescence

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M

Magnesium compounds complex permitt. of arbitrary shape and size dielec. samples, cavity meas. tech., microwave freqs., estim. Santra, M., + , T-MTT Feb 05 718-722 Magnetic anisotropy crossed anisotropy mag. core, toroidal thin-film inductors, integrat. Frommberger, M., + , T-MTT Jun 05 2096-2100 Magnetic cores crossed anisotropy mag. core, toroidal thin-film inductors, integrat. Frommberger, M., + , T-MTT Jun 05 2096-2100 Magnetic devices crossed anisotropy mag. core, toroidal thin-film inductors, integrat. Frommberger, M., + , T-MTT Jun 05 2096-2100 Magnetic devices; cf. Magnetic microwave devices Magnetic field measurement EM field-mapping syst., reson.-suppressed mag. field probe. Jung-Min Kim, + , T-MTT Sep 05 2693-2699 simultaneous elec. and mag. near-field meas., LiNbO3, inverted domain, EO probe. Suzuki, E., + , T-MTT Feb 05 696-701 Magnetic films charactn., microwave range, broad-band method. Vincent, D., + , T-MTT Apr 05 1174-1180 crossed anisotropy mag. core, toroidal thin-film inductors, integrat. Frommberger, M., + , T-MTT Jun 05 2096-2100 thin-film charactn., microwave range, broad-band method. Vincent, D., + , T-MTT Apr 05 1174-1180 Magnetic layered films crossed anisotropy mag. core, toroidal thin-film inductors, integrat. Frommberger, M., + , T-MTT Jun 05 2096-2100 Magnetic materials moisture content, bulk materials, sub-nanosecond UWB pulses, noncontacting determ. Schimmer, O., + , T-MTT Jun 05 2107-2113 unbiased integr. microstrip circulator, mag. nanowired substr. Saib, A., + , T-MTT Jun 05 2043-2049 Magnetic microwave devices thin-film charactn., microwave range, broad-band method. Vincent, D., + , T-MTT Apr 05 1174-1180 Magnetic variables measurement; cf. Magnetic field measurement Magnetism; cf. Magnetic materials Magnetization; cf. Magnetic anisotropy Magnetization processes neg. permitt. and neg. permeab. by of evanescent waveguide Modestheory and expt., simul. Esteban, J., + , T-MTT Apr 05 1506-1514 thin-film charactn., microwave range, broad-band method. Vincent, D., + , T-MTT Apr 05 1174-1180 Magneto-optical effects; cf. Faraday effect Magnetostatic waves anal. software for tunable microstrip filters, magnetized ferrites, expt. validation. Leon, G., + , T-MTT May 05 1739-1744 Masers; cf. Cyclotron masers Materials; cf. Ceramics; Dielectric materials; Magnetic materials; Microwave materials; Optical materials; Quartz; Semiconductor materials Materials testing; cf. Nondestructive testing Mathematical analysis; cf. Bessel functions; Eigenvalues and eigenfunctions; Integral equations; Inverse problems; Numerical analysis; Transforms Mathematics; cf. Conformal mapping; Statistics Matrices general lin. lumped microwave ccts., matrix theory, improved FDTD formulation. Zhenhai Shao, + , T-MTT Jul 05 2261-2266 rect. waveguides, radially symmetric metal insets, fast S-domain modeling. Mira, F., + , T-MTT Apr 05 1294-1303 synthesis of cascaded N-tuplets cross-coupled resonators microwave filters, matrix rotations, anal. tech. Tamiazzo, S., + , T-MTT May 05 1693-1698 Matrix algebra; cf. Jacobian matrices; Matrix decomposition; Transmission line matrix methods Matrix decomposition metamaterial constructed by conductive SRRs and wires, MGS-based algm., propag. property anal. Hai-Ying Yao, + , T-MTT Apr 05 14691476

IEEE T-MTT 2005 INDEX — 42 Maxwell equations coupling 3D Maxwell's and Boltzmann's eqns. for analyzing, terahertz photoconductive switch. Sirbu, M., + , T-MTT Sep 05 2991-2998 FEM solns. of dielec. waveguiding structs., 2D curl-conforming sing. elements. Din-Kow Sun, + , T-MTT Mar 05 984-992 freq. response of SAW filters, FDTD method. King-Yuen Wong, + , TMTT Nov 05 3364-3370 loaded transm.-line neg.-refr.-index metamaterials, periodic FDTD anal. Kokkinos, T., + , T-MTT Apr 05 1488-1495 reduce num. dispers., ADI-FDTD, efficient method. Hong-Xing Zheng, + , T-MTT Jul 05 2295-2301 temporally dispers. dielec., time-domain cavity oscills. supported. Aksoy, S., + , T-MTT Aug 05 2465-2471 Measurement; cf. Distortion measurement; Dosimetry; Frequency measurement; Loss measurement; Noise measurement; Q-factor measurement; Radiometry Measurement errors effect of cable length in vector measurements of very long millimeter wave components. Simonetto, A., + , T-MTT Dec 05 3731-3734 Medical treatment microwave ablation, triaxial antenna. Brace, C.L., + , T-MTT Jan 05 215220 Meetings 15th International Conference on Microwaves, Radar, and Wireless Communications, MIKON (special section). T-MTT Feb 05 425-450 15th International Conference on Microwaves, Radar, and Wireless Communications, MIKON (special section intro.). Modelski, J.W., TMTT Feb 05 425-426 16th Asia-Pacific Microwave Conference, APMC'04 (special section). TMTT Sep 05 2649-2731 16th Asia-Pacific Microwave Conference, APMC'04 (special section intro.). Steer, M.B., T-MTT Sep 05 2649 2004 IEEE MTT-S International Microwave Symposium (special section). T-MTT Jan 05 3-413 2004 IEEE MTT-S International Microwave Symposium (special section intro.). Steer, M., T-MTT Jan 05 3 2004 IEEE Radio Frequency Integrated Circuits Symposium, RFIC (special section). T-MTT Feb 05 451-626 2004 IEEE Radio Frequency Integrated Circuits Symposium, RFIC (special section intro.). Quach, T., T-MTT Feb 05 451-452 2005 International Microwave Symposium, MTT-S05 (special issue). TMTT Nov 05 3264-3678 2005 International Microwave Symposium, MTT-S05 (special issue intro.). Jackson, C., T-MTT Nov 05 3264 34th European Microwave Conference, EuMC (special issue). T-MTT Jun 05 1935-2258 34th European Microwave Conference, EuMC (special issue intro.). Russer, P., T-MTT Jun 05 1935-1936 Memoryless systems EDGE terminal power amps., memoryless digital predistortion, optim. Ceylan, N., + , T-MTT Feb 05 515-522 MESFET circuits relationships between common source, common gate, and common drain FETs. Gao, J., + , T-MTT Dec 05 3825-3831 X-band class-E power amps., EER operation, linearity. Narisi Wang, + , T-MTT Mar 05 1096-1102 MESFETs efficient anal. formulation and sensitivity anal. of neuro-space mapping for nonlin. microwave device modeling. Lei Zhang, + , T-MTT Sep 05 2752-2767 x-band 2-stage high-effic. switched-mode power amps. Pajic, S., + , TMTT Sep 05 2899-2907 Mesh generation (2,4) stencil, optimized FDTD methods. Guilin Sun, + , T-MTT Mar 05 832-842 anal. of dielec. loaded waveguide filters of arbitrary shape, hybrid surface integral-eqn./mode-matching method. Catina, V., + , T-MTT Nov 05 3562-3567 unstructured tetrahedral meshes, transm.-line modeling (TLM). Sewell, P., + , T-MTT Jun 05 1919-1928 Metallization; cf. Integrated circuit metallization Metals Wang's model for RT cond. losses, normal metals, terahertz freqs. Lucyszyn, S., T-MTT Apr 05 1398-1403

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Meteorological radar inform., turbulence, rain by Doppler-polarimetric Radar, retrieval. Yanovsky, F.J., + , T-MTT Feb 05 444-450 Meteorology; cf. Meteorological radar Metropolitan access networks frequency channel blocking scheme in mesh-topology millimeter-wave broad band entrance networks. Sangiamwong, J., + , T-MTT Dec 05 3723-3730 Microactuators dielec. resonator, discrete electromechanical freq. tuning. Panaitov, G.I., + , T-MTT Nov 05 3371-3377 Microassembling; cf. Lead bonding Microelectromechanical devices biol. appls., micromachined probe, in vivo meas. Jung-Mu Kim, + , TMTT Nov 05 3415-3421 high-power MEMS varactors and impedance tuners for mm-wave appls. Lu, Y., + , T-MTT Nov 05 3672-3678 hybrid narrow-band tunable bandpass filter, varactor loaded EM-bandgap CPW. Pistono, E., + , T-MTT Aug 05 2506-2514 low-cost planar probes, broadside apertures for nondestructive dielec. meas. of biol. materials, microwave freqs. Byoungjoong Kang, + , TMTT Jan 05 134-143 phased-array antenna systs., RF MEMS for automotive radar appls., design considerations and technol. assess. Schoebel, J., + , T-MTT Jun 05 1968-1975 Micromechanical devices; cf. Microactuators; Microsensors Microscopy; cf. Electron microscopy Microsensors low-cost planar probes, broadside apertures for nondestructive dielec. meas. of biol. materials, microwave freqs. Byoungjoong Kang, + , TMTT Jan 05 134-143 Microstrip compact dual-fed distrib. power amp. Eccleston, K.W., T-MTT Mar 05 825-831 compact EM-bandgap (EBG) struct. and appls. for microwave ccts. Li Yang, + , T-MTT Jan 05 183-190 distrib. left-handed microstrip lines, effective EM params. Shau-Gang Mao, + , T-MTT Apr 05 1515-1521 EM distribs. demonstrating asymmetry, spectral-domain dyadic Green's fn. for ferrite microstrip guided-wave structs. Krowne, C.M., T-MTT Apr 05 1345-1361 EM field-mapping syst., reson.-suppressed mag. field probe. Jung-Min Kim, + , T-MTT Sep 05 2693-2699 hyperb. transm.-line periodic grids, neg. refr. and focusing. Eleftheriades, G.V., + , T-MTT Jan 05 396-403 lin. inverse space-mapping (LISM) algm., design lin./nonlin. RF and microwave ccts. Rayas-Sanchez, J.E., + , T-MTT Mar 05 960-968 low-loss TFMS, Si substr. up, 220 GHz, fab. and charactn. Six, G., + , TMTT Jan 05 301-305 microstrip-line struct. employing, periodically perforated ground metal and appl., highly miniaturized and low-impedance pass. components fabricated. Young Yun, T-MTT Jun 05 1951-1959 microstrip-line-type PDLC loaded variable phase shifter, increasing speed. Utsumi, Y., + , T-MTT Nov 05 3345-3353 microwave filters, improved stopband, sub-wavel. resonators. GarciaGarcia, J., + , T-MTT Jun 05 1997-2006 modeling microstrip structs., nonuniform grids and perfectly matched layer, compact 2D FDFD method. Jiunn-Nan Hwang, T-MTT Feb 05 653-659 nonintuitive planar structs., fast optim. and sensitivity anal. Cormos, D., + , T-MTT Jun 05 2019-2025 periodically nonuniform coupled microstrip-line filters, harmonic suppression, transm. zero reallocation. Sheng Sun, + , T-MTT May 05 1817-1822 periodically nonuniform coupled microstrip lines-even and odd modes, guided-wave characts. Sheng Sun, + , T-MTT Apr 05 1221-1227 planar microwave components, EM lin. regression models, empirical model gener. techs. Domenech-Asensi, G., + , T-MTT Nov 05 3305-3311 stopband-enhanced and size-miniaturized low-pass filters, high-impedance property of offset finite-ground microstrip line. Sheng Sun, + , T-MTT Sep 05 2844-2850 tunable impedance transformer, transm. line, variable charact. impedance. Hyeong Tae Jeong, + , T-MTT Aug 05 2587-2593 unbiased integr. microstrip circulator, mag. nanowired substr. Saib, A., + , T-MTT Jun 05 2043-2049

IEEE T-MTT 2005 INDEX — 43 virtual transm. line, FDTD, efficient excit. Karkkainen, M.K., T-MTT Jun 05 1899-1903 wide-band finite-element model-order reduction, fast waveguide eigenanalysis. Shih-Hao Lee, + , T-MTT Aug 05 2552-2558 Microstrip antennas anal. of radial waveguides, dielec. resonators, microstrip antennas, spherical multilayer structs., unified approach. Truong Vu Bang Giang, + , T-MTT Jan 05 404-409 branch-line directional coupler in the design of microwave bandpass filters. Gomez-Garcia, R., + , T-MTT Oct 05 3221-3229 broad-band and circ. polarized space-filling-based slot antennas. Ghali, H.A., + , T-MTT Jun 05 1946-1950 highly integr. mm-wave pass. components, 3D LTCC syst.-on-package (SOP) technol. Jong-Hoon Lee, + , T-MTT Jun 05 2220-2229 lossy Foster networks. Kajfez, D., T-MTT Oct 05 3199-3205 low-Q microwave resonator, crit.-points method, accurate charactn. Peng Wang, + , T-MTT Jan 05 349-353 metamaterial-based electronically controlled transm.-line struct., leakywave antenna, tunable radiation angle and beamwidth. Sungjoon Lim, + , T-MTT Jan 05 161-173 microstrip three-port and four-channel multiplexer for WLAN and UWB coexistence. Lai, M.-I., + , T-MTT Oct 05 3244-3250 nonintuitive planar structs., fast optim. and sensitivity anal. Cormos, D., + , T-MTT Jun 05 2019-2025 scan antennas, ferroelec. substr., rigorous anal. and investigs. Yashchyshyn, Y., + , T-MTT Feb 05 427-438 Microstrip arrays compact EM-bandgap (EBG) struct. and appls. for microwave ccts. Li Yang, + , T-MTT Jan 05 183-190 LINC radio transmitters, integr. antenna/power combiner. Gao, S., + , TMTT Mar 05 1083-1088 mm-wave high-effic. multilayer parasitic microstrip antenna array, teflon substr. Seki, T., + , T-MTT Jun 05 2101-2106 Microstrip circuits 2D nonuniform FFT (2-D NUFFT) tech., anal. of shielded microstrip ccts. Ke-Ying Su, + , T-MTT Mar 05 993-999 mm-wave CMOS cct. design. Shigematsu, H., + , T-MTT Feb 05 472-477 nets., orthogonality-based deembedding tech. Spowart, M.P., + , T-MTT Mar 05 938-946 thin-film microstrip and coplanar technols. for reduced-size MMICs, integrat. Hettak, K., + , T-MTT Jan 05 283-291 Microstrip components compact MMIC CPW and asymmetric CPS branch-line couplers and Wilkinson dividers, shunt and series stub loading. Hettak, K., + , T-MTT May 05 1624-1635 ferrite coupled-line circulator, reduced length. Meng Cao, + , T-MTT Aug 05 2572-2579 freq. multiplier, waveguide-based spatial power-combining archit. Belaid, M., + , T-MTT Apr 05 1124-1129 modeling microstrip structs., nonuniform grids and perfectly matched layer, compact 2D FDFD method. Jiunn-Nan Hwang, T-MTT Feb 05 653-659 waveguide polarizers, design tool. Virone, G., + , T-MTT Mar 05 888-894 Microstrip components; cf. Microstrip antennas; Microstrip circuits; Microstrip couplers; Microstrip filters; Microstrip resonators Microstrip couplers miniaturized spurious passband suppression microstrip filter, meandered parallel coupled lines. Shih-Ming Wang, + , T-MTT Feb 05 747-753 thin-film microstrip and coplanar technols. for reduced-size MMICs, integrat. Hettak, K., + , T-MTT Jan 05 283-291 Microstrip filters anal. software for tunable microstrip filters, magnetized ferrites, expt. validation. Leon, G., + , T-MTT May 05 1739-1744 bandpass filters, dual-passband response, design. Jen-Tsai Kuo, + , T-MTT Apr 05 1331-1337 compact bandpass filter, 2 tuning transm. zeros, CMRC resonator. Kam Man Shum, + , T-MTT Mar 05 895-900 compact parallel-coupled microstrip bandpass filters, lumped-element Kinverters. Yo-Shen Lin, + , T-MTT Jul 05 2324-2328 designing microstrip filters utilizing mixed dielectrics, approaches. Semouchkina, E., + , T-MTT Feb 05 644-652 differential 4-bit 6.5-10-GHz RF MEMS tunable filter. Entesari, K., + , TMTT Mar 05 1103-1110

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electronically tunable microstrip bandpass filter, thin-film BariumStrontium-Titanate (BST) varactors. Nath, J., + , T-MTT Sep 05 27072712 future mobile telecomm. systs., high-temp. supercond. filter. Jia-Sheng Hong, + , T-MTT Jun 05 1976-1981 improved coupled-microstrip filter design, effective even-mode/odd-mode charact. impedances. Hong-Ming Lee, + , T-MTT Sep 05 2812-2818 low-loss 2-bit tunable bandpass filters, MEMS DC contact switches. Pothier, A., + , T-MTT Jan 05 354-360 microstrip bandpass filters with multiorder spurious-mode suppression. Chen, C.-F., + , T-MTT Dec 05 3788-3793 miniaturized dual-mode bandpass filter struct., shunt-capacitance perturb. Ming-Fong Lei, + , T-MTT Mar 05 861-867 miniaturized spurious passband suppression microstrip filter, meandered parallel coupled lines. Shih-Ming Wang, + , T-MTT Feb 05 747-753 parallel coupled microstrip filters, floating ground-plane conductor for spurious-band suppression. Velazquez-Ahumada, Md.C., + , T-MTT May 05 1823-1828 periodically nonuniform coupled microstrip-line filters, harmonic suppression, transm. zero reallocation. Sheng Sun, + , T-MTT May 05 1817-1822 quadruplet filters, source-load coupling, design. Ching-Ku Liao, + , TMTT Jul 05 2302-2308 RF inductors and filters, LCP substrs. for Wi-Fi appls., layout-level synthesis. Mukherjee, S., + , T-MTT Jun 05 2196-2210 RF/microwave multifunctional systs., reconfigurable bandpass filter. Fathelbab, W.M., + , T-MTT Mar 05 1111-1116 stopband-enhanced and size-miniaturized low-pass filters, high-impedance property of offset finite-ground microstrip line. Sheng Sun, + , T-MTT Sep 05 2844-2850 suppression of second harmonic, fractal-shaped microstrip coupled-line bandpass filters. Il Kwon Kim, + , T-MTT Sep 05 2943-2948 tapered dual-plane compact EM bandgap microstrip filter structs. Shao Ying Huang, + , T-MTT Sep 05 2656-2664 Microstrip resonators compact bandpass filter, 2 tuning transm. zeros, CMRC resonator. Kam Man Shum, + , T-MTT Mar 05 895-900 designing microstrip filters utilizing mixed dielectrics, approaches. Semouchkina, E., + , T-MTT Feb 05 644-652 highly integr. mm-wave pass. components, 3D LTCC syst.-on-package (SOP) technol. Jong-Hoon Lee, + , T-MTT Jun 05 2220-2229 low-loss 2-bit tunable bandpass filters, MEMS DC contact switches. Pothier, A., + , T-MTT Jan 05 354-360 low-Q microwave resonator, crit.-points method, accurate charactn. Peng Wang, + , T-MTT Jan 05 349-353 microstrip-line-type PDLC loaded variable phase shifter, increasing speed. Utsumi, Y., + , T-MTT Nov 05 3345-3353 miniaturized dual-mode bandpass filter struct., shunt-capacitance perturb. Ming-Fong Lei, + , T-MTT Mar 05 861-867 miniaturized dual-mode ring bandpass filter, perturb. Boon Tiong Tan, + , T-MTT Jan 05 343-348 miniaturized multilayer quasiellipt. bandpass filter, aperture-coupled microstrip resonators. Chi-Feng Chen, + , T-MTT Sep 05 2688-2692 supercond. spiral filters, quasiellipt. charact. for radio astron. Guoyong Zhang, + , T-MTT Mar 05 947-951 Microwave amplifiers 12-GHz SiGe phase shifter, integr. LNA. Hancock, T.M., + , T-MTT Mar 05 977-983 image-rejection CMOS LNA design optim. techs. Trung-Kien Nguyen, + , T-MTT Feb 05 538-547 InP/InGaAs DHBT distrib. amps., modulator drivers for 80-Gbit/s operation, comp. Schneider, K., + , T-MTT Nov 05 3378-3387 LNA protection, watt-level CMOS transceivers, reson. switch. Kuhn, W.B., + , T-MTT Sep 05 2819-2825 noise behavior of microwave amplifiers operating under nonlinear conditions. Escotte, L., + , T-MTT Dec 05 3704-3711 RF front-end characts., spectral regrowth of communs. sigs., impact. Gard, K.G., + , T-MTT Jun 05 2179-2186 very low-noise differential radiometer, 30 GHz for PLANCK LFI. Aja, B., + , T-MTT Jun 05 2050-2062 Microwave amplifiers; cf. Microwave power amplifiers; MMIC amplifiers Microwave antenna arrays PLL-based retrodirective array, anal. and charactn. Fusco, V., + , T-MTT Feb 05 730-738

IEEE T-MTT 2005 INDEX — 44 Microwave antennas protein thermal unfolding and refolding, near-zone microwaves, ultrasensitive detect. Taylor, K.M., + , T-MTT May 05 1576-1586 Microwave antennas; cf. Lens antennas; Microwave antenna arrays Microwave bipolar integrated circuits broad-band power amp., tunable output matching net. Haitao Zhang, + , T-MTT Nov 05 3606-3614 DVB-S appls., Si bipolar technol., monolithic 12-GHz heterodyne receiver. Girlando, G., + , T-MTT Mar 05 952-959 Microwave bipolar transistors InP-InGaAs HBTs, microwave noise modeling. Escotte, L., + , T-MTT Jan 05 415-416 InP-InGaAs HBTs ), microwave noise modeling. Jianjun Gao, + , T-MTT Jan 05 417 Microwave circuits 4-port microwave nets., intrinsic broad-band suppression of commonmode sigs. Fathelbab, W.M., + , T-MTT May 05 1569-1575 compact EM-bandgap (EBG) struct. and appls. for microwave ccts. Li Yang, + , T-MTT Jan 05 183-190 general lin. lumped microwave ccts., matrix theory, improved FDTD formulation. Zhenhai Shao, + , T-MTT Jul 05 2261-2266 Green's fns. for cylindrical enclosures by spatial images method, num. eval. Pereira, F.D.Q., + , T-MTT Jan 05 94-105 high aspect ratio through-wafer interconnect vias, Si substrs., microwave charactn. and modeling. Leung, L.L.W., + , T-MTT Aug 05 2472-2480 lin. inverse space-mapping (LISM) algm., design lin./nonlin. RF and microwave ccts. Rayas-Sanchez, J.E., + , T-MTT Mar 05 960-968 local ground plane, EM anal., deembedding effect. Rautio, J.C., T-MTT Feb 05 770-776 parallel FFT accelerated transient field-cct. simulator. Yilmaz, A.E., + , TMTT Sep 05 2851-2865 ring-hybrid microwave voltage-variable attenuator, HFET transistors. Saavedra, C.E., + , T-MTT Jul 05 2430-2434 software-defined direct conversion receiver, ka-band analog front-end. Tatu, S.O., + , T-MTT Sep 05 2768-2776 wide-band supercond. coplanar delay lines. Yi Wang, + , T-MTT Jul 05 2348-2354 Microwave circuits; cf. Microwave integrated circuits Microwave communication highly efficient Doherty feedforward lin. power amp. for W-CDMA basestation appls. Kyoung-Joon Cho, + , T-MTT Jan 05 292-300 Microwave detectors detect. and localization of mobile phones, large buildings, microwave syst. Hudec, P., + , T-MTT Jun 05 2235-2239 terahertz mixers and detectors, design guidelines. Focardi, P., + , T-MTT May 05 1653-1661 very low-noise differential radiometer, 30 GHz for PLANCK LFI. Aja, B., + , T-MTT Jun 05 2050-2062 Microwave devices 16th Asia-Pacific Microwave Conference, APMC'04 (special section). TMTT Sep 05 2649-2731 16th Asia-Pacific Microwave Conference, APMC'04 (special section intro.). Steer, M.B., T-MTT Sep 05 2649 2005 International Microwave Symposium, MTT-S05 (special issue). TMTT Nov 05 3264-3678 2005 International Microwave Symposium, MTT-S05 (special issue intro.). Jackson, C., T-MTT Nov 05 3264 34th European Microwave Conference, EuMC (special issue). T-MTT Jun 05 1935-2258 34th European Microwave Conference, EuMC (special issue intro.). Russer, P., T-MTT Jun 05 1935-1936 complex pass. devices composed of arbitrarily shaped waveguides, Nystrom and BI-RME methods, CAD. Taroncher, M., + , T-MTT Jun 05 2153-2163 efficient anal. formulation and sensitivity anal. of neuro-space mapping for nonlin. microwave device modeling. Lei Zhang, + , T-MTT Sep 05 2752-2767 even-harmonic C-band modulator. Di Alessio, F.L., + , T-MTT Apr 05 1203-1210 in-line dual- and triple-mode cavity filters, nonresonating nodes. Amari, S., + , T-MTT Apr 05 1272-1279 large-sig. scatt. fns., linearization. Verspecht, J., + , T-MTT Apr 05 13691376

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LV and low-power RF MEMS series and shunt switches actuated by combination of EM and electrostatic forces. Il-Joo Cho, + , T-MTT Jul 05 2450-2457 Microwave devices; cf. Magnetic microwave devices; Microwave amplifiers; Microwave antennas; Microwave circuits; Microwave detectors; Microwave diodes; Microwave filters; Microwave frequency converters; Microwave mixers; Microwave oscillators; Microwave phase shifters; Microwave receivers; Microwave transistors; Superconducting microwave devices Microwave diodes 60-GHz monolithic star mixer, gate-drain-connected pHEMT diodes. Kyung-Whan Yeom, + , T-MTT Jul 05 2435-2440 integr. Si Schottky mixer diodes, cutoff freqs. above 1 THz. Morschbach, M., + , T-MTT Jun 05 2013-2018 Microwave FET integrated circuits on-chip vert. solenoid inductor design for multigigahertz CMOS RFIC. Hau-Yiu Tsui, + , T-MTT Jun 05 1883-1890 Microwave FETs FET self-heating, broad-band charactn. Parker, A.E., + , T-MTT Jul 05 2424-2429 highly efficient Doherty feedforward lin. power amp. for W-CDMA basestation appls. Kyoung-Joon Cho, + , T-MTT Jan 05 292-300 hybrid digital/RF envelope predistortion linearization syst. for power amps. Wangmyong Woo, + , T-MTT Jan 05 229-237 presence of LF dispers. effects, accurate pHEMT nonlin. modeling. Raffo, A., + , T-MTT Nov 05 3449-3459 ring-hybrid microwave voltage-variable attenuator, HFET transistors. Saavedra, C.E., + , T-MTT Jul 05 2430-2434 short-channel AlGaN/GaN heterojunction FETs, 30-GHz-band, 5-W power perform. Inoue, T., + , T-MTT Jan 05 74-80 Microwave filters 12-18-GHz 3-pole RF MEMS tunable filter. Entesari, K., + , T-MTT Aug 05 2566-2571 adaptive models and param. extr., seq. tuning. Pepe, G., + , T-MTT Jan 05 22-31 compact bandpass filter, 2 tuning transm. zeros, CMRC resonator. Kam Man Shum, + , T-MTT Mar 05 895-900 compact planar and vialess composite low-pass filters, folded steppedimpedance resonator, liq.-Crystal-polymer substr. Pinel, S., + , T-MTT May 05 1707-1712 compact super-wide bandpass substr. integr. waveguide (SIW) filters. Zhang-Cheng Hao, + , T-MTT Sep 05 2968-2977 dielec.-filled cavity filters, ultrawide stopband Characteristics, design. Rauscher, C., T-MTT May 05 1777-1786 differential 4-bit 6.5-10-GHz RF MEMS tunable filter. Entesari, K., + , TMTT Mar 05 1103-1110 direct-coupled microwave filters, single and dual stopbands. Cameron, R.J., + , T-MTT Nov 05 3288-3297 dual-passband filters, design techs. Macchiarella, G., + , T-MTT Nov 05 3265-3271 future mobile telecomm. systs., high-temp. supercond. filter. Jia-Sheng Hong, + , T-MTT Jun 05 1976-1981 hybrid narrow-band tunable bandpass filter, varactor loaded EM-bandgap CPW. Pistono, E., + , T-MTT Aug 05 2506-2514 improved stopband, sub-wavel. resonators. Garcia-Garcia, J., + , T-MTT Jun 05 1997-2006 low-loss 2-bit tunable bandpass filters, MEMS DC contact switches. Pothier, A., + , T-MTT Jan 05 354-360 LTCC, metallic resonators, canonical ridge waveguide filters. Ruiz-Cruz, J.A., + , T-MTT Jan 05 174-182 miniaturized microwave pass. filter incorporating multilayer synthetic quasiTEM transm. line. Hsien-Shun Wu, + , T-MTT Sep 05 2713-2720 miniaturized multilayer quasiellipt. bandpass filter, aperture-coupled microstrip resonators. Chi-Feng Chen, + , T-MTT Sep 05 2688-2692 miniaturized spurious passband suppression microstrip filter, meandered parallel coupled lines. Shih-Ming Wang, + , T-MTT Feb 05 747-753 RF inductors and filters, LCP substrs. for Wi-Fi appls., layout-level synthesis. Mukherjee, S., + , T-MTT Jun 05 2196-2210 RF/microwave multifunctional systs., reconfigurable bandpass filter. Fathelbab, W.M., + , T-MTT Mar 05 1111-1116 serial config., 2 finite transm. zeros, LTCC technol., bandpass filter. Chun-Fu Chang, + , T-MTT Jul 05 2383-2388 split-ring resonators and complementary split-ring resonators coupled, planar transm. lines, equiv.-cct. models. Baena, J.D., + , T-MTT Apr 05 1451-1461

IEEE T-MTT 2005 INDEX — 45 supercond. spiral wide bandpass filters, wide upper stopband. Huang, F., T-MTT Jul 05 2335-2339 symmetric composite right/left-handed CPW, appls., compact bandpass filters, modeling. Shau-Gang Mao, + , T-MTT Nov 05 3460-3466 synthesis of cascaded N-tuplets cross-coupled resonators microwave filters, matrix rotations, anal. tech. Tamiazzo, S., + , T-MTT May 05 1693-1698 WDM photonic microwave filters, random errors, stat. anal. Vidal, B., + , T-MTT Aug 05 2600-2603 Microwave frequency converters RF frequency shifting via optically switched dual-channel PZT fiber stretchers. McDermitt, C.S., + , T-MTT Dec 05 3782-3787 Microwave frequency converters; cf. MMIC frequency converters Microwave imaging breast cancer detect., expt. investig. of simple tumor models, tissue sens. adaptive radar. Sill, J.M., + , T-MTT Nov 05 3312-3319 high dielec.-contrast objs., different image-reconstruction approaches, microwave-tomographic imaging. Semenov, S.Y., + , T-MTT Jul 05 2284-2294 particle swarm optimizer for microwave imaging of 2D dielec. scatterers, comput. approach. Donelli, M., + , T-MTT May 05 1761-1776 time reversal, FDTD method for microwave breast cancer detect. Kosmas, P., + , T-MTT Jul 05 2317-2323 vector NWA for microwave imaging, effective usage. Chao-Hsiung Tseng, + , T-MTT Sep 05 2884-2891 Microwave integrated circuits 12-GHz SiGe phase shifter, integr. LNA. Hancock, T.M., + , T-MTT Mar 05 977-983 accurate and scalable RF interconnect model for Si-based RFIC appls. Choon Beng Sia, + , T-MTT Sep 05 3035-3044 act. differential broad-band phase splitter for quadrature-modulator appls. Tiiliharju, E., + , T-MTT Feb 05 679-686 broad-band poly-harmonic distortion (PHD) behavioral models from fast automated simul. and large-sig. vectorial net. meas. Root, D.E., + , TMTT Nov 05 3656-3664 CMOS technol., complementary Colpitts oscillator. Choong-Yul Cha, + , T-MTT Mar 05 881-887 compact intell. RF front-end, reconfigurable RFICs, Si-based technols. Mukhopadhyay, R., + , T-MTT Jan 05 81-93 coupled-slotline-hybrid sampling mixer integr., step-recovery-diode pulse generator for UWB appls. Jeongwoo Han, + , T-MTT Jun 05 1875-1882 fast layout verification of 3D RF and mixed-sig. on-chip structs., largescale broad-band parasitic extr. Feng Ling, + , T-MTT Jan 05 264-273 gain compress. curve, IIP3 estim. Choongeol Cho, + , T-MTT Apr 05 1197-1202 image-rejection CMOS LNA design optim. techs. Trung-Kien Nguyen, + , T-MTT Feb 05 538-547 InGaP/GaAs HBT RF power amps., ESD protection design considerations. Ma, Y., + , T-MTT Jan 05 221-228 inter-chip RF-interconnect, CPW, capacitive coupler, UWB transceiver, perform. Sun, M., + , T-MTT Sep 05 2650-2655 modified ring dielec. resonator, improved mode separation and tunability characts., MIC environ. Srivastava, K.V., + , T-MTT Jun 05 1960-1967 planar microwave components, EM lin. regression models, empirical model gener. techs. Domenech-Asensi, G., + , T-MTT Nov 05 3305-3311 RF switch matrix appls., integr. interconnect nets. Daneshmand, M., + , TMTT Jan 05 12-21 Microwave integrated circuits; cf. MMIC Microwave materials 34th European Microwave Conference, EuMC (special issue). T-MTT Jun 05 1935-2258 34th European Microwave Conference, EuMC (special issue intro.). Russer, P., T-MTT Jun 05 1935-1936 Microwave measurement; cf. Microwave reflectometry Microwave measurements complex permitt. meas., TE11p modes, circ. cylindrical cavities. Zinal, S., + , T-MTT Jun 05 1870-1874 EM field-mapping syst., reson.-suppressed mag. field probe. Jung-Min Kim, + , T-MTT Sep 05 2693-2699 ferroelec. thin-film materials, microwave charactn. Ouaddari, M., + , TMTT Apr 05 1390-1397 liq. crysts., temp.-controlled coaxial transm. line, broad-band microwave charactn. Mueller, S., + , T-MTT Jun 05 1937-1945

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low-cost planar probes, broadside apertures for nondestructive dielec. meas. of biol. materials, microwave freqs. Byoungjoong Kang, + , TMTT Jan 05 134-143 low-noise multiresolution high-dyn. ultra-broad-band time-domain EMI meas. syst. Braun, S., + , T-MTT Nov 05 3354-3363 mag. thin-film charactn., microwave range, broad-band method. Vincent, D., + , T-MTT Apr 05 1174-1180 meas. of scatt. params. of N-ports, multiport method. Rolfes, I., + , T-MTT Jun 05 1990-1996 method, improve VNA calib., planar dispers. media, adding, asymmetrical reciprocal device. Scott, J.B., T-MTT Sep 05 3007-3013 on-wafer charactn. of deep-submicrometer Si MOSFETs, shield-based 3port de-embedding method. Ming-Hsiang Cho, + , T-MTT Sep 05 29262934 on-wafer noise-param. meas., wide-band freq.-var. method. Hu, R., + , TMTT Jul 05 2398-2402 pad-open-short and open-short-load deembedding techs. for accurate onwafer RF charactn. of high-quality passives, comp. Tiemeijer, L.F., + , T-MTT Feb 05 723-729 protein thermal unfolding and refolding, near-zone microwaves, ultrasensitive detect. Taylor, K.M., + , T-MTT May 05 1576-1586 vector NWA for microwave imaging, effective usage. Chao-Hsiung Tseng, + , T-MTT Sep 05 2884-2891 wide-band on-wafer noise-param. meas., improved Y-factor method. Tiemeijer, L.F., + , T-MTT Sep 05 2917-2925 Microwave mixers 90-nm VLSI SOI CMOS technol., high linearity for WLAN, 26.5-30-GHz resistive mixer. Ellinger, F., T-MTT Aug 05 2559-2565 CMOS RF amp. and mixer ccts. utilizing complementary Characteristics of parallel combined NMOS and PMOS devices. Nam, I., + , T-MTT May 05 1662-1671 coupled-slotline-hybrid sampling mixer integr., step-recovery-diode pulse generator for UWB appls. Jeongwoo Han, + , T-MTT Jun 05 1875-1882 PLL-based retrodirective array, anal. and charactn. Fusco, V., + , T-MTT Feb 05 730-738 Microwave mixers; cf. MMIC mixers Microwave oscillators additive and converted noise, gener. of phase noise, nonlin. oscillators, role. Nallatamby, J.-C., + , T-MTT Mar 05 901-906 CMOS technol., complementary Colpitts oscillator. Choong-Yul Cha, + , T-MTT Mar 05 881-887 cylindrical multilayered ceramic resonators, rect. air cavity for low-phase noise K/Ka-band oscillators. El-Tager, A.M., + , T-MTT Jun 05 22112219 low-power ka-band Voltage-controlled oscillator implemented, 200-GHz SiGe HBT technol. Yi-jan Emery Chen, + , T-MTT May 05 1672-1681 push-push/triple-push oscillators for reducing 1/f noise upconversion, design. Jonghoon Choi, + , T-MTT Nov 05 3407-3414 stabil. ccts. for phase-noise reduction, microwave oscillators. Suarez, A., + , T-MTT Sep 05 2743-2751 Microwave oscillators; cf. Microwave parametric oscillators Microwave parametric devices; cf. Microwave parametric oscillators Microwave parametric oscillators model for the mode-splitting effect in whispering-gallery-mode resonators. Bourgeois, P.-Y., + , T-MTT Oct 05 3185-3190 Microwave phase shifters 12-GHz SiGe phase shifter, integr. LNA. Hancock, T.M., + , T-MTT Mar 05 977-983 adaptive duplexer implemented, single-path/multipath feedforward techs., BST phase shifters. O'Sullivan, T., + , T-MTT Jan 05 106-114 microstrip-line-type PDLC loaded variable phase shifter, increasing speed. Utsumi, Y., + , T-MTT Nov 05 3345-3353 planar microwave components, EM lin. regression models, empirical model gener. techs. Domenech-Asensi, G., + , T-MTT Nov 05 3305-3311 self-calibrating SSB modulator. Treyer, D.M., + , T-MTT Dec 05 38063816 shifter utilizing nonreciprocal wave propag. Davis, L.E., T-MTT Jan 05 414 shifter utilizing nonreciprocal wave propag. ). How, H., + , T-MTT Jan 05 414 Microwave power amplifiers 3rd.- and fifth-order baseband component injection for linearization of power amp., cellular phone. Mizusawa, N., + , T-MTT Nov 05 33273334

IEEE T-MTT 2005 INDEX — 46 expt. class-F power amp. design, computationally efficient and accurate large-sig. pHEMT model. Wren, M., + , T-MTT May 05 1723-1731 high-effic. current-mode class-D amps. for wireless handsets, design. TsaiPi Hung, + , T-MTT Jan 05 144-151 high-effic. multistage Doherty power amp. for wireless communs., anal. and design. Srirattana, N., + , T-MTT Mar 05 852-860 highly efficient Doherty feedforward lin. power amp. for W-CDMA basestation appls. Kyoung-Joon Cho, + , T-MTT Jan 05 292-300 hybrid digital/RF envelope predistortion linearization syst. for power amps. Wangmyong Woo, + , T-MTT Jan 05 229-237 InGaP/GaAs HBT RF power amps., ESD protection design considerations. Ma, Y., + , T-MTT Jan 05 221-228 intelligently controlled RF power amp., reconfigurable MEMS-varactor tuner. Dongjiang Qiao, + , T-MTT Mar 05 1089-1095 microwave and wireless power-amp. behavioral modeling approaches, comparative overview. Pedro, J.C., + , T-MTT Apr 05 1150-1163 power amps., RBF neural nets, wide-band dyn. modeling. Isaksson, M., + , T-MTT Nov 05 3422-3428 wireless transmitters, digital subband filtering predistorter archit. Hammi, O., + , T-MTT May 05 1643-1652 x-band 2-stage high-effic. switched-mode power amps. Pajic, S., + , TMTT Sep 05 2899-2907 X-band class-E power amps., EER operation, linearity. Narisi Wang, + , T-MTT Mar 05 1096-1102 Microwave power amplifiers; cf. MMIC power amplifiers Microwave propagation CPW, subterahertz freqs., atten. characts. Jingjing Zhang, + , T-MTT Nov 05 3281-3287 ferrite Faraday rotators, impedance matching considerations. Boyd, C.R., Jr., T-MTT Jul 05 2371-2374 Microwave receivers DVB-S appls., Si bipolar technol., monolithic 12-GHz heterodyne receiver. Girlando, G., + , T-MTT Mar 05 952-959 RF-CMOS and SiGe BiCMOS, WCDMA direct-conversion receiver front-end comp. Floyd, B.A., + , T-MTT Apr 05 1181-1188 software-defined direct conversion receiver, ka-band analog front-end. Tatu, S.O., + , T-MTT Sep 05 2768-2776 Microwave reflectometry moisture content, bulk materials, sub-nanosecond UWB pulses, noncontacting determ. Schimmer, O., + , T-MTT Jun 05 2107-2113 Microwaves metamaterial structures, phenomena, and applications (special issue). TMTT Apr 05 1413-1556 metamaterial structures, phenomena, and applications (special issue intro.). Itoh, T., + , T-MTT Apr 05 1413-1417 Microwave spectroscopy in vivo and ex dielec. spectrosc. of biol. tissues, microwave freqs., precision open-ended coaxial probes. Popovic, D., + , T-MTT May 05 1713-1722 Microwave technology 2004 IEEE MTT-S International Microwave Symposium (special section). T-MTT Jan 05 3-413 2004 IEEE MTT-S International Microwave Symposium (special section intro.). Steer, M., T-MTT Jan 05 3 Microwave technology; cf. Microwave imaging; Microwave propagation Microwave transistors on-wafer noise-param. meas., wide-band freq.-var. method. Hu, R., + , TMTT Jul 05 2398-2402 time-domain large-sig. meas., table-based nonlin. HEMT model extracted. Curras-Francos, M.C., T-MTT May 05 1593-1600 Microwave transistors; cf. Microwave bipolar transistors Military communication advanced multifunction RF concept. Tavik, G.C., + , T-MTT Mar 05 10091020 AMRFC test-bed, high-band digital preprocessor (HBDP). Mazumder, S., + , T-MTT Mar 05 1065-1071 armored vehicle appl., multifunction mm-wave systs. Wehling, J.H., TMTT Mar 05 1021-1025 Military systems; cf. Military communication Millimeter wave amplifiers short stub-matching 77-GHz-band driver amp., attenuator compensating temp. depend. of gain. Chaki, S., + , T-MTT Jun 05 2073-2081 Millimeter wave amplifiers; cf. Millimeter wave power amplifiers

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Millimeter wave antenna arrays mm-wave high-effic. multilayer parasitic microstrip antenna array, teflon substr. Seki, T., + , T-MTT Jun 05 2101-2106 Millimeter wave antennas armored vehicle appl., multifunction mm-wave systs. Wehling, J.H., TMTT Mar 05 1021-1025 compact dual-polarized multibeam phased-array archit. for mm-wave radar. Schulwitz, L., + , T-MTT Nov 05 3588-3594 Millimeter wave antennas; cf. Millimeter wave antenna arrays Millimeter wave bipolar integrated circuits InP DHBT technol., DC-100-GHz freq. doublers. Puyal, V., + , T-MTT Apr 05 1338-1344 mm-wave (Bi)CMOS IC, 30-100-GHz inductors and transformers. Dickson, T.O., + , T-MTT Jan 05 123-133 wide-band SiGe HBT act. mixers, anal. and design. Johansen, T.K., + , TMTT Jul 05 2389-2397 Millimeter wave bipolar transistors g-band (140-220 GHz) and W-band (75-110 GHz) InP DHBT medium power amps. Paidi, V.K., + , T-MTT Feb 05 598-605 Millimeter wave circuits; cf. Millimeter wave integrated circuits Millimeter wave circulators unbiased integr. microstrip circulator, mag. nanowired substr. Saib, A., + , T-MTT Jun 05 2043-2049 Millimeter wave couplers; cf. Millimeter wave directional couplers Millimeter wave devices compact and broad-band millimeter-wave monolithic transformer balanced mixers. Wu, P.-S., + , T-MTT Oct 05 3106-3114 effect of cable length in vector measurements of very long millimeter wave components. Simonetto, A., + , T-MTT Dec 05 3731-3734 high-power MEMS varactors and impedance tuners for mm-wave appls. Lu, Y., + , T-MTT Nov 05 3672-3678 k-band orthomode transducer, waveguide ports and balanced coaxial probes. Engargiola, G., + , T-MTT May 05 1792-1801 mm-wave appls., extremely high-Q factor dielec. resonators. Krupka, J., + , T-MTT Feb 05 702-712 multifunctional RF systems (special issue). T-MTT Mar 05 1005-1116 multifunctional RF systems (special issue intro.). Adler, E.D., + , T-MTT Mar 05 1005-1008 wide-band continuously tunable millimeter-wave signal with an optical external modulation technique. Qi, G., + , T-MTT Oct 05 3090-3097 Millimeter wave devices; cf. Millimeter wave amplifiers; Millimeter wave antennas; Millimeter wave circulators; Millimeter wave diodes; Millimeter wave directional couplers; Millimeter wave filters; Millimeter wave mixers; Millimeter wave oscillators; Millimeter wave phase shifters; Millimeter wave transistors Millimeter wave diodes std. CMOS technol., monolithic integr. mm-wave IMPATT transmitter. Al-Attar, T., + , T-MTT Nov 05 3557-3561 Millimeter wave directional couplers CW 250-GHz gyrotron DNP expts., corrugated waveguide and directional coupler. Woskov, P.P., + , T-MTT Jun 05 1863-1869 Millimeter wave FET integrated circuits 60-GHz broad-band telecomm., resistive HEMT mixers. Varonen, M., + , T-MTT Apr 05 1322-1330 Millimeter wave filters anal. of NRD-guide/H-guide mm-wave ccts., order-reduced vol.-integral eqn. approach. Duochuan Li, + , T-MTT Mar 05 799-812 E-plane filters and diplexers, ellipt. response for mm-wave appls. Ofli, E., + , T-MTT Mar 05 843-851 Millimeter wave imaging radiometric Millimeter-wave detect. via opt. upconversion and carrier suppression. Schuetz, C.A., + , T-MTT May 05 1732-1738 self-collimation, low-index-contrast photonic crysts., mm-wave regime, expt. demons. Zhaolin Lu, + , T-MTT Apr 05 1362-1368 stereoscopic pass. mm-wave imaging/ranging. Luthi, T., + , T-MTT Aug 05 2594-2599 Millimeter wave integrated circuits bridged NRD-guide coupler for mm-wave appls., anal. and design. Duochuan Li, + , T-MTT Aug 05 2546-2551 highly integr. mm-wave pass. components, 3D LTCC syst.-on-package (SOP) technol. Jong-Hoon Lee, + , T-MTT Jun 05 2220-2229 Millimeter wave measurements calibrated 200-GHz waveform meas. Williams, D.F., + , T-MTT Apr 05 1384-1389

IEEE T-MTT 2005 INDEX — 47 on-wafer meas. of differential ccts., mm-wave freqs., pure-mode NWA concept. Zwick, T., + , T-MTT Mar 05 934-937 presence of LF dispers. effects, accurate pHEMT nonlin. modeling. Raffo, A., + , T-MTT Nov 05 3449-3459 Millimeter wave mixers 60-GHz broad-band telecomm., resistive HEMT mixers. Varonen, M., + , T-MTT Apr 05 1322-1330 InP HEMTs and their appls., mm-wave radio-on-fiber systs., phototransistors. Chang-Soon Choi, + , T-MTT Jan 05 256-263 mm-wave (Bi)CMOS IC, 30-100-GHz inductors and transformers. Dickson, T.O., + , T-MTT Jan 05 123-133 wide-band SiGe HBT act. mixers, anal. and design. Johansen, T.K., + , TMTT Jul 05 2389-2397 Millimeter wave oscillators 60-GHz broad-band telecomm., resistive HEMT mixers. Varonen, M., + , T-MTT Apr 05 1322-1330 mm-wave (Bi)CMOS IC, 30-100-GHz inductors and transformers. Dickson, T.O., + , T-MTT Jan 05 123-133 Millimeter wave phase shifters broad-band 180° phase shifters, integr. submillimeter-wave Schottky diodes. Zhiyang Liu, + , T-MTT Sep 05 2949-2955 compact dual-polarized multibeam phased-array archit. for mm-wave radar. Schulwitz, L., + , T-MTT Nov 05 3588-3594 nonuniform plasma layer model for quartz-Si image guide phase shifters, expt. verification. Fickenscher, T., + , T-MTT Jul 05 2375-2382 Millimeter wave power amplifiers g-band (140-220 GHz) and W-band (75-110 GHz) InP DHBT medium power amps. Paidi, V.K., + , T-MTT Feb 05 598-605 Millimeter wave propagation 60-GHz broad-band telecomm., resistive HEMT mixers. Varonen, M., + , T-MTT Apr 05 1322-1330 self-collimation, low-index-contrast photonic crysts., mm-wave regime, expt. demons. Zhaolin Lu, + , T-MTT Apr 05 1362-1368 Millimeter waves frequency channel blocking scheme in mesh-topology millimeter-wave broad band entrance networks. Sangiamwong, J., + , T-MTT Dec 05 3723-3730 low-loss LTCC cavity filters using system-on-package technology at 60 GHz. Lee, J.-H., + , T-MTT Dec 05 3817-3824 millimeter-wave substrate integrated waveguides and filters in photoimageable thick-film technology. Stephens, D., + , T-MTT Dec 05 3832-3838 theory of chained-function filters. Chrisostomidis, C.E., + , T-MTT Oct 05 3142-3151 Millimeter wave technology mm-wave CMOS cct. design. Shigematsu, H., + , T-MTT Feb 05 472-477 Millimeter wave transistors 70-ps SiGe differential RF switch, design and anal. Hancock, T.M., + , TMTT Jul 05 2403-2410 Millimeter wave transistors; cf. Millimeter wave bipolar transistors MIMICs 60-GHz broad-band telecomm., resistive HEMT mixers. Varonen, M., + , T-MTT Apr 05 1322-1330 broad-band 180° phase shifters, integr. submillimeter-wave Schottky diodes. Zhiyang Liu, + , T-MTT Sep 05 2949-2955 gen. surface-vol. integral-eqn. (SVIE) approach for anal. of hybrid planar/NRD-guide IC. Duochuan Li, + , T-MTT Sep 05 2732-2742 InP DHBT technol., DC-100-GHz freq. doublers. Puyal, V., + , T-MTT Apr 05 1338-1344 k- and Q-bands CMOS freq. sources, X-band quadrature VCO. Sangsoo Ko, + , T-MTT Sep 05 2789-2800 mm-wave (Bi)CMOS IC, 30-100-GHz inductors and transformers. Dickson, T.O., + , T-MTT Jan 05 123-133 wide-band SiGe HBT act. mixers, anal. and design. Johansen, T.K., + , TMTT Jul 05 2389-2397 MIMO systems increasing wireless channel capacity, MIMO systs. employing colocated antennas. Konanur, A.S., + , T-MTT Jun 05 1837-1844 Minerals; cf. Diamond; Quartz Minimum shift keying 0.25-ȝm CMOS OPLL transmitter IC for GSM and DCS appls. Peng-Un Su, T-MTT Feb 05 462-471 Mirrors gyrotron phase-correcting mirrors, irradiance moments, synthesis. Shapiro, M.A., + , T-MTT Aug 05 2610-2615 + Check author entry for coauthors

MIS capacitors; cf. MOS capacitors MISFETs power level-depend. dual-operating mode LDMOS power amp. for CDMA wireless base-station appls. Younkyu Chung, + , T-MTT Feb 05 739-746 Mixed analog-digital integrated circuits fast layout verification of 3D RF and mixed-sig. on-chip structs., largescale broad-band parasitic extr. Feng Ling, + , T-MTT Jan 05 264-273 InP/InGaAs DHBT distrib. amps., modulator drivers for 80-Gbit/s operation, comp. Schneider, K., + , T-MTT Nov 05 3378-3387 Mixers microwave mixers, scatt.-param. models and representations. Williams, D.F., + , T-MTT Jan 05 314-321 terahertz mixers and detectors, design guidelines. Focardi, P., + , T-MTT May 05 1653-1661 Mixers (circuits); cf. Microwave mixers; Millimeter wave mixers; Schottky diode mixers; Submillimeter wave mixers; UHF mixers MMIC broad-band power amp., tunable output matching net. Haitao Zhang, + , T-MTT Nov 05 3606-3614 compact and broad-band millimeter-wave monolithic transformer balanced mixers. Wu, P.-S., + , T-MTT Oct 05 3106-3114 compact bandpass filter, 2 tuning transm. zeros, CMRC resonator. Kam Man Shum, + , T-MTT Mar 05 895-900 compact MMIC CPW and asymmetric CPS branch-line couplers and Wilkinson dividers, shunt and series stub loading. Hettak, K., + , T-MTT May 05 1624-1635 coupled slotline mode, finite-ground CPW, unequal ground-plane widths, excit. Ponchak, G.E., + , T-MTT Feb 05 713-717 distrib. preamplifier, cascode FET cells, improved noise anal. Won Ko, + , T-MTT Jan 05 361-371 DVB-S appls., Si bipolar technol., monolithic 12-GHz heterodyne receiver. Girlando, G., + , T-MTT Mar 05 952-959 freq. multiplier, waveguide-based spatial power-combining archit. Belaid, M., + , T-MTT Apr 05 1124-1129 high-power high-effic. SiGe Ku- and Ka-band balanced freq. doublers. Juo-Jung Hung, + , T-MTT Feb 05 754-761 k- and Q-bands CMOS freq. sources, X-band quadrature VCO. Sangsoo Ko, + , T-MTT Sep 05 2789-2800 low-loss TFMS, Si substr. up, 220 GHz, fab. and charactn. Six, G., + , TMTT Jan 05 301-305 lumped-element quadrature power splitters, mixed right/left-handed transm. lines. Kuylenstierna, D., + , T-MTT Aug 05 2616-2621 microstrip-line struct. employing, periodically perforated ground metal and appl., highly miniaturized and low-impedance pass. components fabricated. Young Yun, T-MTT Jun 05 1951-1959 multiple-metal stacked inductors incorporating, extended phys. model, design. Murphy, O.H., + , T-MTT Jun 05 2063-2072 on-chip vert. solenoid inductor design for multigigahertz CMOS RFIC. Hau-Yiu Tsui, + , T-MTT Jun 05 1883-1890 RF sect. of UHF and microwave pass. RFID transponders, design criteria. De Vita, G., + , T-MTT Sep 05 2978-2990 std. CMOS technol., monolithic integr. mm-wave IMPATT transmitter. Al-Attar, T., + , T-MTT Nov 05 3557-3561 surface-passivated high-resist. Si, true microwave substr. Spirito, M., + , T-MTT Jul 05 2340-2347 thermal resist. of FETs, accurate determ. Darwish, A.M., + , T-MTT Jan 05 306-313 thin-film microstrip and coplanar technols. for reduced-size MMICs, integrat. Hettak, K., + , T-MTT Jan 05 283-291 ultra-wideband low-cost phased-array radars. Rodenbeck, C.T., + , T-MTT Dec 05 3697-3703 MMIC; cf. MMIC amplifiers; MMIC frequency converters; MMIC mixers MMIC amplifiers 2.17-dB NF 5-GHz-band monolithic CMOS LNA, 10-mW DC power consumption. Hung-Wei Chiu, + , T-MTT Mar 05 813-824 40-Gb/s wide-band MMIC pHEMT modulator driver amps. designed, real freq. tech. Kerherve, E., + , T-MTT Jun 05 2145-2152 drain-line loss and the S22 kink effect in capacitively coupled distributed amplifiers. Shohat, J., + , T-MTT Dec 05 3767-3773 modified loss-compensation method, 0.35-ȝm SiGe BiCMOS technol., broad-band MMICs. Ming-Da Tsai, + , T-MTT Feb 05 496-505 packaged inductively degenerated common-source low-noise amps., ESD protection, anal. and optim. Sivonen, P., + , T-MTT Apr 05 1304-1313

IEEE T-MTT 2005 INDEX — 48 stabil. of microwave amps., variable termination impedances, necessary and sufficient conds. Olivieri, M., + , T-MTT Aug 05 2580-2586 MMIC amplifiers; cf. MMIC power amplifiers MMIC frequency converters InP DHBT technol., DC-100-GHz freq. doublers. Puyal, V., + , T-MTT Apr 05 1338-1344 sig. gener., control, freq. conversion AlGaN/GaN HEMT MMICs. Kaper, V.S., + , T-MTT Jan 05 55-65 v-band high-order harmonic injection-locked freq.-divider MMICs, wide bandwidth and low-power dissipation. Jinho Jeong, + , T-MTT Jun 05 1891-1898 MMIC mixers 60-GHz monolithic star mixer, gate-drain-connected pHEMT diodes. Kyung-Whan Yeom, + , T-MTT Jul 05 2435-2440 integr. Si Schottky mixer diodes, cutoff freqs. above 1 THz. Morschbach, M., + , T-MTT Jun 05 2013-2018 modified loss-compensation method, 0.35-ȝm SiGe BiCMOS technol., broad-band MMICs. Ming-Da Tsai, + , T-MTT Feb 05 496-505 sig. gener., control, freq. conversion AlGaN/GaN HEMT MMICs. Kaper, V.S., + , T-MTT Jan 05 55-65 MMIC power amplifiers act. predistorter suitable for MMIC implement. Iommi, R., + , T-MTT Mar 05 874-880 broad-band power amp., tunable output matching net. Haitao Zhang, + , T-MTT Nov 05 3606-3614 g-band (140-220 GHz) and W-band (75-110 GHz) InP DHBT medium power amps. Paidi, V.K., + , T-MTT Feb 05 598-605 handset power amp., high effic., low level, load-modulation tech. Joongjin Nam, + , T-MTT Aug 05 2639-2644 multimode J-pHEMT front-end archit., power-control scheme for max. effic. Clifton, J.C., + , T-MTT Jun 05 2251-2258 presence of LF dispers. effects, accurate pHEMT nonlin. modeling. Raffo, A., + , T-MTT Nov 05 3449-3459 RF-driven gate current, DC/RF perform., GaAs pHEMT MMIC power amps., effect. Yeong-Chang Chou, + , T-MTT Nov 05 3398-3406 Mobile antennas LINC radio transmitters, integr. antenna/power combiner. Gao, S., + , TMTT Mar 05 1083-1088 Mobile communication 0.25-ȝm CMOS OPLL transmitter IC for GSM and DCS appls. Peng-Un Su, T-MTT Feb 05 462-471 15th International Conference on Microwaves, Radar, and Wireless Communications, MIKON (special section). T-MTT Feb 05 425-450 15th International Conference on Microwaves, Radar, and Wireless Communications, MIKON (special section intro.). Modelski, J.W., TMTT Feb 05 425-426 baseband-modeled CALLUM archits., spectrum emission considerations. Strandberg, R., + , T-MTT Feb 05 660-669 dual-passband filters, design techs. Macchiarella, G., + , T-MTT Nov 05 3265-3271 GSM/EGSM/DCS/PCS direct conversion receiver, integr. synthesizer. Young-Jin Kim, + , T-MTT Feb 05 606-613 LINC radio transmitters, integr. antenna/power combiner. Gao, S., + , TMTT Mar 05 1083-1088 multifunctional RF systems (special issue). T-MTT Mar 05 1005-1116 multifunctional RF systems (special issue intro.). Adler, E.D., + , T-MTT Mar 05 1005-1008 mutually exclusive data encoding for realization of a full duplexing selfsteering wireless link using a retrodirective array transceiver. Leong, K.M.K.H., + , T-MTT Dec 05 3687-3696 secure high-speed retrodirective commun. link. Goshi, D.S., + , T-MTT Nov 05 3548-3556 Modeling capacitive nonlinearity in thin-film BST varactors. Chase, D.R., + , TMTT Oct 05 3215-3220 metamaterial structures, phenomena, and applications (special issue). TMTT Apr 05 1413-1556 metamaterial structures, phenomena, and applications (special issue intro.). Itoh, T., + , T-MTT Apr 05 1413-1417 Modeling; cf. Integrated circuit modeling Mode locked lasers photoelectronic ADC. Ioakeimidi, K., + , T-MTT Jan 05 336-342

+ Check author entry for coauthors

Mode matching methods anal. of dielec. loaded waveguide filters of arbitrary shape, hybrid surface integral-eqn./mode-matching method. Catina, V., + , T-MTT Nov 05 3562-3567 H-plane contiguous manifold output multiplexers, fictitious reactive load concept, full-wave design. Montejo-Garai, J.R., + , T-MTT Aug 05 2628-2632 input impedance of top-loaded monopole, parallel-plate waveguide by MoM/Green's fn. method. Valero-Nogueira, A., + , T-MTT Mar 05 868873 LTCC, metallic resonators, canonical ridge waveguide filters. Ruiz-Cruz, J.A., + , T-MTT Jan 05 174-182 Modems multistandard WLAN appls., dual-band RF transceiver. Chang, S.-F.R., + , T-MTT Mar 05 1048-1055 MODFET integrated circuits 40-Gb/s wide-band MMIC pHEMT modulator driver amps. designed, real freq. tech. Kerherve, E., + , T-MTT Jun 05 2145-2152 60-GHz broad-band telecomm., resistive HEMT mixers. Varonen, M., + , T-MTT Apr 05 1322-1330 multimode J-pHEMT front-end archit., power-control scheme for max. effic. Clifton, J.C., + , T-MTT Jun 05 2251-2258 sig. gener., control, freq. conversion AlGaN/GaN HEMT MMICs. Kaper, V.S., + , T-MTT Jan 05 55-65 v-band high-order harmonic injection-locked freq.-divider MMICs, wide bandwidth and low-power dissipation. Jinho Jeong, + , T-MTT Jun 05 1891-1898 MODFETs 2.14-GHz Chireix outphasing transmitter. Hakala, I., + , T-MTT Jun 05 2129-2138 60-GHz broad-band telecomm., resistive HEMT mixers. Varonen, M., + , T-MTT Apr 05 1322-1330 60-GHz monolithic star mixer, gate-drain-connected pHEMT diodes. Kyung-Whan Yeom, + , T-MTT Jul 05 2435-2440 efficient anal. formulation and sensitivity anal. of neuro-space mapping for nonlin. microwave device modeling. Lei Zhang, + , T-MTT Sep 05 2752-2767 epitaxial struct., noise fig. of AlGaN/GaN HEMTs. Sanabria, C., + , TMTT Feb 05 762-769 InP HEMTs and their appls., mm-wave radio-on-fiber systs., phototransistors. Chang-Soon Choi, + , T-MTT Jan 05 256-263 presence of LF dispers. effects, accurate pHEMT nonlin. modeling. Raffo, A., + , T-MTT Nov 05 3449-3459 RF-driven gate current, DC/RF perform., GaAs pHEMT MMIC power amps., effect. Yeong-Chang Chou, + , T-MTT Nov 05 3398-3406 small-sig. modeling approach applied, GaN devices. Jarndal, A., + , TMTT Nov 05 3440-3448 time-domain large-sig. meas., table-based nonlin. HEMT model extracted. Curras-Francos, M.C., T-MTT May 05 1593-1600 transistor nonlinearities, noise props. Sungjae Lee, + , T-MTT Apr 05 1314-1321 v-band high-order harmonic injection-locked freq.-divider MMICs, wide bandwidth and low-power dissipation. Jinho Jeong, + , T-MTT Jun 05 1891-1898 wide-band on-wafer noise-param. meas., improved Y-factor method. Tiemeijer, L.F., + , T-MTT Sep 05 2917-2925 Modulation RF sect. of UHF and microwave pass. RFID transponders, design criteria. De Vita, G., + , T-MTT Sep 05 2978-2990 Modulation; cf. Amplitude modulation; Minimum shift keying; OFDM modulation; Optical modulation Modulators; cf. Modems Modules; cf. Multichip modules Moisture measurement content, bulk materials, sub-nanosecond UWB pulses, noncontacting determ. Schimmer, O., + , T-MTT Jun 05 2107-2113 Moment methods 2D nonuniform FFT (2-D NUFFT) tech., anal. of shielded microstrip ccts. Ke-Ying Su, + , T-MTT Mar 05 993-999 anal. of NRD-guide/H-guide mm-wave ccts., order-reduced vol.-integral eqn. approach. Duochuan Li, + , T-MTT Mar 05 799-812 anal. software for tunable microstrip filters, magnetized ferrites, expt. validation. Leon, G., + , T-MTT May 05 1739-1744

IEEE T-MTT 2005 INDEX — 49 complex pass. devices composed of arbitrarily shaped waveguides, Nystrom and BI-RME methods, CAD. Taroncher, M., + , T-MTT Jun 05 2153-2163 coupled-line models from EM simulators and appl., MoM anal., derivation. Farina, M., + , T-MTT Nov 05 3272-3280 coupled lossy transm. lines, multiwavelet-based MoM, full-wave anal. Meisong Tong, + , T-MTT Jul 05 2362-2370 double-delay and SOC EM deembedding, unification. Rautio, J.C., + , TMTT Sep 05 2892-2898 input impedance of top-loaded monopole, parallel-plate waveguide by MoM/Green's fn. method. Valero-Nogueira, A., + , T-MTT Mar 05 868873 metamaterial constructed by conductive SRRs and wires, MGS-based algm., propag. property anal. Hai-Ying Yao, + , T-MTT Apr 05 14691476 microstrip nets., orthogonality-based deembedding tech. Spowart, M.P., + , T-MTT Mar 05 938-946 periodically nonuniform coupled microstrip lines-even and odd modes, guided-wave characts. Sheng Sun, + , T-MTT Apr 05 1221-1227 rect. waveguide, dielec.-filled corrugations supporting backward waves. Eshrah, I.A., + , T-MTT Nov 05 3298-3304 stopband-enhanced and size-miniaturized low-pass filters, high-impedance property of offset finite-ground microstrip line. Sheng Sun, + , T-MTT Sep 05 2844-2850 uniplanar left-handed metamaterials, efficient modeling. Yunchuan Guo, + , T-MTT Apr 05 1462-1468 Monitoring detect. and localization of mobile phones, large buildings, microwave syst. Hudec, P., + , T-MTT Jun 05 2235-2239 Monolithic integrated circuits; cf. MMIC Monopole antennas microwave ablation, triaxial antenna. Brace, C.L., + , T-MTT Jan 05 215220 MOS analog integrated circuits; cf. CMOS analog integrated circuits MOS capacitors CAD techniques suitable for the design of high-power RF transistors. Aaen, P.H., + , T-MTT Oct 05 3067-3074 MOSFETs carrier heating, channel noise, deep-submicrometer NMOSFETs via body bias, expt. study. Hong Wang, + , T-MTT Feb 05 564-570 inverse class-F archit., l-band LDMOS power amps. Lepine, F., + , TMTT Jun 05 2007-2012 microwave on-wafer charactn. of deep-submicrometer Si MOSFETs, shield-based 3-port de-embedding method. Ming-Hsiang Cho, + , TMTT Sep 05 2926-2934 MOS integrated circuits; cf. CMOS integrated circuits Moving average processes deembedding static nonlinearities and accurately identifying and modeling memory effects, wide-band RF transmitters. Taijun Liu, + , T-MTT Nov 05 3578-3587 Multichip modules high-quality solenoid inductor, dielec. film for multichip modules. JongMin Yook, + , T-MTT Jun 05 2230-2234 inter-chip RF-interconnect, CPW, capacitive coupler, UWB transceiver, perform. Sun, M., + , T-MTT Sep 05 2650-2655 Multipath channels adaptive duplexer implemented, single-path/multipath feedforward techs., BST phase shifters. O'Sullivan, T., + , T-MTT Jan 05 106-114 Multiplexing; cf. Wavelength division multiplexing Multiplying circuits; cf. Analog multipliers Multiport circuits 4-port microwave nets., intrinsic broad-band suppression of commonmode sigs. Fathelbab, W.M., + , T-MTT May 05 1569-1575 6-port, communs. receiver. Hentschel, T., T-MTT Mar 05 1039-1047 anal. of NRD-guide/H-guide mm-wave ccts., order-reduced vol.-integral eqn. approach. Duochuan Li, + , T-MTT Mar 05 799-812 comput. time-domain sensitivity of multiport systs. described by reducedorder models, adjoint-based approach. Ahmed, T., + , T-MTT Nov 05 3538-3547 dielec.-filled cavity filters, ultrawide stopband Characteristics, design. Rauscher, C., T-MTT May 05 1777-1786 local ground plane, EM anal., deembedding effect. Rautio, J.C., T-MTT Feb 05 770-776 microstrip nets., orthogonality-based deembedding tech. Spowart, M.P., + , T-MTT Mar 05 938-946 + Check author entry for coauthors

microwave on-wafer charactn. of deep-submicrometer Si MOSFETs, shield-based 3-port de-embedding method. Ming-Hsiang Cho, + , TMTT Sep 05 2926-2934 partially leaky multiport vector net. analyzers, on-wafer calib. algm. Teppati, V., + , T-MTT Nov 05 3665-3671 port discontinuities, full-wave CAD models of multiport ccts., deembedding. Farina, M., T-MTT May 05 1829 port discontinuities, full-wave CAD models of multiport ccts. ), deembedding. Okhmatovski, V.I., + , T-MTT May 05 1829 Multiport networks; cf. Two-port networks Multivariable systems; cf. MIMO systems Multivibrators; cf. Flip-flops Muscles biol. appls., micromachined probe, in vivo meas. Jung-Mu Kim, + , TMTT Nov 05 3415-3421 SAR distribs., 3-layered bio-media, direct contact, water-loaded modified box-horn applicator. Gupta, R.C., + , T-MTT Sep 05 2665-2671 N Negative resistance devices; cf. Tunnel diodes Network parameters; cf. S-parameters Networks (circuits); cf. Amplifiers; Attenuators; Coupled circuits; Discriminators; Distributed parameter networks; Equalizers; Equivalent circuits; Filters; Integrated circuits; Microwave circuits; Oscillators; Phase locked loops; Phase shifters; Printed circuits; VHF circuits Network synthesis; cf. Circuit optimization; Integrated circuit design Neural networks efficient anal. formulation and sensitivity anal. of neuro-space mapping for nonlin. microwave device modeling. Lei Zhang, + , T-MTT Sep 05 2752-2767 RF inductors and filters, LCP substrs. for Wi-Fi appls., layout-level synthesis. Mukherjee, S., + , T-MTT Jun 05 2196-2210 Newton method Chebyshev collocation and Newton-type optim. methods for inverse problem, nonuniform transm. lines. Norgren, M., T-MTT May 05 15611568 complex-permittivity measurement on high-Q materials via combined numerical approaches. Fan, X.C., + , T-MTT Oct 05 3130-3134 high dielec.-contrast objs., different image-reconstruction approaches, microwave-tomographic imaging. Semenov, S.Y., + , T-MTT Jul 05 2284-2294 Newton-Raphson method complex permitt. of arbitrary shape and size dielec. samples, cavity meas. tech., microwave freqs., estim. Santra, M., + , T-MTT Feb 05 718-722 parallel FFT accelerated transient field-cct. simulator. Yilmaz, A.E., + , TMTT Sep 05 2851-2865 Noise; cf. Circuit noise; Phase noise; Random noise Noise measurement InP HBT noise params., noise-fig. meas. syst., direct extr. Jianjun Gao, + , T-MTT Jan 05 330-335 low-noise multiresolution high-dyn. ultra-broad-band time-domain EMI meas. syst. Braun, S., + , T-MTT Nov 05 3354-3363 noise behavior of microwave amplifiers operating under nonlinear conditions. Escotte, L., + , T-MTT Dec 05 3704-3711 on-wafer noise-param. meas., wide-band freq.-var. method. Hu, R., + , TMTT Jul 05 2398-2402 wide-band on-wafer noise-param. meas., improved Y-factor method. Tiemeijer, L.F., + , T-MTT Sep 05 2917-2925 Nondestructive testing in situ S-param. meas. of hermetically encapsulated packages, recursive un-termination method. Pfeiffer, U.R., + , T-MTT Jun 05 1845-1855 Nonhomogeneous media double-neg. metamaterials, pos. future. Nader Engheta, + , T-MTT Apr 05 1535-1556 low-cost planar probes, broadside apertures for nondestructive dielec. meas. of biol. materials, microwave freqs. Byoungjoong Kang, + , TMTT Jan 05 134-143 scan antennas, ferroelec. substr., rigorous anal. and investigs. Yashchyshyn, Y., + , T-MTT Feb 05 427-438 Nonlinear circuits bifurcation control, practical cct. design. Collado, A., + , T-MTT Sep 05 2777-2788 complex enveloped sigs. and appl., feedforward cct. anal., multitone model. Coskun, A.H., + , T-MTT Jun 05 2171-2178

IEEE T-MTT 2005 INDEX — 50 efficient anal. formulation and sensitivity anal. of neuro-space mapping for nonlin. microwave device modeling. Lei Zhang, + , T-MTT Sep 05 2752-2767 efficient nonlin. cct. simul. tech. Dautbegovic, E., + , T-MTT Feb 05 548555 gain compress. curve, IIP3 estim. Choongeol Cho, + , T-MTT Apr 05 1197-1202 linearizing FET low-noise amps., modified derivative superposition method. Aparin, V., + , T-MTT Feb 05 571-581 lin. inverse space-mapping (LISM) algm., design lin./nonlin. RF and microwave ccts. Rayas-Sanchez, J.E., + , T-MTT Mar 05 960-968 power amps., RBF neural nets, wide-band dyn. modeling. Isaksson, M., + , T-MTT Nov 05 3422-3428 RFICs, 2-port lumped-nonlin.-source model, systematic linearity anal. Qingqing Liang, + , T-MTT May 05 1745-1755 Nonlinear distortion; cf. Harmonic distortion; Intermodulation distortion Nonlinear equations gen. Chebyshev filters, asymmetrically located transm. zeros, design. Milosavljevic, Z.D., T-MTT Jul 05 2411-2415 parallel FFT accelerated transient field-cct. simulator. Yilmaz, A.E., + , TMTT Sep 05 2851-2865 Nonlinear filters microwave and wireless power-amp. behavioral modeling approaches, comparative overview. Pedro, J.C., + , T-MTT Apr 05 1150-1163 Nonlinear optics; cf. Optical frequency conversion; Optical phase conjugation Nonlinear programming; cf. Quadratic programming Nonlinear systems microwave and wireless power-amp. behavioral modeling approaches, comparative overview. Pedro, J.C., + , T-MTT Apr 05 1150-1163 model testing, designing multisine excit. Pedro, J.C., + , T-MTT Jan 05 45-54 noise behavior of microwave amplifiers operating under nonlinear conditions. Escotte, L., + , T-MTT Dec 05 3704-3711 Nonlinear systems; cf. Bilinear systems Nonradiative dielectric waveguides anal. of NRD-guide/H-guide mm-wave ccts., order-reduced vol.-integral eqn. approach. Duochuan Li, + , T-MTT Mar 05 799-812 bridged NRD-guide coupler for mm-wave appls., anal. and design. Duochuan Li, + , T-MTT Aug 05 2546-2551 gen. surface-vol. integral-eqn. (SVIE) approach for anal. of hybrid planar/NRD-guide IC. Duochuan Li, + , T-MTT Sep 05 2732-2742 Notch filters image-rejection CMOS LNA design optim. techs. Trung-Kien Nguyen, + , T-MTT Feb 05 538-547 Numerical analysis additive and converted noise, gener. of phase noise, nonlin. oscillators, role. Nallatamby, J.-C., + , T-MTT Mar 05 901-906 AlGaN-GaN devices, thermal resist. calc. Wen-Yan Yin, T-MTT Sep 05 3051-3052 Numerical analysis; cf. Convergence of numerical methods; Curve fitting; Error analysis; Finite difference methods; Interpolation; Iterative methods Numerical stability general lin. lumped microwave ccts., matrix theory, improved FDTD formulation. Zhenhai Shao, + , T-MTT Jul 05 2261-2266 reduce num. dispers., ADI-FDTD, efficient method. Hong-Xing Zheng, + , T-MTT Jul 05 2295-2301 O OFDM modulation frequency planning and synthesizer architectures for multiband OFDM UWB radios. Mishra, C., + , T-MTT Dec 05 3744-3756 Ohmic contacts low-loss 2-bit tunable bandpass filters, MEMS DC contact switches. Pothier, A., + , T-MTT Jan 05 354-360 Optical communication coherent optical vector modulation for fiber radio using electrooptic microchip lasers. Li, Y., + , T-MTT Oct 05 3121-3129 wide-band continuously tunable millimeter-wave signal with an optical external modulation technique. Qi, G., + , T-MTT Oct 05 3090-3097 Optical communication; cf. Optical communication equipment; Optical fiber communication

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Optical communication equipment 10-Gb/s driver amplifier using a tapered gate line for improved input matching. Shohat, J., + , T-MTT Oct 05 3115-3120 40-Gb/s wide-band MMIC pHEMT modulator driver amps. designed, real freq. tech. Kerherve, E., + , T-MTT Jun 05 2145-2152 InP HEMTs and their appls., mm-wave radio-on-fiber systs., phototransistors. Chang-Soon Choi, + , T-MTT Jan 05 256-263 Optical communication equipment; cf. Optical receivers Optical elements; cf. Optical fibers; Optical waveguides Optical fiber communication 10-Gb/s fiber opt. commun. links, 0.18-ȝm CMOS equalization techs. Moonkyun Maeng, + , T-MTT Nov 05 3509-3519 InP HEMTs and their appls., mm-wave radio-on-fiber systs., phototransistors. Chang-Soon Choi, + , T-MTT Jan 05 256-263 Optical fibers injection-locked dual opto-electron. oscillator, ultra-low phase noise and ultra-low spurious level. Weimin Zhou, + , T-MTT Mar 05 929-933 Optical frequency conversion sigs., p-i-n photodiodes, freq. conversion. Malyshev, S.A., + , T-MTT Feb 05 439-443 Optical materials simultaneous elec. and mag. near-field meas., LiNbO3, inverted domain, EO probe. Suzuki, E., + , T-MTT Feb 05 696-701 Optical mixing opt. sigs., p-i-n photodiodes, freq. conversion. Malyshev, S.A., + , T-MTT Feb 05 439-443 Optical modulation radiometric Millimeter-wave detect. via opt. upconversion and carrier suppression. Schuetz, C.A., + , T-MTT May 05 1732-1738 Optical modulation; cf. Electro-optical modulation Optical phase conjugation large-sig. scatt. fns., linearization. Verspecht, J., + , T-MTT Apr 05 13691376 Optical polarization TE/TM propag. and losses of integr. opt. polarizer, accurate modeling. Pierantoni, L., + , T-MTT Jun 05 1856-1862 Optical receivers Si Schottky diode DEMUX cct. for high bit-rate opt. receivers. Jung Han Choi, + , T-MTT Jun 05 2033-2042 Optical refraction double-neg. metamaterials, pos. future. Nader Engheta, + , T-MTT Apr 05 1535-1556 loaded transm.-line neg.-refr.-index metamaterials, periodic FDTD anal. Kokkinos, T., + , T-MTT Apr 05 1488-1495 metamaterials, neg. refr. index, 2D shunt and 3D SCN TLM nets., modeling. So, P.P.M., + , T-MTT Apr 05 1496-1505 Optical signal detection radiometric Millimeter-wave detect. via opt. upconversion and carrier suppression. Schuetz, C.A., + , T-MTT May 05 1732-1738 Optical waveguides TE/TM propag. and losses of integr. opt. polarizer, accurate modeling. Pierantoni, L., + , T-MTT Jun 05 1856-1862 Optical waveguides; cf. Optical fibers Optics; cf. Integrated optics; Physical optics Optimization; cf. Circuit optimization; Genetic algorithms; Simulated annealing Optimization methods (2,4) stencil, optimized FDTD methods. Guilin Sun, + , T-MTT Mar 05 832-842 EDGE terminal power amps., memoryless digital predistortion, optim. Ceylan, N., + , T-MTT Feb 05 515-522 large-sig. diode modeling, alternative param.-extr. tech. Yew Hui Liew, + , T-MTT Aug 05 2633-2638 nonlin. model testing, designing multisine excit. Pedro, J.C., + , T-MTT Jan 05 45-54 optimum load location-expt. approach, multimode cavity effic. optim. Requena-Perez, M.E., + , T-MTT Jun 05 2114-2120 particle swarm optimizer for microwave imaging of 2D dielec. scatterers, comput. approach. Donelli, M., + , T-MTT May 05 1761-1776 software-defined radio systs., sig. path optim. Rykaczewski, P., + , T-MTT Mar 05 1056-1064 waveguide filters, aggressive space mapping, segm. strategy and hybrid optim. algm., fast automated design. Ros, J.V.M., + , T-MTT Apr 05 1130-1142

IEEE T-MTT 2005 INDEX — 51 Optoelectronic devices injection-locked dual opto-electron. oscillator, ultra-low phase noise and ultra-low spurious level. Weimin Zhou, + , T-MTT Mar 05 929-933 Organic compounds; cf. Proteins Oscillators coupled-oscillator arrays without, posteriori tuning, design. Xing Wang, + , T-MTT Jan 05 410-413 RF-CMOS and SiGe BiCMOS, WCDMA direct-conversion receiver front-end comp. Floyd, B.A., + , T-MTT Apr 05 1181-1188 Oscillators; cf. Dielectric resonator oscillators; Injection locked oscillators; Phase locked oscillators P Packaging packaged inductively degenerated common-source low-noise amps., ESD protection, anal. and optim. Sivonen, P., + , T-MTT Apr 05 1304-1313 phased-array antenna systs., RF MEMS for automotive radar appls., design considerations and technol. assess. Schoebel, J., + , T-MTT Jun 05 1968-1975 Packaging; cf. Integrated circuit packaging; Multichip modules Parameter estimation low-loss single-pole 6-throw switch, compact RF MEMS switches. Jaewoo Lee, + , T-MTT Nov 05 3335-3344 microwave filters, adaptive models and param. extr., seq. tuning. Pepe, G., + , T-MTT Jan 05 22-31 on-chip interconnects by double-image Green's fn. method combined, hierarchical algm., param. extr. Wenliang Dai, + , T-MTT Jul 05 24162423 small-sig. modeling approach applied, GaN devices. Jarndal, A., + , TMTT Nov 05 3440-3448 Parameter estimation; cf. Recursive estimation Parametric oscillators; cf. Microwave parametric oscillators Partial differential equations; cf. Laplace equations Particle accelerators; cf. Accelerator RF systems Particle beams; cf. Electron beams Passivation surface-passivated high-resist. Si, true microwave substr. Spirito, M., + , T-MTT Jul 05 2340-2347 Passive circuits 40-Gb/s wide-band MMIC pHEMT modulator driver amps. designed, real freq. tech. Kerherve, E., + , T-MTT Jun 05 2145-2152 fast layout verification of 3D RF and mixed-sig. on-chip structs., largescale broad-band parasitic extr. Feng Ling, + , T-MTT Jan 05 264-273 highly integr. mm-wave pass. components, 3D LTCC syst.-on-package (SOP) technol. Jong-Hoon Lee, + , T-MTT Jun 05 2220-2229 microstrip nets., orthogonality-based deembedding tech. Spowart, M.P., + , T-MTT Mar 05 938-946 Passive filters compact U-shaped dual planar EBG microstrip low-pass filter. Huang, S.Y., + , T-MTT Dec 05 3799-3805 image-rejection CMOS LNA design optim. techs. Trung-Kien Nguyen, + , T-MTT Feb 05 538-547 miniaturized microwave pass. filter incorporating multilayer synthetic quasiTEM transm. line. Hsien-Shun Wu, + , T-MTT Sep 05 2713-2720 Passive filters; cf. Surface acoustic wave filters Passive networks; cf. Passive filters Patient diagnosis; cf. Biomedical imaging Patient treatment; cf. Surgery Performance evaluation multifunctional RF systems (special issue). T-MTT Mar 05 1005-1116 multifunctional RF systems (special issue intro.). Adler, E.D., + , T-MTT Mar 05 1005-1008 Periodic structures; cf. Photonic crystals Permeability crossed anisotropy mag. core, toroidal thin-film inductors, integrat. Frommberger, M., + , T-MTT Jun 05 2096-2100 leaky-wave propag., metamaterial grounded slabs excited by dipole source, effects. Baccarelli, P., + , T-MTT Jan 05 32-44 surface waves, metamaterial grounded slabs, fund. modal props. Baccarelli, P., + , T-MTT Apr 05 1431-1442 Permittivity 3D-connected/nonconnected wire metamaterials, homogenization. Silveirinha, M.G., + , T-MTT Apr 05 1418-1430

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complex permitt. of arbitrary shape and size dielec. samples, cavity meas. tech., microwave freqs., estim. Santra, M., + , T-MTT Feb 05 718-722 cylindrical multilayered ceramic resonators, rect. air cavity for low-phase noise K/Ka-band oscillators. El-Tager, A.M., + , T-MTT Jun 05 22112219 designing microstrip filters utilizing mixed dielectrics, approaches. Semouchkina, E., + , T-MTT Feb 05 644-652 dielec. const., loss tangent, surface resist. of PCB materials, K-band freqs. Egorov, V.N., + , T-MTT Feb 05 627-635 distrib. left-handed microstrip lines, effective EM params. Shau-Gang Mao, + , T-MTT Apr 05 1515-1521 leaky-wave propag., metamaterial grounded slabs excited by dipole source, effects. Baccarelli, P., + , T-MTT Jan 05 32-44 metamaterials, neg. refr. index, 2D shunt and 3D SCN TLM nets., modeling. So, P.P.M., + , T-MTT Apr 05 1496-1505 miniaturized spurious passband suppression microstrip filter, meandered parallel coupled lines. Shih-Ming Wang, + , T-MTT Feb 05 747-753 neg. permitt. and neg. permeab. by of evanescent waveguide Modestheory and expt., simul. Esteban, J., + , T-MTT Apr 05 1506-1514 on-chip interconnects by double-image Green's fn. method combined, hierarchical algm., param. extr. Wenliang Dai, + , T-MTT Jul 05 24162423 permitt. and loss tangent of high-permitt. materials, terahertz freqs., temp. depend. Berdel, K., + , T-MTT Apr 05 1266-1271 scan antennas, ferroelec. substr., rigorous anal. and investigs. Yashchyshyn, Y., + , T-MTT Feb 05 427-438 surface waves, metamaterial grounded slabs, fund. modal props. Baccarelli, P., + , T-MTT Apr 05 1431-1442 tailored and anisotropic dielec. consts., porosity, ceramic components. Xun Gong, + , T-MTT Nov 05 3638-3647 Permittivity measurement biol. appls., micromachined probe, in vivo meas. Jung-Mu Kim, + , TMTT Nov 05 3415-3421 complex-permittivity measurement on high-Q materials via combined numerical approaches. Fan, X.C., + , T-MTT Oct 05 3130-3134 complex permitt. meas., TE11p modes, circ. cylindrical cavities. Zinal, S., + , T-MTT Jun 05 1870-1874 complex permitt. of arbitrary shape and size dielec. samples, cavity meas. tech., microwave freqs., estim. Santra, M., + , T-MTT Feb 05 718-722 dielec. consts. of metallic nanoparticles embedded, paraffin rod, microwave freqs., meas. Yan-Shian Yeh, + , T-MTT May 05 1756-1760 in vivo and ex dielec. spectrosc. of biol. tissues, microwave freqs., precision open-ended coaxial probes. Popovic, D., + , T-MTT May 05 1713-1722 measuring multilayered dielec. plates, open resonator tech. Deleniv, A.N., + , T-MTT Sep 05 2908-2916 Perturbation methods complex permitt. of arbitrary shape and size dielec. samples, cavity meas. tech., microwave freqs., estim. Santra, M., + , T-MTT Feb 05 718-722 ferrite coupled-line circulator, reduced length. Meng Cao, + , T-MTT Aug 05 2572-2579 serial communs., data-depend. jitter. Analui, B., + , T-MTT Nov 05 33883397 Phase conjugation (optical) mutually exclusive data encoding for realization of a full duplexing selfsteering wireless link using a retrodirective array transceiver. Leong, K.M.K.H., + , T-MTT Dec 05 3687-3696 Phase control waveguide polarizers, design tool. Virone, G., + , T-MTT Mar 05 888-894 Phased array radar antenna systs., RF MEMS for automotive radar appls., design considerations and technol. assess. Schoebel, J., + , T-MTT Jun 05 19681975 compact dual-polarized multibeam phased-array archit. for mm-wave radar. Schulwitz, L., + , T-MTT Nov 05 3588-3594 ultra-wideband low-cost phased-array radars. Rodenbeck, C.T., + , T-MTT Dec 05 3697-3703 Phased arrays 24-GHz SiGe phased-array receiver-LO phase-shifting approach. Hashemi, H., + , T-MTT Feb 05 614-626 armored vehicle appl., multifunction mm-wave systs. Wehling, J.H., TMTT Mar 05 1021-1025 coupled-oscillator arrays without, posteriori tuning, design. Xing Wang, + , T-MTT Jan 05 410-413

IEEE T-MTT 2005 INDEX — 52 phased-array antenna systs., RF MEMS for automotive radar appls., design considerations and technol. assess. Schoebel, J., + , T-MTT Jun 05 1968-1975 Phase locked loops 0.25-ȝm CMOS OPLL transmitter IC for GSM and DCS appls. Peng-Un Su, T-MTT Feb 05 462-471 DVB-S appls., Si bipolar technol., monolithic 12-GHz heterodyne receiver. Girlando, G., + , T-MTT Mar 05 952-959 offset-PLL output spur spectrum, systematic anal. Ching-Feng Lee, + , TMTT Sep 05 3024-3034 PLL-based retrodirective array, anal. and charactn. Fusco, V., + , T-MTT Feb 05 730-738 Phase locked oscillators DVB-S appls., Si bipolar technol., monolithic 12-GHz heterodyne receiver. Girlando, G., + , T-MTT Mar 05 952-959 Phase measurement 1D LHM-RHM resonator, expt. realization. Yan Li, + , T-MTT Apr 05 1522-1526 nonlin. device charactn., multitone phase and amplit. meas. Martins, J.P., + , T-MTT Jun 05 1982-1989 Phase modulation baseband-modeled CALLUM archits., spectrum emission considerations. Strandberg, R., + , T-MTT Feb 05 660-669 hybrid digital/RF envelope predistortion linearization syst. for power amps. Wangmyong Woo, + , T-MTT Jan 05 229-237 measuring BER of UWB devices, prod. test tech. Bhattacharya, S., + , TMTT Nov 05 3474-3481 single and multicarrier W-CDMA sigs., LINC digital component separator. Gerhard, W., + , T-MTT Jan 05 274-282 X-band class-E power amps., EER operation, linearity. Narisi Wang, + , T-MTT Mar 05 1096-1102 Phase modulation; cf. Phase shift keying Phase noise 1/f noise and oscillator phase noise, SiGe HBTs, scaling and technol. limitations. Guofu Niu, + , T-MTT Feb 05 506-514 additive and converted noise, gener. of phase noise, nonlin. oscillators, role. Nallatamby, J.-C., + , T-MTT Mar 05 901-906 CMOS technol., complementary Colpitts oscillator. Choong-Yul Cha, + , T-MTT Mar 05 881-887 cylindrical multilayered ceramic resonators, rect. air cavity for low-phase noise K/Ka-band oscillators. El-Tager, A.M., + , T-MTT Jun 05 22112219 DVB-S appls., Si bipolar technol., monolithic 12-GHz heterodyne receiver. Girlando, G., + , T-MTT Mar 05 952-959 injection-locked dual opto-electron. oscillator, ultra-low phase noise and ultra-low spurious level. Weimin Zhou, + , T-MTT Mar 05 929-933 k- and Q-bands CMOS freq. sources, X-band quadrature VCO. Sangsoo Ko, + , T-MTT Sep 05 2789-2800 low-power ka-band Voltage-controlled oscillator implemented, 200-GHz SiGe HBT technol. Yi-jan Emery Chen, + , T-MTT May 05 1672-1681 mm-wave CMOS cct. design. Shigematsu, H., + , T-MTT Feb 05 472-477 multistandard adaptive voltage-controlled oscillators, design. Tasic, A., + , T-MTT Feb 05 556-563 multistandard WLAN appls., dual-band RF transceiver. Chang, S.-F.R., + , T-MTT Mar 05 1048-1055 PLL-based retrodirective array, anal. and charactn. Fusco, V., + , T-MTT Feb 05 730-738 push-push/triple-push oscillators for reducing 1/f noise upconversion, design. Jonghoon Choi, + , T-MTT Nov 05 3407-3414 stabil. ccts. for phase-noise reduction, microwave oscillators. Suarez, A., + , T-MTT Sep 05 2743-2751 transistor nonlinearities, noise props. Sungjae Lee, + , T-MTT Apr 05 1314-1321 Phase shifters phased-array antenna systs., RF MEMS for automotive radar appls., design considerations and technol. assess. Schoebel, J., + , T-MTT Jun 05 1968-1975 Phase shifters; cf. Ferrite phase shifters; Microwave phase shifters; Millimeter wave phase shifters Phase shift keying BPSK to ASK signal conversion using injection-locked oscillators theory, part I. Lopez-Villegas, J.M., + , T-MTT Dec 05 3757-3766 metamaterial-based electronically controlled transm.-line struct., leakywave antenna, tunable radiation angle and beamwidth. Sungjoon Lim, + , T-MTT Jan 05 161-173 + Check author entry for coauthors

software-defined direct conversion receiver, ka-band analog front-end. Tatu, S.O., + , T-MTT Sep 05 2768-2776 Phase shift keying; cf. Quadrature phase shift keying Photocathodes photoelectronic ADC. Ioakeimidi, K., + , T-MTT Jan 05 336-342 Photoconducting devices; cf. Phototransistors Photodetectors radiometric Millimeter-wave detect. via opt. upconversion and carrier suppression. Schuetz, C.A., + , T-MTT May 05 1732-1738 small-signal model for submicrometer nn-i-nn traveling-wave photodetectors. Torrese, G., + , T-MTT Oct 05 3238-3243 TW photodetectors, hybrid drift-diffusion-TLM anal. Pasalic, D., + , TMTT Sep 05 2700-2706 Photodiodes; cf. Avalanche photodiodes; p-i-n photodiodes Photoelectric devices; cf. Photodetectors Photoemissive devices; cf. Photocathodes Photolithography distrib. left-handed microstrip lines, effective EM params. Shau-Gang Mao, + , T-MTT Apr 05 1515-1521 Photoluminescence thermal resist. of FETs, accurate determ. Darwish, A.M., + , T-MTT Jan 05 306-313 Photonic band gap compact U-shaped dual planar EBG microstrip low-pass filter. Huang, S.Y., + , T-MTT Dec 05 3799-3805 Photonic crystals compact U-shaped dual planar EBG microstrip low-pass filter. Huang, S.Y., + , T-MTT Dec 05 3799-3805 Phototransistors InP HEMTs and their appls., mm-wave radio-on-fiber systs., phototransistors. Chang-Soon Choi, + , T-MTT Jan 05 256-263 Physical optics vector NWA for microwave imaging, effective usage. Chao-Hsiung Tseng, + , T-MTT Sep 05 2884-2891 Piecewise linear approximation lin. inverse space-mapping (LISM) algm., design lin./nonlin. RF and microwave ccts. Rayas-Sanchez, J.E., + , T-MTT Mar 05 960-968 Piezoelectric devices dielec. resonator, discrete electromechanical freq. tuning. Panaitov, G.I., + , T-MTT Nov 05 3371-3377 Piezoelectric devices; cf. Crystal filters Piezoelectric resonator filters init. design of LTCC filters, all-capacitive coupling, simplified anal. tech. Rambabu, K., + , T-MTT May 05 1787-1791 serial config., 2 finite transm. zeros, LTCC technol., bandpass filter. Chun-Fu Chang, + , T-MTT Jul 05 2383-2388 p-i-n diodes broad-band power amp., tunable output matching net. Haitao Zhang, + , T-MTT Nov 05 3606-3614 compact dual-polarized multibeam phased-array archit. for mm-wave radar. Schulwitz, L., + , T-MTT Nov 05 3588-3594 p-i-n diodes; cf. p-i-n photodiodes p-i-n photodiodes opt. sigs., p-i-n photodiodes, freq. conversion. Malyshev, S.A., + , T-MTT Feb 05 439-443 Planar transmission lines microwave components, EM lin. regression models, empirical model gener. techs. Domenech-Asensi, G., + , T-MTT Nov 05 3305-3311 Planar waveguides Characteristic eqn. for open nonreciprocal Layered waveguides, different upper and lower half-spaces, appropriate formulation. Rodriguez-Berral, R., + , T-MTT May 05 1613-1623 gen. surface-vol. integral-eqn. (SVIE) approach for anal. of hybrid planar/NRD-guide IC. Duochuan Li, + , T-MTT Sep 05 2732-2742 in-line N-order filters, N real transm. zeros by of extracted poles implemented, low-cost rect. H-plane waveguide, synthesis and design. Montejo-Garai, J.R., + , T-MTT May 05 1636-1642 nonintuitive planar structs., fast optim. and sensitivity anal. Cormos, D., + , T-MTT Jun 05 2019-2025 substr. integr. waveguide, guided-wave and leakage characts. Feng Xu, + , T-MTT Jan 05 66-73 Plasma loaded waveguides neg. permitt. and neg. permeab. by of evanescent waveguide Modestheory and expt., simul. Esteban, J., + , T-MTT Apr 05 1506-1514

IEEE T-MTT 2005 INDEX — 53 Polarization broad-band and circ. polarized space-filling-based slot antennas. Ghali, H.A., + , T-MTT Jun 05 1946-1950 compact dual-polarized multibeam phased-array archit. for mm-wave radar. Schulwitz, L., + , T-MTT Nov 05 3588-3594 effectiveness of wave absorbers, improve DSRC EM environ., express highway. Pokharel, R.K., + , T-MTT Sep 05 2726-2731 k-band orthomode transducer, waveguide ports and balanced coaxial probes. Engargiola, G., + , T-MTT May 05 1792-1801 mutually exclusive data encoding for realization of a full duplexing selfsteering wireless link using a retrodirective array transceiver. Leong, K.M.K.H., + , T-MTT Dec 05 3687-3696 polariz. effects of capacitive RF MEMS switches, temp. study. Papaioannou, G., + , T-MTT Nov 05 3467-3473 surface waves, metamaterial grounded slabs, fund. modal props. Baccarelli, P., + , T-MTT Apr 05 1431-1442 temporally dispers. dielec., time-domain cavity oscills. supported. Aksoy, S., + , T-MTT Aug 05 2465-2471 Poles and zeros 12-18-GHz 3-pole RF MEMS tunable filter. Entesari, K., + , T-MTT Aug 05 2566-2571 4-port microwave nets., intrinsic broad-band suppression of commonmode sigs. Fathelbab, W.M., + , T-MTT May 05 1569-1575 compact bandpass filter, 2 tuning transm. zeros, CMRC resonator. Kam Man Shum, + , T-MTT Mar 05 895-900 determining TE and TM modes, closed waveguides made up of N cylindrical conductors, efficient method. de la Rubia, V., + , T-MTT Feb 05 670-678 gen. Chebyshev filters, asymmetrically located transm. zeros, design. Milosavljevic, Z.D., T-MTT Jul 05 2411-2415 improved coupled-microstrip filter design, effective even-mode/odd-mode charact. impedances. Hong-Ming Lee, + , T-MTT Sep 05 2812-2818 in-line dual- and triple-mode cavity filters, nonresonating nodes. Amari, S., + , T-MTT Apr 05 1272-1279 in-line N-order filters, N real transm. zeros by of extracted poles implemented, low-cost rect. H-plane waveguide, synthesis and design. Montejo-Garai, J.R., + , T-MTT May 05 1636-1642 miniaturized planar Marchand baluns, classes. Fathelbab, W.M., + , TMTT Apr 05 1211-1220 planar duplexers/triplexers by manipulating atten. poles, design methodologies. Ohno, T., + , T-MTT Jun 05 2088-2095 RF/microwave multifunctional systs., reconfigurable bandpass filter. Fathelbab, W.M., + , T-MTT Mar 05 1111-1116 serial config., 2 finite transm. zeros, LTCC technol., bandpass filter. Chun-Fu Chang, + , T-MTT Jul 05 2383-2388 synthesis of cascaded N-tuplets cross-coupled resonators microwave filters, matrix rotations, anal. tech. Tamiazzo, S., + , T-MTT May 05 1693-1698 Portable instruments adaptive power controllable retrodirective array system for wireless sensor server applications. Lim, S., + , T-MTT Dec 05 3735-3743 Powders scan antennas, ferroelec. substr., rigorous anal. and investigs. Yashchyshyn, Y., + , T-MTT Feb 05 427-438 Power amplifiers adaptive predistortion RF power amp., spectrum monitor for multicarrier WCDMA appls. Seung-Yup Lee, + , T-MTT Feb 05 786-793 asymmetrical-cells-based lin. Doherty power Amplifiers-uneven power drive and power matching, optimum operation. Jangheon Kim, + , TMTT May 05 1802-1809 BJT class-F power amp. near transit. freq. Rudiakova, A.N., T-MTT Sep 05 3045-3050 complex enveloped sigs. and appl., feedforward cct. anal., multitone model. Coskun, A.H., + , T-MTT Jun 05 2171-2178 distrib. act. transformer, optimized design. Seungwoo Kim, + , T-MTT Jan 05 380-388 dual-band transmitters, digitally predistorted freq. multipliers for reconfigurable radios. Youngcheol Park, + , T-MTT Jan 05 115-122 EDGE terminal power amps., memoryless digital predistortion, optim. Ceylan, N., + , T-MTT Feb 05 515-522 fully integrated transmitter front-end with high power-added efficiency. Kim, H., + , T-MTT Oct 05 3206-3214 high av.-effic. SiGe HBT power amp. for WCDMA handset appls. Junxiong Deng, + , T-MTT Feb 05 529-537

+ Check author entry for coauthors

LINC amps., Chireix-outphasing combiners, phase-only predistortion. Birafane, A., + , T-MTT Jun 05 2240-2250 multichannel commun. systs., modeling distortion. Gharaibeh, K.M., + , T-MTT May 05 1682-1692 nonlin. device charactn., multitone phase and amplit. meas. Martins, J.P., + , T-MTT Jun 05 1982-1989 power level-depend. dual-operating mode LDMOS power amp. for CDMA wireless base-station appls. Younkyu Chung, + , T-MTT Feb 05 739-746 wide-bandwidth envelope-tracking power amps. for OFDM appls., design. Feipeng Wang, + , T-MTT Apr 05 1244-1255 Power amplifiers; cf. Microwave power amplifiers; Millimeter wave power amplifiers; UHF power amplifiers Power bipolar transistors high av.-effic. SiGe HBT power amp. for WCDMA handset appls. Junxiong Deng, + , T-MTT Feb 05 529-537 InGaP/GaAs HBT RF power amps., ESD protection design considerations. Ma, Y., + , T-MTT Jan 05 221-228 Power combiners freq. multiplier, waveguide-based spatial power-combining archit. Belaid, M., + , T-MTT Apr 05 1124-1129 LINC amps., Chireix-outphasing combiners, phase-only predistortion. Birafane, A., + , T-MTT Jun 05 2240-2250 LINC radio transmitters, integr. antenna/power combiner. Gao, S., + , TMTT Mar 05 1083-1088 power level-depend. dual-operating mode LDMOS power amp. for CDMA wireless base-station appls. Younkyu Chung, + , T-MTT Feb 05 739-746 ring-hybrid microwave voltage-variable attenuator, HFET transistors. Saavedra, C.E., + , T-MTT Jul 05 2430-2434 Power control multimode J-pHEMT front-end archit., power-control scheme for max. effic. Clifton, J.C., + , T-MTT Jun 05 2251-2258 Power conversion modified loss-compensation method, 0.35-ȝm SiGe BiCMOS technol., broad-band MMICs. Ming-Da Tsai, + , T-MTT Feb 05 496-505 Power demand high-power high-effic. SiGe Ku- and Ka-band balanced freq. doublers. Juo-Jung Hung, + , T-MTT Feb 05 754-761 linearizing FET low-noise amps., modified derivative superposition method. Aparin, V., + , T-MTT Feb 05 571-581 modified loss-compensation method, 0.35-ȝm SiGe BiCMOS technol., broad-band MMICs. Ming-Da Tsai, + , T-MTT Feb 05 496-505 multistandard adaptive voltage-controlled oscillators, design. Tasic, A., + , T-MTT Feb 05 556-563 power level-depend. dual-operating mode LDMOS power amp. for CDMA wireless base-station appls. Younkyu Chung, + , T-MTT Feb 05 739-746 Power dividers compact EBG in-phase hybrid-ring equal power divider. Ban-Leong Ooi, T-MTT Jul 05 2329-2334 compact MMIC CPW and asymmetric CPS branch-line couplers and Wilkinson dividers, shunt and series stub loading. Hettak, K., + , T-MTT May 05 1624-1635 harmonics, Wilkinson power divider, dual-band rejection by asymmetric DGS, suppression. Duk-Jae Woo, + , T-MTT Jun 05 2139-2144 power level-depend. dual-operating mode LDMOS power amp. for CDMA wireless base-station appls. Younkyu Chung, + , T-MTT Feb 05 739-746 substr. integr. waveguide technol., 6-port jn. Xinyu Xu, + , T-MTT Jul 05 2267-2273 Power electronics high-power MEMS varactors and impedance tuners for mm-wave appls. Lu, Y., + , T-MTT Nov 05 3672-3678 InP/InGaAs DHBT distrib. amps., modulator drivers for 80-Gbit/s operation, comp. Schneider, K., + , T-MTT Nov 05 3378-3387 Power electronics; cf. Power integrated circuits; Power semiconductor devices Power FETs short-channel AlGaN/GaN heterojunction FETs, 30-GHz-band, 5-W power perform. Inoue, T., + , T-MTT Jan 05 74-80 Power integrated circuits sig. gener., control, freq. conversion AlGaN/GaN HEMT MMICs. Kaper, V.S., + , T-MTT Jan 05 55-65

IEEE T-MTT 2005 INDEX — 54 Power MESFETs 2-stage ultrawide-band 5-W power amp., SiC MESFET. Sayed, A., + , TMTT Jul 05 2441-2449 Power MODFETs expt. class-F power amp. design, computationally efficient and accurate large-sig. pHEMT model. Wren, M., + , T-MTT May 05 1723-1731 Power MOSFETs asymmetrical-cells-based lin. Doherty power Amplifiers-uneven power drive and power matching, optimum operation. Jangheon Kim, + , TMTT May 05 1802-1809 highly efficient Doherty feedforward lin. power amp. for W-CDMA basestation appls. Kyoung-Joon Cho, + , T-MTT Jan 05 292-300 hybrid digital/RF envelope predistortion linearization syst. for power amps. Wangmyong Woo, + , T-MTT Jan 05 229-237 Power semiconductor devices power level-depend. dual-operating mode LDMOS power amp. for CDMA wireless base-station appls. Younkyu Chung, + , T-MTT Feb 05 739-746 Power supplies to apparatus; cf. Accelerator RF systems Power system transmission control ground on power-line communications. Luo, W.Q., + , T-MTT Oct 05 3191-3198 Power transistors; cf. Power bipolar transistors Printed circuit design; cf. Printed circuit layout Printed circuit layout PCB discontinuities, wavelet domain, 2-port equiv. Araneo, R., + , T-MTT Mar 05 907-918 Printed circuits dielec. const., loss tangent, surface resist. of PCB materials, K-band freqs. Egorov, V.N., + , T-MTT Feb 05 627-635 low-loss broad-band planar baluns, multilayered organic thin films. Chen, A.C., + , T-MTT Nov 05 3648-3655 method, improve VNA calib., planar dispers. media, adding, asymmetrical reciprocal device. Scott, J.B., T-MTT Sep 05 3007-3013 RF switch matrix appls., integr. interconnect nets. Daneshmand, M., + , TMTT Jan 05 12-21 Programmable logic arrays; cf. Field programmable gate arrays Proteins protein thermal unfolding and refolding, near-zone microwaves, ultrasensitive detect. Taylor, K.M., + , T-MTT May 05 1576-1586 Pulse amplitude modulation 20-Gb/s 4-PAM backplane serial I/O interconnections, equalization and NEXT noise cancellation. Hur, Y., + , T-MTT Jan 05 246-255 Pulse circuits; cf. Digital circuits; Driver circuits Pulse generation coupled-slotline-hybrid sampling mixer integr., step-recovery-diode pulse generator for UWB appls. Jeongwoo Han, + , T-MTT Jun 05 1875-1882 Pulse modulation; cf. Pulse amplitude modulation Q Q factor CMOS technol., complementary Colpitts oscillator. Choong-Yul Cha, + , T-MTT Mar 05 881-887 compact intell. RF front-end, reconfigurable RFICs, Si-based technols. Mukhopadhyay, R., + , T-MTT Jan 05 81-93 complex permitt. meas., TE11p modes, circ. cylindrical cavities. Zinal, S., + , T-MTT Jun 05 1870-1874 complex permitt. of arbitrary shape and size dielec. samples, cavity meas. tech., microwave freqs., estim. Santra, M., + , T-MTT Feb 05 718-722 cylindrical multilayered ceramic resonators, rect. air cavity for low-phase noise K/Ka-band oscillators. El-Tager, A.M., + , T-MTT Jun 05 22112219 dielec. consts. of metallic nanoparticles embedded, paraffin rod, microwave freqs., meas. Yan-Shian Yeh, + , T-MTT May 05 1756-1760 dielec. helical resonators. Holmes, J.E., + , T-MTT Jan 05 322-329 high-effic. current-mode class-D amps. for wireless handsets, design. TsaiPi Hung, + , T-MTT Jan 05 144-151 high-quality solenoid inductor, dielec. film for multichip modules. JongMin Yook, + , T-MTT Jun 05 2230-2234 lossy Foster networks. Kajfez, D., T-MTT Oct 05 3199-3205 low-loss 2-bit tunable bandpass filters, MEMS DC contact switches. Pothier, A., + , T-MTT Jan 05 354-360 low-Q microwave resonator, crit.-points method, accurate charactn. Peng Wang, + , T-MTT Jan 05 349-353 + Check author entry for coauthors

microwave filters, improved stopband, sub-wavel. resonators. GarciaGarcia, J., + , T-MTT Jun 05 1997-2006 mm-wave appls., extremely high-Q factor dielec. resonators. Krupka, J., + , T-MTT Feb 05 702-712 multilayer on-chip inductors, phys. anal. model. Tong, K.Y., + , T-MTT Apr 05 1143-1149 multiple-metal stacked inductors incorporating, extended phys. model, design. Murphy, O.H., + , T-MTT Jun 05 2063-2072 on-chip vert. solenoid inductor design for multigigahertz CMOS RFIC. Hau-Yiu Tsui, + , T-MTT Jun 05 1883-1890 pad-open-short and open-short-load deembedding techs. for accurate onwafer RF charactn. of high-quality passives, comp. Tiemeijer, L.F., + , T-MTT Feb 05 723-729 Q-factor def. and eval. for spiral inductors fabricated using wafer-level CSP technology. Aoki, Y., + , T-MTT Oct 05 3178-3184 simple systematic spiral inductor design, perfected Q improv. for CMOS RFIC appl. Chih-Yuan Lee, + , T-MTT Feb 05 523-528 substr. integr. waveguide cavity filter, defected ground struct. Yu Lin Zhang, + , T-MTT Apr 05 1280-1287 Q-factor measurement insertion loss function synthesis of maximally flat parallel-coupled line bandpass filters. Chin, K.-S., + , T-MTT Oct 05 3161-3168 Quadratic programming comparison of two optimiz. techniques for the est. of complex permittivities of multilayered struc. using waveguide meas. Baginski, M.E., + , T-MTT Oct 05 3251-3259 Quadrature amplitude modulation act. differential broad-band phase splitter for quadrature-modulator appls. Tiiliharju, E., + , T-MTT Feb 05 679-686 software-defined direct conversion receiver, ka-band analog front-end. Tatu, S.O., + , T-MTT Sep 05 2768-2776 Quadrature phase shift keying 2.14-GHz Chireix outphasing transmitter. Hakala, I., + , T-MTT Jun 05 2129-2138 Quartz mm-wave appls., extremely high-Q factor dielec. resonators. Krupka, J., + , T-MTT Feb 05 702-712 nonuniform plasma layer model for quartz-Si image guide phase shifters, expt. verification. Fickenscher, T., + , T-MTT Jul 05 2375-2382 R Radar 15th International Conference on Microwaves, Radar, and Wireless Communications, MIKON (special section). T-MTT Feb 05 425-450 15th International Conference on Microwaves, Radar, and Wireless Communications, MIKON (special section intro.). Modelski, J.W., TMTT Feb 05 425-426 Radar; cf. Adaptive radar; Doppler radar; Meteorological radar; Phased array radar; Search radar Radar antennas advanced multifunction RF concept. Tavik, G.C., + , T-MTT Mar 05 10091020 Radar equipment; cf. Radar antennas; Radar receivers Radar imaging/mapping breast cancer detect., expt. investig. of simple tumor models, tissue sens. adaptive radar. Sill, J.M., + , T-MTT Nov 05 3312-3319 Radar receivers AMRFC test-bed, high-band digital preprocessor (HBDP). Mazumder, S., + , T-MTT Mar 05 1065-1071 Radiation effects; cf. Biological effects of radiation Radio 2004 IEEE Radio Frequency Integrated Circuits Symposium, RFIC (special section). T-MTT Feb 05 451-626 2004 IEEE Radio Frequency Integrated Circuits Symposium, RFIC (special section intro.). Quach, T., T-MTT Feb 05 451-452 Radio applications; cf. Radiocommunication Radio astronomy supercond. spiral filters, quasiellipt. charact. for radio astron. Guoyong Zhang, + , T-MTT Mar 05 947-951 Radiocommunication 2004 IEEE Radio Frequency Integrated Circuits Symposium, RFIC (special section). T-MTT Feb 05 451-626 2004 IEEE Radio Frequency Integrated Circuits Symposium, RFIC (special section intro.). Quach, T., T-MTT Feb 05 451-452

IEEE T-MTT 2005 INDEX — 55 high-effic. multistage Doherty power amp. for wireless communs., anal. and design. Srirattana, N., + , T-MTT Mar 05 852-860 increasing wireless channel capacity, MIMO systs. employing colocated antennas. Konanur, A.S., + , T-MTT Jun 05 1837-1844 Radio communication equipment 60-GHz broad-band telecomm., resistive HEMT mixers. Varonen, M., + , T-MTT Apr 05 1322-1330 highly integr. mm-wave pass. components, 3D LTCC syst.-on-package (SOP) technol. Jong-Hoon Lee, + , T-MTT Jun 05 2220-2229 Radio equipment; cf. Radio receivers; Radio transmitters; Transceivers Radiofrequency amplifiers; cf. Microwave amplifiers; Millimeter wave amplifiers; UHF amplifiers; VHF amplifiers Radiofrequency filters; cf. Microwave filters; Millimeter wave filters; UHF filters Radiofrequency integrated circuits; cf. Microwave integrated circuits; Millimeter wave integrated circuits; Submillimeter wave integrated circuits; UHF integrated circuits Radiofrequency oscillators; cf. Microwave oscillators; Millimeter wave oscillators; UHF oscillators Radiofrequency spectra multifunctional RF systems (special issue). T-MTT Mar 05 1005-1116 multifunctional RF systems (special issue intro.). Adler, E.D., + , T-MTT Mar 05 1005-1008 Radiofrequency spectroscopy; cf. Microwave spectroscopy; Submillimeter wave spectroscopy Radio interferometry nulling interferometer, single Martin-Puplett interferometer for DSB operation, design and simul. Luthi, T., + , T-MTT Apr 05 1168-1173 Radiometers; cf. Bolometers Radiometry excitable tissues inside human body, focused microwave radiometry, functional noninvasive imaging. Reznik, A.N., T-MTT May 05 18291831 excitable tissues inside human body, focused microwave radiometry ), functional noninvasive imaging. Karanasiou, I.S., + , T-MTT May 05 1831-1832 radiometric Millimeter-wave detect. via opt. upconversion and carrier suppression. Schuetz, C.A., + , T-MTT May 05 1732-1738 stereoscopic pass. mm-wave imaging/ranging. Luthi, T., + , T-MTT Aug 05 2594-2599 Radio propagation FVTD method exploiting, flux-splitting algm., field-based scatt.-matrix extr. scheme. Baumann, D., + , T-MTT Nov 05 3595-3605 Radio receivers 2.4-GHz-band GFSK appls., low-power highly digitized receiver. Bergveld, H.J., + , T-MTT Feb 05 453-461 24-GHz SiGe phased-array receiver-LO phase-shifting approach. Hashemi, H., + , T-MTT Feb 05 614-626 6-port, communs. receiver. Hentschel, T., T-MTT Mar 05 1039-1047 AMRFC test-bed, high-band digital preprocessor (HBDP). Mazumder, S., + , T-MTT Mar 05 1065-1071 compact intell. RF front-end, reconfigurable RFICs, Si-based technols. Mukhopadhyay, R., + , T-MTT Jan 05 81-93 effectiveness of wave absorbers, improve DSRC EM environ., express highway. Pokharel, R.K., + , T-MTT Sep 05 2726-2731 GSM/EGSM/DCS/PCS direct conversion receiver, integr. synthesizer. Young-Jin Kim, + , T-MTT Feb 05 606-613 low-IF receivers, image-rejection down-converter. Sher Jiun Fang, + , TMTT Feb 05 478-487 measuring BER of UWB devices, prod. test tech. Bhattacharya, S., + , TMTT Nov 05 3474-3481 multistandard mobile terminals, fully integr. receivers requirements and archits. Brandolini, M., + , T-MTT Mar 05 1026-1038 Radio spectrum management; cf. Frequency allocation Radio transmitters 2.14-GHz Chireix outphasing transmitter. Hakala, I., + , T-MTT Jun 05 2129-2138 deembedding static nonlinearities and accurately identifying and modeling memory effects, wide-band RF transmitters. Taijun Liu, + , T-MTT Nov 05 3578-3587 EDGE terminal power amps., memoryless digital predistortion, optim. Ceylan, N., + , T-MTT Feb 05 515-522 effectiveness of wave absorbers, improve DSRC EM environ., express highway. Pokharel, R.K., + , T-MTT Sep 05 2726-2731

+ Check author entry for coauthors

effic. of OFDM transmitters, RF/DSP design. Helaoui, M., + , T-MTT Jul 05 2355-2361 LINC amps., Chireix-outphasing combiners, phase-only predistortion. Birafane, A., + , T-MTT Jun 05 2240-2250 LINC radio transmitters, integr. antenna/power combiner. Gao, S., + , TMTT Mar 05 1083-1088 Maximum Output control method for UMTS downlink transmitters, adaptive feedforward amp. Legarda, J., + , T-MTT Aug 05 2481-2486 offset-PLL output spur spectrum, systematic anal. Ching-Feng Lee, + , TMTT Sep 05 3024-3034 single and multicarrier W-CDMA sigs., LINC digital component separator. Gerhard, W., + , T-MTT Jan 05 274-282 X-band class-E power amps., EER operation, linearity. Narisi Wang, + , T-MTT Mar 05 1096-1102 Radiowave propagation; cf. Microwave propagation; Millimeter wave propagation Radiowaves; cf. Microwaves; Millimeter waves Rain inform., turbulence, rain by Doppler-polarimetric Radar, retrieval. Yanovsky, F.J., + , T-MTT Feb 05 444-450 Randomized algorithms; cf. Genetic algorithms Random noise design of 10-Gb/s AGC amp., jitter considerations. Kucharski, D., + , TMTT Feb 05 590-597 low-power ka-band Voltage-controlled oscillator implemented, 200-GHz SiGe HBT technol. Yi-jan Emery Chen, + , T-MTT May 05 1672-1681 noise and oscillator phase noise, SiGe HBTs, scaling and technol. limitations. Guofu Niu, + , T-MTT Feb 05 506-514 push-push/triple-push oscillators for reducing 1/f noise upconversion, design. Jonghoon Choi, + , T-MTT Nov 05 3407-3414 transistor nonlinearities, noise props. Sungjae Lee, + , T-MTT Apr 05 1314-1321 Random noise; cf. Shot noise Rayleigh-Ritz methods; cf. Galerkin method Ray tracing effectiveness of wave absorbers, improve DSRC EM environ., express highway. Pokharel, R.K., + , T-MTT Sep 05 2726-2731 Reactors (electric); cf. Capacitors; Inductors Receivers AMRFC test-bed, high-band digital preprocessor (HBDP). Mazumder, S., + , T-MTT Mar 05 1065-1071 tunable combline filter, continuous control of center freq. and bandwidth. Sanchez-Renedo, M., + , T-MTT Jan 05 191-199 Receivers; cf. Microwave receivers; Optical receivers; Radar receivers; Radio receivers; Repeaters; Submillimeter wave receivers; Transceivers; Transponders Rectangular waveguides bridged NRD-guide coupler for mm-wave appls., anal. and design. Duochuan Li, + , T-MTT Aug 05 2546-2551 current probe transit. from grounded coplanar, substr. integr. rect. waveguides, anal. and design. Deslandes, D., + , T-MTT Aug 05 24872494 in-line N-order filters, N real transm. zeros by of extracted poles implemented, low-cost rect. H-plane waveguide, synthesis and design. Montejo-Garai, J.R., + , T-MTT May 05 1636-1642 mode-matching analysis of a shielded rectangular dielectric-rod waveguide. Wells, C.G., + , T-MTT Oct 05 3169-3177 multifrequency waveguide orthomode transducer. Sharma, S.B., + , TMTT Aug 05 2604-2609 radially symmetric metal insets, fast S-domain modeling. Mira, F., + , TMTT Apr 05 1294-1303 reactance of hollow, solid, and hemispherical-cap cylindrical posts in rectangular waveguide. Roelvink, J., + , T-MTT Oct 05 3156-3160 rough rect. waveguide flanges, pass.-intermodulation anal. Vicente, C., + , T-MTT Aug 05 2515-2525 substr. integr. waveguide, guided-wave and leakage characts. Feng Xu, + , T-MTT Jan 05 66-73 substr. integr. waveguide technol., 6-port jn. Xinyu Xu, + , T-MTT Jul 05 2267-2273 waveguide, dielec.-filled corrugations supporting backward waves. Eshrah, I.A., + , T-MTT Nov 05 3298-3304 Rectangular waveguides; cf. Ridge waveguides

IEEE T-MTT 2005 INDEX — 56 Recursive estimation nondestructive in situ S-param. meas. of hermetically encapsulated packages, recursive un-termination method. Pfeiffer, U.R., + , T-MTT Jun 05 1845-1855 Reduced order systems comput. time-domain sensitivity of multiport systs. described by reducedorder models, adjoint-based approach. Ahmed, T., + , T-MTT Nov 05 3538-3547 wide-band finite-element model-order reduction, fast waveguide eigenanalysis. Shih-Hao Lee, + , T-MTT Aug 05 2552-2558 Reflection waveguide polarizers, design tool. Virone, G., + , T-MTT Mar 05 888-894 Reflectometry; cf. Microwave reflectometry Refraction hyperb. transm.-line periodic grids, neg. refr. and focusing. Eleftheriades, G.V., + , T-MTT Jan 05 396-403 Reliability capacitive RF MEMS, reliab. modeling. Melle, S., + , T-MTT Nov 05 3482-3488 Reliability; cf. Semiconductor device reliability Repeaters adaptive predistortion RF power amp., spectrum monitor for multicarrier WCDMA appls. Seung-Yup Lee, + , T-MTT Feb 05 786-793 Resistors InGaP/GaAs HBT RF power amps., ESD protection design considerations. Ma, Y., + , T-MTT Jan 05 221-228 Resonance EM field-mapping syst., reson.-suppressed mag. field probe. Jung-Min Kim, + , T-MTT Sep 05 2693-2699 Resonator filters compact bandpass filter, 2 tuning transm. zeros, CMRC resonator. Kam Man Shum, + , T-MTT Mar 05 895-900 compact parallel-coupled microstrip bandpass filters, lumped-element Kinverters. Yo-Shen Lin, + , T-MTT Jul 05 2324-2328 init. design of LTCC filters, all-capacitive coupling, simplified anal. tech. Rambabu, K., + , T-MTT May 05 1787-1791 inline filters with arbitrarily placed attenuation poles. Amari, S., + , TMTT Oct 05 3075-3081 microwave filters, adaptive models and param. extr., seq. tuning. Pepe, G., + , T-MTT Jan 05 22-31 miniaturized dual-mode bandpass filter struct., shunt-capacitance perturb. Ming-Fong Lei, + , T-MTT Mar 05 861-867 miniaturized planar Marchand baluns, classes. Fathelbab, W.M., + , TMTT Apr 05 1211-1220 RF low-noise bandpass filter, act. capacitance cct., design. Young-Hoon Chun, + , T-MTT Feb 05 687-695 supercond. spiral wide bandpass filters, wide upper stopband. Huang, F., T-MTT Jul 05 2335-2339 tunable combline filter, continuous control of center freq. and bandwidth. Sanchez-Renedo, M., + , T-MTT Jan 05 191-199 wide tuning-range planar filters, lumped-distrib. coupled resonators. Carey-Smith, B.E., + , T-MTT Feb 05 777-785 Resonator filters; cf. Cavity resonator filters Resonators 1D LHM-RHM resonator, expt. realization. Yan Li, + , T-MTT Apr 05 1522-1526 1st. circ. conds. of turnstile waveguide circulators, finite-element solver, verification. Helszajn, J., + , T-MTT Jul 05 2309-2316 double-neg. metamaterials, pos. future. Nader Engheta, + , T-MTT Apr 05 1535-1556 high-effic. current-mode class-D amps. for wireless handsets, design. TsaiPi Hung, + , T-MTT Jan 05 144-151 hybrid narrow-band tunable bandpass filter, varactor loaded EM-bandgap CPW. Pistono, E., + , T-MTT Aug 05 2506-2514 init. design of LTCC filters, all-capacitive coupling, simplified anal. tech. Rambabu, K., + , T-MTT May 05 1787-1791 LTCC, metallic resonators, canonical ridge waveguide filters. Ruiz-Cruz, J.A., + , T-MTT Jan 05 174-182 microstrip bandpass filters, dual-passband response, design. Jen-Tsai Kuo, + , T-MTT Apr 05 1331-1337 microwave filters, improved stopband, sub-wavel. resonators. GarciaGarcia, J., + , T-MTT Jun 05 1997-2006 model for the mode-splitting effect in whispering-gallery-mode resonators. Bourgeois, P.-Y., + , T-MTT Oct 05 3185-3190

+ Check author entry for coauthors

parallel-coupled line filters with enhanced stopband performances. Fathelbab, W.M., + , T-MTT Dec 05 3774-3781 planar duplexers/triplexers by manipulating atten. poles, design methodologies. Ohno, T., + , T-MTT Jun 05 2088-2095 planar filter design, fully controllable second passband. Chih-Ming Tsai, + , T-MTT Nov 05 3429-3439 split-ring resonators and complementary split-ring resonators coupled, planar transm. lines, equiv.-cct. models. Baena, J.D., + , T-MTT Apr 05 1451-1461 synthesis of cascaded N-tuplets cross-coupled resonators microwave filters, matrix rotations, anal. tech. Tamiazzo, S., + , T-MTT May 05 1693-1698 Resonators; cf. Cavity resonators; Dielectric resonators Ridge waveguides dielec.-filled cavity filters, ultrawide stopband Characteristics, design. Rauscher, C., T-MTT May 05 1777-1786 LTCC, metallic resonators, canonical ridge waveguide filters. Ruiz-Cruz, J.A., + , T-MTT Jan 05 174-182 S Sampling methods; cf. Signal sampling Sapphire dielec. const., loss tangent, surface resist. of PCB materials, K-band freqs. Egorov, V.N., + , T-MTT Feb 05 627-635 electronically tunable microstrip bandpass filter, thin-film BariumStrontium-Titanate (BST) varactors. Nath, J., + , T-MTT Sep 05 27072712 Satellite broadcasting DVB-S appls., Si bipolar technol., monolithic 12-GHz heterodyne receiver. Girlando, G., + , T-MTT Mar 05 952-959 Scattering parameters 2D nonuniform FFT (2-D NUFFT) tech., anal. of shielded microstrip ccts. Ke-Ying Su, + , T-MTT Mar 05 993-999 act. predistorter suitable for MMIC implement. Iommi, R., + , T-MTT Mar 05 874-880 coupled strip-slot guiding structs., full-wave anal. Deleniv, A.N., T-MTT Jun 05 1904-1912 extracting causal time-domain params., iter. methods. Shuiping Luo, + , TMTT Mar 05 969-976 FVTD method exploiting, flux-splitting algm., field-based scatt.-matrix extr. scheme. Baumann, D., + , T-MTT Nov 05 3595-3605 large-sig. diode modeling, alternative param.-extr. tech. Yew Hui Liew, + , T-MTT Aug 05 2633-2638 meas. of scatt. params. of N-ports, multiport method. Rolfes, I., + , T-MTT Jun 05 1990-1996 microstrip lines by virtual transm. line, FDTD, efficient excit. Karkkainen, M.K., T-MTT Jun 05 1899-1903 microstrip nets., orthogonality-based deembedding tech. Spowart, M.P., + , T-MTT Mar 05 938-946 microwave mixers, scatt.-param. models and representations. Williams, D.F., + , T-MTT Jan 05 314-321 microwave on-wafer charactn. of deep-submicrometer Si MOSFETs, shield-based 3-port de-embedding method. Ming-Hsiang Cho, + , TMTT Sep 05 2926-2934 mm-wave (Bi)CMOS IC, 30-100-GHz inductors and transformers. Dickson, T.O., + , T-MTT Jan 05 123-133 mm-wave CMOS cct. design. Shigematsu, H., + , T-MTT Feb 05 472-477 nondestructive in situ S-param. meas. of hermetically encapsulated packages, recursive un-termination method. Pfeiffer, U.R., + , T-MTT Jun 05 1845-1855 nonlin. params. of dispers. APD, pulsed RF meas. and quasiDC opt. excit., extr. Ghose, A., + , T-MTT Jun 05 2082-2087 pad-open-short and open-short-load deembedding techs. for accurate onwafer RF charactn. of high-quality passives, comp. Tiemeijer, L.F., + , T-MTT Feb 05 723-729 PCB discontinuities, wavelet domain, 2-port equiv. Araneo, R., + , T-MTT Mar 05 907-918 periodically nonuniform coupled microstrip-line filters, harmonic suppression, transm. zero reallocation. Sheng Sun, + , T-MTT May 05 1817-1822 periodically nonuniform coupled microstrip lines-even and odd modes, guided-wave characts. Sheng Sun, + , T-MTT Apr 05 1221-1227 SiGe HBTs, small-sig. and HF noise modeling. Basaran, U., + , T-MTT Mar 05 919-928

IEEE T-MTT 2005 INDEX — 57 small-sig. modeling approach applied, GaN devices. Jarndal, A., + , TMTT Nov 05 3440-3448 stopband-enhanced and size-miniaturized low-pass filters, high-impedance property of offset finite-ground microstrip line. Sheng Sun, + , T-MTT Sep 05 2844-2850 substr. removed LV high-speed GaAs/AlGaAs electrooptic modulators, Trail electrodes. JaeHyuk Shin, + , T-MTT Feb 05 636-643 Schottky diode mixers integr. Si Schottky mixer diodes, cutoff freqs. above 1 THz. Morschbach, M., + , T-MTT Jun 05 2013-2018 Schottky diodes 540-640-GHz high-effic. 4-anode freq. tripler. Maestrini, A., + , T-MTT Sep 05 2835-2843 broad-band 180° phase shifters, integr. submillimeter-wave Schottky diodes. Zhiyang Liu, + , T-MTT Sep 05 2949-2955 InGaP/GaAs HBT RF power amps., ESD protection design considerations. Ma, Y., + , T-MTT Jan 05 221-228 Si Schottky diode DEMUX cct. for high bit-rate opt. receivers. Jung Han Choi, + , T-MTT Jun 05 2033-2042 Schottky diodes; cf. Schottky diode mixers Search radar armored vehicle appl., multifunction mm-wave systs. Wehling, J.H., TMTT Mar 05 1021-1025 Semiconductor device modeling efficient anal. formulation and sensitivity anal. of neuro-space mapping for nonlin. microwave device modeling. Lei Zhang, + , T-MTT Sep 05 2752-2767 enhanced high-current VBIC model. Ce-Jun Wei, + , T-MTT Apr 05 12351243 expt. class-F power amp. design, computationally efficient and accurate large-sig. pHEMT model. Wren, M., + , T-MTT May 05 1723-1731 GaInP-GaAs HBT for accurate predict. of phase noise, oscillators, advanced LF noise model. Nallatamby, J.-C., + , T-MTT May 05 16011612 high av.-effic. SiGe HBT power amp. for WCDMA handset appls. Junxiong Deng, + , T-MTT Feb 05 529-537 InP-InGaAs HBTs, microwave noise modeling. Escotte, L., + , T-MTT Jan 05 415-416 InP-InGaAs HBTs ), microwave noise modeling. Jianjun Gao, + , T-MTT Jan 05 417 large-sig. diode modeling, alternative param.-extr. tech. Yew Hui Liew, + , T-MTT Aug 05 2633-2638 microwave on-wafer charactn. of deep-submicrometer Si MOSFETs, shield-based 3-port de-embedding method. Ming-Hsiang Cho, + , TMTT Sep 05 2926-2934 nonlin. params. of dispers. APD, pulsed RF meas. and quasiDC opt. excit., extr. Ghose, A., + , T-MTT Jun 05 2082-2087 presence of LF dispers. effects, accurate pHEMT nonlin. modeling. Raffo, A., + , T-MTT Nov 05 3449-3459 RF-driven gate current, DC/RF perform., GaAs pHEMT MMIC power amps., effect. Yeong-Chang Chou, + , T-MTT Nov 05 3398-3406 SiGe HBTs, small-sig. and HF noise modeling. Basaran, U., + , T-MTT Mar 05 919-928 small-sig. modeling approach applied, GaN devices. Jarndal, A., + , TMTT Nov 05 3440-3448 sub-threshold anal. and drain current modeling of polysilicon TFT, Green's fn. approach. Sehgal, A., + , T-MTT Sep 05 2682-2687 thermal resist. of FETs, accurate determ. Darwish, A.M., + , T-MTT Jan 05 306-313 time-domain large-sig. meas., table-based nonlin. HEMT model extracted. Curras-Francos, M.C., T-MTT May 05 1593-1600 TW photodetectors, hybrid drift-diffusion-TLM anal. Pasalic, D., + , TMTT Sep 05 2700-2706 Semiconductor device noise 1/f noise and oscillator phase noise, SiGe HBTs, scaling and technol. limitations. Guofu Niu, + , T-MTT Feb 05 506-514 carrier heating, channel noise, deep-submicrometer NMOSFETs via body bias, expt. study. Hong Wang, + , T-MTT Feb 05 564-570 design of 10-Gb/s AGC amp., jitter considerations. Kucharski, D., + , TMTT Feb 05 590-597 epitaxial struct., noise fig. of AlGaN/GaN HEMTs. Sanabria, C., + , TMTT Feb 05 762-769 GaInP-GaAs HBT for accurate predict. of phase noise, oscillators, advanced LF noise model. Nallatamby, J.-C., + , T-MTT May 05 16011612 + Check author entry for coauthors

InP HBT noise params., noise-fig. meas. syst., direct extr. Jianjun Gao, + , T-MTT Jan 05 330-335 InP-InGaAs HBTs, microwave noise modeling. Escotte, L., + , T-MTT Jan 05 415-416 InP-InGaAs HBTs ), microwave noise modeling. Jianjun Gao, + , T-MTT Jan 05 417 SiGe HBTs, small-sig. and HF noise modeling. Basaran, U., + , T-MTT Mar 05 919-928 transistor nonlinearities, noise props. Sungjae Lee, + , T-MTT Apr 05 1314-1321 wide-band on-wafer noise-param. meas., improved Y-factor method. Tiemeijer, L.F., + , T-MTT Sep 05 2917-2925 Semiconductor device reliability thermal resist. of FETs, accurate determ. Darwish, A.M., + , T-MTT Jan 05 306-313 Semiconductor devices; cf. Power semiconductor devices; Semiconductor lasers Semiconductor device testing power level-depend. dual-operating mode LDMOS power amp. for CDMA wireless base-station appls. Younkyu Chung, + , T-MTT Feb 05 739-746 Semiconductor diodes; cf. Microwave diodes; Millimeter wave diodes; p-i-n diodes; Schottky diodes; Tunnel diodes; Varactors Semiconductor epitaxial layers epitaxial struct., noise fig. of AlGaN/GaN HEMTs. Sanabria, C., + , TMTT Feb 05 762-769 opt. sigs., p-i-n photodiodes, freq. conversion. Malyshev, S.A., + , T-MTT Feb 05 439-443 substr. removed LV high-speed GaAs/AlGaAs electrooptic modulators, Trail electrodes. JaeHyuk Shin, + , T-MTT Feb 05 636-643 Semiconductor lasers nonuniform plasma layer model for quartz-Si image guide phase shifters, expt. verification. Fickenscher, T., + , T-MTT Jul 05 2375-2382 Semiconductor materials 1/f noise and oscillator phase noise, SiGe HBTs, scaling and technol. limitations. Guofu Niu, + , T-MTT Feb 05 506-514 24-GHz SiGe phased-array receiver-LO phase-shifting approach. Hashemi, H., + , T-MTT Feb 05 614-626 act. differential broad-band phase splitter for quadrature-modulator appls. Tiiliharju, E., + , T-MTT Feb 05 679-686 design of 10-Gb/s AGC amp., jitter considerations. Kucharski, D., + , TMTT Feb 05 590-597 high-power high-effic. SiGe Ku- and Ka-band balanced freq. doublers. Juo-Jung Hung, + , T-MTT Feb 05 754-761 mm-wave (Bi)CMOS IC, 30-100-GHz inductors and transformers. Dickson, T.O., + , T-MTT Jan 05 123-133 modified loss-compensation method, 0.35-ȝm SiGe BiCMOS technol., broad-band MMICs. Ming-Da Tsai, + , T-MTT Feb 05 496-505 Semiconductor technology; cf. Isolation technology Semiconductor thin films; cf. Semiconductor epitaxial layers Sensitivity 1/f noise and oscillator phase noise, SiGe HBTs, scaling and technol. limitations. Guofu Niu, + , T-MTT Feb 05 506-514 comput. time-domain sensitivity of multiport systs. described by reducedorder models, adjoint-based approach. Ahmed, T., + , T-MTT Nov 05 3538-3547 efficient anal. formulation and sensitivity anal. of neuro-space mapping for nonlin. microwave device modeling. Lei Zhang, + , T-MTT Sep 05 2752-2767 multiconductor transm. lines, nonlin. terminations, delay-extr.-based sensitivity anal. Nakhla, N.M., + , T-MTT Nov 05 3520-3530 nonintuitive planar structs., fast optim. and sensitivity anal. Cormos, D., + , T-MTT Jun 05 2019-2025 RF ccts., LCP substrs., stat. anal. and diagnosis methodology. Mukherjee, S., + , T-MTT Nov 05 3621-3630 Sensors; cf. Microsensors; Microwave detectors; Photodetectors Series (mathematics); cf. Volterra series Shielding; cf. Electromagnetic shielding Shot noise 1/f noise and oscillator phase noise, SiGe HBTs, scaling and technol. limitations. Guofu Niu, + , T-MTT Feb 05 506-514 Sigma-delta modulation 2.4-GHz-band GFSK appls., low-power highly digitized receiver. Bergveld, H.J., + , T-MTT Feb 05 453-461 Signal denoising; cf. Image denoising

IEEE T-MTT 2005 INDEX — 58 Signal detection detect. and localization of mobile phones, large buildings, microwave syst. Hudec, P., + , T-MTT Jun 05 2235-2239 Signal detection; cf. Optical signal detection Signal generators gener., control, freq. conversion AlGaN/GaN HEMT MMICs. Kaper, V.S., + , T-MTT Jan 05 55-65 Signal generators; cf. Frequency synthesizers Signal processing multigigahertz parallel bus, transmit preemphasis equalization, perform. anal. and model-to-hardware correl. Beyene, W.T., + , T-MTT Nov 05 3568-3577 nonlin. device charactn., multitone phase and amplit. meas. Martins, J.P., + , T-MTT Jun 05 1982-1989 RF frequency shifting via optically switched dual-channel PZT fiber stretchers. McDermitt, C.S., + , T-MTT Dec 05 3782-3787 secure high-speed retrodirective commun. link. Goshi, D.S., + , T-MTT Nov 05 3548-3556 Signal processing; cf. Array signal processing; Signal sampling Signal reconstruction; cf. Image reconstruction Signal representations complex enveloped sigs. and appl., feedforward cct. anal., multitone model. Coskun, A.H., + , T-MTT Jun 05 2171-2178 Signal sampling calibrated 200-GHz waveform meas. Williams, D.F., + , T-MTT Apr 05 1384-1389 photoelectronic ADC. Ioakeimidi, K., + , T-MTT Jan 05 336-342 Silicon compact intell. RF front-end, reconfigurable RFICs, Si-based technols. Mukhopadhyay, R., + , T-MTT Jan 05 81-93 DVB-S appls., Si bipolar technol., monolithic 12-GHz heterodyne receiver. Girlando, G., + , T-MTT Mar 05 952-959 fast layout verification of 3D RF and mixed-sig. on-chip structs., largescale broad-band parasitic extr. Feng Ling, + , T-MTT Jan 05 264-273 high aspect ratio through-wafer interconnect vias, Si substrs., microwave charactn. and modeling. Leung, L.L.W., + , T-MTT Aug 05 2472-2480 hybrid digital/RF envelope predistortion linearization syst. for power amps. Wangmyong Woo, + , T-MTT Jan 05 229-237 integr. Si Schottky mixer diodes, cutoff freqs. above 1 THz. Morschbach, M., + , T-MTT Jun 05 2013-2018 mm-wave (Bi)CMOS IC, 30-100-GHz inductors and transformers. Dickson, T.O., + , T-MTT Jan 05 123-133 nonuniform plasma layer model for quartz-Si image guide phase shifters, expt. verification. Fickenscher, T., + , T-MTT Jul 05 2375-2382 Schottky diode DEMUX cct. for high bit-rate opt. receivers. Jung Han Choi, + , T-MTT Jun 05 2033-2042 short-channel AlGaN/GaN heterojunction FETs, 30-GHz-band, 5-W power perform. Inoue, T., + , T-MTT Jan 05 74-80 simple systematic spiral inductor design, perfected Q improv. for CMOS RFIC appl. Chih-Yuan Lee, + , T-MTT Feb 05 523-528 sub-threshold anal. and drain current modeling of polysilicon TFT, Green's fn. approach. Sehgal, A., + , T-MTT Sep 05 2682-2687 surface-passivated high-resist. Si, true microwave substr. Spirito, M., + , T-MTT Jul 05 2340-2347 Silicon alloys 12-GHz SiGe phase shifter, integr. LNA. Hancock, T.M., + , T-MTT Mar 05 977-983 1/f noise and oscillator phase noise, SiGe HBTs, scaling and technol. limitations. Guofu Niu, + , T-MTT Feb 05 506-514 24-GHz SiGe phased-array receiver-LO phase-shifting approach. Hashemi, H., + , T-MTT Feb 05 614-626 70-ps SiGe differential RF switch, design and anal. Hancock, T.M., + , TMTT Jul 05 2403-2410 act. differential broad-band phase splitter for quadrature-modulator appls. Tiiliharju, E., + , T-MTT Feb 05 679-686 compact intell. RF front-end, reconfigurable RFICs, Si-based technols. Mukhopadhyay, R., + , T-MTT Jan 05 81-93 crossed anisotropy mag. core, toroidal thin-film inductors, integrat. Frommberger, M., + , T-MTT Jun 05 2096-2100 design of 10-Gb/s AGC amp., jitter considerations. Kucharski, D., + , TMTT Feb 05 590-597 high av.-effic. SiGe HBT power amp. for WCDMA handset appls. Junxiong Deng, + , T-MTT Feb 05 529-537 high-power high-effic. SiGe Ku- and Ka-band balanced freq. doublers. Juo-Jung Hung, + , T-MTT Feb 05 754-761 + Check author entry for coauthors

low-power ka-band Voltage-controlled oscillator implemented, 200-GHz SiGe HBT technol. Yi-jan Emery Chen, + , T-MTT May 05 1672-1681 mm-wave (Bi)CMOS IC, 30-100-GHz inductors and transformers. Dickson, T.O., + , T-MTT Jan 05 123-133 modified loss-compensation method, 0.35-ȝm SiGe BiCMOS technol., broad-band MMICs. Ming-Da Tsai, + , T-MTT Feb 05 496-505 RF-CMOS and SiGe BiCMOS, WCDMA direct-conversion receiver front-end comp. Floyd, B.A., + , T-MTT Apr 05 1181-1188 RFICs, 2-port lumped-nonlin.-source model, systematic linearity anal. Qingqing Liang, + , T-MTT May 05 1745-1755 SiGe HBTs, small-sig. and HF noise modeling. Basaran, U., + , T-MTT Mar 05 919-928 wide-band SiGe HBT act. mixers, anal. and design. Johansen, T.K., + , TMTT Jul 05 2389-2397 Silicon compounds 2-stage ultrawide-band 5-W power amp., SiC MESFET. Sayed, A., + , TMTT Jul 05 2441-2449 complex permitt. of arbitrary shape and size dielec. samples, cavity meas. tech., microwave freqs., estim. Santra, M., + , T-MTT Feb 05 718-722 on-chip interconnects by double-image Green's fn. method combined, hierarchical algm., param. extr. Wenliang Dai, + , T-MTT Jul 05 24162423 short-channel AlGaN/GaN heterojunction FETs, 30-GHz-band, 5-W power perform. Inoue, T., + , T-MTT Jan 05 74-80 Silicon compounds; cf. Quartz Silicon materials/devices dielec. polariz. effects of capacitive RF MEMS switches, temp. study. Papaioannou, G., + , T-MTT Nov 05 3467-3473 freq. response of SAW filters, FDTD method. King-Yuen Wong, + , TMTT Nov 05 3364-3370 Silicon on insulator technology 90-nm VLSI SOI CMOS technol., high linearity for WLAN, 26.5-30-GHz resistive mixer. Ellinger, F., T-MTT Aug 05 2559-2565 Simulated annealing H-plane contiguous manifold output multiplexers, fictitious reactive load concept, full-wave design. Montejo-Garai, J.R., + , T-MTT Aug 05 2628-2632 Simulation; cf. Circuit simulation Skin SAR distribs., 3-layered bio-media, direct contact, water-loaded modified box-horn applicator. Gupta, R.C., + , T-MTT Sep 05 2665-2671 Skin effect differential surface admittance operator, skin effect modeling. De Zutter, D., + , T-MTT Aug 05 2526-2538 transm. lines, guaranteed pass. direct lumped-element modeling. Se-Ho You, + , T-MTT Sep 05 2826-2834 Slot antennas broad-band and circ. polarized space-filling-based slot antennas. Ghali, H.A., + , T-MTT Jun 05 1946-1950 protein thermal unfolding and refolding, near-zone microwaves, ultrasensitive detect. Taylor, K.M., + , T-MTT May 05 1576-1586 terahertz mixers and detectors, design guidelines. Focardi, P., + , T-MTT May 05 1653-1661 Slotline coupled slotline mode, finite-ground CPW, unequal ground-plane widths, excit. Ponchak, G.E., + , T-MTT Feb 05 713-717 dielec. resonator, discrete electromechanical freq. tuning. Panaitov, G.I., + , T-MTT Nov 05 3371-3377 microwave/mm-wave appls., dielec. slab Rotman lens. Jaeheung Kim, + , T-MTT Aug 05 2622-2627 Solar radiation nulling interferometer, single Martin-Puplett interferometer for DSB operation, design and simul. Luthi, T., + , T-MTT Apr 05 1168-1173 Solenoids high-quality solenoid inductor, dielec. film for multichip modules. JongMin Yook, + , T-MTT Jun 05 2230-2234 on-chip vert. solenoid inductor design for multigigahertz CMOS RFIC. Hau-Yiu Tsui, + , T-MTT Jun 05 1883-1890 Solid lasers; cf. Semiconductor lasers S-parameters perfectly matched layer implementation using bilinear transforms. Dong, X., + , T-MTT Oct 05 3098-3105 Spatial variables measurement; cf. Distance measurement

IEEE T-MTT 2005 INDEX — 59 Special issues and sections 15th International Conference on Microwaves, Radar, and Wireless Communications, MIKON (special section). T-MTT Feb 05 425-450 15th International Conference on Microwaves, Radar, and Wireless Communications, MIKON (special section intro.). Modelski, J.W., TMTT Feb 05 425-426 16th Asia-Pacific Microwave Conference, APMC'04 (special section). TMTT Sep 05 2649-2731 16th Asia-Pacific Microwave Conference, APMC'04 (special section intro.). Steer, M.B., T-MTT Sep 05 2649 2004 IEEE MTT-S International Microwave Symposium (special section). T-MTT Jan 05 3-413 2004 IEEE MTT-S International Microwave Symposium (special section intro.). Steer, M., T-MTT Jan 05 3 2004 IEEE Radio Frequency Integrated Circuits Symposium, RFIC (special section). T-MTT Feb 05 451-626 2004 IEEE Radio Frequency Integrated Circuits Symposium, RFIC (special section intro.). Quach, T., T-MTT Feb 05 451-452 2005 International Microwave Symposium, MTT-S05 (special issue). TMTT Nov 05 3264-3678 2005 International Microwave Symposium, MTT-S05 (special issue intro.). Jackson, C., T-MTT Nov 05 3264 34th European Microwave Conference, EuMC (special issue). T-MTT Jun 05 1935-2258 34th European Microwave Conference, EuMC (special issue intro.). Russer, P., T-MTT Jun 05 1935-1936 metamaterial structures, phenomena, and applications (special issue). TMTT Apr 05 1413-1556 metamaterial structures, phenomena, and applications (special issue intro.). Itoh, T., + , T-MTT Apr 05 1413-1417 multifunctional RF systems (special issue). T-MTT Mar 05 1005-1116 multifunctional RF systems (special issue intro.). Adler, E.D., + , T-MTT Mar 05 1005-1008 Spectra; cf. Radiofrequency spectra Spectral analysis noise behavior of microwave amplifiers operating under nonlinear conditions. Escotte, L., + , T-MTT Dec 05 3704-3711 nonlin. model testing, designing multisine excit. Pedro, J.C., + , T-MTT Jan 05 45-54 RF front-end characts., spectral regrowth of communs. sigs., impact. Gard, K.G., + , T-MTT Jun 05 2179-2186 transistor nonlinearities, noise props. Sungjae Lee, + , T-MTT Apr 05 1314-1321 Spectral domain analysis 2D nonuniform FFT (2-D NUFFT) tech., anal. of shielded microstrip ccts. Ke-Ying Su, + , T-MTT Mar 05 993-999 coupled strip-slot guiding structs., full-wave anal. Deleniv, A.N., T-MTT Jun 05 1904-1912 EM distribs. demonstrating asymmetry, spectral-domain dyadic Green's fn. for ferrite microstrip guided-wave structs. Krowne, C.M., T-MTT Apr 05 1345-1361 propag. characts. of cylindrical CPW, finite thickness of conductor, fullwave anal. Yamamoto, H., + , T-MTT Jun 05 2187-2195 Spectroscopy corrections on "Precision open-ended coaxial probes for In Vivo and Ex Vivo dielectric spectroscopy of biological tissues on microwave frequencies" (May 05 1713-1722). Popovic, D., + , T-MTT Sep 05 3053 SPICE PCB discontinuities, wavelet domain, 2-port equiv. Araneo, R., + , T-MTT Mar 05 907-918 Sputter etching high aspect ratio through-wafer interconnect vias, Si substrs., microwave charactn. and modeling. Leung, L.L.W., + , T-MTT Aug 05 2472-2480 Sputtering crossed anisotropy mag. core, toroidal thin-film inductors, integrat. Frommberger, M., + , T-MTT Jun 05 2096-2100 Sputtering; cf. Sputter etching Stability BJT formulation, FDTD framework. Kung, F., + , T-MTT Apr 05 11891196 global stability analysis and stabilization of a class-E/F amplifier with a distributed active transformer. Jeon, S., + , T-MTT Dec 05 3712-3722 submillimeter SIS heterodyne receiver, gain stabil. Battat, J., + , T-MTT Jan 05 389-395 Stability; cf. Circuit stability; Numerical stability + Check author entry for coauthors

Standards; cf. IEEE standards State space methods rect. waveguides, radially symmetric metal insets, fast S-domain modeling. Mira, F., + , T-MTT Apr 05 1294-1303 Static electrification; cf. Surface charging Statistical analysis; cf. Higher order statistics Statistics multichannel commun. systs., modeling distortion. Gharaibeh, K.M., + , T-MTT May 05 1682-1692 RF ccts., LCP substrs., stat. anal. and diagnosis methodology. Mukherjee, S., + , T-MTT Nov 05 3621-3630 WDM photonic microwave filters, random errors, stat. anal. Vidal, B., + , T-MTT Aug 05 2600-2603 Stellar radiation; cf. Solar radiation Stereo vision stereoscopic pass. mm-wave imaging/ranging. Luthi, T., + , T-MTT Aug 05 2594-2599 Stochastic processes particle swarm optimizer for microwave imaging of 2D dielec. scatterers, comput. approach. Donelli, M., + , T-MTT May 05 1761-1776 Stochastic processes; cf. Moving average processes Stripline compact MMIC CPW and asymmetric CPS branch-line couplers and Wilkinson dividers, shunt and series stub loading. Hettak, K., + , T-MTT May 05 1624-1635 ferrite coupled-line circulator, reduced length. Meng Cao, + , T-MTT Aug 05 2572-2579 ultrawide-band tunable true-time delay lines, ferroelec. varactors. Kuylenstierna, D., + , T-MTT Jun 05 2164-2170 Stripline circuits 540-640-GHz high-effic. 4-anode freq. tripler. Maestrini, A., + , T-MTT Sep 05 2835-2843 Strip line circuits; cf. Microstrip circuits Strip line components; cf. Microstrip components; Strip line filters Stripline couplers coupled strip-slot guiding structs., full-wave anal. Deleniv, A.N., T-MTT Jun 05 1904-1912 Strip line couplers; cf. Microstrip couplers Strip line filters quasi-lumped suspended stripline filters and diplexers. Menzel, W., + , TMTT Oct 05 3230-3237 tunable combline filter, continuous control of center freq. and bandwidth. Sanchez-Renedo, M., + , T-MTT Jan 05 191-199 Strip line filters; cf. Microstrip filters Strip line resonators; cf. Microstrip resonators Strip lines quasi-static solutions of multilayer elliptical, cylindrical coplanar striplines and multilayer coplanar striplines with finite dielectric dimensions Asymmetrical case. Akan, V., + , T-MTT Dec 05 3681-3686 Submillimeter wave circuits; cf. Submillimeter wave integrated circuits Submillimeter wave devices coupling 3D Maxwell's and Boltzmann's eqns. for analyzing, terahertz photoconductive switch. Sirbu, M., + , T-MTT Sep 05 2991-2998 terahertz mixers and detectors, design guidelines. Focardi, P., + , T-MTT May 05 1653-1661 Submillimeter wave devices; cf. Submillimeter wave mixers; Submillimeter wave receivers Submillimeter wave integrated circuits 540-640-GHz high-effic. 4-anode freq. tripler. Maestrini, A., + , T-MTT Sep 05 2835-2843 Submillimeter wave measurements hologram-based CATR, 650 GHz, expt. study. Koskinen, T., + , T-MTT Sep 05 2999-3006 Submillimeter wave mixers submillimeter SIS heterodyne receiver, gain stabil. Battat, J., + , T-MTT Jan 05 389-395 Submillimeter wave receivers nulling interferometer, single Martin-Puplett interferometer for DSB operation, design and simul. Luthi, T., + , T-MTT Apr 05 1168-1173 Submillimeter wave spectroscopy permitt. and loss tangent of high-permitt. materials, terahertz freqs., temp. depend. Berdel, K., + , T-MTT Apr 05 1266-1271 Submillimeter wave technology Wang's model for RT cond. losses, normal metals, terahertz freqs. Lucyszyn, S., T-MTT Apr 05 1398-1403

IEEE T-MTT 2005 INDEX — 60 Substrates comparison of two optimiz. techniques for the est. of complex permittivities of multilayered struc. using waveguide meas. Baginski, M.E., + , T-MTT Oct 05 3251-3259 Q-factor def. and eval. for spiral inductors fabricated using wafer-level CSP technology. Aoki, Y., + , T-MTT Oct 05 3178-3184 Superconducting devices; cf. Superconducting microwave devices Superconducting materials; cf. High-temperature superconductors Superconducting microwave devices future mobile telecomm. systs., high-temp. supercond. filter. Jia-Sheng Hong, + , T-MTT Jun 05 1976-1981 Superconducting mixers; cf. Superconductor-insulator-superconductor mixers Superconductivity; cf. High-temperature superconductors Superconductor-insulator-superconductor devices; cf. Superconductorinsulator-superconductor mixers Superconductor-insulator-superconductor mixers submillimeter SIS heterodyne receiver, gain stabil. Battat, J., + , T-MTT Jan 05 389-395 Surface acoustic wave devices adaptive duplexer implemented, single-path/multipath feedforward techs., BST phase shifters. O'Sullivan, T., + , T-MTT Jan 05 106-114 Surface acoustic wave devices; cf. Surface acoustic wave filters Surface acoustic wave filters freq. response of SAW filters, FDTD method. King-Yuen Wong, + , TMTT Nov 05 3364-3370 Surface charging capacitive RF MEMS, reliab. modeling. Melle, S., + , T-MTT Nov 05 3482-3488 Surface collisions; cf. Sputtering Surface mounting local ground plane, EM anal., deembedding effect. Rautio, J.C., T-MTT Feb 05 770-776 Surface phenomena; cf. Surface charging Surgery microwave ablation, triaxial antenna. Brace, C.L., + , T-MTT Jan 05 215220

T

Table lookup EDGE terminal power amps., memoryless digital predistortion, optim. Ceylan, N., + , T-MTT Feb 05 515-522 hybrid digital/RF envelope predistortion linearization syst. for power amps. Wangmyong Woo, + , T-MTT Jan 05 229-237 wireless transmitters, digital subband filtering predistorter archit. Hammi, O., + , T-MTT May 05 1643-1652 Telecommunication; cf. Data communication; Military communication; Mobile communication; Optical communication; Radiocommunication Telecommunication channels; cf. Fading channels; Gaussian channels; Multipath channels Telecommunication equipment; cf. Optical communication equipment; Repeaters Telecommunication networks; cf. Broadband networks Telecommunication switching; cf. Demultiplexing Temperature permitt. and loss tangent of high-permitt. materials, terahertz freqs., temp. depend. Berdel, K., + , T-MTT Apr 05 1266-1271 Testing Ka-band resonant ring for testing components for a high-gradient linear accelerator. Bogdashov, A., + , T-MTT Oct 05 3152-3155 Testing; cf. Electronic equipment testing Thermoresistivity AlGaN-GaN devices, thermal resist. calc. Wen-Yan Yin, T-MTT Sep 05 3051-3052 AlGaN-GaN devices'), 'Thermal resist. calc. Darwish, A.M., + , T-MTT Sep 05 3052-3053 FETs, accurate determ. Darwish, A.M., + , T-MTT Jan 05 306-313 IMD, contact-type MEMS microswitch, determ. Johnson, J., + , T-MTT Nov 05 3615-3620

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Thick film devices millimeter-wave substrate integrated waveguides and filters in photoimageable thick-film technology. Stephens, D., + , T-MTT Dec 05 3832-3838 Thin film circuits low-loss broad-band planar baluns, multilayered organic thin films. Chen, A.C., + , T-MTT Nov 05 3648-3655 microstrip and coplanar technols. for reduced-size MMICs, integrat. Hettak, K., + , T-MTT Jan 05 283-291 Thin film devices electronically tunable microstrip bandpass filter, thin-film BariumStrontium-Titanate (BST) varactors. Nath, J., + , T-MTT Sep 05 27072712 Thin film devices; cf. Thin film circuits; Thin film transistors Thin films capacitive nonlinearity in thin-film BST varactors. Chase, D.R., + , TMTT Oct 05 3215-3220 Thin film transistors sub-threshold anal. and drain current modeling of polysilicon TFT, Green's fn. approach. Sehgal, A., + , T-MTT Sep 05 2682-2687 Time domain analysis anal. of complex EM structs., TLM-G, grid-enabled time-domain TLM syst. Lorenz, P., + , T-MTT Nov 05 3631-3637 comput. time-domain sensitivity of multiport systs. described by reducedorder models, adjoint-based approach. Ahmed, T., + , T-MTT Nov 05 3538-3547 CPW, subterahertz freqs., atten. characts. Jingjing Zhang, + , T-MTT Nov 05 3281-3287 FET self-heating, broad-band charactn. Parker, A.E., + , T-MTT Jul 05 2424-2429 FVTD method exploiting, flux-splitting algm., field-based scatt.-matrix extr. scheme. Baumann, D., + , T-MTT Nov 05 3595-3605 lin. inverse space-mapping (LISM) algm., design lin./nonlin. RF and microwave ccts. Rayas-Sanchez, J.E., + , T-MTT Mar 05 960-968 low-noise multiresolution high-dyn. ultra-broad-band time-domain EMI meas. syst. Braun, S., + , T-MTT Nov 05 3354-3363 multiconductor transm. lines, nonlin. terminations, delay-extr.-based sensitivity anal. Nakhla, N.M., + , T-MTT Nov 05 3520-3530 parallel FFT accelerated transient field-cct. simulator. Yilmaz, A.E., + , TMTT Sep 05 2851-2865 permitt. and loss tangent of high-permitt. materials, terahertz freqs., temp. depend. Berdel, K., + , T-MTT Apr 05 1266-1271 rect. waveguide, dielec.-filled corrugations supporting backward waves. Eshrah, I.A., + , T-MTT Nov 05 3298-3304 temporally dispers. dielec., time-domain cavity oscills. supported. Aksoy, S., + , T-MTT Aug 05 2465-2471 wide-band supercond. coplanar delay lines. Yi Wang, + , T-MTT Jul 05 2348-2354 Time-domain analysis; cf. Finite difference time-domain analysis Time domain reflectometry moisture content, bulk materials, sub-nanosecond UWB pulses, noncontacting determ. Schimmer, O., + , T-MTT Jun 05 2107-2113 Titanium compounds electronically tunable microstrip bandpass filter, thin-film BariumStrontium-Titanate (BST) varactors. Nath, J., + , T-MTT Sep 05 27072712 permitt. and loss tangent of high-permitt. materials, terahertz freqs., temp. depend. Berdel, K., + , T-MTT Apr 05 1266-1271 Tomography high dielec.-contrast objs., different image-reconstruction approaches, microwave-tomographic imaging. Semenov, S.Y., + , T-MTT Jul 05 2284-2294 Traffic control; cf. Automated highways Transceivers adaptive duplexer implemented, single-path/multipath feedforward techs., BST phase shifters. O'Sullivan, T., + , T-MTT Jan 05 106-114 highly efficient Doherty feedforward lin. power amp. for W-CDMA basestation appls. Kyoung-Joon Cho, + , T-MTT Jan 05 292-300 inter-chip RF-interconnect, CPW, capacitive coupler, UWB transceiver, perform. Sun, M., + , T-MTT Sep 05 2650-2655 LNA protection, watt-level CMOS transceivers, reson. switch. Kuhn, W.B., + , T-MTT Sep 05 2819-2825 multistandard WLAN appls., dual-band RF transceiver. Chang, S.-F.R., + , T-MTT Mar 05 1048-1055

IEEE T-MTT 2005 INDEX — 61 software-defined radio systs., sig. path optim. Rykaczewski, P., + , T-MTT Mar 05 1056-1064 Transducers k-band orthomode transducer, waveguide ports and balanced coaxial probes. Engargiola, G., + , T-MTT May 05 1792-1801 multifrequency waveguide orthomode transducer. Sharma, S.B., + , TMTT Aug 05 2604-2609 Transducers; cf. Biomedical transducers Transfer functions E-plane filters and diplexers, ellipt. response for mm-wave appls. Ofli, E., + , T-MTT Mar 05 843-851 harmonics, Wilkinson power divider, dual-band rejection by asymmetric DGS, suppression. Duk-Jae Woo, + , T-MTT Jun 05 2139-2144 inter-chip RF-interconnect, CPW, capacitive coupler, UWB transceiver, perform. Sun, M., + , T-MTT Sep 05 2650-2655 mismatch errors for 400-msamples/s 80-dB SFDR time-interleaved ADC, comprehensive digital correction. Munkyo Seo, + , T-MTT Mar 05 10721082 PCB discontinuities, wavelet domain, 2-port equiv. Araneo, R., + , T-MTT Mar 05 907-918 waveguide polarizers, design tool. Virone, G., + , T-MTT Mar 05 888-894 Transformers distrib. act. transformer, optimized design. Seungwoo Kim, + , T-MTT Jan 05 380-388 mm-wave (Bi)CMOS IC, 30-100-GHz inductors and transformers. Dickson, T.O., + , T-MTT Jan 05 123-133 Transforms general lin. lumped microwave ccts., matrix theory, improved FDTD formulation. Zhenhai Shao, + , T-MTT Jul 05 2261-2266 Transforms; cf. Fourier transforms; Wavelet transforms Transient analysis dielec. polariz. effects of capacitive RF MEMS switches, temp. study. Papaioannou, G., + , T-MTT Nov 05 3467-3473 FET self-heating, broad-band charactn. Parker, A.E., + , T-MTT Jul 05 2424-2429 parallel FFT accelerated transient field-cct. simulator. Yilmaz, A.E., + , TMTT Sep 05 2851-2865 transm. lines, guaranteed pass. direct lumped-element modeling. Se-Ho You, + , T-MTT Sep 05 2826-2834 Transistor circuits; cf. Bipolar transistor circuits Transistors; cf. Bipolar transistors; Field effect transistors; Microwave transistors; Millimeter wave transistors; Phototransistors; Thin film transistors Transition metal alloys; cf. Cobalt alloys; Iron alloys Transition metal compounds; cf. Titanium compounds; Yttrium compounds Transition metals; cf. Copper Transit time devices; cf. IMPATT diodes Transmission line matrix methods anal. of complex EM structs., TLM-G, grid-enabled time-domain TLM syst. Lorenz, P., + , T-MTT Nov 05 3631-3637 FVTD method exploiting, flux-splitting algm., field-based scatt.-matrix extr. scheme. Baumann, D., + , T-MTT Nov 05 3595-3605 metamaterials, neg. refr. index, 2D shunt and 3D SCN TLM nets., modeling. So, P.P.M., + , T-MTT Apr 05 1496-1505 TE/TM propag. and losses of integr. opt. polarizer, accurate modeling. Pierantoni, L., + , T-MTT Jun 05 1856-1862 TLM-based modeling and design exploiting space mapping. Bandler, J.W., + , T-MTT Sep 05 2801-2811 TW photodetectors, hybrid drift-diffusion-TLM anal. Pasalic, D., + , TMTT Sep 05 2700-2706 Transmission lines branch-line directional coupler in the design of microwave bandpass filters. Gomez-Garcia, R., + , T-MTT Oct 05 3221-3229 broad-band suppression of TEM modes, power planes, EM-bandgap layers. Rogers, S.D., T-MTT Aug 05 2495-2505 coaxial waveguide-based left-handed transm. lines, anal., modeling, appls. Salehi, H., + , T-MTT Nov 05 3489-3497 current probe transit. from grounded coplanar, substr. integr. rect. waveguides, anal. and design. Deslandes, D., + , T-MTT Aug 05 24872494 loaded transm.-line neg.-refr.-index metamaterials, periodic FDTD anal. Kokkinos, T., + , T-MTT Apr 05 1488-1495 low-pass filters, defected ground struct., design. Jong-Sik Lim, + , T-MTT Aug 05 2539-2545

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lumped-element quadrature power splitters, mixed right/left-handed transm. lines. Kuylenstierna, D., + , T-MTT Aug 05 2616-2621 microstrip lines by virtual transm. line, FDTD, efficient excit. Karkkainen, M.K., T-MTT Jun 05 1899-1903 miniaturized microwave pass. filter incorporating multilayer synthetic quasiTEM transm. line. Hsien-Shun Wu, + , T-MTT Sep 05 2713-2720 substr. removed LV high-speed GaAs/AlGaAs electrooptic modulators, Trail electrodes. JaeHyuk Shin, + , T-MTT Feb 05 636-643 super-compact multilayered left-handed transm. line and diplexer appl. Horii, Y., + , T-MTT Apr 05 1527-1534 Transmission line theory 3-sect. transm.-line transformer, anal. and design. Chongcheawchamnan, M., + , T-MTT Jul 05 2458-2462 Chebyshev collocation and Newton-type optim. methods for inverse problem, nonuniform transm. lines. Norgren, M., T-MTT May 05 15611568 hyperb. transm.-line periodic grids, neg. refr. and focusing. Eleftheriades, G.V., + , T-MTT Jan 05 396-403 left-handed transm.-line media, nonlin. wave propag. phenom. Kozyrev, A.B., + , T-MTT Jan 05 238-245 modeling microstrip structs., nonuniform grids and perfectly matched layer, compact 2D FDFD method. Jiunn-Nan Hwang, T-MTT Feb 05 653-659 uniplanar left-handed metamaterials, efficient modeling. Yunchuan Guo, + , T-MTT Apr 05 1462-1468 unstructured tetrahedral meshes, transm.-line modeling (TLM). Sewell, P., + , T-MTT Jun 05 1919-1928 Transmission line theory; cf. Transmission line matrix methods Transmitters 0.25-ȝm CMOS OPLL transmitter IC for GSM and DCS appls. Peng-Un Su, T-MTT Feb 05 462-471 baseband-modeled CALLUM archits., spectrum emission considerations. Strandberg, R., + , T-MTT Feb 05 660-669 dual-band transmitters, digitally predistorted freq. multipliers for reconfigurable radios. Youngcheol Park, + , T-MTT Jan 05 115-122 std. CMOS technol., monolithic integr. mm-wave IMPATT transmitter. Al-Attar, T., + , T-MTT Nov 05 3557-3561 wireless transmitters, digital subband filtering predistorter archit. Hammi, O., + , T-MTT May 05 1643-1652 Transmitters; cf. Radio transmitters; Repeaters; Transceivers; Transponders Transponders RF sect. of UHF and microwave pass. RFID transponders, design criteria. De Vita, G., + , T-MTT Sep 05 2978-2990 Traveling wave amplifiers surface-passivated high-resist. Si, true microwave substr. Spirito, M., + , T-MTT Jul 05 2340-2347 Traveling wave tubes Ka-band resonant ring for testing components for a high-gradient linear accelerator. Bogdashov, A., + , T-MTT Oct 05 3152-3155 small-signal model for submicrometer nn-i-nn traveling-wave photodetectors. Torrese, G., + , T-MTT Oct 05 3238-3243 TR devices measuring BER of UWB devices, prod. test tech. Bhattacharya, S., + , TMTT Nov 05 3474-3481 Triboelectricity; cf. Surface charging Tuning dielec. resonator, discrete electromechanical freq. tuning. Panaitov, G.I., + , T-MTT Nov 05 3371-3377 fully integrated transmitter front-end with high power-added efficiency. Kim, H., + , T-MTT Oct 05 3206-3214 InP HBT noise params., noise-fig. meas. syst., direct extr. Jianjun Gao, + , T-MTT Jan 05 330-335 reactance of hollow, solid, and hemispherical-cap cylindrical posts in rectangular waveguide. Roelvink, J., + , T-MTT Oct 05 3156-3160 wide tuning-range planar filters, lumped-distrib. coupled resonators. Carey-Smith, B.E., + , T-MTT Feb 05 777-785 Tuning; cf. Circuit tuning Tunnel diodes large-sig. diode modeling, alternative param.-extr. tech. Yew Hui Liew, + , T-MTT Aug 05 2633-2638 Two-port circuits init. design of LTCC filters, all-capacitive coupling, simplified anal. tech. Rambabu, K., + , T-MTT May 05 1787-1791 meas. of scatt. params. of N-ports, multiport method. Rolfes, I., + , T-MTT Jun 05 1990-1996

IEEE T-MTT 2005 INDEX — 62 method, improve VNA calib., planar dispers. media, adding, asymmetrical reciprocal device. Scott, J.B., T-MTT Sep 05 3007-3013 nonlin. params. of dispers. APD, pulsed RF meas. and quasiDC opt. excit., extr. Ghose, A., + , T-MTT Jun 05 2082-2087 on-wafer meas. of differential ccts., mm-wave freqs., pure-mode NWA concept. Zwick, T., + , T-MTT Mar 05 934-937 PCB discontinuities, wavelet domain, 2-port equiv. Araneo, R., + , T-MTT Mar 05 907-918 periodically nonuniform coupled microstrip-line filters, harmonic suppression, transm. zero reallocation. Sheng Sun, + , T-MTT May 05 1817-1822 Two-port networks high-performance circular TE01-mode converter. Yu, C.-F., + , T-MTT Dec 05 3794-3798

U UHF amplifiers Maximum Output control method for UMTS downlink transmitters, adaptive feedforward amp. Legarda, J., + , T-MTT Aug 05 2481-2486 packaged inductively degenerated common-source low-noise amps., ESD protection, anal. and optim. Sivonen, P., + , T-MTT Apr 05 1304-1313 UHF amplifiers; cf. UHF power amplifiers UHF circuits; cf. UHF integrated circuits UHF devices; cf. UHF amplifiers; UHF filters; UHF mixers; UHF oscillators UHF FETs compact dual-fed distrib. power amp. Eccleston, K.W., T-MTT Mar 05 825-831 UHF filters electronically tunable microstrip bandpass filter, thin-film BariumStrontium-Titanate (BST) varactors. Nath, J., + , T-MTT Sep 05 27072712 miniaturized dual-mode bandpass filter struct., shunt-capacitance perturb. Ming-Fong Lei, + , T-MTT Mar 05 861-867 miniaturized dual-mode ring bandpass filter, perturb. Boon Tiong Tan, + , T-MTT Jan 05 343-348 miniaturized parallel coupled-line bandpass filter, spurious-response suppression. Pedro Cheong, + , T-MTT May 05 1810-1816 RF inductors and filters, LCP substrs. for Wi-Fi appls., layout-level synthesis. Mukherjee, S., + , T-MTT Jun 05 2196-2210 supercond. spiral filters, quasiellipt. charact. for radio astron. Guoyong Zhang, + , T-MTT Mar 05 947-951 tunable combline filter, continuous control of center freq. and bandwidth. Sanchez-Renedo, M., + , T-MTT Jan 05 191-199 UHF frequency conversion InP DHBT technol., DC-100-GHz freq. doublers. Puyal, V., + , T-MTT Apr 05 1338-1344 UHF integrated circuits 2.4-GHz-band GFSK appls., low-power highly digitized receiver. Bergveld, H.J., + , T-MTT Feb 05 453-461 CMOS technol., complementary Colpitts oscillator. Choong-Yul Cha, + , T-MTT Mar 05 881-887 deembedding static nonlinearities and accurately identifying and modeling memory effects, wide-band RF transmitters. Taijun Liu, + , T-MTT Nov 05 3578-3587 InP DHBT technol., DC-100-GHz freq. doublers. Puyal, V., + , T-MTT Apr 05 1338-1344 inverse class-F archit., l-band LDMOS power amps. Lepine, F., + , TMTT Jun 05 2007-2012 packaged inductively degenerated common-source low-noise amps., ESD protection, anal. and optim. Sivonen, P., + , T-MTT Apr 05 1304-1313 RF sect. of UHF and microwave pass. RFID transponders, design criteria. De Vita, G., + , T-MTT Sep 05 2978-2990 UHF measurements EM field-mapping syst., reson.-suppressed mag. field probe. Jung-Min Kim, + , T-MTT Sep 05 2693-2699 liq. crysts., temp.-controlled coaxial transm. line, broad-band microwave charactn. Mueller, S., + , T-MTT Jun 05 1937-1945 low-noise multiresolution high-dyn. ultra-broad-band time-domain EMI meas. syst. Braun, S., + , T-MTT Nov 05 3354-3363 wide-band on-wafer noise-param. meas., improved Y-factor method. Tiemeijer, L.F., + , T-MTT Sep 05 2917-2925

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UHF mixers 0.6-V 1.6-mW transformer-based 2.5-GHz downconversion mixer, +5.4dB gain and -2.8-dBm IIP3, 0.13-ȝm CMOS. Hermann, C., + , T-MTT Feb 05 488-495 UHF oscillators CMOS technol., complementary Colpitts oscillator. Choong-Yul Cha, + , T-MTT Mar 05 881-887 UHF power amplifiers 2.14-GHz Chireix outphasing transmitter. Hakala, I., + , T-MTT Jun 05 2129-2138 2-stage ultrawide-band 5-W power amp., SiC MESFET. Sayed, A., + , TMTT Jul 05 2441-2449 class-F and inverse class-F HBT amps., IMD anal. Ohta, A., + , T-MTT Jun 05 2121-2128 compact dual-fed distrib. power amp. Eccleston, K.W., T-MTT Mar 05 825-831 high-effic. multistage Doherty power amp. for wireless communs., anal. and design. Srirattana, N., + , T-MTT Mar 05 852-860 inverse class-F archit., l-band LDMOS power amps. Lepine, F., + , TMTT Jun 05 2007-2012 Ultrafast optics substr. removed LV high-speed GaAs/AlGaAs electrooptic modulators, Trail electrodes. JaeHyuk Shin, + , T-MTT Feb 05 636-643 Ultra wideband communication frequency planning and synthesizer architectures for multiband OFDM UWB radios. Mishra, C., + , T-MTT Dec 05 3744-3756 Ultra wideband radar ultra-wideband low-cost phased-array radars. Rodenbeck, C.T., + , T-MTT Dec 05 3697-3703 Ultra wideband technology microstrip three-port and four-channel multiplexer for WLAN and UWB coexistence. Lai, M.-I., + , T-MTT Oct 05 3244-3250 V Varactors capacitive nonlinearity in thin-film BST varactors. Chase, D.R., + , TMTT Oct 05 3215-3220 dual-band transmitters, digitally predistorted freq. multipliers for reconfigurable radios. Youngcheol Park, + , T-MTT Jan 05 115-122 electronically tunable microstrip bandpass filter, thin-film BariumStrontium-Titanate (BST) varactors. Nath, J., + , T-MTT Sep 05 27072712 high-power MEMS varactors and impedance tuners for mm-wave appls. Lu, Y., + , T-MTT Nov 05 3672-3678 hybrid narrow-band tunable bandpass filter, varactor loaded EM-bandgap CPW. Pistono, E., + , T-MTT Aug 05 2506-2514 intelligently controlled RF power amp., reconfigurable MEMS-varactor tuner. Dongjiang Qiao, + , T-MTT Mar 05 1089-1095 metamaterial-based electronically controlled transm.-line struct., leakywave antenna, tunable radiation angle and beamwidth. Sungjoon Lim, + , T-MTT Jan 05 161-173 miniaturized dual-mode bandpass filter struct., shunt-capacitance perturb. Ming-Fong Lei, + , T-MTT Mar 05 861-867 RF/microwave multifunctional systs., reconfigurable bandpass filter. Fathelbab, W.M., + , T-MTT Mar 05 1111-1116 tunable combline filter, continuous control of center freq. and bandwidth. Sanchez-Renedo, M., + , T-MTT Jan 05 191-199 ultrawide-band tunable true-time delay lines, ferroelec. varactors. Kuylenstierna, D., + , T-MTT Jun 05 2164-2170 Vectors broad-band poly-harmonic distortion (PHD) behavioral models from fast automated simul. and large-sig. vectorial net. meas. Root, D.E., + , TMTT Nov 05 3656-3664 Very-large-scale integration 90-nm VLSI SOI CMOS technol., high linearity for WLAN, 26.5-30-GHz resistive mixer. Ellinger, F., T-MTT Aug 05 2559-2565 VHF amplifiers 2-stage ultrawide-band 5-W power amp., SiC MESFET. Sayed, A., + , TMTT Jul 05 2441-2449 VHF circuits InP DHBT technol., DC-100-GHz freq. doublers. Puyal, V., + , T-MTT Apr 05 1338-1344 VHF devices; cf. VHF amplifiers; VHF circuits

IEEE T-MTT 2005 INDEX — 63 Voltage control distortion analysis technique based on simulated large-signal voltage and current spectra. Aikio, J.P., + , T-MTT Oct 05 3057-3066 Voltage controlled oscillators 24-GHz SiGe phased-array receiver-LO phase-shifting approach. Hashemi, H., + , T-MTT Feb 05 614-626 bifurcation control, practical cct. design. Collado, A., + , T-MTT Sep 05 2777-2788 CMOS technol., complementary Colpitts oscillator. Choong-Yul Cha, + , T-MTT Mar 05 881-887 compact intell. RF front-end, reconfigurable RFICs, Si-based technols. Mukhopadhyay, R., + , T-MTT Jan 05 81-93 DVB-S appls., Si bipolar technol., monolithic 12-GHz heterodyne receiver. Girlando, G., + , T-MTT Mar 05 952-959 GaInP-GaAs HBT for accurate predict. of phase noise, oscillators, advanced LF noise model. Nallatamby, J.-C., + , T-MTT May 05 16011612 k- and Q-bands CMOS freq. sources, X-band quadrature VCO. Sangsoo Ko, + , T-MTT Sep 05 2789-2800 low-power ka-band Voltage-controlled oscillator implemented, 200-GHz SiGe HBT technol. Yi-jan Emery Chen, + , T-MTT May 05 1672-1681 mm-wave (Bi)CMOS IC, 30-100-GHz inductors and transformers. Dickson, T.O., + , T-MTT Jan 05 123-133 mm-wave CMOS cct. design. Shigematsu, H., + , T-MTT Feb 05 472-477 multistandard adaptive voltage-controlled oscillators, design. Tasic, A., + , T-MTT Feb 05 556-563 offset-PLL output spur spectrum, systematic anal. Ching-Feng Lee, + , TMTT Sep 05 3024-3034 sig. gener., control, freq. conversion AlGaN/GaN HEMT MMICs. Kaper, V.S., + , T-MTT Jan 05 55-65 stabil. ccts. for phase-noise reduction, microwave oscillators. Suarez, A., + , T-MTT Sep 05 2743-2751 Voltage multipliers RF sect. of UHF and microwave pass. RFID transponders, design criteria. De Vita, G., + , T-MTT Sep 05 2978-2990 Volterra equations distortion analysis technique based on simulated large-signal voltage and current spectra. Aikio, J.P., + , T-MTT Oct 05 3057-3066 Volterra series high av.-effic. SiGe HBT power amp. for WCDMA handset appls. Junxiong Deng, + , T-MTT Feb 05 529-537 linearizing FET low-noise amps., modified derivative superposition method. Aparin, V., + , T-MTT Feb 05 571-581 multichannel commun. systs., modeling distortion. Gharaibeh, K.M., + , T-MTT May 05 1682-1692

W Wave equations; cf. Helmholtz equations Waveform analysis; cf. Spectral analysis Wave functions metamaterial constructed by conductive SRRs and wires, MGS-based algm., propag. property anal. Hai-Ying Yao, + , T-MTT Apr 05 14691476 uniplanar left-handed metamaterials, efficient modeling. Yunchuan Guo, + , T-MTT Apr 05 1462-1468 Waveguide antennas; cf. Horn antennas; Microstrip antennas; Slot antennas Waveguide components multiport MEMS-based waveguide and coaxial switches. Daneshmand, M., + , T-MTT Nov 05 3531-3537 on-wafer meas. of differential ccts., mm-wave freqs., pure-mode NWA concept. Zwick, T., + , T-MTT Mar 05 934-937 TE0n-TE0(n+1) ripple-wall mode converters, circ. waveguide, bandwidth studies. Lawson, W., + , T-MTT Jan 05 372-379 Waveguide components; cf. Circulators; Directional couplers; Power combiners; Power dividers; Waveguide couplers; Waveguide filters Waveguide couplers microstrip-line struct. employing, periodically perforated ground metal and appl., highly miniaturized and low-impedance pass. components fabricated. Young Yun, T-MTT Jun 05 1951-1959 narrow-band multimode coupled resonator filters, shorted waveguide-stub coupling mechanism. Arndt, F., T-MTT Jan 05 414-415

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narrow-band multimode coupled resonator filters ), shorted waveguidestub coupling mechanism. Meyer, P., + , T-MTT Jan 05 415 substr. integr. waveguide technol., 6-port jn. Xinyu Xu, + , T-MTT Jul 05 2267-2273 Waveguide couplers; cf. Directional couplers Waveguide discontinuities complex pass. devices composed of arbitrarily shaped waveguides, Nystrom and BI-RME methods, CAD. Taroncher, M., + , T-MTT Jun 05 2153-2163 port discontinuities, full-wave CAD models of multiport ccts., deembedding. Farina, M., T-MTT May 05 1829 port discontinuities, full-wave CAD models of multiport ccts. ), deembedding. Okhmatovski, V.I., + , T-MTT May 05 1829 rect. waveguides, radially symmetric metal insets, fast S-domain modeling. Mira, F., + , T-MTT Apr 05 1294-1303 Waveguide filters 12-18-GHz 3-pole RF MEMS tunable filter. Entesari, K., + , T-MTT Aug 05 2566-2571 aggressive space mapping, segm. strategy and hybrid optim. algm., fast automated design. Ros, J.V.M., + , T-MTT Apr 05 1130-1142 anal. of dielec. loaded waveguide filters of arbitrary shape, hybrid surface integral-eqn./mode-matching method. Catina, V., + , T-MTT Nov 05 3562-3567 anal. of NRD-guide/H-guide mm-wave ccts., order-reduced vol.-integral eqn. approach. Duochuan Li, + , T-MTT Mar 05 799-812 compact super-wide bandpass substr. integr. waveguide (SIW) filters. Zhang-Cheng Hao, + , T-MTT Sep 05 2968-2977 E-plane filters and diplexers, ellipt. response for mm-wave appls. Ofli, E., + , T-MTT Mar 05 843-851 in-line N-order filters, N real transm. zeros by of extracted poles implemented, low-cost rect. H-plane waveguide, synthesis and design. Montejo-Garai, J.R., + , T-MTT May 05 1636-1642 LTCC, metallic resonators, canonical ridge waveguide filters. Ruiz-Cruz, J.A., + , T-MTT Jan 05 174-182 multiple atten. poles, dual-behavior reson. of freq.-selective surfaces. Ohira, M., + , T-MTT Nov 05 3320-3326 narrow-band multimode coupled resonator filters, shorted waveguide-stub coupling mechanism. Arndt, F., T-MTT Jan 05 414-415 narrow-band multimode coupled resonator filters ), shorted waveguidestub coupling mechanism. Meyer, P., + , T-MTT Jan 05 415 symmetric composite right/left-handed CPW, appls., compact bandpass filters, modeling. Shau-Gang Mao, + , T-MTT Nov 05 3460-3466 TLM-based modeling and design exploiting space mapping. Bandler, J.W., + , T-MTT Sep 05 2801-2811 Waveguide filters; cf. Strip line filters Waveguides compact dual-polarized multibeam phased-array archit. for mm-wave radar. Schulwitz, L., + , T-MTT Nov 05 3588-3594 double-neg. metamaterials, pos. future. Nader Engheta, + , T-MTT Apr 05 1535-1556 Waveguides; cf. Circular waveguides; Coaxial waveguides; Coplanar waveguides; Dielectric waveguides; Ferrite waveguides; Helical waveguides; Planar waveguides; Rectangular waveguides; Strip lines; Waveguide discontinuities Waveguide theory accurate waveguide port boundary condition for time-domain FEM. Zheng Lou, + , T-MTT Sep 05 3014-3023 arbitrary waveguides, p-refinement, gen. quadrilaterals, efficient largedomain 2D FEM soln. Ilic, M.M., + , T-MTT Apr 05 1377-1383 distrib. left-handed microstrip lines, effective EM params. Shau-Gang Mao, + , T-MTT Apr 05 1515-1521 EM distribs. demonstrating asymmetry, spectral-domain dyadic Green's fn. for ferrite microstrip guided-wave structs. Krowne, C.M., T-MTT Apr 05 1345-1361 hyperb. transm.-line periodic grids, neg. refr. and focusing. Eleftheriades, G.V., + , T-MTT Jan 05 396-403 propag. characts. of cylindrical CPW, finite thickness of conductor, fullwave anal. Yamamoto, H., + , T-MTT Jun 05 2187-2195 rect. waveguides, radially symmetric metal insets, fast S-domain modeling. Mira, F., + , T-MTT Apr 05 1294-1303 Wavelength division multiplexing photonic microwave filters, random errors, stat. anal. Vidal, B., + , T-MTT Aug 05 2600-2603

IEEE T-MTT 2005 INDEX — 64 Wavelet transforms efficient nonlin. cct. simul. tech. Dautbegovic, E., + , T-MTT Feb 05 548555 PCB discontinuities, wavelet domain, 2-port equiv. Araneo, R., + , T-MTT Mar 05 907-918 Wave mechanics; cf. Wave functions Waves; cf. Magnetostatic waves Welding; cf. Lead bonding Whispering gallery-modes model for the mode-splitting effect in whispering-gallery-mode resonators. Bourgeois, P.-Y., + , T-MTT Oct 05 3185-3190 Wireless LAN 90-nm VLSI SOI CMOS technol., high linearity for WLAN, 26.5-30-GHz resistive mixer. Ellinger, F., T-MTT Aug 05 2559-2565 compact intell. RF front-end, reconfigurable RFICs, Si-based technols. Mukhopadhyay, R., + , T-MTT Jan 05 81-93 effic. of OFDM transmitters, RF/DSP design. Helaoui, M., + , T-MTT Jul 05 2355-2361 image-rejection CMOS LNA design optim. techs. Trung-Kien Nguyen, + , T-MTT Feb 05 538-547 InGaP/GaAs HBT RF power amps., ESD protection design considerations. Ma, Y., + , T-MTT Jan 05 221-228 microstrip three-port and four-channel multiplexer for WLAN and UWB coexistence. Lai, M.-I., + , T-MTT Oct 05 3244-3250 multistandard adaptive voltage-controlled oscillators, design. Tasic, A., + , T-MTT Feb 05 556-563 multistandard mobile terminals, fully integr. receivers requirements and archits. Brandolini, M., + , T-MTT Mar 05 1026-1038

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multistandard WLAN appls., dual-band RF transceiver. Chang, S.-F.R., + , T-MTT Mar 05 1048-1055 RF inductors and filters, LCP substrs. for Wi-Fi appls., layout-level synthesis. Mukherjee, S., + , T-MTT Jun 05 2196-2210 serial config., 2 finite transm. zeros, LTCC technol., bandpass filter. Chun-Fu Chang, + , T-MTT Jul 05 2383-2388 wide-bandwidth envelope-tracking power amps. for OFDM appls., design. Feipeng Wang, + , T-MTT Apr 05 1244-1255 Wireless sensor networks adaptive power controllable retrodirective array system for wireless sensor server applications. Lim, S., + , T-MTT Dec 05 3735-3743 Y Yield estimation stabil. of microwave amps., variable termination impedances, necessary and sufficient conds. Olivieri, M., + , T-MTT Aug 05 2580-2586 Yttrium compounds mm-wave appls., extremely high-Q factor dielec. resonators. Krupka, J., + , T-MTT Feb 05 702-712

Z

Zinc materials/devices freq. response of SAW filters, FDTD method. King-Yuen Wong, + , TMTT Nov 05 3364-3370

EDITORIAL BOARD Editor: M. STEER Associate Editors:A. CANGELLARIS, A. CIDRONALI, M. DO, K. ITOH, S. MARSH, A. MORTAZAWI, Y. NIKAWA, J. PEDRO, Z. POPOVIC, S. RAMAN, V. RIZZOLI, D. WILLIAMS, R. WU, A. YAKOVLEV REVIEWERS M. Abdul-Gaffoor M. Abe R. Abou-Jaoude M. Abouzahra A. Abramowicz L. Accatino R. Achar D. Adam E. Adler M. Adlerstein K. Agarwal D. Ahn H.-R Ahn M. Aikawa C. Aitchison M. Akaike C. Akyel A. Akyurtlu B. Albinsson F. Alessandri A. Alexanian C. Algani W. Ali-Ahmad F. Alimenti B. Allen D. Allsopp D. Allstot R. Alm B. Alpert A. Alphones A. Altintas A. Alvarez-Melcom M. Alzona S. Amari L. Andersen B. Anderson Y. Ando O. Anegawa K.-S. Ang I. Angelov R. Anholt Y. Antar G. Antonini D. Antsos K. Anwar I. Aoki R. Aparicio K. Araki J. Archer P. Arcioni F. Arndt R. Arora U. Arz M. Asai P. Asbeck K. Ashby H. Ashok J. Atherton A. Atia I. Awai K. Aygun S. Ayuz Y. Baeyens T. Bagwell Z. Baharav I. Bahl D. Baillargeat S. Bajpai J. Baker-Jarvis E. Balboni S. Banba J. Bandler I. Bandurkin R. Bansal D. Barataud I. Barba F. Bardati I. Bardi S. Barker D. Barlage J. Barr D. Batchelor B. Bates H. Baudrand S. Beaussart R. Beck D. Becker K. Beilenhoff B. Beker V. Belitsky D. Belot H. Bell T. Benson M. Berroth G. Bertin S. Best W. Beyenne A. Beyer S. Bharj K. Bhasin P. Bhattacharya Q. Bi M. Bialkowski E. Biebl P. Bienstman R. Bierig R. Biernacki S. Bila L. Billonnet T. Bird B. Bishop G. Bit-Babik D. Blackham B. Blalock M. Blank P. Blondy P. Blount D. Boccoli B. Boeck F. Bögelsack L. Boglione R. Boix J. Booske N. Borges de Carvalho V. Boria V. Borich O. Boric-Lubecke E. Borie J. Bornemann R. Bosisio H. Boss S. Bousnina P. Bouysse M. Bozzi E. Bracken P. Bradley R. Bradley T. Brazil G. Brehm K. Breuer B. Bridges L. Briones T. Brookes S. Broschat E. Brown G. Brown R. Brown S. Brozovich S. Bruce

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A. Deutsch Y. Deval T. Dhaene A. Diaz-Morcillo G. D’Inzeo C. Diskus B. Dixon T. Djordjevic M. A. Do J. Doane J. Dobrowolski W. Domino S. Dow C. Dozier P. Draxler R. Drayton A. Dreher F. Drewniak S. Dudorov S. Duffy L. Dunleavy V. Dunn J. Dunsmore A. Dutta D. Duvanaud A. Duzdar S. Dvorak L. Dworsky M. Dydyk L. Eastman J. Ebel R. Egri R. Ehlers T. Eibert H. Eisele B. Eisenstadt G. Eisenstein G. Eleftheriades I. Elfadel S. El-Ghazaly F. Ellinger T. Ellis B. Elsharawy R. Emrick N. Engheta B. Engst Y. Eo H. Eom N. Erickson J. Eriksson C. Ernst M. Eron L. Escotte M. Essaaidi J. Everard G. Ewell A. Ezzeddine M. Faber C. Fager D.-G. Fang N. Farhat M. Farina W. Fathelbab A. Fathy A. Fazal E. Fear R. Feinaugle M. Feldman P. Feldman A. Ferendeci C. Fernandes A. Fernandez A. Ferrero I. Fianovsky J. Fiedziuszko I. Filanovsky P. Filicori D. Filipovic A. Fliflet P. Focardi B. Fornberg K. Foster P. Foster G. Franceschetti A. Franchois M. Freire R. Freund A. Freundorfer F. Frezza R. Fujimoto V. Fusco G. Gabriel T. Gaier Z. Galani I. Galin D. Gamble B.-Q. Gao M. Garcia K. Gard R. Garver G. Gauthier B. Geller V. Gelnovatch P. Genderen G. Gentili N. Georgieva W. Geppert J. Gerber F. Gerecht F. German S. Gevorgian R. Geyer O. Ghandi F. Ghannouchi K. Gharaibeh G. Ghione D. Ghodgaonkar F. Giannini A. Gibson S. Gierkink J. Gilb B. Gilbert B.Gimeno E.Glass A. Glisson M. Goano E. Godshalk J. Goel M. Goldfarb C. Goldsmith P. Goldsmith M. Golio R. Gómez R. Gonzalo S. Goodnick S. Gopalsami A. Gopinath R. Gordon P. Gould K. Goverdhanam J. Graffeuil L. Gragnani B. Grant G. Grau A. Grebennikov B. Green T. Gregorzyk I. Gresham E. Griffin

J. Griffith A. Griol G. Groskopf C. Grossman T. Grzegorczyk M. Guglielmi P. Guillon K.-H. Gundlach A. Gupta K. Gupta R. Gupta F. Gustrau R. Gutmann W. Gwarek R. Haas J. Hacker G. Haddad S. Hadjiloucas C. Hafner M. Hagmann S. Hagness H.-K. Hahn A. Hajimiri D. Halchin A. Hallac B. Hallford K. Halonen R. Ham K. Hamaguchi M. Hamid J.-H. Han A. Hanke V. Hanna V. Hansen G. Hanson Y. Hao L. Harle M. Harris L. Hartin H. Hartnagel J. Harvey H. Hasegawa K.-Y. Hashimoto K. Hashimoto J. Haslett G. Hau S. Hay H. Hayashi J. Hayashi L. Hayden B. Haydl S. He T. Heath J. Heaton I. Hecht G. Hegazi P. Heide E. Heilweil W. Heinrich G. Heiter M. Helier R. Henderson R. Henning D. Heo J. Herren K. Herrick N. Herscovici J. Hesler J. Heston M. Heutmaker C. Hicks R. Hicks A. Higgins M. Hikita D. Hill G. Hiller W. Hioe J. Hirokawa T. Hirvonen V. Ho W. Hoefer R. Hoffmann M. Hoft J. Hong S. Hong W. Hong K. Honjo G. Hopkins Y. Horii D. Hornbuckle J. Horng J. Horton K. Hosoya R. Howald H. Howe J.-P. Hsu Q. Hu C.-C. Huang C. Huang F. Huang H.-C. Huang J. Huang P. Huang T.-W. Huang A. Huber D. Huebner H.-T. Hui A. Hung C. Hung H. Hung I. Hunter J. Hurrell M. Hussein B. Huyart I. Huynen H.-Y. Hwang J. Hwang K.-P. Hwang J. Hwu C. Icheln T. Idehara S. Iezekiel P. Ikonen K. Ikossi K. Inagaki A. Ishimaru T. Ishizaki Y. Ismail K. Itoh T. Itoh F. Ivanek A. Ivanov T. Ivanov C. Iversen D. Iverson D. Jablonski D. Jachowski C. Jackson D. Jackson R. Jackson A. Jacob M. Jacob H. Jacobsson D. Jaeger N. Jaeger N. Jain R. Jakoby G. James R. Janaswamy

Digital Object Identifier 10.1109/TMTT.2005.862200

V. Jandhyala W. Jang R. Jansen J. Jargon B. Jarry P. Jarry A. Jelenski W. Jemison S.-K. Jeng M. Jensen E. Jerby G. Jerinic T. Jerse P. Jia D. Jiao J.-M. Jin J. Johansson R. Johnk W. Joines K. Jokela S. Jones U. Jordan L. Josefsson K. Joshin J. Joubert R. Kagiwada T. Kaho M. Kahrs D. Kajfez S. Kalenitchenko B. Kalinikos H. Kamitsuna R. Kamuoa M. Kanda S.-H. Kang P. Kangaslahtii B. Kapilevich K. Karkkainen M. Kärkkäinen A. Karpov R. Karumudi A. Kashif T. Kashiwa L. Katehi A. Katz R. Kaul S. Kawakami S. Kawasaki M. Kazimierczuk R. Keam S. Kee S. Kenney A. Kerr O. Kesler L. Kettunen M.-A. Khan J. Kiang O. Kilic H. Kim I. Kim J.-P. Kim W. Kim C. King R. King A. Kirilenko V. Kisel A. Kishk T. Kitamura T. Kitazawa M.-J. Kitlinski K. Kiziloglu R. Knerr R. Knöchel L. Knockaert K. Kobayashi Y. Kobayashi G. Kobidze P. Koert T. Kolding N. Kolias B. Kolner B. Kolundzija J. Komiak A. Komiyama G. Kompa B. Kopp B. Kormanyos K. Kornegay M. Koshiba T. Kosmanis J. Kot A. Kraszewski T. Krems J. Kretzschmar K. Krishnamurthy C. Krowne V. Krozer J. Krupka W. Kruppa H. Kubo C. Kudsia S. Kudszus E. Kuester Y. Kuga W. Kuhn T. Kuki M. Kumar J. Kuno J.-T. Kuo P.-W. Kuo H. Kurebayashi T. Kuri F. KurokI L. Kushner N. Kuster M. Kuzuhara Y.-W. Kwon I. Lager R. Lai J. Lamb P. Lampariello M. Lanagan M. Lancaster U. Langmann G. Lapin T. Larsen J. Larson L. Larson J. Laskar M. Laso A. Lauer J.-J. Laurin G. Lazzi F. Le Pennec J.-F. Lee J.-J. Lee J.-S. Lee K. Lee S.-G. Lee T. Lee K. Leong T.-E. Leong Y.-C. Leong R. Leoni M. Lerouge K.-W. Leung Y. Leviatan R. Levy L.-W. Li

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D. Walker V. Walker P. Wallace J. Walsh C. Wan A. Wang B.-Z. Wang C. Wang E. Wang H. Wang J. Wang K.-C. Wang L. Wang T.-H. Wang W. Wang Y. Wang Z. Wang K. Warnick K. Washio T. Watanabe R. Waterhouse R. Waugh D. Webb J. Webb K. Webb R. Webster S. Wedge C.-J. Wei R. Weigel T. Weiland A. Weily S. Weinreb J. Weiss S. Weiss A. Weisshaar C. Weitzel K. Weller T. Weller C.-P. Wen W. Weng M. Wengler S. Wentworth C. Westgate C. Whelan J. Whelehan L. Whicker J. Whitaker P. White S. Whiteley K. Whites W. Wiesbeck G. Wilkins A. Wilkinson D. Williams B. Wilson J. Wiltse P. Winson K. Wong K.-L. Wong T. Wong J. Woo J. Wood G. Woods G. Wrixon B.-L. Wu H. Wu K.-L. Wu R.-B. Wu T. Wu Y.-S. Wu R. Wylde G. Xiao H. Xin H.-Z. Xu S.-J. Xu Y. Xu Q. Xue A. Yakovlev S. Yamamoto C.-H. Yang F. Yang H.-Y. Yang Y. Yang H. Yano H. Yao K. Yashiro S. Ye J. Yeo K. Yeo S.-P. Yeo S.-J. Yi W.-Y. Yin H. Ymeri S. Yngvesson T. Yoneyama C.-K. Yong H.-J. Yoo J.-G. Yook R. York N. Yoshida S. Yoshikado A. Young L. Young G. Yu M. Yu A. Zaghoul K. Zaki J. Zamanillo P. Zampardi J. Zapata J. Zehentner Q.-J. Zhang R. Zhang A. Zhao L. Zhao L. Zhu N.-H. Zhu Y.-S. Zhu Z. Zhu R. Zhukavin R. Ziolkowski H. Zirath A. Zolfaghari T. Zwick