Antennas For Small Mobile Terminals 1630810959, 9781630810955

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Antennas for Small Mobile Terminals

For a complete listing of titles in the Artech House Antennas and Electromagnetics Analysis Library, turn to the back of this book.

Antennas for Small Mobile Terminals Kyohei Fujimoto Koichi Ito Editors

Library of Congress Cataloging-in-Publication Data A catalog record for this book is available from the U.S. Library of Congress. British Library Cataloguing in Publication Data A catalogue record for this book is available from the British Library. Cover design by John Gomes

ISBN 13: 978-1-63081-095-5

© 2018 ARTECH HOUSE 685 Canton Street Norwood, MA 02062

All rights reserved. Printed and bound in the United States of America. No part of this book may be reproduced or utilized in any form or by any means, electronic or mechanical, including photocopying, recording, or by any information storage and retrieval system, without permission in writing from the publisher. All terms mentioned in this book that are known to be trademarks or service marks have been appropriately capitalized. Artech House cannot attest to the accuracy of this information. Use of a term in this book should not be regarded as affecting the validity of any trademark or service mark.

10 9 8 7 6 5 4 3 2 1

Contents 1

Small Antennas for Small Mobile Terminals

1

1.1

Introduction

1

2

Small Antenna Techniques

3

2.1

Introduction

3

2.2

The Definition of Electrically Small

4

2.3

Fundamental Antenna Performance Properties

6

2.4

The Chu Limit

10

2.5

Properties of the Electrically Small Dipole and Loop Antennas

13

Techniques to Design Small Antennas

17

References

18

3

Types of Small Antennas and Small Mobile Terminals

21

3.1

Types of Small Antennas

21

3.2

Types of Small Mobile Terminals

30

2.6

v

Antennas for Small Mobile Terminals

vi

4

Antennas for Internet of Things Applications

33

4.1 4.1.1 4.1.2 4.1.3 4.1.4

Introduction Overview of the IoT Connectivity Technologies and Challenges for the IoT Overview of the CMA Chapter Contribution and Structure

33 33 35 36 38

4.2 4.2.1 4.2.2

Designs of Antennas for the IoT Applications over TVWS with CMA Narrowband Antennas UWB Antennas

38 38 42

4.3

RFID-Based Humidity Sensor

49

4.4

Conclusions

53

References

53

5

Wireless Power Transfer Systems

57

5.1

Introduction

57

5.2

Overview of WPT Via Radio Wave

58

5.3

Beam Efficiency of WPT Via Radio Waves

61

5.4

Requirements of Antenna for WPT Via Radio Waves

63

5.5

Characteristics of Antenna Array at the Receiver for WPT Via Radio Waves

64

5.6

Various Applications of WPT

65

5.7

Conclusions

67

References

70

6

Antennas for Mobile Phones, Including Smartphones

73

6.1

Introduction

73

6.2 6.2.1 6.2.2 6.2.3

Baseline Characteristics Antenna Size and Performance Effect from Chassis Body Effect

74 74 74 74

Contents

vii

6.3 6.3.1 6.3.2

Antenna for Smartphone Antenna Type for Smartphone Tunable Antenna

76 77 82

7

Antennas for Wearable Systems, Including Body-Centric Communication Systems

89

7.1 7.1.1 7.1.2 7.1.3 7.1.4

89 89 90 90

7.1.5

Wearable Antenna and Related Issues Antenna Design for Wearable Devices Multiple Antennas in a Wearable Device Metal Exterior Antennas in a Wearable Device Transparent Conducting Patch Antenna for a Wearable Device Liquid Crystal Antenna for Wearable Antenna

7.2

SAR of a Wearable Device

95

7.3 7.3.1 7.3.2

Wearable Devices Through the Body Area Network Body Area Network Via a 2.4-GHz System Body Area Network Via a Megahertz Frequency Range

98 98 100

7.4

Summary

104

References

105

Antennas for Laptop Computers, Including Information Tags

109

8.1

Antennas for Laptop Computers

109

8.2

Tunable Multiband Antenna for Tablet Computers

115

8.3

Low SAR Antenna for Tablet Computers

119

8.4

Frequency Reconfigurable Antenna for Mobile Terminals Design Examples

120 121

8

8.4.1 8.5 8.5.1 8.5.2

Design of the UHF-RFID Tag UHF-RFID Tag and Its Read Range Example of RFID Tag Antenna Design for Optical Discs References

91 94

125 125 128 129

Antennas for Small Mobile Terminals

viii

9 9.1

9.2 9.2.1 9.2.2 9.2.3 9.3 9.3.1 9.3.2 9.3.3 9.3.4 9.3.5 9.3.6 9.4

Antennas for 5G Millimeter-Wave System Including Some Practical Issues for Mobile Terminals

131

Introductions of the Millimeter-Wave Broadband System and Antennas in the Fifth Generation of the Communication System

131

Millimeter-Wave Antenna Array and Beamforming Technologies Millimeter-Wave Antennas without Beam Steering Millimeter-Wave Antennas with Beam Steering Millimeter-Wave Antenna Integration and Package

136 136 140 142

Antenna Array Spatial Coverage and Body-Shadowing Effect in a 5G Millimeter-Wave Mobile System Coverage of Millimeter-Wave Cellular Systems Antenna Arrays in 5G UE and the Assessment Methodologies User Body Effect on 5G UE Antennas Shadowing Loss in the Outdoor Environment Modeling of Human Body Blockage Array System and Network Planning

143 143 144 147 149 150 152

RF EMF Exposure Standards/Guidelines for 5G Millimeter-Wave User Equipment Power Density Characteristic for Array Antennas Power Constraint Due to Power Density Limits Conservative Power Density Assessment Power Density Assessment Based on Near-Field Reconstruction Algorithms References Selected Bibliography

158 160 168

10

Unmanned Aerial Vehicles

169

10.1

Communications

170

10.2

SAR

171

10.3

Antenna for Collision Avoidance and Direction Finding 173

10.4

Conformal Antenna and Others

173

References

179

9.4.1 9.4.2 9.4.3 9.4.4

152 156 156 157

Contents

ix

11

Antennas for Wireless Medical Devices

183

11.1

Introduction

183

11.2 11.2.1 11.2.2

Antennas for In-Body Medical Devices Stationary Antennas in the Body Nonstationary Antennas in the Body

185 188 191

11.3 11.3.1 11.3.2 11.3.3

Antennas for On-Body or Wearable Devices Narrowband Antennas with Full Ground Plane UWB Antennas with Full Ground Plane Embroidered Antennas for Easy Integration in Clothing References

191 191 195 195 199

12

Electromagnetic Simulation

203

12.1 12.1.1 12.1.2 12.1.3 12.1.4 12.1.5

Electromagnetic Field Simulation Methods Importance of Electromagnetic Simulation Classification of Electromagnetic Simulation Methods Major Electromagnetic Field Simulation Methods The Matrix Computation Method Examples of Electromagnetic Field Simulation

203 203 204 206 219 220

12.2 12.2.1 12.2.2 12.2.3 12.2.4

Design Optimization Definition of Design Optimization Classification of Structural Optimization Major Optimization Methods Topology Optimization Method as a New Optimization Approach Example of Antenna Optimization References

227 227 228 228

13

Evaluation of Small Antenna Performance

237

13.1

Fundamentals of Evaluation

237

13.2 13.2.1 13.2.2 13.2.3

Performance Evaluation Input impedance and Bandwidth Radiation Patterns and Gain Efficiency

239 239 240 241

13.3 13.3.1 13.3.2

Measurement Method of Measurement by Using a Coaxial Cable Method of Measurement by Using Small Oscillator

243 243 245

12.2.5

232 233 235

Antennas for Small Mobile Terminals

x

13.3.3

Method of Measurement by Using Fiber-Optics References

247 252

14

Evaluation of Small Mobile Terminal Antennas

255

14.1

14.1.2

Measurement Value, Used Conditions, and Phantoms for Small Mobile Terminal Antennas 255 Measurement Value: Radiation Efficiency in Free Space, Radiation Efficiency with the Human Body, and Mean Effective Gain 255 Use Conditions 257

14.2 14.2.1 14.2.2

Measurement Environment of Antenna Performance Impedance measurements Radiation Performance Measurements

259 259 260

14.3

Cable Connection Techniques for Evaluating a Small Antenna Embedded in the Product Mobile Terminal

264

Evaluations of 4 × 4 MIMO Antennas in a Cellular Phone Terminal

264

14.5

Evaluations of GPS Antennas

267

14.6

Evaluations of Bluetooth and Wi-Fi Antennas

270

14.7

Evaluations of Digital TV Antennas

273

References

273

About the Editors

275

List of Contributors

278

Index

279

14.1.1

14.4

1 Small Antennas for Small Mobile Terminals Kyohei Fujimoto

1.1

Introduction

Recent wireless systems have become increasingly smaller in dimensions, and hence antennas installed in such systems must follow the miniaturization trend and inevitably be small. As the general requirements of wireless systems have increased, the antennas applied to those systems have also made progress, evolving from simple structures to sophisticated ones. Meanwhile, wireless systems arise not only in communication systems but also in a wide variety of other systems such as control, identification, sensor, data transmission, body-centric communications, and wireless power transmission. Such wireless systems have a large variety of styles to be appropriate for their practical use. Many of those systems are used in mobile status as well as a fixed status. Typical mobile terminals should be small in dimensions and compact in form. They generally are built with a boxlike structure, some of them are handheld and easy to carry and manipulate, and typically they have simple operating functions. It should be noted that progress in the systems has created variation in their styles, resulting in the evolution of significant convenience for practical operations. Almost all of the antennas used for those small mobile terminals are some sort of built-in type. The performance of antennas installed in such mobile terminals should be analyzed as a unified system composed of the antenna 1

2

Antennas for Small Mobile Terminals

and its mobile terminal, because an antenna installed within a terminal does not perform independently and often is highly dependent upon its environmental conditions, including the terminal structure and other nearby hardware made with conducting or dielectric materials. This book is intended to provide readers with the latest information that should be useful for engineers and researchers who are dealing with antennas used in modern mobile terminals for various systems related to communications, sensors, data transmission, wireless power transmission, medical devices, control applications, and so forth. Practicality is intended; readers are provided with less heavily theoretical treatments, with the main emphasis centering on practical applications and on providing useful information for the design and development of small antennas used in mobile terminals with built-in small antennas.

2 Small Antenna Techniques Steven R. Best

2.1

Introduction

This chapter considers the methods and approaches to optimizing the performance properties of the general, electrically small antenna, as much as possible within theoretical limits. For the general antenna, these performance properties include the impedance match to the transmitter and/or receiver, the antenna’s radiation efficiency, operating bandwidth, directivity pattern, and finally, the antenna’s polarization properties. These properties establish the antenna’s gain and effective receiving area. In many practical applications, the electrically small antenna’s directivity pattern and polarization properties are secondary considerations. Oftentimes, optimization of the performance of the electrically small antenna primarily focuses on the impedance match over the required operating band(s) and the antenna’s radiation efficiency. When designing and optimizing the performance of electrically small antennas, it is necessary to understand how the electrical size of the antenna establishes its electrical performance. In this chapter, we begin with defining the electrical size of an antenna that qualifies it to be considered electrically small and why this definition is important. We present a review of basic antenna properties such as impedance, radiation efficiency, quality factor, and bandwidth, with a focus on why these properties are important in understanding the electrically small antenna. We then discuss the electrically small dipole/monopole and the electrically small loop antennas. These fundamental, electrically small antenna types are important to understand since they are the foundation for virtually all 3

4

Antennas for Small Mobile Terminals

electrically small antenna designs. Finally, the chapter continues with detailed discussions on techniques that can be used to optimize the performance properties of the general, electrically small antenna. Techniques are presented for tuning the antenna (achieving self-resonance), implementing an impedance match, establishing high radiation efficiency, and maximizing the antenna’s bandwidth within theoretical limits.

2.2

The Definition of Electrically Small

The most notable early works on the theory and understanding of electrically small antennas are those of Wheeler [1] and Chu [2]. Other relevant and related works of Wheeler and Chu include [3–6]. In [1], Wheeler described the fundamental limitations of small antennas in terms of the radiation power factor, P, rather than quality factor, or Q. Wheeler stated that the small antenna “is one whose maximum dimension is less than the radianlength,” which is λ/2π. Furthermore, he noted that: “An antenna within this limit of size can be made to behave essentially as lumped capacitance or inductance,” which will be evident when discussing the properties of the electrically small dipole and loop antennas. Wheeler also noted the relationship between maximum bandwidth and antenna size: “While the radiation pattern and hence the directive gain of a small antenna remain the same for a smaller size, the radiation resistance decreases relative to the other resistance in the coupling circuit [from the transmitter to antenna]. The resulting reduction in coupling efficiency is one of the principal limitations of the smaller antenna. Another aspect of the same limitation relates to the frequency bandwidth of operation with fixed values of the circuit elements. A smaller antenna with the same reactance and radiation resistance must be tuned more sharply tuned to deliver its available power. Therefore, the reduction of size imposes a fundamental limitation on the bandwidth.” In [2], Chu addressed the relationship between size and bandwidth directly in defining the fundamental limit or lower bound on Q (the Chu limit) for the small antenna in terms of ka, where k is the free space wavenumber, 2π/λ, and a is the radius of a sphere circumscribing the maximum dimension of the antenna. It is well known that, for a fundamental-mode radiator or electrically small antenna transverse-magnetic (TM) mode electric dipole and transverse-electric (TE) mode magnetic dipole, Q and bandwidth are inversely related [7]. In [6], Adler, Chu, and Fano described the maximum dimension of the small dipole radiator as one whose maximum dimension is λ/2π, consistent with Wheeler’s definition. While some may define the small antenna directly in terms of size relative to wavelength (e.g., λ/8 or λ/10), the definitions of Wheeler and Chu are more valid in terms of the antenna’s electrical properties and the lower bounds on

Small Antenna Techniques

5

performance, particularly Q and bandwidth. The λ/2π maximum dimension translates into a ka value of 0.5. The value of ka = 0.5 for the small antenna limit is significant in that it was shown in [8] that at values below ka = 0.5, the radiation resistance and bandwidth of arbitrary dipole and monopole radiators converge to the radiation resistance and bandwidth values of the straight-wire dipole and monopole of the same height and conductor diameter, regardless of conductor shape or total conductor length. From a practical perspective, it is important to distinguish between electrically small and physically small. Electrically small antennas are more commonly used at lower frequencies, where the operating wavelengths are physically large. At higher frequencies, where the wavelengths tend to be physically smaller, there is less of a need for the use of electrically small antennas. Furthermore, at sufficiently high frequencies, antenna directivity and gain requirements are often much higher in order to successfully establish a communication session or link. Electrically small antennas are generally not capable of providing the required high directivity and gain at higher frequencies. While there is no well-defined frequency limit where electrically small antennas may or may not be used, their use is quite common from the very low frequency (VLF) band through the very high frequency (VHF) band and the lower ultrahigh frequency (UHF) band. Because of the small wavelengths, the use of electrically small antennas is not as common above 3 GHz. Other factors to consider when defining the electrical size of an antenna (ka) are the presence of surrounding dielectric (or other) material and any ground plane structure. When the electrically small antenna is covered by, or surrounded by a dielectric (or other) material, the dimensions or extent of this material must be included in the determination of a. When the antenna is mounted on and works in conjunction with a ground plane, the antenna image in the ground plane must be included in the definition of a. For example, in the case of a straight-wire monopole mounted perpendicular to, and at the center of a large ground plane, the circumscribing sphere defined by radius a, encompasses the portion of the monopole above the ground plane as well as the image of the monopole in the ground plane, as shown in Figure 2.1. The significance of defining the electrical size of the antenna is that it gives the engineer a measure of whether or not the antenna is electrically small and it allows the engineer to compare the antenna bandwidth against the Chu limit. In doing so, the engineer has a direct measure of how well the antenna design is optimized. In practice, many small antennas operate within a device (e.g., phone, tablet, or laptop) using the device printed circuit board (PCB) as the radio frequency (RF) ground plane. In these instances, the current distribution on both the antenna and ground plane can be significant in establishing the antenna’s frequency-dependent electrical properties. In this case, the current distribution

6

Antennas for Small Mobile Terminals

Figure 2.1 Depiction of a sphere of radius a circumscribing a straight-wire monopole antenna at the center of a large ground plane. The circumscribing sphere encompasses both the monopole and its image in the ground plane.

on the ground plane must be considered in the definition of the circumscribing sphere that defines the value of ka. The simplest option is to include the extents of the antenna and entire ground plane (PCB) in defining the radius of the circumscribing sphere, a. The downside to this approach is that it may overestimate the electrical size of the antenna, thus overestimating the theoretical limit on achievable bandwidth. Ideally, if the antenna, ground plane and device can be simulated in a three-dimensional (3-D) electromagnetic (EM) simulation tool, the current density over the surface of the ground plane can be determined. This allows the engineer to estimate the extent of the dominant current density over the ground plane, helping to better estimate the value of ka. The extent of the dominant current density over the ground plane may be defined by a level relative to the peak value of current density. For example, the engineer may choose a relative level of −20 dB or −30 dB. At this point, it should be apparent that this approach to determining the value of ka is only an engineering estimate. The significance of the antenna’s electrical size (ka) is that fundamental limits on the performance properties of the small antenna have been defined in terms of ka, most notably the Chu limit or lower bound on Q, and ultimately the upper limit on operating bandwidth. It is most important to understand that, as ka approaches 0 below values of ka = 0.5, these performance limits that cannot be exceeded with a linear, passive, electrically small antenna.

2.3

Fundamental Antenna Performance Properties

Electrically small antennas are used in many wireless device applications. Many of these applications, for example, terrestrial cellular (3G, 4G, and 5G), im-

Small Antenna Techniques

7

planted medical devices, and radio-frequency identification (RFID), operate in environments subject to local multipath scattering. In these applications, the antenna’s directivity pattern and polarization are important but may be secondary considerations when designing and optimizing the performance properties of the electrically small antenna. Generally, when optimizing the performance of the electrically small, wireless device antenna, the antenna’s impedance match and radiation efficiency are the primary considerations. In this section, we review the fundamental performance properties of the general antenna, specifically, the electrically small antenna [9–11]. The frequency-dependent feed-point impedance of the general antenna is given or defined by Z A ( ω) = R A ( ω) + jX A ( ω)

(2.1)

where XA(ω) is the antenna’s total reactance and RA(ω) is the antenna’s total resistance, which includes the radiation resistance, Rr(ω), and the loss resistance, Rl(ω). The frequency dependence of the antenna’s performance properties is defined in terms of the radian frequency ω = 2πf, where f is the frequency in hertz. The electrically small antenna’s total feed-point reactance is primarily established by the self-inductance and self-capacitance within the antenna structure. The self-inductance of a wire or stripline conductor antenna is established by several factors, which include the wire length, wire diameter, and wire shape. The self-capacitance of small antenna is primarily established by the capacitance between the antenna and the ground plane (in the case of a monopolelike antenna) or the capacitance between the two arms of the antenna (in the case of a dipole-like antenna). Generally, when designing the electrically small antenna, the first objective is to tune or self-resonate the antenna such that the total feed-point reactance equals 0. The radiation resistance of the electrically small antenna is primarily established by antenna’s overall height or length relative to the operating wavelength. As ka approaches 0, the radiation resistance of the electrically small antenna also approaches 0. The loss resistance of the electrically small antenna is established by both conductor and dielectric loss. In the case of the wire antenna, the conductor loss resistance is primarily established by the conductor diameter and length. It is important to note that while the antenna’s reactance and radiation resistance scale with size and frequency, conductor loss does not. When designing the electrically small antenna, the second objective is to design the antenna such that the radiation resistance equals the system characteristic impedance or the transmitter/receiver impedance. In most wireless communication systems, this impedance is 50Ω. At the same time, the additional design objective is to minimize the total loss resistance. Together, the radiation and loss resistances establish the antenna’s radiation efficiency.

8

Antennas for Small Mobile Terminals

The radiation efficiency of the general antenna, ηr(ω), is determined from the ratio of the antenna’s radiation resistance to its total resistance as follows: ηr ( ω) =

Rr ( ω) Rr ( ω) = R A ( ω) Rr ( ω) + Rl ( ω)

(2.2)

While no antenna is 100% efficient, the electrically small antenna can be designed to exhibit relatively high efficiency at reasonable values of ka. As ka approaches 0, it becomes increasingly difficult to achieve high or even reasonable values of radiation efficiency for many practical reasons. At small values of ka, the radiation resistance more closely approaches 0, and the conductor loss resistance dominates the total antenna feed-point resistance. At this point, the electrically small antenna is inherently inefficient. With the low radiation resistance at small values of ka, an efficient impedance is difficult to achieve because of the ohmic losses within the matching circuit. This further reduces the realizable radiation efficiency. Furthermore, from a practical perspective, as ka approaches 0, it becomes increasingly challenging and difficult to incorporate a sufficient amount of physical conductor within the small antenna volume to tune or resonate it, particularly as the operating frequency increases. Recently, there have been several works on electrically small antennas addressing both the evaluation of radiation efficiency and defining an upper bound on the radiation efficiency [12–17]. Once the antenna’s feed-point impedance is known, the frequency-dependent impedance mismatch between the antenna and the transmitter or receiver can be determined. The level of impedance mismatch is characterized by the antenna’s frequency-dependent voltage standing wave ratio (VSWR, s) and mismatch loss, ML. The frequency-dependent VSWR is given by s ( ω) = VSWR ( ω) =

1 + Γ ( ω) 1 − Γ ( ω)

(2.3)

where Γ(ω) is the antenna’s frequency-dependent reflection coefficient given by Γ ( ω) =

Z A ( ω) − ZCH Z A ( ω) + ZCH

(2.4)

where ZCH is the characteristic impedance of the transmission line connecting the antenna and the transmitter or the transmitter or receiver impedance. In (2.4), ZCH is assumed to be real. The antenna’s frequency-dependent mismatch loss is given by

Small Antenna Techniques

ML ( ω) = 1 − Γ( ω)

9 2

(2.5)

The distinction between the use of VSWR or ML is that VSWR is used to specify the performance requirement for the antenna’s impedance mismatch, whereas ML quantifies the level of impedance mismatch loss. For example, 10log(ML) is the mismatch loss in decibels, which quantifies how much of the transmitter’s available power is lost due to the mismatch between the transmitter and antenna impedances. Once the antenna’s frequency dependent directivity (D), radiation efficiency and mismatch loss are determined, the antenna’s frequency-dependent realized gain, G, can be determined and is given by G ( ω) = ηr ( ω) ML ( ω)D ( ω)

(2.6)

The definition of gain in (2.5) assumes that the antenna is perfectly polarized; otherwise, (2.5) must include a polarization mismatch loss term. One of the important and unique aspects of the electrically small antenna is that its directivity pattern, and therefore directivity, does not change appreciably with frequency. With decreasing values of ka, the directivity of the fundamental-mode, electrically small antenna approaches a constant of 1.5 or approximately 1.8 dB. Therefore, the maximum achievable gain for the fundamental-mode, electrically small antenna is 1.8 dBi. In practice, this gain cannot be because it requires that the electrically small antenna be 100% efficient and perfectly matched to the transmitter or receiver. Finally, in this section, the remaining performance properties of the small antenna that we discuss are its quality factor (Q) and its fractional matched VSWR bandwidth, both of which are defined for the self-resonant or tuned antenna. This antenna is one whose feed-point reactance is naturally self-resonant (XA = 0) at the frequency ω0, or its reactance is tuned to 0 at ω0 using a single, lossless, series tuning element having a reactance, XS(ω0) = -XA(ω0). The tuned antenna’s impedance is defined as Z 0 ( ω) = R 0 ( ω) + jX 0 ( ω) = R A ( ω) + j [ X A ( ω) + X S ( ω)]

(2.7)

The Q of the tuned antenna is defined by the ratio of reactive energy, W, and accepted power, PA, as [2, 7] Q ( ω0 ) =

ω0W ( ω0 ) PA ( ω0 )

(2.8)

10

Antennas for Small Mobile Terminals

The accepted power, PA, includes the power associated with radiation and the power associated with loss within the antenna structure. The exact Q of the antenna can be approximated from the antenna’s impedance properties and is given by [7] Q ( ω0 ) ≈ Q Z ( ω0 ) =

ω0 Z 0′ ( ω0 ) 2R A ( ω0 )

 X (ω )  ω0 = R A ′ ( ω0 )2 +  X A ′ ( ω0 ) + A 0  ω0  2R A ( ω0 ) 

2

(2.9)

The definition of impedance bandwidth that is inversely related to Q over all frequency is the fractional matched VSWR bandwidth, FBWV(ω0), given by FBWV ( ω0 ) =

ω+ − ω − ω0

(2.10)

where ω+ and ω− are the radian frequencies above and below ω0, respectively, where the VSWR is equal to the value, s, that defines the operating bandwidth. If the desired operating bandwidth is defined by a VSWR of 2:1, the value of s is simply equal to 2. It is significant to note that in defining the fractional matched VSWR bandwidth, the VSWR of the tuned antenna is determined using a characteristic impedance, ZCH, equal to the antenna’s feed-point resistance, RA(ω0). Fractional matched VSWR bandwidth and Q are related as follows [7] Q ( ω0 ) ≈

2 β , FBWV ( ω0 )

β=

s −1 ≤1 2 s

(2.11)

We note that the approximations in (2.9) and (2.11) were derived in [7] under the assumptions that the self-resonant or tuned antenna exhibits a single impedance resonance within its defined fractional matched VSWR bandwidth and that this bandwidth is not too large.

2.4

The Chu Limit

From a practical or operational perspective, the antenna engineer is interested in the antenna’s operating bandwidth, and in many cases, several usable operating bands. That said, engineers designing electrically small antennas will often first characterize the quality factor of the antenna rather than its bandwidth. This is done for primarily three reasons. First, the definition of operating band-

Small Antenna Techniques

11

width can be ambiguous, whereas the definition of Q is not. Second, there is a well-defined theoretical limit on the minimum achievable Q as a function of ka (the Chu limit). Third, there is a well-defined approximate relationship between Q and VSWR bandwidth. Wheeler first addressed the fundamental limits on small antenna performance in 1947 [1]. Wheeler characterized the fundamental bandwidth limit for the small antenna in terms of its radiation power factor, which he defined as the inverse of Q. The maximum achievable power factor for the small antenna is defined as (ka)3, which leads to a lower bound on Q of Q lb −Wheeler = ηr

1 (ka )3

(2.12)

Following Wheeler, the work of Chu [2] and McLean [19] led to what is now referred to as the Chu limit or the lower bound on Q being defined as  1 1 +  Q lb −Chu = ηr  3  (ka ) ka 

(2.13)

At small values of ka, the lower bound of (2.13) approaches the lower bound of (2.12). It is important to note that no linear, passive, electrically small antenna will exhibit a Q that is below this limit. In recent years, there has been a large body of work on the evaluation of antenna Q and the determination of the lower bound on Q for practical electrically small antennas of arbitrary shape [20–27]. In [20], Thal derived Chu’s lower bound on Q accounting for the internal energy inside the spherical volume defined by the circumscribing radius a (the ka sphere). In [2], Chu had assumed that the internal energy inside the ka sphere was 0. The significance of the result in [20] is that it led to a practical increase in the lower bound for the fundamental, electrically small TM-mode and TE-mode dipole antennas. As ka approaches 0, the lower bound on Q for the electrically small TM-mode antenna approaches 1.5 times the lower bound of (2.13), whereas the lower bound on Q for the electrically small TE-mode antenna approaches 3 times the lower bound. This is significant in that the maximum achievable bandwidth for a practical small antenna cannot meet the upper bound on bandwidth derived from the inverse of the Chu limit. The Chu limit stated in (2.13) is not the practical lower bound on Q. In [21–27], numerous approaches were undertaken to define the practical lower bound on Q for small antennas of arbitrary shape. The lower bound of (2.13) is stated for a fundamental mode dipole antenna whose geometry is such that it fully occupies the spherical volume defined by the ka sphere. To optimize

12

Antennas for Small Mobile Terminals

the Q of such an antenna, its conductors must lie on the outside surface of this sphere [28–30]. The practical issue is that wireless devices and small antenna applications very seldomly allow for the use of spherical antenna geometries. Practical small antenna designs are typically cylindrical or planar in shape and they do not fully occupy the available volume within the ka sphere. For this reason, their Qs will not approach the Chu limit as closely as spherical-shaped antennas. The practical lower bound on Q for cylindrical and planar-shaped antennas was addressed by Gustafsson et al. in [21], with antenna design examples given in [30]. Gustafsson et al. provided design guidelines for optimizing the Q of cylindrical and planar shaped antennas as a function of their length to diameter ratio. Using [11, 13] and the work of Gustafsson et al., the practical upper bounds on bandwidth for optimized, fundamental TM-mode cylindrical and planar antennas are given by FBWCVub ≈

1 (ka )3 s − 1 2 ηr 1 + (ka )2 s

(2.14)

FBWPVub ≈

1 (ka )3 s − 1 5 ηr 1 + (ka )2 s

(2.15)

and

respectively. We emphasize that (2.14) and (2.15) are approximations and should only be used as guidelines in the performance analysis of electrically small antennas. A plot of these upper bounds for the lossless antenna, having a bandwidth defined by a 2:1 VSWR, is presented in Figure 2.2. At ka = 0.5, the upper bound on fractional matched VSWR bandwidth is approximately 3.5% and 1.4%, for the cylindrical and planar antennas, respectively. At ka = 0.05, these upper bounds decrease to approximately 0.004% and 0.002%, respectively, illustrating the significant bandwidth limitations with decreasing the antenna size. The lossy antenna will exhibit increased bandwidth in direct proportion to the decrease in radiation efficiency relative to the lossless antenna. Aside from the inherent decrease in radiation efficiency associated with decreasing ka, the major practical challenge at small values of ka is the inherently narrow operating bandwidth. From a practical perspective, the high Q and narrow bandwidth make it challenging to design and manufacture many electrically small antennas for a specific frequency. Furthermore, the narrow bandwidth makes the antenna susceptible to detuning due to manufacturing variations, changes in material properties (e.g., dielectric constant), and variations with installations and changes in the local environment.

Small Antenna Techniques

13

Figure 2.2 Upper bound on the fractional bandwidth for a lossless, electrically small, fundamental TM mode antenna occupying a cylindrical or a planar shape.

2.5

Properties of the Electrically Small Dipole and Loop Antennas

As discussed by Wheeler [1], from a circuit perspective, as ka approaches 0, the electrically small antenna essentially behaves as a lossy capacitor or a lossy inductor. The most fundamental of these electrically small antennas is the straightwire dipole, which is a TM-mode radiator, and the circular loop, which, at very small values of ka, is a TE-mode radiator. Understanding the frequencydependent circuit properties of these two fundamental radiators provides significant insight into their performance properties as well as design techniques suitable for their tuning, impedance matching, and optimizing their operating bandwidth within theoretical limits. The straight-wire dipole and circular loop antennas are depicted in Figure 2.3. The straight-wire dipole has an overall length l and a conductor diameter d. The circular loop has a circumference C and a conductor diameter d. The dipole and loop shown in Figure 2.3 have the same value for the radius of the circumscribing sphere, a, and therefore, the same value of ka at any given frequency. However, they have substantially different conductor lengths, which have a significant impact on their relative resistance and resonant frequency. The feed-point resistance and reactance of the typical, lossy straight-wire dipole and circular loop through values of ka up to 0.5 are presented in Figures 2.4 and 2.5, respectively. From Figure 2.5, we note that the dipole exhibits its first natural resonance, a series resonance, well above ka = 0.5. The dipole

14

Antennas for Small Mobile Terminals

Figure 2.3 Depictions of the straight-wire dipole and circular loop antennas. The straightwire dipole has an overall length, l, and a conductor diameter d. The circular loop has a circumference, C, and a conductor diameter d.

Figure 2.4 of ka.

Resistance of the straight-wire dipole and circular loop antennas as a function

exhibits its first natural resonance at ka ≈ 1.57. The circular loop exhibits its first natural resonance, a parallel resonance, at ka ≈ 0.5. These values of ka for the resonant frequencies of the straight-wire dipole and circular loop are approximations, as is the statement that a dipole is naturally self-resonant when its overall length l ≈ λ/2. From Figure 2.4, we note that the radiation resistance of the straight-wire dipole approaches 0 as ka approaches 0 and remains well below its resonant

Small Antenna Techniques

15

Figure 2.5 Reactance of the straight-wire dipole and circular loop antennas as a function of ka.

resistance through ka = 0.5. For values of ka ≤ 0.5, the radiation resistance of the dipole can accurately be approximated by [11] l  Rrd = 20 π2    λ

2

(2.16)

If the skin depth δ in the dipole conductor is somewhat less than d/2, its loss resistance can be approximated as Rld =

lρ l ≈ 3 πd δ 3 πd

kc µ0 ρ 2

(2.17)

where ρ is the resistivity of the conductor and µ0 is the permeability of free space. It is assumed that the current on the dipole has a triangular dependence over the length of the conductor and the current density decays exponentially from its value at the surface of the conductor. For δ somewhat greater than d/2, the loss resistance of the small dipole can be approximated by Rld = 2l ρ / ( πd 2 ). From Figure 2.4, we note that the radiation resistance of the circular loop approaches 0 as ka approaches 0; however, as ka approaches 0.5, the resistance of the loop is increasing significantly because the operating frequency is approaching the loop’s first natural parallel resonance where the resistance can be thousands or tens of thousands of ohms. Near ka = 0.5, the circular loop does

16

Antennas for Small Mobile Terminals

not exhibit a resistance approximated by the theoretical formulas for small loop antennas. Furthermore, at ka near 0.5, the circular loop does not exhibit the radiation pattern of a fundamental TE-mode (magnetic) dipole. The circular loop will not exhibit a fundamental magnetic dipole pattern until ka approaches approximately 0.1. Below the small antenna limit (ka approaching 0.2), the radiation resistance of the circular loop can be approximated by [11] Rrl

C  = 20 π N    λ 2

4

2

(2.18)

where N is the number of turns in the loop. For the arbitrarily shaped loop, the radiation resistance should be expressed as a function of the loop area, A, as follows Rrl = 320 π 4 N 2 (

A 2 ) λ2

(2.19)

Furthermore, if the small multiturn loop is wound on a ferrite core, its radiation resistance can be expressed as Rrl = 320 π 4 µcr 2 N 2 (

A 2 ) λ2

(2.20)

where µcr is the effective relative permeability of the core given by µcr =

µ fr 1 + Dm ( µ fr − 1)

(2.21)

where µfr is the relative permeability of the unbounded ferrite and Dm is the demagnetization factor, which is related to the core geometry. The value of µfr is the relative permeability of the bulk ferrite material provided by the manufacturer. For a cylindrical core with length, 2l, greater than its radius a, Dm can be approximated by the value of Dm of an ellipsoid, given by 2

 a   2l  Dm =    ln( ) − 1 , l >> a l   a 

(2.22)

Small Antenna Techniques

17

If the skin depth δ in the loop conductor is somewhat less than d/2, the copper loss resistance of the circular loop can be approximated as Rll =

NC ρ NC ≈ πd δ πd

kc µ0 ρ 2

(2.23)

For δ somewhat greater than d/2, the loss resistance of the small circular loop can be approximated by Rll = 4NC ρ / ( πd 2 ). It is assumed that the current is constant around the loop and its density decays exponentially from its value at the surface of the conductor. With decreasing values of ka, the electrically small antenna exhibits decreasing radiation resistance, increasing capacitive reactance (dipole), decreasing inductive reactance (loop), decreasing bandwidth, and decreasing radiation efficiency. The high VSWR associated with the decrease in radiation resistance, and high capacitive reactance in the case of the small dipole, make it impractical to deliver power the small antenna using a practical transmitter. The design challenges associated with the electrically small antenna include: (1) reducing the antenna VSWR to an acceptable value (matching the antenna), which requires the reactance to be tuned close to 0 and the total resistance transformed to a value close to the system impedance (typically 50Ω); (2) making the antenna as efficient as possible within theoretical limits, which requires the radiation resistance to be maximized (within matching impedance limits) and/or the loss resistance to be minimized; and (3) maximizing the bandwidth of the antenna within theoretical limits. Techniques to address these issues will be described in the following sections.

2.6

Techniques to Design Small Antennas

The first goal for the antenna engineer designing an electrically small antenna is to impedance-match it to the transmitter and/or receiver. In the case of a transmit antenna, the VSWR must meet the VSWR specifications of the specific transmitter. In high-power applications, a low VSWR is specified to protect the transmitter. In some wireless applications, the VSWR specification may be less about protecting the transmitter and more about ensuring that it is practical to design an antenna capable of operation at a specific frequency or at multifrequency bands. For the receive-only antenna, the VSWR requirements can generally be relaxed, particularly at lower frequencies (high frequency (HF) and below) where the receiving system is generally externally noise limited. The second design goal in the design of the small antenna includes optimizing the radiation efficiency and operating bandwidth and/or operating the antenna at

18

Antennas for Small Mobile Terminals

multibands. For many applications, the small antenna directivity pattern and polarization properties are not primary concerns.

References [1]

Wheeler, H. A., “Fundamental Limitations of Small Antennas,” Proceedings of the IRE, Vol. 35, December 1947, pp. 1479–1484.

[2]

Chu, L. J., “Physical Limitations of Omni-Directional Antennas,” Journal of Applied Physics, Vol. 10, December 1948, pp. 1163–1175.

[3]

Wheeler, H. A., “The Radiansphere Around a Small Antenna,” Proceedings of the IRE, Vol. 47, August 1959, pp. 1325–1331.

[4]

Wheeler, H. A., “Small Antennas,” IEEE Transactions on Antennas and Propagation, Vol. 23, July 1975, pp. 462–469.

[5]

Wheeler, H. A., “Small Antennas,” Ch. 6 in Antenna Engineering Handbook, Third Edition, R. C. Johnson, (ed.), New York: McGraw-Hill, 1993.

[6]

Adler, R. B., L. J. Chu, and R. M. Fano, Electromagnetic Energy Transmission and Radiation, New York: John Wiley & Sons, 1960.

[7]

Yaghjian, A. D., and S. R. Best, “Impedance, Bandwidth and Q of Antennas,” IEEE Transactions on Antennas and Propagation, Vol. 53, No. 4, April 2005, pp. 1298–1324.

[8]

Best, S. R., “On the Performance Properties of the Koch Fractal and Other Bent Wire Monopoles,” IEEE Transactions on Antennas and Propagation, Vol. 51, No. 6, June 2003, pp. 1292–1300.

[9]

Best, S. R., “Small Antennas,” Ch. 6 in Antenna Engineering Handbook, Fourth Edition, J. L. Volakis (ed.), New York: McGraw-Hill, 2007.

[10]

Best, S. R., “Small and Fractal Antennas,” Ch. 10 in Modern Antenna Handbook, C. A. Balanis, (ed.), New York: John Wiley & Sons, 2008.

[11]

Balanis, C. A., Antenna Theory: Analysis and Design, New York: John Wiley & Sons, 2016.

[12]

Capek, M., et al, “Evaluating Radiation Eficiency from Characteristic Currents,” IET Microwaves, Antennas & Propagation, Vol. 9, No. 1, 2015, pp. 10–15.

[13]

Gustafsson, M., “Eficiency and Q for Small Antennas Using Pareto Optimality,” Proceedings of the IEEE/USNC-URSI International Symposium on Antennas and Propagation, July 2013, pp. 2203–2204.

[14]

Shahpari, M., and D. V. Thiel, “Physical Bounds for Antenna Radiation Eficiency,” https://arxiv.org/abs/1609.01761, 2016.

[15]

Fujita, K., and H. Shirai, “Theoretical Limit of the Radiation Eficiency for Electrically Small Self-Resonant Spherical Surface Antennas,” IEICE Transactions on Electronics, Vol. E100-C, No. 1, January 2017, pp. 20–26.

[16]

Pfeiffer, C., “Fundamental Eficiency Limits for Small Metallic Antennas,” IEEE Transactions on Antennas and Propagation, Vol. 65, No. 4, April 2017, pp. 1642–1650.

Small Antenna Techniques

19

[17]

Pfeiffer, C., “Corrections to ‘Fundamental Eficiency Limits for Small Metallic Antennas’,” IEEE Transactions on Antennas and Propagation, Vol. 65, No. 9, September 2017, p. 4958.

[18]

Thal, H., “Radiation Eficiency Limits for Elementary Antenna Shapes,” IEEE Transactions on Antennas and Propagation, Vol.66, No. 5, May 2018.

[19]

McLean, J. S., “A Re-Examination of the Fundamental Limits on the Radiation Q of Electrically Small Antennas,” IEEE Transactions on Antennas and Propagation, Vol. 44, May 1996, pp. 672–676.

[20]

Thal, H. L., “New Radiation Q Limits for Spherical Wire Antennas,” IEEE Transactions on Antennas and Propagation, Vol. 54, No. 10, October 2006, pp. 2757–2763.

[21]

Gustafsson, M., C. Sohl, and G. Kristensson, “Physical Limitations on Antennas of Arbitrary Shape,” Lund University Report: LUTEDX/(TEAT-7153)/1-36/(2007), July 2007.

[22]

Hansen, P. M., and R. Adams, “The Minimum Value for the Quality Factor of an Electrically Small Antenna in Spheroidal Coordinates - TM Case,” Proceedings of the IEEE/ USNC-URSI International Symposium on Antennas and Propagation, July 2010, pp. 1–4.

[23]

Yaghjian, A. D., and H. R. Stuart, “Lower Bounds on the Q of Electrically Small Dipole Antennas,” IEEE Transactions on Antennas and Propagation, Vol. 58, No. 10, October 2010, pp. 3114–3121.

[24]

Vandenbosch, G. A. E., “Simple Procedure to Derive Lower Bounds for Radiation Q of Electrically Small Devices of Arbitrary Topology,” IEEE Transactions on Antennas and Propagation, Vol. 59, No. 6, June 2011, pp. 2217–2225.

[25]

Thal, H. L., “Q Bounds for Arbitrary Small Antennas: A Circuit Approach,” IEEE Transactions on Antennas and Propagation, Vol. 60, No. 7, July 2012, pp. 3120–3128.

[26]

Kim, O. S., “Lower Bounds on Q for Finite Size Antennas of Arbitrary Shape,” IEEE Transactions on Antennas and Propagation, Vol. 64, No. 1, January 2016, pp. 146–154.

[27]

Thal, H. L., “Q Bounds for Planar and Ellipsoidal Antennas,” IEEE Transactions on Antennas and Propagation, Vol. 65, No. 1, January 2017, pp. 353–358.

[28]

Best, S. R., “The Radiation Properties of Electrically Small Folded Spherical Helix Antennas,” IEEE Transactions on Antennas and Propagation, Vol. 52, No. 4, April 2004, pp. 953–960.

[29]

Best, S. R., “Low Q Electrically Small Linear and Elliptical Polarized Spherical Dipole Antennas,” IEEE Transactions on Antennas and Propagation, Vol. 53, No. 3, March 2005, pp. 1047–1053.

[30]

Best, S. R., “Electrically Small Resonant Planar Antennas: Optimizing the Quality Factor and Bandwidth,” IEEE Antennas and Propagation Magazine, Vol. 57, No. 3, June 2015, pp. 38–47.

3 Types of Small Antennas and Small Mobile Terminals Kyohei Fujimoto

3.1

Types of Small Antennas

The first small antenna used in a wireless system was used in 1901 by Marconi for trans-Atlantic communications. The antenna was a fan type with very big dimensions, measuring 48m high and 68m wide. Although the dimensions are very big, the antenna was a small antenna that can be classified as an electrically small antenna, since it is about one-sixth in dimensions as compared with the operating wavelength 338 m (8.87 kHz) in the low-frequency (LF) bands. In these days, frequencies used for wireless communication systems were mostly in the high-frequency (HF) range. Practical wire antennas used were wire types such as monopole (Figure 3.1(a)), fan type (Figure 3.1(b)), multiwire type (Figure 3.1(c)), and cage type (Figure 3.1(d, e)). In the first radio communication experiment done by Hertz in 1887, he used a simple dipole loaded with a square plate at the end of the dipole (Figure 3.2). He also used a single loop antenna (Figure 3.3). Those antennas were the simplest and most basic ones so they have continuously been used for wireless systems for over 130 years up to recent times. Antenna technologies have made progress along with the progress in the wireless systems; in turn, wireless systems have made progress with the progress of antenna technologies and electronic devices. Wire antennas were evolved to the planar structures and further to three-dimensional structures and integrated structures. 21

22

Antennas for Small Mobile Terminals

Figure 3.1 Wire antenna types: (a) Monopole, (b) fan, (c) multiwire, (d) cage type, and (e) second cage type.

Figure 3.2 Antenna used by Hertz for the irst radio communication.

Figure 3.3 A single loop antenna used by Hertz.

In the meantime, wireless systems were mainly long-range communications and antennas applied to those systems were long wire, where Figure 3.4(a) is the receiver side and Figure 3.4(b) is the transmitter side, and umbrella type is shown in Figure 3.5. Communication systems were not only fixed and mobile stations on the land systems, but also marine station on ships sailing on the

Types of Small Antennas and Small Mobile Terminals

23

Figure 3.4 Antennas used in wireless communications system: (a) receiver, and (b) transmitter.

Figure 3.5 Umbrella type antenna.

ocean and home waters. Antennas on ships are expanded using masts (Figure 3.6). Frequencies used are LF, medium-frequency (MF), and HF bands. As wireless systems made progress to operate in very high-frequency (VHF) regions, antenna dimensions have turned out to be small. Where operating frequencies tend to be higher, from HF to VHF, and ultrahigh frequency (UHF), remarkable changes in antenna structures were observed. Simple wire antennas such as dipole and inverted-L were changed to a folded type (Figure 3.7), Lshape (Figure 3.8), inverted-F shape (Figure 3.9), and T-shape (Figure 3.10). Other distinctive antennas are traveling wave antennas such as spiral (Figure

24

Antennas for Small Mobile Terminals

Figure 3.6 Antennas on a ship.

Figure 3.7 Folded-type antenna.

Figure 3.8 Inverted-L antenna.

Figure 3.9

Inverted-F antenna.

Types of Small Antennas and Small Mobile Terminals

25

Figure 3.10 T-shaped antenna.

3.11), helix (Figure 3.12), meander line (Figure 3.13), and zigzag (Figure 3.14). The features of these antennas are that they are composed with periodical structures and waves travel on that antenna structure with the phase velocity Vp, then the traveling time tp on an antenna with the length Lo (extended length of the periodical structure) (Figure 3.15) is tp = Lo/Vp, which is longer than to = Lo/c

Figure 3.11

Spiral antenna.

Figure 3.12 Helix antenna.

26

Antennas for Small Mobile Terminals

Figure 3.13 Meander line antenna.

Figure 3.14 Zigzag antenna.

Figure 3.15 Antenna composed of periodical structure.

(c: wave velocity in free space): travel time of the wave on the antenna structure with the length Lo, implying that the wave on the antenna structure propagates with longer time, in other words, with slower velocity. The wave in that case is slower than the velocity of light c, Vp = Lo/tp < c = Lo/to, so that the wave is called slow wave (SW). The SW can be observed on periodical structures such as meander lines, spirals, and zigzag lines. Periodical structures are usually used in VHF and UHF regions. The feature of applying the SW concept to antenna structures is significant, since with the SW structure, antennas can have resonance with a shorter length than that without the SW structure, meaning that a small antenna can be composed by applying the SW structure, since on the SW structure, the phase velocity is smaller. Consider an antenna with the length Lo, and a SW antenna with the length Ls (Lo > Ls). Here using the phase constant β, Vp = ωs/βs. So the ratio of Vp to c is expressed as Vp/c = (ωs/βs)/(ωo/βo) 0, magnetic energy is stored in the mode, and if χn < 0, electric energy is stored. The mode is in resonance when χn = 0 [14]. In some practical situations, when an object is not large in terms of wavelengths, most of its modes will have eigenvalues far from 0, which leads to the plot of eigenvalue to frequency having a large vertical range and being difficult to be distinguished near χn = 0. Therefore, the modal significance (Sn) and characteristic angle (an) are introduced to describe the eigenvalue, and they are acquired from: Sn =

1 1 + j χn

an = 180o − tan −1( χn )

(4.2)

(4.3)

According to the previous discussion, when the significance of a mode is 1 or its characteristic angle is 180°, the mode is in resonance.

38

Antennas for Small Mobile Terminals

After being excited, the resultant current on a structure can be decomposed into a linear combination of modal currents based on

J=∑

∫∫ J n  Ein dS *

1 + j χn

Jn

(4.4)

in which Ein is the current produced by excitation weight contributed by the nth mode and the whole coefficient of Jn is named as modal weight (αn). Consequently, the input admittance of an antenna Yin can be calculated by the current Iin and the voltage Vin at the feeding port as follows Yin = ∑

( J nP )2 = ∑Yinn 1 + j χn

(4.5)

where J nP is the modal current at the feeding port. In this chapter, antennas for the IoT devices operating over the TVWS are introduced, designed, fabricated, and tested. The CMA is used to guide the design procedure to fully utilize the limited space and exert effective feeding. 4.1.4

Chapter Contribution and Structure

This chapter first gives an overview of the IoT technology. A variety of its connectivity standards are then discussed and compared, and according to which the sub-1-GHz spectrum has been regarded promising to provide wide-range, energy-efficient, and cost-effective services. Challenges lie in designing low-frequency antennas within limited space while keeping a satisfying performance. Next, the CMA is introduced to help to design and understand the antennas in Section 4.2. Furthermore, in Section 4.3, another antenna, based on the RFID technology, is designed to be a humidity sensor to prompt the application of the IoT on the agriculture. Finally, conclusions are drawn in Section 4.4.

4.2 Designs of Antennas for the IoT Applications over TVWS with CMA 4.2.1

Narrowband Antennas

Printed monopole antennas have been widely adopted in wireless communications during the last decade [20–22] owing to their low profile and simplicities of fabrication and integration. In [20], a miniaturized chip antenna was designed to work at 915 MHz with 20-MHz bandwidth. Through meandering the radiating arm and partially short-circuiting, the dimension of the antenna

Antennas for Internet of Things Applications

39

was reduced by roughly 50% from a simple monopole. Both slots and meander line were incorporated and evaluated in [21] to significantly reduce the operating frequency of the final double E-shaped meander line antenna backed by a defected partial ground plane. The antenna was finally working at 878 MHz with 48.83 MHz with a compact size of 0.137λ0 × 0.217λ0. In [22], an arcshaped monopole was printed on a 60 mm × 60mm FR-4 PCB board (εr = 4.5, tan δ = 0.0002). By placing a magneto-dielectric material above the radiator, the antenna was tuned to operating at the twenty-first channel of the TVWS. However, although the superstrate magneto-dielectric materials help to significantly reduce the antenna dimension and broaden the bandwidth, they are generally too expensive for billions of wireless sensors to be connected. Therefore, in [23], a novel antenna operating at the same channel was proposed with the theoretical guidance from the CMA. As illustrated in Figure 4.2, the ground plane in a typical monopole is transformed into a small disk and the radiating body in the same plane of the ground is bent around it to become a circular arm. Modal current distributions of the first two significant modes are given in Figure 4.3. Based on the illustration, having the current decreasing gradually from the short end of the arm to the open end, mode 1 exhibits a similar current distribution of a typical monopole, and hence it is selected for the mode to be excited. After extensive simulations, the resonance of mode 1 is tuned to f0, for which the wavelength is a quarter of the effective length of the radiating arm. Eigenvalues and modal significances of the first three modes of the proposed f structure with respect to the normalized frequency ( f N = ) are plotted in f0 Figure 4.4. It is noticed that at fN = 1, S1 is equal to 1, and S2 and S3 are almost 0, which indicates that mode 1 is the most significant mode at the resonant frequency. In Figure 4.3(a), the modal currents on the disk are flowing mostly along the left edge, and therefore the ground is modified in three ways, rings 1, 2, and

Figure 4.2 The proposed planar circular radiating body with a disk ground plane transformed from a monopole.

40

Antennas for Small Mobile Terminals

Figure 4.3 The modal current distributions on the proposed radiating body of: (a) mode 1 and (b) mode 2.

3, as depicted in Figure 4.5. In Figure 4.5(a), ring 1 is obtained by removing the center area of the ground, and in Figure 4.5(b, c), besides removing the center part, an aperture is cut on rings 2 and 3. It is observed that the current directions on the ground and the radiating arm have opposite rotation directions for rings 1 and 2, while the directions are the same on ring 3. Subsequently, eigenvalues and far-field radiation patterns of the first mode for the three modified structures are compared with those of the original disk ground as shown in Figure 4.6. It can be seen that eigenvalues of the proposed radiating body with different grounds have similar trend and radiation behaviors, and ring 3 is selected as the final ground structure due to its lower resonant frequency. With the physical insights revealed by the CMA, the proposed structure is printed on Rogers 5880 printed circuit board (PCB) with relative dielectric constant of 2.2 for the ease of fabrication as shown in Figure 4.7(a). A metallic loop paralleled with the proposed radiating body is then printed at the other side of the PCB to generate normal direction equivalent magnetic current to excite the radiating body. After further simulation and optimization, the antenna is tuned to operate at the twenty-first channel of the TVWS, and its radius and height are 33 mm (about 5.2% λ0) and 3.1 mm (0.5% λ0), respectively. The voltage standing wave ratio (VSWR) and total efficiency of the compact and low-profile antenna are presented in Figure 4.7(b). It is observed that the antenna resonates at 474 MHz with a VSWR < 2 bandwidth of 2.2 MHz. The simulated total efficiency is 84%, which indicates that most power fed to the antenna can be radiated.

Antennas for Internet of Things Applications

41

Figure 4.4 Eigenvalues and modal signiicances for the irst three modes of the structure shown in Figure 4.2.

Figure 4.5 Structure with variant grounds and natural current lows of mode 1: (a) ring 1, (b) ring 2, and (c) ring 3.

A prototype of the proposed antenna is then integrated into a sensing node to test its system performance in [24]. As depicted in Figure 4.8(a), the measurement was carried out in an anechoic chamber to eliminate interferences from other electromagnetic waves. The transmitting node with the proposed antenna is 4.5m away from a receiver. When nodes are communicating, the spatial radio spectrum is monitored by a spectrum monitoring system provided by CRFS [25] as plotted in Figure 4.8(b). It can be seen that the peak signal strength appears at 474 MHz, and it is about 65 dBm higher than noises at the rest of the spectrum. Analyses and performance evaluations indicate that the

42

Antennas for Small Mobile Terminals

Figure 4.6 Comparison of radiating bodies with different ground planes: (a) eigenvalues and (b) far-ield radiation patterns.

miniaturized low-profile antenna is promising to be used for compact sensors for the IoT applications. 4.2.2

UWB Antennas

Previous works by those concerned with TVWS achieve a narrowband antenna operating at one of the channels in the TVWS. However, to fully utilize the TVWS, a device is expected to operate at any free channel decided by the spec-

Antennas for Internet of Things Applications

43

Figure 4.7 (a) Structure layout of the proposed circular antenna with a magnetic feeding loop, and (b) VSWR and total eficiency of the proposed antenna.

trum sensing and geolocation database [26], and hence their antennas should cover the whole TV spectrum from 470 MHz to 790 MHz. According to the U.S. Federal Communications Commission (FCC), a radio system is ultrawideband (UWB) if its fractional bandwidth (FBW) is greater than 20% or the absolute bandwidth is wider than 500 MHz [27]. Bandwidth realized by an antenna is inversely proportional to its quality factor (Q):

44

Antennas for Small Mobile Terminals

Figure 4.8 (a) Measurement setup of the communication between sensing nodes and (b) detected signal intensity.

FBW =

1 Q

(4.6)

so the Q should be small to broaden the bandwidth. The fundamental limitation for electrically small antennas were studied by Chu [2] for linear polarized antennas as

Antennas for Internet of Things Applications

Q≥

1 1 + 3 (ka ) ka

45

(4.7)

which has been proven to also be applicable for electrically large antennas [29]. In (4.7), k stands for the wavenumber and a is the radius of the minimum sphere that can wrap the antenna. Therefore, it is even challenging to design compact UWB TVWS antennas. In [30], an UWB antenna was designed to operate over 3.1–12 GHz for IoT applications. By optimizing both the rectangle and the ground with a slit and an L-shaped stub, the antenna achieved a compact size of 175 mm2. When the frequency comes down to the ultrahigh frequency (UHF), in [31], a coplanar waveguide fed antenna was printed on a polyester substrate. Using meandered line as radiator, the antenna reached a measured fractional bandwidth of 26% centered at 864 MHz. The UWB printed monopole antenna operating over the entire UHF TV spectrum was further explored in [32, 33], and the CMA was employed to help to reveal the operating mechanism. As illustrated in Figure 4.9(a, b), a U-shaped radiator is printed on the top of an FR4 board, which is backed by ground with a tapered notch. From Figure 4.9(c), the compact antenna with a size of 231 mm × 35 mm × 0.8 mm achieves an ultrawide impedance matching over 474 to 1,212 MHz with VSWR85%. His research group succeeded in a WPT field experiment using 2.388-GHz waves at a distance of 1 mile using a parabolic antenna with a 26-m diameter. For advanced WPT applications as an alternative to wired power systems, beam direction control and beam-forming technology are important. One advantage of WPT via radio waves is that the receiver can be easily changed. It is easy to control the beam direction with phased array technology. The radio wave technology called retrodirective is applied to detect the direction of the receiver. The combination of the phased array and retrodirective target detecting was often used in previous WPT field experiments [5]. Beam-forming technology is effective in increasing beam efficiency and in decreasing any unexpected radiation for the suppression of interference to conventional wireless systems and for human safety. The Cota system, which is a commercial WPT product for wireless chargers, adopts a beam control system together with a phased array

Figure 5.2 Laboratory experiment of WPT via microwave by W. C. Brown in 1975 [4].

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[6, 7]. The Cota system is a WPT system for multiple users with two methods. One is a time-division WPT. The other is a retrodirective target detecting under multipass circumstance like a room. A WPT system can be used instead of a wired power system as a pointto-point system but similar to a wireless communication system; this system can be used to supply wireless power to multiple users. This results in a very convenient system that can be used to receive wireless power every time and everywhere. Such a system is called a ubiquitous power system. A potential WPT application is the Internet of Things (IoT) and sensor networks. If a low-gain transmitting antenna is used for a ubiquitous power system, the beam efficiency from the transmitting antenna to the receiving antenna is very low, such as in a conventional wireless system, because the transmitted power is diffused. When considering human safety, a 10 W/m2 power density is the limit for such a microwave system, as decided by the International Commission on Non-Ionizing Radiation Protection (ICNIRP) guidelines [8]. This power density is very high compared with conventional wireless communication systems, but it is very low in terms of wireless power. The beam-forming technology with a highgain phased array and/or time-division WPT is effective to increase the beam efficiency. If users do not require much power (e.g., below 1 mW), wireless power can be harvested from broadcasting radio waves. This technology is called energy harvesting or energy scavenging from radio waves. In an energy-harvesting system, there is no special transmitter. We only use a rectenna to receive and rectify the broadcasting wave. The power density of broadcasting waves is usually very weak (below 0.1 mW/m2) and is depended on distance from a transmitter. However, if we only require power in the order of 0.1 mW to drive an integrated circuit (IC) or a sensor, energy harvesting can be a potential wireless power technology. In order to increase the received power, a high gain antenna is required. Radio waves are theoretically diffused omnidirectionally by Maxwell’s equations and their power decreases according to the inverse proportion of the square of the distance traveled by the waves. This is a three-dimensional (3-D) phenomenon. In order to reduce the power of the radio waves, two-dimensional (2-D) WPT and wireless communication were proposed in Japan [9]. This is called surface WPT, wherein a sheet-like waveguide is used for wave propagation. The decrease in radio wave energy in 2-D WPT is inversely proportional to only distance (i.e., not to the square of the distance). In addition, unexpected emission is suppressed because the radio wave propagates inside the sheet. A surface WPT system has been standardized by the Association of Radio Industries and Business (ARIB) as ARIB STD-T113 [10] in 2015 in Japan. The frequency is 2.498 GHz ± 1 MHz and its power is below 30W.

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Figure 5.3 indicates the various WPT systems using radio waves as described in this section.

5.3

Beam Efficiency of WPT Via Radio Waves

The beam efficiency between a transmitting antenna and a receiving antenna of WPT via radio waves in the far field is defined by the following equation. η=

Pr λ2GrGt AA = = r t2 2 Pt ( 4 πd ) ( λd )

(5.1)

where Pr, Pt, Gr, Gt, Ar, At, λ, and d are the received power, transmitted power, receiving antenna gain, transmitting antenna gain, aperture area of the receiving antenna, aperture area of the transmitting antenna, wavelength, and the distance between the transmitting and receiving antenna, respectively. This follows from the Friis transmission equation. The Friis transmission equation assumes plane waves under far-field conditions. If the assumption of the plane wave is broken (e.g., under near-field conditions), the beam efficiency may easily exceed 100%, which is not correct. A definition of the far field, wherein a plane wave can be assumed, is described as follows: d>

2D 2 λ

(5.2)

where D is the diameter (maximum dimension) of the transmitting antenna. This equation can be used to calculate the receiving power at the far field for energy harvesting or for WPT using diffused radio waves. Moreover, it indicates that we should use high-gain antennas, short-distance WPT systems, and/or a higher frequency to increase the beam efficiency and received wireless power. In the near field, where we cannot assume a plane wave, the following equation is used to describe the beam efficiency instead of (5.1) [11–13]: η=

2 Pr = 1 − e −t Pt

(5.3)

where τ2 =

At Ar

( λd )2

(5.4)

Figure 5.3 Various WPT via radio waves: (a) beam-type, (b) ubiquitous-type, (c) energy harvesting, and (d) WPT in closed area (surface WPT).

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This equation is the Friis equation without any assumptions of the plane wave. In order to increase the beam efficiency and received wireless power, we conclude that high-gain antennas, short-distance WPT systems, and/or higher frequency must be used. Therefore, high antenna gain is important for a highefficiency WPT system. π In addition, when we shorten the distance, d < , between the transλ mitting and receiving antenna in the reactive near field, the phenomenon of mutual coupling between the transmitting and receiving antenna occurs. As a result, the antenna impedance changes and the efficiency fluctuates at a distance. This result is the same as that for coupling-type WPT. We can design coupling-type WPT with antennas that are used mainly used for the radiation of radio waves [14]. The beam efficiency described by (5.1) and (5.3) do not include the absorption efficiency of the receiving antenna. When an infinite antenna array is used as the receiving antenna, 100% of the radio wave is received [15–17]. This was experimentally confirmed in WPT via microwaves [18]. In a finite receiving array, an absorption efficiency of 95.1% can be theoretically achieved by finite-difference time-domain (FDTD) simulation with impedance matching of complex conjugate of each radiation impedance of the each antenna element in the receiving array [19].

5.4

Requirements of Antenna for WPT Via Radio Waves

As described in the previous section, a high-gain antenna, which is a large (aperture) antenna, is required to increase the received radio wave power. This is particularly important for ubiquitous-type WPT and energy harvesting. However, the users of WPT require a small antenna for mobile applications. Therefore, for practical applications of WPT, an antenna with high gain and small size is required. In general, this is a high-gain antenna. A Yagi-Uda antenna and a microstrip antenna on a thick dielectric base are considered as high-gain antennas. Thick antennas are used to increase the gain. WPT users may not accept the use of thick antennas in the same way as they may resist the use of large antennas. At Kyoto University, we proposed a high-gain antenna after consideration of the mode combination in a spherical wave antenna [20]. The electromagnetic field of the synthesized spherical wave can be described by a combination of all orders of TM and TE modes. Increasing the maximum order of modes, which is added from mode 1 to the maximum number mode, allows the synthesis of spherical wave asymptotes to half-space plane waves in the z-direction. As a result, the directivity of the proposed antenna becomes narrow and its gain increases irrespective of the size of the antenna. Figure 5.4 indicates the calculated result of the power density of spherical waves in the xz plane wherein

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Figure 5.4 Power density of spherical waves in the xz plane: (a) TM-mode spherical wave of mode number l = 10 and (b) synthesized spherical wave of maximum summed mode number lmax = 10, whose frequency is 5.8 GHz [20].

the TM mode spherical wave of mode number l = 10 is shown in Figure 5.4(a) and the synthesized spherical wave of maximum summed mode number lmax = 10, whose frequency is 5.8 GHz, is shown in Figure 5.4(b). When the summed mode number is included, the directivity of the antenna becomes narrow. This means that the antenna is a high-gain antenna. An unsolved problem of the proposed antenna is how to develop a spherical wave antenna with real physical parameters. This is only possible theoretically. In a real physical antenna, we need some volume of the antenna. In recent years, a dielectric resonator antenna, which includes some modes and for which we can assume the same mode combination, has been under development [21]; however, the technology being used in this antenna is not mature enough for any practical applications. If a super-gain antenna can be developed, the antenna aperture can be expanded and a higher radio wave power can be received. However, the directivity of the antenna becomes narrow with increase in gain. It means that a high-gain receiving antenna must detect the direction of the transmitter, which is inconvenient for users.

5.5 Characteristics of Antenna Array at the Receiver for WPT Via Radio Waves Instead of a super-gain antenna, an array antenna is usually used to increase the antenna gain and aperture area. In general, the relationship between antenna gain and the directivity in an antenna element and an antenna array is the same. However, the relationship between gain and directivity is broken only in the rectenna array for WPT.

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A rectenna comprises an antenna and a rectifying circuit having a diode. It can receive microwave power and rectify it to produce DC power. A general block diagram of a rectenna is shown in Figure 5.5. A lowpass filter is installed between the antenna and rectifying circuit to suppress re-radiation of higher harmonics from the diodes. An output filter is used not only to stabilize the DC, but also to increase the RF-DC conversion efficiency with a combination of harmonics, such as a class-F amplifier. A rectenna is used not only for WPT, but also for energy harvesting of broadcasting waves. Various kinds of rectenna have been developed [22, 23]. When we connect a rectenna as an array, its characteristics are different from that of an array antenna because the rectenna array is connected in the DC domain that includes no phases. Figure 5.6 indicates the beam pattern of an antenna element that is assumed to be a cosine pattern for an array with normal antennas and an array with rectennas, respectively. The maximum power is normalized for each array. The beam pattern becomes narrower when the antenna elements are connected in an array. However, the beam of the rectenna array does not become narrower and is the same as that of one antenna element, and an aperture of the rectenna array becomes much larger than that of one antenna element. This is due to the DC connection in the rectenna array, where phase coupling is not an issue [24].

5.6

Various Applications of WPT

As shown in Figure 5.3, there are various applications of WPT via radio waves and energy harvesting from broadcasting waves. Historically, microwaves in the 2.45 and 5.8-GHz bands were used for a beam-type WPT because of beam focusing with a higher frequency and because of the radio wave regulation wherein the 2.45 and 5.8-GHz bands are on the Industrial, Scientific, and Medical (ISM) band. Presently, the UHF band is considered as a ubiquitous-type WPT, similar to RFID [25, 26] and a 100 GHz band rectenna was developed for various WPT applications [27, 28].

Figure 5.5 Block diagram of rectenna.

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Figure 5.6 Cosine beam pattern of antenna element, array with normal antennas, and an array with a rectenna [24].

Kyoto University in Japan developed the first ubiquitous-type wireless charger for a mobile phone around 2000 (Figure 5.7). A mobile phone could be wirelessly charged under a condition of 10 W/m2 at 2.45-GHz microwave power. Furthermore, the mobile phone can still be used in the room because of the difference between the 2.45-GHz microwave power and the 1.9-GHz

Figure 5.7 Concept of proposed ubiquitous-type wireless charger for a mobile phone and the wireless mobile phone charging experiment [29].

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communication system frequencies. An advanced wireless charger for a mobile phone using WPT was recently manufactured by Ossia Corp. [6]. If IoT or sensor devices require significant wireless power, higher frequency, larger antenna, or shorter distance are required to increase the receiving wireless power, as shown in (5.1) and (5.3). For small sensors, it is difficult to use a large antenna for both the receiver and transmitter. A high-frequency value is better for a small antenna, leading to high beam efficiency; however, it is worse for circuit efficiency in a transmitter and in a rectenna. In order to decrease the distance between the transmitter and receiver, a multicopter assisted wireless batteryless sensing system (WBLS) was proposed and employed as a laboratory experiment in an anechoic chamber (Figure 5.8) at Kyoto University in 2015. The multicopter with a microwave transmitter at 5.8 GHz flies toward wireless and batteryless sensors and transmits wireless power. Information is transmitted from the sensors to a base station without any batteries. Potentially, WBLS applications may rescue victims, such as a WPT-powered sensor at a volcano or at an inspection site of infrastructures (bridges, tunnels, and so forth). In conventional wireless communication systems, suppressing unexpected interference with WPT-assisted sensors or IoT systems is of utmost importance. For WPT, no modulated and continuous radio wave is usually used and its spectrum is a very narrow band. However, its power is higher than that of the wireless communication system. There are some ways to suppress the interference from narrowband and high-power radio waves to modulated low-power wireless system (e.g., frequency division duplex, time division duplex, and space division by beam-forming technologies). Kyoto University proposed a time division duplex WPT (TDD-WPT) system at the same frequency band as that of WPT and wireless communication called ZigBee at the 2.45-GHz band [30]. A scheduled TDD-WPT that can automatically transmit wireless power between wireless communication devices was suggested and examined through an experiment. Figure 5.9 indicates a system image of a TDD-WPT-assisted ZigBee sensor. Figure 5.10 indicates a waveform of the power consumption of an end device and an microwave power transfer (MPT) control signal of 2 seconds during the experiment. We can increase the limit power without any interference to 5 pW/cm2 (CW), 1.91 mW/cm2 (pulse, no scheduling), and 2.61 mW/cm2 (pulse, scheduling).

5.7

Conclusions

A WPT system is a wireless system and a small antenna is required for a WPT receiver for small mobile terminals (e.g., RFID, IoT devices, and a wireless charger for mobile phones). However, there are some differences between the antenna technology for wireless communication and that for WPT. Even if a

Figure 5.8 Proposed multicopter assisted WBLS and its ield experiment in an anechoic chamber in 2015 at Kyoto University.

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Figure 5.9 Image of TDD-WPT assisted ZigBee sensor [30].

Figure 5.10 Waveforms of the power consumption of an end device and MPT control signal of 2 seconds during the experiment [30].

small antenna is used for a WPT system, we need a large aperture antenna, such as a super-gain antenna, because the wireless receiver power is limited to the aperture area of the antenna. A receiving array rectenna for WPT has a large aperture and wide directivity. Only the amplitude of the received radio wave is summed without any phases, which is an interesting characteristic of WPT; this is because the receiver of WPT has a rectifier after the antenna. WPT is sometimes used in the near field (e.g., the beam-type WPT whose beam efficiency reaches almost 100%), and we must consider a sphere wave at the receiver of a WPT in a beam-type WPT system.

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Numerous commercial WPT applications are being developed around the world. The discussion of standardization and new radio wave regulations are ongoing in different countries. For future WPT applications, antenna technologies are most important.

References [1]

Hirayama, H., et al., “EMC Characteristics of Open-End and Short-End Helical Antennas for Coupled-Resonant Wireless Power Transfer,” Proc. of EMC Europe, 2014.

[2]

Brown, W. C., “The History of Power Transmission by Radio Waves,” IEEE Transactions on Microwave Theory and Techniques, Vol. 32, No. 9, 1984, pp. 1230–1242.

[3]

Tesla, N., “The Transmission of Electric Energy Without Wires,” 13th Anniversary Number of the Electrical World and Engineer, March 5, 1904.

[4]

Brown, W. C., “Adapting Microwave Techniques to Help Solve Future Energy Problems,” 1973 G- MTT International Microwave Symposium Digest of Technical Papers 73.1, 1973, pp. 189–191.

[5]

Shinohara, N., “Beam Control Technologies with a High-Efficiency Phased Array for Microwave Power Transmission in Japan,” Proceeding of IEEE, Vol. 101, No. 6, 2013, pp. 1448–1463.

[6]

Ossia Inc., http://www.ossia.com/.

[7]

Hatem, Z., “Wireless Power Transmission System,” U.S. Patent No. 8159364, April 17, 2012.

[8]

ICNIRP EMP guideline, 1998, http://www.icnirp.org/cms/upload/publications/ICNIRPemfgdl.pdf.

[9]

Shinoda, H., et al., “Surface Sensor Network Using Inductive Signal Transmission Layer,” Proc. of Int. Conf. on Networked Sensing Systems (INSS), 2007, pp. 201–206.

[10]

ARIB STD-T113, (in Japanese), http://www.arib.or.jp/english/html/overview/doc/1STD-T113v1_0.pdf.

[11]

Brown, W. C., “Adapting Microwave Techniques to Help Solve Future Energy Problems,” IEEE Transactions on Microwave Theory and Techniques, Vol. MTT-21, No. 12, 1973, pp. 753–763.

[12]

Goubau, G., and F. Schwering, “On the Guided Propagation of Electromagnetic Wave Beams,” IRE Trans. Antennas Propagat., Vol. AP-9, May 1961, pp. 248–256.

[13]

“Applications and Characteristics of Wireless Power Transmission,” Document No. 1A/18-E, Task Group ITU-R WP1A, Reference Question 210/1, ITU Radiocommunication Study Group, October 9, 2000.

[14]

Chen, Q., et al., “Antenna Characterization for Wireless Power-Transmission System Using Near-Field Coupling,” IEEE Antennas and Propagation Magazine, Vol. 54, No. 4, August 2012, pp. 108–116.

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[15]

Diamond, B. L., “A Generalized Approach to the Analysis of Ininite Planar Array Antennas,” Proc. of IEEE, Vol. 56, 1968, pp. 1837–1851.

[16]

Stark, L., “Microwave Theory of Phased Array Antenna - A Review,” Proc. of IEEE, Vol. 62, 1974, pp. 1661–1701.

[17]

Itoh, K., T. Ohgane, and Y. Ogawa, “Rectenna Composed of a Circular Microstrip Antenna,” Space Power, Vol. 6, 1986, pp. 193–198.

[18]

Otsuka, M., et al., “Relation Between Spacing and Receiving Eficiency of Finite Rectenna Array,” (in Japanese), IEICE Trans. B-II, Vol. J74-B-II, No. 3, 1990, pp. 133–139.

[19]

Tsukamoto, Y., et al., “Study on a Dipole Array Antenna with Relector for Non-Leak MPT System to Vehicles,” Proc. of 2015 Asian Wireless Power Transfer Workshop, 2015.

[20]

Matsumuro, T., Y. Ishikawa, and N. Shinohara, “Small-Size Large-Aperture Antenna Using Multilayered Spherical Dielectric Resonators,” Proc. of 7th European Conference on Antenna and Propagation (EuCAP2013), 2013, pp. 2960–2964.

[21]

Matsumuro, T., Y. Ishikawa, and N. Shinohara, “Experiment and Analysis of a Resonant Characteristic of Spherical Dielectric Resonator for Small-Size Large-Aperture Antenna II,” (in Japanese), IEICE tech. Report, WPT2013-34, 2014.

[22]

Shinohara, N., “Rectennas for Microwave Power Transmission,” IEICE Electronics Express, Vol. 10, No. 21, 2013, pp. 1–13.

[23]

Hemour, S., and K. Wu, “Radio-Frequency Rectiier for Electromagnetic Energy Harvesting: Development Path and Future Outlook,” Proc. of IEEE, Vol. 102, No. 11, November 2014.

[24]

Shinohara, N., “4.5.5. Rectenna Array,” in Wireless Power Transfer via Radiowaves (Wave Series), N. Shinohara, (ed.), New York: ISTE/John Wiley & Sons, 2014.

[25]

Fantuzzi, M., D. Masotti, and A. Costanzo, “Simultaneous UHF Energy Harvesting and UWB-RFID Communication,” 2015 IEEE MTT-S Intl. Microw. Symposium (IMS), TH2E-5, 2015.

[26]

Bolos, F., D. Belo, A. Georgiadis, “A UHF Rectiier with One Octave Bandwidth Based on a Non-Uniform Transmission Line,” 2016 IEEE MTT-S Intl. Microw. Symposium (IMS), 2016.

[27]

Weissman, N., S. Jameson, and E. Socher, “W-Band CMOS On-Chip Energy Harvester and Rectenna,” 2014 IEEE MTT-S Intl. Microw. Symposium (IMS), TH2C-5, 2014.

[28]

Hemour, S., C. H. Lorenz, and K. Wu, “Small-Footprint Wideband 94GHz Rectiier for Swarm Micro-Robotics,” 2015 IEEE MTT-S Intl. Microw. Symposium (IMS), WE1F-5, 2015.

[29]

Shinohara, N., et al., Experimental Study on Wireless Power Space, (in Japanese), IEICE Tech. Report, SPS2003-18, 2004, pp. 47–53.

[30]

Shinohara, N., “Simultaneous WPT and Wireless Communication with TDD Algorithm at Same Frequency Band,” in Wireless Power Transfer Algorithms and Applications in Ad Hoc Communication Networks, A. Georgiadis (ed.), New York: Springer, 2016.

6 Antennas for Mobile Phones, Including Smartphones Katsunori Ishimiya

6.1

Introduction

Mobile devices such as a mobile phone, smartphone, and tablet all have a wireless communication system. In order to achieve high-speed data communication, we utilize the multiband of frequency and multimode in different communication system. In this chapter, antenna design guidelines, structure, and technology are explained. The required level of antenna for mobile phone and smartphone is very high, such as performance, multifunction with another device (speaker and vibrator), miniaturization, and cost. Antenna performance must be considered with the human body effect, the effect from other devices closed to the antenna (camera, speaker, battery, and so forth), chassis (material and painting), interference from other antennas, and specific absorption rate (SAR). The environments in which mobile devices are used are under a multiple wave, whose direction is not constant. That is why antenna efficiency is used for a performance measure instead of directional gain, which is used for a typical performance measure for an antenna.

73

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6.2

Baseline Characteristics

6.2.1

Antenna Size and Performance

If antenna size is reduced (downsized), bandwidth and performance will be degraded. Size and performance are trade-off relations. The most effective way to become reliable or improve performance is to increase antenna volume, such as length, width, and height. However, antenna volume is limited due to the design requirements of the mobile device product including length, width, and thickness. Therefore, it is necessary to devise antenna structure and positions, such as the distance and position from other parts, such as a speaker, vibrator, and USB/lightning connector. 6.2.2

Effect from Chassis

A chassis works as an antenna because an antenna current is on printed wiring board (PWB) and chassis is on GND due to the antenna being located in an eclectically small chassis as GND. It is difficult to keep certain performance at a low band (below 1,000 MHz) if the chassis length is less than 1/4 lambda. Further, there are many components in the chassis; an antenna design must be considered. Generally, the main antenna for the cellular system is located at the bottom of the phone and consists of a speaker, microphone, and vibrator as acoustic parts, as an antenna and speaker box. Antenna performance can be affected by these parts easily. The workaround is kept with a distance between the antenna and these parts, but it is difficult to get enough distance since limitation space in a chassis as mentioned. Another workaround is to keep an electrical distance to put an inductor or ferrite beads on a signal line closed to parts. Figure 6.1 shows the cellular main antenna in the smartphone. 6.2.3

Body Effect

The human body is a kind of material that can be regarded as the obstacle and absorber of the radio wave, when the mobile device is used during talking with the phone in the hand, browsing, and keeping in a pocket. Figures 6.2 and 6.3 show the antenna efficiency of free space and talk position (real human head and hand). There are two separate ways to hold the terminal. One of them is the hard hold, which is no space between the terminal and the hand, and the other one is the soft hold, which has space between the terminal and the hand. The resonance frequency is shifted to the lower frequency in 800 to 900 MHz compared between free space and the talk position. There are two types of losses by body effect. Mismatch loss is from a frequency shift of the resonance frequency. Abortion loss is from body loss. For

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Figure 6.1 Cellular main antenna in smartphone.

Figure 6.2 Antenna eficiency (700 to 1,000 MHz).

reducing these body effects, the antenna is designed with a closed loop tuner and should be kept at a distance from the hand. The value of body loss is different between antenna design, type, placement, chassis length, and thickness. Figure 6.4 shows body loss versus chassis length and antenna placement. It was found that in the lower band and the talking position with the hand, the planar inverted-F antenna (PIFA) had advantages with 3 to 4 dB less body loss for a large phone type.

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Figure 6.3

6.3

Antenna eficiency (1,600 to 2,300 MHz).

Antenna for Smartphone

Radio communications systems for smartphone are, for example, Global System for Mobile Communications (GSM), wideband code division multiple access (WCDMA), Long-Term Evolution (LTE), Global Navigation Satellite System (GNSS), wireless local area network (LAN), and Bluetooth. Table 6.1 shows an example of smart phone radio system. Figure 6.5 shows the example of smartphone antenna configuration for the European market. A cellular antenna is divided into two by the frequency range for corresponding to the carrier aggregation (CA), which allows mobile network operators to combine a number of separate LTE carriers. This enables them to increase the peak user data rates and capacity of network and to exploit fragmented spectrum allocations. The antenna for low-band covers is approximately 700 to 1,000 MHz, the antenna for mid-band covers is approximately 1,400 to 2,200 MHz, and the high band covers approximately 2,300 MHz to 3,700 MHz (6,000 MHz including the LTE unsilenced band). Cellular low-band, mid-band, and high-band antenna 1 are located at the bottom of the PCB and cover the frequency for GSM, WCDMA, and LTE. Both cellular low-band antenna and high-band antenna 2 are located at the top of the PCB, cover frequency, and work as RX diversity for WCDMA and LTE

Antennas for Mobile Phones, Including Smartphones

Figure 6.4

77

Body loss comparison between PIFA and monopole.

and RX MIMO for LTE. Cellular mid-band antenna and high-band antenna 3 are located at the top of the PCB and are combined with wireless LAN antenna. Cellular mid-band antenna and high-band antenna 4 are located at the center end. Antennas 3 and 4 are used for higher-order Rx diversity and Rx 4 × 4 MIMO. 6.3.1

Antenna Type for Smartphone

The antennas that are mainly adopted in smartphones are the following four types: 1. Planar inverted-F antenna (PIFA);

78

Antennas for Small Mobile Terminals Table 6.1 Example of Smartphone Radio System (European Market) Radio System GSM WCDMA LTE GNSS Wireless LAN Bluetooth

Band, System, Version 850, 900, 1,800, 1,900 Band 1, 2, 4, 5, 6, 8, 19 Band 1, 2, 3, 4, 5, 7, 8, 12, 13, 17, 19, 20, 26, 28, 29, 32, 38, 39, 40, 41, 42 GPS, GLONASS IEEE 802.11a, 11ac, 11b, 11g, 11n 5.0

Figure 6.5 Example of a smartphone antenna arrangement (Europe).

2. Folded monopole antenna and inverted-F antenna (IFA); 3. Folded loop antenna; 4. Metal bezel antenna.

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PIFA

PIFA is placed on the ground plane (PCB) and consists of a radiator, feeding pin, and ground pin. Some resonance frequencies can be controlled by having some slits and adjusting its length. PIFA was used for the cellular band and placed on the top of the chassis until 2008. Until 2008, there was no demand for a big bandwidth for the low band (only 800 or 900 MHz). Reentry demand is from 700 to 900 MHz. The disadvantage of PIFA is a narrow bandwidth, especially lower frequency. The solution of this disadvantage is to increase the antenna height, which is between the antenna radiator and the ground plane. However, the thickness of the phone is limited, so PIFA has not been fitted since the demand of frequency was increased, and the trend of thickness of the phone became very small after 2009. In recent years, PIFA is used for noncellular band, which is Wi-Fi and Bluetooth. These are higher frequency and easy to get bandwidth compared with lower frequency. Figure 6.6 shows PIFA at the top of the PCB, and Figure 6.7 shows the current distribution of PIFA. 6.3.1.2

Folded Monopole Antenna and IFA

Folded monopole antenna placed outside of ground plane and consists of a radiator and feeding pin or a radiator, feeding pin, and grounding pin. There is no ground plane under the radiator. Therefore, the antenna height can be lower. A folded monopole antenna is placed at the bottom of the chassis as a cellular main antenna and also at the top and side of the chassis as a cellular subantenna (second, third, and fourth as higher-order diversity and 4 × 4 MIMO) and a noncellular antenna, which is GPS, Wi-Fi, and Bluetooth. Its bandwidth is wider than PIFA. Additional resonance can exist if the chassis length is larger than 1/4 lambda in the lower frequency. In recent years, this antenna is more

Figure 6.6 PIFA at the top of the PCB.

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Figure 6.7 Current distribution of PIFA.

widely adopted for the current smartphone. Figure 6.8 shows a folded monopole antenna at the bottom of the PCB, and Figure 6.9 shows the current distribution of the folded monopole antenna. 6.3.1.3

Folded Loop Antenna

The folded loop antenna is placed on both the ground plane (PCB) and outside of the ground plane and consists of a radiator, feeding pin, and ground pin. A loop antenna can have a multiresonance by a folding antenna radiator. In case the length of the antenna radiator is lambda, it works as differential mode and current does not flow on the ground much. This means when the phone is grabbed by the hand, absorption loss is low compared to other antennas. In case the length is 1/2 λ, the ground plane also works as an antenna, so the absorption loss is high, the same as the other antenna type. However, there is no open end that is a high impedance for PIFA and a monopole antenna, so it is a place that is easily affected by the hand and the body at the radiator. The body, especially the hand effect, is low; therefore, mismatch loss can be reduced

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Figure 6.8 Folded monopole antenna at the bottom of the PCB.

Figure 6.9 Current of the folded monopole antenna.

compared to other antennas. In case of the other wavelength, when the loop antenna is devised to fold the radiator to get capacitive coupling with several parts of the radiator, various resonance modes are generated. Figure 6.10 shows a folded loop antenna at the bottom of the PCB, and Figure 6.11 shows the current distribution of a folded loop antenna. 6.3.1.4

Metal Bezel Antenna

Recently, a number of metal bezel phones have increased in the market, especially the high-end and middle-range phone category. A metal bezel is used as cosmetic parts and as part of an antenna on a smartphone. Figure 6.12(a, b) shows examples of antenna configurations with a metal chassis. In Figure 6.12(a), a low-band or mid-band antenna is configured with center of metal bezel which connects the feed points and connects a liquid crystal (LC) (inductor and/or capacitor) to GND. However, IFA or a loop antenna is used for the high band. In Figure 6.12(b), a low-band or mid-band antenna is configured with a center bezel, which connects the feed point and GND through SPxT (single pole double/triple/quadruple throw: x = 2 or 3 or 4) switch with a matching network that is capable of switching between 700 to 800 and 1,400 to 2,200 MHz and 800 to 900 and 1,400 to 2,200 MHz. A side metal bezel is used for a high-band antenna.

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Figure 6.10 Folded loop antenna at the bottom of the PCB.

Figure 6.11 Current distribution of a folded loop antenna.

6.3.2

Tunable Antenna

In the previous section, three different types of antennas for mobile phone were introduced. In this section, the technology of tunable antenna consists of these antennas with switch devices and variable capacitance devices that can increase bandwidth and antenna efficiency. To improve antenna performance, supporting multiband, good antenna efficiency and wider bandwidth, we should use the biggest-size antenna. If the antenna size is not big enough, it is difficult to obtain a certain performance, especially a lower frequency (700 to 1,000 MHz). To solve this issue, a tunable antenna solution is ideal. The typical antenna does not have enough bandwidth at the low band to cover 700 MHz; however, the antenna with turning device extends low-band coverage down to 700 MHz. Figure 6.13 shows a VSWR of an antenna with or without a tunable device.

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Figure 6.12 Example of a metal smartphone antenna coniguration: (a) metal bezel antenna with conventional antenna (IFA or loop antenna); and (b) metal bezel antenna with a band switch solution.

Figure 6.13 VSWR of an antenna.

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Devices such as switch devices and variable capacitance devices that are used are made of gallium arsenide (GaAs), silicon on insulator (SOI), and silicon on sapphire (SOS). For switch devices, single pole double throw (SPDT), single pole triple throw (SP3T), and single pole quadruple pole (SP4T) are mainly used. The harmonics problem and poor isolation between each port were concerns before they were solved by the progress of material technology, design technology, and manufacturing technology. In terms of cost, it is becoming cheaper, so in recent years, it has been adopted in many smartphones. There are two kinds of tuning methods, open loop tuning and closed loop tuning. 6.3.2.1

Open Loop Tuning

Open loop tuning is a nonfeedback solution, and a tunable device can be controlled by trigger, which is timed for band switching. Figure 6.14 shows a block diagram of open loop tuning. Digital baseband IC has a look-up table, which is information when the switch and variable capacitance work by the trigger when the frequency band is changed. Figure 6.15 shows examples of antenna model. Figures 6.16 and 6.17 show examples of an antenna model with an open loop impedance and aperture tuning. A method of attaching a tunable device to a matching circuit to change impedance is called an impedance tuner. A tunable device is implemented in series or shunt in matching circuit. This is easy to implement. A method of attaching a tunable device to an antenna structure is called an aperture tuner. A tunable device is implemented in shunt on extra ground pin. This is difficult to implement, but generally the tuning range is wider.

Figure 6.14 Open loop tuning.

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Figure 6.15 Antenna model.

Figure 6.16 Open loop impedance tuning.

Figure 6.17

Open loop aperture tuning.

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Closed Loop Tuning

Figures 6.18 and 6.19 show closed loop tuning block diagrams for impedance and aperture turning. Closed loop is a feedback solution that automatically adjusts the band and use case variation including the detuned frequency by hand and others. It relies on feedback by mismatch sensor or power detector and needs an algorithm. In recent years, a detector is implemented in radio-frequency integration circuit (RFIC) and micro control unit is implemented in digital baseband IC. Antenna resonant frequency is changing due to external loading effects. A detector (e.g., power detector or dual couplers) is used to monitor the change in reflection coefficient. Then, for impedance matching, a tuning device is used to improve the power transfer between the RF front end and the antenna. For aperture matching, the turning device is used to change the antenna resonant frequency making it back to where it should be, for reaching optimum system performance. Figure 6.20 shows two examples of antenna with closed loop tuning.

Figure 6.18 Closed loop impedance tuning (two examples).

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Figure 6.19 Closed loop aperture tuning (two examples).

Figure 6.20 Examples of antenna with closed loop tuning.

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7 Antennas for Wearable Systems, Including Body-Centric Communication Systems Kun Zhao and Zhinong Ying

7.1

Wearable Antenna and Related Issues

7.1.1

Antenna Design for Wearable Devices

Smart wearable devices (e.g., smartwatch and smart wristband) have drawn dramatic attention from the industry and the society recently. They have become personal devices that are part of the mobile world. Generally, smart wearable devices need to collect information from external sources, such as a cellular network or a paired mobile terminal, which requires that these devices have antennas inside (see Figure 7.1). Due to the limited space, most smartwatches or smart wristbands only have Bluetooth, Wi-Fi, GPS antennas, or a higher-frequency cellular antenna (>1 GHz) inside. The IFA antenna is a popular design that has small volume and low profile [1–4]. The performance of such antenna could be found in [4]. The antenna could perform very well in the body-worn cases. The major challenge of the design is the antenna integration with other components and control the near field and body loss which will be discussed later.

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Figure 7.1 The wearable wireless devices become a part of the mobile world.

7.1.2

Multiple Antennas in a Wearable Device

The data throughput can be further increased with the integration of multiple-input multiple-output (MIMO) technology. A cornerstone in developing MIMO technology relies on designing multiple antennas with low mutual coupling. However, the situation would be more complicated for small form factor device applications and high number of antennas being considered. In [5], a watch-sized, 4-port, dual-band MIMO antenna is proposed. Four radiating elements with heights of 5.4 mm are situated on the edges of a square ground plane with side length of 30 mm. The antennas operate at 2.1 GHz and 4.4 GHz and exhibit reasonably low mutual coupling in average (−15 dB). In [6], it was shown that a pair of antennas operating at the same frequencies can be packed very close together. By introducing a T-shaped slit on the ground plane directly under the radiating elements, the mutual coupling can be suppressed effectively. In Figure 7.2, an evolution of a watch-sized, 8-port MIMO antenna demonstrated for the first time. The total volume of the antenna system is 30 × 30 × 5.4 mm3. There are four three-dimensional (3-D) printed spacers on top of a square ground plane, and each spacer is wrapped by two antennas. The relative permittivity of the spacer used is around 3.1 in the frequency range of 2 GHz to 3 GHz. The T-shaped slit introduced on the ground plane is for suppressing mutual coupling between the two closely packed antennas above. Since the four antenna pairs are rotationally symmetric, Figure 7.3 only shows the performance of S-parameters with respect to Port 1 and Port 2. In the figure, it should be noted that these antennas exhibit some common behavior due to the symmetric structure. The antennas resonate at 2.4 GHz with the worst mutual coupling of −11 dB. 7.1.3

Metal Exterior Antennas in a Wearable Device

Another trend of designing the antenna on the metal bezel of a mobile phone has become popular in recent years. A closed metal ring antenna can be robust

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Figure 7.2 Geometry of the 8-port antenna system suitable for smartwatch application.

to human hand effects due to its more confined E-field and larger physical dimension [7]. Antenna designs using the metal frame of a smartwatch have also been used. For instance, in [8], a closed metal ring antenna prototype is implemented for smartwatch applications. In addition, a cavity mode can also be used if it is a full metal watch in [9]; a cavity-backed annular slot antenna is designed for a full metal watch housing. With the increasing demands of the cellular data network, the lower band of the cellular antenna ( Gthrshold. The maximum solid angle here can be chosen as the surrounding sphere 4π, which is simple and usually useful for evaluating a mobile phone whose orientation can be totally random. It can also be defined by the requirement of communication systems if the desired use scenario does not require a whole spherical coverage. The coverage efficiency and total scan pattern offer an intuitive insight of how large of a solid angle that an array system can cover, which is critical for evaluating and optimizing the array topology in UE. Furthermore, it can also be related to some system-level parameters (e.g., outage probability) under certain assumptions. For instance, if the propagation environment is assumed to be a random LOS with only one incoming ray [93], a loss in the coverage

Figure 9.7 The illustration of the total scan pattern and the coverage eficiency. (After: [87].)

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efficiency can be interpreted as an increment in the outage probability of the communication system [94]. Since the coverage of a single planar array is limited, there is a very high possibility that there are more than two subarrays in a UE to achieve an omnidirectional coverage. As illustrated in Figure 9.8(a), with one subarray covering at least a quarter of the whole sphere, four subarrays can steer their beams to all the directions in space. Meanwhile, the array gain can remain in a higher value in all directions with multiple arrays than only using a single one, as shown in the coverage efficiency results in Figure 9.8(b). Utilizing multiple subarrays in a UE is also necessary to fit for different use cases. The user body will impose a strong

Figure 9.8 (a) The illustration of beam switch between four subarrays in a UE. (b) The coverage eficiency with a different number of subarrays in a UE.

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shadowing loss if the array is blocked by it (as will be discussed in detail later), and therefore a switching mechanism between multiple arrays will be necessary. The coverage efficiency is a useful parameter to characterize the spatial coverage of the UE antenna array in terms of array gain. However, the spherical coverage of a UE antenna system in link budget calculations, is determined not only by the array gain, but also the power level, the system transmission losses, the noise level and the interference level. Recently, the cumulative distribution function (CDF) of equivalent isotropically radiated power (EIRP) and effective isotropic sensitivity (EIS) have been agreed in 3GPP standardization discussions to characterize the spherical coverage of millimeter-wave array systems in UE for transmitting and receiving performance [95]. The CDF of two typical array configurations with 20-dBm conductive power is shown in Figure 9.9. A good UE array system should be able to remain a high EIRP value when the CDF is low, to ensure a proper spherical coverage performance. 9.3.3

User Body Effect on 5G UE Antennas

Since a UE or a mobile phone will mainly operate near a user, the user body effect is critical to the performance of UE antenna systems. The user body effect on a single antenna can be categorized into three aspects: the change of impedance matching, the absorption of radiation power, and the change of radiation pattern. The first two effects will be primary when a user is in the vicinity of the antenna near field, which will lead to a loss in antenna total efficiency and is the major concern in sub-6-GHz frequency bands. However, attributed to the smaller wavelength, the distance between the user body and antenna elements is enlarged in terms of wavelength. Therefore, user body effect on the antenna impedance matching and radiated power is expected to be lower. In [96], linear antenna arrays were fabricated and tested in phone mock-ups, which are shown in Figure 9.10(a). The user body effect on antenna arrays in UE is investigated with real users through 3-D antenna far-field measurements at 15 GHz, which are shown in Figure 9.11(b). The radiation patterns of an embedded element are shown in Figure 9.11. The typical body loss is less than 3 dB [96]. Comparing to their counterpart in the sub-6-GHz bands where the typical value is about 6–12 dB, the user body effect on the antenna’s total efficiency is negligible if the user does not block the element directly. However, the user body will cause a dramatic change in the radiation pattern of the embedded element and result in a pronounced shadowing loss at such a high frequency. In free space, the single element shows that a subhemisphere coverage with peak gain is around 5 to 6 dBi. However, a human body-shaped shadowing region can be observed once the user is included in the measurement in the data mode.

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Figure 9.9 CDF of EIRP for two array topologies in UE. Conductive power = 20 dBm. (After: [96].)

Figure 9.10 The measurement setup of antenna array integrated into a phone mock-up in (a) free space and (b) data mode.

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Figure 9.11 The measurement radiation pattern of the notch-embedded element in (a) free space and (b) data mode.

9.3.4

Shadowing Loss in the Outdoor Environment

The shadowing loss from user’s body will also impose an impact on the propagation channel characterize. The received signal strength of a single element in the UE with and without the user against the distance between the BS and UE is shown in Figure 9.12 [90]. The path-loss model changes dramatically with user body blockage. Therefore, the user body effect will play a decisive role in the channel modeling in future 5G communication systems in the millimeterwave spectrum. A further investigation in Figure 9.13 shows that when the user faces a different orientation, the received signal strength can change rapidly. The fluctuation of received signal strength in the UE can be up to 30 dB lower when

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Figure 9.12 The received signal stregnth of the UE against distance.

Figure 9.13 The received signal strength of the UE in a ixed position with rotated orientation.

the user body blocks the dominant propagation component. In addition, the effect of the user body will depend on the angular spread [90]. The blockage effect can be mitigated if the angle of arrive is more uniformly distributed (e.g., in a rich scattering environment). However, since the millimeter-wave channel is likely to be sparse in general, the user body effect may still be critical in most of the scenarios. 9.3.5

Modeling of Human Body Blockage

Several methods have been proposed to model the human body in millimeterwave frequency. It is widely assumed that a human body can be simplified as perfectly electrically conducting (PEC) cylinder or plate in the millimeter-wave frequency range when evaluating the blockage effect. The shadowing loss can be calculated through the geometrical theory of diffraction (GTD)/the uniform geometrical theory of diffraction (UTD) or the knife-edge diffraction (KED) method (Figure 9.14(a, b)). In [97], the excess loss through a PEC cylinder is

Antennas for 5G Millimeter-Wave System Including Some Practical Issues

Figure 9.14 in 3GPP.

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The illustrations of (a) GTD/UTD, (b) KED [87], and (c) the self-blockage model

assessed through the geometrical theory of diffraction (GTD) at 2.45 GHz, 5.7 GHz, and 62 GHz. In [98], the human body was modeled as a PEC plate at 73 GHz, where the shadowing loss is calculated based on a double knife-edge diffraction (DKED) method. In the 3GPP channel model TR 38.900 [99], the KED method is also adapted to model the human body blockage in the channel. In addition, a static model is also introduced, where the self-blockage from

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the user body is modeled as a 30-dB shadowing loss constantly within a fixed solid angle range (Figure 9.14(c)). In addition, simplified body phantom has also been tested. A water-filled anthropomorphic phantom was used in [100], and the shadowing loss through this phantom is measured with a vector network analyzer (VNA). 9.3.6

Array System and Network Planning

There is no doubted that the shadowing loss from the objects in the channel can be mitigated by using an array system in UE. However, its effect should not be overestimated. A comparison between the received signal strength with a single element in the UE and with an 8 × 1 uniform linear array is carried out in [90], where the dominant eigenmode beamforming scheme is adopted [101]. Even with an array system, the signal strength in NLOS is still below an accepted level since the array gain from the UE is not sufficient to overcome the shadowing loss caused by buildings. The loss of the user body blockage can be more effective compromise. However, the shadowing loss from the user body is expected to change rapidly in real life due to the movement of users, which will increase the complexity of the beamforming algorithm in practical use. Since the shadowing loss caused by the objects in the propagation channel cannot be completely solved by increasing the array gain in the system, it is expected to have a much denser deployment of BS for 5G cellular systems operate in the millimeter-wave spectrum. The concept of the small cell [102] will be necessary for ensuring a good coverage in complicated urban environments. In addition, the BS diversity scheme might be important as well, which can allow the user to switch between multiple BSs that are located in different orientations, to ensure a stable received signal strength of UE as shown in Figure 9.15.

9.4 RF EMF Exposure Standards/Guidelines for 5G MillimeterWave User Equipment EMF exposure level limits have a big impact on the wireless network and device antenna design. People are now quite familiar with the SAR limits for the below-6 GHz device; however, in the millimeter-wave range, there are many different challenges for mobile industry. The recognized adverse effect of RF EMF exposure is a thermal hazard. As an RF dosimetry quantity, the maximum spatial-average localized SAR has a strong correlation with the maximum temperature elevation in tissue for frequencies up to 3 GHz and modest correlation up to 10 GHz [103]. The SAR is a quantitative measure of RF power absorbed in tissue

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Figure 9.15 The illustration of small cell concept and the BS diversity scheme.

SAR =

σ | E ind |2 2ρ

(9.3)

in which Eind is the induced electric field in tissue, and σ and ρ are the electrical conductivity and the mass density of tissue, respectively. In such a frequency range, today, SAR can be easily measured using commercial phantoms and probes [104–106]. As the frequency increases, the energy penetration depth into tissue becomes very small, such as less than 2 mm at 15 GHz and less than 1 mm at 30 GHz. Even the 1-g averaging mass (specified by the FCC) for SAR is too large [107]. For frequencies above 10 GHz, the relationship between the maximum spatial-average SAR and the maximum temperature elevation becomes unstable [108]. Instead, recent thermal modeling studies for 5G UE show that the maximum spatial-average power density can be used as a temperature indicator in the millimeter-wave bands, even though power density is an RF dosimetric quantity in free space [109–111]. Before that, there is a lack of relevant studies supporting power density for UE in the present safety guidelines as follows. There are two major international regulatory guidelines to protect humans from excessive EMFs in RF. One is Guidelines for Limiting Exposure to Time-Varying Electric, Magnetic, and Electromagnetic Fields (Up to 300 GHz), published by the ICNIRP [111]. The other is the IEEE Standard for Safety Levels with Respect to Human Exposure to Radio Frequency Electromagnetic Fields, 3 kHz to 300 GHz [112, 113]. The United States has its own standard issued by

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the FCC [114]. The regulatory guidelines protect against adverse health effects associated with heating. All the regulatory guidelines distinguish the occupational and controlled exposure and general population and uncontrolled exposure. Occupational and controlled exposure limits apply in situations in which persons are fully aware of the potential for exposure and can exercise control over their exposure. General population and uncontrolled exposure limits apply in situations in which the general public may be exposed or in which persons who are exposed as a consequence of their employment may not be fully aware of the potential for exposure or cannot exercise control over their exposure. Above the transition frequencies (3 GHz for the IEEE, 6 GHz for the FCC, and 10 GHz for the ICNIRP), the restriction metric of RF EMF exposure is defined in terms of incident power density (i.e., the free-space Poynting vector): S=

1 Re  E × H *  2

(9.4)

A brief summary of general public and uncontrolled restrictions on power density is listed in Table 9.2. Different safety guidelines have different definitions to calculate power density over a certain amount of the averaging area or use the spatial peak value. Note that the FCC stipulates that the power density should be taken as a spatial-peak value. Recently, the FCC proposed a limit for an average area of 1 cm2 to be consistent with 1-g SAR, although this interpretation of maximum power density was officially been adopted in 2010 [115]. We refer to this as the proposed FCC guidelines. These guidelines are Table 9.2 A Brief Summary of General Public and Uncontrolled Restrictions on Power Density Frequency (GHz) FCC (Present) 6–100 FCC (Proposed) 6–100 ICNIRP 10–300 IEEE 3–30* 30–300

Spatial Peak and Spatial Averaging Peak 1 cm2 1 cm2 20 cm2 Peak 100λ2 Peak 100 cm2

Power Density Limits (W/m2) 10 10 200 10 18.56f 0.699 10 200 10

*To provide a transition in the frequency range of 3 GHz to 6 GHz, compliance may be demonstrated by evaluation of either incident power density or local SAR.

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all presently undergoing their periodic revision and are updating to fit into the new emerging technologies that will operate at the millimeter-wave frequencies. In the plane with separation distance d away from the antenna, the maximum power density averaged over any averaging area, A, as shown in Figure 9.16, in this plane, can be expressed as:

 ∫∫ S ⋅ nˆ dA  A S max (d ) = max  A   all A at d  A 

(9.5)

where nˆ is the unit vector perpendicular to A. For the maximum spatial peak power density, there are two kinds of understandings [116], one is choosing the maximum value of (9.4), that is, p S max (d ) =

max

all points at d

S

(9.6)

The other understanding is to take A → 0 for (9.5); thus,  ∫∫ S ⋅ nˆ dA  p S max (d ) = lim max  A  = max [ S ⋅ nˆ ]  all points at d A →0 all A at d  A 

(9.7)

which one is the correct understanding needs the ongoing standard revision to clarify in the future. Here we adopt (9.7) as the expression for the spatial-peak power density to be consistent with the International Electrotechnical Commission (IEC) TC106 AHG10.

Figure 9.16 Illustration of spatial averaging power density.

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Power Density Characteristic for Array Antennas

For a single antenna, the maximum power density happens very close to the antenna, while for array antennas, especially for large array antennas, the maximum exposure may occur some distance away from the face of the array due to the energy-focusing property of array antennas. For small arrays, the maximum power density may still occur in the immediate vicinity of the array. The location of the energy focus plane associates with the wavelength; therefore, for increasing frequencies, the distance of the maximum power density decreases. The maximum power density also depends on the array factor and the scan angle. For example, for a typical 8 × 1 patch array operating at 15 GHz (see [117, Figure 9.16(a)]), the values and positions of spatial-peak power density fluctuate with the phase shift angle at a plane of 2 mm, 20 mm, and 50 mm above the array (see [117, Figure 9]). 9.4.2

Power Constraint Due to Power Density Limits

As aforementioned, the maximum power density may occur some distance away from the array. Assuming that dc is the compliance boundary, for all distance above dc, the power density values are lower than a given power density limit Slim. The maximum permissible transmitted power, Pmax, can be calculated via the transmitted power Pt Pmax = max d >d c

PtS lim S max (d )

(9.8)

p A where Smax can be S max or S max depending on guidelines. Notice that the ICNIRP and the IEEE do not specify dc. The FCC stipulated that dc = 50 mm in the current standard. However, dc = 50 mm is too far away from the antenna in the context of 5G UE; thus, it has been noticed by the FCC, which proposed to remove it from the standard. However, measuring power density very close to the antennas would lead to much uncertainty because the measurement zone may locate inside the reactive near field. The attained data may not be trustworthy; thus, in practice, the telecommunication industry still may need a reasonable minimum compliance distance to ensure the credibility of measured power density data. In addition, if the dc is chosen to be very small, even if the measured data can be trusted, the maximum power density data could be incredibly high such that Pmax is too low to be used for mobile communication. For example, if dc = 1 mm, Pmax can be as low as −15 dBm for 8 × 1 patch array to be compliant with the current FCC standard at 28 GHz [117]. Reference [118] also showed that the calculated Pmax is significantly lower than the transmitted power levels of present cellular UE (23–33 dBm) for

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small arrays. The way to increase Pmax is either to increase the element number or steer the beam away from human bodies. It was shown that at the transition frequency where the exposure metric changes, the maximum radiated power to meet compliance with ICNIRP and FCC EMF limits, for a device used near bodies, presents a strong discontinuity (in the order of 6 dB for the investigated case) for a canonical dipole [119]. This discrepancy has no scientific basis and is due to inconsistencies in the exposure limits. With a growing interest for utilizing frequency bands in millimeter-wave bands for 5G, it is important that the inconsistencies at the transition frequency from SAR to power density are solved on a timely basis. The relevant standardization organizations and regulatory authorities are addressing this issue in the standard revision. 9.4.3

Conservative Power Density Assessment

To determine the power density in (9.4) through (9.7), knowledge of both the electric and the magnetic fields is required. The electric field is usually measured through dipole probes, and the magnetic field is usually measured through small loop probes. However, measuring both fields would add complexity to the measurement setup and increase the measurement time. Fast and accurate assessment methods are required to simplify the measurement setup and procedures. For a measurement system that provides the magnitude of the electric field only, the time-averaged plane-wave equivalent power density may be determined as: S PW =

1 2 E 2 η0

(9.9)

in which η0 denotes the free space wave impedance. Equation (9.9) gives an accurate estimation of the power density magnitude in the far-field region. For small distances, the near field detected is dominated by the closest antenna element since the other elements are located much further away. Even in the reactive near field, the Poynting vector still represents the directional energy flux density and its direction gives the local direction of wave propagation. For array antennas, a periodic power density pattern will be obtained over the assessment plane where, at least in the regions of maximum exposure, the wave propagation will be essentially perpendicular to the array face. For this case, it is reasonable to approximate the power density based on the tangential components of the electric field parallel to the UE surface. For a measurement system able to measure both the magnitude and phase of the electric field of each antenna element, an embedded measurement ap-

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proach may be utilized where each port is measured separately to allow for accurate field combining in a postprocessing step. With knowledge of the transmitted field from each antenna, conservative field combining methods have been developed where the exposure is maximized by assuming that the fields are totally correlated at all assessment points [120], denoted as the magnitude field-combining method (MFCM) and components field-combining method (CFCM): MFCM: E1 + E2 ≤ E1 + E2 CFCM: E1 + E2 ≤

∑ ( E 1τ

τ= x , y ,z

+ E 2τ

(9.10)

)2

(9.11)

By using two 5G UE mock-ups employing a patch array and a notch array, respectively, measurements of electric field strength magnitude were conducted in a semi-anechoic chamber using a DASY5 near-field measurement system together with an isotropic electric field probe EF3DV3 [121]. The level of conservativeness for the MFCM and CFCM is found to range between 5 and 10 dB, where the largest deviation is found for the ICNIRP exposure metric due to the larger averaging area. The simplest measurement approach is to make use of a scalar measurement system with an isotropic electric field probe. By measuring the electric field excited by each individual port, a conservative estimate of the maximum exposure for all possible phase excitations may be obtained by utilizing either the MFCM or CFCM. Despite its attractive simplicity, this approach suffers from severe drawbacks. The accuracy of MFCM and CFCM is reduced for small test separation distances. Furthermore, the level of conservativeness observed in [121] gives a lower Pmax, which will severely limit the applicability of these approaches for RF EMF compliance assessment of devices with array antennas used in the immediate vicinity of the body. The conservativeness problem of MFCM and CFCM may be alleviated by first conducting field strength measurements for all possible excitations separately, based on (9.10) or (9.11), and then select the maximum RF EMF exposure among all these measurements. However, this would make the measurement much more complicated. 9.4.4 Power Density Assessment Based on Near-Field Reconstruction Algorithms

For many of the systems measuring near-field strengths close to UE, the impact of probes may become quantitatively severe, to the point of jeopardizing the reliability of the results. In these cases, postprocessing algorithms are therefore necessary. The common way to tackle this problem is to take field measure-

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ments far enough from the UE and reconstruct the quantities of interest in proximity of the UE by inverse problems. In 2017, based on inverse problems, which in this context are also called near-field reconstruction algorithms, IEC TC 106 (AHG 10) developed a technical report specifying the state-of-the-art power density measurement techniques and test approaches from 6 GHz to 100 GHz [122]. The reconstructed sources could provide complex field information thus overcome the overconservativeness problem in the previous conservative assessment methods (CFCM and MFCM). As an example (the detailed report can be found in [122, Annex H.3]), a 28-GHz Sony mock-up using a 4 × 1 notch array was tested at Lund University, Sweden, as shown in Figure 9.17. The setup used for the measurement was mounted on an optical table and the planes were measured using a custom-made waveguide probe. The transmitting device under test (DUT) was

Figure 9.17 (Left) Picture of the mock-up used for power density measurements. (Right) The open waveguide probe and alignment system.

Figure 9.18 Distribution of the power density corresponding to the array with zero phaseshift between elements.

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connected through cable adapters to a VNA with the power of −10 dBm. The receiving probe had a similar chain and was connected to a second port of the same VNA. By means of a reconstruction algorithm, the measured E-field was used to determine power density. The 4 cm2 combined averaged power density including the contributions for the four ports was evaluated in the postprocessing step for in-phase combination and was based on the individual measurements of each port. At 27.925 GHz, its maximum on the top evaluation surface, for a distance of 5 mm and an accepted power of 20 mW (5 mW per port), was found to be 7.8 W/m2 based on measurements and 8.3 W/m2 with simulations. The measured power density distribution is shown in Figure 9.18. There are many challenges remaining for investigation such as measurement of multiple antenna array performance in small device and the also the measurements of device power density in efficiency ways.

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10 Unmanned Aerial Vehicles Josaphat Tetuko Sri Sumantyo

This chapter introduces small antennas onboard of unmanned aerial vehicles (UAVs) for communication, synthetic aperture radar (SAR), collision-free guidance control, direction-finding, conformal antenna, and others. Recently, various types of UAVs have been developed for specific applications (disaster monitoring, mapping, environmental observation, infrastructure monitoring) in different formats (fixed wing, rotary wing, balloon) and sizes (small, insect, medium, large-scale like Helios of NASA). Currently, research in UAV development is still evolving [1, 2]. UAVs can fly autonomously or be remotely controlled. Generally, UAV flies by remote control, but presently many types of UAV are flying using autonomous systems. Nowadays, low-cost and small UAVs are being developed for various civilian applications which caused the UAV market to explode. Basically, a UAV is composed by a main body, an engine, a controller, a communication unit, and a sensor unit. Advances in high-density (very large scale integrated) and high-speed processors boosted the technology of autonomous flight of UAVs. Hence, it is now possible for UAVs to fly autonomously by carrying onboard a Global Navigation Satellite Systems (GNSS, that is, GPS, GLONASS, BeiDou) receiver, a gyroscope, and an inertial measurement unit (IMU). In that way, the communication system connects the UAV with a ground station or a pilot to transfer telemetry data and UAV data, flight management data, and several sensor data, interactively. Recently, UAVs have been widely in many civilian applications with the aid of onboard devices as three-dimensional (3-D) digital camera for mapping, hyperspectral camera, infrared camera, communication relay, SAR, and lidar. 169

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The antenna is an important component for flight management and main mission device. Both small-scale and large-scale UAVs need an antenna. Modern UAV needs an antenna for a communication system, GNSS receiver, radar system, and other sensors. When developing antennas for UAV platforms, one needs to consider several parameters, such as size, weight, power consumption, location and field of view (beamwidth) on the platform, bandwidth, isolation and interference, reliability and safety, radome, polarization characteristics (linear polarization, elliptical polarization and linear polarization), signal-to-noise (S/N) ratio, sidelobe level, radiation efficiency, gain, noise temperature, and radar cross-section. Therefore, the development of antennas that are low-profile, lightweight, low-cost, thin, and conformal has become an attractive research area. Several techniques are being developed to tackle these issues, for instance, solid or active antenna types, smart, conformal, fractal, photonics, digital forming, multibeam, and broadband, among others. The location for installing antenna on UAV can be very restricted; for instance, an antenna is installed on the outside of a UAV as a gravity sensor and a magnetometer; a conformal antenna is low-profile and needs a complicated design and fabrication technique; an antenna inside a radome needs the consideration of the inefficiency of volume usage, radome loss, and large field of view. This section discusses antenna for communications, SAR, and other applications.

10.1

Communications

A UAV with a small radius of operation is commonly using direct communication between the UAV and the ground station, but a large radius of a flight operation needs satellite or aircraft to relay the flight, UAV-satellite (SATCOM)/ relay’s aircraft-ground station [3–5]. The UAV onboard antenna for SATCOM communications is shown on Figure 10.1. Data in this communication is UAV flight information as telemetry data (GPS, direction, attitude information or inertial measurement unit (IMU), velocity, altitude, inner temperature) and mission data from various sensor (gravity, SAR, camera, lidar) [6]. The polarization of an antenna uses linear polarization or circular polarization based on the attitude, field of view, and mission of UAV [7–9]. Some antennas need single band, multiband, or broadband [10, 11]. UAV communication uses S, C (WiMAX), X, and Ka-bands [1–3, 12]. Multipolarized antenna was also proposed to reduce interference on UAV communication [13]. Some communication antennas have been adjusted on the shape of the UAV’s body as a main wing or tail wing and a small antenna [14–17].

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Figure 10.1 Antennas for UAV-SATCOM. (© 2005 IEEE [4], © 2012 IEEE [5].)

10.2

SAR

The SAR sensor was first developed for aircraft and the satellite onboard sensor in the 1950s [18]. Particularly, spaceborne SAR has been useful in remotesensing applications, as it can monitor the Earth’s surface for long time frames. However, traditional spaceborne SAR missions have long satellite revisit times, long development times, bulky configuration, and consequently large costs. Alternatively, airborne SAR can be employed in a time-tailored way, but at increased costs as well. Hence, UAV SAR can overcome issues of low development time and costs for a designated application. Particularly, the hardware of a UAV SAR system is composed by a signal generator unit, transmitter, receiver, antenna for transmitter and receiver, motion composition unit (MOCO), and onboard computer (OBC). The SAR sensor signal could employ pulse (chirp pulse) or frequency-modulated continuous wave (FMCW). Various countries have already developed UAV SAR sensors for specific missions, for example, UAVSAR of NASA, NanoSAR of IMSAR (United States), MicroSAR and NuSAR of Brigham Young University (United States), ARBRES of The Universitat Politècnica de Catalunya (Spain), SUMATRA of Fraunhofer (Germany), BUSARD of ONERA (France), SARVANT of OrbiSat (Brazil), CARMSAR of Chinese Academy of Sciences (China), and CP-SAR of Chiba University (Japan) [19–25], as shown in Figure 10.2. Table 10.1 shows UAV SAR sensors in P, L, C, X, snf Ka-band configurations and millimeter-wave, linear, and circular polarization. Here, it is very important to match the antenna characteristics to each frequency and polarization requirements. Based on antenna types, scattering and polarization information of targeted objects can be acquired, and further image analysis and classification techniques can provide accurate ground information [26]. For example, Figure 10.3 shows a SAR image using linear polarization and circular polarization.

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Figure 10.2 Small SAR system and antenna (linear and circularly polarized SAR). (© 2012 EuSAR [21]. © 2012 IEEE [24]. © 2012 EuSAR [25]. © 2016 JIAS [26].)

The very high frequency (VHF) band of a wave is able to penetrate the underground level; therefore, a broadband Vivaldi antenna for VHF radar was installed on the main wing of the UAV [27, 28]. UAVSAR of NASA and CPSAR of Chiba University operates in L-band SAR using phased array antenna and microstrip array antenna, respectively [29, 30], as shown in Figure 10.4(a). As shown in Figure 10.4(b), CP-SAR of Chiba University operated 12 elements of circularly polarized microstrip antenna on one panel [31, 32]. This antenna employed Chebychev synthesis method to suppress the sidelobe of the radiation pattern of an array antenna to improve the quality of S/N of a SAR image [32, 33]. The millimeter wave of a SAR sensor is shown in Figure 10.5: ARBRES of The Universitat Politècnica de Catalunya (Spain, 35 GHz), MISAR of EADS (Germany, 35 GHz), MIRANDA of Fraunhofer (Germany, 35 GHz, 60 GHz,

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Table 10.1 UAV Onboard SAR Country United States

Institution NASA [19]

Name of SAR UAVSAR L

United States

Brigham Young University

MicroSAR NuSAR

Spain

The Universitat Politècnica de Catalunya [20] Fraunhofer [21] ONERA [23] OrbiSat [24] Chinese Academy of Sciences [25] Chiba University [26]

ARBRES

Mode of Observation Stripmap, spotlight, circular and motion target indicator Stripmap, bistatic, continuous, wave design stripmap, monostatic and pulsed design Back-projection

SUMATRA BUSARD SARVANT CARMSAR

Stripmap Stripmap Stripmap Stripmap

CP-SAR

Stripmap

Germany France Brazil China Japan

77 GHz, 94 GHz, or 220 GHz, which employed horn antenna [34–36]. The observation mode of a SAR sensor is stripmap, spotlight, scan SAR, and burst. Each mode could be realized by a phased array antenna as operated by UAVSAR of NASA [37] (Figure 10.6).

10.3

Antenna for Collision Avoidance and Direction Finding

The increasing number of UAV platforms has recently drawn attention towards flight safety and reliability. Algorithms and new techniques for collision avoidance and flight safety are being developed, such as nonlinear model predictive control (NMPC), rule-based technique, force field algorithm, quadratic programming (QP) for collision avoidance, and direction finding [38, 39]. For collision avoidance and direction finding as shown in Figure 10.7, Koch fractal antenna [40] for monopulse collision avoidance radar, vector sensor antenna [41] for direction finding, and modified helix antenna for high-altitude UAV were used to estimate the arrival angle of the wave [42].

10.4

Conformal Antenna and Others

Since each part of the UAV platform can be used for antenna installation, conformal antennas can be developed and customized to fit the shape of the UAV nose, body, main wing, and tail wing of UAV [43–45], for instance, conformal antennas with metal monopole antenna [46] and PIFA [47] for broad beamwidth. Figure 10.8 shows an array conformal antenna on the nose and the front

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Figure 10.3 Scattering based on polarization of antennas. (Credit: Josaphat Laboratory, Chiba University.)

of the UAV (azimuth direction) to realize a large field of view [48]. Figure 10.9 shows a conformal annular slot antenna [49] and a Z-shaped slot array antenna

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Figure 10.4 L-band SAR for UAV.

[50] to reduce the air drag of the UAV’s body and realize an omniradiation pattern. Recently, various UAV have been found in the market, and many antenna types have been developed based on the shape, mission, and size of the UAV platform. For instance, Figure 10.10 depicts the two channels’ array SAR for the hexacopter UAV developed by IMST in Germany [51]. The SAR onboard UAV for mobile satellite communications was also developed as shown

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Figure 10.5 Millimeter-wave SAR and horn antenna for UAV. (© 2007 IEEE [34]. © 2008 EuMA [35]. © 2011 EuRAD [36].)

Figure 10.6 Phased array antenna for UAVSAR. (© 2006 IEEE [37].)

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Figure 10.7 Antenna for collision avoidance. (© 2015 Radioelektronika [40]. © 2006 IEEE [41]. © 2007 PIMRC [42].)

Figure 10.8 Antenna installed on the UAV’s nose. (© 2010 EuCAP [48].)

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Figure 10.9 Antenna installed on the UAV’s body. (© 2010 LAPC [49].)

Figure 10.10 SAR for rotor UAV. (© 2014 IEEE [51].)

Figure 10.11 SAR for GMTI. (© 2015 WDD [52].)

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in Figure 10.11 [52]. This SAR system uses a simple microstrip and slot array antenna.

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Sun, L., et al., “Miniaturized Annular Ring Slot Antenna for Small/Mini UAV Applications,” Progress in Electromagnetics Research C, Vol. 54, 2014, pp. 1–7.

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Sun, L., et al., “Low-Proile, Quasi-Omnidirectional Substrate Integrated Waveguide (SIW) Multihorn Antenna,” IEEE Antennas and Wireless Propagation Letters, Vol. 15, 2016, pp. 818–821.

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11 Antennas for Wireless Medical Devices Karu P. Esselle, Basit A. Zeb, Raheel Hashmi, and Roy B.V.B. Simorangkir

11.1

Introduction

There has been a rapid growth in medical and healthcare devices that incorporate wireless technology. A medical device with wireless capability, hereinafter referred to as a wireless medical device, uses electromagnetic waves and radio-frequency (RF) antennas to collect and transfer rich, often time-critical, data during medical treatments, diagnostic procedures, clinical research, and healthcare delivery. These wireless medical devices are typically placed inside or outside the human body and perform various functions such as: • Transferring data from one device to another and from a device to an external control unit (stationary or mobile) (e.g., a smart phone or nursing station); • Monitoring patients’ health remotely by healthcare professionals such as heart rate, blood glucose levels, and blood pressure; • Device control and programming to troubleshoot device-related issues quickly, easily, and remotely. To achieve these functionalities, a wireless medical device needs compact high-performance antennas. The antenna research and development offer significant advantages to the medical industry, healthcare professionals, and 183

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patients. First, the antennas eliminate the need for physical wires that tether a patient to a medical bed and would otherwise pass through or rest on the body to facilitate monitoring, diagnosis, and therapy. This increases patient mobility and comfort yet provides real-time data connectivity with healthcare professionals (doctors and nurses) around the clock regardless of patients’ location (hospital, home, and office). Second, antennas enable medical devices to be portable and wearable as compared to bulky importable medical equipment currently used in the industry. This accelerates the development and deployment of potentially small medical devices with low cost. In this context, antennas dedicated for medical device connectivity are classified into three categories: (1) in-body antennas that are placed inside the human body such as implantable, ingestible, or injectable antennas; (2) onbody antennas that are temporarily or permanently attached to the exterior of the body or are integrated into wearable garments; and (3) external antennas mounted on monitoring/control equipment such as mobile nursing stations that establish close-range links with a hospital network or long-range fixed wireless access between hospitals and clinics. The antenna design process has several associated challenges such as that the antenna should be compact and fit in small areas, it should be effectively integrated or conformed to the medical device, and its electromagnetic performance needs to be characterized in detail. Antennas for implant medical devices and biotelemetry [1, 2] bring additional challenges in terms of material selection, placement strategies, biocompatibility, and patient safety. The real challenge is how to squeeze the antenna into smaller spaces and still maintain good performance and meet the medical device product specifications on operating power levels, bandwidth, and quality of the wireless interface. The antenna performance is also critically linked to the operating frequency and the selection of a particular frequency band is dictated by the trade-off between the loss through the body, communication speed, and device miniaturization. In general, for medical device connectivity both near-field (inductive coupling) and far-field (RF radiated energy) communication can be used such as the standardized technologies of Wi-Fi, Bluetooth, and near-field communications (NFC). A detailed but nonexhaustive list of medical frequency bands allocated by spectrum regulatory authorities and the types of antennas are given in Table 11.1. It is important to note that, in the past, low-frequency coil antennas have been commonly used to establish inductive links; however, they are gradually replaced by high-frequency antennas to overcome limitations of low-data rates, misalignment of coils, and short communication range and hence are not discussed here. Generally, implanted/on-body antennas use 400-MHz medical implant communications system (MICS) band, 900-MHz/2.4-GHz industrial scientific medical (ISM) bands, or a lower portion of the FCC ultrawideband (UWB)

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band (e.g., 3.1–4.5 GHz). In addition, some body-worn antennas and external antennas mounted on portable nursing stations use mobile cellular networks or WiMAX at 2.5/3.5 GHz. Commonly used antenna types are printed patches, meandered-line dipoles, multilayer PIFAs, helix, and spirals. Stacked multilayer PIFAs are good candidates for implantable devices [3–14] as they can achieve dual-band and triple-band operational 402-MHz MICS and 433, 900, and 2,450-MHz ISM bands in compact size. Helical and spiral antennas are typically suited for ingestible devices such as wireless capsule endoscopes [15–21] due to their ability to produce circular polarization, which helps to compensate the link margin due to the capsule’s orientation inside the body. Miniaturized loop or dipole antennas are used in injectable devices for neuro-stimulation applications [22–24] while UWB circular, elliptical patches or slots are increasingly used for high-data rate applications such as high-resolution breast cancer imaging [25, 26]. Antenna design for wireless medical devices is typically carried out using full-wave three-dimensional (3-D) numerical electromagnetic solvers such as ANSYS HFSS and CST Microwave Studio. Different from free-space antenna simulations, they take into account the analytical models of the human body to study RF attenuation through the biological tissues [27–29] and antenna detuning effects caused by thin biocompatible materials used as coatings and superstrates. The antenna designs are then validated by experiments performed in vitro using phantoms and tissue-mimicking gels that emulate the electrical properties of biological tissues and/or in vivo using animals such as rats and pigs. Some typical bio-accepted materials are Silicone (PDMS) Elastomer Sylgard184 from Dow Corning, MED-1134 from NuSil, and Parylene-C polymer. Often RF properties (permittivity and loss factor) of such biocompatible materials are experimentally characterized before realizing the antenna design because these properties are typically obtained at frequencies well below the lowest medical frequency band (i.e., 400-MHz MICS). For example, the aforementioned biocompatible materials have supplier-provided dielectric constant of εr~2.7–3 typically at 1 MHz. In the following sections, we present and discuss several recent antenna designs for in-body and on-body devices with particular focus on implantable, ingestible, and embroidered antennas for biotelemetry and body-area network applications.

11.2

Antennas for In-Body Medical Devices

To enable wireless transmission thorough the body (i.e., from implanted wireless medical devices to a device outside the body), implanted in-body antennas are required. The device outside the body could be a monitoring hub station

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operating in typical free-space conditions or a wearable hub over the body of the person of interest. Most of the standard antennas can be used for the former type. Antennas suitable for the latter are discussed in Section 11.3. In this section, let us focus on implantable antennas for devices in the body. Implantable medical devices can be broadly classified in to two classes: stationary and moving (relative to the body). A typical example for the former is an artificial cardiac pacemaker [30] and a typical example for the latter is an endoscopy capsule [15]. In principle, similar antennas can be used for both types, but practical issues associated with moving devices could be even more challenging due to variable environment. Let us first consider stationary wireless medical devices. After implantation, tissues are expected to grow around the device to accept and host it. After this period, the implanted device is expected to be nearly stationary relative to the person. The RF subsystem of the device, including the antennas, is designed to operate well in this environment. Often, it is desirable, and sometimes essential, to make these antennas very small. High dielectric constant of body tissues help the antenna designer to make the antenna much smaller than those operating in free space (at the same frequency). Nevertheless, the designer should resist the temptation to minimize the bandwidth of the antenna for the sake of miniaturization. This is because the dielectric properties of tissues surrounding the antenna can vary from person to person and in some cases with time for the same person. The latter is quite common in moving medical devices. Hence, the dielectric loading on the antenna can differ from design parameters, causing the antenna to detune (i.e., shift the resonance frequency). A potential risk of this is the unacceptable impedance mismatch within the system bandwidth and hence the failure of the wireless link. A very successful method to mitigate the above-mentioned problem caused by variation of tissue loading on antenna is preloading the antenna with a dielectric layer. This method is based on two facts: 1. The most sensitive dielectric layer is the one that is adjacent to the key radiating parts (often the metal pieces of the antenna). 2. When this layer is suficiently thick, antenna impedance matching is almost insensitive to body tissues. This method has been successfully applied, in different ways, to achieve reliable performance form both stationary and moving antennas. One example for each application is outlined in the following chapters, as examples.

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Table 11.1 Frequency Bands Used by Antennas for In-Body, On-Body, and External Medical Device Connectivity Frequency Band Wi-Fi 2.4 and 5 GHz

Bluetooth (BLE, 2,404–2,478 low energy) MHz version 4.0

Wireless Medical Telemetry Service (WMTS) MedRadio/ micropower networks (MMN)

608–614 MHz, 1,395–1,400 MHz, and 1,427–1,432 MHz 413–457 MHz [413–419, 426– 432, 438–444, 451–457] MHz

Medical body 2,360–2,400 area networks MHz (MBAN) Medical Implant Communication Services (MICS)

401–402, 402–405, 403.5–403.8 (MITS) and 405–406 (MEDS) MHz ISM 433.1–434.8, 868–868.6, 902.8–928.0 MHz, 2,400–2,483.5 MHz UWB 3.1–5, 6–10, 22–29 (USA) GHz

WiMax

2.3, 2.5, and 3.5 GHz

Typical Antenna Dual-band patch, patch arrays, MIMO antennas PCB mount chip antenna, IFAs monopoles, PIFA

PIFAs, planar meandered dipoles

Planar meandered line dipoles and its variations

Planar monopoles, vertical monopoles, IFAs Stacked PIFA, chip/ PCB mount ceramic antennas

Applications Smart hospital beds, oxygen monitors, infusion pumps, mobile nursing stations Indoor navigation for patients, connectivity between device and smartphone for health data monitoring Implant biotelemetry, intraocular unit for retinal prosthesis, capsule endoscopy

Notes 10 to 150m, up to 1 Mbps

Transmit and relay data for implanted and bodyworn medical devices for diagnostic and therapeutic functions On-body/wearable sensors controlled by a unit in proximity (a few centimeters) of body Two-way biotelemetry between implanted device and external hub for diagnosis and/or therapy

1m, 250 kbps

Low power

2–4m,