Reconfigurable Antenna Design and Analysis 1630817074, 9781630817077

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Reconfigurable Antenna Design and Analysis

For a complete listing of titles in the Antennas and Electromagnetics Analysis Library, turn to the back of this book.

Reconfigurable Antenna Design and Analysis Mohammod Ali

Library of Congress Cataloging-in-Publication Data A catalog record for this book is available from the U.S. Library of Congress. British Library Cataloguing in Publication Data A catalogue record for this book is available from the British Library. Cover design by Charlene Stevens

ISBN 13: 978-1-63081-707-7

© 2021 ARTECH HOUSE 685 Canton Street Norwood, MA 02062

All rights reserved. Printed and bound in the United States of America. No part of this book may be reproduced or utilized in any form or by any means, electronic or mechanical, including photocopying, recording, or by any information storage and retrieval system, without permission in writing from the publisher.   All terms mentioned in this book that are known to be trademarks or service marks have been appropriately capitalized. Artech House cannot attest to the accuracy of this information. Use of a term in this book should not be regarded as affecting the validity of any trademark or service mark.

10 9 8 7 6 5 4 3 2 1

Contents

Preface

1

Introduction to Reconfigurable Antennas

1

1.1

Application Domains

1

1.2

The Basics of Reconfigurable Antennas

2

1.3 1.3.1 1.3.2 1.3.3

xi

Types of Antenna Reconfiguration Frequency Reconfiguration Pattern Reconfiguration Polarization Reconfiguration References

3 4 8 10 10

2

Fundamental Definitions of Antenna Parameters

15

2.1

Radiation Patterns

16

2.2

Input Impedance, Voltage Standing-Wave Ratio, and Return Loss

19

2.3

Bandwidth

21

2.4

Directivity, Gain, and Realized Gain

23

2.5

The Friis Transmission Formula

27

v

vi

Reconfigurable Antenna Design and Analysis

2.6

Polarization

28

2.7

The Wheeler Cap Method of Antenna Efficiency Measurement

30

2.8

Isolation Between Antennas in a System

32

2.9

MIMO Antenna Metrics

33

References

34

3

Overview of RF/Microwave Switches

37

3.1

Introduction

37

3.2

The PIN Diode Switch

38

3.3

The RF MEMS Switch

41

3.4

The Varactor Diode Switch

43

3.5

Other Types of RF/Microwave Switches

44

3.6 3.6.1 3.6.2 3.6.3

Switching Circuits and Their Responses Series Switch Shunt Switch Series-Shunt Switch

44 45 46 46

3.7

Concluding Remarks

47

References

47

4

Basic Antenna Configurations

51

4.1 4.1.1 4.1.2 4.1.3 4.1.4 4.1.5

The Dipole Antenna The Thin-Wire Dipole Antenna Feeding or Exciting a Dipole Antenna Dipole Operation Against a Metal Reflector Dipole Against an EBG Structure Miniaturized Antennas: Meander and Zigzag Dipoles

52 53 59 64 67 68

4.2 4.2.1 4.2.2 4.2.3 4.2.4

The Thin-Wire Monopole Antenna Monopole Input Impedance Monopole Radiation Properties Monopole Properties: Ground Plane Effects The Inverted-L Antenna (ILA)

69 69 70 70 71



Contents

vii

4.3 4.3.1 4.3.2 4.3.3

The Loop Antenna The Small Loop Antenna The Resonant Loop Antenna The Resonant Loop Against a Metal Reflector

72 72 74 75

4.4 4.4.1 4.4.2

The Microstrip Patch Antenna The Rectangular Patch Antenna The Shorted Quarter-Wave Patch Antenna

75 76 83

4.5 4.5.1 4.5.2

The Inverted-F Antenna The Planar Inverted-F Antenna The IFA

84 84 85

4.6

The Slot Antenna

86

References

88

5

Frequency and Polarization Reconfiguration

93

5.1 5.1.1 5.1.2 5.1.3 5.1.4 5.1.5

Frequency Reconfigurable Patch Antennas 95 PIN Diode Reconfigurable Patch Antenna and Arrays 95 PIN Diode Reconfigurable Transmitter Patch Antenna 99 Frequency Reconfigurable U-Slot Patch Antenna Using a Trimmer Capacitor 100 Reconfigurable Patch-Slot Antenna Using PIN Diode 100 Frequency and Pattern Reconfigurable Patch Antenna Using the PIN Diode 104

5.2 5.2.1 5.2.2 5.2.3

Polarization Reconfigurable Patch Antenna Patch Antenna with Switchable Slots (PASS) Circular Polarization Frequency and Polarization Reconfigurable Patch Antenna Frequency and Circular Polarization Reconfigurable Patch Antenna

105

5.3 5.3.1 5.3.2

Reconfigurable Slot Antennas UHF-Band PIN Diode Reconfigurable Slot Antenna Reconfigurable Folded Slot Antenna

113 113 117

5.4 5.4.1 5.4.2

MEMS Reconfigurable Antennas Monolithic MEMS Reconfigurable Patch Antennas Discrete MEMS Frequency Reconfigurable Antenna

118 118 122

5.5

Reconfigurable Dipole, Monopole-Type Antennas

123

105 107 110

viii

Reconfigurable Antenna Design and Analysis

5.5.1 5.5.2 5.5.3

Frequency Reconfigurable Bow-Tie Antenna Reconfigurable Antenna Using a Photoconductive Switch Pattern and Frequency Reconfigurable Yagi Antenna

123

5.6

The Reconfigurable Antenna Aperture (RECAP)

128

5.7

More Examples of Frequency Reconfigurable Antennas 131

5.8

Concluding Remarks References

124 126

135 136

6

Pattern Reconfiguration

141

6.1

The Yagi-Uda Array

142

6.2 6.2.1 6.2.2

Pattern Reconfigurable Yagi-Uda Arrays Reconfigurable Printed Microstrip Yagi-Uda Dipole Array Reconfigurable Yagi-Uda Array

145

6.3 6.3.1 6.3.2 6.3.3

Pattern Reconfigurable Parasitic Dipole Arrays Reactively Controlled Dipole Array Reactively Controlled Monopole Array Switched Parasitic Dipole Array

147 147 148 149

6.4 6.4.1 6.4.2 6.4.3

Reconfigurable Parasitic Patch Array An ESPAR Patch Antenna Array An Aperture-Coupled ESPAR Patch Antenna Array A Frequency and Pattern Reconfigurable Pixelled Monopole Antenna

151 152 152

6.5

Series-Fed Patch Phased Array

158

6.6

Beam Steering Colinear Dipole Array

162

6.7

Other Examples of Pattern Reconfigurable Antennas

164

References

145 146

156

166

7

Basic Scanning Antenna Array Design

169

7.1

Introduction

169

7.2

Mathematical Perspectives

172



Contents

ix

7.3 7.3.1 7.3.2 7.3.3

Phase Shifter Fundamentals Switched Line Phase Shifter Quadrature Phase Shifter or Reflective Phase Shifter Loaded Line Phase Shifter

176 177 178 179

7.4 7.4.1 7.4.2

Linear Microstrip Patch Scanning Array Example Simple Phase Shifter and Beam Scanning Example Beam-Scanning Examples at Additional Angles

179 180 183

7.5

Phase Shifter and Phased Array Examples

186

References

191

8

Switch Biasing and Other Considerations

195

8.1 8.1.1

DC Biasing of PIN Diode Switches Examples of PIN Diode Switch Biasing

196 198

8.2 8.2.1

DC Biasing of Varactor Diode Switches Examples of Varactor Diode Biasing

200 201

8.3 8.3.1

DC Biasing of RF MEMS Switches Examples of RF MEMS Switch Biasing

203 203

8.4

DC Biasing of Other Types of Switches

204

8.5

Concluding Remarks

204

References

205

9

Modeling and Simulation of Reconfigurable Antennas 209

9.1 9.1.1

Reconfigurable Antenna Simulation Consisting of PIN Diodes 210 Examples of PIN Diode Reconfigurable Antenna Simulations 211

9.2 9.2.1

Reconfigurable Antenna Simulation Consisting of Varactor Diodes Examples of Varactor Diode Reconfigurable Antenna Simulations

9.3 9.3.1

Reconfigurable Antenna Simulation Using RF MEMS Switches Examples of MEMS Reconfigurable Antenna Simulations

214 214 216 216

x

Reconfigurable Antenna Design and Analysis

9.4

Other Reconfigurable Antenna Simulation Examples

218

9.5

Concluding Remarks

219

References

219

10

MIMO Reconfigurable Antennas

223

10.1

MIMO Antenna Parameters

224

10.2

Examples of MIMO Antennas

228

10.3

Mutual Coupling and Coupling Reduction Techniques 235

10.4

Conclusion References

238 239

11

Reconfigurable Antennas for Cognitive Radio, Millimeter-Wave 5G, and Other Applications

243

11.1

Reconfigurable Antennas for Cognitive Radio

243

11.2

Millimeter-Wave Reconfigurable Antennas

245

11.3

Reconfigurable Antennas for Sub-6-GHz 5G Applications

254

References

255



About the Author

259



Index

261

Preface The subject of reconfigurable antennas has drawn a great deal of interest among engineers, researchers, graduate students, and scientists primarily because of their attractive properties; that is, a single antenna structure or geometry can be reconfigured in frequency, pattern, or polarization. Reconfigurable antennas find applications in commercial wireless communications, automotive radar, biomedical applications, and many other emerging applications. The purpose of this book is to provide a source for understanding the design and analysis of reconfigurable antennas as they pertain to practical applications. Chapter 1 starts with a basic introduction to reconfigurable antennas and illustrates the differences between frequency, pattern, and polarization reconfiguration. Chapter 2 describes relevant basic antenna parameters in a focused manner so that the reader does not have to consult an antenna textbook while reading this book. Discussions of antenna efficiency measurements using the Wheeler cap method and antenna mutual coupling are presented. Because most reconfigurable antennas attain their operation with the help of electronic switches, Chapter 3 focuses on that subject. Chapter 3 also presents the fundamentals of PIN diode, varactor diode, and radio frequency (RF) microelectromechanical systems (MEMS) switches. Simple analytical formulations on switch insertion loss and isolation calculation are provided. Chapter 4 presents the fundamental building blocks or antennas that may be used to design and build a reconfigurable antenna. Discussions on dipole, monopole, patch, slot, loop, and other antennas are presented along with their feeding methods. Chapter 5 presents a detailed overview of frequency reconfigurable antennas. Polarization reconfiguration is also presented in the same chapter. The topic of frequency reconfiguration is addressed in the context of antenna type (patch, dixi

xii

Reconfigurable Antenna Design and Analysis

pole, slot) and the electronic switches used; for example, PIN diode, RF MEMS (discrete versus monolithic), photoconductive device, and varactor diode. Chapter 6 discusses the various types of pattern reconfigurable antennas that are not phased arrays. For example, the electronically steerable parasitic array radiator (ESPAR) is presented that leverages dipole, monopole, and patch antennas. Chapter 7 presents a brief introduction to phased array or scanning array design, which includes phase shifter design, simple examples of microstrip patch phased array design, and phased array design examples from the literature. Chapter 8 discusses RF switch biasing, and Chapter 9 discusses reconfigurable antenna simulation strategies when they contain practical RF switches. Chapter 10 is focused on multiple-input multiple-output (MIMO) reconfigurable antennas. Fundamentals of MIMO or MIMO parameters are defined first, followed by MIMO antenna design examples from the literature. Finally, Chapter 11 presents reconfigurable antenna design for cognitive radio and millimeter-wave systems. The field of reconfigurable antennas is rapidly moving forward with new publications appearing continuously. Therefore, it is not possible to provide a summary on all available reconfigurable antennas. I have made an attempt to consult as much as of the current literature as possible and to present an up-todate summary on reconfigurable antennas. I would like to acknowledge the authors and coauthors of the papers, books, theses, and reports that I have studied to write this book. It was an intriguing journey to discover so many excellent ideas. The antenna textbooks by R. S. Elliott, C. A. Balanis, W. L. Stutzman, and G. A. Thiele and the microwave engineering textbook by D. M. Pozar have been very helpful. I would also like to thank all my graduate students throughout the years. Some of the examples provided in this book came from their papers and theses. I am grateful to David Michelson from Artech House for initially getting in touch with me a couple of years ago about a book proposal. Subsequently, Rachel Gibson from Artech House was in regular contact with me while the manuscript was being developed. Again, it was David who took over from Rachel and helped to propel it forward to the end. I would like to thank the Artech House production team for doing an excellent job. Finally, I would like to express my gratitude to my wife, my son and daughter in-law, and my daughter and son in-law, all of whom have been constant sources of support and inspiration in this journey.

1 Introduction to Reconfigurable Antennas 1.1  Application Domains Wireless communication, navigation, and radar have seen rapid growth in recent years. Wireless technology is now everywhere in the world. Even in the remotest villages in the developing world, one can see the use of smart phones and other wireless devices. In the developed world, wireless technology can be seen in handheld, wearable (smart watch), and implantable devices. Smart homes are increasingly being equipped with Internet of Things (IoT)-type devices (e.g., Nest from Google and Alexa from Amazon). Wireless routers with multiple-input multiple-output (MIMO) features are also now common and have become necessary to provide support for high-speed video data and gaming. Modern-day automobiles contain a plethora of wireless devices and upcoming self-driving cars will contain even more wireless and radar technologies. The field of medical applications is another area where wireless technology is having a tremendous impact in terms of telehealth, wearable sensors, and implantable wireless sensor technologies. Industrial manufacturing is seeing the integration of IoT devices for sophisticated manufacturing. Wireless systems have been finding applications in remote areas such as the Antarctic to monitor the environment and the polar caps. In essence, wireless applications are numerous and always growing. All of these and other wireless applications require well-performing antennas. While the performance criteria vary from application to application, there is a tremendous need for antennas that can support multiple functions. There is also a need to achieve that with a miniature size so that such antennas can be easily integrated into the housing of a small device. 1

2

Reconfigurable Antenna Design and Analysis

If a single antenna aperture can be modified or reconfigured to perform multiple functions, such an antenna can be called a reconfigurable antenna. The literature contains many examples of reconfigurable antennas, some of which can be found in [1–46]. Many other examples will be referred to and discussed in the next chapters. Some application examples of reconfigurable antennas are provided in Figure 1.1(a).

1.2  The Basics of Reconfigurable Antennas In general, there are three broad classes of reconfigurable antennas: (1) frequency, (2) pattern, and (3) polarization (see Figure 1.1(b)). For frequency reconfiguration, a single antenna geometry can be segmented, and certain of its sections can be activated or deactivated for the antenna to operate in a certain frequency band. For pattern reconfiguration, an antenna may contain multiple elements (some driven, some passive, or all driven), and, by controlling their states and other properties, the radiation pattern in the space can be steered in different directions, thus allowing opportunistic sensing (e.g., for the case of cognitive radio or communications). For polarization reconfiguration, the antenna is reconfigured or manipulated so that antenna polarization changes as function of the reconfiguration states.

Figure 1.1  (a) Application examples of reconfigurable antennas and (b) types of reconfiguration.



Introduction to Reconfigurable Antennas

3

Because the act of reconfiguration generally requires activating or moving parts or sections of an antenna while the other parts or sections remain inactive, some form of an actuation device is needed. Electronic switching devices such as PIN diodes, varactor diodes, and radio frequency (RF) microelectromechanical systems (MEMS), photoconductive switches are the most commonly used devices in reconfigurable antennas. However, examples of using liquid metal switches or liquid metal channels to achieve reconfiguration are available [34–37]. Similarly, the use of phase change materials, like VO2, has also been proposed where the VO2 device changes from a high-resistance structure to a low-resistance structure when heated or biased through electrical currents [38, 39]. Finally, although not common, there has been an idea proposed that antenna structures can be reconfigured through physical mechanical movement [40]. Naturally, the use of a liquid metal switch and mechanical movement will be slower and may only be applicable for cases where the speed of the reconfiguration is not an issue.

1.3  Types of Antenna Reconfiguration Reconfigurable antennas have received widespread attention among researchers and engineers for nearly three decades. As stated before, their application domain may include commercial wireless applications, such as mobile phones, IoT devices, laptops, smart watches, automotive radar, navigation, and medical applications. In general, wherever there are space constraints, it is challenging or even impossible to install a large broadband antenna that can easily have an aperture size that is one wavelength or larger. Broadband antennas such as equiangular spirals, Archimedean spirals, the sinuous antenna, and the axial mode helical antennas may be more suitable for ground-based, marine, space-based, or aircraft platforms where space may not necessarily be a challenge and consistent pattern, gain, and polarization may need to be satisfied over an entire range of frequencies. However, for many other applications, space is a primary challenge. For example, a modern-day smart phone is only about 120 mm long and 50 mm wide and can easily contain many antennas including those to support mobile phone services in the 800–900-MHz, 1,710–1,880-MHz, and 1,850–1,990-MHz Long Term Evolution services in the 700–800-MHz band, Universal Mobile Telecommunications Service (UMTS), Wi-Fi, and GPS, to name a few. While many of these applications may require separate antennas, it is always desirable if a single antenna can be used to provide multiple functionalities at multiple frequency bands. For example, an antenna can be reconfigured for operation at multiple frequency bands, making it a frequency reconfigurable antenna. An antenna may operate in the same frequency band

4

Reconfigurable Antenna Design and Analysis

but may attain different pattern or polarization states, thus being a pattern or polarization reconfigurable antenna. 1.3.1 Frequency Reconfiguration

The act of reconfiguration means that the antenna geometry, components, or material constituents are rearranged (not necessarily by physically moving but through electrical connections) to perform different operations at different times. Antenna frequency reconfiguration is such an example. For instance, consider a dipole antenna containing four electronic switches as shown in Figure 1.2. The dipole is excited at its center. For now, let us not consider issues such as switch biasing, switch types, size, insertion loss, isolation, and material. Let us assume that there are four ideal switches placed as 1, 2, 3, and 4 in Figure 1.2 for the center-fed dipole. Given this scenario, when all switches are on, the dipole operates at a low frequency of f1. When all switches are off, the dipole operates at a high frequency f3. Finally, when switches 1 and 4 are off but switches 2 and 3 are on, the dipole operates at an intermediate frequency, f2. As illustrated in Figure 1.2, the dipole segments are so chosen and the strip width of the dipole is such that these resonant frequencies are discrete, and they do not contain an overlap in their operating frequency bands as can be seen from their S11 plot illustrations. This is an example of a discrete frequency reconfiguration where the frequency bands are quite separate from each other. The dipole example of Figure 1.2 does not necessarily need to be symmetric. If switches 1, 2, and 3 are on while switch 4 is off, the resultant antenna is an asymmetric dipole. It is a similar situation if switches 2, 3, and 4 are on but switch 1 is off.

Figure 1.2  Example of frequency reconfiguration: (a) a frequency reconfigurable dipole antenna, and (b) discrete frequency reconfiguration.



Introduction to Reconfigurable Antennas

5

A second example of frequency reconfiguration is shown in Figure 1.3 where the S11 characteristics show that each of the reconfiguration states overlaps at their band edges assuming the magnitude of S11 less than ‒10 dB as the limit. Although the reconfiguration states themselves are discrete, the resulting operation is continuous frequency reconfiguration where each of the reconfiguration frequency bands is contiguous and has a small overlap. If each resonant mode (f1 through f3) has enough bandwidth, the reconfigurable antenna can cover a wide frequency band. Conversely, if each resonant mode only covers a very narrow bandwidth (say, for example, 2%, which may very well be the case for a miniaturized antenna), the resulting bandwidth is not wideband but could still be significant (10%) for a miniaturized reconfigurable antenna. One must caution and prepare the reader about the practical underpinnings of the above two examples. Although discrete and continuous frequency reconfiguration examples are shown with the help of only S11 versus frequency plots, one must remember that for each reconfiguration frequency band or mode the antenna must satisfy all other performance requirements (e.g., pattern, gain, polarization) to make it practically useful and relevant. The more parameters an antenna must satisfy, the more difficult the design becomes because of obvious challenges associated with them. For cases with discrete frequency reconfiguration where the two frequency bands are further separated from each other, the act of reconfiguration prevents the antenna from collecting noise at the other frequency. In Chapter 5, many examples of frequency reconfiguration for dipoles, slots, and patch type

Figure 1.3  Continuous frequency reconfigurable antenna.

6

Reconfigurable Antenna Design and Analysis

antennas will be provided. As will be shown, the act of frequency reconfiguration without mechanical movement of the antenna requires switches to be placed at appropriate locations along the geometry of the antenna. When the switch is controlled close or open, this changes the geometry and/or dimensions of the antenna. In an RF operation, an ideal switch will present itself as a short circuit in a close position and as an open circuit in an open position. With real electronic switches (e.g., varactor diodes, PIN diodes, MEMS, photoconductive switches, switches made from phase change materials such as VO2), this ideal performance will not occur. Switches in the on position will present insertion loss. In the off or open position, switches will be characterized by their isolation. Both insertion loss and isolation are important metrics to consider when designing a reconfigurable antenna. A first step in a frequency reconfigurable or pattern reconfigurable antenna design using full-wave electromagnetic (EM) simulation may consider ideal switches. Soon thereafter, the antenna simulation models should contain representative switch equivalent circuit models that reflect their realistic insertion loss and isolation at the frequencies of interest. For example, for a PIN diode, a very basic circuit model is a forward resistance in the on state and a small capacitance in the off state, generally obtained from the manufacturer’s datasheet. For reconfiguration operation that spans over large frequency bands, the switches used must have acceptable performance in each frequency band. One switch type may not be possible to use at all frequencies. Except for photoconductive switches, PIN diodes, MEMS, and varactors will require direct current (DC) bias or supply. The DC bias must be designed with care in order to avoid interference. Examples of a frequency reconfigurable folded slot antenna and a patch antenna can be seen in Figures 1.4 and 1.5, respectively.

Figure 1.4  Schematic of the reconfigurable folded slot antenna. The connection between the metallic strip and the ground plane is controlled by the diode’s state. The dimensions are as in Table 5.4(b), except for Ls, which is 21.22 mm. (© 2009 IEEE. Reprinted, with permission, from: [19].)



Introduction to Reconfigurable Antennas

7

Figure 1.5  A PIN diode reconfigurable patch antenna. (After: [30].)

The CPW-fed reconfigurable folded slot antenna contains two PIN diode switches at its two edges. For the folded slot antenna, the first operating frequency is governed by the perimeter of the folded slot. Thus, when the PIN diode switches are off, the slot assumes a longer perimeter, hence a lower frequency of operation, while when the switches are on, the slot assumes a somewhat shorter perimeter, hence a higher frequency of operation. The example given in [19] shows those two frequencies, being 5.2 and 5.7 GHz for an antenna fabricated on RO4003C substrate and reconfigured using PIN diodes. For the reconfigurable patch antenna [30], four PIN diodes are used to reconfigure the antenna from S-band to L-band. When the switches are off, the square patch at the center operates at 2.95 GHz, while when the switches are turned on, the patch combines with the outer ring and operates at the L-band frequency of 1.33 GHz. More details on these two antennas will be provided in Chapter 5. Frequency reconfiguration is also possible and has been studied for pixeled antennas (see Figure 1.6). The reconfigurable aperture (RECAP) antenna proposed in [13] was similar in geometry. The antenna designs proposed in [8, 9] also proposed pixel-type geometries. With the type of geometry shown in Figure 1.6 or similar theoretically, many possibilities for frequency reconfiguration exists as long as an antenna can be practically built and excited. The pixel geometry shown in Figure 1.6 contains 4 by 4 conducting pixels. Assume a microstrip patch type antenna where the pixels construct the patch surface on top of a substrate that is backed by a ground plane. As an example, by connecting pixels 6, 7, 10, and 11, the antenna can operate at a high frequency or mode 1 while by connecting all the pixels the antenna can operate at a low frequency or mode 2. A feed to excite the patch antenna must be designed and positioned properly. Conceptually, many more pixels can be added to extend the frequency reconfiguration in the low-frequency direction. Conversely, if the pixel sizes are reduced, the reconfiguration frequency can be moved in the high-frequency

8

Reconfigurable Antenna Design and Analysis

Figure 1.6  Pixel frequency reconfigurable antenna. (After: [27].)

direction. Pixel antennas have been proposed for both frequency and pattern reconfiguration and we will discuss them in detail in Chapters 5 and 6. 1.3.2  Pattern Reconfiguration

When the radiation pattern of an antenna is reconfigured, the antenna is called a pattern reconfigurable antenna. Similar to frequency reconfiguration, the geometrical arrangements, material properties, and connection schemes between the building blocks can be changed or adjusted to shape or redirect the pattern in desired directions. A simple, easy-to-understand pattern reconfigurable antenna is shown in Figure 1.7, which shows a driven dipole antenna accompanied by two parasitic dipoles each of which contains two switches. The spacing S < 0.25λ ensures enough coupling to the parasitics to allow induced currents in them but not too strong a coupling to present a challenge for the impedance matching for the driven dipole. If switches 1 and 2 are on while switches 3 and 4 are off, the parasitic on the left acts as a reflector for the driven dipole. The parasitic on the right acts as a director. Thus, the pattern is directed to the right. The antenna functions as a classical Yagi-Uda array consisting of one reflector and one director. When switches 3 and 4 are turned on while switches 1 and 2 are turned off, the situation reverses. Another pattern reconfigurable antenna concept called the electronically steerable parasitic array radiator (ESPAR) is illustrated in Figure 1.8 where a driven antenna element is accompanied by a number of parasitic elements. By controlling the distances between the driven and the parasitic elements and



Introduction to Reconfigurable Antennas

9

Figure 1.7  Pattern reconfigurable Yagi-Uda antenna. (After: [12].)

the feed-point impedances of the parasitic elements, the array beam can be steered in space. Harrington [41] described a reactively controlled directive array with one driven dipole and six parasitic dipoles. He demonstrated that the array beam could be steered over an azimuth angle of 60° by adding variable reactances at the bases of the parasitic dipoles. Instead of phased arrays where element-to-element separation of a half wavelength or more is required to reduce the mutual coupling between elements, much smaller element-to-element separation is achieved under the ESPAR idea in portable/wearable wireless devices and systems. Thus, much more tightly coupled, smaller form factor arrays could be developed where the elements actually benefit from the mutual coupling if optimum distances and feed-point terminations are found. The parasitic elements could be controlled using PIN diode switches, MEMS switches, or varactor diodes. Series-fed varactor diode-controlled microstrip patch arrays [42–44] are another class of pattern reconfigurable antenna. Although phased arrays or scanning antenna arrays are another class of antennas, they can also be called pattern reconfigurable antennas in a broad sense as their patterns can be steered in space with the help of electronic phase shifters. Finally, antenna design that leverages the MIMO communication technique [45, 46] are highly beneficial to achieve higher capacity or throughput in a multipath fading communication environment. Although MIMO antennas do not require explicit pattern reconfiguration if signals from multiple antennas are uncorrelated, that MIMO system offers significant performance improvement. Both scanning arrays and MIMO antennas will be discussed in later chapters.

10

Reconfigurable Antenna Design and Analysis

Figure 1.8  Illustrating parasitic beam steering arrays.

1.3.3  Polarization Reconfiguration

Finally, many applications can gain significant benefit if polarization reconfiguration for an antenna can be achieved. Polarization diversity is a desirable feature in wireless communication base stations. Signals often suffer polarization change while traveling in a multipath environment. A system capable of both vertical and horizontal polarization or both right-hand circular polarization (RHCP) and left-hand circular polarization (LHCP) could also provide a significant advantage when the polarization change of the signal of concern is expected. Polarization agility or reconfiguration can similarly be achieved by utilizing the same antenna geometry but by activating certain sections of the radiating structure. Polarization reconfigurable antenna examples available in the literature generally focus heavily on microstrip patch and slot antennas [11, 47–51] for their lower profile and easy integration with the transmitter and receiver system.

References [1] Bernhard, J. T., “Reconfigurable Antennas,” in Encyclopedia of RF and Microwave Engineering, New York: John Wiley & Sons, 2005. [2] Bernhard, J. T., Reconfigurable Antennas, San Rafael, CA: Morgan & Claypool Publishers, 2007.



Introduction to Reconfigurable Antennas

11

[3] Bernhard, J. T., “Reconfigurable Antennas,” in Antenna Engineering Handbook, 4th ed., J. Volakis, (ed.), New York: McGraw-Hill, 2007. [4] Peroulis, D., K. Sarabandi, and L. P. B. Katehi, “Design of Reconfigurable Slot Antennas,” IEEE Transactions on Antennas and Propagation, Vol. 53, 2005, pp. 645–654. [5] Christodoulou, C. G., et al., “Reconfigurable Antennas for Wireless and Space Applications,” Proc. IEEE, Vol. 100, 2012, pp. 2250–2261. [6] Haupt, R., and M. Lanagan, “Reconfigurable Antennas,” IEEE Antennas and Propagation Magazine, Vol. 55, No. 1, February 2013, pp. 49–61. [7] Yang, S., et al., “Frequency-Reconfigurable Antennas for Multiradio Wireless Platforms,” IEEE Microwave Magazine, February 2009, pp. 66–83. [8] Weedon, W. H., and W. J. Payne, “MEMS-Switched Reconfigurable Multi-Band Antenna: Design and Modeling,” Proceedings of the 1999 Antenna Applications Symposium, Monticello, IL, September 15–17, 1999. [9] Weedon, W. H., W. J. Payne, and G. M. Rebeiz, “MEMS-Switched Reconfigurable Antennas,” IEEE Antennas and Propagation Society Int. Symp. Dig., 2001, pp. 654–657. [10] Yang, F., and Y. Rahmat-Samii, “Patch Antenna with Switchable Slot (PASS): Dual Frequency Operation,” Microwave Opt. Technol. Lett., Vol. 31, No. 3, November 2001, pp. 165–168. [11] Yang, F., and Y. Rahmat-Samii, “A Reconfigurable Patch Antenna Using Switchable Slots for Circular Polarization Diversity,” IEEE Microwave Wireless Component Lett., Vol. 12, March 2002, pp. 96–98. [12] Baik, J. W., et al., “Switchable Printed Yagi-Uda Antenna with Pattern Reconfiguration,” ETRI Journal, Vol. 31, No. 3, June 2009, pp. 318–320. [13] Pringle, L. N., et al., “A Reconfigurable Aperture Antenna Based on Switched Links Between Electrically Small Metallic Patches,” IEEE Transactions on Antennas and Propagation, Vol. 52, June 2004, pp. 1434–1445. [14] Nikolaou, S., et al., “Pattern and Frequency Reconfigurable Annular Slot Antenna Using PIN Diodes,” IEEE Transactions on Antennas and Propagation, Vol. 54, February 2006, pp. 439–448. [15] Huff, G. H., and J. T. Bernhard, “Integration of Packaged RF MEMS Switches with Radiation Pattern Reconfigurable Square Spiral Microstrip Antennas,” IEEE Transactions on Antennas and Propagation, Vol. 54, February 2006, pp. 464–469. [16] Anagnostou, D. E., et al., “Design, Fabrication and Measurements of an RF-MEMS-Based Self-Similar Reconfigurable Antenna,” IEEE Transactions on Antennas and Propagation, Vol. 54, No. 2, Pt. 1, February 2006, pp. 422–432. [17] Erdil, E., et al., “Frequency Tunable Microstrip Patch Antenna Using RF MEMS Technology,” IEEE Transactions on Antennas and Propagation, Vol. 55, No. 4, April 2007, pp. 1193–1196. [18] Ali, M., A. T. M. Sayem, and V. K. Kunda, “A Reconfigurable Stacked Microstrip Patch Antenna for Satellite and Terrestrial Links,” IEEE Transactions on Vehicular Technology, Vol. 56, March 2007, pp. 426–435.

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Reconfigurable Antenna Design and Analysis

[19] Anagnostou, D. E., and A. A. Geethan, “A coplanar reconfigurable folded slot antenna without bias network for WLAN applications,” IEEE Antennas and Wireless Propagation Letters, Vol. 8, 2009, pp. 1057–1060. [20] Grau, A., et al., “A Dual-Linearly-Polarized MEMS-Reconfigurable Antenna for Narrowband MIMO Communication Systems,” IEEE Transactions on Antennas and Propagation, Vol. 58, No. 1, January 2010, pp. 4–17. [21] Cetiner, B. A., et al., “RF MEMS Integrated Frequency Reconfigurable Annular Slot Antenna,” IEEE Transactions on Antennas and Propagation, Vol. 58, No. 3, March 2010, pp. 626–632. [22] Besoli, A. G., and F. D. Flaviis, “A Multifunctional Reconfigurable Pixeled Antenna Using MEMS Technology on Printed Circuit Board,” IEEE Transactions on Antennas and Propagation, December 2011, pp. 4413–4424. [23] Islam, M. R., and M. Ali, “Switched Parasitic Body-Worn Array for High Data Rate Wireless Applications,” IEEE Antennas and Wireless Propagation Letters, Vol. 11, 2012, pp. 693–696. [24] Islam, M. R., and M. Ali, “A 900 MHz Beam Steering Parasitic Antenna Array for Body Wearable Wireless Applications,” IEEE Transactions on Antennas and Propagation, Vol. 61, No. 9, September 2013, pp. 4520–4527. [25] Anagnostou, D. E., et al., “Reconfigurable UWB Antenna with RF-MEMs for OnDemand WLAN Rejection,” IEEE Transactions on Antennas and Propagation, February 2014, pp. 602–609. [26] Rodrigo, D., et al., “MEMS-Reconfigurable Antenna Based on a Multi-Size Pixelled Geometry,” 2010 Proceedings of the Fourth European Conference on Antennas and Propagation (EuCAP), Barcelona, April 2010, pp. 1–4. [27] Rodrigo, D., and L. Jofre, “Frequency and Radiation Pattern Reconfigurability of a MultiSize Pixel Antenna,” IEEE Transactions on Antennas and Propagation, Vol. 60, No. 5, May 2012, pp. 2219–2225. [28] Rodrigo, D., B. A. Cetiner, and L. Jofre, “Frequency, Radiation Pattern and Polarization Reconfigurable Antenna Using a Parasitic Pixel Layer,” IEEE Transactions on Antennas and Propagation, June 2014, pp. 3422–3427. [29] Song, S., and R. Murch, “An Efficient Approach for Optimizing Frequency Reconfigurable Pixel Antennas Using Genetic Algorithms,” IEEE Transactions on Antennas and Propagation, February 2014, pp. 609–620. [30] Haider, N., A. G. Yarovoy, and A. G. Roderer, “L/S-Band Frequency Reconfigurable Multiscale Phased Array Antenna with Wide Angle Scanning,” IEEE Transactions on Antennas and Propagation, September 2017, pp. 4519–4528. [31] Panagamuwa, C. J., A. Chauraya, and Y. C. Varadaxglou, “Frequency and Beam Reconfigurable Antenna Using Photoconductive Switches,” IEEE Transactions on Antennas and Propagation, February 2006, pp. 449–454. [32] Chamok, N. H., et al., “High-Gain Pattern Reconfigurable MIMO Antenna Array for Wireless Handheld Terminals,” IEEE Transactions on Antennas and Propagation, October 2016, pp, 4306–4315.



Introduction to Reconfigurable Antennas

13

[33] Wright, M. D., et al., “MEMS Reconfigurable Broadband Patch Antenna for Conformal Applications,” IEEE Transactions on Antennas and Propagation, Vol. 66, No. 6, June 2018, pp. 2770–2778. [34] Mazlouman, S. J., et al., “A Reconfigurable Patch Antenna Using Liquid Metal Embedded in a Silicone Substrate,” IEEE Transactions on Antennas and Propagation, Vol. 59, No. 12, 2011, pp. 4406–4412. [35] Rodrigo, D., L. Jofre, and B. A. Cetiner, “Circular Beam-Steering Reconfigurable Antenna with Liquid Metal Parasitics,” IEEE Transactions on Antennas and Propagation, April 2012, pp. 1796–1802. [36] Alqurashi, K. Y., and J. R. Kelly, “Continuously Tunable Frequency Reconfigurable Liquid Metal Microstrip Patch Antenna,” IEEE Antennas and Propag. Symp. Int. Symp. Dig., 2017. [37] Kelley, M., et al., “Frequency Reconfigurable Patch Antenna Using Liquid Metal as Switching Mechanism,” Electron Lett., Vol. 49, October 2013, pp. 370–371. [38] Anagnostou, D. E., et al., “Ultra-Fast Reconfigurable Antennas with Phase Change Materials,” 2017 International Workshop on Antenna Technology: Small Antennas, Innovative Structures, and Applications (iWAT), Athens, Greece, 2017. [39] Ha, S. D., et al., “Electrical Switching Dynamics and Broadband Microwave Characteristics of VO2 RF Devices,” Journal of Applied Physics, Vol. 113, No. 18, June 2013. [40] Tawk, Y., et al., “Implementation of a Cognitive Radio Front-End Using Rotatable Controlled Reconfigurable Antennas,” IEEE Transactions on Antennas and Propagation, Vol. 59, No. 5, May 2011, pp. 1773–1778. [41] Harrington, R. F., “Reactively Controlled Directive Arrays,” IEEE Transactions on Antennas and Propagation, May 1978, pp. 390–395. [42] Cheng, S., et al., “Compact Reflective Microstrip Phase Shifter for Traveling Wave Antenna Applications,” IEEE Microwave and Wireless Comp. Lett., July 2006, pp. 431–433. [43] Öjefors, E., et al., “Electrically Steerable Single-Layer Microstrip Traveling Wave Antenna with Varactor Diode Based Phase Shifters,” IEEE Transactions on Antennas and Propagation, September 2007, pp. 2451–2460. [44] Cheng, S., “Integrated Antenna Solutions for Wireless Sensor and Millimeter-Wave Systems,” Ph.D. Dissertation, Uppsala University, Sweden, 2009. [45] Kildal, P. S., and K. Rosengren, “Correlation and Capacity of MIMO Systems and Mutual Coupling, Radiation Efficiency and Diversity Gain of Their Antennas: Simulations and Measurements in a Reverberation Chamber,” IEEE Communications Magazine, Vol. 42, No. 12, December 2004, pp. 104–112. [46] Karaboikis, M., et al., “Integrating Compact Printed Antennas onto Small Diversity/ MIMO Terminals,” IEEE Transactions on Antennas and Propagation, Vol. 56, No. 7, July 2008, pp. 2067–2078. [47] Qin, P. -Y., et al., “A Reconfigurable Antenna with Frequency and Polarization Agility,” IEEE Antennas and Wireless Propagation Letters, Vol. 10, December 2011, pp. 1373–1376.

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Reconfigurable Antenna Design and Analysis

[48] Jin, J., F. Yang, and Y. Rahmat-Samii, “A Novel Patch Antenna with Switchable Slots (PASS): Dual-Frequency Operation with Reversed Circular Polarization,” IEEE Transactions on Antennas and Propagation, March 2006, pp. 1031–1034. [49] Nguyen-Trong, N., L. Hall, and C. Fumeaux, “A Frequency- and PolarizationReconfigurable Stub-Loaded Microstrip Patch Antenna,” IEEE Transactions on Antennas and Propagation, November 2015, pp. 5235–5240. [50] Fries, M. K., M. Gräni, and R. Vahldieck, “A Reconfigurable Slot Antenna with Switchable Polarization,” IEEE Microwave and Wireless Components Letters, November 2003, pp. 490–492. [51] Ho, K. M. -J., and G. M. Rebeiz, “A 0.9–1.5 GHz Microstrip Antenna with Full Polarization Diversity and Frequency Agility,” IEEE Transactions on Antennas and Propagation, May 2014, pp. 2398–2406.

2 Fundamental Definitions of Antenna Parameters To understand the performance of an antenna or an antenna array, one requires an understanding of the fundamental antenna parameters. While geometrical and material characteristics can also be referred to as antenna parameters, the phrase “antenna parameters” is used here to refer to performance metrics or parameters that can be used to evaluate an antenna. Antenna textbooks [1–4] provide detailed definitions and explanations of antenna parameters and explain their significance from analytical, numerical, and experimental characterization points of view. Antenna parameter definitions in the literature, by and large, adopt the definitions provided by the IEEE Standard for Definitions of Terms for Antennas [5]. For the sake of brevity and clarity, only the most important antenna parameters that are directly relevant to the design and analysis of reconfigurable antennas will be addressed here. These include radiation pattern, input impedance, voltage standing-wave ratio (VSWR), return loss, bandwidth, gain, efficiency, polarization, and effective aperture. Antenna parameters are discussed in the context of actual practical experimental measurements, measurement challenges, and approaches to solve them and full-wave electromagnetic simulations. Note that different parameters may be needed to be measured or simulated depending on the antenna geometry, size, type, and applications. For example, for a mobile handheld MIMO antenna, one may be more interested in the envelope correlation coefficient (ECC), the diversity antenna gain (DAG), and the mean effective gain (MEG), while for a GPS antenna, one may be interested in RHCP pattern and gain. For narrowband resonant antennas, the VSWR or return loss of an antenna may be one of the most important parameters, while for an electrically large broadband antenna, 15

16

Reconfigurable Antenna Design and Analysis

one may be interested in pattern and gain consistency over a large frequency band spanning more than an octave.

2.1  Radiation Patterns The primary purpose of an antenna is to radiate electromagnetic waves so that a communication, navigation, or radar-type signal can be sent to a receiving antenna at a distance. In a reciprocal sense, for a receiving antenna, its goal is to receive or capture incoming electromagnetic waves from a distant transmitter. The distance in concern here is the far-field distance where the presence of an object will not have the ability to perturb the fields of the antenna. Although ample examples of antenna applications in the near field of the antenna exist (e.g., microwave hyperthermia [6], microwave ablation [7], and RF identification (RFID) antenna [8]), far-field usage and applications are more common. RFID tag coils operating in the near fields of RFID readers have been referred to as antennas, but they are not antennas in the conventional sense. In such cases, the energy transfer (e.g., charging and data transfer) occurs via magnetic coupling where the tag coil must be designed considering the near-field coupling. For all far-field cases of applications, such as mobile wireless communication, Wi-Fi, GPS, and radar, we need to define a parameter called antenna radiation pattern. Antenna radiation patterns are graphical representations of the magnitudes of the radiated far fields (the magnitude of the electric field) or the field intensity (electric field magnitudes squared) as a function of the polar angles, θ and φ. Thus, a pattern clearly shows how the said quantity (field intensity) varies as a function of the angular coordinates. Because radiation patterns are only relevant in the far field, the distance between the antenna and the observation point does not come into question, making the pattern independent of the distance. Patterns can be plotted containing normalized quantities (e.g., electric field, field intensity). Directivity, gain, or realized gain patterns, all of which will be defined later in this chapter, can also be plotted. Patterns are generally plotted using the logarithmic scale. Patterns can be plotted as rectangular plots, but more commonly two-dimensional (2-D) polar plots are used where one of the angles is kept fixed while the other is varied and the variation of the field intensity with that angle is plotted. An antenna pattern can contain one or both components of the radiated fields (e.g., vertical, horizontal, RHCP, LHCP, half-power beamwidth (HPBW), forward-to-backward ratio (F/B)), and the sidelobe-level (SLL) can be obtained from radiation patterns. Although a three-dimensional (3-D) pattern can be plotted that will represent the true and complete nature of the pattern, it rarely reveals some of the most important features of the pattern and is thus not widely used. Instead, 2-D polar plots reveal vital information about the pattern and are therefore



Fundamental Definitions of Antenna Parameters

17

plotted along various planes θ and φ. The angle φ is the azimuthal angle that varies from 0° to 360°. The angle θ is the elevation angle that varies from 0° to 180°. Figure 2.1 illustrates the patterns of a 10-element Dolph-Chebyshev linear antenna array consisting of isotropic elements oriented along the z-axis. Unlike a uniformly excited array, excitation amplitudes of this array are tapered according to a Dolph-Chebyshev amplitude taper [3] in order to achieve low SLL. The solid line and the dashed line represent the patterns when the element-toelement distances are 0.5λ and 0.25λ, respectively. As apparent, these patterns are directional with the main beam pointing broadside, that is, θ = ±90° direction. While a corresponding uniformly excited linear array of isotropic elements will yield –13-dB SLL, the SLL achieved by this particular example is –22 dB. SLL reduction comes at a cost, directivity decreases, and HPBW increases. A complete 3-D pattern can be constructed by taking many φ plane cuts where the angle θ varies from 0° to 180°. That is generally what is done in 3-D pattern measurement systems [9]. Simulation software, such as HFSS [10], FEKO [11], XFDTD [12], and CST [13] also have the same feature to represent both 3-D as well as 2-D pattern cuts. The radiation pattern of an antenna by itself may be less meaningful than when an antenna is considered for a specific application platform. If the application requires a certain pattern (e.g., a beam directed along a certain angular direction), the beamwidth must be within a specified limit, and SLL and back radiation must be below a certain level.

Figure 2.1  Normalized radiation patterns of a 10-element linear Dolph-Chebyshev array of isotropic elements; the amplitude taper reflects –22-dB SLL. The solid line indicates elementto-element spacing of 0.5λ, and the dashed line indicates element-to-element spacing of 0.25λ. (After: [3].)

18

Reconfigurable Antenna Design and Analysis

Consider a vertically oriented half-wavelength-long thin wire dipole antenna; its pattern is uniform in the azimuthal plane or the φ plane. In the elevation or θ plane, the pattern is of the shape of the number 8. That is why such a pattern is called a figure-8 pattern. The elevation pattern of the dipole is the same in any φ cut. The 3-D pattern of a half-wave dipole is in the shape of a torus. Such a pattern is called an omnidirectional pattern (i.e., directional in one plane (θ-plane) but nondirectional in another (φ-plane)). Omnidirectional pattern antennas are greatly desirable where signals may be expected to be received and sensed from any of the 360° azimuthal angle. In the past, before the advent of sectorized antenna design, wireless base stations typically utilized omnidirectional radiation patterns with narrow elevation plane beams to achieve the high gain needed in order to provide coverage over large distances. A classic example of such an array is a collinear array of vertical dipoles [14]. Because the axis of the array is along the vertical direction, assuming a standard separation distance between the elements (i.e., 0.75λ center-to-center distance or less), the E-plane beamwidth of the array will decrease as the array length increases. The H-plane or azimuthal plane beam will remain omnidirectional. For a microstrip patch antenna, operating in its fundamental resonance, the direction of maximum radiation or beam peak is normal or broadside to the patch plane. For the patch antenna, one can easily define an E-plane and an H-plane and plot the patterns in those two principal planes. More details on patch antennas are provided in Chapter 4. Considering a patch antenna fed using a microstrip line, the H-plane is the plane that is orthogonal to the feed line. Therefore, the E-plane is the plane that contains the feed line. If the patch is on the xy-plane (Figure 4.14(a)) with the feed line along the y-axis, the E-plane is the yz-plane or the φ = 90° plane. Therefore, the H-plane is the xz plane or the φ = 0° plane. Array antenna patterns can be defined as broadside, endfire, or scanned. Considering a linear antenna array consisting of isotropic elements a broadside pattern means that the beam peak is directed normal to the array axis. An endfire pattern means that the pattern peak is directed in the same direction as the array axis meaning it could be directed to either one of the two ends of the array axis. A log-periodic dipole array (LPDA) is an endfire array. So is a Yagi-Uda array. By contrast, a collinear dipole array (mentioned earlier), which may be used in a wireless phone base station, is a broadside array. The patterns of an antenna can also specifically indicate the polarization sense of that antenna. For example, an antenna that is predominantly vertically or horizontally polarized may only exhibit that polarization and minimal crosspolarization. A dual-polarized antenna should be capable of generating two orthogonal polarizations. Patterns of circularly polarized antennas must be characterized to evaluate their ability to radiate or receive either RHCP or LHCP



Fundamental Definitions of Antenna Parameters

19

waves. Typically, a directional transmitter antenna of the same polarization and sense is used in measurements (e.g., a conical equiangular spiral antenna or an axial mode helical antenna). With regard to reconfigurable antennas, the specific type of pattern in question will depend on which antenna is being reconfigured and for what parameter. It will also depend on what application the reconfigurable antenna is going to support.

2.2  Input Impedance, Voltage Standing-Wave Ratio, and Return Loss Antennas, whether used in transmit or receive or both, are generally connected to some form of transmission lines. There are exceptions, such as waveguide slot antennas or arrays. For example, an aperture-coupled patch antenna will be excited using an aperture, which, in turn, is excited using a microstrip feed line. A dipole antenna may be fed using a coaxial cable with an integrated balun. The input impedance of an antenna is defined at its input terminals. For instance, the input impedance of a half-wavelength-long dipole antenna made from very thin wire (thin with respect to the wavelength as well as the antenna length to the wire radius ratio being very high) has an input impedance of 73 + j42.5Ω [2]. Typically, the thin-wire dipole antenna is operated at a length slightly shorter than a half-wavelength, which ensures that the reactive impedance is zero and the antenna is in resonance. The dipole antenna size reduction compared to a half-wavelength depends on the length-to-diameter ratio (the Ω parameter where Ω = 2l/a with antenna length, 2l, and wire radius, a), for example, the thicker the conductor, the more the size reduction needed for resonance [2]. The impedance perspective of other antennas will be discussed in detail in later chapters. Note that not all antennas are of the resonant type. For example, dipoles, monopoles, microstrip patches, planar inverted-F antennas (PIFAs), normal mode helical antennas, meander dipole/monopole antennas, and zigzag dipole/monopole antennas are all resonant antennas. Array antennas made from these basic building blocks will also likely be resonant antenna arrays. By contrast, broadband antennas, such as the equiangular spiral, the LPDA, and the axial mode helix, are nonresonant antennas. The impedance of nonresonant antennas is generally purely resistive. From the transmission-line theory, if an antenna is considered as the load, then the reflection coefficient at the load (which is the antenna here) is defined as

ΓL =

ZL − Z0 ZL + Z0

(2.1)

20

Reconfigurable Antenna Design and Analysis

where ZL is the load or antenna impedance and Z0 is the characteristic impedance of the feed transmission line. The VSWR and the return loss are defined as



VSWR =

1 + ΓL 1 − ΓL



Return Loss = −20 log10 Γ L

(2.2)

(2.3)

Many times, modern-day simulation software and vector network analyzers (VNAs) instead plot the magnitude of the S11, which is synonymous to plotting the magnitude of the reflection coefficient as a 20 times 10 based logarithm and without the negative sign shown in (2.3). Thus, for the half-wave, the thin-wire dipole example above, if one considers Z0 = 50Ω, VSWR = 2.2, Return Loss = 6.8 dB, and S11 in dB, is –6.8 dB. A commonly used marker for a well-performing antenna is referred to as an S11 magnitude of –10 dB or less, which is nearly equivalent to a VSWR or 2. To be precise, a VSWR of 2 is an S11 magnitude of –9.57 dB. While the S11 magnitude or VSWR measurement is easy and straightforward, antenna input impedance measurement is not. Because antenna impedance also contains the phase information, one must use caution while measuring it. This is particularly important when antenna impedance matching must be performed. Impedance must be measured at the location where the matching circuit will be placed. If the VNA was calibrated at the end of the VNA cable and subsequently another cable or transmission line was added to it to feed the antenna, then the new additional cable will further move and spread the antenna impedance on the Smith chart. The best is to calibrate the VNA exactly at the point where the antenna will be connected, but that is not always possible because of many reasons (e.g., not having the proper connector or adapter that must be placed at the end of the additional cable). Impedance characterization for a multi-antenna system or array is different than that for a single antenna system. A multi-antenna system can be characterized using either the Z-parameters or the S-parameters. While the Zparameters are more meaningful from an antenna impedance point of view regarding the feeding transmission lines, the S-parameters are the ones that are normally used. For example, for a two-antenna array [3],

V1 = Z 11I 1 + Z 12 I 2

(2.4)



V 2 = Z 21I 1 + Z 22 I 2

(2.5)



Fundamental Definitions of Antenna Parameters

21

where Z11 is the self-impedance of antenna 1 in the absence of antenna 2 or antenna 2 open circuited, and Z12 is the mutual impedance between the two antennas when both are activated. The other Z-parameters (e.g., Z22 and Z21) have similar meanings. The driving point impedance of antenna 1 is defined as [3],

Z 1d = Z 11 + Z 12

I2 I1

(2.6)

In the event that both antennas are identical, the driving point impedance is the sum of the self-impedance and the mutual impedance. If one of the antennas has the same current magnitude but the opposite phase, as is the case for an image dipole, the driving point impedance is self-impedance minus the mutual impedance. Clearly, the driving point impedance could be significantly different from the self-impedance of the antenna in isolation, especially in the presence of strong mutual coupling. The driving point impedance formulation defined in (2.6) could be extended to accommodate many other antenna elements. Note that the VSWR and return loss are dependent parameters that depend on the antenna input impedance as well as the characteristic impedance of the transmission line feeding the antenna. The antenna impedance is a primary parameter that depends on the antenna itself (its geometry, construction, and materials). In many cases, an antenna impedance plot on the Smith chart reveals vitally important characteristics of the antenna. For example, for a broadband well-matched antenna, the entire impedance locus may lie near the center of the Smith chart. However, for a resonant narrowband antenna, the impedance on the Smith chart may illustrate the challenges that might be involved if one intends to enhance the return loss bandwidth of the antenna by adding passive LC matching components.

2.3  Bandwidth The term “bandwidth” pertaining to an antenna indicates that the antenna must meet certain performance requirements within a specified frequency range. This is a very important parameter that is likely governed by the application that the antenna must support. For example, for mobile cellular in the GSM (Global System for Mobile) frequency band, the antenna bandwidth is 880–960 MHz, and that for IEEE 802.11a s 2.4–2.485 GHz. Some common application frequency bands are listed in Table 2.1. Antenna performance characteristics that must meet within the bandwidth may include VSWR or return loss, pattern, gain, and polarization. Depending on the application, some antennas will have far more performance criteria that need to be satisfied within

22

Reconfigurable Antenna Design and Analysis Table 2.1 Some Communication Systems and Their Frequency Ranges Standard GSM GSM GSM UMTS Bluetooth GPS L1 GPS L2 RFID

Frequency Range (MHz) 880–960 1,710–1,880 1,850–1,990 1,920–2,170 2,400–2,484 1,575.42 1,227.60 13.553–13.567

the bandwidth compared to others. Also, for each performance criterion, the metric that governs that performance can be stringent depending on the application. For example, some applications may require only 2-dBi gain, while some others may require greater than 6-dBi gain. Typically, antennas are classified as narrowband or broadband. Other nomenclatures that have been used include wideband, multiband, and ultrawideband. For example, the GSM frequency band indicated above is a narrowband communication system. The bandwidth of a narrowband antenna can be expressed as a percent bandwidth [4]:

BW =

fH − fL ×100 fC

(2.7)

where BW is the bandwidth in percent, fL is the lowest frequency, fH is the highest frequency, and fc is the center frequency within the band. Communication antennas, such as cellular mobile phones, GPS, wireless local area networks (WLAN), and Industrial, Scientific, and Medical (ISM) applications, are narrowband. Note that the bandwidth of narrowband antennas can also simply be specified as the frequency range that they support. Examples of narrowband antenna elements include dipoles, monopoles, patches, slots, and PIFAs. The smaller electrical size of such antennas allows their easy placement and mounting within small handheld wireless devices as well as devices that correspond to IoT applications. Broadband antennas are generally defined in terms of a frequency ratio [4]:

BW =

fH fL

(2.8)



Fundamental Definitions of Antenna Parameters

23

For example, some applications such as ground penetrating radar (GPR) [15, 16] may have an antenna bandwidth requirement of 2:1 or greater. Antennas that can provide 2:1 or greater bandwidths exist. An antenna that can provide 2:1 bandwidth is often referred to as having a bandwidth of 1 octave. In general, large multiwavelength antennas such as equiangular spiral, conical equiangular spiral, Archimedean spiral, and LPDAs can offer broad bandwidths. Although the first target in satisfying the bandwidth requirement for an application is to check if the antenna can meet the VSWR or return loss bandwidth required by the application, other performance metrics are also equally important (e.g., pattern, gain, and polarization). An application may have a requirement for circular polarization within a bandwidth within which the axial ratio at boresight is less than 3 dB. There may be additional requirement for axial ratio within a certain beamwidth also. The challenge for an antenna designer is to consider and address the entire problem as a multivariable optimization problem. Figure 2.2 illustrates the reflection coefficient (dB) versus frequency response of a frequency reconfigurable antenna reported by Hum and Xiong [17]. The antenna frequency is reconfigured from around 2 GHz to 3.6 GHz if we consider reflection coefficient magnitude 1. When expressed in a 10-based logarithm, D will always be a positive quantity greater than zero. We can define the peak directivity of an antenna as [4]:

D=

U max U average

(2.9)

where Umax and Uaverage are the maximum and average radiation intensities. Once the far-field quantities are known (either analytically, through simulations, or measurements), the maximum radiation intensity can be calculated by squaring the maximum of the electric field magnitude. The average radiation intensity is calculated by integrating the square of the electric field magnitude



Fundamental Definitions of Antenna Parameters

25

within the entire theta and phi domains and then dividing that by the total spherical angle, which is 4π. For example, consider the magnitude of the far field of a vertically oriented (z-directed) electrically small (length shorter than one-tenth of the wavelength) dipole antenna E ∝ sin θ



(2.10)

In (2.10), other quantities for the field have been suppressed because they have no role in the calculation of D. As seen in (2.10) and is well known, the antenna fields are symmetric to the angle φ and the peak field magnitude occurs at θ = 90°. Thus, Umax = 1 while 2π π

∫ ∫ sin θ (sin θd θd φ) 2



U average =

0 0

4π

= 2/3

(2.11)

Note that sin(3θ) = 3sinθ – 4sin3θ. For pattern functions that cannot be easily analytically integrated, integration can be performed numerically. For example, one can use one of the built-in numerical integration routines (e.g., “quadl” in MATLAB to perform such integration). For simulated or measured quantities that come as raw data and not as a mathematical function, a simple numerical integration can be performed as described in [3]. From (2.10) and (2.11), the peak D of an electrically small dipole antenna is 1.5 or 1.76 dBi. As stated, all practical antennas will have a peak numeric D greater than 1. Many antennas will have large directivities, such as arrays, horns, and large reflector antennas. Gain is defined as a product of the directivity and antenna efficiency. Note that the IEEE Standard for Definitions of Terms for Antennas does not consider the effect of the reflection coefficient-related mismatch loss when defining gain. Some simulation software, when defining and using the term “gain,” adheres to the IEEE standard. Thus, gain can be expressed as

G = ηD

(2.12)

where η is the antenna efficiency. Efficiency is always 1 or less. Thus, the maximum antenna gain possible is G = D. Common causes of loss in efficiency will make η < 1 and thus G < D. As stated earlier, the decibel directivity of any practical antenna is always greater than 0 but the decibel gain can be negative quite easily because of poor antenna

26

Reconfigurable Antenna Design and Analysis

efficiency. For example, if an antenna has a directivity of 6 yet it is only 6% efficient, then G = 0.36 or –4.4 dBi. Many electrically small antennas at high frequency (HF) and very high frequency (VHF) for radios and aircraft applications can have negative antenna gain because of low antenna efficiency [20]. Low antenna efficiency could be caused by conductor loss or dielectric loss. Conductor loss could occur because of poor conductivity of the material, very small conductor thickness compared to the skin depth at that frequency, poor quality conductor (such as poor sintering of conductive ink) [21], surface roughness in the conductor, or conductor damage. Dielectric loss comes from the host dielectric material in any form, substrate, superstrate or radome, or carrier. While well-known microwave materials or substrates from Rogers Corporation, such as Duroid 5880 (εr = 2.2, tanδ = 0.0004), RO4003C (εr = 3.55, tanδ = 0.0027), and Duroid 6010 (εr = 10.2, tanδ = 0.0023), have very low loss tangents, many other materials have higher loss tangents (e.g., FR4 εr = 4.4, tanδ = 0.02). The loss tangents of some materials may not be even known when an antenna needs to be designed and fabricated in that environment. Materials are not always selected with the antenna design in mind when applications are considered and developed. Other causes for efficiency degradation may include losses due to surface waves, aperture tapers (for aperture antennas), or spillovers (for reflectors). When antennas are operated close to other objects, especially biological objects such as the human head or body, their efficiencies will further deteriorate. In such cases, the objects being in the near field of the antenna will likely absorb a significant amount of power that could not be used for communication. When designing a system consisting of antennas plus transmission lines, the issue of antenna matching with the feed transmission line must be considered. The IEEE definition of “gain” does not consider this, given that this is not a property of the antenna. One can then define a mismatch efficiency considering the reflection coefficient between the antenna and the feed line. The mismatch or reflection efficiency can be expressed as

2

ηRe fl = 1 − Γ L

(2.13)

Thus, we can define a new system-level gain or realized gain, as is used in practical cases

G Re alized = η η Re fl D

(2.14)

As seen in (2.14), the realized gain of an antenna can be further reduced because of lower reflection efficiency. Designers for mobile applications today measure or quantify the total antenna efficiency, which is a product of the two efficiencies in (2.14). Such an efficiency may be expressed in percentage or in



Fundamental Definitions of Antenna Parameters

27

decibels. For example, a 45% efficient antenna has an antenna efficiency of about –3.5 dB. Antenna realized gain can be measured using the gain comparison method [4]. In this method, a standard gain antenna, such as a horn, an LPDA, or a conical spiral (for circular polarization), is needed. The gain of the standard gain antenna must be known and must be accurate. A transmitter with an appropriate input power and transmit antenna is placed within an anechoic chamber. On the receive side, first, the standard gain antenna, followed by the test antenna, is placed, connected, and measured while the transmitter is transmitting. There are two measurements. First, the received power by the standard gain antenna, Ps, is measured followed by the power received by the test antenna, PDUT. Each receive antenna must be aligned properly to ensure that its beam peak is aligned to the beam peak of the transmit antenna. Their polarization should also be matched unless cross-polarized gain measurement is the objective. The gain of the test antenna can then be calculated as [4]

G DUT (dB ) = PDUT (dBm ) − PS (dBm ) + G S (dB )

(2.15)

where GS is the known gain of the standard gain antenna.

2.5  The Friis Transmission Formula The Friis transmission equation for a communication or radar system where a transmit (TX) antenna and a receive (RX) antenna are perfectly aligned both in terms of their radiation pattern and polarization (Figure 2.3) can be derived as follows. A transmit antenna with gain GT and transmit power PT radiates towards a receiver with receive antenna gain GR. The receiver is at a distance of d (m) from the transmitter. The effective isotropically radiated power (EIRP) is defined as

EIRP = PT GT ( w )

(2.16)

The radiated power density, PD, at the location of the receiver antenna is given by

Figure 2.3  Illustration of a transmit-receive antenna situation for the direct line of sight (LOS).

28

Reconfigurable Antenna Design and Analysis

PD =



PT GT (w m2 ) 4πd 2

(2.17)

The effective aperture, Ae, of the receiver antenna is defined as [4] Ae =



λ2G R (m 2 ) 4π

(2.18)

Therefore, the power received by the receiver antenna is given by

 λ  PR = PD Ae = PT GT G R   4 π d 

2

(w )

(2.19)

Note that the gain quantities in (2.19) are numeric, the powers are in watts, and the distance is in meters. For gain quantities that are given in dBi, a simple transformation to convert them to numeric values is required. For example, a gain of 15 dBi is simply 1015/10 = 31.6228. Similarly, if the power quantities are given in dBm (decibel over a milliwatt), then they must be converted to power in watts. For example, 33 dBm of power is 1033/10 milliwatts or 1.995 watts.

2.6  Polarization Antenna polarization is defined in terms of the direction of the electric field vector as the radiated electromagnetic wave travels away from the antenna as function of time. For a linearly polarized antenna, the electric field vector is aligned along a line that may or may not coincide with one of the coordinate axes. Similarly, the electric field vector may trace a circle or an ellipse as the radiated wave propagates away from the antenna as function of time. Polarization is a very important parameter that adds an additional constraint on the system performance. Especially for LOS cases, polarization alignment becomes key in order to minimize system loss or avoid total loss of service. However, for non-LOS mobile cellular communications, the nature of the polarization of the propagating wave changes because of reflection, diffraction, and scattering. Thus, for mobile cellular systems, multiple polarizations in the base stations and sometimes in the handheld phones are used to mitigate loss in performance. Polarization change or polarization reconfiguration can also be used as a feature to improve system performance or capacity. Examples of antennas with various polarizations are illustrated in Figure 2.4.



Fundamental Definitions of Antenna Parameters

29

Figure 2.4  (a) A vertically polarized dipole antenna, (b) a horizontally polarized dipole antenna, (c) a horizontally polarized slot antenna, and (d) a circularly polarized patch antenna excited using a quadrature hybrid.

Most commercial wireless applications, such as mobile telephone, Wi-Fi, and Bluetooth, utilize linear polarization. Given the non-LOS environment, polarization purity is rarely if ever sought after. However, polarization diversity has been found to be beneficial. Dual polarizations are also desired in many applications. For dipole-type antennas, dual polarizations can be easily achieved by orienting the two dipoles in orthogonal orientations. Circular polarization (CP) means that the antenna can generate two components of radiated fields that are in space quadrature, their magnitudes are equal and that the time phase difference between them is 90°. An example could be a crossed dipole antenna, where one dipole is excited using 0° phase while the other is excited using a 90° phase. The phasing can be achieved using an external circuit, such as a quadrature hybrid. Note that CP radiation can be either clockwise or right-hand or counterclockwise or left-hand. These terminologies mean that the electric field vector rotates in a right-hand or left-hand manner and traces a circle when propagating. For cases where either the field magnitudes or the phases do not exactly meet the definition for CP, the antenna is elliptically polarized. Except for a single frequency or vary narrow bandwidth, the CP antenna is actually an elliptically polarized antenna. Because the primary design goal is a CP antenna, the deviation from it is typically specified in terms of the axial ratio measured or computed in the direction of the main beam. To obtain a clearer picture throughout the pattern, spinning linear patterns can be measured (Figure 2.5)

30

Reconfigurable Antenna Design and Analysis

Figure 2.5  Measured spinning linear radiation patterns of a circularly polarized stacked microstrip patch antenna. (© 2013 V. K. Kunda. Reprinted, with permission from: [23].)

for a CP antenna to evaluate the extent of the CP over the entire beam. Also, spinning linear patterns can be taken over the entire bandwidth to see if the CP requirements are met. While narrowband CP antenna design has been proposed that does not require an external circuit for the phasing, most wideband CP antennas that are designed using elemental blocks like patches and dipoles require an external phasing circuit. Quadrifilar helical antennas (QHAs) [22] also require a CP phasing circuit. The four filars of a QHA are excited with phases 0°, 90°, 180°, and 270° phases, respectively. Note that the quadrifilar helix is a narrowband antenna unlike its broadband counterpart, the axial mode helical antenna. Nevertheless, the QHA does radiate a CP beam that is directed towards the tip of the filars. The other advantage of the QHA over CP patch antennas is that it does not require a large ground plane and hence is very conducive for portable satellite terminals. There are also CP antennas, which, by nature of their geometry, are broadband and CP at the same time. An example is the equiangular spiral or the Archimedean spiral antenna. However, such antennas have bidirectional beams that require the use of a reflector to direct the beam in one direction.

2.7  The Wheeler Cap Method of Antenna Efficiency Measurement Antenna efficiency is defined as PRAD/PINPUT where PRAD represents the radiated power and PINPUT represents the input power. Considering the IEEE definition of antenna gain, G, which does not include the losses in a transmission line and the mismatch loss between the antenna and the feed line, one can calculate antenna efficiency by comparing the measured efficiency and directiv-



Fundamental Definitions of Antenna Parameters

31

ity. For example, if D = 6 dBi while G = 4.5 dBi, then the antenna efficiency is –1.5 dB or 70.8%. This method of efficiency measurement requires one to use an anechoic chamber. Wheeler [24] proposed a simple method of efficiency measurement called the Wheeler cap method, which utilizes the measured input impedance of the antenna. Because antenna input impedance can be easily measured using a VNA, this is a simpler method compared to anechoic chamber-type methods. Wheeler suggested his proposed method to be used for electrically small antennas. However, Pozar and Kaufman [25] have used this method to measure the efficiencies of resonant microstrip patch antennas. In the Wheeler cap method, the input impedance of the antenna is measured first without any metal cap. Second, a suitable hemispherical metal cap (Figure 2.6) is used to entirely enclose the antenna and then its input impedance is measured the second time. For example, for printed antennas such as microstrip patches or PIFAs that are on ground planes, the metal cap is placed on the ground plane. The metal cap must make good electrical contact to the ground plane. Let us say that the real part of the measured antenna input impedance with and without the metal cap are RCAP and RNO CAP and then the efficiency can be calculated as [25]

e = 1−

RCAP R NO CAP

(2.20)

Figure 2.6  Illustrating the Wheeler cap on: (a) a monopole antenna and (b) a patch antenna. (After: [25].)

32

Reconfigurable Antenna Design and Analysis

Also note that e =

RRAD where RRAD and RLOSS are the radiation RRAD + RLOSS

and loss resistances, respectively. Pozar and Kaufman [25] stated that if RLOSS occurs in series, then RCAP < RNO CAP. Although a microstrip patch antenna is represented using a parallel resonant circuit as its equivalent circuit, Pozar and Kaufman [25] found that RCAP < RNO CAP even for patches. In that case, (2.20) could be used to calculate the efficiency. Although Wheeler stated that the cap diameter is about one-sixth of the wavelength, Pozar and Kaufman showed that a larger diameter metal caps can be used. For their measurements at 5 GHz, they used a 35-cm-diameter cooking wok. Newman et al. [26] showed that the shape of the metal cap was not a major concern and cubical-shaped metal cubes can be used. In [26], measured antenna efficiencies using the Wheeler cap method of a probe-fed patch, a microstrip line fed patch, and an aperture coupled patch were presented and compared with a radiometric method of efficiency measurement. All three patches were fabricated on 62-mil-thick εr = 2.2 substrate. The measured efficiencies of these three antennas using the Wheeler cap method were 96 ± 2%, 91 ± 2%, and 85 ± 2%. The efficiency of the aperture-coupled patch was lower because of radiation loss through the aperture. These values compared well with the radiometric method of measurement. Pozar and Kaufman [25] indicated that good electrical contact between the metal cap and the ground plane was essential and that the metal cap center needed to be aligned to the patch center as closely as possible.

2.8  Isolation Between Antennas in a System In some applications such as radar, the transmit and receive antennas are separate but could be located close to each other. Ensuring enough isolation between them is quite critical in order to achieve proper functionality. Improved isolation is synonymous to reduced antenna to antenna mutual coupling. Similarly, reduction in mutual coupling between the elements of an array is very important to ensure proper array operation. For MIMO antennas, high isolation between the elements is essential to achieve high MIMO performance, such as EDG and hence MIMO capacity. Mutual coupling between two antennas is generally quantified by their simulated or measured S21 parameter magnitudes when the antennas are wellmatched. The S21 magnitudes plotted as a function of frequency will show how isolated the two antennas are from one another. Thus, S21 magnitude close to 0 dB will indicate extremely poor isolation, while |S21| < –20 dB will indicate excellent isolation. Although the amount of isolation needed will depend on a



Fundamental Definitions of Antenna Parameters

33

particular application, in general, |S21| < –20 dB within the frequency band of interest is considered good for most applications. Mutual coupling between antennas depend on many factors including antenna geometries, their surrounding environment, ground plane, and materials. Given that achieving low mutual coupling between antennas is quite critical, significant efforts have been dedicated to it throughout the years. A careful observation or study is often required to evaluate the trade-offs between a proposed isolation improvement solution and antenna performance. For example, a certain design may yield very high isolation improvement within only a narrow frequency band but not be as effective at other frequencies. An isolation improvement solution may adversely affect the antenna return loss versus frequency characteristics or essentially detune the antenna. In that case, the antenna must be redesigned for it to satisfy the return loss performance or it must accept the poor return loss performance as a trade-off. Another isolation improvement solution may adversely affect the nature of the radiation pattern simply due to its physical appearance, size, and property. A practicing engineer needs to evaluate the potential benefits against its drawbacks. Isolation improvement solutions that are not at the circuit or board level, such as a circulator or isolator (normally used to isolate a receiver from a transmitter), may include the creation of an interruption for the current or fields near the antennas. For example, an absorber block, a metal reflector, an electromagnetic bandgap (EBG) structure [27–36], and a metamaterial structure can be strategically designed and placed between two antennas to improve the isolation between them. Antennas with a ground plane slot on the ground plane can be created to interrupt the current. For example, the mutual coupling for an 8-element mobile handset MIMO antenna array [37] can be seen from Figure 2.7. More details on this antenna geometry can be found in Chapter 10 in Figures 10.4 and 10.5. Due to symmetry, data for only four of the antennas are seen here. Snn represents the input matching quality of the antennas, and Smn represents their mutual coupling. As seen, for each frequency range of operation, the mutual coupling is below 10 dB.

2.9  MIMO Antenna Metrics The primary goal of MIMO antennas is to achieve higher communication capacity and better communication quality. Because multiple antennas with varying radiation patterns and/or polarization can be used, one key performance parameter is that the signals received by the antennas are uncorrelated as much as possible. While more detailed description on MIMO antenna evaluation strategy, especially for portable or wearable devices, will be provided in Chapter

34

Reconfigurable Antenna Design and Analysis

Figure 2.7  Simulated S-parameters of the proposed 8-antenna array. (© 2018 IEEE. Reprinted with permission, from: [37].)

10, several important terms are mentioned here. These include the correlation coefficient, the MEG, and the effective diversity gain (EDG). These parameters can be evaluated using the measured or simulated S-parameters of a multi-antenna system. They can also be measured or simulated using the radiated fields. The former is simpler but is applicable to very low-loss antennas. Once these quantities are evaluated, system-level performance characteristics within given channel conditions or outage probabilities can be calculated. For more details on MIMO antenna parameters, please see Chapter 10.

References [1] Kraus, J. D., and R. J. Marhefka, Antennas, 2nd ed., New York: McGraw-Hill, 1988. [2] Elliott, R. S., Antenna Theory and Design, Revised Edition, New York: IEEE Press, 2003. [3] Balanis, C. A., Antenna Theory Analysis and Design, 4th ed., New York: John Wiley & Sons, 2016. [4] Stutzman, W. L., and G. A. Thiele, Antenna Theory and Design, 2nd ed., New York: John Wiley & Sons, 2013. [5] IEEE, 145-2013 - IEEE Standard for Definitions of Terms for Antennas, 2013; IEEE Revision of IEEE Std 145-1993, 1993. [6] Casey, J. P., and R. Bansal, “The Near Field of an Insulated Dipole in a Dissipative Dielectric Medium (Short Paper),” IEEE Transactions on Microwave Theory and Techniques, Vol. 34, April 1986, pp. 459–463. [7] Mirotznik, M. S., N. Engheta, and K. R. Foster, “Heating Characteristics of Thin Helical Antennas with Conducting Cores in a Lossy Medium. I. Noninsulated Antennas,” IEEE Transactions on Microwave Theory and Techniques, Vol. 41, November 1993, pp. 1878–1886.



Fundamental Definitions of Antenna Parameters

35

[8] Finkenzeller, K., RFID Handbook, 3rd ed., New York: John Wiley & Sons, 2010. [9] Satimo, http://www.satimo.com/. [10] Ansys HFSS, https://www.ansys.com/products/electronics/ansys-hfss. [11] Altair Feko, https://altairhyperworks.com/product/FEKO. [12] Remcom XFDTD, https://www.remcom.com/xfdtd-3d-em-simulation-software. [13] CST Studio Suite, https://www.3ds.com/products-services/simulia/products/cst-studiosuite/. [14] Fujimoto, K., and J. R. James, Mobile Antenna Systems Handbook, 2nd ed., Norwood, MA: Artech House, 2001. [15] Utsi, E. C., Ground Penetrating Radar Theory and Practice, Boston, MA: Elsevier, 2017. [16] Lacko, P. R., et al., “Studies of Ground Penetrating Radar Antennas,” Proc. of the 2nd International Workshop on Advanced Ground Penetrating Radar, May 2003. [17] Hum, S. V., and H. Y. Xiong, “Analysis and Design of a Differentially-Fed Frequency Agile Microstrip Patch Antenna,” IEEE Transactions on Antennas and Propagation, Vol. 58, No. 10, October 2010, pp. 3122–3130. [18] Dyson, J., “The Equiangular Spiral Antenna,” IRE Trans. Antennas Propagat., Vol. AP 7, No. 2, April 1959, pp. 181–187. [19] D. M. Pozar, “A Microstrip Antenna Aperture Coupled to a Microstrip Line,” Electronics Letters, Vol. 21, January 17, 1985, pp. 49–50. [20] Lopez, D. G., M. Ignatenko, and D. S. Filipovic, “Low-Profile Tri-Band Inverted-F Antenna for Vehicular Applications in HF and VHF Bands,” IEEE Transactions on Antennas and Propagation, November 2015, pp. 4632–4639. [21] Tomaszewski, G., et al., “The Influence of Sintering Conditions on the Inkjet Printed Paths Resistance,” Intl. J. Electron. and Telecom., Vol. 62, No. 2, 2016, pp. 135–140. [22] Kilgus, C. C., “Resonant Quadrafilar Helix,” IEEE Transactions on Antennas and Propagation, Vol. 17, No. 3, 1969, pp. 349–351. [23] Kunda, V. K., “Study and Design of a Reconfigurable Stacked Microstrip Patch Antenna for Satellite and Terrestrial Applications,” M.S. Thesis, University of South Carolina, 2004. [24] Wheeler, H. A., “The Radiansphere Around a Small Antenna,” Proc. IRE, August 1959, pp. 1325–1331. [25] Pozar, D. M., and B. Kaufman, “Comparison of Three Methods for the Measurement of Printed Antenna Efficiency,” IEEE Transactions on Antennas and Propagation, Vol. 36, January 1988, pp. 136–139. [26] Newman, E. H., P. Bohley, and C. H. Walter, “Two Methods for the Measurement of Antenna Efficiency,” IEEE Transactions on Antennas and Propagation, Vol. AP-23, July 1975, pp. 457–461.

36

Reconfigurable Antenna Design and Analysis

[27] Lee, J. Y., S. H. Kim, and J. H. Jang, “Reduction of Mutual Coupling in Planar Multiple Antenna by Using 1-D EBG and SRR Structures,” IEEE Transactions on Antennas and Propagation, September 2015, pp. 4194–4198. [28] Naser-Moghadasi, M., et al., “Compact EBG Structures for Reduction of Mutual Coupling in Patch Antenna MIMO Arrays,” Progress in Electromagnetics Research C, Vol. 53, 2014, pp. 145–154. [29] Xie, H. -H., et al., “An Effective Analysis Method for EBG Reducing Patch Antenna Coupling,” Progress in Electromagnetics Research Letters, Vol. 21, 2011, pp. 187–193. [30] Islam, M. T., and M. S. Alam, “Compact EBG Structure for Alleviating Mutual Coupling Between Patch Antenna Array Elements,” Progress in Electromagnetics Research, Vol. 137, 2013, pp. 425–438. [31] Mohajer-Iravani, B., S. Shahparnia, and O. M. Ramahi, “Coupling Reduction in Enclosures and Cavities Using Electromagnetic Band Gap Structures,” IEEE Transactions on Electromagnetic Compatibility, Vol. 48, 2006, pp. 292–303. [32] Islam, M. T., and M. S. Alam, “Design of High Impedance Electromagnetic Surfaces for Mutual Coupling Reduction in Patch Antenna Array,” Materials, Vol. 6, 2013, pp. 143–155. [33] Rajo-Iglesias, E., Ó. Quevedo-Teruel, and L. Inclán-Sánchez, “Mutual Coupling Reduction in Patch Antenna Arrays by Using a Planar EBG Structure and a Multilayer Dielectric Substrate,” IEEE Transactions on Antennas and Propagation, Vol. 56, No. 6, June 2008, pp. 1648–1655. [34] Michailidis, E., C. Tsimenidis, and G. Chester, “Mutual Coupling Reduction in a Linear Two Element Patch Array and Its Effect on Theoretical MIMO Capacity,” Proc. Loughborough Antennas Propag. Conf., 2008, pp. 457–460. [35] Jiang, T., T. Jiao, and Y. Li, “Array Mutual Coupling Reduction Using L-Loading E-Shaped Electromagnetic Band Gap Structures,” Hindawi Publishing Corporation International Journal of Antennas and Propagation, Vol. 2016, Article ID 6731014, August 2016, https://doi.org/10.1155/2016/6731014. [36] Qiu, L., et al., “Transmit-Receive Isolation Improvement of Antenna Arrays by Using EBG Structures,” IEEE Antennas and Wireless Propagation Letters, Vol. 11, 2012, pp. 93–96. [37] Guo, J., et al., “Side-Edge Frame Printed Eight-Port Dual-Band Antenna Array for 5G Smartphone Applications,” IEEE Transactions on Antennas and Propagation, Vol. 66, No. 12, December 2018, pp. 7412–7417.

3 Overview of RF/Microwave Switches 3.1  Introduction RF switches perform key functionalities in communication and radar systems including transmit/receive (T/R) switching, phase shifting, and reconfigurable antenna design and development. In an ideal case, in the on state, the switch allows complete transmission while in the off state, it completely blocks the transmission of the signal. Thus, in the on state, an ideal switch would have 0-dB insertion loss, while in the off state, it would have perfect or infinite decibel isolation. In all practical applications, switches would have nonideal behavior that will make their properties different than those of an ideal switch. Apart from insertion loss and isolation, other properties of switches that would be of interest are power consumption, turn-on voltage, switching time, insertion loss and isolation bandwidth, DC biasing, size, cost, electrostatic discharge (ESD) sensitivity, power handling ability, linearity, and assembly or integration ease. Frequencies of applications will largely dictate and limit the use of discrete switches. For higher frequencies (10 GHz or higher), monolithically integrated switches [1–5] will become typical, while at lower frequencies, discrete switches may be used. The use cases of RF switches for phase shifting or T/R switching applications are fundamentally different from their use in reconfigurable antenna design. This difference arises because the antenna is a radiating structure, while T/R switching or phase shifting is a circuit operation. The reconfigurable

37

38

Reconfigurable Antenna Design and Analysis

antenna being a radiating element, the placement of switches, their properties, sizes, and biasing need to be carefully evaluated when designing such an antenna. It is probably reasonable to make the first design attempt considering ideal switches but soon thereafter all other aspects of the RF switch must be considered. There are myriad switch types that have been studied and reported in the literature with regard to reconfigurable antenna design. The most notable among them are PIN diode [6–12], discrete RF MEMS [13–17], varactor diode [18–20], photoconductive [21], and transistor switches [22]. Switches that are in the development stage include the liquid metal switch [23, 24] and the vanadium dioxide switch (VO2) [25]. A general comparison [26, 27] between three types of RF switches is provided in Table 3.1. As seen, some are currentcontrolled and others are voltage-controlled. Some have fast switching speed and others have high isolation. These and other characteristics would have to be taken into consideration when selecting them for reconfigurable antenna design and development. From a circuit point of view, a switch can be represented using its equivalent circuit parameters. In that case, the insertion loss and isolation can be calculated considering the equivalent circuit of the switch. While modeling and analyzing reconfigurable antennas, incorporating the switch models within the simulation model will yield more representative and accurate results. We will discuss antenna plus switch cosimulation in Chapter 9. The biasing aspects of RF switches will be discussed in Chapter 8.

3.2  The PIN Diode Switch A silicon PIN diode consists of an intrinsic (I) layer sandwiched between a P+ and an N+ layer (see Figure 3.1(a)). When such a diode is forward-biased, the charge carriers (holes and electrons) move into the I layer but they do not immediately recombine. Therefore, the term carrier lifetime, τ, comes into play. Also, the charge stored in the I region (Q) becomes an important parameter. In

Table 3.1 Comparison Between Basic RF Switches Bias Type Voltage PIN diode 5V MEMS 20–90V

Bias Current >10 mA Zero

Bandwidth Narrow Broad

Insertion Loss (dB) 0.5 0.2

Isolation (dB) 10–20 20–40

Switching Power Time ESD Consumption ns Low High High Low µs

Varactor diode

Zero

Narrow

0.5

10–20

ns

0–30V

Source: [20, 26, 27].

Low

Low



Overview of RF/Microwave Switches

39

Figure 3.1  (a) PIN diode schematic. Equivalent circuit of a PIN diode (b) on state and (c) off state. For simplified antenna simulation models, only C for the off state and R for the on state can be considered. (After: [28].)

the RF regime, in the forward bias region, a PIN diode can be modeled using a series resistor and a package inductance. The resistance is given by [28]

R=

W2 ( µn + µP ) × IF τ



(3.1)

where W is the width of the intrinsic region, IF is the forward bias current, µn is the electron mobility, µp is the hole mobility, and τ is the carrier lifetime. In the zero bias or reverse bias regime, the diode exhibits a capacitance that is given by [28]

C=

εA W

(3.2)

where ε is the permittivity of silicon given by the product of the free-space permittivity ε0 = 8.854 × 10–12 F/m and the dielectric constant (εr) of Si and A is the area of the diode junction. The PIN diode [28–31] is a low-cost, easily available switch that has been widely used in reconfigurable antenna design. For lower frequencies (sub-6 GHz or so), discrete PIN diode switches can be easily found in chip form and used on PC board-type environments. A PIN diode (Skyworks SMP 1345, SC79) switch that was used to develop a fabric-based wearable antenna reported in [29] was accompanied by two choke inductors, two DC blocking capacitors, and two resistors. The switch was implemented on thin liquid crystal polymer (LCP) film-type substrate and had an insertion loss between 0.1 to 0.4 dB from 0 to 6 GHz. Its isolation was near 40 dB at very low frequencies, which decreased to 16 dB at 2 GHz and about 8.5 dB at 5 GHz. This particular switch can be represented using a series resistance and a parasitic package inductance in the on state (see Figure 3.1(b)). The series resistance is around 2Ω for 10-mA DC bias current, while it increases to 3.5Ω at

40

Reconfigurable Antenna Design and Analysis

1 mA of bias current. While the series resistance of the PIN diode controls its insertion loss, the diode capacitance (see Figure 3.1(b)) in the off state controls its isolation when it is used as a switch. The off-state capacitance of this diode is 0.17 pF at zero DC bias. In general, if one considers a switch in the series configuration, its insertion loss is given by [28]

IL = 20log10 1 +

Zs 2Z0



(3.3)

where Z0 is the system or transmission line characteristic impedance and Zs is the impedance of the switch. The value of Zs is obtained from the equivalent circuit of the switch in its on state. The insertion loss of the switch is frequencydependent, unless the switch equivalent circuit only contains a resistor and no capacitance or inductance. Equation (3.3) can also be used to calculate the isolation of the switch. In that case, the value of Zs must be calculated from the off-state equivalent circuit of the switch. Considering a 50Ω reference impedance and using only a 3.5Ω resistance in (3.3) result in 0.3-dB insertion loss. For the off state, the use of a 0.17-pF capacitance in (3.3) results in an isolation of 11.9 dB at 2.45 GHz. Isolation deteriorates as the frequency increases. For applications requiring significantly higher isolation, PIN diode switches in series-parallel, Tee, or π configuration can be used. However, unlike a phase shifter design or a T/R switch design, the RF isolation required is governed by the antenna geometry and its reconfiguration needs. Thus, considering 50Ω reference impedance, even 6-dB isolation could be sufficient to design reconfigurable antennas. Especially for reconfigurable antenna design, the element-to-element separation distance also creates a capacitance that works in parallel with the switch capacitance or resistance. Thankfully, any EM simulation software will be able to handle that capacitance by its nature. A microstrip line test circuit of a PIN diode (HSMP 389B) switch is shown in Figure 3.2 along with its measured insertion loss and isolation data. The insertion loss is given by the diode on condition while the isolation is given by the diode off condition. The insertion loss here includes the line loss plus the switch loss. Isolation starts at 12 dB and then gradually decreases to 11 dB at 0.9 GHz. Diodes are nonlinear devices and thus at higher power could emit higher order harmonics and then radiate those through the antenna. For high-power applications, the nonlinear higher order harmonics from PIN diodes need to be carefully investigated.



Overview of RF/Microwave Switches

41

Figure 3.2  Measured insertion loss and isolation of an HSMP 389B PIN diode including the microstrip line. (© 2004 V. K. Kunda. Reprinted, with permission, from: [31].)

3.3  The RF MEMS Switch Microelectromechanical (MEMS) RF switches [32–35] are devices that enable their switching function through actual physical movement of a part or parts. The switches are very small and the moving part(s) are of micrometer dimension. The physical movement of the switch part is made possible with the help of a large enough electric field that applies a force to the moving part or membrane. The large electric field is obtained by applying a large enough DC voltage. The application of the DC bias voltage allows the moving part to either make a close contact or a capacitive connection. Simple schematics of resistive and capacitive RF MEMS switches can be seen in Figure 3.3. For the resistive switch, a DC voltage is applied between contacts #1 and #2, which pulls down the cantilever to make a physical and electrical connection with contact #3. Thus, in the on state, the switch is just a resistance. In the off state, the switch is a capacitance. The capacitive switch shown in Figure 3.3(b) will have its movable membrane pulled down when a DC voltage is applied between contacts #1 and #2. The variable gap between the movable membrane and contact #2 will allow variable capacitance. When

42

Reconfigurable Antenna Design and Analysis

Figure 3.3  Schematics of RF MEMS switches: (a) resistive switch, and (b) capacitive switch.

the membrane is too close to contact #2, the capacitance will be very high or its equivalent RF impedance will be very low, thus making it equivalent to the on state of the switch. For more details on RF MEMS switches, the reader may consult [32, 33]. RF MEMS switches have been proposed and developed for several decades in both discrete and monolithic configurations. An example of a discrete MEMS switch is the rmsw switch from Radant [14, 17]. The switch is activated by applying 90-V DC to the gate terminal, which makes the source and drain contacts to close and connect. With the DC voltage off, the switch is in the off state. The insertion loss of an RF MEMS switch may fall between 0.15 and 0.3 dB for a frequency range of DC to 12 GHz. It may have an isolation of 50 dB at DC that decreases to 17 dB at 6 GHz and then 10 dB at 12 GHz. Thus, such a switch is capable of operation over a broad frequency band. This type of a MEMS switch may be modeled approximately using a 2Ω resistor in the on state and a 40-fF capacitor in the off state. The isolation performances for that hypothetical MEMS switch and a typical PIN diode switch over a broad frequency range are compared in Figure 3.4. As seen, for a frequency of up to 3 GHz, the PIN diode provides 10 dB or more isolation, which falls to about 6 dB at 6 GHz. Isolation for the MEMS switch is greater than 15 dB at 6 GHz. Isolation of the MEMS switch is greater than 10 dB even at X-band (8–12 GHz). Another important difference between MEMS and the PIN diode switch is that while the former is a voltage-controlled device, the latter is a



Overview of RF/Microwave Switches

43

Figure 3.4  Calculated switch isolation for a single PIN diode or MEMS switch. A PIN diode capacitance of 0.17 pF and a MEMS capacitance of 40 fF were considered.

current-controlled device. MEMS switches typically require high voltage, they are more susceptible to ESD, and they typically have low yield.

3.4  The Varactor Diode Switch The varactor diode is a semiconductor device that is operated in the reverse bias regime. Its capacitance changes as a function of the reverse DC bias voltage. With the bias voltage being zero, the diode depletion region is short, which makes the varactor capacitance high; this then translates to a low RF impedance state. As the bias voltage increases, the width of the depletion region increases and hence the capacitance decreases. Because the bias voltage can assume any value between zero and a specific voltage value, many capacitance values can be obtained for the varactor within that bias range. A decrease in capacitance translates into high RF impedance. Unlike PIN diode switches, which has two states, on or off, a varactor switch may assume many intermediate states along with the on and off states. Thus, they find application in analog phase shifter design where continuous phase shifts can be achieved. They also find application in many types of reconfigurable antenna design. An example of frequency reconfiguration using varactor diodes can be seen from Figure 2.2. Examples of more varactor diode reconfigurable antennas can be found in Chapters 5, 6, 8 and 9. The varactor diode can be modeled as a capacitance with a resistance in series with it to represent either the on or off state. For example, the M/A com varactor diode MA46H202 has a capacitance of 0.5 pF at 20V and 8 pF at 0.5V. The measurement of insertion loss at 5 to 6 GHz [36] shows that the switch equivalent circuit should also contain a 3Ω resistor in series. The insertion loss and isolation of a varactor diode switch can also be calculated using (3.3) where the on state will be represented using an appropriate resistance and

44

Reconfigurable Antenna Design and Analysis

the capacitance in the low-voltage situation. The isolation can also be easily calculated using the off-state capacitance. Wherever possible, more than one switching device could be used to achieve optimum insertion loss and isolation for the frequency band concerned, provided that there are spaces to accommodate such switching configurations. The discrete chip inductors and capacitors shown in Figure 3.2 can be replaced with built-in capacitors and inductors or transmission-line sections at higher frequencies.

3.5  Other Types of RF/Microwave Switches Recently, VO2 has emerged as an interesting option for a switch because of its inherent phase change property [25, 37]. The material changes from being an insulator at a low temperature to a conductor at a higher temperature. The temperature increase in the material can be caused by physically raising the temperature of the material or by applying a DC bias current through it. Insertion loss of 3 dB from 0 to 13 GHz and isolation better than 20 dB were reported in [37]. However, as expected, insertion loss is a strong function of the bias current. Insertion loss is 5 dB at 40 mA of DC, which decreases to 3 dB at 80 mA of current. Field-effect transistors on GaAs and GaN technologies have been proposed as switches for a diverse array of applications including phase shifters and circulators [38–42]. In Chapter 7, we will provide an example of a GaAs singlepole double-throw (SPDT) switch that was used to design and build phase shifters for a 2-GHz microstrip patch phased array antenna. An example of a reconfigurable antenna that leverages photoconductive switch will be provided in Chapter 5. In recent years, significant interest has been placed on liquid metal-type antennas. Eutectic gallium, mercury, and gallinstan are liquid at room temperature. Using such materials, antennas have been designed by injecting such liquid within various channels [43, 44]. A review of liquid metal switches that have been proposed is available in [45]. However, there remains challenges and technical barriers in making such switches practical.

3.6  Switching Circuits and Their Responses The RF switching devices described before can be used as a single component to represent a switch or they can be arranged in a specific manner (e.g., series-shunt, pi, Tee-type networks) to meet the desired insertion loss and isolation objectives as long as the other constraints are also satisfied (bandwidth, power consumption, power handling ability, switch size, and footprint). Switch equivalent circuits can be used to perform circuit-based analyses to understand



Overview of RF/Microwave Switches

45

the insertion loss and isolation performances within a specific bandwidth. Although a circuit simulation based on the ABCD parameters of such a network can be easily done and their results converted to S-parameters [46], analytical presentation on simple circuits is valuable in terms of intuitive understanding. In the following, we will present the analytical circuit-based insertion loss and isolation calculation of several basic switching circuits. For a series switch with impedance Zs, its insertion loss and isolation can be calculated as follows. 3.6.1  Series Switch

Consider the circuit shown in Figure 3.5(a). Let us consider that the switch is in the on state. We will develop an expression for the insertion loss with reference to an ideal switch that does not have any loss. For an ideal switch (Zs = 0), the output voltage V01 is given by

V01 =

Vin 2



(3.4)

For a real practical switch, the output voltage V02 is given by

V02 =

Vin Z0 Zs,ON + 2Z0



(3.5)

The insertion loss (IL) is given by

IL = 10log10

V01 V02

2

= 20log10

Zs,ON + 2Z0 2Z0

= 20log10 1 +

Zs,ON 2Z0

(dB)

(3.6)

In (3.5) and (3.6), Zs is the frequency-dependent impedance of the switch in the on state obtained from the switch equivalent circuit.

Figure 3.5  Switch circuit configurations: (a) series and (b) shunt.

46

Reconfigurable Antenna Design and Analysis

Interestingly, the isolation of the switch can also be found from (3.6). To calculate isolation, the switch impedance Zs must represent the switch impedance in the off state. Thus, the isolation is given by

ISO = 10log10

V01 V02

2

= 20log10

Zs,OFF + 2Z0 2Z0

Zs,OFF

= 20log10 1 +

2Z0



(3.7)

3.6.2  Shunt Switch

For a shunt switch as shown in Figure 3.5(b), the switch on state indicates that the device is off, and the switch off state or isolation indicates that the device is in the on state. The voltages are

V01 =



V02 =

Vin 2



Vin Zs,OFF Z0 + 2Zs,OFF

(3.8)



(3.9)

The insertion loss (IL) is given by

IL = 10log10

V01 V02

2

= 20log10

Z0 + 2Zs,OFF 2Zs

= 20log10 1 +

Z0 2Zs,OFF



(3.10)

Z0 2Zs,ON



(3.11)

Therefore, the isolation is given by

ISO = 10log10

V01 V02

2

= 20log10

Z0 + 2Zs,ON 2Zs

= 20log10 1 +

3.6.3  Series-Shunt Switch

For the series-shunt switch configuration shown in Figure 3.6

IL = 20log10

(

Zp,OFF Zs,ON + 2Z0

)

Z0 Zp,OFF + Z + Z0 Zs,ON + Zp,OFF Zs,ON 2 0



(3.12)



Overview of RF/Microwave Switches

47

Figure 3.6  Series-shunt switch configuration.

where the series switch is in the on state and the shunt switch is in the off state.

ISO = 20log10

(

Zp,ON Zs,OFF + 2Z0

)

Z0 Zp,ON + Z02 + Z0 Zs,OFF + Zp,ON Zs,OFF



(3.13)

For switches consisting of multiple devices and organized in Tee, pi, or other configurations, determining analytical equations is not necessarily intuitive. In such cases, it is better to perform circuit simulations using ABCD parameters or use a simulation software such as the Advanced Design System (ADS) from Keysight to determine the insertion loss or isolation versus frequency characteristics.

3.7  Concluding Remarks The choice and use of RF switches in reconfigurable antenna (pattern, frequency, or polarization) design and development are a key element in the entire process. While understanding of the device physics is beneficial, it may not be necessary. However, understanding circuit-level performance and switch placement, assembly, and integration with the antenna building blocks is essential. This chapter attempted to provide an overview of RF switches as they pertain to reconfigurable antenna design and also to some extent phase shifter design. The interested reader may consult [32, 33, 46, 47] for more details.

References [1] Anagnostou, D. E., et al., “Design, Fabrication, and Measurement of an RF-MEMS Based Self-Similar Reconfigurable Antenna,” IEEE Transactions on Antennas and Propagation, February 2006, pp. 422–432. [2] Erdil, E., et al., “Frequency Tunable Microstip Patch Antenna Using RF MEMS Technology,” IEEE Transactions on Antennas and Propagation, April 2007, pp. 1193–1196.

48

Reconfigurable Antenna Design and Analysis

[3] Besoli, A. G., and F. D. Flaviis, “Multifunctional Reconfigurable Pixeled Antenna Using MEMS Technology on Printed Circuit Board,” IEEE Transactions on Antennas and Propagation, December 2011, pp. 4413–4424. [4] Cetiner, B. A., et al., “RF MEMS Integrated Frequency Reconfigurable Annular Slot Antenna,” IEEE Transactions on Antennas and Propagation, March 2010, pp. 626–632. [5] Anagnostou, D. E., et al., “Reconfigurable UWB Antenna with RF-MEMs for On Demand WLAN Rejection,” IEEE Transactions on Antennas and Propagation, Vol. 62, No. 6, February 2014, pp. 602–609. [6] Haider, N., A. G. Yarovoy, and A. G. Roderer, “L/S-Band Frequency Reconfigurable Multiscale Phased Array Antenna with Wide Angle Scanning,” IEEE Transactions on Antennas and Propagation, September 2017, pp. 4519–4528. [7] Hinsz, L., and B. D. Braaten, “A Frequency Reconfigurable Transmitter Antenna with Autonomous Switching Capabilities,” IEEE Transactions on Antennas and Propagation, July 2014, pp. 3809–3813. [8] Ali, M., A. T. M. Sayem, and V. K. Kunda, “A Reconfigurable Stacked Microstrip Patch Antenna for Satellite and Terrestrial Links,” IEEE Transactions on Vehicular Technology, March 2007, pp. 426–435. [9] Peroulis, D., K. Sarabandi, and L. P. B. Katehi, “Design of Reconfigurable Slot Antennas,” IEEE Transactions on Antennas and Propagation, Vol. 53, No. 2, February 2005, pp. 645–654. [10] Anagnostou, D. E., and A.A. Geethan, “A Coplanar Reconfigurable Folded Slot Antenna Without Bias Network for WLAN Applications,” IEEE Antennas and Wireless Propagation Letters, Vol. 8, 2009, pp. 1057–1060. [11] Islam, M. R., and M. Ali, “Switched Parasitic Dipole Antenna Array for High-DataRate Body-Worn Wireless Applications,” IEEE Antennas and Wireless Propagation Letters, Vol. 11, 2012, pp. 693–696. [12] Chamok, N. H., et al., “High-Gain Pattern Reconfigurable MIMO Antenna Array for Wireless Handheld Terminals,” IEEE Transactions on Antennas and Propagation, Vol. 64, No. 10, October 2016, pp. 4306–4315. [13] Weedon, W. H., et al., “MEMS-Switched Reconfigurable Multi-Band Antenna: Design and Modeling,” Proc. of the 1999 Antenna Applications Symp. Allerton Park, Monticello, IL, September 1999. [14] Huff, G. H., and J. T. Bernhard, “Integration of Packaged RF MEMS Switches with Radiation Pattern Reconfigurable Square Spiral Microstrip Antennas,” IEEE Transactions on Antennas and Propagation, Vol. 54, No. 2, February 2006, pp. 464–469. [15] Maciel, J. J., et al., “MEMS Electronically Steerable Antennas for Fire Control Radars,” IEEE Aerosp. Electron. Syst. Mag., November 2007, pp. 17–20. [16] Ho, K. M. -J., and G. M. Rebeiz, “A 0.9–1.5 GHz Microstrip Antenna with Full Polarization Diversity and Frequency Agility,” IEEE Transactions on Antennas and Propagation, May 2014, pp. 2398–2406.



Overview of RF/Microwave Switches

49

[17] Rajagopalan, H., J. M. Kovitz, and Y. Rahmat-Samii, “MEMS Reconfigurable Optimized E-Shaped Patch Antenna Design for Cognitive Radio,” IEEE Transactions on Antennas and Propagation, Vol. 62, No. 3, March 2014, pp. 1056–1064. [18] Behdad, N., and K. Sarabandi, “A Varactor-Tuned Dual-Band Slot Antenna,” IEEE Transactions on Antennas and Propagation, February 2006, pp. 401–408. [19] Luther, J. J., S. Ebadi, and X. Gong, “A Microstrip Patch Electronically Steerable Parasitic Array Radiator (ESPAR) Antenna with Reactance-Tuned Coupling and Maintained Resonance,” IEEE Transactions on Antennas and Propagation, April 2012, pp. 1803–1813. [20] Islam, M. R., and M. Ali, “A 900 MHz Beam Steering Parasitic Antenna Array for Body Wearable Wireless Applications,” IEEE Transactions on Antennas and Propagation, Vol. 61, No. 9, September 2013, pp. 4520–4527. [21] Panagamuwa, C. J., A. Chauraya, and Y. C. Varadaxglou, “Frequency and Beam Reconfigurable Antenna Using Photoconductive Switches,” IEEE Transactions on Antennas and Propagation, February 2006, pp. 449–454. [22] Pringle, L. N., et al., “A Reconfigurable Aperture Antenna Based on Switched Links Between Electrically Small Metallic Patches,” IEEE Transactions on Antennas and Propagation, Vol. 52, June 2004, pp. 1434–1445. [23] Alqurashi, K. Y., and J. R. Kelly, “Continuously Tunable Frequency Reconfigurable Liquid Metal Microstrip Patch Antenna,” IEEE Antennas Propag. Int. Symp., July 2017. [24] Kelley, M., et al., “Frequency Reconfigurable Patch Antenna Using Liquid Metal as Switching Mechanism,” Electronics Letters, Vol. 49, October 2013, pp. 370–371. [25] Anagnostou, D. E., et al., “Ultra-Fast Reconfigurable Antennas with Phase Change Materials,” 2017 International Workshop on Antenna Technology: Small Antennas, Innovative Structures, and Applications (iWAT), Athens, Greece, 2017. [26] Hindle, P., “The State of RF/Microwave Switch Devices,” Microwave Journal, Vol. 53, No, 11, November, 2010, p. 20. [27] Tawk, Y., et al., “Demonstration of a Cognitive Radio Front End Using an Optically Pumped Reconfigurable Antenna System (OPRAS),” IEEE Transactions on Antennas and Propagation, Vol. 60, No. 2, February 2012, pp. 1075–1083. [28] Application Note Design with PIN Diodes Skyworks Solutions Inc. http://www. skyworksinc.com/. [29] Czeresko, P., N. Chamok, and M. Ali, “Fabric Based Beam Steering Wearable Antenna Array,” 12th European Conference on Antennas and Propagation, London, UK, April 2018. [30] Doherty, B., PIN Diode Fundamentals, Micronote Series 701: https://www.microsemi. com/document-portal/doc_download/134814-micronote-701-pin-diode-fundamentals. [31] Kunda, V. K., “Study and Design of a Reconfigurable Stacked Microstrip Patch Antenna for Satellite and Terrestrial Applications,” M.S. Thesis, University of South Carolina, December 2004. [32] Rebeiz, G. M., RF MEMS, Theory, Design and Technology, New York: John Wiley & Sons, 2003.

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Reconfigurable Antenna Design and Analysis

[33] Bahl, I. J., and P. Bhartia, Microwave Solid State Circuit Design, 2nd ed., New York: John Wiley & Sons, 2003. [34] Iannacci, J., “Introduction to MEMS and RF-MEMS: From the Early Days of Microsystems to Modern RF-MEMS Passives,” MDPI Electronics, IOP Publishing Ltd., 2017, pp. 1-1–1-39. [35] Lysenko, I. E., et al., “Analytical Approach in the Development of RF MEMS Switches,” Electronics, Vol. 7, No. 12, 2018, p. 415. [36] Chamok, N. H., “High Gain Pattern Reconfigurable Antenna Arrays for Portable and Body-Centric Wireless Applications,” Ph.D. Dissertation, University of South Carolina, 2016. [37] Ha, S. D., et al., “Electrical Switching Dynamics and Broadband Microwave Characteristics of VO2 RF Devices,” Journal of Applied Physics, Vol. 113, No. 18, June 2013. [38] Ota, Y., et al., “High Isolation and Low Insertion Loss Switch IC Using GaAs MESFETs,” IEEE Transactions on Microwave Theory and Techniques, September 1995, pp. 2175–2177. [39] Marso, M., et al., “AlGaN/GaN Varactor Diode for Integration in HEMT Circuits,” Electronic Letters, Vol. 37, 2001, pp. 1476–1479. [40] Koudymov, A., et al., “Low-Loss High Power RF Switching Using Multifinger AlGaN/ GaN MOSHFETs,” IEEE Electron Dev. Lett., Vol. 23, August 2002, pp. 449–451. [41] Hara, S., T. Tokumitsu, and M. Aikawa, “Novel Unilaterial Circuits for MMIC Circulators,” IEEE Transactions on Microwave Theory and Techniques, October 1990, pp. 1399–1406. [42] Ross, T. N., et al., “Design of X-Band GaN Phase Shifters,” IEEE Transactions on Microwave Theory and Techniques, January 2015, pp. 244–254. [43] Dickey, M. D., et al., “Eutectic Gallium-Indium (EGaIn): A Liquid Metal Alloy for the Formation of Stable Structures in Microchannels at Room Temperature,” Adv. Funct. Mater., Vol. 18, 2008, pp. 1097–1104. [44] Hayes, G. J., et al., “Flexible Liquid Metal Alloy (EGaIn) Microstrip Patch Antenna,” IEEE Transactions on Antennas and Propagation, May 2012, pp. 2151–2156. [45] Sen, P., and C. Jin, “Microscale Liquid-Metal Switches—A Review,” IEEE Transactions on Industrial Electronics, April 2009, pp. 1314–1330. [46] Pozar, D. M., Microwave Engineering, 4th ed., New York: John Wiley & Sons, 2015. [47] Chang, K., I. J. Bahl, and V. Nair, RF and Microwave Circuit and Component Design for Wireless Systems, New York: John Wiley & Sons, 2000.

4 Basic Antenna Configurations Reconfigurable antenna behavior (e.g., frequency, pattern, and polarization reconfiguration) requires one or more antenna building blocks or elements. The function of reconfiguration typically requires the use of electronic, liquid metal, or even mechanical switches. Generally, only narrowband resonant-type antennas such as dipoles, patches, slots, and loops are used for reconfiguration. Broadband antennas such as the equiangular spiral, Archimedean spiral, log-periodic dipole antenna (LPDA), axial mode helix, and horn antennas are generally not candidates for reconfigurable operation because broadband operation and reconfiguration are contradictory. Broadband antennas are electrically larger structures that occupy the space of one or more wavelengths. Because return loss, pattern, gain, and polarization bandwidths of broadband antennas are large, reconfigurable operation is not needed. Reconfigurable operation is more desirable for applications that have space or volume restrictions, thus necessitating the use of narrowband resonant antennas. Such antennas, although not large, can provide multifrequency, multipattern, multipolarization characteristics through a reconfigurable operation. Along with resonant antennas, nonresonant antennas that are electrically small can also be used for a reconfigurable operation if they are impedance-matched and have the gain and pattern characteristics needed to support the application in concern. Although there are myriad antenna types that can be used to achieve a reconfigurable operation, we can only discuss a few of the very basic antenna types, which we will refer to as reconfigurable antenna elements from now on. These include dipoles, monopoles, microstrip patches, planar inverted-F antennas (PIFAs), inverted-F antennas (IFAs), slots, and loops. Within each of these categories, there may be many other types of variations that may adopt geometrical modification, dielectric loading, magnetic loading, and other techniques. 51

52

Reconfigurable Antenna Design and Analysis

An understanding of the geometry and performance characteristics of these basic antenna elements is essential to design and develop reconfigurable antennas. Although the reader can consult a good antenna textbook and study these basic antenna elements, it will be good to provide the fundamental background on these in a succinct manner in one place. In the following sections, we present the geometry, operation principles, and design guidelines for these basic antenna building blocks.

4.1  The Dipole Antenna The dipole antenna is probably one of the most widely studied antenna for its simplicity and wide applicability. In its simplest form, it consists of two pieces of thin conducting wires that have a gap in between them. The antenna is excited at the gap using a time-varying excitation as shown in Figure 4.1. The exact method of feeding and the choice in the use of a transmission line varies depending on the application scenario, the availability of materials, and construction feasibility. The dipole shown in Figure 4.1 is oriented vertically along the z-axis for simplicity of analysis. In practice, it may align or not align with any of the coordinate axes. Two dipoles can coexist near each other as long as their mutual coupling is acceptable or could be used for the benefit of the application. For example, crossed-dipole antennas that are orthogonal to each other and thus have the least amount of coupling are widely used in cases that require polarization diversity (e.g., mobile phone base stations) [1]. Parasitic dipoles (reactively controlled) can be brought very close to driven dipoles to achieve the desired pattern directionality and gain improvement [2]. Dipole antennas can be made using conducting wires with circular cylindrical, square, or other cross-sections. They can also be formed using sheets of metal with very small thickness. When fabricated in printed form, they will generally resemble thin sheets of metal on a substrate. When printed on a substrate,

Figure 4.1  The thin-wire dipole antenna.



Basic Antenna Configurations

53

they may or may not contain a ground plane as a reflector. The presence of a dielectric substrate and/or ground plane requires careful studies that are not easily achievable in simple analytical forms. Dipoles can also be made of fat, cylindrical, hollow conductors. Although dipole antennas can be excited at any point along their lengths, exciting them at their center is the most common. Because the dipole antenna (in free space) is generally operated near its first fundamental resonance (when the antenna length is near half of the wavelength), frequency reconfiguration can be achieved by increasing or decreasing the antenna length. This can be done by placing switches or control elements along the length of the dipole. For pattern reconfiguration, other parasitic dipole elements can be brought near the primary dipole and beams in the desired directions can be formed through parasitic coupling [2, 3]. Polarization reconfiguration will require the use of two orthogonal dipole elements that are switched on and off. The straight wire dipole shown in Figure 4.1 can be bent into various shapes such as meander and zigzag [4] to reduce the antenna resonant length, albeit at the expense of its growth in the width dimension. Dipole antennas, whether reconfigured or not, find applications in mobile wireless communications, aerospace, marine, and IoT applications among many others. Dipoles can also be used as building blocks to form arrays (e.g., Yagi-Uda, LPDA, collinear dipole array). Dipoles can also be used as a feed to the short backfire antenna [1] or corner reflector antenna [5]. As dipoles can be operated in free space, on a substrate, on a grounded substrate, and in a host of other circumstances, we will attempt to describe the most germane cases. Dipole antennas that are either printed on a dielectric substrate with or without ground, made of irregular-shaped conductors (meander, zigzag), or are made of conductors with larger wire diameters or are made of metal strips that are wide require numerical analysis using the method of moments (MoM), the finite element method (FEM), or the finite-difference timedomain (FDTD) method, among others. Thus, all but the very simple wire dipole antenna which is made of extremely thin-wire diameter cannot be analytically studied. However, even if such a wire dipole does not appear to be very practical, its analytical study reveals most of the fundamental characteristics of any dipole antennas. Thus, such a study is often insightful in understanding a whole host of other types of dipole antennas that are mentioned above. 4.1.1  The Thin-Wire Dipole Antenna 4.1.1.1  Current Distribution and Input Impedance

In this section, we will use the term “thin-wire dipole antenna” to refer to an antenna that is made from straight conducting wires and that has a total length to wire diameter ratio that is very high (>500).

54

Reconfigurable Antenna Design and Analysis

Dipole antennas that are extremely short in length (2l < λ/50) have a uniform constant current distribution that can be used to calculate their radiated and near fields. Such dipoles have a very small radiation resistance given by [7]:

Rr = 80 π2 (2l / λ) 2

(4.1)

and a very large capacitive reactance. To make them operational, the capacitive reactance must be tuned out with the help of an inductor. They also have a loss resistance given by [6]:

RLoss = ( 2lRS ) / (2 πa )

(4.2)

where a is the wire radius, Rs is the surface resistance defined asR s = ωµ0 , µ0 2σ = 4π × 10–7 (H/m) is the free-space permeability, σ is the conductivity of the material, ω = 2πf is the angular frequency, and f is the frequency in hertz. Dipoles that are short, but not as small as an extremely short dipole, have a triangular current distribution. Such dipoles are also not or near resonant and their impedance also contains a large capacitive reactance and a small resistance consisting of radiation and loss resistances given by [6]:

Rr = 20 π2 (2l / λ)

(4.3)

RLoss = ( 2lRS ) / (6 πa )

(4.4)

2

and

respectively. For the purpose of this book, our focus is on understanding the characteristics of dipole antennas that operate at or near resonance. The current distribution on such a thin-wire dipole is sinusoidal and can be expressed as [8]:

I (z ′ ) = I 0 sin {k (l − z ′ }

(4.5)

where l is the half-length of the dipole. Considering a dipole that has 2l = 0.5λ, since k = 2π/λ, when l = λ/4, the current has a peak at the feed point where z′ = 0. Note that the current vanishes to zero at either ends of the dipole where |z′| = λ/4. The current is zero at either end for any length of the dipole. However, when 2l = λ, from (4.5), the current is supposed to become zero at the feed point, which does not occur in reality because of a finite feed gap. For 2l = λ, instead of a zero current at the center, there is a very small current that manifests itself as a very high input impedance.



Basic Antenna Configurations

55

A thin-wire dipole that is much shorter or longer than 0.5λ is rarely used as a reconfigurable antenna because for such an antenna the operating impedance bandwidth is very narrow (only a few percent depending on the length-to-diameter ratio). This occurs because the antenna input impedance changes quite rapidly with its electrical length. Note that the thin-wire dipole antenna is, by its nature, a resonant antenna that attains resonance at various electrical lengths. The first resonance occurs when the total dipole length is in the vicinity of 0.5λ. Other resonances occur at longer lengths that can be determined from the exact impedance versus electrical length characteristics plots. However, as mentioned, for most practical purposes, dipoles are used at or near their first fundamental resonance, although one can consider using a dipole of any length and perform impedance matching if the application requires that. In such a case, attention must be paid on the overall bandwidth and efficiency of the antenna. Theoretically, a wire dipole of even an infinitesimal length (length < λ/50) can radiate. However, it will be a very inefficient antenna. The cause for the inefficiency stems from two factors: (1) the small radiation resistance, and (2) mismatch loss between the antenna and the feed transmission line. The latter occurs because the antenna impedance is predominantly reactive with its imaginary part containing a large capacitive reactance. The shorter the antenna, the smaller the input resistance (consists of radiation resistance and loss resistance) and the larger the capacitive reactance. The input impedance of a thin-wire z-directed dipole can be calculated using the induced EMF method [7, 8]:

Z in = −

1 I 0 sin 2 (kl )

l

∫ sin k {l − z ′ } E

z

( ρ = a , z = z ′ ) dz ′

(4.6)

−l

where Ez = − j





r=

ηI 0 4π

 e − jkR1 e − jkR2 e − jkr  + − 2 cos kl ( )   R2 r   R1

ρ2 + z 2 , R1 =

ρ2 + (z − l ) , R 2 = 2

(4.7)

ρ2 + (z + l ) 2

(4.8)

where a is the radius of the dipole conductor, 2l is its length, and η = µ0 / ε0 = 120 π Ω is the intrinsic impedance of free space. Because the integral in (4.6) does not have a closed-form solution a numerical integration routine can be used.

56

Reconfigurable Antenna Design and Analysis

The impedance versus antenna electrical length characteristics of a hypothetical dipole antenna (with l = 75 mm and a = 0.01 mm, a = 0.1 mm) are shown in Figure 4.2. These plots were generated by changing the frequency while keeping the antenna length fixed. As seen, the impedance (resistance or

Figure 4.2  Input (a) resistance and (b) reactance of dipole antennas. The solid line indicates the wire radius of 0.01 mm and the dashed line indicates the wire radius of 0.1 mm. The antenna is a half-wavelength at 2 GHz.



Basic Antenna Configurations

57

reactance) is highly sensitive to the antenna electrical length. The resistance is insensitive to the change in the wire radius, but the reactance is not. Naturally occurring resonance is not a necessary condition to design and build a dipole antenna. For many practical applications because of the design constraints, dipoles may need to be designed when their electrical lengths either are too short to achieve resonance or may be longer than what it is at resonance. In such cases, the antenna’s impedance may need to be matched unless a mismatch is acceptable and tolerable (for example, when the antenna is used as a measurement probe only and not as a communication device). Radiation patterns (described below) must be computed to examine whether they meet the required needs (e.g., the direction of the beam peak(s), directivity, beam shape, beamwidth). Radiation efficiency and mismatch efficiency also must be computed to obtain a complete picture before such an antenna is used in an application. 4.1.1.2  Radiation Pattern and Directivity

The radiated fields of a thin-wire dipole can be easily derived by using its current distribution defined in (4.5). As most antenna textbooks contain extensive derivations of the far fields of a dipole antenna using its current distribution, that topic will not be revisited here. According to [8], the radiated fields for a thin-wire z-directed dipole can be expressed as

E θ = j 60I 0

e j ( ωt −kr )  cos (kl cos θ ) − cos kl    r sin θ  

(4.9)

where the length of the dipole is 2l. For a half-wavelength-long dipole, (4.9) reduces to



 π  cos cos θ   e j ( ωt −kr )   2   E θ = j 60I 0 sin θ r    

(4.10)

In order to generate a pattern plot and compute the directivity, all we need is the quantity within the square bracket. The radiation patterns for antenna lengths of 2l = 0.5λ and 2l = λ have been plotted in Figure 4.3. As can be seen, each pattern has its beam maximum directed normal to the axis of the antenna. The pattern minimum occurs along both directions of the antenna axis as expected because the current vanishes to zero along those directions. If the 2-D pattern is spun around the z-axis, it will generate a 3-D pattern that will look

58

Reconfigurable Antenna Design and Analysis

Figure 4.3  The radiation patterns of a thin-wire dipole antenna; the solid line indicates 2l = 0.5λ, and the dashed line indicates 2l = λ.

like a torus. This is a very attractive pattern (omnidirectional, as described in Chapter 2). It can also be seen that with increasing antenna length, the patterns become more directive as evidenced by the narrowing of the beam widths in Figure 4.3. The half-power beamwidths are 78° and 48° for antenna lengths of 2l = 0.5λ and 2l = λ, respectively. Patterns of dipoles with length longer than 1.5λ have their beam peak shift directions from the θ = 90° direction and contain multiple lobes due to multiple maxima and minima in the current distribution. Note that the pattern of an electrically small dipole or infinitesimal dipole (2l < 0.1λ) is a sinθ pattern with half-power beamwidth of 90°. A summary of the HPBW and directivity of dipole antennas with various lengths is given in Table 4.1. Clearly there is not a substantial difference in

Table 4.1 Radiation Pattern and Directivity Summary of Dipole Antennas Dipole Length 0.05λ

Impedance Nonresonant

HPBW

0.5λ

Near resonance

1λ 1.25λ Source: [7].

90°

Directivity (Numeric) 1.5

Directivity (dB) 1.7

78°

1.65

2.1

Nonresonant

48°

2.4

3.8

Near resonance

40°

3.25

5.1



Basic Antenna Configurations

59

antenna directivity between antennas that are too short or have a length of a half-wavelength. Directivity increases as the length increases beyond that of a half-wavelength. However, the antenna impedance for such cases will contain high reactive components and the resistance will also be high except when the antenna length is in the vicinity of 1.5λ. For this length, the antenna is near its second resonance and the real part of the impedance may not be too high. 4.1.2  Feeding or Exciting a Dipole Antenna

Ideally, a dipole antenna needs to be fed or excited by a balanced transmission line such as a two-wire line. This allows that the currents in the line are balanced, which ensures radiation only from the antenna and not from the transmission line. However, feeding a dipole using a two-wire line is difficult because the characteristic impedance, Z0, of a two-wire line is much higher, typically close to 300Ω. Reducing Z0 requires the lines to come very close to each other, making them impractical for many applications. Also, the two-wire line may radiate itself at higher frequencies. The two-wire line is thus more restricted for use with a specific dipole called the folded dipole antenna, which has an input impedance near 300Ω at resonance. Feeding a dipole antenna other than a folded dipole using a coaxial line requires the use of a device called a balun. The word “balun” stands for balanced to unbalance here. The coaxial line or cable by its nature is an unbalanced transmission line. For example, consider the case illustrated in Figure 4.4(a) and assume that the balun or shorted quarter-wave metal tube or sleeve shown is absent. Without the balun, if one arm of the dipole is attached to the inner conductor of the coax and the other arm of the dipole is connected to the outer conductor of the coax an unbalanced situation is created. Note that the outer conductor is large in diameter and the dipole arm is all but connected at one point on the outer conductor. This causes an unbalance in the feeding arrangement and thus causes unwanted current flow on the outside surface of the outer conductor. For normal operation, there should never be a current flow on the

Figure 4.4  Feeding a dipole antenna using an unbalance coaxial line: (a) bazooka balun, and (b) split-coax balun.

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Reconfigurable Antenna Design and Analysis

outer surface of the outer conductor of a coaxial cable. Here the current flowing on the outer surface of the outer conductor of the cable will cause EM radiation from it. Thus, the resulting antenna pattern will be distorted. 4.1.2.1  Balanced to Unbalanced Transformers or Baluns (Coaxial)

To prevent current flow on the outer surface of the outer conductor of the coaxial cable, a balun is used. There are several types of baluns when fabricated in coaxial forms, for example, the bazooka (Figure 4.4(a)) and the split coax balun (Figure 4.4(b)), both of which do not change the resulting transmissionline impedance. However, there is a 4:1 coaxial balun [6] that can change the impedance to 4 times the impedance of the primary coaxial cable. Thus, if the characteristic impedance of the primary cable is 75Ω, then, at the balun end, the balanced line has an impedance of 300Ω. This type of balun is commonly used to excite a folded dipole antenna. The design, construction, and operation of a bazooka balun can be explained using the illustration shown in Figure 4.4(a). A short, hollow, cylindrical, metal tube of the length of one-quarter of the wavelength at the operating frequency is placed on a metal disk as shown. The metal tube and the disk are soldered together. Alternatively, they can be machined to be made to be a single piece. The disk contains a small hole through which the primary coaxial cable is inserted. The outer conductor of the primary coaxial cable is soldered to the disk. Thus, the disk, the metal tube, and the outer conductor or the primary coaxial cable are all connected at the bottom side. On the top side, the metal tube is open. The metal tube and the outer conductor of the coaxial cable form a transmission line, which is shorted at the bottom. However, because the length of the line is 0.25λ, it presents itself as an open circuit near the antenna feed point, which prevents current flow down the line on the outer surface of the outer conductor of the cable. For the bazooka balun, the two arms of the dipole are connected as shown in Figure 4.4(a). One arm of the dipole is connected to the inner conductor of the cable and the other arm is connected to the outer conductor of the cable. Because the metal tube is 0.25λ long at only one frequency, this balun is only suitable for a narrowband operation. Typically, the balun works well for dipole bandwidth less than 25%. The balun shown in Figure 4.4(b) is called a split coax balun and is probably the easiest to fabricate. Here a short piece of the same coax is used to create an external piece a quarter-wavelength long at the operating frequency, which does not contain the inner conductor or the insulation. Alternatively, a short metal tubing can be used. The outer conductor of the primary coaxial cable and the external metal tubing are placed parallel to each other and are shorted at the bottom as shown. The inner conductor on the dipole side is then bent and soldered to this external piece. One arm of the dipole is connected to the inner conductor and the external piece and the other arm is connected to the



Basic Antenna Configurations

61

outer conductor of the coaxial cable. Here also the shorted quarter-wave section of the metal tubing and the outer conductor of the coax form a separate transmission-line section that prevents current flow below down the cable on the outer surface of the cable, especially below the balun. EM radiation from the cable is thus prevented in either case. This balun can also be used to support operations with bandwidth less than 25%. The Marchand balun introduced by Nathan Marchand is a type of very wideband balun [9] that can provide a much more broadband operation. However, it is not a very simple structure as compared to the other baluns mentioned. Printed versions of the Marchand balun also exist [10–13]. 4.1.2.2  Printed Baluns in Microstrip Forms

There are many types of printed microstrip type baluns that have been proposed in the literature. A microstrip ring hybrid [14] is a very common example to use wherever feasible. The two output ports divide the power equally and with 180° phase difference. The amplitude balance, phase balance, return loss, and insertion loss are properties that an antenna designer must keep in mind when using such baluns. Another example of a printed microstrip type balun that was used to feed a wideband dipole antenna [15] on an electromagnetic bandgap (EBG) structure can be seen in Figure 4.5. This balun shown in Figure 4.5 consists of a 3-dB Wilkinson power divider and a noncoupled-line broadband 180° phase shifter. Figure 4.6(a) displays the simulated and measured return loss and isolation characteristics of

Figure 4.5  Photograph of wideband planar balun. (© 2008 IEEE. Reprinted, with permission, from: [15].)

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Reconfigurable Antenna Design and Analysis

Figure 4.6  (a) Return loss and (b) isolation data of the printed wideband balun. (c) Simulated and measured phase difference between the two balanced ports. (© 2008 IEEE. Reprinted, with permission, from: [15].)

the balun. The measured return losses of the unbalanced port as well as the balanced ports are better than 15 dB from 1.6–2.5 GHz. The simulated and measured amplitude imbalances between the two balanced ports are within 0.5 dB. As apparent from Figure 4.6(c), the phase difference between the two balanced ports of the balun is close to 180°. The measured phase balance is within ±5° of the simulation.



Basic Antenna Configurations

63

Figure 4.6  (continued)

4.1.2.3  Lumped-Element Baluns

Like many RF components, baluns can also be designed and fabricated using lumped-element components (e.g., inductors and capacitors) [16]. An example of a lumped-element balun can be seen in Figure 4.7. The circuit schematic in Figure 4.7(a) shows the inductors and the capacitors as they will be connected on the unbalanced side and the balanced side. The unbalanced side is indicated using number 1 and ground and the balanced side is indicated using number 2. Figure 4.7(b) shows the connection schematic from coaxial to a balanced transmission line. As seen, the two balanced outputs from the balun are connected to the two dipole arms. Values of the lumped element components can be calculated using [16, 17]:

L=

C =



Z IN Z OUT ω



(4.11)

1 ω Z IN Z OUT



(4.12)

where ω = 2πf, ZIN is the characteristic impedance of the balanced line, and ZOUT is the characteristic impedance of the unbalanced line. For example, for a 2.5-GHz operation, the balun may consist of two capacitors (1.27 pF each) and

64

Reconfigurable Antenna Design and Analysis

Figure 4.7  (a) Lumped-element balun circuit, and (b) coaxial to dipole connection schematic using a lumped-element balun.

two inductors (3.2 nH each), as shown in the circuit diagram in Figure 4.7(b) for a 50Ω system. Chip inductors and capacitors can be used to develop such baluns. Care must be taken selecting the inductors and capacitors to ensure that their selfresonant frequencies are much higher than the frequency for which they are being used [18, 19]. Baluns made from lumped-element components can add additional losses due to resistive dissipation in those components. Moreover, such baluns will be narrowband compared to coaxial-type baluns described above. Nevertheless, applications that cannot allow relatively large coaxial-type baluns may find the use of a lumped-element balun to be very attractive. 4.1.2.4  Chip Baluns

Another commonly used balun is the commercially made chip balun [20, 21]. Such devices have very small form factors and are surface-mountable. For example, a surface-mount chip balun is RFXF9503 from minicircuits. The use of chip baluns is warranted where coaxial baluns cannot be accommodated and the loss from such baluns can be tolerated. 4.1.3  Dipole Operation Against a Metal Reflector

Many applications require a dipole antenna to operate against a metal reflector in order to generate a unidirectional beam. For example, the short backfire antenna consisting of crossed dipoles needs to operate against a disk-like reflector for satellite applications [1]. Dipole antennas used in base station applications need to be placed against metal reflectors (flat, corner, or other shape) to suitably direct the beams into a sector or zone. Printed dipole arrays placed on conformal surfaces is another application example. Aside from array analysis,



Basic Antenna Configurations

65

it is quite challenging to find analytical representations or solutions for dipoles that are printed on grounded dielectric substrates. However, it is fairly straightforward to analyze dipoles that may be operating against a reflector but do not contain any dielectric material in their vicinity. Such antennas can be studied and analyzed with the help of image theory. Consider a dipole antenna located at distance h above a perfectly conducting large ground plane shown in Figure 4.8(a). From an application point of view, such a configuration can be created when the dipole (made from either a cylindrical conductor or a metal strip) can be supported by a block of foam, for example. Let us assume that the current distribution on the antenna is sinusoidal as before. The radiated fields for such an antenna can be calculated using the image theory. Applying the image theory, the perfectly conducting large ground plane can be replaced and an image dipole on the opposite side of the ground plane at a distance h can be created (Figure 4.8(b)). The direction of current on the image dipole must be opposite to that of the actual dipole to satisfy the boundary condition. Thus, intuitively, one can clearly see that when h is very small, the two dipoles with opposite directed currents will have their fields destructively interfere with each other creating a very inefficient construct. Only when h is reasonably large (larger than 0.1λ but not too large) is the configuration expected to function well. More detailed analysis of the radiated fields and input impedance are warranted. Note that the radiated fields are now due to the superposition of the fields from the primary dipole and the image dipole. Thus, an array factor effect will be present, and it will depend on the h. The radiated fields of a dipole antenna against a large perfectly conducting metal reflector is given by [8]:

 cos (kl cos θ ) − cos kl  Eθ ∝   sin(kh sin θ sin φ) sin θ  

(4.13)

Figure 4.8  (a) A horizontal dipole antenna against a large conducting ground, (b) application of image theory.

66

Reconfigurable Antenna Design and Analysis

Patterns for h = 0.1λ, 0.25λ, and 0.45λ are plotted in Figure 4.9. As long as the dipole to reflector distance, h, is 0.25λ or less, the radiation pattern is directional and does not contain multiple lobes. The single beam directional pattern persists beyond h = 0.25λ, but at a certain value beyond 0.25λ, the pattern starts to deform. For example, when h = 0.45λ, the pattern clearly shows a –5-dB dip in the upward direction while showing two peaks at ±40° from the upward direction. The input impedance (Z1d) of the dipole against the metal reflector can also be calculated using the image theory. With reference to Figure 4.8(b), from (2.6):

Z 1d = Z 11 + Z 12

I2 I1

(4.14)

As the two currents have equal magnitude but opposite phase,

Z 1d = Z 11 − Z 12

(4.15)

It has been found that input resistance is very small for h 0V, patch edges α are connected to the ground using the shorting posts because the PIN diodes on the alpha edges are on. In this state, the frequency of the y-oriented mode is shifted

Figure 5.15  Schematics of the proposed reconfigurable antenna. (© 2011 IEEE. Reprinted, with permission, from: [9].)



Frequency and Polarization Reconfiguration

109

higher above the x-oriented mode. Considering the x-oriented mode as State I, since V2 > 0V, the varactors in diode group B are now used to tune the frequency of the x-oriented mode. The antenna generates xpolarization (Co-pol.), with y-polarization being the cross-polarization (Cross-pol.). • State II: Frequency tuning is achieved by diode groups B and using the y-oriented mode by applying V1 > 0 and V2 < 0V. In this mode, patch edges β are shorted to the ground. The antenna generates y-polarization (Co-pol.), with x-polarization being the cross-polarization (Cross-pol.). • State III: The 45° linear polarization mode is achieved by applying V1 > 0 and V2 > 0V. The operation of these states are summarized in Table 5.2. Measured and simulated S11 versus frequency plots for State I are shown in Figure 5.16, which illustrates the frequency reconfiguration cases for the xpolarized state. Similar frequency reconfiguration for State II is reported in [9]. The frequency reconfiguration states for State III are slightly different, but for the sake of brevity, those results are not shown here. Results can be seen in [9]. Qin et al [9]. reported simulated and measured E and H-plane radiation patterns for all three states at 1.7 GHz. Patterns look directional like that of a patch antenna. Cross-polarization is below about 18 dB in the E-plane and below about 15 dB in the H-plane. Measured and simulated gains of the antenna are shown in Figure 5.17. Measured antenna gain varies as –2.6 to 6.4 dBi from a frequency of 1.4 to 2.25 GHz. Measured gain is above 0 dBi at the 1.7-GHz reconfiguration frequency. If a gain of 3 dBi or higher is needed, then the antenna has a frequency range of 1.9 to 2.25 GHz or essentially three reconfiguration states (see S11 versus frequency plot). The authors attribute the lower gain at the low frequencies to the reduced antenna size at the low frequencies. They also attribute the lower gain at low frequencies on the high varactor capacitance, which allows increased I2R losses.

Table 5.2 Different Polarization States of the Reconfigurable Antenna PIN Diodes in Group A State I Forward-biased State II Reverse-biased State III Reverse-biased

PIN Diodes in Group B Reverse-biased Forward-biased Reverse-biased

(© 2011 IEEE. Reprinted, with permission. From: [9].)

Polarization x-oriented y-oriented 45°-oriented

Active Varactor Diodes Group B Group A Groups A and B

110

Reconfigurable Antenna Design and Analysis

Figure 5.16  Measured and simulated input reflection coefficients for different bias voltages (varactor diode junction capacitances) for State I. The first and second sets of lines in the inset show the parasitic resonance of the antenna for the bias voltages of 2.2V (1.0 pF) and 3.1V (0.75 pF), respectively. (© 2011 IEEE. Reprinted, with permission, from: [9].)

Figure 5.17  Measured and simulated gains of the antenna. (© 2011 IEEE. Reprinted, with permission, from: [9].)

5.2.3  Frequency and Circular Polarization Reconfigurable Patch Antenna

A frequency and polarization reconfigurable patch antenna with switchable slots (PASS) was introduced by Jin et al. [10]. The antenna consists of a rectangular patch with length, L, and width, W, as shown in Figure 5.18. The patch length is along the x-direction and its width is along the y-direction as



Frequency and Polarization Reconfiguration

111

Figure 5.18  Configuration of the proposed patch antenna with a switchable slot. A switching diode is mounted at (xs; ys) to control the slot configuration. A diagonal coaxial feed located at (xf; yf) excites circular polarizations. (© 2006 IEEE. Reprinted, with permission, from: [10].)

shown. The patch also contains a thin rectangular slot with its length along the x-direction. If the patch is fed along the diagonal, it will excite two orthogonal modes, TMz01 with currents flowing along the y-direction, and the TMz10 with the currents flowing in the x-direction. Notice that the slot also contains an electronic switch, such as a diode. For a patch with nearly a square dimension and containing no slot, where L is only slightly smaller or larger than W, feeding the patch diagonally can generate circular polarization (CP) [11]. The presence of the slot and its control using a switch allow a change in the resonant frequency of the patch where the patch can radiate RHCP for one case and LHCP for another case. The operating frequency can be explained as follows: • TMz01 mode: With the switch on, the patch resonant frequency is nearly the same for the patch without any slot because the current can flow through the switch and the effective length of the patch is still W. With the switch off, the current flow path increases because of the full slot present. Thus, the patch has a resonant frequency smaller than before. • TMz10 Mode: The resonant frequency or operation of this mode is not appreciably affected by the switch on or off status as the slot length is in the same direction (x) as the patch current flow direction.

112

Reconfigurable Antenna Design and Analysis

As stated above, basic CP patch antenna theory shows that, in the absence of any slot or switch if L and W are chosen judiciously, one can generate CP radiation when the antenna is excited along the diagonal. For the reconfigurable antenna proposed in [10], the antenna has the ability to change its frequency and at each frequency the polarization is the opposite of the other (i.e., RHCP in one versus LHCP in the other). For example, with the switch on the slot off, the antenna will assume a low frequency of operation and generate RHCP for the case proposed in [10]. Conversely, with the switch on the slot on, the antenna will assume a high frequency of operation and generate LHCP. To switch between RHCP and LHCP, a scheme shown in Figure 5.19 was proposed. The input RF signal is controlled using a gateway switch that activates one of the two patch antenna feeds. In one case, the antenna can achieve RHCP, and for the other case the antenna can achieve LHCP. This approach allows polarization change in between the same frequency, but without this approach the different polarization states are attained at different frequencies. An antenna was fabricated and tested on a 3.18-mm-thick RT/Duroid 5880 (εr = 2.2) substrate. A photograph of the fabricated antenna can be seen in Figure 5.20. The antenna parameters are: L = 20.5 mm, W = 18 mm, Ws = 1 mm, h = 3.18 mm, and Lg = 60 mm. The location of the probe feed was at (xf, yf) = (3.7 mm, 3.1 mm) and the location of the slot was at (xs, ys) = (0, –6 mm). A Schottky diode (MSS4015) was used as a switch to allow for high-speed switching. However, PIN diode or MEMS switches can also be used.

Figure 5.19  An improved design is suggested to combine frequency and polarization diversities using another coaxial feed located at (–xf; yf) and an RF switch between the two feeds. This antenna has switchable RHCP and LHCP at each operation frequency. (© 2006 IEEE. Reprinted, with permission, from: [10].)



Frequency and Polarization Reconfiguration

113

Figure 5.20  Photograph of a fabricated antenna prototype in Figure 5.18. A Schottky diode is utilized to implement the switch. The slot is cut across the patch, with two 47-pF capacitors soldered at the ends to isolate DC bias while keeping the RF connection. The length of the bias line is λ/4 so that the antenna return loss will not be appreciably affected. (© 2006 IEEE. Reprinted, with permission, from: [10].)

The measured broadside axial ratio versus frequency characteristics for the two polarization states (LHCP and RHCP) are shown in Figure 5.21. The minimum axial ratios measured were around 2 dB at 4.2 and 4.55 GHz with 3-dB axial ratio (AR) bandwidths of 60 and 80 MHz, respectively, at these frequencies. The measured return loss results can be seen from Figure 5.22. The return loss is –10.2 dB at 4.2 GHz and –6.8 dB at 4.55 GHz. The measured spinning linear radiation patterns are shown in Figure 5.23. Patterns generally show good CP characteristics. Some asymmetry in the patterns seen was attributed to the biasing circuit.

5.3  Reconfigurable Slot Antennas 5.3.1  UHF-Band PIN Diode Reconfigurable Slot Antenna

A frequency reconfigurable slot antenna proposed by Peroulis et al. [12] is shown in Figure 5.24. The antenna is a microstrip-fed slot that achieves frequency reconfiguration from 540 MHz to 890 MHz with the help of PIN diode switches. As illustrated, the slot contains three sections. First, it starts with a section in the y-direction, which is then followed by a second section in the x-direction and a third section again in the y-direction. The slot and the

114

Reconfigurable Antenna Design and Analysis

Figure 5.21  Measured results of broadside axial ratio versus frequency. The best broadside axial ratios are observed at 4.55 GHz (LHCP) and 4.20 GHz (RHCP). The associated AR < 3 dB bandwidths are 80 MHz and 60 MHz, respectively. (© 2006 IEEE. Reprinted, with permission, from: [10].)

Figure 5.22  Measured antenna return loss at each operation status. The black triplets show the frequencies where the best broadside axial ratios are observed. (© 2006 IEEE. Reprinted, with permission, from: [10].)

microstrip feed line (2.4 mm wide) were fabricated on a 100-mil-thick Duroid 6010 (εr = 10.2) substrate. Multiple PIN diode switches were integrated into the slot in the shunt configuration to achieve frequency reconfiguration. The



Frequency and Polarization Reconfiguration

115

Figure 5.23  Measured linear spinning patterns in the xz-plane. (a) Switch on, 4.55 GHz, LHCP; and (b) switch off, 4.20 GHz, RHCP. Acceptable AR less than 3 dB is achieved within a 40° beamwidth. The asymmetries in the patterns are attributed to the biasing circuit. (© 2006 IEEE. Reprinted, with permission, from: [10].)

switches are labeled as SW1, SW2, SW3, and SW4, respectively. Using the proposed scheme, multiple reconfigured frequency bands with each band sup-

116

Reconfigurable Antenna Design and Analysis

Figure 5.24  Figure 5.23 is a reworking of Figure 9(a) from [12]. Parameters: l1 = 14.9 mm, l2 = 19 mm, l3 = 3.6 mm, l4 = 31 mm, l5 = 33 mm, l6 = 35.2 mm, and l7 = 11 mm, slot width = 2 mm. (After: [12].)

porting certain bandwidths were achieved. The total reconfigurable bandwidth achieved has a frequency ratio of 1.7:1 in the ultrahigh frequency (UHF) band. As the diodes are implemented in the shunt configuration, their off states correspond to low insertion loss or the RF equivalent of on. Conversely, when the diodes are in their on states, that corresponds to the RF equivalent of off or isolation. Simulation results demonstrated the role of the diode resistance on antenna efficiency. For example, with a single switch, antenna efficiencies were 71.8%, 55.6%, 45.6%, and 33.9% for diode resistances of 0Ω, 1.4Ω, 2.8Ω, and 5.6Ω, respectively. To bias each switch, a 470-nH inductor and 10-pF capacitors were used. Simulated RF performance of a switch showed about –15 to –11-dB insertion loss in the diode on state from 500 MHz to 1 GHz. In the switch off state, the RF insertion loss was negligible. To ensure proper operation, the off states of the switches were implemented by applying –20-V DC to them. In terms of operation, for example, when only SW4 is on, the antenna operates at



Frequency and Polarization Reconfiguration

117

537 MHz, and when SW1 and SW4 are on, the antenna operates at 603 MHz. Antenna frequency reconfiguration from slightly above 500 MHz to nearly 900 MHz was shown with good bidirectional radiation patterns. The measured peak gain at 593 MHz was –1.1 dBi, which corresponded to an efficiency of 47%. Measured slot reconfiguration frequency states as function of switch bias states are illustrated in Table 5.3. The antenna achieves reconfiguration at four discrete frequencies, 537, 603, 684, and 887 MHz. Measured polarization angles at those respective frequencies were 57°, 70°, 55°, and 33°, respectively. 5.3.2  Reconfigurable Folded Slot Antenna

A frequency reconfigurable folded slot antenna was introduced by Anagnostou and Geethan [13] for WLAN applications. Examples of nonreconfigurable folded slot antennas can be found in earlier literature [14–18]. The geometry and switch positions for the antenna can be seen from Figure 1.4. Their proposed folded slot antenna was designed and fabricated using a 0.81-mm-thick RO 4003C substrate (εr = 3.55, tanδ = 0.0027). The substrate and ground plane size were 40 mm by 30 mm. The on resistance or forward bias resistance for the PIN diode was around 1Ω with 15-V DC bias. The perimeter of the folded slot is only about slightly smaller (7% smaller) than the guided wavelength on the substrate [14]. The resonant length of the folded slot can be calculated as Lg =

where

C0 =

C λ0

C0 εavg



(5.1)

with C being the circumference and λ0 is the free-space wave-

length. The parameter εavg is given by

Table 5.3 Measured Resonant Frequencies of the Slot as Function of Switch States Switch SW1 Bias (V) SW2 Bias (V) SW3 Bias (V) SW4 Bias (V) Active slot length

Slot Resonant Frequency 537 MHz 603 MHz –20 (Off) 1.1 (On) –20 (Off) –20 (Off) –20 (Off) –20 (Off) 1.1 (On) 1.1 (On)

684 MHz 0 (On) 1.1 (On) –20 (Off) 1.1 (On)

l1 + l2 + l3 + l4 + l5 + l6

l3 + l4 + l5 + l6 l5 + l6

l2 + l3 + l4 + l5 + l6

887 MHz 0 (On) 1.1 (On) 1.1 (On) 0.2 (On)

Source: [12]. *Switch off means RF on (adjacent slot sections connected) versus switch on means RF off (adjacent slot sections disconnected).

118



Reconfigurable Antenna Design and Analysis εavg =

εr + 1 2



(5.2)

The dimensions of the folded slot antenna were adjusted between initial design and final design. These dimensions are given in Table 5.4. As seen in Figure 1.4, the folded slot contains two PIN diodes at the two edges of the slot. Simulated and measured return loss and gain results are shown in Figures 5.25 and 5.26, respectively. With the diodes on, the effective length of the folded slot is reduced, and the antenna operates at the 5.5-GHz WLAN frequency band. Conversely, when the diodes are turned off, the effective length of the folded slot becomes longer and the antenna operating frequency shifts to 5.14 GHz. Measured return loss data show fairly wideband (greater than 15%) performance for each reconfigured frequency band within 10-dB return loss. The measured gain at 5.25 GHz is 5.2 dB and 5.775 GHz is 5.6 dB. Gain is between 4.5 and 5.2 dB in the 5.2-GHz band and between 4.5 and 5.5 dB in the 5.775-GHz WLAN band.

5.4  MEMS Reconfigurable Antennas 5.4.1  Monolithic MEMS Reconfigurable Patch Antennas 5.4.1.1  L/X-Band Ideal MEMS Frequency Reconfigurable Patch Antenna

Weedon et al. [19] reported simulation studies of MEMS frequency reconfigurable antennas. Their work reported in [20] showed a 3 by 3 square patch reconfigured using MEMS switches as shown in Figure 5.27. Due to the lack of availability of MEMS switches at that time, the antenna configuration was simulated considering an ideal open to represent the off state of a switch and an ideal short to represent the on state of a switch. The experimental prototype of the antenna was also built and tested that included the physical short and the physical open to represent the switch on and off states. The antenna designed using 125-mil-thick Duroid 5880 (εr = 2.2, tanδ = 0.0009) showed frequency Table 5.4 (a) Initial Dimensions of the Folded Slot Antenna. (b) Final Dimensions of the Folded Slot Antenna.

© 2009 IEEE. Reprinted, with permission. From: [13].



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119

Figure 5.25  Measured and simulated return loss response of the reconfigurable folded slot antenna. The measured results are very close to simulated ones. The effect of the DC bias cables is negligible. (© 2009 IEEE. Reprinted, with permission, from: [13].)

Figure 5.26  Measured and simulated maximum gain of the reconfigurable folded slot antenna. The gain response does not change with the reconfigurability, and it exhibits similar values in both bands. (© 2009 IEEE. Reprinted, with permission, from: [13].)

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Reconfigurable Antenna Design and Analysis

Figure 5.27  The MEMS reconfigurable L/X-band patch antenna of [20] on Duroid 5880 substrate. Dimensions in millimeters. (After: [20].)

reconfiguration from L-band to X-band. For the L-band operation, all 3 by 3 patches were activated and for the X-band operation all but the center patch was deactivated. In the L-band, 1.2% impedance bandwidth was achieved versus 7% for the X-band. 5.4.1.2  Monolithic MEMS Reconfigurable Self-Similar Antenna

A MEMS reconfigurable self-similar antenna was introduced in [21] where cantilever MEMS switches were fabricated monolithically with the antenna being fabricated on the same high-resistivity silicon substrate. To turn the MEMS switch on, 40-V DC actuation voltage was used. The measured insertion loss for the switch was 0.2 dB and isolation was around 18 dB at 15 GHz. Notably, measured switch insertion loss was less than 0.4 dB for up to a frequency of 40 GHz. The measured isolation was 15 dB at 27 GHz and 12 dB at 40 GHz. The detrimental effects of conductive DC bias lines on antenna return loss and current distribution were demonstrated. To alleviate this problem, highresistance DC bias lines were fabricated using Al-deposited zinc oxide (AZO), which provided sheet resistance of 10 kΩ/sq. Total measured DC resistance was greater than 200 kΩ. The geometry of the proposed reconfigurable antenna is shown in Figure 5.28. A schematic of the MEMS switch is also shown in the same figure. Measured results from [21] show that, in the switch off state, the antenna operates at 15 GHz. Some differences in the operating frequency range were observed depending on whether the DC probes were up or down. When the switches were turned on, the antenna exhibited two resonances, one at 9.2 GHz and the other at 25.2 GHz. The bandwidth for the first band was 1.6 GHz and the bandwidth for the second band extended from 24.3 to 28.2 GHz. Measured radiation patterns at 8.3, 15, and 25 GHz all show good performance. 5.4.1.3  Monolithic MEMS Reconfigurable Patch Antenna

A MEMS reconfigurable monolithically fabricated patch antenna was introduced in [22]. The antenna structure was fabricated on 0.5-mm-thick Pyrex 7740 glass substrate (εr = 4.6, tanδ = 0.005). The antenna geometry shown



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121

Figure 5.28  Illustrating the RF MEMS reconfigurable self-similar bow-tie antenna from [21]. (After: [21].)

in Figure 5.29 consists of a patch antenna loaded with five MEMS capacitors. The MEMS capacitors can be controlled by applying a variable DC bias voltage, which was applied using a bias tee. Changing the bias voltage as 0V, 11V, and 11.9V results in the antenna operating frequency reconfiguring from 16.05 GHz to 15.75 GHz. The height of the MEMS capacitors decreases from 1.5 µm to 1.4 µm along with an operation around 16 GHz; another resonance around 13 GHz was also observed, which also shifted as a function of the MEMS bias voltage change. Measured radiation patterns showed directional characteristics with both switch-up and switch-down positions. It appears there are two main disadvantages to the proposed approach: (1) the MEMS capacitors and the CPW structure are external to the antenna and thus require significant space, and (2) the range of the frequency reconfiguration is quite limited. 5.4.1.4  Monolithic MEMS Reconfigurable Frequency and Pattern Reconfigurable Patch Antenna

Besoli and Flaviis [23] introduced a reconfigurable pixeled patch antenna for operation in the 4–7-GHz frequency band. The antenna design was studied, fabricated, and tested using monolithic MEMS switches. The antenna consisted of 9 by 9 metallic pixels printed on a 1.5875-mm-thick RO-TMM3

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Reconfigurable Antenna Design and Analysis

Figure 5.29  Illustrating the monolithic MEMS reconfigurable patch antenna of [22]. (After: [22].)

substrate (50.8 mm by 50.8 mm). Each pixel was a square with edge length of 1.43 mm. Each pixel contained a via at its center to provide DC bias to the MEMS switches. The details of the pixel geometry, MEMS, and their biasing can be seen from Figure 3 of [23]. Each via was connected to a Ni-Chrome line to provide the DC bias to the MEMS. A 30-V DC bias was used to activate the MEMS switches. The total patch length for the square patch was 17.43 mm inclusive of all pixels and MEMS. Multifrequency operation within 4.5–7 GHz was achieved. 5.4.1.5  Monolithic MEMS Frequency Reconfigurable Slot and Monopole Antennas

Cetiner et al. [24] reported a MEMS frequency reconfigurable annular slot antenna that consisted of an inner slot and an outer slot. The antenna consisted of two integrated monolithic MEMS switches biased using a high-resistance line. The antenna was reconfigured for operation at 2.4 GHz and 5.2 GHz with gain of 2 and 2.7 dB, respectively, for the two frequencies. Note that the inner slot was responsible for the high-frequency operation and the outer slot was responsible for the low-frequency operation. Anagnostou et al. [25] reported a monolithic MEMS reconfigurable antenna for rejecting on-demand WLAN signals within the 5.15–5.825-GHz frequency band. They showed that their proposed UWB monopole antenna with integrated MEMS switch can reject that frequency band when the MEMS switch is turned on. 5.4.2  Discrete MEMS Frequency Reconfigurable Antenna

A MEMS frequency and polarization reconfigurable patch antenna was reported by Ho and Rebeiz [26]. The antenna was reconfigured for operation in the 0.9–1.4-GHz frequency band. A commercially packaged RF MEMS single-pole double-throw (SPDT) switch from Omron was used. The switches



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123

had insertion loss less than 1 dB and isolation greater than 30 dB for up to 10 GHz. The frequency reconfigurations from 0.9 to 1.4 GHz for vertical, horizontal, LHCP, and RHCP were demonstrated (see Figure 11 in [26]). The highest measured efficiency is 40% (at the highest frequency) and the lowest measured efficiency is about 5% (at the lowest frequency). These results should be understood in the context of the challenges involved (e.g., the design goals are for frequency and polarization reconfiguration within a broad frequency band). It is always difficult to achieve an optimum multiparameter performance for an antenna when reconfiguration is desired over a wide frequency range. There are also other examples of discrete MEMS frequency reconfigurable antennas [27, 28].

5.5  Reconfigurable Dipole, Monopole-Type Antennas 5.5.1  Frequency Reconfigurable Bow-Tie Antenna

A bow-tie antenna was reconfigured in frequency to support operation in the Bluetooth, WiMax, and WLAN frequency bands [29]. The antenna consists of a bow-tie dipole structure with three conducting pieces for each dipole arm. A general simplistic schematic of the reconfigurable antenna without the feed and bias network can be seen in Figure 5.30. For details on the microstrip fed and switch bias networks, the reader should consult [29]. When all switches are off, the parts A and A1 of the bow-tie dipole are activated creating the high-band operation in 4.51 to 6 GHz for the WLAN band operation. When a forward DC voltage of 2.85V is applied, parts A, A1, B, and B1 are activated and the antenna frequency reconfigures to the 2.97–3.71-GHz band for the WiMax operation. Finally, when a reverse bias voltage of 3.8V is applied, all of the antenna parts are activated and the antenna frequency reconfigures to the 2.2–2.53-GHz band for Bluetooth operation. The antenna was fabricated on a 1-mm-thick substrate with εr = 2.65 and tanδ = 0.0015. Six Infineon PIN diodes BAR 50-02 were used that had a forward turn-on voltage of 0.95V and a forward resistance 3Ω. The diode off

Figure 5.30  A simplistic illustration of the frequency reconfigurable bow-tie dipole introduced in [29]. (After: [29].)

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Reconfigurable Antenna Design and Analysis

state capacitance was 0.15 pF. The measured antenna gains were 0.02, 2.34, and 2.8 dBi at 2.4, 3.5, and 5.5 GHz and the measured efficiencies were 53%, 63%, and 71%, respectively, at those frequencies. 5.5.2  Reconfigurable Antenna Using a Photoconductive Switch

A dipole antenna frequency reconfigured using photoconductive switches was introduced by Panagamuwa et al. [30]. To evaluate the effectiveness of the switch, a microstrip-line switch embodiment was fabricated and measured. The microstrip-line measurement scheme for the switch alone when being illuminated using light is shown in Figure 5.31. The measured S-parameter results as function of optical illumination are shown in Figure 5.32. As seen, with no optical illumination, the proposed switch is off, exhibiting isolation between 12 and 22 dB within the frequency range of 1 to 3 GHz. Isolation starts at the high value of 22 dB at 1 GHz and then gradually decreases to about 12 dB at 3 GHz. With 200 mW of optical illumination, the switch is fully turned on, exhibiting a measured insertion loss of 0.68 dB. The proposed optically reconfigured dipole antenna provides an advantage in that the optical fiber used is electromagnetically transparent to the antenna and thus it does not interfere with the antenna radiation pattern. The antenna was fabricated on a 1.17-mm-thick TLY-5 substrate (εr = 2.2), which did not contain any ground plane. A circular balun described in [31, 32] was used to feed the dipole. A photograph of the antenna can be seen in Figure 5.33, which shows the dipole antenna, the two photoconductive switches, the balun, and the CPS feedline. The measured return loss versus optical power characteristics are plotted in Figure 5.34, which shows that with very low optical power, both switches being off, the antenna operates at its highest resonant frequency, 3.15 GHz. As the optical power gradually increases, the antenna operating frequency slowly

Figure 5.31  (a) An example of an optically activated switch in a microstrip transmission line. (b) Experimental setup used for delivering near-infrared light. (© 2006 IEEE. Reprinted, with permission. From: [30].)



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125

Figure 5.32  Measured S-parameters of the switched microstrip line under 0, 10, and 200-mW optical illumination. (© 2006 IEEE. Reprinted, with permission, from: [30].)

Figure 5.33  Photograph of the switched dipole antenna. (© 2006 IEEE. Reprinted, with permission, from: [30].)

shifts from the high-frequency band to the low-frequency band. However, the operation below a 10-dB return loss is only achieved when the optical power is more than 10 mW. For optical power more than 10 mW, the antenna operating

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Reconfigurable Antenna Design and Analysis

Figure 5.34  Measured change in return loss with an increasing optical illumination. (© 2006 IEEE. Reprinted, with permission, from: [30].)

frequency shifts to 2.26 GHz. The gain characteristics plotted in Figure 5.35 show that gain is around 5 dBi at 3.15 GHz when the switches are off. As the optical power increases as 10, 20, 50, and 200 mW, although the antenna operates at the low frequency of 2.26 GHz, gain is lower at a lower optical power. It is only at the 200-mW optical power that the antenna achieves its gain of 3.2 dBi. Gain is greater than 1 dBi for optical power 10 mW or higher. The results of antenna operating frequencies and gain versus optical power extracted from these figures are listed in Table 5.5. The apparent drawback of the proposed reconfigurable antenna seems to be its requirement for a relatively high optical power. The 200-mW optical power requirement may be significant when alternatives such as PIN diode reconfigurable antennas may only require a fraction of that in terms of switch biasing. Reconfigurable antennas using RF MEMS or varactor switches generally do not require any bias current. As stated in the beginning, the proposed scheme in [30] provided an electromagnetically transparent environment for the antenna that could be beneficial for some applications. 5.5.3  Pattern and Frequency Reconfigurable Yagi Antenna

Wahid et al. [33] introduced a reconfigurable Yagi antenna for application in the 2.4-GHz and 5.78-GHz frequencies. The geometry and operation of their proposed antenna can be explained using Figure 5.36. As seen, the antenna array consists of multiple parasitic dipole elements and one driven dipole ele-



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127

Figure 5.35  Change in measured boresight gain with increasing optical illumination. (© 2006 IEEE. Reprinted, with permission, from: [30].)

Table 5.5 Optical Power Versus Antenna Operating Frequencies and Gain Pmw 0 0.2 0.7 2 10 20 50 200

fRES1 (GHz) — — — — 2.26 2.26 2.26 2.26

fRES2 (GHz) 3.15 3.15 — — — — — —

Gain (dBi) 4.6 2.5 — — 1.0 2.1 2.8 3.2

Source: [30].

ment. The elements on the right side of the driven dipole are all directors and the one on the left side is the reflector. There are four nonreconfigurable directors (short size) and four reconfigurable directors. The reconfigurable directors have the locations of their switches identified on them using small rectangles. Understandably, the driven dipole and the reflectors are also reconfigurable and contain their own switches. When all switches are in the off state, the antenna operates at 5.78 GHz and when the switches are in the on state, the antenna operates at 2.4 GHz. The antenna was simulated using IE3D (Zeland Soft-

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Reconfigurable Antenna Design and Analysis

Figure 5.36  Reconfigurable Yagi-Uda array [33], not to scale. (After: [33].)

ware, Freemont, California; subsequently acquired by Mentor Graphics). To represent the switch on and off states, the ideal short and open were used, respectively. Finally, an antenna was also fabricated using 5-mil-thick RT/Duroid 5880 (εr = 2.2, tanδ = 0.0009) substrate, the dimensions and spacings of which at the respective frequencies are listed in Table 5.6. Wahid et al. [33] reported that, at 2.4 GHz, measured directivity, HPBW, and F/B were 7.6 dBi, 50.5°, and 16.6 dB, respectively. The same quantities at 5.78 GHz were 9.8 dBi, 31°, and 12.5 dB, respectively.

5.6  The Reconfigurable Antenna Aperture (RECAP) Pringle et al. [34] introduced a RECAP that was studied, optimized, built, and tested for operation within the 0.9–1.6-GHz frequency range. Although Pringle et al. did not refer to the RECAP as a pixel antenna structure, later literature started to refer to antennas made from many electrically small conductive patches controlled by switches as pixel antennas. Pringle et al. [34] proposed a planar array of electrically small patches interconnected with switches to generate the RECAP using genetic algorithm (GE). Field-effect transistor (FET) switches were used. Figure 5.37 shows the RECAP antenna in monopole configuration where each square metal patch has a side length, l 0.085λ. For cases that fulfill these constraints, the phase of the induced current in the parasitic leads the phase of the current in the driven dipole. Conversely, if only the driven dipole and the parasitic dipoles on Layer 2 are considered, then the parasitics can be made to function as directors as long as d2 > 6 mm or 0.049λ. Note that since the parasitics in Layer 2 are also at a height, their effective distance becomes d2 > 0.095λ to function as directors. The array beam points to φ = 45° when parasitic 3 is on and it works as a director. The implemented array with d1 = 7 mm, d2 = 4 mm and with PIN diode switches (SMP 1345, Skyworks Inc.) has the return loss performance shown in Figure 6.7, which shows operation in the 2.45-GHz frequency band. Figure 6.8 shows simulated array patterns next to an anatomical human body model, indicating beam steering along different angles. Note that Layer 1 was placed 25 mm away from the body. Gain was between 6.9 and 8 dBi. Simulated specific absorption rate (SAR) and temperature rise ∆T for various switching configurations are listed in Table 6.4.



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Figure 6.7  Photographs of the fabricated antenna array and measured free-space input S11 (dB) versus frequency data with switch states as the parameter. Antenna dimensions are given in Figure 6.6, d1 = 7 mm, d2 = 4 mm. (© 2012 IEEE. Reprinted, with permission, from: [10].)

Figure 6.8  Simulated normalized radiation patterns of the array next to an anatomical human body model as a function of various switching states of the array. (© 2012 IEEE. Reprinted, with permission, from: [10].)

6.4  Reconfigurable Parasitic Patch Array Another class of ESPAR antennas includes microstrip patch antennas that contain driven and parasitic elements. Here also, the parasitic elements are controlled using electronic devices to achieve pattern reconfiguration. However, unlike dipole or monopole-type configurations, pattern steering on only one

152

Reconfigurable Antenna Design and Analysis Table 6.4 SAR and peak ΔT at 2.45 GHz due to the proposed array and a driven folded dipole. Separation between the body and the array Layer 1, S=25 mm.

© [2012] IEEE. Reprinted, with permission, from [10].

hemisphere is permitted because patch antennas have ground planes underneath them. 6.4.1  An ESPAR Patch Antenna Array

An ESPAR patch antenna array was introduced by Yusuf and Gong [11]. The array was designed for operation at 3 GHz using 1.6-mm-thick RO3003 substrate (εr = 3, tanδ = 0.0013). A schematic of the array can be seen from Figure 6.9. It consists of three microstrip patch antennas where the center element (i.e., Patch 0) is excited using a source and patches 1 and 2 are terminated with variable capacitors C1 and C2. For example, as can be seen from Table 6.5, when C1 = 0.2 pF and C2 = 2 pF, the array beam points to –20° direction. Reversing those capacitance values (i.e., making C1 = 2 pF and C2 = 0.2 pF) makes the array beam point to +20° direction. For small capacitance values, which means higher reactances, such as C1 = C2 = 0.2 pF, the array beam points in the default 0° direction. For cases with small and moderate capacitances like C1 = 0.5 pF and C2 = 0.2 pF, the beam points close to boresight at an angle of –7°. Further HFSS simulations and measurement results indicated that a combination of low (0.2 pF) and high (1 pF) capacitance or high and low reactance for that matter can allow the beam to be steered to –20° and +20°. 6.4.2  An Aperture-Coupled ESPAR Patch Antenna Array

In [12], an aperture coupled ESPAR patch antenna array was introduced for operation at around 1 GHz with beam-steering capability at –15°, 0°, and 15°. The introduced ESPAR array consists of three aperture-coupled patch antennas (Figure 6.10) of which only the one in the center is actually driven. The two other patches are parasitic. However, they also contain their own coupling



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153

Figure 6.9  A microstrip patch ESPAR array controlled using variable capacitors. All dimensions are in millimeters. (After: [11].)

Table 6.5 Capacitance Versus Beam Direction Estimated from Figure 2 of [11]* Approximate Beam Direction

C1 (pF) 0.2 (265.2)

C2 (pF) 2 (26.5)

2

0.2

+20°

0.2

0.5 (106)

7°

0.5

0.2

–7°

0.2

0.2

0°

–20°

*The numbers in the parenthesis indicate the corresponding reactance values in ohms calculated at 3 GHz.

apertures and microstrip stubs. The driven element at the center is excited using an aperture, which is excited using a 50Ω microstrip feed that lies at the bottom of Substrate 2. Substrate 1 is 62-mil (1.6-mm) thick Duroid 5880 (εr = 2.2, tanδ = 0.0009) and Substrate 2 was 60-mil (1.5-mm) thick RO4003 (εr = 3.55, tanδ = 0.0027). The varactor diodes Dc1, Dc2, Dc3, Dc4 provide capacitances that allow the array beam steering. The other four varactor diodes (e.g., Dt1, Dt2, Dt3, and Dt4)

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Reconfigurable Antenna Design and Analysis

Figure 6.10  A modified schematic of the aperture coupled microstrip patch ESPAR array reported in [12] controlled using varactor diodes. (After: [12].)

allow control of the antenna impedance matching and tuning. The capacitances provided by the Dcn varactor diodes are described as coupling capacitances and are labeled as C_CPL in [12] and those provided by the Dtn diodes are described as compensation capacitances and are labeled as C_CMP. The compensation capacitances are needed to maintain the resonance at the appropriate frequency and obtain a good return loss. The varactor diodes are controlled using DC bias voltages to obtain the proper capacitance. In the reported work [12], Infineon BB857E7902 diodes were used. Some of the approximate values versus reverse bias voltage are listed in Table 6.6. The geometrical details of the array can be seen in Figure 6.11. As seen, each patch is 91 mm by 78 mm. These patches reside on Substrate 1, which is 1.6-mm-thick Duroid 5880. The center patch is driven using a 3.5-mm-wide microstrip feed line etched on the bottom side of Substrate 2, which is 1.5-mmthick RO4003. Each coupling aperture that is on the ground plane of Substrate 2 measures 21 mm by 6 mm. There is a 3-mm gap between any two patches that accommodates the coupling varactor diodes. The compensation varactor diodes lie at the edges of the two parasitic varactor diodes. Note that the microstrip feedline has a 38-mm-long extension stub beyond the aperture center. There are grounding stubs present as well. Simulated beam-steering performance versus varactor diode capacitances can be seen from Table 6.7. To steer the beam to –7°, the Dc3 and Dc4 varactor diode capacitances will have to assume the capacitance values listed for varactor diodes Dc1 and Dc2, and the Dt3 and Dt4 varactor diode capacitances will have to assume the capacitance values listed for varactor diodes Dt1 and Dt2. The same logic applies if one wishes to steer the array beam in the –15° direction. Simulated and measured radiation patterns of the array are shown in Figure 6.12, which show the various beam-steering directions.



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Table 6.6 Reverse Bias Voltage Versus Capacitance of Infineon Varactor Diode BB857E7902 Reverse Bias Voltage (V) 2 4 6 8 10 14

Capacitance (pF) 5 3 2 1.7 1.3 1.0

Source: [12].

Figure 6.11  Dimensions of the radiating structure. (a) Patch layer. (b) Feed layer. (L = 91, W = 78, D = 81, G = 3, S = 38, Ws = 21, Ls = 6, O = 25.5, M = 3.5, and P = 28.) All dimensions are in millimeters. (© 2012 IEEE. Reprinted, with permission, from: [12].)

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Reconfigurable Antenna Design and Analysis

Table 6.7 Simulated Beam Scan Angle Versus Varactor Diode Capacitance Values for the Various Varactors Beam Scan Angle 15°

Array Dc1 and Dc2 Dc3 and Dc4 Dt1 and Dt2 Dt3 and Dt4 Peak Gain Capacitances Capacitances Capacitances Capacitances (dBi) (pF) (pF) (pF) (pF) 6.9 1.5 3 0.5 3.5

7°

7.5

1.7

2.9

1.1

3.2

0°

7.5

2.6

2.6

2.6

2.6

Source: [12].

Figure 6.12  Simulated and measured normalized linear gain patterns for different scanning angles: (a) –15°, (b) –7°, (c) 0°, (d) 7°, and (e) 15°. (© 2012 IEEE. Reprinted, with permission, from: [12].)

6.4.3  A Frequency and Pattern Reconfigurable Pixelled Monopole Antenna

A multisized pixelled reconfigurable monopole antenna was introduced in [13]. Subsequently, further developments were reported in [14, 15]. The configuration and results reported here is a very brief summary of the work reported in [15]. Although both frequency and pattern reconfigurations were reported, we will discuss only the pattern reconfiguration aspect of the proposed antenna. Reconfiguration was achieved through optimization of the geometry considering multiple approaches.



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157

The geometry of the antenna in a schematic form is shown in Figure 6.13. In a simple form, the reconfigurable antenna consists of a monopole configuration consisting of square pixels with two sizes, 4.5 mm (for the bottom 15 pixels) and 18 mm (for the top 6 pixels). While some of the bottom pixels are joined through hard connections, others are controlled using RF switches. As can be seen, there are 11 such hard connections using a metal strips. The switches are numbered as SW1, SW2, and so forth. The details on the biasing of the diodes are available in [15]. The antenna was fabricated on a 0.81-mmthick RO4003 substrate and placed on a metal ground plane. The dimensions of the ground plane were 160 mm by 200 mm. Optimization of the antenna patterns revealed that the antenna can be designed for operation in the 2.45-GHz band with pattern reconfiguration capability at –60°, –30°, 0°, 30°, and 60°. Table 6.8 lists the switching configurations needed to achieve beam pointing in the different directions while maintaining operation at 2.45 GHz. Simulated and measured radiation patterns obtained from [15] are shown in Figure 6.14 and the corresponding geometries are shown in Figure 6.15. There have been many other reported works on pattern reconfigurable pixel antennas some of which can be found in [16–19].

Figure 6.13  Geometry and dimensions of the pixelated reconfigurable monopole antenna reported in [15]. All dimensions are in millimeters. (After: [15].)

158

Reconfigurable Antenna Design and Analysis Table 6.8 Simulated and Measured Beam Directions Versus Switch States for the Pixelated Reconfigurable Monopole Antenna in [15] Switch States 60 SW1 3 SW2 3 SW3 — SW4 — SW5 3 SW6 3 SW7 — SW8 3 SW9 3 SW10 — SW11 — SW12 —

30 3 —

0 3 —

3 3 3 — — —

3 — — — — — — — — —

3 3 3 —

–30 3 3 — 3 — 3 — — 3 3 — 3

–60 — 3 3 3 3 — 3 — — 3 — —

Source: [15].

6.5  Series-Fed Patch Phased Array For wireless base station applications, down-tilting the beam of an array at different angles is beneficial. Series-fed patch phased arrays with varactor-controlled phase shifters (Figure 6.16) have been proposed to support such applications [20–22]. While, ordinarily, the array beam will point towards the horizon, the proposed phased array will allow beam tilting to smaller angles below the horizon provided that the design is able to maintain the required return loss and gain performance. The schematic of a 10-element, series-fed patch array operating at 5.8 GHz reported in [21] is illustrated in Figure 6.16. The array consists of 10 series-fed microstrip patches, each with dimensions L and W. The array contains varactor diode-based phase shifters in between two adjacent patches. In Figure 6.16, only a varactor diode is shown connecting two patches; however, it should mean a varactor diode-based phase shifter. Among the 10 patch elements there are nine identical phase shifters. The microstrip line on the top is terminated with a 50Ω matched load. Note that the configuration can just be flipped where the feed is on the top and the load is on the bottom. Under the scenario shown in Figure 6.16, the default beam direction is along the horizon or X-direction in Figure 6.16. With the application of various varactor diode bias voltages, the integrated phase shifters will allow the beam to tilt down from the X-axis. For a configuration where the feed is on the top and the load is on the bottom, the



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Figure 6.14  Realized gain pattern of the configurations with maximum gain at angular directions θ = –60°, –30°, 0°, 30°, and 60°. (a) f = 2.45 GHz, measurement, and (b) f = 2.45 GHz, simulation. (© 2012 IEEE. Reprinted, with permission, from: [15].)

situation is reversed. If one envisions a reconfigurable situation where the feed and the load are switched back and forth, the beam can be steered up and down along the elevation plane, meaning the XZ plane. The phase shifter used in the array is of the type transmission-type phase shifters (TTPS). A TTPS illustrated in Figure 6.17 and isolated from the array with Ls = 13.3 mm was fabricated and measured. The selected measured insertion loss and phase shift data from [21] can be seen in Table 6.9. The TTPS exhibited excellent return loss (better than 16 dB) and low insertion loss (30 dB). The proposed antenna is capable of generating 0/90 or ±45° polarizations. The measured antenna gain is 4.9 dBi. When studied in a MIMO system configuration, the proposed antenna showed improved diversity gain. Kishor and Hum [20] introduced a pattern reconfigurable chassis-mode MIMO antenna. Their proposed antenna consists of a 4-port configuration where ports 1 and 2 are capacitively excited and ports 3 and 4 contain load impedances maintained by PIN diode switches. There are four metal plates at a height of 6 mm from the chassis. Ports 1 and 2 and their associated metal plates reside along the two corners along a diagonal of the chassis. Port 3 resides next to port 1 and port 4 resides on the same chassis edge as port 1 but remains at the middle of the chassis along the length dimension. The chassis of the board is a 90 mm by 50 mm FR4 PCB (0.762-mm thickness). The plates are made of single-sided, 1.28-mm-thick RO3006 substrates (εr = 6.15, tanδ = 0.0025). Simulation studies and characteristic mode analyses are presented. Kishor and Hum used Skyworks PIN diode SMP1345-079LF to reconfigure ports 3 and 4. The on state of the diode is represented using a small 1.4Ω resistance and an inductance and the off state is represented using a 0.12-pF capacitance and a resistance. The diodes in ports 3 and 4 are turned on and off, respectively, creating two states of operation, state 1 and state 2. In state 1, the diode in port 3 is forward-biased or on and the diode in port 4 is off. In state 2, it is just the opposite. The antenna design is targeted for operation at around 2.3 GHz. The measured 6-dB return loss bandwidths at ports 1 and 2 are 7.2% and 12.6%. The port-to-port isolation at 2.36 GHz is around 19.5 dB. Simulated and measured radiation pattern results for this antenna for the two states are also presented. Antenna efficiencies measured using the Wheeler cap method show efficiencies of 80.5% and 75.9%, respectively, in state 1 and 71.9% and 79.5%, respectively, in state 2. Qin et al. [21] presented a pattern reconfigurable U-slot patch antenna for MIMO systems. Their proposed antenna consists of a patch antenna that is capable of generating patterns that are broadside to the patch as well as nearendfire. The design and operation were achieved at a center frequency of 5.32 GHz. The geometry of the antenna is shown in Figure 10.3, which consists of a microstrip patch antenna printed on a substrate that contains a U-slot. The antenna was fabricated on a 3.175-mm-thick Duroid 5880 substrate (εr = 2.2, tanδ = 0.0009). Although patches with U-shaped slots have been studied by others before, the authors here introduce PIN diode switch enabled pattern reconfiguration. As seen in Figure 10.3, the antenna consists of 8 PIN diode



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Figure 10.3  Schematics of the pattern reconfigurable U-slot antenna. (After: [21].)

switches. For the purpose of the work [21], beam-led PIN diodes (MA4AGBLP912) were used. The on state of the diode represented a 4Ω resistance and the off state represented 0.025-pF capacitance. The antenna configuration has three states of operation as listed in Table 10.1. The detailed dimensions of the antenna, substrate, and other parameters are available in [21]. The patch in state 1 has none of the diodes biased, which means all of the shorting posts are disconnected from the patch resulting in a broadside radiation pattern with the beam peak directed along theta = 0°. In state 2, the diodes labeled M are reverse-biased and those labeled N, are forward-biased. This configuration results in a monopolar-type radiation pattern [22] with the pattern maximum directed in the theta = 30°–60° direction when the pattern in the yz-plane is considered. In state 3, the diode states are as defined in Table 10.1 Table 10.1 The Different States of the Reconfigurable U-Slot Antenna Diodes Labeled M Diodes Labeled N State 1 No bias applied No bias applied State 2 Reverse-biased Forward-biased State 3 Forward-biased

Source: [21].

Reverse-biased

Resulting Radiation Patterns Pattern broadside to the patch plane Monopolar-type pattern with beam maximum along theta = ±30°–60° in the yz-plane Monopolar-type pattern with beam maximum along theta = ±30°–60° in the xz-plane

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and the patterns are once again the monopolar type with the beam maximum in the theta = 30°–60° direction when the pattern in the zx-plane is considered. The measured antenna bandwidth was 6.6% with a center frequency of 5.32 GHz and under S11 < –10 dB. The measured reported antenna gains are around 6 dBi, 2.5 dBi, and 3 dBi for states 1, 2, and 3, respectively. A detailed 2 × 2 MIMO performance study of this antenna was also performed by comparing its performance with a standard 2.2-dBi gain omnidirectional antenna. Two U-slot antennas were placed at one wavelength center-to-center distance and measured in both LOS and non-LOS configurations. As the antenna has three states, nine configurations resulted. The envelope correlation coefficient estimated from the pattern showed the correlation coefficient being below 0.5 for most of the cases except for cases when the combination contained similar cases (e.g., state 1 and state 1 for both antennas 1 and 2 in the MIMO configuration). The proposed configuration provides capacity improvements of 17% and 12% for the LOS and non-LOS cases, respectively, for 10-dB SNR. More detailed description and analysis are available in [21]. Guo et al. [23] introduced an 8-port dual-band MIMO array that can operate in the 3.5-GHz band (3,400–3,600 MHz) and 5-GHz band (4,800– 5,100 MHz) for 5G MIMO mobile phone handsets. Their proposed antennas are mounted on the two longitudinal edges of a mobile phone handset measuring 150 mm by 75 mm. Each antenna consists of a printed monopole and a coupled loop. Only the monopole is directly excited that then excites the loop that is parasitically coupled to it. As shown in Figure 10.4, each antenna occupies a space of 15 mm by 7 mm. The 7-mm dimension is along the height of the phone.

Figure 10.4  Geometry and detailed dimensions of the proposed 8-antenna array (unit: millimeter). (© 2018 IEEE. Reprinted, with permission, from: [23].)



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Antennas 5 through 8 are mirror images of antennas 1 through 4. The gap between two adjacent antenna elements is 10 mm. To reduce the mutual coupling between the adjacent elements, a neutralized line (NL) was added between the two middle gap coupled-loop branches (see Figure 10.5). Simulation results show the antenna operating in the 3.5-GHz and the 5-GHz frequency bands with the S11 bandwidth being satisfied under –10 dB in the low band –6 dB in the high band. The mutual coupling between any two elements is mostly below –10 dB except for antennas 2 and 3 in the low-frequency band. For this case, the mutual coupling is as high has –8 dB. This is without the NL. With the help of the NL, the mutual coupling falls below –10 dB between antennas 2 and 3. The measured results confirm the findings in the simulations. The measured antenna element efficiency is between 40% and 70% in the 3.5-GHz frequency band and between 40% and 80% in the 5-GHz frequency band. ECC values were calculated considering the uniform incident wave environment with balanced polarization, which revealed values between 0.05 and 0.08, which is much better than the ECC guidelines suggested for MIMO antennas with improved performance. Photographs of the proposed MIMO antenna [23] can be seen from Figure 10.5, which show the 8 SMA ports and the NL that is used to reduce the mutual coupling. Calculated MEG results from measured patterns are shown in Figure 10.6, which shows minimal variation in MEG within each frequency band. Guo et al. [23] also calculated the ergodic channel capacity for their proposed

Figure 10.5  Photographs of the fabricated 8-antenna array. (a) Front view. (b) Back view. (c) Enlarged photograph of the gap-coupled loop. (© 2018 IEEE. Reprinted, with permission, from: [23].)

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Figure 10.6  Calculated MEGs from measured complex E-field patterns. (© 2018 IEEE. Reprinted, with permission, from: [23].)

MIMO antenna, assuming an independent and identically distributed Rayleigh fading channel with 20-dB SNR in the propagation environment. Under this condition and under other assumptions, the channel capacities are approximately 37–38.5 bps/Hz for the 3.5-GHz band and 37.5–38 bps/Hz in the 5-GHz band, respectively. These results compare well with the upper bound of 46 bps/Hz for an 8 by 8 MIMO array. A 12-port 4G/5G MIMO antenna for smartphone application was proposed by Dong et al. [24]. Their proposed antenna system consists of 4G and 5G antennas. The 4G antenna module is a 2 by 2 antenna array that supports a number of 4G frequency bands (see Table 10.2). The 4G antenna is capable of operating in three frequency bands (e.g., 691–970 MHz, 1,430–2,740 MHz, and 3,360–3,960 MHz). The 5G antenna module consists of a 10-element monopole antenna MIMO array that can support the 5G application frequency bands listed in Table 10.2. The 5G antenna module operates in the frequency ranges of 3.38–3.66 GHz and 4.33–5.18 GHz. The 12-port antenna system was printed on 0.8-mm-thick FR4 (εr = 4.4, tanδ = 0.02) PCB. The top part of the PCB had a clearance zone where metal was removed to accommodate the 4G antenna. In addition, a vertically mounted top board was also used to accommodate this antenna. There were 10 additional monopole antennas that were printed on the left and right edges of the PCB. To accommodate each monopole, a metal clearance zone of 16.7 mm by 3 mm was created. The authors reported total efficiency between 40% and 80% for the 4G antenna module. The total antenna efficiency for the 5G antenna module is between 70% and 80%. The envelope correlation coefficient (ECC) is generally low except for between antennas 1 and 2 and between antennas 6 and 7.



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Table 10.2 Application Frequency Bands Supported by the MIMO Antenna Proposed in [24] 4G MIMO Antenna Application Frequency (MHz) LTE700 698–784 GSM850 824–894 GSM900 880–960 DCS 1,710–1,880 PCS 1,850–1,990 UMTS 1,920–2,170 LTE2300 2,300–2,400 LTE2500 2,500–2,690 2.4-GHz 2,400–2,485 WLAN 3.5-GHz 3,300–3,700 WiMAX LTE3400 3,400–3,800

5G MIMO Antenna Application Frequency (GHz) 5G 3.4–3.6 GHz 5G 4.8–5.0 GHz — — — — — — — — — — — — — — —

—

—

—

Li et al. [25] introduced an 8-element MIMO antenna array for handset application in the 3.4–3.6-GHz frequency band. The building block for the 8-element array is an open-slot antenna element. Their proposed antenna geometry can be seen in Figure 10.7. There is a 2G/3G/4G antenna on the top left corner of the PCB along with the 8 MIMO antennas. The measured S-parameters for the MIMO antennas are shown in Figure 10.8. Magnitudes of the Snn show the antenna operation within the 3.4–3.6GHz frequency bands. Magnitudes of the Smn show that the mutual coupling between the various pairs is below 17.5 dB in the frequency range of interest. Simulated and measured radiation patterns of antennas 1, 5, 2, 4, and 3 are shown in Figure 10.9. The various patterns show complementary radiation performance directions that are beneficial for MIMO performance. The authors reported measured ECC of