Optical and Wireless Technologies: Proceedings of OWT 2020 (Lecture Notes in Electrical Engineering, 771) 9811628173, 9789811628177

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Lecture Notes in Electrical Engineering 771

Manish Tiwari · Ravi Kumar Maddila · Amit Kumar Garg · Ashok Kumar · Preecha Yupapin   Editors

Optical and Wireless Technologies Proceedings of OWT 2020

Lecture Notes in Electrical Engineering Volume 771

Series Editors Leopoldo Angrisani, Department of Electrical and Information Technologies Engineering, University of Napoli Federico II, Naples, Italy Marco Arteaga, Departament de Control y Robótica, Universidad Nacional Autónoma de México, Coyoacán, Mexico Bijaya Ketan Panigrahi, Electrical Engineering, Indian Institute of Technology Delhi, New Delhi, Delhi, India Samarjit Chakraborty, Fakultät für Elektrotechnik und Informationstechnik, TU München, Munich, Germany Jiming Chen, Zhejiang University, Hangzhou, Zhejiang, China Shanben Chen, Materials Science and Engineering, Shanghai Jiao Tong University, Shanghai, China Tan Kay Chen, Department of Electrical and Computer Engineering, National University of Singapore, Singapore, Singapore Rüdiger Dillmann, Humanoids and Intelligent Systems Laboratory, Karlsruhe Institute for Technology, Karlsruhe, Germany Haibin Duan, Beijing University of Aeronautics and Astronautics, Beijing, China Gianluigi Ferrari, Università di Parma, Parma, Italy Manuel Ferre, Centre for Automation and Robotics CAR (UPM-CSIC), Universidad Politécnica de Madrid, Madrid, Spain Sandra Hirche, Department of Electrical Engineering and Information Science, Technische Universität München, Munich, Germany Faryar Jabbari, Department of Mechanical and Aerospace Engineering, University of California, Irvine, CA, USA Limin Jia, State Key Laboratory of Rail Traffic Control and Safety, Beijing Jiaotong University, Beijing, China Janusz Kacprzyk, Systems Research Institute, Polish Academy of Sciences, Warsaw, Poland Alaa Khamis, German University in Egypt El Tagamoa El Khames, New Cairo City, Egypt Torsten Kroeger, Stanford University, Stanford, CA, USA Yong Li, Hunan University, Changsha, Hunan, China Qilian Liang, Department of Electrical Engineering, University of Texas at Arlington, Arlington, TX, USA Ferran Martín, Departament d’Enginyeria Electrònica, Universitat Autònoma de Barcelona, Bellaterra, Barcelona, Spain Tan Cher Ming, College of Engineering, Nanyang Technological University, Singapore, Singapore Wolfgang Minker, Institute of Information Technology, University of Ulm, Ulm, Germany Pradeep Misra, Department of Electrical Engineering, Wright State University, Dayton, OH, USA Sebastian Möller, Quality and Usability Laboratory, TU Berlin, Berlin, Germany Subhas Mukhopadhyay, School of Engineering & Advanced Technology, Massey University, Palmerston North, Manawatu-Wanganui, New Zealand Cun-Zheng Ning, Electrical Engineering, Arizona State University, Tempe, AZ, USA Toyoaki Nishida, Graduate School of Informatics, Kyoto University, Kyoto, Japan Federica Pascucci, Dipartimento di Ingegneria, Università degli Studi “Roma Tre”, Rome, Italy Yong Qin, State Key Laboratory of Rail Traffic Control and Safety, Beijing Jiaotong University, Beijing, China Gan Woon Seng, School of Electrical & Electronic Engineering, Nanyang Technological University, Singapore, Singapore Joachim Speidel, Institute of Telecommunications, Universität Stuttgart, Stuttgart, Germany Germano Veiga, Campus da FEUP, INESC Porto, Porto, Portugal Haitao Wu, Academy of Opto-electronics, Chinese Academy of Sciences, Beijing, China Junjie James Zhang, Charlotte, NC, USA

The book series Lecture Notes in Electrical Engineering (LNEE) publishes the latest developments in Electrical Engineering - quickly, informally and in high quality. While original research reported in proceedings and monographs has traditionally formed the core of LNEE, we also encourage authors to submit books devoted to supporting student education and professional training in the various fields and applications areas of electrical engineering. The series cover classical and emerging topics concerning: • • • • • • • • • • • •

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More information about this series at http://www.springer.com/series/7818

Manish Tiwari · Ravi Kumar Maddila · Amit Kumar Garg · Ashok Kumar · Preecha Yupapin Editors

Optical and Wireless Technologies Proceedings of OWT 2020

Editors Manish Tiwari Department of Electronics and Communication Engineering Manipal University Jaipur Jaipur, India Amit Kumar Garg Department of Electronics and Communication Engineering Indian Institute of Information Technology (IIIT), Kota (MNIT Campus Jaipur) Jaipur, India

Ravi Kumar Maddila Department of Electronics and Communication Engineering Malaviya National Institute of Technology Jaipur, India Ashok Kumar Department of Electronics and Communication Engineering Government Women Engineering College Ajmer, India

Preecha Yupapin Computational Optics Research Group Advanced Institute of Materials Science Ton Duc Thang University Ho Chi Minh City, Vietnam

ISSN 1876-1100 ISSN 1876-1119 (electronic) Lecture Notes in Electrical Engineering ISBN 978-981-16-2817-7 ISBN 978-981-16-2818-4 (eBook) https://doi.org/10.1007/978-981-16-2818-4 © The Editor(s) (if applicable) and The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022, corrected publication 2022 This work is subject to copyright. All rights are solely and exclusively licensed by the Publisher, whether the whole or part of the material is concerned, specifically the rights of translation, reprinting, reuse of illustrations, recitation, broadcasting, reproduction on microfilms or in any other physical way, and transmission or information storage and retrieval, electronic adaptation, computer software, or by similar or dissimilar methodology now known or hereafter developed. The use of general descriptive names, registered names, trademarks, service marks, etc. in this publication does not imply, even in the absence of a specific statement, that such names are exempt from the relevant protective laws and regulations and therefore free for general use. The publisher, the authors and the editors are safe to assume that the advice and information in this book are believed to be true and accurate at the date of publication. Neither the publisher nor the authors or the editors give a warranty, expressed or implied, with respect to the material contained herein or for any errors or omissions that may have been made. The publisher remains neutral with regard to jurisdictional claims in published maps and institutional affiliations. This Springer imprint is published by the registered company Springer Nature Singapore Pte Ltd. The registered company address is: 152 Beach Road, #21-01/04 Gateway East, Singapore 189721, Singapore

Conference Committee Members

Organizing Committee Patron Prof. Udaykumar R. Yaragatti, Director, MNIT Jaipur General Chairs Prof. Mahendra Mohan Sharma, MNIT Jaipur Prof. Ghanshyam Singh, MNIT Jaipur Prof. Manish Tiwari, Manipal University Jaipur Prof. Preecha Yupapin, TDTU, Vietnam Conveners Dr. Ravi Kumar Maddila, MNIT Jaipur Dr. Prabhat Kumar, NIT, Nagpur Dr. Jitendra Kumar Deegwal, GEC, Ajmer Dr. Dinesh Yadav, Manipal University Jaipur Dr. Sandeep Vyas, JECRC, Jaipur Organizing Secretaries Dr. Kuldeep Singh, MNIT Jaipur Dr. Rajendra Mitharwal, MNIT Jaipur Dr. Sanjeev Kumar Metya, NIT, Arunachal Pradesh Dr. Amit Kumar Garg, IIIT, Kota Dr. Ashok Kumar, GWEC Ajmer IETE Oversight Committee Prof. Deepak Bhatnagar, FIETE Prof. S. K. Bhatnagar, FIETE Mr. K. M. Bajaj, FIETE

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Media and Publicity Committee Dr. Karan Verma, NIT Delhi (Chair) Dr. Arjun Kumar, Bennett University (Co-chair) Dr. Rukhsar Zafar, SKIT, Jaipur Sponsorship Committee Dr. Ashok Kumar, GWEC Ajmer (Chair) Dr. Amit Kumar Garg, IIIT, Kota Dr. Dinesh Yadav, Manipal University Jaipur Dr. Monika Mathur, SKIT, Jaipur Registration Committee Dr. Rahul Kumar Chaurasiya, MNIT Jaipur (Chair) Mr. Nidhish Tiwari, CITM, Jaipur Mr. Nitesh Mudgal, MNIT Jaipur Mr. Ankur Saharia, MNIT Jaipur Ms. Shweta Mittal, Manipal University Jaipur Publication Committee Dr. Amit Kumar Garg, IIIT, Kota Dr. Sarthak Singhal, MNIT Jaipur Dr. Vinay Kanooongo, SKIT, Jaipur. Mr. Bipin Kumar Saw, MNIT Jaipur Hospitality Committee Dr. Mukesh Kumar Gupta, MBM Engineering College, Jodhpur Mr. Narendra Kumar Godara, MNIT Jaipur Mr. Dinesh Bhatiya, RTU Kota Mr. Ankit Agrawal, SKIT, Jaipur Student Volunteering Committee Neha Yaragatti, Manipal University Jaipur Varshali Sharma, Carneige Mellon University, Pittsburg, USA Shishir Kumar Sharma, St. Anselm’s Pink City Sr. Sec. School, Jaipur

International Advisory Committee Prof. Hiroyuki Tsuda, Keio University, Japan Prof. Ali Gharsallah, University of Tunis E. M., Tunisia Prof. Buryy Oleh Anatolievyach, LPNU, Ukraine Prof. Ajoy Kar, HW University, Edinburgh, UK Dr. Suchandan Pal, CEERI Pilani, India

Conference Committee Members

Prof. Kolin Poul, IIT, Delhi, India Prof. Konstantin Kozadaev, BSU, Minsk, Belarus Prof. Mário F. S. Ferreira, University of Aveiro, Portugal Dr. Miklos Veres, HAS, Budapest, Hungary Prof. Sergii Ubizskii, LPNU, Ukraine Prof. Yuri Shpolyanskiy, University of Saint Petersburg, Russia Dr. Yaseera Ismail, UKZN, Durban, South Africa Dr. Aulia M.T. Nasution, ITS, Surabaya, Indonesia Dr. Agus Muhamad Hatta, ITS, Surabaya, Indonesia

Technical Program Committee Dr. Akshay Kr. Rathore, Concordia University, Canada Prof. Takasumi Tanabe, Keio University, Japan Prof. Toshiharu Saiki, Keio University, Japan Prof. Ratnajit Bhattacharya, IIT Guwahati Prof. Fazal Talukdar, NIT, Silchar Dr. Bishnu Prasad Gautam, WAKHOK, Japan Dr. Reza Abdi-Ghaleh, University of Bonab, Iran Dr. Kalpana Dhaka, IIT Guwahati, India Dr. Upena D. Dalal, SVNIT, Surat Dr. Manish Mathew, CEERI Pilani, India Dr. Preetam Kumar, IIT, Patna Dr. C. Periasamy, MNIT Jaipur Dr. Seema Verma, Banasthali University Dr. Narendra Kumar Yadav, JECRC University, Jaipur Dr. Rekha Mehra, GEC, Ajmer Dr. Bramha P. Pandey, GLA University, Mathura Dr. Sanyog Rawat, Manipal University Jaipur Dr. Lokesh Tharani, RTU Kota Dr. Girish Parmar, RTU Kota Dr. Anil Yadav, Amity University, Gurgaon Dr. Nagesh Janrao, Government Polytechnic, Pune

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Our Reviewers

Dr. Surendra Agarwal, Government Women Engineering College, Ajmer, [email protected] Dr. Ashwini Arya, College of Technology Pantnagar, [email protected] Dr. Ramesh Battula, Malaviya National Institute of Technology Jaipur, [email protected] Mr. Sudhir Bhaskar, Indian Institute of Technology (BHU) Varanasi, [email protected] Dr. Jitendra Deegwal, Government Engineering College, Ajmer, Rajasthan, [email protected] Dr. Tarun Dubey, Manipal University Jaipur, [email protected] Dr. Umesh Dwivedi, Amity University, Rajasthan, [email protected] Dr. Ashish Ghunawat, MNIT Jaipur, [email protected] Dr. Mukesh Gupta, MBM Engineering College, Jodhpur, [email protected] Dr. Nikhil Deep Gupta, Manipal University Jaipur, [email protected] Mr. Rajkumar Gupta, AUR, [email protected] Dr. Tawfik Ismail, Cairo University, [email protected] Mrs. Sheilza Jain, YMCA University of Science and Technology, [email protected] Dr. Shruti Jain, JUIT, Waknaghat, [email protected] Dr. Nagesh Janrao, Technical Education, [email protected] Mr. Amit Joshi, Malaviya National Institute of Technology Jaipur, [email protected] Dr. Rajesh Khanna, Thapar University, [email protected] Dr. Arjun Kumar, Intel, [email protected] Dr. Ashok Kumar, Government Mahila Engineering College, Ajmer, India, [email protected] Mr. Umesh Kumar, Ajmer Mahila Engineering College, Ajmer, [email protected] Dr. Brijesh Kumbhani, IIT, Ropar, [email protected] Dr. Ravi Maddila, Malaviya National Institute of Technology Jaipur, [email protected] ix

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Mr. Saurabh Maheshwari, Government Women Engineering College Ajmer, [email protected] Dr. Manish Mathew, CEERI Pilani, [email protected] Ms. Monika Mathur, Rajasthan Technical University, [email protected] Dr. Arka Prokash Mazumdar, Malaviya National Institute of Technology Jaipur, [email protected] Mrs. Rekha Mehra, Government Engineering College, Ajmer, [email protected] Mr. Hari Mewara, Government Engineering College, Ajmer, [email protected] Dr. Satyasai Nanda, Malaviya National Institute of Technology Jaipur, [email protected] Dr. Malaya Nath, National Institute of Technology Puducherry, [email protected] Prof. Lotfi Osman, Higher School of Communication of Tunis—University of Carthage, [email protected] Dr. Vipin Pal, National Institute of Technology Meghalaya, [email protected] Dr. Trilochan Panigrahi, National Institute of Technology, Goa, [email protected] Dr. Girish Parmar, Rajasthan Technical University, [email protected] Mr. C. Periasamy, Malaviya National Institute of Technology Jaipur, [email protected] Mr. Pravin Prajapati, Gujarat Technological University, [email protected] Dr. Sanyog Rawat, Manipal University Jaipur, [email protected] Dr. Chitrakant Sahu, MNIT Jaipur, [email protected] Mr. Sourabh Sahu, Malaviya National Institute of Technology Jaipur, [email protected] Mr. Kapil Saraswat, Indian Institute of Technology Kanpur, [email protected] Dr. Preeta Sharan, The Oxford College of Engineering, Bangalore, [email protected] Dr. Jankiballabh Sharma, Rajasthan Technical University, [email protected] Dr. Neeru Sharma, Jaypee University of Information Technology, [email protected] Dr. Prabhat Sharma, Visvesvaraya National Institute of Technology, [email protected] Dr. Ritu Sharma, MNIT Jaipur, [email protected] Dr. Sumit Srivastava, Manipal University Jaipur, [email protected] Prof. Manisha Upadhyay, Nirma University, [email protected] Mr. Karan Verma, Universiti Teknologi PETRONAS, [email protected] Mr. Pankaj Verma, National Institute of Technology, Kurukshetra, [email protected] Dr. Seema Verma, University Banasthali, Vidyapith, [email protected] Dr. Rajesh Vishwakarma, Jaypee University of Engineering and Technology, [email protected]

Our Reviewers

Dr. Sandeep Vyas, Jaipur Engineering College and Research Centre, [email protected] Dr. Ajay Yadav, Government Women Engineering College Ajmer, [email protected] Mr. Dinesh Yadav, Manipal University Jaipur, [email protected] Dr. Narendra Yadav, JECRC University, [email protected] Dr. Sanjeev Yadav, Government Women Engineering College Ajmer, [email protected] Dr. Rukhsar Zafar, SKIT M&G, [email protected]

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Invited Speakers

Prof. V. Sinha, Chair Professor, E&ICT Academy, MNIT Jaipur Title: A Tutorial on Error Control Coding for Mobile Communication Biography: Prof. Vishwanath Sinha did M.S. (Electronics) and D.Sc. in Electrical Engineering from the University of Ljubljana, Slovenia (erstwhile Yugoslavia). He worked as Professor with Indian Institute of Technology Kanpur, Founder Director of LNMIT, Jaipur, Emeritus Professor with Department of ECE, and Chair Professor for E&ICT Academy of MNIT Jaipur before settling down in Delhi. He has been Visiting Professor, since 1977, in a number of foreign universities/institutions in Germany, Switzerland, USA, Brazil and Yugoslavia. The main areas of his interest are telematics, error control coding, satellite communications and technical education. He is Fellow of Institution of Electronics and Telecommunication Engineers and Institution of Engineers, India, and Senior Member of Institute of Electrical and Electronics Engineers, USA. He is also Life Member of Indian Society for Technical Education. Several awards including S. K. Mitra and J. C. Bose Memorial Awards for best papers presented by him and published in 1976 and 1979 have recognized his rich knowledge and contribution in engineering. He has a long teaching experience at IIT Kanpur Brazilian Space Research Institute and University of Puerto Rico and has supervised 36 Ph.D. theses. He has organized different courses for xiii

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Invited Speakers

engineering colleges, working engineers and scientists and also organized a number of large size national/international conferences. He has to his credit more than 80 papers published/presented in national/international journals/conferences/seminars. His research activities at IIT Kanpur, and Brazilian Space Research Institute made notable contribution in various areas including development of remote tutor, a product which establishes a virtual university and provides online interactive teaching tool, telematics projects, graphics and Indian script terminal project, military communication satellite systems, systems for space telecommunications and modulation techniques. He is endowed with rich administrative experience in various capacities within and outside the teaching institutes.

Prof. Hiroshi Yoshikawa, Nihon University, Japan Title: Computer-Generated Hologram for 3D Display Biography: Hiroshi Yoshikawa received the B.S. degree, the M.S. degree and a Ph.D. from Nihon University, all in Electrical Engineering, in 1981, 1983 and 1985, respectively. He joined the faculty at Nihon University in 1985 where he currently holds the position of Professor of Computer Engineering. From December 1988 to April 1990, he was a research affiliate of MIT Media Laboratory. He is Member of SPIE, OSA, Institute of Television Engineers of Japan (ITE), Optical Society of Japan (OSJ), Institute of Electronics, Information and Communication Engineers (IEICE) and HODIC. His current research interests are in holographic printer, electro-holography, computer-generated holograms, display holography and computer graphics.

Invited Speakers

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Dr. Rocío Borrego-Varillas, Istituto di Fotonica e Nanotecnologie, Milano (Italy) Title: Ultrashort Pulses and Applications to Ultrafast Spectroscopy Biography: Rocío Borrego-Varillas received the M.Sc. and Ph.D. in Physics from the University of Salamanca (Spain). Her research interests are ultrafast nonlinear optics, beam shaping and wavefront sensing, spectroscopy and high-power femtosecond lasers. She has been Visiting Researcher in Prof. Krausz’s group at the MPQ (Germany) and Prof. Lancis’ group at the University Jaume I (Spain). In 2013, she was awarded a Marie Curie fellowship to carry out her postdoctoral research on 2D spectroscopy in the UV under the supervision of Prof. Cerullo at the Politecnico di Milano (Italy). She has been Active Member of OSA since 2009. She has volunteered as a young professional and has participated in grant reviews, youth education material projects as well as a blog contributor. She is also one of the General Chairs of IONS and is involved in the OSA nonlinear optics technical group committee. She is Recipient of the 2016 Ivan P. Kaminow Outstanding Young Professional Prize.

Prof. Preecha Yupapin, Ton DucThang University, Ho Chi Minh City, Vietnam Title: Dual-Quantum Wireless and LiFi Networks for 100 PBits Transmission

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Abstract: Quantum transmission network using the electron density circuit transceiver is designed and used for quantum communications. By using the siliconbased circuit, the plasmonic circuit can be formed and implemented, from which the output signals are the electron density and light power. The transceiver circuit using the same structure can be formed and applied for quantum communication. Both LiFi and wireless transmission platforms can be implemented by using a similar manner transceiver. The high density within the range of 100 Pbit can be achieved, while the security issue is also obtained. More applications such as 3D transmission, teleportation, power transmission and spin wave transmission can also be applied, which can be supplied for the large demand in bits transmission in the future. The related theory and design with the network model in both quantum LiFi and wireless will be given, from which some results will be presented and interpreted for the related applications.

Dr. Yaseera Ismail, University of KwaZulu Natal, Durban, South Africa Title: Quantum Technology for a Quantum Internet Biography: Dr. Ismail is Experimental Physicist working on the development of quantum technology within the quantum research group specifically within the field of quantum communication. She holds a Ph.D. in Physics and had one of her recent publications chosen as editor’s pick highlighting articles of excellent scientific quality. She is Recipient of the 2016 United States TechWomen Emerging Leader award and is 2018 Optical Society of America Ambassador.

Invited Speakers

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Dr. Rajlaxmi Chouhan, Indian Institute of Technology, Jodhpur, India Title: No-reference Image Quality Assessment and Applications Biography: Dr. Rajlaxmi Chouhan received her Ph.D. in August 2015 from the Department of Electronics and Electrical Communication Engineering at the Indian Institute of Technology Kharagpur (India). She is currently Assistant Professor in the Department of Electrical Engineering at Indian Institute of Technology, Jodhpur. Her research interests include image processing, image quality assessment, noise-aided image enhancement and watermarking. She was awarded the IEEE Region 10 (Asia Pacific) WIE Student Volunteer Award in 2014 and the National Award for Best M.Tech. Thesis in Electronics and Electrical Engineering in 2012 by Indian Society of Technical Education, New Delhi. She received the Teaching Excellence Award 2019 at IIT, Jodhpur, and the Late Shri Pralhad P. Chhabria Award for Best Women Professional (Early Career) by HFRC, IEEE India Council and WIE Pune Section in 2019.

Preface

Optical and wireless technologies are advancing at an accelerating rate recently. The traditional approaches to providing high data rates to the masses are transforming and expanding in a way that is beyond our imagination. The challenges in providing uninterrupted data and broadband communications have not changed. Our mission as a technical community is to understand these challenges and find ways to mitigate them. This includes the development and management of appropriate channels, novel devices, new protocols, efficient networks and their integration. Keeping in view the amalgamation of these issues, the proceedings of 4th International Conference on Optical and Wireless Technologies (OWT-2020) is being presented herewith. The conference (OWT-2020) was to be held in the campus of Malaviya National Institute of Technology Jaipur (Institutional Partner) during April 11–12, 2020, but was organized online during October 3–4, 2020, due to outbreak of COVID-19 pandemic and subsequent lockdown as per guidelines of GOI. A total of 150 participants including the invited speakers, contributing authors and attendees participated in the conference. The participants were explored to a broad range of topics critical to our society and industry in the related areas. The conference provided an opportunity to exchange ideas among global leaders and experts from academia and industry in topics like optical materials, optical signal processing and networking, photonic communications systems and networks, all-optical systems, microwave photonics, optical devices for optical communications, nonlinear optics, nanophotonics, software-defined and cognitive radio, signal processing for wireless communications, antenna systems, spectrum management and regulatory issues, vehicular communications, wireless sensor networks, machine-to-machine communications, cellular-WiFi integration, etc. Apart from high-quality contributed paper presented by delegates from all over the country and abroad, the conference participants also witnessed the informative demonstrations and technical sessions from the industry as well as invited talks from renowned experts aimed at advances in these areas. Overall response to the conference was quite encouraging. A large number of papers were received. After a rigorous editorial and review process, 70 papers were invited for presentation during the conference. Among the presented papers, 62 papers were selected for inclusion in the conference proceedings. We are confident that the papers presented in this xix

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proceedings shall provide platform for young as well as experienced professionals to generate new ideas and networking opportunities. The editorial team members would like to extend gratitude and sincere thanks to all contributed authors, reviewers, panelists, local organizing committee members and the session chairs for paying attention to the quality of the publication. We are thankful to our sponsors for generously supporting this event and institutional partner (MNIT Jaipur) for providing all the necessary support and encouragement in this beautiful campus. At last, we pay highest regard to the Irisworld Science & Technology Education and Research (IRISWORLD), a “Not for Profit” society from Jaipur for extending support for financial management of the OWT-2020. Jaipur, India Jaipur, India Jaipur, India Ajmer, India Ho Chi Minh City, Vietnam

Prof. Manish Tiwari Dr. Ravi Kumar Maddila Dr. Amit Kumar Garg Dr. Ashok Kumar Prof. Preecha Yupapin

Acknowledgments

The editors wish to extend heartfelt acknowledgment to all contributing authors, esteemed reviewers for their timely response, members of various organizing committee and production staff whose diligent work puts shape to the OWT-2020 proceedings. We especially thank our dedicated reviewers for their volunteering efforts to check the manuscript thoroughly to maintain the technical quality and for useful suggestions. We also pay our best regards to the faculty members from institutional partners (MNIT Jaipur and Manipal University Jaipur) for extending their enormous assistance during the conference-related assignments, especially to Dr. Dinesh Yadav from Manipal University Jaipur, Dr. Ashok Kumar, Jitendra Deegwal from Government Women Engineering College Ajmer, Mr. Nidhish Tiwari from CIITM Jaipur, Mr. Ramesh Dewanda, Executive Member, Irisworld Society Jaipur, and Sh. Narendra Godara from MNIT Jaipur. Finally, we extend our sincere thanks Springer Nature for agreeing to be our publishing partner. We would specially like to acknowledge the support of Swati Meherishi, Editorial Director and Muskan Jaiswal, Assistant Editor at Springer Nature. Prof. Manish Tiwari Dr. Ravi Kumar Maddila Dr. Amit Kumar Garg Dr. Ashok Kumar Prof. Preecha Yupapin

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Contents

Smart Parking Management System in Smart City . . . . . . . . . . . . . . . . . . . Dhairya Sibal, Aditya Jain, and P. C. Jain

1

Study of Microstrip Antenna Geometry: Effect of Antenna Geometry on Antenna Parameters—A Comprehensive Review . . . . . . . . Brijesh Mishra, Shadab Azam Shiddique, and Vivek Singh

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Effect of Temperature on Incoherently Coupled Dark–Bright Soliton Pair in Photorefractive Crystals . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Aavishkar Katti

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Transmission Analysis of Designed 2D MWC in Hybrid OCDMA System for Local Area Network Application . . . . . . . . . . . . . . . . . . . . . . . . . Madhumita Sarkar, Somali Sikder, and Shila Ghosh

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Optical Code Construction of Balanced Weight Ideal Cross-Correlation Code for Spectral Amplitude Coding Optical CDMA Systems . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Teena Sharma Dual-Band Dual Polarized Circularly Polarized and Linearly Polarized L-Shaped Patch Antenna Loaded with Strip and Square Slot . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Reshmi Dhara, Taraknath Kundu, and Sanjay Kumar Jana

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Analytical Comparison of Various Detection Techniques for SAC-based OCDMA Systems: A Comparative Review . . . . . . . . . . . . . Teena Sharma and M. Ravi Kumar

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Classification of Leaf Disease on Using Triangular Thresholding Method and Machine Learning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Deepak Kumar Saxena, Deepak Jhanwar, and Diwakar Gautam

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Effect of Code and Frequency Index Modulation in MIMO-OFDM-FSO System . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Chinmayee Panda and Urmila Bhanja

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The Structural, Electronic and Optical Properties of Partially Hydrogenated Germanene: A First-Principles Study . . . . . . . . . . . . . . . . . . Routu Santosh and V. Kumar

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Performance Evaluation of Routing Protocols in MANETs with Variation in Pause Time . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 105 Suresh Kumar, Deepak Sharma, and Payal Three-Qubit Implementation of Quantum Fourier Transform for Shor’s Algorithm . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 115 Deepanshu Trivedi, Ankur Saharia, Kamalkishor Choure, Manish Tiwari, Ravi Kumar Maddila, and Ghanshyam Singh Performance Enhancement of Surface Plasmon Resonance (SPR) Structure Using a Sinusoidal Diffraction Grating . . . . . . . . . . . . . . . . . . . . . 123 Manish Jangid, Ankur Saharia, Nitesh Mudgal, Sajai Vir Singh, and Ghanshyam Singh Wide-Band Meander Line Antenna for Ka Band Application . . . . . . . . . . 133 Manan Gupta, Ashok Kumar, Amrita Dixit, and Arjun Kumar RETRACTED CHAPTER: Design of Silicon-on-Insulator Based Mode Splitters with Asymmetrical Variation of Slots . . . . . . . . . . . . . . . . . . 141 Neha Choudhary, Veer Chandra, and Rakesh Ranjan Human Body Monitoring Wearable Antenna . . . . . . . . . . . . . . . . . . . . . . . . . 151 Vani Sadadiwala, Kashish Mahindroo, Vimlesh Singh, Priyanka Bansal, and Sarthak Singhal On Design of Airborne Radomes with Brent’s Method for Radiation Pattern Peak Detection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 163 Romesh Srivastava, Aparna Parameswaran, and Hrishikesh S. Sonalikar Wideband Bandpass Substrate Integrated Waveguide (SIW) Filter for C Band Application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 173 Amrita Dixit, Ashok Kumar, Ashok Kumar, and Arjun Kumar Design of Polarized 2 × 2 MIMO Antenna Using Partially Stepped Ground . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 181 Harsha Prabha Paliwal, Navneet Agrawal, and Suruti Gupta Simulation Studies on Force Sensor Using PDMS Coated Fiber Bragg Grating for Robot-Assisted Surgery . . . . . . . . . . . . . . . . . . . . . . . . . . . 189 Dinesh Lakshmanan and Srijith Kanakambaran Design and Analysis of E Shaped Microstrip Patch Antenna with Defected Ground Structure for Improvement of Gain and Bandwidth . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 195 Ajay Singh, Sunil Joshi, Dhananjay Dashora, Lokesh Lohar, and Harsha Prabha Paliwal

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Microwave Imaging Breast Cancer Detection Techniques: A Brief Review . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 203 Monika Mathur, D. Mathur, G. Singh, S. K. Bhatnagar, Harshal Nigam, and Mukesh Arora Implementation of XOR Gate Using a Nonlinear Polarization Rotation in Highly Nonlinear Fiber . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 211 Vasundhara, Lovkesh, and Surinder Singh Designing of Hybrid Photonic Crystal Fiber for Better Filter Characteristics Using Gallium Nitride . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 219 Shubham Sharma, Sajai Vir Singh, Ankit Agarwal, Nitesh Mudgal, Manish Tiwari, Ravi Kumar Maddila, and Ghanshyam Singh Silicon-On-Insulator Photonics Waveguide Design for Near-IR Evanescent Field-Based Blood Sensor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 229 Veer Chandra, Neha Choudhary, and Rakesh Ranjan Analysis of Upper Aortic Blood Vessels as a Data Communication Channel . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 239 Tanmay Deshmukh, Kauser Husainee, and Prabhat Kumar Sharma A Ground Plane Modified Broadband Circularly Polarized Patch Antenna for Wireless Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 247 G. Anjaneyulu, T. A. N. S. N. Varma, and J. Siddartha Varma Deterministic Two Qubit iSWAP Gate Using a Resonator as Coupler . . . 253 Amit Kumar Sharma and Ritu Sharma Recent Advancement in High Speed and Secure Quantum Key Distribution: A Review . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 259 Kamal Kishor Choure, Ankur Saharia, Nitesh Mudgal, Manish Tiwari, and Ghanshyam Singh Miniaturization and Gain Enhancement of Rectangular Patch Antenna Using CSRR . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 269 Shubhangi Palekar and Neeraj Rao Preventing DoS Attack in VANET by Novel RBS-IP-CHOCK Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 281 Karan Verma, Hemant Kumar Saini, and Ajay K. Sharma Asymmetric 1 × 4 Switch Based on MZI Using Lithium Niobate (LiNbO3 ) for O-Band Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 295 Akanksha Gupta and Kamalesh Kumar Sharma SRR Loaded WLAN Band-Notched UWB MIMO Antenna with Spatial Diversity Characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 303 Summaiyya Saleem, Sarita Kumari, Dinesh Yadav, and Deepak Bhatnagar

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Dispersion Engineered AsSe2 Based Chalcogenide Photonic Crystal Fiber for MIR Region Supercontinuum Generation . . . . . . . . . . . 311 Vaibhav Gupta, Jaiverdahan, Vinay Kanungo, Rukhsar Zafar, Sandeep Vyas, Anand Nayyar, and Ghanshyam Singh Design of a Nanocavity Photonic Crystal Structure for Biosensing Application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 321 Ankit Agarwal, Nitesh Mudgal, Sourabh Sahu, Ghanshyam Singh, and S. K. Bhatnagar AMC-Loaded Compact CPW-Fed Monopole Antenna for Ka-Band Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 331 Ashok Kumar, Amrita Dixit, Ashok Kumar, and Arjun Kumar Design of Compact Microstrip Patch Antenna Using SRR Metamaterial for Wireless Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 339 Swapna Kumari Budarapu, Kanakavalli Harshasri, Y. Ravi Kumar, and B. Rajendra Naik Effect of Number of Users and Number of Clusters Using Distributed Cooperative Spectrum Sensing Over Hoyt Fading Channel . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 349 Yashaswini Sharma, Girraj Sharma, Ritu Sharma, and K. K. Sharma Designing of Low-Cost Energy Harvesting Antenna System for 2.4 GHz ISM Band . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 359 Kirti Kumar Yadav, Tanvi Joshi, Kartik Patidar, and Dinesh Yadav Study of Performance Analysis of Substrate Integrated Waveguide (SIW) Antennas for X-Band Satellite Communication . . . . . . . . . . . . . . . . . 369 Kartik Patidar, Kirti Kumar Yadav, Tanvi Joshi, and Dinesh Yadav CPW-Fed Dual-Sense Cross-Shaped Broadband Circularly Polarized Antenna for Wireless and Satellite Application . . . . . . . . . . . . . . 377 Monika Jangid, M. M. Sharma, and Jaiverdhan Internet of Things Framework for Device Integration in Automation Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 387 Amitkumar Parmar and Amit M. Joshi Frequency Hopping Patterns for Low Probability of Intercept (LPI) Radars Using Costas Arrays . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 401 T. A. N. S. N. Varma and Anjaneyulu Gera Printed Monopole Slot Antenna Inspired by Metamaterial Unit Cell for Wireless Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 413 Swati Sharma and Rekha Mehra

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Reconfigurable Hexa-band Antenna with Defected Ground Structure for 2.1–3.4 GHz Band Applications . . . . . . . . . . . . . . . . . . . . . . . . 425 Shadab Azam Siddique, Brijesh Mishra, Rakesh Kumar Singh, Vivek Singh, and Sanjay Kumar Soni A Compact Design of Square Ring-Shaped Microstrip Monopole Antenna for Wireless Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 435 Ajay Dadhich, Megha Agarwal, J. K. Deegwal, and M. M. Sharma 1D Photonic Crystal Waveguide Based Biosensor for Skin Cancer Detection Application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 443 Sanchit Kundal, Abhinav Bhatnagar, and Ritu Sharma Ka Band Circularly Polarized Antenna with Defected Ground for Close Range Military Radar Target Applications . . . . . . . . . . . . . . . . . . 451 Priya Kaith and M. M. Sharma Design and Simulation of a Photonic Crystal-Based 2-D Octagonal-Shaped Optical Drop Filter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 457 Manish Kumar Pandey, Ritu Sharma, and Manish Jangid Frequency Reconfigurable/UWB Slot-Antenna with Switchable Resonant Function . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 465 M. M. Sharma, Indra Bhooshan Sharma, and Joohi Garg Optimization of Sensing Time for Efficient Spectrum Utilization in NOMA Based Cognitive Radio Networks . . . . . . . . . . . . . . . . . . . . . . . . . . 471 Deepika Rajpoot and Pankaj Verma A CPW-Fed Annular Shape Antenna with Asymmetrical Hexagonal Slot Loaded Defected Ground Plane for Ultra-Wideband Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 483 Ajay Kumar Dwivedi, Brijesh Mishra, Chandrabhan, Shadab Azam Siddique, and Vivek Singh Ultra-Wide Band Microstrip Patch Antenna for Millimetre-Wave Band Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 491 Aditi Chauhan, Utkarsh Jain, Aakash Warke, Manan Gupta, Ashok Kumar, Amrita Dixit, and Arjun Kumar A Review on Attack and Security Tools at Network Layer of IoT . . . . . . . 497 Vidur Agarwal, Preeti Mishra, Sachin Kumar, and Emmanuel S. Pilli A Compact Design of Stub-Loaded Multiband Microstrip Monopole Antenna for WLAN and WiMAX Applications . . . . . . . . . . . . . 507 Ajay Dadhich, J. K. Deegwal, and M. M. Sharma Smart Transportation for Warehouses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 515 T. Jaya Sankar and P. C. Jain

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Mitigating Nonlinear Effects in 16 Channel WDM Radio Over Fiber System with Dispersion Compensation Fiber and Fiber Bragg Grating Combination . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 527 Suresh Kumar, Shagun Singh, and Payal An Overarching Review on Taxonomy of Routing Metric in Concurrence with Trust and Security for CRAHN . . . . . . . . . . . . . . . . . 541 Pooja Ahuja, Preeti Sethi, and Naresh Chauhan Multiband and Wideband MIMO Antenna for X and Ku Band Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 557 Sweta Singh, Brijesh Mishra, Karunesh Shrivastva, Aditya Kumar Singh, and Rajeev Singh Channel Capacity of Underwater Channel Using OCDMA System . . . . . 567 Ajay Yadav, Ashok Kumar, Jitendra Kumar Deegwal, Ghanshyam Singh, and Arjun Kumar Design and Performance Analysis of an Encrypted Two-Dimensional Coding Technique for Optical CDMA . . . . . . . . . . . . . . . 573 Urmila Bhanja Optimization of Physical Parameters of Single-Beam Vibrational Piezoelectric Energy Harvester . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 585 Namrata Saxena, Neha Yaragatti, Shishir Sharma, and Ritu Sharma Retraction Note to: Design of Silicon-on-Insulator Based Mode Splitters with Asymmetrical Variation of Slots . . . . . . . . . . . . . . . . . . . . . . . Neha Choudhary, Veer Chandra, and Rakesh Ranjan

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About the Editors

Dr. Manish Tiwari received Ph.D. in Electronics and Communications Engineering (ECE) in the field of Photonics from MNIT Jaipur, India. Presently, he is Professor in Department of ECE at Manipal University Jaipur. He has been visiting researcher to City University, London under UKIERI project in Microstructured Optical Fibers between 2010 and 2011 and Tsinghua University, Beijing in 2016. Dr. Tiwari has presented talk in PolyU—Hong Kong, KMUT— Bangkok, Kasetsart University—Bangkok, City University—London and several UKIERI workshops. He has also served on panel of experts in various workshops by CSTT, MHRD, Government of India. His current research interest includes Micro/Nano-structure photonic devices, nonlinear optics and photonic crystal fibers. Dr. Ravi Kumar Maddila did his Bachelors (Electronics and Communication Engineering) in 2000 from Utkal University, India. He obtained his Master’s (Opto Electronics and Laser Technology) in 2005 from International School of Photonics, Cochin University of Science and Technology, Kochi, India. His Ph.D. on Optical CDMA codes was awarded by the G. S. Sanyal School of Telecommunications, Indian Institute of Technology (IIT) Kharagpur, India. His research interests include optical communication, integrated optics, visible light communication and daylighting. He has published more than 40 articles in various journals and conferences and is a reviewer of several journals and conferences. His publications have been cited by more than 140 articles. He has guided two Ph.D. scholars, xxix

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About the Editors

more than 15 M.Tech. theses and is currently supervising six Ph.D. scholars. Dr. Amit Kumar Garg is an assistant professor in Department of Electronics and Communications Engineering, Indian Institute of Information Technology, Kota, India. He obtained his Ph.D. from the Malaviya National Institute of Technology (MNIT), Jaipur, India in 2018. He received B.E. and M.Tech. in ECE from the University of Rajasthan and MNIT Jaipur in 2009 and 2013, respectively. His current research interest includes fiber optics and photonics including smart, flexible, energy-efficient and latency-aware optical access networks and high-speed gigabit passive optical networks. He also works on the latest information and communication technologies for developing smart cities, buildings, etc. He is an active member of SPIE and OSA. Dr. Ashok Kumar received his Ph.D. in Electronics and Communication Engineering from Rajasthan Technical University Kota, India (2019). He had completed his M.Tech. degree in Communication Stream from Malaviya National Institute of Technology Jaipur (MNIT), India (2011) and B.E. in Electronics and Communication Engineering from University of Rajasthan, Jaipur, India (2008). Presently, he is working as Assistant Professor in the Department of Electronics and Communication Engineering at Government Mahila Engineering College Ajmer, India. He is the author/co-author of more than 50 research papers published in the refereed SCI/Scopus indexed international/national journals and conferences. His current research interest includes design and applications of microwave planar antennas, circularly polarized antennas, reconfigurable antennas, UWB antennas, VANETs etc.

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Dr. Preecha Yupapin received his Ph.D. degree in electrical engineering from the University of London, UK in 1993. He is currently the full Professor in the Computational Optics Research Group, Advanced Institute of Materials Science and member in the Faculty of Applied Sciences, Ton Duc Thang University, Ho Chi Minh City, Vietnam. He has 650 Scopus index, 15 international book editions. His current research interests are nano-devices and circuits, microring resonator, soliton communication, optical motor, quantum technologies, and quantum meditation.

Smart Parking Management System in Smart City Dhairya Sibal, Aditya Jain, and P. C. Jain

Abstract In today’s world, where we as humans are constantly aiming towards improving machines and giving them a brain of their own, it is about time that our traffic trends and parking systems become as efficient and automated too. The aim of this paper is to acknowledge the increasing traffic, unorganized parking, and the time wasted while searching for a parking spot in an urban high vehicle density area and make an efficient parking system to avoid the same. Our approach includes the live navigation through our android app, which solves the problem of searching for parking lots in the area. Secondly, an android application is used to show the slots available/occupied in a parking lot. Thirdly, nodes at the parking slot help in making the parking organized. Lastly, an OTP-driven convenient payment method is made for fast functioning of parking exit systems. Keywords IoT · Nodes · Cloud · Smart parking · Automation · Navigation · OTP · Database · QR

1 Introduction A smart city aims to improve quality of life for its citizen by harnessing technology to connect infrastructure, resources, and services, making the municipality safer and more sustainable, liveable, workable, and competitive. Smart cities should be able to monitor and automate operations and applications in utilities, buildings, and infrastructure in real time. To meet these challenges, smart cities require a range of communication networks. With a growing number of vehicles on the roads and poor D. Sibal · A. Jain · P. C. Jain (B) Electrical Engineering Department, School of Engineering, Shiv Nadar University, Greater Noida, India e-mail: [email protected] D. Sibal e-mail: [email protected] A. Jain e-mail: [email protected] © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_1

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roadway systems, traffic congestion and not being able to find a parking spot have become major issues. Parking in an urban environment is becoming difficult at times, constant circling round the block looking for a parking place. Drivers spend lot of time looking for parking ads in wasted time, fuel, and CO emission. Smart parking management system can find the vacant parking slot for a vehicle at different public places. Wireless sensors are embedded into parking slots transmitting data on the timing and duration of the free space via local signal processors to a central parking management. Smart parking reduces congestion, decreases vehicle emissions, lower enforcement costs, and reduces driver stress. To improve and better govern parking systems, many automated models/systems have been developed for speed, convenience, and better efficiency for parking. These systems have evolved with time from being offline systems with screens displaying free parking slots to use SMS systems for reserving a slot and RFIDs for entry/exit. The current systems, however, have started using Websites, android applications, IoT, artificial intelligence, and machine learning for creating a better performance for parking systems and making them smart in general. While these smart parking models have many basic things in common as mentioned above, the difference lies in ideation and application of these algorithms and the efficiency of resources used. In the past decade, many papers have improved and built upon earlier ideas for a faster and convenient way to build parking systems, but the chaos, confusion and frustration because of unorganized parking in areas with high vehicle density have been left unaddressed. This is where this paper comes into play by providing an efficient solution for not only faster and convenient parking and payments systems but also for organized and stress-free parking which can be made more and more cost-effective over time. Figure 1 shows the sequential steps in which this model works, and Fig. 2 depicts the prototype of this model with lesser parking slots.

2 Related Work Starting from the paper by Srikanth et al. in 2009 [1] proposes the use of LCD screens to display the number of parking slots available in an area of a parking lot and other relevant information, we can see that the information is limited to that very parking lot. The solution proposed by Wei et al. in 2012 [2] used RFID-based cards that were given to the users, and accordingly, a Website is maintained for a record of free slots available. This approach is time consuming and does not focus on organized parking unlike our approach which focuses on organized parking and involves simply providing an OTP while exiting the parking lot. Orrie et al. [3] proposed an android application-based approach in 2015 similar to ours, but it still failed to include the concept of organized parking in the provided space to avoid congestions. Their payment system also includes RFID system, whereas OTP-based systems are a better approach. The proposal of Aydin et al. in 2017 [4] makes use of a magnetic sensor to find out whether there is a car parked or not; organized parking was not taken into consideration. Rizvi Syed et al. [5] have proposed a method which

Smart Parking Management System in Smart City

Fig. 1 Flowchart of smart parking system

Fig. 2 Prototype model of smart parking system with two slots

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they call ASPIRE. It uses IoT, and in this method significant consideration is given to the driver’s parking preferences such as type of parking sought, the maximum price that the parker is willing to pay, and the maximum time to walk from parking space to the destination that the driver can tolerate. In today’s world what most of the people need is “on the go” parking facilities that are quick and easy to use, and the method proposed in [5] is not suitable to be implemented in many places which requires quick, time-saving solutions. Lin et al. [6] used AI and ML in the smart parking scenario which can be included in our current project for predicting traffic congestion and user behaviour. Athira et al. [7] proposed the use of machine learning to find out free parking slots using CCTV camera footage, but alignment of the camera and accuracy of their algorithm have to be perfect for this method to work; this method cannot be implemented everywhere, and the cost of the cameras exceeds the low-cost sensors that we are using by a great margin. Promy and Islam [8] proposed a manual system of marking a parking slot empty by the user when he/she leaves it and displays it to everyone using an android app. This also allows one user to call another; this approach is a little inconvenient and time consuming. An automated sensor-based approach is more feasible, cost-effective, and convenient. All the papers on this topic in the past show a trend that leads to automation and indicate towards proving that smart parking systems can be made better by the inclusion of AI and ML in the near future.

3 Methodology 3.1 Nodes and Cloud One of the crucial steps in this project is to establish communication between the microcontroller (NodeMCU) and the database (Firebase) which is done by the command: Firebase.begin (FIREBASE_HOST, FIREBASE_AUTH); In our proposed model, two slots are being operated from one microcontroller (Fig. 2). The data from different sensors (ultrasonic and LDR modules) is received and processed by the microcontroller. Based on the data received and the conditions that are defined for the intended working of the model in the microcontroller, a state for the operated slot is selected. If Laser-LDR pairs of slot 1 turn “LOW” and slot 1 ultrasonic sensor turns a distance below the threshold set for the ground clearance level of the vehicles targeted, then a state of “Occupied” is set for slot 1 in the database. If Laser-LDR pairs turn “LOW” and ultrasonic sensor turns a distance more than the threshold, then a state of “Free” is set for slot 1. Lastly, if Laser-LDR pairs turn “HIGH”, a state of “Tripped” is set for slot 1. The above method follows

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for slot 2 as well. The states set for slot 1 and slot 2 are now updated on database by using the command: Firebase.set (firebaseData, “slot1”, slot1);

3.2 Cloud and Android Application First step of using the application involves signing in, which generates a unique UID in the authentication database corresponding to a particular user; this UID is used to store other information of that user in the real-time database. Once the user logs in and opens the “Details” page corresponding to a particular parking lot, the application sends a GET request to the database requesting for the state of the slots in that particular parking lot; the data is returned by the database and is displayed on the details page of the application (Fig. 5a). After parking the vehicle, user scans the QR code corresponding to that slot (a unique QR code is attached with every parking slot), the application again sends a GET request asking the database the state of that particular slot and displays a message accordingly under the QR code scanner in the application (Fig. 6a–c). If the state is “Occupied”, then the application generates a unique random 6-digit OTP. The application then makes a JSON object which contains email, OTP, slot occupied and timestamp of the current user. Then, the application sends a POST request to the database uploading the JSON object previously created on the database. There is a two-way communication between the android application and the database.

3.3 Webpage and Cloud The webpage is made to provide a convenient payment system by simply providing the OTP while exiting a parking lot. The OTP generated by the android application is unique for each user. When the OTP is entered in the search bar provided on the webpage and is submitted, the backend searches the database for that OTP. If the OTP exists, the database returns the user email, timestamp, slot corresponding to that OTP, and the information is displayed on the webpage (Fig. 7). The total amount can be calculated by using the timestamp returned by the database. If the OTP does not exist, it displays “OTP does not match” on the webpage.

4 Results The smart parking system worked with good efficiency and cost minimization with one microcontroller being used for two slots. The first result obtained is in the

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database after the registration/login is done by the user (Fig. 3). The next result obtained will be the real-time navigation to slot selected by the user (Fig. 4).

Fig. 3 User collection in database

Fig. 4 Navigation to parking 1, Shiv Nadar University

Smart Parking Management System in Smart City

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Fig. 5 a When no vehicle is parked at slot 1. b When the vehicle is improperly parked at slot 1. c When the vehicle is properly parked at slot 1

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Fig. 6 a Message below QR scanner when no vehicle is parked at slot 1. b Message below QR scanner when vehicle is not parked properly. c Message below QR scanner when vehicle is parked properly

Fig. 7 Details of the user after searching for the OTP provided

After a user navigates his/her way to the preferred/nearest parking slot and tries to park the vehicle, one of the following three results is obtained: 1. 2. 3.

The application shows “Free” for that slot if no vehicle occupies that slot (Fig. 5a). The application shows “Tripped” for that slot if the vehicle is parked improperly (Fig. 5b). The application shows “Occupied” for that slot if the vehicle is parked properly (Fig. 5c).

The next result is obtained from the QR scanner. When the user reaches the slot, there are following three scenarios possible with their consequent results:

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If the user has not parked the vehicle but scans the QR code or the user scans the OTP before parking the vehicle, the application shows “No vehicle parked at SlotX” (Fig. 6a). If the vehicle is not parked properly (in an organized way), the application shows “Vehicle is not parked properly at SlotX” (Fig. 6b). If the vehicle is parked properly, the application generates an OTP and displays it below the QR scanner (Fig. 6c).

The last result is obtained through the database from the webpage which shows the details of the user after searching for the OTP provided by the user. This would include a time stamp for the time the vehicle was parked to calculate the total payment (Fig. 7).

5 Conclusions With the ever-increasing traffic and pollution, the conventional ways of parking systems are becoming obsolete and time consuming. There is an urgent need of smart, efficient, and effective ways to tackle these problems. The work done in this field by other people observed that the initiative taken up to make the parking systems was smart but lacks the focus of organized parking and quick payment options to save time and avoid parking congestions. This paper has taken up above problems and is able to solve them up to an acceptable level. This paper with the help of Laser-LDR pairs and ultrasonic sensors makes sure that the parking is properly done, and the inclusion of QR codes and OTP has proved to be useful in reducing the time taken while making payment and exiting the parking lot. Other features like navigation to the nearest parking lot and user login system have made the implementation more secure and useful. There are still huge possibilities of improvement in the system by the inclusion of artificial intelligence and machine learning. Features like predicting user behaviour and smooth flow of vehicles in the parking lot without congestion can be implemented using AI and ML.

References 1. Srikanth SV, Pramod PJ, Dileep KP, Tapas S, Patil Mahesh U, Babu Sarat Chandra N (2009) Design and implementation of a prototype smart PARKing (SPARK) system using wireless sensor networks. In: 2009 International conference on advanced information networking and applications workshops 2. Wei L, Wu Q, Mei Y, Wei D, Bo L, Gao R (2012) Design and implementation of smart parking management system based on RFID and internet. In: 2012 International conference on control engineering and communication technology 3. Orrie O, Silva B, Hancke GP (2015) A wireless smart parking system. In: IECON 2015—41st Annual conference of the IEEE industrial electronics society

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4. Aydin I, Mehmet K, Ebru K (2017) A navigation and reservation based smart parking platform using genetic optimization for smart cities. In: 2017 5th International Istanbul smart grid and cities congress and fair 5. Promy N, Islam S (2019) A smart android based parking system to reduce the traffic congestion of Dhaka city. In: 21st International conference on advanced communication technology 6. Athira A, Lekshmi S, Pooja V, Boby K (2019) Smart parking system based on optical character recognition. In: Proceedings of the third international conference on trends in electronics and informatics 7. Lin J, Chen S-Y, Chng C-Y, Chen G (2019) Smart parking algorithm based on driver behaviour and parking traffic predictions. IEEE Access 7 8. Rizvi Syed R, Susan Z, Stephan O (2019) ASPIRE: an agent oriented smart parking recommendation system for smart cities. IEEE Intell Transp Syst Mag

Study of Microstrip Antenna Geometry: Effect of Antenna Geometry on Antenna Parameters—A Comprehensive Review Brijesh Mishra, Shadab Azam Shiddique, and Vivek Singh

Abstract The antenna parameters are dependent on the substrate material, shape and size of the patch antenna. The shape of the antenna is intuitively conceived, and different physical parameters of the antenna can be altered and optimized for a particular application or for particular operating frequencies. In this article, effect of antenna geometry (specifically rectangular, circular, triangular, elliptical, trapezium, pentagonal and hexagonal shapes are encountered) on antenna parameters in terms of antenna size, dielectric materials, resonating band, peak gain, radiation efficiency, simulating tools and their applications is critically reviewed. This review will aid as a guide in the future applications to design/select a transmission-efficient regular shape printed antenna, which needs to be taken to the next level advancement in the field of microwave communication. Keywords Antenna geometry · Review · Antenna parameters · Antenna size · Dielectric material

1 Introduction Microstrip antenna consists of radiating patch on the one side of dielectric substrate and ground plane on the other side of dielectric substrate. Radiating patch can be of any shape, but the regular shapes can be easily analyzed, and its characteristics are easy to understand. Different patch structures such as rectangular, circular, elliptical, triangular, hexagonal, circular ring, disc sector and many more shapes have been presented to enhance or improve the antenna characteristics. New technology of mobile system requires wideband, multiband and UWB band antennas to satisfy mobile and wireless services, and integration of such antennas in small devices is B. Mishra (B) · S. A. Shiddique Department of Electronics and Communication, Madan Mohan Malaviya University of Technology, Gorakhpur, U.P. 273010, India V. Singh Department of Electronics and Communication, Shambhunath Institute of Engineering and Technology, Prayagraj, U.P., India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_2

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highly desirable. Antenna performance can be changed by changing the shape and the size of the antenna. Research activities in last two decades to explore new geometries of antenna for modern communication systems have been enhanced [1]. The wired antennas originally were reported by Joseph Henry in 1842; however, the history of wireless antenna dates back to the period of James Clerk Maxwell who predicted the existence of electromagnetic waves in 1864. In 1885, Thomas Alva Edison received a patent of communication system using vertical top-loaded antenna. Later on, Heinrich Hertz in 1888 generated, transmitted and received electromagnetic waves successfully and validated the theoretical predictions of Maxwell. The Indian scientist Jagdish Chandra Bose conducted experiments on millimeter wavelength transmission in 1895 and developed the first horn antenna in 1897. Marconi demonstrated the ability of radio waves to provide continuous communication with ships sailing the English Channel in 1897. In 1901, he conducted a transatlantic experiment proving the possibility of long-distance (more than 300 km) wireless communication. The concept of microstrip patch antenna (printed antenna) was originally proposed in 1953 by Deschamps [2]. However, practical antennas were implemented by Munson and Howell after 20 years in 1970s [3, 4]. A comprehensive review of theoretical analysis techniques of microstrip patch antenna is reported by Carver in 1981 [5]. In this section, we present a comprehensive review accounting the antenna geometry, the substrate material, the resonating band and fractional bandwidth, the antenna gain, the numerical techniques adopted for the analysis and design and the applications. Information mentioned in the proceeding sections provide a help to select the different shapes of antenna in terms of resonating band, bandwidth, gain, dimensions and materials suitable for various applications in modern communication systems. However, this review article is confined to the shapes of the radiating patch only and structures on the ground plane is omited as it will make the article more voluminous. A detailed review on shape of the ground plane or defected ground structures is omitted as it will make the research paper more voluminous and time consuming for readers.

2 Rectangular Shape Antennas The works describes the reported in [6–15] rectangular-shaped antennas. Detailed information of antenna specifications, parameters and applications is presented. From reported papers, it is observed that antennas [6, 7, 11] resonate at dual frequencies, while antennas [8–10, 12–15] resonate at single frequency. The antenna reported in [9] has the largest planar area, whereas antenna [5] has the minimum planar area and maximum simulated bandwidth of 350 GHz. Minimum simulated bandwidth of 0.034 GHz is reported for [14], and minimum simulated peak gain of 2.36 dBi is reported for the antenna [11]. A minimum measured bandwidth of 0.04 GHz and a maximum measured bandwidth of 2.43 GHz are reported [13]. A minimum measured peak gain of 3.9 dBi [8] and a maximum peak gain of 9.3 dBi [13] are reported. However, antenna [15] has been designed with compact size (1 × 1 mm2 ) by using

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electromagnetic crystal and resonating in (0.6–0.95) THz frequency band. Among the simulated results, it is found that the minimum planar area structure achieves maximum bandwidth and the large structures [6, 14] usually have low bandwidths.

3 Circular Shape Antennas This section depicts the details of circular patch antennas. The antenna reported in [16] has the largest planar area, and antenna reported in [17] has the minimum planar area. Antennas [16, 18–22] and antennas [23–27] have been designed for singleand dual-band operation, respectively, whereas antenna [17] has been designed for triple-band operation. The minimum measured bandwidth of 0.03 GHz is observed in [25], and the maximum measured bandwidth of 7.5 GHz is observed in [23]. The maximum measured peak gain of 9.7 dBi is observed in [26], whereas minimum measured peak gain is observed in [18]. We observe a difference of a factor of 0.244 only between the antennas with maximum [16] and minimum [17] planar areas; however, it is difficult to co-relate antenna bandwidths of [16] and [17] with its size as the substrate materials used are different as well as the operating frequencies are different. The applications of such antenna include wireless communication, remote sensing to RFIDs.

4 Triangular Shape Antennas Antennas [28–41] show single resonance behavior, antennas [42–47] resonates at dual bands, antennas [48, 49] resonates at three frequencies, and antenna [50] has five resonating bands. The authors [28–50] have made efforts to reduce the overall size of antenna; the authors [44] have reported a very compact (12 × 20 mm2 ) dualband antenna for UWB, WLAN and Bluetooth applications. The antenna reported in [38] has the largest planar area (150 × 150 mm2 ). Antenna reported in [44] resonates at dual bands, viz. 2.4–2.52 GHz and 3.2–10.6 GHz, whereas antenna reported in [38] resonates at single band between 0.854 and 1.102 GHz which clearly demonstrates that at the lower frequency ranges the antenna size does not largely affect the resonating frequencies. As the size is increased in [38], we observe the absence of dual band, and further at frequencies beyond 3.0 GHz, we observed larger bandwidth in smaller antenna [44]. The antennas reported in [34, 38, 44] are designed on FR-4 substrate, but the highest impedance bandwidth is observed in the antenna [34]. Antennas [28–50] are designed and fabricated, and their simulated results are verified experimentally. Simulated maximum peak gain of 9.24 dBi and measured maximum peak gain of 7 dBi are reported in [33, 42], respectively. From the perusal of [28–50], the reader can find the applications of antenna in (0.854–12.4) GHz frequency range.

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5 Elliptical Shape Antennas The different microstrip patch antennas of elliptical shape are presented in [51– 57], and comparative overview is shown in terms of antenna performance. Antennas presented in [51–57] have single resonating frequencies except antenna [53] which is resonating at three different frequencies and is useful for C-band wireless applications. A compact (25 × 28 mm2 ) ultra-wideband (3–11) GHz with a notch band (5.1–5.9) GHz antenna [55] has the smallest antenna size as compared to other reported antennas. The antenna reported in [51] has the maximum planar area (300 × 300 mm2 ) among the elliptical shaped antennas reported in [51–57]. A difference of 1.94 GHz is observed between the bandwidths of antennas reported in [51] and [55] with antenna having larger planar area showing larger bandwidth. A maximum/minimum measured peak gain of 6.6 dBi is obtained in antenna [52, 53]. Such antennas can be used in modern wireless communication systems, WiMAX and UWB applications.

6 Trapezium Shape Antennas The antennas reported in [58–64] depict a comparative study of the trapezium shape antenna. The antenna [59] operates in two bands, while the rest of the antennas are designed for single-band operation. Antenna [58] has been found to be the smallest antenna (26 × 20 mm2 ), and antenna [60] has the maximum planar area (60 × 50 mm2 ) among all the antennas reported in [58–64]. The 8.24 dBi measured peak gain is seen in antenna [58], which is maximum in the reported antennas [58–64]. The maximum measured bandwidth of 18.9 GHz is observed in [58], whereas minimum measured bandwidth of 0.39 GHz is observed in [59]. However, it is difficult to establish a correlation in bandwidth and size of the antennas reported in [58, 59] as they are fabricated on different substrate materials and operating at different frequencies.

7 Pentagonal Shape Antennas The measured bandwidth one of the bands of antenna reported in [65] has the minimum bandwidth of 0.38 GHz, whereas antenna reported in [66] has the maximum bandwidth of 12.5 GHz. Bandwidth ratio of antenna with maximum planar area [67] and minimum planar area [68] is calculated as 2.07. However, it is difficult to suggest any relation between the two antennas [67, 68] as the substrate materials are different. But among the antenna fabricated using FR4 substrate [65, 66, 68–72], the antenna reported in [66] has the largest planar area of 3042.10 mm2 , whereas the antenna reported in [68] has the minimum planar area of 625 mm2 . Clearly, the antenna with largest planar area [66] has the measured bandwidth of 12.5 GHz as compared

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to antenna [68] which has measured bandwidth of 4.17 GHz. Therefore, it can be suggested in case of pentagonal shape antenna that a larger bandwidth is observed if the planar area of the radiating patch is large. A maximum measured peak gain of 6.2 dBi and simulated peak gain of 8.7 dBi are observed in [65, 66], respectively. The compact size pentagonal shape antenna is reported [68] for WLAN, WiMAX and HIPERLAN applications.

8 Hexagonal Shape Antennas The antennas [73–83] present a comparative overview of hexagonal shape antennas operating in the frequency range of 1.5–35 GHz. The antenna reported in [77] has the minimum planar area (575 mm2 ), whereas antenna [78] has the maximum planar area (45,000 mm2 ). The antenna [73] reports a maximum measured bandwidth of 32 GHz, whereas minimum measured bandwidth of 0.069/0.07 GHz is reported in references [81, 83]. A measured bandwidth of 9.9 GHz is observed in the antenna [77] with minimum planar area. In hexagonal shape antennas, even the antenna with largest planar area [78] fails to achieve large bandwidth, which is restricted to 0.85 GHz, but we observe a larger bandwidth (9.9 GHz) in case of an antenna with minimum planar area [77]. We further observe in case of hexagonal antennas reported in [73–83] that in general as the size of the antenna increases the bandwidth decreases.

9 Conclusions In the smart world, all the microwave devices are moving toward the miniaturization and economically to be cheap. The shape of the antenna is intuitively conceived, and antenna parameters can be predicted or altered for specific applications. From the above discussion, it is concluded that the circular antenna [17] has very small size (144 mm2 ) with RT/Duroid substrate, triangular antenna [34] has a larger bandwidth (31.94 GHz), circular antenna [20] has a larger gain (12 dBi) and circular antenna [17] has high radiation efficiency as compared to reported papers [6–83]. Most of the patch antennas among [6–83] have been designed and simulated through HFSS, CST and IE3D electromagnetic tool, but finite element method (FEM)-based tool HFSS is currently recommended by researchers for accurate results world-wide. This paper summarized the performance of regular shape antennas up to operating frequency of 35 GHz. The resonating frequency of rectangular, circular and elliptical shape antenna has theoretically been predicted yet. The behavior of microstrip antenna may alter with the introduction of slots, notches, slits, parasitic elements, stacking, defected ground, meandered line, split-ring resonator (SRR), array and the use of different feeding techniques. From these techniques, the effective resonating

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elements (R, L and C) of microstrip patch antenna are optimized and matched with its characteristics impedance for desired applications.

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Effect of Temperature on Incoherently Coupled Dark–Bright Soliton Pair in Photorefractive Crystals Aavishkar Katti

Abstract We study the consequence of temperature variation of the photorefractive crystal on the self-trapping of an incoherently coupled dark–bright soliton pair in biased centrosymmetric photorefractive crystal for the first time. A variation in temperature changes the value of the dielectric constant and the dark irradiance, hence affecting the self-trapping nonlinearity. We find that the soliton width increases with an increase in temperature. Potassium lithium tantalate niobate (KLTN) has been considered as a relevant example to illustrate our results. Keywords Optical solitons · Bright–dark soliton pair · Photorefractive crystal

1 Introduction Photorefractive solitons, first discovered in 1990s, have since been in the focus for continued research [1, 2]. They are interesting because they have a saturating nonlinearity and can be realized at low laser powers. Illumination with a nonuniform light beam results in the development of a space charge field which consequently forms a refractive index waveguide due to the (linear or quadratic) electro-optic effect resulting in a stable self-trapped soliton. When two incoherent solitons with an identical wavelength and polarization state travel parallel to each other, they both create a combined refractive index waveguide which guides both the beams simultaneously resulting in an incoherently coupled soliton pair. There are diverse variants of spatial solitons which have been studied in photorefractive materials like screening solitons [3, 4], screening photovoltaic solitons [5], photovoltaic solitons [6], pyroelectric solitons [7] among others. The fact that the space charge field can be formed by the drift current due to multiple phenomena: an external bias, a photovoltaic field, a transitory pyroelectric field or any combination of these results in an assortment of photorefractive solitons. Similarly, there have been studies about incoherently coupled solitons The author is currently with: Dr. Vishwanath Karad MIT World Peace University, Pune, Maharashtra, 411038, India A. Katti (B) Department of Physics, Banasthali Vidyapith, Newai (Tonk), Rajasthan 304022, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_3

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A. Katti

in all these above-mentioned cases [8–11]. Formation of optical spatial solitons is mediated by the Kerr effect in photorefractive crystals having inversion symmetry [4]. The temperature of the crystal has a profound effect on the dark irradiance and the relative dielectric constant of a centrosymmetric photorefractive crystal, hence affecting the self-trapping nonlinearity [12]. In the present research, we shall study the consequence of varying the temperature of the photorefractive crystal on the spatial profile of a bright–dark soliton pair in biased centrosymmetric photorefractive media. Instead of numerical solutions, here we shall consider the analytical solutions of the bright–dark soliton pair under the approximation of nearly identical peak intensities. The soliton width is an indicator of the self-trapping nonlinearity, and hence, the change in soliton width with respect to temperature will be investigated.

2 Theoretical Model and Discussion We shall consider two collinearly propagating beams along the z-axis in a biased centrosymmetric photorefractive crystal. We assume the diffraction to be along the x-axis and the two beams polarized along the same direction. The two optical beams − → − → are expressed as E 1 = xˆ A1 (x, z) exp(ikz) and E 2 = xˆ A2 (x, z) exp(ikz). The crystal is kept on a metallic plate whose temperature can be varied so as to control the temperature of the crystal. It is also covered with an insulating cover. With the above-mentioned conditions, the dynamical evolution equations for the two optical beams are [8, 13],  ∂ + i ∂z  ∂ + i ∂z

 1 ∂2 k + n A1 (x, z) = 0 2k ∂ x 2 ne  1 ∂2 k + n A2 (x, z) = 0 2k ∂ x 2 ne

(1a)

(1b)

along with 1 2 , n = − n 3e geff 02 (r − 1)2 E sc 2

(2)

where k = k0 n e = (2π/λ0 )n e , ne is the unperturbed refractive index, λ0 is the light beams’ wavelength free space, the quadratic electro-optic coefficient is denoted by geff , 0 is the electrical permittivity in vacuum, the dielectric constant is represented by r , and E sc represents the induced space charge field. The analytical solutions of a dark–bright soliton pair supported in centrosymmetric photorefractive media satisfying Eqs. (1a, 1b) have been found elsewhere [14], U1 = r

1/2

      1 α 1/2 1− ξ sech (2αδ) s exp −i δ 1+δ

(3a)

Effect of Temperature on Incoherently Coupled Dark–Bright …

23

  U2 = ρ 1/2 tanh (2αδ)1/2 s exp[−iαξ ]

(3b)

 I is the overall intensity and is given by I = (n e /2η0 ) |A1 |2 + |A2 |2 along with η0 = (μ0 /0 )1/2 . Also, the dimensionless coordinates used are s = x/x0 , ξ = z/(kx02 ), A1 = (2η0 Id /n e )1/2 U1 , A2 = (2η0 Id /n e )1/2 U2 , α = τ E 0 , δ = (r − ρ)/(1 + ρ), τ = (k0 x0 )2 n 4e geff 02 (r − 1)2 /2. x 0 is the scaling spatial width which is set arbitrarily, r = I I(0) , and ρ = I (∞) . In paraId d electric photorefractives, the temperature of the photorefractive crystal profoundly affects the value of the dielectric constant. The dielectric constant is a function of temperature according to the Curie–Weiss law [12, 15–17], r = r0

T0 − Tc T − Tc

(4)

where r0 is the magnitude of r at T = T 0 and T c is the Curie temperature. Also, the dark irradiance I d is a function of the temperature as [18]  Id (T ) = Id0

T T0

3/2

 exp

Et kB



1 1 − T0 T

 (5)

and, hence, we have the intensity ratio,  −3/2    I (0) T Et 1 1 r (T ) = = r0 exp − − Id (T ) T0 k B T0 T  −3/2    I (0) T Et 1 1 ρ(T ) = = ρ0 exp − − Id (T ) T0 k B T0 T

(6)

(7)

with r0 = II(0) and ρ0 = I (∞) . I d0 d0 The FWHM of the dark–bright soliton pair (3a, 3b) is given by

s = 2 ln(1 +

√

1/2  √ (1 + ρ) 1 1/2 2) = 2 ln(1 + 2) 2αδ (k0 x0 )2 n 4e geff 02 (r − 1)2 E 0 (r − ρ) 

(8) Using (4), (6), and (7) and substituting in the dark–bright soliton pair solution (3a, 3b), the normalized intensity profiles can be obtained for different temperatures of the photorefractive crystal. From (8), we can obtain an explicit dependence of the soliton width on the temperature of the photorefractive crystal. We shall consider a KLTN crystal for illustration of our result taking the following parameters [12, 14], λ0 = 514.5 nm, x0 = 20 µm, E 0 = 5.4 × 105 V/m, r 0 = 10.1,

24

A. Katti

ρ0 = 10, geff = 0.12 m4 C−2 , ne = 2.4, E t ~ 10−19 J, T c = 279.2 K, r0 = 7210 at T 0 = 300 K. For selecting the range of temperatures in which the investigation is to be conducted, we must consider two facts: firstly that the phase transition temperature is T p = 283 K, and hence, the dielectric constant is appreciable near this value of temperature and secondly that the temperature value should be distant enough from the transition temperature in order for the mean-field approximation to be applicable. Hence, we shall take the temperature range of 290–380 K for our investigation. Figure 1 displays the profiles of the soliton components at T = 295 K. Figure 2a shows the bright–dark soliton pair’s normalized intensity profiles at T = 295, 310, and 325 K. Figure 2b shows how the soliton width; i.e., FWHM varies with the temperature. Figure 3 shows the soliton width for bright–dark soliton pair as a simultaneous function of intensity difference, δ and the temperature.

Fig. 1 Dynamical evolution of the bright–dark soliton pair when T = 295 K

Fig. 2 a Normalized intensities of the soliton components of the bright–dark soliton pair at T = 295, 310, 325 K, δ = 0.0091. b Temperature dependence of the soliton pair’s width

Effect of Temperature on Incoherently Coupled Dark–Bright …

25

Fig. 3 FWHM of the soliton pair as a simultaneous function of intensity ratio and temperature

3 Conclusions We have investigated the effect of temperature on bright–dark soliton pair in biased centrosymmetric photorefractive media. The profiles of the soliton pair have been found when the maximum intensity of the bright soliton is slightly greater than that of the dark soliton, i.e., 0 < δ  1. We have considered the effect of temperature on the dielectric constant as well as on the dark irradiance. We have obtained an analytical expression for how the soliton width varies with temperature in a photorefractive crystal. The soliton width increases with increasing temperature which shows a weakening of the self-trapping in case of the dark–bright soliton pair. Our results have potential practical applications for optical switching and routing in addition to bidirectional waveguides and waveguide coupling.

References 1. Stegeman GI (1999) Optical spatial solitons and their interactions: universality and diversity. Science 286(5444):1518–1523. https://doi.org/10.1126/science.286.5444.1518 2. Trillo S, Torruellas WE (eds) (2001) Spatial solitons. Springer series in optical sciences, vol 31. Springer, Berlin 3. Segev M, Valley GC, Crosignani B, Diporto P, Yariv A (1994) Steady-state spatial screening solitons in photorefractive materials with external applied field. Phys Rev Lett 73(24):3211 4. Segev M, Agranat AJ (1997) Spatial solitons in centrosymmetric photorefractive media. Opt Lett 22(17):1299–1301. https://doi.org/10.1364/OL.22.001299 5. Fazio E et al (2004) Screening-photovoltaic bright solitons in lithium niobate and associated single-mode waveguides. Appl Phys Lett 85(12):2193–2195. https://doi.org/10.1063/1. 1794854

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6. Segev M, Valley GC, Bashaw MC, Taya M, Fejer MM (1997) Photovoltaic spatial solitons. J Opt Soc Am B 14(7):1772. https://doi.org/10.1364/JOSAB.14.001772 7. Safioui J, Devaux F, Chauvet M (2009) Pyroliton: pyroelectric spatial soliton. Opt Express 17(24):22209–22216. https://doi.org/10.1364/OE.17.022209 8. Christodoulides DN, Singh SR, Carvalho MI, Segev M (1996) Incoherently coupled soliton pairs in biased photorefractive crystals. Appl Phys Lett 68(13):1763–1765. https://doi.org/10. 1063/1.116659 9. Chen Z, Segev M, Coskun TH, Christodoulides DN, Kivshar YS, Afanasjev VV (1996) Incoherently coupled dark-bright photorefractive solitons. Opt Lett 21(22):1821–1823. https://doi. org/10.1364/OL.21.001821 10. Chun-feng H et al (2001) Incoherently coupled screening-photovoltaic soliton families in biased photovoltaic photorefractive crystals. Chin Phys 10(4):310–313. https://doi.org/10.1088/10091963/10/4/309 11. Keqing L, Yanpeng Z, Tiantong T, Bo L (2001) Incoherently coupled steady-state soliton pairs in biased photorefractive-photovoltaic materials. Phys Rev E Stat Nonlinear Soft Matter Phys 64(5) II:056603/1–056603/9. https://doi.org/10.1103/PhysRevE.64.056603 12. Zhan K, Hou C, Tian H (2010) Effects of the temperature on steady-state bright spatial solitons in biased centrosymmetric photorefractive crystals. Appl Phys B Lasers Opt 100(4):821–826. https://doi.org/10.1007/s00340-010-4067-x 13. Christodoulides DN, Carvalho MI (1995) Bright, dark, and gray spatial soliton states in photorefractive media. J Opt Soc Am B 12(9):1628. https://doi.org/10.1364/JOSAB.12.001628 14. Chun-Feng H, Chun-Guang D, Shi-Qun L (2001) Incoherently coupled bright-dark soliton pairs in biased centrosymmetric photorefractive media. Chin Phys Lett 18(12):1607 15. Crosignani B, Degasperis A, Del Re E, Di Porto P, Agranat AJ (1999) Nonlinear optical diffraction effects and solitons due to anisotropic charge-diffusion-based self-interaction. Phys Rev Lett 82(8):1664–1667. https://doi.org/10.1103/PhysRevLett.82.1664 16. Del Re E, Tamburrini M, Segev M, Della Pergola R, Agranat AJ (1999) Spontaneous selftrapping of optical beams in metastable paraelectric crystals. Phys Rev Lett 83(10):1954–1957. https://doi.org/10.1103/PhysRevLett.83.1954 17. Zhou Z, Li Y, Tian H, Li J, Liu Y, Yang Y (2009) Photocarrier transport in iron-doped potassium lithium tantalate niobate studied by time-of-flight measurement. Opt Commun 282(13):2624– 2627 18. Cheng L, Partovi A (1986) Temperature and intensity dependence of photorefractive effect in GaAs. Appl Phys Lett 49(21):1456–1458. https://doi.org/10.1063/1.97301

Transmission Analysis of Designed 2D MWC in Hybrid OCDMA System for Local Area Network Application Madhumita Sarkar, Somali Sikder, and Shila Ghosh

Abstract This paper explains the design as well the performance analysis of 2D MWC for hybrid OCDMA systems. Code word design has been done for nine users and shown for Fiber Bragg Grating (FBG) based system. Simulated architecture of hybrid system using Opti System simulator for 2D MWC is represented for 5 and 35 km distance to obtain the eye pattern and BER as well. Theoretical analysis shows improved BER value of 10−50 using 2D MWC. This work also reveals the enhancement of performance for LAN applications. Keywords Modified Walsh Code (MWC) · Spectral-amplitude coding-optical code division multiple access (SAC-OCDMA) · Multi-user interference (MUI) · Bit error rate (BER)

1 Introduction Optical CDMA is a broadcast technology, with all information going to all parts of the network, and is a much better option for LAN application. It has advantages such as asynchronous random access, simple management, flexible networking, supporting multiple services while providing some confidentiality of data transmission, and also cost-effective [1]. A hybrid (m × n, w, λa , λc ) wavelength-hopping/time-spreading (WH/TS), multi-wavelength, optical orthogonal code C [2, 3] used in OCDMA is a set of binary {0, 1} m × n matrices where each m × n represents a code word. Here m is related to the number of available wavelengths and n is the code length, i.e., the number of time chips. Each subscriber in a network is assigned a matrix as its own address signature. Spectral-amplitude-coding (SAC) technique is introduced to suppress the influence of multi-user interference (MUI) effect by employing codes with fixed in-phase cross-correlation λ [4, 5]. To eliminate this noise the value of λ should be kept as small as possible. The network capacity and security can be M. Sarkar (B) · S. Sikder B. P. Poddar Institute of Management and Technology, Kolkata 700052, India S. Ghosh St. Thomas’ College of Engineering and Technology, Kolkata 700027, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_4

27

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M. Sarkar et al.

increased by developing a SAC-OCDMA system over WDM scheme [2, 6]. In the present study, in view of getting a high-performance system, we have designed a 2D Modified Walsh Code (MWC) implemented on SAC-OCDMA over WDM (i.e. hybrid) system using our previously originally designed 1D MWC [7, 8] based on FBG encoder/decoder technique for their wavelength selective properties to fulfill fiber’s demand of transmitting high data traffic for long transmission [9, 10]. It has been observed that the performance of our designed 2D MWC is more efficient for a transmission system than 1D MWC, we have designed earlier [7] and other existing 2D OCDMA codes as shown in Figs. 4 and 5 respectively. In this work Erbium-doped fiber amplifier (EDFA) has been used to design the OCDMA-based transmission system. EDFA has more pump energy to amplify many optical signals that multiplexed by its amplification band [11, 12]. The present work in this paper is organized as follows. Section 2 is devoted to the design of 2D MWC for hybrid OCDMA systems. In Sect. 3, we analyze the system performance taking into account intensity noise, shot noise, and thermal noise. Section 3 also deals with the system result analysis. Finally, conclusions are drawn in Sect. 4.

2 Design of 2D MWC for Hybrid OCDMA System Table 1 shows the proposed FBG based encoder system shared by 9 users. The resulting optical signals of each user are directed to the corresponding input ports of the FBG as shown in Fig. 1. Because of the cyclic properties of MWC sequence, code words of the active users are generated in the output ports of FBG. In Table 1, the data bits of users 0, 1, 4, 6, 7 are logical 1, so only the corresponding code words of these users are produced by the encoder [13]. The signal of user 0 is directed to the input port 0 of the encoder, so X 0 = (0 1 0 1 0 0 0 0 1) with central wavelengths of λ2 , λ4 , λ9 appear in output ports 2, 4, and 9 respectively. Due to the cyclic properties of MWC, X 1 = (1 0 1 0 1 0 0 0 0) with central wavelengths of λ3 , λ5 , λ9 also appear in output port 3, 5, 9. Thus 9 users can share the same FBG system as a common encoder. On receiver side, the received signal spectrum S is the summation of all users’ transmitted signal spectrum S = (s0 , s1 , . . . , s N −1 ) =

N −1 

bk xk

(1)

k=0

where bk is the kth user’s data bit and belongs to {0, 1}. In the example of Table 1, S is equal to (1, 1, 0, 1, 0, 1, 0, 1, 1). Table 2 shows the wavelength distribution in the upper branch of FBG. In the output port 0 of FBG, the wavelength chips 2, 4, and 9 of received signal are obtained from input-ports 1, 2, 4, 6, 8, and 9 of the FBG, and the respective photodiode of user obtains SX 0 = 3 unit energy, as shown in Table 2. It can be noticed the receiving power is same for each user. The reverse arrangement

0

1

0

0

0

0

1

0

1

0

1

2

3

4

5

6

7

8

0

1

0

0

0

0

1

0

1

Signature sequence

User No.

1

0

0

0

0

1

0

1

0

0

0

0

0

1

0

1

0

1

0

0

0

1

0

1

0

1

0

0

0

1

0

1

0

1

0

0

Table 1 MWC of length 9 for optical spectral coding

0

1

0

1

0

1

0

0

0

1

0

1

0

1

0

0

0

0

0

1

0

1

0

0

0

0

1

0

1

1

0

1

0

0

1

1

Data bit

0

0

0

0

0

0

0

0

0

0 0 0 0 0 0 0 0 0

λ2 λ3 λ4 λ5 λ6 λ7 λ8 λ9 λ1

λ3

λ2

λ1

λ9

λ8

λ7

λ6

λ5

λ4

Transmitted signals in outputs ports

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

0

λ8

λ7

λ6

λ5

λ4

λ3

λ2

λ1

λ9

Transmission Analysis of Designed 2D MWC in Hybrid OCDMA … 29

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M. Sarkar et al.

Fig. 1 Block diagram of SAC OCDMA using FBG

can be implemented for lower branch of FBG and power can be calculated using same manner.

3 Performance and Result Analysis The spontaneous emission generates an ideally unpolarized thermal light whose detection cause variance of photocurrent which is given by [8]  2 4K T B i = 2eI B + I 2 Bτc RL

(2)

The expression of SNR for designed MWC can be derived as [8] SNR =

2v BK

(3)

For the 2D MWC we have considered “K” as the optical waveforms representing the data and multiple interferers. Pd and Pi are the instantaneous optical powers for data and interferers at the photodetector, respectively. The crosstalk level parameter ξ = Pi /Pd [8]. SNR calculation has been performed by considering thermal noise (TN), shot noise (SN), beat noise (BN) assuming the spacing between the optical channels is significantly wide and the WDM channels have the same speed [6]. In addition, each power spectral component has an identical spectral width and each user gets the same power at the receiver [1, 8]. Here, 2 2 + σint−int SNR = (w Pd + kξ Pd )2 /σsh2 + σth2 + σsig−int

where 2 = 2ξ Pd2 k σsig−int 2 σint−int = 2ξ 2 Pd2 k

(4)

1

1

1

0

0

1

1

0

1

1

2

3

4

5

6

7

8

λ1

λ

0

i/p port

1

0

1

1

0

0

1

1

1

λ2

0

0

0

0

0

0

0

0

0

λ3

1

0

1

1

0

0

1

1

1

λ4

0

0

0

0

0

0

0

0

0

λ5

1

0

1

1

0

0

1

1

1

λ6

0

0

0

0

0

0

0

0

0

λ7

Table 2 Wavelength distribution in FBG based decoder λ8

1

0

1

1

0

0

1

1

1

λ9

1

0

1

1

0

0

1

1

1

8

7

6

5

4

3

2

1

0

o/p port

0

0

1

0

0

0

0

1

0

λ1

 λ2

1

1

0

0

0

0

1

0

1

λ3

0

0

0

0

0

1

0

1

0

λ4

1

0

0

0

1

0

1

0

1

λ5

0

0

0

1

0

1

0

1

0

λ6

0

0

1

0

1

0

1

0

0

λ7

0

1

0

1

0

1

0

0

0

λ8

0

0

1

0

1

0

0

0

0

λ9

1

1

0

1

0

0

0

0

1

3

3

3

3

3

3

3

3

3

Power

Transmission Analysis of Designed 2D MWC in Hybrid OCDMA … 31

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M. Sarkar et al.

σsh2 = 2q(sw Pd + k Pd ξ )Be  BER = 0.5er f c

SNR 8

(5)

Finally, the BER has been calculated using the Eq. (3) as per Refs. [1, 8]. Figure 2 shows comparison of BER performance of 2DMWC and 2D MWC using EDFA for different fiber lengths.BER performance is found better for 2-D MWC using amplifier in the system. The amplifier plays an important role to drive the whole system and to reduce the value of BER for designed MWC as EDFA enhances the transmitted signal power within the communication channel in case of degradation of power. The system can be used for 30–35 km range of distance. Figure 3 shows the theoretical representation of BER for 2-D optical orthogonal code and 2-D MWC and 2-D extended double code. The BER value for 2-D MWC shows better suppression of the signal interferer noise (Table 3). The hybrid system combining the WDM along with OCDMA and employing MWC is simulated using Opti System simulator. The simulation is carried out for maximum 35 km distance. The attenuation of optical fiber is 0.2 dB/km, dispersion is 16.75 ps/nm-km. Figures 4 and 5 illustrate the simulated results of BER performance and eye pattern for 5 and 35 km length of fiber for proposed hybrid OCDMA system.

Fig. 2 BER performance against distance for 1-D and 2-D MWC (with and without amplifier)

Transmission Analysis of Designed 2D MWC in Hybrid OCDMA …

33

Fig. 3 Theoretical BER against number of active users for different 2D OCDMA codes. OOC, MWC, and EDC

Table 3 Comparison of different MWCs with few other OCDMA codes Code

No. of users (K)

Character (L, w, λa , λc )

Cardinality

SNR

MWC

K

(L, 3, 1, 0)

L/9 (for 1D) L 3 /9w2 (for 2D)

SNR =

OOC

K

(L, w, 1, 1)

[(L − 1)/w(w − 1)]

SNR =2

2v BK

4L (K −1)(2L−w 2 )

EDW

K

L = (2n − 1)

2n + 1

SNR = 3

4L (K −1)(L 2 +L−1)

4 Conclusion The system simulation using Opti System shows lower value of BER (10−22 ). The BER value is comparatively low after using EDFA. EDFA has been used as booster in our simulation. The system is quite capable to be used in LAN up to 35 km. The code weight is fixed at 3 which implies simple architecture and implementation. The signal received power is also lower and fixed at 3. The proposed MWC can therefore efficiently suppress the effect of intensity noise and, hence, results in better performance. However, limitation of the code is that the code-length is large with respect to the number of users. In future, the proposed system may be a promising solution for optical access networks offering flexibility, high spectral efficiency, cost effective as well as ensured security for long-distance communication.

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Fig. 4 Eye diagram indicating the performance of 2-D MWC for channel length 5 km

Fig. 5 Eye diagram indicating the performance of 2-D MWC for channel length 35 km

Transmission Analysis of Designed 2D MWC in Hybrid OCDMA …

35

References 1. Prucnal PR (2006) Optical code division multiple access: fundamentals and applications, 1st edn. CRC Press, Taylor & Francis Group 2. Ahmed N, Aljunid SA, Fadil A, Ahmad RB, Rashid MA (2012) Hybrid OCDMA over WDM system for 60 km transmission in optical access networks. Int J Comput Commun Eng 1(2) 3. Sun HS, Wang Y, Anshi ZX (2006) A new family of 2-D optical orthogonal codes and analysis of its performance in optical CDMA access networks. J Light Wave Technol 24(4) 4. Hui Y, Qui K, Yun L (2009) Spectral efficiency of OCDMA systems. Chin Opt Lett 7(3) 5. Bazan MT, David H (2006) Performance analysis of 2-D time-wavelength OCDMA systems with coherent light sources: code design considerations. J Light Wave Technol 24(10) 6. Sarkar M, Sikder S, Ghosh S (2018) Development of architecture for secured data transmission in OCDMA system with designed Modified Walsh Code. In: Proceedings of WOCN’18, West Bengal, India, 2nd–4th Feb 2018 7. Sikder S, Ghosh S (2015) Design and performance analysis of Modified Walsh Code (MWC) in OCDMA system. Int J Innov Adv Comput Sci IJIACS 4(Special Issue). ISSN 2347-8616 8. Sikder S, Sarkar M, Ghosh S (2015) Theoretical analysis and simulation investigation of designed 1-D and 2-D Modified Walsh Code (MWC) in optical CDMA system. In: Proceedings of IOCME, IIT Bhubaneswar, India, Dec 2015 9. Wei Z, Shalaby HMH, Ghafouri-Shiraz H (2001) Modified quadratic congruence codes for fiber Bragg-grating-based spectral amplitude-coding optical CDMA systems. J Lightwave Technol 19 10. Chao-Chin Y, Jen-Fa H, Shin-Pin T (2004) Optical CDMA network codes structured with m-sequence codes over waveguide-grating routers. IEEE Photon Technol Lett 16(2) 11. Islam M, Ahmed N, Ali S, Aljunid S, Ahmad B, Ahmad NB (2016) Hybrid (OCDMA/WDM) system using EDFA for optical access networks. Am Inst Phys 12. Chapman DA, Davies PA, Monk J (2002) Code-division multiple-access in an optical fiber LAN with amplified bus topology: the SLIM bus. IEEE Trans Commun 50(9) 13. Zhang Y, Chen H, Si Z, Ji H, Xie S (2008) Design of FBG en/decoders in coherent 2-D time-wavelength OCDMA systems. IEEE Photon Technol Lett 20(11)

Optical Code Construction of Balanced Weight Ideal Cross-Correlation Code for Spectral Amplitude Coding Optical CDMA Systems Teena Sharma

Abstract In spectral amplitude coding optical code division multiple access (SACOCDMA) codes should be designed with best cross correlation and auto correlation properties. In this paper, we have proposed a new balanced weight ideal cross-correlation (BWICC) code for SAC-OCDMA which provides improved code construction and system performance. The BWICC code has a unitary matrices property and very less overlaps in spectra of various clients due to its ideal crosscorrelation properties among each user. This reduces multiple access interference (MAI) to a considerable extent. Furthermore, proposed code provides moderate code length as compared to other ideal in phase cross-correlation codes. Initially, basic matrix is constructed based on required number of users and code weight value. Number of columns and rows are determined after this. BWICC code is explained as; at first a basic matrix is constructed utilizing the code weight value and the number of users. Finally, rows and columns are found out in the code matrix. Secondly, each diagonal sequences are computed. Proposed code performance analysis is performed in terms of quality factor and BER using simulation. Keywords OCDMA · Spectral amplitude coding · Multiple access interference · Detection · Cross-correlation

1 Introduction Multiple access interference can be eliminated by spectral amplitude coding (SAC) by using efficient detection techniques and proper choice of codes. Most commonly used detection methods are complementary detection technique, direct detection technique, and AND subtraction detection [1]. At the receiving end, multiple acces interference (MAI) is eliminated [2]. T. Sharma (B) Department of Electronis and Communication Engineering, Malaviya National Institute of Technology, Jaipur, India Department of Applied Sciences, University of Quebec in Chicoutimi (UQAC), Chicoutimi, QC G7H 2B1, Canada © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_5

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T. Sharma

Codes orthogonality is the main factor which affects OCDMA performance. SACOCDMA orthogonal codes are multi-diagonal code, modified quadrature congruent code, double weight code, enhanced double weight code, random diagonal code, etc. [3, 4]. These codes show different auto- and cross-correlation features. Since, the cross relationship is not perfect for Walsh Hadamard code, it experiences MAI. OOC and prime codes are larger length code and therefore are bandwidth inefficient which results in lesser number of supportable users [5]. The presented code has moderate code length with benefit of minimum crosscorrelation, and it can be constructed for any weight higher than two. We expected that proposed code gives improved performance due to its efficient code construction algorithm. Balanced weight ideal cross-correlation (BWICC) code algorithm is free from code mapping procedures to obtain higher number of subscribers in the system. Code mapping disadvantage is that cross-correlation increases in system as the number of users increases which results in higher MAI as well as complex receiver and code construction [6]. The remaining paper is compiled as follows: Sect. 2 presents SAC-OCDMA code properties. Section 3 with sub Sects. 3.1–3.3 consists of code design, flowchart, and code construction for odd weight. Section 4 discusses SAC-OCDMA framework with code implementation in optical environment. Newly developed code performance analysis is discussed in Sect. 5. The study is summarized with future possibilities in Sect. 6.

2 Code Construction In an OCDMA environment, the code set (C) is made up of unipolar sequences which is either 0 or 1. An optical code is represented as (N, W, λa , λc ), where N is the code length, W is the code weight, λa is the auto correlation, and λc is the in-phase crosscorrelation. Codes must be designed with low cross-correlation value generally zero or ideal in-phase cross-correlation (1). Multiple user interferences (MUI) are less in system due to low cross-correlation value [7]. To design good properties, SAC codes following are the measures that should be considered [8–10]: 1. 2. 3. 4. 5.

The code length should not change with the active clients, and it should be small, so that more number of users can be accommodated. The cross-correlation factor should be zero reduce PIIN and MAI effects. The code design procedure should be simple. The codes should provide higher users with increased data rate, with minimum BER. Low latency time and increased security should be maintained by the codes.

Considering these points, we have proposed a new code which belongs to ideal cross-correlation code family (Sect. 3).

Optical Code Construction of Balanced Weight Ideal Cross …

39

3 Balanced Weight Ideal Cross-Correlation (BWICC) Codes BWICC code can be designed by following steps:

3.1 The Code Construction Steps Step 1: Number of users (K) is selected based on the system requirement and code weight of code W = 3, 4, …, N is chosen based on the required number of users. Resultant weight (W R ) for base matrix is set to W − 1. Step 2: Calculate the length of the code as: L = (K × WR ) + 2 Step 3: Size of basic balanced matrix is determined using W R : IB = 2 × WR (Balanced) = 2 × (W − 1) Step 4: Basic matrix (I B ) is designed for any number of subscribers as:  IB =

R1 R2



 =

[W/2]1’s [W − 2]0’s [W + 2/3]0’s [W + 1/2]1’s

 2×(W −1)

where R1 and R2 are Row 1 and Row 2 of the I B . Step 5: Weight matrix is designed to increase the code weight which is added after K × L code matrix construction: ⎡ ⎤ 1 0 ⎢ 0 1⎥ ⎢ ⎥ ⎢ ⎥ (1) IW = ⎢ . . ⎥ ⎢ ⎥ ⎣ . .⎦ mn Step 6: Final code is denoted by M of size K × L for K clients. M matrix is constructed in three steps in which first H* is designed. I B is repeated (N − 1) times in M matrix:

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⎡

R ⎢ 1 ··· ⎢ ⎢ R2 R1 ⎢ ⎢ .. ⎢ ∗ H = ⎢ . R2 ⎢ .. .. ⎢ . . ⎢ ⎢ . . ⎣ .. .. ··· ···

. ⎤ · · · · · · · · · .. ⎥ . ⎥ · · · · · · · · · .. ⎥ ⎥ .. ⎥ R1 · · · · · · . ⎥ ⎥ .. . . . . .. ⎥ . . . ⎥ . ⎥ .. ⎥ .. . . . R1 . ⎦ . · · · · · · R2 · · · K ×L

(2)

Empty spaces of matrix H** are filled with null to make final matrix M: ⎡

R1 · · · · · · · · · · · ·

⎢ ⎢ R2 ⎢ ⎢ .. ⎢ . M =⎢ ⎢ . ⎢ .. ⎢ ⎢ . ⎣ .. ···

R1 · · · · · · . . R2 . . . . .. . . . . . . . .. . . . . . . . ··· ··· ···

··· .. . .. . .. . ···

⎤ R2 .. ⎥ . ⎥ ⎥ .. ⎥ . ⎥ ⎥ .. ⎥ .⎥ ⎥ .. ⎥ . ⎦ R1 K ×L

.. . .. .

..

.

R1 R2

(4)

Secondly, H** is designed by repeating the same procedure for K × L times: ⎡

H ∗∗

R1 · · · · · · · · · · · ·

⎢ ⎢ R2 ⎢ ⎢ .. ⎢ . =⎢ ⎢ . ⎢ .. ⎢ ⎢ . ⎣ .. ···

R1 · · · . R2 . . .. . . . . .. . . . . ··· ···

··· ··· .. .. . . .. .. . . .. .. . . ··· ···

.. . .. .

..

.

R1 R2

⎤ R2 .. ⎥ . ⎥ ⎥ .. ⎥ . ⎥ ⎥ .. ⎥ .⎥ ⎥ .. ⎥ . ⎦ R1 K ×L

(3)

The auto-correlation and cross-correlation functions for the BWICC code are stated as: n−1

C[x]C[x] where 0 < m < N −1. Auto-correlation: Ax, x = n=0 n−1

Cross-correlation: C x, y =

C[x]C[y] where 0 < m < N −1.

n=0

3.2 BWICC Code Implementation Flow Diagram Figure 1 depicts how the BWICC code is constructed and implemented.

Optical Code Construction of Balanced Weight Ideal Cross …

41

Start

Enter (K, W) Find WR

L= (K x WR)+2

Define the size of basic matrix IB=2xWR and Construct

Construct H* and H** and Repeat matrix (K-1) Ɵmes Fill the vacant spaces with zeros

Lastly, H matrix is designed

If not ideal cross correlation

END Fig. 1 Implementation of BWICC code

3.3 BWICC Code Construction with Weight 3 Let the number of users (K) is 4 and weight (W ) is 3. Resultant weight W R for base matrix be calculated as W R = W – 1, i.e., 2 in this case. Calculate the length of the code as: L = (K × WR ) + 2 = (4 × 2) + 2 = 10

(5)

Size of basic balanced matrix = I B = 2 × W R (Balanced) = 2 × 2  IB =

R1 R2





[W/2]1’s [W − 2]0’s = [W + 2/3]0’s [W + 1/2]1’s   10 IB = 0 1 2×2

 (6) 2×(W −1)

(7)

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⎡

⎤ 10000001 ⎢0 1 1 0 0 0 0 0 ⎥ ⎥ H =⎢ ⎣0 0 0 1 1 0 0 0 ⎦ 0 0 0 0 0 1 1 0 K ×L ⎡ ⎤ 1 0 ⎢ 0 1⎥ ⎢ ⎥ ⎢ ⎥ Now add weight matrix IW = ⎢ . . ⎥ ⎢ ⎥ ⎣ . .⎦ mn ⎡ ⎤ 1000000110 ⎢0 1 1 0 0 0 0 0 0 1 ⎥ ⎥ Final code becomes H = ⎢ ⎣0 0 0 1 1 0 0 0 1 0⎦ 0000011001

(8)

(9)

(10)

Therefore, codeword for every user is represented with following wavelengths: • • • •

Codeword for user 1 = λ1 , λ8 , λ9 Codeword for user 2 = λ2 , λ3 , λ10 Codeword for user 3 = λ4 , λ5 , λ9 Codeword for user 4 = λ6 , λ7 , λ10 .

4 System Setup Working of the proposed coding scheme and corresponding system architecture is verified by implementation in a highly acknowledged software called Optisystem. The simulation model is developed with reference to the structure of transmitter and receivers modules in Fig. 2. For simulation setup, the proposed code-based system generally consists of a white light LEDs with constant spectrum over a long bandwidth, Mach–Zehnder modulator (MZM), and encoder/decoder sections. Initially, a board band source (BBS) is placed with 30 nm spectral width. It is implemented for four users. Each user has weight 3. The data rate for simulation is 1 Gbps, and spectral chip width is 0.8 nm. Parameters for simulation are as follows: operating wavelength 1550 nm for light source, −10 dBm is transmitted power, and 0.6 is quantum efficiency of LED. In the transmitter, four encoder is placed and code is constructed by selecting particular wavelengths from broadband source. The modulator modulates these wavelengths as per the given data. The modulated data are further transmitted over the optical fiber channel. Fiber length is set to 30 km with dispersion of 18.75 ps/nm-km and attenuation of 0.2 dB/km. At the receiver, each received signal which consists of non-overlapped spectral components are sent to diode detector. Every user code has unique spectrum which can be measured by spectrum analyzer.

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Fig. 2 SAC-OCDMA system setup for BWICC code with W = 3 and N = 4

At the receiver, the responsivity is set to 1 A/W of PIN photo-diode detector, 5 nA is dark current, and thermal noise of 100 × 10−24 W/Hz is taken into account. Direct detection (DD) technique is used for detecting non-overlapping optical pulses. Therefore, phase-induced intensity noise (PIIN) is suppressed by detecting nonoverlapping spectral components [10].

5 Results and Discussion Eye diagram referring to BER and Q factor demonstrates the performance of BWICC in SAC environment (Figs. 3 and 4). The eye diagram clearly shows that for one user we obtained good performance, i.e., BER is 4.30 × 10−13 with a quality factor value of

1 Gbps

1.5 Gbps

2 Gbps

2.5 Gbps

Fig. 3 Eye diagram indicating performance of four users with different data rates (1, 1.5, 2 and 2.5 Gbps)

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Fig. 4 Eye diagram for one user with W = 3 using proposed code at 1 Gbps

7.14 which is desirable for standard error-free transmission case (at 1 Gbps). Figure 3 shows the eye diagram for the proposed setup when four users are communication simultaneously at 1, 1.5, 2, and 2.5 Gbps of data. It is evident from eye openings that the proposed model is able to provide the required performance and support multiple users communicating at large data rates. It is also observed that performance of proposed system decreases with the increase in data rate.

6 Conclusion In this paper, a new code BWICC is designed and implemented for SAC-OCDMA systems. Proposed code construction is simpler compared to existing codes as the code is designed without code mapping compared to other existing techniques of code generation. The codes constructed using code-mapping techniques present limitations on code cardinality. The reason being that code possesses variable crosscorrelation problems with increase in users, which further results into MAI. Codemapping technique is not utilized to construct proposed code; therefore, such problem does not occur which further provides accurate results. Moreover, code can be designed for any weight value more than two. Receiver is implemented with single

Optical Code Construction of Balanced Weight Ideal Cross …

45

photo diode and fewer other components which minimizes receiver complexity and implementation cost with improved system performance. Performance analysis can be further carried out numerically which is the future work to be done for this code. Moreover, its performance comparison can be made with recently developed SAC codes. MAI can be reduced to a much higher extent if proposed code be implemented with complementary subtraction or NAND detection technique.

References 1. Nisar KS (2017) Numerical construction of generalized matrix partitioning code for spectral amplitude coding optical CDMA systems. Optik 130:619–632 2. Sharma T, Maddila RK, Aljunid SA (2019) Simulative investigation of spectral amplitude coding based OCDMA system using quantum logic gate code with NAND and direct detection techniques. Curr Op Photon 3(6):531–540 3. Mostafa S, Mohamed AENA, El-Samie FEA, Rashed ANZ (2017) Performance evaluation of SAC-OCDMA system in free space optics and optical fiber system based on different types of codes. Wireless Pers Commun 96(2):2843–2861 4. Sharma T, Maddila RK (2019) Performance characteristics of the spectral-amplitude-coding optical CDMA system based on one-dimensional optical codes and a multi-array laser. Ukr J Phys Opt 20(2):81 5. Jellali N, Najjar M, Ferchichi M, Rezig H (2017) Development of new two-dimensional spectral/spatial code based on dynamic cyclic shift code for OCDMA system. Opt Fiber Technol 36:26–32 6. Sharma T, Maddila RK (2020) Optical code construction based on enhanced quantum logic gate (EQLG) technique for spectral amplitude coding optical CDMA systems. Wirel Pers Commun 113:2587–2609 7. Kadhim RA, Fadhil HA, Aljunid SA, Razalli MS (2014) A new two dimensional spectral/spatial multi-diagonal code for noncoherent optical code division multiple access (OCDMA) systems. Opt Commun 329:28–33 8. Moghaddasi M, Seyedzadeh S, Glesk I, Lakshminarayana G, Anas SBA (2017) DW-ZCC code based on SAC–OCDMA deploying multi-wavelength laser source for wireless optical networks. Opt Quant Electron 49(12):393 9. Ahmed HY, Zeghid M, Imtiaz WA, Sghaier A (2019) Two dimensional fixed right shift (FRS) code for SAC-OCDMA systems. Opt Fiber Technol 47:73–87 10. Aljunid SA, Fadhil HA, Ahmad RA, Saad NM (2011) Modeling and simulation of multi diagonal code with zero-cross correlation for SAC-OCDMA networks. In: 2nd International conference on photonics, pp 1–5

Dual-Band Dual Polarized Circularly Polarized and Linearly Polarized L-Shaped Patch Antenna Loaded with Strip and Square Slot Reshmi Dhara, Taraknath Kundu, and Sanjay Kumar Jana

Abstract A Dual-band Dual-polarized square slot antenna is presented in this paper. The proposed antenna consists of an L-shaped patch. Two Semi-Hexagonal notches have been etched from the lower boundary of the ground plane in order to increase impedance bandwidth and a rectangular slit at an angle of 45° with respect to center has also been added on the ground plane in order to enhance axial ratio (ARBW) at lower frequency region. By adjusting the size of the antenna, a Dual-band Dual-polarized broadband can be obtained, where we get circular polarization at lower region and linear polarization at higher region. By using this structure simulated Dual-impedance bandwidths are obtained from 1.46 GHz to 2.80 GHz at lower frequency region and from 5.786 GHz to 8.617 GHz at higher frequency region, at center frequencies of 1.998 GHz and 7.228 GHz, which are 67% and 39% respectively Simulated ARBW is 0.17 GHz at resonating frequency of 2.2 GHz which is 7.72%. Proposed antenna could be utilized for FIXED MOBILE, MOBILE-SATELLITE, EARTH EXPLORATION-SATELLITE, SPACE RESEARCH communication at 2.2 GHz circular polarized (CP) frequency band and IEEE 802.11/Wi-Fi, HIPERLAN 5.8 GHz band, for some C-band, ITU (8 GHz) wireless communication application in linear polarized (LP) band. Keywords Dual-band dual-polarized · Square slot antenna · Semi-hexagonal notch · Axial ratio bandwidth

R. Dhara (B) · S. K. Jana Department of Electronics and Communication Engineering, National Institute of Technology Sikkim, Ravangla, South Sikkim 737139, India e-mail: [email protected] S. K. Jana e-mail: [email protected] T. Kundu Department of Chemistry, National Institute of Technology Sikkim, Ravangla, South Sikkim 737139, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_6

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1 Introduction Recently dual-band dual-mode dual-polarized (DBDMDP) antennas have met the multiple purposes of various equipment in wireless communication systems. On the other hand, antennas with Omni-directional radiation are very workable in personal mobile systems and global position system (GPS) enabled devices. Whereas, antennas with unidirectional radiation patterns are becoming a key technology for various communication systems such as global navigation satellite systems, point-topoint communication systems, and satellite communication systems. In a horizontally polarized loop antenna and a vertically polarized monopole antenna were assembled to form omni-directional vertically/horizontally dual-polarized radiation [1]. Various techniques have been used in [2–7] for CP antenna design but the reported IBW as well as ARBW were very small and the structures are very complicated. In Table 1, detail comparison for various structures of antenna designs are shown. In this paper, a DBDMDP antenna is presented. Dual bands, CP in the lower band, and an LP at higher band have been achieved using a square slot antenna design with L-shaped patch. Two semi-hexagonal notches have been etched from the lower boundary of the ground plane in order to increase wide impedance bandwidth and a rectangular slit at an angle of 45° with respect to center have also been added on the ground in order to increase ARBW at lower frequency region. By using this structure simulated impedance bandwidth of 67% ranged from 1.47 to 2.80 GHz with a center frequency at 2.13 GHz, while impedance bandwidth of around 39% cover the higher range from 5.79 to 8.62 GHz, center frequency at 7.23 GHz. Simulated ARBW at center frequency 2.2 GHz is 7.72%. Need for antenna designs giving multiple impedance bands, with different polarization in different band motivated us to focus our work on designing a compact antenna giving CP and LP bands in the Table 1 Comparison table of the proposed antenna

References

Size (mm2 )

IBW (%)

[2]

70 × 70

22

18.2, 1.98

[3]

25 × 25

118

38.20, 6.2

[4]

100 × 100

18.2

4.45, 1.64

18.4

3.5, 2.68

[5]

80 × 80

26.7

6.1, 1.5

11.3

6, 2.6

5.54

5.54, 2.5

[6]

54 × 54

ARBW (%, f c GHz)

4.29

4.29, 6.5 5.6, 1.7

[7]

70 × 70

5.6 32.35

32.35, 2.55

Proposed

60 × 60

62

7.7, 2.2

39

–

Bold signify this is our proposed work compared to other work

Dual-Band Dual Polarized Circularly Polarized and Linearly …

49

same device. Primary objective of this paper was to design a miniaturized, multiband application microstrip antenna having both circular and linear polarizations in different frequency regions. Whatmore: The proposed antenna is designed using theoretical lower resonating frequency 1.4 GHz so that it can cover 1.4/1.5 GHz global IMT band. But after optimization the dimension of the antenna is 60 × 60 × 1.6 mm3 (0.459 λg × 0.459 λg × 0.012 λg where λg is the guided wavelength at lower resonating frequency 1.4 GHz). We know frequency is inversely proportional to antenna dimension. But keeping the same design frequency we got the antenna dimension 20% lower than conventional antenna design procedure. This is good result satisfying the miniaturization criteria. Our proposed antenna gives a CP characteristics in lower frequency region, in addition also gives broad impedance bandwidth at higher frequency region. So in a compact device satisfy both polarization of the wave propagation. To our knowledge, this is one of the best results obtained in comparison to related designs.

2 Antenna Design Geometry of the proposed antenna has been depicted in Fig. 1. Figure 2 enlisting the detailed dimensions. Low cost readily available FR4-epoxy substrate [8, 9] with a thickness of 1.6 mm, εr = 4.4 and tan δ = 0.02 is chosen here to simulate the design. Design guidelines of the antenna are explained in detail in the following section (Fig. 3). As shown in Step 1, Antenna 1 is designed by using a rectangular I-shaped radiating patch and on the opposite side of the substrate a square slot [10] ground is used. There is now extensive assurance for 1427–1518 MHz to become a global IMT band. So our Antenna is designed to cover 1.4/1.5 GHz lower resonating band as well as give wide impedance band within a single device. Antenna 1 has design frequency of 1.4 GHz. Antenna 1 is first designed on 75 × 75 × 1.6 mm3 (0.574 λg × 0.574 λg × 0.012 λg , where λg is the guided wavelength at a resonating frequency 1.4 GHz). After optimization, of dimensions, the designed Antenna 1 is 60 × 60 × 1.6 mm3 (0.459 λg × 0.459 λg × 0.012 λg ) for getting better results at 1.4 GHz Fig. 1 Simulated proposed antenna a top view b bottom view

(a)

(b)

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Fig. 2 Dimension of the simulated proposed antenna

(a)

(b)

(c)

(d)

Fig. 3 Four improvement processes of proposed antenna

design frequency. So 20.03% size is reduced compared to earlier. Now the structure is compact in consideration of earlier design. At first, to clarify the improvement process, four antenna structures’ simulated return losses are defined in Fig. 4. For Antenna 1–4 the simulation results for improvement of ARBW are shown in Fig. 5. For Antenna 1 the simulated IBW is small, i.e., only two small impedance bands and ARBWs are also very poor greater than 15 dB. So in order to increase the IBW and ARBW we have to create perturbation techniques. So that electric field could generate two orthogonal modes with equal amplitude. We added a horizontal rectangular strip in positive Y-direction with previously designed I-shaped radiating strip, which now looked like L-shaped patch as shown in Antenna 2. This structure generates another current patch which increases IBW [11]. The ARBW curve was still poor, came down below 10 dB but still much greater than desired 3 dB. Its performance improved compared to earlier design as horizontal current is increased here but due to large amount of vertical current polarization, it is still linearly polarized. So for further increase horizontal current moved to Step 3, design Antenna 3. Here, two semi-hexagonal notches [12, 13] were etched from the lower boundary of the ground plane which resulted in improvement of ARBW to less than

Reflection Coefficient (dB)

Dual-Band Dual Polarized Circularly Polarized and Linearly …

51

0 -5 -10 -15 -20 -25 (Antenna1) (Antenna2) (Antenna3) (Antenna4)

-30 -35 1

2

3

4

5

6

7

8

Frequency (GHz)

Fig. 4 Simulated reflection co-efficient for proposed antenna

70

(Antenna1) (Antenna2) (Antenna3) (Antenna4)

Axial Ratio (dB)

60 50 40 30 20 10 0 2

4

6

8

Frequency (GHz)

Fig. 5 Simulated axial ratio for proposed antenna

3 dB but unfortunately did not fall within the IBW curve. Here horizontal current is improved in lower region, as shown in Antenna 3. Therefore, to further increase ARBW within IBW, Step 4, design Antenna 4 was envisaged where some more modifications were imparted. A rectangular slit at an angle of 45° with respect to center was added on the square slotted ground for better performance of the modified patch as compared to Step 3 [14]. By using this technique, the AR bandwidth is improved and obtained a CP from 2.11 to 2.28 GHz, having center frequency (f c ) at 2.20 GHz and ARBW is 7.74%. It is the best compared to earlier designs. This CP band may be used for the following purposes. The band 2120–2180 MHz may be used for FIXED MOBILE communication, 2180–2200 MHz may be used for MOBILESATELLITE communication, 2200–2290 MHz may be used for SPACE OPERATION communication (space-to-Earth) (space-to-space), EARTH EXPLORATIONSATELLITE (space-to-Earth) (space-to-space) communication, FIXED (line-ofsight only) communication, MOBILE (line-of-sight only including aeronautical

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(a)

(b)

(c)

(d)

Fig. 6 Simulated surface current distribution of a Ant. 1 b Ant. 2 c Ant. 3 and d Ant. 4

Fig. 7 Simulated surface current distribution of a Ant. 1 b Ant. 2 c Ant. 3 and d Ant. 4

telemetry, but excluding flight testing of manned aircraft) communication and SPACE RESEARCH (space-to-Earth) (space-to-space) communication [15]. The surface current distributions are analyzed in Fig. 6. From Fig. 6a–d we can conclude that the maximum current distribution in (d) flows along the strip. We can also see that due to the 45° strip maximum surface current flows along strips which create the perturbation of electric field in X and Y direction. In Fig. 7a for Antenna 1, we can see that the vector surface current distributions of the rectangular radiating patch in the horizontal direction are zero, as the horizontal currents cancel each other. Therefore, large amount of vertical current produces LP radiation. In Fig. 7b for Antenna 2 the horizontal directed current is very small as the horizontal currents cancel out each so large amount of vertical current could be produced LP. In Fig. 7c for Antenna 3 horizontal currents are improved to give CP at lower frequency zone but not within IBW curve. Finally, Antenna 4 give E x and E y in equal amount but orthogonal to each other which generate a CP at 2.2 GHz frequency.

3 Results and Discussion Figure 8 shows the simulated IBW by varying the length L 1 . From this Fig. 8 we can see that if we increase the length of L 1 , lower resonating frequency shifts towards the lower band region, and if we decrease the length of L 1 higher resonating frequency shifts towards the higher band region and gradually disappears. This happens because as we increase the length of L 1 , parallel capacitive effect increases, due to which

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53

Fig. 8 Simulated return loss against frequency for the proposed monopole antenna with various L 1 length

quality factor also increase and as a result resonating frequency decreases. So we take best optimum value for L 1 = 7 mm. Figure 9 shows the simulated IBW by varying length L. It can be observed that if we increase the length of L, lower resonating frequency shifts towards the lower band region, and if we decrease the length of L, higher resonating frequency shifts towards the lower frequency zone and gradually disappears. This happens because as we increase the length of L, coupling effect between the radiating patch and slotted ground plane has been changed, due to that resonating frequency decreases. So we take best optimum value for L = 40 mm. Figure 10 shows the simulated IBW by varying K 2 . It can be demonstrated that if we increase the width of K 2 , lower resonating frequency change is almost negligible but at K 2 = 3.5 mm gives the best impedance matching and we obtain wide IBW at higher resonating frequency zone at this position. This happens because as we

Fig. 9 Simulated return loss against frequency for the proposed monopole antenna with various L length

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Fig. 10 Simulated return loss against frequency for the proposed monopole antenna with various L 2 length

Reflection Coefficient (dB)

increase the length of K 2 , additional current path length increase, slow-wave factor is increased as a result group velocity decrease, and resonant frequency moves towards the lower value. Figure 11 shows the simulated IBW by varying L 2 . If we increase the length of L 2 , lower resonating frequency change is almost same and higher resonance frequencies are changing but IBW at higher band is almost same. At higher frequency zone L 2 = 9 mm gives the best impedance matching. Figure 12 shows the simulated IBW by varying radius of hexagonal notch R1 . Due to R1 lower IBW is almost same and at higher frequency zone as R1 increases higher resonating frequency shift towards the higher zone. Due to the coupling between the radiating patch and slotted ground plane series capacitive effects are increased, so quality factor decreases and higher resonating frequency shifted to the higher zone [16, 17]. At R1 = 3.5 mm we get wide IBW compared to other values of R1 . This is due to best impedance matching.

0

-10 (K2=2.5mm) (K2=3.0mm) (K2=3.5mm) (K2=4mm) (K2=4.5mm)

-20

2

4

6

8

Frequency (GHz)

Fig. 11 Simulated return loss against frequency for the proposed monopole antenna with various K 2 length

Reflection Coefficient (dB)

Dual-Band Dual Polarized Circularly Polarized and Linearly …

55

0

-10 (R1=3.1mm) (R1=3.2mm) (R1=3.3mm) (R1=3.4mm) (R1=3.5mm)

-20

2

4

6

8

Frequency (GHz)

Fig. 12 Simulated return loss against frequency for the proposed monopole antenna with various R1 length

Reflection Coefficient (dB)

Figure 13 shows the simulated IBW by varying P3 position. If we observe Fig. 13, we can see that at P3 = 11.6 gives the wide IBW and lower zone as well as in higher zone due to best impedance matching at this position. Figure 14 shows the simulated IBW by varying length of R2 . If we carefully analyze Fig. 14, we can see that at higher frequency zone impedance changes are negligible but at lower frequency zone changes are lot [18–20]. This is due to the coupling between the radiating L-shaped patch and rectangular strip on ground plane. At R2 = 49.5 mm, we get a wide IBW at this position compared to others. Figure 15 shows the simulated ARBW by varying length of rectangular strip on ground plane R2 . The best value of ARBW is obtained at R2 = 49.5 mm, ARBW is within the IBW curve at this position at lower frequency. The simulation was performed using ANSYS HFSS 13. From Fig. 16 it’s clear that proposed antenna has the −10 dB impedance bandwidth (IBW) covering the lower frequency zone range from 1.47 to 2.80 GHz, center frequency (f c ) 2.13 GHz, 67% and impedance bandwidth cover the higher frequency zone range from 5.79 to 8.62 GHz, f c 7.23 GHz, 39%. 0

-10

-20

(P3=11.2mm) (P3=11.3mm) (P3=11.4mm) (P3=11.5mm) (P3=11.6mm)

-30 2

4

6

8

Frequency (GHz)

Fig. 13 Simulated return loss against frequency for the proposed monopole antenna with various P3 length

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Fig. 14 Simulated return loss against frequency for the proposed monopole antenna with various R2 length

Fig. 15 Simulated axial ratio against frequency for the proposed monopole antenna with various R2 length

Fig. 16 Simulated reflection coefficient of the proposed antenna

Dual-Band Dual Polarized Circularly Polarized and Linearly …

57

Fig. 17 Simulated VSWR of the proposed antenna

Axial Ratio (dB)

3.5 3.0 2.5 2.0 1.5 1.0 0.5 0.0 2.00

(Proposed Antenna)

2.05

2.10

2.15

2.20

2.25

2.30

2.35

2.40

Frequency (GHz)

Fig. 18 Simulated ARBW of the proposed antenna

Figure 17 shows the simulated VSWR plot for the proposed antenna. VSWR < 2 the entire IBW region, i.e., 1.47–2.80 GHz and 5.79–8.62 GHz, which is very good for practical application. Figure 18 shows the simulated ARBW of the proposed antenna is 170 MHz ranging from 2.11 to 2.28 GHz and f c is 2.2, 7.74% which is within the range of simulated IBW curve. Obtained circular polarized bands can be used for below-mentioned applications. 2110–2120 MHz band can be used for FIXED MOBILE communication; 2120–2180 MHz band can be used also for FIXED MOBILE communication; 2180–2200 MHz band can be used for MOBILE-SATELLITE communication (space-to-Earth); 2200–2290 MHz band can be used for SPACE OPERATION (space-to-Earth) (space-to-space), EARTH EXPLORATION-SATELLITE (spaceto-Earth) (space-to-space), FIXED (line-of-sight only) MOBILE (line-of-sight only including aeronautical telemetry, but excluding flight testing of manned aircraft), SPACE RESEARCH (space-to-Earth) (space-to-space) [15]. Figure 19 shows the real (Resistance) and imaginary (Reactance) parts of the simulated input impedance at 50  microstrip feed line. Within the CP operating band impedance matching are well maintained as the resistance closes to 50  and the reactance is small closes to 0 .

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Fig. 19 Simulated Impedance [resistance (real), reactance (imaginary)] plot for proposed antenna

Fig. 20 Simulated GAIN plot for the proposed antenna

5 0 -5 -10 -15 -20 -25 -30 -35

(LHCP) (RHCP)

-150

-100

-50

0

50

100

150

CP Gain(dBi) at 2.2 GHz

CP Gain(dBi) at 2.2 GHz

Figure 20 shows the simulated GAIN plot, at f c 2.2 GHz for CP band is 11.4 dBi and the gain at center frequency 7.23 GHz for LP band is 6.48 dBi. Figure 21a shows the simulated radiation pattern for the XZ plane (ϕ = 0°) and YZ plane (ϕ = 90°) at 2.2 GHz. The simulated radiation pattern in Fig. 21a, b have

5 0 -5 -10 -15 -20 -25 -30 -35

(LHCP) (RHCP)

-150

Angle Theta (deg)

(a) XZ plane

Frequency= 2.2 GHz

-100

-50

0

50

100

150

Angle Theta (deg)

(b) YZ plane

Fig. 21 Simulated radiation pattern for the proposed antenna in the a XZ (ϕ = 0°) plane and b YZ (ϕ = 90°) planes

Dual-Band Dual Polarized Circularly Polarized and Linearly …

59

shown that the cross-polarization levels are 34 dB lower than co-polarization levels in the broadside direction. Figure 22 shows the radiation pattern for the proposed antenna at ϕ = 0° (XZ plane) and ϕ = 90° (YZ plane) for left-hand circular polarization (LHCP) and righthand circular polarization (RHCP) at f c 2.2 GHz. The polarization is RHCP that can be observed from these figures. From Fig. 23a, it is clear that E-co is greater than E-cross and from Fig. 23b, it is clear that H-co is greater than H-cross. Figure 23a, b have shown that the cross-polarization levels are 16 dB lower than co-polarization levels in the broadside direction (Table 2). 90

90 120

0 -10

60

150

-10

30

150

30

-30

-30 180

180

0

0

-30

-30

-20

-20

0

60

-20

-20

-10

120

0

210

330 240

300

-10 % (LHCP) % (RHCP)

210

0

330 240

300 270

270

(a) XZ plane

(LHCP) (RHCP)

Frequency= 2.2 GHz

(b) YZ plane

Fig. 22 Simulated radiation patterns (LHCP and RHCP) in the a, c for XZ (ϕ = 0°) and b, d for YZ (ϕ = 90°) planes

(a) XZ plane

Frequency= 7.3 GHz

(b) YZ plane

Fig. 23 E-plane and H-plane radiation pattern for the proposed antenna

60 Table 2 Optimal dimension of the proposed antenna

R. Dhara et al. Parameter

Value (mm)

W

60

L

40

L1

11

L2

6

L3

25

K1

5

K2

3.5

R1

6

P3

11

R2

50

Wf

1

H

1

4 Conclusion A dual-band dual-polarized square slot antenna with L-shaped radiating patch is designed in this paper. Two semi-hexagonal notches have been etched from the lower boundary of the ground plane to increase wide impedance bandwidth and a rectangular slit at an angle of 45° with respect to center has also been added on the ground to create coupling between ground plane and patch which increase CP bandwidth at lower frequency zone. By using this structure simulated impedance bandwidths are 67% at f c 2.13 GHz and 39% at f c 7.23 GHz. Also, simulated ARBW is 7.74% at f c of CP band 2.20 GHz, which is within the range of simulated IBW curve. The proposed antenna is suitable for modern wireless communication for dual-band dual-polarized, i.e., CP and LP operation.

References 1. Row J-S (2005) The design of a squarer-ring slot antenna for circular polarization. IEEE Trans Antennas Propag 53(6):1967–1972 2. Sze J-Y, Wong K-L, Huang C-C (2003) Coplanar waveguide-fed square slot antenna for broadband circularly polarized radiation. IEEE Trans Antennas Propag 51(8):2141–2144 3. Pourahmadazar J, Mohammadi S (2011) Compact circularly-polarised slot antenna for UWB applications. Electron Lett 47(15):837–838 4. Bao XL, Ammann MJ (2011) Monofilar spiral slot antenna for dual-frequency dual-sense circular polarization. IEEE Trans Antennas Propag 59(8):3061–3065 5. Bao XL, Ammann MJ (2008) Dual-frequency dual-sense circularly-polarized slot antenna fed by microstrip line. IEEE Trans Antennas Propag 56(3):645–649 6. Shao Y, Chen Z (2012) A design of dual-frequency dual-sense circularly-polarized slot antenna. IEEE Trans Antennas Propag 60(11):4992–4997

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7. Chen Y-Y et al (2011) Dual-band dual-sense circularly polarized slot antenna with a C-shaped grounded strip. IEEE Antennas Wirel Propag Lett 10:915–918 8. Balanis CA (2016) Antenna theory: analysis and design. Wiley. ISBN-978-1-118-64206 9. Pozar DM (1992) Microstrip antennas. Proc IEEE 80(1):79–91 10. Sze JY, Chang CC (2008) Circularly polarized square slot antenna with a pair of inverted-L grounded strips. IEEE Antennas Wirel Propag Lett 7:149–151 11. Zhou SW, Li PH, Wang Y, Feng WH, Liu ZQ (2011) A CPW-fed broadband circularly polarized regular-hexagonal slot antenna with L-shape monopole. IEEE Antennas Wirel Propag Lett 10:1182–1185 12. Dhara R, Sarkar M, Dey TK, Jana SK (2018) A tri-band circularly polarized G-shaped patch antenna for wireless communication application. In: 2018 International conference on computing, power and communication technologies (GUCON). IEEE, pp 992–996 13. Shokri M, Rafii V, Karamzadeh S, Amiri Z, Virdee B (2014) Miniaturised ultra-wideband circularly polarised antenna with modified ground plane. Electron Lett 50(24):1786–1788 14. Paul PM, Kandasamy K, Sharawi MS (2018) A triband circularly polarized strip and SRRloaded slot antenna. IEEE Trans Antennas Propag 66(10):5569–5573 15. https://transition.fcc.gov/oet/spectrum/table/fcctable.pdf 16. Dhara R, Mitra M (2020) A triple-band circularly polarized annular ring antenna with asymmetric ground plane for wireless applications. Eng Rep 2(4):e12150. https://doi.org/10.1002/ eng2.12150 17. Dhara R, Jana SK, Mitra M (2020) Tri-band circularly polarized monopole antenna for wireless communication application. Radioelectron Commun Syst 63(4):213–222 18. Dhara R (2020) Quad-band circularly polarized CPW-fed G-shaped printed antenna with square slot. Radioelectron Commun Syst 63(7):376–385 19. Dhara R, Jana SK, Mitra M (2020) CPW-fed triple-band circularly polarized printed inverted C-shaped monopole antenna with closed-loop and two semi-hexagonal notches on ground plane. In: Optical and wireless technologies. Springer, Singapore, pp 161–175 20. Dhara R, Yadav S, Sharma MM, Jana SK and Govil MC (2021) A circularly polarized quadband annular ring antenna with asymmetric ground plane using theory of characteristic modes. Progress in electromagnetics research, 100, pp.51–68. DOI: https://doi.org/10.2528/PIERM2 0102006.

Analytical Comparison of Various Detection Techniques for SAC-based OCDMA Systems: A Comparative Review Teena Sharma and M. Ravi Kumar

Abstract Spectral amplitude coding-optical code division multiple access (SAC OCDMA) is the most popular technique due to its suitability in asynchronous environment with efficient codes and detection techniques. In this paper, we have presented a comparative analysis of various detection techniques utilized for SACbased OCDMA to improve the system performance and minimize the noises. Each detection technique has a different architecture for detection of information and based on that, amount of multiple access interference (MAI) is eliminated. Further, its selection criteria relies upon multiple factors such as type of optical code and particular application area. In some of the optical codes, very less overlaps in spectra of various clients are present due to its ideal cross-correlation properties among each user; therefore, in this case used detection technique carries fewer optical and electrical components. This reduces MAI to a considerable extent. Furthermore, proposed idea of analyzing various detection technique suggested use of particular detection techniques based on specific application and system parameters. Keywords OCDMA · SAC OCDMA · Cross-correlation code · Detection · Multiple access interference

1 Introduction In OCDMA framework, spectral amplitude coding (SAC) technique got more consideration due to usages of high auto-correlation and low-value cross-correlation-based codes which deals with multiple access interference (MAI) well. Further, simple encoders and decoders consisting of fiber bragg grating (FBG) structures provides easier spectral encoding and decoding when compared to high-speed electronic T. Sharma (B) · M. Ravi Kumar Department of Electronis and Communication Engineering, Malaviya National Institute of Technology, Jaipur, India T. Sharma Department of Applied Sciences, University of Quebec in Chicoutimi (UQAC), Chicoutimi, G7H 2B1 Québec, Canada © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_7

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circuitry utilized in spectral phase decoding [1]. It uses inexpensive incoherent sources of light for spectral encoding [2]. Multiple detection techniques such as direct detection and complementary subtraction detection techniques are utilized at the receiver [3, 4]. Therefore, using codes with best possible auto-correlation and cross-correlation properties, MUI and PIIN are eliminated to a certain extent. Further, detection techniques plays an important role in elimination of MAI in SAC OCDMA by removing overlapping chips [5]. For SAC system, optical communication channel may be either fiber optic or wireless free-space optics. We have focused on detection techniques those utilizing 1-dimensional SAC code. Optical codes should have best cross- and auto-correlation properties to minimize MAI and probability of error. Moreover, codes should possess properties such as flexibility in code weight selection, low bit error rate (BER) and high level of security. The code construction algorithms should be easy and simple, and it should result in less complex and cost-effective architecture implementation [6, 7]. With the code design, selection of detection technique as per the optical code properties plays major role in achieving better quality of signal. Aim of developing good codes can be fulfilled by designing code families, supporting large cardinality with optimal code length so that bandwidth requirement can be minimized and that can also provide flexibility in addressing code sequences [8]. Further, SAC methods have flourished as successful schemes for local area network (LAN), fiber-to-the-home (FTTH), and multimedia applications especially triple play services as it makes use of different constant-weight and variable-weight code families, each with different properties. In direct detection (DD) technique (Fig. 1), each received signal consists of nonoverlapped spectral components, which represented a unique method for individual user required for diode detector [9]. At receiver side, FBGs as filter in combination with single photo diode are used for detection of non-overlapping optical pulses. PIIN is eliminated on detecting non-overlapping spectra [10]. Moreover, single photo

Fig. 1 SAC-OCDMA system architecture using direct detection technique

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Fig. 2 SAC OCDMA system architecture with balanced detection technique

diode reduces receiver complexity and provides improvement in BER performance. In DD technique power splitters, two PIN diodes and subtractor are not required at the receiving side likewise complementary subtraction (CS) or balance detection (BD) technique [11]. System implementation cost is lesser for such system due to use of less components. FBGs are used for encoding and decoding at transmitter and receiver side for balanced and DD techniques. In BD technique, decoder’s output data are further transmitted to two photodiodes (PD1 and PD2), and via an electrical detection process which futher involves subtraction of overlapping chips, integration, sampling of data, and threshold detection to recover the original data bits (Fig. 2). In this process, MAI could be removed completely with the help of balanced photo-detection process by subtraction of overlapping chips [12]. The remaining paper is compiled as follows: Sect. 2 discusses the selection criteria of detection techniques. Types of detection techniques are discussed in Sect. 3 with seven Sects. 3.1–3.7 consists of receiver design for each detection method. Section 4 presents comparative analysis of most popular detection techniques. The study is summarized in Sect. 5.

2 Selection Criteria of Detection Techniques Detection of transmitted signals is quite challenging in presence of system noises and many other performance-affecting parameters for optical CDMA systems. To maintain performance of optical system, detectors should be designed according to code-specific manner to easily decode the transmitter information. Detection techniques placed at the receiver end can effectively suppress MUI if the code is not a zero correlation code. So far, two detection techniques are quite popular. At receiver side of SAC system, either of the two photodiodes is utilized in balanced mode to

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subtract MAI and either one subtractor or single photodiode is used for detection of optical signals. First technique is called complementary subtraction or balance detection and second one is called direct detection (DD) technique. DD detection is generally used with zero cross-correlation codes but can be used for variable cross-correlation codes depending on specific application [13, 14]. In CS, both originally transmitted signal and the compliment of the signal are decoded and compared. Finally, an auto-correlation function is generated, which is useful for retrieving original signal. Cost and complexity of the system hardware increase due to increase in optical and electrical components. Only non-overlapping chips are decoded in the process of DD technique. Therefore, reduced number of components is required. Advanced detection techniques based on operation of logic gate AND named as AND subtraction and modified AND detection technique is introduced in [15], which improved system performance due to reduced number of FBGs utilized in encoding and decoding process.

3 Types of Detection Techniques 3.1 Complementary Subtraction Detection or Balance Detection CS detection technique has been the most commonly used detection technique (Fig. 3). The two different received signals are modulated with data and multiplexed and transmitted over fiber. Combined signals are divided into two different complementary branches of spectral chips at the receiver.

Fig. 3 Complementary subtraction detection technique based SAC OCDMA receiver

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67

A subtractor placed at output finds the correlation difference by subtracting signals received from two branches. The zero value of subtractor output means no cross-correlation between user data so no overlapping. Moreover, the complementary detection technique can be used for the codes those have fixed value of cross-correlation. The drawback of balanced detection is that for SAC systems, most of the codes are generated by using code mapping techniques for increased number of subscribers, but due to mapping, cross-correlation is not always fixed in this case. Consequently, this detection technique does not provide accurate results [15].

3.2 Single Photodiode Detection (SPD) Technique SAC OCDMA receiver with SPD technique is shown in Fig. 4. There are two decoders in this case: The first one has identical filter spectral response matching with intended encoder, and another is subtractive decoder or s-decoder which cancels signals with mismatched signature code sequences or interferers [16]. Frequency bins coming from different interferers are present in s-decoder. Obtained s-decoder output is zero power in case of active user or equal to cross-correlation power for interfering clients. fiber bragg gratings (FBGs) have been used to perform decoding and encoding operations as well as to compensate dispersion. Subtraction output is either equal to code weight power for active client or zero power for interferers. Therefore, before conversion of signals to the electrical domain, interference gets cancelled in the optical domain which further removes MAI and PIIN effects in the optical domain. However, two interference signals differed slightly from each other at the optical subtractor and resulted into less optical power reception at photodiode. The use of single photodiode becomes possible due to suppression of interference signals within optical domain. Shot noise generated at the receiver side and opticalto-electrical conversion processing are reduced with the help of this technique. Any fixed in-phase cross-correlation code could be used to implement SAC with SPD

Fig. 4 SPD technique-based SAC OCDMA

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Fig. 5 SAC OCDMA receiver based on AND subtraction detection technique

detection just by simply modifying spectral distribution of s-decoder, which basically depends on the SAC codes structure [17].

3.3 AND Subtraction Detection Incoming signal is divided into two parts: First part is decoder with same filter structure similar to encoder and second part of the decoder has AND filter structures [18]. Signal between wavelengths is compared by logic AND to detect the same wavelength or overlapping chip. In order to subtract the overlapping code from the intended code sequence, a subtractor is placed before the photo diode. Therefore, overlapping chips are eliminated and photo diode only receives the desired information. Therefore, it can successfully eliminate MUI just like complementary detection but only for the codes having fixed cross-correlation. It provides significant improvement in transmission distance or higher data rates with increased number of subscribers (Fig. 5).

3.4 Modified AND Subtraction Previous studies showed that the performance of modified-AND subtraction detection technique is better for SAC OCDMA systems when compared with AND subtraction and complementary subtraction techniques [19]. The modified-AND subtraction could reduce the effects of both PIIN and MAI. However, it utilizes similar number of filters as used by AND subtraction and less number of filters

Analytical Comparison of Various Detection Techniques for SAC-…

69

Fig. 6 Modified-AND subtraction detection technique-based SAC OCDMA receiver

when compared with complementary subtraction technique. This technique basically divides the decoded signals spectrum by utilizing a parallel structure of filters at the decoder end. The SAC OCDMA receiver showed that the received optical signal is split by splitter 1 into two parts. First part is split into the upper decoding branches, whereas the second part is split to the AND decoder by using an attenuator. The attenuator fortified that for each inactive user, the interference signal showed an equivalent power incident on individual photo-detectors (Fig. 6). The decoder showed a matched spectral response with active user, whereas AND decoder showed the overlapped bins from different interferers. AND operation among active user and interferers could be mathematically represented by these overlapped bins. Two photodiodes such as PD and subtractive-PD are used to compose the photodetector and in order to provide a differential output signal between both decoded signals; they are electrically connected in opposition.

3.5 XOR Subtraction Detection XOR technique is used to suppress MUI for the code signature sequences which have in-phase cross-correlation either 0 or 1, i.e., for the codes which do not have fixed cross-correlation. This technique is used with the code sequences which are less rigorous IPCC constraints. By using this technique, desired signal spectral chips which are in optical form would be filtered; therefore, no MUI between codes is obtained. The reason being that code properties that possess the maximum crosscorrelation between any two code words is 1. This results in suppressed PIIN noise at the receiving end [20]. Thus, system performance is improved by using XOR detection scheme. Advantage of XOR over any other detection technique is that any number of subscribers can be filtered out whether they have fixed cross-correlation

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Fig. 7 XOR subtraction detection technique-based SAC OCDMA transmitter/receiver

or not. Thus, code freedom can be obtained. System performance is improved due to elimination of MUI, intensity noise, and probability of error. The disadvantage of XOR over AND detection is that hardware complexity of overall system is increased slightly. Thus, a trade-off can be set between the code freedom and hardware complexity for better performance. SAC system with XOR detection technique is represented in Fig. 7 which consists of an ON–OFF shift keying modulator to modulate the desired user data. The modulated signal is then sent to FBGs, and desired user’s chip is being attributed with some specific set of wavelengths. FBGs position depends on the chip value, and it is used for only for chip “1.” Its decoder is based on same idea as complementary technique which is consisting of FBGs set with the original weight and its complementary part. Finally, received signal is passed to photo detector. Only codes matched with desired receiver are send to both the photo diodes to get the desired output, and codes that are mismatched pass without getting detected. Moreover, the system performance is highly improved when compared to the existing detection techniques for suppressed MAI and PIIN due to filtering of only wanted chips in the optical domain.

3.6 NAND Subtraction Detection The procedure for obtaining cross-correlation for two-coded sequences is same for AND and NAND detection techniques. MAI could also be removed for both the

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Fig. 8 Receiver based on NAND subtraction detection technique

techniques effectively. However, NAND-based detection gives extra weights during subtraction-based detection procedure. These extra weights result in increased signal power. NAND-based detection scheme results in increased signal-to-noise ratio, and thus, improved system performance for SAC systems. In this technique, only desired optical spectral chip is filtered, and thus, the resultant signal is free from any MUI and PIIN is suppressed at the receiving end. By following all the factors, it can be stated that SAC OCDMA performance is enhanced with NAND based detection [21]. Figure 8 shows that the NAND subtraction received signal has been divided into two blocks. First block decoded by an optical filter FBGs and the same structure as that used in the encoder, whereas the second block decoder also called NAND decoder has a structure of NAND filter [22]. NAND decoder is based on NAND operation of NAND logic and not basically a NAND gate. Overlapping wavelengths or interference from other users are subtracted in optical subtractor to eliminate MAI and finally detected by two photo detectors. Receiver complexity and system cost increase due to subtraction technique, but it provides accuracy in solving MAI-related issues by making use of the complete weight present in the user code sequence and thus suppress MAI. In the NAND subtraction detection, larger code weights are generated which causes signal power to increase thus results in high SNR and improved system performance when compared to AND, complementary, and XOR subtraction.

3.7 Direct Detection Technique DD technique is shown in Fig. 9. It is applicable for codes with zero cross-correlation properties only. Signals can be detected from clean chips only, and no subtractor is required. One pair per user of detector and decoder is required when compared with CS detection process, which used two pairs per user [23]. Thus, numbers of components are reduced in this process. In addition, receiver is cost effective, simple, and reliable due to implementation of DD technique at receiver side of SAC OCDMA system. Number of PIN detectors are just half of that used in complementary detection

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Fig. 9 Receiver based on direct detection technique

process. Moreover, DD technique effectively reduces PIIN due to presence of lesser number of chips at the photo detector [24–27].

4 Comparative Analysis of Detection Techniques We considered two different OCDMA sequences X = (1100) and Y = (0110), and these sequences are modulated using user data and multiplexer. Based on type of detection method, correlation difference between spectral chips is calculated and is illustrated in Table 1.

5 Conclusion In this paper, we have performed a comparative analysis of various detection techniques utilized in SAC-based OCDMA system. Each detection technique-based receiver design is compared with existing techniques. Moreover, for each technique, computation of cross-correlation is done by utilizing two different OCDMA code sequences. This study further provides knowledge of particular detection technique based on specific application. From the analysis, we can say that each detection technique provides a different implementation cost based on number of components utilized and each technique covers a different fiber distance. However, the DD technique is mostly preferred among all the techniques as it uses fewer photo detectors, power splitters, and subtractors; therefore, it provides lowest cost and less complex receiver architecture when compared to existing detection methods. DD provided better performance in terms of PIIN elimination by detecting clean chips only; thus, system performance is improved in this case. Furthermore, full code spectra are transmitted to recover the signature sequence resulting in higher received signal power when DD technique is used.

Z

0

0

1

Z = θX Y − θX Y = 0

θX Y = 1

1

1

λ2 1

0

λ3

Z NAND = θ X Y − θ(X =0  Y )Y

θ(X =1  Y )Y

θ X Y = 0011

1

0

θ X Y = 0011

1

0

θX Y = 1

0

Y

1

θX Y = 1

1

X

λ1

λ4

NAND subtraction

λ3

λ1

λ2

Complementary subtraction

0

0

λ4 1

0

λ2 1

1

λ3

Z AND = θ X Y − θ(X & Y )Y = 0

θX & Y = 1

X & Y = 0010

θX Y = 1

0

0

λ1

AND subtraction

Table 1 Comparison of various complementary, AND-, NAND-, and XOR-based detection techniques

0

1

λ4

λ2 1

0

λ3 1

1

Z XOR = θ X Y − θ(X ⊕Y )Y = 0

θ X ⊕Y = 1

X ⊕ Y = 0101

θX Y = 1

0

0

λ1

XOR λ4 0

1

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References 1. Fouli K, Maier M (2007) OCDMA and optical coding: principles, applications, and challenges. IEEE Commun Mag 45(8):27–34 2. Heritage JP, Weiner AM (2007) Advances in spectral optical code-division multiple-access communications. IEEE J Sel Top Quantum Electron 13(5):1351–1369 3. Kaur N, Goyal R, Rani M (2017) A review on spectral amplitude coding optical code division multiple access. J Opt Commun 38(1):77–85 4. Kavehrad M, Zaccarin D (1995) Optical code-division-multiplexed systems based on spectral encoding of noncoherent sources. J Lightwave Technol 13(3):534–545 5. Sharma T, Maddila RK (2019) Simulative investigation of spectral amplitude coding based OCDMA system using quantum logic gate code with NAND and direct detection techniques. Curr Opt Photonics 3(6): 531–540 6. Nisar KS (2017) Numerical construction of generalized matrix partitioning code for spectral amplitude coding optical CDMA systems. Optik 130:619–632 7. Abd TH, Aljunid SA, Fadhil HA, Ahmad RB, Rashid MA (2012) New approach for evaluation of the performance of spectral amplitude coding-optical code division multiple access system on high-speed data rate. IET Commun 6(12):1742–1749 8. Sharma T, Maddila RK (2020) Optical code construction based on enhanced quantum logic gate (EQLG) technique for spectral amplitude coding optical CDMA systems. Wirel Pers Commun 113:2587–2609 9. Bouarfa A, Kandouci M, Djellab H (2017) A new MIHP code using direct detection for SAC OCDMA system. Int J Electr Eng Inf 9(4):825–833 10. Ahmed HY, Zeghid M, Imtiaz WA, Sharma T, Chehri A, Fortier P (2020) Two-dimensional permutation vectors’ (PV) code for optical code division multiple access systems. Entropy 22(5):576 11. Alhassan AM, Badruddin N, Saad NM, Aljunid SA (2013) Intensity noise mitigation in SAC OCDMA systems using a divided spectrum balanced detection scheme. In: 2013 IEEE Photonics conference, IPC 2013, vol 5, pp 523–524 12. Ahmed N, Aljunid SA, Fadil A, Ahmad RB, Engineering C, Rashid MA (2011) Hybrid OCDMA/WDM system using complementary detection technique for FTTH access networks. In: 2011 IEEE Symposium on industrial electronics and applications, pp 227–230 13. Chen H, Xiao S, Yi L, Shi J, Yang P, Dong Y, Hu W (2012) Complementary decoder based on polarization modulation for the SAC OCDMA PON. IEEE Photonics Technol Lett 24(5):335– 337 14. Mostafa S, El-samie FEA (2017) Performance evaluation of SAC OCDMA system in free space optics and optical fiber system based on different types of codes. Wireless Pers Commun 96(2):2843–2861 15. Al-Khafaji HMR, Aljunid SA, Fadhil HA (2011) Performance enhancement of SAC OCDMA system using modified-AND subtraction detection. In: ICCAIE 2011—2011 IEEE Conference on computer applications and industrial electronics, pp 412–415 16. Al-Khafaji HMR, Aljunid SA, Amphawan A, Fadhil HA (2013) SOA/SPD-based incoherent SAC OCDMA system at 9 × 5 gbps. IEICE Electron Express 10(5):1–6 17. Al-Khafaji HMR, Aljunid SA, Amphawan A, Fadhil HA (2012) Improving spectral efficiency of SAC OCDMA systems by SPD scheme. IEICE Electron Express 9(24):1829–1834 18. Hmud IS, Hasoon FN, Hatim FA, Ghazi Zahid AZ, Shaari S (2010) AND-subtraction detection modified SNR equation for optical CDMA systems in PON applications. Opt Appl 40(1):49–55 19. Anuar MS, Aljunid SA, Arief AR, Junita MN, Saad NM (2013) PIN versus Avalanche photodiode gain optimization in zero cross correlation optical code division multiple access system. Optik 124(4):371–375 20. Ahmed HY, Almaleeh M, Fadhil H, Aljunid SA, Elyass A, Saad NM (2012) Performance analysis of spectral-amplitude-coding optical CDMA systems with new subtract exclusive or detection (SED) using vectors combinatorial (VC) code. Optik 123(15):1352–1359

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21. Mostafa S, Mohamed AENA, Abd El-Samie FE, Rashed ANZ (2015) Performance analysis of diagonal eigenvalue unity (DEU) code using NAND subtraction and spectral direct detection techniques and its use with triple-play-services in SAC OCDMA. Wireless Pers Commun 85(4):1831–1849 22. Ahmed N, Aljunid SA, Fadil A, Ahmad RB, Rashid MA (2013) Performance enhancement of OCDMA system using NAND detection with modified double weight (MDW) code for optical access network. Optik 124(13):1402–1407 23. Garadi A, Djebbari A, Abdelmalik TA (2017) Exact analysis of signal-to-noise ratio for SAC OCDMA system with direct detection. Optik 145:89–94 24. Fadhil HA, Aljunid S, Ahmad R (2009) Performance of random diagonal code for (OCDMA) systems using new spectral direct detection technique. Opt Fiber Technol 15(3):283–289 25. Ahmed HY, Zeghid M, Imtiaz WA, Sharma T, Chehri A (2020) An efficient 2D encoding/decoding technique for optical communication system based on permutation vectors theory. Multimed Syst 1–17 26. Sharma T, Chehri A, Fortier P, Ahmed HY, Zeghid M, Imtiaz WA (2021) Optical code construction of 2D spectral/spatial BIBD codes for SAC-OCDMA systems. Appl Sci 11(783):783 27. Imtiaz WA, Ahmed HY, Zeghid M, Sharma T, Mirza J (2021) Techno-economic analysis of two dimensional optical code division multiple access systems. 53(8):1–15

Classification of Leaf Disease on Using Triangular Thresholding Method and Machine Learning Deepak Kumar Saxena, Deepak Jhanwar, and Diwakar Gautam

Abstract Plant Leaves recognition and estimation is one of the most problematic companies in image processing techniques. This article subtleties techniques and methods are investigated to distinguish and estimate the severity of the disease caused by growth patterns on leaves of plants using threshold triangular strategy. Four set of images obtained from several species of plants and proposed analysis led to identify and measure the extent of the damage caused by disease-causing organisms to the leaves. Experimental research performed using machine learning techniques and final classification result being 97% accuracy. Keywords Division · Thresholding · Picture securing · Leaf sickness · AI

1 Introduction In agriculture this year ended up being breathtakingly more critical than it used to be a couple of years ago, where plants are simply used to support people in a similar way to animals. This is the result of how plants can generate power and different sources of essentiality to improve the living conditions of humanity. Anyway, there are countless such infections that impact plants that can make appreciable harm diverse economies and social demands. It can even cause remarkable regular setbacks. Consequently, it is definitely brighter disease conducive to rest and avoid such losses. Afflictions of the plant can be perceived through a couple of techniques including manual systems-based and PC. Most plant diseases appear as spots on the leaves, which are continuously undoubtedly a human eye. On the other hand, there are a couple of ailments that do not appear on the leaves and others appear in the later stages when they have little damage to the common plants. In such cases, it is recommended that the robotized structures would be the first option of receiving the service condition using a type of apparatus complex content and logic, in a perfect world using D. K. Saxena (B) · D. Jhanwar · D. Gautam Electronics and Communication Engineering, Engineering College, Ajmer, India D. Gautam e-mail: [email protected] © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_8

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the amplification striking focal point and several machines. In some various events, signals must be recognized through electromagnetic techniques, which transmit more images that are not unquestionable to the human eye. Another technique for accomplishing this is through the system known as Remote Sensing Technique (RST) by removing the ends and use of multiple images gets absurd hyperactive. All methods that use the RST approach generally fall into reducing the edge of the image to get ready instruments to achieve their best results. Most of the afflictions that plants are achieved by impact events appear as spots on the leaves of plants. These points make it difficult for these plants to establish their livelihood through techniques for photosynthesis, and green hues that impact sheet is therefore an enormous degree of impact and improving the performance of these plants. In conditions where the ends of disease forms of genuine life, surface stains sheet extend. The plant diseases not simply decrease can still produce the same manner rot variety of such plants and eliminating improvement. Plant diseases, especially the afflictions of the sheet are controlled normally using errors showers, fungicides and pesticides. A segment of these techniques the bounding box is consolidated, scanning time, the color, the support vector machine and neural networks. After all, none of these methods has achieved by several experts has been remarkable. This paper attempts to quantify and assess recognize the severity of illness caused by changes in threshold leaves triangular system. What makes this exceptional technique is the way it is simple, easy to use and provides accurate results.

2 Literature Survey 2.1 Image Processing Methods Rao (2014) showed two main techniques for traffic image. In his article, which is represented as preparing the analog image host. This process change, adjustment and image modification through electrical suggests strategies. A common example of this system is the image made by the TV. The flag of television broadcasts in a kind of tension that moves in the direction of plentifulness wonder image. The writer continues with the second recognized method as the foreground image technical Care. For this circumstance, the image will accept the change or a change in the electronic structure through a device known as a digitizer scanner for further care.

2.1.1

Picture Division

Ballard and Brown (1982) represents the division of the image as the delivery route or break an image in different parts depends on the explicit characteristics. The parties generally conform to something that individuals can, without much of a discrete segment, see and research as individual things. Mechanized the PC as more likely to

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consider not have the option to view the questions and certainly no one else. This is the explanation of the different researchers who have turned out different philosophies and methods of image segmentation. Images more pieces were divided according to the different qualities and characteristics found in the image. These features can be combined hidden information used for histogram showing the cutoffs of pixel information and surface information.

2.1.2

Image Thresholding

Threshold image mode involves producing a (bitonal) image matched by setting an initial step that fills as the base power estimated pixel of the main image. Thresholds of the system are done in grayscale; however, the threshold may be associated special image (true nature) with. The edge of each image estimate or maybe physically or by a calendar or application specific set normally. For this circumstance, all pixels that fall below the containment tip penalty set changed to attenuate bit addresses estimated zero, while some different pixels in the edge value are changed to cope with a white piece of an estimate.

2.1.3

Image Thresholding Counts

There are a number of these experts who have proposed different threshold counts refers degree image. Bit expects this assessment to discuss and set aside a portion of these counts as proposed by specific researchers. Surely, picking up the right to a proper calculation is an inconvenient action. This is a direct result of the range is closer to the view of the different calculations as they expect that contrast with the background image. Ridler and Calvard (1978) coordinate an assessment of plant diseases using threshold frame image. They proposed an estimate using the iterative approach to managing the disease collection of reality revision on the leaves. In their system, which uses an approximate advantage, for example, the average power of the hidden image as the edge ratio.

2.2 Measuring Disease Earnestness on Leaves Pradnya Narvenar et al. (2014) suggests a role in the analysis and leaf disease recognition system using a sequence SGDM methodology. In the evaluation, a system is disclosed techniques for the assessment of foliar diseases. Liu et al. (2009) investigated the methodology for leaf disease of rice depending on the characteristic feel pain threshold tone districts sheet using this method. Zhang (2005) investigated cucumber to choose technical support to diagnose the state of green plants using artificial vision development. The result showed a typical relationship between components fragments and other green sheets with the nitrogen that rapidly could be

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used as diagnostic harvest disorders pointer comparative conditions. Chunhua et al. (2004) use two particular characteristics of cucumber ROB perceive edges deficiency statistical system purge, using the characteristics of the structure Ohta.

2.2.1

Methods of Quantifying Leaf Disease Severity

There are a couple of methods used by the different authorities to assess the reality of leaf disease. One of these methodologies described herein. Leaf plant diseases can be evaluated and considered whether the affected domain or level of disease (the importance of the company is) on the sheet can be evaluated through features such as masking and surface states. Most systems and calculations used to assess the reality of the disease consolidate a division dare to secrete reactions in order to properly focus the characteristics of the process and characteristics to achieve a measure of the severity of the infection. In these techniques and work procedures used by another researcher, it depends on the specific philosophy used to check condition actually included.

2.2.2

Using Neural Network

Neural system is a structure of artificial brains hoping to duplicate the human brain. Bernard Widrow of Stanford College neuronal previously with 1950s mid-neuronal framework used in various fields in enrolling more especially in the structures of the phrase, statement imaging systems, mechanical innovation, medicine, etc. Pydipati et al. (2005) perceived and mounted a pair of diseases in citrus plants using two one of gender strategies. Features are added as a form, concealment, and measure the surface, four different small meeting these characteristics were made, and two control philosophies. Nobis Mahala classifier uses less detachment in his first procedure, along with a rule known as the rule of nearest neighbor to perceive the disease. The subsequent advance was the representation that is used for solving capabilities spiral reason (RBF) neural classifiers count back up the system encourage plants to collect some similar characteristics. They closed by communicating graphical representation carries both procedures identical results.

2.2.3

The Use of Support Vector Machines (SVM)

Models of learning support vector machines are related charges analyze data in order to request and dismemberment of learning objects. They are subject to discriminative classifiers plans describing the limits of decision-making. Youwen et al. (2008) use an estimate based on the statement of the plan to receive a pair of afflictions calculating the impact cucumber leaves. The calculation is used to segment the leaves in sound and ended the territories. This is followed by evacuation characteristics covering, the shape and the image surface. The final application is made by supporting features

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in the SVM. They concluded that through communication, Bolster Vector Machine products favored results in neural frameworks that are based on experimentation. Camargo and Smith (2009) treated the same way of perceiving and controlling the level of defect in cotton plants. Images were incorporated from the stem, leaves and the results of the cotton plant for preliminary. The images were separated using a system that had developed before official threshold was represented in this document. Then I sent a couple of features of unwanted territories. These features are used to support the SVM for the identification and estimation of the unfortunate parts of the photos. Jian and Wei (2010) showed a document using the SVM for diseases of cucumber leaves. They used the procedure section and leaves districts essential undesirable sound threshold. This was followed by the isolation characteristics as concealment, the shape and area of the image for additional ready. These characteristics are maintained in the capacity of the premise SVM coil (RBF), as part of the latter application is developed. In general, attempts to test to measure the amount of pollution caused by developments to impact plants more particularly, the leaves of the plants. Evaluation specifically attempts to: 1. 2. 3.

Perceive and recognize parasites causing disease in plant leaf. Evaluate and measure the reality of parasites caused disease on leaves of plants. Prescribe the sum adjustment and assembly of fungicides for use in the leaves of plants subject to serious disease.

3 Proposed Approach 3.1 Introduction This part of the thinking on the principles, frameworks and methods of reasoning used to achieve results and consequences of the typical investigation. These methods consolidated collection system of the photos above, procedures for the division and thresholding the image, and the calculation to verify the severity of the contaminated area of the sheet through the image attached. Van with equipment and materials used for the task; leaves of plants contaminated tip test, 20 tests, computerized camera, opaque sheet/texture, light structure, PC programming and MATLAB R2015a Form.

3.2 Principles and Systems Three important measures related to the realization of this task. It fuses image acquisition, the division of the image in the region conclusions separator sheet. In any case, a variety of measures such as the differences are also seen in the picture in a variety of structures.

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Image Securing

Leaves of suspected plants are used to complete this task. Processing chamber is used for imaging under controlled conditions with the drilling. The photos were viewed on any JPEG or PNG meeting. First, the road was carried out on the whole although the lighting level (matte texture) with the correct proportion of light sources annihilated. This was to get rid of any reflection of light and get similarly extend anywhere to improve vision and quality of the image to be prepared. Article (sheet) extended properly with the camera drive to ensure that the photo taken contained only leaf and boring establishment as the actual image. B.

Image Division

The division step was used to segregate the image into different locale subject to similar characteristics in the image. These regions show specific and novel qualities from one another and should not meet one another. All areas should represent a component of consistency in the region. The division experienced two distinct stages arranged for pixels sheets hard and fast and pixels in pain leaf. For the purposes behind divider setting, the image structure is changed to RGB gray scale. This can be cultivated by finding the typical three sections that hide the true nature. First get estimates alleged image evacuate red, the color of green and blue pixel by using the numbers are ultimately replaced the first RGB values with new features. The change is made by the ordinary management of the three. If you have a color image as the image shown in Fig. 1 and want to convert grayscale using the average method, the following result is achieved as shown in Fig. 2. One thing is that the normal strategy, does not give accurate results despite the fact that it works splendidly. This is because these figures strategy, the normal of the three colors. Since the three colors have different wavelengths and contribute to the design of the image, which erroneous results. This can be modified by the normal registration depends on the commitment of each shading in the image. Scientifically, this can be communicated as: Fig. 1 Sample true color image

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Fig. 2 Gray scale of Fig. 1

G 1 = (R + G 1 + B)/3 where G1 = dim, R = red, G2 = green and B = blue. Henceforth, Gray = (Red + Blue + Green)/3 Calculation to accomplish this is expressed underneath with respect to every pixel in picture { Red = pixel. Red Green = pixel. Green Blue = pixel. Blue Dim = (Red + Green + Blue)/3 Pixel. Red = Gray Pixel. Green = Gray Pixel. Blue = Gray } C.

Diseased Area Division

To achieve accurate results, there is a need to acquire the single image area of the contaminated area. Arguing that proper thought is not taken, the division probably an immediate consequence of concealing range is not corrected. If the midrib of the leaf (midrib) is shallower than the sheet itself, you can encourage inaccurate

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results. Furthermore, considering the foliar diseases experience different periods of the disease as a result of components, for example, water, light and neutrinos, shows irritated different signs, making the disadvantage methodology division. Given the above factors, it is smarter to change the image to hide their space from RGB space to hide obvious to the human eye. Calculation 1. 2. 3. 4. 5. 6. 7. 8. 9. 10. 11. 12.

Check to guarantee that the client has introduced Image handling Toolbox in MATLAB Load the picture into MATLAB (with full document area) Get the elements of the picture Convert the picture to HSV shading space Calculate the dark pixels Find just the dark that is outside the leaf, not inside the leaf Mask the H, S, and V pictures Plot the histogram of the tone zone Call anything with a tone of somewhere in the range of 0.15 and 0.5 “sound” Call whatever else (that isn’t foundation) “sick” Compute the unhealthy zone portion end

Mathematically, algorithm for estimating diseased severity can be expressed as: DS = DA/TA = P1/P1

(1)

(X, Y ) ∈ DR(X, Y ) ∈ LR = 1/1 (X, Y ) ∈ Rd(X, Y ) ∈ Rl = Pd/Pl

(2)

where DS P DA TA DR LR Pd Pl

Disease severity, Unit pixel value, Diseased leaf area, Total leaf area, Diseased region, Leaf region, Total pixels in diseased area, Total pixels of the leaf.

Each pixel of the image is unvalued therefore it is simpler to determine the diseased portion by accounting the region numbers and relate it to the ratio of the total leaf area as shown in the algorithm above (Figs. 3 and 4).

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START Image Acquisition

Image Segmentation

Leaf Region Segmentation

Disease Region Estimation

Exit Fig. 3 Figure represents all the stages to go through in order to achieve the accurate results of this project

Fig. 4 System shows leaf disease detection of Proteus vulgaris using digital image processing techniques

4 Results and Implementation Color saturation in image processing and graphics used to describe the intensity of the color in a particular image. An image is too bright saturated colors. The more saturated a color, the most vivid than it seems (Fig. 5). Gray is proximal to less saturated region. The answer is ambiguous to say how saturated colors should be present in the image. It depends on the point of view of the image. When there is a high saturation, which resembles indeterminist. It is also

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Fig. 5 Image enhancement

difficult to determine the characteristics of image printing images. However, it is said that a lower saturated image be opaque, while washings images resembles the impression of being smooth. The image (A) becomes saturated form to evaluate the intensity of the color image (Fig. 6). Applicable Sample applied with proposed model and produce relevant outcomes which are comparable with base model.

Fig. 6 System shows leaf detection of Camellia assamica using digital image processing techniques

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Table 1 Comparison of healthy part and affected part of leaf Sample leaf disease

Affected part % Proposed work

SVM model

Proposed work

SVM model

Sample covered

Alternaria alternata

34.22

20.1

65.78

79.9

100

Anthracnose

48

76.63

52

23.37

100

Bacterial blight

33.33

24.8

66.67

75.2

100

Cercospora leaf spot

47.69

32.27

52.31

67.73

100

Affected Part Leaf Area in %

Fig. 7 Plot of affected part of leaf

Healthy part %

100 50 0

Proposed Work SVM Model Disease Type

Healthy Area Leaf Area in %

Fig. 8 Plot of healthy part of leaf

150 100 50 0

Proposed Work SVM Model Disease Type

4.1 Result Comparisons See Table 1; Figs. 7 and 8.

5 Conclusion This work is a method to quantify the severity of organisms’ ailments caused in the sheet. Examination thought about the negative impacts of foliar diseases in plants. Studies have shown that the results of disease leaves of the plant are extraordinary and difficult to handle. However, leaf diseases, more particularly those caused by parasites can be estimated and evaluated to ensure fair use and enough of the right convergence of fungicides to keep out agricultural. Unfortunately, excessive use of synthetic compounds in agricultural products. Plant diseases not only reduce

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your articles, but also have withdrawn their variety and development. The use of pesticides and fungicides in abundance for the treatment of these infections, the risk of accumulation of toxic level is based on horticultural articles and has distinguished itself as a major supporter of pollution of groundwater. Again, farmers acquire a lot of problems because of the costs of these pesticides applied in plants. In this sense, there is a need to take warning more prominent and limit their use to ensure water bodies and human life. The calculation used in this test ends being perhaps fewer complex methods to identify and evaluate leaf diseases in plants. Also, it ends up being really outstanding with regard to accuracy as it works up to 97% accuracy in terms of results.

References 1. Al Bashish D, Braik M, Bani-Ahmad S (2011) Detection and classification of leaf diseases using K-means-based segmentation and neural networks based classification 2. Doudkin AA, Inyutin AV, Petrovsky AI, Vatkin ME (2007) Three level neural network for data clusterization on images of infected crop field. J Res Appl Agric Eng 52(1) 3. Woodford BJ, Kasabov NK, Wearing CH (1999) Fruit image analysis using wavelets. In: Proceedings of the ICONIP/ANZIIS/ANNES 4. Lu C, Ren H et al (2010) Leaf area measurement based on image processing. IEEE, pp 580–582 5. Hu CH, Li PP (2004) Application of computer image processing to extract color feature of nutrient deficiency leaves. Comput Meas Control 9:859–862 6. Kim DG, Burks TF, Qin J, Bulanon DM (2009) Classification of grapefruit peel diseases using color texture feature analysis. Int J Agric Biol Eng 2(3). Open access at http://www.ija be.org. Jenson JR (2003) Digital image processing—a remote sensing perspective, 3rd edn. Prentice-Hall. Castleman KR (1996) Digital image processing. Prentice-Hall 7. Jain AK (1989) Fundamentals of digital image processing. Prentice-Hall. Kondou H, Kitamura H (2002) Shape evaluation by digital camera for grape leaf. Sci Technol Promot Center 586–590 8. Rao KMM et al (1997) Design and fabrication of color scanner. Indian J Technol 15 9. Rao KMM (1996) Image processing for medical applications. In: Proceedings of 14th world conference on NDT, 8th–13th Dec 1996 10. Rao KMM (1995) Medical image processing. In: Proceedings of workshop on medical image processing and applications, 8th Oct 1995. NRSA, Hyderabad-37 11. Wang K, Li SK, Wang CT (2006) Acquired chlorophyll concentration of cotton leaves with technology of machine vision. Acta Agron Sin l:34–40 12. Li, Zhou GM (2009) Research on image feature extraction of crop disease. Trans CSAE 2S:213– 217 13. Babu P, Srinivasa Rao B (2007) Leaves recognition using back propagation neural networkadvice for pest and disease control on crops. Expert Advisory System, IndiaKisan.Net 14. Maliappis KP, Ferentinos HC, Sideridis PAB (2008) Gims: a web based greenhouse intelligent management system. World J Agric Sci 4(5):640–647 15. El Helly M, Rafea A, El-Gammal S. An integrated image processing system for leaf disease detection and diagnosis 16. Marcon M, Mariano K et al (2011) Estimation of total leaf area in perennial plants using image analysis. Rev Bras Eng Agric Ambient 15:96–101 17. Siddiqi MH, Sulaiman S, Faye I, Ahmad I. A real time specific weed discrimination system 18. Tzionas P, Papadakis SE, Manolakis D (2005) Plant leaves classification based on morphological features and fuzzy surface selection technique. In: 5th International conference ON technology and automation ICTA’05, Thessaloniki, Greece, pp 365–370, 15–16

Effect of Code and Frequency Index Modulation in MIMO-OFDM-FSO System Chinmayee Panda and Urmila Bhanja

Abstract This paper analyses the MIMO-OFDM-FSO system using code and frequency index modulation. In this scheme, a joint code and frequency index modulation (CFIM) is used that enhances spectral and energy efficiencies. In weak turbulence condition for a particular spectral efficiency value, a comparative analysis is done by taking conventional OFDM, CFIM scheme with OFDM and MIMOOFDM system in free space. The CFIM-MIMO-FSO scheme exhibits the lowest BER as compared to conventional OFDM and CFIM-OFDM scheme. Additionally, the PAPR reduction is observed by using CFIM-MIMO-OFDM-FSO scheme in free space. Keywords CFIM · MIMO-OFDM · Gamma-Gamma · FSO

1 Introduction Free space optical communication is the system where the optical signal transmitted from transmitter to receiver end through space. The main advantages of FSO are the license free spectrum and larger data rate [1]. The free space optical communication system is affected by rain, snow, fog and atmospheric turbulences. Various modulation techniques are applied to free space for long distance communication. BPSK, QPSK, DPSK, SIM are the prime modulation schemes applied to free space [2]. Orthogonal frequency division multiplexing is another important multiplexing technique which is applied to FSO in order to save the bandwidth and minimize the inter symbol interference (ISI). In MIMO the time diversity, frequency diversity and space diversity schemes are used to reduce the receiver complexity and to increase high throughput. Various researches going on by taking MIMO-FSO system using Gamma-Gamma channel model and lognormal channel model. MIMO system increases the efficiency of the system. Various linear combining techniques such as maximum ratio combining (MRC), equal gain combining (EGC), and selection

C. Panda (B) · U. Bhanja Department of Electronics and Communication Engineering, IGIT, Sarang, Odisha 759146, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_9

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combining (SC) techniques are applied to the receiver section [3]. Experimental evaluations are performed in FSO system by utilizing different coding techniques such as convolution codes, turbo codes, low-density parity-check codes and Reed-Solomon codes [4]. The combination of MIMO-OFDM system be utilized in fifth-generation (5G) communication systems. High throughput with high-speed data obtained by using the combination of MIMO and OFDM [5]. Another modulation technique called index-based modulation associated with orthogonal frequency-division multiplexing (OFDM) is used which enhances the performance of OFDM and in FSO [6]. Another efficient modulation technique named code and frequency index based modulation system where the subcarriers are separated into many blocks and the fraction of subcarriers contained in each block is considered depending on indexing bits [7]. The symbols are transmitted by the activated subcarriers. In this joint modulation technique of code and frequency modulation the code index increases the spectral efficiency and the frequency index reduces BER. The main contributions of this paper are mentioned as below. 1.

2.

The code and frequency index based modulation system (CFIM) are simulated for single input and output system (SISO) and extended to multiple inputs and multiple output (MIMO) system. The CFIM technique is applied to FSO under weak turbulence conditions under Gamma-Gamma channel.

The remaining part of the paper is arranged as follows. Section 1 introduces the CFIM concept, Sect. 2 describes the system model, Sect. 3 describes the channel model and evaluates the PAPR, BER, spectral efficiency, Sect. 4 simulates the result, and Sect. 5 concludes the paper.

2 CFIM-MIMO-OFDM-FSO System Model Figure 1 shows the block diagram of MIMO-OFDM-FSO system. The system shows the CFIM transmitter with M × N subcarriers. Here, a sequence of data is equally divided into M blocks. Since each block of CFIM has the same processing procedure, M th block is considered for easiness. In the transmitter section, all data are split into M blocks starting from d 1 to d m and each consisting of three sub blocks such that d1 = d1t1 + d1t2 + d1t3 . The three sub blocks of d 1 to d m are transmitted as frequency index selection, code index selection and M-ary modulator. The spreading codes C 1 to C m are chosen from the code block section. The code selected data and mary modulated data are given to the spreader as shown in figure and the combined data from frequency index selection section and spreader are given to the IFFT [7]. X 1 IFFT block consists of summation of all code fragments and frequency indexes and similarly X m . Afterward, the remaining procedures are similar to that of MIMOOFDM system. In this work, the channel is assumed to be Gamma-Gamma distributed and the system is analyzed under weak turbulence condition. The signal is received

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Fig. 1 Transmission section of CFIM-MIMO-OFDM-FSO system

by the MIMO antennas, and the receiver section follows the reverse operation of transmitter section.

3 Channel Model with Different Parameter Evaluation 3.1 BER Probability of the System In the CFIM scheme, the blocks of transmitted data are partitioned into three parts where the mapped bits are represented by two blocks and the modulate bits are identified by a single block. The probability of bit error of the CFIM system is calculated by taking the probability of bit error of the mapped bits Bmap and the probability of bit error of the modulated bits Bmod and can be represented by Eq. (1) [7], BCFIM =

t1 t 2 + t3 B mod + Bmap t t

(1)

where Bmod and Bmap are denoted as the number of modulated bits and the number of mapped bits, respectively.

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3.2 Spectral Efficiency The spectral efficiency of CFIM system having M × N subcarriers given by Eq. (2) [8], ξCFIM =

t 1 + t2 + t3 N

(2)

where N number of subcarriers are activated in each block. By increasing the number of subcarriers would minimize the spectral efficiency, whereas by increasing the order of modulation and the number of spreading codes maximize the spectral efficiency. Hence, it would affect the net performance of the system.

3.3 PAPR Evaluation The major disadvantage of multicarrier systems is the high peak to average power ratio (PAPR). The performance of the signal in terms of energy efficiency is distorted by PAPR factor. PAPR reduction saves power which increases the energy efficiency. By activating large number of subcarriers PAPR is increased. A single frequency in each block is activated in the CFIM scheme which leads reduction in PAPR. The PAPR is defined in Eq. (3) [9],   max z(t)2  PAPR =  T 1/Ts 0 s z(t)2 

(3)

where T s is the time span of a CFIM symbol and z (t) is the complete CFIM transmitted signal.

3.4 Gamma-Gamma Model The probability density function (PDF), Df for Gamma-Gamma distribution is given in Eq. (4) [10].   α+β 2(αβ) 2 × s 2 −1 kB αβs (α)(β) α+β

Df( f ) =

(4)

where α and β are the spherical wave constants and k B is the second-order Bessel functions, s denotes the scintillation factor. The optical signal distorted after transmitting through the atmosphere and the channel model is designed by combining the atmospheric attenuation and turbulence conditions.

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4 Simulation Result Figure 2 analyses conventional OFDM system, CFIM-MIMO and CFIM-OFDM system in FSO for a particular value of spectral efficiency ξ = 4 bps. It is observed that MIMO-FSO system in weak turbulence condition (C n 2 = 12 × 10−15 m−2/3 ) with CFIM exhibits BER of 10−4 as compared to conventional OFDM [11].

Fig. 2 BER versus SNR plot CFIM-MIMO-OFDM, CFIM-OFDM and OFDM in weak turbulence condition

Fig. 3 No. of users versus spectral efficiency

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Fig. 4 PAPR reduction in CFIM-MIMO-OFDM-FSO system

Figure 3 shows the spectral efficiency variation with respect to number of users. Spectral efficiency decreases by increasing number of users. Here, the modulation order is fixed at 2, Number of blocks (M) is fixed at 14, and number of subcarriers per block N is fixed at 4. It is observed from Fig. 3 that with 30 users CFIM-OFDM exhibits spectral efficiency of 2.4 bit/Hz and CFIM-MIMO-OFDM depicts spectral efficiency of 2.7 bits/Hz, which is much superior compared to the conventional OFDM-FSO system [12]. Hence, CFIM-MIMO-OFDM system is found to be more spectrally efficient. In Fig. 4, the simulation result shows that CFIM-MIMO-OFDM-FSO scheme exhibits the lowest PAPR reduction as compared to conventional OFDM and CFIMOFDM in FSO for FFT size 64. The probability of being high value of PAPR is reduced in MIMO-OFDM-FSO system using CFIM.

5 Conclusion In this paper, we have analyzed an efficient index modulation based system that remarkably increases the spectral efficiencies of the system with reduced BER. This scheme is based on the combination of code and frequency index modulation. The code index increases the spectral efficiency, while the frequency index provides better BER by minimizing high PAPR values. In weak turbulence condition by using CFIMMIMO-OFDM-FSO System BER of 10−4 is obtained, lowest PAPR is achieved and the system is found to be more spectrally efficient.

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References 1. Ochiai H, Imai H (2001) On the distribution of the peak-to-average power ratio in OFDM signals. IEEE Trans Commun 49(2):282–289 2. Sahu M, Kiran KV, Das SK (2018) FSO link performance analysis with different modulation techniques under atmospheric turbulence. In: Second international conference on electronics, communication and aerospace technology (ICECA), Coimbatore, India 3. Xu F, Khalighi A, Causse P, Bourennane S (2009) Channel coding and time-diversity for optical wireless links. Opt Express 17(2):872–887 4. Liu J, Zhang X, Blow K, Fowler S (2017) Performance analysis of packet layer FEC codes and interleaving in FSO channels. IET Commun 11(13):2042–2048 5. You W, Yi L, Hu W (2013) Reduced complexity maximum-likelihood detection for MIMOOFDM systems. In: 8th International conference on wireless communications, networking and mobile computing, Shanghai, China 6. Alhiga RA, Haas H (2009) Subcarrier-index modulation OFDM. In: IEEE 20th International symposium on personal, indoor and mobile radio communications, Tokyo, Japan 7. Au M, Kaddoum G, Gagnon F, Soujeri E (2017) A joint code-frequency index modulation for low-complexity, high spectral and energy efficiency communications. IEEE Trans Commun 8. Lee WCY (1987) New cellular schemes for spectral efficiency. IEEE Trans Veh Technol 36(4):188–192 9. Bhada S, Gulhanea P, Hiwaleb PAS (2012) PAPR reduction scheme for OFDM. Procedia Technol 4:109–113 10. Ahmadi S (2014) The Gamma-Gamma signal fading model: a survey. IEEE Antennas Propag Mag 56(5):245–260 11. Ajewole BD, Owolawi PA, Odeyemi KO, Srivastava VM (2019) Coded BPSK OFDM-FSO over strong turbulence channel. In: 2019 International conference on advances in big data, computing and data communication systems (icABCD), Winterton, South Africa 12. Bjornson E, Debbah M, Larsson EG (2016) Massive MIMO for maximal spectral efficiency: how many users and pilots should be allocated? EEE Trans Wireless Commun 15(2):1293–1308

The Structural, Electronic and Optical Properties of Partially Hydrogenated Germanene: A First-Principles Study Routu Santosh and V. Kumar

Abstract The structural, electronic and optical properties have been studied at different occupancies of hydrogen atoms upon germanene using first-principles density functional theory calculations. The electronic and optical parameters: energy bandgap (E g ), dielectric constant ε(0), refractive index n(0), absorption coefficient (E α ) and plasmon energy (èωp ) have been calculated for different occupancy of hydrogen for the first time. The calculated values for 100% occupation of hydrogen are in good agreement with the available experimental and reported values. The above-calculated parameters are essential in the fabrication of several optoelectronic devices and applications. Keywords Hydrogenated germanene · First-principles · Structural property · Electronic property · Optical property

1 Introduction In recent years, much attention has been given on graphene-like two-dimensional materials such as silicene and germanene due to their exceptional properties: linear dispersing energy bands, mass-less Dirac fermions behaviour of electrons, high electron mobility, unusual quantum hall effect, spin–orbit coupling, etc. [1–6]. These materials show semi-metallic nature, i.e., zero bandgaps at Dirac point K, which restricts their applications in the field-effect transistors, logic circuits and switching devices [7, 8]. To open bandgap, one of the main techniques used is chemical functionalization. This leads to adsorption of foreign atoms (H, F, Cl, O, B and N), several gas molecules (NH3 , NO, NO2 and O2 ) and alkali earth metals on pristine silicene and germanene and converts into insulators, semiconductors, semi-metals and metals [9–12]. Some theoretical predictions show that hydrogenated silicene shows indirect bandgap, which is not suitable for optoelectronic devices and applications [13–15].

R. Santosh (B) · V. Kumar Department of Electronics Engineering, Indian Institute of Technology (Indian School of Mines) Dhanbad, Dhanbad, Jharkhand 826004, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_10

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The direct optical band gap of 1.59 eV is observed in the case of hydrogenated germanene (germanane), which is demonstrated from topo-chemical de-intercalation of CaGe2 in aqueous HCl [16]. Yao et al. [17] synthesized germanane on Ge2 Pt crystal and observed the bandgap of 0.2 eV for half-hydrogenated germanene and 0.5 eV for fully hydrogenated germanene on scanning and tunnelling spectroscopy (STS). The structural, stability, electronic, magnetic and ferromagnetic properties of hydrogenated germanene nanoribbons studied by first-principle calculations [18– 20]. However, 100% hydrogenation is not possible as some defects may occur while adsorption of hydrogen on germanene. Therefore, authors have taken much interest to calculate several properties of hydrogenated germanene for different occupancies of hydrogen upon germanene. Recently, the authors have studied several properties of hydrogenated, fluorinated and hydrofluorinated graphene using density functional theory (DFT) calculations [21–23]. In this paper, the structural, electronic and optical properties have been studied at different occupancies of hydrogen atoms upon germanene using first-principles density functional theory calculations. The electronic and optical parameters: energy bandgap (E g ), dielectric constant ε(0), refractive index n(0), absorption coefficient (E α ) and plasmon energy (èωp ) have been calculated for different occupancy of hydrogen for the first time. The calculated values for 100% occupation of hydrogen are in good agreement with the available experimental and reported values.

2 Methodology The calculations have performed using first-principle calculations on Cambridge sequential total energy package CASTEP code [24]. The calculations are based on generalized gradient approximation (GGA) parametrized by Perdew-BerkeErnzerhof scheme [25]. The TS method has been used for DFT-D correction in calculation of structural, electronic and optical properties. The geometry optimization was performed for 100 iterations to obtain minimum total energy using Broyden, Fletcher, Goldfarb and Shanno (BFGS) scheme [26]. An ultra-soft pseudopotential representation has been used with the basis set kinetic energy cut-off of 353.70 eV [27]. The maximum tolerance of geometry minimization parameters: total energy convergence is 2 × 10−5 eV/atom, Hellmann–Feynman ionic force is 0.05 eV/Å, max ionic displacement is 2 × 10−3 Å and maximum stress component is 0.1 GPa, respectively.

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3 Results and Discussion 3.1 Structural and Electronic Properties The adsorption of hydrogen on pristine germanene changes the hybridization of germanium from sp2 to sp3 , which changes the structural, electronic and optical properties of germanene. The change of hybridization converts structure of germanene from hexagonal (space group = P63/MMC, 196) to trigonal (space group = P-3M1, 164) which has shown in Fig. 1a, b. The corresponding symmetrical constraints are α = 90°, β = 90° and γ = 120°, respectively. The optimized lattice parameter (a), bond lengths (d Ge–Ge and d Ge–H ) and bond angles (θGe--Ge--Ge and θGe--Ge--H ) are listed in Table 1, along with the reported values. The calculated bond length of Ge–H shows that the bonding between germanium and hydrogen is covalent or semi-ionic, while the bonding between carbon and hydrogen in hydrogenated graphene is covalent [21]. Table 1 shows that the bond length between germanium atoms of hydrogenated germanene is high compared to germanene. This is due to the electronegativity difference between hydrogen and germanene causes the depopulation of bonding orbitals

Fig. 1 a The top view of pristine germanene. The green circles indicate germanium atoms of sp2 hybridization with neighbouring atoms. b The perspective view of hydrogenated germanene in which white circles indicate hydrogen and green circles are germanium

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Table 1 Lattice constant (a), Ge–Ge bond length (d Ge–Ge ), Ge-H bond length (d Ge–H ), Ge–Ge–Ge bond angle (θ Ge–Ge–Ge ), Ge–Ge–H bond angle (θ Ge–Ge–H ), bandgap energy (E g ) and binding energy (E b ) of germanane and germanene Parameters a (Å) d Ge–Ge (Å) d Ge–H (Å)

Germanane

Germanene

This work

Reported

This work

Reported

4.059

3.880a , 3.93e , 3.908f , 3.95d , 4.082c

3.694

4.0j

2.454

2.390f ,

2.401e ,

2.522

2.382j

1.557

1.529d ,

1.56f ,

2.469c

1.563c

–

θGe--Ge--Ge (º) 111.487

108.95f

94.204

θGe--Ge--H (º)

107.372

109.99f

–

E g (eV)

1.010

1.59a , 0.5b , 1.38e , 3.6i , 3.5 g , 1.812f , 1.38d , 0.00 2.81h

E b (eV)

3.95

4.069f , 2.575d

0.00i

–

Experimental values a Ref. [16] and b Ref. [17], reported LDA c Ref. [18], d Ref. [23], e Ref. [13] and j Ref.

[13], reported GGA f Ref. [31], reported GW g Ref. [13] and h Ref. [23], reported HSE i Ref.

[13]

between germanium atoms. The calculated bond angles of hydrogenated germanene are near to diamond bond angle due to sp3 hybridization. The band structure of hydrogenated germanene has been calculated along with high symmetry points (G–M–K–G) with the q-point separation of 0.0151/Å in the Monkhorst pack grid and shown in Fig. 2. Houssa et al. [7] predicts the zero bandgap of germanene at Dirac point ‘K’ and express the linear Dirac-distribution of energy bands. The direct bandgap has been created for hydrogenated germanene at G-point with bandgap value of 1.0 eV without changing of linear distribution in the band structure. The density of states shows the orbital character in the formation of valence and conduction bands of the band structure. The partial density of states (PDOS) and Fig. 2 The calculated band structure of hydrogenated germanene which shows the energy band gap of 1.010 eV at g point

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Fig. 3 The calculated partial density of states (PDOS) of individual atoms and the total density of states (TDOS) of hydrogenated germanene

total density of states (TDOS) of hydrogenated germanene have been plotted in Fig. 3. The energy bands in the lower part of valence band are mainly contributed by H-1s and Ge-4s, and the upper valence band is dominated by Ge-3p and H-1s. The conduction band is primarily due to the influence of H-1s, Ge-3p and Ge-4s state, respectively. The Ge-3p and H-1s orbitals play a significant role in the formation of an energy bandgap between the valence and conduction bands.

3.2 Optical Properties The optical properties are calculated in perpendicular polarization of electric field on the plane of hydrogenated germanene. The momentum matrix elements and the imaginary part of dielectric function ε2 (ω) have been calculated using the relations given by Momida et al. [28]. The calculated ε2 (ω) for different occupancies of hydrogen is shown in Fig. 4a. The conduction in fully hydrogenated germanene started at 2.5 eV and rises rapidly due to the increase of transitions from the upper valence band to lower conduction band. It reaches to a maximum value at 6.8 eV due to the maximum number of transitions from H-1s to Ge-4p states and decreases with the further increase of photon energy. However, as the occupancy of hydrogen decreases, the starting point of conduction also decreases and comes toward the lowenergy region (redshift). This may be due to the decrease in optical band gap of the germanane. The low maximum values of ε2 (ω) for defective occupancy of hydrogen

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Fig. 4 a The imaginary part of dielectric function ε2 (ω) for different occupancies of hydrogen upon germanene. b The real part of dielectric function ε1 (ω) for different occupancies of hydrogen upon germanene. c The refractive index n(ω) for different occupancies of hydrogen upon germanene. d The electron energy loss function L(ω) for different occupancies of hydrogen upon germanene

such as 25, 50 and 75% are due to the lack of symmetry. The pristine germanene has low buckling compared to hydrogenated germanene [14]. Thus, the defect functionalized germanene is the combination of low- and high-buckling structure which is unsymmetrical. While in pristine germanene, the conduction starts nearly from zero point due to semi-metallic nature and reached to a maximum value at 3.5 eV then decreases with the further increase of energy. The real part of dielectric function ε1 (ω) is calculated using the Kramer-Kronig relation [29] and shown in Fig. 4b. The calculated dielectric constant ε(0) for different occupancies of hydrogen is listed in Table 2. The ε(0) may be decreased after the hydrogenation of germanene due to increase in energy bandgap. The refractive index and electron energy loss function have been calculated using the relation given by Sahin et al. [30] and shown in Fig. 4c, d. The calculated Refractive Index and plasmon energy for different occupancies of hydrogen upon germanene are listed in Table 2.

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Table 2 The calculated static dielectric constant ε(0), refractive index n(0) and plasmon energy (èωp ) for different occupancies of hydrogen upon germanene % occupancy of hydrogen

0%

25%

50%

75%

100%

ε(0)

6.52

3.50

3.56

2.97

3.86

n(0)

2.34

1.87

1.94

1.72

1.97

èωp

13.89

13.87

13.83

13.51

16.43

4 Conclusions The calculated lattice constants, bond lengths and bond angles for germanene and hydrogenated germanene are in good agreement with the earlier reported values. After hydrogenation, H-1s orbital hybridized with Ge-3p state which forms strong σ bond and plays a major role in the formation of energy bandgap between the valence and conduction bands. The dielectric constant ε(0) may be decreased after the hydrogenation of germanene due to creation of energy bandgap. This may increase the mobility of electrons in the material. The defects in hydrogenation form an unsymmetrical structure which affects the electronic and optical properties of hydrogenated germanene. The optical parameters: dielectric constant ε(0), refractive index n(0) and plasmon energy (èωp ) for different occupancies of hydrogen upon germanene have been calculated for the first time. The above-calculated values are much crucial in the fabrication of linear and non-linear optoelectronic devices and circuits.

References 1. Cahangirov S, Topsakal M, Aktu E, Ciraci S (2009) Two- and one-dimensional honeycomb structures of silicon and germanium. Phys Rev Lett 102:236804, 1–4 2. Zhuo Z, Wu X, Yang J (2018) Two-dimensional silicon crystals with sizable band gaps and ultrahigh carrier mobility. Nanoscale 10:1265–1271 3. Liu CC, Feng W, Yao Y (2011) Quantum spin hall effect in silicene and two-dimensional germanium. Phys Rev Lett 107:076802 4. Liu CC, Jiang H, Yao Y (2011) Low-energy effective Hamiltonian involving spin-orbit coupling in silicene and two-dimensional germanium and tin. Phys Rev B 84:195430 5. Ezawa M (2012) Valley-polarized metals and quantum anomalous hall effect in silicene. Phys Rev Lett 109:055502 6. Wang Y, Zheng J, Ni Z et al (2012) Half-metallic silicene and germanene nanoribbons: towards high-performance spintronics device. Nano Breif Rep Rev 5:1250037 7. Houssa M, Pourtois G, Afanas’ev VV, Stesmans A (2010) Electronic properties of twodimensional hexagonal germanium. Appl Phys Lett 96:082111 8. Lebègue S, Eriksson O (2009) Electronic structure of two-dimensional crystals from ab initio theory. Phys Rev B 79:115409 9. Pang Q, Zhang C, Li L, Fu Z, Wei X, Song Y (2014) Adsorption of alkali metal atoms on germanene: a first-principles study. Appl Surf Sci 314:15–20 10. Rubio-pereda P, Takeuchi N (2015) Adsorption of organic molecules on the hydrogenated germanene: a DFT study. J Phys Chem C 119:27995–28004

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11. Mortazavi B, Dianat A, Cuniberti G, Rabczuk T (2016) Application of silicene, germanene and stanene for Na or Li ion storage. Electrochim Acta 213:865–870 12. Pang Q, Zhang Y, Zhang J, Xu K (2011) Electronic and magnetic properties of pristine and chemically functionalized germanene nanoribbons. Nanoscale 3:4330–4338 13. Houssa M, Scalise E, Sankaran K, Pourtois G, Afanas VV (2011) Electronic properties of hydrogenated silicene and germanene. Appl Phys Lett 98:223107 14. Voon LCLY, Sandberg E, Aga RS, Farajian AA (2010) Hydrogen compounds of group-IV nanosheets. Appl Phys Lett 97:163114, 13–16 15. Takeda K, Shiraishi K (1989) Electronic structure of Si-skeleton materials. Phys Rev B 39:28– 37 16. Bianco E, Butler S, Jiang S, Restrepo OD, Wind W, Goldberger JE (2013) Stability and exfoliation of germanane: a germanium graphane analogue. ACS Nano 7:4414–4421 17. Yao Q, Zhang L, Kabanov NS et al (2018) Bandgap opening in hydrogenated germanene. Appl Phys Lett 112:171607 18. Wang XQ, Li HD, Wang JT (2012) Induced ferromagnetism in one-side semihydrogenated silicene and germanene. Phys Chem Chem Phys 14:3031–3036 19. Liu J, Yu G, Shen X et al (2017) The structures, stabilities, electronic and magnetic properties of fully and partially hydrogenated germanene nanoribbons: a first-principles investigation. Phys E 87:27–36 20. Xiao P, Fan X, Liu L (2014) Tuning the electronic properties of half- and full-hydrogenated germanene by chlorination and hydroxylation: a first-principles study. Comput Mater Sci 92:244–252 21. Santosh R, Kumar V (2019) The pressure effect on structural, electronic and optical properties of hydrogenated graphene: a first-principle study. J Electron Mater 18:770–778 22. Kumar V, Santosh R (2019) First-principle calculations of structural, electronic, optical and thermodynamic properties of fluorinated graphene. Mater Sci Eng B 246:127–135 23. Santosh R, Kumar V (2019) First-principle calculations of structural, electronic, optical and thermodynamic properties of hydrofluorinated graphene. Solid State Sci 94:70–76 24. Segall MD, Lindan PJD, Probert MJ et al (2002) First-principles simulation: ideas, illustrations and the CASTEP code. J Phys Condens Matter 14:2717–2744 25. Perdew JP, Burke K, Ernzerhof M (1996) Generalized gradient approximation made simple. Phys Rev Lett 77:3865 26. Vanderbilt D (1990) Soft self-consistent pseudopotentials in a generalized eigenvalue formalism. Phys Rev B 41:7892 27. Fischer TH, Almlof J (1992) General methods for geometry and wave function optimization. J Phys Chem C 96:9768–9774 28. Momida H, Hamada T, Takagi Y et al (2007) Dielectric constants of amorphous hafnium aluminates: first-principles study. Phys Rev B 75:195105 29. Kronig RL (1926) On the theory of dispersion of X-rays. J Opt Soc Am 12:547–557 30. Sahin S, Ciftci YO, Colakoglu K, Korozlu N (2012) J Alloys Compd 529:1–7 31. Trivedi S, Srivastava A, Kurchania R (2014) J Comput Theor NanoSci 11:1–8

Performance Evaluation of Routing Protocols in MANETs with Variation in Pause Time Suresh Kumar, Deepak Sharma, and Payal

Abstract With the advancement in technology, there is a need for dynamically changing network applications for efficient, seamless, and last mile connectivity for a cost effective solution. In mobile ad hoc networks (MANETs), the function and location of the randomly connected mobile nodes keep on changing based on the user necessity. The seamless and longer duration network connectivity depends upon mainly on density of nodes, their mobility, speed, pause time, and transmission power. In this research work, three routing protocols (RP), i.e., (i) Ad hoc on demand distance vector (AODV) (ii) dynamic source routing (DSR) (iii) optimized link state routing (OLSR) have been evaluated on a designed MANET network scenario for 80, 100 and 120 nodes at different pause time. Throughput, average jitter, and average MAC delay have been taken as the performance metrics with random waypoint mobility model (RWMM) and constant bit rate (CBR) as the traffic application. With increase in pause time, the OLSR RP shows superior performance in terms of both throughput and average jitter. The average MAC delay of OLSR does not get affected much even with the increase in node density, and thus outperforms the other two RP. Keywords AODV · Average end to end delay (AEED) · CBR · DSR · MANET · OLSR · Packet delivery ratio (PDR)

1 Introduction MANET is a dynamic topology ad hoc network with random node placement, the position of which changes rapidly on need basis. These networks do not possess any pre-existing infrastructure like base stations or access points. Thus, each node S. Kumar (B) · Payal Department of Electronics and Communication Engineering, University Institute of Engineering and Technology Maharshi Dayanand University, Haryana Rohtak, India e-mail: [email protected] D. Sharma Department of Physics and Electronics, A.I.J.H.M. College, Rohtak, Haryana, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_11

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behaves like a router that forwards the progressing data packets to its neighbors. The technique by which the nodes are connected is dependent on transmitted power parameter of the nodes and their location that may vary intermittently in the network. The communication between the nodes is maintained via transmission of packets containing data over a common wireless channel which thereby limits the radio coverage [1, 2]. The nodes in MANETs have limited power capabilities due to which the neighboring nodes get limited in terms of resources. The higher operating frequencies suffer from interference and fading in an urban environment, which uphold the links unreliable. Therefore, effective and accurate power aware routing techniques are required for effective network design in MANETs. The RP is the defined set of regulations and standards that provide a method of communication and mechanism of route selection among the nodes while maintaining high QoS standards [3]. The mobile nature of nodes results in random change of the network topology. Pause time in MANET signifies the duration during which the transmitting mobile nodes remain static at a place while communicating among themselves. As the pause time increases, the static behavior in dynamic mobile nodes increases. The present research work is related to enhancement of transmission range and lifetime, which is certainly a challenging task for researchers. Further the present research paper organization is as follows: The RPs are briefed in Sect. 2 followed by Sect. 3 containing related work. Section 4 describes the simulation parameters in MANET design. The results obtained have been analyzed in Sect. 5. The overall outcome has been summarized in Sect. 6.

2 MANET RP Unipath RP and multipath RP are the two categories of well-known RP the performance of which depends on node density, mobility and behavior of mobile nodes, composite interaction of the protocol mechanisms, and their explicit parameter settings with traffic intensity. The former category of RP can be proactive, reactive, and hybrid. In proactive RP, i.e., DSDV, the nodes provide currently updated routing information about the network topology, and routes are created between them before they are required by the network. The reactive RP, i.e., AODV and DSR on the other hand have no predefined routes and the nodes establish their routes on demand dynamically. Once the route is established, the packets of data are communicated to all or its intermediate neighbor nodes. Hybrid RP is a non-uniform routing protocol which utilizes adaptive and minimal overhead control to optimize the network performance. With the mechanism of route discovery, the scalability of the network is also increased by proactive route management. AODV RP provides broadcast, multicast, and unicast communication in MANETs. The path discovery process begins when a demand is displayed by the source and is terminated only when a route is found or all the routes are already investigated [4]. DSR is an on demand, path creating protocol that maintains path revelation and path preservation functions for effective and error free connectivity

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[5]. Whenever there is data to be transmitted, this RP selects an accessible path from its own caches otherwise a path discovery process is initiated with a path request packet containing origin and end address, besides distinctive identification number. OLSR is a proactive, upgraded type of a pure link state protocol. When the delivery time of control messages is reduced, it uses more reactivity to the topological variance. Hello and topology control are the types of control messages operated with this protocol. Hello messages are operated for revealing data concerning the status of the link and host’s neighbors while to convey data among its own advertised neighbors, the later type is used.

3 Related Work The authors in [6] presented a packet level simulation of ad hoc networks under variable load distribution probabilities for different RP. Simulative results show that the performance of the proposed network improves significantly with incorporation of mobility conditions. The authors in [7] presented a modified AODV for evaluating the performance of designed MANET network for effective protection against black hole and gray hole attack by detecting misbehaving nodes in MANETs. The simulative result shows that the proposed algorithm works efficiently and was successful in minimizing misbehaving nodes effectively. The authors in [8] presented the evaluation of AOMDV, TORA, AODV, DSR, and DSDV RP in MANET under different network scenarios based upon PDR, throughput, and energy consumption. Simulative results show that former RP outperforms other RP being an energy efficient protocol in handling network resources. The authors in [9] presented a comparative evaluation of AODV, DSR, and DSDV routing protocols in MANETs. The simulative results show that AODV outperforms the two RP for all performance metrics. The authors in [10] presented modified versions of AODV and DSR using mean absolute deviation statistical approach and carried out a comparative evaluation for effective protection against wormhole attack in MANET. The simulative results show that the modified DSR protocol outperforms modified version of AODV in terms of performance metrics and was successful in minimizing Wormhole attack in MANET. The authors in [11] evaluated the performance of hierarchical and cluster-based RP in FSO-MANET. Simulative results exhibit that the hierarchical-based optical sphere RP provides improved performance in terms of delay, number of packets dropped and throughput in comparison to cluster based RP. The authors in [12] presented the performance evaluation of AODV, OLSR, and DSDV protocol using PDR, routing overhead, AEED, and packet loss under variable network conditions in MANET. Simulative results show that DSDV and OLSR perform better in terms of AEED, whereas AODV outperforms DSDV in terms of routing overhead. The authors in [13] presented a comprehensive theoretical review of AWSN. The authors in [14] AODV and DSR routing protocol have been evaluated for their performance in AWSN networks using Zigbee traffic application. Simulative results show that AODV outperforms DSR in QoS parameters while using Zigbee as traffic application.

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The authors in [15] evaluated the performance of (i) Generic, (ii) Micaz, and (iii) Micamotes energy conservation models over a designed AWSN network scenario for the two RP, i.e., AODV and DYMO using AEED, Throughput and energy consumed as the performance metrics. With AODV protocol, the Micamotes outperforms the generic and Micaz energy model due to lesser energy consumption both in transmit and receive mode. All the earlier work reported in literature have evaluated the performance of designed network scenarios in isolation using several routing protocols for improved performance at a specified pause time. A few literary works have been reported recently having used variable pause time for AODV, DSR, and DYMO protocol. In this work, we aimed at evaluating the performance of a designed MANET scenario with variable node density in conjunction with random waypoint mobility model at two different pause times of 15 and 30 s using AODV, DSR, and OLSR protocols for optimum performance.

4 Simulation Setup MANET scenario has been created with nodes (80, 100 and 120) randomly placed over 1500 * 1500 m terrain size using QualNet Simulator 7.3.1. The designed MANET performance with three different RP has been evaluated for the performance metrics—Throughput, average jitter, and average MAC delay for an increase in node density. The mobility model used in our network scenario is RWMM and CBR as the traffic application. The data packet size is of 512 bytes with 10 m/s speed per node and data rate of 2 Mbps. Table 1 provides a list of simulation parameters and their values used in the designed network scenario. Table 1 Parameters and values used in simulation Parameter

Value

No. of nodes

80, 100, 120

Terrain size

1500 * 1500 m

MAC protocol

IEEE 802.11

RP

AODV, DSR, OLSR

Model

RWMM

Pause time

15, 30 s

Maximum speed of node

10 m/s

Energy model

MICA motes

Data traffic

CBR

Traffic application

(10–27), (14–11), (46–30), (39–45), (73–12), (13–75), (58–43), (28–52), (16–60), (37–25)

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The performance metrics chosen for evaluating performance are throughput, average jitter, and MAC delay. Throughput: It is defined as the average rate of data successfully delivered using suitable communication channel and expressed in bits/s or packets/s. It is desired that the throughput remains higher. Average Jitter: It is defined as the alteration in delay of various data packets that reaches the terminating nodes and a measure of volatility in latency in delivery of packets from source to destination. It is measured in seconds. For optimum performance of a network, the jitter has to be very less. Higher jitter might prompt the buffers to underflow or overflow and may lead to algorithm collapse. MAC Delay: It is defined as the medium access delay for a packet including the time spent in collisions as well as the time spent in back off process. The dropped packet poses problem in delay calculation. MAC delay is calculated as the time elapsed between the moment a packet is put to service and its successful transmission or drop. Mica Mote Energy Model: The Mica mote energy model uses second generation energy efficient WSN nodes utilizes TinyOS operating system at each node and incorporates an Atmel family based 8 bit microcontroller operating at a frequency of 4 MHz. The nodes also incorporate a 10 bit ADC for digitizing the output. The radio of MICA mote can transmit upto 40,000 bits per second over a distance of hundred meters.

5 Result and Discussion

Throughput in bits/sec

In this present research, we have varied the pause time of nodes to create two different scenarios with 15 and 30 s pause time for evaluating network performance. The graph depicting throughput with nodes varying from 80 to 120 at a pause time of 15 s is shown in Fig. 1. The value of throughput for AODV, DSR, and OLSR protocol increases as the number of nodes vary from 80 to 120 nodes. From Fig. 1, it is also evident that the average throughput increases with node density as AODV, DSR, and OLSR Variation of Throughput with Nodes AODV DSR OLSR

15000 10000 5000 0 80 Nodes

100 Nodes

120 Nodes

Nodes Fig. 1 Variation of throughput with nodes at a pause time of 15 s

S. Kumar et al.

Throughput in bits/sec

110

Variation of Throughput with Nodes 14000 12000 10000 8000 6000 4000 2000 0

AODV DSR OLSR

80 Nodes

100 Nodes

120 Nodes

Nodes Fig. 2 Variation of throughput with nodes at a pause time of 30 s

protocols show an increase of 44.48%, 37.59%, and 45.51%, respectively, when the node density increases from 80 to 120 nodes. The bar plots in Fig. 2 show the variation in throughput at pause time of 30 s. At 30 s pause time, AODV, DSR, and OLSR protocols show an increase of 44.44%, 37.59%, and 48.66%, respectively, when the node density increases from 80 to 120 nodes. From Figs. 1 and 2, the value of throughput is higher at pause time of 30 s as compared to 15 s. With increase in pause time, the static behavior of nodes increases, and they remain in their place for longer duration, thereby allowing an error free seamless transmission. OLSR provides higher throughput for both the pause time scenarios and outperforms AODV and DSR. Further OLSR shows approximately 4% increase in throughput with the increase in pause time form 15 to 30 s. Figures 3 and 4 depict the variation of average jitter with increasing node density for pause time scenarios of 15 and 30 s. The bar chart in Fig. 3 shows the variation of average jitter for increasing node configuration for a pause time of 15 s. From Fig. 3, the values of average jitter for AODV, DSR, and OLSR vary from 0.16, 0.17, and 0.059 for 80 nodes to 0.22, 0.24, and 0.08 s for 120 nodes, respectively. At 15 s pause time, AODV, DSR, and OLSR protocols show an increase of 43.75, 41.17%, and 35.59%, respectively, in Variation of Average Jitter with Nodes Average JiƩer

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average jitter when the node density increases from 80 to 120 nodes. It is evident from the results that with node density, the average jitter increases and OSLR provide minimum jitter in comparison to both the RP, i.e., AODV and DSR. Figure 4 shows the variation of average jitter for a pause time of 30 s. At 30 s pause time, AODV, DSR, and OLSR protocols show an increase of 23.80, 29.16%, and 17.24%, respectively, in average jitter when the node density increases from 80 to 120 nodes. However, OSLR provides minimum jitter in comparison to AODV and DSR. With increase in pause time from 15 to 30 s, the average jitter increases thereby limiting the performance. However, in both scenarios of different pause time, the OLSR RP shows optimum performance. The variation of MAC delay with node density for pause time of 15 and 30 s is shown in Figs. 5 and 6, respectively. Variation of MAC delay with Nodes

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At a pause time of 15 s, the average MAC delay varies from 0.009, 0.0042, and 0.0039 to 0.007, 0.004, and 0.0037 for increase in node density from 80 to 120 nodes for AODV, DSR, and OLSR protocol, respectively. AODV, DSR, and OLSR protocols show variation of 22.22%, 14.7%, and 5.4%, respectively, in average MAC delay when the node density increases from 80 to 120 nodes. It is evident from the results that with node density, the MAC delay decreases and OSLR provides minimum MAC delay in comparison to AODV and DSR protocol. At pause time of 30 s, the average MAC delay varies from 0.014, 0.004, and 0.004 to 0.0012, 0.0039, and 0.0038 for increase in node density from 80 to 120 nodes for AODV, DSR, and OLSR, respectively. From Figs. 5 and 6, it is evident that the average MAC delay decreases with increase in number of nodes. However, the MAC delay increases with increase in pause time from 15 to 30 s as the increase in pause time tend to increase static node behavior. From the results, it is further evident that the designed MANET scenario works in consonance with the theoretical results. The average throughput and average Jitter increases and average MAC delay decreases for each of AODV, DSR, and OLSR routing protocol with increase in number of nodes from 80 to 120 nodes. Further with increase in pause time from 15 to 30 s, the static behavior of nodes increases, and they remain in their place for longer duration there by allowing an error free seamless transmission and increase in average throughput for all three routing protocols. However, OLSR provides higher throughput in both the pause time scenarios and outperforms AODV and DSR. Again in both pause time of 15 and 30 s, the OLSR RP shows optimum performance and provides minimum average jitter in comparison to DSR and AODV protocol. Also from the results, it can be further concluded that with increase in node density, OLSR provides minimum average MAC delay. The average MAC delay of OLSR has a very small variation in comparison to DSR and AODV protocol. From all the above results, it can be conclude that the OLSR RP performs efficiently in comparison to AODV and DSR protocol.

6 Conclusion The present research involves the performance evaluation of designed MANET scenario for AODV, DSR, and OLSR RP with variation in node density and pause time. The simulation results reveal that the OLSR RP outperforms AODV and DSR in throughput and average jitter evaluation. When it comes to average MAC delay, the OLSR does not get effected much and performs better even with the increase in node density. The OLSR RP works optimally with a 13,150 throughput, 0.17 s average Jitter, and 0.0036 s MAC delay for 120 nodes at a pause time of 30 s. This signifies that pause delay can be kept small or large based on the desired network application and amount of data to be transferred for analysis. This simulation work will facilitate the hardware designers in selecting the components for various sub-modules of the nodes to be used in deploying field network.

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References 1. Toh CK (2002) Ad hoc mobile wireless networks. Prentice Hall Publishers 2. Sharma D, Kumar S (2015) A comprehensive review of routing protocols in heterogeneous wireless networks. Int J Enhanced Res Manage Comput Appl 4(8):105–121 3. Odeh A, Fattah EA, Alshowkan M (2012) Performance evaluation of AODV and DSR routing protocols in MANET networks. Int J Distrib Parallel Syst (IJDPS) 3(4):13–22 4. Khurana S, Kumar S, Sharma D (2017) Performance evaluation of congestion control in MANETs using AODV, DSR and ZRP protocols. Int J Adv Res Comput Sci Softw Eng 7(6):398–403. https://doi.org/10.23956/ijarcsse/V7I6/0218 5. Gupta AK, Sadawarti H, Verma AK (2013) Performance analysis of MANET routing protocols in different mobility models. Int J Inf Technol Comput Sci 5(6):73–82 6. Zirimite R, Bhate M (2018) MANET performance analysis using different distribution load. J Anal Comput (JAC) 11(1):1–9 7. Joshi D, Velani N (2018) Study of modified routing protocols in MANET. Int J Sci Res Comput Sci Eng Inf Technol 3(1):1621–1624 8. Kute BV, Kharat MU (2013) Analysis of quality of service for the AOMDV routing protocol. ETASR Eng Technol Appl Sci Res 3(1):359–362 9. Cana E (2013) Comparative performance simulation of DSDV, AODV and DSR MANET protocols in NS2. Int J Bus Technol 2(1):24–31. https://doi.org/10.33107/ijbte.2013.2.1.04 10. Majumder S, Bhattacharyya D (2019) Comparative study between modified DSR and AODV routing algorithms to improve the PDF due to wormhole attack in MANET. Int J Sci Res Rev 8(1):1095–1102 11. Kavitha B, Jawahar A (2018) Performance evaluation of hierarchical routing protocol with multiple transceivers and cluster based routing protocol in FSO MANET. TAGA J Graphic Technol 14:1169–1178 12. Kumari N, Gupta SK, Choudhary R, Agrwal SL (2016) New performance analysis of AODV, DSDV and OLSR routing protocol for MANET. In: 3rd international conference on computing for sustainable global development (INDIA Com), pp 33–35 13. Mor K, Kumar S, Sharma D (2018) Ad-Hoc wireless sensor network based on IEEE 802.15.4: theoretical review. Int J Comput Sci Eng 6(3):220–225. https://doi.org/10.26438/ijcse/v6i3. 220225 14. Mor K, Kumar S (2018) Evaluation of QoS metrics in ad-hoc wireless sensor networks using Zigbee. Int J Comput Sci Eng 6(3):92–96. https://doi.org/10.26438/ijcse/v6i3.9296 15. Kumar S, Dhull K, Sharma D, Payal DS (2019) Evaluation of AODV and DYMO routing protocol using generic, Micaz and Micamotes energy conservation models in AWSN with static and mobile scenario. Scalab Comput Pract Exp 20(4):653–661. https://doi.org/10.12694/scpe. v20i4.1584

Three-Qubit Implementation of Quantum Fourier Transform for Shor’s Algorithm Deepanshu Trivedi, Ankur Saharia, Kamalkishor Choure, Manish Tiwari, Ravi Kumar Maddila, and Ghanshyam Singh

Abstract Quantum computers are capable of very fast computation as compared to the classical counterpart. Problems impossible for the classical computer are efficiently solved on a quantum computer. Shor’s factoring algorithm (SFA) calculates the prime factors of a given number exponentially quicker than the available classical algorithm. The paper deals with a vivid explanation of the methodology and various other future possibilities related to the development of the SFA. The paper also emphasizes the three-qubit realization of the quantum Fourier transform on the IBM Q experience. Keywords Shor’s factoring algorithm · Quantum computing · Quantum Fourier transform

1 Introduction In the early 1980s, quantum prototype for the turing machine was proposed by physicist Paul Benioff that lead to the beginning of quantum computing [1]. Later, Richard Feymann and Yuri Manin suggested that quantum computers can outperform classical computers [2]. Then, Shor came up with a quantum algorithm for factoring integer with the capacity to decrypt all possible secured systems [3]. The classical computer works on the classical bits- 0 and 1, while the quantum computer makes use of qubits—|0 and |1. Various objects used as a qubit (electrons, protons, and nucleus). Researchers are using outermost electrons in phosphorus as qubits. Quantum computing makes use of two basic phenomena of quantum mechanics— quantum superposition and quantum entanglement [4].

D. Trivedi (B) · A. Saharia · K. Choure · R. K. Maddila · G. Singh Department of Electronics and Communication Engineering, MNIT Jaipur, Jaipur, India e-mail: [email protected] M. Tiwari Department of Electronics and Communication Engineering, Manipal University Jaipur, Jaipur, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_12

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Quantum superposition refers to the uncertainty of the particle to be in several states at once. For example, an electron can be either in ground state or in the excited state. By the principle of superposition, the electron is in the state which is a linear combination of both the states b0 |0 + b1 |1 where b0 and b1 are the coefficients can be complex numbers which are adding to 1. Quantum entanglement is interpreted as the exchange of quantum information between two particles at a distance. It means that when the particles are separated, the quantum states of each particle is dependent of the state of other particle and cannot be defined independently.  of the well-known application of the quantum computers. It takes SFA is one O (log N )3 time to factor a number faster than its classical equivalent. The error introduced due to the use of physical qubits and large number of gates, and the algorithm is still far away from the real-time implementation. The most difficult case of factorization is when a number is the product of two odd primes which are equal in length. This is the outline of RSA cryptosystem, which uses a public key N, the product two large odd primes. RSA cryptography is based on the fact that it is difficult to factor a very large number. In order to crack the RSA cryptosystem, Shor proposed a quantum factoring algorithm which is polynomial in time.

2 Basic Concepts of Shor’s Algorithm Kitaev replaced a fully coherent QFT by the semi-classical quantum Fourier transform (sc-QFT) in the Shor’s algorithm. In sc-QFT, each time one of the qubits of the period register is measured [5]. The measurement on the second qubit is determined by the result of measurement on the first qubit. So the 2 log2 N qubits required for the period register can be replaced by a single qubit. Hence, the number of qubits is now reduced to execute the Shor’s algorithm. For example, for N = 15, 21, 35, the number of qubits required is n = 5, 6, 7. The scalable algorithm has been realized with an ion-trap quantum computer that provides success probabilities above 90%. In 2017, WANG Yahui et al. proposed a quantum algorithm capable of breaking the public key cryptosystem like RSA [6]. It has some essential outlines like—(1) without factoring a number the plaintext can be recovered from the ciphertext, (2) even order of the elements to be avoided, (3) with better probability of success than Shor’s algorithm, (4) equal complexity compared to Shor’s algorithm. The algorithm proposed by Peter Shor works iff (if and only if) the period is even. If the period is odd, the factors cannot be found, and the algorithm is relaunched using distinct a values for the function a x mod N = 1. For the square coprimes, the factors can be found using odd orders [7]. This somehow increases the possibility of success by considering odd orders. The rate of success of the algorithm can be improved by avoiding square coprimes rather than to consider the odd orders. Earlier author considered factoring 21 with the coprime four giving order three. In spite of odd order,

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 3  the factors are successfully calculated by the algorithm, 3 = gcd 4 2 + 1, 21 , and  3  7 = gcd 4 2 − 1, 21 . As coprime is square, the factors o are integers, but coprimes do not always serve the purpose. For example, factoring 21 with coprime 16. The paper analyzes the role of odd orders in factoring a number, and it should not be ignored directly. The recent research in SFA, Amico et al. [8] discussed the implementation of the compiled form of SFA for the specific case of N = 15, 21, and 35 on the ibmqx5 superconducting chip. Using the sc-QFT, the algorithm is implemented using small number of physical bits as compared to a large qubits required using the coherent quantum Fourier transform. The similarity between the theoretically obtained values (distribution of phase) and experimentally calculated values gives the quantitative measure for which square of statistical overlap is used. Nene et al. [9] presented the simulation of the algorithm on MATLAB using quantum computing function tool box. Development and commercialization of quantum computer are still far away; so, the paper produced a standardized method for the implementation of SFA on a classic computer. The analysis of the periodicity of the function upto 3-digit of N is presented where the result of simulations are collaborated with the theoretical results. Vivid description about the methodology and mathematical analysis of the different parts of Shor’s algorithm is clearly explained by Loceff [10]. It also deals with the basic concepts of quantum mechanics and explanation of other quantum algorithm like Simon’s algorithm, Deutsch’s algorithm, and quantum teleportation that leads to clear understanding of the SFA.

3 Methodology The algorithm composed of two parts: • Classically processing the problem by changing the factoring problem to period finding problem. • A quantum algorithm to find the period of the function responsible of quantum speedup. Steps involved in the process of SFA are [11]: Step 1 Step 2 Step 3

Step 4 Step 5

Choose a random integer ‘a’ such that (a < N) Compute the gcd(a, N ), the greatest common divisor of N. This can be done using Euclidean algorithm. If gcd(a, N ) = 1, signifies there is non-trivial factors of N. Then the function f (x) = a x mod (N ) is used to find the unknown number ‘r’ which gives the period. If the period ‘r’ found to be odd, go back to Step 1; If the period ‘r’ found to be even, then go to Step 6

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Fig.1 General circuit of Shor’s factoring algorithm. Reproduced from [10]

Step 6

 r   r  gcd a2 + 1, N and gcd a2 − 1, N are the non-trivial factors of N. The process is now over.

In order to find the factor of a number, we need to find the power x of integer a for which the function a x mod N = 1 where a is some random number which is less than N and exponent x is the order or period of a (Fig. 1). Two quantum registers are required for the implementation of the algorithm. First register, known as the period register or A-register for storing period values. Second register, known as the computational register or B-register, is used to store the result of the modular exponentiation function (MEF) given by a x mod N [10]. Depending on the number N, the size of the  registers varies.  The  qubits in a period register should be in the range log2 N 2 ≤ n p ≤ log2 2N 2 while n q = log2 N qubits in the computational register. Two separable states are prepared |00..0 p |00..1q where the notation p and q signifies the period register and computational register [8]. All possible x values are stored in the period register which gives the approximate value of the period. When |00..0 p passes through the multi-dimensional Hadamard gate it results in  Q−1 np √1 x=0 |x p which is the equal superposition of all qubits, where Q = 2 . Q As soon as the n-qubit passes through the Hadamard gate, the concept of quantum parallelism comes into effect which suggests that if we apply the unitary transformation to all the possible 2n p inputs, it will produce the superposition of the results of applying f to them in parallel [10]. After the first step, the qubits are now in the tensor product which when passed through the uniform transformation function (U f ) result in  Q−1 x √1 x=0 |x p |a mod N q . Now, to find the period, QFT is used. As a result of Q QFT, interference between different possible states occurs, and it produces different superposition states as the output [12]. This interference either makes the signal stronger or weaker depending upon the type of interference depending upon their phase and amplitude. Now both the registers are measured, but the order of measurement is the trick. How? Let us see…. If we measure the A-register, then the B-register would collapse into its normalized partner, | f (x)n .

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If we measure the B-register first, then the A-register would also disintegrate due to the property of entanglement, but in this case that would be Nr = m, not one, pre-images x for every f (x) values. Now we choose to measure the B-register first, and later on, the A-register is measured. The measurement of A-register after the B-register would result in one of the m values, x0 + jr , but we have no way to extract r (period) from the measurement, so we do not measure A-register yet. Now, apply QFT to the period register or the A-register and measure the qubits later [10]. The measurement obtained in the quantum part is classically processed contributing to the final part of the algorithm. The period value r can be found using the continued fraction algorithm [13] or get direct estimation of the period value by running the algorithm several times [8]. The algorithm proposed by Shor’s in 1994 known as the factorization algorithm can be implemented using n q = log2 N qubits in the computational registers that are used for the MEF and n p = 2 log2 N qubits in the period registers for QFT. Thus, the entire algorithm would require a total of 3 log2 N qubits which is still a challenge for present quantum computer is N is large [5].

4 Three-Qubit Implementation of QFT In order to find the period of the function f , the function values are calculated at every interval or points (x1 , x2, . . . xn ) simultaneously. When measured, it will give one of the possible values and neglect all others by the property of entanglement. QFT differs as it operates on superposition state and produces different superposition state as the output [12]. The component interferes constructively or destructively depending on their amplitude and phase. The three-qubit implementation of QFT is implemented on IBM Q experience as shown in Fig. 3. If the QFT operates on any basis state alone, the output is the superposition of all the possible states. Change of phase of various states can be seen using QFT. QFT makes use of two gates-Hadamard gate (single qubit) and the controlled rotation gate (two-qubit) as shown in Fig. 2. QFT is an essential part of Shor’s algorithm. On the other hand, the MEF can also be implemented using the IBM Q experience platform for the complete implementation of the Shor’s factoring algorithm.

5 Conclusion Due to various properties of quantum mechanics like quantum superposition, entanglement, and parallelism, the various public-key cryptographic systems (RSA, ECC, etc.) are no longer secure and can be breached shortly. The new cryptographic system needs to be introduced for cyberspace security shortly. Since Shor proposed a

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Fig. 2 Three-qubit QFT implementation using Hadamard gate and phase controlled gate. Reproduced from [10]

Fig. 3 Superposition of all states of three-qubit input as the result of QFT; x-axis-state and y-axisprobability

factoring quantum algorithm in 1994, various techniques of implementing different versions of the algorithm have been suggested. Also, various other facts like the consideration of odd periods are taken into use and cannot be denied straight away. New quantum algorithm breaking cryptography without factoring is also studied. Though, reducing number of qubits is still a challenge in the field of quantum. Acknowledgements We acknowledge the open-source platform of IBM.

References 1. Benioff P (1980) The computer as a physical system: a microscopic quantum mechanical hamiltonian model of computers as represented by Turing machines. J Stat Phys 22(5):563–591 2. Feynman RP (1982) Simulating physics with computers. Int J Theor Phys 21(6/7):467–488 3. Shor PW (1997) Polynomial-time algorithms for prime factorization and discrete logarithms on a quantum computer. SIAM J Comput 26(5):1484–1509 4. Sharma AK, Ghunawat A (2019) A review on linear optics quantum computing. IEEE Conf Proc 5. Monz T, Nigg D, Martinez EA, Brandl MF, Schindler P, Rines R, Wang SX, Chuang IL, Blatt R (2016) Realization of scalable Shor’s algorithm. Science 351(6277):1068–1070 6. Yahui W, Songyuan Y, Huanguo Z (2017) A new computing algorithm for computing RSA ciphertext period. Wuhan Univ J Nat Sci 22(1):068–072 7. Lawson T (2015) Odd orders in Shor’s factoring algorithm. Quantum Inf Process 14(3):831–838

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8. Amico M, Saleem ZH, Kumph M (2019) Experimental study of Shor’s algorithm using IBM Q experience. Phys Rev A 100(1) 9. Nene MJ, Upadhyay G (2016) Shor’s algorithm for quantum factoring. Adv Comput Commun Technol:325–331 10. Loceff M (2015) A course in quantum computing 1:610–650 11. Zhang W, Xu C, Li F, Feng J (2007) A period-finding method for Shor’s algorithm. Int Conf Comput Intell Secur:778–780 12. Ekert A, Jozsa R (1996) Quantum computing and Shor’s factoring algorithm. Rev Mod Phys 68(3):733–753 13. Vazirani U (2004) Shor’s factoring algorithm. CS 294-2, Fall 2004. http://www.cs.berkeley. edu/~vazirani/f04quantum/notes/lec9.pdf

Performance Enhancement of Surface Plasmon Resonance (SPR) Structure Using a Sinusoidal Diffraction Grating Manish Jangid, Ankur Saharia, Nitesh Mudgal, Sajai Vir Singh, and Ghanshyam Singh

Abstract In this paper, a sinusoidal diffraction grating is being used for designing a Surface Plasmon Resonance (SPR) structure. The operation of the structure has been examined by using the wavelength-interrogation technique. We have considered the reflected amplitude and absorption dip of the SPR response curve as design parameters for designing this structure. On the performance comparison of gold (Au)based over silver (Ag)-based SPR structure, although the Ag-based SPR structure gives better results. Due to the poor chemical stability of silver, a thin film of gold is used over it which enhances the performance of the proposed bimetallic SPR structure. This can be used in bioscience for observing the variation of refractive index in an analyte. Keywords Surface Plasmon Resonance · Sinusoidal grating · FDTD simulation · Refractive index · Oxidation · And reflectivity

1 Introduction Nowadays, SPR-based biosensor attracts more attention. In 1902, when a polarized light was illuminated by wood onto a diffraction grating, Surface Plasmon Waves (SPW) were observed for the first time [1]. A coupling method for Surface Plasmon excitation based on attenuated total reflection (ATR) was proposed [2]. Kretschmann [3] reported the well-known method named as Kretschmann configuration of ATR coupling for SPR sensors. Later on, the grating-based SPR sensors came into the picture due to their easy designing, miniaturization, and integration features. In 1983, the first application of SPR-based biosensor in gas sensing received more attention from researchers [4]. Sensing a small variation in the refractive index(RI) provides M. Jangid (B) · A. Saharia · Nitesh Mudgal · G. Singh Department of Electronics and Communication, Malaviya National Institute of Technology Jaipur, Jaipur 302017, India e-mail: [email protected] S. V. Singh Department of Electronics and Communication Engineering, Jaypee Institute of Information Technology Noida, Noida 201309, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_13

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a better way to characterize gases, chemical molecules, and living cells, showing advantages of real-time and label-free detection [5–8]. Because the SPW vector is greater than the free space vector, so to fulfill the requirement of high index profile for excitation of SPR, we need the high index optical structure like a prism [9, 10], patterned nanostructures like diffraction grating [11, 12] or optical fibers [13–18]. In the past two decades, study interests in SPRs have been emerging very fastly, and a large number of practical applications particularly on biosensing are in progress to develop, for example, affinity, kinetics of bio-molecular interaction, specificity, concentration of an analyte, etc. [19] (Fig. 1). The phenomenon of an SPR is caused when a p-polarized light vector is guided into a smooth, rough, or grating thin metal surface like silver, gold, and aluminum, etc. (Fig. 2). The free electrons near such surface collectively oscillate, and an SPW propagates at metal–dielectric interface. At the resonant condition, most of the part of light is coupled, and only small amounts of light loss is observed; a sharply reflected signal is being observed in intensity profile. The required e-field components corresponding to SPW should be p-polarized light because the metal layer and oscillations of e-field vector both are the same in-plane. These resonance characteristics of SPR spectra (sensogram) provide information about surface (metal-dielectric) interactions in a real-time monitoring system [22]. The SPR sensors can be categorized based on wavelength-interrogation, angle-interrogation, intensity-interrogation, phase-interrogation, and polarizationinterrogation techniques [23].

Fig. 1 Concept of SPR sensors. Reproduced from [20]

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Fig. 2 a Condition for SPR occurrence, b TM electromagnetic wave guided between metal and dielectric interface. c Evanescent field and SPP distribution. Reproduced from [21]

For the evaluation of SPR sensor performance, metal’s dielectric constant plays a vital role. Here, the performance of gold (Au) and silver (Ag)-based SPR structure was compared and found that silver-based SPR structure has better performance over the gold-based structure. In the end, to eliminate oxidation problem that arose in silver-based structure, a thin gold film is layered on its surface.

2 Mathematical Modeling The condition of SPR condition occurs only at the boundary of two different mediums having opposite dielectric constants in nature [24]. A simple SPR contains a metal   + imetal and a dielectric layer with layer and has a permittivity of metal = metal   a permittivity diel = diel + idiel [8]. SPW has a property of an evanescent wave containing maximum magnetic field intensity at the metal–dielectric interface and decaying into both the medium [6]. SPW is a transverse-magnetic (TM) wave in nature, guided metal and dielectric interface [25]. The Surface Plasmon’s   between Wave vector K Sp can be expressed for the metal–dielectric interface as K Sp

2π = λ



diel metal diel + metal

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where dielectric constant for metal is metal and for the dielectric layer is given by diel . The incident light vector is generally less than SPW vector, which is defined as n sin θx (Refractive Index of an analyte is n x and the angle in the analyte kα = 2π λ x

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is θx ). In the coupling mechanism of grating, the diffracted light beam vector (kβ ) is the sum of kα and the grating wave vector [7, 26]. kβ = kα + γ

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where diffraction order γ is an integer number and P is grating constant. So, now SPR excitation condition can be expressed as kβ = ksp , which is written as λ n x sin θx n x + γ ± = P



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Here, diel is replaced by n 2x , the sign ‘ ± ’ shows the propagation of Surface Plasmon along the positive or negative direction. Silver and gold are preferably used metals for the designing of SPR sensors since they exhibit a narrow resonance curve [27]. Based on these two metals, SPR structure’s performance is analyzed. Lorentz–Drude equation [28] is used to calculate the value of metal . metal (ω) = 1 −

ϕ 2p ω(ω − i0)

(4)

√ where ω = 2πc and ϕ P = f 0 ω P is the plasma frequency, f 0 is the oscillator λ strength, and 0 is the damping constant. The values of f 0 , ω P , and 0 are calculated from [29, 30].

3 Designing and Simulation A schematic for designing the proposed SPR structure using sinusoidal grating is shown in Fig. 3. Grating thickness (g), constant/period (p), and substrate thickness (t) are being considered as constructor parameters for designing the proposed SPR structure. The wavelength-interrogation-based performance of SPR sensor can be calculated by resonance wavelength shift and absorption dip of reflected signal in the SPR response spectra. The amplitude of reflected signal and resonance wavelength shifts and must be significantly large for observing a distinctive and high precision signal. Figure 4a states the reflection curve of silver (Ag)-based SPR structure for grating constants p1 = 800 nm and p2 = 1000 nm. The other parametric values of the structure are grating thickness (g) = 50 nm, substrate thickness (t) = 40 nm. Here, the wavelength of 633 nm is being used as a guiding light signal. In Fig. 4a, we observed that the grating period/constant (p) increases from 800 to 1000 nm, and the resonance

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Fig. 3 Schematic diagram of a sinusoidal diffraction grating-based SPR sensor g1 = 50 nm

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(b)

Fig. 4 a Reflection curve of silver (Ag) based SPR structure for grating constants p1 = 800 nm and p2 = 1000 nm. b Reflection curve of silver (Ag) based SPR structure for grating thicknesses g1 = 50 nm and g2 = 100 nm

wavelength’s amplitude decreases. So, we selected the grating constant p = 800 nm as an optimal value. It was observed that as the grating constant was increased, the reflectance dips of the SPR response curve also rose. For better performance of SPR structure, we need a larger reflectance dip of the SPR response curve. Similarly, Fig. 4b displays the reflection curve of silver (Ag)-based SPR structure for grating thicknesses g1 = 50 nm and g2 = 100 nm. It is clear that at 50 nm grating thickness SPR, we got the smallest SPR dip along with the smallest reflected amplitude at g = 30 nm. But when g = 100 nm, the SPR dip observed was wider. Here, it is clear that for optimal performance, the value of grating thickness should lie between 30 and 100 nm. So, when the grating thickness (g) is 50 nm, the reflected amplitude is the highest, and the SPR dip is also very small

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Fig. 5 Reflection curve of silver (Ag) based SPR structure for substrate thicknesses t1 = 40 nm and t2 = 80 nm

t2 = 80 nm

1 0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0

600

620

640

660

680

Wavelength (nm)

1 0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0

600

p1 = 800 nm

620

p2 = 1000 nm

640

660

Wavelength (nm)

(a)

680

TM Refelctive amplitude

TM Refelctive amplitude

This is the condition, where the maximum light of incident wave is being coupled. Therefore, based on the above consideration, we selected the grating thickness (g) = 50 nm as optimal value. Figure 5 illustrates the reflection curve of silver (Ag)-based SPR structure for grating substrate thickness t1 = 40 nm and t2 = 80 nm. The light is launched onto the grating surface vertically. By considering the substrate thickness (t) below 40 nm, the reflected amplitude observed was small as compared to the 40 nm thickness. Similarly, on increasing the substrate thickness beyond the 40 nm, there was no variation observed in reflected amplitudes of the SPR dips. Hence, the substrate thickness (t) = 40 nm provides better performance for proposed structure Figure 6a states the reflection curve of gold (Au)-based SPR structure for grating constants p1 = 800 nm and p2 = 1000 nm. The parametric values of the structure are the same as a silver-based sensor, grating thickness (g) = 50 nm, and substrate thickness (t) = 40 nm. As the grating period/constant was increased from 800 to 1000 nm, the resonance wavelength decreased. It was observed that as the grating constant was increased, the reflectance dips of the SPR response curve also rose. Since we needed a larger reflectance amplitude, so we selected grating constant (p) = 800 nm as optimal value. Figure 6b displays the simulated reflection spectra of gold (Au)-based SPR structure for grating thicknesses g1 = 50 nm and g2 = 100 nm.

1 0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0

600

g1 = 50 nm

620

g2 = 100 nm

640

660

680

Wavelength (nm)

(b)

Fig. 6 a Reflection curve of gold (Au) based SPR structure for grating constants p1 = 800 nm and p2 = 1000 nm. b Reflection curve of gold (Au)-based SPR structure for grating thicknesses g1 = 50 nm and g2 = 100 nm

Performance Enhancement of Surface Plasmon Resonance …

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t1 = 40 nm TM Refelctive amplitude

Fig. 7 Reflection curve of gold (Au)-based SPR structure for substrate thicknesses t1 = 40 nm and t2 = 80 nm

t2 = 80 nm

1 0.8 0.6 0.4 0.2 0

600

620

640

660

680

Wavelength (nm)

TM Refelctive amplitude

1 0.8 = 1.31

0.6

= 1.315 = 1.32

0.4

= 1.325

0.2

= 1.33

0

600

610

620

630

640

650

660

670

680

TM Refelctive amplitude

It was observed that below 50 nm and on and beyond 100 nm grating thickness, the SPR response curve did not meet the requirement. At grating thickness (g) = 50 nm, the maximum incident light coupled with SPW Figure 7 illustrates the reflection curve of gold (Au)-based SPR structure for grating substrate thickness t1 = 40 nm and t2 = 80 nm. The best performance of Au-based structure was observed at substrate thickness (t) = 40 nm. Here, we observed that the gold-based structure has a wider resonance dip than a silver-based structure But we require a sharp resonance dip as per the requirement of a high-performance SPR sensor. So, we suggested a bimetallic gold (Au)–silver (Ag)-based structure and simulated under the same parametric values. The wavelength SPR curves of the Ag-based sensor and Au-based sensor are illustrated corresponding in Fig. 8a, b. It clearly witnessed that as the RI of the analyte n a increases, the reflectance dips move gradually to the leftward. It can be evidenced that the reflectance dip value of gold-based sensor analogous to the resonance wavelength is higher than that of silver-based sensor. Since at resonant wavelength, silver offers the lower reflectance dip compared with gold, so we consider that silver-based sensor exhibits superior resonance performance. We got the sensitivities of 890.26 nm/RIU for silver-based and 646.74 nm/RIU for gold-based sensor. It clearly states that the silver-based sensor has larger sensitivity than gold-based sensor. But, there is problem 1 0.8 0.6 0.4 0.2

= 1.31 = 1.315 = 1.32 = 1.325 = 1.33

0

600 610 620 630 640 650 660 670 680

Wavelength (nm)

Wavelength (nm)

(a)

(b)

Fig. 8 a Wavelength SPR curves for silver-based SPR structure. The RI of an analyte ranging from 1.31 to 1.33. b Wavelength SPR curves for gold-based SPR structure. The RI of an analyte ranging from 1.31 to 1.33

M. Jangid et al.

TM Refelctive amplitude

130 1 0.9 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 0

600

610

= 1.31

= 1.315

= 1.325

= 1.33

620

630

640

650

= 1.32

660

670

680

Wavelength (nm)

Fig. 9 Wavelength SPR curves for bimetallic SPR structure. The RI of an analyte ranging from 1.31 to 1.33

associate with silver metal, i.e., oxidation. This kind of problem decreases the durability and reliability of such SPR structure. That’s why, a bimetallic gold (Au)–silver (Ag)-based structure is suggested and simulated under the same parametric values. In designing perspective, a bimetallic structure is construct with an Ag-layer of thickness 47 nm and an Au- layer (as a protective layer) of thickness 3 nm over it, which practically enhances the durability of this SPR sensor. The SPR curve for bimetallic-based sensor illustrates in Fig. 9. The bimetallic SPR curve is almost similar to the Ag-based SPR structure as depicted in Fig. 8a. The sensitivity of bimetallic SPR sensor is 842.28 nm/RIU. This value is small to some extent that of Ag-based sensor. Results show that there is no significant difference in the sensitivity of the silver-based SPR structure compared to the bimetal SPR structure. Both sensitivities values are quite similar. Here the proposed bimetallic SPR structure is better in reliability point of view as compared to silver SPR structure. Here we observed minimum reflectivity of proposed SPR structure that indicates that maximum light is being coupled and also enhances the efficiency of SPR structure.

4 Conclusion A practical simulation of a SPR-based sensor is done in this research work to witness the impact of silver and gold thin metal’s film on the performance parameters of the proposed structure. The investigation was performed using a reflectivity graph which provide the information about the shift in Plasmon dip of coupled optical signal at a particular angle of incidence. Subsequently, a numerical investigation was performed that reflected the influence of structural parameters on the performance analysis of proposed SPR biosensor. The effect of sinusoidal grating on the output of the proposed SPR was assessed by varying the grating thickness, grating constant, and thickness of the substrates. For optimized values of grating’s design parameters,

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the shift of the Plasmon wavelength was determined. The main feature of this SPR biosensor is the use of sinusoidal bimetallic (gold–silver) grating to improve the quality of the SPR signals. The RI dynamic range of this sensor ranges from 1.31 to 1.33 and has the highest sensitivity (842.28 nm/RIU). We have ensured minimum possible reflectivity with maximum sensitivity in this proposed biosensor.

References 1. Jamil NA, Menon PS, Said FA, Tarumaraja KA, Mei GS, Majlis BY (2017) Proc IEEE Reg Symp Micro Nanoelectron RSM 2017:112–115 2. Wood RW (1902) On a remarkable case of uneven distribution of light in a diffraction grating spectrum. Lond Edinb Dublin Philos Mag J Sci 4:396–402 3. Otto A (1968) Excitation of nonradiative surface plasma waves in silver by the method of frustrated total reflection. Zeitschrift für Physik A Hadrons and Nuclei 216:398–410 4. Kretschmann E (1996) Determination of optical constants of metals by excitation of surface plasmon sensing. Sens Actuators B Chem 35:212 5. Liedberg B, Nylander C, Lunström I (1983) Surface plasmon resonance for gas detection and biosensing. Sens Actuators 4:299–304 6. Homola J, Yee SS, Gauglitz G (1994) surface plasmon resonance sensors: review. Sens Actuators B 54:3–15 7. Wijaya E, Lenaerts C, Maricot S, Hastanin J, Habraken S, Vilcot JP, Boukherroub R, Szunerits S (2011) Surface plasmon resonance-based biosensors: from the development of different SPR structures to novel surface functionalization strategies. Curr Opin Solid State Mater Sci 15:208–224 8. Homola J (2008) Surface plasmon resonance sensors for detection of chemical and biological species. Chem Rev 108:462–493 9. Tong L, Wei H, Zhang S, Xu H (2014) Recent advances in plasmonic sensors. Sensors 14:7959– 7973 10. Kretschmann E, Raether H (1968) Radiative decay of non-radiative surface plasmons excited by light. Z Naturforsch 23A:2135–2136 11. Otto A (1968) Excitation of nonradiative surface plasma waves in silver by the method of frustrated total reflection. Z Phys A Hadron Nuclei 216:398–410 12. Cullen DC, Brown RG, Lowe CR (1987) Detection of immuno-complex formation via surface plasmon resonance on gold-coated diffraction gratings. Biosensors 3:211–225 13. Jory MJ, Vukusic PS, Sambles JR Development of a prototype gas sensor using surface plasmon resonance on gratings. Sens. Actuators B Chem 14. Bhatia P, Gupta BD (2011) Surface-plasmon-resonance-based fiber-optic refractive index sensor: sensitivity enhancement. Appl Opt 50:2032–2036 15. Bhatia P, Gupta BD (2013) Surface plasmon resonance based fiber optic refractive index sensor utilizing silicon layer: effect of doping. Opt Commun 286:171–175 16. Singh S, Mishra SK, Gupta BD (2013) Sensitivity enhancement of a surface plasmon resonancebased fiber optic refractive index sensor utilizing an additional layer of oxides. Sens Actuators A Phys 193:136–140 17. Tabassum R, Gupta BD (2015) Performance analysis of bimetallic layer with zinc oxide for SPR-based fiber optic sensor. J Lightw Technol 33:4565–4571 18. Tabassum R, Gupta BD (2017) Influence of oxide overlayer on the performance of a fiber optic SPR sensor with Al/Cu layers. IEEE J Sel Top Quantum Electron 23:81–88 19. Usha SP, Gupta BD (2017) Performance analysis of zinc oxide-implemented lossy mode resonance-based optical fiber refractive index sensor utilizing thin-film/nanostructure. Appl Opt 56:5716–5725

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20. Homola J, M Piliarik (2006) Surface plasmon resonance (SPR) sensors. Ser Chem Sens Biosens 4:45–67 21. Reports on progress in physics. IOPscience 75:036501 (2012) 22. Liedberg B, Lundstrom I, Stenberg E (1993) Principles of biosensing with an extended coupling matrix and surface plasmon resonance. Sens Actuators B Chem 11(1–3):63–72 23. Homola J (2003) Anal Bioanal Chem 377:528 24. Schasfoort RB (2017) Handbook of surface plasmon resonance. Royal Society of Chemistry, Cambridge, UK 25. Homola J, Koudela I et al (1999) Surface plasmon resonance sensors based on diffraction gratings and prism couplers: sensitivity comparison. Sens Actuators B 54:16–24 26. Homola J (1997) On the sensitivity of surface plasmon resonance sensors with spectral interrogation. Sens Actuators B Chem 41:207–211 27. Byun KM, Kim SJ et al (2007) Grating-coupled transmission-type surface plasmon resonance sensors based on dielectric and metallic gratings. Appl Opt 46:5703–5708 28. Ung B, Sheng Y (2007) Interference of surface waves in a metallic nanoslit. Opt Expr 15:1182– 1190 29. Raki´c AD, Djuriši´c AB, Elazar JM, Majewski ML (1998) Optical properties of metallic films for vertical-cavity optoelectronic devices. Appl Opt 37:5271–5283 30. Ung B, Sheng Y (2007) Interference of surface waves in a metallic nanoslit. Opt Express 15:1182–1190

Wide-Band Meander Line Antenna for Ka Band Application Manan Gupta, Ashok Kumar, Amrita Dixit, and Arjun Kumar

Abstract A wide band meander line antenna with coplanar waveguide feed has been presented for Ka band application. It consists of a ladder-shaped feed to provide a large fractional bandwidth of 51.6% ranging from 23 to 39 GHz. Also, over the resonating band we obtain a peak gain of 7.1 dB at 37.5 GHz and a gain of 5.45 dB at the central frequency of 29.11 GHz. The proposed antenna has been simulated on Ansoft HFSS using Rogers RO4003 as the substrate with dimensions 22 × 19 × 1.52 mm3 , which has a relative permittivity (εr ) of 3.55. The proposed monopole antenna can provide broadband characteristics and stable gain radiation performance for Ka band applications. The simulated results have been validated with CST Microwave studio, both the results of Ansys HFSS 19.1 and CST are close to agreement with each other. Keywords 5G · Ka band · Meander line · Millimeter wave (mm Wave)

1 Introduction Wireless communication has become a necessity and has an ever-increasing demand in the market. The rapid development in this technology has increased the demand of efficient antenna designs, i.e., designs which are compact, provide high gain and large bandwidth and have low-maintenance cost. This explosion in the communication industry has provoked an intense research in the 5G communication field [1, 2]. The fifth-generation cellular network (i.e., 5G) can be operated in two different spectrums, one being the present LTE band (ranging from 600 MHz to 6 GHz), and the other in the high-frequency band (ranging from 24 to 86 GHz), the latter falling in a spectrum known as the Ka band also known as millimeter wave band ranging from 26.5 to 40 GHz. The millimeter wave band provides high-data rates (ranging up-to tens of Gbps) and larger bandwidths when compared to the low-frequency range and thus has become an important part of 5G communication. Though at the same time M. Gupta (B) · A. Kumar · A. Dixit · A. Kumar Department of ECE, Bennett University, Greater Noida, India e-mail: [email protected] © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_14

133

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intensive research has found some defects in the Ka band such as penetration loss and propagation losses in free space, but they are substantiated by other parameters such as higher antenna gains and long-range links (> 300 m) [3]. Another important factor of 5G application in Ka band is low latency (almost up-to 1 ms), i.e., the delay before a transfer of data begins following an instruction of its transfer is lowered [4, 5]. Hence, the use of 5G will start the extensive exploitation of the untouched Ka band. Microstrip patch antennas are being used widely due to their compact size, easy to use and installation nature. Larger bandwidths and higher gain are the important characteristics in any antenna designed for 5G. A microstrip patch antenna for 28 GHz frequency band with 6.72 dB gain and 1.1 GHz bandwidth was reported in [6]. One of the drawbacks of microstrip patch antennas, i.e., low bandwidth, has been rectified in the design proposed in this paper [7]. The antenna proposed in this paper has been designed for a 5G network operating in the Ka band. It is a CPW fed antenna with a two-step discontinuity and a meandered patch. Rogers RO4003 has been used here as the substrate as it is more suitable for radiation in high frequencies. This design was simulated on Ansys HFSS 19.1.

2 Proposed Antenna Design In this segment of the paper, the geometry of the proposed CPW feed meander line antenna has been discussed. As shown in the following figure, the antenna consists of a CPW feed with a DGS slot in it. The meander line is fed by a ladder shaped feed. Rogers RO4003, of dielectric constant 3.55, has been used as the substrate to obtain good radiation characteristics of the antenna in Ka band. The substrate size used here is 22 × 19 × 1.52 mm3 and the height of the conducting material i.e. copper is 0.035 mm. Figure 1 depicts the proposed antenna design. The 50  feed impedance is given by taking feed width W f . All the geometric parameters are obtained by simulation on Ansys HFSS and the final geometric parameters are listed in Table 1.

2.1 Parametric Analysis 2.1.1

Effects of variation in Ld

The parametric analysis of S 11 parameters for different values of L d is shown in Fig. 2. From Fig. 2, we can notice variation in L d has a negligible effect on the upper band in the S 11 graph, but the lower band shifts to the lower side as L d is increased. Based on analysis L d = 2.2 mm is chosen for better impedance matching.

Wide-Band Meander Line Antenna for Ka Band Application

135

Fig. 1 Geometry of proposed meander line antenna

2.1.2

Effects of Variation in W d

Figure 3 depicts the variation of S 11 parameters for different values of W d . As we can notice change in W d influences the − 10 dB bandwidth and the return loss. From Fig. 3 we can notice as W d increases the upper band in the S 11 graph gradually shifts to the lower side due to defected ground effect (DGS). On a broader perspective, we notice that bandwidth and W d are inversely proportional. Based on the analysis, we choose W d = 0.2 mm as the most appropriate dimension.

136 Table 1 Parameters of proposed antenna

M. Gupta et al. Parameter name

Value (mm)

Ls

22

Ws

19

L1

3.9

L2

4.4

L3

4.2

L4

4

L5

3

L6

6.5

L7

0.5

L8

4.6

W2

0.4

W3

0.6

Lg

9

Wg

8.25

Ld

2.2

Wd

0.2

Wf

1.5

Hg

0.035

Fig. 2 Effects of L d on antenna performance

3 Result Analysis In this paper, a CPW fed meandered line antenna was designed for 5G communication Rogers RO4003 substrate, for easy transmission at high frequencies. The antenna resonates at central frequency of 29.11 GHz with a gain of 5.45 dBi and gives a

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Fig. 3 Effects of W d on antenna performance

bandwidth of 15.81 GHz and a maximum return loss of 27.0 dB. The simulated S parameter (S 11 ) and peak gain are shown in Fig. 4. The previously published work for Ka band is depicted in Table 2.

4 Conclusion A wideband meander line antenna with simple structure has been presented. With the DGS slot in the ground plane a wide fractional bandwidth of 51.6% was achieved. The ladder-shaped feed helped us to achieve a good impedance matching. This antenna has good radiation characteristics at 28 GHz frequency band, so it is potential candidate for 5G communication in Ka band applications (26.5–40 GHz). Over the operating frequency range, a stable gain is achieved as 6.15 dBi at 24 GHz, 5.45 dBi at 29 GHz and 5.85 at 31.5 GHz.

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Fig. 4 Simulated results a return loss S11, b peak gain (dBi) plot Table 2 Performance comparison of the proposed antenna with existing antenna technology References Size (mm2 ) Gain (dBi) Frequency (GHz) Return loss (dB) Bandwidth (GHz) [2]

3.25 × 3.2

–

[3]

11 × 6.6

–

28

− 20.3

1.25

[6]

5.5 × 4.5

6.72

28

− 18.5

1.1

[7]

16 × 13

4.06

28

− 20

–

[8]

41.3 × 46

This work 22 × 19

28/37.9

− 17.35/− 22

1.485/0.22

13.0

28.4

− 30

11.8

5.45

29

− 27.0

15.81

Wide-Band Meander Line Antenna for Ka Band Application

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References 1. Imran DM, Farooqi M, Khattak MI, Ullah Z, Khan MI, Khattak MA, Dar H (2018) Millimeter wave microstrip patch antenna for 5G mobile communication. In: 2018 international conference on engineering and emerging technologies (ICEET). IEEE, pp 1–6 2. Mohana SP, Veeramani R (2017) Multiband microstrip patch antenna for 5G wireless applications using MIMO techniques. In: 2017 first international conference on recent advances in aerospace engineering (ICRAAE). IEEE, pp 1–5 3. Chen Y, Jian R, Ma S, Mohadeskasaei SA (2017) A research for millimeter wave patch antenna and array synthesis. In: 2017 26th wireless and optical communication conference (WOCC). IEEE, pp 1–5 4. EL_Mashade MB, Hegazy EA (2018) Design and analysis of 28GHz rectangular microstrip patch array antenna. WSEAS Trans Commun 17:1–9 5. Park J-S, Ko J-B, Kwon H-K, Kang B-S, Park B, Kim D (2016) A tilted combined beam antenna for 5G communications using a 28-GHz band. IEEE Antenn Wirel Propag Lett:1685–1688 6. Goyal RK, Modani US (2018) A compact microstrip patch antenna at 28 GHz for 5G wireless applications. In: 2018 3rd international conference and workshops on recent advances and innovations in engineering (ICRAIE). IEEE, pp 1–2 7. Neha K, Sunil S (2018) A 28-GHz U-slot microstrip patch antenna for 5G applications. Int J Eng Dev Res 1:363–368 8. Yoon N, Seo C (2017) A 28-GHz wideband 2× 2 U-slot patch array antenna. J Electromag Eng Sci 17(3):133–137

RETRACTED CHAPTER: Design of Silicon-on-Insulator Based Mode Splitters with Asymmetrical Variation of Slots

ER

Neha Choudhary, Veer Chandra, and Rakesh Ranjan

D

C

H

A

PT

Abstract Due to rapid increase in the demand of internet data rates, it is difficult to satisfy the requirement using the single-mode fiber. Therefore, the multimode fiber can replace the single-mode fiber, by treating each mode as an individual channel, to increase the capacity. Due to intermodal dispersion, the multimode fiber has not been considered in the past for data transmission. Nonetheless, by using each mode of multimode fiber separately, we can increase the overall data rate. However, it is a challenging task to separate the individual mode from a multimode fiber, without mixing, and utilizing them as different signal channels. In this paper, we are demonstrating a mode splitter using the coupled waveguides with slots. By introducing a slot in the waveguide, a desired coupling length ratio of 2:1 between the fundamental and higher modes can be obtained. Also, the asymmetrical variations of slot, and its impact on the coupling length ratio have been demonstrated.

R AC

TE

Keywords Mode splitter · Coupling length ratio · Asymmetrical directional coupler

1 Introduction

R ET

In order to fulfill the exponentially increasing demand of data rate, the optical fiber/waveguide technology based approaches, such as multicore/multimode

The original version of this chapter was retracted: The retraction note to this chapter is available at https://doi.org/10.1007/978-981-16-2818-4_63

N. Choudhary · V. Chandra (B) · R. Ranjan Department of Electronics and Communication Engineering, Optical Fiber Communication and Photonics Laboratory, National Institute of Technology Patna, Patna, Bihar 800005, India e-mail: [email protected] N. Choudhary e-mail: [email protected] R. Ranjan e-mail: [email protected] © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022, corrected publication 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_15

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fiber/waveguide have significant potential to achieve the requirement [1–3]. In multicore fiber technology, due to the presence of closely packed cores, the crosstalk can deteriorate the performance of fiber technology, which usually worsen with the increase in the number of cores in the same cladding region [2]. On the other hand, the few/multimode based multiple-input-multiple-output (MIMO) approach can be the alternative solution to address the rapidly growing demand of information carrying capacity of the optical systems and networks. [3]. For the multimode based optical technology, various devices, such as filters, splitters, couplers, interconnects, etc. [4, 5] are essentially required to investigate the different data signal channels as different modes of few/multimode waveguide/fiber [6]. Signals can be coupled from the multimode fiber to the splitter by using the grating coupler [7], and taper [8] techniques. However, utilizing each of the modes as an independent channel is a challenging task [9]. To undertake this issue, the design of symmetrical mode splitter has been proposed in [10]. The authors have used the symmetrical directional coupler as a mode splitter and evaluated the performance of the proposed structure in terms of coupling length ratio (L R ), which should be equal to 2, to avoid the dispersion phenomena for the efficient reception of the different signal channels at the output. The work presented in [10] has been extended in the current work by designing the asymmetrical mode splitter, in anticipation of improvement in the device performance in terms of different waveguide parameters, while maintaining the coupling length ratio as L R = 2. The asymmetrical structure of the mode splitter has been realized by creating the unsymmetrical directional coupler. Here, in this paper, one of the slots of the symmetrical directional coupler has been shifted by a slot offset, represented by ‘a’ or ‘Δa’, to achieve the design of the asymmetrical directional coupler. The main aim of this work is to achieve the coupling length ratio of 2, by which dispersion can be minimized.

2 Design of Symmetrical and Asymmetrical Directional Coupler

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Directional coupler is the most common device to split or combine the light in the photonic systems. It consists of the parallel waveguides that can be used to separate the different modes in such a way that each mode can act as an independent transmission channels [11, 12]. The design of mode splitter depends on the coupling length ratio, which is the ratio of coupling length for the fundamental mode and the same for the higher mode. Moreover, the coupling length (L C ) is usually dependent on the dimension of the waveguide, including the separation between the waveguides [10], and it can be expressed as in Eq. (1), LC =

π βe − βo

(1)

RETRACTED CHAPTER: Design of Silicon-on-Insulator Based Mode …

143

where, βe and βo are the propagation constants of the even and odd modes, respectively, and it can be calculated by using the Eq. (2) below, β=

2π n λ

(2)

where, n is the effective index and λ is the operating wavelength. Further, the length of the device (L) can be calculated as [10], ij

L = m.L 11 C = n.L C

ER

(3)

PT

where, m and n are the odd and even integer numbers, respectively, or vice-versa, and hence, the coupling length ratio (L R ) can be given by, L 11 C

LR =

ij

A

LC

ij

(4)

TE

D

C

H

where, L 11 C is the coupling length of the fundamental mode and L C is the coupling length of the higher order modes. Figure 1 shows the schematic diagram of a mode y y splitter utilizing the directional coupler, where, H11 and H21 are, respectively, the fundamental and higher order mode, launched at the port 1 of the directional coupler. y Considering m as odd, let m = 1, for the H11 mode, and n as even, let n = 2, for the y y y H21 mode, then, the H11 should appear in the cross-port, and H21 in the bar-port, as illustrated in Fig. 1.

R AC

Side-I

Side-II

y

H 11

y

H 21

Port 1

Bar-port

y

R ET

H 21

y

H 11 Cross-port

Z=0

Fig. 1 Schematic diagram of a mode splitter [10]

Z=L

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2.1 Symmetrical and Asymmetrical Directional Couplers

R ET

R AC

TE

D

C

H

A

PT

ER

The design of symmetrical mode splitter using the symmetrical directional coupler, whose properties mainly depend on the coupling length ratio has been presented in [10] and its cross-sectional view has been provided in Fig. 2. From figure, it is clear that each of the two parallel arms of the symmetrical directional coupler, having equal dimension with total height of ‘H’, is consisting of one slot of height ‘t’, at the middle. The separation between the two arms is ‘S’, the width of slot is ‘g’, and the total width of each arm is ‘W ’. Usually, the symmetrical coupler without slots provides a high value of the coupling length ratio, however, the main objective of the current work is to obtain a lower value of coupling length ratio to minimize the dispersion phenomena. In order achieve it, one slot has been introduced in the middle y of the waveguide. By introducing the slot, the modal properties of fundamental (H11 ) mode are strongly affected, however, it has a very less effect on the modal properties y of the second mode, H21 [10]. As, H y is the dominant field, hence, the effect of slot can be observed by the variations of H y along the x-direction. The input light is provided at a port in silicon core and light get coupled to another waveguide in the coupler, after a certain length known as, coupling length, as shown in Fig. 1. The refractive indices of silicon core, silica buffer, and air cladding are considered, respectively, as 3.47638, 1.44427, and 1.0, at the operating wavelength of 1550 nm. Further, the authors [10] have observed that the L R value decreases by increasing both the slot height as well as the slot width, and the desired coupling length ratio of 2:1 has been realized with g = 150 nm, t = 150 nm, W = 850 nm, S = 100 nm, and slot offset, a = 11 nm. The slot offset is basically represents the shift (left/right) in slot position to realize the variations in the coupling length ratio (L R ) and to achieve the desired ratio, L R = 2. Moreover, in this paper, the work presented in [10] has been extended by considering the asymmetrical directional coupler/mode splitter, to achieve the ratio, L R = 2:1, with the anticipation of reduced slot offset, slot width, etc. The asymmetrical structure, as depicted in Fig. 3, has been realized by shifting the position of one of the slot, left or right, for some range of slot offset (0.1–17 nm). Further, the asymmetrical structure has been simulated using the COMSOL Multiphysics platform,

t

g

Air

W

Slot 1

Slot 2

H

S Si core

Si core

SILICA BUFFER

Fig. 2 Cross-sectional view of a symmetrical directional coupler

RETRACTED CHAPTER: Design of Silicon-on-Insulator Based Mode …

a

t

Air

g

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Slot 1

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H

S Si Si core core

Si core

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Fig. 3 Cross-sectional view of the directional coupler with the asymmetrical shift of slot

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by considering similar waveguide parameters, such as H = 300 nm, S = 100 nm, W = 850 nm, g = 135 nm and 150 nm, and t = 150 nm. As the presence of slot mainly affects the magnetic field distribution of fundamental mode, the coupling length ratio also depends on the geometry of the slot. The authors in [10] have illustrated that the ideal L R value of 2 can be realized at the slot offset, Δa = 11 nm, when the slots of symmetrical coupler have been shifted along right side. While for the other offset values, it is difficult to realize L R = 2. Similar to its symmetrical counterpart, the presence of the slot in asymmetrical waveguide mainly affects the fundamental mode and again, it has very marginal effect on the higher order modes, as observed in terms of the variations of magnetic field along the arc length/width (W ) of the waveguide. The variations in magnetic field along the width of the arms of directional coupler have been shown in Figs. 4a, b, respectively, in the absence (i.e., g = 0), and in the presence of the slot, in the mid of arm of the directional coupler. These variations have been obtained through the COMSOL Multiphysics simulations for the fundamental mode propagation. Therefore, the presence of a slot causes the discontinuity of the magnetic field in the middle of the waveguide. Similarly, for the higher order mode

Fig. 4 Magnetic field variations along arc length/width of coupler for fundamental mode, H11 a without slot, b with slot

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Fig. 5 Magnetic field variations along arc length/width of coupler for higher order mode, H 21 a without slot, b with slot

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3 Simulation Results

D

C

H

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(i.e., H 21 ), Fig. 5 illustrates the variations of magnetic field with respect to the arc length/slot width for both the cases of g = 0 nm, and g  = 0 nm. From this figure, it has been observed that the higher order mode is affected very marginally by the presence of slot, and the field was close to zero, near in the middle of the slot. Further, by shifting the slot on both, right and left sides, by ‘Δa’, the variations of L R values have been observed.

R ET

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The simulations of slot waveguide have been performed mainly with two values of slot gaps, i.e., g = 135 nm, and g = 150 nm, to analyze the performance of asymmetrical directional coupler, and hence, the asymmetrical mode splitter. The impacts on the variations in L R values have been observed for the above two values of slot gaps, in the asymmetrical mode splitter, which have been discussed below. Case-I: Considering the Slots 1 and 2, with the parameters as, W = 850 nm, S = 100 nm, H = 300 nm, t = 150 nm, and g = 150 nm. By shifting the Slot 1 only, towards the right or left side, the variations in L R with respect to the slot offset ‘Δa’ have been observed, and plotted in Fig. 6. The solid (blue) line graph is obtained, when Slot 1 was shifted right side, and the dash (red) line graph was obtained, when the Slot 1 was shifted toward left. From the figure, it is clear that when the Slot 1 was shifted towards the right side, the L R values are approximately close to the optimized value of L R (= 2), and the ratio of 2:1 has been realized at Δa = 14.1 nm. Further, when the Slot 1 was shifted towards the left side, the coupling length ratio decreases and always remains less than 2, as shown in Fig. 6. In the anticipation of reduction in the value of slot offset with the optimized value of L R , i.e., 2:1, the similar kind of analysis has been done with another value of slot width, and presented below as Case-II.

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Fig. 6 Variations in coupling length ratio, of L R , with respect to slot offset, ‘a’, for g = 150 nm

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Fig. 7 Variations in coupling length ratio, L R , with respect to slot offset ‘a’, for g = 135 nm

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Case-II: Considering the Slot 1 again, to shifting it left/right, with the slot width of g = 135 nm, while, other parameters are the same as that in the Case-I. The variations of coupling length ratio with respect to the slot offset, ‘Δa’ has been observed and illustrated in Fig. 7. In this particular case, when the slot was shifted towards the right side by ‘Δa’, the L R values are observed as ≥ 2 (but close to 2) with the increasing values of ‘Δa’. Here, the L R ratio of 2:1 have been realized at a = 3.1 nm and a = 4.1 nm. Moreover, when the slot was shifted towards the left side, the L R value decreases as the value of ‘Δa’ increases and remains less than 2, as shown in Fig. 7. Further, the shifts in Slot 2 only, towards the left side can provide the desired coupling length ratio of 2:1.

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4 Discussion and Comparison

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In the current work, we had designed the asymmetrical directional coupler to obtain the desired coupling length ratio of 2:1. Also, the variations in coupling length ratio with respect to the offset value ‘Δa’, have been investigated for the presented asymmetrical directional coupler. The directional couplers without the slots, usually have the high-coupling length ratio, therefore, a slot has been introduced in the middle of the waveguide that causes to reduce the coupling length ratio. Mainly, the fundamental mode propogations are affected by the introduction of slot, while, the higher modes are affected very marginally. In this work, basically the symmetrical directional coupler is modified into asymmetrical directional coupler by shifting the position of one slot at a time, towards the right/left side. Here in our design, the ideal L R value of 2 has been achieved at Δa = 14.1 nm, for g = 150 nm, and to obtain the L R value of 2 at the smaller slot offset, we had changed the slot width as g = 135 nm. With this reduced slot width, more optimized result in terms of the coupling length ratio of 2:1 has been obtained for the asymmetrical mode splitter, as described in just previous section (Case-II), where the slot offset values have been reduced to 3.1 nm and 4.1 nm. Whereas in case of symmetrical coupler, the authors in [10] have obtained the coupling length ratio of 2:1 for the slot offset value of 11 nm, with g = 150 nm, and after an offset value of 11 nm, the L R becomes less than 2. Therefore, in comparison to symmetrical directional coupler, the designed asymmetrical directional coupler can achieve the desired value of coupling length ratio (2:1) with the significantly reduced slot offsets of 3.1 and 4.1 nm. Hence, the obtained coupling length ratio is decently optimized at g = 135 nm, in comparison to g = 150 nm.

5 Conclusion

R

ET R

In order to use each mode as an individual channel for data transmission, the mode splitter has been designed using the asymmetrical directional coupler, to get an optimized value of coupling length ratio, i.e., 2:1, which is also beneficial to avoid the dispersion phenomena. For the design of a mode splitter, an asymmetrical shifts in waveguide slots have been shown, which was used to get a desired value of L R (= 2). From the results, it has been observed that as the Slot 1 was shifted towards the right side, the L R value of 2 was achieved, whereas, when the Slot 1 was shifted towards the left side, the L R value decreases and becomes less than 2. Similarly, when we shift the Slot 2 towards the left side, the L R ratio of 2:1 was achieved and when the Slot 2 was shifted toward the right side, L R value decreases, and remains always less than 2. Hence, it can be concluded that as the distance between slots decreases, the coupling length ratio of 2:1 can be realized, and conversely, when the distance between slots increases, the L R value decreases, and achieved as, less than 2.

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Acknowledgements The authors gratefully acknowledge National Institute of Technology Patna, and Science and Engineering Research Board, Department of Science and Technology, Government of India for providing COMSOL Multiphysics simulation software, used in the current simulation work.

References

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1. Yuan L, Liu Z, Yang J (2006) Coupling characteristics between single-core fiber and multicore fiber. Opt Lett 31:3237–3239 2. Fini JM, Zhu B, Taunay TF, Yan MF (2010) Statistics of crosstalk in bent multicore fibers. Opt Express 18:15122–15129 3. Yu CP, Liou JH, Chiu YJ, Taga H (2011) Mode multiplexer for multimode transmission in multimode fibers. Opt Express 19:12673–12678 4. Xu X, Ma L, He Z (2018) 3D polymer directional coupler for on-board optical interconnects at 1550 nm. Opt Express 26:16344–16351 5. Kyriakis-Bitzaros ED, Haralabidis N, Lagadas M, Georgakilas A, Moisiadis Y, Halkias G (2001) Realistic end-to-end simulation of the optoelectronic links and comparison with the electrical interconnections for system-on-chip applications. J Lightwave Technol 19(10):1532– 1542 6. Hada SL, Rahman BMA (2015) Design concepts of a novel mode splitter for multimode communication systems. In: 6th international conference on computers and devices for communication (CODEC), Kolkata, pp 1–4 7. Loh T-H, Wang Q, Zhu J, Ng K-T, Lai Y-C, Huang Y, Ho S-T (2010) Ultra-compact multilayer Si/SiO2 GRIN lens mode-size converter for coupling single-mode fiber to Si-wire waveguide. Opt Express 18:21519–21533 8. Jia L, Zhou H, Liow T-Y, Song J, Huang Y, Xiaoguang Tu, Luo X, Li C, Fang Q, Mingbin Yu, Lo G (2015) Analysis of the polarization rotation effect in the inversely tapered spot size converter. Opt Express 23:27776–27785 9. Zhu B, Taunay TF, Yan MF, Fishteyn M, Oulundsen G, Vaidya D (2010) 70-Gb/s multicore multimode fiber transmissions for optical data links. IEEE Photonics Technol Lett 22(22):1647– 1649 10. Hada SL, Rahman BMA (2019) Design of compact mode splitters using identical coupled waveguides with slots. OSA Continuum 2:848–861 11. Pan C, Rahman BMA (2016) Accurate analysis of the mode (de)multiplexer using asymmetric directional coupler. J Lightwave Technol 34(9):2288–2296 12. Lu Z, Yun H, Wang Y, Chen Z, Zhang F, Jaeger NAF, Chrostowski L (2015) Broadband silicon photonic directional coupler using asymmetric-waveguide based phase control. Opt Express 23:3795–3808

Human Body Monitoring Wearable Antenna Vani Sadadiwala, Kashish Mahindroo, Vimlesh Singh, Priyanka Bansal, and Sarthak Singhal

Abstract This paper is an academic study on multiband wearable antenna. Proposed antenna designed on denim fabric as substrate on which circular patch is radiating element. Multiband characteristics of wearable antenna are achieved for frequency of 2.7, 4.1, 7 and 9.3 GHz. Designed wearable antenna has benefits of multiband and wideband characteristics over early reported wearable antenna. This antenna is fully compatible with human body and hence can easily be fitted with woven fabric. Keywords Wearable · Multiband · Monitoring · Textile

1 Introduction Wearable antennas design offers the possibilities of communication, patient monitoring, storage and energy harvesting. The planar structure and the materials for flexible construction are the requirement of a wearable antenna. There has been a continuous growth from the past few years in the human body application [1–6]. For design of wearable biomedical system, low-profile compact antennas are very important. Along with the gradual improvement in this system of wireless communication in the modern biomedical field, there is a major role of the implanted antenna in communicating with the external devices. The telecommunication systems that will be integrated within the garments, when fully developed will have the ability of modifying and then demanding for attention if or when required which leads to V. Sadadiwala · K. Mahindroo Aeronautical Engineering, FET, MRIIRS, Faridabad, India V. Singh (B) · P. Bansal Electronics and Communication Engineering, FET, MRIIRS, Faridabad, India e-mail: [email protected] S. Singhal Electronics and Communication Engineering, MNIT, Jaipur, India e-mail: [email protected]

© The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_16

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minimization of hospital resources and labour. The wearable devices for taking in account the different biological signals are gathering huge interest these days. The use of textile material as a substrate material for the antenna design is increasing rapidly. The ability to sense and communicate data may be seen in the new generation of clothing [7]. To get better results, these wearable textile antennas need to be fine/thin, low-maintenance, sturdy, lightweight, low cost and also simply integrated in the radio frequency circuit. The planar structure antennas are the preferable sort of antennas which allows excellent antenna integration with RF circuits. The biomedical electronics play a very important role for medical treatment, diagnosis, and in discipline academics [8–13]. The antennas can be inserted in the human bodies or they can simply be placed on top of the torso which is a skin fat muscle for the formation of a biomedical electronic system which use to communicate between the medical devices and the external instruments in low ranged biotelemetry usage. Other than this there are several visible benefits of wearable systems in healthcare in terms of implanted devices which is economical accepted as well as equally admissible [14–18].

2 Characteristics of Wearable Antenna Properties of the fabric are mainly determined by structure of yarn used in weaving of that fabric as well as properties of component fibres. The density and thickness of fabrics change with change in pressures on yarn. Thus, it become essential to understand that how density and thickness of a fabric influences the behaviour of antenna. For solving our purpose, we need to study about the basic parameters of antenna/dielectric substrate which in turn require study of basic properties of fabric [7, 19–22].

2.1 Dielectric Properties The most important characteristics of any material are named as dielectric constant. Permittivity is a complex valued parameter which is constitutive parameter of dielectrics. Dielectric properties of any material are a dependent variable of frequency, surface roughness, temperature, homogeneity of material, purity and the moisture content in the material. The behaviour of the material is described by the relative permittivity which is tested under specified electric field frequency and orientation. The textile fibres are porous, rough, heterogeneous and have air filled spaces in between. Thu,s accurate measurement of the dielectric constant poses a challenge and a vast range of experimental methods have been used, for instance, MoM segment method, cavity perturbation method, free space method, resonance method

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and transmission line method which is the simplest and promising one. Fabrics are mostly porous in nature because of the presence of air in the structure itself, thus the dielectric constant of fabric material is very low. This low-dielectric constant provides an additional benefit of increase in spatial wave which in turn allow the development of high-gain antenna [23, 24].

2.2 Thickness of Fabric Thickness of dielectric fabric is another important parameter because it is a deciding factor for bandwidth, input impedance and resonance frequency of the antenna. Height (thickness) of dielectric material is actually a judgemental factor which defines bandwidth and efficiency antenna. Thickness of substrate is also a deciding factor for the size of antenna because a substrate has more height of dielectric material has relatively small value of permittivity which result increase in size of patch and a thin substrate material will result in decrease in size of patch [22].

2.3 Surface Resistivity of Fabric Another important parameter required to study characteristic of a fabric is surface resistivity, it defined by ratio of DC voltage across per unit length to the surface current per unit width. For a perfect antenna design the surface resistance must have a constant value over the entire surface area of the antenna but there may be some heterogeneities present in the fabric which result in discontinuities in electric current [25, 26]. Conductive fabrics perform better than coated fibres because of the presence of discontinuities in material that increases the surface resistance of coated fabrics. Knits have far higher anisotropy than woollens and this anisotropy can be further increased with the knit deformation. There is also some difference between different faces of fabrics and this must be considered, whilst positioning the conductive fabrics in the planar assemblage.

2.4 Moisture Content in Fabric The moisture content of a fabric is in equilibrium with humidity and temperature of air of the surroundings. The sensitivity of the material towards moisture content explained by regain property of material. There are many textile material having different moisture contents absorption ratio but have same relative humidity conditions [7, 27–30]. When the textile fibres trap water they swell axially, which tightens

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the fabric. This affects the dimension stability of fabric and so of the antenna which influences its behaviour. Water absorption is an exothermic reaction which releases heat. Thus, amount of heat released is more for high water absorbing fibres. This property in turn affects the electromagnetic properties of the fabric. Therefore, any change in surrounding climate alters the relative content of environmental water or air which influences the antenna performance. This process is termed as hydeophobicity. Regain factor and hydrophobicity are two important factor of the fabric which has dominant characteristics when antenna is wear on the skin, as fabric absorbs moisture from the skin. So low-regain materials are preferable to be used as substrates [8, 9, 31].

2.5 Mechanical Deformation Human body has curves all over that also show that curvature on the human body results super positioning of bending in arbitrary directions. But antenna made from fabric material possess outstanding elasticity and flexibility, thus these materials adapt to the surfaces very well. Textile fabrics bend and deform thus changing electromagnetic properties which in turn influences the antenna performance. The resonance frequency, bandwidth of antenna, thickness, permittivity all these parameters are influenced by the elongation of dielectric fabric. As fabric materials possess elasticity, it is challenging to super position various materials for various layers of antenna without folds. Non-wovens and wovens fabric provide accuracy of higher degree in terms of geometric structure for the frame of the antenna [4, 5, 24, 29].

3 Steps of Design of Wearable Antenna Wearable antenna has deformable characteristics of textile material. The mechanical stabilization of patch as well as dielectric substrate both is necessary to maintain the desired antenna characteristics. Then, a major problem arises are connect between different layers of antenna so that electrical properties of patch is not affected at all [20]. Good results are obtained by connecting different layers using the adhesive sheets and conductive fabrics with a thermal adhesive face. The relative permittivity of the substrate and surface resistance of the patch is not significantly changed as the adhesive stays at the interface of the materials. An alternative technique is connection with seams that is also a tough task [13, 32–34]. The seam should be done by taking care of wrinkles and the stitch must be passed through the patch, the substrate, and the ground plane of the antenna, but this type of stitches may cause electrical shots between them. The positioning of the textile components should be taken care of as there is considerable difference between back and front face in terms of density and

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roughness of the conductive elements. Putting connections at antenna terminal are also a critical task in case of wearable and flexible antennas as these connections need to be mechanically robust [21, 35, 36].

4 Proposed Antenna A wearable antenna is proposed in this paper is shown in Fig. 1. Designed antenna is fabricated on textile as substrate of Z top = 3 mm, relative permittivity of Jeans fabric used as substrate is 1.7 and loss tangent of 0.002. Antenna is simulated for copper tap of 1-mm thickness. Proposed geometry is show in Fig. 1a–c. Details of proposed antenna are given in Table 1. Radiating patch formed by subtracting three circles from radiating circular patch for increase effective area. Feed length and width of 5 × 49 mm2 to achieve 50  characteristic impedance. To achieve resonant frequency radius of radiating patch dimensions are predicted by [35]:

a: Top View

b : Back View

c: With Waveguide Port

Fig. 1 Proposed antenna structure

Table 1 Proposed antenna dimensions Antenna structure

Patch diameter (mm)

Inner circle diameter (mm)

Lower circle diameter (mm)

Partial ground (mm2 )

Feed length (mm)

Feed width (mm)

Substrate (mm2 )

Dimension

27

4

2

50 × 42

5

49

60 × 80

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Ractual =  1+

2h πεr f



 ln

f πf 2h

+ 1.7726

 1/2

8.791 × 109 √ f r εr

  1/2 πR 2h ln = R 1+ + 1.7726 π εr R 2h

Frequency( f ) = Reffective

5 Result and Discussion Proposed wearable antenna is simulated by CST microwave studio simulator. Figure 2 shows S11 parameter of simulated antenna structure. This show that proposed antenna behave as multiband antenna for centre frequency 2.64, 4.14, 6.89 and 9.31 GHz, with gain of − 13, − 40, − 30 and − 18 dB. The proposed antenna analyzed for bending effect along the y-axis with the cylindrical radius of 10 mm for patch as well as substrate material for bending of 30°, 60° and 90°. Antenna S11 parameter is compared with actual radius and bending radius which show small shift in resonance frequency appears due to bending of radiating patch as in Fig. 3. The result shows good arrangement resonance frequency and bandwidth of antenna between without bending and with bending. The proposed antenna far field pattern is show omnidirectional pattern in Fig. 4. The small variation in far field pattern appears during bending this pattern is almost omnidirectional as Fig. 4a–c. Distribution of current in radiating patch without bending is shown in Fig. 5a–d.

Fig. 2 S11 parameter of proposed antenna structure

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Fig. 3 Bending effect of proposed antenna

Table 2 shows simulated result of proposed wearable antenna. At the 2.64 GHz bandwidth of antenna is 570 MHz, for 4.14 GHz bandwidth is 1170 MHz, 6.98 GHz bandwidth is 1110 MHz and for 9.31 GHz it is 700 MHz.

6 Conclusion The proposed antenna is claiming human body monitoring in different telecommunication application. This wearable antenna work as multiband antenna in WBAN system such as passive sensor, radiolocation antenna for 2.64, 4.14 GHz work as fixed satellite mobile, 6.98 GHz as passive sensors (satellite) and aeronautical navigation, aeronautical military systems, active sensors (satellite), shipborne, land and airborne surveillance for 9.31 GHz frequency. This proposed antenna is compact in size, high efficiency and omnidirectional radiation pattern for resonance frequency with suitable specific absorption rate.

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a. Far Field Parameter of proposed antenna without Bending

b. Far Field Parameter of proposed antenna with Bending of 300

c. Far Field Parameter of proposed antenna with Bending of 600

d. Far Field Parameter of proposed antenna with Bending of 900

Fig. 4 Far field parameter of proposed antenna with and without bending effect

a:2.64GHz

b:4.14GHz

Fig. 5 Current distribution in proposed antenna

c:6.98GHz

d:9.31GHz

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Table 2 Simulated results of proposed wearable antenna Antenna geometry

F L (GHz)

F H (GHz)

Centre frequency (GHz)

Reflection coefficient (dB)

Bandwidth

Without bending

2.32

2.89

2.64

− 13

570 MHz

3.646

4.816

4.14

− 40

1170 MHz

6.472

7.58

6.98

− 30

1110 MHz

Bending at 30°

Bending at 60°

Bending at 90°

9.30

10

9.31

− 18

700 MHz

2.323

2.899

2.66

− 14

580 MHz

3.646

4.816

4.14

− 40

1170 MHz

6.472

7.58

6.98

− 30

1110 MHz

9.316

10

9.31

− 18

700 MHz

2.39

2.998

2.72

− 16.13

600 MHz

3.799

4.609

4.19

− 21

810 GHz

5.97

7.912

7.22

− 20

1940 MHz

9.208

10

9.71

− 33

792 MHz

2.503

3.088

2.81

− 20.15

594 MHz

3.979

4.573

4.27

− 16

590 MHz

5.76

7.939

7.37

− 26

2179 MHz

9.28

10

9.63

− 38

720 MHz

References 1. Lee HS, Kim JG, Hong S, Yoon JB (2005) Micromachined CPW-fed suspended patch antenna for 77 GHz automotive radar applications. In: Wireless technology-2005, The European conference, pp 249–252 2. Nikolaou S, Ponchak GE, Papapolymerou J, Tentzeris MM (2006) Conformal double exponentially tapered slot antenna on LCP for UWB applications. IEEE Trans Antenn Propag 54(6):1663–1669 3. Koulouridis S, Kiziltas G, Zhou Y, Hansford DJ, Volakis JL (2006) Polymer–ceramic composites for microwave applications: fabrication and performance assessment. IEEE Trans Microw Theory Techn 54(12):4202–4208 4. Numakura D (2007) Flexible circuit applications and materials—printed circuit 5. Galehdar A, Thiel DV (2007) Flexible, light-weight antenna at 2.4 GHz for athlete clothing. IEEE Trans Antenn Propag:4160–4163 6. Halonen E, Kaija K, Mantysalo M, Kemppainen A, Osterbacka R, Bjorklund N (2009) Evaluation of printed electronics manufacturing line with sensor platform application 7. Li Y, Zhang Z, Feng Z, Khaleel HR (2014) Innovation in wearable and flexible antennas. WIT Press 8. Scarpello ML, Kazani I, Hertleer C, Rogier H, Ginste DV (2012) Stability and efficiency of screen-printed wearable and washable antennas. IEEE Antenn Wirel Propag Lett 11:838–841 9. Wang Z, Zhang L, Bay Ram Y, Volakis JL (2012) Embroidered conductive fibers on polymer composite for conformal antennas. IEEE Trans Antenn Propag 60(9):4141–4147 10. Khaleel HR, Al-Rizzo HM, Rucker DG, Mohan S (2012) A compact polyimide-based UWB antenna for flexible electronics. IEEE Antenn Wirel Propag Lett 11:564–567 11. Kaufmann T, Fumeaux C (2013) Wearable textile half-mode substrate-integrated cavity antenna using embroidered vias. IEEE Antenn Wirel Propag Lett 12:805–808

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12. Monti G, Corchia L, Tarricone L (2013) UHF wearable rectenna on textile materials. IEEE Trans Antenn Propag 61(7):3869–3873 13. Casula GA, Montisci G, Mazzarella G (2013) A wideband PET inkjet-printed antenna for UHF RFID. IEEE Antenn Wirel Propag Lett 12:1400–1403 14. Lakafosis V, Rida A, Vyas R, Yang L, Nikolaou S, Tentzeri MM (2010) Progress towards the first wireless sensor networks consisting of inkjet-printed; paper-based RFID-enabled sensor tags. Proc IEEE 98(9):1601–1609 15. Hertleer C, Van Laere A, Rogier H, Van Langenhove L (2010) Influence of relative humidity on textile antenna performance. Textile Res J 80(2):177–183 16. Kirsch NJ, Vacirca NA, Kurzweg TP, Fontecchio AK, Dandekar KR (2010) Performance of transparent conductive polymer antennas in a MIMO ad-hoc network. In: Wireless and mobile computing; networking and communications, pp 9–14 17. Virtanen J, Bjorninen T, Ukkonen L, Sydanheimo L (2010) Passive UHF inkjet-printed narrowline RFID tags. IEEE Antenn Wirel Propag Lett 9:440–443 18. Koo TW, Kim D, Ryu JI, Seo HM, Yook JG, Kim JC (2011) Design of a label-typed UHF RFID tag antenna for metallic objects. IEEE Antenn Wirel Propag Lett 10:1010–1014 19. Alqadami ASM, Jamlos MF (2014) Design and development of a flexible and elastic UWB wearable antenna on PDMS substrate, pp 27–30 20. Singh N, Singh AK, Singh VK (2015) Design and performance of wearable ultrawideband textile antenna for medical applications. Microw Opt Technol Lett 57(7):1553–1557 21. Zahran SR, Gaafar A, Abdalla MA (2016) A flexible UWB low profile antenna for wearable applications. In: international conference on “computing for sustainable global development, pp 1931–1932 22. Singh V, Bansal P, Singhal PK (2018) Microstrip line antenna fabrication material. Int J Eng Technol 7(2.8):340–344 23. Hertleer C, Rogier H, Vallizzi L, Van Langenhove L (2009) A textile antenna for offbody communication integrated into protective clothing for firefighters. IEEE Trans Antenn Propag:919–925 24. Kaija T, Lilja J, Salonen P (2010) Exposing textile antennas for hash environment. In: Proceedings of military communications conferences, pp 737–742 25. Grupta B, Sankaralingam S, Dhar S (2010) Development of wearable and implantable antennas in the last decade: a review in proceedings of mediterranean microwave symposiums, pp 251– 267 26. Kim Y, Kim H, Yoo HJ (2010) Electrical characterization of screen-printed circuits on the fabric. IEEE Trans Adv Packaging 33(1):196–205 27. Kaivanto EK, Berg M, Salonen E, de Maagt P (2011) Wearable circularly polarized antenna for personal satellite communication and navigation. IEEE Trans Antenn Propag 59(12):4490– 4496 28. Liu N, Lu Y, Qiu S, Li P (2011) Electromagnetic properties of electro-textile for wearable antennas application. Front Electron Eng China 6:553–566 29. Ha SJ, Jung CW (2011) Reconfigurable beam steering using a microstrip patch antenna with a U-slot for wearable fabric applications. IEEE Antenn Wirel Propag Lett 10:1228–1231 30. Zhang L, Wang Z, Psychoudakis D, Volakis JL (2012) Flexible textile antennas for body-worn communication. In: Proceedings of IEEE international workshop on antenna technology, pp 205–208 31. Khaleel HR, Al-Rizzo HM, Rucker DG (2012) Compact polyimide-based antennas for flexible displays. J Display Technol 8(2):91–97 32. Wang H, Zhang Z, Li Y, Feng Z (2013) A dual-resonant shorted patch antenna for wearable application in 430 MHz band. IEEE Trans Antenn Propag 61(12):6195–6200 33. Chahat N, Zhadobov M, Muhammad SA, Le Coq L, Sauleau R (2013) 60-GHz textile antenna array for body-centric communications. IEEE Trans Antenn Propag 61(4):1816–1824 34. Shi SR, Wu YM, Lu PP, Wu Q (2014) Investigation on the radiation features of the flexible antenna at 5.8 GHz. Antenn Propag:1496–1500

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On Design of Airborne Radomes with Brent’s Method for Radiation Pattern Peak Detection Romesh Srivastava, Aparna Parameswaran, and Hrishikesh S. Sonalikar

Abstract In this paper, a one-dimensional Brent’s method based on inverse parabolic interpolation is proposed for detection of antenna radiation pattern peak. The computational efficiency of Brent’s method is compared to that of the golden section search method. It will be shown that the Brent’s method requires less number of function evaluations to reach the desired accuracy and hence superior. The Brent’s method is then incorporated into 3D ray tracing method to compute the performance parameters of streamlined airborne radome and optimize the radome wall thickness. The resulting radome design is found to have a satisfactory performance for airborne applications. Keywords Radome · Antenna · Boresight error · Power transmission · 3D ray tracing · Inverse parabolic interpolation · Brent’s method · Golden section search.

1 Introduction Radome is a cover used for antennas for the protection from severe operating environment [1]. In airborne vehicles, streamlined radomes are used to protect scanning antennas. Although radome should be transparent to electromagnetic radiation from the underlying antenna, practical radomes degrade antenna performance by causing insertion loss, introducing boresight errors and cross polarization [2]. For the radome to be useful in military applications, very stringent specifications on power transmission and boresight error performance must be achieved. Also, the mechanical and structural requirements of radome sometime contradict the electromagnetic requirements. As a result, the design of airborne radomes is active topic of research. The design of airborne radome requires repeated calculation of power transmission and boresight error performance over a wide antenna scan angle range. Calculation of these parameters depends on accurate angular position and value of the radiation R. Srivastava · A. Parameswaran · H. S. Sonalikar (B) Goa Campus, BITS Pilani, Zuarinagar, Goa, India e-mail: [email protected] URL: https://universe.bits-pilani.ac.in/goa/Hrishikeshs/profile © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_17

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pattern peak. Therefore, it is essential to choose fast and efficient method to reduce computation time and cost. A simple iterative method suggested in [3] provided a good improvement in computation time over a non-iterative method. Another method based of golden section search (GSS) was proposed in [4]. This method significantly reduced the number of function evaluations required. The GSS method does not make any assumptions about the smoothness of a function and can handle uncooperative functions effectively. However, antenna radiation patterns are usually smooth functions of angles in vicinity of the maximum. Also, the shape of the main lobe of radiation pattern resembles that of a parabola. This observation suggests that a method based on inverse parabolic interpolation can be more effective in detection of radiation pattern peak. This is the motivation behind the present work. In this paper, a one-dimensional Brent’s method based on inverse parabolic interpolation is implemented to numerically compute the position of radiation pattern maximum. The computational efficiency of Brent’s method is compared to that of the GSS method in terms of number of function evaluations and computation time. It will be shown that the Brent’s method is superior to the GSS method as it requires less number of function evaluations to reach the desired accuracy. The Brent’s method is then incorporated in the 3D ray tracing method to compute the radome performance parameters and optimize the wall thickness of the radome. The resulting radome design is found to have a satisfactory performance for airborne applications. Presentation of this work is organized as follows. Section 2 describes the antennaradome configuration considered in this work. Section 3 presents the numerical methods used. Results and discussion are presented in Sect. 4. Finally, Sect. 5 presents concluding remarks.

2 Antenna and Radome Configuration In this work, a tangent ogive radome with a constant thickness monolithic wall is considered. The geometry of radome with enclosed antenna is shown in Fig. 1. x  , y  , z  represent axes of antenna coordinate system with x  serving as the antenna axis. x, y, z represent axes of radome coordinate system with x serving as the radome axis. An array antenna with a circular aperture of diameter 0.2 m is enclosed within the radome. The antenna operates at a frequency of 10 GHz and is mounted on an EL/AZ gimbal with zero rotational offsets. Electromagnetic waves with vertical polarization originate from the 112 source points on the antenna aperture placed half-wavelength apart. The optimized radome wall thickness of 6.9192 mm is considered to ensure maximum power transmission. The dielectric constant and loss tangent of the material used for the radome wall is 5.515 and 0.0003, respectively.

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Fig. 1 Side view of tangent ogive radome enclosing antenna

3 Numerical Methods 3.1 3D Ray Tracing Method A 3D ray tracing method is used for analysis of the antenna radome system [2]. The ray tracing methods are well understood and useful for computationally efficient analysis of electrically large objects such as airborne radomes enclosing aperture antennas. In this method, the EM waves radiated by the antenna are considered to have properties like high frequency optical rays. These rays intersect with the radome wall and undergo a reflection as well as transmission. In this work, a MATLAB program based on geometrical optics in transmitting mode is developed to implement the 3D ray tracing method. The program goes through the following stepwise procedure: (1) Coordinate transformation, (2) construction of ray and electric field vectors, (3) obtain points of intersection of ray and radome, (4) obtain angle of incidence, and (5) obtain effective transmission coefficient [5]. Finally, the sum port radiation pattern of the antenna is computed as [2]: M S=

i=1

Fa e− j sin θ(yi cos φ+zi sin φ) Ti . M i=1 Fa

(1)

Here, M is both the number of source points as well as the number of rays originating from these source points. (yi , z i ) are the coordinates of ith source point in the antenna coordinate system. Ti is the transmission coefficient for the ith ray with θ and φ being the angles of spherical coordinate system in the antenna space. Fa is a cosine aperture field distribution.

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The radiation pattern of a radome enclosed antenna is computed as a function of spherical coordinates θ and φ. To compute the radiation pattern in the elevation plane x − y, φ = 0 and θ is varied. To compute the radiation pattern in the Azimuth plane x − z, φ = π/2 and θ is varied. Therefore, the radiation pattern is calculated as a function of a single variable θ . As the antenna is enclosed by the radome and scanned in the elevation and Azimuth plane, the peak of the radiation pattern is determined numerically. Once the radiation pattern peak is detected, the radome performance parameters can be obtained. The power transmission (PT) and the boresight error (BSE) of the radome is given by [6]:   PT = 20 log10 (|Speak |) and BSE = angle |Speak | ,

(2)

respectively. Here, |Speak | represents the peak magnitude of sum port radiation pattern, whereas angle(|Speak |) is the angular location of sum pattern peak. The accuracy of the radiation pattern peak determines the accuracy of BSE and PT.

3.2 Brent’s Method In this paper, we compare the computational efficiency of the golden section search (GSS) method with that of the Brent’s method for the detection of the sum port radiation pattern peak [7]. In both the methods, the interval of angle θ bracketing the maximum value of radiation pattern is reduced in successive iterations until a value satisfying the desired precision is obtained. The description of the golden section search method for radiation pattern peak detection is given in our earlier work [4]. As the focus of the present work is on Brent’s method, it will be described in step-wise manner as follows. • Step 1: The initial interval (θstart , θend ) bracketing the maximum of the function is chosen. Another point is chosen at the midpoint of the interval (θstart , θend ). It is referred to as θb1 . • Step 2: Sum pattern is evaluated at θstart , θend , and θb1 . Let these function values be denoted as f (θstart ), f (θend ), and f (θb1 ). • Step 3: A second point is also chosen within the interval (θstart , θend ). This point is referred to as θb2 . This point is the abscissa of the maximum of the parabola which goes through θstart , θend , and θb1 . Hence, it is calculated as θb2 = θb1 −

1 · F(θb1 , θstart , θend ) 2

(3)

where F is given by F≡

θb21 ,start ·  f b1 ,end − θb21 ,end ·  f b1 ,start θb1 ,start ·  f b1 ,end − θb1 ,end ·  f b1 ,start

.

(4)

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In Eq. 4, θb1 ,start = θb1 − θstart , θb1 ,end = θb1 − θend ,

(5)

 f b1 ,end = f (θb1 ) − f (θend ),  f b1 ,start = f (θb1 ) − f (θstart ).

(6)

• Step 4: Value of θb2 is compared with that of θb1 . If θb2 < θb1 , then the new bracketing interval is determined by the following steps. • If f (θb2 ) > f (θb1 ) then θend,new = θb1 , f (θend,new ) = f (θb1 ),

(7)

θb1 ,new = θb2 , f (θb1 ,new ) = f (θb2 ).

(8)

• If f (θb2 ) < f (θb1 ) then θstart,new = θb2 , f (θstart,new ) = f (θb2 ).

(9)

• If f (θb2 ) = f (θb1 ) then • If f (θend ) > f (θstart ) then

• else,

θstart,new = θb2 , f (θstart,new ) = f (θb2 ),

(10)

θend,new = θb1 , f (θend,new ) = f (θb1 ),

(11)

θb1 ,new = θb2 , f (θb1 ,new ) = f (θb2 ).

(12)

On the other hand, if θb2 > θb1 , then the new bracketing interval is calculated as, • If f (θb2 ) > f (θb1 ) then θstart,new = θb1 , f (θstart,new ) = f (θb1 ),

(13)

θb1 ,new = θb2 , f (θb1 ,new ) = f (θb2 ).

(14)

• If f (θb2 ) < f (θb1 ) then θend,new = θb2 , f (θend,new ) = f (θb2 ).

(15)

• If f (θb2 ) = f (θb1 ) then • If f (θend ) > f (θstart ) then θstart,new = θb1 , f (θstart,new ) = f (θb1 ),

(16)

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θb1 ,new = θb2 , f (θb1 ,new ) = f (θb2 ),

(17)

θend,new = θb2 , f (θend,new ) = f (θb2 ).

(18)

There might arise a case when θb2 is equal to θb1 . In such a situation, it is essential to choose a different value for θb1 . To ensure the minimum number of function evaluations per iteration, function evaluation at θb2 is done only after it is confirmed that θb2 is not equal to θb1 . • Step 5: Repeat steps 3 and 4 until |θend − θstart | < tol

(19)

where tol is the error tolerance set by the user. • Step 6: When the desired precision is achieved, the value of θ at which the sum pattern is maximum is chosen as the final value θ peak . • Step 7: The value of PT and BSE is calculated as PT = 20 log10 (|Speak |) = 20 log10 (|S(θ peak )|), BSE = θ peak .

(20)

4 Results and Discussion To numerically compute the position of radiation pattern peak, the initial interval of θ bracketing the maximum value of radiation pattern should be obtained first. Figure 2a shows the phase distribution on antenna aperture at the antenna scan angle values of 0◦ , 15◦ , 30◦ , 45◦ , 60◦ and 75◦ in the azimuth plane. At the 0◦ scan angle, the antenna axis coincides with the radome axis. Therefore, the view of the radome as seen from the antenna is symmetric. As the antenna is scanned at different angles, the view of the radome as seen from the antenna becomes asymmetric because different rays emanating from the antenna undergo different phase delays in asymmetric manner. At higher antenna scan angle such as 75◦ , the radome looks almost flat as seen from the figure. Figure 2b shows the magnitude of complex sum patterns of the antenna obtained at the same scan angles in the Azimuth plane. For the scan angle of 0◦ , the symmetric aperture phase distribution results in the symmetric sum pattern with maximum at θ = 0◦ . As a result, the BSE at this scan angle is zero. However, for antenna scan angles other than 0◦ , the position of the sum pattern peak is shifted due to asymmetric aperture phase distribution and results in a nonzero value of BSE. These initial results show that 1. the parabolic shape of the sum pattern may be exploited for the faster peak detection using Brent’s method and 2. the initial bracketing interval of θ ranging from −4◦ to 4◦ is suitable for the antenna radome configuration considered.

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Fig. 2 a Antenna aperture phase distribution at different antenna scan angles. b Antenna sum patterns as a function of angle θ at different antenna scan angles Table 1 Comparison between computational efficiencies of the Brent’s method and the golden section search method Error tolerance Brent’s method Golden section search method T, s Function T, s Function evaluations evaluations 1e−3 1e−4 1e−5 1e−6 1e−7 1e−8 1e−9

0.027295 0.039686 0.043379 0.048827 0.062792 0.074595 0.088715

3 6 7 8 11 14 17

0.056813 0.078841 0.100769 0.123095 0.142273 0.161977 0.184874

11 16 21 26 30 35 40

Table 1 shows the comparison between the Brent’s method and the golden section search method in terms computation time and the number of function evaluations required to compute BSE and PT from the antenna sum pattern at a single antenna scan angle. As the error tolerance is reduced, both the processing time and the number of function evaluations increase. It can be observed that for any given error tolerance value, the number of function evaluations required by the Brent’s method is substantially less compared to the GSS method. As a result, the Brent’s method requires lesser processing time and hence is more efficient. For the successively decreasing values of error tolerance, the increase in the number of function evaluations for the GSS method is 5, whereas for the Brent’s method, this number is 1–3. Figure 3 shows the variation of BSE as a function of the antenna scan angle for different values of the error tolerance. A very crude approximation dictated by the

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(b)

Boresight error, mrad

Boresight error, mrad

(a) 1E-1

60

40

20

1E-3

0.5 0 -0.5 -1 -1.5 -2

0 20

40

60

80

0

20

40

60

80

Antenna scan angle, deg

(c)

(d)

0.5

1E-4

0 -0.5 -1 -1.5 -2 0

20

40

60

80

Antenna scan angle, deg

Boresight error, mrad

Boresight error, mrad

0

Antenna scan angle, deg

0.5

1E-5

0 -0.5 -1 -1.5 -2 0

20

40

60

80

Antenna scan angle, deg

Fig. 3 BSE as a function of antenna scan angle for an error tolerance of a 1 × 10−1 , b 1 × 10−3 , c 1 × 10−4 , d 1 × 10−5

initial bracketing interval of θ is obtained when the error tolerance is 1 × 10−1 . As the error tolerance is reduced to 1 × 10−3 and 1 × 10−4 , a significant improvement in the BSE prediction is obtained with only minor irregularities. At the error tolerance of 1 × 10−5 and lesser, a very good BSE performance prediction is obtained. A similar result was observed in the computation of PT. Table 2 shows the total computation time required to compute radome performance parameters when the Brent’s method and the GSS method are incorporated as the last step in the 3D ray tracing procedure. Therefore, this computation time includes the time required by the other steps in the 3D ray tracing procedure starting from coordinated transformation and ending at detection of radiation pattern peak. The antenna scan angle is varied from 0◦ to 90◦ in steps of 1◦ , and the processing time is listed for different values of error tolerance. The superiority of the Brent’s method can be clearly observed as it takes less processing time to predict radome parameters as a function of antenna scan angle.

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Table 2 Total time required for computation of radome parameters as a function of antenna scan angle Error tolerance Brent’s method, s GSS, s 28.3668 28.7151 29.3538 29.3644 30.3065 30.9544 31.8731

30.5363 32.4071 34.4534 36.4046 37.8499 39.8722 41.7301

0.5

Boresight error, mrad

Power transmission, dB

1e−3 1e−4 1e−5 1e−6 1e−7 1e−8 1e−9

-0.05 -0.1 -0.15 -0.2

0 -0.5 -1 -1.5 -2

-0.25 0

20

40

60

80

Antenna scan angle, degrees

0

20

40

60

80

Antenna scan angle, degrees

(a)

(b)

Fig. 4 Variation in a PT and b BSE as a function of antenna scan angle

After adopting the Brent’s method in the 3D ray tracing procedure, the optimized wall thickness of the constant thickness tangent ogive radome was computed at the antenna scan angle which causes the maximum angle of incidence at the radome wall. Figure 4a, b shows the PT and BSE performance of the optimized radome, respectively. The insertion loss caused by the radome is less than 0.3 dB, and the magnitude of BSE is less than 2.5 mrad over the entire range of antenna scan angles. Therefore, the radome can be considered to have a satisfactory performance for higspeed airborne applications. The computations reported above were carried out on a personal computer with Intel CORE i5 processor at the approximate clock frequency of 2100 MHz. The same software configuration was used for accurate comparison of the GSS method and Brent’s method for sum pattern peak detection.

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5 Conclusion This paper provides a comparison between the computational efficiency of the Brent’s method and the previously reported golden section search method for the detection of antenna radiation pattern peak. The work establishes the superiority of the Brent’s method in terms of required computation time and function evaluations. The paper also shows that the Brent’s method is suitable to be incorporated in the 3D ray tracing procedure for efficient analysis and design of electrically large antenna radome system for airborne applications.

References 1. Shavit R (2018) Radome electromagnetic theory and design. Wiley Online Library 2. Kozakoff DJ (2010) Analysis of Radome-Enclosed antennas, 2 edn. Artech House 3. Sonalikar HS (2019) Efficient computation of Radome boresight error using iterative method for antenna sum pattern. In: Proceedings of international conference for convergence in technology. IEEE 4. Aparna A, Sonalikar HS (2019) Fast computation of Radome EM parameters with golden section search method for radiation pattern peak detection. In: 2019 IEEE international conference on electronics, computing and communication technologies (CONECCT). IEEE, pp 1–5 5. Nair RU, Jha RM (2009) Electromagnetic performance analysis of a novel monolithic radome for airborne applications. IEEE Trans Antennas Propag 57:3664–3668 6. Xu W, Duan BY, Li P, Hu N, Qiu Y (2014) Multiobjective particle swarm optimization of boresight error and transmission loss for airborne radomes. IEEE Trans Antennas Propag 62(11):5880–5885 7. Press WH, Teukolsky SA, Vetterling WT, Flannery BP (1988) Numerical Recipes in C, 2nd edn. Cambridge University Press

Wideband Bandpass Substrate Integrated Waveguide (SIW) Filter for C Band Application Amrita Dixit, Ashok Kumar, Ashok Kumar, and Arjun Kumar

Abstract This paper presents a substrate integrated waveguide (SIW) filter using electromagnetic band gap (EBG) structure. The periodic EBG structure is etched on the top metal surface of SIW cavity. These periodic structures create a slow wave effect on the filter performance to achieve wide pass band at lower frequency in a small compact size. In the proposed design, Rogers 4350 is used as a dielectric material with the permittivity(εr ) of 3.48 and thickness 1.524 mm. The simulated results obtained by HFSS 19.1 has a broadband from 3.25 to 6.94 GHz with the bandwidth of 3.38 GHz in C band used for satellite communication. The insertion loss is less than 0.5 dB and return loss is better than 18 dB. The size of filter is 48 × 10 mm2 . The fractional bandwidth (FBW) of proposed filter is 68%. Keywords Substrate integrated waveguide (SIW) filter · Wideband bandpass (WB-BPF) filter · Electromagnetic band gap (EBG)

1 Introduction In present time modern communication systems for mobile and satellite applications required a high-performance RF/microwave filters with some important characteristics in terms of weight, cost, insertion loss, quality factor and power handling capability and it is a challenging task to meet all these requirements. SIW technique is growing candidate to fulfil all these requirements in past few decades. SIW technique is a transition of non-planar technology to planar technology. Using this technique, various RF components can be designed such as antenna[1], filter [2], diplexer [3, 4] power divider [5] etc. Microwave bandpass filter (BPF) is the most A. Dixit (B) · A. Kumar · A. Kumar School of Engineering and Applied Sciences, Bennett University, Greater Noida, India A. Kumar e-mail: [email protected] A. Dixit · A. Kumar · A. Kumar · A. Kumar Department of Electronics and Communication Engineering, Government Women Engineering College, Ajmer, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_18

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important component in wireless communication systems because it can control the spectrum of signals and reduce the interference problem. U.S. federal communications commission (FCC) authorized the unlicensed use of ultra-wideband (UWB, from 3.1 to 10.6 GHz) for a variety of applications so wideband band pass filters are highly required in wireless communication systems. To design a wideband bandpass filter using SIW technique can be done with the help of various topologies to achieve required band of frequency. These topologies are electromagnetic band gap structures (EBGs) [6], defected ground structures (DGSs) [7–9], perforated sections (PSs) [10, 11] or different types of resonators including split ring resonators (SRRs) [12–15]. For the band pass behaviour of SIW filter, we can combine its characteristics as a high pass and a stop band characteristic as discontinuity. A bandpass filter with fractional bandwidth greater than 20% is known as wideband bandpass filter. In this paper, the fractional bandwidth is 68%, and it is in C band that can be used for satellite communication. The SIW filter is based on dielectric filled waveguide shown in Fig. 1 with the difference of metal via as the side wall of waveguide to confine the electric field within the waveguide. The cut-off frequency of dielectric filled waveguide is shown in Eq. (1) [17]. c fc = 2π

 mπ 2 a

+

 nπ 2 b

(1)

For the fundamental mode TE10 the frequency is fc =

c 2a

(2)

The width of dielectric filled waveguide is calculated by Eq. (3) ad a=√ εr

Fig. 1 Dielectric filled waveguide with SIW configuration. [16]

(3)

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Fig. 2 Top and side view of proposed SIW filter with its dimension

From the Ref. [18] the SIW width is calculated by Eq. (4) as = a −

d2 0.95s

(4)

In order to reduce, the radiation loss and return loss due to gap between via the following condition are required for SIW filter [18] shown in Eqs. (5) and (6), where d is the diameter of via and s is the pitch, i.e. distance between two via. λg 5

(5)

s ≤ 2d

(6)

d≤

2 Proposed Wideband SIW Filter Configuration As shown in Fig. 2 the SIW transmission line structure is a basic requirement of proposed wide band bandpass filter, which is constructed using a series of metallic via in the side structure with metal top and bottom. SIW structure is a transition of non-planar technology to planar technology. In this paper, SIW with EBG structure have been mixed to provide wideband filter at low frequencies. The present design is in the form of transmission line structure in which EBG structure is created by etching the periodic structure on the top metal layer of substrate. Figure 2 shows the top and side view of proposed design with all the parameters. All dimensions are shown in Table 1. The dielectric substrate used in the designing of filter is Rogers 4350 with the dielectric constant εr = 3.48, a loss tangent tan δ = 0.004 and the thickness is 1.524 mm. The overall dimension of filter is 48 × 10 mm2 .

3 Evolution of Proposed Design In the above design, the loss less transmission line shown as β = ω(LC)1/2 , where L and C are the series inductance and shunt capacitance. In this EBG structure, a slow

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Table 1 All dimensions of proposed SIW WB-BPF S. No.

Parameter

Dimension (mm)

S. No

Parameter

Dimension (mm)

1

lg

48

8

lf

3

2

wg

10

9

wf

1.1

3

hg

0.07

10

l ebg

8.5

4

lp

26.5

11

webg

8

5

wp

10

12

l stub

5.5

6

hs

0.508

13

wstub

1

7

d/s

0.25/0.8

14

Circle-rad

1

wave effect is generated due to increase in series inductance and shunt capacitance. As the evolution of design is started with the simple structure without any horizontal strips so the losses are more. When the horizontal strips are added in the design better response can be achieved because the periodic EBG structure create a simple L-C parallel resonant circuit shown in Fig. 3. The value of capacitance and inductance can be controlled by the number of horizontal thin lines and gap between the lines which can produce a stop band with a certain width. This stop band produces a wide pass band at low frequency. For improving the roll-off rate an open stub is used and finally a circle is etched in the tapered feed section to reduce the losses and increasing the bandwidth. In Fig. 4, the evolution of proposed filter is shown. The result of all designs is compared in Fig. 5 as return loss and Fig. 6 as insertion loss. It is easily identified that in a simple design without strips and open stub losses are very high and roll off rate in also very poor. Fig. 3 L-C parallel resonant circuit

Fig. 4 Evolution of proposed SIW Filter

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0

Fig. 5 Return loss of all designs

S 11 (dB)

-10 -20 -30 S11 for simple design S11with horizontal strips S11with open stub

-40 -50

Fig. 6 Insertion loss of all designs

2

3

4 5 6 7 8 9 10 Frequency (GHz)

0

S 21 (dB)

-10 -20 -30 S21 for Simple design

-40 -50

S21with Horizontal strips S21with Open stub

2

3

4 5 6 7 8 9 Frequency (GHz)

10

4 Result and Discussion A wideband SIW BPF was designed in this paper with the help of periodic EBG structure, the initial specifications are shown in Table 1. Figure 7 shows the final simulation result of proposed design in which the insertion loss is less than 0.5 dB and the return loss is better than 18 dB. The proposed design is simulated using HFSS 19.1. The proposed design has a group delay of 0.46 ns in the passband which is very small as compared to previous published work shown in Fig. 8. The result of proposed design is compared with the published work for the validation in Table 2. It is observed in this table that the fractional bandwidth (FBW) is 68% for the proposed design, whilst in other designs it is maximum 61%. The insertion loss is also very low for proposed design.

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Fig. 7 Return loss and insertion loss of proposed design

S-Parameter (dB)

0 -10 -20 -30 -40

S11- Parameter S21 - Parameter

-50 2

Fig. 8 Group delay of proposed design

3

4 5 6 7 Frequency (GHz)

8

Group Delay (nS)

2.0 1.5 1.0 0.5 0.0 -0.5 Group Delay

-1.0 3

4 5 6 Frequency (GHz)

7

Table 2 Comparison between the designed SIW WB- BPF with other SIW BPF implemented using EBG, DGS and resonators References Type-No. of Design cell method

f O (GHz) FBW (%) IL (dB) RL (dB) Size (λO × λO )

[6]

EBG-11

Periodic structure

12.2

[19]

DGS-4

[20]

61.2

1.3

10

0.52 × 1.24

Band pass 5.88 resonator

5.1

1.38

15

0.42 × 2.05

DGS-2

Band pass 4.9 resonator

9.2

1.1

18

0.39 × 0.39

[21]

DGS-3

Stop band 10 resonator

23

1.2

20

0.22 × 0.45

[11]

PS-5

Band pass 3.65 resonator

17.7

1.31

18

0.21 × 0.98

This Work

EBG-3

Periodic structure

68

0.5

16

0.11 × 0.52

4.95

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5 Conclusion A wideband bandpass substrate integrated waveguide (SIW) filter using periodic electromagnetic band gap (EBG) structure has been designed and simulated in this paper. The proposed design has a broad band extends from 3.25 GHz to 6.94 GHz with the bandwidth of 3.38 GHz and a centre frequency of 5 GHz. The group delay is 0.46 nS in the passband. Due to the mixed feature of EBG and SIW a compact wideband band pass filter is designed with the fractional bandwidth (FBW) of 68%. This passband is coming in C band used for satellite communication.

References 1. Bozzi M, Georgiadis A, Wu K (2011) Review of substrate-integrated waveguide circuits and antennas. IET Microw Antenn Propag 5(8):909–920 2. Deslandes D, Wu K (2003) Single-substrate integration technique of planar circuits and waveguide filters. IEEE Trans Microw Theory Tech 51(2):593–596 3. Zheng SY, Su ZL, Pan YM, Qamar Z, Ho D (2018) New dual-/tri-band bandpass filters and diplexer with large frequency ratio. IEEE Trans Microw Theory Tech 66(6):2978–2992 4. Qu L, Zhang Y, Liu J, Fan Y (2019) Three-state SIW diplexer with independently controllable centre frequencies. Electron Lett 55(9):548–550 5. Moznebi A-R, Afrooz K, Danaeian M, Mousavi P (2019) Four-way filtering power divider using SIW and eighth-mode SIW cavities with ultrawide out-of-band rejection. IEEE Microw Wirel Compon Lett 29(9):586–588 6. Hao ZC, Hong W, Chen JX, Chen XP, Wu K (2005) Compact super-wide bandpass substrate integrated waveguide (SIW) filters. IEEE Trans Microw Theory Tech 53(9):2968–2976 7. Chu H, Shi XQ (2011) Compact ultra-wideband bandpass filter based on SIW and DGS technology with a notch band. J Electromagn Waves Appl 25(4):589–596 8. Liu C, An X (2017) A SIW-DGS wideband bandpass filter with a sharp roll-off at upper stopband. Microw Opt Technol Lett 59(4):789–792 9. Xu S, Ma K, Meng F, Seng Yeo K (2015) Novel defected ground structure and two-side loading scheme for miniaturized dual-band SIW bandpass filter designs. IEEE Microw Wirel Compon Lett 25(4):217–219 10. Coves A, Torregrosa-Penalva G, San-Blas AA, Sánchez-Soriano MA, Martellosio A, Bronchalo E, Bozzi M (2016) A novel band-pass filter based on a periodically drilled SIW structure. Radio Sci 51(4):328–336 11. Silvestri L, Massoni E, Tomassoni C, Coves A, Bozzi M, Perregrini L (2017) Substrate integrated waveguide filters based on a dielectric layer with periodic perforations. IEEE Trans Microw Theory Techn 65(8):2687–2697 12. Feng W, Yao S, Shen J, Cao R (2015) Wideband balun bandpass filter based on substrate integrated waveguide and CSRRs. Progr Electromegn Res 53:115–119 13. Li W, Tang Z, Cao X (2017) Design of a SIW bandpass filter using defected ground structure with CSRRs. Active Passive Electron Compon 14. Zhang X-C, Zhi-Yuan Yu, Jun Xu (2007) Novel band-pass substrate integrated waveguide (SIW) filter based on complementary split ring resonators (CSRRs). Progr Electromagn Res 72:39–46 15. Zhang H, Kang W, Wu W (2018) Miniaturized dual-band differential filter based on CSRRloaded dual-mode SIW cavity. IEEE Microw Wirel Compon Lett 28(10):897–899 16. Noura A, Benaissa M, Abri M, Badaoui H, Vuong TH, Tao J (2019) Miniaturized halfmode SIW band-pass filter design integrating dumbbell DGS cells. Microw Opt Technol Lett 61(6):1473–1477

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17. Montgomery CG, Dicke RH, Purcell EM (1987) Principles of microwave circuits 18. Cassivi Y, Perregrini L, Arcioni P, Bressan M, Wu K, Conciauro G (2002) Dispersion characteristics of substrate integrated rectangular waveguide. IEEE Microw Wirel Compon Lett 12(9):333–335 19. Zhang YL, Hong W, Wu K, Chen JX, Tang HJ (2005) Novel substrate integrated waveguide cavity filter with defected ground structure. IEEE Trans Microw Theor Technol 53(4):1280– 1286 20. Shen W, Yin WY, Sun XW (2011) Compact substrate integrated waveguide (SIW) filter with defected ground structure. IEEE Microw Wirel Compon Lett 21(2):83–85 21. Wang C, Wang Z, Huang YM (2018) Size-miniaturized half-mode substrate integrated waveguide bandpass filter incorporating E-shaped defected ground structure for wideband communication and radar applications. In: International conference of advance communication technology (ICACT), pp 12–16

Design of Polarized 2 × 2 MIMO Antenna Using Partially Stepped Ground Harsha Prabha Paliwal, Navneet Agrawal, and Suruti Gupta

Abstract A 2 × 2 dual-band MIMO antenna is proposed for present wireless applications using partially stepped ground and polarization diversity techniques. The proposed design combines the horizontally and vertically polarized radiating elements. Using partially stepped ground (PSG) and orthogonally placed antenna elements, the effect of mutual coupling between radiators is reduced. The substrate of size 60 × 60 mm2 is mandatory to design the whole configuration. The resonated frequency bands extend from 3.39 to 7.91 GHz and 9.88 to 15 GHz with SWR < 2. The proposed 2 × 2 antenna has adjacent and diagonal ports with isolation 67.62 dB and gain up to 3.52 dB for operating frequencies 5.3 and 11.8 GHz. An envelope correlation coefficient ≤ 0.01 has been achieved in the proposed antenna band. Keywords MIMO · PSG · ECC · Polarization diversity · Isolation

1 Introduction In fast growing wireless technologies large bandwidth, high isolation, reliability, and high-data rate are the major objectives for indoor and outdoor applications of the MIMO system [1]. As per requirements of low cost and planner designs, the variety of antenna structures has been innovated in the technical market. The single port antenna is unable to achieve high-data rate, required bandwidth, spectral efficiency and capacity for the presently available 45, 5G data. Also, co-channel interference degrades the performance of a single antenna system. To overcome the drawbacks of the single input single output (SISO), multiple input multiple output (MIMO) antenna system has been introduced [2]. MIMO technology can improve the performance of wireless communication systems by increasing the channel capacity as well as reducing the adverse effect due to multipath fading. MIMO technology makes use of multipath to increase the throughput of the channel with every pair of antennas added to the system design. Due to strong mutual coupling among the radiating elements, it is difficult to place multiple radiators together within the substrate, while maintaining H. P. Paliwal (B) · N. Agrawal · S. Gupta Department of ECE, College of Technology and Engineering (CTAE), Udaipur, Rajasthan, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_19

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high isolation [3]. 4G/5G technologies also used MIMO as a core because of high gain, efficiency, link reliability, and large bandwidth. One of the major challenges to employ MIMO technology in portable devices in small size with low-mutual coupling. The mutual coupling or correlation between antenna elements in MIMO through surface wave propagation, the common surface current should be minimized either by antenna design or by the introduction of features on the ground plane to inhibit the current flow [4]. Various methods have been introduced to improve the isolation characteristics of the MIMO antenna for a wireless application like diversity techniques, decoupling structures, metamaterial structures, neutralization line structures, etc. An envelope correlation coefficient (ECC) is evaluated for the uncorrelated channels to find whether the channel is good, bad, or good enough. In the real propagation world, even in the line of sight communication, finding a truly uncorrelated channel is impossible. The ECC varies from 0 to 1, and the finding formula is given below [5]: |S ∗11 S12 + S ∗21 S22 |2   ρe =  1 − |S11 |2 − |S21 |2 1 − |S22 |2 − |S22 |2 Most of the described MIMO antenna systems have complex structures or having low impudence bandwidth and isolation. In the proposed dual band, MIMO antenna with polarization diversity achieved high port to port isolation between diagonal elements.

2 Antenna Geometry Design Methodology The proposed MIMO antenna is designed for 5.3 and 11.8 GHz frequencies having dimensions of 60 × 60 mm2 size on the FR4 substrate with 1.524 mm thickness, the loss tangent of 0.002 and dielectric constant 4.4. The inbuilt optimization technique (PSO) of the computer simulation tool (CST) is applied to the design parameters to obtain the best performance parameters. The essential parameters for best goals were obtained after 1500 iterations. The optimized value of considered designing parameters are shown in Table 1. The performance parameters are reported in terms of isolation, gain, VSWR, ECC, and return loss. By using a diamond shaped slot in patch gives high isolation and return loss < 10 dB. Table 1 Optimized values of considered design parameters (mm) Parameters

Pw1

Pw2

Pw3

D1

D2

A

b

f1

C

Values

14

2.8

7.7

8.4

7

3.7

2

6.3

0.8

Parameters

F2

Fw1

Fw2

E

S1

S2

Wg1

Wg2

Values

5.2

3

2.86

0.5

2

6

10

18

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To design the proposed MIMO antenna, first we design the single multi-cut patch and partially stepped ground. The partially stepped ground is used to control the degrading factor of return loss and gain. The stepped ground structure used to reduce and change the distribution of the surface current, consequently mutual coupling is also reduced and it will help to increased isolation. While designing step 1 shows the value of − 29.04 dB at 5.3 GHz frequency. The proposed MIMO antenna resonates at 5.3 GHz, but isolation between the ports is below 10 dB. When we design step 2 the MIMO antenna does not show any changes in the result. But finally after designing the last step we got higher isolation. Then the single antenna is converted into a 2 × 1 MIMO antenna, but it degrades the isolation and radiation efficiency [6]. The 2 × 1 MIMO configuration with a dimension of as shown in Fig. 1a, b front and back of antenna, respectively. By using polarization diversity the 2 × 1 MIMO configuration converted into 2 × 2 MIMO antenna is shown in Fig. 2. The dimension of the polarized antenna is the same as the dimension shown in Fig. 1. Diversity improves the gain parameter of any MIMO system. Polarization diversity improves isolation between radiators is 67.62 dB and gives the proper correlation between radiators.

3 Simulation Results and Discussion The simulated S-parameters or reflection coefficient of the proposed MIMO antenna is shown in Fig. 3. In this MIMO antenna system, all four ports are identical and symmetric to each other. The return loss and isolation of the proposed MIMO antenna design are maintained using orthogonally placed partially stepped ground and radiating elements [7]. We got a return loss of approximately 25.054 and 28 dB with a bandwidth of 4.524 and 5.12 GHz. From the simulation results of s parameters S11 = S22 = S33 = S44, and S12 = S21 = S34 = S43, and S13 = S31 = S24 = S42, and = S14 = S41 = S23 = S32. The radiation efficiency of designed MIMO antenna above 70% which is calculated by CST antenna simulation software. The diversity behavior and spectral efficiency are explained in terms of gain, 2D-3D radiation pattern and antenna efficiency. It explains the far-field property of the MIMO antenna. The achieved moderate gain is 3.59 dB at 5.3 GHz frequency and 3 dB at 11. 5 GHz frequency. The 3D omni radiation pattern of the gain parameter is shown in Fig. 4 The impedance mismatch between radiators is explained by VSWR. It is the function of the reflection coefficient which explains the power imitation factor of the radiating element. High VSWR shows the radiator mismatching, in these conditions more power, is reflected back toward the transmitter side radiator [8]. When VSWR is < 2, it shows the perfect match between radiators. It is a proportion of the peak amplitude and minimum amplitude of a standing wave. The acceptable level of VSWR for most of the wireless applications should not be more than 2.5 dB and it should be 1 dB ideally which is shown in Fig. 5.

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(a) Front view of proposed 2x1 MIMO antenna

(b) Back view of proposed 2x1 MIMO antenna Fig. 1 Schematic view of proposed 2 × 1 MIMO antenna

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Fig. 2 Schematic view of proposed 2 × 2 polarized MIMO antenna

Fig. 3 Simulated S11 and S12 parameters of 2 × 2 designed MIMO antenna

The performance of all the scattering parameters of the proposed MIMO antenna is reported in terms of the envelope correlation coefficient. The lower value of ECC justified the meaning of a lower correlation between antenna elements and vice versa. The gain and ECC show the diverse behavior of designed 2 × 2 circularly polarized MIMO antenna [9]. It shows the decorrelation property of the antenna. The information about coupling at ports is described by only the isolation parameters S12 and S21. Lowering the value of ECC means less correlation between radiating elements, while higher values of it show the negative impact on scattering parameters. The 0 performance of the antenna is good when ECC varies between 0 and 1. The ECC also gives changes in 0 result when we used circular polarization. The value of

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Fig. 4 3D gain of designed MIMO antenna at 5.6 and 11.5 GHz frequency

VSWR1/abs

25 20 15 10 5 0 0

2

4

6

8

10

12

14

16

Frequency/GHz

Fig. 5 VSWR of designed MIMO antenna

ECC must be less than 0.5 for mobile applications. The simulated value of ECC is 0.01 which is shown in Fig. 6. The current traveling from antenna is specially depend on types of antenna. There are many types of antenna resonant antenna, monopole antenna, dipole antenna, array antenna. The microstrip antenna is also one of the antenna structures in which maximum current present at center and zero current present at edge. The current distribution of proposed antenna is showing the flow of current from feedline to patch. In Fig. 7, the current distribution of port 1 at frequency 5.3 GHz and at another frequency 11.8 GHz is shown. The current flow from all port is equal and same as port 1. The comparison of the proposed MIMO antenna with the existing MIMO antennas is given in Table 2. The proposed design contains the best results in comparison with Ref. [5]. In the proposed design, the isolation and bandwidth are high therefore the proposed design suits the modern wireless environment.

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Envelope Correla on Coefficient 0.16 0.14 0.12 0.1 0.08 0.06 0.04 0.02 0 0

2

4

6

8

10

12

14

16

Frequency/GHz

Fig. 6 ECC of designed MIMO antenna

Fig. 7 a Current distribution of proposed MIMO antenna of port 1 at frequency 5.3 GHz and at frequency 11.8 GHz, b current distribution of proposed MIMO antenna of port 1 at frequency 11.8 GHz Table 2 The comparison of proposed and other MIMO antennas References

Size (mm2 )

No. of ports

Frequency band Gain (dB) (GHz)

Isolation (dB)

ECC

[5]

75 × 75

4

4.96–5.5

2.4

− 20

0.03

[10]

93 × 45

4

2.7–3.3

5.83

− 28

0.03

Proposed

60 × 60

4

3.39–7.915 and 9.88–15

3.59

− 67.62

0.01

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4 Conclusion A dual band polarized MIMO antenna with the partially stepped ground was proposed for 5.6 and 11.5 GHz wireless standards using four multi-cut optimized patches. This paper shows polarized antenna parameters for proper frequency range, isolation, compactness, ECC, and gain. The designed MIMO antenna covers 3.39–7.915 and 9.88–15 GHz frequency bands with more than 67dB isolation between radiators. Using S parameters and far-field patterns the result is verified. The ECC, of the antenna is within the acceptable threshold limits, is 0.01 which is demanded to upgrade the data rate through spatial multiplexing in wireless communication channels.

References 1. Malviya L, Panigrahi RK, Kartikeyan MV (2018) Offset planar MIMO antenna for omnidirectional radiation patterns. Int J Microw Comput-Aided Eng:1–9 2. Sharawi MS, Numan AB, Aloi DN (2013) Isolation improvement in a dual band element MIMO antenna system using capacitively loaded loops (clls). Prog Electromagn Res 134:247–266 3. Agusta RP, Wijanto H, Syihabuddin B, Prasetyo AD (2019) The 4×4 hybrid L-slotted rectangular microstrip antenna for dual band WiFi communication. IOP Conf Ser Mater Sci Eng 620:1–8 4. Zhai G, Chen ZN, Qing X (2015) Enhanced isolation of a closely spaced four-element MIMO antenna system using metamaterial mushroom. IEEE Trans Antennas Propag 63:3362–3370 5. Malviya L, Panigrahi RK, Kartikeyan MV (2016) A 2×2 dual-band MIMO antenna with polarization diversity for wireless applications. Progr Electromagn Res C 61:91–103 6. Nadeem I, Choi DY (2018) Study on mutual coupling reduction technique for MIMO antennas. IEEE, pp 2169–3536 7. Konkyana FALB, Sudhakar BA (2019) A review on microstrip antennas with defected ground structure techniques for ultra-wideband applications. In: International conference on communication and signal processing (ICCSP), Chennai, India, pp 0930–0934 8. Ghimire J, Choi KW, Choi DY (2019) Bandwidth enhancement and mutual coupling reduction using a notch and a parasitic structure in a UWB-MIMO Antenna. Hindawi Int J Antenn Propag:1–9 9. Umayah E, Srivastava V (2019) Comparative view of return loss, VSWR, gain, and efficiency of cylindrical surrounding patch antenna with frequency shift. Int J Electr Electron Eng Telecommun:352–357 10. Paliwal HP, Agrawal N, Joshi S (2017) Dual band UWB pentagonal antenna with partial ground. J Emerg Technol Innov Res 4:25–31 11. Malviya L, Chouhan S (2019) Multicut four port shared radiator with stepped ground and diversity effect for WLAN application. Int J Microw Wirel Technol:1–9 12. Agrawal T, Srivastava S (2017) High gain microstrip MIMO antenna for wireless applications. Int J Microw Wirel Technol 12:74–81 13. Sharawi M, Hassan A, Khan MU (2017) Correlation coefficient calculations for MIMO antenna systems: a comparative study. Int J Microw Wirel Technol 9:1–14 14. Agrawal N, Paliwal HP, Fegade M (2019) A 2 × 1 dual-band MIMO antenna using complementary split ring resonator for wireless applications. In: 4th international conference on communication systems BKBIET Campus, Pilani, Rajasthan, India. ISBN: 978-93-89107-48-7

Simulation Studies on Force Sensor Using PDMS Coated Fiber Bragg Grating for Robot-Assisted Surgery Dinesh Lakshmanan

and Srijith Kanakambaran

Abstract We investigate the performance of a cantilever-type force sensor based on PDMS-coated fiber Bragg gratings for robot-assisted surgical applications. The cross section area of PDMS matrix and the location of the FBG sensor inside it are optimized through simulations. The sensitivity of the proposed sensor is obtained to be around 0.5 mN. Keywords Force sensor · Fiber Bragg grating · PDMS

1 Introduction Brain tumor is one of the leading reasons for cancer-related deaths. MRI guidance is performed on such patients in order to locate the residual portion of the tumor in the brain. The use of a robot in this procedure to assist neurosurgeon is being preferred in order to facilitate the complete removal of tumor in the brain [1]. However, this procedure demands the nature of the probe in the robotic arm to be biocompatible and immune to MRI. The conventional piezoelectric-based force sensors are highly susceptible to electromagnetic interference (EMI). Fiber Bragg grating (FBG)-based optical sensor is an attractive alternative due to its immunity to EMI owing to its dielectric property [2]. In addition, the other advantages offered by FBG are lightweight, small size, and ease of multiplexing. In robot-assisted surgical applications, the FBG sensor must be encapsulated in a biocompatible material which is inert to human skin. One possible solution for this is polydimethylsiloxane (PDMS) which is a silicon-based organic polymer. This material has excellent temperature stability and is chemically inert and biocompatible. PDMS coating has been shown to enhance temperature sensitivity of the FBG sensor [3]. The use of such sensors to measure the vertical compression loads and D. Lakshmanan · S. Kanakambaran (B) Department of Electronics and communication Engineering, Indian Institute of Information Technology, Design and Manufacturing Kancheepuram, Chennai 600127, India e-mail: [email protected] © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_20

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Fig. 1 3 × 3 PDMS structure with fiber

transverse loads have also been reported [4, 5]. PDMS-coated FBG sensors have been demonstrated to monitor vital signs in the human body as well [6]. In this paper, we propose a cantilever type force sensor based on PDMS coated FBG. The optimum dimension of the coating and the location of the FBG sensor are investigated.

2 Simulation Setup The simulation studies were performed using ANSYS software. The optical fiber with 125 µm diameter is made with silica glass and is enclosed in a PDMS matrix (Fig. 1). The properties of the materials are listed in Table 1. Three different cross sections of the PDMS matrix are considered − 3 mm × 3 mm, 5 mm × 5 mm, and 10 mm × 10 mm. The 3D model is created in ANSYS Space claim. One of the lateral sides is assigned as fixed support boundary condition. Force is then applied on the top surface of the structure with values varying from 0.005 to 0.1 N. After assigning the necessary boundary conditions, the longitudinal strain experienced by the fiber is monitored at various distances from the free end. The reflectivity curve of FBG given by Eq. 1 is simulated using MATLAB [7]. R(l, λ) =

κ 2 sinh2 (sl) β 2 sinh2 (sl) + s 2 cosh2 (sl)

(1)

where R(l, λ) is the reflectivity, which is a function of the grating length l and wavelength λ. s is given by s 2 = κ 2 − β 2 , and κ denotes the coupling coefficient. The de-tuning wave vector is given by Eq. 2, β = β −

π 

(2)

Force Sensor Using PDMS Coated FBG for Robot-Assisted Surgery Table 1 Elastic properties of materials Elastic parameters PDMS Young’s modulus Poisson ratio Bulk modulus Shear modulus Density

8.7e5 Pa 0.499 1.45e8 Pa 2.9e5 Pa. 0.97 kg/m3

191

Silica glass 7.3e10 Pa 0.155 3.53e10 Pa 3.16e10 Pa 2190 kg/m3

where β is given by β = 2π n 0 /λ and  is the grating period of the FBG. Neglecting the temperature effects, the shift in Bragg wavelength due to strain may be expressed as n2 (3) λ = λ B [1 − [ p12 − ν( p12 + p11 )]] z 2 where p11 and p12 are components of the strain optic tensor, ν is the Poisson’s ratio, z = δl/l [7]. A typical FBG exhibits 1.2 pm shift in the Bragg wavelength for a change of 1 μ strain applied on the grating. The force applied on the robotic arm results in strain changes on the instrument which results in shift in the Bragg wavelength according to Eq. 3. For simulations, the average refractive index n 0 was chosen as 1.448, the grating period was chosen as 534 nm and the length of the grating as 10 mm. The shift in Bragg wavelength corresponding to the different strain is then incorporated in Eq. 1 to simulate the reflection spectra under different strain conditions.

3 Results and Discussion The longitudinal strain observed at different locations in the fiber for structures of various cross section of PDMS matrix is shown in Figs. 2a, 3a. It may be observed that the largest strain response is obtained in the 3 mm × 3 mm case. As expected, as the cross section area of the PDMS matrix increases, the strain that gets transferred to the fiber will be lesser. This is why the strain response of 10 mm × 10 mm structure shown in Fig. 3a appears to the flat and almost four to five orders lesser in magnitude. Another interesting thing to note is that the strain response increases as the point of observation on the fiber moves away from the free end of the structure. This may be explained as follows. At the free end of the structure, the deformation of the PDMS will be maximum and the structure will bend. The bending will also translate to the fiber. However, since the fixed support is farther away, there is not much resistance offered to bending of the fiber. As a result, the longitudinal strain will not be high. On the other hand, at a location much closer to the fixed end, the fiber will be now constrained to bend and as a result would yield a larger longitudinal strain. This phenomena is observed in both Fig. 2a, b.

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)

100

0.005N 0.01N 0.03N 0.05N 0.1N

80

Strain(

Strain(

)

150

60

40

50 20

0

0 1

2

3

4

5

1

Distance from the free end(mm)

2

3

4

5

Distance from the free end(mm)

(a)

(b)

Fig. 2 Longitudinal strain for various forces at different positions on the fiber in PDMS matrix of cross section a 3 mm × 3 mm, b 5 mm × 5 mm 1000

150 0.005N 0.01N 0.03N 0.05N 0.1N

)

600

3x3 5x5 10x10 100

Strain(

Strain(p )

800

400

50 200

0

0 1

2

3

4

Distance from the free end(mm)

(a)

5

1

2

3

4

5

Distance from the free end(mm)

(b)

Fig. 3 a Longitudinal strain for various forces at different positions on the fiber in 10 mm × 10 mm PDMS matrix. b Longitudinal strain for various structures at a constant applied force of 0.1 N

Figure 3b shows the response of different cross section of PDMS to a constant force of 0.1 N. As explained previously, the 3 mm × 3 mm structure gives the highest value of strain for a given force. Hence, this is selected to be optimum cross section for further simulations. The shift in Bragg wavelength is calculated for various forces for the optimized 3 mm × 3 mm structure. A strain sensitivity of 1.2 pm/μ is considered. From the simulated values of strain, the shift in the Bragg wavelength is estimated. The reflection spectrum for various applied forces is shown in Fig. 4a. The sensitivity of the proposed structure is evaluated through simulations. The optimized structure of 3 mm × 3 mm is considered. The strain is monitored at 5 mm from the free end for various applied forces. The results are shown in Fig. 4b. Considering the use of a spectrum analyzer with a resolution of 1 pm, the force sensitivity is around 0.54 mN.

Force Sensor Using PDMS Coated FBG for Robot-Assisted Surgery 1

1.2

0.6

1.15 1.1

(pm)

0N 0.005N 0.01N 0.03N 0.05N 0.1N

0.8

Reflectivity

193

1.05 1

0.4

0.95 0.2

0 1549

0.9

1549.5

1550

1550.5

1551

Wavelength(nm)

(a)

1551.5

1552

0.85 4.5

5

5.5

Force(N)

6

6.5 10-4

(b)

Fig. 4 a Reflection spectra of FBG at various applied forces in 3 mm × 3 mm structure. b Sensitivity of the proposed sensor

4 Conclusion The performance of cantilever-type PDMS coated FBG sensor for force-sensing applications is investigated through simulations. A cross-sectional area of 3 mm × 3 mm of the PDMS matrix is found to be optimum for getting the highest strain response. The FBG sensor needs to be placed preferably closer to the fixed end to enhance the sensitivity. Future work includes analyzing the force feedback provided by the human tissues on the robotic arm. Experimental studies are underway to confirm the simulation results.

References 1. Rahman N, Deaton NJ, Sheng J, Cheng SS, Desai JP (2019) Modular FBG Bending Sensor for Continuum Neurosurgical Robot. IEEE Robotics and Automation Letters 4(2):1424–1430 2. Saccomandi P, Caponero MA, Polimadei A, Francomano M, Formica D, Accoto D, Schena E (2014) An MR-compatible force sensor based on FBG technology for biomedical application. In: 2014 36th annual international conference of the IEEE engineering in medicine and biology society, EMBC 2014. Institute of Electrical and Electronics Engineers Inc, pp. 5731–5734. https://doi.org/10.1109/EMBC.2014.6944929 3. Park C, sub, Joo, K. I., Kang, S. W., & Kim, H. R. (2011) A PDMS-coated optical fiber Bragg grating sensor for enhancing temperature sensitivity. J Opt Soc Korea 15(4):329–334. https:// doi.org/10.3807/JOSK.2011.15.4.329 4. Niu L, Chan C, Wah K, Chen L (2015) Bent optical fiber Bragg grating embedded in pdms for vertical compression load sensor. In: Workshop on specialty optical fibers and their applications, OSA technical digest (online). Optical Society of America, 2015, paper WT4A.38 5. Yang R, Yu YS, Zhu CC, Xue Y, Chen C, Zhang XY, Sun HB (2015) PDMS-coated S-tapered fiber for highly sensitive measurements of transverse load and temperature. IEEE Sensors Journal 15(6):3429–3435. https://doi.org/10.1109/JSEN.2015.2388490

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6. Nedoma J, Fajkus M, Cubik J, Kepak S, Martinek R, Vanus J, Jaros R (2018) SMART medical polydimethylsiloxane for monitoring vital signs of the human body. In: 2018 IEEE 20th international conference on e-health networking, applications and services, Healthcom 2018. Institute of Electrical and Electronics Engineers Inc. https://doi.org/10.1109/HealthCom.2018.8531190 7. Onoufriou A, Kalli K, Pureur D, Mugnier A (2006) Fibre Bragg gratings. https://doi.org/10. 1007/3-540-31770-86

Design and Analysis of E Shaped Microstrip Patch Antenna with Defected Ground Structure for Improvement of Gain and Bandwidth Ajay Singh, Sunil Joshi, Dhananjay Dashora, Lokesh Lohar, and Harsha Prabha Paliwal

Abstract A design of an E shaped microstrip patch antenna (MSA) with defected ground structure is presented in this paper. The dimension of this antenna is 30.33 × 25.34 mm2 . The proposed antenna uses Fr-4 substrate material with dielectric constant 4.4 and substrate height 1.6 mm. The defected ground structure (DGS) performance is used to improve relevant antenna parameters. The performance parameters of the designed antenna such as return loss, voltage standing wave ratio, gain, directivity, radiation pattern, and impedance are simulated using ANSoft HFSS. This antenna has resonated at frequencies 5.6 GHz with return loss 30.06 dB, impedance bandwidth 1.39 GHz, and maximum achieved gain is 9.94 dB. The proposed antenna is used for WLAN, RADAR, and Mobile phones applications etc. Keywords E-shape · DGS · Gain · Bandwidth and FR-4

1 Introduction There are several features of microstrip antenna like low profile, low cost, portability, and easy to fabrication. The MSA consists of radiating patch on one side of a dielectric substrate, which has a ground on the other side. The measure limitation of conventional patch antenna has narrow bandwidth and low gain. The development of MSA is the aim to provide to future communication like high data transfer rate with high reliability at anytime and anywhere around the universe. The miniaturization of the antenna has with improved bandwidth, and gain is very important issue. Many researchers are developing new MSA designs to improve the bandwidth and gain of patch antenna using different methods designed such as incorporating of high permittivity, by proximity feed, multilayer, using array antenna, and materials. [1–3]. In [4], multi-layered dielectric substrate with stripping of partial substrate has been designed for gain and bandwidth improvement. In [5], E shaped MSA for ISM band has been optimized with 36% bandwidth, and maximum gain has achieved

A. Singh (B) · S. Joshi · D. Dashora · L. Lohar · H. P. Paliwal ECE Department, College of Technology and Engineering (MPUAT), Udaipur, Rajasthan, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_21

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8.95 dB. In [6], wideband E shaped patch antenna using shorting pin with reduced sized has proposed. This paper presents an analysis of E shaped microstrip patch antenna with defected ground structure. The DGS is used for improving the bandwidth and gain as well directivity. The antenna is excited with 50  inset feed line. The operating frequency range of this designed antenna is 4.67–6.06 GHz with impedance bandwidth 1.39 GHz is achieved.

2 Design Procedure The E shaped microstrip patch antenna, having dimension 30.33 × 25.34 mm2 with thickness of substrate 1.6 mm, has been designed. Figure 1a, b shows the front view and back view geometry of the antenna. This antenna has been designed to operate for C band application. The length and width of patch are calculated from Eqs. (1) and (2) with operating frequency 5.6 GHz. So, this antenna is suitable with this operating frequency according to antenna dimensions equation. The width and length of ground surface are calculated with the help of the equations shown below [7]. The size of patch is obtained by equations shown below.

εf f =

Wp =

c √ 2 f r (εr + 1)/2

(1)

Lp =

c − 2l √ 2 f r εf f

(2)

εr + 1 εr − 1 + {1 + 12h/W p}0.5 2 2

(3)

      Wp Wp l = 0.412 (ε f f + 0.3) + 0.264 /(ε f f − 0.258) + 0.8 1 h h

(4)

where c is speed of light, fr is operating frequency, l is extension length, and εff is the effective dielectric constant. The width and length of ground dimension are obtained by equation shown below. W g = 6h + W p

(5)

Lg = 6h + L p

(6)

where h is the thickness of substrate, Wp is width of the patch, and Lp is length of the patch.

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Fig. 1 a Front view of proposed antenna. b Back view of proposed antenna

Table 1 shows the detailed description of this antenna geometry. The patch layer consists of two arms in an E shaped. The left side arms width is 3.8 mm, and right-side arm width is 4.5 mm. With this different width value of arms, return loss of the antenna increases. The width of center arm of the patch is 9 mm has been taken to this proposed antenna. The fr-4 dielectric material has been used as substrate materials with permittivity of 4.4. The 30.33 × 4.5 mm2 dimension defected ground plane has been incorporated in this designed antenna for improving gain and bandwidth of antenna. The 50  inset feed line dimension 8 × 3 mm2 is used in this patch antenna. Table 1 Dimensions of proposed antenna parameters

Parameters

Dimension (mm)

Substrate width W

30.33

Substrate length L

25.34

Substrate height h

1.6

Ground width Wg

30.33

Ground length Lg

4.5

Width of patch Wp

20.73

Length of patch Lp

15.74

Length patch slot Sp

12

Width of patch Wp1

9

Width of left arm S1

3.8

Width of right arm S2

4.5

Feed length FL

8

Feed width Fw

3

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3 Results and Discussion The proposed E shaped model is designed and simulated using HFSS for C band application successfully. The detailed discussion of the antenna characteristics is discussed in this section. For practical application, the vswr should be ≤ 2, and this corresponds to return loss of −10 dB is acceptable. Table 2 shows the various parameter obtained by the simulation software ANSoft HFSS.s The proposed antenna has resonated at 5.6 GHz having return loss −30.06 dB which is applicable for C-Bband application is shown in Fig. 2. The resonated frequency bands extend from 4.67 to 6.06 GHz having bandwidth 1.39 GHz bandwidth with SWR < 2 by using two rectangular slots of different sizes on patch which converts the patch into E-shape. This kind of patch geometry gives the minimum return loss. It also improves the radiations efficiency of designed antenna which is up to 85%. VSWR is a measure how much power is delivered to an antenna. This does not mean that antenna radiates all power it receives. VSWR measures the potentials power of radiators. Figure 3 shows that the variation of frequency versus voltage standing wave ratio. VSWR ≤ 2 is maintained at operating frequency range from 4.67 to 6.06 GHz. Table 2 Simulated resultant antenna parameters Resonating frequency

Return loss (dB)

VSWR

GAIN (dB)

Bandwidth

Directivity

5.6 GHz

−30.06

1.06

9.94

(4.67–6.06 GHz) 1.39 GHz

10.03

Fig. 2 Return loss versus frequency

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Fig. 3 VSWR versus frequency

Figure 4 shows that the variation of frequency versus gain. For appropriate designed antenna, gain should be minimum 6 dB. From Fig. 4, it clearly indicates the gain of this antenna is 9.94 dB achieved at operating frequency 5.6 GHz. The gain is varying from 1.62 to 10.32 dB at resonant 4.67 to 6.06 GHz. In this antenna geometry, we use defected ground and slit slot in feedline for best impedance matching and improve the gain and radiation pattern of the designed antenna. Figure 5 shows that the variation of directivity versus frequency. The directivity of this designed antenna is achieved 1.91–10.56 dB from resonating frequency 4.67– 6.06 GHz. Figure 6 shows that impedance plot for the feed line. It is clearly shown in this figure. 50  impedance matched for operating frequency range. Figure 7a–c shows the radiation pattern of E field and H field for 4.7, 5.6, and 6 GHz frequency. These antenna radiation patterns are almost radiating to the particular direction. So, this antenna radiation pattern is in directional radiation pattern.

Fig. 4 Gain versus frequency

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Fig. 5 Directivity versus frequency

Fig. 6 Impedance versus frequency

4 Conclusions The E shaped microstrip patch antenna using defected ground structure with improved gain and bandwidth has been successfully designed and simulated using HFSS software. The impedance bandwidth and gain have achieved 139 and 9.94 dB with directivity 10.03 dB at operating frequency 5.6 GHz for Fr-4 substrate material. The 50- impedance has maintained at resonating frequency between 4.67 and 6.06 GHz, which shows maximum power transmission between sources to load. All the observed parameter satisfied for this designed antenna. This antenna is suitable for satellite communications, phone, radar system, and commercial wireless LAN. Due to demand of higher data transfer for future communication with high reliability, this antenna meets all requirements.

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Fig. 7 a E field (φ = 0) and H field (φ = 90) for frequency 4.7 GHz. b E field (φ = 0) and H field (φ = 90) for frequency 5.6 GHz. c E field (φ = 0) and H field (φ = 90) for frequency 6 GHz

References 1. Pozar DM, Kaufan B (1987) Increasing the bandwidth of microstrip antenna for coupling. Electron Lett 23:368–369 2. Luke KN, Mak Chow YL, Lee KF (1998) Broadband microstrip patch antenna. Electron Lett 1442–2143 (1998) 3. Yang HY, Alesotoulous NG (1987) Gain enhancement methods for print antenna through multiple superstrates. IEEE Trans Anten Propag 35(7):862–863 4. Rao N, Kumar DV (2016) Gain and bandwidth enhancement of microstrip patch antenna using partial substrate removal in multiple layered dielectric substrate. In: PIER Symposium proceeding, pp 285–289 (2016) 5. Sahil AM, Ali K (2016) Design and optimization of E shaped variable antenna in ISM band. In: International Workshop on Computational Intelligence (2016) 6. Padra ACC, Bullah G, Serafini P, de Salles AAA (2009) Shorting application in wideband E shaped patch antenna. In: International microwave and optoelectronics conference, SBMO/IEEE, pp 229–234 (2009)

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7. Singh A, Singh GP, Manvendra PPC, Kumar M, Saxena R (2017) Analysis of V slot multiband microstrip patch antenna for S, C and X Bands. Int J Eng Trends Technol (IJETT) V48(6):331– 334. ISSN:2231-5381 8. Paliwal HP, Agrawal N, Joshi S (2017) Dual band UWB pentagonal antenna with partial ground. J Emerg Technol Innov Res 4:25–31

Microwave Imaging Breast Cancer Detection Techniques: A Brief Review Monika Mathur, D. Mathur, G. Singh, S. K. Bhatnagar, Harshal Nigam, and Mukesh Arora

Abstract The medical field experiencing a rapid development in the implantable medical devices for the convenience of the patients. The implants in the human bodies are an incorporation of many new technologies, i.e., wireless, nano/micro, IOT and sensing technologies. These are the advanced techniques for human body health monitoring, vital life gathering auxiliary and diseases detecting and supervisory. Further, breast cancer is a serious disease, and it is notified that it is the largest prominent reason of cancer death for women in the world. Early cure of this disease save the 80% life of women. Many techniques are available in the medical platform to detect as well as to control this cancer. In latest, RF implantable/wearable antennas are found popular for this purpose. This paper presents a review of the many microwave imaging approaches available for detection of the breast cancer. It includes the review of different implantable/wearable antennas available for the clinical measurement of the breast cancer detection. The different aspects to be taken care for the implantable are the addition of this review paper. Keywords Implantable medical devices · Wearable antennas · Health monitoring · Breast cancer detection · Cancer death · Microwave imaging

1 Introduction RF and biosensors play a major role for designing future gadgets that can monitor various physiological data inside the human body; this can help in diagnosis of virus [1], prevention of body and improving the patient’s life quality and reducing the chances of patient being hospitalized. The various monitoring devices used inside the body may include temperature monitors [2], blood glucose monitors [3], deep M. Mathur (B) · S. K. Bhatnagar · H. Nigam · M. Arora Electronics and Communication Department, SKIT, Jaipur, India D. Mathur Electronics and Communication Department, RTU, Kota, India G. Singh Electronics and Communication Department, MNIT, Jaipur, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_22

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brain simulators [4], breast cancer detectors [5], blood pressure monitors [6] and many more. The sensors inside the body need to link with the external devices for data monitoring and recording, so these communication links need to be established. The link can be a wired link where the sensors and monitors are attached with the body, but if a longer range of communication is required a wireless link is preferred. The wireless link can be inductive or RF link, inductive links were the most prevalent for implanted medical devices [7, 8], but they have the disadvantage of limited range and less sensitivity, so RF based links are highly required for future biotelemetry communications. The establishment of RF links requires the need of antennas but the task of designing antenna for these kinds of applications is a highly challenging task. The various challenges for implantable antennas may include small size, compatibility with body tissues, high bandwidth, and high gain with low-effective radiated power [9]. The wearable antennas suffer challenges such as power efficiency as they are battery operated devices, radiation loss due to movement of human body and some tissues beneath the device, physical obstructions, and interference from other wireless devices, size factor, heat indulgence and radiation from EM wave [10]. Further, important thing for designing implantable is the operating frequency for antennas. MICS band (Medical implant communication services) (402–405 MHz) is allocated for medical applications as per the recommendations from ITU-R SA, 1346, but with an EIRP limit of −16 dBm [11], also ISM band (Industrial, Scientific and Medical band) it includes 433–434.8 MHz band, 902–928 MHz band, 2.4– 2.5 GHz band and 5.725–5.875 GHz band. These are the widely used bands for medical applications [12, 13]. Breast cancer is a major disease mostly finding in the women. Data suggests [14] that the projection of cancer in to the breast will touch the number 1,797,900 till the year 2020. And from this number around 41,070 will be suspected to convert into the death. Medical science doing many affords to reduce the numbers of death. Pre cure is also need of this disease. To find the solution of any problem, prior knowledge of that area is essential. This paper covers the whole area and important things related to the breast cancer detection. This is partitioned into three portions. First part presents a review about all the techniques available for detecting the tumor in breast. Second part extant the RF antenna techniques reported for detection of tumor in the breast part and in the last section different measurement methods from the RF antenna is discussed. To detect the tumor in the body many imaging techniques are available, e.g., MRI, mammography, CT-Scan, etc. But for the particularly detection of breast cancer, mammography, is most popular method. Different techniques having its own benefits and hitches. Microwave imaging techniques are here discussed in detail the Part I of this paper. In this platform, the implantable or wearable devices can be categorized into two types: Diagnostic and therapeutic. The diagnostic type devices only able to detect the diseases and therapeutic devices are for the controlling the diseases. The detection from RF antenna and implantable sensory devices are nowadays being popular diagnostic method. Many antenna structures are reported in

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last years for this purpose. Section-II includes different techniques that are used for detecting the breast cancer. Section-III is the summary and comparison about some antenna reported and also a brief on what is in the future perspectives for this. Prior to implant the device/antenna in the human body some essential issues should be taken care. That is discussed in the last section. Breast tumor from the implantable /wearable devices is mainly detected from: (a)

Change in the permittivity of the human tissues. Usually when the breast tissues are healthy, their permittivity is 36 and the permittivity of tumor is 50. Hence the resultant permittivity increases when tumor detected in the breast. So microwave energy scattered into the body is higher in the presence of tumor as compared to healthy body. The effective dielectric constant (permittivity) is defined as ε∗ = ε −

(b)

(1)

Here ε* is the effective permittivity, ε is the permittivity of the tissue, and is the σ conductivity of the test area ε0 = 1.85 × 10–12 F/m. Change in the temperature of the human body. The temperature in the body increases when the tumor induced in the body. The power and temperature are defined as the given equation: T =

(c)

jσ ωε0

P K Xf

(2)

So change in received power will be measured and able to detect the existence of the tumor. Change in the value of the conductivity of the human body. The conductivity increases in the presence of the tumor.

2 Microwave Imaging Techniques to Detect the Breast Tumor There are many methods for detecting the breast tumor. Microwave imaging techniques [15] are popular now days. A brief review of the microwave imaging methods is here presented below:

2.1 Microwave Imaging Microwave imaging is a technique in which the microwave range energy/power is transmitted via antenna/cavity, to the body area under test and reflected energy

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is received via antenna and cavity. The difference between both the energies will define the presence of any disease. There are two types of microwave imaging system: Passive and active. Passive system is also called microwave radiometry. In this type of imaging system, the electromagnetic field is measured which is emitted by the warm body. In other word in passive imaging system the electromagnetic field will vary with the variation in temperature. This is the indication of the presence of the tumor. Due to the tumor, temperature is increases [16, 17], because the tumor affected cells are metabolically dynamic and produces extra heat. This is called passive imaging because the electromagnetic field is affected by the change in body’s temperature. Active imaging may be classified in to three parts: (a) Microscopy, (b) Tomography, (c) Hybrid Method. Further, the tomography is classified time domain system, frequency domain system and combined system. Also the hybrid system is classified as microwave thermal acoustic system and ultrasound guided microwave imaging. A microscopy technique is developed by the near field wave and tissue interface method. In this method, microwave cavity resonator is applied as open ended transmitter to the targeted position under the skin and the change in the frequency of resonance of this open ended resonator will indicate the presence of tumor. This is calculated by the relation of partial change in resonant frequency and change in the dielectric constant [18].  2    − ε  E 0  dv V0 ω − ω0  =    2  ,  ω0   2  dv V0 ε  E 0  + μ H0 

(3)

Here ω is the original frequency and ω0 is the cavity resonator frequency, E 0 , H 0 are the fields of cavity, and ν is the volume of cavity (3). In frequency domain tomography inverse scattering techniques is used. In which the received electromagnetic fields are measured at different locations and difference of incident field and scatter field is the indication of presence of tumor, if there is a change in the value of permittivity. And in time domain tomography radar principle is used. In this time delay between the radiated and received signal is the indication of the signal loss. Many array elements are placed as receiver and they make 2D or 3D images. In combined technique of frequency and time domain allocate to recreate the high-resolution images and gives the overall complex permittivity of the human body. It is also a relatively low-cost method. This method is introduced for overcoming the disadvantages of both the techniques. In this technique as time domain method RADAR principle is used for find out the scatters spatial location and dimensions and frequency domain method is used for recreate the dielectric properties of the target. In hybrid methods for induced-thermal acoustic methods acoustic sensors are used. In this method, microwave power is first applied to the breast this absorbed power is proportional to electric conductivity by the relation:

Microwave Imaging Breast Cancer Detection …

Pv =

1   2 W σ E  3 2 m

207

(4)

Here E is the electric field intensity of human tissue. The absorption of the power will stimulate the tissues and it induces the thermo acoustic waves which are sensed by the acoustic sensor. The latest technique is microwave imaging guided by ultrasound, in which image reform is shown by the process ultrasonography. The process ultrasound is used to take the information about the breast structure and numerical analysis is done by the electromagnetic analysis.

3 Summary of Reported RF Antennas for the Detection of Breast Cancer Implanted antennas are designed on different materials. This is kept between substrate and superstrate material as shown in Fig. 1. The antenna bandwidth is enhanced by the loading of the superstrate. It decreases the resonant frequency and consequently the resonant resistance, and enhances the impedance bandwidth of the antenna. The designed antennas are of various shapes (Fig. 2a–d). The whole antenna is wrapped by some biocompatible material with a thickness of 0.1 mm. Comparison of different structure is shown in Table 1. In future dual band operation of the antennas is needed, for the long life of the battery. Battery less devices are also the future aspects of the implantable. Circularly polarized, differentially fed antennas will also make the devices for available for more testing area. Capsule antennas are the latest for the research in this area of work. Fig. 1 In body placement of Antenna

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Fig. 2 Different Structures for implantable antennas. a [3], b [19], c, [9], d [20]

4 Needs to Be Taken Care of Implantable in Human Body (a)

(b)

(c)

Specific Absorption rate (SAR) value should be taken as per standard for the safety of patient. The IEEE C95.1-1999 suggest the SAR 1.6 W/kg for any 1 g of tissue and IEEE C95.1-2005 standard suggest 2 W/kg over 10 g of tissues. Miniaturization of the devices is the basic requirement for the implantable. The miniaturization could be done by using high-dielectric material for designing the antenna, by changing the current track of the patch, by using high-operating frequency, by introducing the inductive loading for impedance matching. The temperature in the body due to the power absorbed in the body should not increase more than 1–2 °C across the implantable device.

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Table 1 Comparison table of the different reported antennas [3]

[19]

[9]

[20]

Size of antenna

22.5 × 22.5 × 1.27 mm3

27 × 14 × 1.27 mm3

13.2 × 15.8 × 0.635 mm3

10 × 10 × 2.54 mm3

Type

Dual band

Dual band

Dual band

Triple band

Operating frequency band

Dual band MICS (402–405 MHz) and ISM band (2.4–2.48 GHz)

433.9 MHz and 542.4 center frequency in MICS band

402–405 MHz MICS and 2.4–2.5 GHz ISM band

MICS band at 402 MHz, ISM at 433 MHz and 2.4 GHz

Bandwidth

20.4% MICS and 4.2% ISM band

31 MHz for lower band and 29 MHz for upper band

96 MHz for lower band and 530 MHz for upper band

86 and 113 MHz in MICS and 60 and 70 MHz ISM band

Implementing technique

Using skin mimicing gel, antenna is placed between skin and fat layers

Antenna Antenna placed implemented on beneath the three layer tissue surface of skin model

Implemented on a test tissue which reduced the steps required to prepare a skin mimicing gel and saved time

(d)

(e)

The biocompatibility for long term is essential for implantable devices. For designing the antenna biocompatible materials, e.g., Teflon; Ceramic alumina etc. may be used. The encasing of the antenna/device should be into the lowloss biocompatible material otherwise a coating on the outer case may be done for long term operation of device. As the implantable device will work as the receiver/transmitter, in order to make the wireless communication possible from the external device (measurement device), S/N0 ratio of the implantable device should be more than the required S/N0 for the communication.

5 Conclusion In this paper, a brief review of microwave imaging techniques for the breast cancer detection has been presented. Next, briefly explained, about the three parameters, which are responsible to quote the presence of the tumor. Some important designs of antenna for the breast cancer have been discussed and compared. Future aspects for designing the antenna for this purpose have also defined. At last section discussed some safety issues that should be considered for designing the implantable. Acknowledgements This work is the part of the CRS project grant funded by TEQUIP-III, RTU, Kota. The authors are thankful to Swami Keshvanand Institute of Technology, Management and Gramothan, Jaipur, RTU, Kota and Malviya National Institute Jaipur for their support for carry out the research.

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References 1. Loktongbam P, Solanki LS (2017) A brief review on implantable antennas for biomedical application. Int Adv Res Sci Eng 6(5) 2. Scanlon WG, Evans NE, Mc. Creesh ZM (1997) RF performance of a 418 MHz radio telemeter packaged for human vaginal placement. IEEE Trans Biomed Eng 44(5):427–430 3. Karacolak T, Hood AZ, Topsakal E (2008) Design of a dual-band implantable antenna and development of skin mimicking gels for continuous glucose monitoring. IEEE Trans Microw Theory Tech 56(4):1001 4. Hout S, Chung J-Y (2019) Design and characterization of a miniaturized implantable antenna in a seven-layer brain phantom. IEEE Access 7:162062–162069 5. Song H, Kubota S, Xiao X, Kikkawa T (2016) Design of UWB antennas for breast cancer detection. In: Proceedings of international conference on electromagnetics in advanced applications (ICEAA) 6. Shults MC, Rhodes RK, Updike SJ, Gilligan BJ, Reining WN (1994) A Telemetryinstrumentation system for monitoring multiple subcutaneously implanted glucose sensors. IEEE Trans Biomed Eng 41(10):937–944 7. Tang Z, Smith B, Schild JH, Peckham PH (1995) Data transmission from an implantable biotelemeter by load-shift keying using circuit configuration modulator. IEEE Trans Biomed Eng 42(5):524–528 8. Valdastri P, Menciassi A, Arena A, Caccamo C, Dario P (2004) An implantable telemetry platform system for in vivo monitoring of physiological parameters. IEEE Trans Inform Technol Biomed 8(3):271–278 9. Duan Z, Guo Y-X, Je M, Kwong D-L (2014) Design and in vitro test of a differentially fed dualband implantable antenna operating at MICS and ISM bands. IEEE Trans Antennas Propag 62(5):2430–2439 10. Sabria SF, Mohd. Samb S, Kamardinb K, Mohd. Daudb S, Sallehb NA (2015) Review of the current design on wearable antenna in medical field and its challenges. J Teknol 78(6–2):111– 117 11. Rec.ITU-RRS.1346 (1998) Int. Telecommun. Union, Geneva, Switzerland 12. Xia W, Takahashi M, Ito K (2009) Performances of an implanted cavity slot antenna embedded in the human arm. IEEE Trans Antennas Propag 57(4):894–899 13. Haga N, Saito K, Takahashi M, Ito K (2009) Characteristics of cavity slot antenna for body-area networks. IEEE Trans Antennas Propag 57(4):837–843 14. Malvia S, Bagadi SA, Dubey US, Saxena S (2017) Review article: epidemiology of breast cancer in Indian women. Asia-Pacific J Clin Oncol 1–7 15. Kajal MM (2018) A review on the microwave breast imaging techniques. Opt Wirel Technol 56. https://doi.org/10.1007/978-981-13-6159-3_56 16. Tipa R, Baltag O (2006) Microwave thermography for cancer detection. Rom J Phy 51(3– 4):371–377 17. Zhurbenko V (2011) Challenges in the design of microwave imaging systems for breast cancer detection. Adv Electr Comput Eng 11(1):91–96 18. Pozar DM (1998) Microwave engineering, 2nd edn. Wiley, Hoboken 19. Duan Z, Guo Y-X, Xue R-F, Je M, Kwong D-L (2012) Differentially fed dual-band implantable antenna for biomedical applications. IEEE Trans Antennas Propag 60(12) 20. Huang F-J, Lee C-M, Chang C-L, Chen L-K, Yo T-C, Luo C-H (2011) Rectenna application of miniaturized implantable antenna design for triple-band biotelemetry communication. IEEE Trans Antennas Propag 59(7)

Implementation of XOR Gate Using a Nonlinear Polarization Rotation in Highly Nonlinear Fiber Vasundhara, Lovkesh, and Surinder Singh

Abstract In this paper, XOR gate is implemented based on nonlinear polarization rotation, initiated by Kerr effect in a single highly nonlinear fiber (HNLF) having length of 5 km and bit rate of 20 Gbps. Two cross-polarized inputs and polarization controller are used to control the Azimuth and ellipticity angle for the different input powers. Quality factor is analyzed versus different input power, extinction ratio, Azimuth, and ellipticity angle over various length of an optical fiber. Erbium-doped fiber amplifier (EDFA) is used for its nonlinear property and fast response time for the generation of XOR logic output. Keywords Nonlinear fiber optics · EDFA · Polarization rotation · Optical fiber communication

1 Introduction The problem of speed limitations of electronic devices is needed to be solved by using an all optical signal processing network. The optical logic gate plays a crucial role to tackle this issue. These gates are used in the number of applications such as bit error monitoring, switching triode, optical communication, designing of half adder, ultra-high speed pattern generation and recognition, and data encoding and decoding circuits. Optical logic gates are implemented by considering nonlinear effects in semiconductor optical amplifier (SOA), highly nonlinear fiber (HNLF) [1, 2]. In SOA, slow recovery time reduces the speed of operation but EDFA provides fast output response. Various techniques are used for achieving the XOR function based on the utilization of integrated Mach–Zehnder interferometer based on SOA, Vasundhara (B) · S. Singh Department of Sant, Longowol Institution of Engineering and Technology, Longowal, Punjab 148106, India Lovkesh Department of Electronics and Communication, Punjabi University, Patiala, Punjab, India S. Singh Indian Institute of Information Technology, Una, Himachal Pradesh, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_23

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but SOA suffer from noise and speed factor. In another way, the cost of the system also increases which required multiple SOA to perform the XOR logic operation. In [3], the author demonstrated XOR and XNOR logic gates based on optical signal in a 1 km HNLF at 10 Gbps using an erbium-doped fiber amplifier (EDFA). Author mentioned that a cross-polarization enhances a nonlinear depolarization which gives rise to cross-talk between the polarization division multiplexing (PoIDM) subchannels [4]. In [5], the author analyzed an optical sampling technique rely on rotation of a nonlinear polarization using a single SOA. Further work [6] presents all optical networks based on a contention detection circuit at a data rate of 120 Gbps using the four-wave mixing (FWM) in an HNLF. This paper analyzed an optical XOR logic gate using a nonlinear polarization rotation produced by a Kerr effect in a single 5 km highly nonlinear fiber by combining two different input light waves at ň1 and ň2 which enhances birefringence in an HNLF; therefore, the polarization state of third lightwave at ň3 is rotated. Logic 0/Logic1 state of the two input light waves is decided by the induced birefringence ON/OFF condition of the two input light waves which is decided by the induced birefringence. Getting an XOR logic gate after passing through a polarizer by mixing of two inputs. Obtaining an extinction ratio of 26 dB on a 5 km length of HNLF, we obtain the XOR logic gate at the input of 10 Gb/s. Considering the different input powers of different lengths, we implement the XOR function utilizing the EDFA and HNLF. Optical signal processing depends on the functioning of the XOR logic gate.

2 Simulation Setup A schematic figure of XOR logic function is shown in Fig. 1. In this setup, the polarization state of both the wavelengths ň1 and ň2 are kept orthogonal to one other. The inputs are positioned at an angle of 45° with respect to the third CW input signal at ň3 . The polarizer is aligned orthogonal to the original state of polarization of the input signal having wavelength ň3 . When both inputs of the XOR gate are OFF, output will not be present after passing through the polarizer. If anyone of the input is ON, then the Kerr effect is generated. It produces a difference in the refractive index

Fig. 1 Technique behind of optical XOR (IEEE photonics technology letters, p. 1232)

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between polarization positioned parallel along with ň1 and the one which positioned perpendicular to the ň1 (polarization alignment of ň2 ). Further, the polarization condition of ň3 is rotated due to the difference in the refractive index by applying one of the inputs is ON. In this case, polarizer is not orthogonal to ň3 and some part of the signal cross and an output signal at ň3 will be present at the detector end. When both the inputs are ON, birefringence produced by ň1 and ň2 is canceled out and provides a total zero polarization rotation value of the third input light wave of wavelength ň3 . By this, the polarization state of ň3 is positioned with the same angle with respect to the polarizer. Based on the above principle, XOR operation is implemented at the input of ň1 and ň2 . Figure 2 depicts a simulation setup. Input data pattern 101,011 is applied at ň1 , and another data pattern 101,000 is considered at ň2 . These two optical signals are fed to the Mach–Zender modulator. XOR gate performance is analyzed at different input power, optical fiber length, and different Azimuth and ellipticity angles. The input wavelength of ň1 , ň2 , and ň3 is 1548 nm, 1550 nm, and 1554 nm. The two inputs are collaborated by a polarization beam combiner (PBC), and the resultant output is combined with a third input wavelength (ň3 ). The coupling ratio for this setup is 50:10 and the length of HNLF is 5 km. Further, the nonlinear coefficient is fixed 9.1 W−1 km−1 , and 0.45 dB/km is considered as a fiber loss value with a reference wavelength of 1552 nm. Polarization mode dispersion (PMD) is taken below 1 ps for HNLF. The polarization controller (PC3) is used to adjust the polarization state of ň3 which is perpendicular to the polarization state of ň2 . Polarization state of ň3 is rotated to 45° using a polarization waveplate by 22.6°. PC4 is required to maintain the polarization perpendicular to the direction of the polarizer, when both the inputs (ň1 , ň2 ) are off. Optical filter having a bandwidth of 0.6 nm is taken to obtain the desired signal (Tables 1 and 2).

Fig. 2 Simulation setup of XOR gate implementation

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Table 1 Parameters used of HNLF Fiber type

Fiber length (km)

Attenuation (db/km)

Dispersion (ps/nm/km)

Dispersion slope (ps/nm2 /km)

Eff area (μm2 )

Differential group delay (DGD) (ps/km)

HNLF

3 km

0.45

16.75

0.075

80

0.5

Table 2 Parameters used of EDFA

Amplifier type

Noise figure (dB)

Noise center freq. (THz)

Noise bandwidth (THz)

EDFA

3

193.4

13

3 Results and Discussion XOR performance is analyzed using an optisystem simulator on different input optical power. The optical input power of the two inputs is kept same first and then varying. The input power of ň3 is nearby 3 dBm having a wavelength of 1554 nm. Figure 3 shows the effect of input power versus the quality factor. P1 is of wavelength 1548 nm, and P2 is of the wavelength of 1550 nm. With an increase in the input power of both of the inputs, the quality factor increases. As the fiber having length 2 km has higher quality factor value as 3.6, then a fiber having a length of 2.9 km has a quality factor of 2.9. So as the length of the fiber and power increases, the quality factor is also gets affected. Extinction ratio (ER) is defined as the fraction of two optical power signal produced by an optical source. ER = P1/P2, P1 is the optical power when the light wave is present, and P2 is the optical power when the light 3.8

Quality Factor

3.6

3.4

3.2

3.0

2.8 -20

-15

-10

-5

Input Power (dBm)

Fig. 3 Quality factor versus input power

0

5

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Fig. 4 Description of Kerr effect (2011 Fourth International Conference on Intelligent Computation Technology and Automation, p 401

wave is off. Figure 4 shows that the ER value increases and quality factor increases and then decreases and finally remains constant. Polarization rotation-based Kerr effect is used to enhance the performance of the logic gate. A schematic diagram of a Kerr effect is given in the figure in which the probe and the pump input optical beams are polarized linearly at an angle of 45° to one another at the input end. The function of a polarizer is to stop the probe transmission when pump signal is absent.

3.1 High Power Requirement for a Nonlinearity Main issue of fiber nonlinearity is its high power requirement. With the increase in the power, quality factor enhances for all the pump power levels. At a minimum input signal power, quality factor generally distorted due to the low value of amplified spontaneous emission (ASE). As the low input power does not let the photons to excite to higher level, so less energy level causes minimum quality factor. Analyzing this variation for three input powers. (1) P1 = P2 = −21 dB. (2) P1 = P2 = −20 dB. (3) P1 = P2 = −19 dB. For less input power, a quality factor value is more. Figure 5 shows that with the increase in an ellipticity angle, quality factor decreases over the different lengths of an optical fiber. We analyze this value for different values of the ellipticity angle from 0 to 45° having a different length of 2, 2.1, 3 km. Figure 6 shows that with the increase in an Azimuth angle, quality factor decreases over the different lengths of an optical fiber. We analyze this value for different values of Azimuth angle from 0 to 80° having different lengths of 2, 2.1, 3 km (Fig. 7). Azimuth angle (phi) describes the direction of the axis when the intensity of the transmitted beam is the largest, while the lowest intensity is evaluated when the axis (phi) of the polarimeter is perpendicular to this direction. The angle (phi) indicates the principal axis of the polarization ellipse of the light beam. Single linear polarizer is far enough to measure the Azimuth and ellipticity angle. The Azimuth angle will be corresponding to the polarization angle yielding the maximum transmission of light passing through it (Table 3) (Fig. 8). As the Azimuth and ellipticity angle value increases, Q-factor decreases because according to the definition these angle are measured at the peak intensity of power where the chances of noise is more and subsequently bit error ratio (BER) is more and we all are aware that Q-factor is inversely proportional to the BER, so BER is

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Quality Factor

3.2 0.6 3.1 0 .0 3 4 2.9 0.2 2.8 2 .0 0 10

20

30

40

50

Extinction Ratio (dB)

Fig. 5 Quality factor versus extinction ratio

50

Quality factor

40

30

20

10

0 0

10

20

30

40

50

Ellipticity angle (degrees)

Fig. 6 Quality factor versus ellipticity angle

more and Q-factor is less. Criteria for better describing the polarization state are by considering a Stokes parameter. Orientation of Azimuth and ellipticity angle (ψ, χ) defines the Stokes parameter as by these equations.[9] ψ = 1/2 arctan(s2 /s1 )

(1)

χ = 1/2 arcsin(s3 )

(2)

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50

Length=2km Length=2.1km Length=3km

40

Quality Factor

217

30 20 10 0 0

20

40

60

80

100

Azimuth angle(degrees)

Fig. 7 Quality factor versus azimuth angle Table 3 Comparison between implementation of a logic gate by a different author

Parameter

Fok et al. [7]

Singh et al. [8]

Proposed design

Data rate (Gbps)

10

10

20

Interactive medium

HNLF

SOA

EDFA

Nonlinear effect

FWM

XPoIM

XPoIM

Quality factor (dB)

10

33

50

Fig. 8 Description of Azimuth and ellipticity angle (2007 Proc. R. Soc. A, p. 3378)

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s1 , s2 , and s3 are the Stokes parameters.

4 Conclusion We have analyzed a new approach of designing an XOR logic gate based on the rotation of a linear polarizer. The Q-factor is also analyzed at different Azimuth and ellipticity angle over the various fiber lengths. Extinction ratio can be obtained from the range of 26 dB. This design can be further extended to perform various signal processing operations in an optical signal network. When pump signal is present, state of polarization (SOP) of the probe signal changes and being rotated due to the birefringence caused by the pump and a part of the probe signal cross through and output present after the polarizer. Here, XOR signal is our pump signal and being amplify by EDFA. Output power and extinction ratio (ER) of XOR gate amend considerably since the Kerr effect and the fiber nonlinearity using HNLF are directly proportional to each other. Quality factor of 50 is achieved in this work. Acknowledgements The authors would like to thank Science & Engineering Research Board, New Delhi, for their funding under Core Research Grant vide sanction no: File No. EMR/2017/004162 dated: 01-11-18 and DST (International Bilateral Cooperation Division), New Delhi, for their funding to Indo-Russian joint project vide sanction no: INT/RUS/RFBR/P-312 dated: 11.03.2019.

References 1. Yu C, Christen L, Luo T, Wang Y, Pan Z, Yan L-S, Willner AE (2005) All-optical gate using polarization rotation in single highly nonlinear fiber. IEEE Photon Tech Lett 17:1232–1234 2. Wei CC, Huang MF, Chen J (2005) Enhancing the frequency response of cross-polarization wavelength conversion. IEEE Photon Technol Lett 17(no. 8):1683–1685 3. Li L, Jian Wu, Qiu J, Bingbing Wu, Kun Xu, Hong X, Li Y, Lin J (2010) Reconfigurable all-optical logic gate using four-wave mixing (FWM) in HNLF for NRZ-PolSK signal. Opt Commun 283(19):3608–3612 4. Winter M, Setti D, Petermann K (2010) Cross-polarization modulation in polarization-division multiplex transmission. IEEE Photon Technol Lett 22(8):538–540 5. Zhang S, Liu Y, Liu Y, Dorren HJS (2012) All-optical sampling exploiting nonlinear polarization rotation in a single semiconductor optical amplifier. Opt Commun 285(6):1001–1004 6. Singh S, Singh D, Sharma V, Singh S, Ngo QM (2019) Design of all optical contention detection circuit based on HNLF at the data rate of 120 Gbps. Opt Fib Technol 52:101958 7. Fok MP, Prucnal PR (2010) Polarization effect on optical XOR performance based on four-wave mixing. IEEE Photon Technol Lett 22(15):1096–1098 8. Singh S, Ye X, Kaler RS (2013) All optical wavelength conversion based on cross polarization modulation in semiconductor optical amplifier. J Lightwave Technol 31(11):1783–1792 9. Said Y, Rezig H, Bouallegue A (2008) Evaluation of the cross-polarization modulation impact on the SOA behavior: using to perform all-optical signal processing. In: MELECON 2008—the 14th IEEE mediterranean electrotechnical conference, pp. 867–871. IEEE

Designing of Hybrid Photonic Crystal Fiber for Better Filter Characteristics Using Gallium Nitride Shubham Sharma, Sajai Vir Singh, Ankit Agarwal, Nitesh Mudgal, Manish Tiwari, Ravi Kumar Maddila, and Ghanshyam Singh

Abstract In today’s era of fast and miniaturized technology, photonic crystal fibers (PCFs) are quite popular in the field of optical sensing network systems. In addition to the advantage of having minimum dispersion in the network, PCFs have minimum power consumption, lower confinement losses, and higher birefringence than optical fibers. These attributes prove to be suitable for filtering applications. A hybrid photonic crystal fiber (HPCF), having five layered air holes of elliptical shape and a ring waveguide at core, has been proposed having Gallium Nitride (GaN) as the wafer material, which achieves a power of 0 dB at 1.03 μm and a maximum power of nearly 0.15 W, i.e., −8.23 dB at 1.55 μm. For hybrid, transverse electric (TE), and transverse magnetic (TM) mode, it supports narrow band gaps of 0.095822– 0.130652, 0.0958–0.1365, and 0.095822–0.136521, respectively. Proposed design is easy to fabricate and provides better results than the already done research in the past. Keywords Photonic band gap · PWE · Photonic crystal fiber · FDTD · Elliptical air holes

S. Sharma (B) · N. Mudgal · R. K. Maddila · G. Singh Department of Electronics and Communication Engineering, MNIT, Jaipur, India e-mail: [email protected] A. Agarwal Swami Keshavanand Institute of Technology, Management and Gramothan, Jaipur, India S. V. Singh Department of Electronics and Communication Engineering, Jaypee Institute of Information Technology, Noida, India M. Tiwari Department of Electronics and Communication Engineering, Manipal University Jaipur, Jaipur, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_24

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1 Introduction In recent past, optical systems utilization in communication field have been increased at an alarming rate as compared to other technologies in view of low dispersion and fast communication being provided by optical fiber. Optical fibers find its applications in designing of optical devices such as tunable wavelength division multiplexer, filters, waveguides, delay lines, etc. Among different types of fiber viz; multicore, polarization maintaining, photonic crystal, etc., Photonic crystal fiber (PCF) has emerged as the most convenient technology to be implemented in various applications of telecommunication and non-telecommunication sectors. PCFs are highly flexible micro-structured fiber and hole—assisted fiber whose working principal is based on modified total internal reflection (MTIR) in view of photonic crystal properties. PCF is considered to be deemed next generation optical fiber. Classical optical fibers have few shortcomings when it comes to its structural properties. Classical optical fibers have core surrounded by cladding which needs greater and skilled precision at the time of manufacturing. In photonic crystal fibers, microstructure holes are being arranged periodically which affects the photons when in motion. Total internal reflection is the basic concept behind guiding of light into the waveguide or it guiding can be achieved by photonic band gap concept. Pure silica is generally used as core and air holes’ act as cladding, running along the length of the fiber. The structure provides the core to be safe from the introduction of cladding holes. It needs lesser precision, and it is less complicated than manufacturing of classical optical fiber. Therefore, it is easier to fabricate as compared to optical fiber. What makes PCF advantageous over classical fibers is the flexibility in changing the lattice structure, change in diameter of air holes, number of air holes and distance between them. By having an ease in changing the various attributes of PCFs, it becomes easier to alter the design as per the required performance. Due to which they are best suited candidate for nanotechnology. Various research has been done in past to improvise the propagation properties of PCFs. This paper is divided into four sections. Introduction illustrated in the above paragraphs comprises Sect. 1. Section 2 comprises of design and methodology. In Sect. 3, results of the simulation are elaborated and discussed. Lastly, conclusions made from the discussions of Sect. 3 are summarized in Sect. 4.

2 Design and Methodology Various parameters, which change the properties of PCF, are shape and size of cladding air holes, pitch of air holes, pattern arrangement, azimuth angle, and filling. Based on the concept of finite difference time domain (FDTD), OptiFDTD is used to design the structure.

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Fig. 1 Design of the proposed HPCF [1]

In the proposed HPCF design, a ring waveguide is being introduced at the core and array of elliptical air holes as cladding. It consists of symmetric elliptical holes structures, and all elliptical holes have air with refractive index of 1. A two-dimensional hexagonal array has a scale of Ai + Bj + C k with values A = 11, B = 1, C = 12 (i, j, k are integer values). A lattice dimension of 2.3 μm is being represented as  pitch. The proposed design has fixed pitch; other suggested designs also have variable pitch. Pitch is defined as distance between centers of two adjacent cladding holes [1]. In wide sense, PCF is periodic dielectric medium with less loss and material’s refractive index used for the medium plays an important role. A ring waveguide, having air as material profile, is being used at the core, and Gallium Nitride is used as wafer material that shows non-dispersive characteristics. For major and minor axis, radius r 1 and r 2 with values 0.7 and 0.3 μm is being considered for all cladding air holes having elliptical shape. The outer and inner radius has values of 1 and 0.4 μm (width 0.6 μm) for ring waveguide (Fig. 1). To represent effective refractive index neff , relation as given in Eq. (1) is used. n eff = β0 keff where β 0 = propagation constant. k eff = wave number.

(1)

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The effective refractive index of the material is being calculated using Sellmeier’s equation. n 2eff

−1 =

 i



B I λ2   2 λ −ki2

 (2)

Here, neff = effective refractive index. λ = operating wavelength (μm). and BI and k i are Sellmeier’s coefficients. Table 1 shows the Sellmeier’s coefficients used in Eq. (2). Figure 2 shows the wavelength versus refractive index using Eq. (2). Using MATLAB. The wavelength range used here is 0.2–1.6 (μm). For input signal source, continuous point source of 1.55 μm wavelength is being used to the ring waveguide. For x and z axis, boundary conditions are periodic boundary condition (PBC), and type of boundary condition used for axis y is anisotropic perfectly matched layer (APML). The medium created by PML boundary condition has no incident Table 1 Sellmeier’s coefficients of proposed material

Fig. 2 Wavelength versus effective refractive index curve

Sellmeier’s coefficient BI

k i (μm)

B1 = 1.4313493

k 1 = 0.0726631

B2 = 0.65054713

k 2 = 0.1193242

B3 = 5.3414021

k 3 = 18.028251

Designing of Hybrid Photonic Crystal Fiber … Table 2 Simulation material parameters of proposed HPCF

Parameters

223 Typical value

Shape

Hexagonal

No. of layers

5

Pitch ()

2.3 (μm)

Size of mesh

x = 0.09 (μm) y = 0.09 (μm)

No. of mesh cells

X = 243 Z = 281

Length of wafer

25.3 (μm)

Width of wafer

21.91 (μm)

Material of wafer

n = 2.31

Step size

930

wave reflection. APML is an absorbing type boundary condition. Table 2 shows the parameter proposed for HPCF. Most important property of PCF is photonic band gap which occurs because of Bragg scattering interference in periodic dielectric structure and is dependent on wavelength and mesh constant.

3 Results and Discussion The performance of proposed HPCF is evaluated on the benchmark of plots of electric field E y component, magnetic field H x component, frequency domain analysis of structure, and bandgap diagram for TE, TM, and hybrid mode, respectively. The normalized power of ~0.15 W is achieved. Figures 3 and 4 illustrate the improving electric field E y component and magnetic field H x component of TE field of the HPCF structure. The electric field component E x enhances from 0.3803 to 0.5453 V/μm at the core of proposed structure. Figure 5 illustrates the frequency domain analysis of the proposed structure. Figures 6, 7, and 8 illustrate photonic band diagram for TE, TM, and hybrid polarization. The photonic band diagram of TM, TE, and hybrid modes is plotted between Eigen vector range and normalized frequency (a/λ). The photonic band gap supports hybrid, TE, and TM mode in the range of 0.095822–0.130652, 0.0958– 0.1365, and 0.095822–0.136521.

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Fig. 3 Electric field component E y of TE field

Fig. 4 Magnetic field component H x of TE field

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Fig. 5 Frequency analysis at various observation point

Fig. 6 Bandgap diagram for TE mode

4 Conclusion In this paper, a five-layered photonic crystal fiber having air holes of elliptical shape, and a ring waveguide at the core has been proposed. The proposed design has been analyzed by FDTD method. The photonic bandgap of 0.0958–0.1365, 0.095822– 0.136521, and 0.095822–0.130652 for TE, TM, and hybrid mode will be supported by the proposed design. The minimum power of 0 dB is achieved at 1.03 μm, and the maximum power of nearly 0.15 W (i.e., −8.23 dB) is achieved at 1.55 μm wavelength,

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Fig. 7 PWE bandgap diagram of TM mode

Fig. 8 PWE bandgap diagram of hybrid mode

and these values are present at nearly 290.7 THz and 96.70 THz and for maximum and minimum power in the frequency domain, respectively. By introducing the GaN as the substrate material can proved to achieve the results that are being achieved by the complex PCF designs (i.e., introducing octagonal and decagonal geometries). The novelty of the work is the Gallium Nitride (GaN) being used as the substrate which has yielded much better results than the existing one.

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References 1. Vyas AK (2019) Elliptical air holes based photonic crystal fiber for narrow Band gap and peak power at 1.55 micrometre wavelength. Optik 184:28–34 2. Amirouche A, Bouridah H, Beghoul MR, Boutaoui N (2016) Study of two dimensional photonic crystal nanocavities based on Gallium Nitride (GaN). Optik-Int J Light Electron Opt 127(5):2708–2714 3. Liu Y, Jing X, Li S, Zhang S, Zhang Z, Guo Y, Wang J, Wang S (2018) High sensitivity surface plasmon resonance sensor based on D-shaped photonic crystal fiber with circular layout. Opt Fiber Technol 46:311–317 4. Arif MFH et al (2019) A nonlinear photonic crystal fiber for liquid sensing application with high birefringence & low confinement loss. Sens Bio-Sens Res 22:100252 5. Arif MFH, Biddut MJH (2017) Enhancement of relative sensitivity of photonic crystal fiber with high birefringence & low confinement loss. Optik 131:697–704 6. Arif MFH, Ahmed K, Asaduzzaman S, Azad MAK (2016) Design & optimization of photonic crystal fiber for liquid sensing applications. Photon Sens 6(3):279–288 7. Ademgil H, Haxha S (2015) PCF based sensor with high sensitivity, high birefringence & low confinement losses for liquid analyte sensing applications. Sensors 15(12):31833–31842 8. Saitoh K, Koshiba M (2003) Single-polarization single-mode photonic crystal fibers. IEEE Photon Technol Lett 15(10):1384–1386

Silicon-On-Insulator Photonics Waveguide Design for Near-IR Evanescent Field-Based Blood Sensor Veer Chandra, Neha Choudhary, and Rakesh Ranjan

Abstract Silicon-on-insulator-based photonics waveguides, namely Ridge, Rib, and Slot waveguides are explored for the blood sensing application in near-IR region. The absorption-based sensing principle has been adopted in the upper cladding/evanescent region of the waveguide. The finite element method-based simulation results, obtained through the COMSOL multiphysics software, have illustrated that the slot waveguide has the optimum performance for the absorption-based sensor applications. Keywords Silicon-on-insulator · Photonics waveguide · Evanescent field · Sensor

1 Introduction Silicon-on-insulator (SOI) based photonic sensors have found wide applications in many areas of real-life requirements, such as in medical, environmental, etc. Due to its CMOS compatibility and small chip area requirement, the SOI-based sensors are usually preferred, and can be integrated with on-chip control and processing circuits to build a chip-scale sensing system. The SOI waveguide-based sensors are popular because of its low propagation loss and ease of fabrication. The working principle of such photonic sensors is mainly based on the evanescent field absorption [1]. As the light is propagating through the SOI based photonics waveguide, it may undergo to the total internal reflection at core-cladding boundary, and some fraction of the light may penetrates towards the cladding region. The SOI based evanescent field sensors have been investigated intensively in [2, 3]. By etching the upper portion of V. Chandra (B) · N. Choudhary · R. Ranjan Optical Fiber Communication and Photonics Laboratory, Department of Electronics and Communication Engineering, National Institute of Technology Patna, Patna, Bihar 800005, India e-mail: [email protected] N. Choudhary e-mail: [email protected] R. Ranjan e-mail: [email protected] © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_25

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cladding region, the cladding surface sensing performance can be achieved, as the guided light in the core is directly related to the evanescent area over the surface. These kinds of sensor have been demonstrated using ring resonators, etc., in near-IR spectrum [3–5]. The authors have utilized the refractive index deviation due to the presence of different sample, with various concentrations of the sample, placed in the etched cladding/evanescent field region. Further, in order to increase the sensitivity of the sensor, a film (of TiO2 ) is coated on device surfaces. Silicon-on-sapphire (SOS) waveguides based evanescent field sensors have been investigated extensively in [6– 10]. These waveguides can be utilized to provide an inexpensive and robust sensor solution for different sensing applications, with faster response and larger reusability. In the current work, three different photonic waveguides, namely Ridge, Rib, and Slot have been considered, in order to search their optimum performance for some sensing applications, such as photonic blood sensor in near-IR region. The photons present in the upper cladding/evanescent region interact with the surrounding blood sample placed over the etched cladding region, which results in power attenuation/absorption depending on the types/concentration of blood sample.

2 Theory and Design Parameters 2.1 Absorption-Based Sensing The sensing phenomena based on the absorption property of materials are usually preferred over the effective refractive index-based sensing. As, in photonic waveguides, the change in the effective refractive index has lesser impact, as compared to the change in the evanescent field, which mainly caused by the absorption phenomena. Absorption is depending on the sample taken for sensing, materials used for sensor design and operating wavelength. The percentage of evanescent field in upper cladding region changes during the interaction with the molecules of the sensing sample. Therefore, the photonic sensors are required to be designed with waveguides having comparatively large evanescent field and, hence, high sensitivity. The sensitivity can be calculated by Eq. (1) [10], Sensitivity(s) = −ηE Lexp(−ηEC L − αwvg L)

(1)

where, η, E, C, αwvg , and L are the evanescent field fraction, absorption coefficient of blood sample, concentration of blood sample, intrinsic waveguide loss, and length of the waveguide, respectively. The evanescent field fraction is defined as the amount of power in upper (etched) cladding/evanescent region, interacting with the sensing environment.

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2.2 Photonic Waveguide In the SOI based photonic waveguide sensor, the sensing phenomena is depending on the extent of evanescent field detected during the interaction with the environment, which includes the sample (gas) [11]. The variations, due to the insertion of blood sample, in the evanescent filed or effective refractive index, can be identified by determining the power of light intensity at the output of the corresponding photonic waveguides, by using some suitable IR (Infrared) detector. Hence, to achieve higher sensitivity, one can use the suitable photonic waveguide, with the high evanescent field in upper cladding region. In addition to this, the sufficient amount of light through the core region may also be required for some of the photonic waveguides. The SOI-based different waveguide structures that are used in this paper for sensing applications are shown in Fig. 1. Where, ‘w’ is the width of the upper high-index region of the Ridge and Rib waveguides, ‘H’ is the height of the high-index layer, which is deposited on the low-index region (SiO2 ), as shown in Figs. 1a, b, respectively. Figure 1c shows the slot waveguide having arm width ‘w,’ height ‘H,’ and slot gap ‘d’ of slot region. The material used for the core is silicon, and upper cladding is blood sample, having the refractive indices, respectively, of approximately 3.45 and 1.28 [12] at the operating wavelength of 2.25 µm (i.e., in the Near-IR region).

Fig. 1 Structures of a Ridge waveguide, b Rib waveguide, and c Slot waveguide for sensing purpose

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3 Simulation Results For mode characterization, all the simulations have been performed using the finite element method (FEM) based COMSOL multiphysics simulation software. During the simulation for all the three photonic waveguides, the scattering boundary condition has been applied to address the losses during the propagation of light. In this paper, only fundamental mode propagation constraint has been considered to search the applicability of the three major photonic waveguides for blood sensing application at the wavelength of 2.25 µm.

3.1 Mode Field Distribution For the Rib and Ridge photonic waveguides, the height ‘H’ is considered as 0.22 µm; while for slot waveguide, the height ‘H’ is considered as 0.25 µm. Initially, the width ‘w’ is assumed as 0.5 µm for both Ridge and Rib waveguides; while for slot waveguide, width ‘w’ of both the arms/high-index regions is taken as 0.3 µm, with the slot gap ‘d’ of 80 nm. The mode field distributions of all the three different waveguides have been shown in Fig. 2. The optical field intensity is mainly confined in the core region for Ridge and Rib waveguides, as depicted in Figs. 2a, b. While, Fig. 2c shows the same for the slot waveguide, where, the light confinement is mainly in the slot region. This is mainly due to the discontinuity of electric field in the high refractive index regions (i.e., Si); whereas, the optical field intensity in the high refractive index regions is extremely low. The performance of these photonic waveguides in blood sensing applications has been investigated in terms of real part of effective refractive index ‘Re(n eff )’ and evanescent field (%) by varying the width ‘w’ of the Ridge and Rib waveguides. Moreover, for the slot waveguide, the performance analysis has been done by varying both slot gap ‘d’ and arm width ‘w.’ For the validation of the obtained results, it is compared with the results obtained with air as cladding region. Similar mode distributions have been obtained with the other considered values of ‘w’ and ‘d’ for all the three waveguide structures, which are used for the further analysis of various modal characteristics.

3.2 Effective Refractive Index (ERI) The effective refractive index (ERI) depends on the waveguide geometry and materials used. Here, width ‘w’ has varied from 0.5 to 1 µm for both Ridge and Rib waveguides, to estimate the real part of effective refractive index, as shown in Figs. 3a, b, respectively. The effective refractive index starts increasing as ‘w’ increases. This is mainly due to the fact that with the increase in ‘w,’ more optical field starts getting confined inside the core. The figures clearly show that the presence of blood sample,

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Fig. 2 Mode field distributions of a Ridge waveguide, b Rib waveguide, and c Slot waveguide

(a)

(b)

Fig. 3 Effective refractive index for the different width in the a Ridge waveguide, and b Rib waveguide

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Fig. 4 Effective refractive index for the different width in the Slot waveguide

having higher refractive index than air, in the upper cladding region, causes to increase the values of Re(n eff ) for different considered widths. For the analysis of effective refractive index in slot waveguide, the two different slot gaps, 80 and 140 nm, have been considered with the variations in arm width (w) from 0.3 to 0.8 µm, as shown in Fig. 4. The figure clearly demonstrates that in slot waveguide, the difference in ERI has significantly larger values as compared to other two waveguides, which results in high sensitivity with slot waveguide. The maximum achievable difference in ERI values is approx. 0.15 at slot gap (d) of 80 nm for slot waveguide; while, at width (w) of 0.50 µm, the maximum differences in ERI, respectively, for Ridge and Rib waveguides are 0.14 and 0.05. Therefore, slot waveguide has the optimum performance for the blood sensing applications.

3.3 Evanescent Field The evanescent field is also dependent on the waveguide geometry and materials used. Here, again the width has been varied from 0.5 to 1 µm for both Ridge and Rib waveguides to compute the percentage of evanescent field as shown in Figs. 5a, b, respectively. With the increasing widths, the percentage evanescent field starts decreasing, as confinement in core region increases. For both Ridge and Rib waveguides, the figures clearly show that insertion of blood sample increases the evanescent field percentage, as the blood sample has higher value of refractive index than air. Similarly, for the analysis of evanescent field in slot waveguide, the slot arm width (w) has been varied from 0.3 to 0.8 µm for two different slot gaps of 80 and 140 nm, as shown in Fig. 6. In case of slot waveguide, it clearly shows the significant increase in percentage evanescent field as compared to other two waveguides. The maximum value of evanescent field has been observed as approx. 46%, at slot gap of 80 nm and width (w) of 0.25 µm. The highest achievable difference in evanescent field values in comparison to air is approx. 25% at slot arm width (w) of 0.3 µm and slot gap (d) of 140 nm. While, at width (w) of 0.50 µm, the maximum percentage differences

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(b)

Fig. 5 Evanescent field variations for the different widths in the a Ridge waveguide, and b Rib waveguide

Fig. 6 Evanescent field variations for the different width in the slot waveguide

can be achieved as 6%, and 2% for Ridge, and Rib waveguides, respectively, with respect to air in upper cladding region. Hence, in terms of the percentage evanescent field also, the slot waveguide has the most favorable performance.

4 Result and Discussion A comparative study has been done for three photonic waveguide structures, namely Ridge, Rib, and Slot waveguides for blood sensing applications, which are designed for a wavelength in near-IR region (at 2.25 µm). The larger the evanescent field in the upper cladding region, the higher the sensitivity of the photonic sensor [13]. Using Eq. (1), the sensitivity can be determined for different waveguides. Hence, the sensitivity of the waveguide sensor is dependent on the percentage of the evanescent field. The absorption coefficient of blood is 25.137 cm−1 [12] at the operating wavelength of 2.25 µm. Table 1 shows the percentage evanescent field and the maximum

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Table 1 Estimated parameters for different photonic waveguides S. No.

Type of waveguide

Evanescent field (%)

ERI (Max.)

1

Ridge waveguide

33.27

2.29 (at w = 1 µm)

2

Rib waveguide

27.10

2.32 (At w = 1 µm)

3

Slot waveguide

45.63

2.30 (At d = 0.8 µm, and w = 0.8 µm)

achievable value of ERI for all the three considered photonic waveguides. From the table, it is clear that the slot waveguide has the maximum field in the evanescent region (approx. 46%), and hence, it can provide the highest sensitivity for sensing application point of view.

5 Conclusion In this work, the photonic waveguides, such as Ridge, Rib, and Slot waveguides are explored and designed for the evanescent field-based sensing applications in near-IR region, for the blood samples. The basic principle of operation of such photonic sensor is based on the evanescent field absorption by the material to be sensed. As, the slot waveguide with the suitable waveguide dimension can provide the highest evanescent field in the upper cladding region, it can provide the optimum sensing of the blood samples, in comparison with the other two waveguides, i.e., Ridge and Rib waveguides. This kind of analysis may be extended for some other sensing/image detection applications, such as for thumb impression imaging, etc. Acknowledgements The authors gratefully acknowledge National Institute of Technology Patna, and Science and Engineering Research Board, Department of Science and Technology, Government of India for providing COMSOL Multiphysics simulation software, used in the current simulation work.

References 1. Rifat AA, Ahmmed R, Bhowmik BB (2019) SOI waveguide-based biochemical sensors. Chapter 16 2. Fabricius N, Gauglitz G, Ingenhoff J (1992) A gas sensor based on an integrated optical Mach-Zehnder interferometer. Sens Actuators, B Chem 7:672–676 3. Yebo NA, Lommens P, Hens Z, Baets R (2010) An integrated optic ethanol vapor sensor based on a silicon-on-insulator microring resonator coated with a porous ZnO film. Opt Express 18:11859–11866 4. Robinson JT, Chen L, Lipson M (2008) On-chip gas detection in silicon optical microcavities. Opt Express 16:4296–4301 5. Soref RA, Emelett SJ, Buchwald WR (2006) Silicon waveguided components for the long-wave infrared region. J Opt A: Pure Appl Opt 8(10):840

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6. Tien E, Huang Y, Gao S, Song Q, Qian F, Kalyoncu SK, Boyraz O (2010) Discrete parametric band conversion in silicon for mid-infrared applications. Opt Express 18:21981–21989 7. Baehr-Jones T, Spott A, Ilic R, Spott A, Penkov B, Asher W, Hochberg M (2010) Silicon-onsapphire integrated waveguides for the mid-infrared. Opt Express 18:12127–12135 8. Li F, Jackson SD, Grillet C, Magi E, Hudson D, Madden SJ, Moghe Y, Brien CO, Read A, Duvall SG, Atanackovic P, Eggleton B, Moss D (2011) Low propagation loss silicon-on-sapphire waveguides for the mid-infrared. Opt Express 19:15212–15220 9. Wang Z, Liu H, Huang N, Sun Q, Wen J, Li X (2013) Influence of three-photon absorption on Mid-infrared cross-phase modulation in silicon-on-sapphire waveguides. Opt Express 21:1840–1848 10. Huang Y, Kalyoncu SK, Zhao Q, Torun R, Boyraz O (2014) Silicon-on-sapphire waveguides design for mid-IR evanescent field absorption gas sensors. Opt Commun 313:186–194 11. Mere V, Selvaraja SK (2016) Germanium-on-glass waveguides for Mid-IR photonics. In: 13th international conference on fiber optics and photonics. OSA Technical Digest, Optical Society of America 12. Rowe DJ, Smith D, Wilkinson JS (2017) Complex refractive index spectra of whole blood and aqueous solutions of anticoagulants, analgesics and buffers in the mid-infrared. Sci Rep 7(7356) 13. Butt MA, Khonina SN, Kazanskiy NL (2018) Silicon on silicon dioxide slot waveguide evanescent field gas absorption sensor. J Mod Opt 65:174–178

Analysis of Upper Aortic Blood Vessels as a Data Communication Channel Tanmay Deshmukh , Kauser Husainee , and Prabhat Kumar Sharma

Abstract This paper presents the analysis of blood flow through the upper aortic blood vessel of human cardiovascular system using laminar flow model. The model is solved as a boundary value problem (BVP) in pressure conditions, and the velocity distribution of the blood flow is determined. The communication is established by transmitting particles through blood stream to remotely located nano-surgical equipment. For the analysis of communication, particle tracing model is employed and transmission probabilities of injected particles reaching remote site are evaluated. The variation of probability with respect to time in which it reaches the receiver of the equipment is used to determine a threshold time for detection of total particles which defines the bit duration and establishes synchronization. The further scope to communication is discussed in detail. Keywords Molecular communication · Laminar flow · Particle tracing · Transmission probabilities · COMSOL multiphysics · Cardiovascular system

1 Introduction In the complex medical procedures, which require nano-surgical robots, data communication is usually established using the applications of RF engineering. But some particular frequency waves may have adverse effects on pre-adolescent children, pregnant women, elderly humans, patients, etc. So, the data communication can be established using the applications of emerging research field of molecular communication [1]. The approach is to inject a biodegradable solute into the blood stream from the transmitter side, and the total number of particles detected at the receiver sensor nodes of robots placed at various outlets in a given time determines the data sent in the form of a symbol which can be exclusively designed for medical purposes [2].

T. Deshmukh (B) · K. Husainee · P. K. Sharma Department of Electronics and Communication Engineering, Visvesvaraya National Institute of Technology, Nagpur, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_26

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2 Structure and Properties of Vessel Under Analysis In order to carry out the analysis of a blood vessel as a digital communication channel, the characteristics of a particular section of the cardiovascular system have to be considered [3]. In this model, we have studied the upper section of the aorta along with its single inlet from the left ventricle of the heart and five outlets to the body as shown in Fig. 1. These inlet and outlets are placed inside the tissues which are particularly called as the ‘cardiac muscles,’ which help to keep the vessels non-deformable. Hence, the blood fluid characteristics inside the vessels do not change. The length of the vessel is 21.2 cm, the inlet radius is 2.8 mm, and the radius of all the outlets is 2.5 mm. The material properties are given in Table 1.

Fig. 1 Velocity distribution inside the upper aortic blood vessel

Table 1 Material properties

Parameters

Value

Blood density

1060 kg/m3

Blood dynamic viscosity

0.005 Pa s

Arterial density

960 kg/m3

Muscle density

1200 kg/m3

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3 Laminar Flow Model 3.1 Assumptions for Fluid Characteristics Before Modelling • Blood is contemplated as a ‘Newtonian fluid’ which has the relation that the viscous stress is directly proportional to the strain developed. • The orientation of blood vessels in space and the effect of gravity have been neglected. • Blood vessels are assumed to be non-deformable components cylindrical in shape. They do not change their shape as blood flows through them. • Properties of blood before and after addition of oxygen from lungs are considered to be as it is.

3.2 Approach to Model Solution The Laminar flow model has been used for the simulation whose equations are defined as given in Eqs. (1) and (2) ρ(u fluid . ∇)u fluid = ∇ . [− p I + K ] + F

(1)

ρ∇ . (u fluid ) = 0

(2)

where u fluid , μ, p, and ρ are the fluid velocity, kinematic viscosity, pressure, and fluid density, respectively [4]. Also, K = μ(∇u fluid + (∇u fluid )T ) The equations can be solved by simulation in COMSOL multiphysics using the boundary values of either pressure or velocity. The approach used is to model the blood pressure by a function of heart beat as given in Table 2 and plotted in Fig. 2. Table 2 Mathematical equation of heart beat function

Start

End

Function

0

0.5

sin(π t)

0.5

1.5

3 2

−

1 2

   ∗ cos 2π t − 21

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Fig. 2 Heart beat function

3.3 Simulation Result The simulation results of the laminar flow model give the distribution of velocity of fluid inside the aortic blood vessel which is variable in space as well as time.

4 Particle Tracing for Fluid Flow 4.1 Particle Properties The velocity distribution determined from the laminar flow model is further used for the particle tracing. A certain number of particles are released as liquid droplets of the particle solution from the inlet 1 into the blood vessel with the properties as given in Table 3. The assumptions made for particle are: • The particles have perfect spherical geometry. • The particles have smooth surfaces. Table 3 Particle properties

Parameters

Value

Particle density

2200 kg/m3

Particle diameter

1 μm

Dynamic viscosity of droplet

0.885 mPa*s

Surface tension of droplet

0.0729 N/m

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• The particles are made of homogeneous material.

4.2 Analysis of Particle Tracing for Fluid Flow The drag force applied on the particles released from inlet 1 follows the Stokes law of fluid mechanics which is gives by Eqs. (1) and (2), FD =

1 mPV τP

(3)

τP =

ρ P D 2P 18 μ

(4)

where F D is the drag force applied on the particle and mp , v, ρ p , Dp are the mass, velocity, density, and diameter of the particle, respectively. Also, μ is the dynamic viscosity of the droplet containing particles [5]. The distribution of the particle trajectories is determined by computing the model for above properties in the COMSOL multiphysics particle tracing module. The distribution further gives the transmission probabilities at each outlet at each time instant. Probabilities are evaluated for each outlet with time step of 0.2 s till 5 s. Such distribution of probabilities is a great source of data to analyse the blood vessel as a communication channel. This probability distribution matrix works as good as the channel transition matrix of the wireless channel to estimate the performance of the channel or to set the parameters for establishing the communication. The number of particles released from the inlet 1 is varied in the range of 1000 to 1,000,000. Samples of probability matrix are taken at multiple instances in this range, and it is observed that the probability matrix does not vary significantly. Thus, the average probability matrix is computed from these samples. With respect to the total number of particles injected from the inlet 1, the average transmission probabilities of the number of particles received at each outlet at the given instances of time are shown in Fig. 3. The density of the particle also has the impact on the trajectories, but the injection of the particles in the blood stream has conditions on the density of material as it may disrupt the natural working of the heart and cardiovascular system. Thus, the scope of variation of density in the desired application is very limited.

5 Application in the Communication Scheme The transmission probabilities of the particles received at the various receiver sensor nodes of the nano-surgical robots at various instants of time converge to a constant value as shown in Fig. 3. From the variation, the instant of time after which all the

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0.35

transmission probability

0.3 0.25 Outlet 1

0.2

Outlet 2

0.15

Outlet 3

0.1

Outlet 4

0.05

Outlet 5

0 0 -0.05

1

2

To

3

4

5

6

Ɵme (sec)

Fig. 3 Transmission probabilities of particles received at the respective outlets

probabilities converge can be taken as a threshold time ‘T o ’. For data communication to be successful, the receiver sensors have to detect and make a decision on the basis of the count of the total particles detected by each of the sensor nodes at various outlets up to the defined threshold time ‘T o ’ for the combined system [6]. The estimation of the initial number of transmitted particles injected can be made by dividing the count of the total particles detected at an outlet with the transmission probability defined at that outlet at time ‘T o ’. The bit duration of both the transmitter and the receiver side should be equal to ‘T o ’ defined for the combined system of receivers at outlets. Thus, the local synchronization is established using the transmission probabilities. If all the sensor nodes are placed in the same outlet, where the medical operation is required, the threshold time can be particularly set for that probability variation curve only and the communication can be made faster as well as efficient. For sensors with higher detection range and accuracy, multilevel logic can be applied; i.e., the molecular communication in the blood vessel can be established by mapping different numbers of particles to different symbols in the communication model [7, 8].

6 Conclusion The concepts of molecular communication have been used to establish communication with nano-surgical robots used in complex medical operations. For the purpose of analysis of the blood vessel as a digital communication channel, an aortic vessel of the human cardiovascular system with its actual material parameters has been studied.

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After making some required assumptions of fluid characteristics, the laminar flow model equations have been solved using the boundary condition of blood pressure by modelling the heart beat function to get the velocity distribution of the aortic vessel. This distribution was used to trace particles by simulation and determine the variation of transmission probabilities of particles at the various receiver sensor nodes of the nano-surgical robots with respect to time. The variation gives the threshold time which is used to define the bit duration and establish synchronization for successful communication and for further scope for multilevel logic as well.

References 1. Doustali M, Zoofaghari M, Arjmandi H (2019) Diffusive molecular communication in partially blocked cylindrical environment. In: 27th Iranian Conference on Electrical Engineering (ICEE), Yazd, Iran, 2019, pp 1704–1709 2. Felicetti L, Femminella M, Reali G, Gresele P, Malvestiti M, Daigle JN (2014) Modeling CD40based molecular communications in blood vessels. IEEE Trans Nanobiosci 13(3):230–243 3. Navidbakhsh M, Monshizadeh H, Javidi M, Rahmani S (2013) Blood flow simulation in a stenotic vessel surrounded by biological tissue. In: 2013 20th Iranian conference on biomedical engineering (ICBME), Tehran, pp 22–26 4. Filipovic N, Isailovi´c V, Ðuki´c T, Ferrari M, Kojic M (2012) Multiscale modeling of circular and elliptical particles in laminar shear flow. IEEE Trans Biomed Eng 59(1):50–53 5. Guo H et al (2019) Extreme-scale stochastic particle tracing for uncertain unsteady flow visualization and analysis. In: IEEE transactions on visualization and computer graphics, vol 25, no 9, pp 2710–2724, 1 Sept. 2019 6. Chouhan L, Sharma PK, Varshney N (2019) Optimal transmitted molecules and decision threshold for drift-induced diffusive molecular channel with mobile nanomachines. IEEE Trans Nanobiosci 18(4):651–660 7. Felicetti L, Femminella M, Reali G (2013) Establishing digital molecular communications in blood vessels. In: 2013 first international black sea conference on communications and networking (BlackSeaCom), Batumi, pp 54–58 8. Kabir MH, Riazul Islam SM, Kwak KS (2015) D-MoSK modulation in molecular communications. IEEE Trans Nano Biosci 14(6):680–683

A Ground Plane Modified Broadband Circularly Polarized Patch Antenna for Wireless Applications G. Anjaneyulu, T. A. N. S. N. Varma, and J. Siddartha Varma

Abstract A compact monopole with circular polarization is obtained in this work. CP performance is achieved by modifying the ground plane and also introducing the slots will help achieving the lower frequency range in a compact structure. The simulated results show an impedance bandwidth (IBW) of 101% from 2.33 to 7.16 GHz and 25% AR bandwidth from 4.14 to 5.66 GHz. The simulated patch is a compact one with an area of 27 mm × 18 mm. the presented antenna is useful in the 2.6 GHz ISM band and other C band satellite and wireless applications. Keywords Circular polarization · Monopole · RHCP · C band

1 Introduction A very useful and effective part of the modern communication systems are circularly polarized antennas. Because of their versatility in polarizing in every direction, they are often preferred in the most communication systems. Circularly polarized antennas provide more flexibility in the course of multipath interference and multipath fading due to tall buildings and different objects in the surroundings. It will provide enhanced system performance by improving the weather penetration, spectral efficiency, and polarization diversity in radio propagation environment. In context to all the advantages mentioned above, the modern communication systems prefer using circularly polarized antenna for wireless and satellite applications. CP can be achieved with the help of single feed or dual feed mechanism, but the advantage of single feed over dual feed is its simplicity in structure [1, 2], which is more compact than the dual feed structures. Several designs with single feed CP patch antennas having square and circular shape microstrip patch antennas are designed over the years. In the recent years, several circularly polarized antennas with various shapes and compact sizes for G. Anjaneyulu (B) · T. A. N. S. N. Varma MVGR College of Engineering, Chintalavalasa, India J. Siddartha Varma Lendi Institute of Engineering and Technology (A), Vizianagaram, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_27

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specific applications are also being reported. In these designs, the hexagonal and circular shaped slot antennas along with the microstrip stub [3] is designed to achieve wideband CP performance for dual band applications. A square slot on ground plane and a cross-shaped patch [4] for broadband circularly polarized antenna is designed. In [5], L shaped radiator with a vertical probe on a ground and similarly square slot [6] fed with a L shaped coupling strip are designed to achieve circular polarization. In [7], a relatively small D shaped patch with modified ground is used to realize CP. A coplanar fed slot antenna with a ring reflector [8] is used to have a circular polarization. In [9], a Vivaldi antenna cut by half and mirrored by metal cavity is used and a truncated corner patch [10] is used to have CP. A ring slot fed by strip line hybrid coupler [11], a multiple circular sector patch [12] antenna, and a patch using a pair of slots [13] is used to achieve circular polarization. In [14], a CPW fed patch having two T shape feed lines and reverse L shaped ground strips are used to realize CP. A simple monopole patch antenna with effectively planned perturbations on the ground is designed. Two slots are introduced on the ground plane along with some ground modification is done to achieve circular polarization. Here the monopole is acting as primary radiator. An improved impedance bandwidth (IBW) and axial ratio bandwidth (AR-BW) are observed by incorporating the slots on the ground plane. This antenna can be used in the various wireless applications in ISM band and IEEE C band. Further, the antenna design and results are discussed in the next section, and conclusions are provided in the final section.

2 Design Methodology The patch antenna is excited with the help of a 50  microstrip feed line on a FR4 dielectric substrate material having a relative permittivity value of 4.4 and thickness of 1.6 mm. the structure of the proposed circularly polarized antenna is shown in Fig. 1. The main radiator is a monopole structure at the required frequency range, and the ground plane on the other side of the dielectric substrate is modified to achieve circular polarization. Two different sized slots are also introduced on the ground plane. The dimension and placement of the slots are carefully parameterized to achieve the better possible performance of the patch antenna. The optimized values of the simulated antenna are given in Table 1.

3 Simulated Results Discussion The simulation of the propounded antenna has been carried out using the ANSYS HFSS. The proposed patch antenna is the resultant structure of the various iterations and changes to have an optical antenna performance. The reflection coefficient versus the frequency plot is shown in Fig. 2a, and it can be seen that the minimum S11 is at

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Fig. 1 Structure of proposed antenna

Table 1 Optimized values of the proposed structure

Design variable

Value (mm)

Design variable

Value (mm)

ls

27

wf

3

ws

18

l

20

a

6

e

2

b

1

f

4

c

2

g

0.4

d

8

h

1.6

− 34 dB at 2.6 GHz and the IBW of 4.82 GHz from 2.36 to 7.16 GHz. One can also observe that it is operating in a wide bandwidth range. The entire bandwidth range is also having good impedance matching. The AR versus frequency plot is seen in Fig. 2b. The minimum AR value at the center frequency of axial ratio bandwidth is at 0.10 dB. The AR-BW is about 1.25 GHz from 4.41 to 5.66 GHz. The simulated gain of the antenna operating in the CP band is shown in Fig. 4a. It has an average peak gain of 2.2 dBi. The radiation pattern at 4.68 GHz, which is center frequency of AR bandwidth in phi = 0° and phi = 90° planes, is shown in Fig. 3. The LHCP and RHCP plots are shown in the radiation pattern. There is better isolation

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Fig. 2 a S11 versus frequency. b AR versus frequency

Fig. 3 Radiation pattern at Phi = 0° and Phi = 90°

between the both polarizations. The radiation efficiency of the patch antenna in the entire bandwidth is shown in Fig. 4b, and it is around 85%. The designed antenna has the applications in the ISM band and other wireless applications.

4 Conclusion A new simple CP patch antenna on a FR4 substrate is designed in this work. A monopole antenna with the modified ground plane and two slots is used to achieve

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Fig. 4 a Gain of the proposed antenna. b Radiation efficiency

the circular polarization. CP is achieved with the extension of the ground plane and by introducing slots. It has an impedance bandwidth of 101 and 25% fractional AR bandwidth. The propounded antenna is having a right-hand circular polarization, and radiation efficiency is around 85% over entire bandwidth range. This design is useful for 2.6 GHz ISM band and other wireless applications in C band. This wideband performance of this antenna is useful for many satellite applications also. Acknowledgements This work (Major project) was funded by SERB, the Department of Science and Technology (DST), Government of India, Sanction No. EEQ/2016/000396 and Order No. SERB/F/8020/2017-18.

References 1. Wong KL (2002) Compact dual-frequency and dual-polarized microstrip antennas, vol 3 2. Kumar G, Ray KP (2003) Broadband microstrip antennas. Artech House 3. Nasimuddin AA, Jeevanandham N (2016) Circularly polarized slot antennas with wideband performance. In: Asia-Pacific Microwave Conference Proceedings, APMC, vol 2, pp 6–8 4. Chou CC, Lin KH, Su HL (2007) Broadband circularly polarised cross-patch-loaded square slot antenna. Electron Lett 43(9):40–41 5. Bisharat DJ, Liao S, Xue Q (2017) Wideband unidirectional circularly polarized antenna with l-shaped radiator structure. IEEE Antennas Wirel Propag Lett 16:12–15 6. Florencio Díaz R, Rodríguez Boix R, Carrasco Yépez F, Encinar Garcinuño J, Barba Gea M, Pérez Palomino G (2014) Broadband reflectarrays made of cells with three coplanar parallel dipoles. Microw Opt Technol Lett 56(3):748–753 7. Kunwar A, Gautam AK, Kanaujia BK, Rambabu K (2018) Circularly polarized D-shaped slot antenna for wireless applications. Int J RF Microw Comput Eng 29(1):1–10 8. Yuan J, Li Y, Xu Z, Zheng J (2019) A compact CPW-fed low-profile wideband circularly polarized slot antenna with a planar ring reflector for GNSS applications. Int J Antennas Propag

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9. Hu YJ, Qiu ZM, Yang B, Shi SJ, Yang JJ (2015) Design of novel wideband circularly polarized antenna based on vivaldi antenna structure. IEEE Antennas Wirel Propag Lett 14(5):1662–1665 10. Yang SLS, Lee KF, Kishk AA (2008) Design and study of wideband single feed circularly polarized microstrip antennas. Prog Electromagn Res 80:45–61 11. Qing XM, Chia YWM (1999) Circularly polarized circular ring slot antenna fed by stripline hybrid coupler. Electron Lett 35(25):2154–2155 12. Trinh-Van S, Yang Y, Lee KY, Hwang KC (2018) Broadband circularly polarized slot antenna loaded by a multiple-circular-sector patch. Sensors (Switzerland) 18(5):1–12 13. Sitompul PP, Sumantyo JTS, Kurniawan F, Nasucha M (2019) Axial ratio and gain enhancement of a circular-ring slot antenna using a pair of asymmetrical rectangular slots and a parasitic patch for a radio beacon on a nanosatellite. Aerospace 6(4) 14. Saini RK, Dwari S (2016) A broadband dual circularly polarized square slot Antenna. IEEE Trans Antennas Propag 64(1):290–294

Deterministic Two Qubit iSWAP Gate Using a Resonator as Coupler Amit Kumar Sharma and Ritu Sharma

Abstract In this paper, a two Qubit deterministic iSWAP gate is designed and simulated using a resonator as coupler. Fidelity and concurrence of the designed two Qubit iSWAP gate have been calculated. The behavior of this gate is universal. The feasibility of the scheme by observing variation in fidelity and concurrence with different system parameters are assessed and reported. Keywords Quantum computing · Qubits · iSWAP gate · Fidelity · Linear optics quantum computing (LOQC) · Quantum Dot (QD)

1 Introduction In recent years, quantum computing has emerged as very interesting field of research. Lot of researchers all over the world are working in this field to investigate about quantum phenomena and its use for quantum computing. For implementation of quantum computer many approaches have been investigated. Nuclear magnetic resonance (NMR), ion-trap, quantum dot and linear optics are the most efficient and developed techniques [1]. Richard Feynman and Yuri Manin coined the term quantum computing in early 1980s. They explained that certain quantum properties such as Entanglement and Superposition can be used for computation and encryption purpose that is more efficient than classical computing. In 1984, quantum key distribution was developed by C Bennett that is used for cryptography purpose [2]. Then in 1992 first quantum algorithm was developed by Deutch [3]. In 1993, teleportation was demonstrated by Bennett [4]. For integer factorization peter Shor proposed an algorithm in 1994 [5]. Grover proposed an algorithm for data base search in 1996 [6]. Optical simulation of quantum gates was demonstrated in 1998 by Cerf [7]. KLM protocol was developed in 2001 to implement different quantum gates using linear optics [8–10]. Deterministic two Qubit gate using a QD spin was proposed in 2013 by Wang [1, 11]. Time bin Qubit concept was introduced by Takesue in 2018 [12]. Universal Quantum cloners A. K. Sharma (B) · R. Sharma ECE Department, MNIT, Jaipur, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_28

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based CNOT gate was designed in 2019 by Gueddana [13]. Recently, deterministic Fredkin gate for photonic Qubits has been designed and analyzed by considering QD spontaneous emission noise (vacuum noise) and side band leakage [14]. Linear optics quantum computing (LOQC) can be used to design scalable quantum computer using linear components (beam splitter, phase shifter etc.). Photons are used as qubits in LOQC approach so faster quantum communication and decoherence can be achieved. In recent years, significant effort has been done by researchers to implement quantum gates, such as Fredkin, SWAP, Cphase, CNOT, Toffoli gate and so on. However, Gates designed using LOQC approach are probabilistic. For example CNOT gate success probability is 0.25. SWAP gate can be designed using three CNOT gates, so probability of success for SWAP gate is 0.0156. Recent research efforts have shown that cavity quantum electrodynamics (QED) has potential to design deterministic and efficient quantum gate design. In this scheme, the atoms are used as qubits and gate operation are achieved by coupling via interacting with the cavity photon [1]. Figure 1 details the swap gate that is designed using three CNOT gates. Swap gate is universal gate that can be used to design any quantum circuit with help of one Qubit gates. The matrix representation for ideal Iswap gate U is [2] ⎡

1 ⎢0 iSWAP[U] = ⎢ ⎣0 0

⎤ 0 00 0 i0 ⎥ ⎥ i 0 0 ⎦ 0 0 1

(1)

The swap gate is used to swap the Qubits for the given basis |00 ,|01 ,|10  and 11. When swap gate is operated on any basis it will swap the Qubits. Iswap gate is universal quantum gate which can be used with single Qubit gates to implement any quantum circuit or algorithm. The ISWAP gate is designed using resonator as a coupler for Qubits. In this paper to describe our system, the Heisenberg equations of motions are used. The Heisenberg equations of motions [3] Fig. 1 Swap gate cascades three quantum Controlled-Not gates [2]

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√ da ks k = − i(wc − w) + + a − gσ− − ka in dt 2 2  dσ− γ = − i(w X − w) + σ− − gσz a dt 2

 dσz + = 2g σ+ a + a σ − − γ (1 + σz ) dt √ a out = a in + ka 





(2)







(3)





(4)



(5)

where w, wc , w X are the frequencies of incoming photon, cavity and the transition, respectively. g is the coupling strength, k is the cavity field decay rate, and ks is the side leakage rate of the cavity field. γ/2 is the total QD dipole decay rate and γ /2 is the spontaneous emission rate. Iswap gate can be designed by using CNOT gate. Schematic of CNOT gate designed using linear optical component and QD cavity system is shown in Fig. 2 [1]. Initially, two photons 1 and 2 are prepared in states |ψ ph 1 > = α|R1 > +β|L1 > and |ψ ph 2 > = δ|R2 > +γ √ |L2 > and QD electron spin is initialized in state |ψ s > = (|↑s > −|↓s >)/ 2. Photon 1 is control photon and photon 2 is target photon. Half wave plate (HWP) will work as Hadamard gate for right and left circular polarized quantum states. Switch (SW) will control the flow of photons it will first allow photon 1 and after some time delays it will allow photon 2. c-PBS is polarizing beam splitter which transmit right circular polarized photon (|R >) and reflect left circular polarized photon (|L >). Before photon 2 entered in QD cavity a Hadamard

Fig. 2 CNOT gate designed using quantum dot inside double sided optical cavity

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operation is performed on electron spin and after interaction Hadamard operation is applied again.

2 Results and Discussion Dynamics of iSWAP gate designed using cavity system can be found by solving master equation. Although, analytical solution of master equation is very difficult, and quantum optics toolbox in MATLAB can be used to found density matrix of cavity qubit system.   1 + 1 + dρ + = −i[H J C , ρ] + (K + K S ) a ρa − a a ρ − ρa a dt 2 2     1 1 + γ σ − ρσ + − σ + σ − ρ − ρσ + σ − + γ ∗ σ z ρσ z − ρ 2 2 



























(6)

Parameters are same as in Eqs. (1), (2), (3) and (4). HJC is the driven Jaynes– Cummings Hamiltonian. 



√ + + + HJC = (wc − w)a a + (w X − w)σ+ σ− + ig σ+ a − a σ− + i ka in a − a 













(7) To measure the performance of the iSWAP gate, the fidelity has been introduced and defined as [1]. Fidelity is measurement of faithfulness of quantum gate. It is the ability of quantum gate to produce exact output of ideal quantum gate. It can be calculated as defined in Eq. (8) FC N O T = ψ0 |U + ρt U |ψ0 

(8)

where the over line is average over all input states ψ0 , U is the ideal iSWAP gate, and ρt = |ψt  ψt |, with ψt the final state after the iSWAP gate operation. It has been observed from Fig. 3 that Fidelity is strongly correlated with coupling strength “g” and cavity field decay rate “k”. The maximum fidelity F = 54.7% is observed at g = 0.001 and k = 0.005 for iSWAP gate. However for CNOT gate maximum fidelity F = 93.74% has been reported at g = 0.0125 and k = 0.0025 [1]. But to the best of our knowledge the fidelity and concurrence are not available so far in literature. Further, fidelity is also affected by decoherence time of cavity and qubits. At output port of two Qubit quantum gates both qubits must appear at same time. Time delay between two Qubit will produce incorrect output. So concurrence is performance parameter defined for quantum gates which indicate synchronization of output Qubits. It can be seen from Fig. 4 that concurrence also depends on both

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0.5

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0.4 0.45 0.4 0.35

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5 4

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Fig. 3 Fidelity of iSWAP gate versus the coupling strength g and cavity field decay rate k

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Fig. 4 Concurrence of iSWAP gate versus the coupling strength g and cavity field decay rate k

normalized coupling strength “g” and cavity field decay rate “k”. The maximum value of concurrence C = 0.474 is observed at g = 0.02 and k = 0.

3 Conclusions The result reported in this paper shows that coupling strength “g” and cavity field decay rate “k” greatly affect the concurrence and fidelity of iSWAP gate. The desired

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fidelity and concurrence can be achieved by optimizing the system parameters which opens promising prospects for quantum computing and communication. Further decoherence of cavity also effects the fidelity. Decoherence can also be modeled which will provide more practical value of fidelity.

References 1. Wang H-F, Wen J-J, Zhu A-D, Zhang S, Yeon K-H (2013) Deterministic cnot gate and entanglement swapping for photonic qubits using a quantum-dot spin in a double-sided optical microcavity. Phys Lett A 377(40):2870–2876 2. Bennett C, Brassard G (1984) Quantum cryptography: public key distribution and coin tossing. In: Proceeding of IEEE International Conference on Computers, Systems, and Signal Processing, p 175 3. Deutsch D, Jozsa R (1992) Rapid solutions of problems by quantum computation. In: Proceedings of the Royal Society of London, vol 439, pp 553–558 4. Bennett C, Brassard G, Crepeau C, Jozsa R, Peres A, Wootters W (1993) Teleporting an unknown quantum state via dual classical and EPR channels. Phys Rev Lett 70:1895–1899 5. Shor PW (1994) Algorithms for quantum computation: discrete logarithms and factoring. In: Proceeding of 35th Annual IEEE Symposium on Fundamentals of Computer Sciences, pp 124–134 6. Grover LK (1996) A fast quantum mechanical algorithm for database search. In: Proceedings of 28th Annual ACM Symposium on the Theory of Computing, pp 212–219 7. Cerf NJ, Adami C, Kwiat PG (1998) Optical simulation of quantum logic. Phys Rev A 57:R1477-1480 8. Knill L, Milburn (2001) A scheme for efficient quantum computation with linear optics. Nature 409:46–52 9. Glancy S, Lo Secco JM, Vasconcelos HM, Tanner CE (2002) Imperfect detectors in linear optical quantum computers. Phys Rev A 65:062317 10. Kok P, Munro WJ, Nemoto K, Ralph TC, Dowling JP, Milburn GJ (2007) Linear optical quantum computing with photonic qubits. Rev Mod Phys 79:135–174 11. Wang H-F, Zhu A-D, Zhang S, Yeon K-H (2013) Optically controlled phase gate and teleportation of a controlled-NOT gate for spin qubits in a quantum-dot–microcavity coupled system. Phys Rev A 87:062337 12. Lo H-P, Ikuta T, Matsuda N, Honjo T, Takesue H (2018) Entanglement generation using a controlled-phase gate for time-bin qubits. Appl Phys Exp 11:092801 13. Gueddana A, Gholami P, Lakshminarayanan (2019) Can a universal quantum cloner be used to design an experimentally feasible near-deterministic CNOT gate? Quantum Inform Proces 18:221 14. Kang M, Heo J, Choi S et al (2020) Optical Fredkin gate assisted by quantum dot within optical cavity under vacuum noise and sideband leakage. Sci Rep 10:5123 15. Nielsen MA, Chuang IL (2000) Quantum computation and quantum information. Cambridge University Press 16. Wei H-R, Deng F-G (2014) Universal quantum gates on electron-spin qubits with quantum dots inside single-side optical microcavities. Opt Express 22:593–607 17. Hu CY, Munro WJ, O’Brien JL, Rarity JG (2009) Proposed entanglement beam splitter using a quantum-dot spin in a double-sided optical microcavity. Phys Rev B 80:205326 18. Wei H-R, Deng F-G (2013) Scalable photonic quantum computing assisted by quantum-dot spin in double sided optical microcavity. Opt Express 21:17671–17685

Recent Advancement in High Speed and Secure Quantum Key Distribution: A Review Kamal Kishor Choure, Ankur Saharia, Nitesh Mudgal, Manish Tiwari, and Ghanshyam Singh

Abstract The quantum communication system will arguably play a major role in all future application in which quantum key distribution would be an important tool. Quantum key distribution is a quantum cryptography protocol which is getting popularity because of its high speed and secured communication system. Many researchers have proposed novel design and ideas regularly. In this manuscript, we are providing a brief review of the lately schemes proposed by the researchers for high speed and secure quantum key distribution in yesteryears. Keywords Quantum communication · Quantum key distribution · Discrete variable QKD · Continuous variable QKD · Secure key rate

1 Introduction Quantum communication is a modern era of speedy and secured communication where transmission of quantum bits takes place from sender to receiver. In quantum communication, the quantum information is constituted of Qubit in the case of 2D Hilbert spaces. As compared to classical communication, quantum communication provides far more possibility which cannot be easily achieved through classical communication [1–3]. Quantum technologies are still in the developing stage and rising very rapidly to transform conventional communication. Quantum key distribution, which is a quantum cryptography method, permits two distant users to share symmetric keys for their communication, and security of their communication is bound by the basic laws of quantum physics [3, 4]. These laws of quantum physics make unattainable for a third user to measure these quantum states without altering communication and generating errors [5]. The conventional cryptography protocols are not much reliable in terms of security for quantum computing [4]. QKD enables

K. K. Choure (B) · A. Saharia · N. Mudgal · G. Singh Department of ECE, Malaviya National Institute of Technology, Jaipur, Rajasthan, India M. Tiwari Department of ECE, Manipal University, Jaipur, Rajasthan, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_29

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secure communication by sharing random encoded secured keys through transmitting single photons [6] and uses securely shared bit strings for the cryptography process like encrypted secure communication and message authentication [7, 8]. The wheel for QKD starts rolling in the 80s, and the earliest set of rules for quantum key distribution introduced in 1984 [1] ignited the continuous development on the QKD channels: optical fibre and free-space communication in last few decades. Some of the popular QKD protocols are the BB84 [1], the E91 protocol [2], the BBM92 [9], the COW protocol [10, 11], the six-state protocol (SSP) [12], time phase coding scheme [12, 13] and the DPS protocol [14, 15]. In an optical fibre, polarization variation is the main concern during propagation which is compensated by QKD through quantum states in time phase domain. There are several other important protocols for the prevention of insecurities in devices: measurement device-independent (MDI) QKD, decoy QKD and other advance proposed schemes of QKD for long-distance quantum communication [16–20]. Many of the countries like China, Switzerland and South Africa have tested QKD application in optical fibre communication network [21–23], and China has also established a 2000 km long Beijing-Shanghai QKD link network [24]. Researchers have also proposed QKD for long-distance intercontinental quantum satellite communication [25–27]. In this paper, we have tried to summarize and review the recent advancement done by the several researchers across the globe for the quantum key distribution system on parameters of speed of secure key rate (speed), transmission range and stability.

2 Basic Principle of QKD The QKD protocols are bound by the quantum mechanics laws that provide the unconditional security to the QKD system. Figure 1 represents the principle of QKD in a simple way. Here the QKD scheme is deployed to make secure transmission between the Alice (sender) and the Bob (receiver), while Eve (eavesdropper) is third person. The information transferred between Alice and Bob is in the mode of quantum bits via quantum link. The quantum link can be an optical fibre cable or the free-space link. There is one more channel, i.e. public channel which gives information about the process of qubit transmission. The sender Alice transmits the information along with secret key generated from quantum states which were shared with the Bob at the receiver end. In between the communication of Alice and Bob if any eavesdrop activity occurred by the Eve, it can be easily detectable [28]. Basically two types of QKD protocol are—the discrete variable quantum key distribution protocol (DVQKD) and the continuous variable quantum key distribution protocol. Among both, the protocol the DV-QKD was the first QKD protocols which were introduced earlier than the other. The first DV-QKD was BB84 where encoding is done through the single-photon polarization; while in continuous variable QKD, encoding is done through photon packets with the help of Gaussian modulation technique using the position or momentum quadrature of coherent quantum states. To overcome the technological limits of DV-QKD, the CV-QKD was introduced [28].

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Fig. 1 Block diagram of QKD system. Reproduced from Nurhadi and Syambas [28]

2.1 Discrete Variable Quantum Key Distribution Discrete variable quantum key distribution was the first QKD protocol. In DV-QKD system, single-photon polarization is used to send the information to the receiver. The polarization of the single-photon is done through the two bases that result in four polarization state. The two bases are rectilinear basis corresponding to horizontal and vertical polarization, and the other one is a diagonal basis which to diagonal polarization of the photon. At the receiver end, Bob tried both the bases randomly to extract the information. The basic information is shared over the quantum channel, and the processing of the secure key is shared over the public channel. Figure 2 explains the principle of DV-QKD more simply. The quantum states are non-orthogonal which makes the eavesdropper not feasible. The quantum states generated at the transmitter are sent to the receiver. If the receiver chooses the correct basis as send by transmitter, then the information can be retrieving, and while if at receiver chooses the any other basis, it results in difference in information [29].

2.2 Continuous Variable Quantum Key Distribution In the CV-QKD system, the data encoding is done through the Gaussian modulation to make it as continuous variables. Gaussian modulated continuous variable states from

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Fig. 2 Block diagram of DV-QKD system. Reproduced from Gyongyosi and Bacsardi [29]

the Alice are sent to Bob over the quantum channel. At the receiver, Bob detected the information through the homodyne or heterodyne detection. After the measurement done at the homodyne and the heterodyne detection, the parameter estimation is done by Alice and Bob to retrieve the information. Through reconciliation process, privacy amplification and error correction are implemented for the Gaussian data, Fig. 3 explains the principle of CV-QKD more simply [29].

Fig. 3 Block diagram of CV-QKD system. Reproduced from Gyongyosi and Bacsardi [29]

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3 Challenges in Quantum Key Distribution Besides the many advantages and advancement of QKD, still there are several challenges to be deal with, for an ideal quantum key distribution system. These challenges are at both the hardware and the software level. Some of the main challenges are achieving the high key rate, large transmission distance range and to provide high compact, robust quantum key distribution system at low cost [30].

4 Recent Advancement in Quantum Key Distribution to Overcome the Challenges • Philip Sibson et al. have proposed and demonstrated silicon photonics-based integrated device for high-speed quantum key distribution. The silicon photonicsbased integrated device not only enhances the physical ability, but also helps for high-speed modulation. They have tried to overcome the limitation of highspeed modulation of quantum states with the introduction of silicon photonics. They have achieved 1 GHz clock-rate, 1.1% quantum bit error rate, 329 kbps estimated asymptotic secret key rate after the successful implementations on protocol time-bin encoded BB84 state preparation and measurement, polarization encoded BB84 and 1.72 GHz clock-rate, 1.01% quantum bit error rate, 916 kbps estimated asymptotic secure key rate pulse modulation for COWQKD to cover a distance of 20 km long fibre. The proposed integrated chip has been successful in providing the high speed, less complexity and robustness to the system [31]. • Zhen Qu and Ivan B.Djordjevic have proposed a high speed and secure mechanism on the basis of Kramers–Kronig scheme for the free-space optical continuous variable quantum key distribution system. They have proposed a novel approach of Kramers–Kronig scheme to achieve the increased speed and successfully experimentally got the secure key rate for eight state continuous variable quantum key distribution protocols and 10-channels wavelength division multiplexing. They secure key rate of 2.1 Gb/s which is achieved experimentally at the mean transmittance in presence of less turbulence and experimentally verified that 0.6 minimum transmittance is required for the guaranteed secure transmission [32]. • Alasdair B. Price et.al. use the integrated photonics to get a compact and practical solution for high-speed QKD with wavelength division multiplexing (WDM). Through the proposed scheme, they not only got the fast key rate, but also fulfil the requirements of network user through system flexibility. Through the experiment demonstration, authors have achieved the 1.11 Mbit/s of secret key rate for 20 km long mimic fibre, using a couple of transmitter and receiver pairs in the system [33]. • Heasin Ko et.al. have proposed a model to eliminate the side channel effects due to the multiple lasers and experimentally getting the improved key rate from 856 kbps to 1.037 Mbps with the help of the temporal filtering technique [34].

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• Fabian Laudenbachet.al. have demonstrated a reception method for CV- QKD stand on pilot-assisted coherent intradyne with the true local oscillator. In their approach, a pilot tone of strong optical amplitude is taken into consideration. The quantum data is multiplexed in both frequency, and polarization due which the higher symbol rate can be achieved through the proposed scheme. Experimentally, their approach gives the symbol rate of 250 Mbps and with a low excess noise cover up to 40 km of transmission distance [35]. • Qiong Li et al. have proposed a novel approach for high-speed discrete variable quantum key distribution applying a fast Fourier transform (FFT) based on adaptive field-programmable gate array for privacy amplification. In this proposed model, the authors designed a remodel 2D fast Fourier transform-based Toeplitz privacy amplification hardware which helps to reduce the complexity of the system. On the other hand, to improve the processing speed, a real-value-based FFT acceleration process and a fast read/write balanced matrix transposition method are used. The authors experimentally got the throughput of 116 Mbps (input block length n = 1 M) through this proposed scheme, and the adaptivity property of this scheme to compression ratio also opens its gate for many of the quantum key distribution [36]. • Huaxing Xu et al. have proposed a scheme to provide the stability which is immune to an environmental disturbance in the quantum channel, high efficiency and high speed in terms of key rate of QKD using photonic integrated phase decoder. To achieve the high key rate, stability and the high-efficiency author’s uses sagnac configuration-based orthogonal polarizations exchange reflector Michelson interferometer. This proposed scheme is easily practical realizable, and all the optical components can be easily fabricated which makes it very attractive for use in quantum key distribution application [37]. • Ririka Takahashi et al. designed a key management mechanism for high-speed quantum key distribution system. From their proposed mechanism, they got the satisfactory throughput of 414 Mb/s for the local key manager, 185 Mb/s for OTP tunnel manager, 85 Mb/s for global key manager and more than 900 Mb/s for key providing web API, by implementing this key management mechanism in software and evaluating by emulating quantum key distribution key generation [38]. • Heasin Ko et al. proposed a free-space BB84 QKD system with high key rate, with the help of the miniaturized silicon integrated chip and the micro-opticsbased module. Here, they have made the use of effective noise filtering systems in the temporal domain, spectral domain and spatial domain in their developed model and experimentally got the 701.22 kbps secure keys rate in the free-space medium covering distance 275 m in daylight [39]. • Wei Geng et al. have proposed silicon photonic transceiver-based stable quantum key distribution system. They have proposed a time-bin protocol through which the system is getting stable as well as the secure key rate is also increasing. Experimentally, they have shown that with use of silicon photonic transceiver, the secure speed of 85.7 kbps is achieved at a quantum bit error rate of 0.84% for

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covering a mimic fibre of 20 km under slow-varying reference frame misalignment [40]. • Zhang et al. have proposed a silicon chip-based model for CV-QKD. They have tried to incorporate all the optical components on the silicon photonic chip through which they are getting a secured speed of 0.14 kbps for a distance of 100 km in fibre. The proposed system is stable, smaller in size, low cost and easy to install with fibre which made it possible for various quantum key distribution system [41].

5 Conclusion In this paper, we have given a short review of the recent advancement of the QKD. After explaining the general working of the QKD system, the main focus of the paper was to summarize the recent development done by the researchers to gain the highspeed secure key rate along with the increasing distance of transmission. Through this review, we have found that a great job has been done not only on the integrated silicon photonic chip, but also on some protocol level schemes which provide the compactness and robustness to the system that will help in many QKD applications.

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13. Bruss D, Erdelyti G, Meyer T, Riege T, Rothe J (2007) Quantum cryptography: a survey. ACM Comput Surv 39(2), Article 6 14. Waks E, Takesue H, Yamamoto Y (2006) Security of differential- Phase-Shift quantum key distribution against individual attacks. Phys Rev A 73 15. Scarani V, Acin A, Ribordy G, Gisin N (2004) Quantum cryptography protocols robust against photon number splitting attacksfor weak laser pulse implementations. Phys Rev Lett 92 16. Lo H-K, Ma X, Chen K (2005) Decoy state quantum key distribution. Phys Rev Lett 94(23):230504 17. Wang X-B (2005) Beating the photon-number-splitting attack in practical quantum cryptography. Phys Rev Lett 94(23):230503 18. Lo H-K, Curty M, Qi B (2012) Measurement-device-independent quantum key distribution. Phys Rev Lett 108(13):130503 19. Liao S-K, Cai W-Q, Liu W-Y, Zhang L, Li Y, Ren J-G, Yin J, Shen Q, Cao Y, Li Z-P, Li F-Z, Chen X-W, Sun L-H, Jia J-J, Wu J-C, Jiang X-J, Wang J-F, Huang Y-M, Wang Q, Zhou Y-L, Deng L, Xi T, Ma L, Hu T, Zhang Q, Chen Y-A, Liu N-L, Wang X-B, Zhu Z-C, Lu C-Y, Shu R, Peng C-Z, Wang J-Y, Pan J-W (2017) Satellite-to-ground quantum key distribution. Nature 549(7670):43–47 20. Lucamarini M, Yuan ZL, Dynes JF, Shields AJ (2018) Overcoming the rate–distance limit of quantum key distribution without quantum repeaters. Nature 557(7705):400–403 21. Chen T-Y, Liang H, Liu Y, Cai W-Q, Lei Ju, Liu W-Y, Wang J, Yin H, Chen K, Chen Z-B, Peng C-Z, Pan J-W (2009) Field test of a practical secure communication network with decoy-state quantum cryptography. Opt Express 17:6540–6549 22. Stucki D, Legré M, Buntschu F, Clausen B, Felber N, Gisin N, Henzen L, Junod P, Litzistorf G, Monbaron P, Monat L, Page J-B, Perroud D, Ribordy G, Rochas A, Robyr S, Tavares J, Thew R, Trinkler P, Ventura S, Voirol R, Walenta N, Zbinden H (2011) Long-term performance of the SwissQuantum quantum key distribution network in a field environment. New J Phys 13:123001 23. Mirza A, Petruccione F (2010) Realizing long-term quantum cryptography. J Opt Soc Am B 27:A185–A188 24. Qiu J (2014) Quantum communications leap out of the lab. Nature 508:441 25. Bedington R, Arrazola JM, Ling A (2017) Progress in satellite quantum key distribution. npj Quantum Inf 3:30. https://doi.org/10.1038/s41534-017-00315 26. Bonato C, Tomaello A, Da Deppo V, Naletto G, Villoresi P (2009) Feasibility of satellite quantum key distribution. Published 30 April 2009 IOP Publishing and Deutsche PhysikalischeGesellschaft 27. Liao S, Yong H, Liu C et al (2017) Long-distance free-space quantum key distribution in daylight towards inter-satellite communication. Nat Photon 11:509–513. https://doi.org/10. 1038/nphoton.2017.116 28. Nurhadi AI, Syambas NR (2018) Quantum key distribution (QKD) protocols: a survey. In: 2018 4th international conference on wireless and telematics (ICWT), Nusa Dua, pp 1–5. https:// doi.org/10.1109/ICWT.2018.8527822 29. Gyongyosi L, Bacsardi L, Imre S (2019) A survey on quantum key distribution. Infocomunn J XI(2) 30. Diamanti E, Lo HK, Qi B et al (2016) Practical challenges in quantum key distribution. npj Quantum Inf 2:16025. https://doi.org/10.1038/npjqi.2016.25 31. Sibson P, Kennard JE, Stanisic S, Erven C, O’Brien JL, Thompson MG (2017) Integrated silicon photonics for high-speed quantum key distribution. Optica 4:172–177 32. Qu Z, Djordjevic IB (2018) High-speed free-space optical continuous variable-quantum key distribution based on Kramers–Kronig scheme. In: IEEE Photon J 10(6):1–7, Art no. 7600807 33. Price AB, Sibson P, Erven C, Rarity JG, Thompson MG (2018) High-speed quantum key distribution with wavelength-division multiplexing on integrated photonic devices. In: Conference on Lasers and Electro-Optics (CLEO), San Jose, CA, pp 1–2 34. Ko H, Choi B-S, Choe J-S, Kim K-J, Kim J-H, Ju Youn C (2018) High-speed and highperformance polarization-based quantum key distribution system without side channel effects caused by multiple lasers. Photon Res 6:214–219

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Miniaturization and Gain Enhancement of Rectangular Patch Antenna Using CSRR Shubhangi Palekar and Neeraj Rao

Abstract Rectangular patch antennas are widely used microstrip antennas. In this paper, a 45 mm × 45 mm rectangular patch antenna is designed on FR4 substrate with height 1.6 mm resonating at frequency 8.6 GHz. A periodic array of left handed split ring resonators is embedded with this antenna. The return loss, radiation patterns, and input impedances are studied. The gain in both cases with and without metamaterial surface antenna is compared. It is observed that when the antenna is loaded with metamaterial structure, there is a gain enhancement of 1.02 dBi. Also the resonance frequency is down shifted, and miniaturization is obtained by 9.5%. The antenna can be used for X band applications like radio frequency and ranging (RADAR) and satellite communication. Keywords Microstrip patch antenna · Split ring resonator (SRR) · Gain · Antenna miniaturization

1 Introduction Microstrip antennas are widely used in wireless communications. They are simple and inexpensive to manufacture, low profile, and easy to integrate with microwave devices. They show versatility in resonance frequency and radiation pattern when patches with different shapes are selected. However, there are some disadvantages of microstrip antennas. They are low efficient have narrow bandwidth and generate surface waves. Surface waves absorb some power of total available radiation power, and thus, there is degradation of antenna performance in terms of gain, bandwidth, and radiation pattern [1, 2]. Some techniques have been proposed to enhance the bandwidth and gain of antenna. Enhancement of gain and bandwidth of rectangular microstrip antenna by loading slots is presented in [3]. Multiple substrate layer is one of the technique S. Palekar (B) · N. Rao Department of Electronics and Communication VNIT, Nagpur, India N. Rao e-mail: [email protected] © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_30

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improve to gain [4]. A better impedance matching and higher gain of (DRA) dielectric resonator antenna with metasurface lens (ML) are presented in [5]. A rounded bow tie antenna is embedded with seven slabs of 2 × 7 split ring resonator (SRR) unit cells. This SRR is analogous to metalens as they act as mu near zero (MNZ) media. Using this SRR array, there is gain enhancement of 5–6 dBi compared to conventional bow tie antenna [6]. Gain enhancement of microstrip patch antenna using negative index metamaterial (NIM) is presented in [7]. Electromagnetic band gap (EBG) structure is another method to enhance the gain of microstrip antenna. Electrons cannot occupy certain energy bands in EBG. Those are called forbidden bands. Electromagnetic waves, having frequency inside the forbidden band, cannot propagate through EBG structure. Mushroom-like structure is one of the EBG structure. In mushroom type EBG structures, metallic patches are connected to the ground through cylindrical vias. The EBG structures without vias are called as uniplanar compact electromagnetic band gap (UC-EBG) [8–12]. Metamaterials are man-made materials with electromagnetic properties not easily found in nature. Left handed metamaterials show negative ε (permittivity) and negative μ (permeability) and thus negative RI (Refractive index). The size of periodic array of metasurface is much less than the wavelength of incident radiation. Metamaterial properties of structure depend on their geometry and orientation and not on the materials used for the design [13–16]. In this paper, the gain of rectangular patch antenna with and without metamaterial surface is compared. Observing the simulation results, we can say that there is gain enhancement of 1.02 dBi with metasurface loaded compared to conventional patch antenna. Sections 2 and 3 describe the antenna design. Simulated results are shown in Sections 4 and 5 is about the conclusion.

2 Antenna Design A microstrip antenna consists of three parts. A radiating patch is mounted on a dielectric substrate over a metallic ground. The resonance frequency of interest here is 8.6 GHz. Substrate used for designing is FR 4 (lossy) with dielectric constant 4.3 and loss tangent 0.025. Height of substrate is 1.6 mm. The dimensions of rectangular patch are calculated by the following equations [16].  c ω= 2 fr

2 εr + 1

(1)

  1 2h − 2 εr + 1 εr − 1 εeff + 1+ 2 2 ω L = 0.412h

εeff + 0.3 εeff − 0.258

h ω

+ 0.264 h ω

+ 0.8

(2)

(3)

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Fig. 1 Conventional patch antenna

L eff = L + 2L

(4)

where w = width of patch L = length of patch h = height of substrate L = extension in length εr is dielectric constant of substrate and εeff is its effective value. The length and width of patch are found to be 10.714, 7.719 mm, respectively. Width of microstrip line is 3.13 mm. It is excited with 50  waveguide port, and the antenna is tuned to resonate at frequency 8.6 GHz. 45 mm × 45 mm × 1.6 mm size antenna is shown in Fig. 1. The scattering parameters and radiation pattern of antenna are studied with simulation results.

3 Split Ring Resonator Design Split ring resonators are symmetrically placed on the substrate of dielectric constant 2.2 and height 0.508 mm. This left handed structure is referred from ‘METAMATERIAL’ by Jiang [15]. Top view and bottom view of SRR are shown in Figs. 2 and 3,

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Fig. 2 Top view of SRR

Fig. 3 Bottom view of unit SRR

respectively. This structure is simulated with CST solver, and its ε and μ are extracted from scattering parameters. Figures 4 and 5 describe S parameters and metamaterial property of structure. It resonates at frequency 8.6 GHz and shows negative ε and negative μ over frequency range 8.45–8.6 GHz. This unit cell is periodically repeated. An array of 9 × 5 such unit cells is placed over a ground at height of 0.508 mm. A rectangular patch is placed above this

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Fig. 4 S parameters of CSRR structure

Fig. 5 Metamaterial property of CSRR showing negative ε and μ

metasurface at height 1.6 mm above the ground. Antenna embedded with metasurface is shown in Figs. 6 and Fig. 7. Exciting it with 50  microstrip line, simulation results are observed.

274 Fig. 6 Metasurface at height 0.508 mm above the ground

Fig. 7 Patch at height 1.6 mm above the ground

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Fig. 8 Comparison of S parameters of antenna with and without metamaterial

4 Simulated Results 4.1 Scattering Parameter The return loss of conventional patch antenna and metasurface loaded antenna are plotted in Fig. 8. Antenna without metamaterial resonates at 8.6 GHz with reflecting coefficient −29.17 dB. After loading metamaterial, it resonates at frequency 7.78 GHz with reflection coefficient −24 dB. Thus, there is downshift in resonance frequency, and antenna is miniaturized by 9.5%

4.2 Radiation Pattern and Input Impedance 3D radiation patterns of the conventional patch antenna and metasurface loaded antenna are compared in Figs. 9 and 10. Gain of conventional patch antenna is 3.92 dBi at 8.6 GHz and that of metamaterial loaded antenna is 4.94 dBi at 7.8 GHz. This shows that there is enhancement in gain by 1.02 dBi. Gain curves of both antennas are compared in Fig. 11. Radiation efficiency of an antenna signifies the total radiated power out of total supplied power. It is the ratio of gain to the directivity [16] Radiation efficiency(η) = Gain/Directivity

(5)

When gain and directivity are measured in dBi, efficiency (η) becomes, η(dB) = Gain(dBi) − Directivity(dBi)

(6)

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Fig. 9 3D radiation pattern of conventional antenna

Fig. 10 3D radiation pattern of metasurface loaded antenna

when directivity of an antenna is more than its gain, the radiation efficiency comes out to be a negative value. It is made sure that the antenna and port are matching at 50  to transfer maximum power and avoid voltage standing waves. Variation of input impedance of antenna with frequency is shown in Figs. 12 and 13 with the help of smith chart. Real value of input impedance of antenna signifies that the voltage and current are in phase, and

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Fig. 11 Comparison of antenna gain with and without metamaterial

Fig. 12 Impedance view of conventional antenna

the antenna is resonant. Imaginary part of input impedance represents non-radiated power stored in the near field of antenna [16]. From figures, it is clear that the input impedance of antenna is approximately matched to the input impedance of transmission line with real value close to 50  and imaginary value near to 0 .

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Fig. 13 Impedance view of metasurface loaded antenna

5 Conclusion CST is a helpful tool for designing of antennas. In this paper, microstrip patch and SRR are designed with two different materials to enhance the parameters of antenna and improve its performance. The simulation results show that antenna with metamaterial surface shows enhancement in gain by 1.02 dBi over conventional patch antenna. The resonating frequency is down shifted, and miniaturization of antenna is obtained by 9.5%. The designed antenna is applicable in X band for satellite communication and radio detection and ranging (RADAR) applications. Acknowledgements We would like to thank Dr. Dinesh Kumar faculty member of microwave and communication group for allowing us to use CST microwave studio.

References 1. Chen Z (2018) A new H-slot coupled microstrip filter-antenna for modern wireless communication systems. 978-1-5386-1851-6/18/31.00 2018©IEEE 2. Deshmukh AA (2016) Space fed ring microstrip antenna array with stacked rectangular microstrip antenna feed. 978-1-5090-3646-2/16/31.00 ©2016IEEE 3. Prasad RK (2016) Gain and bandwidth enhancement of rectangular microstrip antenna by loading slot. In: 2016 1st international conference on innovation and challenges in cyber security (ICICCS 2016) 4. Ineneji CN (2015) Gain enhancement in microstrip patch antenna using the multiple substrate layer method. 978-1-4799-4874-1/14/31.00 ©2015 IEEE 5. Kannan K (2017) Boresight gain enhancement of a dielectric resonator antenna using a metasurface lens. 978-1-5386-0646-9/17/31.00 ©2017 IEEE

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6. Dadgarpour A (2016) Gain enhancement of planar antenna enabled by array of split-ring resonators. IEEE Trans Antennas Propag 64(8):3682–3687 7. Singh AK (2017) A negative index metamaterial lens for antenna gain enhancement. 978-15386-0465-6/17/31.00 ©2017 IEEE 8. Helena Margaret D, Subasree MR, Susithra S, Keerthika SS, Manimegalai B (2014) Comparison of compact EBG structures on the mutual coupling reduction of antenna arrays. Int J Fut Comput Commun 3(2):76–79 9. Zhang J (2017) A wide band-gap slot fractal UC-EBG based on moore space-filling geometry for microwave application. IEEE Antennas Wirel Propag Lett 16:33–37 10. Martn F (2003) Split ring resonator-based left-handed coplanar waveguide 11. Jafargholi A (2019) Mutual coupling reduction in an array of patch antennas using CLL metamaterial superstrate for MIMO applications. IEEE Trans Antennas Propag 67(1):179–189 12. El Ayachi M (2008) Planar low profile and gain enhancement of printed antennas using EBG structures. 978-1-4244-2042-1/08/25.00 ©2008 IEEE 13. Gudarzi A (2012) Gain enhancement and miniaturization of microstrip antennas using MTM superstrates. 978-1-4673-0479-5/12/31.00 ©2012 IEEE 14. Denidni TA (2014) Stepped impedance resonator technique for metamaterial miniaturization. 978-1-4799-2225-3/14/31.00 ©2014IEEE 15. Jiang X-Y (2012) METAMATERIAL 16. BalanisC. Antenna theory analysis and design, 2nd edn.

Preventing DoS Attack in VANET by Novel RBS-IP-CHOCK Model Karan Verma, Hemant Kumar Saini, and Ajay K. Sharma

Abstract Vehicular ad-hoc networks (VANETs) have suffered from many risks in the past, such as security privilege and authentication because attackers/hackers always try to disturb the network and break the communication services. The series of attacks that have broken secured communication are network jammer, source sink, and MAC crack, and these attacks are on as denial of service (DoS) attacks. The proposed method is based on the master chock filter concept for the filtration of packets during busy traffic. The IP-CHOCK protocol is implemented along with routing protocol employed in the VANET. The performance of the proposed protocol was evaluated using network simulator NS-2.34. The protocol includes blocking the source IP originator by the DoS attacks and checking the prevention of TCP/UDP flooding and IP sniffing attacks. The evaluation of the protocol was based on the mobility node’s interaction and utilization of bandwidth. The simulation results were analyzed with two different scenarios; first scenario includes 20 mobile nodes and second scenarios with 50 mobile nodes. These two scenarios were performed on the highway transportation condition. The results showed that the throughput, packet delivery ratio, packet loss time, and packet receiving ratio have improved as compared to random IP-trackback protocol. Hence, indicates that the proposed protocol is efficient for intelligent transportation system in VANET. Keywords Reference broadcast synchronization (RBS) · Ad-hoc on-demand distance vector (AODV) · Denial of service (DoS) attacks · Internet protocol (IP)

K. Verma (B) National Institute of Technology Delhi, New Delhi, India e-mail: [email protected] H. K. Saini Modern Institute of Technology and Research Centre, Alwar, India A. K. Sharma National Institute of Technology Jalandhar, Jalandhar, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_31

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1 Introduction RBS using IP-CHOCK technique, which is used to synchronize a set of receivers with one another. In this model, nodes broadcast reference beacons to their neighbors. The synchronize feature used in the ensemble was incoming traffic volume, number of source IP addresses [1], and the incoming new source IP addresses. Using the RBS using IP-CHOCK technique is to find reference and without reference nodes, to generate a wide variety of attacks and benign traffic origination from a wide range of valid IP addresses. The RBS using IP-CHOCK has accurately estimated the prevention of DoS attacks due to its stable reference broadcast synchronization process.

2 RBS Using IP-CHOCK Model The simplest form of RBS is one broadcast beacon and n receivers. The timing packet is broadcasted to receivers n which record the time the packet was received as per local clocks. Two receivers then exchange timing data which enables the calculation of the offset, thus rendering sufficient data to retain the local timescale. The RBS is then expanded from its simplest form to broadcasting to n receivers for synchronization between n receivers, but where (> n) may require more than one broadcast. Increased broadcast increases the precision of the synchronization. RBS differs from the traditional sender–receiver synchronization by using receiver-to-receiver synchronization. The reference beacon is broadcasted across all nodes, and once received, the receivers note the local time and exchange timing data with neighboring nodes after which these same nodes are able to calculate their offset (see Fig. 1) • A transmitter broadcasts a reference packet to receivers. • Each receiver records the local time of reception. • Receivers exchange respective observations. The precision of synchronization can be increased by sending more than one reference broadcast: • • • •

A transmitter broadcasts m reference packets. Each of the n receivers records the local time of reception. Receivers exchange respective observations. Each receiver I computes its phase offset to any other receiver j as the mean of phase offsets implied by each pulse received by both nodes, i and j. That is, given ∀i ∈ n, j ∈ n : offset[i, j] =

m  1  T j,k − Ti,k m k=1

(1)

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Fig. 1 Flowchart showing detection scheme using proposed reference broadcast synchronization (RBS) system

where n: the number of receivers, m: the number of reference broadcasts, and T(r,b) : r  s clock on reception of broadcast b. This basic scheme does not account for clock skew (0), for which instead of averaging phase offsets for multiple observations; RBS performs a least-squares linear regression. This offers a fast, closed-form method of finding the best-fit line through phase-error observations over time. The frequency and phase of local clock nodes with respect to the remote node can be recovered from the slope of the line and its intercept with the y-axis. The fundamental property of RBS is that a broadcast message is used to synchronize a set of receivers with each another. Doing so eliminates both sending and access times from the critical path as these parameters are typically the greatest source of error and largest contributors to latent non-determinism. In addition, the minimized operating system modification that reads a clock interrupt at reception time is optimized. Therefore, the critical path length in RBS only includes time from injection of the packet into the channel to the last clock’s reading it. As depicted in Fig. 2, RBS is only sensitive to the difference in propagation time between a pair of receivers [2]. In this method, a RBS-based IP-CHOCK (master filter) sent by the node is called the reference node. And those nodes accepting the joint query from the reference node generate tokens with the help of the chock filter.

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Timescale

RBS

Fig. 2 Comparison of traditional synchronization systems to RBS

1. 2.

3.

4.

Receivers compare respective clocks and calculate relative phase offsets. This timing is based on when the node received the reference beacon. The timing packet is broadcasted to receivers who record its reception time on their local clocks. Receivers then exchange their timing data and calculate the offset. If the measured time interval is within the range of offset values, the next hop node is considered legitimate. In cases where the time interval exceeds the offset value, the next hop node is set aside as malicious. Each token has an offset value, clock, and distance.

New schemes to reduces false alarm rate in chock filters For the process of false alarm reduction, a vector method was designed to count changes of (M). The value of the M-Counter is stored in the vector with the index value of an incoming IP address. The value of the suspicious alarm threshold generated by the maximum hop count time duration threshold is basically a check-point for the generation of the alarm. The value of the alarm is estimated by subtraction. The size of the RBV vector is simply subtracted from the size of the matrix, representing the maximum limit for acceptance of a packet. After obtaining the threshold value (δi ), a counter check is done for the maximum number of frequent changes for ‘turn-ons’ and turn-offs.’ During this period, the maximum change of frequent values for the M—counter is computed, and a suspicious alarm for the spoofed packet is generated [3]. • Frequent Vector Value When the counter value of filter (M) is turned on and off, the frequency of counting (e) begins and it becomes a necessity to insert (e) into the RBV data structure. When the value of (e) generates an alarm message, it is then removed from the RBV [4] data structure (see Fig. 3).

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Fig. 3 Flowchart of the ‘insert and remove’ mobile packet using reference broadcast synchronization (RBS)

Let us assume that: e = the frequent change value of the M—counter and H(x) = the index of incoming and outgoing IP addresses. RBV = reference CHOCK vector. T = time duration hop of the frequent counter value. SA = Suspicious alarm. To generate the value of the suspicious alarm: SA =

F(e)  i=0

(h f (e) × ti )

(2)

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Now, the spoofed packet is captured, and a suspicious alarm is generated; it then removes the frequent element (e) value from the RBV vector. The vector value is a bit value in the IP-CHOCK filter of the proposed model. This value shows attack variation in a network. Initially, the vector value is (0), and the change in the reference node reaction value is (0). The frequency of 0 and 1 is counted. The behavior of the mobile node in VANET is measured. This IP-CHOCK is a hybrid model of the well-known reference broadcast method (RBS) and clock synchronization. In addition, it uses symmetric key cryptography for data privacy. Algorithm 1 : Insert mobile node into RBS approach input: RBV, e, T output: for each incoming mobile packet node of counter e 1 for i Devices > Functions > Variables. System hierarchy is shown in Fig. 2. For home automation system, the devices could be switches, Variac, sensors, etc. The functions are switching, speed limit alert, automatic speed control by sensing sensor value, reading the data, threshold alert, etc. The variables would be value of the sensors.

3.2 Gateway Gateways are registered on the Web platform. It is inbuilt software and hardware of the system which is directly connected to the Internet. For inbuilt gateway, wireless and wired connection module is required. Single gateway platform is shown in Fig. 3. In multisystem gateway, multiple systems will be connected to Internet. The communication between system and gateway is done by RF or Zigbee or Wi-Fi or any such protocol.

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Fig. 2 System hierarchy

Fig. 3 Gateway in the System

As shown in Fig. 4, the separate gateway has three hardware modules: (1) Wi-Fi module, (2) RF/Zigbee module, and (3) controller. The system developer and gateway developer companies can be different in this situation. Gateway is registered on the IoT platform. The main goal of gateway is to verify system and pass the data between IoT platform and system. Fig. 4 Separate gateway

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Fig. 5 IoT platform

3.3 IoT Platform IoT platform will collect data from all the gateways and store it in its own database. Different IoT platforms are available in market (Exosite, Carriots, IBM IoT, etc.). IoT platform provides easy way to manage and communicate with gateways. Gateway manufacturer must have to register their gateway product (inbuilt as well as separate) on the supported IoT platform [15]. Different protocols are used in IoT for communication between Web and gateway, and the representation is defined in Fig. 5.

3.4 Proposed Common Interface Platform (Madhuboon) In IoT, it is about taking decisions by learning from other devices and to give commands to actuators for controlling the devices. The Madhuboon platform is divided into four sub-blocks: Each block of Madhuboon is explained below.

3.4.1

System Management Web Portal

It is a Web portal for system manufacturer or developer. This would make the platform usable by system (hardware/software) manufacturer, a system manufacturer must have to register their systems on the platform. On the registration they have to provide manufacturer id, secret key, system id, and model number. All these details will be also programmed in hardware. This platform provides facility to make functionality of system customization from user to user (even if hardware is same). As shown in Fig. 6, the application server with MQTT client/REST API client will collect data from IoT platform. This application server will divide the data and save the data on the database according to systems, which are registered earlier on System Management Web Portal. Hence, Madhuboon platform is system-oriented. It saves data according to the systems.

3.4.2

Application Server with MQTT Client or REST API Client

The third-party IoT platform collects data from every gateway, which is registered on the IoT platform. Application server with MQTT or REST API client will collect data from IoT platform and manage according to the systems that are registered on System Management Web Portal. It is kind of communication channel between Madhuboon and IoT platforms.

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Fig. 6 Difference between IoT platform and SMWP

3.4.3

Database

There are two types of databases used in market: (1) SQL and (2) NoSQL. Two methods to host database are (1) database on server and (2) database on cloud. Different platforms like IBM-Cloudant, Google-Cloud, and Mongo DB provides cloud-based database.

3.4.4

Web Server + Application Server

This is front end for the Madhuboon users. Web server will host Web site, while application server will provide different services like database management, request management, push notifications, e-mail services, etc. There are some cloud-based platforms for Web application hosting. Some also provide mobile application support, e.g., IBM Bluemix, Google App Engine, and Amazon Web Services all provide Platform as a Service (PaaS) cloud platform.

4 Specification and System Implementations The functional parts of the whole system is divided in two parts: i.e., hardware part and software part. Hardware part consists of two approaches: (i) controllerbased (MSPF5430 from Texas Instruments, Raspberry Pi) and (ii) FPGA-based (Spartan, Virtex or Zynq) [5]. For software part, there are mainly two approaches: (i) application-based (MQTT, REST API) and (ii) application + Web-based (Amazon

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Fig. 7 System block diagram (physical structure)

Web Engine, IBM Bluemix) [16]. The prototyping system is developed to validate the proposed system with following steps: 1. 2. 3.

System realization by implementing the dedicated hardware. Feature realization by adding the new device/node in the network through gateway. Interfacing using REST API/MQTT protocols with any cloud platform.

The conceptual diagram for the implementation is shown in Fig. 7 which has main components such as hardware, gateway, Interface1 and Interface2, and cloud platform.

4.1 System Realization with Dedicated Hardware for the Gateway The realization of the proposed concept is done using LabVIEW. The main function is to access the data of the devices which are connected to the gateway by using some interface with it, that interfacing is done with help of LabVIEW software. NI LabVIEW framework plan programming coordinates with almost any equipment from any vendor in one environment and spares advancement time with advantageous elements and a steady programming system over all equipment. It also finds the association with everything from stand-alone instruments to secluded stages utilizing LabVIEW to get, examine, and introduce your information in an important and expert way. The software and hardware parameters and features are defined as follows:

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Software Part 1. 2. 3.

IAR Embedded Workbench: For simulation and debugging of MSP430 Microcontroller HKSCADA: For data acquisition and control LabVIEW: For graphical user interface (GUI) and control.

Hardware Part 1. 2. 3.

MSP430 Microcontroller: For protocol development, analog-to-digital conversion (ADC), communication using UARTs, etc. RS232 and RS485 cables, connectors, and supporting hardware Sensors and signal conditioning circuits.

These protocols are used: (1) MODBUS: Application layer protocol used in SCADA for communication among devices to control and to acquit required data. (2) RS232 and RS485: Physical layer protocols in wired communication used in SCADA for communication among devices to control and to acquire required data. In Fig. 8, we have shown the block diagram for the hardware, and the main part of that is MSP430 F5419 controller board which acts as a gateway. Gateway is connected to many other sensors and devices. These devices can be added and removed from the existing network by configuring some specific parameters. By doing this we have done the prototyping of the gateway which is master in the network and provide communication to all other network devices. The output is shown in Fig. 9 from the LabVIEW GUI.

Fig. 8 Implementation of proposed system

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Fig. 9 System realization LabVIEW block diagram

4.2 Steps for LabVIEW Implementation 1.

2. 3. 4. 5. 6.

Create serial interface between LabVIEW and computer’s COM port using LabVIEW VISA function which includes serial parameters like baud rate, resource name, parity check, stop bit, etc. Prepare analog configuration page to configure analog parameters. Parse the analog string from microcontroller according to channel index. Create analog display page to display real-time analog values with their respective tags from microcontroller via serial communication. Display all channels’ analog tags with their respective values in tabular form and log them in spreadsheet. Create graphical chart of all the channels’ analog values.

The block diagram shows how the parameters for the new device are used in the system to generate the output. Result for this will be recognized in the LabVIEW GUI as below. Figure 10 shows that when we want to add new device, in simple case we

Fig. 10 GUI for the device configuration

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Fig. 11 GUI for the output of the configured device

need to enter some parameter values in this GUI which is front end of the LabVIEW. Parameters we need to enter for the particular device are as below: device Id, baud rate, timeout, data bits, stop bits, parity, and input/output. After adding these values, we can start the implementation of the LabVIEW code which will show the values of the sensors connected to the device whose device id is entered in the parameters. As defined in Fig. 11, it shows the output from the device “3002” (as per the device id in Fig. 9) which notifies that the particular device is having three temperature sensors configured with it and they are not in the operating mode means they are in an ideal state. The work done till now is for the access of the IoT node through gateway, now we are going for the important feature in that, inserting the new device (IoT node) in the network through gateway. We have done this through the simulation environment again using the LabVIEW. The main condition is that the new device should be configured and activated in the same communication protocol which is earlier adopted by the hardware network, i.e., here we used I2C to connect the devices with gateway so new device has to be configured and capable of the I2C communication then we will follow the steps shown. As we can see in Fig. 12, the new device block is there, which will take all the required parameters using the GUI (Fig. 10) inserted by user. The IoT block will connect the device with the gateway in firmware manner, i.e., the code will be changed to get the data from the new device. Furthermore, all blocks are to configure the gateway with parameters of new device. Figure 13 is the GUI for the user to enter the new device in gateway configuration, as all parameters are same as when we see the device behavior through the first execution step. By doing the same steps, the performance has been evaluated with different sensors (temperature sensors, humidity sensors, accelerometer) to verify the functionality of the proposed framework.

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Fig. 12 Adding new node (device)—LabVIEW block diagram

Fig. 13 Front end for the addition of new device

5 Conclusion and Future Work The paper presents the common framework for seamless integration of IoT devices. The system can accept new devices by having their configuration added up in the database of the system (memory of the gateway). The usage of publicly shared devices is possible because of the availability of the proposed common platform. Even group

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sharing like family members accessing the home IoT nodes and many more will be much easier by the same framework. The involvement of the vast features of gateway will reduce the complexity of the whole system because of the low power, low hardware complexity because the overall Internet connection of the network will be taken care of by gateway. In addition to that, the software and cloud platform are being made efficient and more reliable. There is future scope to address privacy and security issues for the proposed framework.

References 1. Yadav G, Mangla SK, Luthra S, Rai DP (2019) Developing a sustainable smart city framework for developing economies: an Indian context. Sustain Cities Soc 47:101462 2. Rahman MA, Hossain MS, Hassanain E, Muhammad G (2018) Semantic multimedia fog computing and IoT environment: sustainability perspective. IEEE Commun Mag 56(5):80–87 3. Akan OB, Andreev S, Dobre C (2019) Internet of things and sensor networks. IEEE Commun Mag 57(2):40–40 4. Xu Y, Ren J, Wang G, Zhang C, Yang J, Zhang Y (2019) A blockchain-based nonrepudiation network computing service scheme for industrial IoT. IEEE Trans Ind Inf 15(6):3632–3641 5. Jain P, Joshi AM, Mohanty SP (2020) iGLU: an intelligent device for accurate noninvasive blood glucose-level monitoring in smart healthcare. IEEE Consum Electron Mag 9(1), 35–42 6. Pancholi S, Joshi AM (2019) Improved classification scheme using fused wavelet packet transform based features for intelligent myoelectric prostheses. IEEE Trans Ind Electron 7. Paul PV, Saraswathi R (2017) The internet of things—a comprehensive survey. In: 2017 international conference on computation of power, energy information and communication (ICCPEIC). IEEE, pp 421–426 8. Tang S, Shelden DR, Eastman CM, Pishdad-Bozorgi P, Gao X (2019) A review of building information modeling (BIM) and the internet of things (IoT) devices integration: present status and future trends. Autom Constr 101:127–139 9. Joshi A, Mishra V, Patrikar R (2015) Real time implementation of integer DCT based video watermarking architecture. Int Arab J Inf Technol (IAJIT) 12 10. Singh KJ, Kapoor DS (2017) Create your own internet of things: a survey of IoT platforms. IEEE Consum Electron Mag 6(2):57–68 11. Bellavista P, Cardone G, Corradi A, Foschini L (2013) Convergence of MANET and WSN in IoT urban scenarios. IEEE Sens J 13(10):3558–3567 12. Arasteh H, Hosseinnezhad V, Loia V, Tommasetti A, Troisi O, Shafie-Khah M, Siano P (2016) IoT-based smart cities: a survey. In: 2016 IEEE 16th international conference on environment and electrical engineering (EEEIC). IEEE, pp 1–6 13. Aazam M, Zeadally S, Harras KA (2018) Deploying fog computing in industrial internet of things and industry 4.0. IEEE Trans Ind Inf 14(10):4674–4682 14. Rachakonda L, Mohanty SP, Kougianos E, Sundaravadivel P (2019) Stress-lysis: a DNNintegrated edge device for stress level detection in the IoMT. IEEE Trans Consum Electron 65(4):474–483 15. Mulligan CE, Olsson M (2013) Architectural implications of smart city business models: an evolutionary perspective. IEEE Commun Mag 51(6):80–85 16. Thakral S, Goswami D, Sharma R, Prasanna CK, Joshi AM (2016) Design and implementation of a high speed digital FIR filter using unfolding. In: 2016 IEEE 7th Power India International Conference (PIICON), pp 1–4. https://doi.org/10.1109/POWERI.2016.8077361.

Frequency Hopping Patterns for Low Probability of Intercept (LPI) Radars Using Costas Arrays T. A. N. S. N. Varma and Anjaneyulu Gera

Abstract Low probability of intercept (LPI) radars are used in tracking the targets at low distances by avoiding detection by hostile elements like passive radar detection equipment. Frequency agility and frequency selection are some of the important parameters of LPI radars. The LPI radar typically uses frequency shift keying (FSK) or phase shift keying (PSK). FSK has advantages like large bandwidth and high peak to side lobe ratio in both Doppler domain and autocorrelation domain. The frequency sets of FSK signals which are constructed with continuous wave can be obtained by Costas arrays. With suitable design of the frequency sequence, the Costas waveform will have good range and Doppler resolution properties consistent over the entire signal duration and bandwidth. In the present work, authors have generated Costas arrays (frequency step sequences) for lengths 2–27 and evaluated their performance in both autocorrelation domain and Doppler domain and reported the best sequences suitable for LPI radar applications. Keywords Low probability of intercept (LPI) radars · Costas array · Frequency hopping · Range resolution · Doppler resolution

1 Introduction Radar is an electromagnetic system used to find the presence of target and its range. It operates by radiating energy into space and detecting the echo signal reflected from a target. Low probability of intercept (LPI) radar is a special class of radar systems with certain performance characteristics that make them avoid detection by radar detection equipment like digital intercept receivers or radar warning receivers. There are different methods to reduce the profile of the LPI radar that include ultrawideband, frequency agility or frequency hopping, low power and choosing antennas with low side lobes. The present work is focussed on selecting proper frequency hopping pattern for the job. T. A. N. S. N. Varma (B) · A. Gera Electronics and Communication Engineering Department, MVGR College of Engineering (A), Vizianagaram, AP, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_43

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Costas waveforms are one kind of pulse compression waveforms having aspects of both phase coded and stepped frequency pulse bursts waveforms. A Costas waveform is similar to a polyphase waveform in that it is a single pulse waveform, divided into N sub-pulses. Ideal ambiguity function is of a shape like thumbtack can be obtained by proper design of the frequency sequence. Good range and Doppler resolution properties consistent with overall signal duration and bandwidth can be obtained by Costas arrays. Today these Costas arrays have found applications not only in radar, Sonar but in computer graphics, mobile communications, data mining and for matching patterns of electromagnetic and ferromagnetic surfaces for clamping purposes [1, 2]. These codes are very useful in spread spectrum communication systems, where the objective may be to achieve either jamming resistance or frequency diversity for a selectively fading channel. Some other advantages using these Costas codes are they are individually identifiable at the receiver among a set of Costas waveforms, improve active Sonar performance and its performance is superior to conventional waveforms such as continuous wave and frequency-modulated pulses and the range Doppler side lobe peaks are well controlled so that ideal ‘thumbtack’ ambiguity function behaviour is closely approximated. Selecting a useful permutation is a key deployment concern for suitable application. Open literature on details of waveforms is often scarce because waveforms are frequently proprietary or classified.

2 Formulations To improve the target detection performance of frequency hopping Sonar systems, Costas introduced special permutations in 1965 [3]. Later Gilbert devised construction methods for certain orders. These permutations found applications in RADAR by Golomb in 1984 [4]. Till date, there exist only four algebraic methods for the systematic construction techniques of Costas permutations, known as Welch construction, Lempel construction, Golomb construction and Taylor’s construction [5]. The frequency sequence of a Costas waveform is known as Costas array. A Costas array is a square matrix of size N which can take only two values, ‘0’ and ‘1’. Each row is filled with one ‘1’s and (N − 1) ‘0’s, so for each column also has one ‘1’ and remaining elements as zeros. N rows of an array signify N frequencies spaced f apart, and N columns signify N adjoining sub-pulses (“bits”), each of duration t b . A ‘1’ in the (i, j) element of an array indicates that during the jth time interval, the ith frequency is transmitted [6]. The Costas array for the frequency sequence [2 4 8 5 10 9 7 3 6 1] is shown in Fig. 1. Costas arrays [7] are square arrangements of dots and blanks (or 1s and 0s, respectively) that satisfy the following two constraints:

Frequency (KHz)

Frequency Hopping Patterns for Low Probability of Intercept …

10 9 8 7 6 5 4 3 2 1

403

1 1 1 1 1 1 1 1 1 1 1 2 Time

3

4

5

6

7

8

9

10

Fig. 1 Costas array for the frequency sequence [2 4 8 5 10 9 7 3 6 1]

1. 2.

Every row and column contain exactly one dot: a Costas array is, therefore, a permutation array. All vectors connecting pairs of dots must be distinct.

To date, mainly two approaches (Welch and Golomb) have been followed to identify Costas arrays. These specific constructions provide examples of Costas array for many different values of n. One has to do exhaustive search, to find all N × N Costas arrays to identify the best patterns in every length [8]. In the present work, exhaustive study of Costas arrays has been for lengths N ≤ 27. In the exhaustive search method of producing Costas codes for a given length N. Order the complete set of integers from 1 to N and produce permutations. There will be N! permutations (ordered sequences) for any length N, and for each of such permutation we will compute the difference triangle and check the Costas condition. The difference triangle from the ordered sequence shall have no repeated terms in any row. That is, first row is formed by taking the differences between adjacent numbers. All differences in that row must be unique. Second row is formed by taking the differences between next adjacent terms. This must also be free of repeated values and so forth. Those sequences which satisfied above said Costas conditions are known as Costas sequences or Costas codes or Costas arrays (Fig. 2). Even though, for value N, there are N! possible sequences, only few sequences which satisfy the difference triangle properties are considered to be as Costas sequences. For example, N = 6, there are 6! = 720 possible sequences of values, but only 116 of them will yield a Costas array. For some lengths, we do not find any Costas arrays (e.g. N = 32, 33). The enumeration results show the patterns which satisfy the Costas criteria out of N! permutations (Table 1). Welch Construction To find the Costas signal for a sequence of length N using Welch construction, we use the following procedure.

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Fig. 2 Difference triangle for a N F = 10 Costas sequence

Let p be an odd, prime number. Then N = p − 1. Let g be the primitive root of p. A primitive root of p is defined as the value g such that the sequence of powers g1, g2, g3, … g p − 1 modulo p generates every integer from 1 to p − 1. Note that the primitive root of p is not the same as a prime factor of p or N, though a primitive root may be a factor of N. For instance, the primitive root of 41 is 6, which is neither a factor of p = 41 or of N = 40. For the Costas code of length N, create an N × N matrix, and label the columns and rows, respectively, as follows: j = 0, 1, 2, 3, . . . , ( p − 2) i = 1, 2, 3, 4, . . . , ( p − 1) We will place a ‘1’ or ‘dot’ in the matrix location (i, j) if and only if i = (g)j modulo p. The dot or 1 in location (i, j) as defined above indicates that we will use the frequency fi in that part of the Costas coding sequence, where we define fi as follows: fi = fo + i f ; i = 1, N where fo is some constant frequency and fo >> f . f is obtained from dividing the frequency span of the transmitted radar pulse by N. Ambiguity Function (AF) The ambiguity function (AF) is a significant tool for determining the target resolution capability of a radar system and is used broadly for signal processing in various applications like radio astronomy, sonar, communications and optics. The range and radial velocity of a moving target can be estimated by delay time and Doppler frequency shift, where the range is proportional to the round-trip travel time, i.e. the delay time τ, of the radar signal while the radial velocity is proportional to the Doppler frequency shift ν. In simple terms, it is time response of matched filter to a given finite energy signal received with a delay and Doppler shift [9–12].

Frequency Hopping Patterns for Low Probability of Intercept … Table 1 Enumeration results of Costas arrays for different array sizes

N

405

C(N)

1

1

2

2

3

4

4

12

5

40

6

116

7

200

8

444

9

760

10

2160

11

4368

12

7852

13

12,828

14

17,252

15

19,612

16

21,104

17

18,276

18

15,096

19

10,240

20

6464

21

3536

22

2052

23

872

24

200

25

88

26

56

27

196

C(N) = number of Costas arrays of size N

The ambiguity function that is given as  ∞      |χ (τ, ν)| =  u(t)u ∗ (t + τ ) exp(j2π νt)dt    −∞

χ (τ, ν) is the two-dimensional correlation function in delay and Doppler. The peak to side lobe ratio (PSLR) can be found from the ambiguity plot

(1)

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PSLR (dB) = 20 log10

MAX(χ (τ )) χ (0, 0)

where χ (0,0) is the main lobe level and MAX(χ (τ )) is maximum side lobe level among all side lobes.

3 Result Analysis In this paper, authors have studied the performance of all Costas signals which were generated by exhaustive search up to length 27 by drawing the ambiguity plot’s for these signals and comparing their peak to side lobe ratio (PSLR) (both in ambiguity domain and autocorrelation domain) and reported best sequences at each and every length in autocorrelation and Doppler domain. Authors also generated Costas arrays using one of the systematic construction technique, namely Welch construction and find the best sequences among them at every length and compared these results with best sequences of exhaustive search. The best sequence at respective lengths in autocorrelation and Doppler domains along with their peak to side lobe ratios (PSLR) is found out. The ambiguity plots of the best sequences obtained through exhaustive search are plotted below in Doppler domain and autocorrelation domain (Figs. 3 and 4). Comparing Table 2 with Table 4 and Table 3 with Table 5, the best Costas sequences (by exhaustive search) and best Welch Costas sequences have same performance for few lengths (i.e. for 2–4), and for rest all other lengths, the best Costas

Fig. 3 Ambiguity plot of best sequence at length 20 in autocorrelation domain

Frequency Hopping Patterns for Low Probability of Intercept …

407

Fig. 4 Ambiguity plot of best sequence at length 5 in Doppler domain

sequences (by exhaustive search) have superior performance over best Welch Costas sequences. From Table 2, the best Costas sequence of all Costas arrays below the length 28 in autocorrelation domain is [7 12 15 16 8 18 5 2 6 20 1 14 10 3 9 17 19 13 4 11] with peak to side lobe ratio 14.8371. From Table 3, the best Costas sequence of all Costas arrays below the length 28 in Doppler domain is [4 5 2 1 3] with peak to side lobe ratio 8.5777.

4 Conclusions Costas signals are generated for lengths 2–27 using Welch construction method and exhaustive search method. The best sequences in both the autocorrelation domain and Doppler domain for each length in two methods are reported. The ambiguity plots for best sequences are plotted in autocorrelation domain and Doppler domain. It can be observed from the data generated in Tables 2, 3, 4 and 5 that the Costas signal with code [7 12 15 16 8 18 5 2 6 20 1 14 10 3 9 17 19 13 4 11] for N = 20 is the best sequence among the generated codes in autocorrelation domain with peak to side lobe ratio 14.8371, and signal with code [4 5 2 1 3] for N = 5 is the best sequence among the generated codes in Doppler domain with peak to side lobe ratio 8.5777. The best Costas sequences generated by exhaustive search method and best Costas sequences generated by Welch construction have same performance for few lengths (for 2–4), but for the remaining lengths (N = 5–27), the Costas sequences generated

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T. A. N. S. N. Varma and A. Gera

Table 2 Best sequences and their corresponding PSLRs for different lengths in autocorrelation domain (ACD) S. No.

Costas sequence length Best sequence

PSLR

1

2

21

11.6503

2

3

312

6.7408

3

4

4213

6.8576

4

5

51342

8.8097

5

6

613425

8.5189

6

7

6342571

7.9331

7

8

74356281

8.7131

8

9

837146529

8.6316

9

10

2 9 10 1 7 4 8 3 6 5

7.9223

10

11

2 7 9 3 11 6 5 1 4 10 8

11

12

4 12 10 9 6 2 7 1 11 3 5 8

12.0470

12

13

1 7 2 9 10 12 4 8 13 6 5 3 11

12.4506

13

14

2 8 10 6 11 5 14 7 4 3 1 9 12 13

12.9195

14

15

7 14 11 1 12 6 5 8 3 15 13 9 2 4 10

12.7547

15

16

5 10 4 12 2 15 14 16 8 9 13 6 3 1 11 7

13.0607

16

17

8 10 15 5 4 2 17 11 3 12 7 14 1 13 16 9 6

13.6584

17

18

5 12 4 10 18 2 14 7 1 17 16 3 13 11 15 6 8 9 13.6040

18

19

7 13 9 4 12 2 16 19 8 18 1 6 10 17 15 14 11 3 13.8179 5

19

20

7 12 15 16 8 18 5 2 6 20 1 14 10 3 9 17 19 13 14.8371 4 11

20

21

8 17 7 4 14 9 20 3 15 11 16 1 18 19 21 5 13 2 14.2160 6 12 10

21

22

11 6 12 4 18 1 9 21 22 13 15 20 5 2 17 16 19 14.4764 3 7 14 10 8

22

23

3 5 17 12 22 7 6 10 21 19 13 2 15 23 14 20 1 14.4251 18 11 8 4 9 16

23

24

7 22 15 10 14 19 20 8 4 6 5 24 21 1 13 16 3 12 23 9 17 11 2 18

14.2034

24

25

3 21 9 15 24 11 14 12 1 5 22 19 13 18 20 2 23 7 8 4 16 6 25 10 17

14.6670

25

26

5 20 19 10 2 15 21 8 17 7 12 1 25 11 23 18 3 14.6083 24 26 6 4 14 22 16 9 13

26

27

1 8 22 18 16 5 23 17 14 19 12 20 26 25 7 10 14.6128 11 27 3 15 2 21 13 24 9 4 6

11.7278

Frequency Hopping Patterns for Low Probability of Intercept …

409

Table 3 Best sequences and their corresponding PSLRs for different lengths in Doppler domain (DD) S. No.

Costas sequence length Best sequence

PSLR

1

2

21

6.8770

2

3

231

6.6647

3

4

1423

7.6218

4

5

45213

8.5777

5

6

354126

8.1519

6

7

5637214

8.0413

7

8

16583742

7.1309

8

9

573649128

7.0691

9

10

5 8 4 9 2 6 7 1 3 10

6.2830

10

11

2 8 11 1 10 6 7 4 3 5 9

6.5438

11

12

8 1 4 10 12 2 9 3 7 5 6 11

6.0597

12

13

3 8 7 9 6 12 2 10 11 4 13 1 5

6.2271

13

14

9 14 5 7 3 4 8 11 10 2 13 6 12 1

6.3097

14

15

6 12 10 14 4 13 2 5 7 1 15 11 3 8 9

6.2196

15

16

12 2 8 9 14 6 1 10 13 4 11 15 3 16 5 7

6.5920

16

17

11 4 12 10 2 7 1 15 14 3 5 17 8 9 13 16 6

6.8504

17

18

8 14 18 11 3 10 5 2 12 15 17 16 6 7 1 13 4 9

6.6672

18

19

12 16 7 2 8 9 19 1 15 3 5 18 10 17 13 6 11 14 4 7.1279

19

20

16 8 5 20 6 19 1 13 14 17 2 7 11 4 18 12 3 10 9 6.5834 15

20

21

10 13 4 17 16 1 11 3 9 20 15 8 5 19 6 18 2 21 7 6.8332 12 14

21

22

9 21 15 3 8 1 16 20 11 12 10 13 19 2 22 17 7 4 14 6 5 18

6.7317

22

23

7 18 10 12 21 20 4 2 3 15 19 22 8 1 17 23 5 13 9 14 11 6 16

6.5447

23

24

15 7 12 19 1 18 4 14 23 6 5 24 21 8 2 3 17 20 22 10 16 11 9 13

6.4453

24

25

16 25 22 20 7 23 5 12 14 24 19 8 13 9 21 1 18 17 11 4 15 3 6 10 2

6.3789

25

26

13 14 18 15 7 19 25 16 23 8 10 20 3 26 12 17 6 6.6531 24 22 9 2 11 5 21 1 4

26

27

13 17 23 9 24 16 18 12 2 7 10 22 19 15 4 20 21 6.7829 3 11 6 25 8 1 27 26 5 14

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T. A. N. S. N. Varma and A. Gera

Table 4 Best sequences among Welch Costas sequences and their corresponding PSLR’s for different lengths in autocorrelation domain (ACD) S. No.

Length Costas sequence

PSLR

1

2

12

11.6503

2

3

132

6.7408

3

4

1342

6.8576

4

5

43512

6.6273

5

6

154623

7.1109

6

8

26387514

7.3476

7

9

137498625

7.3928

8

10

1 6 3 7 9 10 5 8 4 2

7.0982

9

11

5 9 7 8 1 11 6 2 4 3 10

9.1141

10

12

1 6 10 8 9 2 12 7 3 5 4 11

11

15

6 14 2 3 10 8 11 15 9 1 13 12 5 7 4

12.0524

12

16

1 12 8 11 13 3 2 7 16 5 9 6 4 14 15 10

11.3270

13

17

9 4 11 5 2 10 14 16 17 8 13 6 12 15 7 3 1

11.7537

14

18

1 10 5 12 6 3 11 15 17 18 9 14 7 13 16 8 4 2

12.0867

15

21

13 11 6 5 14 2 18 12 20 17 21 8 10 15 16 7 19 3 9 1 4

12.7756

16

22

1 5 2 10 4 20 8 17 16 11 9 22 18 21 13 19 3 15 6 7 12 14

13.1212

17

26

2 6 14 1 4 10 22 17 7 16 5 12 26 25 23 19 11 24 21 15 3 8 18 9 13.2687 20 13

18

27

17 4 2 24 14 8 16 15 26 21 18 22 7 27 10 23 25 3 13 19 11 12 14.2956 1 6 9 5 20

9.4912

Table 5 Best sequences among Welch Costas sequences and their corresponding PSLRs for different lengths in Doppler domain (DD) S. No.

Length Costas sequence

PSLR

1

2

12

6.4977

2

3

231

6.6647

3

4

1342

6.1735

4

5

43512

6.9014

5

6

154623

7.2827

6

8

26387514

5.3818

7

9

137498625

4.7855

8

10

1 8 9 6 4 10 3 2 5 7

4.7408

9

11

1 3 7 2 5 11 10 8 4 9 6

4.1617

10

12

1 2 4 8 3 6 12 11 9 5 10 7

4.3607

11

15

2 8 9 12 4 14 10 15 13 7 6 3 11 1 5

4.6645 (continued)

Frequency Hopping Patterns for Low Probability of Intercept …

411

Table 5 (continued) S. No.

Length Costas sequence

PSLR

12

16

1 3 9 10 13 5 15 11 16 14 8 7 4 12 2 6

4.8503

13

17

12 16 11 3 13 10 9 15 17 5 1 6 14 4 7 8 2

5.5826

14

18

1 13 17 12 4 14 11 10 16 18 6 2 7 15 5 8 9 3

5.5437

15

21

6 2 20 8 16 3 4 11 14 12 21 15 19 1 13 5 18 17 10 7 9

5.2323

16

22

1 5 2 10 4 20 8 17 16 11 9 22 18 21 13 19 3 15 6 7 12 14

5.5497

17

26

2 6 14 1 4 10 22 17 7 16 5 12 26 25 23 19 11 24 21 15 3 8 18 9 5.4263 20 13

18

27

18 12 14 23 20 21 11 24 10 5 26 19 2 27 9 15 13 4 7 6 16 3 17 6.3548 22 1 8 25

by exhaustive search method were found to have superior performance over Costas sequences generated by Welch construction method. From the results, it can be concluded that Costas arrays produced has good peak to side lobe ratio in both Doppler domain and autocorrelation domain are very useful in LPI radar applications and eventually lead to high performance.

References 1. Güleç F, Afacan E (2015) Usage of Costas arrays in low probability of intercept radars. In: 2015 23rd signal processing and communications applications conference (SIU), Malatya, pp 347–350 2. Afacan E (2017) A new search methods for Costas arrays by using difference triangle analysis. In: Progress in electromagnetic research symposium (PIERS), St. Petersburg, 22–25 May 2017, pp 456–461 3. Costas JP (1984) A study of class of waveforms having nearly ideal range Doppler ambiguity properties. Proc IEEE 72(8):996–1009 4. Golomb SW, Taylor H (1984) Constructions and properties of Costas arrays. Proc IEEE 72(9):1143–2263 5. Drakakis K (2006) A review of Costas arrays. J Appl Math 2006:1–32 6. Nathanson FE. Radar design principles signal processing and the environment, 2nd edn 7. Drakakis K (2010) Some results on the degrees of freedom of Costas arrays. In: 2010 44th annual conference on information sciences and systems (CISS), Princeton, NJ, pp 1–5 8. Beard JK, Russo JC, Monteleone M, Wright M (2007) Costas array generation and search methodology. IEEE Trans Aerosp Electron Syst 43(2):522–538 9. Correll B (2019) More new structural properties of costas arrays. In: 2019 IEEE radar conference (RadarConf), Boston, MA, pp 1–6 10. Jedwab J, Yen L (2018) Costas cubes. IEEE Trans Inf Theory 64(4):3144–3149 11. Chang C-F, Bell MR (2003) Frequency-coded waveforms for enhanced delay-Doppler resolution. IEEE Trans Inf Theory 49(11):2960–2971 12. Nusenu SY, Wang Z, Wang W (2016) FDA radar using Costas sequence modulated frequency increments. In: 2016 CIE international conference on radar (RADAR), Guangzhou, pp 1–4

Printed Monopole Slot Antenna Inspired by Metamaterial Unit Cell for Wireless Applications Swati Sharma and Rekha Mehra

Abstract Negative index metamaterial unit cell-loaded slot antenna fed with microstrip line is fabricated and analyzed in this paper. The proposed triple-band antenna structure is constructed on flame retardant-4 substrate (εr = 4.3). Circularshaped split-ring resonator acts as radiating element which creates multiple resonances at 1.68, 2.32, and 3.9 GHz. Proposed antenna structure yields −10 dB impedance bandwidth of 2.38% (1.66–1.70) GHz, 17.54% (2.08–2.48) GHz, and 3.07% (3.84–3.96) GHz. Negative index properties of metamaterial unit cell corroborated by applying Nicolson-Ross-Weir (NRW) approach. Simulated and measured results of return loss and radiation patterns are in a match and have been verified. The proposed antenna can be applied for Bluetooth, wireless local area network (WLAN), and a higher range of worldwide interoperability for microwave access (Wi-MAX). Keywords Metamaterials · NRW approach · Slot antenna · SRR · WLAN · Wi-MAX

1 Introduction As wireless communication is growing rapidly, the concept of metamaterial-inspired is widely used for designing miniaturized and multiband antennas. The multiband antenna system supports the incorporation of different wireless standards in a single antenna unit. Literature shows that numerous narrowband, wideband, ultra-wideband (UWB), dual-band, and multiband antennas have been developed during the last decades [1–4]. Multiband antennas are more preferable than ultra-wideband antennas as its ability to diminish the effect of the electromagnetic interference (EMI), which exist due to other nearby communication systems. In literature, several low-profile, dual-band antennas are proposed for wireless applications, which are based on a split ring and complementary split-ring resonators [5, 6]. These sub-wavelength resonators (SRR S. Sharma (B) · R. Mehra Department of Electronics and Communication Engineering, Government Engineering College Ajmer, Ajmer, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_44

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and CSRR) are the key components of metamaterials, which were introduced by the Russian physicist Victor Veselago in the year 1968. Since then, it is researched and utilized by the researchers of microwave and photonics for improving the parameters of the antenna [7]. Metamaterial properties lie in its structure rather than its composition. The structural properties of metamaterial influence the propagation directions of the electromagnetic waves and hence new resonant frequencies are generated [8–10]. Triple-band printed slot antenna inspired with circular-shaped SRR is designed, fabricated, and analyzed here. The slot antenna is used in this design as it is known to provide superior isolation between the radiating element and feed network [11, 12]. A square-shaped loop slot is designed in the ground plane and a circular-shaped split-ring resonator is designed at the top of the right arm of slot loop, are playing the role of radiating elements for this antenna. The dimensional parameters of split-ring resonator and its placement are chosen in such a way so that it can provide additional bands to the original slot resonance at lower frequencies. The circular shape of the split-ring resonator is considered here as it produces the wider bandwidth compare to [13], which used square-shaped SRR. The paper is organized into four sections. In Sect. 2, the stepwise design of metamaterial-inspired antenna is proposed. In Sect. 3, measured results are compared with simulated results, and in Sect. 4, conclusions are given.

2 Metamaterial-Inspired Antenna Design 2.1 Transformation of the Proposed Antenna Figure 1 depicts the transformation of the proposed antenna structure in two configurations. Confg_A shows a slot antenna fed by a 50  microstrip line. The slot loop and four rectangular slots at the top of the slot loop are engraved at the ground surface. It shows dual-band characteristics and resonates at 2.17 GHz with 340 MHz impedance bandwidth and 4.16 GHz with 160 MHz impedance bandwidth. In confg_B, a circular-shaped SRR is loaded at the right arm of the slot loop, on the other side of the substrate, which creates lower-order resonance with shifts in other exiting resonating frequencies due to capacitance effect of SRR and produces Fig. 1 Transformation of the proposed antenna

Printed Monopole Slot Antenna Inspired by Metamaterial Unit …

415

W

LB

Top layer metallization

WB

Bottom layer metallization WS SRR

LS

LOff

L

Slot loop

Lf

g

C1

X

g

a

d

Feed Line

C2

b

Wf Enlarged view of SRR

Fig. 2 Schematic diagram of proposed antenna loaded with a split-ring resonator

Table 1 Parameters of the proposed antenna and circular SRR Parameters

Values (mm)

Parameters

Values (mm)

Parameters

Values (mm)

L

55

Lf

23

X=g

0.9

W

42

Wf

1.5

LB

3

Ls

25

L off

4

WB

5

Ws

24

a

5

b

4

C1

0.5

C2

0.4

s

0.3

a triple-band antenna. The schematic diagram of the proposed antenna is displayed in Fig. 2. The antenna parameters are listed in Table 1.

2.2 Parametric Analysis of Metamaterial Structure The equivalent circuit model is used to determine the behavior of the SRR structure. It consists of an LC tank circuit, where inductance (L) of SRR is due to the metallic ring and capacitance (C) is due to the slit gap and the slots between metal rings. Resonating frequency f 0 of the circular SRR is given by [14] and can be calculated as  1 1 (1) f0 = 2π LeqCeq Here, Leq is the equivalent inductance and Ceq is the equivalent capacitance of the LC tank circuit.

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The structure of an SRR and its equivalent circuit is shown in Fig. 3. Parameters of SRR structure are evaluated here by using Nicolson-Ross-Weir (NRW) approach. Figure 4 shows the waveguide setup inside which metamaterial structure (SRR) is placed, so that reflection (S 11 ) and transmission coefficient (S 21 ) of SRR can be calculated. In Fig. 4, Y-Plane, and Z-Plane are portrayed as perfect electric boundary (PEB) and perfect magnetic boundary (PMB), respectively. SRR is positioned between these waveguide ports, as proposed by [15]. The reflection and transmission coefficient obtained from the waveguide setup is shown in Fig. 5, which reflects that the resonance of SRR occurs at 3.1 GHz, which is the same as calculated by Eq. (1). The S 11 and S 21 parameters obtained are then utilized to calculate magnetic permeability and electric permittivity by using the following equations [15]: μr ≈

2 · c(1 − v2 ) ω · d · i(1 + v2 )

(2)

2 · S11 · c · i ωdi

(3)

εr ≈ μr + where Fig. 3 SRR and its equivalent circuit model

Fig. 4 Metamaterial structure (SRR) between waveguide ports

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Fig. 5 Simulated reflection and transmission coefficients graph of SRR

ω d c v1 v2

v1 = S21 + S11

(4)

v2 = S21 − S11

(5)

ω = k0 c

(6)

Angular frequency in radian (ω = 2π f ) Thickness of the substrate Speed of light (3 × 108 ) m/s Voltage maxima Voltage minima.

By subsituting the simulated S-parameters of circular split-ring resonator in above formulas, the graphs between frequency versus permeability and frequency versus permittivity are plotted as shown in Fig. 6 and Fig. 7, respectively. The real and imaginary plots of permeability and permittivity are shown. The negative value of permeability and permittivity is observed around the 3.1 GHz due to the stop band behavior of SRR; hence, it proves that the proposed SRR structure exhibits the metamaterial properties.

3 Result and Analysis The simulated |S11 | characteristics of the evolution of the proposed antenna are shown in Fig. 8. It is seen here that Confg_A, has dual-band resonance at 2.17 and 4.16 GHz along with an impedance bandwidth of 330 MHz (1.99–2.36 GHz) and 160 MHz (4.08–4.24 GHz), respectively. In Confg_B, the circular split-ring resonator changes the path of current flow, which gives rise to a lower-order resonance at 1.68 GHz with an impedance bandwidth of 40 MHz (1.66–1.70 GHz). A shift in higher-order resonance has been observed, and it resonates at 3.9 GHz with an impedance bandwidth of 120 MHz

418

Fig. 6 Permeability versus frequency graph

Fig. 7 Permittivity versus frequency graph

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Fig. 8 Simulated S 11 characteristics of the proposed antenna comparing two deferent configurations

(3.84–3.96 GHz). The shift has also been observed in the second band, and it resonates at 2.32 GHz with an increased bandwidth of 400 MHz (2.08–2.48 GHz). The picture of the fabricated antenna is displayed in Fig. 9. Return loss (S 11 ) characteristics of the fabricated prototype are measured with Rohde and Schwarz ZVB 8 vector network analyzer, measurement setup shown in Fig. 10a. The measurements of radiation patterns are performed using an anechoic chamber are shown in Fig. 10b. Simulated and measured results are compared and the relation is illustrated in Fig. 11. The numerical comparison is shown in Table 2. A little difference has been observed in simulated and measured values of |s11 | at lower frequencies due to the manufacturing tolerance of substrate and soldering of Fig. 9 Picture of the fabricated proposed antenna (front view and back view)

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(a)

(b)

Fig. 10 Image of measurement setup of the fabricated prototype

Fig. 11 Comparison of return loss characteristics (S 11 ) obtained from the simulation and measurement of the proposed antenna Table 2 Numerical comparison of obtained results from simulation and measurements of the proposed antenna Parameters

Simulated

Resonating frequency 1.68 (GHz)

Measured 2.32

3.9

1.91

2.5

3.86

−27.28

−17.59

−33.2

−26.76

−19.10

Return loss (dB)

−27.83

Band (GHz)

1.66–1.70 2.08–2.48 3.84–3.96 1.79–2.06 2.33–2.58 3.79–3.94

Bandwidth (MHz) FWB%

40 2.38%

400 17.54%

120 3.08%

270 14%

250 10.2%

150 4.05%

Printed Monopole Slot Antenna Inspired by Metamaterial Unit …

1.68 GHz

2.32 GHz

421

3.9 GHz

Fig. 12 Surface current distribution at resonating frequencies

the SMA connector, whereas at higher frequency it is in a good match. A tripleband resonance at 1.91 GHz, 2.5 GHz and 3.86 GHz, with impedance bandwidth of 270 MHz (1.79–2.06 GHz), 250 MHz (2.33–2.58 GHz), and 150 MHz (3.70– 3.94 GHz), respectively, is achieved by the measurement, which is useful for 4G LTE, WLAN, Bluetooth, and Wi-MAX applications. The surface current distribution of proposed antenna at 1.68 GHz, 2.32 GHz, and 3.9 GHz is represented in Fig. 12a–c, respectively. In Fig. 12a, distribution of current is intense around the split rings of SRR, in Fig. 12b, current is intense at the feed line and left corner of the slot loop, and in Fig. 12c, the current is intense at the entire slot loop along with the fissure of SRR. It is observed that the metamaterial structure (SRR) changes the flow of current hence bandwidth is enhanced of this antenna. The simulated and measured two-dimensional polar plots of antenna radiation pattern at resonating frequencies are shown in Fig. 13. The desired polarization of wave and orthogonal radiation radiated by an antenna in E-plane and H-plane is clearly shown here. At phi (φ) = 0◦ the antenna radiates in omnidirectional pattern and at phi (φ) = 90◦ , i.e., in H-plane, and antenna radiates in bi-directional pattern at 1.68, 2.32, and 3.9 GHz. The frequency versus gain plot is shown in Fig. 14 where the simulated gain is of 0.5 dBi, 2.93 dBi, 2.61 dBi, and measured gain of 1.09 dBi, 2.9 dBi, and 2.6 dBi is observed at 1.68 GHz, 2.32 GHz, and 3.9 GHz, respectively.

4 Conclusion A monopole slot antenna inspired by a circular-shaped split-ring resonator for tripleband behavior is fabricated and analyzed here. The metamaterial structure (SRR) is used to produces a new resonating band along with the existing slot antenna resonating bands with little shift. The circular shape of SRR encourages wider bandwidth at resonating bands. SRR structure is analyzed by applying the NRW method to verify its metamaterial property. An omnidirectional and bi-directional radiation pattern at three different frequencies in and azimuthal plane and elevation plane is simulated, measured, and illustrated here, respectively. The proposed antenna is

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1.68 GHz

2.32 GHz

3.9 GHz Fig. 13 2D far-field radiation pattern at resonating frequencies

suitable for various applications such as 4G LTE, WLAN, Bluetooth, and Wi-MAX applications due to its multiband and wideband characteristics.

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Fig. 14 Gain versus frequency plot

Acknowledgements The authors would like to convey their sincere thanks to the Center for Applied Research in Electronics (CARE-Lab), Indian Institute of Technology (IIT) Delhi, India, for providing necessary facilities for measurement in an anechoic chamber and Veermata Jijabai Technological Institute, Mumbai (VJTI) for measurement on VNA for the accomplishment of this research work.

References 1. Daniel RS, Pandeeswari R, Raghavan S (2018) Dual-band monopole antenna loaded with ELC metamaterial resonator for Wi-MAX and WLAN applications. Appl Phys A 124 2. Rajasekhar NV, Kumar DS (2018) Metamaterial-based compact UWB planar monopole antennas. Microw Opt Technol Lett 60:1332–1338 3. Sharma S, Mehra R (2018) Dual-band planner slot antenna loaded with split ring resonators for WLAN/Wi-Max applications. In: International conference on smart city and emerging technology (ICSCET), IEEE, Mumbai 4. Daniel RS, Pandeeswari R, Raghavan S (2017) Offset-fed complementary split ring resonators loaded monopole antenna for multiband operations. AEU Int J Electron Commun 78:72–78 5. Wang H, Si LM, Lv X (2017) A compact dual-band patch antenna using metamaterial structures. In: International applied computational electromagnetics society symposium (ACES), China. IEEE, pp 1–2 6. Antoniades MA, Eleftheriades GV (2012) Multiband compact printed dipole antennas using NRI-TL metamaterial loading. IEEE Trans Antennas Propag 60:5613–5626 7. Zhu J, Antoniades MA, Eleftheriades GV (2010) A compact tri-band monopole antenna with single-cell metamaterial loading. IEEE Trans Antennas Propag 58:1031–1038 8. Buell K, Mosallaei H, Sarabandi K (2006) A substrate for small patch antenna providing tunable miniaturization factor. IEEE Trans Microw Theory Tech 54:135–146 9. Alu A, Bilotti F, Engheta N, Vegni L (2007) Subwavelength, compact, resonant patch antennas loaded with metamaterials. IEEE Trans Antennas Propag 55:13–25 10. Rosaline SI, Raghavan S (2016) Design of split ring antennas for WLAN and WiMAX applications. Microw Opt Technol Lett 58:2117–2122

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11. Basaran SC, Olgun U, Sertel K (2013) Multiband monopole antenna with complementary split-ring resonators for WLAN and WiMAX applications. Electron Lett 49:636–638 12. Daniel RS, Pandeeswari R, Raghavan S (2018) A compact metamaterial loaded monopole antenna with offset-fed microstrip line for wireless applications. AEU Int J Electron Commun 83:88–94 13. Sarkar D, Saurav K, Shrivastava KV (2014) Multi-band microstrip-fed slot antenna loaded with split ring resonator. Electron Lett 50:1498–1500 14. Saha C, Siddiqui JY (2009) Estimation of the resonance frequency of conventional and rotational circular split ring resonators. In: IEEE applied electromagnetics conference (AEMC), Kolkata. IEEE 15. Ziolkowski RW (2013) Design, fabrication, and testing of double negative metamaterials. IEEE Trans Antennas Propag 51:1516–1529

Reconfigurable Hexa-band Antenna with Defected Ground Structure for 2.1–3.4 GHz Band Applications Shadab Azam Siddique, Brijesh Mishra, Rakesh Kumar Singh, Vivek Singh, and Sanjay Kumar Soni

Abstract This paper presents the design and analysis of the reconfigurable hexa-band microstrip patch antenna for wireless applications. Antenna is designed over the FR4 epoxy substrate (16 × 33 mm2 ) with defected ground structure (16×7.5 mm2 ) and three PIN switch diodes. The proposed antenna operates at one dual band (2.1 and 3.4 GHz) and four single bands (3.3, 3.2, 3, and 2.8 GHz) with 193 MHz, 400 MHz, 628 MHz, 647 MHz, 409 MHz, and 413 MHz bandwidth, respectively. For the proposed antenna, a maximum peak gain of 9.2 dB at 2.1 GHz resonating frequency has been achieved. Proposed antenna covers the wide range (2.1–3.4 GHz) band of reconfigurability with a maximum of 1.6 frequency ratio. The multiband behavior of the proposed antenna makes it best choice for wireless applications over single-band antenna. Keywords Frequency reconfigurable · FR4 epoxy · PIN diodes · Hexa-band · Impedance bandwidth

1 Introduction Previous developments have shown the advancement of mobile and portable communication systems focused on more compact and stable antenna structures. Miniaturized structure is preferred over conventional structures to be built for practical mobile handsets. However, limited space have made it complex to introduce various frequency bands such as GSM, GPS, UMTS, WLAN, Wi-Fi, and WiMAX with high efficiency when all the electronic components are integrated on the same terminal [1, 2]. Hence, the frequency reconfigurable technology is introduced in order to minimize the size of microstrip patch antenna and to achieve multiband operation S. A. Siddique · B. Mishra (B) · S. K. Soni Department of Electronics and Communication, Madan Mohan Malaviya University of Technology, Gorakhpur, Uttar Pradesh, India R. K. Singh · V. Singh Department of Electronics and Communication, Shambhunath Institute of Engineering and Technology, Prayagraj, Uttar Pradesh, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_45

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with the use of single antenna only [3–5]. Reconfigurable antennas are of two major types; physically reconfigurable antennas which has low insertion loss and high reliability but difficult to tune and hard to apply in fast time-varying control system [6] and electrically reconfigurable antennas which has high switching speed, simple structure and low price made the electrically reconfigurable antenna most desirable in the wireless applications [5, 7–10]. Reconfigurable antennas can be designed for frequency, radiation pattern, polarization, and beam steering adjustment with several switches like RF-MEMS [7], liquid crystal switches [8], photoconductive switches [9], stepper motor [10], and RF diodes [5]. The PIN diodes are the most preferable switch in the history of reconfigurable antennas due to its efficient switching time and low insertion loss. Reconfigurable microstrip patch antennas have gained special demand in the field of wireless application systems. Several designs with the aforesaid applications have been recently published in repute journals are enumerated in Table 1. In [3], a frequency reconfigurable antenna is proposed with simple antenna geometry capable of configuring its frequency to operate either in multiband mode (2.4, 3.5, and 5.2 GHz) or wideband mode ranging (2–6 GHz) suitable for the use in cognitive radio and wideband communication system. Better gain (2.45 dB) is observed at lower frequencies (2.4 and 3.5 GHz) as compared to gain (0.24 dB) at higher frequency (5.2 GHz). In [11], a compact (15 × 28.5 mm2 ) frequency reconfigurable antenna is designed with the use of two PIN diodes in the radiating element covering the 0.82–1.03, 1.9–2.4, 2.09–2.82, 1.6–2.09, and 2.6–2.8 GHz frequency bands for mobile and wireless applications with peak gain varies from (−2.19 to −0.79 dB) and efficiency of 34–48% in low band (820–960 MHz), peak gain of (−1.87 to 3.59 dB) and efficiency of 42–73% in middle band (1710–2170 MHz) and peak gain of (0.87–3.59 dB)and efficiency of 75–84.86% at higher band (2300–2690 MHz) suitable for multiband mobile handset devices. In [12], a low-profile high-gain frequency reconfigurable patch antenna is designed with the ability to change the working frequency continuously ranging from 2.45 to 3.55 GHz maintaining a broadside and stable radiation pattern at all the operating modes. Sufficient gain rising from 4.25 to 7.9 dB with the increasing working frequency from 2.45 to 3.55 GHz which is observed in the proposed model makes it suitable for cognitive radio and other wireless communication systems. In [13], a compact (20×22 mm2 ) printed monopole antenna is proposed with inverted T-shaped stub and two C-shaped slots in the radiating patch for obtaining triple band notch function. Pin diodes are used to achieve reconfigurable characteristics and are applicable for Bluetooth, LTE, UWB, and other wireless applications in the range (8–12 GHz) in X-band and satellite communication in Ku-band. Bandwidth ranging from 2.285 to 19.35 GHz is observed with gain ranging from 3.65 to 5.65 dBi without notch band, whereas for notch bands gain decreases to 21.36 dBi at 3.58 GHz, 27.33 dBi at 5.35 GHz, and 12.98 dBi at 7.42 GHz. Sufficiently good radiation efficiency is observed for bands without notches as compared to notched bands. In [14], a printed multiband frequency reconfigurable patch antenna is designed in which reconfigurability is achieved by designing RLC based DC line circuit providing DC biasing to the three PIN diodes used as a switch. Comparatively lower peak gain

Reconfigurable Hexa-band Antenna with Defected Ground Structure …

427

is observed with stable radiation pattern and is suitable for nine different 4G LTE frequency bands (0.9, 1.4, 1.5, 1.6, 1.7, 1.8, 2.6, 3.5 GHz, and WLAN band 2.5 GHz). In this paper, a compact (16×33 mm2 ) hexa-band frequency reconfigurable microstrip patch antenna is designed for the wireless applications. The overall size (mm2 ) of the proposed antenna is larger than the antenna reported in [11, 13] with the factor of 1.23 and 1.2, respectively, whereas smaller than the antenna reported in [3, 12, 14] with the factor of 14.67, 9.28, and 5.68, respectively. The proposed model is operated at six different resonating frequencies obtained by switching action of three PIN diodes fabricated in slot (1×2 mm2 ) created between two conducting strip. Proposed antenna is compared on the basis of size, substrate, number of diodes, number of resonating bands, resonating frequency, impedance bandwidth and gain, with previously reported papers in Table 1. In the proposed work, authors have gained significant improvement in antenna parameters as compared to antennas [3, 11–14] reported in Table 1. Antenna geometry, methodology, result and discussions, and conclusion of the proposed work elaborated in the succeeding sections.

2 Antenna Design and Configuration The proposed antenna is designed over the FR4-epoxy substrate (16×33 mm2 ) with the dielectric constant of 4.4, loss of tangent tan (δ) of 0.002 and antenna height of 1.6 mm. The design constructed with the microstrip feed line (3×13 mm2 ) of 50  characteristics impedance and three PIN diodes (D1, D2, and D3) to make the antenna reconfigurable. For the switching action, PIN diodes (Skyworks SMP 1345079LF) with the dimension of (1×2 mm2 ) are used. Defected ground structure (DGS) (16×7.5 mm2 ) is etched on the ground plane of dielectric substrate to improve the antenna bandwidth. The positions of diodes are intuitively identified on the radiating patch to provide wide range of frequency reconfigurability. The geometry of front view and back view of the proposed antenna with dimensions of each conducting strip are well labelled in Fig. 1a, b, respectively.

3 Result and Discussion The proposed antenna has been analyzed and simulated by means of high frequency structure simulator (HFSS v.17) EM tool. Performance of the proposed antenna has been carried out in terms of surface current distribution, return loss, VSWR, radiation pattern, and antenna gain. The proposed antenna consists of three PIN diode switches (cf. Fig. 1a) D1, D2, and D3 (Skyworks SMP 1345-079LF) and resonates at six different frequencies 2.1, 3.4, 3.3, 3.2, 3, and 2.8 GHz by switching action. Diode-lumped equivalent circuit consists of series combination of inductor (0.7 nH) and resistor (1 ) in forward bias and series combination of inductor (0.7 nH) and parallel combination of resistor

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Table 1 A comparative study of the proposed antenna and previously reported antennas Ref. Year Antenna Substrate No. of RF IBW Gain size (mm2 ) used diodes (GHz) % (dB) /bands [3]

2018

88×88 = 7744

[11]

2017

15×28.5 = 427.5

[12]

2018

70×70 = 4900

[13]

2019

20×22 = 440

[14]

2018

50×60 = 3000

[*]

NA

16×33 = 528

r = 4.3 tan = 0.025 h= 1.6 mm r = 4.4 tan(δ) = 0.02 h= 0.8 mm r = 2.2 tan(δ) = 0.002 h = 1 mm r = 2.33 tan(δ) = 0.012 h= 0.787 mm r = 4.4 tan(δ) = 0.002 h= 1.6 mm r = 4.4 tan(δ) = 0.002 h= 1.6 mm

2/3

2.4, 3.5, 5.2

6.3, 5.7, 14.4

2.45 2.33 0.24

2/5

0.9, 1.9, 2.1, 2.4, 2.5

23.3, 26.3, −1.87 34.7, 20.48 to 3.59

2/6

2.42, 2.57, 2.69, 2.91,

3% at each RF

3.14, 3.56 3.86, 5.48, 7.60

18.1, 17.8 6.97

3/3

3/9

0.9, 1.4, 1.5, 1.6, 1.7, 1.8,

4.26, 4.94 4.98, 5.92 6.91, 7.9 5.88

NR

2.3–4.7

9.2, 11.8, 19, 20.2

9.2, 2.1 2.2, 2.24

13.6, 14.8

2.15, 3.1

2.6, 3.5, 2.5 3/6

2.1, 3.4, 3.3, 3.2, 3, 2.8

Legends [*]—Proposed, NR—Not reported, RF—Resonating frequency, IBW—Impedance bandwidth, NA—Not applicable

(1 M) and capacitor (0.17 pF) in reverse bias (cf. Fig. 1c). However, the antenna performance has been studied at these six resonating frequencies to validate the antenna reconfigurability. Surface current distributions of the proposed antenna presented in Fig. 2a–f at 2.1 GHz, 3.4 GHz, 3.3 GHz, 3.2 GHz, 3 GHz, and 2.8 GHz, respectively. From the perusal of Fig. 2, it can be safely concluded that the antenna operates in the T M10 mode due to its unidirectional surface current distributions (along x-axis) behavior. Surface current distribution strength at 2.1 and 3.4 GHz of state 010 is high as

Reconfigurable Hexa-band Antenna with Defected Ground Structure … 16

16

429 Lumped RLC diode

11.3

6.5

Strip

2 mm

D3

Strip

2

15

16.6

1mm

D2 10

1

L

R

33

33

D1

millimeter scale

Diode ON Battery

8.5 L

millimeter scale

R

13 7.5

Diode ON C

3

a

Battery

b

c

Fig. 1 Antenna geometry a front view, b bottom view, c lumped equivalent of diode

Fig. 2 Surface current distribution of the proposed antenna at a 2.1 GHz, b 3.4 GHz, c 3.3 GHz, d 3.2 GHz, e 3 GHz, f 2.8 GHz

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Table 2 Performance analysis of the proposed antenna in (2.1–3.4 GHz) frequency range States Resonating Resonance Bandwidth Gain Return loss frequency band (GHz) (GHz) (MHz)/ % (dB) (S11 ) (dB) 010 100 101 110 111

2.1 3.4 3.3 3.2 3.0 2.8

2.0–2.193 3.189–3.589 3.027–3.655 2.952–3.599 2.815–3.224 3.0–2.587

193/9.2 400/11.8 628/19 647/20.2 409/13.6 413/14.8

9.203 2.0998 2.192 2.2454 2.1599 3.0677

< −23.593 < −24.117 < −38.359 < −31.321 < −40.230 < −23.114

compared to other proposed resonating frequencies (cf. Fig. 2a–f) which is also in conformity of Fig. 5. Performance of the proposed antenna in terms of return loss is illustrated in Fig. 3a and enumerated in Table 2. The return loss characteristic of the proposed antenna has been analyzed in the five different states like state 010 (D3-OFF, D2-ON, D1OFF), state 100 (D3-ON, D2-OFF, D1-OFF), state 101 (D3-ON, D2-OFF, D1-ON), state 110 (D3-ON, D2-ON, D1-OFF), and state 111 (D3-ON, D2-ON, D1-ON) and omitted rest three states (state 000, 001 and 011) from the discussion due to the same behavior. In state 010, antenna resonates at two different frequencies (2.1 and 3.4 GHz), while in state 100, 101, 110, and 111, antenna resonates at 3.3, 3.2, 3.0, and 2.8 GHz frequencies, respectively. The impedance bandwidth of 193 MHz (2.0– 2.193 GHz), 400 MHz (3.189–3.589 GHz), 628 MHz (3.027–3.655 GHz), 647 MHz (2.952–3.599 GHz), 409 MHz (2.815–3.224 GHz), and 413 MHz (3.0–2.587 GHz) are observed, respectively. The return loss of the proposed antenna is less than −23 dB (< −23 dB) in all the states. The VSWR characteristic of the antenna for all states is presented in Fig. 3b. Proposed states have less than 2 VSWR bands which are in the conformity of low loss of electromagnetic power. Figure 4 represents the comparison of simulated E- and H-field patterns of the proposed antenna and analyzed at 2.1 GHz, 3.4 GHz, 3.3 GHz, 3.2 GHz, 3 GHz, and 2.8 GHz in Fig. 4a–f, respectively. Figure 4 depicts the dipole and omnidirectional like radiation pattern in E-plane and H-plane, which illustrates that the proposed antenna design is most suitable candidate for the wireless applications. A consistent omnidirectional radiation pattern is observed in H-plane field whereas bidirectional radiation pattern is observed in E-plane field. Gain of the antenna is the key factor to decide the EM wave transmission distance. High gain offers long distance microwave communication for modern devices. The 3D gain plots of the proposed antenna are presented in Fig. 5a–f at 2.1 GHz, 3.4 GHz, 3.3 GHz, 3.2 GHz, 3 GHz, and 2.8 GHz, respectively. A peak gain of the proposed antenna of 9.2 dB, 2.09 dB, 2.19 dB, 2.24 dB, 2.15 dB, 3.06 dB at 2.1 GHz, 3.4 GHz, 3.3 GHz, 3.2 GHz, 3 GHz, and 2.8 GHz are, respectively, observed. The proposed antenna has a maximum peak gain in state 010 (D1-OFF, D2-ON, D3-OFF) of 9.2 dB at 2.1 GHz which validates the results obtained in Figs. 4a and 5a.

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a

431

b

Fig. 3 a Return loss plot of the proposed antenna, b VSWR plot of the proposed plot

a

b

c

d

e

f

Fig. 4 Radiation pattern of the proposed antenna at a 2.1 GHz, b 3.4 GHz, c 3.3 GHz, d 3.2 GHz, e 3 GHz, f 2.8 GHz

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Fig. 5 3D gain plot of proposed antenna at a 2.1 GHz, b 3.4 GHz, c 3.3 GHz, d 3.2 GHz, e 3 GHz, f 2.8 GHz

4 Conclusion In this paper, a novel and compact (16×33 mm2 ) frequency reconfigurable antenna with five switchable states is presented. Antenna covers a wide range (2.1–3.4 GHz) of reconfigurabilty with six resonating bands. The proposed antenna has a maximum impedance bandwidth of 20.2% at 3.2 GHz resonance frequency in state 101 which is greater than the antennas [3, 12–14] reported in Table 1, a maximum peak gain of 9.2 dB at 2.1 GHz resonance frequency in state 010 as greater than the all reported antennas enumerated in Table 1 and good return loss (< −40 dB) in state 110. Antenna [11] has larger bandwidths and small size as compared to proposed antenna, but gain (−1.87 to 3.59) and number of resonating bands (5) are comparatively low. However, performance of the proposed antenna makes it more suitable element for wireless communication systems. Acknowledgements Research described in this paper was financially supported by The World Bank and National Project Implementation Unit (NPIU), MHRD, India, under the TEQIP III project scheme of Collaborative Research Scheme (CRS) of Project Entitled “Development of IoT controlled frequency/pattern reconfigurable MIMO antenna for harvesting systems.”

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References 1. Mishra B, Singh V, Singh R (2018) Gap Coupled Dual-Band Petal Shape Patch Antenna for WLAN / WiMAX Applications. Advances in Electrical and Electronic Engineering 16(2):185– 198 https://doi.org/10.15598/aeee.v16i2.2416 2. Singh V et al (2016) A Compact Quad-Band Microstrip Antenna for S and C-Band Aapplications. Microwave and Optical Technology Letters 58(6):1365–1369. https://doi.org/10.1002/ mop.29799 3. Idris, I. H. et al. (2018) ‘A Multi to Wideband Frequency Reconfigurable Antenna’, International Journal of RF and Microwave Computer-Aided Engineering, 28(4), p. e21216 (1–9). https:// doi.org/10.1002/mmce.21216 4. Vamseekrishna A et al (2019) An IoT Controlled Octahedron Frequency Reconfigurable Multiband Antenna for Microwave Sensing Applications. IEEE Sensors Letters 3(10):1–4. https:// doi.org/10.1109/LSENS.2019.2943772 5. Shah IA et al (2019) Design and Analysis of a Hexa-Band Frequency Reconfigurable Antenna for Wireless Communication. AEU-International Journal of Electronics and Communications 98:80–88. https://doi.org/10.1016/j.aeue.2018.10.012 6. Tawk, Y. et al. (2017) ‘Physically Reconfigurable Antennas: Concepts and Automation’, in 2017 IEEE International Symposium on Antennas and Propagation & USNC/URSI National Radio Science Meeting. IEEE, pp. 419–420. https://doi.org/10.1109/APUSNCURSINRSM. 2017.8072252 7. Deng, Z. and Yao, Y. (2011) ‘Ka Band Frequency Reconfigurable Microstrip Antenna Based on MEMS Technology’, In: Wan X. (eds) Electrical Power Systems and Computers. Lecture Notes in Electrical Engineering, vol 99. Springer, Berlin, Heidelberg pp. 535–541. https://doi. org/10.1007/978-3-642-21747-0-67 8. Shi, H. et al. (2018) ‘Radiation Pattern Reconfigurable Waveguide Slot Array Antenna Using Liquid Crystal’, International Journal of Antennas and Propagation, pp. 1–9. https://doi.org/ 10.1155/2018/2164065 9. Zhao, D. et al. (2014) ‘Low-Power Optically Controlled Patch Antenna of Reconfigurable Beams’, International Journal of Antennas and Propagation, pp. 1–6. https://doi.org/10.1155/ 2014/978258 10. Costantine, J., Tawk, Y. and Christodoulou, C. G. (2016) ‘Reconfigurable Antennas’, in Handbook of Antenna Technologies. Singapore: Springer Singapore, pp. 1737–1772. https://doi. org/10.1007/978-981-4560-44-3-61 11. Zhang, X. et al. (2017) ‘A Frequency Reconfigurable Antenna for Multiband Mobile Handset Applications’, International Journal of RF and Microwave Computer-Aided Engineering, 27(9), p. e21143 (1–8). https://doi.org/10.1002/mmce.21143 12. Cai Y-M et al (2018) A Low-Profile Frequency Reconfigurable Grid-Slotted Patch Antenna. IEEE Access 6:36305–36312. https://doi.org/10.1109/ACCESS.2018.2850926 13. Sharma, M., Awasthi, Y. K. and Singh, H. (2019) ‘Compact Multiband Planar Monopole Antenna for Bluetooth, LTE, and Reconfigurable UWB Applications Including X-Band and Ku-Band Wireless Communications’, International Journal of RF and Microwave ComputerAided Engineering, 29(6), p. e21668 (1–11). https://doi.org/10.1002/mmce.21668 14. Chattha HT et al (2018) Frequency Reconfigurable Patch Antenna for 4G LTE Applications. Progress In Electromagnetics Research M 69:1–13. https://doi.org/10.2528/PIERM18022101

A Compact Design of Square Ring-Shaped Microstrip Monopole Antenna for Wireless Applications Ajay Dadhich, Megha Agarwal, J. K. Deegwal, and M. M. Sharma

Abstract A compact design of a square ring-shaped microstrip monopole antenna for wireless application is presented in this paper. The antenna has square shaped ring monopole in which U- and T-shaped stubs are connected in opposite directions inside the ring. The antenna is designed on an FR-4 (lossy) substrate (h = 1.6 mm ∈r = 4.4). The antenna is compact with dimensions of 18 × 34.5 × 1.6 mm3 (0.127λ0 × 0.243λ0 × 0.011λ0 ), where λ0 is the free space wavelength at the lowest resonating frequency, i.e., 2.12 GHz. The designed antenna resonates at 2.12 GHz (2.04– 2.19 GHz), 2.59 GHz (2.43–2.77 GHz), 4.19 GHz (4.14–4.275 GHz), 5.69 GHz (5.52–5.965 GHz), and 10.35 GHz (10.01–11.31 GHz) frequencies. The proposed antenna is suitable for Bluetooth/WLAN/Wi-MAX/X-Band wireless applications. Keywords CPW · CST · Wi-Max · WLAN X-Band

1 Introduction The recent growth due to the high demand for wireless communication devices increased more advancements of multifunctional operational ability (WiMax/WLAN/Bluetooth, etc.) in the single-antenna device. Developing this multifunctional, lightweight, and compact antenna device is the principal motivation for scientists in the past some decades. The microstrip line feed antenna is the right candidate for this application. General methods for making antenna device compact are cutting slots on the patch as well as on the ground, shorting pins, and by increasing the electrical length with the modification of antenna design structure; this may give radiation pattern deterioration, backpropagation, low bandwidth, etc.

A. Dadhich (B) · M. Agarwal · J. K. Deegwal Government Engineering College Ajmer, Nh-8, Badiliya Circle, Ajmer, India e-mail: [email protected] M. M. Sharma MNIT, Jaipur, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_46

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WLAN, which is an augmentation to wired LAN, has incredible growth in modern technologies. It has frequency ranges for the antenna to operate 2.40– 2.484 GHz, 5.15–5.35 GHz, 5.725–5.85 GHz, WiMAX operates at 2.5–2.69 GHz, 3.4–3.69 GHz and 5.25–5.85 GHz, Bluetooth working range is 2.40–2.48 GHz, and X-Band operates in 8–12 GHz. Much research has been carried out for dual and multiband antenna to meet modern challenges. Different types of geometry have been described in the literature [1–11]. Although many design/shapes have been reported, all above antenna operates in the triple band with the considerably large dimension. The proposed antenna is compact and operates in five resonating bands. In this paper, the squared ring-shaped microstrip monopole antenna is presented in which U- and T-shaped stubs are embedded inside the ring-shaped monopole. The proposed antenna resonates at five frequencies which are 2.12, 2.59, 4.19, 5.69, and 10.35 GHz with return loss of −20.14, −16.77, −16, −15.65, and −17.255 dB, respectively. The proposed antenna is compact and has a good radiation pattern and wide bandwidth of 1293 MHz at 10.35 GHz. The proposed antenna may be used for Bluetooth, WLAN, WIMAX, and X-Band applications. Table 1 epitomizes a review of the previously reported literature to the proposed article.

2 Antenna Design Figure 1 epitomizes the physical geometry of the developed antenna. A 50  microstrip feed line is used to a square ring radiator. The size of the substrate is (L 1 ×W1 ), i.e., (34.5×18 mm2 ) and the ground plane is (L g ×Wg ), i.e., 8.7×18 mm3 . The antenna on the FR4 (∈r = 4.4 and h = 1.6 mm) substrate is analyzed, and the parameters of the designed antenna are recorded in Table 2. Antenna has square ring-shaped monopole in which U and T stubs are connected horizontally in opposite direction inside the square ring-shaped monopole. The dimensions are optimized to find desired resonant frequencies and bands. Antenna is resonating at five resonating frequencies, i.e., 2.12/2.59/4.1/5.69/10.35 GHz with return loss of −20.14 dB, −16.77 dB, −16 dB, −15.65 dB, and −17.255 dB, respectively, which covers Bluetooth, WI-Max, WLAN and X-Band wireless applications.

3 Results and Discussions The proposed structure is simulated on CST-2017 for 1–12 GHz frequency range. S 11 , bandwidth, surface current, and radiation patterns in x–z and y–z plane for respective frequencies are observed and analyzed.

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Table 1 Summary of designed antenna with previously reported antennas Published Antenna Antenna size work response L × W × h (Electrical size) (in mm3 )

Operating frequency range (GHz)

Gain (dBi)

Bandwidth (MHz)

1

Triple band

30 × 65 × 1.6 (0.245λ0 × 0.53λ0 × 0.01λ0 )

2.375–2.525 3.075–3.8 5.0–6.9

1.1–1.5 4.6–5.6 2.0–3.6

150 725 1900

2

Triple band

20 × 38.5 × 0.8 (0.166λ0 × 0.32λ0 × 0.006λ0 )

2.10–2.49 3.22–4.3 4.89–6.12

2.7–3.2 3.1–3.5 2.8–3.3

390 400 500

3

Triple band

23 × 36 × 1 (0.187λ0 × 0.29λ0 × 0.008λ0 )

2.35–2.59 3.31–3.93 5.07–6.35

2.67–3.19 240 2.05–2.24 620 2.66–3.48 1280

4

Triple band

28 × 32 × 1 (0.23λ0 × 0.266λ0 × 0.008λ0 )

2.29–2.88 3.26–3.88 4.17–6.07

3.8–4.4 4.0–4.65 1.9–3.5

590 620 1900

5

Triple band

33 × 50.9 × 0.8 2.8–3.0 (0.319λ0 × 0.492λ0 × 0.007λ0 ) 3.3–3.5 3.0–3.8

2.32 1.21 −6

200 200 800

6

Triple band

26 × 30 × 0.8 (0.212λ0 × 0.25λ0 × 0.006λ0 )

2.33–2.55 3.0–3.88 5.15–5.9

1.08–1.39 220 2.35–3.49 880 2.46–3.28 750

7

Triple band

23 × 38 × 1.6 2.28–2.56 (0.184λ0 × 0.304λ0 × 0.013λ0 ) 3.29–4.21 5.05–5.91

1.48–1.96 280 2.1–3.22 920 2.63–3.56 860

8

Triple band

40 × 40 × 0.8 2.35–2.58 (0.325λ0 × 0.325λ0 × 0.007λ0 ) 3.25–4.0 4.95–5.9

−0.3 0.9 3.8

230 750 950

9

Triple band

40 × 45 × 1.6 (0.28λ0 × 0.315λ0 × 0.011λ0 )

1.87 2.90 4.13

120 20 110

10

Triple band

37 × 40 × 1.58 2.3–2.75 (0.302λ0 × 0.327λ0 × 0.013λ0 ) 3.18–3.19 5.06–6.15

2.5 0.824 2.02

450 100 1090

11

Triple band

24 × 30 × 0.79 (0.2λ0 × 0.25λ0 × 0.006λ0 )

1.33–2.52 210 1.35–2.43 260 1.25–2.62 650

Proposed work

Penta band

18 × 34.5 × 1.6 2.04–2.19 (0.127λ0 × 0.243λ0 × 0.011λ0 ) 2.43–2.77 4.14–4.27 5.52–5.965 10.01–11.316

2.02–2.14 4.26–4.28 5.45–5.56

2.50–2.71 3.37–3.63 5.20–5.85

2.26 2.16 2.25 4.25 6.95

λ0 refers to the wavelength in free space at the lowest resonating frequency

149 334 134 450 1293

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Fig. 1 Design of proposed antenna with U- and T-shaped stubs a front view and b bottom view (ground) Table 2 Physical values of the proposed antenna

Parameter

Value (in mm)

Parameter

Value (in mm)

Lg

8.7

Wg

18

L1

34.5

W1

18

L2

12.9

W2

3

L3

3

W3

4

L4

16

W4

14.6

L5

8

W5

9

L6

1.5

W6

6

L7

3

W7

1.3

L8

1.4

W8

3.8

L9

4.6

W9

1.5

L 10

4

h

1.6

L 11

7

S

1.8

L 12

6

εr

4.4

L 13

6.8

L 14

7.4

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Fig. 2 Representation of S11 (dB) against frequency (GHz)

The simulated S 11 of the proposed antenna is illustrated in Fig. 2. The antenna is resonating at five operating bands. For S11 < −10 dB five resonating frequencies with operating bands are 2.12 GHz (2.04–2.19 GHz), 2.59 GHz (2.43–2.77 GHz), 4.19 GHz (4.1–4.27 GHz), 5.69 GHz (5.52–5.965 GHz), and 10.35 GHz (10.01– 11.31 GHz) with return loss of −20.14, −16.77, −16, −15.65, and −17.255 dB, respectively, as shown in Fig. 1. Real and imaginary parts of simulated input impedance concerning frequency are shown in Fig. 3. At each resonating frequency, imaginary part of impedance is near to zero and real part Re(Z) is 50 . This is the evidence for the resonance at the operating frequencies. Surface current distribution of proposed antenna at respective resonating frequencies is shown in Fig. 4. Figure 5 shows the simulated co- and cross-polarization radiation pattern in XZand YZ-plane at resonating frequencies. Good radiation pattern is observed at all resonating frequencies.

4 Conclusion A novel and compact design of square ring-shaped microstrip monopole antenna is presented. Antenna has ring-shaped monopole and U- and T-shaped stubs are connected inside the ring of monopole. Antenna resonates at five operating frequencies 2.12 GHz (2.04–2.19), 2.59 GHz (2.43–2.77), 4.19 GHz (4.1–4.27), 5.69 GHz (5.52–5.965), and 10.35 GHz (10.01–11.31) with return loss of −20.14, −16.77, −

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Fig. 3 Frequency versus input impedance plot

Fig. 4 Surface current distribution of proposed antenna at five resonance frequencies a 2.12 GHz, b 2.59 GHz, c 4.1 GHz, d 5.69 GHz, e 10.35 GHz

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Fig. 5 Radiation pattern of proposed antenna in both plane at a 2.12 GHz, b 2.59 GHz, c 4.1 GHz, d 5.69 GHz

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16, −15.65, and −17.255 dB, respectively. The achieved bandwidth for five bands is 149, 334, 134, 450, and 1293 MHz. Antenna shows good surface current distribution and radiation pattern for all resonating frequencies, real and imaginary impedance, and reflection coefficient is observed and analyzed for 1–12 GHz. Proposed structure is suitable for WLAN/WiMAX/Bluetooth/X-Band applications.

References 1. Li J, Guo J, He B, Zhang A, Liu QH (2016) Tri-band CPW-fed stub loaded slot antenna design for WLAN/WiMAX applications. Frequenz 70(11–12):521–526 2. Sim CYD, Chen HD, Yeh CH, Lin HL (2015) Small size triple band monopole antenna with a parasitic element. Microw Opt Technol Lett 57(2):342–348 3. Tang Z, Liu K, Yin Y, Lian R (2015) Design of compact triband monopole antenna for WLAN and WiMAX applications. Microw Opt Technol Lett 57(10):2298–2303 4. Liu G, Liu Y, Gong S (2016) Compact tri-band wide-slot monopole antenna with dual-ring resonator for WLAN/WiMAX applications. Microw Opt Technol Lett 58(5):1097–1101 5. Goswami SA, Karia D (2017) A compact monopole antenna for wireless applications with enhanced bandwidth. Int J Electron Commun 72:33–39 6. Yang X, Kong F, Liu X, Song C (2014) A CPW-fed triple-band antenna for WLAN and WiMAX applications. Radioengineering 23(4):1086–1091 7. Xu Y, Zhang C, Yin Y, Yang Z (2015) Compact triple-band monopole antenna with inverted-L slots and SRR for WLAN/WiMAX applications. Prog Electromagn Res Lett 55:1–6 8. Sun XL, Zhang J, Cheung SW, Yuk TI (2012) A triple-band monopole antenna for WLAN and WiMAX applications. In: The 2012 IEEE international symposium on antennas and propagation (APSURSI), Chicago, IL, pp 1–2 9. Gupta A, Chaudhary RK (2016) A compact planar metamaterial tripleband antenna with complementary closed-ring resonator. Wirel Pers Commun 88(2):203–210 10. Kundu A, Bhattacharjee AK (2015) Design of compact triple frequency microstrip antenna for WLAN/WiMAX applications. Microw Opt Technol Lett 57(9):2125–2129 11. Kumar A, Jhanwar D, Sharma MM (2017) A compact printed multistubs loaded resonator rectangular monopole antenna design for mutiband wireless systems. Wiley Int J RF Microw Comput Aided Eng 21147:1–10

1D Photonic Crystal Waveguide Based Biosensor for Skin Cancer Detection Application Sanchit Kundal, Abhinav Bhatnagar, and Ritu Sharma

Abstract In this letter, a new biosensing technique for cancer cell detection based on 1D photonic crystal is reported. The proposed structure is 1D photonic crystal with a gradient index lens in the middle of structure and two fluid layers around the lens. The shape of lens is also varied to study the effect on the transmission and reflected power. The study reported in this paper may prove to be interesting in early detection of cancer cell and its early treatment. By exposing, a gaussian wave light in a selected frequency band output power profiles are obtained. By changing the refractive index of fluid and shape and size of GRIN lens, skin cancer cells of various types can be detected. Keywords GRIN lens (gradient index lens) · RIU (refractive index unit) · PBG (photonic bandgap)

1 Introduction Cancer is one of the deadliest diseases whose detection if not done in earlier stage can lead to severe consequences. If the disease diagnosed in later stage it can also spread to other parts of the body. Skin cancer is one of them in which scientists are paying more attention so as to detect it at earlier stage. For such purpose, scientists are trying to use the optical properties of biosensors having very high sensitivity. The alternative layers of low- and high-refractive index materials lead to the formation of photonics crystal which have the properties of photonic band gap (PBG). Band of frequencies that cannot penetrate through the structure are called as PBG. Photonic S. Kundal (B) · A. Bhatnagar · R. Sharma Department of Electronics and Communication, MNIT, Jaipur, Rajasthan 302017, India e-mail: [email protected] A. Bhatnagar e-mail: [email protected] R. Sharma e-mail: [email protected] © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_47

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crystals periodicity can be in three directions and based on that they are categorised into different dimensional structures like one, two and three. For designing of a sensor, various studies are done based on 1D photonic crystals and photonic crystal slabs [3]. The cladding region of biosensors has the cavity for cancer cell where the infiltartion of the sample is done [4, 8]. When the condition for phase is matched the optical power gets transfer to the silica core from the cancerous cell [11]. Based on the sample, if there is defelction in the refrative index it will leads to the shifting of loss spectrum. Refractive index depends on the intensity of the light. Intensity profile has ring-shaped whose radius can be change in accordance with the refractive index. By utilizing the scanning slit beam profilers the variations can be observed. In this letter 1D photonic crystal with GRIN lens in center of structure is proposed. It had microfluidic channels around it. Different solutions can flow in the microfluidic channels and the change in their refractive index can be monitored in real time. When various fluids are passed through these channels it will lead to the variation in the refractive index which can be observed practically.

2 Proposed Model and Theory By using optical waveguide properties of 1D photonic crystals with a GRIN lens in the centre of structure a very sensitive biosensor is designed [4, 5]. Central GRIN lens is applied with two fluidic channels across it. The refractive index of these fluids can cause the change in the output transmission and reflected power profiles. So by monitoring the change in output power of monitor we can detect various cancer cell types and their charactertics. The proposed structure is ((AB)n FDF(AB)n ), where A and B are dielectric layers with high- and low-refractive indices, respectively. F is the microfluidic channel and D is GRIN lens. Here ‘n’ is the number of A, B layers [1]. The schematic of the structure is showed for n = 3 in figure [1, 4, 7]. Here, various structures with different lens shape of rectangular, star and hexagonal shaped have been observed as shown in Figs. 1, 2 and 3, respectively.

3 Results and Discussions It has been observed that the best output can be achieved for the structure having (AB)3FDF((AB)3), where layer A is zinc selenide (ZnSe) with refractive index of 2.6 and layer B is SiO2 with refractive index of 1.5. Here, F is the microfluidic channel and D is the GRIN lens. Frequency of gaussian source falls around 531 THz. Refractive index of different cancer and normal cell is studied in detail. The database given in Table 1 [2, 5, 6, 10] shows the refraction index of both normal and cancer cells. It can be observed from Table 1 [6, 7, 9] that the difference in refractive index of normal and cancer cell is very small. To capture this change, the sensor design must be highly sensitive.

1D Photonic Crystal Waveguide Based Biosensor for Skin Cancer … Fig. 1 Diagram of 1D photonic crystal waveguide based biosensor. The green layer (A) is ZnSe and blue layer (B) is SiO2 . Two fluidic channels (F) of black colour, which are around the central RECTANGULAR SHAPED GRIN lens (D) of colour purple and red layer is silicon substrate

Fig. 2 The green layer (A) is ZnSe and blue layer (B) is SiO2 . Two fluidic channels (F) of black colour, which are around the central STAR SHAPED GRIN lens (D) of colour purple and red layer is silicon substrate

Fig. 3 The green layer (A) is ZnSe and blue layer (B) is SiO2 . Two fluidic channels (F) of black colour, which are around the central HEXAGONAL SHAPED GRIN lens (D) of colour purple and red layer is silicon substrate

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Table 1 The detailed input refractive index database for basal, hela and jurkat cancer and normal cells Basal Normal 1.36 Basal Cancer 1.38 Hela Normal 1.368 Hela Cancer 1.392 Jurkat Normal 1.376 Jurkat Cancer 1.39

Fig. 4 Output transmission power of different cancer cells of rectangular lens

3.1 Rectangular Lens The basic structure has rectangular lens with two fluid channels applied across it. When fluid channels of normal and cancer BASAL cells are applied, 0.038837 mW difference in the peak is observed in output transmitted power. When fluid channels of normal and cancer HELA cells are applied, 0.043325 mW difference in the peak is observed in output transmitted power. When fluid channels of normal and cancer JURKAT cells are applied, 0.024678 mW difference in the peak is observed in output transmitted power. The results are shown in Figs. 4 and 5 depict the output transmitted and reflected power.

3.2 Star Shaped Lens The basic structure has a star shaped lens with two fluid channels applied across it. When fluid channels of normal and cancer BASAL cells are applied, 0.04599 mW difference in the peak is observed in output transmitted power. When fluid channels of normal and cancer HELA cells are applied, 0.057592 mW difference in the peak is observed in output transmitted power. When fluid channels of normal and cancer

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Fig. 5 Output reflected power of rectangular lens

Fig. 6 Output transmitted power of star shaped lens of various cancer cells

JURKAT cells are applied, 0.033847 mW difference in the peak is observed in output transmitted power. The results are shown in Figs. 6 and 7.

3.3 Hexagonal Lens The basic structure has a Hexagonal lens with two fluid channels are applied across it. When fluid channels of normal and cancer BASAL cells are applied [11], 0.038837 mW difference in the peak is observed in output transmitted power. When fluid channels of normal and cancer HELA cells are applied, 0.043324 mW difference in the peak is observed in output transmitted power. When fluid channels of normal and cancer JURKAT cells are applied, 0.024678 mW difference in the peak is observed in output transmitted power. The results are shown in Figs. 8 and 9.

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Fig. 7 Output reflected power of star shaped lens of various cancer cells

Fig. 8 Output transmitted power of hexagonal lens of various cancer cells

4 Comparison Table 2 shows the comparison in peak output differences of various cancer cells. By observing the comparison between the output power of the various lens, a waveguide with star shaped lens senses with maximum efficiency.

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Fig. 9 Output reflected power of hexagonal lens of various cells Table 2 The comparison of peak output power for basal, hela and jurkat cancer and normal cells Rectangular lens Basal 0.038837 Rectangular lens Hela 0.043325 Rectangular lens Jurkat 0.024678 Star shaped lens Basal 0.04599 Star shaped lens Hela 0.057592 Star shaped lens Jurkat 0.033847 Hexagonal lens Basal 0.033837 Hexagonal lens Hela 0.043324 Hexagonal lens Jurkat 0.024678

5 Conclusion In this letter, 1D photonic crystal based biosensor is designed that has the capability of monitoring of refractive index changes of the fluid in two microfluidic channels. This structure has a GRIN LENS in the centre with two microfluidic layers surrounded by a lens. By adding normal and cancer cells in the fluid layer the various changes in transmitted and reflected power at the output monitor of the sensor can be detected. Different cells always produce a unique spectrum. Thus, the proposed design offers an advantage of early and successful detection of cancer cells so that early cure of patients can be done.

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References 1. Bayat F, Ahmadi-Kandjani S, Tajalli H (2016) Designing real-time biosensors and chemical sensors based on defective 1-d photonic crystals. IEEE Photonics Technol Lett 28(17):1843– 1846 2. Chiavaioli F, Trono C, Giannetti A, Brenci M, Baldini F (2014) Characterisation of a label-free biosensor based on long period grating. J Biophotonics 7(5):312–322 3. Fan X, White IM, Shopova SI, Zhu H, Suter JD, Sun Y (2008) Sensitive optical biosensors for unlabeled targets: a review. Anal Chim Acta 620(1):8–26 4. Lee M, Fauchet P (2007) Two-dimensional silicon photonic crystal based biosensing platform for protein detection. Opt Express 15:4530–4535 5. Natesan A, Raja G, Sharma M, Kumar D (2018) Photonic crystal fiber based refractive index sensor for early detection of cancer. IEEE Sens J 1 6. Santi S, Musi V, Descrovi E, Paeder V, Francesco J, Hvozdara L, Wal P, Lashuel H, Pastore A, Neier R, Herzig H (2013) Real-time amyloid aggregation monitoring with a photonic crystalbased approach. ChemPhysChem 7. Schmitt K, Schirmer B, Hoffmann C, Brandenburg A, Meyrueis P (2007) Interferometric biosensor based on planar optical waveguide sensor chips for label-free detection of surface bound bioreactions. Biosens Bioelectron 22(11):2591–2597 8. Shamah S, Cunningham B (2011) Label-free cell-based assays using photonic crystal optical biosensors. Analyst 136:1090–1102 9. Wang X, Flueckiger J, Schmidt S, Grist S, Fard ST, Kirk J, Doerfler M, Cheung KC, Ratner DM, Chrostowski L (2013) A silicon photonic biosensor using phase-shifted Bragg gratings in slot waveguide. J Biophotonics 6(10):821–828 10. Yablonovitch E (2002) Photonic crystals: semiconductors of light. Sci Am 285:47–51, 54 11. Yaroslavsky AN, Patel R, Salomatina E, Li C, Lin C, Al-Arashi M, Neel V (2012) High-contrast mapping of basal cell carcinomas. Opt Lett 37(4):644–646

Ka Band Circularly Polarized Antenna with Defected Ground for Close Range Military Radar Target Applications Priya Kaith and M. M. Sharma

Abstract In this paper, a circularly polarized ultra-compact Ka band microstrip patch antenna has been designed for close range military target applications. The proposed antenna operates in Ka band over frequencies ranging from 24.92 to 33.32 GHz. The designed antenna has radiation efficiency of 94.7% and has a considerably good gain of 4.25 dBi with single element and can find its usage in close range military radar targeting applications. For uplink space satellite communication, 27.5 and 31 GHz are used; hence, by implementing the array of this proposed single element, the gain could be further enhanced and antenna could be used for the space satellite applications as well. Keywords Ka band · Military radar applications · Space satellite communication · Microstrip · Patch antenna · Wideband

1 Introduction Use of satellite technology has spread out its wings in almost all the spheres of today’s day-to-day functioning. Whether, its radio communication to broadcasting; weather forecasting to astronomy; or mapping, implementation of satellite technology has got its roots deeply rooted everywhere. Due to advent of increase in number of applications using satellite technology in the lower spectrum of the electromagnetic band, there is a continued congestion in lower frequency bands. Also, size of the devices is pretty enlarged to be able to work at lower frequencies, and therefore, there has been an emerging shift toward the usage of higher frequency bands. Using higher frequency bands for satellite communication is accompanied by various inherent advantages such as compact size of the devices, which means that designed devices P. Kaith (B) · M. M. Sharma Department of Electronics and Communication Engineering, Malaviya National Institute of Technology Jaipur, Jaipur, Rajasthan, India e-mail: [email protected] M. M. Sharma e-mail: [email protected] © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_48

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will be lightweight and could be easily integrated for space applications. Further, frequency reuse factor at such high frequency increases immensely, and hence, there is better allocation and utilization of the frequency bands. Ka band is the new star in making in the world of space satellite communication (ranging from 26 to 40 GHz), with communication satellites uplink in 27.5 GHz band. The development of planar structures operating in Ka bands has been recognized as significant milestones in future wireless communication systems, including satellite or radar communication [1, 2]. Further, the researchers have reported various approaches to achieve single band/wideband/broadband/dual-band CP antenna using different approaches, i.e., dual or synchronous feed, fractal structure, multilayer reflector antennas, ground plane, or radiating patch loaded with different stubs/slits/strips [3–7]. In our work, we have tried to explore Ka band and have designed a simple circularly polarized microstrip antenna for close range military radar target applications and whose array implementation may lead to its usage in space satellite communication. Proposed antenna has a defected ground and a two-step stepped inset feed for better impedance matching. Simulation results show operating impedance bandwidth ranging from 24.92 to 33.32 GHz and ARBW less than 3 dBi for theta = 90 and phi = 0 for few range of operating frequencies. The work has been showcased in four sections in this paper. Section 1 was about introduction to the proposed work, as illustrated in above paragraphs. Section 2 demonstrates the designing of antenna. Section 3 comprises of results and discussions, and Sect. 4 summarizes the conclusions derived out of the results and discussions.

2 Designing of Antenna The proposed antenna is designed and simulated in CST MWS software using 60 mil (1.524 mm) RO3003 as substrate material whose dielectric constant is 3, tangent loss is 0.0010, and conductor thickness of 0.035 mm. The size of proposed antenna is 10 × 10 mm2 and has a circular slot cut of 4 mm radius in the center of the ground. Antenna is fed by a two-step stepped feed having widths of 4.2 mm, 1.9 mm, and 0.85 mm, respectively. Stepped feeds provide better impedance matching and hence results in proper transfer of the power and energy. A circular patch of 2 mm radius is fed by the stepped feed. Figure 1 shows the dimensions and geometry layout of the front and back view of proposed antenna. Table 1 comprises of the optimized parameters used in designing of proposed antenna.

3 Results and Discussions On the benchmark of plots of overall gain, efficiency, S parameter (S 11 , dB), broadband circular polarization (3-dB axial ratio bandwidth (ARBW)), radiation patterns,

Ka Band Circularly Polarized Antenna with Defected Ground …

453

W

R

Rg

L

H3 W3

H2

W2

H1

W1

Front View

Back View

Fig. 1 Design of the proposed antenna

Table 1 Optimized parameters of proposed antenna

Parameter

Value (mm)

Parameter

Value (mm)

L

10

W1

4.2

W

10

W2

1.9

Rg

4

W3

0.8

R

2

H1

2.5

H2

1.5

H3

0.85

performance of proposed antenna is evaluated. Simulation of proposed antenna design results in an impedance bandwidth ranging from 24.92 to 33.32 GHz with peak gain of 4.25 dBi, which increases with increase in frequency in entire operating frequency range and an excellent radiation efficiency of 94.7% is displayed. Figure 2 depicts the impedance bandwidth (IBW) plot for the simulated design. Figure 3 illustrates the plot of gain over frequency, which is fairly good and is increasing with increase in frequency as per the basics of antenna theory. Proposed antenna provides peak gain of 4.25 dBi. Figure 4 displays the plot of radiation efficiency (in percentage) versus frequency, which is having maximum radiation efficiency of 94.7% and is quite excellent. Figure 5 displays the farfield radiation patterns for broadband left and right polarization at 27.5 and 29.5 GHz, which are little distorted due to very high operating frequency range. Figure 6 demonstrates that axial ratio bandwidth plot for directivity at phi = 0 and theta = 90 is having few of its minimum values below the bar of 3 dBi, thereby showcasing the behavior of circular polarization between the frequency range 27 and 33 GHz.

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S-Parameter (S11, dB)

0

-10

-20

-30

-40

24

26

28

30

32

34

32

34

32

34

Frequency (GHz)

Fig. 2 Impedance bandwidth plot for proposed antenna 5

Gain (dBi)

4 3 2 1 0

24

26

28

30

Frequency (GHz)

Fig. 3 Maximum gain over frequency plot 100%

% Efficiency

80% 60% 40% 20% 0%

24

26

28

30

Frequency(GHz)

Fig. 4 Plot of radiation efficiency versus frequency plot

Figure 7 displays the surface current distribution of the proposed antenna at 29 GHz at various instantaneous phases, viz. 0, 90, 180, and 270°, respectively. From the surface current distribution, it is observed that the resultant current vector rotates

Ka Band Circularly Polarized Antenna with Defected Ground …

LHCP

LHCP 30 60 90

5 -5 -15 -25 -35

0

RHCP

30

30 60

60

90

90

120

5 -5 -15 -25 -35

0 60 90 120

150

150

RHCP

30

120

120 150

455

150

180

180

(a)

(b)

Fig. 5 Farfield plot for left and right polarization radiation patterns at a 29.5 GHz and b 27.5 GHz, respectively 10

dBi

8 6 4 2 0

24

26

28

30

32

34

36

38

40

Frequency(GHz)

Fig. 6 ARBW plot less than 3 dBi for theta = 90 and phi = 0

Fig. 7 Surface current distribution at 29 GHz at phase 0°, 90°, 180°, and 270°, respectively

in clockwise at different time phase instant. This rotation signifies that proposed antenna’s LHCP rotation in +z-direction.

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4 Conclusion To summarize, an ultra-compact design of microstrip patch antenna with two-step stepped feed and a defected ground has been analyzed. Proposed antenna operates in Ka band, and it is designed for close range military radar target applications. Simulated impedance bandwidth of proposed antenna ranges from 24.92 to 33.28 GHz and has good gain characteristics of 4.25 dBi and displays an excellent efficiency of 94.7%. ARBW characteristics for directivity at phi = 0 has been showcased by the proposed antenna in the simulation results, thereby displaying circular polarization characteristics. As a part of future work, by implementing the array of the proposed antenna, gain can be increased and hence it can be used for space satellite communications as well, where uplink frequency is at 27.5 and 31 GHz. Due to its lightweight, compact size, wide IBW and 3-dB ARBW, array implementation of proposed antenna is a suitable candidate for space satellite communication applications as well. Acknowledgements One of the authors, Priya Kaith, is highly thankful to Indian Navy for sponsoring her M. Tech. course from the prestigious MNIT Jaipur, Rajasthan, India.

References 1. Densmore A, Jamnejad V, Wu T-K, Woo K (1993) K/Ka band antenna system for mobile satellite service. https://doi.org/10.1109/APS.1993.385387 2. Kandwal A (2017) Compact dual band antenna design for Ku/Ka band applications. Adv Electromagn 6(4):1–5. https://doi.org/10.7716/aem.v6i4.450 3. Leong SC, Sun R-T, Yip PH (2015) Ka band satellite communications design analysis and optimization. DSTA Horizons 4. Shavit R, Joffe R, Falek E (2013) Planar Ka band antenna for satellite communication based on metamaterial technology. In: 2013 IEEE international conference on microwaves, communications, antennas and electronic systems (COMCAS2013), Tel Aviv, pp 1–3. https://doi.org/ 10.1109/COMCAS.2013.6685235 5. Silva JS, García-Vigueras M, Debogovi´c T, Costa JR, Fernandes CA, Mosig JR (2017) Stereolithography-based antennas for satellite communications in Ka-band. Proc IEEE 105(4), 655–667. https://doi.org/10.1109/JPROC.2016.2633898 6. Sharmila D, Purnachandra Rao M, Subbarao PSV, Nagakishore Bhavanam S (2019) Design, simulation and fabrication of multiband antenna using HFSS. Int J Recent Technol Eng (IJRTE) 8(2). ISSN: 2277-3878 7. Jaiverdhan, Singhal S, Sharma MM, Yadav RP (2020) Epsilon shaped circularly polarized strip and slot loaded ultra-wideband antenna for Ku band and K band. Int J RF Microw Comput Aided Eng e22142 8. Wong K-L (2002) Compact and broadband microstrip antennas. Wiley 9. Balanis CA (2016) Antenna theory analysis and design. Wiley, Hoboken, NJ 10. CST Inc (2017) CST microwave studio suite

Design and Simulation of a Photonic Crystal-Based 2-D Octagonal-Shaped Optical Drop Filter Manish Kumar Pandey, Ritu Sharma, and Manish Jangid

Abstract A 2-D photonic crystal (PhC) based octagonal-shaped optical channel drop filter (CDF) resonance structure is suggested and simulated. The structure comprises of four ports. Port ‘A’ acts as an input and ‘B’, ‘C’ and ‘D’ are taken out as output ports. The plane wave expansion (PWE) method is used to evaluate the Photonic bandgap (PBG) as well as the distributions of electric fields. When the applied optical signal lies in the PBG range and also equal to the resonance wavelength of the structure then it enabled the structure to behave as a filter; otherwise, it would be performed as a normal waveguide. In the proposed structure for the wavelengths 1200–1600 nm, it behaves as a normal waveguide and for 1650–1700 nm. Wavelengths behave like a drop filter. By using scatterer and coupling rods in the proposed structure, we obtained 98–99% of drop efficiency for the wavelengths 1650 and 1700 nm. Keywords Photonic crystal fiber · Waveguide · Resonance wavelength · Photonic bandgap · 2-D PhC

1 Introduction The generation of photonic bandgaps occurs in PhCs due to periodicity phenomena that limit the light propagation at particular frequency ranges. On based upon these photonic bandgaps, a large number of optical devices (active and passive devices) that leads the contribution in the field of optical integration significantly. For practical applications in an optical system, various optical devices like lasers, PhCs-based fiber and channel drop filters (CDFs) are being researched and developed. Channels add/drop filter is a very compact and highly integrated device, which is very useful for DWDM (dense wavelength division multiplexing) in optical communications systems. In optical drop filters, ring resonators are key components to drive its functioning properly. M. K. Pandey (B) · R. Sharma · M. Jangid Department of Electronics and Communication, Malaviya National Institute of Technology Jaipur, Jaipur 302017, India e-mail: [email protected] © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_49

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The working principle of photonic crystals in optical systems is similar to the semiconductor that used in an electronic system. In the PBG frequency ranges, applied light waves will not be able to propagate through the photonic crystal structure and would experience total internal reflection (TIR). Therefore, by creating different defects in the photonic crystal structure, different optical devices can be designed. Actually, by creating defects in PhCs, light can propagate only in specific directions, forms different optical devices. As known, in optical integrated circuits, all the blocks should function suitably in optical frequencies. Photonic crystals are of the most confident ways of designing the optical components. PhCs attract great attention due to their strong photon confinement effects and PBGs, which make them appropriate for designing of different optical components like filters, optical logic gates, etc. can be considered based on photonic crystals. In an optical system, filters play a key role in different devices. Therefore, considering photonic crystals-based filter plays great importance which has been studied intensely in recent years. By designing defects in circular/closed shaped, characteristics of resonators can be increased and thus the accuracy and precision of filters. In such structures, resonance wavelengths of filters can be tuned and chosen as per PBGs and resonance frequencies of resonators. Therefore by designing photonic crystal filters with closed path shaped defects, the benefits of a structure can be utilized efficiently and can be designed appropriately. The proposed structure is being analyzed using the plane wave expansion (PWE) method in R Soft. PWE is an important computational technique that can be used to solve Maxwell’s equations. It is mostly used to determine the bandgaps of photonic crystal structures in specific geometries. Many different types of research were reported in this area. In [1–3], an optical channel drop filter based on 2-D photonic crystal had been investigated with ring resonators and had exhibited acceptable optical properties. In [4], PhC-based highquality PBG heterostructures were realized and also characterized. In [5], PhC-based nanocavities with point defects are fabricated that possess a high-quality factor. In [6], a highly efficient design of a two-channel wavelength de-multiplexer in the visible region is presented with finite-difference time-domain simulations. In [7], investigating the effects of structural parameters on the optical characteristics of add-drop filters. In [8], design and simulation of high sensitive photonic crystal waveguide sensor. In [9], a novel ultrahigh birefringence dual-core tellurite glass photonic crystal fiber is designed in this paper. We analyze the characteristics of birefringence and coupling at the two communication windows by using the finite element method. In [10], In this paper, an all-optical switch has been proposed based on a photonic crystal T-type ring resonator with very low switching power consumption and small switching time. In [11], an ultra-narrow band channel drop filter based on embedded photonic crystal ring resonator with distributed coupling for optical wavelength division multiplexing was designed. In [12], a novel optical ring-shaped de-multiplexer based on photonic crystals was reported. In [13] a new configuration of an add-drop filter is designed. The proposed structure indicated high efficiencies in filtering applications [14], and another filter based on photonic crystal was designed and analyzed. The proposed filter was based on symmetry matching

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between the defect and waveguide modes in a square lattice photonic crystal. In [15], a defect plasma photonic crystal-based multichannel filter was suggested. The structure can be a good candidate for multichannel filters for EM waves. In another research considering hexagonal-shaped defect [15], channel drop filter was designed. The proposed structure was appropriated for Coarse Wavelength Division Multiplexing.

2 Designing Methodology The proposed structure consists of 30 * 23 Si rods of a 2-D square lattice PhC along with air as background dielectric material. The designing parameters of the structure are: Rod radius (r) = 0.15b, lattice constant (b) = 0.6 µm and refractive index of Si rods (n) = 3.5. Figure 1 shows a detailed configuration of the structure with four input–output ports. Here, port A taken as input port and other three ports B, C and D are taken as an output port. By creating an octagonal-shape resonator in between two waveguides, at corresponding resonant frequency energy can be coupled from one waveguide to another waveguide. In the proposed structure some scatterer rods are added to reduce the effect of back reflections. These rods are shown in light green color in Fig. 1. The diameter of scatterer rods is adjusted in order to minimize back reflections. The optimized radius of the scattering rod is set to r  = 0.5 * r and some coupling rods (depicted in blue color in Fig. 1) are also used in the structure to achieve maximum drop efficiency. When the signal is launched to port ‘A’, at particular resonance

Fig. 1 Suggested structure of optical channel drop filter

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frequency the signal will be coupled to the octagon resonator and would pass to port C and port D that functions as a drop filter otherwise the pass-through port B without coupling and behave as a waveguide.

3 Simulation and Results Figure 2 illustrates the PBG regions for TE/TM modes of the proposed structure. Here we observed that PBG regions for TE mode lie between 0.35 µm < λ < 0.48 µm and in 0.638 µm < λ < 0.67 µm. These wavelengths regions exist in optical NIR and IR regions. So these can be used for optical communication systems. Figure 3 shows the propagation of light of wavelength 1400 nm. Here we observed that when scatterer and coupling rods are not used then it behaves like a normal waveguide but when they are employed then it doesn’t work as a normal waveguide and light gets absorbed by the structure. Figure 4 shows the propagation field strength of light with wavelength 1500 nm. Here we observed that when scatterer and coupling rods are not used, then it behaves as normal waveguide and maximum light propagates through the same waveguide but, when scatterer and coupling rods are employed then no field propagates to any of the output ports which employs that addition of scatterer and coupling rods affects the flow of light through defects. Figure 5a shows the propagation of light of wavelength 1600 nm. In this we can observe as the wavelength of input light increases some part of the light is getting coupled from one waveguide to another waveguide as we can see in Fig. 5b. Figure 6a shows the propagation of light of wavelength 1650 nm. Here it is observed more than 98% of light coupled from one waveguide to another and no light propagates through the same waveguide. Figure 6b exhibits 40–50% of light is getting coupled at the same time around 80% of light is propagated through the same waveguide resulting in a poor drop in wavelength without the use of the scatterer and Fig. 2 TE/TM photonic bandgap of structure

Design and Simulation of a Photonic Crystal-Based 2-D …

461

Fig. 3 Schematic representation of pulse propagation and transmitted powers in different ports of CD filter a with coupling and scatterer rods at λ = 1.4 µm, b without coupling and scatterer rods

Fig. 4 Schematic representation of pulse propagation and transmitted powers in different ports of CD filter a with coupling and scatterer rods at λ = 1.5 µm, b without coupling and scatterer rods

coupling rods. So here it can be observed that the addition of scatterer and coupling rods improves the filtering efficiency of the structure. Figure 7a shows the propagation of light of wavelength 1700 nm. Here it is observed more than 98% of light coupled from one waveguide to another and no light propagates through the same waveguide. Figure 7b exhibits 70–80% of light is getting coupled at the same time around 80% of light is propagated through the same waveguide resulting in a poor drop in wavelength without the use of the scatterer and coupling rods. So here also it can be observed that the addition of scatterer and coupling rods improves the filtering efficiency of the structure.

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Fig. 5 Schematic representation of pulse propagation and transmitted powers in different ports of CD filter a with coupling and scatterer rods at λ = 1.6 µm, b without coupling and scatterer rods

Fig. 6 Schematic representation of pulse propagation and transmitted powers in different ports of CD filter a with coupling and scatterer rods at λ = 1.65 µm, b without coupling and scatterer rods

Here, Table 1 shows that the output field strength of channel drop filter is much higher when coupling and scatterer rods are added in comparison with coupling and scatterer rods are not added. As we observed that without using coupling and scatterer rods we were getting drop efficiency of around 60–65% only. But, on applying coupling and scatterer rods; we obtained a drop efficiency of 98–99% and a forward gain of 98–99% at port D.

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Fig. 7 Schematic representation of pulse propagation and transmitted powers in different ports of CD filter a with coupling and scatterer rods at λ = 1.7 µm, b without coupling and scatterer rods

Table 1 Output field strength comparison of channel drop filter with and without coupling and scattering rods Input signal’s wavelengths (in nm) Port A

Field strength (in %) without coupling and scatterer rods

Field strength (in %) with coupling and scatterer rods

Port B

Port D

Port C

Port B

Port D

Port C

1400

90–95

0

0

5–10

0

0

1500

90–95

0

0–5

0–5

0–2

0–2

1600

45–50

35–40

25–30

5–10

0–5

0–2

1650

85–90

45–50

35–40

0–1

98–99

0–5

1700

85–90

80–85

0–5

0–1

98–99

0–5

4 Conclusion The paper comprised a 2-D PhC-based octagonal-shaped structure for a new optical channel drop filter. A configuration of 30 * 23 Si rods with air as background material is being used to construct the proposed structure. For simulations, the PWE method has been used. The structure has been analyzed for two different TE PBG regions, i.e., 0.35 µm < λ < 0.48 µm and in 0.638 µm < λ < 0.67 µm. The filter performance quite significant in TE PBG wavelength regions. The proposed structure shows a drop in the efficiency of almost 100% with the use of scatterer and coupling rods for the wavelengths 1650–1700 nm. So the proposed structure can be utilized as a channel drop filters in WDM for an optical communication system.

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References 1. Li L, Liu GQ (2013) Photonic crystal ring resonator channel drop filter. Optik 124:2966–2968 2. Kee CS, Ko DK, Lee J, Lim H (2006) Functional optical filters based on two-dimensional photonic crystals. J Kor Phys Soc 48:978–981 3. Georgeta Popescu D (2013) Two-dimensional photonic crystal with different symmetries for waveguides and resonant cavities applications. UPB Sci Bull Ser A 75(3):237–252 4. Liu GQ, Hua HH, Liao YB, Wang ZS, Chen Y, Liu ZM (2011) Synthesis and photonic bandgap characterization of high-quality photonic crystal heterostructures. Optik 122:9–13 5. Tekeste MY, Yarrison-Rice JM (2006) High-efficiency photonic crystal based wavelength demultiplexer. Opt Express 14:7931–7942 6. Ren HL, Jiang C, Hu WS, Gao MY, Wang JY (2006) Design and analysis of two-dimensional photonic crystals channel filter. Opt Commun 266:342–348 7. Rafiee E, Emami F (2016) Investigating the effects of structural parameters on the optical characteristics of add-drop filters. Optik 127:1690–1694 8. Goyal AK, Pal S (2015) Design and simulation of high sensitive photonic crystal waveguide sensor. Optik 126:240–243 9. Fan ZK, Li SG (2014) Analysis of the polarization beam splitter in two communication bands based on ultrahigh birefringence dual-core tellurite glass photonic crystal fiber. Opt Commun 333:26–31 10. Mansouri-Birjandi MA, Ghadrdan M (2013) All-optical ultra-compact photonic crystal switch based on nonlinear microring resonators. Int Res J Appl Basic Sci 4(4):972–975 11. Almasian MR, Abedi K (2016) A proposal for optical WDM using embedded photonic crystal ring resonator with distributed coupling. Phys E 79:173–179 12. Rafiee E (2017) Design of a novel all-optical ring shaped de-multiplexer based on twodimensional photonic crystals. Optik 140:873–877 13. Zhang J et al (2018) A novel photonic crystal ring resonator configuration for add/drop filtering. Photonics Nanostruct 30:14–19 14. Zhang T et al (2018) Photonic crystal filter based on defect mode and waveguide mode symmetry matching. Opt Commun 428:53–56 15. Saranya D et al (2018) Design and analysis of multi-channel drop filter using dual L defected hexagonal photonic crystal ring resonator. Digit Commun Netw (in press)

Frequency Reconfigurable/UWB Slot-Antenna with Switchable Resonant Function M. M. Sharma, Indra Bhooshan Sharma, and Joohi Garg

Abstract A frequency reconfigurable/UWB slot antenna with switchable resonant function is presented. The ultra-wideband characteristic from 2 to 13 GHz is obtained. The inverted U-shaped ring slot is induced in ground plane and embedding a PIN diode (D = 0.5 × 0.5 mm2 ) in this slot, a switchable band-resonance function is obtained. By changing the states of PIN diode, the antenna exhibits different characteristics. Good radiation pattern is observed in the desired frequency band. The simulated results of the proposed antenna are good candidate for ultra-wideband and cognitive radio application. Keywords Circular UWB antenna · Rectangular slot · Inverted U-ring shape slot · C-band · X-band

1 Introduction Recently, reconfigurable ultra-wideband (UWB) antenna for 3.1–10.6 GHz application is designed due to its many inbuilt virtue such as high data rate, lightweight, wide bandwidth, omni-directional pattern and low profile. Ultra-wide frequency bands have many narrow frequency bands for different-different applications (i.e., 3.4–3.69/5.475–5.725 GHz WiMAX, 5.15–5.35/5.725–5.825 GHz WLAN, 7.25– 7.75 GHz X-band downlink satellite communication. 8.025–8.4 GHz ITU band, 8.5–9.5 GHz deep space communication, 11.7–12.2 GHz satellite Ku downlink band communication and 8–12 X-band), which is achieved by reconfigurable antenna with single antenna at a time. Thus, one antenna works multi-functional operation as a M. M. Sharma · I. B. Sharma (B) · J. Garg Department of Electronics and Communication Engineering, Malaviya National Institute of Technology Jaipur, Jaipur 302017, India e-mail: [email protected] M. M. Sharma e-mail: [email protected] J. Garg e-mail: [email protected] © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_50

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multiple antenna in same antenna at same time instant. As above describe application band in UWB operating narrow, dual and wide frequency band is responsible for cognitive application. The many bands operated UWB antennas have been investigated and designed in the literature [1–5]. Different types of slots and radiating stubs are used to obtain planar UWB antennas. Many single to dual band and vice versa types antenna have been investigated and designed in the literature. Nowadays, huge demand requires to design such type antenna for improvement in further characteristics and performance of UWB antenna [6–9]. In this paper, A frequency reconfigurable/UWB slot-antenna with switchable resonant function is introduced. The resonating frequency bands are mainly dependent on the location and size of rectangular slot as well as u-ring inverted shape slot in the backside as ground plane. In addition, single and dual switchable resonating frequency operations are obtained by PIN diode with suitable position in the slot.

2 Antenna Design The antenna structure consists of micro-strip feed to circular patch, and ground plane with rectangular slot (1.75 × 6 mm2 ), and inverted U-shaped ring slot in the ground plane. The FR-4 dielectric substrate is used with 4.3 relative permittivity, 0.025 loss tangent and 1.53 mm thickness. The designed antenna is shown in Fig. 1 and has a simple geometry and overall dimension is 40 × 30 × 1.53 mm3 . For impendence matching, a gap ‘g’ is inserted between ground plane and radiating patch. The final parameter of proposed antenna has been illustrated in the given list which are L g =

Fig. 1 View (front and backside) of the proposed antenna

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467

12 mm, W f = 3.1 mm, g = 1 mm, W = 30 mm, L = 40 mm, W 2 = 1.5, L 2 = 4, g2 = 0.5, and L 3 = 2.5, R = 9 mm.

3 Results and Discussions All results of designed antenna have been analyzed as well as simulated by the CST MWS suit [10]. The S11 (return loss) curve versus frequency graph is shown in Fig. 2. The ultra-wideband characteristic is obtained for Simple Monopole antenna (SMA) design as first step in term of S11 (return loss) from 2 to 13 GHz. In the second step, rectangular center slot inserting at top of ground plane. Due to this perturbation, S11 result switch UWB frequency band to dual resonating frequency band, i.e., 2.94– 6.75 GHz and another operating frequency band 7.83–11.28 GHz which is almost cover the lower UWB bandwidth range. By inserting the inverted ring U-shaped-slot in the ground plane and embedding a PIN diode (D = 0.5 × 0.5 mm2 ) in this slot, a switchable band-resonance function is obtained. The antenna exhibits different characteristics by analyzing different states of PIN diode. For PIN diode OFF condition, the designed antenna is a natural ultra-wideband antenna and provides an impedance-bandwidth 3–10.8 GHz. When PIN diode is in ON condition, the antenna provides an operating frequency 2.97–12 GHz with dual band-resonance function 2.97–4.84 GHz and 6.05–12 GHz to avoid the interference from WiMAX (i.e., 3.4– 3.69 GHz), C-band (i.e., 3.8–4.2 GHz downlink), INSAT FSS (i.e., 4.4–4.8 GHz), and another resonating band is responsible for X-band satellite communication (i.e., 7.25–7.75 GHz for downlink), ITU 8-GHz band (i.e., 8.025–8.4 GHz), X-band for deep space communication (i.e., 8.5–9.5 GHz), satellite Ku-band communication (i.e., 11.7–12.2 GHz, downlink) and (i.e., 8–12 GHz) for X-band. Figure 3 has the surface current (distribution) for operating frequencies 3.4 and 6.7 GHz. The current density at 3.4 GHz in figure shows more current is distributed at the edge of inverted U-shaped slot which is responsible for the first resonance. The second resonance depends on center rectangular slot edge at 6.7 GHz.

Fig. 2 S11 curve versus frequency for different antenna design step

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Fig. 3 Surface current (distribution) plot of proposed reconfigurable antenna at resonating frequencies 3.4 and 6.7 GHz

Figure 4 realized gain comparison graph against frequency for all the four step design of proposed antenna. It is observed that realized gain characteristic of antenna is satisfactory for preferred frequency sort. The maximum value of realized gain is 2.9 dBi achieved at 4.5 GHz frequency. Minimum realized gain is obtaining−1.41 dBi for rejection band. Figure 5 exhibits the xz-field and yz-field radiation pattern in terms of cross as well as co-polarization separately. The plot is shown for 3.4 and 4.6 GHz at the resonant frequency band when the diode is ON condition. Cross polarization should be less then co-polarization for an effective antenna. The desired polarization component direction is same with respected to co-polarization component. On the other hand, cross-polarization component has 90° phase shift.

Fig. 4 Realized gain plot versus frequency of reconfigurable antenna

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Fig. 5 Simulated xz-plane and yz-plane with (cross and co) polarization-radiation plot of reconfigurable antenna at two resonating frequencies a 3.4 GHz, b 4.6 GHz

4 Conclusions A frequency reconfigurable/UWB slot-antenna with switchable resonant function is presented. The resonating frequency bands are mainly dependent on the location and size of rectangular slot as well as inverted shape ring slot in the backplane. In addition, single and dual switchable resonating frequency operations are obtained by PIN diode with suitable position in the slot. The proposed antenna structure performance is analyzed by many simulated results, i.e., return loss, radiation pattern, surface current and gain. Reconfigurable circular-disk UWB is a good candidate antenna with switchable operating bands for UWB applications.

References 1. Ebadzadeh SR, Nourinia J, Ghobadi Ch (2014) Extremely UWB/multiresonance monopole antenna with dual band-notch function. Microw Opt Technol Lett 56(11):2628–2630

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2. Ojaroudi N, Ojaroudi M (2013) A novel design of reconfigurable small monopole antenna with switchable band-notch and multi-resonance functions for UWB applications. Microw Opt Technol Lett 55(3):652–656 3. Li Y, Li W (2014) A circular slot antenna with wide tunable and reconfigurable frequency rejection characteristic using capacitance loaded split-ring resonator for UWB applications. Wireless Pers Commun 78(1):137–149 4. Ojaroudi N, Ghadimi N, Ojaroudi Y, Ojaroudi S (2014) A novel design of microstrip antenna with reconfigurable band rejection for cognitive radio applications. Microw Opt Technol Lett 56(12):2998–3003 5. Kumar A, Sharma IB, Sharma MM (2016) Reconfigurable circular disc monopole UWB antenna with switchable two notched stop bands. In: International IEEE India conference, INDICON 6. Al-Husseini M, Costantine J, Christodoulou CG, Barbin SE, El-Hajj A, Kabalan KY (2012) A reconfigurable frequency-notched UWB antenna with split-ring resonators. In: Proceedings of Asia-Pacific microwave conference, pp 618–621 7. Ojaroudi M, Ghadimi N (2014) Reconfigurable band-notched small square slot antenna with enhanced bandwidth for octave-band, multiresonance applications. Microw Opt Technol Lett 56(8):1960–1965 8. Badamchi B, Norrinia J, Ghobadi C, Shahmirzadi AV (2014) Design of compact reconfigurable ultra-wideband slot antenna with switchable single/dual band notch functions. IET Microw Antennas Propag 8(8):541–548 9. Kumar A, Sharma IB, Saraswat RK, Sharma MM (2016) Dual band-notched circular disc monopole UWB antenna with switchable five notched stop bands. In: Proceedings of the Asia-Pacific microwave conference 10. CST Microwave Studio (2013) User’s manual. www.cst.com

Optimization of Sensing Time for Efficient Spectrum Utilization in NOMA Based Cognitive Radio Networks Deepika Rajpoot and Pankaj Verma

Abstract The Cognitive Radio (CR) network provides the solution to the spectrum deficiency problem by enhancing the spectrum utilization. In recent years, Nonorthogonal multiple access (NOMA) has also gained significant interest as a popular access technique for 5G networks. It also helps to improve the user’s quality of service requirement by allocating the available resources dynamically. The simultaneous wireless information and power transfer (SWIPT) is a technique for the transfer of wireless information and power simultaneously for the power-limited wireless networks. In this paper, we are focusing on the optimization of the sensing time in NOMA based CR networks with SWIPT for maximizing the throughput of the available spectrum. Keywords Cognitive radio · NOMA · Spectrum utilization · Optimization

1 Introduction As a spectrum sharing system, the CR can be utilized to enhance the spectrum utilization of wireless networks. Here, the secondary user (SU) is permitted to access the unused spectrum of the primary user (PU). But, the SU is not permitted to disrupt the ordinary PU’s communication. The SU can access the allocated PU’s band only when the PU is absent and as PU is detected SU has to leave the channel [1, 2]. The PU’s absence or presence is observed with the help of spectrum sensing algorithms. The energy detection is the widely used spectrum sensing algorithm due to its simple implementation. It is based on the comparison of PU’s energy statistics signal with a fixed threshold [3–6]. The effectiveness of the above energy detection algorithm is based on false alarm probability) and the probability of detection (Pd ) where D. Rajpoot (B) · P. Verma Department of Electronics and Communication Engineering, National Institute of Technology, Kurukshetra, Haryana, India e-mail: [email protected] P. Verma e-mail: [email protected] © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_51

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the former represents faithful detection of an unused channel while later indicates accurate detection of PU presence. In [7], a sensing-throughput trade-off strategy was suggested to find the best sensing time to enhance SU’s throughput. For the optimal sensing duration, the SU’s throughput has to be maximized under the constraint of PU protection. To fulfill the above requirement, a multiple channel CR was proposed [8], which enhances SU’s throughput by enabling the SU to access several idle subchannels simultaneously. Later on, to enhance the SU’s throughput, the power optimization of multiple channels was proposed but this can be achieved only in the absence of PU [9]. In all the ideas, when PU was detected as absent, the SU still cannot utilize the spectrum. Therefore, to overcome the above drawbacks, NOMA has been proposed to enhance the spectrum efficiency of 5G communications. In the NOMA, many users can be combined on identical subchannel by involving superposition coding at the transmitter and successive interference cancellation (SIC) at the receiver [10]. Therefore, when PU is not absent, the SU can still access the spectrum to enhance the throughput by using NOMA. Also, if the non-orthogonal property of NOMA is combined with SWIPT techniques, the spectrum efficiency can be improved. The SWIPT enables the transmission of the power and data concurrently by introducing basic changes in the receiver design. Thus, the application of NOMA and SWIPT can help in the improvement of spectrum utilization [12]. In [13], the author has formulated a new objective function by the name of ‘Spectrum Utilization’ that considers the spectrum used by both PU and SU for the optimization of threshold and sensing duration and it indicates the comparison between three different optimization techniques. In this paper, we have considered the same spectrum utilization function for optimizing the sensing time with an objective of maximizing the possible throughput of the network. The rest of the paper is organized as follows: In Sect. 2, we present the system model of CR network and the spectrum utilization of the NOMA based CR network. Section 3 describes the proposed model for the optimization algorithm for sensing time and maximum throughput. Section 4 provides the simulation results and the conclusion is given in Sect. 5.

2 System Model The system model of CR-NOMA network is shown Fig. 1. In this figure, we consider the CR-NOMA network which combined of N = 3 secondary users, a PU and a base station (BS). In this, PU and SU harvest the energy first after this PU and SU transmit independent information to BS. Besides, SU produce the interference with PU’ s data transmission when spectrum sensing is performed by SU. Figure 2 shows the frame structure of overlay CR-NOMA network in which the frame is split into two distinct time slots τ and 1 − τ . Downlink sub slot τ is utilized for SWIPT and spectrum sensing while uplink sub slot 1 − τ is utilized for transmitting the SU’s data. During the downlink sub slot, PU and SU collect energy and PU transmits its data concurrently by using harvest-then-transmit protocol.

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Fig. 1 Network design of CR-NOMA [12]

Fig. 2 Frame design of CR network [12]

In [11], the author has proposed the harvest-then-transmit protocol for wireless powered communication network where BS transmit wireless energy to all users in downlink sub-slot while users broadcast their independent information to the BS in the uplink sub-slot using their independently harvested energy by Time-divisionmultiple-access (TDMA). So, the harvested energy during the downlink sub-slot can be written as: εibs = τ Pbs

(1)

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where, Pbs represents the BS transmit power at the transmitter. Then the transmit power for downlink sub slot can be expressed as: PT =

τ Pbs 1−τ

(2)

This network has applied the uplink NOMA to the secondary network during the sub slot 1 − τ . In the Uplink NOMA, more power is allocated to the nearest user while less power is allocated to the farthest user that is depend on the quality of service requirementthe hence BS employs SIC at the receiver side while all users can be combined on the identical sub channel by applying superposition coding at the transmitter side. The user who has good channel states will be decrypted first and users that are decrypted after it will be behaved as noise or interference. The h n represents user’s channel gain and assume that the users are arranged as h 1 > h 2 > · · · . . . h n while n(t) is a white gaussian noise which has zero mean and noise density N0 (W/Hz). Therefore, the SNR and possible throughput for the nth user of the NOMA can be written as [12]: SNRn =

Kn = W

N 

log2 1 +

n=1

K ns = W

N  n=1

K np

1+ 

Pn γn N j=n+1

1+

Pn γn N



j=n+1

Pn γn N

log2 1 +

(3)

Pj γ j 

(4)

Pj γ j 

Ps gs + j=n+1 P j γ j   Ps h s = W log2 1 + N0 W

(5)

(6)

Theorem: the closed form solution of the optimal power distribution for uplink NOMA based CR network can be given as [12]: P1 = PT −

N 

Pn

(7)

n=1

Pnmax ≥ Pnmin =

1 2 γn

N j=n+1 R j (W =1)



Rn 2 (W =1) − 1 , ∀n ∈ {1, 2, ....N − 1}

(8)

where, Pnmin is the minimum power needed to fulfill the Quality of Service (QoS) requirement for user n and Pnmax is the maximum transmit power for the nth user. When we replace the value of PT with Eq. (2), the throughput of the SU can be framed as:

Optimization of Sensing Time for Efficient Spectrum …

Rsum



⎞ N − n=1 Pn γ1 ⎠ = log 2⎝1 +  ps gs + Nj=n+1 P j γ j   N  Pn γn + log 2 1 +  ps gs + Nj=n+1 P j γ j n=2 ⎛

475

τ Pbs 1−τ

(9)

where, K np is the possible throughput of the PU while K n and K ns is the possible throughput of the SU at the absence and presence of the PU, ps is the transmission power of PU, gs is the normalized channel gain of PU, the normalized channel condition of the nth user is given by γn = Nh0nW , W is the transmission bandwidth of the sub band and N is the number of users. In this model, SUs essential to detect the PU’s licensed spectrum intermittently. PU’s radio signal is received by BS independently at the time of spectrum sensing and received signal at SU is tested for two hypotheses H 0 and H 1. When the PU is present, the received signal can be written as: H1 : y(n) = s(n) + u(n)

(10)

When the PU is not present, the received signal can be written as: H0 : y(n) = u(n)

(11)

where u(n) is represented as a white gaussian noise with zero mean and variance σu2 .Similarly, s(n) is a random variable that is represented as PU signal with zero mean and variance σs2 . In the Signal to Noise Ratio (SNR), signal and noise does not σ2 depend each other. γ = σs2 is represented as SNR of PU under the hypothesis H 1 . u For deciding of the existence of PU, the i-th CR uses the following test statistic [12]: T (y) =

M 1  |y(n)|2 M i=1

(12)

Traditional energy detector computes the energy T (y) related with received signal and corelate it with a predefined threshold (Yth ) to choose among the two hypotheses. For M number of samples, H 1 will be decided by Neyman-Pearson criteria if, p(y/H1 ) > Yth p(y/H0 )

(13)

For a given threshold Yth , The performance metrics pd and p f will be given as:

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⎞ 2 Y − σ th u ⎠ p f (Yth , τ ) = Q ⎝  2 4 σ M u ⎛ ⎞ 2 Yth − (1 + γ )σu ⎠ pd (Yth , τ ) = Q ⎝  1 (1 + 2γ )2 σu4 M ⎛

(14)

(15)

Q(x) is a standard gaussian complementary distribution function and it is given as: 1 Q(x) = √ 2π

∞

 t2 dt exp − 2 

(16)

x

When we select an appropriate probability of detection (pd ), pf in terms of pd can be expressed as: pf = Q



(1 + 2γ )Q −1 ( pd ) + γ



τ fs

(17)

pd in terms of p f can be expressed as:  pd = Q

√

     1 Q −1 p f − γ τ f s 1 + 2γ

(18)

2.1 Objective Function-Spectrum Utilization In order to understand the formulation of the term spectrum utilization, we need to understand the general frame design used in CR networks. The general frame structure is shown Fig. 2. Here, T represents the total frame time, τ is the time slot used by SUs for sensing and slot (T − τ ) is used for transmission of the data by SUs if the band is detected as idle. The term spectrum utilization considers the spectrum used by both PUs and SUs at appropriate times. This term can be written mathematically [13] as:    U (τ ) = (T − τ )(1 − θ) 1 − p f 1 − Pp   + (T − τ )(1 − pd ) )θ 1 − Pip + θ τ + θ (T − τ ) pd where, θ represents the spectrum occupancy and it is given as:

(19)

Optimization of Sensing Time for Efficient Spectrum …

θ=

α α+β

477

(20)

α and β represents the mean occupation and idle time per frame. Pp is the probability of interference in perfect sensing case. The probability of wrong decisions is assumed as zero in perfect sensing. But there may still be interference between PU and SU transmission when PU reappears in between two sensing instants. The mathematical expression is given as [13]: Pp = 1 −

   T −τ β 1 − exp T −τ β

(21)

In imperfect sensing, where there is a miss detection, the SU starts transmission on the band in the presence of PU, which results in the collision of data transmitted by SU and PU concurrently. This will result in data loss and thus reduced spectrum utilization. This probability is defined as probability of interference in imperfect sensing Pip . This is represented mathematically as [13]:    T −τ α 1 − exp Pip = T −τ α

(22)

From the definition of spectrum utilization ∅(τ ) =

U (τ ) T

(23)

where, the spectrum is utilized either by a PU or SU can be written as U (τ ) = USU + UPU

3 Proposed Optimization Algorithm When PU is not active, the false alarm will not be generated by SU. The  throughput probability for this case is (1 − θ ) 1 − p f and  for this case  the  is R0 possibleuthroughput (τ ) = ((T − τ )/T )(1 − θ ) 1 − p f 1 − Pp K SU . When PU is present and it is not detected by SU. The throughput for this case is   R1 possible throughput (τ ) = ((T − τ )/(T ))(1 − pd ) 1 − Pip θ K PSU . Where K P SU and K SU is the achievable throughput of the secondary network on the absence and power presence of PU, Ps is the received power of SU, Pp is the interference of 

s and PU and the noise power is represented by N 0, thus K PSU = log2 1 + N0P+P p  Ps K SU = log2 1 + N0 . When we apply the NOMA on the secondary network in Fig. 1, the achievable throughput of CR network is replaced by throughput of the NOMA network which is given in (4) and (5). After adding the throughput of the

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SU in the absence and presence of the PU into (19), the possible throughput of the NOMA based CR network can be written as: ∅possible throughput (τ ) =

+

     (T − τ )(1 − θ) 1 − p f 1 − Pp K n + (T − τ )(1 − pd ) )θ 1 − Pip K ns T

←−−−−−−−−−−−−−−−−−−−−−−−−−−−−−−−−−−−−−−−−−−−→ Achievable throughput of the SU by NOMA based CR network

(θτ + θ(T − τ ) pd K np T

(24)

←−−−−−−−−−−−−−−−−−−−−−→ Achievable throughput of the PU

3.1 Proposed Algorithm for Sensing Time Optimization Tmin ← 0, Tmax ← 1 and for convenience, we assumed that T = 1 s. Step 1

General purpose search technique for to find a maximum or minimum of unimodal function.    ∅possible throughput (τ ) ← (1 − τ )(1 − θ ) 1 − p f 1 − Pp K n   (a) + (1 − τ )(1 − pd ) )θ 1 − Pip K ns + (θ τ + θ (1 − τ ) pd K np (b) τ ← decide the range of τ values

Step 2

Function rewritten to work in MATLAB (a)

Step 3

f ← @(τ ) ∅possible throughput (τ )

To decide the lower and upper value (a)

τlow ← Tmin and τup ← Tmax

Step 4

To compute the golden ratio and difference of upper and lower bound. √   (a) g ← 5 − 1/2 and d ← g × τup − τlow (b) τ1 ← τlow + d and τ2 ← τup − d

Step 5

f or i ← 0 to 20 ← do (a) (b) (c) (d)

Step 6

if ( f (τ1 ) > f (τ2 )) do (a) (b) (c)

Step 7

Initialize f (τ2 ) and f (τ1 ) if ( f (τ2 ) > f (τ1 )) do   τup ← τ1 , τ1 ← τ2 and d ← g × τup − τlow τ2 ← τup − d else   τlow ← τ2 , τ2 ← τ1 and d ← g × τup − τlow τ2 ← τup − d else end if , end if and end for

Out put ← the optimal solution

Optimization of Sensing Time for Efficient Spectrum …

(a)

479

τmax ← τup , f (τ )max ← f (τup )

4 Simulation Results The parameters for system analysis are listed below: • • • •

Transmission band width of Sub band, W = 1 Hz f s = 12 MHz, γ = − 10 dB θ = 0.5, α = 0.5, β = 0.5 PT = 20 dBm, pd = 0.9

In this section, the performance analysis of the proposed scheme to obtain the maximum possible throughput of the network has been explained. Figure 3 shows the maximum possible throughput as the function of the sensing time. It has been observed that the optimal value of τ obtained is 0.3200 s, and the maximum possible throughput ∅possible throughput (τ ) = 4.3649 bps/Hz is obtained in Fig. 3. We also show the several effects of transmission power Pbs and possible throughput as the τ varies in Fig. 3. Figure 4 also shows the maximum possible throughput with the presence of interference probability. In this case, it has been observed that τ obtained is 0.4850, and the maximum spectrum utilization ∅possible throughput (τ ) = 2.8688. Figure 5 shows the relation between possible throughput and channel gain. In this figure, we can see

Fig. 3 Possible throughput versus sensing time with θ = 0.5, ps = 20, gs = 2.5, γ = 0.1, Pp = 0, Pip = 0 and frame duration T = 1 s

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Fig. 4 Possible throughput versus sensing time with θ = 0.5, ps = 30, gs = 3.5, γ = 0.1, Pp = 0, Pip = 0 and frame duration T = 1 s

Fig. 5 Possible throughput versus channel gain

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that as gs increases, the possible throughput also increases. The effect of possible throughput on the interference probability in both perfect and imperfect sensing has been shown in Fig. 4. Thus the conclusion is drawn that interference probability has an impact on sensing time. When we consider the interference probability, sensing time will increase which decreases the spectrum utilization of the system.

5 Conclusion In this paper, we have optimized the sensing time in NOMA based cognitive radio networks with the objective of maximizing the maximum possible throughput. The mathematical expression for the achievable throughput is formulated for 3 SUs and 1 PU for NOMA based CR networks. We have also considered the interference probability when there are sensing errors like miss detection and false alarm while formulating the objective function. The optimal sensing time obtained 0.3200 s. It has also been observed that the maximum possible throughput on the channel is also improved when NOMA is used.

References 1. Mitola J (2001) Cognitive radio for flexible mobile multimedia communications. Mobile Netw Appl 6:435–441 2. Ghasemi A, Sousa ES (2008) Spectrum sensing in cognitive radio networks: requirements, challenges and design trade-offs. IEEE Commun Mag 46:32–39 3. Shen J, Liu S, Wang Y, Xie G, Rashvand HF, Liu Y (2009) Robust energy detection in cognitive radio. IET Commun 3:1016–1023 4. Choi W, Song MG, Ahn J, Im GH (2013) Soft combining for cooperative spectrum sensing over fast-fading channels. IEEE Commun Lett 18:193–196 5. Liu X, Jia M, Tan X (2013) Threshold optimization of cooperative spectrum sensing in cognitive radio networks. Radio Sci 48:23–32 6. Liu X, Jia M (2017) Joint optimal fair cooperative spectrum sensing and transmission in cognitive radio. Phys Commun 25:445–453 7. Liang YC, Zeng Y, Peh ECY, Hoang AT (2008) Sensing-throughput trade-off for cognitive radio networks. IEEE Trans Wireless Commun 7:1326–1337 8. Fan R, Jiang H (2010) Optimal multi-channel cooperative sensing in cognitive radio networks. IEEE Trans Wireless Commun 9:1128–1138 9. Liu X, Li F, Na Z (2017) Optimal resource allocation in simultaneous cooperative spectrum sensing and energy harvesting for multichannel cognitive radio. IEEE Access 5:3801–3812 10. Higuchi K, Benjebbour A (2015) Non-orthogonal multiple access (NOMA) with successive interference cancellation for future radio access. IEICE Trans Commun 98:403–414 11. Ju H, Zhang R (2014) Throughput maximization in wireless powered communication networks. IEEE Trans Wireless Commun 13:418–428 12. Song Z, Wang X, Liu Y, Zhang Z (2019) Joint spectrum resource allocation in NOMA-based cognitive radio network with SWIPT. IEEE Access 7:89594–89603 13. Verma P, Singh B (2018) Joint optimization of sensing duration and detection threshold for maximizing the spectrum utilization. Digital Sig Process 74:94–101

A CPW-Fed Annular Shape Antenna with Asymmetrical Hexagonal Slot Loaded Defected Ground Plane for Ultra-Wideband Applications Ajay Kumar Dwivedi, Brijesh Mishra, Chandrabhan, Shadab Azam Siddique, and Vivek Singh Abstract In this report, we present a compact printed antenna with a defected ground surface for ultra-wideband (UWB). It consists of an asymmetrical hexagonal slot loaded defected ground plan. CPW-fed annular shape radiating patch with two rectangular shape parasitic patches in order to enhance the gain and impedance bandwidth. The proposed antenna is operating between 6.2 and 19.8 GHz, which covers the frequencies of partial C-band, X-band, and Ku-band. The simulation, parametric analysis, and optimization are executed by using HFSS software. The maximum radiation efficiency and peak gain of the proposed antenna are observed 93.3% and 9.34 dB, respectively. Keywords Defected ground plan · X-band and Ku-band · Radiation efficiency · Parasitic patches · UWB

1 Introduction With the staggering evolution of printed circuit-based microwave communication systems and standards, the requirement of antennas of miniaturized size, large bandwidth with optimum radiation characteristics, and with ease of fabrication is increasing. Since the last few years, several research papers have been published in the domain of ultra-wideband (UWB) technology [1–7], which has triggered an upturn in the progress of printed antennas. However, patch antennas have limitations of their applicability because of their narrowband and poor gain. These limitations can be overcome by the modification A. K. Dwivedi Indian Institute of Information Technology Allahabad, Prayagraj, Uttar Pradesh 211015, India B. Mishra · S. A. Siddique Department of Electronics and Communication, Madan Mohan Malaviya University of Technology, Gorakhpur, Uttar Pradesh, India Chandrabhan · V. Singh (B) Department of Electronics and Communication, Shambhunath Institute of Engineering and Technology, Prayagraj, Uttar Pradesh, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_52

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Table 1 Comparative description of antennas with a proposed antenna for UWB applications References

Area (mm)2

BW/BW range (GHz)

Peak gain (dB)/radiation efficiency (%)

[15]

80 × 12 = 960

1.62/(13.36–14.98)

9.0/NR

[16]

46 × 52 = 2392

12.2/(2.8–15)

1.55/88

[17]

30 × 30 = 900

9.0/(3.0–12.0)

5/NR

[18]

45 × 40 = 1800

11.70/(2.3–14)

5.25/NR

[19]

29 × 31 = 899

7.60/(3.0–10.6)

5.50/NR

[20]

47 × 47 = 2209

11.58/(2.18–13.76)

8.00/NR

[21]

60 × 60 = 3600

10.22/(2.67–13.0)

4.20/NR

Proposed antenna

30 × 30 = 900

13.60/(6.2–19.8)

9.34/93.3

NR not reported

of the geometry of conducting patch by cutting the notches and slots of regular and irregular shape on radiating patches [8–10], by using the gap coupled parasitic patches [11, 12], replacing solid ground structure by defected ground structure and by stacking of substrates of different dielectric constants [13, 14]. In the proposed article, a miniaturized coplanar waveguide fed annular ring shape printed antenna with a defected ground surface (DGS) is discussed for UWB applications. The operating frequency range of the designed antenna varies from 6.2 to 19.8 GHz which makes the antenna suitable for partial C to Ku-band applications. This antenna comprises a coplanar waveguide (CPW) fed annular shape conducting patch, two parasitic patches of rectangular shape, and an asymmetrical hexagonal slot loaded defected ground plane. This work aims to propose a simple, planar, and compact patch antenna for ultra-wideband operations. To ensure compactness and simplicity a CPW-feed technique with coplanar electromagnetically coupled parasitic patches is used. In Table 1, the comparative analysis is carried out between the designed antenna and reported antennas for ultra-wideband applications with regards to the area of the antenna, operating bandwidth, radiation efficiency, and peak gain. From the perusal of Table 1, it is found that the area of the designed antenna is compact by the index of 1.06, 2.65, 2, 2.45, and 4 from the antennas reported in Refs. [15, 16, 18, 20, 21], respectively, and has an equal area to the antennas mentioned in Refs. [17, 19]. However, other antenna parameters of the proposed prototype are optimum in comparison with the antennas mentioned in Table 1.

2 Designing of Antenna A simplified layout of the designed antenna is presented in Fig.1a–c. Low cost, durable, and commercially available FR4 epoxy substrate (εr of 4.4 and tanδ of 0.02) has been used for the designing of the proposed antenna.

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Fig. 1 Layout of designed patch antenna: a front view, b rearview, c lateral view, d fabricated antenna front view, e fabricated antenna back view

The area of the designed antenna is 30 × 30 mm2 , and it has a simple configuration that consists of an annular shape radiating patch with outer radius and inner radius of R1 and R2, respectively. It also consists of two rectangular shape parasitic patches with the height of A1 and width of A2. A coplanar waveguide of height A3 and width A4 is used to feed the antenna. The purpose of using coplanar waveguide feed is to achieve impedance matching and other important feathers. A hexagonal asymmetrical slot loaded defected ground is used in the antenna. Detail dimensions are presented in Table 2. A stepwise analysis is carried out to obtain the optimum structure. Five subsequent steps have been shown in Fig. 2a, and its corresponding simulated |S11 | (dB) plot is presented in Fig. 2b. It is clear from the figure that the antennas 1, 2, 3, and 4 behave like ultra-wideband antenna with some band-notch characteristics while antenna 5 covers the entire UWB region of 6.2–19.8 GHz with better return loss (S11< − 10 dB) and with revamped impedance matching. Therefore, antenna 5 is considered Table 2 Parameters and corresponding values of the antenna Parameters

L

W

A1

A2

A3

A4

A5

Units (mm)

30

30

10

4

16

2

18.4

Parameters

R1

R2

H1

H2

H3

H

Units (mm)

6

4

17.8

19.6

26.8

1.6

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Fig. 2 a steps to obtain the proposed antenna. b |S11 | plot of corresponding antennas

as a proposed antenna. It is also worth mentioning that the incorporation of the asymmetrical hexagonal slot on the ground plane increases the operating BW. Parametric analysis is laid down to find out the effect of variation in dimensions of various parameters like the inner radius of the ring (R2), the position of asymmetrical hexagonal slot on the ground plan (A5), and height of rectangular patches (A1). Variation of the inner radius of the annular ring (R2) from 3 to 5 mm is presented in Fig. 3a. From the close investigation of Fig. 3a, it is clear that the designed antenna behaves like a dual-band antenna for R2 = 3 mm while for R2 = 4 mm and 5mm, and it behaves like an ultra-wideband antenna. However, as the radius of the inner annular ring is increases beyond 4 mm, the bandwidth is decreasing. Therefore, R2 = 4 mm is selected as the ideal value for BW of GHz to 19.8 GHz, which covers the whole desired ultra-wideband rage with good impedance matching. Figure 3b shows the plot for the variation of the position of asymmetrical hexagonal slot (A5) from 17.4 to 19.4 mm. From the perusal of Fig. 3b, it is stated that the Variation of the inner radius of the ring (R2)

Variation of the position of asymmetrical hexagonal slot (A5)

Fig. 3 |S11 | plot for different values of a R2. b A5 , c A1

Variation of the length of parasiticpatches (A1)

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proposed prototype is behaving like a UWB antenna for A5 = 18.4 mm. When we are shifting the slot in an upward direction (A5 = 19.4 mm), the proposed antenna is showing dual-band characteristics. When we are shifting the slot in a downward direction (A5 = 17.4 mm), the whole bandwidth is shifted toward the lower frequency range, which makes the antenna unsuitable for Ku-band applications. Figure 3c shows the variation of length (A1) of parasitic patches from 6 to 10 mm. Slight variations are observed in return loss (S11) values for different values of A1, but it is noticeable from Fig. 3c that the optimum value of return loss is obtained for A1 = 10 mm for the whole desired ultra-wideband range of 6.2–19.8 GHz. Outcomes, Experimental Validation, and Discussions Investigation of the proposed structure is carried out in terms of different antenna parameters by simulating the design on HFSS and simulated results are experimentally validated by VNA E5071C. Distribution of surface current of excited antenna is shown in Fig. 4a, b for the resonating frequencies of 9.4 GHz and 17 GHz, respectively. Maximum surface density of 148.14 A/m and 93.14 A/m is observed for the frequencies 9.4 GHz and 17 GHz, respectively. At both the frequencies, maximum current is flowing through the CPW feed attach with annular ring structure and parasitic patches. However, the orientation of the current vectors in both cases is different due to which proposed antenna generates the different resonating modes. The proposed antenna offers ultra-wideband of 6.2–19.8 GHz which meets the requirements of partial C-band, X-band, and Ku-band as shown in Fig. 5. Figure 6a, b represent the 3-dimensional polar plot for the designed antenna at resonating frequencies of 9.4 GHz and 17 GHz, respectively. The 3-D polar plot provides useful information about the distribution of the radiated electric field in the far-field region. Figure 7 illustrates the gain and radiation efficiency plot for the designed antenna. From the close examination of Fig. 7, it is clear that the peak gain of the antenna is 9.34 dB while radiation efficiency varies from 93.3 to 73.3% for the operating bandwidth range of 6.2–19.8 GHz.

Fig. 4 Surface current distribution of the proposed antenna at a 9.4 GHz, b 17 GHz

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Fig. 5 Measured and simulated |S11| plot

Fig. 6 3-D polar plot of the proposed antenna at a 9.4 GHz, b 17 GHz

3 Conclusion A novel coplanar waveguide fed annular shape patch antenna with defected ground surface is discussed for ultra-wideband applications, and parametric analysis is also carried out to investigate the effect of change of dimension of various parameters on the characteristics of the antenna. The introduction of a defected ground surface increases the impedance bandwidth of the antenna. The proposed antenna is well suited for various wireless applications

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Fig. 7 Gain (dB) and radiation efficiency plot for the proposed antenna

Acknowledgments The research described in this paper was financially supported by The World Bank and National Project Implementation Unit (NPIU), MHRD, India under the TEQIP III project scheme of Collaborative Research Scheme (CRS) of Project Entitled “Development of IoT controlled frequency/pattern reconfigurable MIMO antenna for harvesting systems”.

References 1. Gautam K, Yadav S, Kanaujia BK (2013) A CPW-fed compact UWB microstrip antenna. IEEE Antennas Wirel Propag Lett 12:151–154 2. Kamakshi K, Ansari JA, Singh A, Aneesh M, Jaiswal AK (2015) A novel ultrawideband toppled trapezium-shaped patch antenna withpartial ground plane. Microw Opt Technol Lett 57(8):1983–1986 3. Shakib MN, Moghavvemi M, Mahadi WNL (2015) A low-profile patch antenna for ultrawideband application. IEEE Antennas Wirel Propag Lett 14:1790–1793 4. Shrivastava MK, Gautam AK, Kanaujia BK (2014) A novel A-shaped monopole-like slot antenna for ultrawideband applications. Microw Opt Technol Lett 56(8):1826–1829 5. Shrivastava MK, Gautam AK, Kanaujia BK (2014) An M-shaped monopole-like slot UWB antenna. Microw Opt Technol Lett 56(1):127–131 6. Mishra B (2019) An ultra compact triple band antenna for X/Ku/K band applications. Microw Opt Technol Lett 61(7):1857–1862 7. Mishra B, Singh V, Singh RK, Singh N, Singh R (2018) A compact UWB patch antenna with defected ground for Ku/K band applications. Microw Opt Technol Lett 60(1):1–6 8. Singh V, Mishra B, Narayan Tripathi P, Singh R (2016) A compact quad-band microstrip antenna for S and C-band applications. Microw Opt Technol Lett 58(6):1365–1369 9. Singh V, Mishra B, Singh R (2018) Dual-wideband semi-circular patch antenna for Ku/K band applications. Microw Opt Technol Lett 10. Singh V, Mishra B, Dwivedi AK, Singh R (2018) Inverted L-notch loaded hexa band circular patch antenna for X, Ku/K band applications. Microw Opt Technol Lett 60(8):2081–2088

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11. Singh V, Mishra B, Singh R (2019) Anchor shape gap coupled patch antenna for WiMAX and WLAN applications COMPEL. Int J Comput Math Electr Electron Eng 38(1):263–286 12. Mishra B, Singh V, Singh R (2018) Gap coupled dual-band petal shape patch antenna for WLAN/WiMAX applications. Adv Electr Electron Eng 16(2) 13. Mishra B, Singh V, Singh R (2017) Dual and wide-band slot loaded stacked microstrip patch antenna for WLAN/WiMAX applications. Microsyst Technol 23(8):3467–3475 14. Mishra B, Singh V, Dwivedi AK, Pandey AK, Sarwar A, Singh R (2017) Slots loaded multilayered circular patch antenna for Wi-Fi/WLAN applications, pp 49–59 15. Chen Yu, Hong W, Kuai Z, Wang H (2012) Ku-band linearly polarized omnidirectional planar filtenna. IEEE Antennas Wirel Propag Lett 11:310–313 16. Kumar SS, Rao GS, Pillalamarri R (2015) Rectangular slotted microstrip line fed compact printed antenna with etched ground plane for UWB communications. Microsyst Technol 21(10):2077–2081 17. Kumar R, Praveen Naidu V, Kamble V (2015) A compact asymmetric slot dual band antenna fed by CPW for PCS and UWB applications. Int J RF Microw Comput Eng 25(3):243–254 18. Bitchikh M, Aksas R, Azrar A, Kimouche H (2013) A 2.3-14 GHz UWB planar octagonal antenna with modified ground plane. Microw Opt Technol Lett 55(3):479–482 19. Kumar A, Shanmuganantham T (2014) A CPW fed octagonal patch UWB antenna with WiMAX band-notched characteristics. In: International conference on information communication and embedded systems (ICICES2014), 2014, pp 1–5 20. Sharma A, Khanna P, Singh AK, Kumar A (2018) CPW—fed dodecagon ring shape antenna for ultra wideband application. Int J Ultra Wideband Commun Syst 3(4):201 21. Pourahmadazar J, Ghobadi C, Nourinia J, Felegari N, Shirzad H (2011) Broadband CPW-fed circularly polarized square slot antenna with inverted-L strips for UWB applications. IEEE Antennas Wirel Propag Lett 10:369–372

Ultra-Wide Band Microstrip Patch Antenna for Millimetre-Wave Band Applications Aditi Chauhan, Utkarsh Jain, Aakash Warke, Manan Gupta, Ashok Kumar, Amrita Dixit, and Arjun Kumar

Abstract This paper presents the design for a compact microstrip patch antenna that operates in the Ka band with dimensions (13 × 13) mm2 and is applicable for 5G communication. The antenna resonates at a central frequency of 34.2 GHz, providing a gain of 7.5 dB. It comprises of a partial ground structure in order to provide a large bandwidth ranging from 24.1 to 49.9 GHz. This antenna has been simulated on Ansys HFSS 19.1 using Rogers RO4003, of dielectric constant 3.55, as the substrate. Keywords 5G · Ka band · Millimetre Wave (mmWave) · Partial grounds · Ultra-wide band · Ansys HFSS

1 Introduction Since its introduction in the 1970s, wireless communication has come a long way with its incredibly rapid advancements, slowly transforming the society into a fully connected network [1]. Consequently, this increases the demand for antennas that yield better performances in terms of the antenna size, gain, bandwidth, cost and data rate [2]. Hence, 5G or the fifth generation of wireless communication is proposed in order to obtain high data rate. Millimetre-wave band or the Ka band, that occupies the frequency spectrum from around 30–300 GHz in the electromagnetic spectrum, is suitable for 5G as it can provide a relatively larger absolute bandwidth. And most of it still needs extensive exploration [3]. It provides a bandwidth that is almost ten times more than what the 4G or the fourth-generation cellular band offers, and a higher bandwidth is required in order to reduce the amount of path loss [4, 5]. Small and low-profile antennas are optimum for mobile communication and microstrip patch antennas qualify this criterion. However, they have their own merits and demerits over A. Chauhan (B) · U. Jain · A. Warke · A. Kumar Department of Physics, School of Engineering and Applied Sciences, Bennett University, Greater Noida, India M. Gupta · A. Dixit · A. Kumar Department of Electronics and Communication Engineering, School of Engineering and Applied Sciences, Bennett University, Greater Noida, India e-mail: [email protected] © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_53

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the conventional antennas. Microstrip patch antennas are less expensive, low profile as aforementioned, can operate at multiple frequencies and are easy to fabricate; however, gain and efficiency are significantly low with larger losses [6]. A few of the techniques useful in order to overcome the shortcomings posed by the microstrip patch antennas have been discussed in [2]. Several designs have been explored for further enhancements in the field of 5G wireless communication. A small microstrip patch antenna resonating at millimetre-wave frequency (i.e. 28 GHz) was presented in [5], while antenna arrays were used in [7, 8]; ideal for 5G applications. In this research, a simple patch antenna for millimetre-wave band application has been proposed for 5G communication. High-frequency material, Rogers RO4003, has been used as a substrate and the antenna results have been validated using Ansys HFSS.

2 Proposed Antenna Design In this section, the structural geometry and configuration of the design have been presented. The proposed antenna has been designed on a Rogers RO4003 substrate of thickness 1.52 mm and a relative permittivity (εr ) of 3.55 compatible for radiation in high frequencies while both the ground and the patch are 0.035 mm thick, and the conducting material used here is PEC. The dimensions of the substrate are 13 mm × 13 mm × 1.52 mm, and Fig. 1 presents the geometric parameters of the antenna while their corresponding lengths have been listed in Table 1. The structure of the ground is one of a partial ground, as shown in Fig. 1, and it plays a very crucial role in increasing the bandwidth of the antenna.

Fig. 1 Geometry of the proposed antenna

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Table 1 Parameters of the proposed antenna Parameter

Value (mm)

Parameter

Value (mm)

Parameter

Value (mm)

L1

4.25

Lg

2.95

W5

1.53

L2

3.25

Ws

13

W6

1.64

L3

2.9016

Wf

0.7

W7

1

L4

0.25

D1

0.6

W8

1.25

L5

1.25

W1

9

W9

0.25

L6

0.25

W2

0.75

W 10

0.5

L7

2.53

W3

3.435

W 11

0.565

L8

0.5

W4

1.25

W 12

0.75

Ls

13

W 13

0.25

Fig. 2 Effects of L7 on antenna performance

2.1 Parametric Analysis 2.1.1

Effects of Variation in L7

The analysed results for variations in L7 have been shown in Fig. 2. We notice that the change has a slight difference in the return loss but a significant variation in the −10 dB bandwidth. Based on this, L7 = 2.53 mm has been chosen since this parametric value provided the best bandwidth and return loss.

2.1.2

Effects of Variation in D1

Figure 3 depicts the variation of S11 parameters for different values of D1. As we can notice, change in D1 influences the return loss. From Fig. 4, we notice that as D1

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Fig. 3 Effects of D1 on antenna performance

increases, the upper band in the graph slightly shifts to the lower side. However, we choose D1 = 0.6 mm as the most appropriate dimension due to the best impedance matching it offers.

3 Result Analysis In this paper, an ultra-wide band antenna using Rogers RO 4003 as the substrate and PEC as the conducting material was designed using the concept of partial grounds in order to obtain a fractional bandwidth of 69.72% from 24.01 to 49.9 GHz. The antenna provides a gain of 7.5 dB at the central frequency, i.e. 34.2 GHz and a peak gain of 8.01 dB, and a maximum return loss of 28.42 dB. The simulated results of the S11 parameters and the gain are reported in Fig. 4. Some of the previously published work on the millimetre-wave band has been shown in Table 2. In this table, it is clearly visible that the proposed design has a wide bandwidth and compact size.

4 Conclusion A wide band microstrip patch antenna with a compact structure has been proposed in this paper. With the help of using partial grounds, a fractional bandwidth of 69.72% was achieved while varying the length L7 helped match the impedance to 48.9 . Due to its good radiation characteristics and an exceptionally large bandwidth from

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Peak Gain (dBi)

(a) 10 9 8 7 6 5 4 3 2 1 0 25

30

35 40 Frequency (GHz)

45

50

(b) Fig. 4 Simulated results a return loss S11 , b peak gain plot Table 2 Comparison of proposed design with the previous work References Size (mm2 )

Gain (dB) Frequency (GHz) Return loss (dB) Bandwidth (GHz)

[2]

6 × 6.25

6.9/7.4

38/54

15.5/12

1.94/2.05

[5]

5.5 × 4.5

6.72

28

18.25

1.1

[9]

41.3 × 36

13

28.4

30

11.8

[10]

15.8 × 13.1 4.06

28

20

-

[11]

20 × 20

12.48

28

30

2.4

This work

13 × 13

7.5

34.2

28.42

25.8

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24.1 to 49.9 GHz with return loss better than 25.8 dB, this antenna can be used for 5G communications in the millimetre-wave band spectrum.

References 1. Baldemair R, Dahlman E, Parkvall S, Selen Y, Balachandran K, Irnich T, Fodor G, Tullberg H (2013) Future wireless communications. In: 2013 IEEE 77th vehicular technology conference (VTC Spring). IEEE, pp 1–5 2. Imran D, Farooqi MM, Khattak MI, Ullah Z, Khan MI, Khattak MA, Dar H (2018) Millimeter wave microstrip patch antenna for 5G mobile communication. In: 2018 international conference on engineering and emerging technologies (ICEET). IEEE, pp 1–6 3. Lodro Z, Shah N, Mahar E, Tirmizi SB, Lodro M (2019) mmWave novel multiband microstrip patch antenna design for 5G communication. In: 2019 2nd international conference on computing, mathematics and engineering technologies (iCoMET). IEEE, pp 1–4 4. Center DMCRD (2015) Samsung electronics “5G Vision”. White paper (online) 5. Goyal RK, Modani US (2018) A compact microstrip patch antenna at 28 GHz for 5G wireless applications. In: 2018 3rd international conference and workshops on recent advances and innovations in engineering (ICRAIE). IEEE, pp 1–2 6. Kaushal A, Tyagi S (2015) Microstrip patch antenna its types, merits demerits and its applications 7. Ojaroudiparchin N, Shen M, Pedersen GF (2015) A 28 GHz FR-4 compatible phased array antenna for 5G mobile phone applications. In: 2015 international symposium on antennas and propagation (ISAP). IEEE, pp 1–4 8. Hong W, Baek K, Lee Y, Kim YG (2014) Design and analysis of a low-profile 28 GHz beam steering antenna solution for future 5G cellular applications. In: 2014 IEEE MTT-S international microwave symposium (IMS2014). IEEE, pp 1–4 9. Yoon N, Seo C (2017) A 28-GHz wideband 2 × 2 U-slot patch array antenna. J Electromagn Eng Sci 17(3):133–137 10. Neha K, Sunil S (2018) A 28-GHz U-slot microstrip patch antenna for 5G applications. Int J Eng Dev Res 6(1):363–368 11. Haraz OM, Elboushi A, Alshebeili SA, Sebak A-R (2014) Dense dielectric patch array antenna with improved radiation characteristics using EBG ground structure and dielectric superstrate for future 5G cellular networks. IEEE Access 2:909–913

A Review on Attack and Security Tools at Network Layer of IoT Vidur Agarwal, Preeti Mishra, Sachin Kumar, and Emmanuel S. Pilli

Abstract Internet of things (IoT) is one of the emerging areas which connects billions of the devices across the world through Internet. Security and privacy in such a technological era is one of the major challenges. The IoT devices are vulnerable to dangerous attacks at various layers such as application layer, network layer, and perception layer because of their simple architecture and less secure network. In this paper, we mainly focus on the network layer security of IoT. We provide a detailed taxonomy of various attack and security tools at the network layer of IoT. We also provide a comparative analysis of these tools. We hope that our work will be helpful to the researchers working in the area of IoT security to gain better understanding about various existing tools. Keywords Security and privacy · Attack tools · Security tools

1 Introduction The worldwide expansion of the Internet and its availability to more than 4 billion end users has given exponential rise to the Internet of things (IoT) [1]. The IoT has made the lives of people more comfortable and easy, leading to reduced human intervention. These smart things connected to the Internet has simplified almost all the important aspects of daily necessity such as connected car, smart refrigerator, IP cameras, and healthcare systems. Since the last decade, almost half of the population of the world is using the Internet because of the advantages provided by it. The IoT devices are vulnerable to attacks because of their simple architecture and less secure network. To perform various attacks, there exits attack tools such as Aircrack-ng [2], V. Agarwal (B) · S. Kumar Graphic Era Deemed to be University, Dehradun, India P. Mishra Doon University, Dehradun, India E. S. Pilli Malaviya National Institute of Technology, Jaipur, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_54

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AirSnort [3], Kismet [4], Cain and Abel [5], Low Orbit Ion Canon (LOIC) [6], and High Orbit Ion Canon (HOIC), which target specific layer of IoT. These tools may target an IoT device and try to break its password or gain illegitimate access to the information related to these devices. In order to deal with various attacks, various security tools have been developed. However, there is no security tool that can detect all possible IoT attacks. Various security solutions have been developed for detecting attacks at specific layer of IoT. IoT has been expanded along with industry 4.0 which provides many security tools such as Cotopaxi [7] and SSLyze [8] to protect, manage and monitor the IoT-based attacks. Perception layer [9], network layer [10], and application layer [11] are the three layers in IoT. In this paper, we mainly focus on the network layer [9] of IoT which handles network communications. It is the Layer-3 in the open systems interconnection model (OSI) which manages host and network addressing, managing subnetworks, and internetworking, mapping different addressing schemes and protocols. This layer is also responsible to route the packets from source to destination. This paper provides a detailed taxonomy of various attack and security tools at the network layer of IoT. A comparative discussion of these tools is also provided to have a better understanding of tools. The knowledge of attack tools is very helpful for researchers to create the attack datasets. In addition, knowledge about security tools is very helpful for understanding the existing solutions. The major contributions of the work can be summarizing as follows: • To provide a taxonomy and detailed description of various network layer attack tools. • To provide a taxonomy and detailed description of various network layer security tools. • To provide the comparative description of various network layer attack and security tools. The rest of the paper is organized as follows: Sect. 2 provides the details of the related work, carried out in the field of IoT security. Section 3 gives the taxonomy of attack and security tools. Sections 4 and 5 provide a detailed description of various attack and security tools, respectively. Finally, Sect. 6 concludes the paper with future research directions.

2 Related Work Recently, the issues of privacy and security for the IoT have attracted a lot of research interests. During the early stage of IoT, some security issues have become very popular such as privacy, information security, trust, physical security, and network layer security. However, due to the rapid growth of several security issues, there are some tools available to provide a secured framework to the IoT devices.

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Zhao [12] discussed various network layer security problems, i.e., communication network problems, compatibility problems, privacy disclosure, and cluster security problems. Peng [13] discussed different network layer security technologies such as key management strategy and information encryption technology so that transmission of information can be secured. According to the authors, there are two types of encryption point-to-point encryption and end-to-end encryption. Various application domains like data mining, machine learning, and deep learning techniques have been widely adopted in the domain of IoT security and IoT traffic analysis. The major key challenges are the data privacy problem, insecure data transfer and storage, and the encryption of messages. According to the best of our knowledge, there is no survey available on attack and security tools at the network layer. Not only, we try to categorize different tools for the network layer in IoT on their attacks as well as security nature but also arranged them on further subdivisions also.

3 Taxonomy of Attack and Security Tools at Network Layer In this section, a taxonomy of various attack and security tools at the network layer of IoT has been discussed and also shown in Fig. 1. Network layer tools are classified into two categories, i.e., attack tools and security tools. We have also compared them on the basis of tool’s type, expertise requirement, language used, cost of service, year, and the most important, i.e., interface used by tools as shown in Tables 1 and 2, respectively. If we do not have a security plan, then our network and data are vulnerable to many attacks. The tools which cause any disruption or damage to a network are placed in the attack category whereas the tools used to detect or prevent malicious activities are placed in security tools [14] category. These tools are widely used for attacking and security testing purposes. According to the taxonomy discussed, the attack tools can be used to trigger network layer attacks. On the other hand, the security tools are used to detect this malicious behavior on the network layer. There are various uses of security tools such as they enable communicating nodes to encrypt messages and prevents eavesdropping by third parties. They provide assurance that a received packet was actually transmitted by the actual party and confirms that the packet has not been altered or otherwise. They allows secure exchange of keys also. These security tools follow some specific procedures for detection of specific patterns of threats and attacks. In the next section, we discuss these attack and security tools with their functionalities.

4 Attack Tools at Network Layer In this section, we have discussed various network layer [9] attack tools that are capable of compromising IoT devices as shown in Table 1. These tools are used to

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Fig. 1 Taxonomy of attack and security tools Table 1 Taxonomy of network layer attack tools Attack tools on network layer Tool name

Type of tool

Expertise required

Language

Cost of service

Interface

Year

Aircrack-ng [2]

WEP and WPA-PSK key cracker

Yes

Python

Free

Both

2006

Kismet [4]

Wireless network detector

No

C++

Free

Both

2019

Cain and Abel [5]

Password cracking

Yes

Python

Free

GUI

2014

coWPAtty [15]

Password cracker

Yes

C

Free

CLI

2004

NetStumbler [16]

Password cracker

NA

NA

NA

GUI

2008

Ferm Wi-Fi cracker Crack and recover [17] WEP/WPA/WPS keys

Yes

Python

Paid

Both

NA

Commview for-Wi- Packer analyzer Fi [18]

No

NA

Paid

GUI

1998

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perform various types of attacks like identity spoofing [19], blackhole attack [20], Sybil attack [21], wormhole attack [22], smurf attack [23], and hijacking attack [24]. All the necessary information regarding system requirements, availability, interface, etc., has been discussed. These tools also have been classified on the basis of type of tools, environment of tool, etc. Here, we are discussing some of the attacking tools that may cause disruption or damage.

4.1 Aircrack-Ng Aircrack-ng [2] is a free software application written in C. It operates under Macintosh Operating System (macOS), Linux, Windows, Berkeley Software Distribution (OpenBSD), and FreeBSD. It is used for testing, monitoring, attacking, and cracking wired equivalent privacy (WEP) and Wi-Fi protected access (WPA). Having a simple attacking mechanism, it can be used to recover the password and for monitoring and collecting packets also.

4.2 Kismet Kismet [4] is an open-source passive tool written in C++ which is free to use and does not interact with the network. Without sending any loggable packets, it is used for detecting wireless LANs, wireless access points, wireless clients, for monitoring network traffic, as a wardriving tool, and as an intrusion detection system also. It constantly changes from one channel to another channel non-sequentially in a userdefined sequence with a default value that leaves big holes between channels.

4.3 Cain and Abel Cain and Abel [5] is a great tool for network penetration testing and password recovery for Microsoft Windows. It captures the data when it is transmitted to the network and cracks passwords that are encrypted using various password cracking techniques, perform crypt-analysis attacks as well as discover wireless keys by analyzing wireless protocols.

4.4 CoWPAtty CoWPAtty [15] is a Linux-based network tool having a command-line interface (CLI) that is used to implement an accelerated and brute-force dictionary attacks

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on vulnerable wireless network systems. It is good at cracking weak WPA-PSK passwords and can also be used by hackers to acquire passwords to wireless networks.

4.5 Fern Wi-Fi Cracker Fern Wi-Fi Cracker [17] is an attacking software program written in Python. It cracks and recovers WEP/WPA/WPS keys, runs other network-based attacks on wireless or ethernet based networks, supports automatic access points, and possesses an internal MITM engine also. After a successful attack, it saves the key in the database.

4.6 NetStumbler NetStumbler [16] also known as Network Stumbler is a window tool that finds out open Wi-Fi networks, rogue access points, network misconfigurations, and areas having poor connectivity, etc., during war driving and war walking kinds of activities. It is also used to detect the cause of wireless interference and to aim directional antennas for long-haul WLAN links.

4.7 CommView for Wi-Fi CommView for Wi-Fi [18] is a graphical user interface (GUI)-based software that analyzes packets and can monitor wireless 802.11 a/b/g/n networks. It helps us to view and examine packets, pinpoint network problems, and troubleshoot software and hardware. We can prefer a non-wireless CommView edition if we want to analyze traffic on our machine.

5 Security Tools at Network Layer IoT devices are more vulnerable due to a lack of standardizing security frameworks and protocols. There are various security-related problems or hacking issues with IoT services. So, there are some tools that can overcome the security issues generated by attackers. In this section, we have discussed several security tools that are capable to provide security at the network layer (as shown in Table 2). These tools also have been classified on the basis of type of tools, environment of tool, etc. We try to summarize all the necessary information, requirement, and system configuration for the security tools.

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Table 2 Taxonomy of network layer security tool Security tools on network layer Tool name

Type of tool

Expertise required

Language

Cost of service Interface

Year

COTOPAXI [7]

Security testing

Yes

Python

Free

CLI

2019

SSLyze [8]

Analyze the SSL configuration

No

Python

Free

CLI

NA

Tls Prober [25] Penetration testing

No

NA

Free

Both

2017

Acunetix [26]

Penetration testing

Yes

Javascript

Paid

GUI

2005

Snort [27]

Intrusion detection & prevention system

Yes

C

Free

Both

2020

Nessus [28]

Proprietary vulnerability scanner

Yes

NASL

Paid

Both

2019

Retina [29]

Vulnerability scanner

No

NA

Free

CLI

1998

5.1 COTOPAXI COTOPAXI [7] is a set of tools for security testing of IoT devices using specific network IoT/industrial Internet of things (IIoT)/machine-to-machine (M2M) protocols. It works only with Python and is predestined to be used only for authorized security testing. It checks the availability of network services for supported IoT protocols at given IPs and port ranges (“service ping”) and identifies a software used by a remote network server (“IoT software fingerprinting”) based on feedback for given messages using a machine learning classifier.

5.2 SSLyze SSLyze [8] is a CLI tool, written using Python library that analyzes the SSL configuration of a server by connecting to it. It is a fast and comprehensive tool that helps organizations and testers to identify misconfigurations affecting their SSL/TLS servers. It allows various security testing such as weak cipher suites, insecure renegotiation, ROBOT, Heartbleed, and performance testing such as session resumption and supports StartTLS handshakes on extensible messaging and presence protocol (XMPP), simple mail transfer protocol (SMTP), lightweight directory access protocol (LDAP), Internet message access protocol (IMAP), post office protocol (POP), remote desktop protocol (RDP), and file transfer protocol (FTP).

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5.3 TLS Prober TLS Prober [25] requires Python. It allows fingerprinting to determine the TLS implementation in use on a server and analyzes the behavior of a server by sending a range of probes, and after that, it compares the responses with a database of known signatures. TLS Prober analyzes differences in behavior that are inherent in the implementation and are not affected by configuration changes. No knowledge of the server configuration is required, and it does not rely on the supported cipher suites also.

5.4 Acunetix Acunetix [26] is the network security scanner written in C++ and is one of the fastest Web security tools. It helps one to scan Internet protocol (IP) address ranges to discover open ports and other security vulnerabilities specific to network devices, and it is integrated with the Open Vulnerability Assessment System (OpenVAS) open-source tool. Using only one dashboard, one can manage our Web and network vulnerabilities.

5.5 Snort Snort [27] is an open-source intrusion detection system that is based on captures library packet (libpcap) and is widely used in Transmission Control Protocol (TCP)/IP traffic sniffers and analyzers. Having three modes of operation, sniffer, packet logger, and network intrusion detection, it monitors network traffic and analyzes it against a user-defined rule set.

5.6 Nessus Nessus [28] is open source, i.e., free to see and modify the source as we wish. It is a remote security scanning tool that discovers any vulnerabilities that malicious hackers could use to gain access to any computer which is connected to a network. After detecting the possibility of being attacked, it is able to suggest the best way we can reduce vulnerability. The plugs that are available within it are often specific to detect a common virus or vulnerability.

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5.7 Retina Retina [29] is a network vulnerability scanning tool that generates a full brief network vulnerability report. A retina network scanner can scan multiple networks. It secures our network properly as it works against all the critical vulnerabilities. The scanner first builds a scan list from the address group and discovery options. After building a scan list, it finds out all the open, closed, and filtered ports. It tells us about the OS on the target system and also accesses the vulnerability of each port and their respective services.

6 Conclusion Internet of things security is key research area that has attracted a considerable number of researchers from all around the world. There are many challenges in the manufacturing of IoT devices such as fulfillment of customers expectations, keeping IoT hardware updated, overcoming connectivity issues, waiting for governmental standards and regulations. However, security is one of the biggest concern in the IoT devices. More research efforts are required to be in place to solve the problem. This paper provides a detailed taxonomy of various attack and security tools at network layer. Further, we have described some of those tools and provided a comparative analysis among them. We have classified tools on the basis of attack and security types. In future, we would like to use some of these attack tools to create a publicly available data for some of the IoT attacks.

References 1. Lee GM, Crespi N, Choi JK, Boussard M (2013) Internet of things. In: Evolution of telecommunication services, pp 257–282 2. Aircrack-ng, Thomas d’Otreppe de Bouvette. http://sourceforge.net/projects/airsnort/ (2006) 3. AirSnort, Blake Hegerle. http://sourceforge.net/projects/airsnort/, 8/10/2001 4. Kismet, Kershaw M (dragorn) http://www.kismetwireless.net/download.shtml, 2019-09 5. Cain and Abel, Massimiliano Montoro. https://www.hackingtools.in/free-download-cain-andabel/, 07/04/14 6. abatishchev, LOIC. https://sourceforge.net/projects/loic/. Last Accessed 28 Nov 2019 7. COTOPAXI, Botwicz J. https://www.blackhat.com/asia-19/arsenal/schedule/index.html# cotopaxi-iot-protocols-security-testing-toolkit-14325, 26-03-19 8. SSLyze, iSECPartners. https://tools.kali.org/information-gathering/sslyze 9. Hill GR, Chidgey PJ, Kaufhold, F, Lynch T, Sahlen O, Gustavsson M, Janson M, Lagerstrom B, Grasso G, Meli F (1993) A transport network layer based on optical network elements. J Lightwave Technol 11:667–679 10. Khattak HA, Shah MA, Khan S, Ali I, Imran M (2019) Perception layer security in Internet of Things. Future Gener Comput Syst 100:144–164 11. Ben-Itzhak Y (2007) Application-layer security method and system. In: Google Patents

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12. Zhao K, Ge L (2013) A survey on the internet of things security. In: Ninth international conference on computational intelligence and security, pp 663–667 13. Peng S, Shen H (2012) Security technology analysis of IoT. In: Internet of Things, pp 401–408 14. Zhang Z-K, Cho MCY, Wang C-W, Hsu C-W, Chen C-K, Shieh S (2014) IoT security: ongoing challenges and research opportunities. In: 2014 IEEE 7th international conference on serviceoriented computing and applications, pp 230–234 15. coWPAtty, Wright J. http://sourceforge.net/projects/cowpatty/, 22/03/2004 16. NetStumbler. http://www.stumbler.net/, 04/2004 17. Ferm wifi cracker. http://www.fern-pro.com/downloads.php 18. Commview for wifi, TamoSoft. http://www.tamos.com/products/commwifi/ (1998) 19. Chen Y, Trappe W, Martin RP (2007) Detecting and localizing wireless spoofing attacks. In: 4th annual IEEE communications society conference on sensor, mesh and Ad Hoc communications and networks, pp 193–202 20. Sun B, Guan Y, Chen J, Pooch UW (2003) Detecting black-hole attack in mobile ad hoc networks, detecting black-hole attack in mobile ad hoc networks 21. Douceur JR (2002) The Sybil attack. In: International workshop on peer-to-peer systems, pp 251–260 22. Choi S, Kim D, Lee D, Jung J (2008) WAP: Wormhole attack prevention algorithm in mobile ad hoc networks. In: IEEE international conference on sensor networks, ubiquitous, and trustworthy computing, pp 343–348 23. Kumar S (2007) Smurf-based distributed denial of service (DDoS) attack amplification in internet. In: Second international conference on internet monitoring and protection, p 25 24. Gill RS, Smith J, Looi MH, Clark AJ (2005) Passive techniques for detecting session hijacking attacks in IEEE 802.11 wireless networks. University of Queensland 25. Tls-prober. https://en.kali.tools/all/?tool=1433, 11-02-17 26. Acunetix. https://www.acunetix.com/penetration-testing/?utm_source=softwaretestinghelp& utm_medium=listing&utm_campaign=pentest (2005) 27. Snort, Roesch M. https://www.snort.org/, 06-01-20 28. Nessus, Tenable, Inc. https://www.tenable.com/products/nessus/nessus-professional, 29-0219 29. Retina, eEye. https://www.hackingarticles.in/retina-a-network-scanning-tool/ (1998)

A Compact Design of Stub-Loaded Multiband Microstrip Monopole Antenna for WLAN and WiMAX Applications Ajay Dadhich, J. K. Deegwal, and M. M. Sharma

Abstract A compact design of a multiband microstrip monopole antenna for wireless application is proposed. The antenna has asymmetrically shaped monopole and a quarter wavelength long inverted L-shaped stub used to resonate at 2.45 GHz and hook-shaped stub used to resonate at 3.24/5.5 GHz. By optimizing the dimension of these stubs and placing these stubs at optimized spacing resonance in 2.5/3.5/5.5, bands are achieved. The antenna is designed on FR-4 (lossy) substrate (h = 1.6 mm, dielectric constant = 4.3, and loss tangent = 0.02). Antenna is compact with dimensions of 32 × 38 × 1.6 mm3 (0.26λ0 × 0.31λ0 × 0.013λ0 ), where λ◦ is the free space wavelength at lowest resonating frequency, i.e., 2.41 GHz. The antenna resonates at 2.41 GHz (2.31–2.55 GHz), 3.24 GHz (2.93–6.75 GHz), and 8.49 GHz (8.1– 10.371 GHz) frequencies with a return loss of −18.6, −24.75, and −14.7 dB, respectively. The proposed antenna is a good candidate for 2.5/3.5/5.5 band wireless applications. Keywords CST · WiMAX · WLAN X-band

1 Introduction The recent development of modern wireless communication increased the demand for multifunctional operational ability (WiMAX/WLAN/Bluetooth, etc.,) from a single device. This multifunctional ability, low-weight, along with compactness in the antenna device is the primary motivation for researchers in the past two and three decades. Microstrip antenna is an excellent candidate for this application. Compactness can be achieved by cutting slots on patch and ground, use of shorting pins, and by increasing the electrical length with the modification of antenna

A. Dadhich (B) · J. K. Deegwal Government Engineering College, Ajmer Nh-8, Badiliya circle, Ajmer, India e-mail: [email protected] M. M. Sharma MNIT, Jaipur, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_55

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design structure. This may give radiation pattern deterioration, backpropagation, low bandwidth, etc. WLAN (wireless local area network), which is an addendum to wired LAN, has tremendous growth in modern technologies. It has frequency ranges for the antenna to operate at 2.40–2.484/5.15–5.35/ 5.725–5.85 GHz, WiMAX (Worldwide Interoperation for Microwave Access) operates at 2.5–2.69/3.4–3.69/5.25–5.85 GHz; Bluetooth working range is 2.40–2.48 GHz, whereas X-band operates at 8–12 GHz. Many research has been carried out for dual- and multiband antenna to meet the modern challenges, and different types of geometry have been described in literature—CPW-fed antenna [1], a monopole antenna with parasitic element [2], dualring resonator monopole antenna [5], L-shaped slot and C-shaped loaded and DGS annular antenna [7], coplanar waveguide-fed compact printed antenna with two rectangular metallic loops in front and a slit square-ring on backside [6], a compact CPW-fed tri-band monopole antenna with employing inverted L-slot and split ring resonator [7], microstrip-fed square monopole antenna with two branch and short stub [8], compact composite metamaterial-loaded planar antenna [9], inverted Land H-shaped slotted rectangular antenna [10], and a multi-stubs inverted L-shaped antenna and T-shaped stub with inverted long L-shaped monopole antenna [11]. All these are triple-band antennas which work for wireless applications [12–14]. A comparative study of proposed work with the previously reported literature is given in Table.1. Although many designs/shapes have been reported, all above, antenna operates in the triple band with the considerably large dimension. The proposed antenna is compact and operates in triple resonating bands with high-impedance bandwidth. In this paper, the antenna has asymmetrically shaped monopole. A quarter wavelength long inverted L-shaped stub is used to resonate at 2.45 GHz, and hook-shaped stub is used to resonate at 3.24/5.5 GHz. The proposed antenna resonates at 2.41 GHz (2.31–2.55 GHz), 3.24 GHz (2.93–6.75 GHz), and 8.49 GHz (8.1–10.371 GHz) frequencies with a return loss of −18.6, −24.75, and −14.7 dB, respectively. The proposed antenna is compact, has a good radiation pattern, and wide bandwidth of 1293 MHz at 10.35 GHz. The proposed antenna may be used for Bluetooth, WLAN, WiMAX, and X-band applications.

2 Antenna Design Figure 1 points up the configuration of the developed antenna. A 50--feed line applied to sizable L-shaped stub and hook-shaped monopole consists of partial ground. The dimension is 32 × 38 × 1.6 mm3 (0.26λ0 × 0.31λ0 × 0.013λ0 ) on the FR4 lossy substrate (r = 4.4) with thickness h = 1.6 mm. The parameters of the antenna are enumerated in Table 2.

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Table 1 Comparison of designed antenna with other recently reported antennas Reported Multiband Antenna volume (W × L × h article response mm3 )

Operating Gain frequency (dBi) range (GHz)

Bandwidth (MHz)

1

Triple band

30 × 65 × 1.6 (0.245λ0 × 0.53λ0 × 0.013λ0 )

2.375–2.525 1.1–1.5 3.075–3.8 4.6–5.6 5.0–6.9 2.0–3.6

150 725 1900

2

Triple band

20 × 38.5 × 0.8 (0.116λ0 × 0.32λ0 × 0.006λ0 )

2.10–2.49 3.22–4.3 4.89–6.12

390 400 500

4

Triple band

23 × 36 × 1 2.35–2.59 (0.187λ0 × 0.294λ0 × 0.008λ0 ) 3.31–3.93 5.07–6.35

2.67–3.19 240 2.05–2.24 620 2.66–3.48 1280

5

Triple band

28 × 32 × 1 (0.23λ0 × 0.266λ0 × 0.008λ0 )

3.8–4.4 4.0–4.65 1.9–3.5

590 620 1900

7

Triple band

33 × 50.9 × 0.8 2.8–3.0 (0.319λ0 × 0.492λ0 × 0.007λ0 ) 3.3–3.5 3.0–3.8

2.32 1.21 −6

200 200 800

8

Triple band

26 × 30 × 0.8 (0.212λ0 × 0.25λ0 × 0.006λ0 )

2.33–2.55 3.0–3.88 5.15–5.9

1.08–1.39 220 2.35–3.49 880 2.46–3.28 750

9

Triple band

23 × 38 × 1.6 2.28–2.56 (0.184λ0 × 0.304λ0 × 0.013λ0 ) 3.29–4.21 5.05–5.91

1.48–1.96 280 2.1–3.22 920 2.63–3.56 860

11

Triple band

40 × 40 × 0.8 2.35–2.58 (0.325λ0 × 0.325λ0 × 0.007λ0 ) 3.25–4.0 4.95–5.9

− 0.3 0.9 3.8

230 750 950

12

Triple band

40 × 40 × 1.6 (0.28λ0 × 0.315λ0 × 0.011λ0 )

1.87 2.90 4.13

120 20 110

13

Triple band

37 × 40 × 1.58 2.3–2.75 (0.302λ0 × 0.327λ0 × 0.013λ0 ) 3.18–3.19 5.06–6.15

2.5 0.824 2.02

450 100 1090

14

Triple band

24 × 30 × 0.79 (0.20λ0 × 0.25λ0 × 0.006λ0 )

2.50–2.71 3.37–3.63 5.20–5.85

1.33–2.52 210 1.35–2.43 260 1.25–2.62 650

Proposed Triple work band

32 × 38 × 1.6 mm3 (0.26λ0 × 0.31λ0 × 0.013λ0 )

2.31–2.55 2.93–6.75 8.1–10.37

2.26 2.16 2.25

2.29–2.88 3.26–3.88 4.17–6.07

2.02–2.14 4.26–4.28 5.45–5.56

λ0 is the wavelength in free space at the lowest resonating frequency

2.7–3.2 3.1–3.5 2.8–3.3

225 3818 2219

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Fig. 1 Design of proposed antenna, 1 front view, 2 bottom view

Table 2 Parameters of proposed antenna

Parameter

Value (in mm)

Parameter

Value (in mm)

H

32

g

0.9

H1

9.1

P1

16.2

H2

6

P2

4.5

H3

2.5

P3

12

H4

2.1

P4

5

H5

1

P5

2.5

H6

4

Pg

18

P

38

Wf

2

3 Results and Discussions The proposed antenna simulated on CST-2017 for 1 to 14 GHz frequency range; S11 , impedance, surface current, and radiation patterns in XZ and YZ plane for respective frequencies observed and analyzed. The simulated S11 of the proposed antenna manifested in Fig. 2 shows that antenna is resonating at five operating bands. For S11 ≤ 10 dB, three resonance with a wide bandwidth of 225 MHz (2.31–2.55 GHz), 3818 MHz (2.93–6.75 GHz), and 2219 MHz (8.1–10.37 GHz) with a return loss of −18.6, −24.75, and −14.7 dB at 2.41, 3.24, and 8.49 GHz observed, respectively. The real and imaginary part of the simulated input impedance of the proposed antenna concerning frequency is shown in Fig. 3. At each resonating frequency, imaginary part of the impedance is near to zero, and the real (impedance(R)) is approximately 50 . This is the evidence for the resonance at the operating frequencies.

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Fig. 2 Representation of S11 (dB) against frequency (GHz) for the proposed antenna

Fig. 3 Frequency versus input impedance plot of proposed antenna

The surface current of the proposed antenna at respective resonating frequencies is demonstrated in Fig. 4. At 2.45 GHz, antenna is resonating through the length of sizeable inverted L-shaped stub, at 3.24 GHz antenna shows high radiation at the length of hook-shaped stub, and at 8.49 GHz surface current is maximum on the length of hook-shaped stub and small L-shaped stub as shown in Fig. 4.

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Fig. 4 Surface current distribution of proposed antenna at a 2.45 GHz, b 3.24 GHz, c 8.49 GHz

Figure 5 shows the simulated co- and cross-polarization radiation pattern for the proposed antenna in the XZ and YZ plane at five resonating frequencies. A suitable radiation pattern is observed at all resonating frequencies.

4 Conclusion A novel and compact design of a multiband microstrip monopole antenna is presented. The antenna has quarter wavelength long inverted L-shaped stub to resonate at 2.45 GHz and hook-shaped stub to resonate at 3.24/5.5 GHz. Antenna resonates at three operating frequencies 2.41 GHz (2.31–2.55 GHz), 3.24 GHz (2.93– 6.75 GHz), and 8.49 GHz (8.1–10.371 GHz) frequencies with a return loss of −18.6, −24.75, and −14.7 dB, respectively. The achieved bandwidths for three bands are 255, 3818, and 2219 MHz. Antenna has magnificent surface current distribution and radiation pattern for all resonating frequencies, real and imaginary impedance, and reflection coefficient is observed and analyzed for 1–14 GHz. The proposed antenna is worthy of wireless application in WLAN/WiMAX/Bluetooth/X-band.

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Fig. 5 Radiation pattern in both planes at a 2.45 GHz, b 3.24 GHz, c 8.49 GHz

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References 1. Li J, Guo J, He B, Zhang A, Liu QH (2016) Tri-band CPW-fed stub loaded slot antenna design for WLAN/WiMAX applications. Frequenz 70(11–12):521–526 2. Sim CYD, Chen HD, Yeh CH, Lin HL (2015) Small size triple band monopole antenna with a parasitic element. Microw Opt Technol Lett 57(2):342–348 3. Dadhich A, Deegwal JK (2018) Multiband microstrip patch antenna with rectangular slots on patch for blue-tooth and c-band applications. Int J Eng Technol 7(1.2):191–193. ISSN: 2227-524X. https://doi.org/10.14419/ijet.v7i1.2.9064 4. Tang Z, Liu K, Yin Y, Lian R (2015) Design of compact triband monopole antenna for WLAN and WiMAX applications. Microw Opt Technol Lett 57(10):2298–2303 5. Liu G, Liu Y, Gong S (2016) Compact tri-band wide-slot monopole antenna with dual-ring resonator for WLAN/WiMAX applications. Microw Opt Technol Lett 58(5):1097–1101 6. Dadhich A, Deegwal JK, Sharma MM, Yadav S (2019) CPW fed monopole microstrip antenna for multiband wireless applications. In: IEEE Indian conference on antennas and propogation (InCAP). Ahmedabad, India, pp 1–4. https://doi.org/10.1109/InCAP47789.2019.9134677 7. Goswami SA, Karia D (2017) A compact monopole antenna for wireless applications with enhanced bandwidth. Int J Electron Commun 72:33–39 8. Yang X, Kong F, Liu X, Song C (2014) A CPW-fed triple-band antenna for WLAN and WiMAX applications. Radio Eng 23(4):1086–1091 9. Xu Y, Zhang C, Yin Y, Yang Z (2015) Compact triple-band monopole antenna with inverted-L slots and SRR for WLAN/WiMAX applications. Prog Electromagn Res Lett 55:1–6 10. Dadhich A, Deegwal JK, Sharma MM (2020) Study and design of slotted antenna with inset feed for multiband wireless application. Test Eng Manage J 83:23465–23472. ISSN: 0193-4120 11. Sun XL, Zhang J, Cheung SW, Yuk TI (20122) A triple-band monopole antenna for WLAN and WiMAX applications. In: The 2012 IEEE international symposium on antennas and propagation (APSURSI). Chicago, IL, pp 1–2 12. Gupta A, Chaudhary RK (2016) A compact planar metamaterial tripleband antenna with complementary closed-ring resonator. Wirel Pers Commun 88(2):203–210 13. Kundu A, Bhattacharjee AK (2015) Design of compact triple frequency microstrip antenna for WLAN/WiMAX applications. Microw Opt Technol Lett 57(9):2125–2129 14. Kumar A, Jhanwar D, Sharma MM (2017) A compact printed multistubs loaded resonator rectangular monopole antenna design for mutiband wireless systems. Int J RF Microw ComputAided Eng 21147:1–10

Smart Transportation for Warehouses T. Jaya Sankar and P. C. Jain

Abstract Smart transportation for warehouses focuses on moving of the products more secure and efficient. The paper focuses on different parts such as maintaining the safety of the products and the container, keeping track of the products being loaded, tracking the location of the container, alerting in the case of an accident, and also using efficient communication system in the case of an emergency. The first part of the paper deals with the safety of the products which involves keeping track of the surrounding parameters such as temperature, humidity, and fire (in case a fire broke out) by utilizing temperature, humidity sensor, and the fire sensor and also the accident detection using accelerometer and gyroscope. This part also deals with the information of the products using RFID and using conditional statements to verify whether the product is allotted to a specific container. The second part focuses on the communication system with which all the required alerts can be transmitted in the form of SMS and emails with the help of third-party APIs. Using the data from the emails, a software platform is used to automate the task of storing that data in Google sheets. The second part also deals with location tracking of the container using Google maps, which helps the person in-charge to keep track of the locations the container has been to. The final part deals with the implementation of the dashboard. Keywords ITS · Warehouse · RFID · Sensors · Location tracking

1 Introduction Smart transportation is one of the most important applications in Internet of Things (IoT) that refers to the integration of modern technologies and management strategies in transportation systems. These technologies aim to provide innovative services T. Jaya Sankar (B) · P. C. Jain (B) Department of Electrical and Engineering, School of Engineering, Shiv Nadar University, Greater Noida, Uttar Pradesh, India e-mail: [email protected] P. C. Jain e-mail: [email protected] © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_56

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relating to different modes of transport and traffic management and enable users to be better informed and make safer and smarter use of transport networks. Smart transportation is a part of the intelligent transportation system. Intelligent transportation system (ITS) is defined as a system in which information and communication technologies (ICT) are applied in the field of road transport, including infrastructure, vehicles and users, and in traffic management and mobility management, as well as for interfaces with other modes of transport. The evolution of intelligent transportation system is providing a growing number of technology solutions for transportation managers as they seek to operate and maintain the systems more efficiently and improve performance. Now-a-days e-commerce industry is booming. Various companies have been investing a lot and maintaining various warehouses for the products. Most of the products are usually imported from other countries and we need transportation in order to get these products into the warehouses. There could be a hassle when working with the moving and organizing of the products at such a large scale. The smart transportation is a subset of the intelligent transport system (ITS). The idea basically deals with the proper selection of methods that are the part of ITS and bringing together a group of ways which provides for a better implementation of robust transportation methods for the products. Smart transportation has been divided into various parts that tries to group a variety of ways that help in efficiently transporting of the products from one location to the another keeping in the mind the safety of the products themselves. Basically, it involves utilization of various sensors that together form a safety net for the products, a couple of third-party APIs (Application Programming Interface) that helps in efficient messaging systems, and radio frequency identification (RFID) for keeping the track of the products that are being dealt with. This paper has conceptually implemented a smart transportation for warehouses. Section 2 discusses about the status of smart transportation, while Sect. 3 discusses about implementation of RFID and sensors, communication and APIs, and dash board. Section 4 discusses results obtained after implementation, and finally Sect. 5 concludes the paper.

2 Related Work Smart transportation plays a central role in supply chains affecting every part of the process from planning and procurement to logistics and lifecycle management. This leads to more sales, helping business to grow. The physical movement of goods is a critical link in supply chain and more and more business are relying on smart transportation to manage above function. Smart transportation uses technology to help business plan, execute, and optimize the physical movement of goods, both incoming and outgoing, and making sure the shipment is complement. It provides visibility into day-to-day transportation in operations, and ensuring the timely delivery of goods. Smart transportation provides track and trace services, enabling real time information

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exchange among carriers, distributors, warehouses, and customers. Smart transportation reduces costs for the business and the end customers, simplification of supply chain process improvement in visibility in transit, faster delivery times. One of the papers [1] mainly focuses on how to manage products effectively and resources efficiently in a warehouse. It stresses on the need for improved monitoring, tracking, controlling systems and how traditional methods cannot sustain the increasing dynamicity of the market. It proposes the development of an IoT system that helps the person in-charge to have more control and monitor the operations in real time and with the help of Wi-Fi, RFID, sensors, and cloud computing a robust system could be developed for gathering and sharing information that help in managing the processes. The proposed scheme talks about the use of RFID to track the products effectively and how it helps in reducing the errors in product distribution. On the whole, this paper states that the sensors, actuators, and RFID are used to collect the required data and with the help of wireless network, these data are then sent to the cloud to analyse and simplify the data so that it can be easily understood by the users. Another paper [2] focuses on creating an IoT based intelligent Transport system that intends to utilize the progressed and capable communication systems. It focuses on the importance of wireless sensor networks and how when sensors are utilized properly, and they lead to a better and efficient ways of transportation. The mainly focused components in this case are the accelerometer, GSM, GPS, RFID reader, and temperature sensor. The third paper utilized the accelerometer for getting the information on the accident and when paired up with a GSM and GPS, the required message gets transmitted along with the location details and also helps in the case of location tracking. It also talks about the importance of the RFID system and how it helps the person responsible to be authenticated in order to gain specific control wherever required in a transport system.

3 Implementation The implementation of smart transportation is done by utilizing a network of sensors, RFID module, Raspberry Pi, an IoT platform and a couple of third-party APIs. This paper has been divided into three major parts. One comprising of all the required sensors to be able to collectively work together for obtaining the useful data along with the RFID module that deals with the verification of the products. The second part deals completely with the communication methods such as sending data to IoT platform for monitoring, using Google maps, API for location tracking, and using Twilio API and Mailjet API for messaging service that is used in the case of sending out SMS and email alert to the authorized individual. The third part involves making a dashboard website where one can keep track of the location along with monitoring the data. Figure 1 shows the block schematic of smart transportation for warehouses.

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Fig. 1 Block diagram smart transportation

3.1 Sensors and RFID Implementation This part mainly focuses on the hardware that includes sensors and the RFID module. Figure 2 shows the complete hardware utilized. It consists of RC522 RFID reader module connected with Arduino Uno processor, DHT11 temperature and humidity sensor, SEN16 flame detector, and a MPU6050 six-axis motion detector for accident detection, connected with Raspbery Pi 3b + mini processing unit. DHT11 is utilized in order to get the temperature and humidity values. For product management, RC522 RFID module helps in removing hurdles like products mismatch etc. With every container having its own set of unique IDs, only those products that have the IDs in that range and are identified by the RFID reader can be loaded into the container [3]. In case a fire broke out, the SEN16 flame sensor identifies it and send out alarm [4]. The MPU6050 helps in the case of accident. With inbuilt accelerometer and gyroscope, it becomes easier to set a stable range and by any chance if the values cross the defined range, it denotes that the container has met with an accident [5].

3.2 Communication Methods Involving APIs This part deals with the communication methods such as sending data acquired from the DHT11 sensor to the IoT platform for monitoring, using APIs for location

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Fig. 2 Final setup including sensors and RFID

tracking and for injecting a messaging service that is used in the case of sending out an alert to an authorized individual. ThingSpeak is an IoT analytics platform service that allows us to aggregate, visualize and analyze live data streams in the cloud. It provides instant visualizations of data posted by our devices. The implementation of the DHT11 sensor has been done in the first part. Now in order to visualize the data, we need an IoT platform and that’s where the ThingSpeak platform comes in. By utilizing the ThingSpeak’s custom library for sensor data transfer, the DHT11 temperature and humidity sensor data are read and graphs can be visualized. Google Maps Javascript API is utilized in order to work with the concept of location tracking. It lets us customize maps with our own content and images for display on web pages and mobile devices. In other words, it helps us add better interactivity. To visualize the map, we need a website. HTML is an integral part of the web. It not only helps in creating website layout but also provides a lot of custom functions involving the GPS (Global Positioning System). Every device is GPS enabled, and using html Geo-location API, one can grab the location from the browser and work on it. The simple and effective form of an alert is the SMS. Twilio platform is used to create a robust system of sending out alerts in the case of the sensor values crossing the threshold like gyroscope and accelerometer values change during an accident, temperature is dropping, fire broke out, we need to trigger an alert. So, in order to send email alerts that can help in future use, we utilize Mailjet platform. It removes the hassle of sending out emails by heavy code and provides a beautiful dashboard to keep track of all the emails sent. We might need the data from the emails in order to study and improve the performance. So, using an online automation platform called Integromat, we can now

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connect multiple services to work together. In this case, the focus is on Gmail and Google sheets. By defining what kind of emails one wants Integromat to scrap the data from, it then stores the acquired data into Google sheets. These sheets can be sent around to different departments for deeper insights.

3.3 Dashboard Implementation This part completely deals with the admin side which involves creating a dashboard website for monitoring sensor values and keeping track of the container’s location. Dashboard is a useful concept that helps us to keep track of multiple things in a single place. The dashboard is built using html, css, and some JavaScript. Since location tracking part already implemented, we just need to import that code into the dashboard. For the sensor values, the ThingSpeak provides us with a wonderful feature of exporting the graphs from their own website in the form of i-frames. The i-frame is an html document embedded inside another html document on a website. In this case, we display the graphs in our dashboard, but behind the scenes, ThingSpeak itself grabs the sensor data and updates it. So, the only important thing in this section is to design and website layout and styling it. The flow chart of smart transportation shown in Fig. 3 completely explains the process of all the sensors, modules and software platforms that are involved in implementing the smart transportation.

4 Results The smart transportation for warehouses can be implemented by effectively grouping together various sensors, the concept of RFID, micro-controllers and a few third-party APIs that help us in automating various tasks. Results given below explain the flow of smart transportation. (a)

(b)

The first part deals with the products. Every container has its own set of products to be loaded. Figure 4 shows that if the product UID is among the set of UIDs defined in the container code, the product can be loaded. If not, display an error (second item UID 64 in Fig. 4). For the location tracking of the container, we have utilized the Google Maps API with pre-defined locations. This has been done by simulation of moving to different places without actually traveling. Figure 5 shows the tracking. The location name in red color denotes the current location and if it turns to green color, and it means that the container has passed that location. For every location, a marker is placed on the map.

Smart Transportation for Warehouses

Fig. 3 Flow chart of smart transportation

Fig. 4 Serial monitor display the product messages

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Fig. 5 Display of locations container traveled

(c)

(d)

(e)

(f)

DHT11 sensor keeps track of the temperature and humidity levels in the container. In order to monitor these in the form of graphs, we have introduced the ThingSpeak IOT platform (Fig. 6). By connecting the Raspberry Pi to the internet, we have been successful in transferring the sensor data to ThingSpeak. A dashboard is needed where the person in-charge can keep track of the location of the container as well as monitors the graphs generated above. Using html, css, and Javascript a basic dashboard website designed, which displays the map (Fig. 7), the locations, and also the graphs which are pulled from the ThingSpeak website. The sensors such as the SEN16 and the MPU6050 have been utilized for fire detection and accident detection, respectively. With these sensors along with DHT11, one needs an alerting system in the case of a fire or an accident or the temperature value beyond the limit. We have utilized the Twilio messaging API and the Mailjet API. With Twilio, we send an alert SMS (Fig. 8) to the person in-charge and in the case of Mailjet, we send custom messages in the form of an email (Fig. 9). The above step helps in solving the current problem. But in the long run, we need to store the data so that it can be analyzed and will provide better insights to improving the transportation methods. For example, there might be cases of faulty sensors that send out a random alert frequently. This problem might be corrected at present, but if this data is stored and studied, more efficient

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Fig. 6 Temperature and humidity graphs on ThingSpeak platform

Fig. 7 Final dashboard with graphs and map

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methods can be introduced later. Figure 10 shows integration of Gmail with Google sheets on Integromat. Integromat scraps the data from these emails and then stores the data into custom Google sheets (Fig. 11).

Fig. 10 Integrating Gmail with Google on Integromat

Fig. 11 Google sheet with emails data

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5 Conclusions With the concept of smart transportation discussed in this paper, the large e-commerce companies can now create a simple yet effective and organized way to transport the products, while also keeping the safety of the products in mind. With dual communication systems such as the SMS and email in place, companies can address the problems fast and also store the data consistently. This method provides insights into the stored data that can be worked on and improved later in future. Acknowledgements The first author is grateful to Prof. S. Sen, Director, School of Engg., and Prof. Dinkar Prasad, Head, EE Dept., and Associate Director, School of Engineering., Shiv Nadar University, G. Noida (UP), India for providing necessary resources and infrastructure to complete this project work and second author for their encouragement, and permission to publish this paper.

References 1. Accorsi R, Manzini R, Maranesi F (2014) A decision-support system for the design and management of warehousing systems. Comput Ind 65(1):175–186 2. Lee SJ, Tewolde G, Kwon J (2014) Design and implementation of vehicle tracking system using GPS/GSM/GPRS technology and smartphone application. In: IEEE world forum on Internet of Things (WFIoT), Seoul, Mar 2014 3. Hsu CI, Shih HH, Wang WC (2009) Applying RFID to reduce delay in import cargo customs clearance process. Comput Ind Eng 57(2):506–519 4. Jedermann R, Behrens C, Westphal D, Lang W (2006) Applying autonomous sensor systems in logistics—combining sensor networks, RFIDs and software agents. Sens Actuators A 132(1):370–375 5. Jacob B, Feypell-de La Beaumelle V (2010) Improving truck safety: potential of weight inmotion technology. IATSS Res 34(1):9–15 6. Prabha C, Sunitha R, Anitha R (2014) Automatic vehicle accident detection and messaging system using GSM and GPS modem. Int J Adv Res Electr Electron Instrument Eng 3(7):10723– 10727

Mitigating Nonlinear Effects in 16 Channel WDM Radio Over Fiber System with Dispersion Compensation Fiber and Fiber Bragg Grating Combination Suresh Kumar, Shagun Singh, and Payal

Abstract The exponentially rising data volume and the advancement in multimedia technologies such as ultra-high definition online video streaming requires enormous bandwidth with reduced latency for seamless service delivery. Radio–over-Fiber (RoF) with wavelength division multiplexing (WDM) technology is a very promising approach to be used in conjunction with wireless access networks. However, the integration of optical fiber and wireless communication brings nonlinear effects due to higher number of users, introduction of signal noise, unwanted frequencies, low quality of signals, etc. In this paper, a 16 channel each of 10 Gbps data rate WDM based RoF system is analyzed for optimum performance at variable input power using dispersion compensation fiber (DCF) and fiber bragg grating (FBG) combination at different channel spacing of 50 and 100 GHz. The efficacy of the system is compared with traditional WDM-RoF system using bit error rate (BER), quality factor (QFactor) and eye diagrams as performance metrics using Optisystem simulator. It has been found that the system offers optimum performance at a power level of – 5 dBm for all the evaluation parameters. Also for larger channel spacing, i.e., 100 GHz, the network offers optimum performance in comparison to 50 GHz. Keywords Cross-phase modulation (XPM) · DCF · Four wave mixing (FWM) · FBG · Mach Zehnder modulator (MZM) · RoF self-phase modulation (SPM)

1 Introduction The currently rising bandwidth necessities have led the technologists to move to optical communications. RoF technology is one of the best means that facilitates cost-effective and trustworthy communication. In RoF technology, the modulated RF signal is transmitted uplink and downlink between central station (CS) and base station (BS) [1, 2]. This transmission is done through an optical fiber link. The optical

S. Kumar (B) · S. Singh · Payal Department of ECE, UIET, MDU Rohtak, Haryana, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_57

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Fig. 1 Nonlinearities in an optical fiber

fiber can introduce various nonlinearities in the transmission [3]. FSO is another highspeed communication technology that transmits optical signal through air at much lesser costs and is termed as fiber optics without fiber [4]. In a communication system, nonlinear effects play both generative and degenerative roles. Nonlinear effects on one hand enhance the fiber performance and offer innovative applications such as fiber lasers, multiplexers, De-multiplexers etc., while on the other hand it limits the optical fiber communication. Fiber nonlinearities can be categorized under two classes. The first category is the index related category which includes the nonlinearities caused due to intensity dependent disparities in refractive index of optical fiber termed as Kerr Effect. The Kerr effect can further be categorized into different nonlinearities namely, SPM, XPM and FWM. The second category of nonlinearities is scattering related, which is caused by inflexible stimulated scattering which are classified as SRS and SBS [5, 6]. Figure 1 represents the classification of nonlinearities in an optical fiber. SPM—As the name suggests, it denotes self-induced phase shift which is induced by the optical field at the time of its transmission in the fiber. When light beam propagates through optical fiber, it leads to high intensity of light in core which further results in high-refractive index. This variation in refractive index results in phase changes with respect to time. Nain A. et al. investigated the performance of MZM and optical phase modulator (OPM) under SPM effect in RoF system [7]. XPM—In XPM, one wavelength of light can have an effect on the phase of another wavelength of light. In this the nonlinear refractive index of a light signal beam is reliant on its own intensity as well as the intensity of other beams propagating through the fiber. It translates the power variations of one wavelength channel into phase variations of other channels. Nain A. et al. studied the influence of XPM crosstalk in SCM based RoF systems. It was found that the crosstalk increases with the rise in transmission distance and modulation frequency [8]. FWM—It takes place when wavelength channels are placed very near each other. Channel spacing and fiber vary the FWM products. The wavelengths that propagate through the fiber simultaneously produce a new wavelength (also called as idler) by the effect of FWM and this resultant wavelength is different than any of the input wavelengths. f x yz = f x + f y + f z (x, y = z)

(1)

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where, f x , f y ,f z = frequency of input signals and f xyz = frequency of resultant signal. The equations below represent the high-order dispersion parameters of FWM in terms of propagation constant ‘b’ by Taylor series; d2 b db 1 + (ω − ω0 )2 2 dω 2 dω 3 4 1 1 3d b 4d b + (ω − ω0 ) + + ··· − ω ) (ω 0 6 dω3 24 dω4

b = b0 + (ω − ω0 )

(2)

db Now, dω = δ, where δ is the propagation delay per optical length. We will replace the values in Eq. (2);

dδ dω 2 3 π3 2π 2 3d δ 4d δ + + ··· + f ( f ) ( ) 3 dω2 3 dω3

b = b − b0 = 2π [( f )δ + π ( f )2

(3)

Therefore, by (4), second order dispersion is given by; b2 =

λ2 dδ λ2 dδ = = F0 dω 2π c dλ 2π c

(4)

Third order dispersion is given by;   2 d2 δ λ2 dδ 2d δ λ = + 2λ dω2 dλ2 dλ (2π c)2 2   λ λ2 F1 − 2λF0 = (2π c)2

b3 =

(5)

where, f = channel spacing, F 0 = fiber chromatic dispersion [9]. The resultant FWM signal power is expressed as: PF (L) =

2  Px (0)Py (0)Pz (0) −αL 1 − eαL 1024π 2 e . n (D ). X π 4 λ2 c 2 α2 A2e f

(6)

where L = fiber length; λ = wavelength; c = speed of light; D = degrading factor; X = nonlinear susceptibility; Px Py Pz = power of input signal with frequency f x , f y and f z, respectively; Aef = effective area of optical fiber core; α = loss coefficient; n = refractive index. When input channels increase, the FWM sideband products also increase. When an equal amount of power is given as input to all the channels and phase matching is maintained then the following equation holds true:

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PFWM = (DF γ L)2 Pi2 (mW) P0

(7)

where γ denotes non-linear coefficient, Pi is the input power and DF is the degeneracy factor, which is defined such that DF = 1, when x = y but 2 when x = y, i.e., different frequency channels. At high optical power levels, the FWM effects become significant. Kathpal and Garg studied 8/32 RoF-BF system and concluded that the FWM effect decreases with decrease in power level and increase in channel spacing [10]. Pham T. V. et al. investigated the performance of hybrid Optical fiber/Free Space optics (FSO) backhaul downlink over FWM impaired WDM-Passive Optical Network (PON) taking into consideration other physical impairments too. The results revealed that the FWM effects and BER can be reduced by employing amplifier or Avalanche Photodiode (APD) at the receiver [11]. Garg, A. K. and Janyani, V. proposed a WDM–TDM (Time Division Multiplexed) PON that proved to be flexible, bandwidth scalable and energy proficient. It was capable of providing services to the users at higher data rates and reducing the consumption of power in cases of low traffic [12]. Garg, A. K. and Janyani, V. reviewed existing energy efficient approaches and proposed an energy-efficient hybrid WDM–TDM PON architecture which allows dual rate transmission based on traffic load. The architecture is much flexible to line card failure and energy efficient to support next generation optical networks [13]. Scattering induced effects—SRS plays a vital role in WDM fiber communication system. The photons of the optical beam on interacting with the molecular vibrations of the fiber and the photons of other light signals result in scattering of light. The wavelength of the resultant light is longer than the other light signals [14]. Nain, A. et al. concluded that SRS induced crosstalk varies with variation in modulation frequency and optical power, whereas it remains almost unchanged with variation in transmission length [15]. SBS causes scattering of light in backward direction. This effect can be minimized by keeping the input power lesser than the threshold or by increasing the spectral line of the source. SBS effects are also dependent upon the bandwidth of the input signal and on the optical fiber [16]. The present work aims at mitigating the nonlinearities in the propagation path with the design of a 16Ch (160 Gbps) WDM-RoF system with DCF and FBG combination. The remaining work is structured as follows. The designed simulation schematic is described in Sect. 2. The results are explained in Sects. 3 and 4 presents the paper conclusion.

2 Simulation Design The schematic representing WDM-RoF system using DCF and FBG combination is depicted in Fig. 2. The transmitter section comprises of sixteen RF signals that are

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Fig. 2 Block diagram of WDM-RoF system with DCF and FBG combination

combined using WDM multiplexer and are transmitted over optical fiber at 50 km transmission distance. Each transmitter includes a pseudo-random bit sequence (PRBS) generator that generates binary data at 10 Gbps data rate which is given to return to zero (RZ) pulse generator for base band signal generation. The RZ coding format has been used due to its increased fairness to fiber nonlinear effects. It is then multiplexed over RF sinusoidal source. The MZM provides optical modulation using high-frequency optical carrier signal generated by CW laser. For capacity enhancement of the system, the modulated output optical signals from different CW lasers at different operating wavelengths are combined using WDM multiplexer. The multiplexed signal propagates through the single-mode fiber (SMF). The design incorporates Erbium Doped Fiber Amplifier (EDFA) due to its capability of providing the necessary level of amplification to the optical signal with minimum amount of noise. In order to compensate for link losses and dispersion, DCF and FBG are used in the layout schematic. The amplified signal goes through DCF which offers equal and opposite dispersion of − 80 ps/nm/km in order to make chromatic dispersion zero. FBGs are made by exposure of fiber core to intense UV light which causes an increase in refractive index of fiber core, which produces a fixed modulation index known as grating [17]. It is a low-cost filter that reduces chromatic dispersion and does wavelength selection. The apodized and linearly chirped FBG (LCFBG) has been used in the system design for reducing side lobe levels and ripples in group delay response. For improved receiver sensitivity and required SNR, a pre amplifier is used before the receiver. The signal after passing using WDM demultiplexer is received by 16 remote stations. Each receiver comprises of an Avalanche Photodiode detector (APD) for optical to electrical conversion, fourth order low-pass Bessel filter for noise removal, 3R regenerator for recreation of original electrical signal and BER analyzer for visualizing the results. The optical power meter has also been used to examine the impact of nonlinearities. The simulation layout is designed in Optisystem 16.1. Figure 3 depicts the designed layout of 16 Ch (160 Gbps) WDM-RoF System with DCF and FBG combination, transmitter subsystem and receiver subsystem.

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(a) Designed layout

(b) Transmitter subsystem

(c) Receiver Subsystem

Fig. 3 Simulation layout of 16Ch (160 Gbps) WDM-RoF system with DCF and FBG combination

3 Results and Discussion The designed 16-Channel WDM-RoF network has been analyzed with variation in input transmission power using performance metrics: Q factor, BER and Eye diagrams. The two different network configurations (i) simple optical fiber link (ii) optical fiber with combination of DCF-FBG at two different channel spacing scenarios of 50 and 100 GHz are taken into consideration. Figure 4 shows the variation of Q Factor with variation in input power for three different channels (channel 1, 8 and 16) at a channel spacing of 50 GHz. The input power is varied from 5 to – 15 dBm in a step size of 5 dBm. The value of Q Factor

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8 6 4 2 0 5

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Power (in dBm) Fig. 4 Variation of Q factor with input power at 50 GHz channel spacing

shows a decreasing trend with increase in number of input channels. Further, with variation in input power from 5 to – 15 dBm, the value of Q Factor first increases from (7.52, 7.30 and 6.51) at 5 dBm to (8.66, 8.24 and 8.18) at – 5 dBm for channel 1, 8 and 16, respectively. The Q Factor decreases to (5.26, 5.15 and 5.14) for channel 1, 8 and 16, respectively as the input power further decreases from – 5 to – 15 dBm. Figure 5 shows the BER curves corresponding to variation in input power. The variation in BER is in synchronization with Q Factor variation. With decrease in input power, the Q Factor increases and BER reduces up to optical power of – 5 dBm below which the system performance is deteriorated. As the power decreases further, the FWM power generated to the existing channel power decreases which in turn reduces the SNR. Figure 6 depicts a bar chart showing the variation in Q Factor for three different channels at 100 GHz channel spacing.

Fig. 5 Variation of BER with power at channel spacing of 50 GHz

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Fig. 6 Variation of Q factor with power at channel spacing of 50 GHz

With variation in input power from 5 to – 15dBm, the value of Q Factor first increases from (7.66, 7.43 and 6.98) at 5 dBm to (9.79, 9.61 and 8.12) at -5dbm for channel 1, 8 and 16, respectively. The Q Factor decreases to (6.31, 5.20 and 5.12) for channel 1, 8 and 16, respectively, as the input power further decreases from – 5 to – 15 dBm. Figure 7 shows that with increase in channel spacing, a decrease in BER is observed from 2.30E – 18 to 5.51E – 23 for channel 1 and consequently for other channels at the optimum power level of – 5dBm. From Figs. 4 and 6, the value of Q Factor increases at a particular value of input power with increase in channel spacing from 50 to 100 GHz. At input power of – 5 dBm, an improvement of up to 16.62% is obtained in the Q Factor. Figure 8 depicts a bar chart showing the variation in Q Factor values for channel 1, 8 and 16 for a hybrid combination of DCF-FBG at a channel spacing of 50 GHz. The input power is varied from 5 to – 15 dBm. The value of Q Factor first increases from (7.63, 7.48 and 6.82) at 5 dBm to (11.54, 11.25 and 10.41) at – 5 dbm for channel 1, 8 and 16, respectively. The Q Factor decreases to (8, 7.33 and 7.28) for

Fig. 7 Variation of BER with power at channel spacing of 100 GHz

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channel 1, 8 and 16, respectively, as the input power further decreases from – 5 to – 15 dBm. Figure 9 shows the variation of BER with input power. The variation in BER is in synchronization with the variation in Q Factor, the value of BER first decreases with variation in input power then increases. Figure 10, depicts a bar chart showing the variation of Q Factor for three different channels at a channel spacing of 100 GHz. From Fig. 11, the value of Q Factor decreases with increase in number of channels. Further as the input power is varied from 5 to – 15 dBm, the value of Q Factor first increases from (7.89, 7.83 and 7.67) at 5 dBm to (12.24, 11.63 and 11.50) at – 5 dbm for channel 1, 8 and 16, respectively. The Q Factor decreases to (8.48, 7.57 and 7.56) for channel 1, 8 and 16, respectively as the input power further decreases from – 5 to – 15 dBm.

Fig. 9 Variation of BER with power at channel spacing of 50 GHz

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Fig. 11 Variation of BER with power at channel spacing of 100 GHz

Figure 11 shows BER versus input power at 100 GHz channel spacing. The variation in BER is in synchronization with the variation in Q Factor, and the value of BER first decreases with variation in input power then increases. From Figs. 8 and 10, with increase in channel spacing from 50 to 100 GHz the value of Q Factor increases at a particular value of input power. At input power of – 5 dBm, an improvement of upto 10.47% is obtained in the Q Factor. With increased amount of nonlinearities, the propagation path losses increase. This reduces the Q Factor and increases the error rate. These results have been justified by studying the eye diagrams of both the conventional and DCF-FBG based ROF system. Eye diagrams are used for evaluating the joint effects of nonlinearities and interference during propagation in a communication system. Figures 12 and 13 show the eye diagrams at a channel spacing of 100 GHz and input power of -5dBm for conventional and DCF-FBG based ROF system, respectively.

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Fig. 12 Eye diagrams at channel 1, 8 and 16 for conventional WDM based RoF System at – 5dBm input power and 100 GHz channel spacing

Fig. 13 Eye diagrams at channel 1, 8 and 16 for WDM based RoF system with DCF and FBG at – 5dBm input power and 100 GHz channel spacing

The eye diagrams obtained in Fig. 12 have smaller eye opening as compared to those obtained in Fig. 13 due to the impact of nonlinearities in the propagation path such as third-order harmonics, FWM etc. The eye height for channel 1, 8 and 16 is 0.0019, 0.00057, and 5.76E – 5, respectively. It subsequently reduces with increase in number of channels due to increased interference. With the use of hybrid combination of DCF and FBG in the conventional WDM Based RoF system, larger eye openings are obtained with eye height 2.16, 2.10, 2.09 corresponding to channel 1, 8 and 16, respectively. Larger the eye opening, best is the SNR. From Fig. 12, the eye diagrams are thicker which shows that the amount of distortion is larger. The clear eye diagrams of Fig. 13 give a clear indication that the impact of nonlinearities has been mitigated.

4 Conclusion In the present work, a 16Ch (160 Gbps) WDM-RoF system has been evaluated for optimum performance using Q Factor and BER with varying input power at channel spacing of 50 and 100 GHz and is compared with traditional RoF system. In order to

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mitigate the non-linearities, DCF-FBG hybrid combination has been used. From the qualitative analysis, with the reduction in input power, the Q Factor first increases up to input power of – 5 dBm and there after it decreases with input power, whereas the BER first decreases up to – 5 dBm and then increases thereafter with further variation in input power. The results also depicted that at a given input power, the Q Factor increases with increase in channel spacing from 50 to 100 GHz and an improvement of up to 16.52% is obtained. The WDM-RoF system with hybrid combination of DCF-FBG performs efficiently at input power of -5dBm and provides an optimum BER performance with an improvement of up to 41.42% in the value of Q Factor as compared to conventional WDM based RoF system. On further analysis of eye diagrams, the hybrid combination of DCF-FBG provides larger eye openings with eye height 2.16, 2.10, 2.09 for channel 1, 8 and 16, respectively. With considerable increase in channel spacing (50–100 GHz) and a simultaneous reduction in channel input power in the designed system, the effect of nonlinearities has been sufficiently mitigated at an optimum power of – 5 dBm.

References 1. Singh S, Kumar S, Payal (2020) Radio over fiber communication system: lateral shift in cellular communication. Int J Emerg Technol 11(2):731–734 2. Kumar S, Sharma D, Payal SR (2020) Performance analysis of radio over fiber link using MZM external modulator. In: Luhach A, Kosa J, Poonia R, Gao XZ, Singh D (eds) First international conference on sustainable technologies for computational intelligence ( ICTSCI-2019), vol 1045. Springer, Singapore ASIC, Jan 2020. ISBN: 9789811500282. https://doi.org/10.1007/ 978-981-15-0029-9_18 3. Payal KS (2018) Nonlinear impairments in fiber optic communication systems: analytical review. In: Futuristic trends in network and communication engineering (FTNCT-2018), vol 958. Chapter 3, Springer, Singapore, CCIS, 28–44 Jan 2019. ISSN 1865-0929. https://doi.org/ 10.1007/978-981-13-3804-5_3 4. Willebrand HA, Ghuman BS (2001) Fiber optics without fiber. IEEE Spectr 38(8):40–45 5. Ferreira M (2017) Impact of nonlinearities on fiber optic communications. Preprints 2017, 2017120175.https://doi.org/10.20944/preprints201712.0175.v1 6. Amari A, Dobre OA, Venkatesan R, Kumar OS, Ciblat P, Jaouën Y (2017) A survey on fiber nonlinearity compensation for 400 Gb/s and beyond optical communication systems. IEEE Commun Surv Tutorials 19(4):3097–3113. https://doi.org/10.1109/COMST.2017.2719958 7. Nain A, Kumar S (2018) Performance investigation of different modulation schemes in RoF systems under the influence of self phase modulation. J Opt Commun 39(3):343–347. https:// doi.org/10.1515/joc-2016-0155 8. Nain A, Kumar S, Singla S (2017) Impact of XPM crosstalk on SCM-based RoF systems. J Opt Commun 38(3):319–324. https://doi.org/10.1515/joc-2016-0045 9. Bhatia R, Sharma AK, Saxena J (2016) Improved analysis of four wave mixing with sub-plank higher-order dispersion parameters in optical communication systems. Optik 127(20):9474– 9478. https://doi.org/10.1016/j.ijleo.2016.07.035 10. Kathpal N, Garg AK (2020) Analysis of radio over fiber system for mitigating four-wave mixing effect. Dig Commun Netw 6(1):115–122. https://doi.org/10.1016/j.dcan.2019.01.003 11. Pham TV, Nguyen TV, Nguyen NT, Pham TA, Pham HT, Dang NT (2019) Performance analysis of hybrid fiber/FSO Backhaul downlink over WDM-PON impaired by four-wave mixing. J Opt Commun 41(1):91–98. https://doi.org/10.1515/joc-2017-0127

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12. Garg AK, Janyani V (2018) Resilient, bandwidth scalable and energy efficient hybrid PON architecture. Telecommun Syst 67(4):687–698. https://doi.org/10.1007/s11235-017-0369-1 13. Garg AK, Janyani V (2017) Adaptive bandwidth mechanism using dual rate OLT for energy efficient WDM–TDM passive optical network. Telecommun Syst 66(4):657–670. https://doi. org/10.1007/s11235-017-0316-1 14. Kaur G, Singh ML, Patterh MS (2010) Effect of fibre nonlinearities in a WDM transmission system. Optik 121(10):889–896. https://doi.org/10.1016/j.ijleo.2008.09.035 15. Nain A, Kumar S, Singla S (2017) Performance estimation of WDM radio-over-fiber links under the influence of SRS induced crosstalk. Proceeding of International conference on intelligent communication, control and devices. Springer, Singapore, pp 279–284 16. Ruzbarsky J, Turan J, Ovsenik L (2016) Stimulated brillouin scattering in DWDM all optical communication systems. In: 2016 26th international conference Radioelektronika (RADIOELEKTRONIKA). IEEE, pp 395–398. https://doi.org/10.1109/RADIOELEK.2016. 7477354 17. Ahlawat D, Arora P, Kumar S (2019) Performance evaluation of proposed WDM optical link using EDFA and FBG combination. J Opt Commun 40(2):101–107. https://doi.org/10.1515/ joc-2018-0044

An Overarching Review on Taxonomy of Routing Metric in Concurrence with Trust and Security for CRAHN Pooja Ahuja, Preeti Sethi, and Naresh Chauhan

Abstract Cognitive radio technology is introduced to solve the problem of spectrum paucity by providing efficient, dynamic and opportunistic use of frequency bands. Wireless devices with cognitive capability refer as cognitive radio users and when these devices topologically arranged in ad hoc manner without any centralized entity termed as cognitive radio ad hoc network (CRAHN). Ironically, however CRAHN deals with major issues like network topology, frequency band change, and unpredictable PU behavior and spectrum variation temporally and spatially. In this paper, we present survey of recent routing protocols in CRAHN. Initially, routing challenges in CRAHN is addressed. Thereafter, we present the different routing metric which is classified into seven categories like end-to-end Delay, hop count, PDR, reliability, throughput, energy efficient and location centric. After surveying above routing protocols, we have seen that reactive routing protocol is more popular among all. This survey also focuses on routing protocols that have considered two key issues for improving Path trustworthiness and data security while communication. It also shed light on the open issues for designing and implementing the routing protocols in CRAHN. Keywords Cognitive radio network · Trust · QOS · Data security · Routing metrics

1 Introduction Pervasiveness of wireless technology and mobile devices in this new era where availability of bandwidth is expected everywhere and it is much needed to have seamless flow of information. However, due to huge increase in usage of wireless technologies (e.g., personal area networks, body area network, sensors networks, etc.) which are operating on unlicensed bands, like ISM band caused congestion in this band. Initially, wireless applications are operable on Fixed Spectrum Access P. Ahuja (B) · P. Sethi · N. Chauhan Department of Computer Engineering, J.C. Bose YMCA University of Science and Technology, Faridabad, Haryana, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_58

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(FSA) regulations which further lead to spectrum paucity problem. This initiated the use of alternative policy known as Dynamic Spectrum Access that allow efficient use of spectrum. DSA [1] assigns spectrum to more than one user, users with higher priority in using spectrum know as Primary Users (PU) or licensed users and other users which access the spectrum only when the primary users are not using known as Secondary Users (SU). This leads to effective usage of spectrum. In this perspective, SU is enabled to sense the radio environment and these characteristics of user are known as cognitive radio or CR users. CR users are designed with cognitive capability like sensing the on /off behavior of PU. Cognitive radio network is intelligent wireless network which contain more than one CR users. Moreover, these types of network improve end-to-end performance by reconfiguring various communication parameters. It can be further classified as infrastructure-based and infrastructure less networks. In CRN, each CR user belongs to different network and posses different cognitive capability. Furthermore, dynamic use of spectrum band also affects overall network performance. Thus, building a new routing protocol which supports proper functionality of CRN is challenging task. Initially, the studies in cognitive radio networks mainly focuses on the physical layer for signal processing and MAC layer for spectrum management and network layer for routing. Nowadays, an effort is been done to design a cross-layer [2] and control framework. Figure 1 represents the functionalities of three basic layers known to be Physical layer, MAC layer and Network Layer. However, there are a number of works which propose routing protocol for CRN. It has been revealed that due to distinct nature of CRN many researchers explore variety of methods that are well suited for CRN by adding some features to existing ones and creating the newer version. In this paper, various routing protocols are classified and critically reviewed on the basis of routing metric and type of underlying routing protocol adapted for communication like reactive, proactive and hybrid. In addition to this, some open issues also addressed like trusted route selection and data security for communication in CRN. Here an effort is made to understand the challenging nature of CRN and comparative study is performed and encountered various issues in order to design a complete and feasible routing protocol which in cooperates all the above-mentioned factors. In the following section, routing in cognitive radio ad hoc networks (CRAHN) is elaborated and its subsection performs the review of various existing routing protocols with regard to routing metric and compared qualitatively. Followed by extensive survey on trust-based and secured routing protocols in CRN. The next section describes future and open challenges in CRN for routing. Finally, this paper is concluded in last section.

2 Routing in Cognitive Radio Ad hoc Networks CRAHN belongs to infrastructure less class of cognitive radio network which implies there is no centralized entity. Due to which each CR user have to cooperate between

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Fig. 1 Functionalities of different layer in CRN

them in an ad hoc manner in order to exchange information like PU presence and network topology. The main concern of cognitive radio communication is to refrain SU to interrupt PU transmission and ensures that route should consider PU avoidance and end-to-end communication. Routing in CRAHN [3] means it should consider the characteristics of both cognitive radio network and as well as ad hoc networks. These characteristics are PU avoidance which is must for CRN. In addition to that node mobility, control channel and unidirectional link are characteristics of ad hoc network. The major requirement for designing routing protocol for CRAHN is SU should be enabled to sense the available spectrum immediately in real-time manner. Furthermore, adapt to self-configuration nature as no centralized body is present and all nodes exchange information between them via distributed mode or through common control channel. Another factor which needs consideration is set of available spectrum bands varies from node to node. Thus, raising the issue of using common control channel. Besides, above-mentioned issues, nowadays researchers are also exploring

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that how efficient and accurate route can be selected between source and destination end. In this context, trust mechanisms are performed on each node to detect the malicious or suspicious behavior. Likewise, for secure data communication validity of each CR node is obtained verifying security credential issued by authorized nodes. The following section presents the comprehensive review of various routing protocols in CRAHN in terms of routing metric, trust and secured routing schemes for communication in CRAHN. Figure 2 depicts the categorization of routing protocols.

Fig. 2 Categorization of routing protocols in CRAHN

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2.1 Comprehensive Review on Routing Protocol Based on Routing Metric Classification This section presents the review done on various routing protocols in CRAHN based on routing metric classification in terms of basic routing mechanism used, i.e., reactive, proactive or hybrid, and we also categorize the reviewed routing protocols into seven categories: end-to-end delay, hop count, reliability, PDR, throughput, energy efficiency and location-based. Routing metric helps the routing protocol to make better routing choice. Therefore, routing metric also indicates the quality of chosen routing path.

2.1.1

End-to-End Delay

End-to-end delay is defined as the delay in the overall time taken by packet to reach from source to destination. End-to-end delay is the summation of propagating delay and processing delay. Here processing delay incorporates route discovery latency and queuing delay. Routes with minimum end-to-end delay are considered to be better routes. Some routing protocols reviewed here considered end-to-end delay as prime routing metric factor in selecting best route. Minimum channel switch routing (MCSR) Meghanathan et al. [4], designed routing protocol which find the routing path that have minimum no of channel switches. As far as channel switches are concerned it takes considerable amount of time and energy. At the same time, it lowers the end-to-end delay which reduces the interference with neighboring nodes and incurs fewer path transitions. Delay motivated on-demand routing (DORP) Cheng et al. [5], presents delaybased approach that combines many delay metrics to efficiently select the minimum end-to-end delay route. They are used to evaluate the commutative delay of the path. DORP inherits the basic procedure of AODV for route formation. Improved Ant Routing Song et al. [6], proposed routing protocol which applies Swarm intelligence into CRN routing to find out the best route which has minimum delay from source to destination. STOP-RP Zhu et al. [7], comes with the framework of route metric which considers both CR user’s Quality of Service (QOS) requirement and statistical PU activities. It adapts tree-based proactive routing and on-demand recovery with spectrum adaptive route recovery method for resuming communication. Moreover, this method shows that the average end-to-end delay decreases as the no. of gateway nodes increases.

2.1.2

Throughput

Throughput is defined as the average rate of successful packets delivery per second. Here, are some routing protocols that have target of achieving maximum throughput.

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Spectrum Aware Routing protocol (SPEAR) Sampath et al. [8], proposes a multihop distributed channel assignment and routing algorithm. It supports high throughput packet transmission. The main idea behind SPEAR is to integrate the end-to-end optimization of flow-based approaches with the flexibility of link-based approach to address spectrum heterogeneity. Spectrum aware mesh routing in CRN (SAMER) Pefkianakis et al. [9], proposes the routing solution for cognitive mesh network and opportunistically route traffic where spectrum availability and quality both meet and it shows better results in terms of achieving higher end-to-end performance by reducing delays. It also makes long term stability by not deviating from shortest hop count path. Therefore, maximizing the aggregate throughput.

2.1.3

Energy Consumption

As far as mobile devices are concerned all are battery operated so energy consumption is an important metric. Above stated concept applies to CRN as CR user has to continuously monitor the spectrum for data transmission. Therefore, it is considered to be critical factor for routing in cognitive radio network. In this context, we reviewed various routing protocols below whose prime concern is energy efficiency in cognitive radio network. Delay and Energy-based Spectrum Aware Routing protocol (DESAR) Rehman et al. [10], it presented the combination of delay and energy aware metrics for the selection of the efficient path. DESAR is reactive protocol that is based on the AODV protocol. Firstly, it represents the delay metric as the combination between switching delay, back off delay and queuing delay. Secondly, the energy consumed by a node at time t depends upon the no. of packets transmitted and received by the nodes. This protocol works well as far as performance is concerned. Energy Aware routing protocol for CRAHN Zhang et al. [11], makes use of the relative energy metric to select the optimal route thereby minimizing the average total power consumption and prolonging network lifetime. This type of protocol is designed to find an energy efficient route for the session using selected set of nodes, channel transmitting power. It also proposes the metric called energy weight of the link. This concept of weight factor presents uneven energy consumption and make sure that all nodes more or less have same energy consumption patterns. This protocol contributes in the appropriate and justified selection of relay nodes, avoiding the nodes with less energy as compared to threshold value which in turns helps to avoid network partitioning. Low latency and energy-based routing (L2ER) Rehman et al.[12], proposes a multi-metric on-demand and reactive routing protocol for CRAHN. The protocol tends to avoid tradeoff situation between the route with shortest end-to-end delay and route with a high residual energy but significant end-to-end delay. L2ER chooses path with maximum energy and minimum delay.

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Location-Based

In current scenarios, all the wireless devices are GPS or network location enabled due to omnipresent nature of it. CRN gather location information from FCC Geolocation Database. With this in mind, here we explored some location centric routing protocols used in cognitive radio networks. Location-based routing protocol (LAUNCH) Habak et al. [13], it is designed to track location information to guide discovery process. LAUNCH makes use of greedy decision to select the next hop neighbor that satisfies condition like: (a) it is close to the destination. (b) It has the minimum expected delay. Furthermore, besides handling the PU activities when it is active, it also selects the long-lived route that is expected to be stable during communication. SEARCH Chowdhury et al. [14], it is designed for mobile multi-hop CRN based on greedy location-based routing protocol. This protocol makes routing and channel selection decision while avoiding the regions with PU activity. The focused objective of this proposed approach is evaluating when the coverage region of the PU should be circumvented, when changing the channel is preferred option. SEARCH operates in 2 basic modes greedy forwarding and choosing minimum no. of hops to the destination and PU avoidance. IP-based location-based spectrum aware routing protocol IPSAG Badoi et al. [15], it uses the IP notion of piggybacking control information as header in the data packets instead of sending them separately, to avoid the control overhead. The selected next hop is the closest neighbors to the destination that has at least one common available channel and satisfies certain SNR threshold.

2.1.5

Link Reliability

A routing solution that produces stable routes in CRN is highly desirable as it one of the main challenges of CRN due to the PU activities. Unstable routes will lead to frequently finding new re-routing events which consumes the network resources and degrades its performance. Link Reliability can be captured in routing mechanism or implicitly. Protocols like STOD-RP [7], SEARCH [14], Coolest Path [16].

2.1.6

Hop Counts

The number of hops a data packet has taken toward reaching its destination. Hop count is the basic measurement of distance in a network between source and destination. Hop count H signifies that source is H hops distant from destination. Hop count as the metric is identified by various routing protocol such as multipath routing and spectrum access SEARCH [14], SAMER [9] and CAODV [17]. Table 1 summarizes the different metric used for cognitive radio routing. In fact, all metric has evolved from traditional ones used in ad hoc networks such as delay, hop count, anenergy efficiency. Moreover, moving toward CRN specific metric like

No

Reactive

Reactive

Reactive

Proactive

Reactive

Reactive

Reactive

Reactive

Hybrid

SAMER [9]

CAODV [17]

MASAR [18]

DSAR [19]

MRSA [20]

DORP [5]

Ant routing [6]

IPSAG [15]

Ai-sorp [21]

Reactive

Reactive

Reactive

L2ER [12]

DESAR [10]

Energy aware routing protocol [11]

STOD-RP [7] Hybrid

Yes

Reactive

MCSR [4]

No

Yes

No

No

No

D2CARP [22] Reactive

Yes

Yes

No

Yes

No

No

Yes

No

Common control channel

Protocol name Reactive/proactive/hybrid

Yes

Yes

Yes

Yes

Yes

Yes

Yes

Yes

Yes

Yes

Yes

Yes

Yes

Yes

Yes

Spectrum awareness

Table 1 Routing metric used by routing protocols in CRAHN

√

–

√

√

– √

–

–

√

√

√

√

– √

End-to-end delay √

–

–

–

–

–

–

–

√

–

–

–

–

–

–

–

Hop count

Performance metrics

–

–

–

–

–

–

–

–

–

–

–

–

–

–

–

Reliability

–

–

–

–

–

– √

–

–

–

-

–

– √

–

–

√

– √

–

–

√

√

√ √

√

√

Throughput

– √

PDR

–

–

√

– √

–

–

–

–

–

–

–

–

–

–

Energy efficiency

–

–

–

–

–

–

–

√

–

–

-

√

√

(continued)

Location-based

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No

Reactive

Hybrid

Reactive

Reactive

Reactive

Reactive

Proactive

Reactive

LAUNCH [13]

CRM-IC [23]

SEARCH [14]

COOLEST [16]

WHAT [24]

Local coordination [22]

SAOR [25]

SPEAR [8]

No

Yes

No

Yes

Yes

No

No

Common control channel

Protocol name Reactive/proactive/hybrid

Table 1 (continued)

Yes

Yes

Yes

Yes

Yes

Yes

Yes

Yes

Spectrum awareness PDR

Throughput

Energy efficiency

√

√

√

√

–

√

√

–

–

–

–

–

– √

–

– √

–

–

√

–

–

√

– √

–

√

–

–

–

–

√

√

–

–

–

√

√

–

–

–

–

–

–

–

–

–

–

–

–

–

–

– √

√

Reliability

End-to-end delay √

Hop count

Location-based

Performance metrics

An Overarching Review on Taxonomy of Routing Metric … 549

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spectrum awareness, use of common control channel, route reliability and location centric are used. However, one factor which being considered is type of routing mechanism used, i.e., reactive, proactive or hybrid for CRN communication. In fact, different routing techniques combine more than one metric or use hybrid metric to achieve different goals or to choose the best path when more than one routes falls under same criteria, with respect to the above-mentioned comparative review in Table 1. It revealed that adaption of on-demand routing protocol is more than others routing mechanisms. Due to some reasons like energy conversation, route setup as when required and helps to increase the network lifetime. AODV is probably more suitable for cognitive radio network than others. In addition to stated complexities in CRN routing, we could not find out any routing protocol with specific QOS support. Furthermore, to run real-time applications QOS support is advantageous to secondary user. By defining the QOS, spectrum management becomes more efficient.

2.2 Extensive Review on Trust and Secured Routing Protocols in CRN This section explored various routing protocols in cognitive radio network that have been put forward the concept of trust evaluation and data security during communication. Many researchers believe that trust and security are useful for finding accurate and trusted route between source end and destination end. In fact, due to ad hoc and dynamic nature of CRN, it is confronted by various attack behavior that can be inside attacks or outside attacks. For example, Selective forwarding attack malicious nodes make themselves in the path of data transmission through exchanging false information. They may refuse or discard the transmission which impacts the overall network performance. Here, an effort has been made to resolve the inside attack raised by malicious user by deploying trust management framework in CRN. Furthermore for making the secure data communication or resolving outside attacks cryptographic approach can be implemented for authenticating the source and destination. A trust-based technique for secure spectrum access in CRN Kar et al. [26], it proposes the trust-based mechanism to identify malicious and comprised nodes which have been authenticated to ensure secure communication. It also detects the malicious nodes that violate opportunistic access policy. The measures used by this technique are to evaluate the history of earned trust during previous transmission and use this to judge current and future allocations. Moreover, it provides incentive for fair usage and SU with highest priority gets access. This approach will able to detect and identify malicious nodes. A multi-factor trust management scheme for secure spectrum sensing in CRN Kar et al. [27], this approach focuses on specific security attack called spectrum data falsification attack. In this attack malicious internal member of the network reports false sensing results in sensing spectrum. This scheme proposes the sensing repudiation methods for identifying these attacks. Sensing repudiation involves multiple

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decisions like history, active factor, incentives and consistency factor which are able to detect malicious and suspicious users. It results in higher accuracy and false alarm rate. Security management-based trust determination in CRN Li et al. [28], it proposes novel penalty mechanism based on cognitive trust value. It uses six functions: authentication, interactive, configuration, trust value collection, storage, update and punishment. It uses the data fusion center and cluster head placed in the hierarchal architecture to measure the trust value of CR users. In addition to this misbehaving node is punished by fusion center by decreasing their trust value and thus able to detect malicious and suspicious node. An omnipresent formal trust model (FTM) for Pervasive Computing environment Haque et al. [29], it presented a recommendation protocol that provides a multi-hop recommendation capability and flexible behavioral model to handle communication. It has made an attempt to simulate human trust, its augmented model with both time and distance aging of trust value. Zhang et al. [30], designed trust-based secure routing model specific to selective forwarding attack. It helps to monitor forwarding behaviors, trust of nodes are constructed to identify malicious nodes. In its routing decision phase nodes trust are used to construct available path trust and do delay measurements that helps to select best route. It gives sticker punishment to malicious nodes. This model shows good performance in terms of throughput and end-to-end delay. Rajaraman et al. [31], focused on calculating the indirect trust mechanism, in which each node monitors the forwarding behavior of its neighbors in order to detect whether any node behaves selfishly and does not forward the packets it receives. This approach uses link state routing protocol based on indirect trust which forms the shortest route and finds the most trustworthy route among all. Khasweh et al. [32], proposes the 2-level authentication scheme for communication in CRN. Before joining the network, a CR node validate by obtaining the security credentials from authorized source. It proposed scheme relies on public and symmetric key cryptography. It encrypts the data between the communicating entities in order to improve network security in terms of resource availability and accessibility. It mitigates DOS, man in the middle and reflection attacks. Its performance shows less computation and communication requirements. Alahmadi et al. [33], it proposes the AES-assisted DTV scheme, in which an AES encrypted reference signal is generated at TV transmitter and used the sync bits of the DTV data frame. This enabled the sharing of secret key between transmitter and receiver. In addition, when combined with the analysis on the auto connection of the received signals, the presence of the malicious node is detected accurately whether PU is present or not. This scheme is used to detect primary user emulation attack. Therefore, AES-assisted DTV scheme, the primary user as well as malicious users can be easily detected with high accuracy under primary use emulation attacks. It has been revealed from above-mentioned Table 2, in which qualitative comparison of different trust and security-based routing schemes explored on basis of inside and outside attacks. In this context, application of trust mechanism in collaboration with data security improves path reliability, packet delivery ratio, route load

Trusted route selection

Yes

Yes

Yes

Yes

Yes

No

No

Protocols

Kar and Sethi [27]

Li [28]

Haque [29]

Zhang [30]

Rajaraman [31]

Khasawneh [32]

Alahmadi [33]

Yes

Yes

No

No

No

No

No

Data security

Yes

Yes

Yes

Yes

Yes

Yes

Yes

Malicious node

No

No

No

Yes

No

Yes

Yes

Selective forwarding attack

Inside and outside attacks

No

No

No

No

No

Yes

Yes

Spectrum data falsification

Table 2 Qualitative comparison of trust and secured routing protocols used in CRN

Yes

No

No

No

No

No

No

Primary user emulfication

No

Yes

No

No

No

No

No

Modification attack

No

Yes

No

No

No

No

No

Man in the middle

No

Yes

No

No

No

No

No

Reflection attack

–

Visual C++

MATLAB

MATLAB

Omnet++

MATLAB

MATLAB

Tools used

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balancing and makes secure data communication. However, it has been seen that there is very less work done on making the framework which helps to choose both trusted route and also ensures secured data transmission.

3 Future and Open Issues From the above-reviewed protocols, it is clearly visible that significant efforts have been made to design efficient, effective and secured routing protocol for CRN. However, still there exists some loophole and limitation that need considerable amount of contribution.

3.1 QOS Specific Requirements QOS means the ability to provide differentiated service to different users or applications as per their demanded requirements. QOS is suitable for real-time and elastic application since delay and capacity plays vital role in it. Moreover, with QOS support, the routing protocol assigns the path and spectrum band purely on its routing metric. This led to efficient use of available resources.

3.2 Path Reliability Path reliability or link stability refers as a generalized concept taken into consideration, it means that to explore many routes at the same time. Thus, introducing multiple paths is very important due to dynamic nature of CRN. Level of reliability that is required by the cognitive radio network shows the basic connectivity between the nodes and all routes should be disjoint and not close to each other. Moreover, selecting non-close routes makes them less vulnerable to PUs’ activities at the same time.

3.3 Trusted Route with Data Security Most of the studies tended to imbibe trust and security as prime factors while designing routing protocol for CRN. Moreover, after reviewing various protocols, we have found that a mass of studies focuses on the attacker detection but very few are studied how to address those attackers while they were detected some of them used penalty and incentive methods to deal with security problem in CRN. It has been observed that CR nodes are more prone to security attacks and threat. Another

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important area of concern is providing secure communication between source end and destination end. Some studies are available that are using authentication methods like symmetric key cryptography and other approaches. Moreover, there is no significant work done on using both like considering trusted route with data security as well. However, still there is need of routing protocol that integrate spectrum awareness with specific QOS requirement and also applies data security with trusted route formations.

4 Conclusion Much work on the routing protocols in CRAHN has been carried out. Yet, there are still some critical issues that need to be considered like routing protocol should consider specific QOS requirements from SU. In addition to this, routing protocol should also ensure data security with trusted route selection. Moreover, as all wireless devices are battery operable so energy preservation should also be taken care of. The above observation indicates the need for deriving a routing protocol that integrates mentioned requirements.

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11. Zhang Y, Song F, Deng Z, Li C (2013, Mar) An energy-aware routing for cognitive radio ad hoc networks. In: 2013 IEEE third international conference on information science and technology (ICIST). IEEE, pp 1397–1401 12. Rehman RA, Kim BS (2014) L2ER: low-latency and energy-based routing protocol for cognitive radio ad hoc networks. Int J Distrib Sensor Netw 10(9):963202 13. Habak K, Abdelatif M, Hagrass H, Rizc K, Youssef M (2013, Jan) A location-aided routing protocol for cognitive radio networks. In: 2013 international conference on computing, networking and communications (ICNC). IEEE, pp 729–733 14. Chowdhury KR, Felice MD (2009) Search: a routing protocol for mobile cognitive radio ad-hoc networks. Comput Commun 32(18):1983–1997 15. B˘adoi CI, Croitoru V, Prasad R (2010, June) IPSAG: an IP spectrum aware geographic routing algorithm proposal for multi-hop cognitive radio networks. In: 2010 8th international conference on communications. IEEE, pp 491–496 16. Huang X, Lu D, Li P, Fang Y (2011, June) Coolest path: spectrum mobility aware routing metrics in cognitive ad hoc networks. In: 2011 31st international conference on distributed computing systems. IEEE, pp 182–191 17. Cacciapuoti AS, Calcagno C, Caleffi M, Paura L (2010, Oct) CAODV: Routing in mobile ad-hoc cognitive radio networks. In: 2010 IFIP wireless days. IEEE, pp 1–5 18. Abedi O, Berangi R (2013) Mobility assisted spectrum aware routing protocol for cognitive radio ad hoc networks. J Zhejiang Univ Sci C 14(11):873–886 19. Abedi O, Berangi R (2014) Beaconless dynamic spectrum-aware routing protocol for cognitive radio ad hoc networks. Arab J Sci Eng 39(5):3941–3952 20. Wang X, Kwon TT, Choi Y (2009, Sept) A multipath routing and spectrum access (MRSA) framework for cognitive radio systems in multi-radio mesh networks. In: Proceedings of the 2009 ACM workshop on Cognitive radio networks, pp 55–60 21. Gong L, Deng S, Tang W, Li S (2008, Oct) Anti-intermittence source routing protocol in distributed cognitive radio network. In: 2008 4th international conference on wireless communications, networking and mobile computing. IEEE, pp 1–6 22. Najafi B, Keshavarz-Haddad A, Jamshidi A (2014, Sept) A new spectrum path diversity routing protocol based on AODV for cognitive radio ad hoc networks. In: 7th international symposium on telecommunications (IST’2014). IEEE, pp 585–589 23. Zhong Z, Wei T (2010, Oct) Cognitive routing metric with improving capacity (CRM-IC) for heterogeneous ad hoc network. In: 2010 international conference on information, networking and automation (ICINA), vol 1. IEEE, pp V1-271 24. Chen J, Li H, Wu J (2010, May) WHAT: a novel routing metric for multi-hop cognitive wireless networks. In: The 19th annual wireless and optical communications conference (WOCC 2010). IEEE, pp 1–6 25. Rozner E, Seshadri J, Mehta Y, Qiu L (2009) SOAR: simple opportunistic adaptive routing protocol for wireless mesh networks. IEEE Trans Mob Comput 8(12):1622–1635 26. Kar S, Sethi S, Sahoo RK (2018) A trust-based technique for secure spectrum access in cognitive radio networks. In: Progress in computing, analytics and networking. Springer, Singapore, pp 9–18 27. Kar S, Sethi S, Sahoo RK (2017) A multi-factor trust management scheme for secure spectrum sensing in cognitive radio networks. Wireless Pers Commun 97(2):2523–2540 28. Li J, Feng Z, Wei Z, Feng Z, Zhang P (2014) Security management based on trust determination in cognitive radio networks. EURASIP J Adv Signal Process 2014(1):48 29. Haque MM, Ahamed SI (2007, July) An omnipresent formal trust model (FTM) for pervasive computing environment. In: 31st annual international computer software and applications conference (COMPSAC 2007), vol 1. IEEE, pp 49–56 30. Zhang G, Chen Z, Tian L, Zhang D (2015) Using trust to establish a secure routing model in cognitive radio network. PloS One 10(9) 31. Rajaram S, Karuppiah AB, Kumar KV (2014) Secure routing path using trust values for wireless sensor networks. arXiv:1407.1972

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Multiband and Wideband MIMO Antenna for X and Ku Band Applications Sweta Singh, Brijesh Mishra, Karunesh Shrivastva, Aditya Kumar Singh, and Rajeev Singh

Abstract In this article, a two-port MIMO microstrip patch antenna (27 × 23 × 1.6 mm3 ) for X and Ku band applications is presented. Mushroom shape MIMO antenna structure shows multiband (8.1–8.8, 9.8–10.5, 12–12.8, 13.4–14, 14.8– 15.5 GHz) and wideband (13.5–15.8 GHz) characteristics from port-1 and port-2 respectively. Defected ground structure provides better isolation between the two antenna elements. Antenna performance in terms of return loss, gain, isolation and radiation pattern is investigated. Diversity gain of the proposed antenna is verified with the help of ECC (envelope correlation coefficient) plot. Isolation is also discussed and presented along with simulated results. Keywords Asymmetric MIMO · Multiband · Wideband · Diversity gain · Envelope correlation coefficient

1 Introduction The MIMO antenna plays a vital role in the field of modern wireless communication system due to its special features. A simple microstrip patch can be converted into multiple-input and multiple-output (MIMO) patch antenna [1, 2]. Conventional microstrip patch antennas exhibit narrow bandwidth and low gain performance. Many techniques are known to increase gain, return loss and bandwidth like using slots, parasitic patches, substrate integrated waveguide (SIW) structure and switches (e.g. PIN diode, varactor diode, RF MEMS switches, GaAs Switches) [3]. Abundance of literature is available for wireless communication bands like L, S and C therefore scientists have started to work on higher frequency bands like X and Ku bands [3, 4]. Advanced wireless communication system such as 4G technology demands fast and higher data rate for transmission and reception for which higher diversity S. Singh (B) · K. Shrivastva · A. K. Singh · R. Singh Department of Electronics and Communication Engineering, University of Allahabad, Prayagraj, India B. Mishra Department of Electronics and Communication Engineering, MMMUT, Gorakhpur, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_59

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gain in broadside direction is desirable. MIMO antenna system is capable to provide higher diversity gain with trade-off between reliability and data rate. Researchers are trying to design compact MIMO antenna to provide high isolation between ports. Small-sized MIMO antenna however degrades the performance of antenna elements. Several techniques have been reported [5–7] to reduce mutual coupling in MIMO antenna system such as diversity technique, neutralization line, meta-surface, and reflector. Additionally, defected ground structure and electromagnetic bandgap structures are very simple and powerful techniques to enhance the isolation between antenna elements as reported in [8–10]. A compact antenna with pattern diversity and band rejection characteristics is reported in [11–13] for UWB MIMO applications. A triple-band notch loaded compact MIMO antenna with defected ground structure and split ring resonator (SRR) is reported for wideband applications [14]. Owing to the work reported above, it can be fairly concluded that the symmetrical designs have been used to reduce mutual coupling in MIMO antenna. Asymmetrical U-shaped MIMO antenna is also reported in [15] for 4G LTE system with improved isolation. Multiband antennas are generally preferred over single band antenna as they can be utilized for the different RF frequencies and for different wireless communication systems. However, use of multiband MIMO antennas is very rare [16]. In this paper, we propose a design of asymmetric mushroom shaped 2 × 2 MIMO patch antenna over FR-4 as substrate for X and Ku band applications. Mushroom shaped MIMO antenna is designed and simulated by means of electromagnetic tool High Frequency Structure Simulator (HFSS) and shows multiband characteristics (8.1–8.8, 9.8–10.5, 12–12.8, 13.4–14, 14.8–15.5 GHz) from port-1 and wide band (13.5–15.8 GHz) characteristics from port-2. Evolution of antenna design result and discussion and conclusion of the proposed work is presented in foregoing sections.

2 Evolution of Antenna Design Figure 1a shows the systematic growth of the proposed mushroom shaped MIMO patch antenna. In step-1, conventional mushroom shape antenna is used as shown in Fig. 1a. But it fails to provide usable band in the frequency range 8–16 GHz as shown in Fig. 1c. Further, a small circular slot (1.5 mm) is introduced at the centre position of the mushroom shape structure as depicted in step 2. Figure 1c portrays the effect of step-2 and a band of 8.1–8.8 GHz is observed. In step 3, a rectangular slot (1 × 4 mm2 ) is etched in the design of step-2 which introduces the multiband characteristics. Return loss characteristics of the structure of step-3 show four different bands, i.e. 8.1–8.8, 10–10.5, 12–12.6 and 14.8–15.5 GHz. Finally, an additional rectangular slot (27 × 2 mm2 ) on the ground plane is etched (cf. Fig. 1c) to provide the isolation between the ports as shown in structure of step 4. After modification in design of step-3 a multiband with five bands viz. 8.1–8.8, 9.8–10.5, 12–12.8, 13.4–14 and 14.8–15.5 GHz and a wideband of 13.5–15.8 GHz from port-1 and port-2 are observed respectively in Fig. 1c. Figure 1b shows top and

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Step 1

Step 2

Step 3

Step 4

(a)

(b)

(c)

Fig. 1 a Evolution of the proposed mushroom shape antenna. b Top and bottom view of the proposed antenna. c Simulated reflection coefficient versus frequency plot for four antenna configurations of (a)

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Table 1 Dimensions of the proposed antenna Parameters

Lp1

Lp2

Lp3

Wp1

Wp2

Wp3

S1

S2

Value (mm)

11

5

22

9

4

10

1.5r

1×4

bottom view of optimized (proposed) mushroom shape MIMO antenna. The physical dimensions (mm) of the proposed antenna are enumerated in Table 1.

3 Results and Discussion The performance of the proposed antenna is studied in terms of surface current distribution, return loss, diversity gain, envelope correlation coefficient (ECC) and radiation pattern. Distribution of current in the proposed MIMO antenna is observed with the surface current distribution plot. With port 1, we observe the current distribution at five different resonant frequencies, i.e. 8.4, 10.2, 12.4, 13.8 and 15.2 GHz as shown in Fig. 2a–e. The maximum surface current densities at 8.4, 10.2, 12.4, 13.8 and 15.2 GHz are 66.6, 64, 68.87, 60.18 and 81.99 A/m respectively. Similarly surface current distribution at resonating frequency 14GHz corresponding to port 2 is 81.99 A/m (cf. Fig. 2f). From Fig. 2a–f. Surface current distribution at patch and ground is observed simultaneously it is clear that strong current is concentrated at port 1 for all resonating frequencies and it flows towards the edges of the patch and ground which makes antenna highly radiated entity. This asymmetric design can be used for multiband (8–16 GHz) and wideband (13.5–15.8 GHz) applications with two ports as suggested. Figure 3 shows the combined plot of |S11|, |S22| and gain with variation of frequency. Table 2, illustrates the performance of port 1 and 2. Five bands having impedance bandwidths of 8.3, 6.8, 4.9, 4.4 and 4.6% with peak gain 1.5, 2.1, 3.2, 3.5 and 4.5 dB respectively is observed at port-1. A wideband with an impedance bandwidth of 15.7% and peak gain 3.4 dB is observed at port 2. Figure 4a shows the graph of ECC. The proposed design consists of defected ground structure which enhances the isolation between two ports of the antenna. ECC and isolation the commonly used parameters to explain the efficiency of MIMO microstrip patch antenna. Relation between diversity gain and ECC can be understood with Eqs. (1) and (2) as given below. From the equations, it is fairly concluded that the lower the ECC value, higher the diversity gain and higher data transfer rate. The relation between diversity gain and ECC are given in Eqs. (1)–(2) as reported in [17].  Diversity Gain = 10 1 − ECC2

(1)

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Fig. 2 Surface current distribution at port 1 for a 8.4 GHz, b 10.2 GHz, c 12.4 GHz, d 13.7 GHz, e 15.2 GHz, f at port 2 at 14.0 GHz

ECC = 

|S ∗ S + S ∗ S |  2 11  212 21  2 22  2   1 − S  − S  1 −  S  −  S  11

21

22

(2)

12

efficiently and when ECC value is close to zero, the efficiency of MIMO antenna increased as the mutual coupling and signal interruptions between antenna elements are minimized.

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Fig. 3 Simulated S11, S22 and gain versus frequency plot for the proposed mushroom shaped MIMO antenna

Table 2 Port characteristics of the proposed MIMO antenna Bands Band frequency Resonant |S11| (dB) Impedance BW Gain (dBi) (GHz) frequency (GHz) (in %) 8.1–8.8

8.4

− 18.7

8.3

1.5

2

9.8–10.5

10.2

− 17.4

6.8

2.1

3

12–12.8

12.4

− 20.0

4.9

3.2

4

13.4–14

13.8

− 17.4

4.4

3.5

Port 1 1

5

14.8–15.5

15.2

− 16.8

4.6

4.5

Port 2 1

13.5–15.8

14.0

− 23.6

15.7

3.4

Envelope correlation coefficient (ECC) for the proposed MIMO mushroom shaped antenna shows a maximum value of 0.07, which is near to zero which is indicative of minimum mutual coupling effect between antenna elements leading to higher data rates and diversity gain for the proposed antenna. A comparative overview of two-port mushroom shape MIMO antenna in terms of antenna size, number of ports, number of bands, ECC, peak gain for the common X and Ku band is presented in Table 3. The Proposed antenna is compact (27 × 23 mm2 ) as compared to antennas [14, 16, 18] reported in Table 3. A better isolation loss (≥ − 20 dB) is observed as compared to antenna [14] (≥ − 15 dB) and [16] (≥ − 12.5 dB) and same as antenna [16] (≥ − 20 dB).

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Fig. 4 a ECC, b isolation and diversity gain

ECC is shown in Fig. 4a and isolation and diversity gain is shown in Fig. 4b. The effect of mutual coupling between any of the two ports of MIMO can be understood with the two scattering parameters S12 or S21. The scattering parameter S12/S21 provides the isolation between the ports for the two-port MIMO antenna system [6]. The isolation value is less than approximate − 20 dB for the entire desired frequency spectrum. Diversity Gain response for the proposed MIMO antenna ranges from

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Table 3 A comparative overview of proposed antenna References

Antenna size (mm2 )

Ports

No. of bands

[14]

36 × 22

2

[16]

40 × 40

2

ECC

Gain (dB)

Isolation (dB)

Operating bands

Triple band 0.005

5

≥ − 15

C, X, Ku

Multiband

Not reported

1–25

≥ − 12.5 C, X, K, Ku

[18]

27.4 × 27.4 3

Multiband

≈0

5.5–11

≥ − 20

Ku

Prop Ant.

27 × 23

Port 1

0.07

6

≥ − 20

X, Ku

2

Multiband Port 2 Wideband

9.979 to 9.999 dB. From Fig. 4a, b, it can be understood that ECC, isolation and diversity gain values lie in their acceptable limits. The simulated E-plane and H-plane for the far-field radiation pattern (dB) at all resonating frequencies are presented in Fig. 5a–f. Figure 5a–e shows radiation pattern at the resonant frequencies for port 1, i.e. for 8.4, 10.2, 12.4, 13.8 and 15.2 GHz respectively and Fig. 5f shows radiation pattern of resonant frequency for port 2, i.e. 14 GHz. Figure 5 illustrates the orientation of simulated antenna which lies in xz plane at all resonating frequencies. Phi component is dominant in E-plane (φ = 0) as well as H-plane (φ = 90).it is observed that due to identical orientation of antenna elements radiation pattern is almost same for all resonating frequencies and it is ideally suitable for omnidirectional radiation pattern

4 Conclusions The antenna design discussed above reveals the novelty in terms of structure, antenna performance and applications. The MIMO antenna has two ports at different location with different resonating frequencies. With the single antenna, multiband and wideband characteristics have been achieved. Isolation and ECC values between two ports are in acceptable range for the use of different ports without any interference.

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Fig. 5 Radiation pattern in E-plane and H-plane of port 1 at a 8.4 GHz, b 10.2 GHz, c 12.4 GHz, d 13.8 GHz, e 15.2 GHz and of port 2 at, f 14 GHz

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References 1. Balanis CA (1992) Antenna theory. A review. Proc IEEE 80(1):7–23 2. Kumar G, Ray KP (2003) Broadband microstrip antennas. Artech House 3. Singh V, Mishra B, Singh R (2019) Dual wide band semi circular shape microstrip patch antenna for Ku/K band applications. Microw Opt Technol Lett 61:1–7 4. Singh V, Mishra B, Singh R. A compact and wide band microstrip patch antenna for X-band applications. In: Second international conference in computing and communication engineering 5. Malviya L, Panigrahi R, Kartikeyan M (2017) MIMO antennas with diversity and mutual coupling reduction techniques, a review. Int J Microw Wirel Technol 9:1763–1780 6. Alibakhshikenari M, Virdee BS, See CH, Abd-Alhameed RA, Falcone F, Limiti E (2019) Surface wave reduction in antenna arrays using metasurface inclusion for MIMO and SAR systems. Radio Sci 54:1067–1075 7. Roshna TK, Unni D, Sajitha VR, Vasudevan K, Mohanan P (2015) A compact UWB MIMO antenna with reflector to enhance isolation. IEEE Trans Antennas Propag 63:1873–1877 8. Tang X, Yao Z, Li Y, Zong W, Liu G, Shan F (2020) A high performance UWB MIMO antenna with defected ground structure and U-shape branches. Int J RF Microw Comput Aided Eng e22270:1–14 9. Khandelwal M, Binod K, Santanu D Bandwidth Enhancement and Cross-polarization suppression in ultra-wideband microstrip antenna with defected ground plane. Microw Opt Technol Lett 56:2141–2145 10. Alnaiemy Y, Elwi TA, Nagy L (2019) Mutual coupling reduction in patch antenna array based on EBG structure for MIMO applications. Periodica Polytechnica Electr Eng Comput Sci:1–11 11. Ibrahim A, Machac J, Shubair R (2017) Compact UWB MIMO antenna with pattern diversity and band rejection characteristics. Microw Opt Technol Lett 59:1460–1464 12. Srivastava G, Mohan A (2016) Compact MIMO slot antenna for UWB applications. IEEE Antennas Wirel Propag Lett 15:1057–1060 13. Tang X, Yao Z, Li Y, Zong W, Liu G, Shan F (2020) A high performance UWB MIMO antenna with defected ground structure and U-shape branches. Int J RF Microw Comput Aided Eng 14. Patchala K, Rao YR, Prasad AM (2020) Triple band notch compact MIMO antenna with defected ground structure and split ring resonator for wideband applications. Heliyon 6(1) 15. Chen I-F, Peng C-M, Bao J-J, Chen C-C. 2 × 2 MIMO asymmetric u-shaped nonuniform slot antenna for the 4G LTE metal-housing mobile handsets. Int J Antennas Propag:1–9 16. Srivastava K, Kumar A (2016) MIMO based multi band antenna for wireless communication in C-Band, X-Band, K-Band and Ku Band, WiSATS. Lecture notes of the institute for computer sciences, social informatics and telecommunications engineering, vol 186 17. Garg P, Jain P (2020) Isolation improvement of MIMO antenna using a novel flower shaped metamaterial absorber at 5.5 GHz WiMAX band. IEEE Trans Circ Syst II Exp Briefs 67(4):675– 679 18. Satam V, Nema S (2018) Compact, high gain, linearly polarized diversity antenna for Ku band applications. Microw Optical Technol Letter 60:2843–2849

Channel Capacity of Underwater Channel Using OCDMA System Ajay Yadav, Ashok Kumar, Jitendra Kumar Deegwal, Ghanshyam Singh, and Arjun Kumar

Abstract In this research work, channel capacity of two-dimensional (2D) optical code division multiple access (OCDMA) system is evaluated in the underwater wireless optical communication channel. The various water categories which have been investigated for capacity evaluation are pure sea, clear ocean, and coastal water. Also, lognormal probability density function is considered for studying the impact of weak turbulence in the underwater channel. For unity channel capacity, the transmitted power should be greater than 15 dBm for all the water types. The channel capacity increases with rise in transmitted power and decreases with increment in number of users. The channel capacity is lowest in coastal water and highest in pure sea. In clear ocean, channel capacity is intermediate between coastal water and pure sea. Keywords UWOC · 2D · OCDMA · OCFHC/QCC · HL

1 Introduction For high data rate applications in underwater channel, acoustic communication is not enough which has been used for several decades. In comparison with acoustic communication, underwater wireless optical communication (UWOC) system consists of more bandwidth, enhanced security and less latency [1]. Optical code division multiple access (OCDMA) is a suitable choice for multi-user optical communication system since it provides secure communication. In OCDMA-based A. Yadav (B) · A. Kumar Department of Electronics and Communication Engineering, Bennett University, Greater Noida, Uttar Pradesh 201310, India A. Kumar · J. K. Deegwal Department of Electronics and Communication Engineering, Government Mahila Engineering College, Ajmer, Rajasthan 305002, India G. Singh Department of Electronics and Communication Engineering, MNIT, Jaipur, Rajasthan 302017, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_60

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UWOC system, each user is allocated an optical code sequence called codeword. An optical codeword is a sequence of “1” and “0”. The optical codeword with dimensions of time and wavelength is a part of two-dimensional (2D) code family. From the numerous 2D code, one-coincidence frequency hopping code/quadratic congruence code (OCFHC/QCC) is considered for the channel capacity calculation of UWOC OCDMA system [2]. No research paper had earlier evaluated the channel capacity of 2D OCFHC/QCC-based OCDMA system in the underwater channel. The underwater channel is influenced by attenuation (due to absorption and scattering) and turbulence which restricts the UWOC systems for short distance communication. Absorption and scattering deteriorate the optical signal in the form of attenuation. The irregular variations in refractive index of water lead to turbulence in the underwater channel [3]. Turbulence affects the optical signal by fading in the underwater wireless optical communication channel. Further, the performance of UWOC OCDMA system is affected by beam divergence due to decrease in power density at the receiver. Beer-Lambert’s law is generally used for studying the effect of attenuation on the channel path loss. The various water categories which have been studied are pure sea, clear ocean, and coastal water. The water which is 3.5% saline is known as pure sea irrespective of its source (sea or ocean). The density of pure sea water is more than fresh water and water which is not contaminated (also known as pure water). The water in the Tongue of the Ocean (TOTO), Bahamas islands falls in the other water category which is clear ocean. TOTO is a wide oceanic gorge which divides the isle of Andros and New Providence. The last category is coastal water like in San Pedro Channel, California, US. The coastal water is the water adjacent to seashore. In addition, lognormal probability density function (pdf) describes the impact of weak turbulence in the UWOC channel [4]. Multiple user interference (MUI) also influences the various parameters of OCDMA system like probability of error, channel capacity, and throughput. To reduce the influence of MUI on channel capacity, two hard limiters (HLs) are used. HL reduces the impact of MUI by eliminating interference pulses when information is retrieved at the receiver. The various sections of the paper are as follows: Sect. 2 details about the UWOC OCDMA system where transmitter, channel, and receiver are described. The impact of beam divergence, attenuation, turbulence, and various water categories on channel capacity is studied in Sect. 3. The mathematical analysis results are explained in Sect. 4.

2 OCDMA System The message from the source is in the form of information bits. On–off keying (OOK) is used for modulation of these information bits as shown in Fig. 1. The optical correlator assigns optical codeword to the modulated bits. Wavelength division multiplexer, fiber-optic delay lines, and wavelength division demultiplexer are part of optical correlator. N × 1 coupler is used for combining the output from the correlator 1 to correlator N. The combined signal is transmitted using the optical lens (optical antenna). The various types of water attenuate the transmitted optical

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Wireless Optical Channel Optical lens Source 1

Optical lens

Encoder 1 N×1 Coupler

Source N

Encoder N

Threshold detector 1

Photodetector 1

HL 1

HL 1

Decoder 1

1×N Coupler Threshold detector N

Photodetector N

Decoder N

HL N

HL N

Fig. 1 2D OCDMA system architecture

signal depending upon the attenuation coefficient. Due to variation in temperature and salinity levels of water, the optical signal strength varies with time and space, i.e., turbulence in the water affects the optical signal. At the receiver side, the optical lens (antenna) receives the optical signal which is further divided using 1 × N coupler. Double HLs are used at the receiving end which eliminate the MUI in the OCDMA system. In the decoder, the optical signal is correlated with code sequence of the intended user. This results in detecting the autocorrelation peak. The impact of MUI is decreased by first HL before the decoder, and it is decreased to larger extent by the second HL. The optical signal is converted to electrical signal by the photodetector. This is followed by the threshold detector which detects the presence of “1” and “0.”

3 Channel Modeling In underwater communication, the optical signal amplitude deteriorates (due to absorption and scattering), and power density decreases due to beam divergence. In the channel, the link loss factor β can be calculated using Beer-Lambert’s formula which is given by [2] β=

π



A ϕs L 2

2 e

−γ l

(1)

Here, A is the area of optical receive lens (antenna), φ s is the angle of divergence (in radian), l is the channel length (meters), and γ is the attenuation factor (m−1 ), respectively. The value of γ for various water categories is shown in Table 1 [8]. The underwater turbulence also affects the optical signal. In weak underwater turbulence, the received optical signal is characterized by lognormal pdf. The physical underwater surroundings are well simulated using lognormal pdf which is given as [2]

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f Y (y) = 

1 2π σY2 y

  exp −

ln y + σY2 /2

2 

2σY2

(2)

Here, Y is the random variable which is defined as f Y (y) = d/ dy(P(Y ≤ y)) where P(Y ≤ y) is the probability that Y ≤ y. In the weak turbulence regime, log-amplitude variance σ Y 2 = 0.16 [1]. The channel capacity is analyzed in presence of interfering users, scintillation (turbulence), and various water categories. Chip synchronous and slot asynchronous system is considered for the channel capacity evaluation. In the analysis, the number of photons which are falling on avalanche photo-diode (APD) is counted. In a codeword, the positions of the sequence where “1” is present is called mark, and for “0,” it is called space. So, a codeword consists of marks (for “1”) and spaces (for “0”). The number of marks can be represented by an interference vector K = (K 1 , K 2 , …, K w ). Also, the code weight (w) is equal to the number of marks. Due to usage of two hard limiters (HLs), MUI can be mitigated to a large extent. The intensity of the pulse above threshold Th is limited to wαs im (|K|). |K| represents the number of marks of the codeword. The algorithm for generating OCFHC/QCC is detailed in [5]. The total probability of error (PE ) can be calculated as given in [6–8]. This research work is extension of the work done in [8]. The channel capacity C of the 2D OCDMA system can be evaluated which is given by C = 1 − Hb (PE )

(3)

where H b (PE ) is binary entropy function which is defined as Hb (PE ) = −P E log2 PE − (1 − PE )log2 (1 − PE )

(4)

4 Results and Discussion The channel capacity C of 2D OCDMA system with change in transmitted power PT and number of users N is detailed in this section. The specifications which have been considered for the channel capacity evaluation are demonstrated in [8]. Pure sea, clear ocean, and coastal water are the various categories which have been studied in this research work. For a four-user OCDMA system with link length of 20 m and receiver aperture Dr = 5 cm, C increases with increase in transmitted power PT as shown in Fig. 2. The increment in quantity of photons with transmitted power increases C. When the transmitted power is 15 dBm, capacity equals unity irrespective of water type.

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571

0

10-1

10

-2

10

-3

Pure sea Clear ocean Coastal water

-5

0

5

10

15

Transmitted power, PT (dBm)

Fig. 2 Channel capacity versus transmitted power for various water categories

C decreases with increase in number of users irrespective of water categories as demonstrated in Fig. 3. The increment in number of users substantially increases MUI which decreases C. On comparison, C is very high for pure sea than the coastal water and clear ocean. The main reason for the use of 2D OCDMA-based underwater communication system is protected communication in the channel. Thus, the various performance degrading factors which have been considered in the analysis decreases the channel 10

0

10

-1

10

-2

Pure sea Clear ocean Coastal water

4

6

8

10

12

14

16

18

Number of users, N

Fig. 3 Channel capacity versus number of users for various water categories

20

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capacity of the underwater OCDMA-based communication system. However, this analysis shows that OCDMA-based underwater communication system can play an important role when secure communication is required amid submarines, battleships, and boats.

5 Conclusion The channel capacity of 2D OCDMA underwater communication system has been calculated in different water categories. Pure sea, clear ocean, and coastal water are the various water categories which are studied. When the number of users is four and transmitted power is greater than 15 dBm, channel capacity is unity irrespective of the water category since with increase in transmitted power the number of photons falling on the photodetector increases. This leads to unity channel capacity. Channel capacity of underwater OCDMA communication system is highest with pure sea as compared to clear ocean and coastal water. OCDMA-based underwater communication system can play an important role when secure communication is required amid submarines, ships, navy battleships, aircraft carriers, and boats.

References 1. Jamali VH, Salehi JA (2015) On the BER of multiple-input multiple-output underwater wireless optical communication systems. In: 4th international workshop on wireless optical communications, Istanbul, pp 26–30 2. Yadav A, Kar S, Jain VK (2017) Performance analysis of wireless OCDMA multi-user system based on new 2-D code in presence of atmospheric turbulence and various weather conditions. In: 9th international conference on communication systems and networks (COMSNETS), Bengaluru, pp 109–115. https://doi.org/10.1109/COMSNETS.2017.7945365 3. Elamassie M, Miramirkhani F, Uysal M (2019) Performance characterization of underwater visible light communications. IEEE Trans Comm 67(1):543–552. https://doi.org/10.1109/ TCOMM.2018.2867498 4. Jamali MV, Nabavi P, Salehi JA (2018) MIMO underwater visible light communications: comprehensive channel study, performance analysis, and multiple-symbol detection. IEEE Trans Veh Techn 67(9):8223–8237. https://doi.org/10.1109/TVT.2018.2840505 5. Yadav A, Kar S, Jain VK (2018) Performance analysis of a new OCFHC/QCC vis-a-vis synchronous PC/OOC code using photon count approach. IET Optoelectron 13(2):77–84. https:// doi.org/10.1049/iet-opt.2018.5090 6. Yadav A, Kar S, Jain VK (2018) Performance enhancement of double hard-limited 2D OCDMA system using aperture averaging and spatial diversity. IET Commun 13(5):583–593. https://doi. org/10.1049/iet-com.2018.5787 7. Yadav A, Kar S, Jain VK (2017) Performance of 1-D and 2-D OCDMA systems in presence of atmospheric turbulence and various weather conditions. IET Commun 11(9):1416–1422. https:// doi.org/10.1049/iet-com.2016.1008 8. Yadav A, Kumar A (2019) Performance analysis of under water 2D OCDMA system. In: International conference on optical and wireless technologies (OWT2019), Jaipur

Design and Performance Analysis of an Encrypted Two-Dimensional Coding Technique for Optical CDMA Urmila Bhanja

Abstract Recently, security plays an important role in wired and wireless optical communication. An eavesdropper or a jammer can intercept the data using sophisticated equipment. An eavesdropper can also tap the data. Hence, security in optical communication network plays a significant role and needs to be addressed. In this paper, security is enhanced by incorporating an encryption module to the existing 2D MDPHC encoder circuit referred in this work as integrated multi diagonal prime hop code (IMDPHC). The data in the OCDMA network are first encrypted and then encoded to prevent the attack by an eavesdropper or by a jammer. Different types of attacks are analyzed theoretically for the novel encryption circuit. In this work, the bit error rate (BER) performance of the 2D IMDPHC is also analyzed at different data rates. Keywords 2D W/T code · MDPHC · Multiple access interference (MAI) · Encryption · Eavesdropper · Bit error rate

1 Introduction The performance of optical code division multiple access (OCDMA) is limited by the presence of various noise sources. Multiple access interference (MAI) is one of the most dominating noise sources [1]. When the number of simultaneous active users increases, the system performance degrades with the increasing noise sources. For a better cardinality and improved bit error rate (BER) performance, various types of two-dimensional (2D) coding techniques are proposed by different authors [1– 6]. Recently, the authors also have also addressed 32-bit 2D OCDMA codes using a technique based on the folding of Golomb ruler, which shows the poor BER output for a small number of simultaneous active users [5]. In [6], the authors have proposed an enhanced double weight (EDW) code, which exhibits satisfactory BER results only for a smaller range of fiber distance. In this work, the BER is evaluated for the 2D MDPHC code as addressed by authors in [1] and compared with an existing 64-bit U. Bhanja (B) Department of Electronics and Communication Engineering, IGIT, Sarang, Odisha 759146, India © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_61

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2D code. The performance of the 2D MDPHC known in this work as integrated multi-diagonal prime hop code (imdphc) is evaluated through simulation for various data rates, users and received powers. Furthermore, the existing code is integrated with the encryption circuit to evaluate the BER performance. Currently, data security plays a significant factor with the growing applications of network usage in different fields. An optical code division multiple access (OCDMA) technology provides the greatest security contrast to the other techniques. Sometimes, the security provided by the OCDMA fails because of use of intelligent optical tapping devices in various parts of the networks [7]. To augment the security level of one-dimensional (1D) OCDMA code, two-dimensional (2D) and three-dimensional (3D) OCDMA codes exist in the literature. Few authors have suggested that the security can be enhanced by the use of the code word switching, intelligent coding techniques and by using different modulation formats such as differential phase shift keying (DPSK) OCDMA, coherent code shift keying optical code division multiple access (CSK-OCDMA) techniques, which do fail sometimes against different types of attacks [7]. Recently, many authors also have reported various alternative techniques that need all-optical processing such as optical steganography techniques [7], optical chaos-based communication [7], optical scrambling techniques [7] and optical encryption technique [8–11] to provide security. Fok et al. have addressed an all-optical encryption scheme with interleaved waveband switching modulation using four-wave mixing (FWM) technique to achieve Exclusive-OR (XOR) operation [8]. Natalie et al. have reported variable two code keying encryption and decryption techniques using nonlinear optical loop mirror-based Exclusive-OR (XOR) gates [9]. Recently, Wang et al and Chang et al. have addressed the security enhancement with OCDMA code swapping technique and multimode keying encryption technique based on electro-optic (E/O) switch XOR gate, respectively, [10, 11]. However, the encryption techniques are complex and all-optical in nature, which are addressed in the literature. Moreover, the authors have not taken the side effect of the nonlinear phenomena such as four-wave mixing (FWM), switch losses and delay in switches which affect the quality of the signal at the receiver and hence deteriorate the bit error rate (BER). Therefore, in this work, the 2D IMDPHC is combined with a novel encryption technique to enhance the confidentiality level of messages in an OCDMA network. In the proposed encryption and decryption circuits, various binary operations such as Exclusive-OR (XOR), binary combiner and binary splitter are implemented with the optisystem software version 14. The main contribution of the proposed technique is simple design compared to other existing encryption techniques [8–11]. To increase the security level, the encryption circuit is made complex with different binary operations. The binary operations are executed with eight random keys. In this work, the security performance of the proposed encryption circuit is described assuming a single active user, and an eavesdropper is assumed to be equipped with a sophisticated device. The significance of the proposed circuit is that the same circuit can be used for a larger data set. The rest of the paper is organized as follows: Sect. 2 presents briefly the existing 2D MDPHC code referred in this paper as IMDPHC. Section 3 describes the proposed security model. Sections 4

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and 5 depict the noise model and confidentiality analysis of the proposed system. Section 6 describes the performance analysis results, and the paper is concluded in Sect. 7.

2 The 2D MDPHC CODE The two-dimensional MDPHC code referred in this paper as IMDPHC is formed by integrating the multi-diagonal code (MD) and prime hop code (PC) [1]. The 2D MDPHC code with X wavelengths and a temporal code length of Y, W as the code weight, and λa , λc the auto and cross-correlation values, respectively, is represented as (X × Y, W, λa ,λC ) [1]. The 2D code is generated by integrating two one dimensional codes and is described by the authors in [1]. Modified PC code is obtained from the prime hop code (PC) as described by the authors in [1]. MD code is obtained by concatenating even and odd number of diagonal matrices [1]. A MDPHC code is obtained by crossing PC and MD code with identical users. Authors in [1] have described the code for 25 users for simplification purpose. Optical pulses are positioned as per the MD code sequence and expanded to create a code length of 125 bits for 25 users [1]. The cross-correlation for the MDPHC remains at zero for all the users, and hence, multiple access inference among several users is minimized [1].

3 Proposed Security Model and Description The transceiver circuit diagram is shown in Fig. 1. Random 128-bit binary data is applied to a BER test set and an encryption circuit. The function of the BER test set is to compare the original data with the received decrypted and decoded data to estimate the errors. Four different data sets each of 32 bits are encrypted using simple EXOR gates with eight different random keys. The encryption circuit is depicted in Fig. 2a. The binary combiner takes four inputs and gives 128-bits random data. The random data are converted to electrical and optical domain by using an NRZ pulse generator and Mach–Zehnder modulator, respectively. The data is encoded by the existing 2D MDPHC encoder prior to transmission. An eavesdropper or a malicious user at the end of fiber receives the encrypted and encoded information, which differs from the original 128-bit data. The decoder and decryption process do the reverse operation of the encoder and the encryption process to get back the original 128-bit data. BER test set at the receiver estimates the error by comparing the original and the received data. The photodiode (PD) detector converts the optical signal to the electrical signal, which is fed to a low-pass electrical Bessel filter to obtain the original signal. The filter output is fed to a recovery circuit to get the 128-bit binary original data. The data

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Fig.1 Block schematic of the transceiver circuit [1]

recovery circuit converts the electrical signal into binary original data. The splitter at the receiver after receiving the binary data divides it into four 32-bit data. Each datum of the 32 bits is made to pass through the decryption circuit and a binary combiner circuit, respectively, to obtain 128 bits of original data, which is fed to a BER test set. The binary splitter and combiner circuit are generated using a C++ programming module that is compatible with the optisystem software. In Fig. 2a, Data_in_1, Data_in_2, Data_in_3 and Data_in_4 represent the four randomly generated data, each of the 32 bits. E_Data_op_1, E_Data_op_2, E_Data_op_3 and E_Data_op_4 represent encrypted data, each of the 32 bits, generated from the encryption circuit shown in Fig. 2a. As shown in Fig. 2b, R_Data_IN_1, R_Data_IN_2, R_Data_IN_3 and R_Data_ IN_4, respectively, represent four encrypted data; D_Data_OP_1, D_Data_OP_2, D_Data_OP_3 and D_Data_OP_4 represent the four decrypted data outputs. As shown in Fig. 3, the 32-bit keys, Key 1 to Key 8, are generated randomly and are the inputs to the key generation circuit; the generated output keys of the key generation circuit, K1–K8, are used as keys for both the circuits.

4 Noise Model for the Proposed System In the proposed system, Gaussian approximation is used for the calculation of SNR. Signal to noise ratio (SNR) for the proposed system is expressed below in Eq. (1) considering various types of noise sources like thermal noise, phase induced intensity noise and photodetector shot noise [1, 3, 12].

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Fig. 2 a Encryption circuit. b Decryption circuit

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Fig. 3 Key generation circuit

Psig SNR = = PPIIN + Pshot + Pther

 Pr s 2 M B2 Pr2 M 2 vs2 (MN−1)2

+

eBPr Ms(MN−1)

+

4K B Tn B RL

(1)

All the parameters used in Eq. (1) are identical and have the same meaning as expressed by the authors in [1, 3, 12].

5 Confidentiality Analysis Many authors have addressed the confidentiality issue in OCDMA that exists in literature [7, 11, 13, 14]. In this paper, security is enhanced by integrating the 2D encoding technique with the encryption technique. Ciphertext only attacks (COA): A set of ciphertexts is obtained by an eavesdropper. In the proposed circuit, keys cannot be extracted as the circuit utilizes multilevel operations as explained briefly in Sect. III, unlike other existing approaches in the literature [7–11]. Known plain text attack (KPA): In the proposed encryption circuit, the keys are generated randomly and made to swap with other keys and hence provide robust security.

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Chosen plain text attacks (CPA): The multi-level operation of an encryption circuit makes difficult to obtain any set of keys with the knowledge of only the plain texts. Energy detection attack in OOK-OCDMA: In the proposed encrypted OCDMA technique, the average energy of the data for a certain period of time duration is estimated by a malicious user or by an eavesdropper at the transmitter end. The data obtained from the computed average energy by the eavesdropper are not same as the transmitted data. Security against Brute-force attack: Computational complexity of a Brute-force attack grows exponentially with increasing key size. In general, 128 bits key requires (2128 − 1) calculations in a processor. The proposed circuit can be extended for larger sets of data and keys, which provides computationally larger security against Brute-force attack [15].

6 Results and Discussion The proposed circuit is simulated using optisystem software ver.14.0. Figures 4, 5 and 6 are simulated without the encryption technique, and the BER performance of the 2D IMDPHC code is shown in Figs. 4, 5 and 6 for different numbers of simultaneous users, data rate and fiber range. The eye diagram for 22 users is shown in Fig. 4, which is simulated for a 10 Gbps data rate. Figure 5 exhibits the BER performance of the existing 2D IMDPHC code, which is found to be better for a 2.5 Gbps data

Fig. 4 Eye diagram for 22 users for a fixed data rate of 10 Gbps

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Fig. 5 Comparison of BER for different values of received power

rate compared to the existing 64-bit 2D code for various received powers [16]. It is observed that the 2D IMDPHC code achieves an acceptable level of BER even at a lower level of received power of − 21 dBm. Received power determines the transmission range covered by the fiber. In this experimental result, to validate the experiment, the software is run ten times, and the margin of error is found to be ± 0.0715935 for eight users and 0.028923348 for ten users, respectively, for a 2.5 Gbps data rate of 95% of the time. It is also seen from the experiment that the BER value degrades for an increase in the number of simultaneous active users irrespective of received powers. Figures 6 and 7 depict the simulation results for the integrated 2D IMDPHC encrypted circuit to measure the BER performance and confidentiality of the system. Figure 6a, b represents the output in time domain at the input end and at the fiber end, which is accessed by an eavesdropper, respectively, at a date rate of 1.25 Gbps. These figures are both identical with a delay of few nano-seconds only. As it is observed, Fig. 6b is the time-shifted and attenuated version of Fig. 6a. Hence, an eavesdropper can easily estimate the data with the help of a bandlimited photodetector and amplifier. However, in this work, the data obtained by an eavesdropper are the integrated encrypted-encoded data. Figure 7 exhibits the 3D plot of average BER, while fiber length varies from 1 to 50 km and with a variation of amplifier gain from 5 to 25 dB in steps of 5 dB each. However, the circuit does not give the desired BER at a fiber length beyond 50 km. Table 1 depicts the simulation parameters that are used for the simulation purpose.

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Fig.6 (a) Input data (encrypted version-time domain) at 1.25 Gbps. b Eavesdropper data at the fiber end at 1.25 Gbps

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Fig. 7 3D plot average BER, gain and fiber length for 1.25 Gbps data rate

Table 1 Simulation parameters

Receiver noise temperature

300 K

Receiver load resistor

1030 

Input laser power

0 dBm

Fiber length

10–50 km

Data rate

1.25–10 Gbps

Chromatic dispersion

17 ps/nm/km

Fiber attenuation

0.2 dB/km

PMD coefficient

0.05 ps/sqrt(km)

Amplifier gain

20 dB

Noise figure

4 dB

Carrier frequencies use

193.1–193.8 THz

7 Conclusion This paper presents the design and system performance of IMDPHC for an OCDMA system. The BER degrades with increasing number of active users because of multiple access interference. However, the BER performance of the 2D IMDPHC code is better compared to that of the 2D matrix code in terms of received power [16]. The BER performance of IMDPHC code yields better results than the existing 2D matrix techniques with a number of simultaneous active users [16]. Furthermore, a novel encryption technique with binary splitter, combiner and XOR operations is addressed, which is integrated with the 2D IMDPHC code that provides security for different types of attacks. Additionally, the circuit renders zero BER up to a fiber length of

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50 km while the gain variation is kept fixed between 5 and 25 dB. However, the circuit fails to render an acceptable value of BER when the fiber length exceeds 50 km. This experiment implies that the circuit can work in local area network (LAN), access network and metropolitan area network (MAN) environments while providing the required BER and with minimum MAI.

References 1. Bhanja U, Singhdeo S (2020) Novel encryption technique for security enhancement in optical code division multiple access. Photon Netw Commun 39(3):195–222 2. Salehi JA (1989) Code division multiple-access techniques in optical fiber networks. I. Fundamental principles. IEEE Trans Commun 37(8):824–833 3. Shivaleela ES, Selvarajan A, Srinivas T (2005) Two-dimensional optical orthogonal codes for fibre-optic CDMA networks. J Light Wave Technol 23(18):647–654 4. Hernandez VJ, Mendez AJ, Bennett CV, Lennon WJ (2005) ‘Simple robust receiver structure for gigabit ethernet O-CDMA using matrix codes. J Light Wave Technol 23(10):210–206 5. Jyoti V, Kaler RS (2011) Design and implementation of 2-dimensional wavelength/time codes for OCDMA. J Optik 122:851–857 6. Menon PS, Zahid AJG, Mandeep JS, Shaari S (2012) Realization of 2-D OCDMA network using EDW code. J Optik 123:1385–1389 7. Jyoti V (2014) Security enhancement in optical code division multiple access network. PhD thesis, Thapar University 8. Fok MP, Prucnal PR (2009) All-optical encryption for optical network with interleaved waveband switching modulation. Opt Soc Am 34(9):1315–1317 9. Kostinski N, Kravtsov K, Prucnal PR (2008) Demonstration of an all optical OCDMA encryption and decryption system with variable two-code keying. IEEE Photon Technol Lett 20(24):2045–2047 10. Wang Z, Huang YK, Deng Y, Chang J, Prucnal PR (2009) Optical encryption with OCDMA code swapping using all-optical XOR logic gate. IEEE Photon Technol Lett 21(7):411–413 11. Chang WH, Yang GC, Chang CY, Kwong WC (2015) Enhancing optical CDMA Confidentiality with Multi code keying Encryption. J Lightwave Technol 33(9):1708–1718 12. Arief AR, Aljunid SA, Anuar MS, Junita MN, Ahmad RB (2012) Mitigation of multiple access interference using two-dimensional modified double weight codes for optical code division multiple access systems. Opt Eng 51(6) 13. Goldberg S, Menendez RC, Prucnal PR (2007) Towards a cryptanalysis of spectral phase encoded optical CDMA with phase scrambling. Proc Int Conf Opt Fiber Commun 14. http://www.scielo.br/scielo.php?pid=S2179-10742013000200011&script=sci_abstract 15. https://en.wikipedia.org/wiki/Brute-force_attack 16. Monga H (2014) Performance evaluation of optical code division multiple access system. PhD thesis, Thaper University

Optimization of Physical Parameters of Single-Beam Vibrational Piezoelectric Energy Harvester Namrata Saxena, Neha Yaragatti, Shishir Sharma, and Ritu Sharma

Abstract This paper presents the cantilever-type vibrational piezoelectric energy harvester for wireless applications. The physical dimensions of the structure are optimized using parametric sweep analysis for beam length, width and thickness. The modal analysis is carried out for the computation of the resonance frequency. The stationary analysis is performed for the evaluation of displacement, von Mises stress, electric potential and total electric energy at 1 g (1 g = 9.8 m/s2 ) to 5 g input acceleration. The structure presents in this paper has low structure volume, stress with reasonable electric potential and total electric energy. This structure is appropriate for implementation as low-frequency energy harvester. Keywords V-PEH · Piezoelectric energy harvester · Cantilever beam · Stationary analysis

1 Introduction The continuous downscaling of the dimensions of modern microelectronic devices from micrometer to nanometer range leads to the power consumption from microwatt to nanowatt. Generally, chemical fuel cells have been employed to power these electronic devices. With the reduction in energy dissipation levels, the advancement of self-sustaining devices that uses energy harvesting as the best option for the realistic power source started increasing. This motivates the researchers to discover new N. Saxena (B) · R. Sharma Department of Electronics and Communication Engineering, MNIT Jaipur, Jaipur, Rajasthan, India e-mail: [email protected] R. Sharma e-mail: [email protected] N. Yaragatti Department of Mechanical Engineering, Manipal University Jaipur, Jaipur, Rajasthan, India S. Sharma Department of Computer Engineering, PennState University Park, Pennsylvania, USA © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_62

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energy harvesting techniques [1–3]. Nowadays, a lot of research is going on across the globe to explore new cantilever-type vibration energy harvesters that can be easily fabricated using MEMS fabrication technology to energize wireless sensor nodes and gadgets. These energy harvesters can generate unremitting energy in the range from hundreds of microwatt to few milliwatts. The ambient mechanical vibrations are effortlessly available in the environment which would establish energy harvesting as a better option at remote places where substitution of battery is not feasible [4–6]. The prominent ambient energy sources are biogas, solar, sound, thermoelectric, mechanical vibrations and wind [7, 8]. The four basic transduction mechanisms available for the harvesting of ambient mechanical vibrations are electromagnetic, magnetostrictive, electrostatic and piezoelectric [9–11]. Because of the complexity in implementation of the planar magnets and the necessity of minimum number of coils, the electromagnetic energy harvesters suffer several complicacies in manufacturing using MEMS technology [3, 5, 12]. Normally, the magnetostrictive type of transducers is appropriate for the harvesting of highfrequency vibrations due to its inherit advantages like no depolarization setback, very-high coupling coefficient and high flexibility. On contrary to this, the magnetostrictive type of energy harvesters suffers from nonlinearities and complexity in fabrication using MEMS technology and the requisite of pickup coils inhibits its implementation for wireless applications. The electrostatic transducers have parasitic capacitances and the high output impedance which leads to the confinement of the output current and consequently degrades the harvester efficiency. These generators work on the principal that the capacitance varies with the displacement from the ambient vibrations. These transducers can be fabricated using MEMS technology but they also require the application of external voltage for their proper operation [13, 14]. The piezoelectric type of energy harvesters has various benefits over other transduction mechanisms such as its simple design, working principal and the ease of fabrication using MEMS technology which makes these harvesters as the best choice for the ambient mechanical energy harvesting applications [15]. The piezoelectric energy harvesters can be realized using cantilever type of structure, i.e., fixed from one end and free to vibrate from another end to which the proof mass is connected [3, 5, 16]. The cantilever type of energy harvesters can be modeled as the spring-mass damper system to attain a fixed resonance frequency. The resonance frequency depends on the harvester’s dimensions (width, length and thickness) of the cantilever beam and the attached seismic-mass. In order to achieve optimum potential, the resonant frequency of the structure should be tuned according to the frequency of the mechanical vibrations. The main objective of this work is to design the single-beam cantilever-type piezoelectric energy harvester for the purpose of low-frequency ambient mechanical energy harvesting. The designing of the structure is obtained by using polycrystalline silicon for the proof mass and the structural layer of the beam, SiO2 (silicon dioxide) as the insulation layer between silicon substrate and the bottom electrode, gold is used for the top and the bottom electrode and ZnO is used as the piezoelectric material as it has high piezoelectric coefficient and electromechanical coupling coefficient [17]. In this paper, the optimization of physical parameters such as beam

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length, width and thickness is performed in order to obtain the optimized dimensions for low-frequency vibrational piezoelectric energy harvester (V-PEH). The work is categorized among three types of analyses. First, the modal analysis for the computation of eigenfrequency of the structure. Second, the parametric sweep is done to achieve the optimized dimensions and third, the stationary analysis is done for the computation of von Mises stress, electric potential and total electric energy of the structure.

2 Mathematical Modeling Most of the cantilever structures can be modeled as the spring-mass damper system that generates maximum potential when the structure’s resonant frequency corresponds to the frequency of the ambient mechanical vibrations. As the one end of the cantilever is fixed and the other end is free for the translational motion, it is also termed as the fixed-free structure. The resonance frequency of the fixed-free cantilever structure is defined by Eq. (1) [16] 1 fr = 2π



k m

(1)

where m is a mass of the attached proof mass. The spring constant (k) of the cantilever is given by the Eq. (2) k=

3Ewt 3 Ewt 3 3E I = = l3 12l 3 4l 3

(2)

where E is the Young’s modulus, I is the moment of inertia, l is the length, w is the width, and t is the thickness of the cantilever beam. The resultant formula for the resonant frequency of the single-beam cantilever is derived as Eq. (3) 1 fr = 2π



Ewt 3 4l 3 m

(3)

It shows that the resonant frequency depends on the cantilever dimensions and the mass of the proof mass.

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3 Design Parameters of Cantilever Beam In this paper, a single-beam cantilever-type vibrational piezoelectric energy harvester is designed and simulated using a FEM-simulator. In this structure, one end is fixed, and other end is free to vibrate to which the proof mass is attached at the bottom of the beam. The parametric sweep for the physical dimensions of the beam is performed to achieve the optimized dimensions. For the proper operation of the energy harvester, the mechanical and the electrical boundary conditions are applied to the structure. In the mechanical boundary condition, the one end of the cantilever beam is fixed, i.e., restrict its movement. In the electrical boundary condition, the terminal voltage is applied at the bottom layer of the top electrode, and the ground is applied to the top layer of the bottom electrode. Some of the important material properties are listed in Table 1. The dimensional view of the single-beam energy harvester structure is shown in Fig. 1. The length and width of the beam are 2500 and 2000 µm. The thickness of structural layer, i.e., silicon is 15 µm, insulation layer (SiO2 ) is 0.1 µm, and both the bottom and the top gold electrode is 0.2 µm each. The length of the proof mass is 1000 µm, width is kept same as of the beam, i.e., 2000 µm, and the thickness of the proof mass is 340 µm. These optimized parameters are obtained by performing parametric sweep for beam length, width and thickness. where ξ T is 3 × 3 dielectric constant matrix at constant stress, i.e., in stress-charge form. Table 1 Material properties Material properties

Materials Silicon

SiO2

ZnO

Gold

Density (kg/m3 )

2320

2200

5680

19,300

Relative permittivity (ξ T )

4.5

4.2

{8.5446, 8.5446, 10.204}

6.9

Fig. 1 Dimensional view of the single-beam piezoelectric energy harvester

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Fig. 2 First eigen frequency of the single-beam piezoelectric energy harvester

4 Result and Discussion The results obtained by various analysis are reported in this section:

4.1 Modal Analysis The modal analysis is performed for the computation of the resonant frequency of the energy harvester out of first-six frequency modes. As shown in Fig. 2., the first resonance frequency is considered as the eigenfrequency of the structure as it generates maximum displacement, piezoelectric potential and its deformation which is suitable for the proper operation of the energy harvester. Figure 2 shows that the single-beam cantilever-type piezoelectric energy harvester vibrates at the low frequency of 879.06 Hz at 1 g input acceleration.

4.2 Parametric Sweep Analysis The parametric sweep is performed in order to obtain the optimized physical dimensions of the structure. For this, the effect on the resonance frequency with the variation of all the three physical parameters, i.e., beam length, width and thickness is analyzed and is depicted in Fig. 3a–c, respectively. From Fig. 3a, it can be seen that

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Fig. 3 Effect of variation of a beam length, b beam width and c beam thickness with the first eigenfrequency of the single-beam piezoelectric energy harvester

the resonance frequency decreases with the increment in the beam length. Figure 3b shows that the resonance frequency increases in accordance with the beam width and beam thickness, respectively. These variations can also be justified by Eq. (3) for the resonance frequency.

4.3 Stationary Analysis The V-PEH produces maximum displacement in the ± Z direction. The stationary analysis is performed to compute displacement, von Mises stress, electric potential and the total electric energy of the single-beam piezoelectric energy harvester. This is performed by varying the input acceleration from 1 to 5 g. Figure 4a shows the displacement generated along with the arc length. It reveals that there is zero displacement at the fixed end and maximum displacement at the free-end of the energy harvester. Figure 4b shows that there is maximum stress at the fixed end of the cantilever beam and negligible in the area where proof mass is attached as it is the rigid portion of the structure. Figure 5 shows the variation of electric potential with respect to the arc length of the structure. It shows that maximum potential is generated at the fixed end of

Fig. 4 Variation of a total displacement, b von Mises stress with the arc length of the single-beam piezoelectric energy harvester

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Fig. 5 Variation of electric potential with the arc length of the single-beam piezoelectric energy harvester

the cantilever beam structure. The negative sign shows that the energy harvester vibrates in − Z direction. Figure 6 shows the variation of electric potential and total electric energy with the input acceleration. The harvester generates maximum electric potential of 27.1 mV and total electric energy of 0.494745 pJ at 1 g and 135.6 mV and 1.2368 pJ at 5 g input acceleration, respectively. Fig. 6 Variation of electric potential and total electric energy with the input acceleration of the single-beam piezoelectric energy harvester

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5 Conclusion This paper presents the single-beam cantilever-type V-PEH that resonates at the frequency of 879.06 Hz which is appropriate for low-frequency energy harvesting applications. The physical dimensions of the beam are optimized using parametric sweep analyses. The energy harvester can be implemented at various wireless applications as it produces the maximum displacement, von Mises stress, electric potential and total electric energy of 0.4 µm, 1.9×105 N/m2 , 27.1 mV and 0.494745 pJ, respectively, at 1 g input acceleration and 2 µm, 9.58 × 105 N/m2 , 135.6 mV and 1.2368 pJ at 5 g input acceleration, respectively, with the structure volume of 7.675 × 108 μm3 . Acknowledgements The authors would gratefully acknowledge DRDO, New Delhi for providing the financial assistance to this project.

References 1. Song HC et al (2017) Ultra-low resonant piezoelectric MEMS energy harvester with high power density. JMEMS 26:1226–1234 2. Uddin MdN et al (2016) Design and analysis of a T-shaped piezoelectric cantilever beam at low resonant frequency using vibration for biomedical device. Asian J Sci Res 9:160–166 3. Saxena N et al. Design and performance analysis of different structures of MEMS PVDF-based low-frequency piezoelectric energy harvester. ICTSES’18 LNEE 607:173–182 4. Domínguez-Pumar M et al (2016) MEMS technologies for energy harvesting. Springer International Publishing, Switzerland 5. Iqbal A, Mohd-Yasin F (2015) Comparison of seven cantilever designs for piezoelectric energy harvester based on Mo/AlN/3C-SiC. In: RSM’15 IEEE regional symposium on micro and nanoelectronics 6. Yu H et al (2014) A vibration-based MEMS piezoelectric energy harvester and power conditioning circuit. Sensors 14:3323–3341 7. Aktakka EE (2011) Thinned-PZT on SOI process and design optimization for piezoelectric inertial energy harvesting. In: Transducers’11 8. Zhang J et al (2015) Analysis of frequency characteristics of MEMS piezoelectric cantilever beam based energy harvester. In: Symposium on piezoelectricity, acoustic waves, and device applications (SPAWDA) 9. Li H et al (2014) Energy harvesting from low frequency applications using piezoelectric materials. Appl Phys Rev 1:041301 10. Saxena N et al (2018) Design and performance analysis of different structures for low-frequency piezoelectric energy harvester. Nanoelectron Circuits Commun Syst NCCS LNEE 642:47–56 11. Kim SB et al (2013) Comparison of MEMS PZT cantilevers based on d 31 and d 33 modes for vibration energy harvesting. JMEMS 22:26–33 12. Guo T et al (2017) Optimized structure design of a bridge-like piezoelectric energy harvester based on finite element analysis. In: ICEMS’17 20th international conference on electrical machines and systems 13. Kim H-U et al (2009) Piezoelectric microgenerators-current status and challenges. IEEE Trans Ultrason Ferroelectr Freq Control 56:1555–1568 14. Kim HS et al (2011) A review of piezoelectric energy harvesting based on vibration. Int J Precis Eng Manuf 12:1129–1141 15. Priya S, Inman DJ (2009) Energy harvesting technologies, 1st edn. Springer, New York

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16. Liu C (2012) Foundation of MEMS, 2nd edn. Pearson Education Limited 17. Saxena N et al (2020) Enhancement in structural, morphological and optical features of thermally annealed zinc oxide nanofilm. Indian J Pure Appl Phys 58:642–648

Retraction Note to: Design of Silicon-on-Insulator Based Mode Splitters with Asymmetrical Variation of Slots Neha Choudhary, Veer Chandra, and Rakesh Ranjan

Retraction Note to: Chapter “Design of Silicon-on-Insulator Based Mode Splitters with Asymmetrical Variation of Slots” in: M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_15 The Editors have retracted this conference paper “Design of Silicon-on-Insulator Based Mode Splitters with Asymmetrical Variation of Slots” because of substantial overlap with an earlier article published by others (1). Although the authors have cited the earlier study (1) and state that their work was an extension of it, the Editors concluded that because the design of the mode splitter reported is not original this conference paper is redundant. All Authors agree to this retraction. (1) Hada SL, Rahman BMA (2019) Design of compact mode splitters using identical coupled waveguides with slots. OSA Continuum 2:848–861

The retracted version of this chapter can be found at https://doi.org/10.1007/978-981-16-2818-4_15 © The Author(s), under exclusive license to Springer Nature Singapore Pte Ltd. 2022 M. Tiwari et al. (eds.), Optical and Wireless Technologies, Lecture Notes in Electrical Engineering 771, https://doi.org/10.1007/978-981-16-2818-4_63

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