Multiphase Hybrid Electric Machines: Applications for Electrified Powertrains [1st ed. 2022] 3030804348, 9783030804343

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Ahmad S. Al-Adsani Omid Beik

Multiphase Hybrid Electric Machines

Applications for Electrified Powertrains

Multiphase Hybrid Electric Machines

Ahmad S. Al-Adsani • Omid Beik

Multiphase Hybrid Electric Machines Applications for Electrified Powertrains

Ahmad S. Al-Adsani Department of Electrical Engineering Technology, Public Authority for Applied Education and Training (PAAET) College of Technological Studies (CTS) Kuwait City, Kuwait

Omid Beik Department of Electrical and Computer Engineering McMaster University Hamilton, ON, Canada

ISBN 978-3-030-80434-3 ISBN 978-3-030-80435-0 https://doi.org/10.1007/978-3-030-80435-0

(eBook)

© Springer Nature Switzerland AG 2022 This work is subject to copyright. All rights are reserved by the Publisher, whether the whole or part of the material is concerned, specifically the rights of translation, reprinting, reuse of illustrations, recitation, broadcasting, reproduction on microfilms or in any other physical way, and transmission or information storage and retrieval, electronic adaptation, computer software, or by similar or dissimilar methodology now known or hereafter developed. The use of general descriptive names, registered names, trademarks, service marks, etc. in this publication does not imply, even in the absence of a specific statement, that such names are exempt from the relevant protective laws and regulations and therefore free for general use. The publisher, the authors, and the editors are safe to assume that the advice and information in this book are believed to be true and accurate at the date of publication. Neither the publisher nor the authors or the editors give a warranty, expressed or implied, with respect to the material contained herein or for any errors or omissions that may have been made. The publisher remains neutral with regard to jurisdictional claims in published maps and institutional affiliations. This Springer imprint is published by the registered company Springer Nature Switzerland AG The registered company address is: Gewerbestrasse 11, 6330 Cham, Switzerland

Preface

Development of road vehicles through electrified powertrains without compromising their power capability, efficiency, performance, reliability, safety, driving range, and cost has been the center of attention in academia and industry. This book is a userfriendly reference and attractive subject for researchers and undergraduate and graduate engineers who are interested in multiphase, permanent magnet and hybrid machine topologies with a specific application in electric and hybrid electric vehicles. This book begins with an overview and principals of classical electric machine operation, magnetic modeling, and characteristics of materials. Various classical electric machine topologies, including brushed DC, and different AC machines are discussed in Chap. 1. Chapter 2 discusses hybrid electric machine concept together with a review of different hybrid machine topologies, indicating their operational philosophy, advantages, and disadvantages. Chapter 3 presents a hybrid permanent magnet (HPM) machine topology that is selected and analyzed in terms of its geometry, excitation field technique, backEMF, and developed electromagnetic torque for both no-load and on-load operations. The HPM topology consists of two different synchronous machines, a permanent magnet (PM) and a wound field (WF) machine that are coupled on the same rotor shaft, rotate with the same speed, and share the same multiphase stator. Chapter 4 discusses an overview of multiphase electric machines. AC and rectified DC output voltage waveforms of three- and nine-phase systems with associated power electronics are presented. A comparison of three- and two ninephase machine winding, rectification characteristics, and losses for both HPM and PM machine topologies is presented in Chap. 4. Chapter 5 presents an overview of electric vehicles (EVs) and hybrid electric vehicles (HEVs), their powertrains, and on-board energy sources. Different battery technologies are discussed, and in the case of HEVs the feasibility of disconnecting the internal combustion engine (ICE) from the electric drivetrain is studied. Driving range, fuel economy, and emissions are evaluated over different driving cycles, and at different vehicle powertrain hybridization ratios (HR) in Chap. 5. v

vi

Preface

In Chap. 6, a dynamic model of vehicle powertrain that includes a HPM generator integrated into an ICE in an SHEV while considering a load demand is presented. The ICE/HPM generator output power control scheme is modeled while maintaining ICE efficiency within its optimal region. Several operating scenarios for the HPM generator excitation scheme are assessed, and the HPM generator is characterized utilizing a 32-phase brushless excitation scheme. In addition, different cases, such as normal, boost, and buck functionality of HPM machine operation, are analyzed, and a choice of the most appropriate operation mode has been selected to regulate the total back-EMF via a WF excitation current control. Dr. Al-Adsani wishes to express his sincere gratitude toward his wife, and the authors extend special thanks to Dr. Nigel Schofield at the University of Huddersfield for his valuable inputs and to the team at Springer for their care during the book production. Kuwait City, Kuwait Toronto, ON, Canada

Ahmad S. Al-Adsani Omid Beik

Contents

1

2

General Electric Machine Theory . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.1 Magnetic Circuit Principles . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.1.1 Magnetic Field Distribution and Flux Density . . . . . . . . . 1.1.2 Ferromagnetic Materials and Magnetization Curves . . . . . 1.2 Electric Machine Fundamentals . . . . . . . . . . . . . . . . . . . . . . . . . . 1.3 Overview of Classical Electric Machine Topologies . . . . . . . . . . . 1.3.1 Brushed DC and AC Machines . . . . . . . . . . . . . . . . . . . . 1.3.2 Brushless AC Machines . . . . . . . . . . . . . . . . . . . . . . . . . 1.3.3 Switch Reluctance Machines . . . . . . . . . . . . . . . . . . . . . 1.4 WF and PM Synchronous Machine Excitation Fields . . . . . . . . . . 1.4.1 Magnetic Flux Path Representation of WF Synchronous Machines . . . . . . . . . . . . . . . . . . . . . . . . . . 1.4.2 Magnetic Flux Path Representation of PM Synchronous Machines . . . . . . . . . . . . . . . . . . . . . . . . . .

1 1 1 3 5 6 7 10 11 13

Hybrid Electric Machine Concept . . . . . . . . . . . . . . . . . . . . . . . . . . 2.1 History . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.2 Hybrid Electric Machine Classification . . . . . . . . . . . . . . . . . . . . 2.3 Different Hybrid Machine Topologies . . . . . . . . . . . . . . . . . . . . 2.3.1 PM Synchronous Machine with Claw Pole Field Excitation (PSCPF) . . . . . . . . . . . . . . . . . . . . . . . 2.3.2 Toroidal-Stator Transverse-Flux Machine (TSTFM) . . . . 2.3.3 Hybrid Excitation Synchronous Machine (HESM) . . . . . 2.3.4 Synchronous Permanent Magnet Hybrid AC Machine (SynPM) . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.3.5 Consequent Pole Permanent Magnet Hybrid Excitation Machine (CPPM) . . . . . . . . . . . . . . . . . . . . . 2.3.6 Field Controlled Torus-NS (FCT-NS) Machine . . . . . . . 2.3.7 Dual-Rotor Machine . . . . . . . . . . . . . . . . . . . . . . . . . . 2.3.8 Imbricated Hybrid Excitation Machine (IHEM) . . . . . . .

. . . .

17 17 17 18

. . .

18 21 23

.

26

. . . .

27 29 32 33

13 14

vii

viii

Contents

2.3.9

.

37

.

38

. .

39 43

Hybrid Permanent Magnet Machine Design . . . . . . . . . . . . . . . . . . . 3.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.2 Machine Volume Envelope Consideration . . . . . . . . . . . . . . . . . . 3.2.1 PM Machine Dimensions . . . . . . . . . . . . . . . . . . . . . . . . 3.2.2 PM Machine Stator Winding Layout . . . . . . . . . . . . . . . . 3.2.3 Stator Winding Fill Factor and Resistance . . . . . . . . . . . . 3.2.4 Finite Element Method Program . . . . . . . . . . . . . . . . . . . 3.2.5 Machine Back-EMF Prediction . . . . . . . . . . . . . . . . . . . . 3.2.6 PM Machine Analysis Via EMC Model . . . . . . . . . . . . . 3.3 WF Machine . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.3.1 WF Rotor Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.3.2 WF to PM Split Ratio . . . . . . . . . . . . . . . . . . . . . . . . . . 3.3.3 Comparative Analysis of WF Rotor Designs . . . . . . . . . . 3.4 HPM Machine Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.4.1 Torque Prediction and Saturation . . . . . . . . . . . . . . . . . . 3.4.2 Synchronous Inductance and Winding Resistance . . . . . . 3.5 HPM Machine Final Design Model Analysis . . . . . . . . . . . . . . . . 3.5.1 Rotor PM Demagnetization . . . . . . . . . . . . . . . . . . . . . . 3.5.2 Core Loss Prediction . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.6 HPM Machine Thermal Model . . . . . . . . . . . . . . . . . . . . . . . . . . 3.6.1 General Principle of the Lumped Parameter Method . . . . . 3.6.2 Conduction Heat Transfer . . . . . . . . . . . . . . . . . . . . . . . . 3.6.3 Convection Heat Transfer . . . . . . . . . . . . . . . . . . . . . . . . 3.6.4 Radiation Heat Transfer . . . . . . . . . . . . . . . . . . . . . . . . . 3.6.5 HPM Machine Thermal Model . . . . . . . . . . . . . . . . . . . . 3.7 Comparison Between PM and Four HPM Machine Topologies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.8 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

45 45 45 47 47 50 52 55 56 61 62 64 65 69 69 73 75 75 77 81 81 82 84 86 86

Multiphase HPM Generator Systems . . . . . . . . . . . . . . . . . . . . . . . 4.1 Overview on Multiphase Machines . . . . . . . . . . . . . . . . . . . . . . 4.1.1 Multiphase Windings Principles . . . . . . . . . . . . . . . . . . 4.1.2 Rectified Voltage due to Three- and Nine-Phase HPM Generator Systems . . . . . . . . . . . . . . . . . . . . . . . 4.2 Nine-Phase HPM Generator Parameters . . . . . . . . . . . . . . . . . . . 4.2.1 Nine-Phase Winding Layout and Back-EMF . . . . . . . . . 4.2.2 Back-EMF and Torque Waveform Harmonics Prediction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

95 95 96

2.4 3

4

Series Double Excited Synchronous Machine (SDESM) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.3.10 Switch Reluctance Machine with Stator Field Assistance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.3.11 Dual-Stator Hybrid Excited Synchronous Wind Generator (DSHESG) . . . . . . . . . . . . . . . . . . . . . Summary of Surveyed Literature on HPM Machines . . . . . . . . . .

. . .

90 94

. 96 . 101 . 101 . 103

Contents

4.3

4.4

4.5

4.6

ix

4.2.3 Synchronous Inductance Prediction . . . . . . . . . . . . . . . . 4.2.4 Construction of HPM Machines Prototype . . . . . . . . . . . 4.2.5 Resistance and Inductance Measurements . . . . . . . . . . . Analysis Models . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.3.1 General dq Mathematical Model of HPM Generator . . . . 4.3.2 Simulation Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . Three- and Nine-Phase HPM Generator System Studies . . . . . . . 4.4.1 Impact on Synchronous Inductance and Rectifier . . . . . . 4.4.2 System Sensitivity to Generator Synchronous Inductance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.4.3 DC-Link Voltage Quality . . . . . . . . . . . . . . . . . . . . . . . Loss Audit of Generator Systems . . . . . . . . . . . . . . . . . . . . . . . . 4.5.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.5.2 Core Loss Prediction . . . . . . . . . . . . . . . . . . . . . . . . . . 4.5.3 Passive and Active Converter Loss for HPM and PM Generator Systems . . . . . . . . . . . . . . . . . . . . . . Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

. . . . . . . .

107 111 114 115 115 118 119 121

. . . . .

126 130 134 134 136

. 136 . 141

5

Electric and Hybrid Electric Powertrains . . . . . . . . . . . . . . . . . . . . . 5.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.2 Overview of EVs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.2.1 EV Powertrain Configuration . . . . . . . . . . . . . . . . . . . . . 5.2.2 Battery Technology . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.3 Overview of HEVs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.3.1 HEV Powertrain Configurations . . . . . . . . . . . . . . . . . . . 5.4 Vehicle Driving Cycles . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.5 Series Hybrid Electric Vehicle (SHEV) . . . . . . . . . . . . . . . . . . . . 5.5.1 ZEBRA Battery . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.5.2 Internal Combustion Engine (ICE) . . . . . . . . . . . . . . . . . 5.5.3 Engine-Mounted Multiphase HPM Generator . . . . . . . . . . 5.6 Electric Vehicle Range Extender . . . . . . . . . . . . . . . . . . . . . . . . . 5.6.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.6.2 Literature Review of EV Range Extender Studies . . . . . . . 5.7 ICE/HPM Generator Range Extender in SHEVs . . . . . . . . . . . . . . 5.7.1 Vehicle Traction Machine Torque . . . . . . . . . . . . . . . . . . 5.7.2 Hybridization Ratio . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.7.3 Range Extender Sizing in SHEV Powertrain . . . . . . . . . . 5.7.4 Study Cases . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.8 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

143 143 143 145 146 146 148 151 151 154 156 159 159 159 160 162 163 165 165 169 170

6

Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.2 HPM Machine Back-EMF Control Strategy . . . . . . . . . . . . . . . . 6.2.1 Control Strategy Analysis . . . . . . . . . . . . . . . . . . . . . . . 6.2.2 DC-link Design Options . . . . . . . . . . . . . . . . . . . . . . . .

171 171 173 174 178

. . . . .

x

Contents

6.3

6.4

6.5

HPM Machine Output Power Control . . . . . . . . . . . . . . . . . . . . . 6.3.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.3.2 HPM Generator Operating Scenarios . . . . . . . . . . . . . . . . 6.3.3 Energy Loss Prediction for Two Driving Cycles . . . . . . . . 6.3.4 Solving Final Choice with Full Simulation Model . . . . . . 6.3.5 Thermal Analysis Results of the Investigated HPM Machine . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . HPM Machine Characterization Using Brushless Excitor . . . . . . . . 6.4.1 32-Phase Brushless Excitation Scheme . . . . . . . . . . . . . . 6.4.2 Performance Curves . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.4.3 Efficiency Mapping . . . . . . . . . . . . . . . . . . . . . . . . . . . . Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

181 181 184 187 189 191 194 197 200 204 206

References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 207 Index . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 221

Chapter 1

General Electric Machine Theory

1.1

Magnetic Circuit Principles

In electric machines, a magnetism phenomenon is utilized to build an electromotive force (EMF) to supply an electric load, as in generators, or to drive a mechanical load as in motors. In electric devices, four basic principles can describe how magnetic field is used [1]: (i) In a current-carrying conductor, a magnetic field is produced around that conductor. (ii) In transformer action, a time changing magnetic field induces a voltage in a coil if it passes on it. (iii) In generator action, a moving conductor in the presence of a magnetic field induces a voltage and hence current flows through that conductor. (iv) In motor action, a current-carrying conductor in the presence of a magnetic field has an electromotive force induced on it.

1.1.1

Magnetic Field Distribution and Flux Density

The magnetic field that is created by current-carrying conductors based on Ampere’s right-hand rule, as in Fig. 1.1a, shows a right hand with the thumb pointing in the direction of current flow, while the magnetic field is rotating in the direction of the other fingers. Notice in Fig 1.1b that the symbol ⨂ denotes a cross-sectional view of the conductor carrying the current into the paper, while the symbol ⦿ denotes the current flow out of the paper. A magnetic field intensity (H ) is characterized as an effort a current is putting into establishing a magnetic field. The field intensity due to excitation DC current (I), which passes in a coil with (N ) turns through magnetic circuit path length (Lc), is calculated in (1.1) [2]. The strength of a magnetic field flux density (B) is governed by H and core material, as in (1.2). μ represents the magnetic © Springer Nature Switzerland AG 2022 A. S. Al-Adsani, O. Beik, Multiphase Hybrid Electric Machines, https://doi.org/10.1007/978-3-030-80435-0_1

1

2

1 General Electric Machine Theory

Fig. 1.1 Magnetic field direction due to a currentcarrying conductor coil (a) Current-carrying conductor coil and right-hand rule (b) Magnetic flux direction for in and out of the page current directions

permeability of the material. A convenient way of representing the magnetizability of materials is by comparing material permeability to the permeability of free space (μo ¼ 4π  107 H/m), which is referred to as relative permeability (μr). Note in the magnetic circuit that the flux density is governed by the flux (φ) and the crosssectional area (A) of the medium that it is passing through, as in (1.4). NI Lc

ð1:1Þ

B ¼ μH μ μr ¼ μo φ B¼ A

ð1:2Þ

H¼

ð1:3Þ ð1:4Þ

By considering the simple core shape in Fig. 1.1a, the magnetic path in the core has a uniform shape, which has a reluctance value that depends on the path length, core permeability, and cross-sectional area as in (1.5). The reluctance in the magnetic circuit is like the resistance in the electric circuit, where one governs the flow of flux and the flow of current, respectively, as in Fig. 1.2. In the magnetic circuit, the coil has NI quantity that is called magnetomotive force (MMF). However, for permanent magnet (PM) materials, the MMF is calculated as in (1.7), where Hc is the PM core field intensity and Lc is the PM core length (thickness).

1.1 Magnetic Circuit Principles

3

Fig. 1.2 Electric and its magnetic circuit analogy (a) Electric circuit (b) Magnetic circuit

R ¼

1.1.2

L μA

ð1:5Þ

MMF ¼ NI

ð1:6Þ

MMF ¼ H c Lc

ð1:7Þ

Ferromagnetic Materials and Magnetization Curves

When the magnetic circuit is divided into sections of materials that easily allow flux line path to be formed, this is called ferromagnetic materials (FERMMs). In FERMMs, there are atoms, and each atom has its own magnetic moment direction, which is separated by a domain wall in each crystal boundary, as in Fig. 1.3a. These magnetic moments tend to align in the same direction over domains containing many atoms when they are subjected to a magnetic field intensity [2]. As H increases further and further, more domain directions will align until all the domains are in the same direction and when the material is magnetized to the maximum extent (saturation region), as shown in Fig. 1.3a. Here, if the majority of the domains are in the same directions after the applied field is removed, the material is said to be permanently magnetized. Another important phenomenon that occurs in FERMMs is called hysteresis. Hysteresis is described by referring to a typical B–H curve in Fig. 1.3a. When a current flows through a coil warped around a ferromagnetic core, MMF will then be created. As the MMF increases, so does H until the core saturates, which is presented by point o to point a in Fig. 1.3a. Now, if the current decreased to zero, the MMF and hence field intensity will go to zero. However, flux density will not go to zero, which is presented by point a to point b in Fig. 1.3a. Here, the core remains magnetized, even though the applied current and field intensity have gone to zero. The magnetism that remains in the core is called residual magnetism, and this effect creates a permanent magnet. If the applied current is reversed and slowly increases in the negative direction, the flux density will be driven to zero, as presented by point b to c in Fig. 1.3a. The negative field intensity needed to drive B to zero is called coercive force, as presented by point c in Fig. 1.3a. As the current

4

1 General Electric Machine Theory

Fig. 1.3 General hysteresis loops for FERMMs (a) B versus H hysteresis loop showing path major points (b) Soft ferromagnetic materials (c) Hard ferromagnetic materials

is made more negative, the core will eventually saturate and the flux density will have a polarity opposite to that in the original case, as presented by point d in Fig. 1.3a. Finally, if the current is reduced to zero and then made positive again, the curve will join up with the original curve passing through points e and f [2], where this closed loop joining points a, b, c, d, e, and f is called a hysteresis loop. Thus, a PM material is typically a metal alloy, which after being subjected to field intensity retains a substantial residual flux density (Br). In order to reduce the flux density to zero, an H field direction opposite in sense to the original magnetizing field must be applied. This impressed field magnitude must have a value (Hc) known as the coercive force. Here, materials magnetism can be categorized as permanent or temporary based on their ability to be magnetized and hold their magnetism or their magnetism

1.2 Electric Machine Fundamentals

5

Fig. 1.4 Different cuts of PM materials available in the market [3] (a) Ferrite (b) Neodymium– Iron–Boron (NdFeB) (c) Samarium–Cobalt (SmCo) (d) Aluminum–Nickel–Cobalt (AlNiCo) (Magnetic materials: Goudsmit Magnetics, the Netherlands) Fig. 1.5 Hard and soft permanent magnet demagnetization curves

vanishes as the DC supply source of the conductor that carries the current is turned off [1]. Magnetic materials are relatively easy to magnetize since their relative permeability values are high. FERMMs are classified as soft, in which the most common magnetic materials include steels, iron, nickel, aluminum, cobalt, and rareearth elements. Fig. 1.3b illustrates the expected hysteresis loop behavior for the soft FERMMs. Hard FERMMs, which have the expected hysteresis loop as in Fig. 1.3c, comprise the permanent magnet materials such as alnico, the alloys of cobalt with rare-earth elements such as samarium, copper–nickel alloys, chromium steels, and other metal alloys. Fig. 1.4 shows different permanent magnet materials with special cuts found in the market. In Fig. 1.5, the B–H demagnetization curves for several soft and hard magnetic materials with different grades are illustrated. The PM material grade is a number, which is specified after material type, to show different curves for the same material based on cost, magnetic performance, and operational temperature resistance.

1.2

Electric Machine Fundamentals

Electric machines are considered electromechanical power converters, such that they convert mechanical power into electrical, as in generators, and convert electrical into mechanical energy, as in motors. For generators, a source of mechanical power is

6

1 General Electric Machine Theory

Fig. 1.6 Fundamental concepts associated with electrical/mechanical systems

required to rotate the machine shaft (prime mover), via applied torque (T ), at a fixed or variable speed (ω) to develop an electromotive force (voltage difference) at the machine terminals (v) and vice versa for motor action as illustrated in Fig. 1.6. Hence, the electric field is considered the coupled medium between generation and motor output quantities. The electric machines differ in their internal magnetic source type, construction, and operation. In this chapter, the source of the magnetic field in general electric machines and their stator and rotor geometry are discussed.

1.3

Overview of Classical Electric Machine Topologies

Electric machines consist of two major parts: stator and rotor. Stator is the stationary part that does not move during operation, while the rotor is free to move and it can be the inner or outer part of the machine. Both stator and rotor parts are made of FERMMs that are discussed in the previous section. The stator accommodates the alternating current (AC) conductors in slots that are cut on the inner periphery and in some machines topologies in the outer periphery of the rotor structure. The coupling between stator and rotor fields can be increased by selecting a low reluctance material, which increases the flux density through the machine’s active parts. The classification of various electrical machine topologies focuses on the machines with or without commutators together with synchronous and asynchronous AC machines types, as shown in Fig. 1.7. The utilization of these machine topologies in the industry is subjected to meet the designer target in terms of efficiency, power density, and cost while their usage ranges from light- to heavy-duty loads.

1.3 Overview of Classical Electric Machine Topologies

7

Fig. 1.7 Classical electric machine common classifications

1.3.1

Brushed DC and AC Machines

Direct current (DC) machines have essential features that made them continually find application because of the relative simplicity and flexibility of their drive systems compared with AC machines. In brushed DC machine topology, as in Fig. 1.8, having a higher number of stator salient poles causes core saturation; hence, two, four, and six poles are common. Their field winding is placed on the salient stator, and the armature winding is placed on the round rotor. Through the field winding, a DC current is applied to produce the flux, which presents the major component in the general induced voltage formula. Such that, the induced voltage (e) in a conductor of length (l) moving with linear velocity (v) in a non-time-varying magnetic flux density is given in (1.8) [1, 4]. A unidirectional terminal voltage can be applied through a brush and mechanical commutator assembly. For a single coil in DC machines, a commutator action is to provide a full-wave rectification, and by

8

1 General Electric Machine Theory

Fig. 1.8 Radial view for two and four salient pole brushed DC machines topology. (a) Two salient pole (b) Four salient pole

assuming sinusoidal flux distribution, the voltage waveform between brushes can be transformed to a DC or average voltage (Ea) value between brushes as in (1.9) [4], where ω represents the machine rotational speed.

Ea ¼

1 π

e ¼ Blv

Z

π 0

2 ωNφ sin ωtd ðωt Þ ¼ ωNφ π

ð1:8Þ ð1:9Þ

DC machine working principle lays on the current flow through a coil within a magnetic field, and then a magnetic force is produced to generate a torque that rotates the rotor through four field excitation design topologies to display a wide variety of volt-ampere or speed-torque characteristics for both dynamic and steady-state operation [4]. In DC generators, the field excitation topologies are called (i) separately excited, (ii) shunt, (iii) series, (iv) cumulative compound (adds shunt and series effect), and (v) differential compounded (subtract shunt and series effect) generator [4]. Generally, these DC generator schemes are compared by their terminal voltage regulation. Unlike DC motors, which are compared based on their speed regulation capability. DC motors are driven from DC power supply. Unless otherwise specified, the input voltage to a DC motor is assumed to be constant because that assumption simplifies the analysis of motor comparison. Also, DC motors have five field excitation topologies: (i) separately excited, (ii) shunt, (iii) series, (iv) compound, and (v) permanent magnet [1]. However, brushed AC machines, as in Fig. 1.9, differ from DC machines in their armature winding location. Their armature windings are almost always located on the stator, while their field windings are located on the rotor. Generally, there are two magnetic fields presented: magnetic field from rotor circuit DC current excitation and another magnetic field from stator circuit. The interaction of these two magnetic fields produces a torque in the machine, just like two PMs near each other that will experience a torque that causes them to line up. The rotating magnetic field from the

1.3 Overview of Classical Electric Machine Topologies

9

Fig. 1.9 Radial view for two and four salient pole brushed three-phase AC machine topology. (a) Two pole (b) Four pole

rotor field windings of an AC machine induces a three-phase set of AC voltages, which are shifted by 120 electrical, into the stator armature windings calculated as in (1.10). Conversely, a three-phase set of currents in the stator armature windings produces a rotating magnetic field, which interacts with the rotor magnetic field, producing torque in the machine [1]. Hence, the relationship between electrical angle (θe) and the mechanical angle (θm) for AC machines with a number of poles (P) is given in (1.11). Similarly, the relationship between electrical frequency ( fe) and the mechanical frequency ( fm) of magnetic field rotation is given in (1.12). Note that it is also possible to relate the electrical frequency in hertz to the resulting mechanical speed (nm) of the magnetic fields in revolutions per minute (RPM) as in (1.13). ea ðtÞ ¼ ωNφ sinðωtÞ eb ðtÞ ¼ ωNφ sinðωt  120o Þ ec ðtÞ ¼ ωNφ sinðωt  240o Þ

ð1:10Þ

P θ e ¼ θm 2

ð1:11Þ

P f 2 m

ð1:12Þ

fe ¼

10

1 General Electric Machine Theory

fe ¼

nm P 120

ð1:13Þ

There are two rotor types, salient and nonsalient (round), in wound field (WF) synchronous generators. The rotors are subjected to changing magnetic fields, and it is constructed of thin laminations to reduce eddy current losses. Rotor DC field winding can be supplied by DC source through slip rings and brushes as in Fig. 1.9, or it can be through a special DC source mounted directly on the shaft of the synchronous generator. Slip rings and brushes are applied for small synchronous machines because no other methods are cost-effective [5]. On the other hand, large generators and motors and brushless exciters are used to supply the DC field current to the machine. A brushless exciter is a small AC generator with its field circuit mounted on the stator and its armature circuit on the rotor [4]. By controlling the small DC field current of the exciter generator, the rotor DC field winding of the main WF synchronous generator is regulated.

1.3.2

Brushless AC Machines

An induction machine (IMs) is one in which AC current is supplied to the stator directly and to the rotor by induction or transformer action. As in the synchronous machine, the stator winding is like the synchronous generator discussed in the previous section. When excited from a balanced three-phase source, it produces a magnetic field in the air-gap rotating at synchronous speed as determined by the numbers of poles and the applied stator voltage frequency. In IM topology, there are two rotor types: squirrel-cage and wound rotor [4]. In this section, only the squirrelcage rotor is considered, as shown in Fig. 1.10. Compared with the wound rotor type,

Fig. 1.10 Radial view for brushless three-phase squirrel-cage IM machine topology

1.3 Overview of Classical Electric Machine Topologies

11

Fig. 1.11 Radial view for round and salient four-pole brushless three-phase PM machine topology. (a) Salient PM rotor (b) Nonsalient PM rotor

the squirrel-cage rotor winding does not require slip rings and brushes; however, it consists of conducting bars embedded in slots in the rotor iron core and shortcircuited at each end by conducting end rings. The squirrel-cage motor is substantially a constant speed motor having a few percent drops in speed (slip) from no load to full load. Different classes of squirrel-cage machines are presented in the literature based on the effective resistance of the rotor-cage circuit [4]. Such that, the effect of using these rotor-cage classes dictates machine torque-speed characteristics. Hence, the extreme simplicity and raggedness of the squirrel-cage construction are exceptional advantages of this type of IM. As for the brushless machine types, permanent magnet AC machines or brushless PM machines are occasionally built to operate as synchronous machines with rotating field winding replaced by a PM. Fig. 1.11 illustrates the brushless threephase PM synchronous machine having either salient or nonsalient PM rotor type. The flux paths due to a four-pole PM AC machine that links stator phase coils with rotor magnetic field are shown in Fig. 1.11a. Knowing that, if a constant torque is exerted on the shaft to run the machine at a constant speed, this provides generator action. On the other hand, if the three-phase winding is excited using a semiconductor control switching pattern, then the machine is operating as a motor.

1.3.3

Switch Reluctance Machines

In terms of electric machine construction, switch reluctance machines (SRMs) are considered a simple and rigid machine type. Their excitation winding is placed in the salient or nonsalient stator only, where they always have salient magnetic rotor shape. They operate using generated flux linkage due to stator applied current; its path between stator and rotor tries to generate maximum torque through a tendency

12

1 General Electric Machine Theory

Fig. 1.12 Radial view for brushless SRM topology showing two stator poles to rotor poles ratio. (a) 4/2 SRM (b) 6/4 SRM

to align rotor with the stator-produced flux linkage [4], as shown in Fig. 1.12. For the control, the rotor position sensing is required in order to properly energize the stator phase windings to produce torque. The SRM needs to be designed such that the stator winding inductance varies with rotor position, while the stator core of SRM requires high permeability magnetic material. The torque characteristics of SRM are governed by the saliency of stator and rotor, which enhances the difference between maximum and minimum inductances [4]. In SRMs, the torque is proportional to the magnitude of the phase current and does not depend on its direction. Hence, unidirectional current can be used to supply the stator winding through solid-state switches. Therefore, for motor drive, only half of the solid-state switches are required to energize the stator phase through a single current direction, which reduces the control electronics by half compared with the other machine drive system [4], such as in brushless PM machines. The zero torque position in the SRM cannot be presented if the ratio between the stator poles (SP) to rotor poles is not an integer. For instance, SP/P for 6/4 SRM is 1.5, and hence there will not be a simultaneous alignment of stator phase inductance. However, in some instances, a SRM with an integer pole ratio is desirable; in this case, the elimination of zero torque is attained by constructing the machine with an asymmetric rotor [4]. Therefore, the rotor pole width is made wider than that of the stator. In general, when a given phase is excited, the torque is such that the rotor is pulled to the nearest position of maximum flux linkage. As excitation is removed from this phase and the next phase is excited, the rotor is then pulled to a new maximum flux-linkage position [4].

1.4 WF and PM Synchronous Machine Excitation Fields

1.4

13

WF and PM Synchronous Machine Excitation Fields

As discussed in the previous sections, the wound field (WF) and PM rotor types of AC synchronous machines provide rotating magnetic fields that produce the threephase set of voltages in the stator coils as given in (1.10). The excitation field in the WF rotor type is supplied by the DC voltage source through slip rings and brushes, as in Fig. 1.9. While the PM rotor type does not need that, it instead requires spatial arranging of soft or hard PM material, which can be accommodated on the rotor core in common ways known as surface-mounted magnets, inset magnets, buried magnets with radial magnetization, and buried magnets with circumferential magnetization [5]. Note that, in this book, surface-mounted magnet rotor type is chosen for the synchronous PM machine topology, as in Fig. 1.11.

1.4.1

Magnetic Flux Path Representation of WF Synchronous Machines

By Ampere’s law, the current in a coil of wire wrapped around a ferromagnetic material core produces a magnetic flux in the core. In a magnetic path representation, the reluctance is the counterpart of electrical circuit resistance, and its unit is ampereturns per weber (A•t/Wb), while the MMF in magnetic path representation is analogous to EMF voltage in an electrical circuit and its unit is ampere-turns (A•t). The magnetic path representation translates the magnetic field behavior within electric machine parts to a simplified manner, which otherwise is complex to analyze for machine design process, as will be seen in detail in Chap. 3. Assume that a concentrated stator winding is employed when the number of stator slots is equal to the number of rotor poles. By considering a sectional view of a WF synchronous machine, here the flux does not behave in a simple manner since there are different ferromagnetic rotor and stator materials in addition to air-gap and different cross-sectional flux path areas. Therefore, different cross-section path reluctances are calculated using (1.5). Machine active parts are the parts through which magnetic flux is passing causing generation of EMF voltage and electromagnetic torque. There are nine different reluctances and one rotor coil MMF, which is calculated as in (1.6), in the considered machine section. The flux-linkage path reluctances are represented by left side stator yoke (R sy1), right side stator yoke (R sy2 ), stator tooth (R st ), stator tooth tip (R stt ), air-gap (R g ), rotor tooth tip (R rtt ), rotor tooth (R rt ), left rotor yoke (R ry1 ), and right rotor yoke (R ry2 ), as illustrated in Fig. 1.13. Note, air-gap reluctance is very large compared with the other core sections’ reluctances due to very low air permeability value. Hence, Kirchhoff’s voltage law (KVL) can then be used to calculate the magnetic flux linkage.

14 Fig. 1.13 Magnetic flux path representation of a WF synchronous machine section

Fig. 1.14 Magnetic flux path representation of a PM synchronous machine section

1 General Electric Machine Theory

1.4 WF and PM Synchronous Machine Excitation Fields

1.4.2

15

Magnetic Flux Path Representation of PM Synchronous Machines

Given the same concentrated winding assumptions for the PM machine topology as in WF machines, different cross-section path reluctances are calculated using (1.5). There are eight different reluctances and one rotor PM MMF, which is calculated as in (1.7), in the considered machine section. For the stator and air-gap of the PM machine section, the flux-linkage path reluctances are similar to those found in the WF machine case, while the PM rotor reluctances are represented by rotor PM (R m), left rotor yoke (R ry1 ), and right rotor yoke (R ry2 ), as illustrated in Fig. 1.14. Again, KVL can be used to calculate the magnetic flux linkage that will be shown in detail in Chap. 3.

Chapter 2

Hybrid Electric Machine Concept

2.1

History

As the demand for less expensive and more efficient electrified powertrain grows, the need for optimized electric machines becomes more apparent. An interesting electric machine topology that leads to simplified powertrains is hybrid excitation electric machines. In hybrid excitation machines, there exist two magnetic fields. This provides a flexible field control capability with an acceptable power density and without the need for an expensive power converter control system. Different methods of hybrid excitation field regulation topologies, including a PM combined with a WF excitation, have been considered in the literature [6–15, 16–39]. By combining PM and WF excitation, here referred to as hybrid PM (HPM) machine, the advantages of both PM and WF synchronous machines are utilized. The HPMs can be classified based on their magnetic excitation field paths (series or parallel) and based on their place in the machine stationary, rotary or both parts, as in Fig. 2.1.

2.2

Hybrid Electric Machine Classification

For HPM machine topologies, there are at least two excitation field sources that provide the net machine excitation. In general, a PM source provides the main excitation, and a wound field component acts to regulate the machine flux distribution either by boosting or by weakening the PM field depending on the direction of the wound field DC excitation current. The DC field winding may be placed on the rotor part of the machine as the PMs [22, 23, 33, 39], which necessitates slip rings and brushes or an exciter, or on the stator [17–22, 24, 26, 28, 31, 37–39].

© Springer Nature Switzerland AG 2022 A. S. Al-Adsani, O. Beik, Multiphase Hybrid Electric Machines, https://doi.org/10.1007/978-3-030-80435-0_2

17

18

2 Hybrid Electric Machine Concept

Fig. 2.1 General classification of HPM machines

Fig. 2.2 Cross sections of the permanent magnet synchronous machine with claw pole field excitation (PSCPF) [16]

2.3

Different Hybrid Machine Topologies

For the HPM machines to operate as a variable voltage generator, the range of air-gap flux density variation has to be designed to match the anticipated application requirements. A number of HPM machine topologies have been reported in the literature in recent years. The reported HPM machine topologies will be reviewed next. An assessment will be made for each topology with a view of arriving at a topology that will be studied in the following chapters.

2.3.1

PM Synchronous Machine with Claw Pole Field Excitation (PSCPF)

PM synchronous machine with claw pole field excitation (PSCPF) is briefly discussed by Zhao and Yan [16], where machine components and the associated flux linkages are detailed; Fig. 2.2 shows the machine cross section. The PSCPF is composed of two parts, one called the main part and the other the assistant part. Both

2.3 Different Hybrid Machine Topologies

19

Fig. 2.3 Flux path of the (PSCPF) machine as reported in [16]

Fig. 2.4 Simplified construction figure of HESG as reported in [17] (a) Axial section view (b) Radial section view

parts of the machine share one common stator. Referring to Fig. 2.2, the assistant part is composed of components 2–5; these represent the claw pole structure. The field winding is placed on the stator; therefore, slip rings and brushes are not required. When current flows through the field winding (component 5), the magnetic path of the DC flux is through the inner cylinder of component 3 (axial); the bottom of component 3 (radial); the outer cylinder of component 3 (axial); the air-gap δ1 (radial); plane magnet pole (axial); the main air-gap δ (radial); stator iron core (radial); air-gap δ (radial); claw pole magnet pole 2 (radial); magnetic shaft (axial); air-gap δ2 (radial); and inner cylinder of component 3. The magnetic path of the PM is through the claw pole magnet pole; air-gap δ (radial); stator iron core; air-gap δ (radial); claw-plane magnetic pole; PM (N pole); and rotor iron core and PM (S pole), as illustrated in Fig. 2.3. Zhao and Yan also discussed an improved PSCPF machine, referred to as the hybrid excitation synchronous generator (HESG), as illustrated in Fig. 2.4. It is

20

2 Hybrid Electric Machine Concept

Fig. 2.5 A new type hybrid excitation claw pole synchronous machine (HECPSG) components [40] Fig. 2.6 HECPSG machine assembly [40]

basically a similar structure to that of the PSCPF, the dissimilarity being that the latter has clapboard inserts that are made of nonmagnetic material. The clapboard introduces an air-gap and thus reduces the coupling between the PM and wound field excitation, making the two fields independent of each other. For both the PSCPF and HESG designs, the PM and wound field excitations act independently; that is, they are magnetically in parallel. In 2007, Chao-hui et al. [18] presented a study of a new HPM machine based on the HESG topology called the hybrid excitation claw pole synchronous generator (HECPSG). The structure of the HECPSG is shown in Figs. 2.5 and 2.6. The stator of the HECPSG consists of multiphase windings. The claw poles of the rotor are magnetized by a cylindrical wound coil and a cylinder-shaped permanent magnet, which is axially magnetized. The flux under one pole pair consists of two

2.3 Different Hybrid Machine Topologies

21

Table 2.1 HESPSG advantages and disadvantages Advantages (1) Good flux weakening [18] (2) The structure of the claw pole is helpful to arrange more magnet poles when the rotor diameter is relatively small [18] (3) Slip rings and brushes are not required [18]

Disadvantages (1) Rotor structure is relatively complex*

Field path Parallel

Note: Deduced by the author (*) Reported in the literature [18]

parts: one is produced by the permanent magnets and the other produced by the coil exciting current [18]. The magnetic field from one claw pole passes through the air-gap and stator core and back to another claw pole. No detailed discussion is given for the interaction between the PM and winding fields, that is, potential for demagnetization, heating effects, and reaction effects. Furthermore, the contribution from each field source to the stator-induced back-EMF is not discussed. Table 2.1 summarizes the advantages and disadvantages of the HECPSG topology.

2.3.2

Toroidal-Stator Transverse-Flux Machine (TSTFM)

Spooner et al. [19]. discussed hybrid excitation of AC and DC machines for rail traction and engine-mounted generators. Transverse-flux AC synchronous machines are excited by means of a simple DC coil mounted on the stator, as shown in Fig. 2.7a. Consequently, they are naturally brushless, they are reported to have low rotor losses (since the rotor has no permanent magnet poles), and they are mechanically suited to very high speed. However, the authors do not consider highfrequency losses that may occur in the solid rotor poles. The basic machine crosssection schematic is illustrated in Fig. 2.7a, consisting of two stator sections joined by a soft-magnetic outer casing and separated by the field coil. The rotor has two similar sections, one in each stator section and mutually displaced in space, in this case by 180 mechanical. Each rotor section has a salient structure, Fig. 2.7b. The field coil DC current establishes a set of north poles on rotor Sect. 1 and a set of south poles on rotor Sect. 2, as illustrated in Fig. 2.7b. Each stator coil encloses both stator core sections and experiences alternate north and south rotor poles as the rotor turns. The flux-linking of a stator coil is equivalent to that in a conventional radial field machine design of half the total core length [19] since there are empty spaces between the rotor soft-magnetic iron poles. A major problem for designers is the provision of sufficient magnetic material to carry flux between the two rotor sections. Furthermore, there is a substantial leakage flux when the stator sections are faced by the large effective air-gap of the “empty” or high reluctance rotor sections. Fixing magnets in the empty spaces of each rotor section, as shown in Fig. 2.7c, provides a pole opposite to those established by the field winding and enhances the

22

2 Hybrid Electric Machine Concept

Fig. 2.7 Transverse-flux machine components as reported by Spooner et al. [19] (a) Machine cross section (b) Machine soft-magnetic rotor (c) Machine rotor with saliency and permanent magnets

mechanical rotational symmetry (balance). The flux that passes through the machine shaft due to the permanent magnets is subtracted from that due to the excitation field current and so makes possible a greater flux-per-pole for each rotor section. The required field current can thus be reduced from the design of Fig. 2.7b, and leakage flux is also reduced [19]. Thus, transverse-flux machine arrangements appear to be an attractive option for small- and medium-size generators [19]. Spooner et al. [40] presented a rotary toroidal version of the transverse-flux hybrid excitation machine, based on the work of Evans and Eastham transverseflux AC machine topology. The machine construction is illustrated in Fig. 2.8, showing a toroidal wound stator core of multiphase windings, DC field winding located inside the toroidal core, and two rotating discs with alternate permanent magnet and soft-magnetic poles. The flux-linkage paths throughout the machine parts due to both the PM’s and stationary field coil are illustrated in Fig. 2.9. If the two rotor poles are only provided by PMs, the flux path can be traced from one rotor plate containing north pole magnets, crossing the air-gap into the toroidal stator, and then traveling circumferentially across the second air-gap into the south magnet pole on the opposite plate, through the plate into the shaft and back to the first plate to

2.3 Different Hybrid Machine Topologies

23

Fig. 2.8 Toroidal transverse-flux machine reported by Spooner et al. [19]

close the loop at the north pole [20], as shown in Fig. 2.9a. A modification to the design of Fig. 2.9a has soft-magnetic poles between the respective north and south PM poles, as illustrated in Fig. 2.11b [19], resulting in additional flux paths. Thus, flux from the north pole on the right-hand side plate crosses to the stator but then comes back to the same rotor disc via the soft iron pole [20], as shown in Fig. 2.9b. In this case, flux does not generally pass through the rotor shaft. However, during the operation of the machine, flux travels through both paths, subject to reluctance variation in the shaft. Finally, there is a third flux path due to the field excitation coil that drives flux through the rotor shaft, rotor plate, iron poles, air-gap, stator, and the second iron poles on the opposite disc [20], as illustrated in Fig 2.9b, c for both strengthening and weakening modes, respectively. The toroidal transverse-flux machine configurations are brushless machines generating an AC output that is modified by the DC field winding excitation current [21]. For both transverse-flux topologies illustrated in Figs. 2.7, 2.8 and 2.9, the main PM field and moderating wound field are magnetically in parallel, their advantages and disadvantages being noted in Table 2.2.

2.3.3

Hybrid Excitation Synchronous Machine (HESM)

Naoe and Fukami discussed the structure of a hybrid excitation synchronous machine (HESM) [22]. The machine has a conventional AC stator and a two-part rotor construction where each part is separated by an air-gap. One rotor part has PM excitation and the other part wound field excitation. Each rotor part is independent of the other and, in the case reported, is of radial field design. The HESM is illustrated schematically in Fig. 2.10. The flux produced by the field winding is designed not to pass through the PMs because of their large reluctance, thus keeping the field

24

2 Hybrid Electric Machine Concept

Fig. 2.9 Flux paths in the toroidal transverse-flux machine, as reported by Spooner et al. [19]. (a) Flux paths due to PMs alone; without rotor iron poles (b) Flux paths due to both PMs and DC field excitation in strengthening mode; with rotor iron poles (c) Flux paths due to both PMs and DC field excitation in weakening mode; with rotor iron poles

winding MMF low [22]. Hence, the machine air-gap flux can be modified by the field winding current direction and magnitude. The PM and rotor wound field excitation sources are magnetically in parallel. Table 2.3 summarizes the advantages and disadvantages of the HESM topology.

2.3 Different Hybrid Machine Topologies

25

Table 2.2 Advantages and disadvantages of toroidal-stator transverse-flux machine topologies Advantages (1) Control is relatively simple [19] (2) The short axial length makes this machine suitable for directly mounting to an engine shaft replacing, in part, the flywheel [19] (3) Slip rings and brushes are not required [19]

Disadvantages (1) The magnetic path of the electrical excitation is relatively large, which necessitates relatively high excitation MMF* (2) Mechanics are complex* (3) The design magnetic field of the toroidal is restricted by the machine diameter [19]

Field path Parallel

Note: Deduced by the author (*) Reported in the literature [19] Fig. 2.10 Structure of the HESM with a two-part rotor [22]

Table 2.3 Advantages and disadvantages of the HESM Advantages (1) Machine is simple in structure [22] (2) Short magnetic path [22] (3) The air-gap flux can be easily controlled by the field current [22]

Disadvantages (1) Slip rings and brushes exist, which increases complexity and maintenance costs*

Field path Parallel

Note: Deduced by the author (*) Reported in the literature [22]

2.3.4

Synchronous Permanent Magnet Hybrid AC Machine (SynPM)

The synchronous permanent magnet hybrid AC machine (SynPM) was presented by Xiaogang and Lipo [23]. The machine is a combination of four PM poles and two wound field excitation poles on the same rotor, as illustrated in Fig. 2.11. The PM poles provide the major part of air-gap flux, while the wound field excitation poles act as a flux regulator to adjust the air-gap flux distribution. By appropriate connection of the stator coils and rotor winding excitation, the net phase flux linkage and hence back-EMF may be weakened or strengthened. Considering one of the stator

26

2 Hybrid Electric Machine Concept

Fig. 2.11 Cross section of the of SynPM machine reported by Xiaogang and Lipo, showing one phase belt of the stator winding [40]

Fig. 2.12 Back-EMF of one coil of the phase belt winding [40]

phase belt coils, the coil back-EMFs for the three excitation modes are as shown in Fig. 2.12, while Fig. 2.13 illustrates the corresponding open circuit flux lines due to positive, zero, and negative DC field currents. A phase belt is formed by connecting three coils of the same phase in series, as shown in Fig. 2.11; thus, the resulting phase back-EMFs for the cases of positive, zero, and negative field winding current are as shown in Fig. 2.14. Slip rings and brushes are required for this machine topology. For the machine discussed, excitation produces around 67% of the total air-gap flux [23]. The flow of the flux is radial for both PM and DC field windings, which are magnetically acting in parallel. Table 2.4 summarizes the advantages and disadvantages of the SynPM topology.

2.3 Different Hybrid Machine Topologies

27

Fig. 2.13 Flux lines of the six-pole SynPM machine presented by Xiaogang and Lipo [40]. (a) Full positive field current (b) Zero field current (c) Full negative field current

2.3.5

Consequent Pole Permanent Magnet Hybrid Excitation Machine (CPPM)

Tapia et al. discussed a consequent pole permanent magnet hybrid excitation machine [24–25]. The machine combines fixed PM excitation with variable flux via a field winding fixed in the stator. The machine is similar to the transverse-flux machine reported by Spooner et al. [19]. However, Tapia et al. discussed a greater number of design options and discussed the design in greater depth. The machine consists of a rotor divided into two sections, each section having radially magnetized surface-mounted permanent magnets interleaved with laminated iron poles, as illustrated in Fig. 2.15a. The magnetization of each rotor section is shifted 1-polepitch with respect to the other section. The stator is composed of two laminated tooth sections inside a solid outer soft magnet yoke. A conventional three-phase AC winding is located in slots around the periphery of the inner stator diameter, and a circumferential field winding is placed between the two stator sections, as illustrated in Fig. 2.15a. The field winding is excited by DC current. For no field current, the machine excitation is due to the rotor PMs alone and is essentially radial, each PM linking with a consequent soft iron pole on the same machine half. When excited with positive current, the flux generated by the field winding flows in a direction such that it adds to the PM flux and the flux closes its path in the same half stator, as illustrated in Fig. 2.15b. If the field current is negative, the direction of the air-gap flux is as shown in Fig. 2.15c. Figure 2.15d shows further views of the CPPM components. The stator and rotor yokes provide a low reluctance path for the axial flux, which is considered an important feature of the machine operation. The current of the field winding is externally controlled in order to provide variable excitation.

28 Fig. 2.14 Example of resultant coil and phase back-EMF for different field winding excitation conditions [40]. (a) BackEMF of one circuit with full positive excitation (b) BackEMF of one circuit with zero excitation (c) Back-EMF of one phase with full negative excitation

2 Hybrid Electric Machine Concept

2.3 Different Hybrid Machine Topologies

29

Table 2.4 Advantages and disadvantages of the SynPM Advantages (1) Machine is comparatively simple in structure* (2) In addition, it is easy to fabricate short magnetic paths. A high power density is suggested, but no data are quoted*

Disadvantages (1) Slip rings and brushes exist [23] (2) The combination of four-pole or two-pole field flux in field weakening, with the six-pole stator flux, will result in a number of space and time-harmonic components and undesirable torque pulsations and vibration [23] (3) At high speed, when the field is weakened, a high iron loss in the stator might appear*

Field path Parallel

Note: Deduced by the author (*) Reported in the literature [23]

Fig. 2.15 Consequent pole PM hybrid excitation machine (CPPM) [40]. (a) Magnetic structure of the CPPM machine [25] (b) Magnetizing effect of the field flux (c) Demagnetizing effect of the field flux (d) 3 kW CPPM prototype

30

2 Hybrid Electric Machine Concept

Fig. 2.16 Field controlled Torus-NS machine (FCT-NS) [40]. (a) Machine components (b) Stator assembly (c) Rotor assembly

2.3.6

Field Controlled Torus-NS (FCT-NS) Machine

Aydin et al. discussed an axial flux machine designed to improve the flux weakening operation of the previously reported axial flux, toroidal PM machines [26]. The machine is essentially an axial field version of the CPPM and was referred to as the field controlled Torus-NS (FCT-NS) machine. The machine construction consists of two outer rotor discs carrying axially magnetized permanent magnets alternatively placed with slotted magnetic iron pole pieces. There are two slotted stator cores, an inner and outer core, realized by tape wound laminations inserted with multiphase AC windings and a DC field winding between the stator inner and outer cores, as illustrated schematically in Fig. 2.16. Variations on the FCT-NS design were presented by Lipo and Aydin [27, 28]. Figure 2.17 shows the main flux direction of a two-pole portion of the FCT machine at the average diameter [26] (a); rotor flux directions (b); air-gap flux directions (c); and operating principle of the FCT machine (d) for zero (i), positive (ii), and negative (iii) field current. Figure 2.17e shows the FCT stator and rotor components. Figure 2.18 illustrates schematics of the single-rotor-single-stator topology (a); the NN- and NS-type double-rotor-single-stator (b and c); doublestator-single-rotor (d); and multistage (e) concepts. Figure 2.19 illustrates the hardware of the NN-type FCAFPM machine as reported in the literature. The CPPM and variants are all parallel permanent magnet and wound field magnetic designs. Table 2.5 summarizes the advantages and disadvantages of the CPPM and variants as reported in [24–29].

2.3.7

Dual-Rotor Machine

Amara et al. proposed a dual-rotor machine that is composed of two rotors placed together (one wound and the other with PMs) inside the same stator assembly, as shown in Fig. 2.20. The design employs juxtaposed magnetic circuits that, according

2.3 Different Hybrid Machine Topologies

31

Fig. 2.17 Field controlled Tours-NS type (FCT-NS) [40]. (a) Main flux direction of the FCT machine [26] (b) Rotor flux directions [26] (c) Air-gap flux directions (d) Operating principle (e) FCT rotor and stator components

to the authors, avoids the risk of PM demagnetization [30]. Flux weakening is achieved via excitation of the wound rotor to create a flux opposite to that created by the rotor PMs [30]. The design is similar in form to the HESM presented in Sect. 2.3.3 [22] but having slightly different rotor topologies.

32

2 Hybrid Electric Machine Concept

Fig. 2.18 Reported combinations of the FCAFPM machines [40]. (a) Single-rotor-single-stator (b) NN-type double-rotor-single-stator (c) NS-type double-rotor-single-stator (d) Double-stator-singlerotor (e) Multistage

Fig. 2.19 NN-type FCAFPM machine reported in [40]. (a) Stator view pre-impregnation (b) Complete stator assembly (c) Rotor assembly

2.3.8

Imbricated Hybrid Excitation Machine (IHEM)

Amara et al. also proposed an imbricated hybrid excitation machine (IHEM), as illustrated in Fig. 2.21. The rotor is composed of two magnetically isolated parts, one containing the PM excitation, and the other is used to direct flux created by an

2.3 Different Hybrid Machine Topologies

33

Table 2.5 Advantages and disadvantages of CPPM and variants as reported in [24–29] Advantages (1) Control of the CPPM is convenient (2) The air-gap flux can be controlled without affecting the magnetization characteristics of the PMs (3) A wide range of air-gap flux control can be obtained with a low DC excitation field ampere-turn requirement (4) Slip rings and brushes are not required

Disadvantages (1) Additional DC winding in the stator reduces the power density, such that the additional air-gap surface associated with this winding does not participate in the energy conversion process

Field path Parallel

Fig. 2.20 Dual-rotor machine [40]

Fig. 2.21 Imbricated hybrid excitation machine (IHEM) [40]. (a) Machine cross section (b) Rotor structure

excitation coil that is located on either the rotor or the stator, the latter case avoiding all sliding contacts. The stator is composed of two identical parts linked by a yoke, as shown in Fig. 2.21a. The main goal of this design was to ensure that the flux created by the excitation winding does not pass through the PM; hence, the possibility of demagnetization is greatly reduced [30].

34

2 Hybrid Electric Machine Concept

Furthermore, Vido et al. proposed two improved versions of the IHEM [31], the (i) homopolar and (ii) bipolar hybrid excitation synchronous machines, HHESM and BHESM, respectively, as illustrated in Fig. 2.22. Cross-sectional schematics of both prototypes are shown in Fig. 2.22. The rotors consist of three parts, one a solid core, one part laminated core, and a set of permanent magnets. The schematics show an axial cut of the stator and rotor for both prototypes, which are six-pole pairs. The two machine rotors have the same dimensions. By comparing the two topologies, it can be observed that the lateral permanent magnets are not present in the BHESM prototype [31]. The various flux paths created by excitation coils, lateral PMs (side magnets), and azimuth PMs may be divided into two categories: homopolar and bipolar flux paths. The homopolar flux path represents a flow of flux through machine parts in axial and radial directions. The bipolar design has flux paths in either radial or axial direction. Therefore, the flux generated by the field DC coils has only one path, which is homopolar in nature, as shown in Fig. 2.23a. Moreover, the homopolar path for the lateral PMs can be observed in Fig. 2.23a. The flux generated by the PMs has two distinct paths, one of which is bipolar, as shown in Fig. 2.23b, c, which creates north and south poles under the active parts [32]. The flux path generated by the azimuth PMs is primarily oriented perpendicular to the axial direction of the machine [32]. A portion of the flux generated by the lateral PMs is oriented in the axial direction of the machine via the rotor flux collector, as shown in Fig. 2.23c. In other words, the fluxes created by either the PMs or the wound field excitation that exhibits a homopolar path only give rise to one type of pole (either north or south), depending on the direction in which the magnets are magnetized and the polarity of current in the DC field coils [32]. Flux only passes once through the air-gap under the active part, and then it returns first via the stator end shields and then via the rotor flux path, as illustrated in Fig. 2.23d [32]. Figure 2.24a shows flux paths created by the DC field coils for the BHESM design. This bipolar configuration passes through two annular excitation coils. Each coil acts in one polarity of pole [31]. The flux created by an excitation coil goes through active parts and an air-gap (homopolar path). Figure 2.24b shows the bipolar flux path created by PMs, where this bipolar flux passes through active parts and air-gap, creating north and south poles. Figure 2.24c shows the PM leakage flux path, which is not through the active parts and hence does not contribute to torque production [31]. Figure 2.25 shows homopolar flux paths created by PMs, as reported in [31]. For homopolar hybrid excitation machines, the total flux passing through the stator windings exhibits a DC component, while for bipolar hybrid excitation machines, the total flux passing through the armature windings does not have a DC component [31]. Thus, although air-gap flux control is effective for both the HHESM and BHESM machines, the DC current excitation efficiency is better for the HHESM because of the solid rotor core parts [31]. For the HHESM operating with enhanced excitation flux, magnetic saturation occurs when the magnetic pole in which the DC excitation is acting is saturated, even if the other pole is still not saturated [32]. However, for the BHESM, magnetic saturation occurs only when both magnetic poles are saturated, from which the authors conclude that the BHESM has a wider excitation flux variation [32]. The efficiency of the hybrid

2.3 Different Hybrid Machine Topologies

35

Fig. 2.22 Homopolar and bipolar hybrid excitation synchronous machines [40]. (i) Schematic (i) Schematic (ii) Prototype rotor details (ii) Prototype rotor details (iii) Prototype stator and rotor (iii) Rotor laminations (a) First prototype machine (HHESM) (b) Second prototype machine (BHESM)

36

2 Hybrid Electric Machine Concept

Fig. 2.23 HHESM various flux paths due to deferent excitations [40]. (a) Homopolar flux path due to DC coils (b) Bipolar flux path (azimuthal magnets) (c) Bipolar flux path due to PMs (d) Homopolar flux path due to PMs

excitation is better for the HHESM than it is for the BHESM design because of the leakage flux path, as shown in Fig. 2.24c [31], which does not contribute to torque production. Table 2.6 summarizes the advantages and disadvantages of the IHEM topology.

2.3.9

Series Double Excited Synchronous Machine (SDESM)

Fodoren et al. present the series double excited synchronous machine (SDESM) that has series excitation circuits [33, 34]. The parallel excitation circuit reported in some of the previously presented topologies suffer from the drawback of construction complexity [33]. The main advantage of the SDESM appears in applications where

2.3 Different Hybrid Machine Topologies

37

Fig. 2.24 BHESM various flux paths created by different excitations [40]. (a) Flux paths created by DC field coils (b) Bipolar flux path created by a rotor PM (c) Leakage flux path created by PM

Fig. 2.25 Homopolar paths of fluxes created by PM [40]. (a) First homopolar path (b) Second homopolar path Table 2.6 IHEM advantages and disadvantages Advantages (1) The flux created by the excitation winding does not pass through the PM [30] (2) The risk of magnet demagnetization is reduced [30] (3) Slip rings and brushes are not required [30]

Disadvantages (1) Machine is complex* (2) Substantial amount of magnetization leakage*

Field path Parallel

Note: Deduced by the author (*) Reported in the literature [30]

the electric drive operates under partial loads for most of the time [33]. Fodoren et al. presented a design procedure, prototype, and test results for a SDESM design in [33]. The proposed SDESM design has the field excitation winding fixed on the rotor in a series magnetic configuration with the surface PMs, as shown in Fig. 2.26. The

38

2 Hybrid Electric Machine Concept

Fig. 2.26 SDESM design presented by Fodoren et al. [40]. (a) SDESM cross section, basic principle (b) SDESM cross section, actual design

Fig. 2.27 Hardware of SDESM [40]. (a) SDESM rotor (b) Test bench

stator is that of a commercial induction motor, while the rotor was constructed as shown in Fig. 2.27a. The three-phase stator winding is single-layer, with three slotsper-pole-per phase [33]. Table 2.7 summarizes the advantages and disadvantages of the SDESM topology as reported in [33].

2.3 Different Hybrid Machine Topologies

39

Table 2.7 SDESM advantages and disadvantages [33] Advantages (1) Construction is simple (2) Good flux weakening with reduced demagnetization risk (3) Reduced iron losses in extended speed operating region

Disadvantages (1) Rotor sliding contacts to rotor DC excitation field

Field path Series

Fig. 2.28 Cross section and actual switch reluctance motor with field assistance [40]. (a) Motor/ generator assembly (b) Cross section of the motor/generator (c) Motor/generator hardware

2.3.10 Switch Reluctance Machine with Stator Field Assistance Afjei et al. presented a new configuration of switch reluctance machine with stator field assistance that represents a hybrid generator topology, albeit with no PMs, as illustrated in Fig. 2.28 [36, 37]. This machine design was intended to be utilized in a hybrid vehicle motor/generator unit. The proposed hybrid machine consists of two

40

2 Hybrid Electric Machine Concept

stator and two rotor sections placed on both sides of the field coil assembly, as illustrated in Fig. 2.28b [36]. Here, the magnetic flux produced by the field coil travels through the guide and shaft to the rotor poles and then to the stator poles, finally closing through the motor housing [36]. A variant of the Afjei design that incorporates PMs was presented by Chau et al. for small wind applications [38]. This hybrid design has a unique structure that, which it is claimed, contributes to simplified mechanical manufacturing and magnetic fixing. The machine design is illustrated in Fig. 2.29 [38]. Due to the extra air-bridge that is in parallel with each PM, an amplification of the effect of the flux in the reinforcing mode is achieved, as illustrated in Fig. 2.29b, where the field winding MMF is opposing the PM MMF. The PM flux leakage will increase, causing an amplification of the effect of the flux weakening, as shown in Fig. 2.29b [38]. Thus, as with a proper design of the air-bridge width, a wide flux regulating range can be obtained by virtue of a small DC field excitation [38]. The PM and rotor wound field excitation sources are magnetically in series. Table 2.8 summarizes the advantages and disadvantages of the PM brushless hybrid generator topology.

2.3.11 Dual-Stator Hybrid Excited Synchronous Wind Generator (DSHESG) Liu et al. presented a novel dual-stator hybrid excited synchronous wind generator (DSHESG), as illustrated in Fig. 2.30 [39]. The proposed generator stator is composed of outer stator, inner stator, and field winding. The rotor consists of the PMs, claw poles, rotor yoke, and cup rotor [39]. There are two independent parallel magnetic circuits in the DSHESG due to PMs and the DC excitation coil. Here, the series PM magnetic circuit consists of PMs, air-gap, cup rotor, and laminated stator core. The rotor series magnetic circuit consists of claw poles, air-gap, laminated core of the outer stator, and bracket of the field winding. It is claimed that this topology overcomes some of the previous hybrid machine topologies’ weakness, for example, the benefit of the two independent magnetic circuits, which reduces the leakage flux problems, and there is a reduced risk of PM demagnetization. The PM and rotor wound field excitation sources are magnetically in parallel. Table 2.9 summarizes the advantages and disadvantages of the DSHESG topology.

2.4

Summary of Surveyed Literature on HPM Machines

The review of publications has highlighted a number of notable issues. The published HPM machine designs are, in general, complicated and have particular weaknesses in their design, for example, excessive flux paths that lead to high leakage (magnetomotive force loss) [21, 26, 33] and PM demagnetization

2.4 Summary of Surveyed Literature on HPM Machines Fig. 2.29 Cross section, magnetic field distributions, and actual prototype PM brushless hybrid generator [40]. (a) Generator cross section (b) Magnetic field distributions under different DC excitations (c) Prototype generator

41

42

2 Hybrid Electric Machine Concept

Table 2.8 PM brushless hybrid generator advantages and disadvantages Advantages (1) Wide flux regulating capability with small DC field excitation [38] (2) The rotor has neither PMs nor field windings, which offers high mechanical integrity [38] (3) Slip rings and brushes are not required [38]

Disadvantages (1) There is a risk of magnet demagnetization* (2) There is a risk of stator teeth saturation* (3) There is a substantial amount of flux leakage*

Field path Parallel

Note: Deduced by the author (*) Reported in the literature [38]

Fig. 2.30 Detailed structure and magnetic circuits of DSHESG [40]. (a) Structural 3D FEA model (b) Structure cross section (c) Magnetic circuit of PM (d) Magnetic circuit of DC field winding (e) Actual prototype

2.4 Summary of Surveyed Literature on HPM Machines

43

Table 2.9 DSHESG advantages and disadvantages Advantages (1) Good field control capability [39] (2) Reduced leakage flux problems [39] (3) Low risk of magnet demagnetization [39]

Disadvantages (1) Machine complex assembly* (2) Slip rings and brushes exist*

Field path Parallel

Note: Deduced by the author (*) Reported in the literature [39]

[33]. Table 2.10 collects data for the different hybrid excitation machines reported in the literature review. Note that Table 2.10 is not complete due to the lack of some information in the published papers. The open-circuit back-EMF regulation capability that has been reported for some of the HPM machine designs is generally based on experimental or finite-element analysis (FEA) results. The variation of opencircuit back-EMF due to the machine DC field current of the reported HPM machine topologies varies from 42 to 175% relative to the machine back-EMF with zero DC field excitation. However, Table 2.10 does not give a clear picture regarding the best machine topology for vehicle application; thus, in order to get an adequate and fair comparison, the machine mass, volume, and thermal limits should be considered along with their performance and back-EMF regulation capabilities. Furthermore, the power rating of some of the actual HPM machine prototypes discussed in the literature ranged from 0.65 to 10 kW, as detailed in Table 2.10. Some of the publications highlighted the area of interest for HPM machine topologies, for example, traction, wind power, and vehicle systems.

–

HESM SynPM

CPPM

Note: Experimental (+) FEA (*)

DSHESG

Switch reluctance mot. With field assistance generator Brushless PM hybrid generator

HHESM BHESM SDESM

300 +600 –

–

( 0.8) to (0.8)

–

0 to 1

86+ –

–

Unclear

24.5+

53+

Unclear

–

–

(0) to (24)

30+

70+

–

( 7) to (5)

IHEM

24+ 125+

18+ 50+

54+

0 –

( 18) to (18) ( 0.7) to (1.8)

44+

25+

– –

–

1200

500

25+

– –

–

138+

Unclear

–

100+

42+ 175+

87*

98+

50+

– –

–

– 27+

Control range (%) 66* –

No-load back-EMF regulation capability Bucking Boosting mode (%) mode (%) 33* 33* – –

–

–

FCTPM

–

– –

– –

TSTFM

300

+3000

(0) to (2)

Topology HECPSG TSTFM

DC field Ampere-turns (A-t) – –

DC field current range (A) ( 20) to (20) –

Table 2.10 Main particulars of HPM machines presented in research publications

–

–

–

5.5

3

1.34 –

10

3

3

– –

5

Rating (kW) 0.65 10

Wind power

Wind power

Vehicles

–

– Traction system –

–

–

–

– –

Area of interest – Fixed on an engine shaft –

Liu [39]

Aydin [26, 29] Aydin [28] Amara [30] Vido [31, 32] Fodoren [33] Afjei [36, 37] Chau [38]

Tapia [25]

References Zhao [18] Spooner [19] Novinschi [20] Naoe [22] Xiaogang [23] Tapia [24]

44 2 Hybrid Electric Machine Concept

Chapter 3

Hybrid Permanent Magnet Machine Design

3.1

Introduction

The multiplicity of machine designs reviewed in Chap. 2 creates a certain level of difficulty when selecting the most appropriate HPM machine design topology since each author declared some degree of novelty but did not reference their design against any benchmark solution. It was anticipated from previous research that many future HPM machine applications would be direct engine-mounted having a large outer diameter to active axial length aspect ratio and possible facility for some through-shaft element (for flywheel or multiple geared outputs) necessitating an essentially “donut” shaped volume envelope constraint [41–43]. Consequently, an available 3-kW, surface magnet mounted, brushless permanent magnet (PM) machine having the above volumetric attributes was chosen as a benchmark PM design from which to develop a suitable HPM generator in this book since the average power demand of 3 kW is typical of small 1.0 to 1.2 ton urban vehicles. This chapter discusses the design and analysis of the benchmark brushless PM machine and testing thereof to validate the calculation models and procedures. The developed tools were then used to analyze and design WF rotors within the same stator constraints and with an adequate axial length split ratio of the brushless PM machine, from which a preferred rotor design solution was chosen. Thus, a machine that combined the brushless PM and chosen WF design features was proposed. This HESM structure, as discussed in Chap. 2 (Sect. 2.3.3), was compared along with three other HPM machine topologies (all designed around the benchmark brushless PM machine) to attempt a comparative analysis of the four competing topologies. Conclusions are presented, and the chosen HPM machine design solution is subsequently adopted for further study in Chaps. 4, 5 and 6.

© Springer Nature Switzerland AG 2022 A. S. Al-Adsani, O. Beik, Multiphase Hybrid Electric Machines, https://doi.org/10.1007/978-3-030-80435-0_3

45

46

3.2

3

Hybrid Permanent Magnet Machine Design

Machine Volume Envelope Consideration

The proposed and investigated hybrid generator, in this book, is designed as a direct engine-mounted machine having a large outer diameter to active axial length aspect ratio similar to the engine-PM machine system layout shown in Fig. 3.1 [44]. The radial flux hybrid machine topology consists of two different synchronous machines: PM and WF machines coupled on the same rotor shaft. Based on the application volume envelope constraint, the design ensures (i) similar stator winding configuration and geometry for both PM and WF sections, (ii) similar air-gap flux-density waveform for both sections of the machine, and (iii) HPM machine adequate rating and axial length based on buck/boost capability of the WF rotor. Consequently, using available laminations, a three-phase, 3 kW, surface magnet mounted, brushless PM machine having the set volumetric attributes, as will be discussed in the next subsections, is designed as a benchmark machine from which to develop a suitable HPM generator rating to meet a 3–4 kW average power demand for small (1.0–1.2 ton) vehicles [45].

Fig. 3.1 Direct engine-mounted PM synchronous generator layout [44]

3.2 Machine Volume Envelope Consideration

3.2.1

47

PM Machine Dimensions

As stated earlier, direct engine coupling with a large outer diameter to active axial length aspect ratio and possible facility for some through-shaft element (for flywheel or multiple geared outputs) necessities a “donut” shaped machine volume envelope. Consequently, an available 3-kW, surface magnet brushless permanent magnet (PM) machine having the required volumetric attributes was chosen as a benchmark PM machine design. The benchmark brushless PM machine is illustrated in Fig. 3.2, showing main dimensions and a photograph of the machine assembly. Table 3.1 lists the main specification of the benchmark PM machine.

3.2.2

PM Machine Stator Winding Layout

The stator winding details of the benchmark PM machine are given in Table 3.2. A phase belt (i.e., number of slots per pole per phase) for the three-phase PM stator Dso hsy Wst hstt DPMri

Wstt hry

hst τst

τPMrp

Winding

Stator back-iron

Tooth hgPM LM

Rotor back-iron

LPMa

Magnet

(a)

(b) Fig. 3.2 Benchmark PM Machine (a) Main Dimensions (b) Machine Assembly

48

3

Hybrid Permanent Magnet Machine Design

Table 3.1 Main design dimensions of benchmark brushless PM machine Stator outer diameter (Dso) Stator inner diameter (Dsi) Stator yoke thickness (length) (hsy) Stator tooth width (Wst) Stator tooth tip width (Wstt) Stator tooth thickness (hst) Stator tooth tip thickness (hstt) Spacing between stator teeth (at tooth tip) (τst) No. of stator slots (S) Half stator slot area (Ass) Rotor outer diameter (DPMro) Rotor inner diameter (DPMri) Rotor yoke thickness (length) (hry) Magnet thickness (LM) Magnet material Spacing between rotor poles (at pole tip) (τPMrp) Number of rotor poles (P) Air-gap thickness (hgPM) Active axial length (LPMa)

230 167.2 13 9.2 12.4 16.5 1.9 2.42

mm mm mm mm mm mm mm mm

36 60.6 165.6 130 15 2.8 NdFeB 1.8 32 0.8 25

–
1026 m2 mm mm mm mm 37 MGOe mm – mm mm

Table 3.2 Stator winding details of benchmark brushless PM machine No. of phases (n) No. stator slots/rotor poles (nsp) Coils per pole per phase (ncpp) Slot pitch (Sρ) Slot span (Sγ) Pole pitch (Pρ) PM pole span (PPMγ ) Coil pitch (Cρ) Coil span (Cγ ) Conductor diameter (Dc) (mm) No. of parallel conductors per single turn (nPc) No. of stator winding layers (nl) No. of turns per coil (nstc) No. of series coils per phase (nsc) No. of turns per phase (nst)

3 36/32 0.375 160 elec. ¼ 10 mech. 1 180 elec. ¼ 11.25 mech. 10.11 mech. 180 elec. 1 slot 0.56 12 2 6 12 36

winding is connected as 3/8; hence, a concentric winding with 36 coils wound around stator teeth is realized. Each phase has four sets of three concentrated coils connected in series as illustrated in Fig. 3.3a. Note that the middle coil of the three concentrated coils is wound opposite to the other two coils, which results in the largest amplitude back-EMF in the three-phase winding. In Fig. 3.3b, the

3.2 Machine Volume Envelope Consideration

49

Fig. 3.3 Three-phase PM machine stator lamination and phase (A) winding scheme (a) A phase winding configuration (b) Phase back-EMF configuration vectors

Phase A winding start

Phase A winding start

Phase AA Phase winding end winding end

(a)

E1 2 2

Stator slot numbers

Stator slot numbers

3 3 4 10 11 11

12 12 13 19 20 20

3-phase winding

21 21 22 28 29 29

kw1= 0.9455

30 30 31

(b) fundamental winding factor (kw1) is calculated using the methods described in [46], which incorporates both the slot pitch and distribution factor in one equation. Therefore, by numbering the stator slots using vector U, the phase back-EMF vectors (Ei) and kw1 are calculated as follows in (1, 2, 3, 4) [46].

50

3

Hybrid Permanent Magnet Machine Design

! E i ¼ signðU ðiÞÞ:e

kw1

jπP S jU ðiÞj

ð3:1Þ

  nl S=3  1  X ! ¼ E  nl S=3  i¼1 i 

ð3:2Þ

U ¼ ½1  2  2 3 3  4 10  11  11 12 12  13 19  20  20 21 21  22 28 29  29 30 30  31  36   X 2
3 !  E i  ¼ e   i¼1 

jπP S



1  2e πP

jπP S

ð3:3Þ

πP

πP

πP

πP

πP

πP

πP

þ 2e2j S  e3j S þ e9j S  2e10j S þ 2e11j S  e12j S þ e18j S πP

πP

πP

πP

πP

πP

2e19j S þ 2e20j S  e21j S þ e27j S  2e28j S þ 2e29j S  e30j S Þ

ð3:4Þ

where P, S, and nl are the number of poles, slots, and stator layers. The three-phase fundamental winding factor determines the machine back-EMF voltage waveform peak. Note that, from machine stator winding design point of view, as the fundamental winding factor approaches 1, a better back-EMF voltage utilization is achieved.

3.2.3

Stator Winding Fill Factor and Resistance

A mechanical or electrical slot fill factor is an important parameter in machine stator winding design. The mechanical slot fill factor accounts for the conductor wire with its insulation layer, where the electrical slot fill factor accounts for the conductor wire without the insulation layer. Filling the slot between stator teeth with a desired number of conductors and with an adequate conductor cross-sectional area, as in Fig. 3.4, contributes to compact and optimized machine design through minimizing the coil resistance, which in turn decreases the developed heat and core loss during operation. The three-phase back-EMF voltage generation is governed by rotational speed, flux, and number of turns, as in (1.10); hence, the required number of turns can be calculated by specifying the desired speed and phase voltage, assuming a constant flux lines passing in the stator teeth due to either rotor PM or wound field DC excitation. Winding space in machine slot design goes through an iterative process to find a slot area that would accommodate the required number of turns with a certain conductor cross-section area that produce a minimum resistance. However, for a given stator lamination, the mechanical or the electrical slot fill factor is obtained by knowing the following:

3.2 Machine Volume Envelope Consideration

51

Fig. 3.4 Schematic for electrical fill factor determination of a stator slot

(i) (ii) (iii) (iv) (v) (vi)

Slot-liner (insulation paper) thickness. Single conductor diameter with varnish coat (wire with insulation layer) (Dc). Parallel or single conductor area with varnish coat (Ac). Effective conductor diameter (wire without insulation layer) (Dec). Parallel or single effective conductor area (Aec). Slot winding area (half of the stator slot area with slot-liner) (Ass).

A stranded wire is recommended for stator winding of small electric machines to achieve the required Ac with bendable conductors, which introduces the number of parallel conductors per turn (nPc) term, in the effective or noneffective conductor area calculation, as in (3.5). By using machine AutoCAD drawing, the slot winding area can simply be determined, and both the mechanical and electrical slot fill factors (kmf, kef) are respectively calculated as in (3.6). Ac ¼ 0:5πD2c nPc Aec ¼ 0:5πD2ec nPc

k mf ¼

nstc Ac Ass

k ef ¼

nstc Aec Ass

ð3:5Þ

ð3:6Þ

Note, as the effective conductor diameter per single turn increases, the coil resistance decreases. Using the effective stator slot fill factor, the phase winding resistances (Rs) for the benchmark PM machine are calculated using (3.7) and (3.8),

52

3

Hybrid Permanent Magnet Machine Design

where τrewr is the approximated stator mean end winding radius and Lsc is the copper length per stator coil. Lsc ¼ nstc ð2L þ 2πτsewr Þ

Rs ¼

3.2.4

ð3:7Þ

nsc nstc Lsc σkef Ass

ð3:8Þ

Finite Element Method Program

There are a number of tools that have been developed over the years to investigate and predict electric machine behavior in linear and nonlinear modes. Here, rotary machine analysis is achieved by utilizing the finite-element numerical techniques [47]. FEMM is a commercial software package that can solve several low-frequency electrostatic and electromagnetic problems in a 2D plane. In this chapter, the various machine geometries are analyzed using their equivalent 2D FE models. Before embarking on actual machine hardware assembly and testing, FEMM is used to analyze the proposed machine construction, changes to geometry, parameters, injected currents, coil number of turns, and so on to obtain the adequate variables, which satisfy the desired mode of operation. In addition, while the selected HPM machine topology design meets certain design criteria, such as low saturation and demagnetization risks, a quantitative comparison of HPM machine topologies considered most likely to be applicable for high-volume automotive manufacture is investigated within the same volumetric and slot current density constraints of that of the reference PM machine design. Magnetostatic problems are problems in which the fields are time-invariant. In this case, the field intensity (H ), flux density (B), and current density (J ) must obey (3.9) and (3.10) [42]: J ¼Δ
H

ð3:9Þ

Δ∙B ¼ 0

ð3:10Þ

subject to a constitutive relationship between B and H for each material [48]: B ¼ μH

ð3:11Þ

If the material is nonlinear (saturating iron or permanent magnet material), the permeability μ is a function of B as in (3.12) [48]:

3.2 Machine Volume Envelope Consideration

μ¼

53

B H ðBÞ

ð3:12Þ

The FEMM field solver converges such that (3.9), (3.10) and (3.11) are satisfied via a magnetic vector potential approach. Flux density is written in terms of the vector potential, Av, as in (3.13) [48]: B ¼ Δ
AV

ð3:13Þ

This definition of B always satisfies (3.10). Then, (3.9) can be rewritten as [48]  J ¼Δ


 1 Δ
AV μðBÞ

ð3:14Þ

For a linear isotropic material and assuming the Coulomb gauge (Δ.Av ¼ 0), (3.14) reduces to [48]:   1 2 Δ AV J¼ μ

ð3:15Þ

FEMM retains the form of (3.14) so that magnetostatic problems with a nonlinear B–H relationship can be solved. The advantage of using the vector potential formulation is that all the conditions to be satisfied have been combined into a single equation. If Av is found, B and H can then be deduced by differentiating Av [48]. Hence, the FEMM solver technique is based on the division of the volume or domain in which this equation is valid into smaller volumes or domains or so-called finite elements. Within each element, a simple polynomial is used to approximate the solution via some iterative methods such as conjugate-gradient and Newton–Raphson methods. The procedure for numerical computation of magnetic field problems using FEMM is divided into three steps: preprocessing, processing, and postprocessing. In the preprocessing mode, the machine outline is drawn (in AutoCAD in this case) and then uploaded into the FEMM solver. The machine material properties are defined, boundary conditions assigned, and mesh generation achieved. In the processing mode, the relevant Maxwell’s equations are used to solve the problem and to obtain the field distribution in the analyzed domain of the electric machine. In the postprocessing mode, calculations of characteristics, as well as parameters of the analyzed electric machine, are obtained. Figure 3.5 illustrates the benchmark brushless PM 2D machine model used in FEMM, showing the peripheral boundary condition (a), the mesh detail (b), and defined material (c). Material specification in FEMM is via blocks. Each block has a defined name, a material assigned from the FEMM library, and an associated current or circuit if relevant referring to Fig. 3.5c. The PM machine consists of five material areas:

54

3

Hybrid Permanent Magnet Machine Design

Fig. 3.5 Benchmark brushless PM machine 2D FEA model (a) Boundary conditions (b) Mesh detail (c) Material definitions

(i) (ii) (iii) (iv)

Air: for all air-gaps inside the machine and around it. US Steel Type 2-S, 0.018-inch thickness: for the stator part of the machine. Carpenter electrical iron: for the rotor part of the machine. NdFeB 37 MGOe: sintered neodymium–iron–boron permanent magnet material used for the machine rotor permanent magnets. (v) Copper: used for the stator phase windings.

Having set up the machine problem at the preprocessing stage, the FEMM solves the magnetic field equations and produces results, for example, the distribution of

3.2 Machine Volume Envelope Consideration

55

magnetic flux in the machine. Postprocessing of the solved field solution can be carried out to further analyze machine performance. To consider machine rotation, it is necessary to produce multiple field solutions from which the machine back-EMF can be determined. In FEMM, a Lua script code can be written to perform multiple field solutions. An example of the Lua script tasks are listed below: Start by opening the desired file in FEMM. Apply a “For-loop” function that is controlled by a specified rotor angular step. Solve and analyze the model. Save the analyzed data in a new file. Rotate the desired parts of the machine by angular step. Solve and analyze the new rotated model. Loop back to (d) or exit the for loop function when the desired final angle value is reached. (h) Stop the analysis. (a) (b) (c) (d) (e) (f) (g)

Hence, it is relatively straightforward to obtain the desired data from the FEMM solver for stationary and rotating magnetostatic fields and thus predict the electric machine characteristic behavior for different operating conditions.

3.2.5

Machine Back-EMF Prediction

The benchmark PM machine back-EMF versus rotor position is obtained by manipulating the FEMM postprocessing data that was collected for different rotor positions for machine no-load and varying load test cases. A Matlab script file was written to collect and manipulate the FEMM data. The proposed stator winding layout for a quadrant of the benchmark PM machine is shown in Fig. 3.6. Here, flux φ passing through a single coil is predicted at discrete rotor position angles over two electrical cycles, via the FE solver line integration function based on (3.16) [49]. Referring to Fig. 3.6, Av1 and Av1’ are the magnetic vector potentials of two coil sides. The previous step is applied for three consecutive stator coils. Based on the coil number of turns and rotor speed (ω), the generated φ and rotor position angle data arrays produced by FEA, as in Fig. 3.7a, are then substituted into (3.16) and (3.17) to predict the back-EMF waveforms for each of the three consecutive coils and single phase at no-load, as shown in Fig. 3.7. φcoil ¼ ðAV1  AV10 ÞL

ð3:16Þ

  dφcoil EMF ¼ ωnstc dθr

ð3:17Þ

56

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Hybrid Permanent Magnet Machine Design

Fig. 3.6 One quarter of the benchmark PM machine stator winding layout

where dθr is the difference in rotor position angle and ω is the angular electrical speed in rad/sec as calculated by (3.18): ω ¼ 2π f e

ð3:18Þ

where fe is the machine electrical frequency that is calculated by (3.19) and Ns is the rotor mechanical speed in rpm. fe ¼

3.2.6

PN s 120

ð3:19Þ

PM Machine Analysis Via EMC Model

Magnetic fields are present around current-carrying conductors and also exist around magnetized objects such as permanent magnet material [50]. The magnetic circuit quantities are represented by magnetic flux, reluctance, and magnetomotive force. These magnetic quantities can be thought of by an analogy to electrical circuit quantities, such that the electrical circuit analysis rules can be applicable to equivalent magnetic circuits (EMCs) [51]. The equivalent magnetic circuit (EMC) representation based on the lumped parameter method gives an approximate to the machine field solution since it represents a lumped, and somewhat idealized, section of the machine magnetic circuit. Nevertheless, the lumped parameters solution gives fast field calculation that can consequently speed up iterative design procedures. In this section, an EMC model is developed for one-quarter of the benchmark brushless PM machine (due to symmetry) since the solution thereof can be compared with the FEA model and measured data are discussed in the preceding section. Having gained confidence in the PM machine EMC model, the model will then be developed to

3.2 Machine Volume Envelope Consideration

57

Coils flux-linkage (mWb)

0.4

Coil 1 flux-linkage Coil 2 flux-linkage Coil 3 flux-linkage

0.3 0.2 0.1

0 -0.1

0

100

200

300

400

500

600

700

800

-0.2 -0.3 -0.4

Rotor angle (elec. deg.)

Three coils back-EMF (V)

(a) 10 8 6 4 2 0 -2 0 -4 -6 -8 -10

100

200

300

400

500

600

700

800

(+) Coil 1 EMF (--) Coil 2 EMF (+) Coil 3 EMF Rotor angle (elec. deg.)

Three coils and phase back-EMF (V)

(b) 100 80 60 40 20 0 -20 0 -40 -60 -80 -100

100

200

300

400

500

600

700

800

(+) Coil 1 EMF (--) Coil 2 EMF (+) Coil 3 EMF Phase EMF

(c) Fig. 3.7 Benchmark PM machine open-circuit back-EMF waveforms at 3000 rpm (a) Flux-linkage waveforms for three consecutive stator coils (b) Back-EMF waveforms for three consecutive stator coils (c) Back-EMFs per phase and for three consecutive stator coils

design a WF rotor within the benchmark PM machine stator. These two machines, (i) surface magnet and (ii) wound field rotor, both having the same stator magnetic and winding designs, will then be combined to form the prototype HPM machine (HESM topology). The lumped parameter method has a number of limitations [51]: (a) Only one value of relative permeability μ, from FEMM databases, is used for each soft magnetic section, as in (3.20), (3.21) and (3.22) [51]:

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Hybrid Permanent Magnet Machine Design

ΔB ¼ μ0 μr ΔH

ð3:20Þ

μri ¼

μi ¼ 2141 μ0

ð3:21Þ

μrs ¼

μs ¼ 1834 μ0

ð3:22Þ

μ¼

where ΔH is the difference in magnetic field intensity, μri is the relative permeability of iron, and μrs is the relative permeability of steel. (b) Soft magnetic material saturation is not included, although it can be accounted for by setting a maximum flux density for each section. (c) Leakage reluctance is not included. The considered flux density is normal to the surface flux density (Bn), which is calculated by (3.23) [51]. The flux passing through each part of the magnetic circuit is calculated by (3.24) [51] along with the magnetomotive force due to both the permanent magnets and wound field poles (WF rotor case) by (3.25) and (3.26) [51]: φ Ac

ð3:23Þ

MMF R

ð3:24Þ

Bn ¼

φ¼

MMF ¼ H c LM

ð3:25Þ

MMF ¼ n f I f

ð3:26Þ

where MMF is the magnetomotive force, R is the component reluctance, Hc is the material coercivity, nf is the excitation field number of turns, and If is the DC excitation field current. Furthermore, (3.26) represents the MMF of a wound coil, thus, by using (3.27) [51], the reluctance of each part of the machine may be calculated and the relevant EMC model is constructed. R ¼

h μWL

ð3:27Þ

where h is the material thickness, W is the width, and L is the axial length relevant to the flux path. The lumped parameter model reluctance definition was given in

3.2 Machine Volume Envelope Consideration

59

Fig. 3.8 Lumped parameter model for benchmark PM machine. (a) Different sections for the EMC model (b) Reluctances for one section enclosing a stator tooth (c) EMC representation

Chap. 1. Due to machine stator winding symmetry, calculation of the reluctance paths of the benchmark PM machine EMC model, based on the lumped parameter method of one-quarter of the machine cross section, is developed and illustrated in Fig. 3.8. The respective machine sections forming the EMC network of MMFs and reluctances are shown in Fig. 3.8a, flux path for a rotor pole and stator tooth is shown in Fig. 3.8b, and a circuit representation of the EMC model is shown in Fig. 3.8c. Assuming a clockwise flow of magnetic flux in each loop of the EMC model, Fig. 3.7c, and applying KVL, a set of equations is derived as in (3.28) and (3.29). φL ¼ R1 :MMF

ð3:28Þ

60

3

Magnet ic f lux (mW b)

0.4

Hybrid Permanent Magnet Machine Design

FEA Analytical using EMC

0.3

Tooth No. 9

0.2 0.1 0

0

10

20

30

40

50

60

70

80

-0.1 -0.2 -0.3 -0.4

Tooth No. 1 Rotor angle (mech. deg.)

Fig. 3.9 Magnetic flux comparison for the PM machine sections shown in Fig. 3.8a Table 3.3 Comparison of benchmark PM machine flux density Section No. (tooth body) 1 2 3 4 5 6 7 8 9

Rotor angle (mech. deg.) 0 10 20 30 40 50 60 70 80

φmb ¼

Flux density (T) EMC FEA 1.27 1.34 1.29 1.22 0.96 0.93 0.57 0.56 0.18 0.19 0.18 0.19 0.60 0.57 1.16 1.22 1.29 1.34

X

# φLi  " φLj

EMC and FEA flux density difference (%) 5.4 5.5 4.0 0.5 6.1 5.6 5.8 5.4 4.1

ð3:29Þ

i, j

where φL is the vector of loop fluxes (n
1), R is the matrix of reluctances (n
n), MMF is the vector of MMFs (n
1), φmb is the vector of resultant fluxes through each circuit branch (m
1), and the loop magnetic flux #φLi and "φLj represent the downward and upward magnetic flux directions, respectively, passing through two neighboring loops, i and j. Solving Eqs. (3.28) and (3.29), the flux flowing in each part of the benchmark PM machine EMC model is computed and compared with that of FEA, as illustrated in Fig. 3.9, for the section of the machine shown in Fig. 3.8a, b. The circled areas in Fig. 3.9 show magnetic flux for stator teeth. Note that the graphs in Fig. 3.9 are for one static solution at a fixed rotor angle; however, flux in each part of the machine changes as the rotor rotates. Table 3.3 lists the flux density in the benchmark PM machine tooth body in different sections as specified in Fig. 3.8a. The maximum difference between the results obtained from EMC and FEA is 6.1%, and it occurs in Sect. 5 where the flux leakage is highest for the presented static field solution. Note

3.3 WF Machine

61

Fig. 3.10 Comparison of benchmark PM machine predicted and measured results of the coil and phase back-EMFs at 250 rpm

that the machine operates on the knee region of the steel B–H curve and the highest flux density occurs in the tooth body that is 1.34 T. The saturation effect is small due to machine geometry that results in low flux leakages and hence not considered in the EMC modeling. To further confirm the validity of the FEA model, back-EMF measurements are obtained from the laboratory prototype benchmark PM machine, as in Fig. 3.2b, and compared with those of analytical results as illustrated in Fig. 3.10. To form a stator winding for the benchmark PM machine, coils are wound around one stator tooth as well as three consecutive teeth that form one-quarter of a full phase winding. The analytical, FEA, and measured results show a good agreement, hence confidence in the applied method, tools, and model accuracy.

3.3

WF Machine

The benchmark PM machine is used as the base of the design and hence forms the PM section of the HPM generator. To ensure the design is within the application volume constraint, the same stator geometry and winding configuration are considered for both PM and WF sections. To design a WF rotor, the lumped parameter model for the benchmark PM machine is modified to model flux paths on a wound field rotor. The WF rotor is designed to have a similar pole number (3.32) and shape as for the benchmark PM machine. However, there is a slight difference between rotor pole spacing due to practical limitations on WF rotor coils placement. To ensure similar stator flux levels, the WF rotor is designed to develop a close nominal air-gap flux density and stator flux linkage at full WF excitation current as the PM rotor does with an unexcited stator.

62

3.3.1

3

Hybrid Permanent Magnet Machine Design

WF Rotor Design

The same steps used to develop the PM section EMC model are applied to generate a suitable WF rotor design. The equivalent magnetic circuit model of one of the WF machine designs is illustrated in Fig. 3.11c. The flux flowing in each part of the WF machine EMC model is computed analytically and compared with that of FEA, as illustrated in Fig. 3.12a. Subsequently, for nf¼100 turns and If¼ 3 A, which gives a rotor slot current density of 4.9 MA/m2, the EMC model is adjusted in an iterative process to investigate the impact of key rotor magnetic circuit dimensions on overall design, the driving criteria being to achieve the required air-gap flux density and

Fig. 3.11 Lumped parameter model for WF machine. (a) Different sections for the EMC model (b) Reluctances for one section enclosing stator and rotor tooth (c) EMC representation

3.3 WF Machine

63

Fig. 3.12 WF machine stator teeth flux comparison via FEA and EMC along with stator tooth iterative process using different air-gap thicknesses. (a) Magnetic flux comparison for sections shown in Fig. 3.11a (b) Stator tooth flux for different air-gap thicknesses and flux density

stator flux linkage subject to minimizing rotor excitation MMF, saturation, slot current density, and rotor copper losses. Figure 3.12b shows flux variation of WF stator tooth for different air-gap thicknesses (δ) with corresponding air-gap normal flux densities (Bn). For a WF air-gap thickness of δ ¼ 0.35 mm, the stator tooth peak magnetic flux will be equitable to that of the benchmark PM machine; however, it results in a flux density of 1.34 T, which is outside the defined boundary. Therefore, the WF section air-gap is increased to 0.4 mm, resulting in a peak stator tooth flux density of 1.195 T. Final WF rotor geometry is shown in Fig. 3.11a, b. Hence, for HPM machine operation stability, the air-gap flux density waveforms for the PM and WF parts are plotted via FEA for one quadrant, as previously shown in Figs. 3.8a and 3.11a, to check their symmetry, as shown in Fig. 3.13. This

64

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Hybrid Permanent Magnet Machine Design

Fig. 3.13 Air-gap flux density waveforms due to PM and WF machines via FEA

symmetry in the machine air-gap flux density waveforms will minimize the possibility of adding more air-gap MMF harmonic effects to the HPM machine backEMF, which increases total harmonic distortion (THD) due to the addition of PM and WF machine elements back-EMF waveforms. The impact of stator slot opening with the rotor poles width at different locations is shown through the nonuniform air-gap flux density waveforms. Note that it is recognized in the literature that cogging torque harmonics can be reduced when nonoverlapping winding, minimal slot opening, and the right combination of pole and slot numbers with adjusted rotor pole width are achieved [52, 53].

3.3.2

WF to PM Split Ratio

In the previous section, the WF machine radial air-gap length is selected using the EMC model to produce a phase flux linkage level as for the benchmark PM machine axial length at a fixed rotor speed of 3000 rpm, the WF to PM generator split ratio can be deduced to meet certain operation constraints. The operating philosophy in SHEVs using an internal combustion engine/HPM generator scheme acts as an axillary power system to provide average power for the vehicle powertrain where the main battery system handles the transients power demand. By adopting this HPM machine operating philosophy in SHEVs, the size of the engine and hence both fuel consumption and emission are then reduced. The speed variations during transients and dynamics are to be adjusted via the WF excitation of the HPM generator. Hence, an expected vehicle DC-link voltage variation of 30% due to vehicle transients and dynamics has been reported in the literature [46]. Therefore, the HPM generatorrectified output voltage is required to follow these voltage variations. The second

3.3 WF Machine

65

Fig. 3.14 Magnetic flux due to axial length variation via the EMC model of the WF machine

iterative process is implemented here to come up with the adequate split ratio between PM and WF machines’ axial lengths (WF/PM) based on the operation philosophy of the final HPM generator design. The active length has a great influence on the machine weight and output voltage; hence, it should be selected carefully. Thus, by reducing the axial length of the WF machine element in several steps and observing the stator teeth peak magnetic flux, the optimized WF machine axial length based on operational philosophy is identified via the EMC model. The second iterative process results via the EMC model are shown in Fig. 3.14, where the adequate WF machine axial length is found to be 10 mm. Thus, this WF DC excitation of the HPM generator final design will provide the required 30% buck/ boost back-EMF voltage capability in the SHEV powertrain. This HPM generator will operate without demagnetizing the PM machine magnet during back-EMF buck mode since the magnetic linkage between the HPM generator rotors is magnetically decoupled.

3.3.3

Comparative Analysis of WF Rotor Designs

Having developed the lumped parameter model for the WF rotor and stator, the major rotor magnetic dimensions were varied to investigate their impact on the design, the driving criteria being to achieve the required air-gap flux density while minimizing rotor excitation MMF and saturation. Several WF machine rotor designs (with identical stators) were identified, the results for which are detailed in Table 3.4. Here, the design that can produce maximum air-gap normal flux density (Bgn) with the lowest DC excitation current, rotor slot current density (Jrs), field copper loss (Pfc), and DC excitation field voltage (Vf) will present the best design choice to be used in the final HPM machine design.

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Hybrid Permanent Magnet Machine Design

Table 3.4 Dimension details for WF machine rotor designs WF rotor design parameters Rotor outer diameter (DWFro) Rotor inner diameter (DWFri) Rotor tooth width (Wrt)

1 166.75 109 8.54

2 166.75 100 9.2

Units mm mm mm

14.2 17.5 2.2 15 1.8

3 166.75 109 7.6 max. 11 min. 13 16.5 0.8 11.5 3.1

Rotor tooth tip width (Wrtt) Rotor tooth thickness (hrt) Rotor tooth tip thickness (hrtt) Rotor yoke thickness (hry) Spacing between rotor teeth (at tooth tip) (τrt) Pole number (P) Active axial length (LWFa) Half rotor slot area (Ars) Air-gap thickness (hgWF)

13.2 27 1.3 5 3.2 32 10 59.3 0.45

32 10 36.3 0.45

32 10 41.7 0.45

– mm
1026 m2 mm

mm mm mm mm mm

Note that in the FEA and lumped parameter models, the WF rotor designs consist of three materials: (a) Air: for all air-gaps inside and around the machine. (b) US Steel Type 2-S, 0.018-inch thick laminations: for the stator and rotor parts of the machine. (c) Copper: for both field excitation winding circuits on the rotor and phase winding circuits on the stator. Figure 3.15 illustrates FEA results for the three WF rotor designs satisfying the above design criteria. To better inform the choice of WF rotor design, some analysis was carried out utilizing (3.30) (3.31), (3.32), (3.33) and (3.34) and considering two test cases: Case (1): Calculating the WF copper loss and required supply voltage Pfc and Vf respectively, when the rotor slot current density Jrs is kept constant at 3.7 MA/m2. Here, field current If and rotor tooth coil turns nf are varied to choose the most appropriate number for nf. Case (2): Calculating the air-gap flux density Bgn, and Jrs, Pfc, and Vf when nfis kept constant and If varies. The analysis test cases will support the decision on the adequate excitation field number of turns and the maximum allowable DC excitation current value for the selected WF rotor design. nf ¼

J rs Ars If

ð3:30Þ

3.3 WF Machine

67

Fig. 3.15 FEA results for three WF designs considered for detailed study. (a) Design 1 (b) Design 2 (c) Design 3

Lfc ¼ n f ð2LEFa þ 2πτrewr Þ

Rf ¼

Pn f Lfc σkP Ars

Pfc ¼ I 2f R f

ð3:31Þ

ð3:32Þ

ð3:33Þ

68

3

Hybrid Permanent Magnet Machine Design

Fig. 3.16 Case (1) results with a vertical line intersection at 35.9 V; 100 turns Table 3.5 Case (1) results when Jrs constant at 3.7 MA/m2

WF rotor design 1 2 3

Bgn (T) 0.374 0.153 0.183

Table 3.6 Summary of the case (2) results when nf equals 100 turns WF rotor design 1 2 3

If (A) 4.2 5 6

Jrs (MA/m2) 7.07 13.8 14.4

V f ¼ I fRf

Pfc (W) 288.0 666.4 834.7

Vf (V) 68.8 133 139

Bgn(max) (T) 0.666 0.529 0.682

ð3:34Þ

where τrewr is the rotor mean end winding radius, Lfc is the copper length per excitation field coil, kp is the slot fill factor, σ is the conductor conductivity, and Rf is the wound rotor field coil resistance. The results of test case (1) illustrate that Vf for all wound field rotor designs equals 35.9 V when nf equals 100 turns, but with different copper losses, as shown in Fig. 3.16. Furthermore, Table 3.5 compares air-gap maximum flux density values when Jrs equals 3.7 MA/m2. Thus, test case (1) shows that design 1 presents the preferable rotor design geometry. The results of test case (2) are summarized in Table 3.6 and Fig. 3.17 and demonstrate that WF rotor design 1 is considered the best choice of the three designs. The maximum air-gap flux density for all of the WF machine designs is around 0.67 T; however, the slot current density and copper losses at this air-gap flux density are high for designs 2 and 3. Thus, WF rotor design 1 represents the best choice since it can realize Bgn(max) with the lowest values of If, Jslot, Pfc, and Vf.

3.4 HPM Machine Parameters

69

Fig. 3.17 Air-gap flux versus slot current density for the three WF rotor designs

Thus, rotor design 1 was adopted as the most suitable solution and used in the design of a WF machine having the same stator cross section as the benchmark brushless PM design, but with a 10-mm as opposed to a 25-mm active axial length. Applying full rotor field excitation, the open-circuit back-EMFs of this WF machine design are illustrated in Fig. 3.18. Note that Fig. 3.7 is repeated so that the back-EMFs due to the 25-mm PM machine and the 10-mm WF machine may be compared. The choice of active axial lengths for the PM and WF rotor machines simply comes from a consideration of the contribution of the WF machine backEMF to that of the PM machine back-EMF.

3.4

HPM Machine Parameters

The FEA is used again in this section to solve the machine electromagnetic field and hence determine the machine parameters in the presence of complex magnetic circuit geometry and nonlinear material properties.

3.4.1

Torque Prediction and Saturation

Electromagnetic torque developed in the machine stator-to-rotor air-gap can be obtained for each rotor angular increment by using the FEMM solver Maxwell’s stress tensor integral routine [48]. This method is a standard FEA technique based on prescribing a force per unit area produced by the electromagnetic field on a defined surface [48]. Torque is computed by integrating the stress tensor along a line running through the center of the air-gap between the rotor and the stator, as illustrated in

70

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Hybrid Permanent Magnet Machine Design

Fig. 3.18 WF machine open-circuit back-EMF waveforms at 3000 rpm. (a) Flux-linkage waveforms for three consecutive stator coils (b) Back-EMF waveforms for three consecutive stator coils (c) Back-EMFs per phase and for three consecutive stator coils

Fig. 3.19 [47]. The same contour is used for the WF machine. For both PM and WF stators, current densities are defined in the three-phase stator winding slots as per the phasor relationship illustrated in Fig. 3.20. With the WF current set at a normal maximum of 4.2 A, as will be clarified later, torque as a function of rotor position can

3.4 HPM Machine Parameters Fig. 3.19 FEMM model of PM machine section, highlighting the circumferential contour line taken for Maxwell stress integration

Fig. 3.20 Phasor relationship of stator currents

Fig. 3.21 Predicted torque versus rotor angle for the HPM machine sections

71

72

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Hybrid Permanent Magnet Machine Design

Fig. 3.22 Magnetic saturation characteristics of the PM and WF HPM machine parts. (a) PM peak torque characteristic (b) WF peak torque characteristic

be calculated, as illustrated in Fig. 3.21. The HPM electromagnetic capability is determined by gradually increasing the three-phase stator current magnitudes until the developed peak torque per ampere saturates, as illustrated in Fig. 3.22 for the (a) PM and (b) WF machine sections. Based on the stator current density of both the PM and WF machine parts, the peak power of each part can be calculated individually by (3.35), where ωs is the mechanical angular speed of the rotor in rad/sec. The results show that the peak power that can be achieved from the PM part equals 5311 W and 2357 W for the WF part due to the relatively high rotor current density (7.07 MA/m2). However, the peak torque and power capabilities will be duty rated due to the machine internal losses, such as iron and copper losses. Thus, the machine thermal power rating is less, as will be shown later.

3.4 HPM Machine Parameters

73

PRated ¼ T d ωs

3.4.2

ð3:35Þ

Synchronous Inductance and Winding Resistance

The HPM generator self- and mutual-inductances are predicted using 2D FEA models and an empirical factor applied to account for end-winding. To consider the contribution from the self- and mutual-inductance components, the series coils are reduced to a single effective coil, and the inductances are calculated for WF and PM sections separately. For the three-phase machine configuration, the spatial displacement between the single stator coils is 120 electrical, as illustrated in Fig. 3.23. The machine magnetic field is then solved with and without the PM or WF rotor excitation. The self-inductance of this single effective coil, which represents the equivalent phase coil, can be calculated by (3.36) [54]. For the case of the PM part of the HPM machine, the rotor PM pieces are changed to air in the FEA model, DC current is injected in the phase coil for which the self- and mutualinductances are calculated, and the other phase currents are set to zero. The FEA results for the self-inductance is based on (3.36), while the mutual-inductance between, for example, phases (A) and (B), is based on (3.37) [54]:

Fig. 3.23 Single coil representation of threephase HPM machine

74

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Hybrid Permanent Magnet Machine Design

Table 3.7 HPM generator inductances calculated from FEA with and without rotor

Item Three-phase

Stator PM WF

WF rotor inductance (with stator) Item Three-phase a

Lf (mH) 68

Mutual inductances between phases as a % of LSelf MAC LSelf (μH) MAB 146.0 7.5 7.5 (123)a (9.1)a (9.1)a 178.7 4.9 4.9 (49.3)a (9.1)a (9.1)a Mutual inductances between WF and stator phases as a % of Lf MfA MfB MfC 2.7 0.56 3.4

Values without rotor

Table 3.8 HPM generator measured inductances and resistance

Stator section PM WF WF rotor (with stator) Item Threephase

Inductance (μH) (with rotor) EndSelf-winding winding FEA Measured 2Lend 146.0 175 29.0 178.7 308 129 68
103 91
103 23
103

LSelf ¼

λ e ne λ t ¼ I in I in

M¼

λel I in

Resistance (mΩ) Predicted Measured 40.84 50.00 20.95 31.00 13.06
103 11.35
103

ð3:36Þ

ð3:37Þ

where λe is the total phase flux linkage, ne is the number of turns per phase, λt is the flux linkage per turn per phase, Iin is the injected phase current and λel is the fluxlinking with the coupled coil of interest, for example, the effect of phase A excitation on phase B. A similar procedure can be followed to obtain the self- and mutualinductances for the other phases. Table 3.7 lists predicted inductances via FEA while Table 3.8 compares the predicted and measured inductances and stator per-phase and WF rotor resistances. Note that the stator inductances are calculated for two cases: (i) with the presence of the PM and WF rotor sections and (ii) without rotor sections. Since the inclusion of the rotor reduces the reluctance of flux paths, the inductances are higher than the inductances calculated without rotors as seen from Table 3.7. The 2D FEA is used to calculate the machine inductances as it concludes more accurate results compared to the EMC model due to the inclusion of flux leakages. Predictions of synchronous inductance do not include end-winding effects due to the 2D FEA limitations. The end-winding effect on inductance can be calculated from 3D FEA proposed in [55]

3.5 HPM Machine Final Design Model Analysis

75

or from a series of measurements made on machines of varying axial length. The latter is used in this chapter where actual winding self-inductances of the assembled HPM machine stators were measured during the machine assembly and included in the self-inductance as follows: Lself m ¼ 2Lend þ Lself FEA

ð3:38Þ

where Lend is the end winding inductance and Lself  m and Lself  FEA are measured and predicted self-inductances, respectively. Thus, as the machine axial length increases, the impact of the end-winding inductance on total inductance reduces, as seen in Table 3.8.

3.5

HPM Machine Final Design Model Analysis

The final design specification of the proposed three-phase HPM machine is illustrated in Table 3.9. The predicted HPM machine rotor PM demagnetization along with core and copper losses will be analyzed based on this final design specification.

3.5.1

Rotor PM Demagnetization

The hard magnetic material used on the PM rotor is sintered neodymium–iron–boron (NdFeB) that has a linear second-quadrant B–H characteristic at 20  C [51]. However, at higher temperatures, say 120  C, the second-quadrant characteristic becomes nonlinear and there is potential for recoil working of the material. In terms of recoil working, an important area to consider is demagnetization or partial demagnetization due to the thermal operating environment of the permanent magnet. Figure 3.24 illustrates a permanent magnet material showing a typical open-circuit load line and how the second-quadrant characteristic becomes nonlinear with elevated temperature. N35H is close to the design material specification used for the prototype PM machine; thus, 120  C operation could be problematic. To check the potential for rotor demagnetization, the FEMM post-processor file information was analyzed to calculate the B–H working point of each element defined in the rotor FEA mech. Figure 3.25 illustrates one of the rotor PMs with its internal mesh elements (triangles). Each of these elements has a flux density value that varies with the element and rotor position. The flux density value for each rotor PM element can be projected on the PM material demagnetization curve to ensure that they fall within the desired operating limits or not. In this analysis, each material specified in the 2D FEMM machine model contains many numbers of elements. Each element is produced by three nodes, as shown in Fig. 3.25b, and each node has an X, Y coordinate and vector

76

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Hybrid Permanent Magnet Machine Design

Table 3.9 HPM generator final design specifications Item Stack length (mm) Clearance between PM and WF rotors (mm) Yoke outer diameter (mm) Yoke inner diameter (mm) Yoke thickness (mm) Stator tooth-body width (mm) Stator tooth-body length (mm) Stator tooth-tip width (mm) Stator tooth-tip length (mm) Stator slot opening (mm) Stator slot area (mm2) Number of (stator slots/rotor poles) Number of stator turns per coil Number of rotor turns per coil Air-gap thickness (mm) Shaft outer diameter (mm) Shaft outer diameter (mm) Magnet thickness (mm) Magnet length (mm) Rotor yoke outer diameter (mm) Rotor yoke inner diameter (mm) Rotor tooth-body width (mm) Rotor tooth-body length (mm) Rotor tooth-tip width (mm) Rotor tooth-tip length (mm) Rotor slot opening (mm) Rotor slot area (mm2)

HPM stator section PM section 25 5 230.4 204.4 16.5 9.2 16.7 12.4 1.9 2.45 121.2 (36/32) 6 100 0.8 130 40 2.8 14.4 160 130 –

WF section 10

0.4

122 109.2 8.54 27 13.2 1.3 3.2 59.3

potential. Equations (3.39), (3.40), (3.41) and (3.42) [40] are solved using data from the FEMM solver.     2Av ¼ X j Y k  X k Y j þ ðX k Y i  X i Y k Þ þ X i Y j  X j Y i

ð3:39Þ

     X k  X j Avi X j  X i Avk ðX i  X k ÞAvj BX ¼ þ þ 2Av 2Av 2Av

ð3:40Þ

3.5 HPM Machine Final Design Model Analysis

77

Fig. 3.24 Rotor permanent magnet material characteristics [56]

 BY ¼

    Y j  Y k Avi Y i  Y j Avk ðY k  Y i ÞAvj þ þ 2Av 2Av 2Av

jBj ¼

qffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffi B2X þ B2Y

ð3:41Þ

ð3:42Þ

Figure 3.25c illustrates the rotor permanent magnet operating points plotted on the material second-quadrant B–H characteristic and for a typical full-load operating point. Note that this analysis is further used to assess the HESM and DESM topologies.

3.5.2

Core Loss Prediction

The prediction of iron losses in rotary machines has been discussed in many papers [57–61]. The most commonly used iron loss formulas divide the losses into hysteresis, eddy current, and excess losses. Li et al. discuss the separation of iron losses general equation into two equations to calculate iron losses [59], such that one of the equations relates to the existence of alternating magnetization and the second one is related to the rotary magnetization iron losses. In addition, Li et al. calculate iron losses via four different methods showing that there were no significant disparities between the analytically obtained iron losses [59]. Boglietti et al. presented the wellknown iron loss model; however, some of the new coefficients in their equation

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Fig. 3.25 PM pole in FEMM post-processing mode. (a) PM pole showing FEA mesh triangles (b) Stylized close up of a PM pole illustrating element coordinates (c) Typical PM operating points at full-load current

depend on the knowledge of the chemical and physical characteristics of the considered magnetic material [59]. In this section, no separation of these iron loss elements will be applied and iron loss will generally be predicted by polynomial regression used to curve fit actual iron loss data of United Laminated steel material (0.47 mm, 26 gauge) [62], obtained at several frequencies. The manufacturer’s measured iron loss curves present the iron losses in watts per kilogram versus magnetic flux density, as illustrated in Fig. 3.26. Therefore, the general equation that represents those iron loss curves is to be obtained in order to predict the iron loss curve at any operating frequency. Here, two main steps have been applied via polynomial regression before solving the final iron loss equation, as in (3.43). Iron loss due to current harmonics is not included in the following loss prediction steps.

3.5 HPM Machine Final Design Model Analysis

79

Fig. 3.26 Manufacturer’s loss data for United Laminated steel (0.47 mm) [62]

Piron ðBÞ ¼ k e B3m þ kh B2m þ k a Bm þ kd

ð3:43Þ

where the Piron is the iron loss, Bm is the peak flux density, and ke, kh, ka, and kd are the polynomial coefficients (frequency-dependent terms). Hence, the first step in this derivation was to get the approximate loss curve via polynomial regression for each of the manufacturer’s iron loss curves as in (3.43). The second step has been applied by using the approximate obtained curve equations to plot the different ke coefficients via their respective frequencies. Here, again an equation via polynomial regression is derived as in (3.44), where in this equation ke is a function of frequency and ae2, ae1, and ae0 are the regression coefficients. ke ð f Þ ¼ ae2 f 2 þ ae1 f þ ae0

ð3:44Þ

By the second step procedure, kh, ka, and kd are presented as in (3.45), (3.46) and (3.47). kh ð f Þ ¼ ah2 f 2 þ ah1 f þ ah0

ð3:45Þ

ka ð f Þ ¼ aa2 f 2 þ aa1 f þ aa0

ð3:46Þ

k d ð f Þ ¼ ad2 f 2 þ ad1 f þ ad0

ð3:47Þ

Table 3.10 presents the obtained regression coefficient values that are used to get an approximate value for ke, kh, ka, and kd at different frequencies and then be used to solve (3.43), as in Fig. 3.27, for different flux density peak values.

80 Table 3.10 Coefficients of Eqs. (3.44), (3.45), (3.46) and (3.47)

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Hybrid Permanent Magnet Machine Design

Regression coefficients ae2 ae1 ae0 ah2 ah1 ah0 aa2 aa1 aa0 ad2 ad1 ad0

0.000100 0.004100 0.287300 0.000040 0.000200 0.874300 0.000010 0.025900 1.076700 0.000007 0.003000 0.148000

Fig. 3.27 Curve fitting for HPM machine core loss prediction

The last step of this HPM machine iron loss prediction is to find the mass of each stator section, such as stator back iron, tooth, and tooth tip, as shown in Fig. 3.28, since the iron loss curves are given in W/kg. Then, the approximated iron loss curves are used to find the iron loss of each part depending on the peak value of its flux density waveform. Finally, the loss from the different stator parts is summed for both HPM machine parts (PM and WF stators) to get the total machine iron loss at the specified rotor speed. Table 3.11 details the different stator section masses of the HPM machine parts, their calculated iron loss, and the total mass and iron loss. Thus, by (3.44), Fig. 3.27, and Table 3.11, the predicted HPM machine iron losses at 3 krpm and with no-load and rated WF excitation equal 411.8 W.

3.6 HPM Machine Thermal Model

81

Fig. 3.28 Stator sections for iron loss calculations with FEA results at no-load condition (PM part)

Table 3.11 Different stator and rotor sections masses and core losses for the fundamental harmonic only of three-phase PM and HPM machine at no-load HPM machine

Machine section Back-iron

Teeth

Teeth-tip

Total

3.6 3.6.1

PM stator LPMa (25 mm) Mass Core (kg) Losses (W) 1.748 34.60 (19.8 W/ kg) 0.968 191.20 (197.5 W/ kg) 0.148 13.80 (93.2 W/ kg) 2.865 239.60

WF stator LWFa (10 mm) Mass Core (kg) Losses (W) 0.700 8.85 (12.6 W/ kg) 0.387 38.50 (99.4 W/ kg) 0.059 2.96 (49.8 W/ kg) 1.146 50.30

WF rotor LWFa (10 mm) Mass Core (kg) Losses (W) 0.139 28.80 (207.2 / kg) 0.472 90.50 (192.2 / kg) 0.055 2.39 (43.2 W/ kg) 0.665 121.90

WF Total LWFa (10 mm) Core Losses (W) 37.65

129.00

5.35

172.20

HPM Machine Thermal Model General Principle of the Lumped Parameter Method

Heat dissipation in all energy generation and conversion machines arise as a consequence of losses and need to be assessed since they may severely affect machine integrity and lifetime. As NdFeB magnet materials are temperature

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sensitive, it is necessary to accurately predict temperature distribution to prevent damage and demagnetization. Thus, the importance of thermal management within machines has led to a wider range of work devoted to the development of electric machine thermal models [63–71]. Therefore, detailed thermal modeling of the HPM machine integral parts is required since the knowledge of thermal behavior in different cases and conditions can prevent overheating and improve the utilization of the machine. In this section, a general description of the thermal model for both PM and WF machine sections will be developed via commercial software along with thermal analysis results. As with electromagnetic analysis, there are two main methods used for thermal modeling: lumped parameter (analytical) and the finite-element method [63– 67]. Finite-element methods have some undesirable deficiencies due to the computation times, 2D cross-section simplification, and loss of accuracy [70]. Thus, a commercial software package, the Motor-CAD [66] program, which is based on the lumped parameter method, will be utilized in this thesis to obtain the equivalent thermal model and predict the temperature on the machine sensitive parts, based on natural convection cooling for both PM and WF machine sections.

3.6.2

Conduction Heat Transfer

The principle of the lumped parameter method consists of dividing the machine into basic thermal elements that represent a combination of conduction, convection, and radiation heat transfer processes. Heat is transferred in electric machines by means of conduction in solid and lamination parts and by convection and radiation between surfaces that are in contact with air. The determination of thermal constraints is an essential prerequisite for estimating these temperature distributions in order to ensure that constraints imposed, for example, by insulating and PM materials, are not

Fig. 3.29 Illustrative lumped parameter thermal networks for three heat flow directions

3.6 HPM Machine Thermal Model

83

violated. Here, the lumped parameter method uses T-equivalent circuit, as shown in Fig. 3.29, to represent the relationship between the heat flow and the main component temperature in the axial, radial, and circumferential directions, which have been used for the thermal analyses of some electric machines [63, 64]. In each network, one of the terminals represents the main temperature θm of the component at which any internal heat generation (u) or thermal storage capacitance (C) is introduced. The other two terminals represent the appropriate surface temperature of the component. For each network, there is a central node that gives the main temperature of the component if there were no internal heat generation or storage [63]. The values of the thermal resistances in each network are derived from the independent solutions of the heat conduction equations in the axial, radial, and circumferential directions. These thermal resistances are given in terms of the dimensions of the general cylindrical shape and the axial, radial, and circumferential thermal conductivities Ka, Kr, and Kc as shown in (3.48), (3.49), (3.50), (3.51), (3.52), (3.53), (3.54), (3.55) and (3.56) [63, 64]. R1a ¼

L  2πK a r 21  r 22

ð3:48Þ

R2a ¼

L   2πK a r 21  r 22

ð3:49Þ

R3a ¼

L   6πK a r 21  r 22

ð3:50Þ



 3 2 2r 22 ln rr12 1 4 5 1  2 R1r ¼ 4πK r L r 1  r 22   2 3 r1 2 2r ln r2 1 4 1  2   15 R2r ¼ 4πK r L r 1  r 22 2 R3r ¼



 2

1 4 2 r1 þ r2 8πK r L

 3 4r 21 r 22 ln rr12  5   2 r 1  r 22

ð3:51Þ

ð3:52Þ

ð3:53Þ

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2π

 

ð3:54Þ

 

ð3:55Þ

2π   6LK c ln rr12

ð3:56Þ

R1c ¼

2LK c ln

R2c ¼

R3c ¼

2π 2LK c ln

r1 r2

r1 r2

Hence, the negative values of the interconnected thermal resistances R3a, R3r, and R3c show that the main temperature θm, which is related to all three networks, has to be lower than the temperature given by the central node. The total thermal capacitance of the cylinder geometry is shown in (3.57) [63].   C ¼ ρcΡ π r 21  r 22 L

ð3:57Þ

where ρ is the material density, cΡ is the material specific heat, r1 is the cylinder outer radius, r2 is the cylinder inner radius, and L is the axial length.

3.6.3

Convection Heat Transfer

There are two types of convection heat transfer related to electric machines, natural convection, and forced convection. When heat is removed by means of ventilation or circulating liquid inside the machine, it is called forced convection heat transfer. Natural convection means that neither an external blower nor any coolant liquids exist [72]. Natural convection heat transfer is a primary function of the temperature difference between the component and the internal or external air. These types of convections occur via the air next to the heated body. For example, electric machine convection heat transfer can occur in the air-gap, endcap air, and at its surface. The thermal resistance Rc that can be utilized to model the convection heat transfer at the frame, air-gap, and endcap air is calculated as in (3.58) [63–71]. Rc ¼

ðT s  T a Þ 1 ¼ Q hAca

ð3:58Þ

where Ta presents the ambient temperature, Ts is the emitting surface, Q is the heat dissipation, Aca is the surface area in contact with the ambient, and h is the heat transfer coefficient.

3.6 HPM Machine Thermal Model

85

The main difficulty in the previous equation lies with the calculation of the heat transfer coefficient, which depends on many variables such as the temperature difference between the heated component and air, the component geometry, and the property of the component surface. However, accurate values for the previous variables are very hard to obtain since electric machines are constructed in different manners and shapes. Therefore, the previous obstacle is resolved by using a proven empirical formulation (correlation), which is provided by Motor-CAD that gives a good approximate value for the heat coefficient for any convection surface in the machine [66]. Natural and forced convection from simple electric machine surfaces, such as cylinders, flat plates, and more complex surface structures such as open and closed channels of various shapes and sizes, can be found in [73]. Such correlations are usually based on empirical dimensionless analysis. Some of the dimensionless numbers like Nusselt (NNu), Reynolds (NRe), Grashof (NGr), Prandtl (NPr), Taylor (NTa), and Rayleigh (NRa) are used as in (3.59) and (3.60) [66, 73]. For natural convection, the typical form of correlation is N Nu ¼ aðN Gr N Pr Þb

ð3:59Þ

and for forced convection, the typical form is N Nu ¼ aðN Re Þb ðN Pr Þc

ð3:60Þ

where a, b, and c are constants giving in the Motor-CAD correlation. For various electric machine structures, the magnitude of NRe is used to judge if there is laminar or turbulent flow in forced convection systems. Similarly, the NRa number, which is equal to NGrNPr, is used in natural convection systems [66, 73]. In addition, the NTa number can be used to judge if there is a laminar, vertex, or turbulent flow in the air regions around the rotating rotor since the case is different with these regions [63, 71, 72]. However, some adjustments have to be made to the heat transfer coefficients, which are related to the air regions around the rotor, where stator or rotor slot opening is not included in the NTa number. Empirical results obtained by Gazley [63, 66] suggest that if the flow is laminar and the slots are in the rotor side only, then there is a decrease by 10% in the heat transfer compared to the smooth air-gap, and if the slots are in both sides, stator and rotor, then the decrease is by 20%. However, if the flow is turbulent and the slots are in the stator side only, then there is an increase by 10% in the heat transfer compared to the smooth air-gap, and if the slots are in the rotor side only, then the increase is by 20% [66]. Turbulent flows give enhanced heat transfer, and they add resistance to the heat flow path in a forced convection system [66]. In addition, if there is a mixed heat transfer due to a combination of both natural and forced convection, then (3.61) is required to compensate for this heat flow mixture, in which the machine orientation determines whether the sign between the natural and forced heat coefficient is positive or negative [66, 72]:

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h3MIXED ¼ h3Forced  h3NATURAL

ð3:61Þ

where a positive sign is used for assisting and transverse flow and a negative sign is for the opposing flows. Each surface requires one value of heat transfer coefficient for stationary case (hs) and another heat transfer coefficient value for the rotating case (hr) [63]. The heat transfer coefficient between the rotating body and air-gap can be defined in terms of the dimensionless NNu number, air-gap length Lg, and the thermal conductivity of air Kair, as in (3.62). h¼

3.6.4

N Nu K air Lg

ð3:62Þ

Radiation Heat Transfer

Radiation heat transfer can be described as the transfer of heat from a surface due to energy transfer by electromagnetic waves [74]. Such that, (3.63) [74] describes the radiation resistance of the lumped circuit: RR ¼

1 Aco hR

ð3:63Þ

where Aco is the component surface area and hR is the heat transfer coefficient. The heat transfer coefficient is calculated from [74].   σ s ε1 F 12 T 41  T 42 hR ¼ T1  T2

ð3:64Þ

where σ s is the Stefan–Boltzmann constant, ε1 is the emissivity of surface 1, F12 is the view factor (how well surface 2 is viewed by surface 1), T1 is the absolute temperature of surface 1, and T2 is the absolute temperature of surface 2.

3.6.5

HPM Machine Thermal Model

The Motor-CAD software is used to model and solve the machine equivalent thermal network by inputting the desired machine geometric data using the graphical radial and cross-section editor, along with material names and predicted machine losses. The thermal model of the HPM machine, which does not account for the circumferential heat flow, is dealt with as two individual thermal machine models, which

3.6 HPM Machine Thermal Model

87

Fig. 3.30 PM machine steady-state thermal analysis results via Motor-CAD (a) Radial cross section (b) Axial cross section

represent the PM and WF parts, due to the limitation imposed by the program. This thermal program limitation has minor impact on the predicted temperatures of the HPM machine parts due to the separation distance between the PM and WF sections

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Fig. 3.31 WF machine steady-state thermal analysis results via Motor-CAD. (a) Radial cross section (b) Axial cross section

of the machine. The thermal parameters such as conduction, convection, radiation, and thermal resistances are calculated by the program, and hence the machine thermal model is created. For the housing outer cooling by natural convection case, the thermal analysis results for both machines (PM and WF) at steady state are illustrated in Figs. 3.30 and 3.31, where the transient temperature curves are shown in Figs. 3.32 and 3.33. The thermal analysis results were satisfactory for the PM machine; however, the WF

3.6 HPM Machine Thermal Model

89

Fig. 3.32 PM machine thermal transient analysis curves via Motor-CAD

Fig. 3.33 WF machine thermal transient analysis curves via Motor-CAD

machine thermal analysis showed an elevated temperature, especially in the rotor, such as 189.7  C at the rotor winding and 194.2  C at the rotor tooth. These high temperatures are due to the relatively high rotor copper losses caused by the rated DC excitation current, which in this analysis is 4.2 A. However, the required DC excitation current to regulate the selected HPM machine-rectified DC-link voltage when the machine system is placed in a hybrid EV powertrain is actually discrete and less than the rated value, as will be discussed in Chap. 5, and this in turn will decrease the rotor copper losses. For example, if the rated excitation current decreased by half the copper losses will be 75% less than the rated copper losses, and an acceptable steady-state machine internal part temperatures will be gained. Hence, the applied thermal analysis is a good starting point to predict the HPM machine distributed temperatures before embarking with the machine actual testing at different operating conditions. Thus, different excitation current values are analyzed next to monitor the decrease in temperature in the rotor tooth, which present the hottest rotor spot, while

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Fig. 3.34 WF machine rotor tooth thermal transient analysis curves for different excitation currents via Motor-CAD

maintaining the machine parts core losses constant as for the worst case (rated excitation current). Figure 3.34 shows the decrease in the WF rotor tooth temperature due to the decrease in the excitation current, where the temperature decreased from 194 to 85  C for If equals 4.2 to 0.5 A, respectively. Note that the tooth temperature did not decrease beyond 85  C due to the fixed core losses applied to the MotorCAD, which have been predicted previously at rated speed and excitation current condition.

3.7

Comparison Between PM and Four HPM Machine Topologies

The discussion of the HPM machine has so far been centered on the HESM topology, that is, a two-part machine having separate PM and WF sections. The electromagnetic and the thermal analysis tools discussed in the preceding sections were used to assess four contending HPM machine topologies, as detailed in Chap. 2, and compare them against the benchmark brushless PM machine previously discussed. The selected topologies are illustrated in Fig. 3.35 along with the reference PM machine. The machines are compared for the same volume envelope and in terms of their maximum phase back-EMF when the DC field excitation current equals zero and when it is varied with both positive and negative flux paths, and whether the permanent magnets are at a risk of demagnetization due to the total field weakening by the wound field excitation, as summarized in Table 3.12. The volumes and component masses for the different HPM machines are calculated and compared in Table 3.13. In all of the selected HPM machine topologies, the magnetic flux path is radial except the CPPM design, which has both radial and axial flux paths when it is operated with zero or negative excitation currents and a radial flux path when

3.7 Comparison Between PM and Four HPM Machine Topologies

91

Fig. 3.35 PM and several HPM machine models used in the analysis. (a) PM (b) HESM (c) DESM (d) SynPM (e) CPPM Table 3.12 Comparison of maximum back-EMF per phase for the selected HPM topologies and the reference PM machine Machine type PM HESM Excitation If ¼ +4.2 A If ¼ 4.2 A If ¼ 0 A % variation of EMFmax/phase PM demagnetization Rotor excitation

DESM

CPPM

SynPM

70.21 54.50 64.55 24.3 Some Yes

30.23 21.99 26.10 31.57 No No

(65.39) 54.53 32.24 (65.39) 44.53 24.4 No Yes

EMFPk/phase (V)

70 0 No No

60 30 45 66.7 No Yes

Note: Rotor Jrs ¼ 7.07 MA/m2; nf ¼ 100 turns; speed ¼ 3 krpm; and machine axial length ¼ 0.04 m

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Table 3.13 Total volume and mass for the different HPM machines

Copper (kg) Iron (kg) PMs (kg) Total mass (kg) Total volume (1023 m3)

Machine type PM HESM 0.652 0.967 10.89 9.923 0.382 0.239 11.93 11.23 1.48 1.45

DESM 1.406 9.579 0.382 11.37 1.45

CPPM 0.917 10.85 0.167 11.81 1.48

SynPM 0.841 10.58 0.312 11.93 1.48

Fig. 3.36 Back-EMF waveform for SynPM and the PM machine. (a) SynPM back-EMF waveforms (b) PM back-EMF waveform

operating with positive excitation. The analysis results demonstrate that all of the HPM topologies are capable of producing similar back-EMF waveforms that may be varied, to a greater or lesser extent, by the secondary source of field excitation. However, the topologies demonstrate different peak amplitudes depending on the excitation current magnitude and polarity except for the SynPM. Excitation of the SynPM field creates a modulated back-EMF waveform due to asymmetry in the air-gap field distribution, as illustrated in Fig. 3.36a and presented in Table 3.12 where the data in parenthesis relates to the PM element of back-EMF and the lower values to the peak of the modulated period. In the SynPM design, some of the rotor poles consist of permanent magnets, B, C, and D in Fig. 3.35d, with an iron tooth third pole, E in Fig. 3.35a. Therefore, the resultant SynPM back-EMF waveform is modulated by the rotor excitation field. This feature is undesirable in the application considered since it leads to a modulation of the machine back-EMF and hence the connecting rectifier DC output. This modulation feature only arises because of the stator winding and pole number combination constraint adopted for the comparison study. For example, if the stator had a conventional short-pitch winding and the rotor pole number was set to a six-pole pair, the rotor field coil would buck or boost the PM-induced back-EMF as opposed to modulating the back-EMF, as illustrated in Fig. 3.36a. However, this

3.7 Comparison Between PM and Four HPM Machine Topologies

93

Fig. 3.37 Comparison between PM flux density distribution points for the HESM and DESM designs when If ¼ 4.2A (field weakening) and at full stator field current. (a) Typical HESM PM operating points at full-load current (b) Typical DESM PM operating points at full-load current

change in pole number would require deviation from the benchmark PM volumetric constraints. Furthermore, the rotor field excitation has a demagnetizing effect at high buck operation, as with the DESM topology of Fig. 3.35c. Hence, the SynPM topology was not considered for further study. By comparing the four topologies in terms of their back-EMF production without excitation current, the DESM shows the highest back-EMF value, but it gives the lowest voltage regulation at 24%, with full excitation current. The HESM flux linkage has short radial paths that minimize leakage flux and hence it gives a high voltage regulation region with rotor excitation at 66%. In terms of rotor permanent magnet demagnetization, the HESM appears to better withstand demagnetization compared to the DESM, as illustrated in Fig. 3.37, showing the PM working points for both the HESM (a) and DESM (b) superimposed on the PM B–H characteristic. This feature is essential because the rotor excitation flux path does not pass through the permanent magnet pieces. The obvious drawback of the HESM and DESM designs is the DC rotor excitation via brushes and slip rings. However, compact exciters would mitigate this problem. The CPPM has no slip rings and brushes, but it has the lowest backEMF and highest leakage flux components. Thus, the HESM presents the most attractive design among the analyzed topologies since it has the highest back-EMF regulation capability, no PM demagnetization risk from the rotor excitation field, low leakage flux, and low part (stator and WF rotor teeth) saturation risk. Moreover, the HESM mass is similar to the other topologies.

94

3.8

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Hybrid Permanent Magnet Machine Design

Conclusion

This chapter has discussed the analysis tools and models developed to assess HPM machine topologies and design the WF rotor of a HESM topology that will form the base machine in subsequent chapters. The FEA analysis was performed using FEMM and used to numerically evaluate several electric machines reported in Chap. 2 against a benchmark PM electromagnetic and volumetric design. Furthermore, several WF rotor designs have been analyzed, from which it was found that the rotor slot area is a key constraining factor. Hence, WF design 1 represents the best design geometry among the proposed designs. The magnetic circuit representation of PM and WF test machines was obtained, and the magnetic flux was calculated by using a lumped parameter equivalent magnetic circuit model and checked by using 2D FEA. The analytical and numerical results for both machines were successfully compared and showed a good correlation. In addition, PM and WF test machines have been investigated for their backEMF waveforms (no-load case) and power rating capabilities (on-load case) for a specified stator slot current density. In addition, several HPM topologies have been analyzed for the same volume envelope. The validity of the FEA results for the PM was confirmed by actual machine measurements. Moreover, the Motor-CAD lumped parameter thermal network software was used to predict the transient temperature rise in some of the HPM machine parts, such as back-iron, stator windings, and WF rotor winding. The predicted results illustrate the temperature rise in the important machine parts due to the natural convection case. Thus, combining both PM and WF machines in one machine housing, as with the HESM, will make the most of their respective advantages.

Chapter 4

Multiphase HPM Generator Systems

4.1

Overview on Multiphase Machines

The concept of multiphase machines is not new, and the roots of multiphase machine modeling can be traced back to 1959 [75] with many papers published to date [76– 137]. Levi et al. and Bojoi et al. have presented a comprehensive survey of existing and future applications for multiphase machines [78–80], reviewing machines found in various applications ranging from electric shipboard generation and ship propulsion, hybrid electric vehicles, wind generation and more electric aircraft, to highpower industrial applications where, for a variety of reasons, a three-phase machine does not satisfy the desired specification requirements, particularly with regard to hardware implementation or fault criticality. Greater efficiency, higher torque density with reduced torque pulsations, lower per-phase power handling requirements, increased fault tolerance, and improved noise characteristics are the major advantages of multiphase machines over their three-phase counterparts, as reported in [81]. The modeling and optimization of multiphase machines in terms of their power density and harmonic content is the subject of considerable interest, since the designs are not standard “off-the-shelf” machines [82, 83]. Levi et al. discuss the degree of freedom merits that are presented by the choice of phase numbers greater than three, an example being multiphase power electronic converters that can be utilized to control independent machines within a multi-motor drive system, or to enhance the torque production above that of standard three-phase machines by injecting higher stator current harmonics [78, 84]. In drive system applications, Singh et al. presented a straightforward approach to developing an indirect field oriented control scheme for a six-phase induction machine with an arbitrary displacement between the two three-phase winding sets [77]. In addition, high level drive control via a space vector pulse width modulation algorithm for a multilevel, multiphase, voltage source converter was presented by Lopez et al. [81]. The losses and pulsating torque in multiphase induction machines have been described by Williamson and Smith in [82]. In addition, Brisset et al. have presented a comparative study of different © Springer Nature Switzerland AG 2022 A. S. Al-Adsani, O. Beik, Multiphase Hybrid Electric Machines, https://doi.org/10.1007/978-3-030-80435-0_4

95

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4 Multiphase HPM Generator Systems

configurations of nine-phase, concentrated winding, axial flux, permanent magnet synchronous generators for direct-drive wind turbines [83]. Moreover, Arwyn et al. have investigated a high phase number flux-switching permanent-magnet brushless machine via 2D FEA. They presented a comparison of the electromagnetic performances of three-, four-, five-, and six-phase machine variants within the specific context of an aerospace application [85]. Arwyn concluded that the high phase number offered more torque density over conventional three-phase machines along with lower sensitivity to the level of rotor saliency [85].

4.1.1

Multiphase Windings Principles

To realize near-sinusoidal back-EMF waveform for the AC machines, more slot per pole per phase is required, and hence, as the phase number increases, it becomes gradually difficult to achieve that. However, matching a quasi-rectangular backEMF waveform by stator current waveform provides enhanced torque production. The multiphase machines are divided into symmetrical or asymmetrical machines based on their electrical displacement between two adjacent phases. A symmetrical multiphase machine is attained once the electrical angular displacement between any two stator phases equal 2π/n; this can always be achieved if the phase number (n) is odd prime number, such as 5, 7, 11, .... Where the number of phases is an even or an odd and not a prime number, such as 4, 6, 8, 9, 10, ..., the multiphase machine phase winding can be realized in a different way [78]. In this way, the machine windings can be seen as the number of sets having a phases each. Such that, the number of phases equals a ∙ k, where for multiphase (above three) machines a ¼ 3, and k ¼ 2, 3, 4, 5, . . ., and the electrical angular displacement between the first phases of two consecutive sets is π/n electrical degree, leading to asymmetrical multiphase machine [80]. In the literature, many multiphase machine designs and applications have been reported, as illustrated in Tables 4.1 and 4.2. It is shown in the tables that different electric machine types with more than three-phase winding schemes can be utilized in several industrial applications.

4.1.2

Rectified Voltage due to Three- and Nine-Phase HPM Generator Systems

An interesting feature of the design studied in Chap. 3 was the stator tooth and rotor pole combination. The 36-slot, 32-pole combination was primarily a design feature to help mitigate against cogging torque. However, the HPM-rectified output voltage of the three-phase winding scheme must be conditioned to remove the harmonics introduced by the passive rectification stage. Traditionally, this is facilitated via electrolytic capacitors and line inductors placed on the DC side of the output of the

4.1 Overview on Multiphase Machines

97

Table 4.1 References of multiphase electric machine types and applications Application Ship propulsion

Shipboard generation Electric and hybrid electric vehicles

Wind generation

More electric aircraft

Machine type IM Brushless PM Synchronous WF Synchronous WF Brushless PM Synchronous WF Brushless hybrid PM Brushed hybrid PM IM IM Brushless PM Brushless DC Brushless DC Brushless PM Brushed hybrid PM Brushless axial flux PM Brushless PM Brushless PM IM Switch reluctance Switch reluctance Brushless PM Brushless PM Brushless DC

Number of phases 15 12 9 6 12 6 9 9 6 and 9 6 5 5 4 and 5 12 9 9

References [89] [86, 87] [90] [91] [88] [92] [94] [93, 95, 96] [76] [97] [98] [99–101] [102] [114] [103–109, 115] [83, 111, 112]

6 and 9 6 6 6 4 3, 4, 5, and 6 4 5

[113] [110] [116] [119] [118] [85] [117] [120, 121]

Table 4.2 References of general multiphase machines modeling and analysis Machine type IM IM IM IM Brushed hybrid PM Brushless PM Brushless PM Brushless axial flux double rotor PM Brushless PM Brushless PM Switch reluctance

Number of phases 4, 5, 6, 9, and 15 3, 4, 6, and 12 6 5 9 7 and 9 7 7 5 2, 3, and 5 5

References [82] [128] [122, 123] [124, 125] [46] [136] [133, 134] [137] [131, 132] [135] [129, 130]

98

4 Multiphase HPM Generator Systems

Fig. 4.1 Three- and nine-phase HPM generator with power circuit converter. (a) HPM model. (b) Three-phase. (c) Nine-phase

full-bridge rectifier (AC/DC). The move to higher phase numbers reduces the level of ripple voltage on the generated DC output voltage and thus reduces the required capacitor and/or inductor passive device rating, as will be demonstrated later. Thus, for the chosen machine topology, an evaluation can be made and benefits can be quantified. Figure 4.1 illustrates the HPM machine showing the wound field rotor and PM rotor cross sections schematically (a), and the three-phase (b) and ninephase (c), equivalent circuits when connected to a simple full-bridge rectifier and resistive load. In terms of the machine phase peak back-EMF voltage (EMFPk), the

4.1 Overview on Multiphase Machines

99

general form of the average DC-link voltage for loss-less full-bridge rectification is given by [138]: V DClink

4 ¼ ð2π=nÞ

Z

π=n

EMF Pk cos ðωt Þd ðωt Þ ¼ EMF Pk

0

  2n π sin π n

ð4:1Þ

which, for a three-phase system, simplifies to V DClinkð3phaseÞ ¼

pffiffiffi 3 3 EMF Pk ¼ 1:654EMF Pk π

ð4:2Þ

and for the nine-phase system, (4.1) solves to V DClinkð9phaseÞ ¼

6:1564 EMF Pk ¼ 1:96EMF Pk π

ð4:3Þ

This average DC output voltage is derived from the machine AC voltage waveform, as illustrated in Fig. 4.2 for the three- and nine-phase cases, from which the DC voltage ripple can be defined: V r ¼ V DClinkð max Þ  V DClinkð min Þ

ð4:4Þ

From Fig. 4.2, it can be seen that the DC-voltage ripple reduces with increased phase number and has a frequency of 2n times that of the machine fundamental frequency [139]. Moreover, the move to a higher phase number distributes current across a larger number of rectifier legs providing the designer with an opportunity to use rectifiers having lower power loss (but similar peak current) distributed around the periphery of the machine, leading to the potential for improved packaging and cooling. With the advent of higher temperature rectifier devices, for example, silicon carbide (SiC) [140], the rectifiers could potentially be mounted within the machine housing. However, at high operating frequency, the machine synchronous inductance, LS, can have a significant effect on output power and voltage regulation that might lead to an incorrect decision regarding the feasibility of utilizing the HPM generator with a passive rectification stage. If a DC voltage is to be generated by a three-phase machine with a sinusoidal back-EMF via a passive, full-bridge, three-phase rectifier, then (4.2) is valid only for the no-load case. Under load conditions, there will be voltage drops due to the diode reverse recovery time (trr) causing commutation overlap, and due to machine synchronous inductance, LS. If a linear rise of DC generated current (IDC-link) from zero to the rated value is assumed or a constant di/dt, the influence of the machine operating frequency, synchronous inductance, and load current in terms of DC-link output voltage drop can be calculated from [138]:

100

4 Multiphase HPM Generator Systems

Fig. 4.2 AC L-L voltage and rectified DC output for three- and nine-phase machines at no-load. (a) Three-phase. (b) Nine-phase

4.2 Nine-Phase HPM Generator Parameters

V dr ¼ 2 n f e LS I DClink

101

ð4:5Þ

Therefore, limitations are imposed on the DC output power and voltage regulation for both three- and nine-phase configurations; hence, LS is evaluated for both topologies.

4.2

Nine-Phase HPM Generator Parameters

Having studied the individual coil and phase back-EMFs, as in Chap. 3, the possibility of connecting the stator as a nine-phase winding became apparent. Importantly, the stator lamination, rotor magnetization, and other key magnetic dimensions did not have to be changed; hence the relative merits of a three-phase versus a nine-phase winding topology could be evaluated fairly. Thus, a general investigation into multiphase machine topologies is undertaken via a direct three- to nine-phase stator winding comparison study.

4.2.1

Nine-Phase Winding Layout and Back-EMF

As discussed in Chap. 3, the WF rotor of the HPM generator is fed from an external DC source via slip rings and brushes. The WF rotor lamination has 32 slots, and the WF DC winding is formed of 32 series coils woundaround the rotor teeth. Here, the phase belt (i.e., number of slots per pole per phase) for the nine-phase HPM stator winding is 1/8, and with subsequence phases being displaced by 40 electrical, hence a concentric winding with 36 coils wound around stator teeth is realized. For the three-phase winding, as in Chap. 3, each phase has four sets of three concentrated coils connected in series while for the nine-phase winding, there are two design variations where four coils are connected in series, as illustrated in Fig. 4.3. Note that the middle coil of the three concentrated coils is wound opposite to the other two coils, which results in the largest amplitude back-EMF in the proposed three-phase winding. The fundamental winding factor (kw1) is calculated again using the methods described in [46], which incorporates both the slot pitch and distribution factor in one equation. Therefore, by numbering the stator slots using vector U, the phase back-EMF vectors (Ei) and fundamental winding factor (kw1) are calculated for both nine-phase winding designs using Eqs. (4.5, 4.6, 4.7, 4.8, 4.9, and 4.10).

kw1

  nl S=9  1  X ! ¼ E  nl S=9  i¼1 i 

ð4:6Þ

102

4 Multiphase HPM Generator Systems

Fig. 4.3 Two nine-phase HPM machine stator lamination and phase (A) winding schemes. (a) Nine-phase winding design 1. (b) Nine-phase winding design 2. (c) Design 1 (A) phase back-EMF configuration vectors. (d) Design 2 (A) phase back-EMF configuration vectors

U design 1 ¼ ½1  2  2 3 19  20  20  21  36  2 9    X ! πP πP πP πP πP πP  Ei  ¼ e j S 1  e j S þ 2e2j S þ e18j S  2e19j S þ e20j S   i¼1   36  2 9  X !  E i  ¼ e   i¼1 

U design 2 ¼ ½1  2 jπP S



1e

jπP S

πP

10  11 πP

19  20 πP

ð4:7Þ ð4:8Þ

28  29 πP

πP

ð4:9Þ πP

þ e9j S  e10j S þ e18j S  e19j S þ e27j S  e28j S



ð4:10Þ

The nine-phase winding design 2, Fig. 4.3b, results in 1.53% and 4.20% higher fundamental winding factor and hence back-EMF as compared to design 1 and the

4.2 Nine-Phase HPM Generator Parameters

103

Fig. 4.4 Normalized back-EMF waveforms of the three- and nine-phase HPM machines

previous three-phase windings. Therefore, nine-phase winding design 2 is adopted as the preferable HPM generator winding scheme. The higher phase voltage translates to a better power density given a fixed load current. In addition, going to a higher number of phases, the rectified output voltage has lower ripple, better quality, and hence lower requirements for filtering. To numerically evaluate the HPM machine design performance, a reliable model is required. FEA is used to solve the machine electromagnetic field and hence determine the machine terminal parameters in the presence of complex magnetic circuit geometry and nonlinear material properties, as discussed in Chap. 3. These parameters are then used in a Matlab/ Simulink SimPower model to simulate the machine and nonlinear effects of the diode rectification schemes. Phase back-EMF waveforms for the coil distributions shown in Fig. 4.3 were calculated for the three- and nine-phase designs, as illustrated in Fig. 4.4, showing the back-EMF coefficient waveforms with normalized 1.0 p.u. magnitudes and electrical rotor angle. The utility of Fig. 4.4 is simply to illustrate the waveform shape since it is envisaged that the more trapezoidal EMFs could yield more torque and hence power capability than a winding designed to realize a sinusoidal backEMF. The sinusoidal back-EMF is considered since such a machine leads to easier analytic performance estimation.

4.2.2

Back-EMF and Torque Waveform Harmonics Prediction

In this section, 2D FEA analyses are carried out on the back-EMF waveform at no-load for three- and nine-phase generator winding schemes for the PM, WF, and the HPM machine. For normalized back-EMF waveforms, a fast Fourier transform (FFT) is applied on expected operating scenarios in HEV powertrain to investigate

104

4 Multiphase HPM Generator Systems

Fig. 4.5 Amplitudes of normalized back-EMF harmonic orders due to different WF excitation modes. (a) Three-phase HPM machine. (b) Nine-phase HPM machine

Fig. 4.6 Predicted buck/boost capabilities for back-EMF and torque waveforms of the nine-phase HPM machine. (a) back-EMF at no-load. (b) electromagnetic torque

the THD based on the fundamental, third, fifth, and seventh harmonic orders. Figures 4.5a and b show harmonic spectrum for normalized back-EMF voltage waveform at no-load for three WF excitation scenarios. These scenarios are characterized as (i) boost (If ¼ +3A), (ii) buck (If ¼ 3A), and (iii) normal (If ¼ 0A) operation modes, as shown for the nine-phase machine in Fig. 4.6a. The FFT analyses have shown an improvement in the third and fifth harmonic amplitude at buck mode for three- and nine-phase winding schemes compared to buck and normal operation modes. In addition, the nine-phase winding scheme showed larger third and fifth harmonic amplitude compared to the three-phase winding; hence, larger back-EMF waveform harmonic distortion is expected in the nine-phase HPM machine scheme.

4.2 Nine-Phase HPM Generator Parameters

105

Table 4.3 Predicted back-EMF THD for three- and nine-phase PM, WF, and HPM machines Machine type Three-phase PM Three-phase WF Three-phase HPM (boost) Three-phase HPM (buck) Nine-phase PM Nine-phase WF Nine-phase HPM (boost) Nine-phase HPM (buck)

EMF (rms) V 0.79 0.77 0.78

EMF1/ EMF (%) 96.3 96.6 96.4

0.80

96.2

0.84 0.80 0.83 0.85

EMF3/ EMF1 (%) 7.79 5.88 7.35

EMF5/ EMF1 (%) 0.94 0.35 0.74

EMF7/ EMF1 (%) 0.05 0.28 0.07

THD (%) 7.85 5.90 7.39

8.60

1.32

0.14

8.71

95.5 96.2 95.7

12.8 8.51 11.8

2.35 1.38 1.51

0.07 0.59 0.10

13.0 8.67 11.9

95.1

14.5

3.88

0.32

15.0

Further analysis of the normalized back-EMF waveforms is illustrated in Table 4.3. The three-phase PM part of the HPM machine has a higher back-EMF THD than the three-phase WF machine part. The three-phase HPM machine backEMF at buck mode showed a higher THD compared with all of the three-phase machine scenarios, as shown in Table 4.3. Moreover, degradation in the THD presented by the more trapezoidal back-EMF waveform in the nine-phase PM part compared with the three-phase and the nine-phase WF part is noticeable. At normal mode scenario, the THD is 12.98%, and at boost mode, the THD drops to 11.9%; however, at buck mode, the THD increased to 15.01%. Thus, the nine-phase HPM machine operating at buck mode scenario provides the highest THD for the normalized back-EMF waveform among all of the scenarios. Similarly, the electromagnetic machine torque waveform is analyzed via 2D FEA at rated load for the three- and nine-phase winding schemes. Here, a stator slot current density of 3.4 MA/m2 is chosen based on the allowable machine thermal limit at rated load. The HPM machine with three WF excitation scenarios is applied on the three- and nine-phase winding schemes. As shown in Fig. 4.6b, a 36% buck/ boost torque capability is achieved for the HPM machine. To investigate the electromagnetic torque THD, an FFT is applied on the expected HEV powertrain operating scenarios. The second, fifth, eighth, and so on harmonics are the negative sequence harmonics that create a negative torque, where the fundamental, fourth, seventh, tenth, and so on harmonics present a positive torque sequence. Furthermore, the multiple of third harmonics, such as the third, sixth, ninth, and so on contribute to the zero sequence, which results in a circulating current in the machine and needs to be accounted for in unbalanced systems. Figure 4.7 shows the harmonic spectrum for normalized electromagnetic torque at rated load for three- and nine-phase HPM machines. The FFT shows that at normal operation (zero WF excitation), some of the nine-phase harmonic amplitudes are less than three-phase winding scheme, such as the second, fourth, and fifth harmonics. Hence, the buck mode harmonics are found to be dominant in the three- and

106

4 Multiphase HPM Generator Systems

Fig. 4.7 Amplitudes of normalized electromagnetic torque harmonic orders due to different WF excitation modes. (a) Three-phase HPM machine. (b) Nine-phase HPM machine Table 4.4 Predicted electromagnetic torque THD for three- and nine-phase PM, WF, and HPM machines Machine type Three-phase PM Three-phase WF Three-phase HPM (boost) Three-phase HPM (buck) Nine-phase PM Nine-phase WF Nine-phase HPM (boost) Nine-phase HPM (buck)

T (rms) Nm 0.70 0.69 0.70 0.70 0.70 0.69 0.70 0.71

T1/T (%) 87.2 87.1 87.2 87.1 96.8 96.7 96.8 96.8

T3/T1 (%) 0.30 2.93 0.98 1.18 0.31 1.69 0.67 0.45

T5/T1 (%) 0.55 0.33 0.33 1.04 0.12 0.04 0.10 0.17

T7/T1 (%) 0.04 0.16 0.11 0.15 0.19 0.09 0.14 0.31

THD (%) 0.89 4.03 1.24 2.63 0.51 4.01 1.23 2.04

nine-phase HPM machines. It is also noticed that there is a small increase in the second and eighth spatial harmonics in the nine-phase winding scheme in the buck mode, where the other harmonics decrease compared to three-phase winding. The FFT showed an improvement in the nine-phase HPM machine harmonic contents over three-phase in boost mode except for the second harmonic, which is responsible of the excitation of the second mechanical deformation (distorting) and therefore it is significant [141]. Table 4.4 illustrates the predicted torque THD for the PM, WF, and HPM machines.

4.2 Nine-Phase HPM Generator Parameters

4.2.3

107

Synchronous Inductance Prediction

The calculation of machine synchronous inductance (LS) based on stator winding schemes has been discussed in the literature [142–147]. The most common method for analytically predicting the self- and mutual-inductances based on magnetic fluxlinkage is presented by the winding function method [142]. This method uses a Fourier series to predict a winding function for the machine phase winding configuration and the number of poles. This function is utilized in common flux-linkage, self-, and mutual-inductance equations [144]. El-Refaie and Thomas have discussed a closed-form technique for analyzing surface PM machines equipped with fractional-slot concentrated windings as per the machine being studied. The technique takes advantage of some established analytical methods to analyze a wide range of fractional-slot concentrated winding configurations [145]. Furthermore, the impact of winding layer number on surface PM machine synchronous inductance was investigated by El-Refaie and Thomas [146]. Hence, based on the comparison between self- and mutual-inductances via analytic, FEA, and measured values, as presented by El-Refaie et al. [147], it can be concluded that FEA prediction can give satisfactory results if the end winding effect is properly taken into account. In this book, the HPM machine self- and mutual-inductances for the PM and WF machines are predicted with 2D FEA models, and an empirical factor is applied to account for end-winding inductances. To consider the contribution from the self- and mutualinductance components, the series phase coils of both the three- and nine-phase machines can be reduced to a single effective coil. The machine magnetic field is then solved without the PM or WF rotor excitation. The self-inductance of this single effective coil, which represents the equivalent phase coil, can be calculated by (4.11). For the case of the PM part of the HPM machine, the rotor PM pieces are changed to air in the FEA model, DC current is injected in the phase coil for which the self- and mutual-inductances are calculated, and the other phase currents are set to zero. The FEA results for the self-inductance is based on (4.11), while the mutualinductance between, for example, phases (A) and (B), is based on (4.12) [148]: LSelf ¼

λ e ne λ t ¼ I in I in

M¼

λel I in

ð4:11Þ ð4:12Þ

where λe is the total phase flux-linkage, ne is the number of turns per phase, λt is the flux-linkage per turn per phase, Iin is the injected phase current, and λel is the fluxlinking with the coupled coil of interest, for example, the effect of phase A excitation on phase B. A similar procedure can be followed to obtain the self- and mutualinductances for the other phases. Table 4.5 presents the self- and mutual-inductances for both the PM and WF sections of the HPM machine stators, as predicted by FEA. Note that calculations are presented with and without the respective rotors since

108

4 Multiphase HPM Generator Systems

Table 4.5 Comparison of three- and two nine-phase HPM generator inductances calculated from FEA with and without rotor

Item Threephase nstc ¼ 6

Stator PM

Ninephase nstc ¼ 14

PM

Ninephase nstc ¼ 18

WF

WF PM WF

WF rotor inductance (with stator) Item Lf (mH) 68 Threephase nstc ¼ 6 68 Ninephase nstc ¼ 14 68 Ninephase nstc ¼ 18 a

Mutual inductances between phases as a % of LSelf MAD MAE MAC MAG MAF MAH LSelf (μH) MAB 146.0 7.5 7.5 Not applicable (123)a (9.1)a (9.1)a 178.7 4.9 4.9 (49.3)a (9.1)a (9.1)a 190.5 6.0 5.8 5.9 31.4 5.8 (140.4)a (1.9)a (0.7)a (0.94)a (45)a (0.7)a 272.7 10.9 10.0 10.3 18.5 10.0 (56.1)a (1.9)a (0.7)a (0.94)a (45)a (0.7)a 317.7 6.0 5.8 5.9 31.4 5.8 (233)a (1.87)a (0.71)a (0.94)a (49.7)a (0.71)a 331.0 10.9 10.0 10.3 18.9 10.0 (92.7)a (1.87)a (0.71)a (0.94)a (47.3)a (0.71)a Mutual inductances between WF and stator phases as a % of Lf MfA

MfB

MfC

2.67

0.56

2.02

2.60

MAI

6.0 (1.9)a 10.9 (1.9)a 6.0 (1.86)a 10.9 (1.87)a

MfH

MfI

3.39

MfD MfE MfG MfF Not applicable

1.48

0.29

0.89

1.89

0.29

1.48

1.90

0.38

1.14

2.40

0.38

1.90

Values without rotor

early tests to confirm the FEA predictions were carried out on stator assemblies as opposed to full machine constructions. To consider the machine synchronous inductance, the voltage equation for one phase of a three-phase machine winding, as derived from first principles, is given by [149]: vA ¼ RA iA þ

dLSelf iA dM AB iB dM AC iC þ þ þ EMF A dt dt dt

ð4:13Þ

If the machine self- and mutual-inductances are not a function of their respective phase currents, or rotor position, which is the case for normal operation of the machine studied, (4.13) simplifies to

4.2 Nine-Phase HPM Generator Parameters

vA ¼ RA iA þ LSelf

109

diA di di þ M AB B þ M AC C þ EMF A dt dt dt

ð4:14Þ

For balanced winding construction and phase current excitation, that is, M AB ¼ M AC ¼ M CA ¼ M

ð4:15Þ

and iA þ iB þ iC ¼ 0 diA diB diC þ þ ¼0 dt dt dt thus,   diA diB diC ¼ þ dt dt dt

ð4:16Þ

Hence, (4.14) reduces to vA ¼ RA iA þ ðLSelf  MÞ

diA þ EMF A dt

di vA ¼ RA iA þ LS A þ EMF A dt

ð4:17Þ

where LS is usually termed the phase synchronous inductance. It can be shown that for an ideal, balanced sinusoidal winding distribution excited with balanced sinusoidal phase currents, the phase synchronous inductance becomes [149]:    2π 1 ¼  LSelf  LL M ¼ LSelf  LL cos 3 2

ð4:18Þ

where LL is the leakage inductance and hence, for classical AC (sinusoidal) machine analysis, LSð3phaseÞ ¼

3 LSelf  LL 2

ð4:19Þ

However, for the machine of interest, the phase windings are formed from concentrated coils; hence, the self- and mutual-inductances are calculated via FEA to yield LSð3phaseÞ ¼ 1:075LSelf

ð4:20Þ

Thus, the mutual coupling and hence inductances have less significance for the concentrated winding of interest. For the nine-phase machine configuration, the

110

4 Multiphase HPM Generator Systems

Fig. 4.8 Single coil representation of the three- and nine-phase HPM machines

spatial displacement between the single stator coils is 40 electrical, as illustrated in Fig. 4.8, which also shows the three-phase coil schematic for completeness. For the nine-phase machine, the method for calculating the synchronous inductance can be as per the three-phase case, but extending the equations to account for the mutual couplings of all coils. Thus, with the same assumptions and expanding (4.14), the phase voltage equation is now diA di di þ M AB B þ M AC C dt dt dt diD diE diF þ M AD þ M AE þ M AF dt dt dt di di di þ M AG G þ M AH H þ M AI I þ EMF A dt dt dt

vA ¼ RA iA þ LSelf

ð4:21Þ

where by symmetry MAB ¼ MAI, MAC ¼ MAH, MAD ¼ MAG, MAE ¼ MAF and (4.21) can be rewritten as   diA diB diI þ M AB þ vA ¼ RA iA þ LSelf dt dt dt     diC diH diD diG þ M AC þ þ M AD þ dt dt dt dt   diE diF þ M AE þ þ EMF A dt dt

ð4:22Þ

4.2 Nine-Phase HPM Generator Parameters

111

Again, it can be shown that for an ideal, balanced sinusoidal winding distribution excited with balanced sinusoidal phase currents, the phase synchronous inductance becomes    2π M AB ¼ LSelf  LL cos 9    4π M AC ¼ LSelf  LL cos 9    6π M AD ¼ LSelf  LL cos 9    8π M AE ¼ LSelf  LL cos 9

ð4:23Þ ð4:24Þ ð4:25Þ ð4:26Þ

hence, for classical AC (sinusoidal) machine analysis, LSð9phaseÞ ¼

9 L  LL 2 Self

ð4:27Þ

However, for the machine of interest, the phase windings are found from concentrated coils; hence, the self- and mutual-inductance values are again calculated via FEA to yield LSð9phaseÞ ¼ 1:67LSelf

ð4:28Þ

As with the three-phase winding, the mutual couplings have less significance than the phase self-inductance to the total synchronous inductance value.

4.2.4

Construction of HPM Machines Prototype

The construction process of the HPM machine prototype based on the final stator windings layout, as in Figs. 3.3a and 4.3a, and design specifications as in Table 3.9, is illustrated in Fig. 4.9. Once the two stators with different axial thickness are soldered, as in Fig. 4.9a, they are wound in the three- and nine-phase configuration. For each HPM winding configuration, two stators with 25 and 10 mm axial length are wound as a three-phase configuration, and the other two stators are wound as a 14-turn nine-phase configuration. Based on the calculated lengths and parallel conductors per-turn for the three- and nine-phase configuration, the windings of the different phases are prepared for the hand winding, as shown in Fig. 4.9b. One of the important steps that are required before winding is preparing the slot liners to protect the stator winding from damaging the insulation layer. Figure 4.9c shows the four wound stators of three- and nine-phase HPM machines.

112

4 Multiphase HPM Generator Systems

Fig. 4.9 HPM machine stator fabrication with both winding configurations. (a) Welded stator pack. (b) Preparation of stator windings. (c) Four wound stators

The HPM machine rotor consists of three parts: shaft, PM rotor lamination, and WF rotor lamination. The PM rotor back-iron was carefully prepared to ensure a clean and smooth surface for the permanent magnets to be glued on to get a similar PM rotor assembly as in Fig. 3.2b. As for the WF rotor lamination design is highlighted in the dark dotted circle in Fig. 4.10a. The construction process is similar to that of the stator. For the WF rotor, the DC field coils were professionally handwound to ensure a high packing factor. Even so, 80 turns could be fitted in and not

4.2 Nine-Phase HPM Generator Parameters

113

Fig. 4.10 Assembly of 10-mm WF rotor of the HPM machine section. (a) WF rotor iron sheet. (b) Finalized WF rotor

Fig. 4.11 HPM machine final prototype assembly. (a) Rotor with plain shaft. (b) Machine with finalized rotors. (c) Machine viewed at no coupling side

100 turns giving a packing factor of 0.48. Figure 4.10b shows the finalized WF rotor after the vacuum pressure impregnation (VPI) and backing process. Figure 4.11 shows the HPM machine rotor while it is placed inside the machine for the plain shaft and overall rotor-designed hardware viewed at the WF rotor. Finally, two holes

114

4 Multiphase HPM Generator Systems

were drilled into the HPM machine housing to assure adequate access to the stator phase windings of both stators along with the fixation of brushes and slip rings on the machine shaft, as illustrated in Fig. 4.11c.

4.2.5

Resistance and Inductance Measurements

As mentioned previously, the predictions of synchronous inductance for the threeand nine-phase stator winding configurations do not include winding end effects due to the 2D FEA limitations. Hence, some consideration is given since the machine axial length to diameter ratio is relatively small. Different analytical approximation methods that predict end-winding inductance have been reported in [147] and [150– 153], for example. However, these techniques tend to be empirical and associated with some uncertainty. The most appropriate method to determine end-winding effects is 3D FEA [152] or from a series of measurements made on machines of varying axial length. The latter was considered in this book, where actual winding self-inductances of the assembled HPM machine stators were measured via an Agilent 4284, a precision LCR meter having a measuring bandwidth of 20 Hz to 1 MHz. The measured inductance results can be characterized as in (4.29). LSelf m ¼ 2Lend þ L

ð4:29Þ

where LSelf-m is the measured self-inductance, Lend is the end-winding inductance, and L is the self-inductance determined from 2D FEA. Note L equals LPMa for the PM stator and LWFa for the WF stator. Thus, as the machine axial length increases, the impact of the end-winding inductance on total inductance reduces, as shown in Table 4.6. Furthermore, for the three-phase machine case, (4.20) has also been verified by measuring the line-to-line inductance, which includes both self- and mutual-line-inductance values, and dividing this value by 2, and then subtract the self-inductance from this value in order to obtain the line mutual-inductance that will Table 4.6 Comparison of three- and nine-phase HPM generator measured inductances and resistance

Stator Item section PM Threephase WF nstc ¼ 6 PM Ninephase WF nstc ¼ 14 WF rotor (with stator)

Inductance (μH) (with rotor) EndSelf-winding winding FEA Measured 2Lend 146.0 175 29.0 178.7 308 129

Resistance (mΩ) Predicted Measured 40.84 50.00 20.95 31.00

190.5 272.7

217 481

78.56 55.05

90.00 70.00

68  103

91  103

13.06  103

11.35  103

26.5 208 23  103

4.3 Analysis Models

115

Table 4.7 Mutual-inductance calculation based on line-to-line inductance measurements of the three-phase machine LSelf (measured) (μH) 175

2(LSelf +M ) (measured) (μH) 372

PM part of the HPM machine (LPMa ¼ 0.025 m) Three-phase (nsct¼6)

M (measured) 0.063LSelf

M (predicted) 0.075LSelf

be added to the self-inductance to get the line synchronous inductance as in Table 4.7. The disparity between FEA and measured inductance values was predicted, and it showed satisfactory results. Furthermore, sensitivity analysis has been carried out, as in Sect. 4.4.2, to account for and predict the impact of different LS values on the analyzed system outcomes. Moreover, the stator winding resistance (Rs) was calculated for the three-phase HPM machine stators in Chap. 3, as given in Table 3.8. In this section, the resistance for the 14-turn nine-phase scheme is predicted and measured, as given in Table 4.6. Again, the shown results illustrate the comparison between the three- and nine-phase stator resistances and predicted and measured WF rotor resistance, along with the difference between the predicted and measured data of the total stator winding resistances, which is due to the mean end winding approximation and measurement error.

4.3

Analysis Models

In this section, a generalized mathematical model is given to support the classical sinusoidal waveform analysis only of the HPM generator along with simulation dynamic models of three- and nine-phase HPM generator systems.

4.3.1

General dq Mathematical Model of HPM Generator

The HPM generator is operating at all times to supply the vehicle DC-link with constant power at driving mode and charge the battery at vehicle stop mode. The generalized mathematical model of the HPM generator, considered here, is presented in the dq coordinates with the rotor excitation field on the d-axis. In the d coordinate system, the steady-state voltage equation is expressed as in (4.30) [154]:

V d  Vq



¼

Rs

ωLq

ωLd

Rs



Id Iq

λHPM¼ λPM  λWF

 þ ωλHPM

 0 1

ð4:30Þ ð4:31Þ

116

4 Multiphase HPM Generator Systems

Fig. 4.12 Equivalent steady-state circuits of the HPM machine in the dq axes. (a) d-axis. (b) q-axis

λWF ¼ kex I f

ð4:32Þ

where Id, Iq, Vd, Vq, Ld, and Lq are the d and q components of armature currents, terminal voltages, and armature synchronous inductances, respectively, λHPM is the amplitude of the total excitation flux-linkage in the reference d-axis, λPM is the amplitude of the PM excitation flux-linkage in the reference d-axis, λWF is the amplitude of the WF excitation flux-linkage in the reference d-axis, kex is the wound field excitation constant, and Rs is the armature resistance. Figure 4.12 represents the equivalent circuits in dq axes of the HPM machine. Hence, by introducing the steady-state phasor results, as in (4.33, 4.34, 4.35, 4.36, 4.37, and 4.38) [35], the phasor diagram of the HPM generator for two operating cases is illustrated in Fig. 4.13. V s ∠δ þ ðRs þ jX s ÞI s ∠δ ¼ EMF d ∠0 λs ¼ λd þ jλq



ð4:33Þ ð4:34Þ

4.3 Analysis Models

117

Fig. 4.13 HPM generator phasor diagram in steadystate region for three operating cases. (a) Phasor diagram with zero wound field excitation mode from no-load to rated resistive load. (b) Phasor diagram in the voltage regulation mode at constant resistive load

λq ¼ L q I q

ð4:35Þ

λd ¼ Ld I d þ λHPM

ð4:36Þ

 jX s I s ¼ jω Ld I d þ jLq I q

ð4:37Þ

118

4 Multiphase HPM Generator Systems

EMF d ¼ jωλs ¼ jωλHPM

ð4:38Þ

where Xs equals 2πfeLS. In this book, the dynamics of the HPM generator variables are characterized initially by these two operating cases. Figure. 4.13a presents the HPM generator phasor diagram due to a gradual increase in a resistive load with zero wound field excitation, as will be investigated in Sect. 4.4. For the induced back-EMF regulation mode at constant resistive load, the HPM generator phasor diagram is illustrated in Fig. 4.13b. δ in Eq. (4.33) and Fig. 4.13 presents the angle of the phase voltage Vs and phase current Is with respect to the reference back-EMF vector.

4.3.2

Simulation Model

The HPM generator system’s dynamic model consists of two major sections. The first section generates the machine back-EMF waveforms. Here, by using normalized sinusoidal and trapezoidal back-EMF waveforms, obtained analytically and via FEA data, the control over machine back-EMF frequency and amplitude for all of the machine phases becomes relatively simple, as in (4.39) and (4.40), where ωnorm is the normalized angular speed at 3 krpm. EMF φ1 ¼ ωðλPM  λWF Þ cos ðωnorm t Þ ⋮

⋮

⋮

ð4:39Þ

⋮

EMF φn ¼ ωðλPM  λWF Þcos ðωnorm t 

2π Þ n

ð4:40Þ

The second section is presented by the line resistance, synchronous inductance, and passive rectification stage that link machine terminal voltage with a resistive load bank, which is modeled simply by utilizing SimPower built-in blocks with certain modification to their internal values, where these predicted internal values of the phase resistance, synchronous inductance, diode internal resistance, and diode voltage drop along with an adequate analysis time step will account for more accurate system behavior. The three- and nine-phase machine dynamic models are based on the following assumptions. • The three-phase stator winding is based on a benchmark assumption that utilizes a distributed winding to produce a sinusoidal back-EMF, so the space harmonics of the magnetic field of the stator are neglected and on the FEA trapezoidal backEMF waveforms based on the actual machine with concentrated windings. • The nine-phase stator winding is based on the FEA trapezoidal back-EMF waveform. • The iron losses in the stator and rotor are neglected. • Nonlinear effects of saturation and hysteresis are neglected.

4.4 Three- and Nine-Phase HPM Generator System Studies

119

Fig. 4.14 Three-phase HPM generator with power circuit converter dynamic model

• The stator and rotor winding resistances are not a function of frequency or temperature. Figures 4.14 and 4.15 illustrate the dynamic model of the three- and nine-phase HPM generator systems that are adopted in this book to analyze the system outcomes, such as DC-link ripple, voltage regulation, and average output power due to different internal parameters and loading conditions, which are considered the main issue of this chapter.

4.4

Three- and Nine-Phase HPM Generator System Studies

In this section, a comparison of three- and nine-phase HPM generators with different back-EMF waveforms and stator number of turns for the same slot current density and machine geometry will be presented along with the quality and regulation of both the machine and DC-link voltage terminals. Note that the analysis undertaken in this chapter is for the case of zero WF excitation. Simoes and Vieira present a general form of output power calculation of any machine [99]: n Po ¼ T

Z

T

EMF ðt Þiðt Þdt ¼ nK p EMF Pk I Pk

ð4:41Þ

0

where EMF(t) and EMFPk are the instantaneous phase back-EMF and its peak value, respectively, similarly i(t) and IPk are the phase current and its peak value, T is the period of the back-EMF waveform, and Kp is the back-EMF waveform factor that

120

4 Multiphase HPM Generator Systems

Fig. 4.15 Nine-phase HPM generator with power circuit converter dynamic model

accounts for variation from a pure sinusoidal to one that is more trapezoidal. Furthermore, (4.41) shows that a high number of phases will allow high power density only if Kp factor is presented by the more trapezoidal back-EMF waveform. This general equation is investigated by implying the above assumption for different machine configurations without any rectification stage, as shown in Fig. 4.16b. Then the passive rectification stage was added to the systems to predict output power at rated conditions, as shown in Fig. 4.16c. Finally, the active rectification stage, as shown in Fig. 4.16a, is considered. For the active rectification stage, as shown in Fig. 4.16a, different techniques of forced commutation are discussed in [138], and the authors’ view is that the most suitable one for this application is the sinusoidal-pulse-width modulation (SPWM). This AC–DC conversion technique overcomes the third-order harmonic problem, which presents the lowest order harmonic in the phase angle, extinction angle, and symmetrical angle control techniques. However, due to the high switching frequency of the SPWM technique, the higher order harmonics would have higher amplitudes, which could easily be filtered out [138]. The diode losses in the passive rectifier and the switching losses in the active rectifier case are investigated in Sect. 4.5.2.

4.4 Three- and Nine-Phase HPM Generator System Studies

121

Fig. 4.16 Multiphase generators without and with power circuit converters. (a) With active rectifier (b) Without passive rectifier (c) With passive rectifier

4.4.1

Impact on Synchronous Inductance and Rectifier

The three- and two sets of nine-phase HPM generator configurations with and without passive power circuit rectifier are analyzed for sinusoidal and trapezoidal back-EMF waveforms, for the same stator slot current density and with a machine geometry that will give similar stator copper losses, based on (4.42, 4.43, 4.44, and 4.45) by assuming that LS ¼ LSelf for all of the considered stator winding configurations. Psc is the stator copper losses, Jss is the stator current density, 3 and 9 denote three- and nine-phase configurations. Psc3 ¼ 3I 2rms3 Rs3 ns3 I rms3 Ass

ð4:43Þ

rffiffiffiffiffiffiffiffiffiffiffiffi Psc3 ¼ 9Rs9

ð4:44Þ

J ss3 ¼

I rms9

ð4:42Þ

122

4 Multiphase HPM Generator Systems

Table 4.8 Comparison of different HPM machine configurations at rated Psc when LS ¼ LSelf and without applying excitation current to the wound field part Without passive rectifier

Winding configuration Three-phase (sinusoidal) (nsct ¼ 6) Three-phase (trapezoidal) (nsct ¼ 6) Nine-phase (trapezoidal) (nsct ¼ 14) Nine-phase (trapezoidal) (nsct ¼ 18)

Phase EMFrms (V) 61.5

Phase Irms (A) 23.7

69.6

23.7

Machine output (Po) (average) (W) Case (a) Case (b) (LS ¼ 0) (LS ¼ LSelf)

With passive rectifier Machine System load output (Po) PDC-link (average) (W) (average) (W) Case (c) Case (c) (LS ¼ LSelf) (LS ¼ LSelf)

4388

3043

2826

2574

4914

3773

3490

3229

58.2

9.73

5083

4037

3339

3077

74.8

7.87

5283

4085

3393

3140

J ss9 ¼

ns9 I rms9 Ass

ð4:45Þ

The results illustrated in Table 4.8 show the predicted improvement by the trapezoidal back-EMF waveform that is offered by the fractional slot HPM machine stator winding configuration, over the three-phase machine with sinusoidal backEMF, for four cases. For Table 4.8, where the machine synchronous inductance is 1.0LSelf, both nine-phase schemes with a trapezoidal back-EMF give more machine output power than the three-phase trapezoidal back-EMF waveform when the passive rectification stage is not included in the system. However, once the passive rectification stage is added to the system, the machine terminal and DC-link average output power of the three-phase configuration with a trapezoidal back-EMF waveform becomes slightly greater than the 18-turn ninephase machine configuration. Moreover, due to the passive diode switching function, the average output power drops by 7.1% and 7.5% for both three-phase machine back-EMF waveforms (sinusoidal and trapezoidal), respectively, where for the nine-phase machine configuration (14- and 18-turn), the drop is 17%. In addition, the rectification stage efficiency for all machine configurations is around 92.5%. Thus, it is known that there is no benefit in power output to be gained by increasing the machine phase number above three for sinusoidal phase back-EMF waveform. Figures 4.17 and 4.18 demonstrate different phase back-EMF, terminal voltage, and current waveforms for the four analyzed HPM machine configurations without a rectification stage, at rated conditions only and with a rectification stage for

4.4 Three- and Nine-Phase HPM Generator System Studies

123

Fig. 4.17 Phase voltages and current waveforms of the six-turn three-phase HPM generator system at different loading conditions (LS ¼ LSelf). (a) Sinusoidal EMF3-phase without rectifier (b) Sinusoidal EMF3-phase with rectifier (c) Trapezoidal EMF3-phase without rectifier (d) Trapezoidal EMF3phase with rectifier

increments from minimum to rated resistive load power demands. The generator phase terminal voltage, as in Fig. 4.17b, shows clearly the three-phase rectifier six-step switching states that are changed every sixth of a cycle (V levels are VDC-link/3 and 2VDC-link/3), resulting in a nonsinusoidal generator phase current, as in Fig. 4.17b. For the nine-phase generator configuration, there are four possibilities of equivalent load configurations that can be obtained if an active rectification stage is utilized based on the generated IGBT’s gate signals [155]. However, for the passive nine-phase rectification stage, which is utilized in this thesis, the 4–5 load configuration is generated via the rectification stage (18 diodes) and a maximized phase current is achieved [155]. Therefore, Figs. 4.18b and (d) show clearly the 4–5 configuration effect (nine-phase rectifier 18-step switching states that are changed every 18th of a cycle) in the phase terminal voltage amplitude due to 4VDC-link/9 and 5VDC-link/9 levels. Furthermore, it is shown that as the phase current increases, due to the load power demand, the clear effect of the machine LS on the output power

124

4 Multiphase HPM Generator Systems

Fig. 4.18 Phase voltages and current waveforms of the 14- and 18-turn nine-phase HPM generator system at different loading conditions (LS ¼ LSelf) (a) Trapezoidal EMF9-phase (without rectifier and nstc ¼ 14) (b) Trapezoidal EMF9-phase (with rectifier and nstc ¼ 14). (c) Trapezoidal EMF9-phase (without rectifier and nstc ¼ 18) (d) Trapezoidal EMF9-phase (with rectifier and nstc ¼ 18)

rating becomes dominant, which imposes a limitation on the system rating for all of the machine configurations. Moreover, the detailed results for the case with passive rectifier are presented in Table 4.9. Here, the nine-phase HPM machine configuration with two sets of tooth number of turns, 14- and 18-turn, are investigated. To assure an adequate comparison between the three- and nine-phase HPM machine systems at rated conditions, it is desirable to have their DC-link output voltages at similar levels. The results in Table 4.9 illustrate that the 14-turn nine-phase machine gives a similar average DC-link voltage to the three-phase with sinusoidal back-EMF waveform case, but with less ripple value, as in Fig. 4.19. For DC-link % ripple improvement, both nine-phase configurations give similar improvements in % ripple value. The increase in output power compared to the three-phase with sinusoidal back-EMF is 19.5% and 21.9% for nine-phase machine configurations with 14- and 18-turn, respectively. However, for the three-phase machine configurations, the trapezoidal back-EMF waveform has an output DC-link voltage ripple that is

4.4 Three- and Nine-Phase HPM Generator System Studies

125

Table 4.9 Comparison of three- and nine-phase HPM generator system data for LS ¼ LSelf

Machine phase current (Irms) Phase EMFPk Phase voltage (VPk) Peak VDC-link Average VDC-link DC-link voltage ripple (Vr) % VDC-link ripple/average Ripple voltage (VDC-link) frequency ( fr) Average electromagnetic power (Po) Average DC-link power (PDC-link) Ratio of DC-link power nsc nPc Conductor length per tooth (PM stator) Conductor length per tooth (WF stator) Rs LS based on (LS ¼ LSelf)

Units A V V V V V

Three-phase (nstc¼6) (sinusoidal) 23.7 86.9 58.4 84.5 80.8 11

Three-phase (nstc¼6) (trapezoidal) 23.7 89.9 81.5 113.3 102.2 14.37

p.u.

0.136

0.14

kHz

4.8

4.8

Nine-phase (nstc¼14) (trapezoidal) 9.735 69.9 56.2 81.44 80.6 1.94 0.024 14.4

Nine-phase (nstc¼18) (trapezoidal) 7.87 89.9 68.5 103.4 102.2 2.4 0.024 14.4

W

2826

3490

3339

3393

W

2575

3229

3077

3140

p.u. – – m

1.00 12 12 0.6044

1.254 12 12 0.6044

1.195 4 5 1.403

1.219 4 4 1.8

m

0.4214

0.4214

0 .9834

1.2600

Ω mH

0.081 0.37

0.081 0.37

0.160 0.71

0.245 1.16

Note: Excitation current ¼ 0 A; axial length ¼ 40 mm; mass ¼ 11.2 kg; volume ¼ 1.45  103 m3; copper area per single conductor ¼ 2.46176  107 m2; WF rotor turns per tooth ¼ 100; total conductor length of the WF rotor part ¼ 7.5 m; stator fe ¼ 800 Hz; Jss ¼ 2.4 MA/m2; Psc ¼ 136.5 W; and rotor speed ¼ 3 krpm

worse than the three-phase machine with a sinusoidal back-EMF; however, it provides the highest DC-link average output power. Hence, the results suggest that to achieve the most average DC-link output power, the three-phase configuration with trapezoidal back-EMF presents the best choice among the other compared cases. Furthermore, the nine-phase machine with 14-turn will be compared with the threephase machine configuration with sinusoidal back-EMF due to the similarity in DC-link voltage level at rated conditions as shown in Fig. 4.20. The trapezoidal back-EMF three-phase machine is to be compared with the 18-turn nine-phase machine, as shown in Fig 4.20.

126

4 Multiphase HPM Generator Systems

Fig. 4.19 DC-link voltage waveforms for the three- and nine-phase HPM configurations at rated conditions (LS ¼ LSelf)

Fig. 4.20 Voltage regulation of HPM machine at different loads

4.4.2

System Sensitivity to Generator Synchronous Inductance

In this section, two more comparisons via sensitivity analysis based on LS are carried out. The first comparison is between sinusoidal back-EMF three-phase machine and trapezoidal back-EMF nine-phase machine with 14-turn per tooth configuration. The second comparison is between the trapezoidal back-EMF three-phase machine and the 18-turn nine-phase machine configurations. The comparison is performed for the same stator slot current density and machine geometry and for a machine synchronous inductance value of LSelf, which presents the sensitivity analysis benchmark based on the three-phase machine. In other words, the sensitivity analysis investigates the impact of four LS9 values (0.5, 0.75, 1, and 1.25LSelf) with respect to LS3 of

4.4 Three- and Nine-Phase HPM Generator System Studies

127

Fig. 4.21 Output power ratio curves for three-phase machine with sinusoidal back-EMF versus 14-turn nine-phase machine

LSelf on the output power ratio, machine terminal voltage, and DC-link voltage of the three- and nine-phase machine systems at different operating conditions. For the first comparison, in both the three- and nine-phase systems, degradation in the output power is expected at higher loads with higher values of machine selfinductance due to the reactive voltage drop presented by this inductance. The presented machine stator winding schemes offer lower machine self-inductance value, which presents an advantage in this application, if they were to be compared with over-lapping or single layer distributed stator winding schemes. Figure 4.21 shows the effect of different LS9 values with respect to one value of LS3, for different load conditions, on the output power ratio between the sinusoidal back-EMF threeand 14-turn trapezoidal back-EMF nine-phase configurations. The system power ratio illustrates that the advantage of the 14-turn nine-phase configuration with trapezoidal back-EMF over the three-phase with sinusoidal back-EMF waveform is directly related to the values of both configurations’ synchronous inductances, where the advantage offered by the nine-phase scheme may become a disadvantage with higher values of LS9 in the proposed HPM machine system. Furthermore, Table 4.10 illustrates the HPM machine terminal and DC-link voltage regulation due to different LS values for the nine-phase configuration (14-turn). Here, better voltage regulation capability in both machine terminal and DC-link voltage is gained by the nine-phase machine with a trapezoidal back-EMF at rated conditions, as shown in Table 4.10 and Fig. 4.22. Furthermore, the sensitivity analysis showed the expected voltage regulation capability as the synchronous inductance increases or decreases in the nine-phase machine with respect to a single value of the three-phase machine synchronous inductance. The second comparison, as shown in Fig. 4.23, presents the effect of different LS9 values with respect to one value of LS3, for different load conditions, on the output power ratio between the trapezoidal back-EMF three-phase machine and the 18-turn

Machine Three-phase (sinusoidal) (nsct¼6) nine-phase (trapezoidal) (nsct¼14) 78.9

43.2

15.8

24.7

1.25LSelf –

Machine terminal voltage regulation (%) 0.5LSelf 0.75LSelf 1.0LSelf – – 56.5 34.6

45.1

67.3

DC-link voltage regulation (%) 0.5LSelf 0.75LSelf 1.0LSelf – – 77.0

Table 4.10 Voltage regulation of three- and nine-phase machine systems for the first comparison case

110

1.25LSelf –

128 4 Multiphase HPM Generator Systems

4.4 Three- and Nine-Phase HPM Generator System Studies

129

Fig. 4.22 Different terminal and DC-link voltage regulation curves of the three-phase with sinusoidal back-EMF versus the 14-turn nine-phase HPM machines. (a) Terminal voltage regulation (LS3 ¼ 1.0LSelf). (b) DC-link voltage regulation (LS3 ¼ 1.0LSelf)

trapezoidal back-EMF nine-phase machine configurations. The system power ratio illustrates that the previous advantage, as illustrated in Fig. 4.21, offered by the 14-turn nine-phase machine with more trapezoidal back-EMF over the sinusoidal back-EMF three-phase machine disappears and the power ratio degrades in this second comparison. Furthermore, Table 4.11 illustrates the HPM machine terminal and DC-link voltage regulation due to different LS values for the nine-phase configuration (18-turn). The voltage regulation capability in both machine terminal and DC-link voltage is almost equal at rated conditions, as shown in Fig. 4.24. The sensitivity analysis also showed the predicted voltage regulation capability as the synchronous inductance increases or decreases in the nine-phase machine with respect to a single value of the three-phase machine synchronous inductance.

130

4 Multiphase HPM Generator Systems

Fig. 4.23 Output power ratio curves for three-phase machine with trapezoidal back-EMF versus 18-turn nine-phase machine

4.4.3

DC-Link Voltage Quality

In many systems with high DC-link voltage ripple, a smoothing (filter) capacitor is introduced to eliminate or minimize DC-link ripple, as illustrated in Fig. 4.25. The filter capacitor is the most vulnerable and one of the most expensive components in the drive system [156], where its life is affected by the ripple current and it has a much shorter life compared to the other components. This poses a clear drawback if it is to be used in a harsh environment or where maintenance access is denied or difficult. In Table 4.12, the voltage ripple equals 14.37 V and 2.4 V for the three- and ninephase (18-turn) machines with a trapezoidal back-EMF waveform, respectively. For the three-phase HPM machine case, to reduce ripple from 14.37 to 2.4 V, the smoothing capacitor (CS) would need to be 1.3498 mF using equation (4.46) [157], where the IDC-link equals 31.1A and fr equals 4.8 kHz. However, this calculated value requires adjustment by an iterative process that has been applied via the numerical model, which suggested a new value for CS to achieve this reduction in ripple, where the new capacitor value based on the dynamic model prediction equals (1.35/52.1) mF and the RMS current (ICrms) passing through it equals 1.362 A. C 3ϕ ¼

I DClink Vγ f γ

ð4:46Þ

Hence, an equal DC-link voltage ripple between three- and nine-phase machines was achieved. A suitable commercial capacitor was found online [158]. Based on the available capacitors and the SimPower model DC-link voltage levels, a 200-V rating capacitor (twice the DC-link voltage) was initially suggested. Based on the electrolytic capacitor datasheet, the capacitor internal resistance (ESR) at 105  C, ripple

Machine Three-phase (trapezoidal) (nsct¼6) Nine-phase (trapezoidal) (nsct¼18) 87

44.6

14.5

23.8

1.25LSelf –

Machine terminal voltage regulation (%) 0.5LSelf 0.75LSelf 1.0LSelf – – 43.2 30.0

40.3

62.3

DC-link voltage regulation (%) 0.5LSelf 0.75LSelf 1.0LSelf – – 64.4

Table 4.11 Voltage regulation of the three- and nine-phase machine systems of the second comparison case

108

1.25LSelf –

4.4 Three- and Nine-Phase HPM Generator System Studies 131

132

4 Multiphase HPM Generator Systems

Fig. 4.24 Different terminal and DC-link voltage regulation curves of three-phase with trapezoidal back-EMF versus 18-turn nine-phase HPM machines. (a) Terminal voltage regulation (LS3 ¼ 1.0LSelf). (b) DC-link voltage regulation (LS3 ¼ 1.0LSelf)

frequency, and RMS current were accounted for in the DC-link capacitor selection process. To calculate the DC-link capacitor losses (PCap-loss) and rating (PCap-rating) and check if they satisfy the design requirements, (4.47) and (4.48) were utilized.  PCaploss ¼

I Crms ncp

2 ESR ncp

ð4:47Þ

4.4 Three- and Nine-Phase HPM Generator System Studies

133

Fig. 4.25 DC-link voltage waveform with and without smoothing capacitor effect for the three-phase trapezoidal back-EMF HPM machine system at rated conditions

Table 4.12 The encountered iterative process for selecting the DC-link capacitors Selected voltage level (V)

200

350

Capacitor (μF) 68 33 22 15 10 68 33 22 15

ESR (Ω) 1.46 3 4.45 6.13 9.75 1.05 2.25 3.22 7.45

ICrmsM (A) 0.71 0.42 0.31 0.26 0.17 0.91 0.56 0.44 0.24

ncp 1 2 3 4 6 1 2 3 6

PCap-loss (W) 2.707 2.781 2.750 2.841 3.013 1.947 2.086 1.990 2.302

PCaprating ¼ ðI CrmsM Þ2 ESR ncp

PCap-rating (W) 0.736 1.058 1.283 1.658 1.690 0.869 1.411 1.870 2.575

Satisfy requirements No No No No No No No No Yes

ð4:48Þ

where ICrmsM is the datasheet RMS capacitor current and ncp is the number of capacitors in parallel. Table 4.12 illustrates the results of the iterative process for choosing the adequate number of DC-link capacitors, which are to be connected in parallel and satisfy the design requirements. Thus, the results suggest that six capacitors of 15μF designed to handle 350 V are to be connected in parallel in the DC-link terminal of the three-phase system case to reduce the DC-link voltage ripple. However, more investigation on some of the alternatives to reducing this voltage ripple by using an LC filter, with different DC-link series inductor (LDC-3) and capacitor (CS-3) values, for the three-phase case, is carried out. By adding LDC in the three-phase system simulation models, as shown in Fig. 4.26, the prediction of this inductance in the three-phase case has been done, based on the best and optimum combination of both CS-3 and LDC-3 values. Therefore, an investigation is applied on

134

4 Multiphase HPM Generator Systems

Fig. 4.26 SimPower models for three-phase HPM generator systems with DC-link inductance

Fig. 4.27 Impact of DC-link inductance on DC-link voltage ripple in three-phase HPM machine system

the three-phase HPM generator dynamic model with four different capacitor values and a range of inductance values, as illustrated in Fig. 4.27. Due to the harsh environment at which CS is to be placed in, a decrease in the capacitor requirements by substituting an adequate inductor to achieve the desired quality of DC-link output will improve the powertrain system robustness in HEVs. Note that the addition of this DC-link inductance improves the robustness of the vehicle powertrain while degrading DC-link voltage regulation. However, this degradation in voltage is not a significant drawback since it can be compensated for, for example, by increasing the number of stator turns.

4.5 4.5.1

Loss Audit of Generator Systems Introduction

The proposed nine-phase HPM generator system in this book is an alternative to the three- or nine-phase PM generator with an active converter, which provides a

4.5 Loss Audit of Generator Systems

135

Table 4.13 Comparison of HPM rectifier system with PM active conversion

Item Generator topology Stator winding No. of IGBTs (bridge) No. of diodes (bridge) Control scheme Control complexity Phase current measurement Rotor angle measurement Slip rings and brushes Shaft speed Active switching losses (p.u.) PM axial length (p.u.) WF axial length (p.u.) Total machine volume (p.u.) Active converter volume (p.u.) Passive rectifier volume (p.u.)

HPM connected to passive rectifier WF-PM

PM connected to VSC PM

HPM connected to passive rectifier WF-PM

PM connected to VSC PM

Three-phase –

Three-phase 6

Nine-phase –

Nine-phase 18

6

6

18

18

WF control using voltage regulator Simple

Vector control Complex

WF control using voltage regulator Simple

Vector control Complex

–

–

2

Current sensors Encoder/ resolver –

2

Current sensors Encoder/ resolver –

Fixed –

Variable 1.0

Fixed –

Variable 1.0

0.75

1.0

0.75

1.0

0.3

–

0.3

–

1.05

1.0

1.05

1.0

–

1.0

–

1.0

1.0

1.0

1.0

1.0

–

–

simplified power conversion scheme, specifically by eliminating the active switching devices and reducing the associated measurement devices (phase current and rotor angle sensors) and control hardware. Table 4.13 shows a general comparison of the HPM generators using a passive rectification system with PM generators using an active voltage source converter (VSC) for both three- and nine-phase stator winding configurations. Hence, this section investigates some of the undiscussed losses that appear in the HPM generator system with passive rectification and the PM generator system with active rectification. The losses of interest are mainly due to the stator core and passive and active switches. An assessment of these losses is computed as a feasibility study of using a HPM generator with a passive rectifier system as an alternative to a PM generator with an active rectifier system.

136

4.5.2

4 Multiphase HPM Generator Systems

Core Loss Prediction

Assuming similar core loss for three- and nine-phase winding configurations, the used core loss for HPM machines is conducted previously as in Chap. 3 for the 25-mm PM and 10-mm WF machines. For completeness, the same calculation method was used to assess the core losses for a PM machine having 40 mm axial length, as in Table 4.14. The HPM machine iron loss at 3 krpm with rated WF excitation is 411.8 W. The same volume of PM machines with an axial length of 40 mm has predicted a loss of 383.4 W.

4.5.3

Passive and Active Converter Loss for HPM and PM Generator Systems

For a HPM generator with passive rectification, a simple diode loss equation is used in the SimPower dynamic model to calculate the rectification stage losses. The average diode conduction loss across the switching period (Tsw ¼ 1/FS) is given by (4.49) [159], where IDav is the average diode current, IDrms is the rms diode current, uD0 is the maximum forward voltage drop, and rD is the diode internal resistance. PCD ¼

1 T SW

Z 0

T SW 

uD0 :iD ðt Þ þ r D :i2D ðt Þ dt ¼ uD0 :I Dav þ r D :I 2Drms

ð4:49Þ

The datasheet of a three-phase bridge rectifier 36MT series is used to get the typical values for uD0 and rD. Hence, Table 4.15 illustrates the passive rectification stage losses in three- and 18-turn nine-phase HPM generator configurations with trapezoidal back-EMF at rated conditions. Predicting the losses in the semiconductor active power switches, such as IGBTs, is complicated if they are to be compared with the passive ones. Some papers have discussed active semiconductor switch loss calculations [160–163] with minor differences in their assumptions. The proposed IGBTs loss equations by Casanellas [162] are used to predict semiconductor power losses when switching sinusoidal currents. The loss mechanisms in an IGBT are • Turn-on losses (PSW-on) where energy is required to activate the device; the rise time (tr) for the device is related to the gate resistor (RG), which limits the gate charge. • Conduction losses (PIGBT); that is, when the device is conducting, there is a voltage drop across the P–N junction (VCO) and a proportional relationship with the collector current (Icc) and an ohmic loss component. • Power losses in the reverse biased diode related to (VF) (PDIODE).

Machine section Back-iron Teeth Teeth-tips Total

HPM machine PM stator (LPMa ¼ 25 mm) Mass Core losses (kg) (W) 1.748 34.6 0.968 191.2 0.148 13.8 2.865 239.6

PM machine

Stator (LPMa ¼ 40 mm) Mass Core losses (kg) (W) 2.797 55.4 1.549 305.9 0.238 22.1 4.584 383.4

WF stator (LWFa ¼ 10 mm) Mass Core losses (kg) (W) 0.7000 8.9 0.3874 38.5 0.0594 3.0 1.1458 50.3

WF rotor (LWFa ¼ 10 mm) Mass Core losses (kg) (W) 0.139 28.8 0.471 90.5 0.055 2.4 0.665 121.9

Core losses (W) 37.7 129.0 5.4 172.2

WF total (10 mm)

Table 4.14 Different stator and rotor section masses and core losses for the fundamental harmonic only of PM and HPM machine at no-load

4.5 Loss Audit of Generator Systems 137

138

4 Multiphase HPM Generator Systems

Table 4.15 Losses due to passive rectification stage in HPM generator system Electrical power losses (W) Losses per diode Smoothing capacitor losses Total rectification stage losses

Three-phase system 18.44 2.302 (six capacitors of 15 μF) 110.64

Nine-phase system 3.49 – 62.82

• Turn-off losses (PSW-off) related to the fall time (tf) of the device and collector current. • Reverse recovery losses (PRR) where the diode must recover from a conducting to blocking state—this is relational to the charge (QRR) and the time (trr). While these characteristics of the IGBT can be used to model its losses, many parameters change with junction temperature. The model developed by Casanellas considers temperature and operationally varying characteristics and provides linear approximations for use in modeling. The model produced by Casanellas has been verified via calorimetric test and is considered to give accurate results to within 5–10% [163]. Power silicon losses with sinusoidal current control can thus be calculated from the turn-on losses, which are estimated using (4.50) [162]: PSWon

2  I 1 ¼ V DClink t rN F S cm 8 I cn

ð4:50Þ

The conduction losses, including the third harmonic sine modulation that allows full DC-link utilization, are estimated using (4.51) [162]: PIGBT ¼

pffiffiffi pffiffiffi    3 1 2 3 V  V co 2 M index cos ðθÞ : cen þ M index cos ðθÞ  I cm 45π 8 9π I cn

þ PIGBT 3rdharmonic pffiffiffi   3 1 M index cos ðθÞ :V co I cm þ ¼ 2π 12

ð4:51Þ

The diode conduction losses, including the third harmonic sine modulation that allows full DC-link utilization, are estimated using (4.52) [162]:

4.5 Loss Audit of Generator Systems

139

Table 4.16 Parameter definitions and typical values for semiconductor loss calculations Definition Rated collector–emitter forward voltage drop Rated collector–emitter forward voltage drop Collector current rating Collector current Modulation index Switching frequency Phase angle Reverse recovery charge IGBT rated fall time at rated current Diode recovery fall time at rated current (Icn)

PDIODE

Parameter Vcen Vco Icn Icm Mindex FS cos(θ) Qrrn tfn trrn

Units V V A A – Hz Rads C ns ns

1200 V 2.00 1.00 400 NA 1 10000 NA 1.60 n 350 250

pffiffiffi pffiffiffi    3 1 2 3 V  V co 2 M index cos ðθÞ : cen  M index cos ðθÞ þ ¼ I cm 45π 8 9π I cn þ PIGBT 3rdharmonic pffiffiffi   3 1 M index cos ðθÞ :V co I cm ¼  2π 12 ð4:52Þ

The turn-off losses are estimated using (4.53) [162]:  PSWoff ¼ V DClink I cm t fN F S

1 I þ cm 3π 24I cn

 ð4:53Þ

The diode reverse recovery losses are estimated using (4.54) [162]: " Prr ¼ F s V DClink Qrrn

 2 !  # 0:38 I cm I cm 0:8 I cm þ 0:05 0:28 þ þ 0:015 þ I cm t rrn π I cn π I cn I cn ð4:54Þ

The parameters trn, Fs, Icm, Icn, Mindex, cos(θ), Vcen, Vco, and tfn and their typical values, as obtained from the device datasheet (CM400DY-24NF), are defined in Table 4.16. IGBT power loss per device is calculated at 10, 15, and 20 kHz switching frequency and then multiplied by the total number of devices, as shown in Table 4.17. Table 4.18 illustrates the comparison between passive and active rectification stage losses for the three- and nine-phase systems. The results suggest that all systems have similar efficiency. Table 4.18 predicts that the total system loss in the three-phase case with active rectification is less than that with passive rectification by 5%, where the total system loss for the nine-phase case with active rectification is greater than that with passive rectification by 5.8%. Therefore, for a

Definition Turn-on loss IGBT conduction loss Reverse biased diode loss Turn-off loss Reverse recovery loss Losses per IGBT Total rectification stage loss 0.3 0.98 4.8 86.42

0.92 2.03

12.93 77.59

PSW-off PRR

IGPTloss RecStageloss

0.65

1.99

PDIODE

Nine-phase system (W) 0.00316 1.9

FS ¼ 10 kHz Three-phase system (W) 0.0287 5.91

Power loss components PSW-on PIGBT

15.45 92.7

1.38 3.04

0.65

FS ¼ 15 kHz Three-phase system (W) 0.04305 1.9

5.93 106.75

0.45 1.46

0.65

Nine-phase system (W) 0.00474 1.9

Table 4.17 Calculated loss due to active rectification stage in PM generator system for several switching frequencies

17.97 107.81

1.84 4.06

0.65

FS ¼ 20 kHz Three-phase system (W) 0.0574 1.9

7.06 127.07

0.6 1.95

0.65

Nine-phase system (W) 0.00633 1.9

140 4 Multiphase HPM Generator Systems

4.6 Conclusion

141

Table 4.18 Comparison between passive and active rectification stage losses of three- and ninephase machine systems Loss types Copper Core Rectification DC-link capacitor Total Loss types Copper Core Rectification DC-link capacitor Total

Three-phase HPM machine with passive rectification losses (W) 136.5 411.8 110.64 2.302 (six capacitors of 15 μF)

Three-phase PM machine with active rectification losses (W) 136.5 383.39 107.81 (FS ¼ 20 kHz) –

661.24 Nine-phase HPM machine with passive rectification losses (W) 136.5 411.8 62.82 –

627.7 Nine-phase PM machine with active rectification losses (W) 136.5 383.39 127.07 (FS ¼ 20 kHz) –

611.16

646.96

more reliable system to be implemented in the HEV powertrain, the nine-phase HPM machine with a passive rectification stage is a good candidate for vehicle applications when compared with the other systems.

4.6

Conclusion

In this chapter, an overview of different multiphase machines and their applications are discussed. Based on a similar stator slot current density and machine geometry, a comparison between three- and nine-phase HPM machine configurations along with their DC-link voltage quality and regulation is investigated. SimPower is used to analyze and predict the HPM generator system’s behavior at both the machine and DC-link terminals while the WF excitation is set to zero. In Table 4.9, it is shown that the highest output power is achieved by the trapezoidal six-turn three-phase HPM machine configuration. Both nine-phase configurations, with 14- and 18-turn, with trapezoidal EMF waveform produce a very low DC-link ripple value with less DC-link output power when compared with the three-phase configuration with trapezoidal back-EMF, when LS ¼ LSelf for all of the considered stator winding configurations. Furthermore, the impact of different values of LS9 (with respect to a single value of LS3) has been investigated via sensitivity analysis, and it showed the imposed limitation on the system output DC-link voltage and power due to different values of LS9. Passive and active rectification stage losses in the semiconductor power switches, such as diodes in the passive and IGBTs in the active cases, for three- and nine-phase HPM and PM machines are investigated. The overall predicted losses and efficiency

142

4 Multiphase HPM Generator Systems

of both systems, HPM with passive and PM with active switches, and different winding configurations, have been compared to judge the feasibility of utilizing HPM machine with a passive rectification stage as an alternative to a PM machine with active rectification stage in a HEV powertrain system. Thus, in terms of system robustness, reliability, and cost, the nine-phase HPM machine with a separate DC excitation source seems to be a suitable configuration that can be considered a replacement of the PM machine system in a HEV powertrain.

Chapter 5

Electric and Hybrid Electric Powertrains

5.1

Introduction

Worldwide concerns over increasing energy consumption and pollution due to transportation systems are a primary motivation for alternatively fueled or moreelectric vehicles. Electric vehicles (EVs) are set to play an increasingly predominant role in the automotive market since they address the energy and environmental impact of an expanding road transport population by offering a more energy-efficient and less-polluting powertrain alternative to conventional internal combustion engine (ICE) vehicles. However, when compared with conventional gasoline or dieselfueled vehicles, all-electric vehicles are disadvantaged in driving range because of the relatively low energy storage capacity of the onboard batteries, excessive battery mass and refueling thereof, and recharging time, associated with existing electrochemical battery technologies. For example, a 2.5-ton ICE urban vehicle will have a typical range of 400 km [95] compared with 126 km for the equivalent all-electric vehicle [164]. On the other hand, hybrid electric vehicles (HEVs) address the vehicle range issue while offering improved fuel economy and emission reduction when compared with conventional ICE vehicles, but only when the vehicle has intermittent or transient duty operation.

5.2

Overview of EVs

The driving range of electric vehicles depends mainly on the stored energy on an onboard high-voltage battery system and vehicle gross weight. The driving distance is also affected by driving behaviors, vehicle architecture, road conditions, type of battery used, and vehicle age. Many companies now are producing all-electric vehicles, an example of which is listed in Table 5.1 for the year 2020, including

© Springer Nature Switzerland AG 2022 A. S. Al-Adsani, O. Beik, Multiphase Hybrid Electric Machines, https://doi.org/10.1007/978-3-030-80435-0_5

143

144

5 Electric and Hybrid Electric Powertrains

Table 5.1 Top EV models in the year 2020 [165] Vehicle model Hyundai Kona Electric Nissan Leaf Mini Cooper SE Electric BMW i3 Kia Niro EV Chevrolet Bolt Tesla Model Y Hyundai Ioniq Electric Jaguar I-Pace Tesla Model X Audi e-tron Porsche Taycan Tesla Model 3 Tesla Model S

Minimum (p.u.) cost 0.36

Max. driving range (miles) 258

Wheel-drive architecture Front

0.31 0.29

226 110

Front Front

0.43 0.38 0.35 0.48 0.32 0.67 0.77 0.75 1.00 0.37 0.72

153 239 259 216 170 234 351 204 201 322 402

Rear Front Front Rear Front All Rear All Rear Rear Rear

Note: 1 p.u. cost equals 103,500 US dollars

Fig. 5.1 Several EV models (a) Nissan Leaf [166] (b) BMW i3 [167] (c) Chevrolet Bolt [168] (d) Tesla Model S [169] These vehicle photos are not sponsored, endorsed, or promoted by Nissan North America, Inc., BMW North America, LLC, General Motors Co. (GM), Tesla, Inc., or their affiliates

5.2 Overview of EVs

145

their expected driving range, wheel-drive architecture, and per-unit price. Figure 5.1 shows pictures of four selected vehicle models.

5.2.1

EV Powertrain Configuration

In this book, the powertrain presents the system that propels the vehicle, which is significantly different between electric and conventional ICE-powered vehicles. For example, it is estimated that a conventional ICE vehicle powertrain has 2000 components, where EVs have substantially fewer powertrain components [170]. Hence, the EV powertrain system is expected to require less labor to produce

Fig. 5.2 EV powertrain components (a) Simplified powertrain configuration (b) Detailed powertrain components [171]

146

5 Electric and Hybrid Electric Powertrains

compared with conventional gasoline or diesel engine vehicles [170]. Figure 5.2 presents a simplified and detailed EV powertrain component.

5.2.2

Battery Technology

Battery technology is a key element in EV and HEV powertrains. The specific power, specific energy, pack size, charging time, cycle life, and cost per kW are the major constraints that dictate the most adequate battery technology to be used and developed for future EV and HEV powertrains. For more than 100 years, the specific power capability of lead-acid battery technology has continually been improved, but their specific energy has slightly increased, typically between 30 and 50 Wh/kg. This imposes a threshold on their continuous use in the automotive industry. Moreover, the lead-acid battery, an example of which is shown in Fig. 5.3a, has a short cycle life compared with competing technologies [172]. An alternative battery technology is nickel–metal hydride (Ni-MH), as shown in Fig. 5.3b, which has an energy density that falls between 60 and 120 Wh/kg. This technology has been used in Toyota Prius; however, it is not the best choice for EVs. A significant performance improvement can be gained by utilizing sodium–nickel chloride (Na-Ni-Cl) battery technology, which is commonly referred to as ZEBRA battery technology. It is considered safe and low-cost with a long cycle life since it can be discharged completely without degrading its lifetime expectancy. The ZEBRA battery shown in Fig. 5.3c has a specific energy that falls between 60 and 120 Wh/kg. On the other hand, the ZEBRA battery has great potential to be used in association with other energy sources, such as ICE or supercapacitors in HEV powertrains. The most promising battery technology is the lithium-ion (Li-ion), where its energy density is between 100 and 265 Wh/kg. However, the Li-ion battery, as shown in Fig. 5.3d, suffers from material availability and overcharging safety issues and higher price compared with the previously mentioned battery technologies. Table 5.2 illustrates a brief comparison in terms of specific energy and power, energy and power density, cycle lifetime, and energy and power capacity cost between different battery technologies.

5.3

Overview of HEVs

Driving range and battery charging time are the major EV barriers. However, HEVs and plug-in HEVs alleviate some of the drawbacks experienced with EVs in terms of range while providing an intermediate powertrain concept in the move from conventional to an all-electric rational [177]. The major difference between HEVs and plug-in HEVs is the onboard battery size with respect to the ICE size. The plug-in HEV powertrains have a larger and more powerful battery pack compared with the

5.3 Overview of HEVs

147

Fig. 5.3 Different EV and HEV battery technology (a) Lead-acid battery [173] (b) Ni-MH battery pack [174] (By Claus Ableiter—Own work, CC BY-SA 4.0) (c) ZEBRA battery [175] (Von Claus Ableiter—Eigenes Werk, CC BY-SA 4.0) (d) Li-ion battery pack [176] (By Claus Ableiter—Own work, CC BY-SA 4.0) Table 5.2 Comparison of different battery technologies [172] Specification Specific energy (Wh/kg) Energy density (Wh/L) Specific power (W/kg) Power density (W/L) Cycle lifetime (cycles) Energy capacity cost ($/kWh) Power capacity cost ($/kW)

Lead-acid 30–50 50–80 75–300 10–400 100–1000 150–400 175–600

NiMH 60–120 140–300 250–1000 80–300 500–1200 150–1500 150–1500

ZEBRA 100–120 150–180 150–200 220–300 2500+ 100–200 150–300

LI-ion 100–265 250–730 250–340 100–210 400–1200 500–2500 175–4000

148

5 Electric and Hybrid Electric Powertrains

Table 5.3 Top HEV models in 2020 [178] Vehicle model Honda Insight Toyota Avalon Hybrid Hyundai Ioniq Hybrid Toyota Prius Prime Hybrid Toyota Prius V Honda Accord Hybrid Kia Optima Hybrid Toyota Camry Hybrid Lincoln MKZ Hybrid Toyota Prius C Hyundai Sonata Hybrid Ford Fusion Energi Chevrolet Volt Toyota Prius Ford Fusion Hybrid

Minimum (p.u.) cost 0.22 0.36

Fuel economy in (mpg) (city/ highway) 55/49 43/44

Wheel-drive architecture Front All

0.22

55/58

Front

0.27

55/53

Front or all

0.23 0.23

54/50 48/47

Front or all Front

0.28 0.26

21/30 44/47

Front Rear

0.35 0.22 0.27

20/31 48/44 50/54

Front or all Front Front

0.34 0.39 0.24 0.27

109/97 43/42 54/50 43/41

Front Front Rear or all Front

Note: 1 p.u. cost equals 103,500 US dollars

onboard ICE power rating, where the HEV has a smaller and less power rating battery pack compared with the ICE power rating. Fuel consumption or fuel economy plays a key factor in future road vehicle production since now many countries are using gasoline vehicle fuel economy new standards to direct their automotive industry to meet certain miles per gallon (mpg) numbers. Table 5.3 lists various commercial HEVs, their fuel economy, per-unit price, and wheel-drive architecture, while Fig. 5.4 shows pictures of four hybrid EV models. In addition to the improvement in the driving range, which can be governed by fuel tank size, HEVs and plug-in HEVs showed a clear cost advantage over EVs while generating less emission for similar ICE-powered vehicles’ performance.

5.3.1

HEV Powertrain Configurations

The surveyed hybrid EVs, as in Table 5.3, might have powertrain components similar to what is shown in Fig. 5.5. The common simplified hybrid EV powertrain configurations that consist of two energy sources, such as battery and ICE, are illustrated in Fig. 5.6. The parallel hybrid electric vehicle format, as in Fig. 5.6a, is

5.3 Overview of HEVs

149

Fig. 5.4 Several HEV recent models [178] (a) Honda Insight (By Kevauto—Own work, CC BY-SA 4.0) (b) Toyota Prius V (By Vauxford—Own work, CC BY-SA 4.0) (c) Lincoln MKZ Hybrid (By Kevauto—Own work, CC BY-SA 4.0) (d) Hyundai Sonata Hybrid (By Chu—Own work, CC BY 4.0) These vehicle photos are not sponsored, endorsed, or promoted by American Honda Motors Co., Inc., Toyota Motors North America, Inc., Ford Motor Co., Hyundai Motor America, or their affiliates

Fig. 5.5 Detailed plug-in HEV powertrain components [179]

generally the favored scheme for most vehicle suppliers to date since it allows them to continue to use their existing powertrain components while accommodating an additional power input, thus requiring lower investment and minimizing system risk. However, the series hybrid electric vehicle, as illustrated in Fig. 5.6b, can offer more flexibility where the ICE energy source is replaced by alternatively fueled combustion engines (using petroleum products, methanol, hydrogen) or fuel cell systems [180]. Here, the ICE is mechanically disconnected from the powertrain and is hence independent of vehicle road wheel speed. Energy conversion is then from the

150

5 Electric and Hybrid Electric Powertrains

Fig. 5.6 Different simplified hybrid EV powertrain configurations (a) Parallel hybrid (b) Series hybrid (c) Series-parallel hybrid (power-split hybrid)

onboard fuels’ chemical energy to kinetic energy via the ICE, and then from kinetic to electrical energy via an engine-mounted or -coupled generator, Fig. 5.6b [181]. In Fig. 5.6c, another possible powertrain implementation is shown, where three drive modes, such as all-electric, hybrid, and engine drive, can be realized. There is, therefore, an interest in onboard auxiliary power units that would serve as an energy input to the vehicle powertrain for suburban or highway driving.

5.5 Series Hybrid Electric Vehicle (SHEV)

5.4

151

Vehicle Driving Cycles

The fuel economy and emission certification tests can be applied onboard or off-board standard light-duty and heavy-duty vehicles. These tests are derived from the total fuel consumption of fuel over approved driving cycle using a dynamometer and gaseous particular matter portable emission measurement system known as AVL PEMS. In some labs, such as for the off-board economy and emission test, the AVL PEMS can remain on a cart and placed where the sample line can reach the vehicle exhaust pipe for laboratory testing, as shown in Fig. 5.7. For example, in the United States, a highway driving cycle (HWFET) is used for the driving test, as shown in Fig. 5.8a. In Europe, a new European driving cycle (NEDC) is used. The NEDC consists of four urban (ECE15) cycles and one suburban cycle, as shown in Fig. 5.8b. In Japan, two standard drive cycles, known as 10–15 mode and JC08 cycles, are used for driving fuel economy and emission measurement tests [183].

5.5

Series Hybrid Electric Vehicle (SHEV)

The previously discussed hybrid EV powertrain configurations, illustrated in Fig. 5.6, tend to combine the engine start and generator functionalities into one machine, a philosophy of reduced component count. However, the start and generation operational requirements are quite different. Here, the machine has to provide high peak power transients during starting and engine braking. More importantly, the starter/alternator requires a voltage source inverter for active power conversion between starter/alternator and the ICE; hence, the power converter needs to have active switching and high silicon VA rating. However, most of the machine operation is in the steady power generation mode, which is usually over a limited speed Fig. 5.7 Actual laboratory AVL PEMS [182]

152

5 Electric and Hybrid Electric Powertrains

Fig. 5.8 US and European standard driving cycles [95] (a) US highway driving cycle (HWFET) (b) NEDC driving cycle

range and at lower power levels than for the system starting transient. Thus, if the starting and steady power generation are realized by two machines, one a traditional starter and the second a lower power generator operating over a constrained speed range, the installed machines could be simpler in form and the power electronic conversion requirements reduced. Compared with series HEVs, the parallel HEV format is generally the favored electrified scheme to date for most vehicle suppliers since it allows the original equipment manufacturers (OEM) to utilize their existing products and gradually increase their investments over time. However, with the series HEV topology, the combustion engine is disconnected from the system, as shown in Fig. 5.6b, its size is reduced, and its rotational speed can be made dependent on or independent of the

5.5 Series Hybrid Electric Vehicle (SHEV)

153

Fig. 5.9 Different ICE/generator machine and power conversion system implementations for auxiliary power unit in SHEVs. (a) ICE/PM machine and passive power converter (b) ICE/PM machine and active power converter (c) ICE/HPM machine and passive power converter

vehicle road speed, essentially to provide a fixed or variable speed to the generator. Therefore, in the series hybrid electric vehicle (SHEV) format, the battery pack can provide the peak power for the vehicle DC-link while the combustion engine, powered from fuel tank, provides the vehicle average power using an onboard generator. Here, there are a number of ways in which the output voltage of the PM (or other machine technology) generator may be facilitated, as illustrated schematically in Fig. 5.9. The PM generator output (three-phase in this case) can be passively rectified, and the output voltage magnitude can be regulated by speed control of the ICE prime mover, Fig. 5.9a. However, voltage transients are usually much faster than the control dynamics of ICEs, and hence an active rectification scheme is usually required and the ICE speed effectively fixed, as shown in Fig. 5.9b. This adds power electronic complexity but gives additional control functionality. Alternatively, the ICE can be used as the prime mover to the HPM machine, discussed in Chaps. 3 and 4, the output of which is connected to a simple passive rectification stage, as illustrated in Fig. 5.9c, showing the HPM machine (again three-phase) in the powertrain system schematic. As discussed in previous chapters in this book, the hybrid PM (HPM) machine is predominantly a permanent magnetic machine, but with some facility to control its output open-circuit voltage via a secondary field excitation source. Hence, the hybrid terminology refers to the hybridization of permanent magnet and wound field excitation in one machine package. The function of HPM machines in SHEV powertrains is illustrated schematically in Fig. 5.10. Here, the ICE acts as the prime mover to an electrical three-phase HPM generator that is specified to match the vehicle terminal voltage transients, as shown in Fig. 5.10, while maintaining constant HPM power output to the vehicle DC-link.

154

5 Electric and Hybrid Electric Powertrains

Fig. 5.10 Series HEV powertrain schematic with a three- or nine-phase HPM generator supplying power to the vehicle DC-link

Whether the generator is a HPM topology or not, the quality of the generated DC voltage and power will impact other system components in the vehicle powertrain, in particular, the requirement for additional DC-link capacitance at the input to the traction machine (TM) power electronic converter (traction drive), which is predominantly a voltage source inverter. The three-phase HPM machine terminal AC voltage is converted to DC via a passive rectification stage, shown as AC/DC in Fig. 5.10. The HPM wound field excitation modifies the machine output voltage due to the permanent magnets, again as illustrated in Fig. 5.10, via a simple excitation control loop, as will be discussed in more detail in Chap. 6.

5.5.1

ZEBRA Battery

Battery-specific energy (Wh/kg) depends on the battery technology, where the discharge and charge efficiency does not only depend on battery technology but

5.5 Series Hybrid Electric Vehicle (SHEV)

155

Table 5.4 ZEBRA battery parameters [95] Capacity (Ah) Rated energy (kWh) Max. open circuit voltage (V) Max. Regen. voltage (V) Max. discharge current (A) Cell type and No. of cells No. of strings Specific power (W/kg) Peak power (kW) at 2/3rd open-circuit voltage; for 30 s or until max temperature rated 335  C Cooling

32 17.6 575 670 112 ML3C 216 1 178.5 32 Forced air

also on the battery state-of-charge (SOC). Note, SOC is a key quantity to measure the amount of electric energy stored in or released from the battery during charging and discharging, respectively, and it is calculated based on (5.1). SOC ¼

QA QT

ð5:1Þ

where QA is the actual battery charge and QT is the total battery charge. Usually, the battery discharge efficiency is at its highest value at high SOC values, and it decreases with the SOC. The ZEBRA battery considered here is capable of tolerating 5%–10% cell failure and discharging to full capacity without degradation [184]. The ZEBRA battery has passed various tests, including crash test at 50 km/h, short circuit, vibration, underwater, and external fire tests [184– 187]. Compared to Lithium-ion (Li-ion), the ZEBRA battery has a comparable specific energy; however, its specific power is lower [184]. The SOC for the ZEBRA battery considered here varies between 0.1 and 1, while the battery cell open-circuit voltage varies between 2.32 V and 2.66 V. Table 5.4 lists specifications of a 216-cell ZEBRA battery while Fig. 5.11 shows the battery open-circuit voltage curve over a driving cycle. There are three regions of battery operation, which are specified as Region 1: the battery system has the highest charge (575 V), Region 2: the battery voltage is at its nominal value (555 V), and Region 3: the battery voltage decreases to minimum operating voltage from its nominal value. Furthermore, Fig. 5.12 shows the battery terminal voltage transient and SOC of Region 2 in Fig. 5.11. Note, during operation, a battery management interface (BMI) supervises the battery status, that is, voltage, current, and SOC, and ensures that all the battery parameters are within the normal operating states.

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5 Electric and Hybrid Electric Powertrains

Fig. 5.11 Characteristics of ZEBRA battery [95]

Fig. 5.12 Region 2 of Fig. 5.11 terminal voltage and SOC for ZEBRA battery during NEDC [95]

5.5.2

Internal Combustion Engine (ICE)

Generally, ICEs demonstrate high efficiency and provide a low specific fuel consumption over a relatively small region of their engine power-speed characteristics, as shown Fig. 5.13. They also demonstrate particularly high fuel consumption and emissions during transient engine operation. Hence, ICE operating point can be adjusted by controlling its output power at a predefined fixed speed. The selection of an adequate operating point assures an efficient and optimized ICE operation in SHEVs. Here, a benchmark powertrain is used to model vehicle physical parameters, required traction torque, and the energy demand, while a Toyota Prius 43 kW engine actual test data [188] is used to model the ICE. Note that although the Prius peak demand may reach 43 kW, the average power for a Toyota Prius over New European Driving Cycle (NEDC) is around 3.2 kW [188]. The Toyota Prius considered in this section is a 1.5-ton vehicle. Figure 5.14 shows that the scaled-down engine used carbon dioxide (CO2) emission data of the Toyota Prius 43 kW engine based on both

5.5 Series Hybrid Electric Vehicle (SHEV)

157

Fig. 5.13 Efficiency map of a typical ICE [189]

Fig. 5.14 Toyota Prius engine emission for different power and speed [95]

an actual taxi engine and the calculated peak power requirements calculated for the proposed driving cycles’ emissions based on CO2 for an average power varying from 0.5 kW to 3.2 kW and for a speed range of 477 RPM (50 rad/s) to 4770 RPM (500 rad/s). As seen, the emissions are lowest when the ICE operates at 3000 RPM (314 rad/s) while delivering 3–3.2 kW power. Therefore, in the analyzed SHEV powertrain, the ICE is to be operated at its optimum speed of 3000 RPM to deliver a fixed power to the HPM generator system. Furthermore, the SHEV fuel consumption (fuel economy) based on ICE output power (PICE) and efficiency (ηICE) can be calculated as

158

5 Electric and Hybrid Electric Powertrains

VFC ¼

FCR  PICE  t EC  ηICE

ð5:2Þ

where FCR represents the fuel consumption rate, t is the instantaneous driving time, and EC is the energy content of gasoline in consumed fuel rate. The VFC is used to calculate the mass of the SHEV fuel tank (in kg). Note that the mass is based on the maximum capacity of the vehicle fuel tank.

5.5.3

Engine-Mounted Multiphase HPM Generator

In the SHEV considered here, the vehicle is powered by two sources of energy, a battery (ZEBRA) system as a main power source, and an ICE driving a HPM generator as an auxiliary power source. In the scheme shown in Fig. 5.15, the ICE is mechanically decoupled from the main SHEV powertrain using a multiphase HPM generator. Therefore, mechanically decoupling the ICE from the vehicle powertrain (and hence the road speed) and operating the engine at a fixed speed and output power offers the potential for reduced engine size, fuel consumption, and emissions. As discussed in the previous chapters, the HPM generator has two rotor elements: a PM rotor with fixed excitation and a WF rotor with adjustable excitation. At a given speed, the PM rotor induces a fixed stator voltage while the WF rotor induces a variable but controlled stator voltage. Therefore, the HPM generator’s total output voltage, which is the sum of induced PM and WF voltages, is controlled over a prescribed range. The SHEV DC-link voltage may vary from its nominal value as the vehicle accelerates/decelerates. The DC-link voltage profile shown in Fig. 5.15 is measured when the vehicle operates on the NEDC driving cycle. As observed, the voltage transiently exceeds its nominal value (555 V) during vehicle deceleration. However, when the vehicle accelerates, the voltage significantly drops (to 350 V) due to a sudden increase in the current required for the traction motor to drive the vehicle to the acceleration. The higher current flow results in a higher equivalent impedance voltage drop, hence transiently reducing the DC-link voltage. The stronger the acceleration/deceleration, the higher the voltage drop/increase. The HPM generator is interfaced to this variable voltage DC-link; therefore, its rectified output voltage is required to follow the variations. Traditionally, the ICE provides a variable speed to the shaft, hence facilitating the HPM generator output voltage control. However, as previously discussed, the ICE speed is maintained fixed to maintain an optimum efficiency. Therefore, as the electrical system dynamics are much faster than those of the ICE, the HPM generator output voltage is controlled by the WF rotor excitation to match the HPM generator rectified voltage to the DC-link voltage.

5.6 Electric Vehicle Range Extender

159

Fig. 5.15 The SHEV powertrain components and DC-link voltage variation

5.6 5.6.1

Electric Vehicle Range Extender Introduction

To extend the range of all-electric vehicles, a larger battery may be used. However, alternatively, this can be achieved by using auxiliary power sources such as ICE or fuel cell together with the main battery, as shown in Fig. 5.16. Various energy management and control strategies for onboard and off-board range extender functions for SHEV powertrain topology have been studied. In the following section, a literature review of EV range extender studies will be discussed along with a detailed evaluation of an ICE/HPM generator that is used as an onboard range extender in the SHEV powertrain.

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5 Electric and Hybrid Electric Powertrains

Fig. 5.16 Gasoline-powered ICE and fuel cell technologies for EV range extension (a) Lotus range extender engine-generator system [190] (b) ALP-5 fuel cell range extender [191]

5.6.2

Literature Review of EV Range Extender Studies

Different control strategies for HEVs to optimize the operation of SHEVs for fuel economy and emissions are investigated in [192–204]. The approach presented in [192] optimizes the SHEV fuel consumption in the presence of three energy sources: an ICE-generator set, a battery, and ultra-capacitors. The authors in [192] offer a torque distribution strategy between the traction motors in the SHEV that minimizes energy usage. The study in [193] conducted on a sports class SHEV utilizes the battery system to reduce the power demand instead of the battery acting as a primary energy source. A control strategy proposed in [193] minimizes the size of the battery system while considering the driving range. In [194], the results show that by a proper selection of battery size and operating the SHEV on combination of ICE and on-board energy storge the fuel consumption is minimized. A study of cost map for a bus utilizing SHEV powertrain is presented in [195], where the proposed control strategy optimizes an ICE-generator set while targeting the desired emission. The approach in [195] is based on a two-step algorithm for size reduction and demand determination. A power distribution control for the SHEV energy sources, that is, a hybrid thermostat strategy, is proposed in [196], where a thermostat strategy is combined with traditional power flow to improve efficiency and achieve target performance. Optimization of a small ICE in terms of fuel consumption and emission by maintaining a fixed ICE speed and a variable output torque is discussed in [197]. However, the downsized ICE in [197] does not operate at its maximum efficiency. A two-step strategy to maximize the fuel economy of a SHEV powertrain configuration is proposed in [198], where a genetic algorithm is used to determine component sizing and a dynamic programming is employed for supervisor control optimization. Analysis of a 10-kW ICE-PM generator system supplying a constant DC-link power for a small HEV is presented in [199, 200], while [201] proposes an

5.6 Electric Vehicle Range Extender

161

Fig. 5.17 Electric vehicle with an off-board range extender on a trailer (courtesy of EP Tender) prototype [203]

Fig. 5.18 Plug-in Nissan e-NV200 van with a fuel cell range extender [205] This vehicle photo is not sponsored, endorsed, or promoted by Nissan North America, Inc., or their affiliates

optimized control for a range extension in SHEVs based on power demand prediction. A simulation powertrain platform, as in [202], combines two energy sources: an energy-dense ZEBRA and a power-dense supercapacitor. The ZEBRA battery is specified to fulfill the vehicle range requirements, while the supercapacitor provides the peak power for acceleration and regenerative braking. A control approach is proposed in [203], where it manages the energy between a battery and a rental range extender to minimize the end-use incurred costs. The range extender comprises mainly an ICE coupled with an electric generator. Furthermore, [203] investigates different operational scenarios and presents benefits of convex optimization compared with dynamic programming. The authors in [204] propose a hydrogen-generator system that can use a mix of gasoline and hydrogen as a range extender for EV powertrain. An example of an actual off-board and onboard EV range extender functions is shown in Figs. 5.17 and 5.18. Figure 5.17 shows an actual off-board ICE-generator system used as a range extender for EV, while Fig. 5.18 shows a plug-in HEV that has an onboard fuel cell energy source.

162

5.7

5 Electric and Hybrid Electric Powertrains

ICE/HPM Generator Range Extender in SHEVs

The structure of a SHEV powertrain with an ICE/HPM generator system is shown in Fig. 5.19b, while Fig. 5.19a presents the nine-phase HPM generator prototype that has previously been developed in Chap. 4. A 2.5-ton vehicle [185] is chosen as a benchmark system based on which two powertrain configurations are studied: (i) an all-electric powertrain utilizing a high energy density battery system acting as a standalone energy source, here a commercially available ZEBRA battery technology, and (ii) a SHEV powertrain featuring a primary source, that is, a ZEBRA battery and an ICE/HPM generator system acting as an axillary energy source. Figure 5.19 shows a schematic view of the SHEV powertrain under study. The ICE considered here is based on available data from the

Fig. 5.19 Series hybrid electric vehicle powertrain (a) Nine-phase HPM generator prototype (b) System configuration

5.7 ICE/HPM Generator Range Extender in SHEVs

163

Toyota Prius engine [188]. In the SHEV, Fig. 5.19, the ICE is used as a constant energy source and, together with the HPM generator, acts as a range extender to the SHEV powertrain. Therefore, the ICE operates at a fixed speed and spins a HPM generator, the output of which is rectified and connected to the vehicle DC-link.

5.7.1

Vehicle Traction Machine Torque

The future of electrified drivetrain lies on the new landscape that considers compatibility, reliability, efficiency, costs, and high torque density, a major requirement of old and new types of electric machine topologies along with their drive schemes. Brushless DC machines with nonsinusoidal (more-trapezoidal) back-EMF waveform are considered one of the established candidates for the growing field of traction motor drive system [206–208]. In general, motor control is achieved via either sensorless [209, 210] or sensored [208, 211, 212] control algorithm. Sensored types incorporate the use of encoders, resolvers, or Hall effect sensors to always detect rotor position with respect to the stator to achieve a proper voltage source converter (VSC) commutation, while the sensorless control removes any sensored components from the design. Note that it is hard to have a sensorless position control, where torque control is difficult to be applied sensorlessly. Therefore, a simplified brushless DC machine with an innovative motor drive system is of interest to many researchers. Here, for the upcoming analysis, the prime mover for the vehicle is a three-phase brushless DC motor acting as a traction machine, with an integrated gear reduction and differential drive to the vehicle back-axle. The efficiency of the gear system is assumed to be constant, while the brushless DC traction machine efficiency is calculated from the machine efficiency map. The brushless DC motor is controlled via a three-phase VSC, the DC supply to which is provided by the onboard battery pack, an ICE/HPM generator combination, using conventional vector control. Moreover, the traction machine torque equation can be extracted from the road vehicle kinematics, which is used to estimate the dynamic tractive requirement of the vehicle drivetrain. According to the basic principles of physics, any type of vehicle motion can be determined by analyzing the forces acting on it in the direction of motion, as illustrated in Fig. 5.20. Hence, the vehicle resultant force governs the motion based on Newton’s second law. Here, when the vehicle moves, it encounters different resistive forces that try to retard its motion, such as tire to road power loss or rolling resistance (Fr), aerodynamic resistance or aerodynamic drag (Fd), resistive force related to road gradient or uphill resistance (Fg), and transient force required to accelerate or retard the vehicle (Fa). F r ¼ K r mg cos θ

ð5:3Þ

164

5 Electric and Hybrid Electric Powertrains

Fig. 5.20 Forces acting on a vehicle while in motion

1 F d ¼ ρC d A f v2 2

ð5:4Þ

F g ¼ mg sin θ

ð5:5Þ

Fa ¼ m

dv dt

ð5:6Þ

where Kr is the rolling resistance coefficient, m is the mass of the vehicle, θ is the road gradient, g is the gravitational constant, ρ is the air density, Cd is the drag force coefficient, Af is the vehicle frontal area, and v is the vehicle linear velocity. Therefore, the net traction force is defined in Eq. (5.7), and based on the equation of motion, the road wheel torque equation can be calculated as in (5.8), where Jw, ωw, and rw are the wheel inertia, angular velocity, and mean radius, respectively, and df is a distribution factor proportion to torque distribution on the vehicle rear axle. F net ¼ F r þ F d þ F g þ F a

T w ¼ Jw 

dωw þ d f  r w  F net dt

ð5:7Þ

ð5:8Þ

The output torque of the traction machine can be modified if a gear stage is included in the drivetrain system, which is related to the road wheel torque by the

5.7 ICE/HPM Generator Range Extender in SHEVs

165

total transmission gear ratio (ntg), transmission efficiency (ηt), and the machine rotor inertia (Jm). By combining these components into (5.8) yields a general expression for traction machine torque: Tm ¼ Jm 

dωm 1 T þ ntg ηt w dt

ð5:9Þ

Furthermore, the wheel and traction machine angular velocity can be expressed in terms of vehicle linear velocity as ωw ¼

v rw

ωm ¼ ntg

v rw

ð5:10Þ

ð5:11Þ

Hence, the traction machine torque equation can be expressed in terms of vehicle linear velocity by substituting (5.7), (5.8), (5.10), and (5.11) into (5.9) [213] as shown in (5.12). Note that the parameters included in Eq. (5.12) depends on the vehicle type, and the required mechanical power (Pm) to satisfy the prescribed driving cycle is given in (5.13).  Tm ¼

ntg ηt rw



 þ

Jw ntg ηt r w



 þ

d f rw m ntg ηt



Pm ¼ T m ωm

5.7.2

dv d f r w þ ðF þ F g þ F d Þ ð5:12Þ dt ntg ηt r ð5:13Þ

Hybridization Ratio

In the SHEV powertrain, the relative size of the battery and the ICE is an important design criterion as it defines how strongly the energy sources are combined. Therefore, for the scheme presented in Fig. 5.18, a hybridization ratio is characterized to measure the power share between the battery and the ICE/HPM generator. This power share depends on the driving cycle peak and average power requirements. One of the design scenarios in the SHEV is the prevision of a steady power by ICE/HPM generator system while the dynamic power is provided by the battery. The ICE in this case is smaller, produces less emission, and is more efficient as it is designed to provide a constant power at its optimum operating region. Therefore, in the SHEV scheme presented in Fig. 5.18, the ICE is set to supply a constant vehicle average power, spinning the HPM generator at a constant speed, while the ZEBRA

166

5 Electric and Hybrid Electric Powertrains

battery supplies the peak power as the vehicle accelerates. In the event of vehicle idling or deceleration, the ICE/HPM generator system is used to charge the battery. A constant power source that provides the vehicle energy, E (Wh), during a driving cycle with T (hrs) duration is calculated as Pconst ¼

E T

ð5:14Þ

For the SHEV configuration, the constant power source is the maximum allowable ICE-HPM generator output power, hence, Pconst ¼ PICE=HPM ð max Þ

ð5:15Þ

Therefore, the minimum battery power required to provide vehicle total demand is calculated as Pbatð min Þ ¼ Ppeak  PICE=HPM ð max Þ

ð5:16Þ

where Ppeak is the peak power demand. Thus, the hybridization ratio for the SHEV is calculated as HR ¼

Ppeak  PICE=HPM ð max Þ  100 Pbat

ð5:17Þ

Using equation (5.17), a percentage power share between the vehicle battery and the ICE/HPM generator is defined.

5.7.3

Range Extender Sizing in SHEV Powertrain

Searching for the adequate ICE/HPM generator size depends on the electric vehicle’s desired driving range. A sequential algorithm can be implemented in a simulation platform to size the primary power source, here the ZEBRA battery, and axillary power unit, here the ICE/HPM generator, in the SHEV powertrain. The algorithm consists of two sections: (i) an all-electric and (ii) a hybrid model that searches for an adequate onboard energy source and a hybridization ratio subject to the energy source’s output power (kW), mass (kg), and a desired extended range (km), as illustrated in Fig. 5.21. The input to the platform includes driving cycles, vehicle physical parameters, and profile of electrified powertrain components for any type of road vehicle. This repetitive process is intended for the hybrid mode only, which is utilized to obtain an adequate average power percentage share (X) that lies between (0 and 100%) of the maximum allowable ICE/HPM generator fixed power output, as illustrated inside the red dotted line in Fig. 5.21. Note that the average vehicle power demand over a specified driving cycle is set as a maximum allowable power to be

5.7 ICE/HPM Generator Range Extender in SHEVs

167

Fig. 5.21 Energy management algorithm for EV driving range extension Table 5.5 Example of power requirements for a 2.5-ton vehicle over two driving profiles Driving cycle profile NEDC HWFET

Average power (kW) 18 20

Peak power (kW) 63.2 62.0

Cycle times (s) 400 765

provided by the axillary onboard power unit. The fuel consumption and emissions are calculated based on power demand over the selected driving range and the calculated hybridization ratio. Note that the ICE/HPM generator constant power is maintained via a HPM generator excitation field control function, which will be discussed in detail in Chap. 6. Here, for two driving cycle profiles, such as NEDC and HWFET, there is a wide disparity in the peak-to-average power requirements for a 2.5-ton vehicle, as summarized in Table 5.5. For the NEDC suburban driving cycle, as shown in Table 5.5, the vehicle peak power demand is 63.2 kW. Using the developed simulation platform, the energy consumed during the driving cycle and time is calculated and, based on Eq. (5.14), an 18-kW constant power source is set to produce all that energy. Hence, the minimum battery power calculated based on Eq. (5.16) is

168 Table 5.6 Two different HPM generator ratings

5 Electric and Hybrid Electric Powertrains Parameters PM flux per single turn (mWb) Axial length (mm) Mass (kg) Volume (mm3) Phase number of turns Parallel conductor per turn Phase synchronous inductance (mH) Stator resistance (Ω) Rotor resistance (Ω)

3 kW 0.25 40 10.7 1.5 54 1 2.3 2.5 16.3

14.5 kW 1.22 176 52.1 6.9 11 8 0.43 0.17 34.4

Note: Rotor turns ¼ 80 turns, number of poles ¼ 32, speed ¼ 3000 RPM, peak EMF voltage ¼ 269 V, specific power ¼ 280 W/kg, and power density ¼ 2.1 MW/m3 (for lamination volume excluding casing) Fig. 5.22 Hybridization ratio between pure all-electric and pure gasoline powertrain

45.2 kW. Therefore, to switch from an all-electric to a SHEV powertrain, the maximum battery power required is 63.2 kW (i.e., a total of two 216-cell ZEBRA batteries listed in Table 5.4), while the minimum battery power is 45.2 kW. Therefore, the ICE/HPM generator is set as the constant power source whose size varies from zero in an all-electric to 18 kW for the SHEV powertrain. Similarly, for a 62-kW vehicle peak power in HWFET driving cycle (Table 5.5), a 20-kW constant power source, ICE/HPM generator, is calculated to be sufficient to produce all the required energy while the minimum battery power is calculated as 42 kW. Therefore, to turn the benchmark 2.5-ton pure EV into a SHEV, an axillary ICE/HPM generator unit is set to provide an average fixed power while spinning at a fixed speed of 3000 RPM. As discussed, this will result in ICE minimum fuel consumption and maximum efficiency. Table 5.6 lists the parameters for 3-kW and 14.5-kW HPM generator ratings, which are calculated for 95% and 77% powertrain hybridization rations, respectively, as illustrated in Fig. 5.22.

5.7 ICE/HPM Generator Range Extender in SHEVs

5.7.4

169

Study Cases

To examine the ICE/HPM generator system, two case studies are considered here. Case 1 is composed of two scenarios that compare a pure EV and a SHEV with a hybridization ratio (HR) of 95% based on the 3-kW ICE/HPM generator system over both NEDC and HWFET driving cycles. Case 2 is composed of two scenarios that compare a pure EV and a SHEV with a HR of 77% based on a 14.5-kW ICE/HPM generator system that is scaled from the 3-kW prototype unit. Case 2 attains the desired minimum range of driving between some destinations without passing the allowable maximum ICE/HPM generator system output power. Note that the ZEBRA battery size in the pure EV in Case 1 is calculated based on a mass equivalent to the total weight of 3-kW ICE/HPM generator and full fuel tank. For Case 2, the same approach is taken, albeit for a 14.5-kW ICE/HPM generator. The analysis results of the suburban driving cycle case studies are shown in Table 5.7. The results illustrate that by utilizing the proposed SHEV configuration compared with all-electric powertrain, the driving range can be extended by 18% and 23% for HWFET and NEDC cycles, respectively, in Case 1, and the driving range can be extended by 34% and 52% for HWEFT and NEDC, respectively, in Case 2. However, using the combination of Case 1, the vehicle could not reach the proposed destination, but the energy used was better than the pure battery scenario. In Case 2, where equal energy storage mass for both scenarios was taken into account, the new rating of the ICE/HPM generator system in the vehicle powertrain reached the specified destination and passed it by 102 km with less CO2 emission, compared with the European Commission proposal 2015–2021, which is Table 5.7 Comparison between all-electric (pure EV) and SHEV over two driving cycles Cases Scenarios (1–2–3-4) Driving cycle Range (km) Fuel use (l) Fuel use (l/km) Fuel use (m/g) SOC CO (g/km) CO2 (g/km) HC (g/km) HC (g/km)

Case 1

Case 2

Pure EV (ZEBRA battery) HWFET NEDC

SHEV (ZEBRA battery and 3.0kW ICE/HPM) HWFET NEDC

Pure EV (ZEBRA battery) HWFET NEDC

SHEV (ZEBRA battery and 14.5kW ICE/HPM) HWFET NEDC

103 0 0

89.3 0 0

121.8 1.57 0.013

110.3 1.72 0.016

154.3 0 0

124.2 0 0

361.9 21.7 0.06

437.5 30.9 0.071

0

0

220.7

181.1

0

0

47.1

40.0

0.1 0 0 0 0

0.18 0 0 0 0

0.1 0.56 25.3 0.136 0.265

0.149 0.65 30.14 0.159 0.31

0.1 0 0 0 0

0.24 0 0 0 0

0.1 1.87 86.5 0.456 0.89

0.1 2.20 102 0.537 1.05

Note: The energy content of gasoline in 1 liter ¼ 32.2 MJ and the gram to liter conversion factor ¼ 737.22

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5 Electric and Hybrid Electric Powertrains

Fig. 5.23 Battery terminal voltage for all-electric and hybrid mode for a 14.5-kW ICE/HPM generator over NEDC case (a) All-electric mode (b) Hybrid mode

120–130 g/km [214]. The impact on the driving range due to the proposed ICE/HPM generator size in SHEVs is shown in Fig. 5.23. Such that, Fig. 5.23a presents the battery terminal voltage for all-electric and Fig. 5.23b presents the hybrid mode of Case 2 for the NEDC suburban driving scenario. (Fig. 5.23).

5.8

Conclusion

In this chapter, an overview of different EVs and HEVs based on powertrain configuration, per-unit cost, fuel economy, driving range, and wheel architecture is presented. Various battery technologies used in electric and hybrid electric vehicle powertrains are discussed. Electric vehicle range extension, fuel economy, and emissions are evaluated over HWFET and NEDC driving cycles at different vehicle powertrain hybridization ratios, which are characterized by the ICE/HPM generator and ZEBRA battery system in SHEV configuration. It is shown that by employing a 14.5 kW ICE/HPM generator system, as discussed in Sect. 5.7.4, the requirements of emissions lower than the 2015–2021 European Commission proposal are achieved. The battery energy can be used more efficiently, and an improved battery life is suggested by virtue of the reduced transient battery loadings.

Chapter 6

Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

6.1

Introduction

Generally, ICEs demonstrate a low specific fuel consumption over a relatively small region of their engine power-speed characteristics. They also demonstrate particularly high fuel consumption and emissions during transient engine operation. Therefore, mechanically decoupling the ICE from the vehicle powertrain and operating the engine at a fixed speed and output power offers the potential for reduced engine size, fuel, and emissions. For hybrid EV topologies, a significant saving in energy can be realized by leveling of the vehicle energy demand via the inclusion of a peak power buffer in the vehicle powertrain, downsizing the ICE power capability and operating the ICE over the most optimum region of the engine power-speed characteristics. Therefore, ICE acts as the prime mover to an electrical generator specified to satisfy the vehicle average power demands, while a battery system supplies the main peak power demands. Hence, one of the control objectives is to keep the ICE/HPM generator output power constant while minimizing speed variation via regulation of the HPM wound field excitation. Recent studies have shown that brushless permanent magnet generators with a secondary source of excitation are an interesting topology for implementation in hybrid EV powertrain [94–96, 106–108, 215]. For HEVs, the use of dual power and energy sources requires a detailed analysis of the drivetrain in order to optimize component specifications and energy management strategies [216]. In particular, the vehicle powertrain DC-link voltage may typically have a 30% voltage variation during acceleration and deceleration due to vehicle power transients, an example of which is illustrated in Fig. 6.1 [217, 218]. Hence, a form of voltage management is required for the ICE/electrical generator system, thus the interest in HPM generators that can be designed to match the 30% voltage variation. It is widely recognized in the literature that, in terms of higher efficiency and power density, PM machines with an active rectification stage represent a better candidate for use in SHEVs than classical wound rotor machines with passive © Springer Nature Switzerland AG 2022 A. S. Al-Adsani, O. Beik, Multiphase Hybrid Electric Machines, https://doi.org/10.1007/978-3-030-80435-0_6

171

172

6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

Fig. 6.1 Schematic of a series hybrid EV powertrain and typical DC-link voltage variation

rectification [36, 37]. Afjei et al. present a new configuration of a switch reluctance motor with field assistance that represents a hybrid generator topology [36, 37]. A major drawback of the active rectification scheme is the cost of power electronics, typically accounting for two-thirds of the total generator system cost. Yoo et al. discuss a power flow management scheme for a series hybrid electric vehicle, proposed mainly for military applications [219]. The powertrain discussed by Yoo et al. comprises a diesel engine linked to an interior PM (IPM) generator and passive rectification stage, lead-acid battery, and supercapacitor bank. The primary voltage source is the engine-IPM generator, designed with a terminal voltage equal to the rated DC-link voltage. The output of the generator is controlled by varying the engine speed via the engine throttle. The lead-acid battery provides additional energy storage, while the supercapacitor is used to supply short-term power fluctuations [219]. The improvement of DC-link voltage variation by replacing part of the PM machine excitation with different wound field configurations has attracted some research efforts in recent years. For example, Bin He et al. discuss the average

6.2 HPM Machine Back-EMF Control Strategy

173

modeling and control of a diesel-based auxiliary power unit (APU) that consists of a diesel engine, wound field synchronous generator, and three-phase passive rectifier stage [220, 221]. The diesel APU provides the primary energy source for the vehicle while the battery acts as a transient power buffer. In the Bin He papers [220, 221], the vehicle DC-link voltage is supplied and regulated during power transients by the diesel engine fuel injection and synchronous generator field excitation. However, fuel consumption and pollutant emissions will not be minimized for this system due to the varied engine operating speed. A different operating philosophy is considered in this chapter. Here, it is assumed that the ICE is operated within a very tight speed envelope delivering a constant power of 3 kW to the vehicle powertrain. An average power demand of 3 kW is typical of small, 1.0–1.2 ton, urban vehicles and normally supplements the battery energy or charges the battery when the vehicle is at rest. Here, the battery technology considered is ZEBRA [222, 223], and the vehicle is assumed to operate over repetitive NEDC and ECE-15 driving cycles, these being automotive industry standard cycles for vehicle powertrain performance evaluation [224]. Note that this choice of battery technology or vehicle powertrain is not claimed to be the most optimum or appropriate solution but is considered purely to investigate the HPM generator excitation requirements and operating philosophy. The choice of 3 kW arises from the average power required to drive a typical family 1.5-ton vehicle over the ECE-15 driving cycle [225]. The HPM machine wound field is modified to facilitate the control of the total generated back-EMF. This chapter proposes a guideline for the method and implementation of a suitable control strategy and investigates the HPM machine stator coil number of turns, where design decisions posed by higher system DC-link voltages will be discussed. Given the available data for higher voltage ZEBRA batterypowered vehicles (i.e., 587 VDC), the initial vehicle system study is based on such a high voltage vehicle powertrain. Furthermore, characterization and operational envelope of the multiphase ICE/HPM generator system at low DC-link voltage winding (18-turn) will be discussed in detail over NEDC driving cycle.

6.2

HPM Machine Back-EMF Control Strategy

Dynamic models that can be used to analyze the system behavior of an ICE/generator system, power electronic converter, and controller scheme in HEV applications have been widely reported in the literature [226, 227]. This chapter utilizes a mixture of machine data obtained by FEA, vehicle DC-link voltage variation during driving cycle operation, and Matlab/Simulink vehicle simulation data [202] to link the generator dynamic model with real data collected by a vehicle data acquisition system. Part of the control of this dynamic system is subjected to input and output constraints, such as the ICE/HPM generator, which is controlled via an adaptive means of field excitation adjustment to present the overall system dynamic behavior in offline simulation.

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

6.2.1

Control Strategy Analysis

To deduce the HPM machine total back-EMF control strategy, the Matlab/Simulink model presented in Chap. 4 and illustrated in Fig. 4.15 is expanded to include the vehicle DC-link voltage variation during dynamic driving and the WF back-EMF regulation function via a flux-linkage feedback loop. An overview of the expanded model is illustrated in Fig. 6.2, showing the HPM generator, rectifier stage, and variable DC-link voltage potentials, for the case of VRDC > VDC-link, which supports the derivation of the system power flow equation. When delivering a constant power into the vehicle DC-link, the rectified DC voltage (VRDC) due to the back-EMF of the HPM generator has to be slightly greater than the DC-link voltage to overcome the DC-link resistance (RDC-link) and other voltage drops while facilitating the desired DC power flows. The HPM machine back-EMF control strategy calculations are divided into three stages. The first stage calculates the rectified output voltage for the required power output to the vehicle DC-link. The initial calculation steps are V RDC ¼ I DC RDClink þ V DClink

I DC ¼

V RDC  V DClink RDClink

PDCD ¼ I DC V DClink

Fig. 6.2 Simplified schematic diagram of HPM generator system

ð6:1Þ

ð6:2Þ

ð6:3Þ

6.2 HPM Machine Back-EMF Control Strategy

V RDC ¼ V DClink þ

175

RDClink PDCD V DClink

ð6:4Þ

where IDC is the DC current and PDCD is the desired DC output power. To account for any offset between the desired and demand DC-link output power, an error signal is added to (6.4):  V RDC ¼

R P V DClink þ DClink DCD V DClink

 þε

or  V RDC ¼

PDCD RDClink þ V 2DClink V DClink

 þε

ð6:5Þ

where the error signal is given by ε ¼ ðPDCD  PDCM Þk

ð6:6Þ

and PDCM is the actual DC output power and k is the error tuning gain. Furthermore, the second stage should provide the required excitation current value based on the previously obtained VRDC. Finally, the third stage utilizes the required excitation current, as specified in (6.7), to calculate the total HPM machine flux-linkage per phase, and this in turn is substituted into the phase back-EMF equation (4.39) in Chap. 4. λHPM ¼ λPM  λWF

var

¼ λPM  λWF

rated

If If

var rated

ð6:7Þ

where λPM and λWF _ var are fixed PM and variable WF flux-linkages, respectively, λWF _ rated is the rated WF flux-linkage, and If _ rated and If _ var are rated and variable WF currents, respectively. The model of Fig. 6.2 is now represented by the open-loop single line diagram illustrated in Fig. 6.3, as applied to the 18-turn, 9-phase HPM machine that was described in Chap. 4 and the model analyzed over incremental time periods representing a few electrical periods of the HPM machine operation, where the DC-link voltage is assumed constant. This analysis facilitates the development of the HPM machine design and control philosophy while providing benchmark data and waveforms. Note that in order to investigate the full system response characteristic, the DC-link voltage was varied over a much wider range than the 30% variation expected during vehicle operation, as illustrated in Fig 6.4, where the system DC-link voltage is varied from 40 to 250 V. The utility of this approach will be demonstrated later in this chapter where different voltage ranges, winding turns, and excitation schemes are considered against the options for field excitation control.

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

Fig. 6.3 HPM generator open-loop control system block diagram

Fig. 6.4 DC-link voltage and peak phase back-EMF with respect to the WF excitation current at constant DC-link output power (3 kW)

It was shown in Chap. 4 that the maximum DC-link power, which can be delivered by the HPM machine system at a fixed speed and hence fixed back-EMF with zero wound field excitation and at the thermal limit, was 3.15 kW. Control is applied to the system model to ensure that the HPM maintains a constant DC-link output power of 3 kW with minimal DC-link voltage ripple. Figure 6.4 illustrates the total (PM and WF) peak phase back-EMF and the DC-link voltage variation at this constant power of 3 kW for the 18-turn, 9-phase HPM machine with respect to wound field excitation current. Because of the fixed power output requirement, the field excitation current options are constrained, as illustrated by Fig. 6.4, where, for this specific machine, the HPM machine peak phase back-EMF can only be regulated over a field current range from 0.4 to 4.2 A. This result is a function of the

6.2 HPM Machine Back-EMF Control Strategy

177

Table 6.1 Operational points data of Fig. 6.4 Operational points (a) (b) (c) (d) (e) (f) (g) (h)

If (A) 6.83 2.36 0 0.4 0 2.46 5.04 6.6

VDC-link (V) 40 55 76 95 114 160 200 225

Vterm (peak) (V) 23.9 32.1 44.1 54.6 64.9 90.5 112.7 125.7

Back-EMF (peak) (V) 138.5 106.7 89.9 87.0 89.9 107.4 125.8 136.9

Is (peak) (A) 28.5 20.7 14.9 11.9 10.2 8.2 7.2 6.7

machine stator turns and DC-link power flow demand, as will be demonstrated later. As can be noted from Fig. 6.4, the back-EMF boost mode represents the dominant operating mode for this choice of HPM machine winding power demand and operating speed. Data for the points illustrated in Fig 6.4 are given in Table 6.1. When the DC-link voltage is low, the DC-link current must be high to maintain the same output power. Therefore, the field excitation current and hence total machine back-EMF are high since the total machine back-EMF has to compensate for the machine reactive voltage drop and the voltage drops in the rectification scheme. Essentially, the HPM machine is operating at what equates to a poor lagging power factor, as illustrated by Fig. 6.5a showing machine terminal voltage, total back-EMF, and phase current when generating into a DC-link voltage of 40 V. This generating condition is also identified on Fig. 6.4 as Point (a). As the DC-link voltage is varied from 40 to 225 V, that is, Points (a) to (h) on Fig. 6.4, the HPM phase current reduces and hence the machine and rectification voltage drops reduce. Referring to Figs. 6.4, 6.5, and 6.6 illustrate phase terminal voltage back-EMFs and current as the DC-link voltage varies from 40 VDC, Fig. 6.5a, to 225 VDC, Fig. 6.6h, and the HPM machine field excitation is varied from a maximum positive current, to a maximum negative current of 0.4 A and again to a maximum positive current, to maintain the fixed generated power demand of 3 kW. Figure 6.7 illustrates the same waveforms for points (a), (d), and (h), but with magnified axes to show the phase current detail and respective phase relationships more clearly. As can be seen, the boost mode at higher DC-link voltages results in the phase current becoming discontinuous, Fig. 6.7h. Note, the simulation results of Figs. 6.5 and 6.6 are not in steady state until  0.4 ms, hence the asymmetry of the waveforms during this time period. The simulation settles at around 0.4 ms, as inferred from the resulting waveforms.

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

Fig. 6.5 Phase voltage and current waveforms for four DC-link voltage levels for the 18-turn, 9-phase HPM machine at constant DC-link output power (3 kW). (a) VDC-link ¼ 40 V, If ¼ 6.83 A (b) VDC-link ¼ 55 V, If ¼ 2.36 A. (c) VDC-link ¼ 76 V, If ¼ 0 A (d) VDC-link ¼ 95 V, If ¼ 0.4 A

6.2.2

DC-link Design Options

The result of Fig. 6.4 presents a number of control options if only a 30% variation in the vehicle system DC-link voltage is required. The questions to be resolved are • Should the PM component of the HPM output voltage be reduced by the WF back-EMF component (i.e., bucked)? • Should it be increased by the WF back-EMF component (i.e., boosted)? • Should some element of buck-boost be considered for design? Therefore, having considered the analysis of the 18-turn machine over a much wider range of DC-link voltages, machines having the same lamination and PM design but with increased turns per phase were analyzed over wide voltage ranges, but still delivering the required 3 kW at 3 kRPM. Table 6.2 gives the machine winding electrical parameters for winding designs from 18 to 67 turns. Figure 6.8 illustrates the trend in DC-link voltage and peak phase back-EMF characteristics with respect to wound field excitation current for HPM machines having turns per phase ranging from 18 to 67. The results for the 18-turn machine are as per Fig. 6.4, while the results for turns from 62 to 67 come from winding design optimization for a vehicle system having a nominal battery open-circuit DC-link voltage of

6.2 HPM Machine Back-EMF Control Strategy

179

Fig. 6.6 Phase voltage and current waveforms for four DC-link voltage levels for the 18-turn, 9-phase HPM machine at constant DC-link output power (3 kW). (e) VDC-link ¼ 113.5 V, If ¼ 0 A (f) VDC-link ¼ 160 V, If ¼ 2.46 A. (g) VDC-link ¼ 200 V, If ¼ 5.04 A (h) VDC-link ¼ 225 V, If ¼ 6.6 A

600 V. The results for the 53-turn design are included as a check to verify the trend in the characteristic curves from 18- to 67-turn. As with the 18-turn machine design, the options to buck, boost, or buck-boost are determined by matching the HPM machine output to the required specified DC-link variation by choice of an appropriate number of turns per phase. Having simulated open-loop characteristics for the HPM as a function of winding turns (Fig. 6.8), a curve fitting function based on a sixth-order polynomial regression method is applied to the characteristics to calculate the desired nonlinear high order equations that represent the required wound field excitation current for a specified DC-link power demand, HPM machine parameters, and DC-link voltage variation. The open-loop model of Fig. 6.3 is now closed and includes a block with a defined curve fit function and a limiter block that sets the upper and lower limits to the DC-link voltage excursions. The excitation current calculation then inputs to the flux-linkage calculation block, as illustrated in Fig. 6.9, showing the complete HPM generator closed-loop control system block diagram. The second stage of the HPM system analysis procedure now calculates the desired HPM-rectified DC output voltage, VRDC, from the difference in actual and demand power, and system DC-link voltage, VDC-link, as illustrated in Fig. 6.9.

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

Fig. 6.7 Magnified views of Figs. 6.5a and d and Fig. 6.6h illustrating waveform phase relationships. (a) VDClink ¼ 40 V, If ¼ 6.83 A (d) VDC-link ¼ 95 V, If ¼ 0.4 A (h) VDC-link ¼ 225 V, If ¼ 6.6 A

6.3 HPM Machine Output Power Control

181

Table 6.2 HPM machine winding electrical parameters as a function of stator turns nstc (turns) Rs (Ω) LS (mH) λPM (mWb-turn)

18 0.245 1.16 4.468

53 1.466 9.67 13.155

62 2.006 13.23 15.389

63 2.039 13.73 15.637

65 2.167 14.64 16.134

67 2.373 15.49 16.630

Fig. 6.8 DC-link voltage and peak phase back-EMF with respect to the WF excitation current for varying turns per phase and constant DC-link output power (3 kW) (a) DC-link voltage versus WF excitation current (b) Back-EMF versus WF excitation current

6.3 6.3.1

HPM Machine Output Power Control Introduction

It is well known that the dynamic behavior of vehicle power requirements presents some control issues to the operational philosophy of hybrid EVs. Therefore, analysis of the vehicle powertrain is required in order to optimize component specifications and develop and understand the impact of varied energy management strategies. To

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

Fig. 6.9 HPM generator closed-loop control system block diagram

investigate the function of a HPM generator operating in a hybrid EV powertrain and thus be in a position to optimize or improve the machine electromagnetic design, case study designs must be undertaken. Here, the primary energy source of the hybrid powertrain is a ZEBRA battery having maximum and minimum terminal voltage levels of 585 and 405 V, respectively. The ICE-driven HPM generator system is used as a constant energy input to the vehicle powertrain DC-link, essentially acting in a range extending function. The electrical system dynamics are much faster than those of the ICE. Thus, the function of the HPM generator excitation is to match the generator output voltage to the varying DC-link voltage of the vehicle system while delivering a fixed power input to the DC-link. Constant ICE/HPM generator output power therefore requires control of the HPM generator field excitation current, which is calculated by feedback signals that monitor the DC-link voltage and current variations, HPM generator output power, and the desired output power, as previously discussed. Thus, the feedback signals provide continuous control over the field current that forces the generator terminal voltage to be larger than the DC-link voltage by a certain small value to maintain constant generator output power. As a result of this auxiliary hybrid power source control, an extended vehicle range will be achieved while maintaining maximum ICE efficiency. In the case studied, vehicle range is extended by more than 50%, for both repetitive NEDC and ECE-15 driving cycles, by the addition of a 3-kW range extender to the vehicle powertrain [164, 213]. The vehicle is a 2.4-ton electric vehicle operating as a public taxi in urban city areas. The range extender is a downsized ICE driving the HPM. The additional range could be realized by a larger ICE/HPM generator combination, as discussed previously in Chap. 5. For the ZEBRA battery technology, the battery open-circuit voltage, as illustrated in Fig. 6.10, shows full SOC (1.0 p.u.) (a), a gradually reducing region from around 0.95 to 0.2 SOC (b), a reduction in open-circuit voltage at 0.2 SOC, which is only

6.3 HPM Machine Output Power Control

183

Fig. 6.10 ZEBRA battery open-circuit voltage versus SOC characteristics [40]

Fig. 6.11 Battery terminal voltage during repetitive NEDC driving cycles (a) All-electric case (b) Hybrid case

characteristic of this battery technology (c), and the reducing voltage to fully discharged (0.2 to 0.0 SOC) (d) [40]. Figure 6.11 illustrates the vehicle system DC-link voltage variation during repetitive NEDC cycling showing the case for all-electric operation (a) and for series hybrid operation (b), or range extender mode where the range extender adds an additional constant 3 kW to supplement the battery energy. Similarly, Fig. 6.12 illustrates the vehicle system DC-link voltage variation during repetitive ECE-15 cycling showing the case for all-electric operation (a) and for series hybrid operation (b), or range extender mode as before. In both figures, the battery terminal voltage reductions are caused by acceleration current demands and the consequential voltage drop due to the battery internal resistance. Likewise, the peaks are due to regenerative current braking. Note that the mean voltage envelope is indicative of the ZEBRA battery open-circuit voltage characteristic, as shown in Fig. 6.10 [213].

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

Fig. 6.12 Battery terminal voltage during repetitive ECE-15 driving cycles (a) All-electric case (b) Hybrid case

Figure 6.13 illustrates examples of the 3 kW HPM generator during part of the duty cycles of Fig. 6.11, showing the rectified DC-link output voltage (a), the generator DC-link current output (b), and the rotor excitation current (c).

6.3.2

HPM Generator Operating Scenarios

A number of operating scenarios can be considered for the ICE/HPM generator system control. To investigate and understand the functionality of the ICE/HPM generator in a series hybrid EV powertrain, different operating scenarios relating to the battery terminal voltage levels are considered. Five case studies are made based on predefined HPM machine open-circuit back-EMF voltages with zero wound field excitation. With regard to the battery terminal voltage of the vehicle system, as illustrated in Fig. 6.11b, five operating scenarios are considered, as illustrated in Fig. 6.14. For each of the five scenarios, the vehicle system voltage varies from 420 VDC [40], requiring a predominantly boosting function from the WF back-EMF component, to 588 VDC [40], where the WF back-EMF function is essentially bucking that of the PM back-EMF. To determine the intermediate voltage levels of interest, and hence the HPM system operational philosophy, a further three regions (that are comparable to the battery open-circuit voltage characteristic previously illustrated in Fig. 6.10) were chosen for study, as illustrated in Fig. 6.14, showing the HPM PM back-EMF voltage levels for the five operating scenarios considered. There is therefore a decision to be made as to the operating voltage of the HPM generator with zero WF excitation, that is, whether the field current excitation increases or decreases the net terminal voltage of the generator to match the DC-link voltage variation at constant output power, as illustrated schematically in Fig. 6.14 by five-scenario concepts.

6.3 HPM Machine Output Power Control Fig. 6.13 HPM generator system variables variation for 3 kW at 3000 RPM (a) DC-link voltage (b) Generator DC-link current (c) Rotor excitation current

185

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

Fig. 6.14 Five operating scenarios of the HPM generator with respect to the DC-link voltage levels in volts

6.3 HPM Machine Output Power Control

6.3.3

187

Energy Loss Prediction for Two Driving Cycles

The HPM generator system model is solved using ZEBRA battery terminal voltage variation from repetitive NEDC and ECE-15 driving cycles provided from [213]. Key simulation data for the NEDC and ECE-15 driving cycles are presented in Tables 6.3 and 6.4, respectively. These tables summarize the HPM machine energy loss over the full battery discharge regime from 1.0 to 0.0 SOC. It is assumed that the machine iron, winding, and frictional losses are constant for the scenarios considered and that the machine copper losses form the dominant energy loss mechanism. In terms of HPM generator system energy losses, the results presented in Tables 6.3 and 6.4 show that scenarios 3 and 4 produce the lowest total HPM machine energy losses over the full battery discharge regime. This is further illustrated in Fig. 6.15, showing the minimum energy loss solution for both the NEDC and ECE-15 driving cycles. Referring to Tables 6.3 and 6.4, scenario 1 shows the highest average rotor copper and total energy losses; however, the stator copper loss is less than that for the other scenarios due to the stator winding turns. For scenarios 2, 3, 4, and 5, loss of the HPM wound field excitation by any means during vehicle operation would result in (high) fault current being generated into the vehicle system DC-link. Therefore, the HPM machine and rectifier stage should be fuse protected to prevent such a system fault. Alternatively, although scenario 1 produces the highest energy loss in the HPM generator system, it represents the most robust in terms of WF failure, a decision ultimately for the vehicle system designer. Note that, as discussed, the data in Tables 6.3 and 6.4 are obtained from the simplified simulation model. However, data for the excitation current (ΔIf) obtained from the detailed simulation model are Table 6.3 HPM machine rotor excitation options for repetitive NEDC driving cycles Parameters Voltage due to PM (V) DC voltage range due to WF (V) (ΔVDC ¼ 167V) Excitation voltage range ΔVf (V) Excitation current range ΔIf (A)

RMS excitation current (A) RMS phase current (A) Average stator copper loss (W) Average rotor copper loss (W) Stator energy loss (kJ) Rotor energy loss (kJ) Total energy loss (kJ)

Scenario 1 420 167 0 +46 0 2.624 (2.939) 0 (0.01396) 1.565 1.956 50.49 39.97 571.5 443.7 1015.2

Scenario 2 515 72 95 +19.2 26.6 0.9 (0.9207) 1.25 (0.98) 0.424 1.852 61.89 2.93 687.0 32.5 719.5

Scenario 3 535 52 115 +16 29.5 0.7574 (0.75268) 1.353 (1.0324) 0.411 1.844 62.41 2.75 692.8 30.5 723.2

Scenario 4 555 32 135 +8.6 37.5 0.375 (0.42828) 1.63 (1.138) 0.523 1.825 64.94 4.47 720.8 49.6 770.4

Scenario 5 575 +12 155 +3.5 42.5 0.1257 (0.117) 1.795 (1.2487) 0.654 1.796 68.85 6.99 764.2 77.6 841.9

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

Table 6.4 HPM machine rotor excitation options for ECE-15 driving cycles Parameters Voltage due to PM (V) DC voltage range due to WF (V) (ΔVDC ¼ 199V) Excitation voltage range ΔVf (V) Excitation current range ΔIf (A)

RMS excitation current (A) RMS phase current (A) Average stator copper loss (W) Average rotor copper loss (W) Stator energy loss (kJ) Rotor energy loss (kJ) Total energy loss (kJ)

Scenario 1 420 +168 31 45 9 2.56 (2.939) 0.52 (0.706) 1.682 1.925 48.9 46.2 988.3 933.7 1922.0

Scenario 2 515 73 126 18.5 37 0.86 (0.9208) 1.75 (1.247) 0.369 1.821 59.86 2.226 1209.8 45.0 1254.8

Scenario 3 535 53 146 15 40.5 0.7 (0.7527) 1.9 (1.2535) 0.319 1.816 60.54 1.659 1223.5 33.5 1257.1

Scenario 4 555 33 166 7.89 47 0.35 (0.42828) 2.03 (1.2762) 0.368 1.797 62.95 2.204 1272.3 44.6 1316.8

Scenario 5 575 +12 187 2.5 51.5 +0.15 (0.1169) 2.18 (1.3067) 0.503 1.766 66.63 4.127 1346.6 83.4 1430.0

Fig. 6.15 Total energy losses for the NEDC and ECE-15 driving cycles

also presented, for comparison, as shown in parentheses. Although there is some inconsistency in the excitation current range between the actual and simplified HPM generator model, this inconsistency can be considered a worst-case calculation; hence, the obtained results are considered acceptable regarding the choice of best operating scenario. Thus, for the lowest energy losses, scenarios 2 and 3 present the preferable operating scenarios. However, scenario 3 was selected due to the lower rotor energy loss compared with scenario 2. Figure 6.16 illustrates the advantage of scenario 3 over the other scenarios in terms of average wound rotor excitation power loss, while Fig. 6.17 illustrates the HPM generator wound field excitation current variation during repetitive NEDC driving cycles for the five operating scenarios.

6.3 HPM Machine Output Power Control

189

Fig. 6.16 HPM machine rotor power loss for the five operating scenarios and two driving cycles (a) Repetitive NEDC driving cycles (b) Repetitive ECE-15 driving cycles

6.3.4

Solving Final Choice with Full Simulation Model

To finalize the HPM machine design study, the 63-turn HPM design is analyzed at fixed DC-link voltage levels using the full dynamic model as a cross-check of the simplified model and to illustrate actual phase voltage and current waveforms. Figure 6.18 illustrates the total (PM and WF) peak phase back-EMF for the 9-phase, 63-turn HPM machine and the DC-link voltage variation while delivering a constant power of 3 kW with respect to wound field excitation current. As can be noted from Fig. 6.18, the back-EMF bucking mode represents the dominant

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

Fig. 6.17 HPM generator WF excitation current variation for repetitive NEDC driving cycles

operating mode for this choice of HPM machine winding, power demand, and operating speed. Data for the points identified in Fig. 6.18 are given in Table 6.5 for completeness. Again, because of the fixed power output requirement, the field excitation current options are constrained. For this specific machine and the

6.3 HPM Machine Output Power Control

191

Fig. 6.18 DC-link voltage and peak phase back-EMF with respect to the WF excitation current for 63-turn case and a constant DC-link output power (3 kW) Table 6.5 Operational point data of the 63-turn HPM machine Operational points (a) (b) (c)

If (A) 0.753 0 1.27

VDC-link (V) 589 530 360

Vterm (peak) (V) 309.5 282.9 202.9

Back-EMF (peak) (V) 333.3 314.5 282.8

Is (peak) (A) 2.14 2.33 3.34

considered driving cycles, the HPM machine peak phase back-EMF can only be regulated over a field current range from 1.27 to 0.753 A. Referring to Figs. 6.18 and 6.19 illustrates phase terminal voltage, back-EMFs, and current as the DC-link voltage varies from 589 VDC, Fig. 6.19a, to 360 VDC, Fig. 6.19c, and the HPM machine field excitation is varied from a maximum positive current, to a maximum negative current of 1.27 A, and then again to a maximum positive current of 0.753 A, to maintain the fixed generated power demand of 3 kW. Figure 6.20 illustrates the same waveforms for points (a), (b), and (c), but with magnified axes to show the phase current detail and respective phase relationships more clearly. Note that, as before, the simulation results of Fig. 6.19 are not in steady state until  0.4 ms, hence the asymmetry of the waveforms during this time period. The simulation settles at around 0.4 ms, as inferred from the resulting waveforms.

6.3.5

Thermal Analysis Results of the Investigated HPM Machine

As a final check on the machine design solution, the general thermal analysis of Chap. 4 is now resolved for the stator and rotor losses specified to the vehicle application considered and the 9-phase, 63-turn HPM machine. The method utilized in Chap. 4 to predict dynamic and steady-state temperature distributions is used

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

Fig. 6.19 Phase voltage and current waveforms for eight DC-link voltage levels for the 63-turn, 9-phase HPM machine at constant DC-link output power (3 kW) (a) VDC-link ¼ 589 V, If ¼ 0.753 A (b) VDClink ¼ 530 V, If ¼ 0 A (c) VDC-link ¼ 360 V, If ¼ 1.27 A

again to predict the machine steady-state temperature based on the NEDC driving cycle loads. The steady-state thermal analysis results are detailed in Table 6.6 and illustrated in Figs. 6.21 and 6.22, showing much lower temperature distributions. Hence, the thermal analysis results are considered to be satisfactory and acceptable.

6.3 HPM Machine Output Power Control Fig. 6.20 Magnified views of Fig. 6.19 that illustrates waveform phase relationships (a) VDClink ¼ 589 V, If ¼ 0.753 A (b) VDC-link ¼ 530 V, If ¼ 0 A (c) VDC-link ¼ 360 V, If ¼ 1.27 A

193

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

Table 6.6 HPM machine thermal modal results Node name Ambient Housing Rotor lamination pole Rotor magnet Rotor lamination yoke Rotor surface Rotor copper Stator surface Stator yoke Stator tooth Stator tooth tip Stator winding average

6.4

PM section Temperatures ( C) 25.0 30.1 – 41.4 40.7 41.7 – 55.0 49.3 45.3 55.1 55.3

WF section Temperature ( C) 25.0 28.4 81.3 – 76.8 80.1 79.5 52.8 42.1 51.2 52.6 55.7

HPM Machine Characterization Using Brushless Excitor

Excitation systems, which are generally referred to as schemes that provide DC voltage and current to the wound field of electric generators, are either rotating or static [228, 229], where the rotating schemes can be brushed or brushless [230]. Due to issues of brush maintenance, commutators, and slip rings, the brushless exciters are more attractive and offer improved reliability and better performance [231, 232]. Brushless excitation schemes can be implemented using the concept of contactless energy transfer as rotating transformers [233–235]. In these schemes, there usually exist two sets of windings, stationary and rotating, and the power from the stationary winding is induced to the rotating section. Both single- and threephase rotary transformers have been proposed [235–237]. Here, the brushless DC excitation schemes are used to provide the field current and voltage to the wound field of hybrid excitation machines where applicable [33, 238, 239]. In [239], two-stage brushless excitation scheme for a hybrid excitation synchronous generator is proposed where the input power of the exciter is supplied from the output of the main generator. Dynamics and control of the exciter contribute to the system stability and operation and, as such, there have been studies focused on this area [240, 241]. This section discusses the characterization of the HPM generator in SHEVs for low DC-link voltage (85–125 V) using a brushless exciter scheme. This is achieved by (i) integration of HPM generator and brushless exciter in one machine housing as shown in Fig. 6.23, (ii) proposing a control strategy for the integrated HPM generator–exciter system in a SHEV, (iii) materializing the conceptual study in [241] by proposing a practical design for brushless exciter suitable for integrated system, (iv) characterizing the system using the previously discussed control strategy while the SHEV operates under NEDC, and (v) providing an analysis of the

6.4 HPM Machine Characterization Using Brushless Excitor Fig. 6.21 Steady-state temperature distribution for PM section of 63-turn HPM generator (a) Radial cross section (b) Axial cross section

195

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

Fig. 6.22 Steady-state temperature distribution for WF section of 63-turn HPM generator (a) Radial cross section (b) Axial cross section

6.4 HPM Machine Characterization Using Brushless Excitor

197

Fig. 6.23 Sketch of the HPM generator with the brushless excitation scheme (a) Exciter and HPM generator in one machine housing (b) Cross-sectional view of the HPM generator and brushless exciter rotor/stator laminations

integrated system by presenting operating performance curves, operational envelope, and efficiency mapping under prescribed operating conditions.

6.4.1

32-Phase Brushless Excitation Scheme

The brushless exciter rotor has 32 poles where the stator is constrained to laminations with 36 slots, as shown in Fig 6.23, which have to accommodate a DC winding supplied from the rectified output of the HPM generator via a buck-boost converter. The saturation of steel core constraints the air-gap flux density. Therefore, when considering the magnetic design of a brushless exciter with a given lamination, the

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

saturation of steel core constraints the air-gap flux density. As the number of magnetic poles reduces, the flux density in the back-iron increases bringing the operating point closer to saturation. In addition, for the brushless exciter, the number of poles is ultimately limited by the number of stator teeth since the wound field rotor acts as secondary winding. Therefore, to study a different number of poles (minimum 2 and maximum 36) and in order to avoid steel saturation, two different pole combinations that is, 18- and 36-poles for the brushless exciter are chosen. To form 18-pole configurations, two different winding arrangements for the stator laminations can be considered: (i) Design 1: 18 concentric coils wound around every other stator tooth (single layer) and (ii) Design 2: distributed winding with every other tooth having two coil sides (double layer). However, for the 36-pole configurations (Design 3), there are 36 coils wound around each stator tooth, forming a double layer concentrated winding. The exciter air-gap flux density and stator slot current density with respect to stator excitation current are presented in Fig. 6.24a. Therefore, the excitation current is chosen such that the air-gap flux density approaches saturation, equating to an excitation current of 300 Ampere-turns (A-t) and corresponding to a peak air-gap flux density of 0.89 T, 0.74 T, and 0.94 T and a stator slot current density of 2.49 (A/mm2), 4.97 (A/mm2), and 4.97 (A/mm2) for Design 1, Design 2, and Design 3, respectively. Here, Design 3, that is, 36-pole with 36 concentrated stator coils connected in series is selected as the brushless exciter stator configuration. Note that the HPM generator is a 32-pole machine, and this is not to be confused with the 36-pole brushless exciter. Figure 6.24b illustrates the rotor coil flux-linkage and induced back-EMF per turn per unit axial length versus electrical angle of rotation for a stator excitation current of 300 A-t and a rated speed of 3000 RPM. Note that the more trapezoidal voltage waveform in Fig. 6.24b is beneficial as it results in lower rectified voltage ripple at the output of exciter rotating rectifier, hence reduced filtering requirements. The brushless exciter rotor is constrained to laminations with 32 slots, the same as HPM WF, into which an AC winding has to be accommodated. A multiphase design approach for the brushless exciter rotor winding is considered for high power density and a high-quality DC at the output of rotating rectifier and to minimize/eliminate passive capacitances and improved reliability. Assuming there are m number of rotor phases, the number of slots per pole per phase, that is, phase belt, to accommodate 36 poles would be 32/(36 m). No matter how many phases are chosen, the phase belt ratio is always less than unity, resulting in 32 concentrated coils wound on each rotor tooth. However, due to the symmetry of the magnetic design, teeth that are 180 degrees mechanically displaced from each other experience the same flux density in the same direction. Given the same number of coil turns, back-EMFs of diametrically opposite rotor coils have the same magnitude and phase shift. Therefore, for 32 rotor coils, a maximum of 16 phases can be achieved. To form 16 phases, a series or parallel connection of rotor coils can be realized. For the series connection, as illustrated in Fig. 6.25a, two coils are series wound with one connection taken to the main star point and the other to a 16-leg rotating rectifier. In the parallel connection shown in Fig. 6.25b, one end of each coil is connected to

6.4 HPM Machine Characterization Using Brushless Excitor

199

Fig. 6.24 Brushless exciter characteristics (a) Air-gap flux density and stator slot current density study (b) Per-unit rotor flux linkage and EMF at 3000 RPM

the main star point and the other ends are taken to a 32-leg rotating rectifier. Note that although there are 32 coils and a 32-leg rotating rectifier, the winding is still a 16-phase winding arrangement. The benefit of the 16-phase parallel connection with a 32-leg rotating rectifier is minimized interconnections, which improves the manufacturing process, an important feature for mass production. Thus, in the

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

Fig. 6.25 Two brushless excitor rotor winding designs (a) 16-phase series configuration (b) 16-phase parallel configuration Table 6.7 HPM generator brushless exciter specifications

32-phase brushless exciter Stator outer diameter (mm) Rotor outer diameter (mm) Air-gap thickness (mm) Axial length (mm) No. of stator slots No. of rotor slots No. of poles No. of turns per rotor coil (p.u.) No. of turns per stator coil (p.u.) No. of exciter rotor phases Conductor diameter (mm) Conductor area (mm2) Copper area (mm2)

230.4 122 0.4 4 36 32 36 73 50 16 0.56 0.25 35.62

upcoming analysis, the 16-phase parallel option is the adopted brushless exciter rotor winding design, and this specification is illustrated in Table 6.7.

6.4.2

Performance Curves

The multiphase HPM generator system is designed to deliver a rated power of 3 kW at 3000 RPM, a detailed design of which is given in Chap. 4. The nominal voltage of the vehicle battery system interfaced to the DC-link, Fig. 5.19, in this analysis becomes 120 V at the open circuit but varies around 30% from its rated voltage, that is, from 85 to 125 V during load transients throughout the driving cycle as shown in Fig. 6.26a. The HPM generator is designed to interface this varying DC-link voltage by modulation of the machine WF element. As discussed, the

6.4 HPM Machine Characterization Using Brushless Excitor

201

Fig. 6.26 Control strategy of the multiphase HPM generator via brushless exciter (a) Vehicle DC-link voltage variation (b) Control block diagram schematic

HPM generator WF element facilitates the control of the total generated back-EMF and hence machine rectified output DC voltage. The HPM generator is required to maintain a constant power input to the vehicle DC-link during the entire driving cycle. Indeed, the control objective is to maintain the ICE/HPM generator system output power constant at a given value while minimizing speed variation via regulation of the HPM wound field excitation. Thus, a DC-link voltage increase beyond the nominal value requires a WF current increase at a given speed, a scenario of boosting capability. Similarly, in a buck functionality, the WF current is decreased to reduce the machine output voltage. A general schematic diagram of the HPM generator system along with its feedback control is illustrated in Fig. 6.26b. The brushless exciter stator is fed from the output of the HPM generator via a DC/DC converter. Here, a WF excitation current linear function that uses interpolationexportation is utilized as a lookup method to produce a duty cycle (D) for the brushless exciter DC/DC converter. By controlling the duty cycle, the rectified

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

Fig. 6.27 Brushless HPM generator performance curves at 3000 RPM for different output power

voltage at the output of the exciter rotating rectifier is adjusted to provide the desired WF current. However, in order to characterize the system under a prescribed control strategy, the HPM machine performance curves are deduced for a rated power range of 0.5–4.2 kW with an extended DC-link voltage variation, that is, beyond the actual voltage of 85–125 V. The HPM generator is capable of delivering 3.2 kW continuously; however, the 4.2 kW rated power is chosen as the upper limit due to the limitation on machine current density and thermal performance under a short period of time. Figure 6.27 presents the brushless HPM generator performance curves for different output power, DC-link voltage, and WF excitation current. Utilizing the previously described control strategy while maintaining a maximum stator current density (S.C.D.), a minimum DC-link voltage is derived for each power curve at a fixed speed of 3000 RPM. The dotted line in Fig. 6.27 denotes the performance curve limit due to the maximum stator current density. For an excitation WF current limit of 4 A, chosen to limit the teeth and back-iron saturation and the WF rotor slot current density, the 0.5-kW and 1-kW performance curves present a linear behavior as the DC-link voltage varies. However, for powers greater than 1 kW, the curves follow a nonlinear cubic function behavior. This is because at some WF excitation currents, there exist two DC-link voltages that result in the same output power. For instance, for an excitation current of 1 A, both 52 V and 104 V DC-link voltages result in the same rated power of 2.5 kW. As observed from Fig. 6.27, for rated output powers greater than 2.5 kW, the WF excitation current is always positive, hence simplifying requirements for the DC/DC power electronics converter, Fig. 6.25b, that otherwise would be necessary to reverse the WF excitation current. Furthermore, as the power increases, the minimum DC-link voltage limit is increased while the maximum DC-link voltage limit is reduced. For instance, the minimum DC-link voltage for a rated power of 2.5 kW is 52 V while this value for the 3 kW is 60 V. This is due to the effect of the machine synchronous inductance and hence power factor effect that is worsened as the power

6.4 HPM Machine Characterization Using Brushless Excitor

203

Table 6.8 Comparison of HPM machine system characteristics at 3000 RPM Item Region Underrated

NCR

PCR

P (kW) 0.5 1 1.5 2 2.5 3 3.2 3.5 4 4.2

If (A) 4 4 2.7 1.3 0 1.1 1.6 2.15 3.15 3.5

DC-link voltage Percentage (%) Base (V) Buck () Boost (+) 124 37 38 114 44 46 106 71 49 97.2 58 52 72.3 29 98 79.4 23 73 82 19 63 87 19 51 91 9 34 95 11 24

HPM WF back-EMF Percentage (%) Buck () Boost (+) 33.3 33.3 33.3 33.3 22 33.3 11 33.3 0 33.3 0 33.3 0 33.3 0 33.3 0 33.3 0 33.3

Note: Maximum excitation field current is 4 A.

Fig. 6.28 HPM WF excitation current and DC-link voltage at 3000 RPM for a range of HPM generator output power

is increased. Note that these curves are not to be confused with the V-curves of the conventional wound field synchronous machines. Table 6.8 lists the brushless HPM generator buck and boost capability at different WF current and machine output power. For each power level, a base DC-link voltage is considered where the WF current is zero. However, if the performance curve lies only in the first quadrant, the base DC-link voltage is defined at minimum WF current, that is, the cube base of the curve. It is seen from Table 6.8 that in all cases except for 4.2 kW, the range of DC-link voltage variations, that is, buck-boost capability, is larger than the backEMF variations. Therefore, for a rated power of 3 kW, the HPM generator’s maximum nine-phase back-EMF boost capability is 33.3% while the DC-link voltage can be 23% and 73% bucked and boosted, respectively.

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

Figure 6.28 shows base, minimum, and maximum DC-link voltage for a range of power, that is, 0.5–4.2 kW at 3000 RPM along with the required WF excitation current for each power. Three regions are recognized: (i) nominal continuous power rating (NCR) 2.5–3.2 kW, (ii) peak continuous power rating (PCR) 3.2–4.2 kW, and (iii) underrated power region 0.5–2.5 kW. As it is observed from Fig. 6.28, the highest DC-link boost capability of the HPM generator occurs in the NCR region. Indeed, for the 2.5 kW, 3 kW, and 3.2 kW, the DC-link boost is 98%, 73%, and 63%, respectively, as seen from Table 6.8. Therefore, if the HPM generator output power is maintained in the NCR region, the electric vehicle DC-link voltage is allowed to sustain the highest variation. In the NCR region, the field current is always positive, hence simplifying the requirements for the brushless exciter DC/DC converter, Fig. 6.26b. Hence, the 3-kW demand average power lies in the NCR region, a preferred control region due to the highest boost capability for the HPM machine. Therefore, the brushless HPM generator system is controlled to deliver 3 kW at a rated speed of 3000 RPM.

6.4.3

Efficiency Mapping

In this section, efficiency maps of the HPM generator system is investigated for a 1.0-ton SHEV with an average power demand of 3 kW under NEDC driving cycle and for a 85–25 V DC-link voltage variation. The operational envelopes are constrained by the WF excitation current, prime mover speed, and average DC-link power. Here, the HPM generator iron loss is predicted using the curve fitting method, as detailed in Chap. 3 (Sect. 3.5.3), which is expanded to include higher speed losses. The manufacture’s iron loss curves combine hysteresis, eddy current, and excess losses and are given in watts per kilogram, that is, loss density for different fundamental flux densities. Figure 6.29 compares the predicted and

Fig. 6.29 Predicted and measured iron loss versus flux density at different speeds

6.4 HPM Machine Characterization Using Brushless Excitor

205

Fig. 6.30 HPM generator torque-speed and efficiency mapping. Note that the efficiencies include the passive rectification stage losses (a) Efficiency map for different HPM generator speed at 3 kW (b) Torque-speed characteristics

measured iron loss density for a range of speeds (0.5–7.0 kRPM). The comparison shows a good agreement, hence the accuracy of the loss analysis. Figure 6.30a illustrates HPM generator efficiency (including the passive rectification stage) for different delivered power. As seen, higher positive WF currents result in improved system efficiency making the boosting functionality of the HPM generator system a favorite control strategy. Figure 6.30b presents the HPM generator torque-speed characteristics along with efficiencies at different machine speeds. As seen from Fig. 6.30b, for the speeds between 2000 and 3500 RPM, the HPM generator and rectification system results in the highest efficiency with an output power of 3 kW while satisfying WF current constraints. As previously discussed, the

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6 Operation and Characterization of Multiphase HPM Generator in SHEV Powertrain

ICE speed is maintained within a tight envelope that results in optimum engine efficiency. The HPM generator system operating speed region is therefore within the ICE speed envelope.

6.5

Conclusion

This chapter discusses the operation of the HPM machine in a series hybrid EV powertrain, where a dynamic has been developed to include the DC-link voltage variation during dynamic driving and the WF back-EMF regulation function via a flux-linkage feedback loop. The HPM machine back-EMF control strategy to supply a constant DC-link power demand has been discussed, along with a justification of the desired wound field excitation current regulation function for different machine stator winding turns. The analysis has highlighted the importance of choosing the most suitable number of stator winding turns for a given variation of the vehicle system terminal voltage. Moreover, five operating scenarios for the HPM machine in a series hybrid EV powertrain have been investigated. The five analyzed operating scenarios gave a decision regarding the most appropriate operating scenario that can be adopted for minimal HPM machine with regard to total energy and rotor copper loss (scenario 3) or system failure modes (scenario 1). For low DC-link voltage (80–125 V), characterization of the multiphase HPM generator system is also addressed utilizing a 32-phase brushless excitation scheme. Different cases, such as normal, boost, and buck functionality of HPM machine operation, are analyzed, and a choice of the most appropriate operation mode has been selected to regulate the total back-EMF via a WF excitation current control. Several system power performance curves have been generated, and the most suitable continuous machine power rating range is concluded. The selected NCR power region supports the proposed machine safe continuous operation, which is 70% of the PCR power region with 74% efficiency. The dynamic model simulation study illustrates the buck/boost percentage regulation capability of the HPM machine back-EMF and vehicle DC-link based on the DC excitation field voltage and current limits.

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Further Reading 242. O. Beik, N. Schofield, A Brushless Exciter Design for a Hybrid Permanent Magnet Generator Applied to Series Hybrid Electric Vehicles (7th IET International Conference on Power Electronics, Machines and Drives (PEMD 2014), Manchester, 2014) 243. O. Beik, A. Dekka, M. Narimani, A New Modular Neutral Point Clamped Converter with Space Vector Modulation Control (2018 IEEE International Conference on Industrial Technology (ICIT), Lyon, 2018), pp. 591–595 244. O. Beik, A. Al-Adsani, Wind Turbine Productivity and Wind Energy Assessment: An Ontario Case Study (2020 IEEE Electric Power and Energy Conference (EPEC), Edmonton, AB, Canada, 2020), pp. 1–5

References

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245. O. Beik, R. Yang, A. Emadi, Design of a 5-Phase IPM Machine for Electric Vehicles (2018 IEEE Transportation Electrification Conference and Expo (ITEC), Long Beach, CA, 2018), pp. 168–172 246. O. Beik, Design optimization of a PM motor: A practical approach for mass production. IEEE Transactions on Energy Conversion 35(4), 1849–1858 (2020) 247. D. Schumacher, O. Beik, A. Emadi, Standalone integrated power electronics system: Applications for off-grid rural locations. IEEE Electrification Magazine 6(4), 73–82 (2018) 248. O. Beik, A. Emadi, Toward integrated digital aircraft control systems: For electrified-system optimized size, cost, and efficiency. IEEE Electrification Magazine 5(4), 46–52 (2017) 249. N. Schofield, X. Niu, O. Beik, Multiphase Machines for Electric Vehicle Traction (2014 IEEE Transportation Electrification Conference and Expo (ITEC), Dearborn, MI, 2014), pp. 1–6 250. O. Beik, Operational envelope of an electric transmission auxiliary system. in IEEE Transactions on Energy Conversion 35(3), 1513–1521 (2020) 251. A. Dekka, O. Beik, M. Narimani, Modulation and voltage balancing of a five-level seriesconnected multilevel inverter with reduced isolated direct current sources. IEEE Trans. Ind. Electron. 67(10), 8219–8230 (2020) 252. M.R. Aghamohammadi, A. Beik-Khormizi, Small Signal Stability Constrained Rescheduling Using Sensitivities Analysis by Neural Network as a Preventive Tool (IEEE PES T&D 2010, New Orleans, LA, 2010), pp. 1–5 253. A.B. Khormizi, A.S. Nia, Damping of Power System Oscillations in Multi-Machine Power Systems Using Coordinate Design of PSS and TCSC (2011 10th International Conference on Environment and Electrical Engineering, Rome, 2011), pp. 1–4 254. A. Al-Adsani, N. Schofield, Comparison of Three and Nine-Phase Hybrid Permanent Magnet Generator Designs (The 35th Annual Conference of the IEEE Industrial Electronic Society (IECON 2009), Porto, Portugal. , 3–5 Nov. 2009, 2009) 255. A. Al-Adsani, A. Jarushi, N. Schofield, Operation of a Hybrid PM Generator in a Series Hybrid Electric Vehicle (The 35th Annual Conference of the IEEE Industrial Electronic Society (IECON 2009), Porto, Portugal, 2009), pp. 3898–3903. , 3–5 Nov. 2009 256. A. Al-Adsani, A.M. Jarushi, N. Schofield, An ICE/HPM Generator Range Extender in a Series Hybrid Electric Vehicle (5th IET International Conference on Power Electronics, Machines and Drives (PEMD 2010), Brighten, UK, 2010), pp. 1–6. , 19–21 April 2010 257. N. Schofield, A. Al-Adsani, Operation of a Hybrid PM Generator in a Series Hybrid EV Power-Train (The 7th IEEE Vehicular Power and Propulsion Conference (VPPC’11), Chicago, USA, 2011), pp. 1–6. , 6–9 Sept. 2011 258. N. Xin, N. Schofield, A. Al-Adsani, Comparison of 3- and 9-Phase Brushless PM Drive Systems for Aerospace Actuation (6th IET International Conference on Power Electronics, Machines and Drives (PEMD 2012), Bristol, UK, 2012), p. 13., , 27–29 March 2012 259. O. Beik, An HVDC off-Shore Wind Generation Scheme with High Voltage Hybrid Generator (Ph.D. dissertation, McMaster University, 2016) 260. N. Schofield, O. Beik, Wind Turbine High Voltage Generator and System (The 15th International Workshop on Large-Scale Integration of Wind Power into Power Systems, Vienna, Austria, 2016) 261. O. Beik, Electrified Landing Gear Adaptor (US Patent US 2020/0377062 A1, Dec 03, 2020, 2020) 262. M.R. Aghamohammadi, A. Beik Khormizi, M. Rezaee, Effect of Generator Parameters Inaccuracy on Transient Stability Performance (2010 Asia-Pacific Power and Energy Engineering Conference, Chengdu, 2010), pp. 1–5 263. A.B. Khormizi, M. Hajati, M.R. Aghamohammadi, An analytical method to parameter estimation by dynamic performance evaluation. International Journal of Energy and Power (IJEP) 2(1), 1–7 (2013)

Index

A AC L-L voltage, 100 Afjei design, 40 Alternating current (AC), 6 Ampere’s law, 13 AVL PEMS, 151 Axillary ICE/HPM generator unit, 168

B Back-EMF, 101–106 Battery management interface (BMI), 155 Battery technology, 146, 147, 170 Bipolar hybrid excitation synchronous machines, 35 Brushed DC and AC machines, 7–10 Brushless AC machines, 10, 11, 163 Brushless DC traction machine efficiency, 163 Brushless excitor, characterization using, 194, 197 efficiency mapping, 204–206 performance curves, 200–204 32-phase brushless excitation scheme, 197–200

C Carbon dioxide (CO2) emission data, 156 Classical electric machine topologies, 6 brushed DC and AC machines, 7–10 brushless AC machines, 10, 11 switch reluctance machines, 11, 12 Commercial HEVs, 148 Conduction heat transfer, 82–84 Conduction losses, 136

Consequent pole permanent magnet hybrid excitation machine (CPPM), 27, 29, 33 Convection heat transfer, 84–86 Conventional gasoline/diesel-fueled vehicles, 143

D DC excitation, 194, 206 DC-link voltage, 130, 132–134, 158, 159, 172 Diode conduction losses, 138 Direct current (DC) machines, 7–10 dq mathematical model, 115–118 Driving cycle profiles, 167 Dual-rotor machine, 30, 33 Dual-stator hybrid excited synchronous wind generator (DSHESG), 40, 43

E Efficiency mapping, 204–206 Electric machine fundamentals, 5, 6 Electric machine theory classical electric machine topologies, 6 brushed DC and AC machines, 7–10 brushless AC machines, 10, 11 switch reluctance machines, 11, 12 electric machine fundamentals, 5, 6 ferromagnetic materials and magnetization curve, 3–5 magnetic circuit principles, 1 magnetic field distribution and flux density, 1–3

© Springer Nature Switzerland AG 2022 A. S. Al-Adsani, O. Beik, Multiphase Hybrid Electric Machines, https://doi.org/10.1007/978-3-030-80435-0

221

222 Electric machine theory (cont.) WF and PM synchronous machine excitation fields, 13 magnetic flux path representation, 13–15 Electric vehicles (EVs), 172, 204 automotive market, 143 battery technology, 146, 147 companies, 143, 144 driving range, 143 high-voltage battery system, 143 powertrain configuration, 145 vehicle gross weight, 143 Electrified powertrain components, 166 Energy conversion, 149 Energy management algorithm, 167 Energy management and control strategies, 159 Engine-mounted multiphase HPM generator, 158 Equivalent magnetic circuit (EMC) model PM machine analysis via, 56–61 European Commission proposal, 170 EV models, 144 EVs powertrain configuration, 145 EVs range extender auxiliary power sources, 159 battery system, 160 control strategies, 160 energy sources, 160 hybrid thermostat strategy, 160 ICE coupled electric generator, 161 ICE/HPM generator, 159 off-board and onboard functions, 161 operational scenarios, 161 requirements, 161 simulation powertrain platform, 161 supervisor control optimization, 160 EVs vs. SHEV, 169 Excitation systems, 194

F Fast Fourier transform (FFT), 103 Ferromagnetic materials (FERMMs), 3–5 Field controlled torus-NS (FCT-NS) machine, 30, 33 Finite element method program, 52–55 Flux density, 1–3 Fuel cell range extender, 161

Index H Highway driving cycle (HWFET), 151, 152 Homopolar hybrid excitation synchronous machines, 35 HPM generator, 158 HPM generator excitation field control function, 167 HPM generator open-loop control system block diagram, 176 HPM generator ratings, 168 HPM machine, 153 HPM machine back-EMF control strategy, 173 control strategy analysis, 174–178 DC-link design options, 178, 179, 181 HPM machine final design model analysis, 75, 76 core loss prediction, 77–81 rotor PM demagnetization, 75–78 HPM machine output power control, 181–185 operating scenarios, 184, 186 solving final choice with full simulation model, 189–191, 193 thermal analysis, 191, 194–196 two driving cycles, energy loss prediction for, 187–189 HPM machine thermal model, 86, 88, 89 HPM machine wound field, 173 HPM machines, 18 HPM machines prototype construction of, 111–113 HPM topology, 154 HWFET driving cycle, 168 Hybrid electric machine concept classification, 17 history, 17 surveyed literature on, 40, 43, 44 topologies, 18 consequent pole permanent magnet hybrid excitation machine, 27, 29 dual-rotor machine, 30, 33 dual-stator hybrid excited synchronous wind generator, 40, 43 field controlled torus-NS machine, 30, 33 hybrid excitation synchronous machine, 23, 25 imbricated hybrid excitation machine, 32, 34, 36, 37 PM synchronous machine with claw pole field excitation, 18, 19, 21 series double excited synchronous machine, 36, 38, 39

Index switch reluctance machine with stator field assistance, 39, 40, 42 synchronous permanent magnet hybrid AC machine, 25, 26, 28, 29 toroidal-stator transverse-flux machine, 21–25 Hybrid electric vehicles (HEVs) battery charging time, 146 commercial, 148 fuel consumption/fuel economy, 148 models, 148 plug-in HEVs, 146 powertrain components, 148, 150 topology, 152 vehicle range issue, 143, 146 Hybrid EV powertrain configurations, 151 Hybrid excitation field regulation topologies, 17 Hybrid excitation synchronous generator (HESG), 19 Hybrid excitation synchronous machine (HESM), 23, 25 Hybrid permanent magnet machine design, 45 HPM machine final design model analysis, 75, 76 core loss prediction, 77–81 rotor PM demagnetization, 75–78 HPM machine parameters synchronous inductance and winding resistance, 73–75 torque prediction and saturation, 69, 71, 72 machine thermal model conduction heat transfer, 82–84 convection heat transfer, 84–86 HPM machine thermal model, 86, 88, 90 lumped parameter method, principle of, 81, 82 radiation heat transfer, 86 machine volume envelope consideration, 46 finite element method program, 52–55 machine back-EMF prediction, 55–57 PM machine analysis via EMC model, 56–61 PM machine dimensions, 47, 48 PM machine stator winding layout, 47–50 stator winding fill factor and resistance, 50, 51 PM and four HPM machine topologies, 90–93 WF machine, 61 rotor design, 62–64

223 WF rotor designs, comparative analysis of, 65–70 WF to PM split ratio, 64, 65 Hybrid PM (HPM), 153 Hybrid terminology, 153 Hybrid thermostat strategy, 160 Hybridization ratio (HR) calculation, 166 constant power source, 166 dynamic power, 165 ICE/HPM generator, 166 minimum battery power, 166 power share measurement, 165 ZEBRA battery, 165

I ICE energy source, 149 ICE/HPM generator fixed power output, 166 ICE/HPM generator range extender, SHEVs algorithm, 166 benchmark system, 162 case studies, 169 HWFET and NEDC cycles, 169 hybridization ratio, 165 ICE, 162 sizing, 166 structure, 162 vehicle traction machine torque, 163 ICE/HPM generator system output power, 169 ICE/PM machine, 153 Imbricated hybrid excitation machine (IHEM), 32, 34, 36, 37 Induction machine (IMs), 10 Interior PM (IPM) generator, 172 Internal combustion engine (ICE), 143 actual test data, 156 driving cycles emissions, 157 efficiency, 156, 157 FCR, 158 operating point, 156 SHEV fuel consumption, 157

L Lead-acid battery technology, 146 Lithium-ion (Li-ion), 146 Loss audit multiphase HPM generator systems, 134, 135

224 Loss audit (cont.) core loss prediction, 136, 137 passive and active converter loss, 136, 138–141 Lumped parameter method, 59, 65, 81, 82

M Machine back-EMF prediction, 55–57 Magnetic circuit principles, 1 magnetic field distribution and flux density, 1–3 Magnetic field distribution, 1–3 Magnetization curves, 3–5 Matlab/Simulink SimPower model, 103 Motor-CAD, 85, 86 Motor control, 163 Multiphase HPM generator, 171 brushless excitor, characterization using, 194, 197 32-phase brushless excitation scheme, 197–200 efficiency mapping, 204, 205 performance curves, 200–204 HPM machine back-EMF control strategy, 173 control strategy analysis, 174–178 DC-link design options, 178, 179, 181 HPM machine output power control, 181–185 operating scenarios, 184, 186 solving final choice with full simulation model, 189–191, 193 thermal analysis, 191, 194–196 two driving cycles, energy loss prediction for, 187–189 HPM machine wound field, 173 hybrid EV powertrain, 172 power flow management scheme, 172 vehicle DC-link voltage, 173 Multiphase HPM generator systems, 95, 96 analysis models general dq mathematical model, 115–118 simulation model, 118, 119 loss audit, 134, 135 core loss prediction, 136, 137 passive and active converter loss, 136, 138–141 multiphase windings principles, 96, 97 nine-phase HPM generator parameters, 101 back-EMF and torque waveform harmonics prediction, 103–106

Index HPM machines prototype, construction of, 111–113 nine-phase winding layout and backEMF, 101–103 resistance and inductance measurements, 114, 115 synchronous inductance prediction, 107–111 rectified voltage, 96, 98–100 three- and nine-phase HPM generator system studies, 119–121 DC-link voltage quality, 130, 133, 134 synchronous inductance and rectifier, impact on, 121–126 system sensitivity, 126–128, 130, 131 Multiphase windings principles, 96, 97

N NEDC driving cycle, 158 NEDC suburban driving cycle, 167, 170 Neodymium–iron–boron (NdFeB), 75 New European driving cycle (NEDC), 151, 156 Nickel–metal hydride (Ni-MH), 146 Nine-phase HPM generator parameters, 101 back-EMF and torque waveform harmonics prediction, 103–106 HPM machines prototype, construction of, 111–113 nine-phase winding layout and back-EMF, 101–103 resistance and inductance measurements, 114, 115 synchronous inductance prediction, 107–111 Nine-phase HPM generator prototype, 162 Nine-phase HPM generator system studies, 119–121 DC-link voltage quality, 130, 132–134 synchronous inductance and rectifier, impact on, 121–126 system sensitivity, 126–131 Nine-phase HPM generator systems, 96, 98–100 Nine-phase stator winding, 118 Nine-phase winding layout, 101–103

O Off-board ICE-generator system, 161 Off-the-shelf machines, 95 Original equipment manufacturers (OEM), 152 Output power ratio curves, 127

Index

225

P Parallel HEV format, 152 Parallel hybrid electric vehicle format, 148 Passive and active rectification stage losses, 141 Performance curves, 200–204 Permanent magnet machine dimensions, 47, 48 Plug-in HEV powertrain components, 146, 149 PM and four HPM machine topologies, 90–93 PM machine analysis via EMC model, 56–61 PM machine stator winding layout, 47–50 PM synchronous machine with claw pole field excitation (PSCPF), 18, 19, 21 PM synchronous machines magnetic flux path representation of, 14, 15 Power flow management scheme, 172 Power losses, 136 Powertrain hybridization rations, 168

Simulation model multiphase HPM generator systems, 118, 119 Squirrel-cage machines, 11 Standard drive cycles, 151 Starter/alternator, 151 State-of-charge (SOC), 155 Stator winding fill factor and resistance, 50, 51 Steady power generation mode, 151 Suburban driving cycle, 169 Switch reluctance machine with stator field assistance, 39, 40, 42 Switch reluctance machines (SRMs), 11, 12 Synchronous inductance and winding resistance, 73–75 Synchronous inductance prediction, 107–111 Synchronous permanent magnet hybrid AC machine (SynPM), 25, 26, 28, 29

R Radiation heat transfer, 86 Rectified voltage, 96, 98–100 Reverse recovery loss, 138 Revolutions per minute (RPM), 9 Rotor PM demagnetization, 75–78

T 32-phase brushless excitation scheme, 197, 198, 200 Thermal model, HPM conduction heat transfer, 82–84 convection heat transfer, 84–86 HPM machine thermal model, 86, 88, 90 lumped parameter method, principle of, 81, 82 radiation heat transfer, 86 Three-phase HPM generator system studies, 119–121 DC-link voltage quality, 130, 133, 134 synchronous inductance and rectifier, impact on, 121–126 system sensitivity, 126–128, 130, 131 Three-phase HPM generator system studies synchronous inductance and rectifier, impact on, 124 Three-phase HPM generator systems, 96, 98–100 Three-phase HPM machine, 48, 73, 75 Three-phase HPM machine terminal AC voltage, 154 Three-phase stator winding, 118 TM power electronic converter, 154 Toroidal-stator transverse-flux machine (TSTFM), 21–25 Torque prediction and saturation, 69, 71, 72 Torque waveform harmonics prediction, 103–106 Total transmission gear ratio, 164 Traction machine (TM), 154

S Sensorless control, 163 Series double excited synchronous machine (SDESM), 36, 38, 39 Series HEV powertrain schematics, 154 Series hybrid electric vehicle (SHEV) battery pack, 153 excitation control loop, 154 HPM generator, 158 HPM machine, 153 ICEs, 153, 156 PM output voltage, 153 ZEBRA battery (see ZEBRA battery technology) Series hybrid electric vehicle powertrain, 162 SHEV DC-link voltage, 158 SHEV fuel consumption, 160 SHEV powertrain configuration, 160 SHEV powertrain topology, 159 Simplified hybrid EV powertrain configurations, 148, 150 Simplified powertrain configuration, 145 SimPower, 141 SimPower model, 130, 134

226 Turn-off losses, 136, 138 Two driving cycles energy loss prediction for, 187, 188, 190

U US and European standard driving cycles, 152

V Vacuum pressure impregnation (VPI), 113 Vehicle DC-link voltage, 173 Vehicle driving cycle AVL PEMS, 151 fuel consumption, 151 NEDC, 151 Vehicle linear velocity, 165 Vehicle traction machine torque angular velocity, 165 driving cycle, 165 electric machine topologies, 163 equation, 163, 164 forces, 163, 164 motor control, 163 Newton’s second law, 163 output torque, 164 three-phase brushless DC motor, 163 vehicle linear velocity, 165 Voltage source converter (VSC), 163

Index W WF machine, 61, 62, 69, 72, 82 rotor design, 62–64 WF rotor designs, comparative analysis of, 65–70 WF to PM split ratio, 64, 65 WF rotor designs comparative analysis of, 65–69 WF synchronous machine excitation fields, 13 magnetic flux path representation, 13, 14 WF synchronous machines magnetic flux path representation of, 13, 14

Z ZEBRA battery technology, 182 BMI, 155 characteristics, 156 commercial, 162 comparable specific energy, 155 Na-Ni-Cl battery technology, 146 operation regions, 155 parameters, 155 SOC, 155, 156 specific energy, 146 specifications, 155 tests, 155