Electromagnetic Compatibility of Electric Vehicle [1st ed. 2021.] 9789813361652, 9813361654

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Key Technologies on New Energy Vehicles

Li Zhai

Electromagnetic Compatibility of Electric Vehicle

Key Technologies on New Energy Vehicles

Key Technologies on New Energy Vehicles publishes the latest developments in new energy vehicles - quickly, informally and with high quality. The intent is to cover all the main branches of new energy vehicles, both theoretical and applied, including but not limited: • • • • • • • • • • •

Control Technology of Driving System Hybrid Electric Vehicle Coupling Technology Cross Disciplinary design optimization technology Single and Group Battery Technology Energy Management Technology Lightweight Technology New Energy Materials and Device Internet of Things (IoT) Cloud Computing 3D Printing Virtual Reality Technologies

Within the scopes of the series are monographs, professional books or graduate textbooks, edited volumes, and reference works purposely devoted to support education in related areas at the graduate and post-graduate levels. To submit a proposal or request further information, please contact: Dr. Mengchu Huang, Senior Editor, Applied Sciences Email: [email protected] Tel: +86-21-2422 5094

More information about this series at http://www.springer.com/series/16377

Li Zhai

Electromagnetic Compatibility of Electric Vehicle

Li Zhai National Engineering Laboratory for Electric Vehicles Beijing Institute of Technology Beijing, China

ISSN 2662-2920 ISSN 2662-2939 (electronic) Key Technologies on New Energy Vehicles ISBN 978-981-33-6164-5 ISBN 978-981-33-6165-2 (eBook) https://doi.org/10.1007/978-981-33-6165-2 Jointly published with China Machine Press, China The print edition is not for sale in China Mainland. Customers from China Mainland please order the print book from: China Machine Press. © China Machine Press 2021 This work is subject to copyright. All rights are reserved by the Publishers, whether the whole or part of the material is concerned, specifically the rights of translation, reprinting, reuse of illustrations, recitation, broadcasting, reproduction on microfilms or in any other physical way, and transmission or information storage and retrieval, electronic adaptation, computer software, or by similar or dissimilar methodology now known or hereafter developed. The use of general descriptive names, registered names, trademarks, service marks, etc. in this publication does not imply, even in the absence of a specific statement, that such names are exempt from the relevant protective laws and regulations and therefore free for general use. The publishers, the authors, and the editors are safe to assume that the advice and information in this book are believed to be true and accurate at the date of publication. Neither the publishers nor the authors or the editors give a warranty, express or implied, with respect to the material contained herein or for any errors or omissions that may have been made. The publishers remain neutral with regard to jurisdictional claims in published maps and institutional affiliations. This Springer imprint is published by the registered company Springer Nature Singapore Pte Ltd. The registered company address is: 152 Beach Road, #21-01/04 Gateway East, Singapore 189721, Singapore

Preface

Electromagnetic compatibility (EMC) is a key technology of electric vehicles (EV), which is very important to ensure the safe driving of new energy vehicles and reduce or avoid vehicle failures. With the development of EV electrification, intellectualization, networking, and sharing, the requirements of high bandwidth, high real time, high security, and high reliability are put forward for high-efficient wire-controlled electronic and electrical architecture. EMC has become an important index affecting the electromagnetic safety and functional safety of intelligent networked EV. This book covers the EMC of EV parts and vehicles, and there are eight chapters in this book which include the EMC of new energy vehicles, electromagnetic interference (EMI) prediction and suppression of motor drive systems, EMI prediction and suppression of DC-DC converters, electromagnetic safety and EMC of wireless charging systems, signal integrity and EMC of vehicle controller, EMC of battery management system, vehicle electromagnetic radiation diagnosis and suppression, etc. This book can provide readers a general understanding of the EMC of new energy vehicles. Due to space limitations, although it is impossible to carry out in-depth analysis of each part of the content, EMC analysis methods, modeling simulation methods, testing methods, and rectification methods are given. This book can be used as a reference for graduate students, senior undergraduates, and engineering/technical personnel in vehicle-engineering-related majors. In addition, this book acts as a reference and support for engineers to solve EMC problems for the EMI prediction, suppression, and EMC optimization design of new energy vehicles. The author has been engaged in the EMC research of EV since 2006, and has presided over or participated in the National Natural Science Foundation of China, National Key Research and Development Projects, National Defense Key Pre-research Projects, Equipment Pre-research Field Fund Projects, Shared Technology Projects, Enterprise Cooperation Projects, etc. The publication of this book was supported by these scientific research project funds. In the process of writing this book, the graduate students of author’s team have made significant contributions, part of which comes from their research results. The author would like to thank the postgraduates who participated in the writing and collation work: Hu Guixing, Lv Mengyuan, Gao Runze, Dong Mingcheng, Zhang Xinyu, Li Guangzhao, Feng Huiyuan, Song Chao, Lin Liwen, Yang Sipeng, Cao v

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Yu, Zhang Tao, Qin Huanyao, Zhong Guangyuan, Li Xiang, Mok Yuh Ming, Hou Yu Han, Zhang Xueying, etc. The author would like to thank colleagues from the National Engineering Laboratory of Electric Vehicles of Beijing Institute of Technology for their help and valuable comments and suggestions. Author would also like to thank Dr. Gao Xinjie of Beijing New Energy Automobile Co., Ltd.; Dr. Fan Sen of Zhengzhou Yutong Bus Co., Ltd.; and Dr. Liu Haiming of China Automotive Technology Research Center Co., Ltd. for their valuable suggestions, and thanks all domestic and foreign experts and peers in the field of vehicle EMC for their selfless guidance and help. Due to the author’s limited academic level and writing experience, omissions and errors are inevitable, and feedbacks for corrections are welcome. Beijing, China August 2020

Li Zhai

Contents

1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.1 Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.2 EMC Problems of EVs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.2.1 EMC Problems of Motor Drive System . . . . . . . . . . . . . . . . 1.2.2 EMC Problems of DC-DC Converter System . . . . . . . . . . . 1.2.3 EMC Problems of Wireless Charging System . . . . . . . . . . . 1.2.4 EMC Problem of Vehicle Controller . . . . . . . . . . . . . . . . . . . 1.2.5 EMC Problems of Battery Management System (BMS) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.2.6 Vehicle EMC Requirements . . . . . . . . . . . . . . . . . . . . . . . . . . 1.3 Content Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2 Electromagnetic Compatibility Foundation of New Energy Vehicles . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.1 Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.2 EMC of New Energy Vehicle . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.2.1 Electrical Infrastructure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.2.2 Electromagnetic Compatibility . . . . . . . . . . . . . . . . . . . . . . . . 2.3 Theoretical Basis of Electromagnetic Compatibility . . . . . . . . . . . . . 2.3.1 Electromagnetic Interference Source . . . . . . . . . . . . . . . . . . . 2.3.2 Coupling Path . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.4 EMC Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.4.1 High Performance Electrical and Electronic Architecture Modeling . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.4.2 EMC Modeling and EMC Design . . . . . . . . . . . . . . . . . . . . . 2.5 EMI Suppression Measures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2.5.1 EMI Suppression Technology . . . . . . . . . . . . . . . . . . . . . . . . . 2.5.2 EMI Suppression of Key Components . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

1 1 3 3 10 13 18 23 26 27 28 35 35 37 37 38 47 47 49 53 54 55 58 58 62 63

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3 EMI Prediction and Suppression of Motor Drive System . . . . . . . . . . . 3.1 Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.2 EMI Mechanism of Motor Drive System . . . . . . . . . . . . . . . . . . . . . . . 3.2.1 Conducted Emission Test of Motor Drive System . . . . . . . . 3.2.2 IGBT EMI Source . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.2.3 EMI Coupling Path . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.3 EMI Modeling of Motor Drive System . . . . . . . . . . . . . . . . . . . . . . . . 3.3.1 Modeling and Simulation of EMI in Motor Drive System . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.3.2 Co-simulation of Motor Drive System . . . . . . . . . . . . . . . . . 3.4 Spice Equivalent Circuit Modeling of Electromagnetic Emission of Electric Drive System Based on Measurement . . . . . . . 3.4.1 Overview of Modeling Methods . . . . . . . . . . . . . . . . . . . . . . . 3.4.2 Construction of the Equivalent Circuit . . . . . . . . . . . . . . . . . 3.4.3 Validation of the Complete Model . . . . . . . . . . . . . . . . . . . . . 3.4.4 System Resonance Suppression Method . . . . . . . . . . . . . . . . 3.4.5 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.5 EMI Power Filter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.5.1 Design of High-Voltage Input Port Filter . . . . . . . . . . . . . . . 3.5.2 Design of Filter Circuit Based on Resonance Suppression . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.5.3 Design of EMI Filter for HV DC Power Supply of Motor Controller . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.5.4 Design of Air-Core Inductor EMI Filter for High-Voltage DC Power Supply . . . . . . . . . . . . . . . . . . . . 3.6 Electromagnetic Disturbance Test . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.6.1 Electromagnetic Disturbance Test Platform . . . . . . . . . . . . . 3.6.2 Conducted Emission-Voltage Method . . . . . . . . . . . . . . . . . . 3.6.3 Conducted Emission-Current Probe Method . . . . . . . . . . . . 3.6.4 Coupling of LV Power Lines to HV Power Lines . . . . . . . . 3.6.5 Conducted EMI Suppression . . . . . . . . . . . . . . . . . . . . . . . . . 3.6.6 Voltage and Current of Motor Drive System DC and AC Power Lines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3.6.7 The Influence of Speed and Torque on Conducted Disturbance Voltage and Current . . . . . . . . . . . . . . . . . . . . . . 3.6.8 Electromagnetic Radiation Test . . . . . . . . . . . . . . . . . . . . . . . 3.6.9 Conducted Disturbance Voltage DM and CM Separation Test . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 EMI Prediction and Suppression of DC-DC Converter . . . . . . . . . . . . . 4.1 Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.2 Conducted Emission Test of DC-DC Converter . . . . . . . . . . . . . . . . . 4.2.1 Isolated Full-Bridge HV/LV Voltage DC-DC Converter Structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

65 65 66 66 69 72 80 80 111 115 115 116 121 121 137 138 139 147 155 163 172 172 173 175 179 180 184 185 188 191 191 193 193 194 194

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4.2.2 Test Setup for the Conducted EMI . . . . . . . . . . . . . . . . . . . . . 4.3 EMI Prediction of DC-DC Converter . . . . . . . . . . . . . . . . . . . . . . . . . . 4.3.1 EMI Source . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.3.2 The Conducted Emission High-Frequency Equivalent Circuit of DC-DC Converter System . . . . . . . . . 4.3.3 EMI Coupling Path . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.3.4 Conducted EMI Prediction Through Simulation . . . . . . . . . 4.4 EMI Filter of High Voltage Power Supply . . . . . . . . . . . . . . . . . . . . . . 4.4.1 Design of High Voltage Input Port EMI Filter . . . . . . . . . . . 4.4.2 Filter Circuit Design Based on Resonance Point Suppression . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.5 Radiated Emission . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.5.1 Theoretical Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4.5.2 Radiated Emission Measurement . . . . . . . . . . . . . . . . . . . . . . 4.6 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 Wireless Charging System Electromagnetic Safety and Electromagnetic Compatibility . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.1 Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.2 Wireless Charging System Structure . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.2.1 System Structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.2.2 System Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.2.3 System Performance Simulation . . . . . . . . . . . . . . . . . . . . . . 5.3 Principle of the Wireless Charging . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.3.1 Transmission Power . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.3.2 Efficiency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.3.3 Offset Characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.3.4 Effect of Offset on Efficiency . . . . . . . . . . . . . . . . . . . . . . . . . 5.4 Magnetic Field Distribution . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.4.1 Vehicle Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.4.2 Distribution of Magnetic Field Test Points . . . . . . . . . . . . . . 5.4.3 Coupling Coil Mutual Inductance at Offset . . . . . . . . . . . . . 5.4.4 Magnetic Field Distribution . . . . . . . . . . . . . . . . . . . . . . . . . . 5.4.5 Magnetic Induction Strength Measurement . . . . . . . . . . . . . 5.5 Modeling and Suppression of Power Line Conducted Electromagnetic Interference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.5.1 Interference Source . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.5.2 Electromagnetic Interference Mechanism . . . . . . . . . . . . . . . 5.5.3 Conducted Interference Prediction . . . . . . . . . . . . . . . . . . . . . 5.5.4 EMI Filter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.5.5 Conducted Disturbance Voltage Measurement . . . . . . . . . . . 5.5.6 Harmonics of Public Power Supply Lines . . . . . . . . . . . . . . 5.5.7 Harmonic Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

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5.5.8

Harmonics of Traditional Rectifier Filter Booster Circuit System . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.5.9 Harmonics Using PFC Circuit System . . . . . . . . . . . . . . . . . 5.5.10 System Harmonic Suppression Method . . . . . . . . . . . . . . . . 5.6 Electromagnetic Radiation of Secondary Side Circuit of Wireless Charging System . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5.6.1 Radiation Emission System Modeling . . . . . . . . . . . . . . . . . . 5.6.2 Radiated Emission from Secondary Charging Cable . . . . . . 5.6.3 Radiated EMI Suppression . . . . . . . . . . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 Signal Integrity and Electromagnetic Compatibility of Vehicle Controller . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.1 Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.2 Function and Structure of Vehicle Controller . . . . . . . . . . . . . . . . . . . 6.3 Hardware EMC Design of Vehicle Controller . . . . . . . . . . . . . . . . . . . 6.3.1 Power Line Electromagnetic Immunity . . . . . . . . . . . . . . . . . 6.3.2 PCB Power Integrity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.3.3 Signal Integrity Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.4 Power Distribution Network Equivalent Circuit . . . . . . . . . . . . . . . . . 6.4.1 Equivalent Circuit of Chip Power Distribution Network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.4.2 Lumped Parameter Model of Power Distribution Network for Ethernet Chip . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.5 PCB Decoupling Capacitor Optimization Method . . . . . . . . . . . . . . . 6.5.1 The Impedance of the PCB Power Distribution Network of the Vehicle Controller . . . . . . . . . . . . . . . . . . . . . 6.5.2 PDN Decoupling Capacitor Optimization . . . . . . . . . . . . . . . 6.5.3 Impedance Verification in Frequency Domain . . . . . . . . . . . 6.6 Modeling and Analysis of PCB Board Signal Integrity . . . . . . . . . . . 6.6.1 Vehicular Ethernet Signal and CAN Bus Signal Spectrum . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6.6.2 Reflection Modeling and Simulation . . . . . . . . . . . . . . . . . . . 6.6.3 Crosstalk Modeling and Simulation . . . . . . . . . . . . . . . . . . . . 6.7 PCB Plate Electromagnetic Radiation . . . . . . . . . . . . . . . . . . . . . . . . . 6.7.1 Electromagnetic Radiation Mechanism . . . . . . . . . . . . . . . . . 6.7.2 PCB Electromagnetic Radiation . . . . . . . . . . . . . . . . . . . . . . . 6.8 Conducted EMI Suppression of Power Line of Ethernet Vehicle Controller . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 Electromagnetic Compatibility of Battery Management System . . . . . 7.1 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7.2 BMS System Function and Structure . . . . . . . . . . . . . . . . . . . . . . . . . . 7.3 PCB Decoupling Capacitor Power Distribution Network . . . . . . . . . 7.3.1 Optimal Design Method of Decoupling Capacitor . . . . . . . 7.3.2 Decoupling Capacitor Placement . . . . . . . . . . . . . . . . . . . . . .

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7.3.3 Analysis of Power Plane Resonance . . . . . . . . . . . . . . . . . . . 7.4 Clock Signal . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7.4.1 Electromagnetic Interference Mechanism . . . . . . . . . . . . . . . 7.4.2 Near Field Scanning Prediction . . . . . . . . . . . . . . . . . . . . . . . 7.4.3 Clock Interference Signal Suppression . . . . . . . . . . . . . . . . . 7.4.4 BMS System Electromagnetic Radiation Emission Test . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7.5 DC-DC Chip Switching Noise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7.5.1 Switching Noise Mechanism . . . . . . . . . . . . . . . . . . . . . . . . . 7.5.2 Electromagnetic Interference Model Prediction . . . . . . . . . . 7.5.3 DC-DC Interference Suppression Method . . . . . . . . . . . . . . 7.6 Anti-interference Analysis of CAN Bus . . . . . . . . . . . . . . . . . . . . . . . . 7.6.1 Electromagnetic Interference of CAN Bus Circuit . . . . . . . 7.6.2 Design of Anti-interference Circuit of CAN Bus . . . . . . . . . 7.7 PCB Electromagnetic Radiation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 Measurement, Diagnosis and Suppression of Vehicle Electromagnetic Radiation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8.1 Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8.2 Electromagnetic Radiation of Passenger Car . . . . . . . . . . . . . . . . . . . . 8.2.1 Vehicle Radiation Emission Test . . . . . . . . . . . . . . . . . . . . . . 8.2.2 Vehicle Radiation Emission Suppression . . . . . . . . . . . . . . . 8.3 Electromagnetic Radiation of Commercial Vehicles . . . . . . . . . . . . . 8.3.1 Vehicle Electromagnetic Radiation Test . . . . . . . . . . . . . . . . 8.3.2 Diagnosis of Electromagnetic Disturbance Source . . . . . . . 8.3.3 Vehicle Radiated Electromagnetic Interference Suppression . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8.4 EMI Diagnosis Method of Whole Vehicle . . . . . . . . . . . . . . . . . . . . . . 8.5 EMI Measurement on Real Vehicle . . . . . . . . . . . . . . . . . . . . . . . . . . . 8.5.1 Frequency Domain Characteristics of EMI . . . . . . . . . . . . . . 8.5.2 Time Domain Characteristics of EMI . . . . . . . . . . . . . . . . . . References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

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Chapter 1

Introduction

1.1 Overview Electromagnetic compatibility (EMC) is a common technology of electric vehicle (EV), which is very important to ensure the safety of EV and reduce or avoid faults. The increasingly strict standards and regulations of EMC for EV and the fierce competition in the product market have put forward urgent requirements for EMC of EV. In recent years, EMC problems caused by EVs have gradually attracted people’s attention. International well-known automobile enterprises (Toyota, Ford, GM, BMW, etc.) have listed EMC technology as the key technology of EVs. To solve the problem of EMC and break through the key technology of EMC has become the key to successfully enter the market of EVs. At present, the research work on EMC of electric vehicles mainly focuses on the following aspects: • Aiming at the electromagnetic interference (EMI) problems in the actual operation of vehicles, the rectification technologies such as filtering, shielding and grounding of parts and systems of EVs are studied. • According to the standards and regulations, the electromagnetic radiation emission test of the whole vehicle is carried out, and the rectification and control are carried out to solve the problem of electromagnetic radiation exceeding the standard caused by the operation of electric drive system. • According to the standard regulations, the conducted and radiated emission tests of high voltage components of EV is carried out, and aiming at the problem of electromagnetic emission exceeding the standard, EMI diagnosis and rectification are carried out. • EMI can be predicted and suppressed by using EMC theory analysis and modeling simulation. At present, scholars and technicians have made some achievements in the EMC test and rectification of electric vehicles. However, due to the lack of accurate and © China Machine Press 2021 L. Zhai, Electromagnetic Compatibility of Electric Vehicle, Key Technologies on New Energy Vehicles, https://doi.org/10.1007/978-981-33-6165-2_1

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1 Introduction

effective simulation model and prediction method, the research on electromagnetic interference mechanism and mathematical characterization is insufficient, which cannot effectively analyze, predict and suppress electromagnetic interference in the product design stage. The prominent problems are as follows: (1) There are many types of interference sources, including narrow-band interference sources (such as clock, crystal oscillator, microprocessor and digital logic integrated circuit in vehicle electronic control unit) and broadband interference sources (such as motor and ignition system). Specific examples include IGBT power module of motor controller, MOSFET power module of DC-DC converter, clock circuit of vehicle controller unit, etc. The time-domain and frequency-domain characteristics of interference source signals are different. All kinds of interference source signals act on the vehicle high-voltage system and low-voltage system at the same time. At present, the linear ideal interference source is usually used in the modeling of interference source, which is quite different from the actual interference source. (2) Electromagnetic interference coupling path is complex. High voltage and low voltage components are arranged in the limited space of the vehicle, and the location and length of the components and cables will affect the conduction and radiation coupling path. The lumped circuit model is often used to model the EMI propagation path, which ignores or simplifies the influence of parasitic parameters and distributed parameters on EMI, which makes the EMI path analysis incomplete or incorrect. Therefore, the EMI noise signal cannot be accurately and effectively characterized, predicted and suppressed. (3) Sensitive equipment is diversified, such as intelligent sensors such as radar and camera, safety electronic controllers such as ABS and ESP, vehicle controller unit (VCU), battery management system (BMS), various radio receiving equipment, etc. During the actual operation of the vehicle, the high-voltage power system such as motor drive system will produce EMI to sensitive equipment such as intelligent sensors, electronic controllers and actuators through highvoltage cables, on-board CAN bus network and space. At the same time, intelligent sensors and vehicle wireless communication equipment will also produce radiation disturbance. (4) Dynamic changes of vehicle load conditions. There are many operating conditions of electric vehicles, such as starting, accelerating, constant speed, over speed, idling, braking, etc., and its load conditions change dynamically. The electromagnetic disturbance characteristics measured in laboratory cannot reflect the electromagnetic emission of real vehicle. (5) Functional security is ignored. EMC is analyzed and studied only according to the standards and regulations, and the influence of low frequency and radio frequency EMI generated by key systems such as electric drive system, intelligent sensor and vehicle wireless communication equipment on the functional safety of traction, braking and steering system is not fully considered. Therefore, the research on the mechanism, prediction and suppression methods of EMI from EVs is of great significance to improve the electromagnetic compatibility,

1.1 Overview

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reliability and safety of vehicles, as well as the design, manufacture and application of new energy vehicles.

1.2 EMC Problems of EVs Unlike conventional engine vehicles, there are a large number of high-voltage components in EV (Masih et al. 2019), such as: drive motor, DC-AC inverter, DC-DC converter, and on-board charger (AC-DC Converter), high voltage power battery, etc. In addition, there are also many low-voltage electrical components such as battery management systems (BMS), vehicle control units (VCU), and Telematics BOX (TBOX) (Zhai et al. 2017). Therefore, the electromagnetic environment of EV has become more complicated, and EMC has become more and more important. In order to protect the on-board and off-board receivers from radio interference of EV, international standards SAE J551-5, CISPR12 and CISPR25-2016 MOD set the limits for electromagnetic emission from EVs and on-board voltage components (National Radio Interference Standardization Technical Committee 2018). EMC has become one of the key technologies for EVs. CISPR25-2016 MOD specifies the limits and methods of measurement for conducted emission in 150 kHz–108 MHz and radiated emission in 150 kHz–2.5 GHz band for high and low voltage components of EVs. According to the standard ISO7637-4 《Road Vehicles—Electrical disturbance by conduction and coupling—Part 4: Electrical transient conduction along shielded high voltage supply lines only》 , The electrical transient conducted emission and immunity of high voltage components of electric passenger vehicles and commercial vehicles along the shielded high voltage power line are tested and evaluated.

1.2.1 EMC Problems of Motor Drive System Motor drive system is a key component of EV, which is supplied by DC power from 100 V to 800 V and typically uses power semiconductor devices (such as IGBTs) to implement pulse width modulation (PWM) control to regulate the three-phase AC voltage of the motor. The switching frequency of IGBT of the motor inverter is several kHz to several tens of kHz. The fast switching of the power semiconductor device produces a high current change rate di/dt and a voltage change rate of du/dt, forming EMI source signal, which propagates outward through the parasitic parameters of the internal components of the motor controller and the external high and low voltage lines, resulting in the unwanted conducted and radiated emissions. These emissions not only interfere with the on-board and off-board receivers, but also interferes with the high-voltage (HV) and low-voltage (LV) components of the vehicle through the conduction coupling path, and even affect the safety of the whole vehicle. In particular, the conducted EMI generated by the PMSM drive system not only induces

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1 Introduction

the radiated emission of its own system to exceed the limits defined by standards, but also causes the radiation emission of the whole vehicle to fail to meet the EMC requirements. There are three main methods to suppress the EMI of the motor drive system: PWM control strategy optimization, system structure optimization, installation of EMI filter (Baidy et al. 2013; Bishnoi et al. 2013). In previous research, optimization control strategies such as RSPWM and NSPWM are proposed to reduce common-mode (CM) interference of the motor inverter system (Liu and Liu 2014; Gao 2014). However, this method has some limitations in reducing both CM EMI and DM EMI effectively. There are also some studies on suppressing CM EMI by optimizing the inverter topology and motor’s stator winding structure (Cai et al. 2018; Zhu et al. 2001). However, these method needs to redesign the system, which makes the design more complex and more expensive. Installation of filters is an effective way to suppress conducted EMI, including active filters, passive filters and hybrid filters. Active filters are a hot topic of EMI suppression (Qiang et al. 2014). However, due to the limited bandwidth of the operational amplifier circuit of active power filter, it is difficult to effectively suppress EMI in wide frequency range of motor drive system for EV. Therefore, adding passive filter is the most commonly used and convenient method to suppress conducted interference in motor drive system. Passive filters are generally composed of differential mode inductor, differential mode capacitor, common mode choke, common mode capacitor, common mode transformer, etc. The CM and differential mode (DM) conducted interference can be effectively suppressed through reasonable combination of passive components such as capacitance, inductance and resistance (Zhai et al. 2016; Yang et al. 2012; Chen et al. 2012). The EMI filter of electric vehicle motor controller is different from that of industrial motor controller. The input DC power supply of motor controller ranges from 200 V to 800 V, and the current reaches hundreds of amperes. At present, there is no special theory and method for the design of the wide-frequency-band and highcurrent EMI filter of HV power supply system for EVs from 150 kHz to 108 MHz with high insertion loss. The conventional design method of industrial motor filter is usually used to suppress the conduction EMI of motor inverters for EVs, which is only suitable for EMI suppression of AC power line below 30 MHz. Some EMI filters are also installed at the motor inverter output port to suppress radiated emission from the long three-phase cable connected to the motor (Zhang 2016; Yao 2014). However, the capacitance of the filter is easy to resonate with the inductance of motor winding, resulting in three-phase current or voltage imbalance, which makes the motor unable to run normally. Therefore, EMI filter for EV is usually added to DC input port of motor inverter instead of AC output port. On the other hand, different from the stable and single operation condition of industrial drive motor, the operation condition of motor drive system of EV is complex and dynamic, which makes the source impedance of EMI noise from the motor inverter change with the change of motor load, and is no longer as stable as that of industrial motor. Therefore, the constant impedance of EMI noise source is adopted in the design method of industrial motor filter, which is not suitable for EMI suppression

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under variable noise source impedance of motor inverter for EV. So, it is often necessary to modify the filter parameters by multiple EMC tests before passing the standard test. This traditional industrial motor filter design method not only has a long design cycle and high cost, but also the filter designed is often large and heavy, which not only will cause over design, but also is difficult to integrate into the motor controller to meet the requirements of vehicle environmental test. ➀ EMI of Motor drive system The fast switching of power devices (such as IGBT) is the main cause of EMI in electric drive system. Electromagnetic disturbance forms differential mode (DM) interference and CM interference through the electromagnetic coupling path. Due to the different system structure and electrical and mechanical characteristics, the formation mechanism of EMI in industrial motor drive system is not suitable for EMI analysis of EVs. At present, the electromagnetic radiation test of electric vehicle is mainly based on the standard SAEJ551-5-2012 to measure the whole vehicle electromagnetic field intensity from 150 kHz to 108 MHz. In order to protect the vehicle receiver from the interference of high voltage components such as motor drive system, the EMI characteristics of high voltage components are described by testing the conducted disturbance voltage, conducted disturbance current and radiated electromagnetic field strength of high voltage DC power lines. Therefore the limit requirements and measurement methods for the electromagnetic emission of the HV and LV components of EVs in the frequency band of 150 kHz–108 MHz was defined in the international standard CISPR25-2016. However, through a large number of tests, it can be seen that the components without EMC design can hardly meet the requirements of the standard limit level 3. Therefore, the mechanism, prediction and suppression method of EMI in the PMSM drive system are very important for the EMC for EV, especially for conducted electromagnetic emission, which can cause radiated emission. Although the motor controller has adopted EMI suppression measures, the electromagnetic disturbance measurement results still can not meet the standard limit requirements, as shown in Fig. 1.1. EMI test of motor drive system for EV includes conducted emission test and radiated emission test. The conducted disturbance measured by line impedance stabilization network (LISN) and current clamp is a mixed result of CM interference and DM interference, while radiated disturbance measured by antenna is the sum of electromagnetic field vector superposition. The mechanism of CM interference and DM interference can only be qualitatively analyzed by conducted and radiated emission test (Baidy et al. 2013; Bishnoi et al. 2013), but it cannot reflect the characteristics of EMI in the dynamic operation of EVs under multiple working conditions, nor can it analyze the influence of system components and internal elements of motor controller on EMI, so there are great limitations. The modeling and simulating of EMI source and propagation path can analyze various EMI of motor drive system of EV under multi working conditions. Therefore, the research on the prediction and suppression methods of EMI based on modeling

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1 Introduction

Fig. 1.1 Conducted disturbance voltage of high voltage positive line of a motor controller

and simulating is becoming more and more important, and has become an important technical way to analyze and predict EMI mechanism (Liu and Liu 2014). ➁ EMI Modeling and Simulation Many scholars have carried out modeling and simulation research on CM interference and DM interference (Gao 2014; Cai et al. 2018). The modeling and simulation of CM interference mainly focuses on the influence of distributed capacitance between inverter and ground, distributed capacitance between cables and ground, distributed capacitance between motor winding and housing on CM interference (Zhu et al. 2001; Qiang et al. 2014). The modeling and simulation of DM interference mainly focuses on the influence of parasitic parameters of electric drive system on DM interference (Zhai et al. 2016). The simulation model of electromagnetic emission is mainly composed of EMI source model and propagation path model. The characteristic parameters of propagation path directly affect the DM interference path and CM interference path, and then affect the total EMI response characteristics. At present, there are many researches on electromagnetic emission model of system components, but few on Modeling and Simulation of conducted and radiated EMI of the whole electric drive system. Many switching states of power devices in three-phase PWM inverter in motor controller are equivalent to ideal interference sources (Chen et al. 2012). The influence of parasitic parameters and nonlinear working characteristics of power devices on interference source signals is not considered. The electromagnetic emission model of electric drive system is composed of power battery model, DC and AC power cables model, motor model and power inverter (such as IGBT module, DC link module, radiator, chassis, metal bar, etc.)

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(Zhang 2016). The power battery model is mainly used to study the influence of distributed parameters between battery and vehicle chassis on EMI (Wang et al. 2015). DC and AC power cables are mostly modeled by transmission line theory (Wang et al. 2015). According to the amplitude frequency characteristics of motor winding port impedance, many scholars have constructed high frequency equivalent circuit model of motor (Zhang 2014).A study is presented which develops a low-frequency parameter-based model of induction machines using DM and CM impedance measurements (Zhang 2014). For the inverter model, C. jettanasen, B. revol and J. espina used two port equivalent circuit method and linear matrix equivalent method to analyze and predict electromagnetic interference (Jettanasen and Ngaopitakkul 2010; ReVol et al. 2011), but did not consider the influence of parasitic parameters of inverter internal circuit on EMI. J. Lai and Huang et al. established the high frequency equivalent circuit model of inverter (Lai et al. 2004), analyzed the influence of high frequency parasitic parameters on EMI characteristics, and proposed that the extraction of parasitic parameters is the key to establish the inverter EMI model. However, due to the simplified interference source and incomplete parameter extraction, this circuit model is only applicable to conducted EMI simulation below 10 MHz. Therefore, the reasonable establishment and optimization of inverter model has increasingly become the core problem of electromagnetic interference modeling of electric vehicle electric drive system, which needs to be solved urgently. ➂ Advantages and problems of system behavior level simulation modeling In the modeling and simulation of lumped circuit of the inverter system, it is difficult to obtain the physical parameters of circuit elements, and the power inverter circuit of motor controller is relatively complex, which makes the time-domain simulation of lumped circuit long, difficult to converge, and can only predict conducted emission at low frequency, and the prediction accuracy is poor (Bondarenko et al. 2015; Zhu et al. 1999). Compared with the time domain simulation, although the frequency domain simulation has the advantages of fast calculation and easy convergence, it is difficult to ensure the accuracy of EMI prediction due to the simplification of the model and the difficulty in extracting parasitic parameters. System behavior level simulation modeling can solve the above problems in lumped circuit modeling and simulation. However, the simulation modeling method based on Thevenin and Norton equivalent circuits of the system with two or three ports is mostly adopted by researchers (Bishnoi et al. 2014), which can only analyze the EMI of DC port or AC port of motor controller, but cannot analyze the interference between ports (such as EMI from AC output port to DC input port). Jettanasen proposed a two port equivalent circuit simulation model to predict the total EMI of the system. However, due to the simplification of EMI source and inverter model, it is only suitable for simulation below 10 MHz (Jettanasen and Ngaopitakkul 2010). As far as the simulation modeling of inverter is concerned, due to its complexity, the inverter modeling based on system behavior level is the difficulty of EMI modeling of electric drive system (Bondarenko et al. 2015).

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1 Introduction

➃ Modeling and Simulation of EMI in power inverter While full-wave models are accurate, applying full-wave models to complex systems like a power inverter is difficult, as they require substantial computational time and memory. More importantly, such full-wave models are often a “black box” that does not directly show which parts of the system are responsible for a particular EMI problem or how to solve the problem without additional simulations and effort. Models of inverters also often require nonlinear elements that cannot be modeled easily with a full-wave solver and should be considered using circuit analysis. An equivalent SPICE (Simulation Program with Integrated Circuit Emphasis)-based model which includes the system parasitics is a better approach, since it can give a straightforward correlation between system geometry and parasitic circuit elements and the resulting common-mode (CM) currents (Zhai et al. 2017; National Radio Interference Standardization Technical Committee 2018; Baidy et al. 2013; Bishnoi et al. 2013; Liu and Liu 2014; Gao 2014; Cai et al. 2018; Zhu et al. 2001; Qiang et al. 2014; Zhai et al. 2016; Yang et al. 2012; Chen et al. 2012; Zhang 2016; Yao 2014; Wang et al. 2015; Zhang 2014; Jettanasen and Ngaopitakkul 2010; ReVol et al. 2011; Lai et al. 2004; Bondarenko et al. 2015; Zhu et al. 1999; Bishnoi et al. 2014; Witting et al. 2006; Toure et al. 2013; Ardon et al. 2010). Model-order reduction (MOR) techniques may help provide an equivalent circuit representation, but these circuits are purely functional and, like the S parameters, do not provide significant physical insight into the inner workings of the device. In, equivalent SPICE circuit elements are determined from z-parameters found from 3-D full-wave models. While the resulting circuit is useful, modeling the complete power inverter (whose precise geometry may be unknown) requires substantial time and effort. A SPICE-based model can be obtained from schematics, harness information, and system layout (e.g., the IGBT, heatsink, and enclosure geometry). Several methods are available for extracting parasitic SPICE parameters from a complex geometry. Many of these methods are based on 3-D finite-element analysis or the partial element equivalent circuit (PEEC) method. The output from finite-element analysis is not typically a simple SPICE circuit but a black box measure of circuit characteristics, for example, the S parameter values between two ports. The PEEC approach provides a SPICE model of parasitics in terms of RLGC matrices, but may require hundreds or thousands of elements to represent even a simple geometry, which is too many for an intuitive understanding of how the circuit works. Parasitics may also be obtained through measurements or a combination of measurements and full-wave simulations. One approach is to use time-domain reflectometry (TDR) and transmission line theory to extract parasitics (Ardon et al. 2010). A measurement based inverter modeling method proposed by M, Reuter, the measured scattering parameters are equivalent to common mode and differential mode impedance. Su et al. Proposed an EMI modeling method based on the measurement of common mode impedance and differential mode impedance of three-phase AC motor (Reuter et al. 2013). However, this kind of method regards the inverter as a “black box” and does not provide the

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parasitic parameters of the parasitic circuit of the internal components of the inverter (Witting et al. 2006). ➄ EMI suppression method of inverter system The EMI suppression methods of three-phase PWM inverter include soft switching technology, optimizing PWM control algorithm and installing filters on power input and output cables. Due to the limited EMI suppression effect of soft switching technology and optimized PWM control algorithm, filtering technology is a common method for EMI suppression of motor inverter. Using full wave modeling method, the inverter is equivalent to a “black box”. The interference source and propagation path inside the inverter are not known, so the filter and shielding can only be added outside the inverter and cable, and the interference path can be cut off outside the inverter. In the research of EMI suppression outside the inverter, Akagi designed an inverter EMI filter to suppress the common mode voltage at the motor side, the leakage current from the motor bearing to the ground and the common mode leakage current from the inverter to the ground (Bishnoi et al. 2012). S. Wang and H. bishnoi et al. Designed an EMI filter to suppress the common mode current of inverter and motor to vehicle chassis (Gong and Ferreira 2014; Gang et al. 2015). Gong proposed a design method of EMI CM filter to suppress conducted CM interference and DM interference generated by SiC JFETS of inverter (Di and Sarlioglu 2016). M. Reuter and D. Piazza et al. Proposed a EMI suppression method by inserting damping resistance in series between inverter and vehicle chassis or between motor and vehicle chassis, which can suppress the CM current generated by series resonance (Reuter et al. 2013). Due to the high power and high current of electric vehicle motor drive system, the EMI filter designed is large in volume and takes up limited space in the vehicle. In order to effectively suppress EMI of electric vehicle motor drive system, the resonance effect of inverter parasitic components must be considered. In the early stage of product design and development, the internal circuit of inverter is optimized for EMC and EMI suppression design. It’s remarkable that this method not only increases the volume and weight of the system, but also induces new EMI. In addition, the resonance effect of parasitic components in the inverter is ignored, so the EMI cannot be effectively suppressed. The above research methods can only suppress EMI below 30 MHz, while the power cables of EV will produce conducted interference from 150 kHz to 110 MHz. The existing filter for industrial inverter cannot meet the standard limit requirements. Based on SPICE modeling method, Natalia et al. proposed a measurement-based model of the electromagnetic emissions from a power inverter, established the relationship between the geometric dimensions of the internal components of the inverter and the parasitic circuit parameters. By establishing the transmission characteristics (s parameters) and port impedance characteristics of the two port network of inverter, the causes of resonance were analyzed to determine the parasitic components in the inverter response for resonance, the idea of adding RC filter in the DC cables bar and CM ferrite choke at the AC cables bar to suppress EMI is proposed (Bondarenko et al. 2015).

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This book focuses on: considering the parasitic parameters of power semiconductors, the high frequency equivalent circuit model of motor inverter system is established to predict conducted EMI, which provides a simulation platform for predicting conducted EMI. Several EMI filter design methods for high-voltage DC ports of EV motor controllers are proposed, as follow: • • • •

A wide-band EMI filter based on engineering method. The EMI filter based on the resonance peaks suppression. EMI filter for high voltage DC power line using magnetic ring. EMI filter for high voltage DC power line using air core.

To reduce the conducted emission of 150 kHz–108 MHz frequency band to meet the standard limit requirements. The EMI characteristics of the motor drive system under typical working conditions are predicted by using the method of modeling, simulation and experiment.

1.2.2 EMC Problems of DC-DC Converter System Unlike conventional internal combustion engine vehicles (ICEV), LV power supply of 12 V or 24 V is need to be provided by the HV/LV DC-DC converter in EVs. As a key component of EVs, the HV/LV DC-DC converter in EV is used to convert hundreds of volts of HV DC of power battery into LV DC to charge on-board LV batteries, and supply power to LV on-board electrical components (Gang et al. 2015). The DC-DC converter typically uses power semiconductor devices (such as IGBTs, MOSFETs, etc.) to implement pulse width modulation (PWM) control to regulate the output low voltage (Di and Sarlioglu 2016; Hegazy et al. 2012). Fast switching on and off of the power semiconductor devices generates high current change rate di/dt and voltage change rate du/dt, which forms electromagnetic disturbance source (Safaee et al. 2015; Hasan and Graham 2018) and propagates outward through the parasitic parameters of the internal components and the external HV and LV lines of the HV/LV DC-DC converter, forming coupling paths, causing conducted and radiated disturbances (Grobler and Gitau 2015). It would not only interfere with the on-board and off-board receivers, but also interfere with the on-board HV and LV components through the conducted interference coupling paths, such as motor controller (DC-AC inverter), Battery Management System(BMS), Vehicle Control Unit (VCU), etc., and even affect the safety of the vehicle. In particular, the conducted EMI generated by the HV/LV DC-DC converter may cause the radiated emission of the HV/LV DC-DC converter to exceed the standard limits, and even cause the radiated emission of the vehicle to fail to meet the requirements of the EMC standards. Therefore, the generation mechanism, prediction and suppression methods of HV/LV DC-DC converter conducted EMI are very important

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for the EMC of EVs. In order to finally ensure the EMC of EV and make the electromagnetic emission of the whole vehicle meet the standard requirements, the international standard CISPR25 and the Chinese standard GB/T18655 stipulate the limit requirements and measurement methods for the conducted emission of 150 kHz– 108 MHz and the radiated emission of the 150 kHz–2.5 GHz of HV and LV parts of EVs. A large number of test results show that the products without EMC design can hardly meet the requirements of the standard limits level 3. Figure 1.2 shows the conducted emission of the LV power lines when the HV/LV DC-DC converter is working. It can be seen that the conducted disturbance voltage cannot meet the requirements of level 1. The electromagnetic radiation of the DC-DC converter will make the vehicle radiation emission unable to meet the standard limit requirements, as shown in Fig. 1.3. Although DC-DC converters are also used in switching power supply for industrial applications, a lot of research has mainly studied the EMI analysis and filter design of PCB level DC-DC converters (Huynh et al. 2016; Ales et al. 2014; Ali et al. 2012a, b). However, the topology, voltage type and level, and load characteristics of DC-DC converters for switching power supply are different from those of HV/LV DC-DC converters for EVs. In addition, the applicable EMC standards are also different. Therefore, the EMI generation mechanism and suppression method of switching power supplies are not suitable for the HV/LV DC-DC converters of EVs. In the field of EVs, most of the research of EMI from DC-DC converter is mostly focused on LV PCB board-level DC-DC converters(Grobler1 and Gitau 2017; Zhao

Fig. 1.2 Conducted emission of DC-DC converter

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Fig. 1.3 The impact of DC-DC converter on vehicle radiated emission

et al. 2019). Based on the dual-port network theory, the effects of PCB parasitic parameters, DC-DC switching techniques and layout topology optimization on the EMI from DC-DC converter circuit are analyzed in (Zhai et al. 2018; Gu 2016). However, there are few studies on the EMI mechanism and suppression methods of HV/LV DC-DC converters for EVs. At present, the research on EMC of HV/LV DC-DC converters focuses on the formation mechanism of conducted EMI and its suppression methods. Modeling, simulation and experimental measurements are usually used in the research of the EMI from the DC-DC converter of EVs (Ma 2014). Gu Long studied interference source, interference coupling paths of the low power isolated full-bridge DC-DC converter for EV with input voltage of 120–160 V. However, the influence of high frequency parasitic parameters on EMI paths has not been fully considered (An et al. 2013). On the basis of considering the influence of parasitic parameters of switching devices, cables and transformers, Lingyuan Ma and Zongyu An established a conducted EMI prediction model for low-power HV/LV DC-DC converter system of hybrid electric vehicles, and only qualitative analysis of conducted path was made(Laour et al. 2017). Previous studies on conducted EMI of the DC-DC converter for EV are mostly based on port network theory and black box theory. The input port and the output port of the converter are equivalent to EMI sources, and the relationship between EMI sources and EMI noise from power lines is analyzed. However, the effects of internal circuit parasitic parameters on EMI noise is not analyzed (Zongyu 2013; Pahlevaninezhad et al. 2013). Ideal trapezoidal wave is usually equivalent to interference source, and parasitic parameters of switching devices such as MOSFET are not taken into account, so it can not reflect the signal characteristics of real EMI source. Although some scholars have studied the effects of parasitic parameters such as the lead inductance of MOSFET and the distributed capacitance of transformer on

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common mode interference (Zhai et al. 2018; Ferber and Vollaire 2013), no quantitative analysis of the conducted EMI of the system has been made(Ishtiyaq and Morten 2014). At present, the high-frequency equivalent circuit of conducted EMI of DCDC converters have not fully consider the influence of parasitic parameters. Then, it is impossible to determine the parasitic components and coupling paths inside the DC-DC converter that are responsible for EMI. Therefore, it can’t be realized to guide the EMC design of the internal circuit of the DC-DC converter in the period of the stages of product design (Makda and Nymand 2014). At present, the EMI suppression of the HV DC power supply still adopts the industrial classic filter design methods, and its EMI suppression frequency band is lower than 30 MHz (Kovacevic et al. 2013). At present, the design theory and method of EMI filters for high-voltage DC power systems in the frequency band of 150 kHz– 108 MHz are lacking. When the conducted emission of the DC-DC converter appears to exceed the standard, an EMI filter is usually installed at the DC HV input port of the HV/LV DC-DC converter. several rectifications are needed to make the HV/LV DC-DC converter pass the standard test. This type of filter can only be installed in the later stages of product design, which results in high costs and long cycles in research and development, and it is not easy to implement in engineering. In addition, the verification of the EMI filter insertion loss usually adopts the offline measurement method, and the EMI filter is not added to the actual DC-DC converter system for on-load operation to verify the actual effective insertion loss of the filter module (Tan et al. 2013). This book focuses on the descriptions: considering the parasitic parameters of power semiconductors, the high-frequency equivalent circuit model of the zerovoltage switching (ZVS) DC-DC converter is established to predict the conducted EMI; using the established high-frequency equivalent circuit model, the transfer functions of CM interference and DM interference at the key frequency points are established to predict HV power line conducted EMI and radiated EMI and to determine the main component parameters that affect the formation of EMI. An effective HV port wide-band conducted interference suppression method is proposed which can reduce electromagnetic emissions in the 150 kHz–108 MHz frequency band to meet the standard limit requirements; a PCB board-level filter circuit design method based on the resonance point conducted emission suppression is proposed which can be implemented in the controller, is small in size, low in cost, high in efficiency and can be realized in different stages of product development.

1.2.3 EMC Problems of Wireless Charging System The principle of wireless charging is related to wireless power transfer (WPT). With the development of wireless charging technology for electric vehicles, there are many functions and uses. According to the classification of charging power level, see Table 1.1. According to the motion state of the vehicle during charging, it can be divided into static charging and dynamic charging (see Fig. 1.4), and long-distance

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Table 1.1 WPT systems with different power levels Type

WPT1

WPT2

WPT3

WPT4

WPT5

WPT6

Power

3.7 kW

3.7–7.7 kW

7.7–22 kW

22–30 kW

30–60 kW

60 kW

Fig. 1.4 Static charger and dynamic charger

Fig. 1.5 Short distance charging, midway charging and long distance charging

charging, intermediate charging and short-distance charging can be divided according to the purpose of charging vehicles (see Fig. 1.5). In addition to the efficiency and power requirements of WPT, electromagnetic safety and electromagnetic compatibility are also important issues. Long time exposure to strong magnetic fields seriously endangers human health (Ding et al. 2015), and also affects nearby electronic equipment. In addition, electromagnetic noise through the power line will interfere with other on-board equipment, will also pollute the power grid, and affect the office, home and other power grid equipment. Therefore, in order to improve the safety and reliability of electric vehicles, we must focus on two aspects of the wireless charging system of electric vehicles: electromagnetic field (EMF) which affects human safety and electromagnetic interference (EMI) which affects the safety of electrical equipment, it includes: ➀ Ground side: Influence of harmonics and conduction voltage on public power grid ➁ Between ground and vehicle body: Effects of coupling field leakage on organisms ➂ Vehicle side: Harm of electromagnetic emission of system to vehicle components ➃ Vehicle electromagnetic radiation.

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Table 1.2 EMC standards for wireless charging system Issuing unit or country

Standard No

Standard name

ICNIRP

ICNIRP2010

ICNIRP Guidelines for Limiting exposure to Time-Varying Electric and Magnetic Fields (2010 version)

Europe

IEC 61980-1

Electric vehicle wireless power transfer (WPT) systems - Part 1: General requirements

IEC 61980-2

Specific requirements for communication between electric road vehicle (EV) and infrastructure

IEC 61980-3

Specific requirements for the magnetic field wireless power transfer systems

SAE J2954

Wireless Power Transfer for Light-Duty Plug-in/Electric Vehicles and Alignment Methodology

ISO 19363

Electrically propelled road vehicles—Magnetic field wireless power transfer—Safety and interoperability requirements

Japan

JASO TP 13002

Measurement methods for electromagnetic field of vehicles with regards to human exposure

China

GB/T 37132-2018

General requirements and test methods of electromagnetic compatibility for wireless power transmission equipment

U.S.A

(1) Domestic and international standards for wireless charging system Domestic and foreign electric vehicle WPT technical standards and regulations define the measurement methods and limits of electromagnetic field and electromagnetic emission, and put forward higher technical and safety requirements for EV with WPT function, shown in Table 1.2. (2) Electromagnetic field (EMF) safety The EMF of electric vehicle wireless charging system is concerned about the low frequency electromagnetic field emission (1–400 kHz) of coupling coil. Although the electric field strength between the coils decreases with the increase of the distance from the coils, there may still be electromagnetic fields around the vehicle body that are not conducive to the health of the organism. With the improvement of wireless transmission power, the electromagnetic field of human and animals in the exposed area will also increase, especially in special cases such as the coupling coil offset, which will produce high-intensity electromagnetic field, and human body exposed to high-intensity electromagnetic field for a long time will cause certain harm to human sensitive organs. For the magnetic coupling resonant wireless charging system, the research of electromagnetic field mainly focuses on two aspects.

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1 Introduction

One aspect is the magnetic field distribution when the coupling coil is aligned. When the coupling coils are aligned, Wang Q et al. Studied the electromagnetic field distribution characteristics around the coupling coil and compared the magnetic field distribution under different charging modes, such as constant current charging mode and constant voltage charging mode (Wang et al. 2018). The compensation circuits with different profiles, different spatial arrangements and different topologies are studied, and the changes of transmission efficiency and EMF are compared. Chen w et al. Proposed a method based on cost-effectiveness equivalent equation to compare the transmission efficiency and effective magnetic field area of rectangular, hexagonal and circular coil structures (Chen et al. 2016). Cho Y proposed the effects of horizontal and vertical winding on efficiency and electromagnetic field distribution (Cho et al. 2018). The topology of the compensation circuit and discusses the optimization of transmission efficiency and suppression of electromagnetic leakage under different structures is studied in reference (Esteban and Sid 2015; Rozman and Fernando 2017). Many scholars have studied and analyzed the magnetic field distribution characteristics of the coupling coil (Christ 2013; Zeng et al. 2017). For example, Hikage simulation predicted the magnetic field distribution generated by the coupler, especially the impact on human medical implants and EMF suppression measures (Alicia et al. 2018). In order to reduce the leakage magnetic field, some methods such as adding shielding case, optimizing coil structure and ferrite layout, impedance matching and magnetic field reverse elimination are proposed. On the other hand, the magnetic field distribution of the coupling coil offset. In the charging process of electric vehicle equipped with wireless charging system, the coil lateral deviation will be caused by incorrect parking position, and the coil will roll due to vibration. Many previous literatures have studied the magnetic field distribution of the coupling coil of wireless charging system and the safety of human electromagnetic field under different offset distances. The electromagnetic field distribution of 18kw rectangular coil with 75 mm longitudinal offset and 120 mm lateral offset and described the magnetic field distribution of human body model near the coupling coil under offset (Cai et al. 2018). Tommaso Campi et al. studied the magnetic field distribution of a 7.7 kW circular coupling coil under alignment and maximum offset. The magnetic field safe area of 22 kW circular coil with offset of 100 mm and 200 mm is analyzed in reference (Lee et al. 2018; Hikage et al. 2017). Lei Zhao et al. considered the offset of coupling coil in three directions (Zhao et al. 2019). The magnetic field distribution in the case of alignment and offset of coupling coils is analyzed in references (Lee et al. 2018; Hikage et al. 2017). Santis V also analyzed the distribution of electric and magnetic fields at different locations and their effects on drivers. A wireless charger based on a novel tightly coupled resonance method, which reduces the harmonic content of the primary and secondary coil currents and the leakage magnetic field between the coil pairs, thus improving the transmission efficiency are proposed in reference (Valerio et al. 2018; Hwang et al. 2017). Some researchers have studied two-way wireless energy transmission. Active switches are used to replace uncontrollable diodes, and PWM phase-shifting control method is used to drive the switches, which reduces the harmonic content and ringing phenomenon, and improves the transmission efficiency of the system (Dai et al. 2017; Zeng et al.

1.2 EMC Problems of EVs

17

2017; Mahmoud and Mahmoud 2017; Moon et al. 2015). Based on the human body high-resolution model and body model, the magnetic field distribution around the wireless charging vehicle and the evaluation of human electromagnetic safety are studied (Hui et al. 2014). Most scholars only pay attention to the change of mutual inductance and coupling coefficient when the coupling coil is offset, and do not pay attention to the influence of coupling coil current amplitude and phase on magnetic field distribution. Previous studies only focus on the changes of power or electromagnetic field during migration, and do not describe the relationship between power changes and electromagnetic field distribution. In addition, in the previous study of magnetic field distribution, there is no detailed measurement method, and there is no comprehensive measurement of magnetic field in different areas of the vehicle according to the standard requirements. (3) Electromagnetic interference (EMI) of wireless charging system The wireless charging system needs a higher resonant frequency when it works. The high-speed on-off of power switching devices produces high voltage change rate (du/dt) and current change rate (di/dt), which leads to conducted electromagnetic interference and radiated electromagnetic interference. Electromagnetic interference will not only affect the on-board wired sensitive equipment and wireless receiving equipment inside and outside the vehicle, but also affect the power supply quality of the power grid by connecting the power line of the public power grid. The conducted EMI research of wireless charging system mainly includes the establishment of conducted EMI prediction model and suppression method. Hongseok Kim and other Korean scholars obtained the spectrum distribution of low-order current harmonic and voltage harmonic of wireless charging system through theory and experiment (Chung et al. 2017). Heyuan Qi et al. studied the conducted electromagnetic interference of series resonant wireless charging system by constructing the circuit model of series compensation structure system (Ding et al. 2014). Lin Liwen of Beijing University of technology has established a conductive high-frequency equivalent circuit for the wireless charging system of 3.7 KW electric vehicles, analyzed the formation mechanism of common mode interference and differential mode interference, and designed common mode filter and differential mode filter respectively (Lin 2018). Although this over design method can suppress EMI better, there are many filter elements, which not only increases the size, weight and cost of the filter, but also causes unwanted resonance. Cao Yu et al. proposed a design method of priority common mode interference suppression filter for wireless charging system based on sensor function method (Zhai et al. 2018). These filtering methods usually use 50  instead of source impedance and load impedance, and do not consider the actual source impedance and load impedance changes with frequency, resulting in unsatisfactory EMI suppression or new resonance points. The suppression methods of radiated interference sources in wireless charging system include optimizing the layout rules of inverter PCB board, adding damping circuit and optimizing PWM driving pulse. Nguyen et al. reduced the ringing loop by optimizing the PCB layout rules of the controller and reduced the parasitic inductance

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1 Introduction

to reduce the ringing amplitude (Nguyen et al. 2018). Literature (Puyal et al. 2018) uses RC damping circuit on PCB to reduce radiated electromagnetic interference noise. These methods are only suitable for PCB low-power DC-DC converters. When the power is high and the current is large, there will be a certain amount of energy loss. In addition, adding damping elements will affect the high frequency parameters of the circuit, causing additional conducted interference and radiation emission. H Kim et al. proposed a selective harmonic reduction method to reduce the radiation emission of WPT system, but did not consider the overall system efficiency and battery charging mode (Kim et al. 2017). Sunkyu et al. of Korea studied the influence of radiation emission of WPT system on input ADC converter through transmission cable (Kong et al. 2016). Yu Ya of Zhengzhou University used frequency jitter method, chaotic modulation method and periodic modulation method, and applied these three spread spectrum technologies to suppress radiated EMI in wireless charging system (Ya 2016). Zheng Wei et al. of Hubei University of technology used filters to reduce conducted interference, so as to suppress radiated interference (Zheng 2013). This book analyzes the power and efficiency of the coupling device, as well as the anti-offset characteristics by establishing the wireless charging system model of the circular coupling coil with two-sided LCC topology. Then, through modeling, simulation and measurement methods, the electromagnetic field distribution of the coupling coil to it and the offset is described. Finally, the modeling and suppression of conducted EMI in DC power line of wireless charging system, harmonic wave suppression method of power line in public power grid, and electromagnetic radiation of secondary circuit of wireless charging system are described.

1.2.4 EMC Problem of Vehicle Controller With the development of intelligent driving vehicles and unmanned vehicles, automotive electronic and electrical architecture is changing. The information collected and processed by the vehicle controller of two-wheel and four-wheel distributed drive vehicles, four-wheel hub-driven pure electric vehicles, hybrid vehicles, and even centralized electric drive vehicles is increasing, and the vehicle controller and central controller are proposed higher communication bandwidth and speed requirements (Jin et al. 2015). Ethernet will gradually replace most other buses. Figure 1.6 shows a future vehicle domain control network architecture, which is a hybrid network structure in which the backbone network uses Ethernet, and the subsystem network uses Ethernet or traditional vehicle network (Ferber and Krahenbuhl 2013). With the development of Ethernet and domain control, the hardware circuit of the vehicle controller has undergone changes. Ethernet communication circuits have been added on the basis of traditional CAN bus communication. Car Ethernet uses two-wire twisted-pair cable, which has a high speed and will bring new electromagnetic compatibility problems. Therefore, on the one hand, the power integrity and signal integrity of the vehicle controller PCB need to be considered,

1.2 EMC Problems of EVs

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CAN/Ethernet

CAN/Ethernet

Airbag

Radar

Collision sensor

Domain controller Chassis control system

Domain controller Security system

Backbone network Ethernet

Ethernet Car Ethernet Switch Network

Ethernet

Ethernet Information system

Body control system

Domain controller

Domain controller

Seat

Car window

Ethernet

Rearview mirror

Door lock

Image/ language

Camera

Fig. 1.6 Vehicle domain control network architecture

and on the other hand, the electromagnetic emission and electromagnetic susceptibility issues of Ethernet need to be considered. In addition, the electromagnetic immunity of the Ethernet vehicle controller needs to be considered. The electromagnetic environment of the vehicle controller is relatively harsh. It not only faces mechanical environmental problems such as rain and vibration, but also needs to resist the influences of the conduction interference and radiation interference caused by the operation of the motor drive system, high and low voltage DC-DC converters and other high voltage components. The EMC problems of in-vehicle Ethernet are mainly electromagnetic susceptibility, immunity and ESD. The EMC-related issues of vehicle controllers mainly include: power integrity, signal integrity, electromagnetic emission and electromagnetic immunity of PCB board-level. On the one hand, with the increase of vehicle controllers collecting and processing signals, as well as the application of Ethernet, higher requirements are placed on chip speeds. Which leads to severe switching noise and challenges the power distribution network design of the vehicle controller. On the other hand, high-speed Ethernet signals are prone to reflection under the condition of impedance mismatch, resulting in a series of signal integrity problems such as ringing and common mode noise, which may affect signal recognition in severe cases. In addition, because Ethernet has a higher rate, the spatial coupling of electromagnetic fields produces crosstalk interference to adjacent signal lines.

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1 Introduction

➀ Power integrity The performance of Power Distribution Network (PDN) directly affects system performance such as system reliability, signal-to-noise ratio and bit error rate, as well as EMC performance. Too high impedance of the board-level power supply channel and too large simultaneous switch noise (SSN) will cause serious power integrity problems (such as reference level error caused by synchronous switching noise, excessive DC voltage drop, heat, etc.).Which will have a fatal impact on the stability of devices and systems. Serious PDN design defects will also cause electromagnetic radiation and conduction disturbances at resonance points with higher impedance. Power Integrity (PI) design is to reduce the board-level PDN impedance through reasonable planar capacitors, discrete decoupling capacitors, planar partitions, and electromagnetic bandgap (Electrical Bandage Gap, EBG) applications to ensure that the power supply requirements of the chip are met, control the synchronize switching noise and reduce electromagnetic interference emission. In previous studies, there are many ways to reduce PDN impedance, including decoupling capacitors, planar structures, and embedded capacitors. In practical applications, the main method is to add decoupling capacitors. In order to have a decoupling effect in sufficient bandwidth, decoupling capacitors include chip-level, package-level and board-level decoupling capacitors. The planar decoupling method has a good decoupling effect in the range of several hundred MHz–1 GHz, but it is not suitable for the decoupling of the middle and low frequency bands and high frequency bands, and it will increase the noise coupling of the PDN. Therefore, on the basis of the planar de-rigid method, decoupling capacitors are also needed to help reduce the PDN switch. Compared with power planes, embedded capacitors and EBG, the decoupling capacitor method is the most flexible low-impedance solution for PCB power distribution networks. The value of the decoupling capacitor, the selection method of the package, the quantity and the installation position of the capacitor are the research focus. In the frequency band above the self-resonance point of the decoupling capacitor, the location of the capacitor is particularly critical (Chu et al. 2013). Jun Fan et al. modeled and analyzed the influence of the connection distance between the patch capacitor and the power/ground plane in the multilayer PCB on the PDN (Fan et al. 2002). The fast optimization algorithm of decoupling capacitor is also a research hotspot. Kai-Bin Wu et al. used a genetic algorithm to calculate the optimal layout, capacitance and quantity of decoupling capacitors, which improved the accuracy of decoupling capacitor selection (Wu et al. 2005). Krishna Bharath et al. used genetic algorithms combined with a high-efficiency PDN simulator based on the multilayer finite difference method to optimize the decoupling capacitance of the multilayer PCB. Combined with the design method based on the target impedance, the hybrid genetic algorithm is applied to the design of the PDN decoupling capacitor network, and the type and number of decoupling capacitors required are optimized (Wang et al. 2015). In addition to the research on single-chip power integrity, some scholars have carried out modeling and decoupling design for multi-chip multi-input PDN

1.2 EMC Problems of EVs

21

(Zhang 2016; Qin and Li 2013). Through the theoretical derivation of the multiport network and accurately capture the current distribution characteristics of the PDN, a distributed modeling method suitable for multi-chip multi-input PDN and a complex PDN whole board decoupling scheme are proposed. On the premise of meeting the PDN design requirements, the combination with the least number of decoupling capacitors and the optimal target impedance are selected. Using the optimized frequency domain target impedance method, the selection scheme of the type and number of decoupling capacitors is given for the multi-chip power distribution network model. In addition to keeping the PDN low impedance, the method to improve the quality of the PDN is to use EBG to isolate the noise (Shen et al. 2017), but the actual application cost is higher. Moreover, this method is more suitable for power distribution network optimization of systems of several GHz and above, and the application benefit in vehicle controllers is small. ➁ Signal integrity Signal integrity problems include signal reflection, crosstalk, and delay, which are problems faced by high-speed signals. The reflection of the crystal signal in the vehicle controller may cause signal misjudgment and secondary triggering. Severe reflection and crosstalk may cause noise oscillation, and electromagnetic radiation is formed in the PCB where the impedance is high. The communication rate of the traditional vehicle controller is low, and the signal integrity problem is not obvious. After the introduction of Ethernet, higher-speed automotive Ethernet signals may produce reflections or crosstalk to adjacent signal lines during transmission. Ethernet is a differential wiring structure. In the PCB, when the wiring structure of the differential signal is asymmetry, a common mode current will be generated. Common mode current is usually an important source of conduction interference and radiation interference in circuits. As the rate of differential signals such as CAN bus and Ethernet in the vehicle controller continues to increase, the electromagnetic radiation of common mode current will also be much larger than that of previous VCU circuit boards (Paul 1989). Therefore, in terms of Ethernet wiring design, special attention should be paid to the symmetrical design of the differential line. If the Ethernet differential pair routing design on the PCB is unreasonable, crosstalk and reflection will easily occur, as shown in Fig. 1.7. C. Paul et al. studied the coupling characteristics of differential transmission lines, and proposed that the differential line vias are the main cause of circuit asymmetry. Xiaomin Duan et al. proposed that the asymmetry of the differential signal routing on the PCB connector pins will produce common mode interference signals. Usually the PCB asymmetric structure is unavoidable, and the asymmetry and impedance mutation of the differential wiring can be reduced by means of termination or improvement of the wiring (Hockanson et al. 1994). Chen Jianhua and others used the method of π-type termination and T-type termination to suppress the common mode current generated by the asymmetry of the differential circuit structure, and proposed that the common mode noise of the differential signal is determined by frequency, loop area, wire length, medium thickness, and dielectric

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1 Introduction

Fig. 1.7 Crosstalk and reflection phenomenon of PCB Ethernet trace

constant, differential current and common mode current (Chen 2010). Wei-Da Guo et al. proposed a planar spiral wiring scheme to reduce the asymmetry of differential wiring (Guo et al. 2006; Shiue et al. 2006). Celina Gazda et al. proposed a scheme to suppress common mode noise over a wide frequency band by tightly coupling the differential microstrip line (Gazda et al. 2010). In addition to improving the structure of the differential trace, some scholars have adopted compensation measures such as installing capacitors or inductances at the turn of the differential trace to suppress common mode noise (Chang et al. 2012). There are also researchers designing filters for differential signals to get a good common-mode rejection ratio (Liu et al. 2008; Wu et al. 2009; Naqui et al. 2012). ➂ Electromagnetic radiation Under the action of common mode interference source and differential mode interference source, the trace on the PCB board may be equivalent to an effective transmitting/receiving antenna. Therefore, improper layout may significantly increase the electromagnetic radiation of the PCB board. The common mode interference current and differential mode interference current on the PCB can pass through the power

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23

line and signal line outside the vehicle controller to form electromagnetic radiation. In addition, the gap of the vehicle controller housing and the wiring harness connector may also leak electromagnetic fields and form radiation. These electromagnetic radiations may cause the radiation emission of the vehicle controller to fail to meet the EMC standard limit requirements, and even interfere with the on-board and outdoor radio receiving equipment. ➃ Electromagnetic immunity problem Chinese Standard GB/T 21437.2 “Electrical Disturbance Caused by Conduction and Coupling on Road Vehicles Part 2 Electrical Transient Conduction Along Power Lines” and GB/T 21437.3 “Electrical Disturbance Caused by Conduction and Coupling on Road Vehicles Part 3 Except Power Electrical transient emission of wires outside the line through capacitive and inductive coupling” stipulates the immunity requirements of vehicle components to power lines and electrical transient interference of wires other than power lines. Chinese standard GB/T199512005 “Test Method for Electrical Disturbance from Electrostatic Discharge of Road Vehicles” specifies the level of electrostatic discharge. The standards for immunity of on-board components also include Chinese standard GB/T 33012.4-2016 Vehicle immunity to narrowband radiated electromagnetic energy-Large Current Injection (BCI) method (1 MHz–400 MHz) and GB/T 33012.2-2016 Vehicles to narrowband radiation Immunity of electromagnetic energy-outside vehicle radiation source method (10 kHz–18 GHz). In addition, anti-lightning and anti-surge measures should be considered in the circuit design stage. According to standard requirements, electromagnetic immunity design and test verification of vehicle controllers are required. The vehicle controller low-voltage power supply system should be designed to suppress overvoltage, reverse overvoltage, surge current and transient voltage pulse. Vehicle controller hardware usually uses anti-reverse diodes, thermistors, transient diodes, common mode chokes, π-type filters, capacitors, etc. to form anti-interference hardware circuits. This book focuses on the function and structure of the Ethernet-based vehicle controller, as well as the hardware electromagnetic compatibility design, including electromagnetic emission and electromagnetic susceptibility design, power distribution network (PDN) equivalent circuit modeling and decoupling capacitor optimization method, modeling and Analysis of PCB Signal Integrity Based on Ethernet, PCB electromagnetic emission, as well as the method of suppressing the conduction disturbance of the power line of the Ethernet vehicle controller.

1.2.5 EMC Problems of Battery Management System (BMS) BMS has the following functions:Real time monitoring or calculation of a series of battery related parameters, such as battery voltage, temperature, working current, battery power; management of battery charging or discharging according to relevant

24

1 Introduction

parameters such as environmental state and battery state; balancing function between individual batteries. The topology of BMS hardware is categorized into two types: centralized and distributed. Centralized BMS is to centralize all functions in one controller, which is more suitable for occasions where the battery pack capacity is relatively small. Distributed BMS separates the main control board from the slave control board, sometimes separates the low-voltage and high-voltage parts, so as to increase the flexibility of system configuration, and adapt to modules and battery packs with different capacities and specifications. BMS hardware circuit mainly includes the main control chip and its interface circuit, battery parameter acquisition module, fault circuit, equalization circuit, contactor safety protection circuit, communication circuit, etc. With requirements of the real-time performance and safety of battery parameters acquisition, higher requirements of EMC design of BMS hardware circuit are put forward. Synchronous switching noise (SSN) will be generated when the transistors in high-speed digital chip of the BMS PCB are turned on and off quickly, which can cause power integrity problems. Similar to the VCU power integrity design, the effective way to suppress SSN is to add decoupling capacitors outside the IC to meet the target impedance requirements of the power distribution network (PDN). For the current BMS controllers of electric vehicle, adding appropriate decoupling capacitors to the PCB is considered an efficient and cost-effective method. Crosstalk is another very serious problem faced by BMS PCB. The narrow-band interference generated by high-speed data lines and clocks is coupled to the adjacent signal line through mutual capacitance and mutual inductance between PCB traces, resulting in signal distortion of signal lines. The higher the transmission speed of high-speed data lines, the greater the influence of crosstalk and reflection, impedance matching becomes more and more important. In literature (Wu and Kong 2013), a process that uses matched termination resistors to reduce overshoot and undershoot voltages and electromagnetic noise of reflected signals is proposed. In literature (Song et al. 2010), a passive equalizer structure which can reduce the reflection and crosstalk problems during 16 GB/s high-speed data transmission is introduced. In literature (Pantic-Tanner et al. 1998), the crosstalk problem of signal line caused by synchronous switching noise in power line is studied, and a band-pass filter is designed to reduce the signal line crosstalk noise voltage value. In literature (Xu and Wang 2015), a shielded magnetic ring structure and grounding mode for the interference of the narrowband interference generated by the clock signal on other adjacent signal circuits is designed. In literature (Iida et al. 2014), the influence of crosstalk of parallel signal lines on vehicle FM radio is studied, and the crosstalk voltage of the signal lines is suppressed by comparing the ground plane segmentation of different structures. These literatures reduce the signal line reflection and crosstalk effects by designing shielding structures and reducing the signal line return path impedance. The radiation effect of the differential mode loop antenna and the common mode biostatic antenna caused by the synchronous switching noise can also be suppressed by adhering a conductive plane under the PCB board. It is very important to study the suppression method of common mode current and differential mode current by multilayer conductive plane.

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Fig. 1.8 Resonance peak of electric field intensity caused by periodic pulse signal of clock

In addition, clock circuit and DC-DC module are the main EMI sources of PCB. The resonant peak of electric field intensity caused by clock periodic pulse signal is shown in Fig. 1.8. In the early stage, many scholars conducted research on EMI suppression of vehicle low-voltage electronic controllers. Literature (Mihali and Kos 2006) and literature (Matsuda et al. 2009) used random pulse width modulation and spread spectrum clock modulation to suppress EMI interference generated by DC-DC module MOSFET. However, the effect of this method is limited, and other absorption or filtering circuits are needed to help suppress EMI interference. Literature (Fu et al. 2013) and literature (Pouiklis et al. 2013) respectively proposed common mode choke absorption circuit and bypass capacitor filter circuit to reduce EMI interference. Literature (Berzoy et al. 2015) designed a shielding ring installed around the PCB board power switch to absorb the leakage current generated by the power switch. It is also possible to reduce the coupling path of interference by optimizing the layout of PCB components. At present, BMS controllers of electric vehicles generally use CAN network for communication. Literature (Chen et al. 2009) studies the radiation emission of the traditional CAN bus controller harness, and studies the shielding effectiveness and grounding mode of shielded twisted pair CAN wires. Literature (Lim 2014) studied the relationship between the connection method of CAN contact and the reflection effect of CAN signal, and proposed a method to reduce the reflection. Literature (Fontana and Hubing 2015) studied the influence of electrical fast transient noise on CAN transceivers, and showed that port impedance symmetry is an important factor for CAN transceivers to suppress common mode noise. This book first proposes a method to select the decoupling capacitors for the BMS hardware PCB power distribution network; propose a method to suppress the clock EMI by analyzing the frequency spectrum characteristics of the clock signal and the EMI coupling path;

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1 Introduction

design a switching noise filter by analyzing the switching noise frequency characteristics and coupling path generated by the MOSFET; propose an anti-interference design method for CAN bus circuit through simulation analysis of CAN bus reflection characteristics. Finally, the electromagnetic radiation of the electric vehicle BMS PCB is simulated and predicted by using the SIwave software.

1.2.6 Vehicle EMC Requirements With the development of communication technology, network technology, wireless technology, and power electronic technology, the electromagnetic environment is becoming more and more complex (Lin 2019; Gong 2019). The EMC of electric vehicles marks the reliability and safety of the operation of vehicles and their auxiliary equipment. Electric vehicle EMC allows compatibility between the electromagnetic storage relationship between various electric power, electronic equipment or subsystems of vehicles and the surrounding electromagnetic environment. The prediction and simulation of electric vehicle EMC plays an important role in improving the operation quality and ensuring the safety of EV (Huang et al. 2013; Gao et al. 2013; Ding and Liu 2013; Liu et al. 2018). The operation of electric vehicle will cause interference to the radio receiving and transmitting equipment of the vehicle and living environment around the vehicle. Therefore, the Chinese national standards GB/T18387 and GB/T14023 respectively stipulate the limits of electromagnetic emission of vehicles in the frequency range of 150 kHz–30 MHz and 30 MHz–1000 MHz. In order to meet the electromagnetic emission requirements of the vehicle, the EMC design of the vehicle must be carried out. EMC design plays an important role in EMC and electrical safety of electric vehicle. Domestic and foreign standardization committees and some well-known automobile enterprises have formulated the vehicle EMC standard, which provides design specifications and basis for EMC design. Through the analysis of EMI source, EMI conduction and radiation coupling paths existing on the vehicle, EMI suppression methods such as shielding, filtering and grounding, and vehicle layered design method are used for EMC comprehensive design to meet the requirements of EMC standard limits. EMC test verification is required for vehicle EMC design and product certification. The EMC test content of electric vehicle mainly includes vehicle radiated emission test and radiated immunity test. The main standards are shown in Table 1.3. In order to improve the pass rate of vehicle EMC test certification, it is very important to predict and evaluate the EMI sources in the early stage. Therefore, it is necessary to perform EMI diagnosis and prediction technology to quickly and accurately locate the source and determine the level of EMI.

1.3 Content Introduction

27

Table 1.3 Vehicle EMC Standards Serial Test items number

Regulations/Standards National standard

International standard

1

Magnetic and electric field strength from electric vehicles (150 kHz–30 MHz)

GB/T 18387-2017

CISPR36/CD /

SAE J551-5

2

Measurement of vehicle radio disturbance characteristics for the protection of off-board receivers (30 MHz–1000 MHz)

GB 14023-2011

CISPR12

ECE R10

/

3

Measurement of vehicle radio disturbance characteristics for the protection of on-board receivers (150 kHz–2.5 GHz)

GB/T 18655-2018

CISPR25

ECE R10

SAE J551-41

4

Vehicle Electromagnetic Immunity—off-vehicle radiation sources (10 kHz~18 GHz)

GB/T ISO11451-2 33012.2-2016

ECE R10

SAE J551-11

5

Vehicle Electromagnetic Immunity–Bulk Current Injection–—Bulk current injection (1 MHz–400 MHz)

GB/T ISO11451-4 33012.4-2016

ECE R10

SAE J551-13

6

Vehicle Electromagnetic / Immunity—Power Line Magnetic Fields (60 Hz–30 kHz)

/

7

Vehicle Electromagnetic / Immunity–On-Board Transmitter Simulation-(1.8 MHz–5.85 GHz)

ISO11451-3

/

SAE J551-12

8

Vehicle Electromagnetic Immunity—Electrostatic Discharge

GB/T 19951-2005

ISO 10605

/

SAE J551-15

9

Measurement of electromagnetic GB fields of vehicle with regard to 37130-2018 human exposure (10 Hz–400 kHz)

IEC 62764/CD

/

/

EU American regulations standard

SAE J551-17

1.3 Content Introduction Chapter1 Chapter 2 Chapter 3 Chapter 4 Chapter 5 Chapter 6 Chapter 7 Chapter 8

Introduction. EMC of new energy vehicle. Prediction and suppression of EMI in motor drive system. Prediction and suppression of EMI in DC-DC converter. Electromagnetic safety and EMC of wireless charging system. Signal integrity and EMC of vehicle controller. EMC of BMS. Diagnosis and suppression of vehicle electromagnetic radiation.

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1 Introduction

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Hikage T, Yamagishi M, Shindo K et al (2017) Active implantable medical device EMI estimation for EV-charging WPT system based on 3D full-wave analysis. In: Asia-Pacific international symposium on electromagnetic compatibility, pp 87–89 Hockanson DM, Drewniak JL, Hubing TH et al (1994) Investigation of fundamental EMI source mechanisms driving common-mode radiation from printed circuit boards with attached cables. IEEE Trans Electromagn Compatib 38(4):557–566 Huang X, Lei J, Lai Z et al (2013) Electromagnetic radiation emission correction for electric vehicle below 30 MHz. Safety EMC (4):28–30, 47 Hui S, Zhong W, Lee CK (2014) A critical review of recent progress in mid-range wireless power transfer. IEEE Trans Power Electron 29:4500–4511 Huynh H, Joo S, Kim S (2016) An experimental study of EMI reduction of DC-DC converter with frequency hopping technique. In: IEEE electrical design of advanced packaging and systems (EDAPS) Hwang K, Cho J, Kim D et al (2017) An autonomous coil alignment system for the dynamic wireless charging of electric vehicles to minimize lateral misalignment. Energies 10(3):315–334 Iida M, Maeno T, Fujiwara O (2014) Effect of ground pattern size on FM-band crosstalk between two parallel signal traces of printed circuit boards for vehicles. Electr Eng Jpn 186(1):11–17 Ishtiyaq AM, Morten N (2014) Common mode noise generation and filter design for a hard switched isolated full-bridge forward converter. In: Annual conference of the IEEE industrial electronics society, pp 1312–1317 Jettanasen C, Ngaopitakkul A (2010) Minimization of common-mode conducted noise in PWM inverter-fed AC motor drive systems using optimized passive EMI filter. Lect Notes Eng Comput Sci 2181(1):1249–1252 Jin HK, Seo SH, Hai NT et al (2015) Gateway framework for in-vehicle networks based on CAN, FlexRay, and ethernet. IEEE Trans Veh Technol 64(10):4472–4486 Kim H, Jeong S, Kim D H et al (2017) Selective harmonic elimination method of radiation noise from automotive wireless power transfer system using active rectifier. In: 2016 IEEE 25th conference on electrical performance of electronic packaging and systems (EPEPS). IEEE, 2017, pp 161–164 Kong S, Bae B, Kim DH et al (2016) Electromagnetic interference on analog-to-digital converters from high-power wireless power transfer system for automotive charger. In: IEEE international symposium on electromagnetic compatibility. IEEE, pp 370–373 Kovacevic IF, Friedli T, Musing AM et al (2013) 3-D Electromagnetic modeling of parasitics and mutual coupling in EMI filters. IEEE Trans Power Electron 29(1):135–149 Lai J, Huang X, Chen S et al (2004) EMI characterization and simulation with parasitic models for a low-Voltage high-current AC motor drive. IEEE Trans Ind Appl 40:178–185 Laour M, Tahmi R, Vollaire C (2017) Modeling and analysis of conducted and radiated emissions due to common mode current of a buck converter. IEEE Trans Electromagn Compat 59(4):1260–1267 Lee WS, Kim JH, Cho SY et al (2018) An improved wireless battery charging system. Energies 11(4):791–802 Lim H (2014) Time domain analysis of the physical wiring for a star-like topology in high-speed controller area networks. Proc Inst Mech Eng 228(12):1491–1501 Lin L (2018) Electromagnetic interference modeling and suppression in wireless charge system of electric vehicles. Beijing Institute of Technology, Beijing Lin C (2019) Handbook of Electric—vehicle design of pure electric vehicle. China Machine Press, Beijing Liu S, Liu W (2014) Progress of relevant research on electromagnetic compatibility and electromagnetic protection. High Voltage Eng 40(6):1605–1613 Liu W, Tsai C, Han T et al (2008) An embedded common-mode suppression filter for GHz differential signals using periodic defected ground plane. IEEE Microwave Wirel Compon Lett 18(4):248–250 Liu H, Wu Y, Zhang G et al (2018) Review on electrical performance and test methods of traction cables in electric vehicles. Auto Electric Parts 7:10–13

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Shiue GH, Guo WD, Lin CM et al (2006) Noise reduction using compensation capacitance for bend discontinuities of differential transmission lines. IEEE Trans Adv Packag 29(3):560–569 Song E, Cho J, Kim J et al (2010) Modeling and design optimization of a wideband passive equalizer on PCB based on near-end crosstalk and reflections for high-speed serial data transmission. IEEE Trans Electromagn Compat 52(2):410–420 Tan W, Cuellar C, Margueron X et al (2013) A high frequency equivalent circuit and parameter extraction procedure for common mode choke in the EMI filter. IEEE Trans Power Electron 28(3):1157–1166 Valerio DS, Tommaso C, Silvano C et al (2018) Assessment of the induced electric fields in a carbonfiber electrical vehicle equipped with a wireless power transfer system. Energies 11(3):684–692 Wang Q, Zhang F, Peng H et al (2015a) Permanent magnet synchronous AC motor EMI model based on vector-fitting. Trans China Electrotech Soc 30(6):77–78 Wang B, Du J, Tian X et al (2015b) Design of decoupling capacitor network based on hybrid GA. Appl Electron Technique 41(7):146–149 Wang Q, Li W, Kang J et al (2018) Electromagnetic safety evaluation and protection methods for a wireless charging system in an electric vehicle. IEEE Trans Electromagn Compat 61(6):1913– 1925 Witting T, Schuhmann R, Weiland T (2006) Model order reduction for large systems in computational electromagnetics. Linear Algebra Appl 415(2–3):499–530 Wu J, Kong D (2013) Signal integrity analysis of high-speed data acquisition card. Instrum Technique Sensor 12:93–96 Wu KB, Shiue GH, Wu RB (2005) Optimization for the locations of decoupling capacitors in suppressing the ground bounce by genetic algorithm. Piers Online 1(4):411–415 Wu S, Tsai C, Wu T et al (2009) A novel wideband common-mode suppression filter for gigahertz differential signals using coupled patterned ground structure. IEEE Trans Microw Theory Tech 57(4):848–855 Xu J, Wang S (2015) Investigating a guard trace ring to suppress the crosstalk due to a clock trace on a power electronics DSP control board. IEEE Trans Electromagn Compat 57(3):546–554 Ya Y (2016) Research on maximum power transfer and suppressing electromagnetic interference level for wireless power transfer system. Zhengzhou University, Zhengzhou Yang L, Shan CL, Xiao YJ (2012) Common and differential mode EMI noise for muti-converter system. In: IEEE PES Asia-Pacific power and energy engineering conference (APPEEC), Shanghai, China, 2012, pp 5.27–5.29 Yao M (2014) EMC Characteristics analysis and the model research of electric vehicle traction battery. Chongqing University Zeng H, Liu Z, Hou Y et al (2017) Optimization of magnetic core structure for WPT coupler. IEEE Trans Magn 53(6):1–4 Zhai L, Zhang X, Li G (2016) Effect of distributed parameters on conducted EMI in electric vehicle motor drive system. Trans Beijing Inst Technol 36(9):935–939 Zhai L, Lin X, Zhang et al (2017) The effect of distributed parameters on conducted EMI from DC-Fed motor drive systems in electric vehicles. Energies 10(1):1–17 Zhai L, Zhang T, Cao Y et al (2018a) Conducted EMI prediction and mitigation strategy based on transfer function for a high-low-voltage DC-DC converter in electric vehicle. Energies 11(5):1028–1044 Zhai L, Cao Y, Lin L, Zhang T, Kavuma S (2018b) Mitigation conducted emission strategy based on transfer function from a DC-Fed wireless charging system for electric vehicles. Energies 11(3):477–493 Zhang F (2014) Study on the conducted electromagnetic interference of electric automotive motor drive system. Chongqing University Zhang X (2016) Electromagnetic interference and suppression in power inverter system of electric vehicles. Beijing Institute of Technology Zhang Y (2016b) A new distributed modeling and decoupling method for multi-chip power delivery network. Electron Sci Technol 29(7):132–135

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Zhao L, Thrimawithana D, Madawala U et al (2019) A misalignment-tolerant series-hybrid wireless ev charging system with integrated magnetics. IEEE Trans Power Electron 34(2):1276–1285 Zheng W (2013) Study of electromagnetic compatibility based on the wireless energy transmission system. Hubei University of Technology, Hubei Zhu H, Hefner AR, Lai J (1999) Characterization of power electronics system interconnect parasitics using time domain reflectometry. IEEE Trans Power Electron 14(4):622–628 Zhu HB, Lai JS, Allen R et al (2001) Modeling-based examination of conducted EMI emissions from hard-and soft-switching PWM inverters. IEEE Trans Ind Appl 37(5):1383–1393 Zongyu QW (2013) Parameter extraction of conducted electromagnetic interference prediction model and optimisation design for a DC–DC converter system. IET Power Electron 6(7):1449– 1461

Chapter 2

Electromagnetic Compatibility Foundation of New Energy Vehicles

2.1 Overview Energy vehicles are developing towards electric, intelligent and networked. The intelligent network connected electric vehicle has put forward the requirements of high bandwidth, high real-time, high security and high reliability for the wire control efficient electronic and electrical architecture (Lin 2019). The intelligent driving system, Internet of vehicles system and electric drive system of intelligent network connected electric vehicle have the characteristics of many sensors, many antennas, many cables, complex signal types, wide working frequency band (DC, low frequency and radio frequency ultra-wideband), high transmitting power, high receiving sensitivity, complex electromagnetic frequency spectrum and overlapping working frequency bands, etc. Electric vehicles make the electromagnetic environment extremely bad, which brings new problems of electromagnetic compatibility (EMC) and electromagnetic interference (EMI), as shown in Fig. 2.1. Due to the electromagnetic interference caused by electromagnetic coupling between electronic and electrical system hardware, the system will have serious failure and function degradation. Automotive functional safety standard ISO 26262 takes electromagnetic compatibility as the key element of functional safety evaluation of electronic and electrical system(Gong 2019). Therefore, EMC safety has become an important scientific issue that affects the functional safety of intelligent networked electric vehicles. In this complex electromagnetic environment, there are mutual interference between electric vehicles and electronic equipment in other environments, and even serious self-disturbance and mutual interference between electric vehicles and their own equipment, as well as the electromagnetic environment incompatibility caused by civil communication and radio frequency equipment, which have a great impact on the normal operation of electric vehicles. The electromagnetic compatibility between electric vehicle equipment and systems, between vehicles, between vehicles and the environment has become an important factor restricting the development of electric vehicles and restricting the effectiveness of electric vehicle electronic equipment. At

© China Machine Press 2021 L. Zhai, Electromagnetic Compatibility of Electric Vehicle, Key Technologies on New Energy Vehicles, https://doi.org/10.1007/978-981-33-6165-2_2

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2 Electromagnetic Compatibility Foundation of New Energy Vehicles

Fig. 2.1 EMC problems existing in new energy vehicles

the same time, it also reflects the research on electromagnetic compatibility of electric vehicles, including not only the electromagnetic problems of on-board electronic equipment, but also the impact of vehicle electromagnetic system on external electromagnetic environment. To assess the capacity of electric vehicles, it is necessary to predict and analyze the electromagnetic compatibility between equipment and systems in the design, manufacture, test and commissioning, and actual operation and application of electric vehicles, and timely take practical and effective protective measures. For the scientific of electric vehicle design scheme and the effectiveness of operation process, and to ensure the efficiency of electronic equipment, the existence of new energy vehicles is shown in Fig. 2.1. The EMC problem is extremely

2.1 Overview

37

important. Therefore, the importance and complexity of EMC of intelligent network connected electric vehicle is increasingly prominent. All kinds of electronic equipment on electric vehicles form an integrated electromagnetic system through mutual coupling. The operation safety of electric vehicles depends not only on the electromagnetic performance of single vehicle electronic equipment or system, but also on the overall electromagnetic performance of electric vehicles. If the EMC problem between electric vehicle equipment and subsystems can’t be solved reasonably, the receiver noise level will increase, the data bit error rate will increase, and the command communication will not be smooth, which will not only cause the electromagnetic interference characteristics between the electric vehicle and other surrounding vehicles, but also cause the situation of “electronic equipment self-disturbance and self-chaos” inside the vehicle.

2.2 EMC of New Energy Vehicle 2.2.1 Electrical Infrastructure Electric vehicle is a complete electromagnetic system, which can be divided into high-voltage electrical system and low-voltage electrical system according to power supply level, and wired equipment and wireless equipment according to whether there are connecting cables, as shown in Fig. 2.2. The high-voltage system of electric vehicles is usually provided with high-voltage DC power supply by power battery or generator set. The voltage level is 60–1500 V, which is usually shielded. Low voltage system power supply is usually 12 V and 24 V, usually unshielded. According to the working characteristics of the equipment, the high voltage components are divided into DC-AC inverter, DC-DC converter and AC-DC rectifier. The driving motor system belongs to DC-AC inverter, and the power supply voltage is DC 36 V ~ 750 V. High voltage DC-DC converter includes bidirectional and unidirectional converters. The charging system belongs to AC-DC rectifier, including vehicle charger, charging pile (station) and wireless charging system. In addition, the high voltage system components include power batteries and electric heaters. The communication between high voltage and low voltage equipment mainly adopts vehicle can bus. With the development of intelligent networking technology, Ethernet and Internet of vehicles are gradually applied. All kinds of electric and electronic equipment of electric vehicle are arranged in relatively independent functional area of vehicle, and become the basic working unit of electric vehicle system, such as automobile radar, communication system, sensor, air conditioning, monitoring module, driving motor, power conversion module, etc. Besides wireless or wired information exchange, there are also some electrical work units contact, on the basis of meeting specific functional requirements, to constitute the executive system of electric vehicles. In addition, there are some electromagnetic coupling relations in various power and electronic equipment, which are mainly formed by

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2 Electromagnetic Compatibility Foundation of New Energy Vehicles

Fig. 2.2 Electrical architecture of new energy vehicles

the transmission of cables and connectors and the emission and reception of space electromagnetic radiation energy, and the coupling energy changes with the change of unit equipment spacing and operating environment conditions. In order to analyze and predict the electromagnetic compatibility of the whole electric vehicle, each work unit is defined as the “subsystem” in the EMC analysis between systems. Combined with the electric vehicle power and electronic equipment architecture, the electromagnetic characteristics of the vehicle power and electronic equipment are analyzed, modeled and predicted.

2.2.2 Electromagnetic Compatibility The research on electromagnetic compatibility characteristics of electric vehicle mainly includes vehicle radiation characteristics, electromagnetic sensitivity and anti-electromagnetic interference characteristics. Combined with the electric vehicle power and electronic equipment architecture, through the analysis and modeling of the electromagnetic characteristics of power and electronic equipment, the electromagnetic model of the whole vehicle electromagnetic system is established, and based on the vehicle body electromagnetic model, the radiation characteristics and electricity of the vehicle electromagnetic system are given. According to the characteristics of magnetic interference, electromagnetic sensitivity and anti-interference

2.2 EMC of New Energy Vehicle

39

characteristics of electromagnetic system, it is the development trend to design electromagnetic compatibility simulation and prediction software platform of electric vehicle system based on electromagnetic simulation software. (1) EMC problem of high voltage power system integration The high voltage system of new energy vehicles mainly includes: drive motor system, DC-DC converter, electric air conditioner, electric power steering motor controller motor, vehicle charging system, etc., as shown in Fig. 2.3. Especially with the rapid development of new energy bus, in order to meet the requirements of high-power density, lightweight, high efficiency and high reliability of electric drive system, high voltage power integrated controller has become the main development trend. EMC problem and thermal management problem are prominent problems of high voltage power integrated controller. High voltage power integrated controller adopts high current and high voltage power electronic devices (such as IGBT). Its fast on-off produces high-power electromagnetic interference, which not only affects the electromagnetic compatibility of the electric drive system, but also leads to the electromagnetic emission level of the new energy vehicle higher than that of the traditional vehicle. High power electromagnetic interference not only affects its own function, but also affects other vehicle parts and even the normal operation of electrical equipment in the surrounding environment. For example, electromagnetic interference generated by motor drive system is not only coupled to DC-DC converter electric air conditioner, electric power steering motor controller motor, vehicle charging system through high-voltage cable, but also coupled to low-voltage power supply system through high-voltage and low-voltage

Fig. 2.3 High-voltage power integrated controller

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2 Electromagnetic Compatibility Foundation of New Energy Vehicles

wiring harness, which affects the normal operation of low-voltage equipment such as VCU and BMS. In addition, the electromagnetic interference generated by the highvoltage system can also interfere with on-board wireless equipment (such as sensors, antennas, etc.) through space coupling, and can also interfere with the external radio receiving equipment. (2) EMC problems of intelligent network connected electric vehicle The working frequency of intelligent network connected electric vehicle is different, from DC signal to 400 Hz, it is mainly power control signal; digital signal is generally in hundreds of Kbps; WIFI and GPS are in L and S band; radar signal is in ultrasonic frequency, millimeter wave band (30–77 GHz) or even optical band, and electromagnetic noise can cover the whole frequency. Therefore, the research and analysis of UWB EMI noise mechanism and EMC optimization design should be combined with UWB noise environment. In the intelligent network connected electric vehicle, the high-voltage power system will produce electromagnetic interference to ADAS system (intelligent sensor, electronic controller, actuator and other sensitive equipment) through the electromagnetic coupling channel of high-voltage cable, vehicle Ethernet and can bus network, as shown in Fig. 2.4. The electric vehicle equipped with a new generation of vehicle safety system, also known as advanced driver assistant system (ADAS), can be completed by the driver and the auxiliary driving system. In order to realize the ADAS function, there are several kinds of sensors used to realize the automatic driving of electric vehicles at present: image sensor, laser radar, millimeter wave radar, ultrasonic radar and biological sensor. Intelligent sensors and vehicle wireless communication equipment will also produce radio frequency electromagnetic interference, which will affect the safety of traction, braking and steering functions. (3) EMC regulations for new energy vehicles For the EMC of new energy vehicles, EMC regulations mainly include vehicle electromagnetic radiation standards and electromagnetic emission and electromagnetic sensitivity standards of high-voltage components. In addition, new energy vehicles Fig. 2.4 The impact of the high-voltage system of intelligent connected electric vehicles

2.2 EMC of New Energy Vehicle

(a) Position of the electric field antenna

(b) Radial direction and position of the magnetic field loop antenna 

41

(c) Transverse direction and position of the magnetic field loop antenna

Fig. 2.5 Measurement layout for EMI from EV

also need to meet the EMC standards for road vehicles. The following focuses on the new EMC standards for new energy vehicles only. (1) Vehicle radiated emission—Limits for electromagnetic field emission intensity of electric vehicles (150 kHz–30 MHz) According to the Chinese standard GB/T 18387-2017, the test site is a shielding room equipped with absorbing materials or an outdoor test site meeting the requirements of the standard. Electric field antenna and magnetic field antenna are used to measure the electric field intensity and magnetic field intensity respectively at four locations outside the vehicle. The antenna is 3 M ± 0.3 m away from the nearest part of the vehicle, It is located on the transverse and longitudinal centerline of the vehicle, and the measurement arrangement is shown in Fig. 2.5. When the vehicle is fully loaded at low speed (16 km/h) and high speed (70 km/h), peak electric field scanning and magnetic field peak scanning shall be conducted on the maximum emission side of the vehicle, and the emission limit shall not exceed the specified limit value (National Technical Committee of Auto Standardization 2017). (2) Conducted and radiated emissions from high voltage components In order to protect the on-board receiver from the conducted and radiated emission generated by high-voltage components in the vehicle, “Vehicles, boats and internal combustion engines—Radio disturbance characteristics—Limits and methods of measurement for the protection of on-board receivers” (CISPR 25) specifies the test method of high voltage power supply system with internal shielding for electric and hybrid electric vehicles, including conducted emission measurement of highvoltage components and modules on shielded high-voltage (HV) power lines, radiation emission measurement methods of high-voltage components and modules, and high-voltage and low-voltage coupling test methods. The test procedure and limit value are the preventive control of vehicle radiation emission. However, the test of parts and components does not represent the vehicle test, and the exact relationship between the two depends on the installation position of the parts, the length and layout of the wiring harness, the grounding position and the antenna position (CSBTS/TC79 2018).

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In the frequency range of 150 kHz–108 MHz, the conducted disturbance voltage and current of high-voltage components such as motor controller and DC-DC converter on the shielded high-voltage (HV) power line are measured. Taking the motor controller as an example, the test layout of conducted emission voltage method is shown in Fig. 2.6. The radiation emission test frequency range is 150 kHz–2.5 GHz, and the radiation emission measurement arrangement of motor system is taken as an example, as shown in Fig. 2.7. (3) Transient immunity test along shielded high voltage line ISO/TC 22/sc32/WG 3 (Sub Committee on Electrical and electronic components and general systems) formulated ISO 7637-4 “Road Vehicles—Electrical disturbance by conduction and coupling—Part 4: Electrical transient conduction along shielded high voltage supply lines only”, which tests and evaluates the electrical transient conduction of on-board electric drive systems and high-voltage components on new energy passenger vehicles and commercial vehicles (ISO/TC 22/SC 2020). Suitable for DC60V-1500 V. Three kinds of internal pulse generated by DC high voltage module are given: voltage ripple (pulse A), sinusoidal pulse (pulse B) and low frequency sinusoidal disturbance (pulse C). The waveforms and parameters are shown in Fig. 2.8. The transient disturbance pulse is produced on the high voltage power supply cable when various high voltage load equipment is turned on and off. In order to test the transient voltage immunity of high voltage equipment (voltage ripple 3–300 kHz), pulse A shall be applied between HV and HV− HV cables, and between HV+ and ground, HV+ and ground respectively. Pulse B (sinusoidal pulse 1–10 MHz) and pulse C (low-frequency sinusoidal pulse 3–300 kHz) respectively simulate high-frequency oscillation and low-frequency oscillation signals generated by fast switching of power devices, and are applied between HV+ and ground, HV+ and ground respectively. In addition, Audi, BMW, Mercedes Benz, Porsche, Volkswagen and other German United Automobile Enterprise Company standard LV 123 “Electrical Characteristics and Electrical Safety of High-Voltage Components in Road Vehicles requirements and tests” specifies the electrical performance and safety specification of new energy vehicles in detail, which is mainly divided into three parts: high voltage power supply characteristic test, high voltage safety regulation test and high voltage accessories test. The test method of the high-voltage system parts, such as high-voltage battery system, motor inverter, electric air-conditioning compressor, power transmission oil pressure pump, DC/DC high-low voltage converter, vehicle charger, etc., are described. The test waveform is shown in Fig. 2.9. (4) Transient immunity test along shielded high voltage line The Chinese standard GB/T36282 “Electromagnetic compatibility requirements and test methods of drive motor system for electric vehicles” covers the electromagnetic radiation emission (in the frequency range of 30 MHz–1000 MHz), including broadband electromagnetic radiation and narrow band electromagnetic radiation) and electromagnetic immunity (electromagnetic radiation immunity in the frequency range

2.2 EMC of New Energy Vehicle

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1. EUT

16. Power line filter

2. Ground plane 3. Low relative permittivity (εrİ1.4) thickness 50mm (a nonconductive support can be used for the electric motor) 4. 50Ω load

17. Fiber optic feed through

5. LV harness

20. Measuring equipment

6. HV lines (HV+, HV-)

21. High quality coaxial cable e.g. double shielded (50Ω)

18. Bulk head connector 19. Stimulating and monitoring system

7. LV load simulator

22. Optical fiber

8. Impedance matching network (optional)

23. Ground straps

9. LV AN

24. Electric motor

10. HV AN

25. Three phase motor supply lines 26. Mechanical connection (e.g. nonconductive)

11. LV supply lines 12. HV supply lines 13. LV power supply 12V/24V/48V(Should be placed on the plane)

27. Filtered mechanical bearing

14. Additional shielding box 15. HV power supply (should be shielded if placed inside the shielded enclosure)

29. Shielded enlosure

Fig. 2.6 Conducted emission test layout of motor controller

28. Brake or propulsion motor

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2 Electromagnetic Compatibility Foundation of New Energy Vehicles

1. EUT

14. Additional shielded box

2. Reference ground plane

15. HV power supply˄should be shielded if placed inside ALSE˅

3. Low relative permittivity support˄εr≤1.4˅thickness 50mm

16. Power line filter

4. Ground straps

17. Fibre optic feed through

5. LV harness 6. HV lines˄HV+ǃHV-˅

18. Bulk head connector

7. LV load simulator 8. Impedance matching network˄optional˅

20. Measuring instrument 21. High quality coaxial cable e.g. double shielded (50
)

9. LV AN

22. Optical fibre

10. HV AN

23. Biconical antenna

19. Stimulating and monitoring system

11.LV supply lines

24. RF absorber material

12.HV supply lines

25. 50
load

13.LV power supply 12V/24V/48V˄should be placed on the reference ground plane˅

26. Motor three-phase line

27. Mechanical connection (e.g. non-conductive)

28. Filter bearing

29. Brake or propulsion motor

Fig. 2.7 Machine system radiation emission-double cone antenna

2.2 EMC of New Energy Vehicle

Fig. 2.8 Electrical transient test pulse along the shielded voltage line

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2 Electromagnetic Compatibility Foundation of New Energy Vehicles 8 >9@

U [V]

E 20V/ms



F

D U:20V/ms

2s tr1

2s

th1

tr1

Ttest=10 minutes

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Fig. 2.9 Electrical characteristics and electrical safety test waveforms of high-voltage components of road vehicles

of 20 MHz–2000 MHz) (National Technical Committee of Auto Standardization 2018). The transient immunity and electrostatic discharge immunity of low voltage power line are tested, and the test methods and limits are specified. (5) Measurement of vehicle electromagnetic field exposure relative to human body The Chinese standard GB/T 37130-2018 specifies the low-frequency magnetic field emission of the vehicle environment in which the human body is located, with the frequency range of 10 Hz–400 kHz. Vehicle measurement can be carried out on indoor dynamometer and outdoor flat and dry road (CSBTS/TC79 2018). The measuring positions of static state and running state of M-class passenger cars and commercial vehicles (buses) are shown in Fig. 2.10. In the charging state, the charging interface

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a) Passenger cars Front passenger seat

Central passenger seat

Central passenger station Front passenger station

Rear passenger station

Rear passenger station

Driver position

Footwell seat

Central passenger seat

Rear passenger seat

b) Commercial vehicles Fig. 2.10 Example of the measurement position of the static state and driving state of M class passenger cars and commercial vehicles (buses)

area is the contact area of the measuring probe, and the charging cable within 0.5 m after the charging interface.

2.3 Theoretical Basis of Electromagnetic Compatibility 2.3.1 Electromagnetic Interference Source The electromagnetic interference sources of electric vehicles can be roughly divided into three categories (Zhai 2012), namely on-board interference sources, natural interference sources and man-made interference sources, as shown in Fig. 2.11.

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Fig. 2.11 Classification of EMC sources

The natural interference source refers to the electromagnetic interference caused by natural phenomena; the artificial interference source refers to the electromagnetic interference generated by the external artificial devices of the automobile. The following is mainly from the new energy vehicle on-board interference sources to analyze. The on-board interference sources mainly refer to the electromagnetic interference generated by various electrical systems on the vehicle. The on-board interference sources mainly include motor drive system, power battery system, power electronic devices (DC-DC converter, AC-DC rectifier, DC-AC inverter) and electric auxiliary system. The fast transient of voltage and current will produce radiation and noise, especially the fast rectification, motor starting, and high voltage radiation of power electronic devices will cause conduction and radiation interference with high field strength. (1) Interference source of high voltage system The main reason of EMI in high voltage system is the fast turn-on and off power devices, which produce high current change di/dt and voltage change du/dt. High voltage power devices mainly include drive motor controller IGBT, high and low voltage DC-DC converter and MOSFET of charger. The current level of power devices: 200 A to 900 A; voltage level 400 V to 1200 V; switching frequency 10 kHz to 100 kHz. IGBT trapezoidal wave and its spectrum are shown in Fig. 2.12. In Chap. 4, the EMI generated by IGBT and MOSFET is introduced in detail. (2) Interference source of low voltage system Electromagnetic interference sources of low-voltage controllers such as VCU and BMS are mainly digital chip high-speed switch signal, low-voltage power dc-dc chip and clock signal. The waveform and spectrum of clock signal are shown in Fig. 2.13.

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49

Fig. 2.12 a IGBT trapezoidal waveform and spectrum distribution trapezoidal wave, b Spectrum distribution

Fig. 2.13 Clock signal waveform and spectrum distribution

In addition, the power integrity and signal integrity of the low voltage controller PCB must also be paid attention to. The relevant contents will be introduced in Chaps. 6 and 7.

2.3.2 Coupling Path There are two transmission ways of vehicle electromagnetic interference: one is conducted emission, and electromagnetic interference noise is transmitted through high and low voltage cable bundles and metal connectors, such as metal chassis and metal connectors; the other is radiation transmission, and electromagnetic interference noise is transmitted through space in two ways: electromagnetic induction and electromagnetic radiation. Therefore, electromagnetic interference coupling can be divided into conduction coupling and radiation coupling. According to the electromagnetic interference mode, it can be divided into differential mode interference and common mode interference.

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(1) Conduction coupling There are many vehicles wiring harness, so the cable is an important carrier of conductive coupling, which is usually the longest part of the system, and can even be equivalent to an antenna to receive and transmit electromagnetic noise. Because of the assumption of the specific wave length of the cable, the coupling between circuits can be expressed by the lumped capacitance and inductance between conductors. Then we can analyze the circuit with the common network theory. Three types of coupling are considered below. Conduction coupling can be divided into common impedance coupling, capacitive coupling and inductive coupling. Common impedance coupling Multiple electronic and electrical components in the vehicle use the same vehicle lowvoltage power supply or high-voltage power battery for power supply. The internal impedance of the power supply and the impedance of the power line they share become the common impedance of these components. If multiple components are grounded with the same ground wire, the impedance of the ground wire will also become the common impedance of these components. Circuit common coupling is the most common conduction coupling mode, in which at least two coupled circuits exist. The circuit shown in Fig. 2.14 is a typical conduction coupling mode through grounding common impedance. The grounding point and grounding mode of the vehicle will affect the circuit coupling. (2) Capacitive coupling Capacitive coupling usually exists between cable bundles, between cables and metal car bodies, and between controller metal chassis installed in different spaces. Capacitive coupling is the result of electric field interaction between circuits. Figure 2.15a) shows the capacitive coupling between a pair of parallel conductors, and its equivalent circuit is shown in Fig. 2.15b). The voltage U 1 on conductor 1 is the interference source, and conductor 2 is regarded as the affected circuit or receiver. The noise voltage U 2 generated between conductor 2 and ground can be expressed as follows: Fig. 2.14 Common ground coupling impedance circuit

2.3 Theoretical Basis of Electromagnetic Compatibility

(a) Coupling model

51

(b) Equivalent circuit

Fig. 2.15 Capacitive coupling between two conductors

U2 = jω RG2 CU1

(2.1)

Formula 2.1 shows that the noise voltage U 2 is directly proportional to the frequency ω of the noise source, the resistance RG2 of the affected circuit to ground, the mutual capacitance C between conductor 1 and conductor 2, and the voltage U 1 . Assuming that the voltage and frequency of the noise source cannot be changed, the receiver circuit can operate at a lower resistance level, or the mutual capacitance C can be reduced. The capacitance C can be reduced by properly changing the direction of the conductor, shielding or physically separating the conductor. (3) Inductive coupling There is usually inductive coupling between cable bundles, especially between high and low voltage harness. Inductive coupling is that the conductor with alternating current will produce alternating magnetic field around it, and then produce induced electromotive force in the surrounding closed circuit. The inductive magnetic coupling between the two circuits is shown in Fig. 2.16. I 1 is the current in the interference circuit, M is the mutual inductance between the two circuits, and the noise voltage U N is

Fig. 2.16 Inductive coupling between two circuits

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VN = jωM I1 = M

di 1 dl

(2.2)

If the current flows in the twisted pair rather than through the ground plane, M can be reduced by physical separation of the circuit or by twisting the power cord. The area of the receiver circuit can be reduced by placing the conductor closer to the ground plane (if the return current is through the ground plane) or by using two conductors twisted together (if the return current is on a pair of conductors rather than on the ground plane) to reduce the mutual inductance M. (2) Radiation coupling The combination of electric and magnetic fields is called electromagnetic coupling or radiation. For the near-field analysis, we usually consider the electric field and magnetic field respectively, while for the far-field analysis, we consider the electromagnetic field. When the sensitive part is in the far-field region of the electromagnetic disturbance source, the electromagnetic disturbance is coupled to the sensitive part in the form of space electromagnetic wave, which is called radiation coupling. 1. Radiation coupling between antenna and antenna Antenna coupling is receiving electromagnetic wave through antenna. All kinds of antennas are the most effective equipment for electromagnetic radiation, and wiring, structural parts, components and components will play the role of transmitting antenna and receiving antenna as long as they meet the radiation conditions. In practical engineering, there are a lot of antenna coupling in vehicles. For example, long power lines, signal lines, control lines, input and output leads have antenna effect, which can receive electromagnetic interference and form antenna radiation coupling. 2. Coupling of field and cable Many electromagnetic interferences occurs through the coupling path of electromagnetic field to cable conductor, and its coupling mechanism is relatively complex, and the interference propagation path is also relatively hidden, including inductive coupling, high-frequency radiation field, coupling of slot leakage field to conductor, etc. multiple coupling combinations of field to wire may exist as shown in Fig. 2.17. 3. Conductor to conductor coupling The coupling interference between conductors in the cable is one of the most common interference coupling modes, which is a typical interference mode for EMC analysis between internal equipment of the system. The statistical results of various electromagnetic interference coupling paths are shown in Fig. 2.18. The coupling between conductors is the most serious, accounting for 60%, which is one of the problems that must be taken seriously in EMC design.

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Fig. 2.17 Various combinations of field to line coupling

Fig. 2.18 Statistical results of coupling paths of electromagnetic interference

2.4 EMC Design The precise numerical simulation of electromagnetic coupling between electronic equipment or systems of electric vehicles requires strict simulation platform due to the huge amount of unknown (up to several billion orders of magnitude), and the simulation time is long, so it is difficult to realize the optimization experiment. Therefore, this paper studies and discusses a rapid simulation method, prediction software and evaluation standard of electromagnetic compatibility system of general electric vehicle, so as to achieve good electromagnetic performance of electric vehicle Compatibility has important practical significance. As shown in Fig. 2.19, according to the requirements of vehicle EMC, the framework of vehicle EMC design index Fig. 2.19 EMC design content

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system is constructed, the index quantitative combination model is established, the membership function of vehicle system level EMC measurement index is established, the multi-objective optimization design method is proposed to solve the problem of index quantitative distribution, and the theory of vehicle system level EMC quantitative design is established.

2.4.1 High Performance Electrical and Electronic Architecture Modeling High performance electronic and electrical architecture modeling of intelligent electric vehicle mainly includes electrical function modeling and electromagnetic interference noise modeling. The structure of behavioral EMC and EMI noise model of vehicle system is shown in Fig. 2.20. The geometric model of vehicle electromagnetic compatibility, emission element model of vehicle system behavior level interference source, electromagnetic coupling element model, sensitive element model, EMI noise source multi factor model are established, and the EMI noise prediction method based on the model is established to reveal the formation mechanism of vehicle EMI noise. According to the system function and EMC requirements, the system is decomposed by layers, and the subsystem model, system model and vehicle three-dimensional model are established, and the electromagnetic interference noise

Fig. 2.20 Behavioral electromagnetic compatibility modeling of vehicle system

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Fig. 2.21 Research method of electromagnetic interference mechanism

of subsystem, system, vehicle interior and exterior is predicted, and the EMC design method and evaluation method of the system are constructed. The research method of EMI mechanism is shown in Fig. 2.21.

2.4.2 EMC Modeling and EMC Design As shown in Fig. 2.22, based on the measurement results of electromagnetic emission, the electromagnetic emission elements of typical interference sources are mathematically characterized, and the equivalent circuit model of electromagnetic interference source based on Ibis and spice is established. By measuring the impedance characteristic parameters and transmission characteristic parameters (S parameters) at both ends of the harness, the cable bundle model and electromagnetic coupling element model are established by using transmission line theory and port network theory. By measuring the S parameters and transient characteristics of typical sensitive devices, the sensitive element model is established by using port network theory. By cascading the interference source model, electromagnetic coupling model and sensitive equipment model, a multi-level model of system level collaborative analysis, design and evaluation is obtained. (1) Behavioral EMI noise modeling of vehicle system The mathematical representation of behavior level interference source electromagnetic emission element model, electromagnetic coupling element model, conducted electromagnetic sensitive element model, radiated electromagnetic sensitive element model and external electromagnetic environment EMI model of intelligent electric vehicle system is described. The electromagnetic interference relationship among key systems such as vehicle automatic driving system, vehicular communication system, high voltage power

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Fig. 2.22 The electromagnetic compatibility modeling and electromagnetic compatibility design scheme of the whole vehicle

system and other key systems is established. According to the characteristics of launch and sensitivity of the equipment, the electromagnetic compatibility classification and classification method of vehicle system is put forward. The system behavioral level interference source model, coupling channel model and sensitive device model are established respectively. The electromagnetic emission model of high-voltage power devices, high-frequency digital circuits, communication antennas, gateways and other equipment is established; the transmission line model of high-voltage and low-voltage wire harness, bus communication harness, electromagnetic coupling model between cable bundles, electromagnetic radiation model of cable bundle are established, and the system behavior of sensitive equipment such as ADAs sensor and communication receiving equipment is established. This paper proposes a new interface between electromagnetic model of on-board equipment and radio wave propagation loss model of vehicle body, and studies optimization method of vehicle body model and radio wave propagation loss model. The formation mechanism of electromagnetic interference noise at system level is revealed. (2) Prediction model and evaluation method of vehicle system level EMI noise The main research contents are as follows: EMI noise prediction model at system level, EMC measurement index system architecture, safety measurement index function, correlation degree mathematical modeling of EMC influencing factors, quantitative evaluation method of vehicle electromagnetic compatibility, etc. The electromagnetic field distribution inside the vehicle, the current and electromagnetic field distribution on the surface of the vehicle shell and the electromagnetic radiation characteristics outside the vehicle are studied under the typical

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environment. The emission characteristics, electromagnetic interference characteristics, electromagnetic sensitivity and electromagnetic interference prediction models of vehicle electromagnetic system are studied. The prediction models of electromagnetic interference and noise generated by the transmission and radiation of vehicle cable bundles and antenna radiations at the ports of sensitive devices such as ADAS sensors, communication receiving devices and low-voltage controllers are studied. The prediction model for the influence of electromagnetic radiation of high-voltage system on human safety inside and outside the vehicle; the prediction model for predicting the electromagnetic interference noise generated by electromagnetic radiation of the whole vehicle at the ports of V2I, V2V and V2G network equipment; the prediction model for predicting the electromagnetic interference noise generated by the external radiation sources on the ports of sensitive equipment in the vehicle. Through the simulation and prediction, the frequency points or frequency bands which affect the intensive EMI noise inside and outside the vehicle are obtained. The quantitative model, correlation degree model and evaluation model of EMC influencing factors are established. The framework of EMC measurement index system of intelligent electric vehicle is proposed. The quantitative model of EMC evaluation index, the definition of safety measurement index and the function of safety measurement index are established, and the quantitative evaluation method of vehicle electromagnetic compatibility is proposed. The EMI noise prediction method based on network resonance tracing is proposed, and a high-precision broadband EMI prediction and safety assessment expert system is established. (3) Layered optimization design method for whole cycle EMC of vehicle and EMI suppression method for dynamic tracking measurement The EMI noise suppression method of intelligent electric vehicle (EV) based on layered, frequency band and region are established. The EMI suppression method is designed from three aspects: reducing the emission of electromagnetic interference source, cutting off the coupling channel and improving the immunity of sensitive equipment. Based on the hierarchical simulation model of vehicle high-voltage power supply network, low-voltage power supply network, Ethernet communication network and can bus communication network system, the electromagnetic radiation, electromagnetic noise, crosstalk value, cost and weight are minimized under the multi constraint conditions of air electromagnetic shielding and less than sensitive threshold Light is multi-objective optimization. Through top-down index allocation and quantitative design, the optimal layout design of high-voltage and low-voltage power supply network, Ethernet cable bundle, sensor and antenna are completed, and the optimal quantitative design indexes such as cable bundle topology, shielding scheme and grounding layout are obtained. Combined with the influence of vehicle structure and material medium characteristics on electromagnetic field strength distribution, the optimization results of vehicle EMC index are obtained.

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2.5 EMI Suppression Measures 2.5.1 EMI Suppression Technology (1) Filtering technology In the field of electromagnetic compatibility, filtering is a method or technology to extract useful signal components from the signal mixed with noise or interference, in order to reduce electromagnetic noise. The filter with filtering function can attenuate the transmission energy of a certain frequency range very little, making the energy pass through easily; but it has a great attenuation to the transmission energy of another frequency range, thus suppressing the energy transmission. EMI filter is divided into differential mode filter and common mode filter, which can be divided into the following types: (1) Reflective filter Reflective filters usually consist of reactance elements such as inductors and capacitors (ideally, these elements are lossless) to provide low series impedance and high shunt impedance in the passband of the filter, while providing large series impedance and small shunt impedance in the stopband of the filter. This kind of filter does not consume energy but reflects the energy of unnecessary frequency components back to the signal source to achieve the purpose of suppression. There are four kinds of filters: low pass filter, high pass filter, band-pass filter and band stop filter. (2) Absorption filter The principle of absorption filter is to put the energy loss of unwanted interference frequency components in the filter (convert it into heat energy), instead of reflecting. Therefore, this kind of filter is also called lossy filter, including active filter. (3) Power line EMI filter The power line filter of electric vehicle electronic equipment is divided into high voltage filter and low voltage filter. When selecting and using power line EMI filter, the most important characteristic parameters are rated voltage, rated current, insertion loss, leakage current, impedance matching, working environment conditions (temperature, etc.), as well as volume, quality and reliability. Power supply EMI filters for motor controllers, high and low voltage DC-DC converters and wireless chargers are described in Chaps. 3, 4 and 5. (4) Signal line EMI filter The main function of signal line filter is to solve the problem of space electromagnetic interference, such as the equipment radiates strong electromagnetic interference to space, or the equipment is sensitive to the electromagnetic interference in space. The

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coupling between the signal cable and the power line cable leads to the phenomenon that the conducted emission of the power line exceeds the standard at high frequency, which is caused by the high frequency interference on the signal line coupling to the power line through space. This phenomenon occurs because the signal cable is equivalent to a very efficient radiation and receiving antenna. (2) Shielding technology Electromagnetic shielding is the metal isolation between two space areas to control the induction and radiation of electric field and magnetic field from one area to another. From the point of view of the whole system, shielding the noise source is more effective than shielding the receiver. (1) Electrostatic shielding After the shield is grounded, the interference current flows into the earth conductor cavity through the outer layer of the shield. When there is no other charged body, there is no charge inside the conductor and on the inner surface of the conductor, and the charge is only distributed on the outer surface of the conductor. (2) Electromagnetic shielding A method of near-field electrical shielding is to add a well-grounded metal plate between the inductive source and the inductor to short-circuit the parasitic capacitance of the induction source to ground. By suppressing the parasitic capacitance coupling, the purpose of electric field shielding can be achieved. In the far field, according to Maxwell’s equation, the direction of electric field and magnetic field is perpendicular to each other, but the phase is the same. In the form of electromagnetic wave, energy radiates to the surrounding in space, so it is necessary to design a shielding body to shield the electromagnetic wave. (3) Magnetic field shielding ➀ Static magnetic field: The magnetic field produced by electromagnet or DC coil distributes magnetic line or flux in space. The magnetic line of force mainly concentrates on the magnetic circuit with low magnetoresistance (high permeability). The shielding of magnetic field mainly uses materials with high permeability, such as iron, nickel steel, etc. the magnetic line of force will be “closed” in the shielding body to play the role of magnetic shielding. ➁ Low frequency alternating magnetic field: The principle of magnetic shielding is the same as that of static magnetic shielding. High magnetic conductivity material is used as shielding body to confine magnetic field in shielding material, such as ferromagnetic material. ➂ High frequency magnetic field shielding: The anti-magnetic field generated by the eddy current induced on the shielding shell plays the role of repelling the original magnetic field. The larger the eddy current, the better the shielding effect. Good conductor materials should be selected, such as copper, aluminum or copper

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silver plating. The higher the frequency, the better the magnetic shielding effect. In addition, due to the skin effect, the eddy current only flows on the surface of the material, so only a thin layer of metal material is needed to shield the high-frequency magnetic field. ➃ Alternating electromagnetic field shielding: Generally, the material with high conductivity is used as the shielding body, and the shielding body is grounded. The expression of electromagnetic shielding is as follows: S = 20 lg

E0 dB E1

(2.3)

S = 20 lg

H0 dB H1

(2.4)

where E 0 and E 1 are the electric field strength before and after shielding, and H 0 and H 1 are the magnetic field strength before and after shielding respectively. (3) Grounding Grounding is one of the main methods to reduce unnecessary noise and form safety system. The correct use of grounding technology can solve most of the noise problems. A good grounding system can not only ensure safety, but also prevent unnecessary electromagnetic interference and emission without increasing product cost. Grounding will only affect the common mode noise, not only to suppress the 50/60 Hz frequency and its harmonics, but also to reduce the high frequency common mode noise. (1) Safely The purpose of safe grounding is to make the equipment and the earth have a low impedance current path, to ensure personal safety and equipment safety. The effectiveness of grounding mainly depends on the grounding resistance. The smaller the resistance, the better. The size of grounding resistance is related to grounding device and environmental conditions. (2) Signal ground According to the signal or power return path, signal ground is called “signal ground” or “power ground” to define the type of current they carry and distinguish them from “safe ground”, which usually does not carry current. There are single point grounding, multi-point grounding and hybrid grounding. ➀ Single point grounding. The ground wire of all circuits connected to the same point of common ground wire is called single point grounding, which can be divided into two categories: series single point grounding and parallel single point grounding. Single point grounding is used for low frequency circuits, from DC to about 20 kHz.

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By controlling the grounding topology, the grounding current flows to the desired place. ➁ Multi point grounding. Multipoint grounding refers to that all points in the equipment (or system) that need to be grounded are directly connected to the nearest grounding plane (nearby grounding), which can make the length of grounding wire the shortest. Multipoint grounding is used for high frequency (above 100 kHz) and digital circuits. The multi-point grounding system minimizes the grounding noise voltage. However, the problem of common impedance coupling is easy to occur in multi-point grounding. Increasing the thickness of PCB ground plane has no effect on its high frequency impedance. It is the inductance rather than the grounding resistance that determines the grounding impedance. Due to skin effect, high frequency current only flows on the surface of the ground plane. Any printed circuit board containing high frequency or digital logic circuit must have good low inductance grounding. The ground can be a ground plane or a grounding grid on a double-sided plate. The ground plane provides a low inductance loop for the signal current and allows the use of constant impedance transmission lines for signal interconnection. PCB multi-point grounding does not mean that the power supply needs multi-point grounding. ➂ Hybrid grounding The so-called hybrid grounding requires designers to analyze the work of each part of the system, and only directly ground those points that need to be grounded nearby, while the other points are grounded by single point. When the signal frequency covers a wide frequency range, hybrid grounding may be a solution. Video signals are a good example; signal frequencies can range from 30 Hz to tens of megahertz. Hybrid grounding is a kind of system grounding structure with different grounding modes at different frequencies. It is used as single point grounding at low frequency and multi-point grounding at high frequency. Figure 2.23 shows a common hybrid grounding system in which points requiring high-frequency grounding are connected to the ground plane through a bypass capacitor. Fig. 2.23 Mixed grounding system Ground current

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2.5.2 EMI Suppression of Key Components (1) EMI source components It can be seen from Fig. 2.24 that the vehicle has both switching power supply equipment such as motor system, DC/DC converter and vehicle charger, as well as instrument and navigation electrical components with high-frequency characteristics. The specific summary is as follows: (1) Motor components: Drive motor, oil pump, air pump, wiper motor, warm air motor, radiator fan, ventilator, etc. (2) Relay components: Flash relay, air conditioning compressor, etc. (3) Internal control circuit components: Wiper controller, air conditioning controller, electronic road sign, etc. (4) The internal parts with microprocessor: Vehicle controller, battery management system, reversing monitoring system, monitoring host, vehicle audio and video, electronic clock, combination instrument, vehicle terminal, etc. (5) Components with power conversion devices: DC/DC, DC/AC motor controller, two in one, five in one, etc. (6) Parts with antenna: GPS, GPRS, radio, Car Navigator, etc. The most important thing is that these components share low-voltage or part of them share high-voltage power supply, and the electromagnetic environment is complex, so it is difficult to analyze the correlation between parts EMC and vehicle EMC. (2) EMI suppression method Integrated design: Integration of multi-functional circuits and systems, such as the integration of bi-directional DC/DC up/down converter, DC/DC low-voltage charger, vehicle battery fast charger, generator and motor inverter unit and other functional circuits are integrated to reduce cable consumption and improve vehicle EMC performance. Other integrated assembly manufacturing technologies, such as

Fig. 2.24 On-board electronic control components of electric vehicles

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Fig. 2.25 EMC zoning prediction and layered design

integrated design of film capacitor and laminated bus, integrated design of power cable and sensor, etc. Partition prediction and layered design: For the vehicle EMC protection technology, the partition prediction and layered design are adopted according to the high and low voltage system, DC and AC system, vehicle interior and chassis. The design is shown in Fig. 2.25. The layered design of grounding, filtering and shielding technology will be introduced in the EMC design of key parts in the following chapters.

References CSBTS/TC79 (2018) Vehicles, boats and internal combustion engines—Radio disturbance characteristics—Limits and methods of measurement for the protection of on-board receivers: Chinese standard GB/T 18655-2018. Standards Press of China, Beijing CSBTS/TC79 (2018) Measurement methods for electromagnetic fields of vehicle with regard to human exposure: GB 37130-2018. Standards Press of China, Beijing Gong J (2019) Handbook of electric vehicle—drive motor and power electrics. China Machine Press, Beijing ISO/TC 22/SC 32 (2020) Electrical and electronic components and general system aspects. Road Vehicles—Electrical disturbance by conduction and coupling—Part 4: Electrical transient conduction along shielded high voltage supply lines only: ISO7637-4. ISO, 2020 Lin C (2019) Handbook of electric—vehicle design of pure electric vehicle. China Machine Press, Beijing National Technical Committee of Auto Standardization (2017) Limits and test method of magnetic and electric field strength from electric vehicles: Chinese standard GB/T 18387-2017. Standards Press of China, Beijing National Technical Committee of Auto Standardization (2018) Electromagnetic compatibility requirements and test methods of drive motor system for electric vehicles: Chinese standard GB/T36282-2018. Standards Press of China, Beijing Zhai L (2012) Fundamentals of vehicle EMC. China Machine Press, Beijing

Chapter 3

EMI Prediction and Suppression of Motor Drive System

3.1 Overview Unlike conventional engine vehicles, there are a large number of high-voltage components in EV, such as: drive motor, DC-AC inverter, DC-DC converter, and on-board charger (AC-DC Converter), high voltage power battery, etc. In addition, there are also many low-voltage electrical components such as battery management systems (BMS), vehicle control units (VCU), and Telematics BOX (TBOX) (Zhong and Lipo 1995). Therefore, the electromagnetic environment of EV has become more complicated, and electromagnetic compatibility (EMC) has become more and more important. In order to protect the on-board and off-board receivers from EV radio interference, international standards SAE J551-5, CISPR12 and CISPR25 set the limits for electromagnetic emission from EVs and on-board voltage components (Ran and Gokani 1998; Skibinski et al. 1997). EMC has become one of the key technologies for EVs. PMSM drive system typically uses power semiconductor devices (such as IGBTs) to implement pulse width modulation (PWM) control to regulate the three-phase AC voltage of the motor. The fast switching of the power semiconductor device produces a high current change rate di/dt and a voltage change rate of du/dt, forming EMI source signal, which propagates outward through the parasitic parameters of the internal components of the motor controller and the external high and low voltage lines, resulting in the unwanted conducted and radiated emissions (Chen et al. 2003). These emissions not only interfere with the on-board and off-board receivers, but also interferes with the high-voltage (HV) and low-voltage (LV) components of the vehicle through the conduction coupling path, and even affect the safety of the whole vehicle (Lai et al. 2006). In particular, the conducted EMI generated by the PMSM drive system not only induces the radiated emission of its own system to exceed the limits defined by standards, but also causes the radiated emission of the whole vehicle to fail to meet the EMC requirements. Therefore, the mechanism, prediction and suppression method of conducting EMI in the PMSM drive system are very important for the EMC for EV. In order to control the electromagnetic emission of the © China Machine Press 2021 L. Zhai, Electromagnetic Compatibility of Electric Vehicle, Key Technologies on New Energy Vehicles, https://doi.org/10.1007/978-981-33-6165-2_3

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3 EMI Prediction and Suppression of Motor Drive System

whole vehicle and finally ensure the EMC of the whole EV, the international standard CISPR25-2016 proposed the limit requirements and measurement methods for the electromagnetic emission of the HV and LV components of EVs in the frequency band of 150 kHz–108 MHz. Through a large number of tests, it can be seen that the components without EMC design can hardly meet the requirements of the standard limit level 3. (1) A high-frequency equivalent circuit model of inverter system for PMSM considering parasitic parameters of power semiconductor and system is established to predict conducted EMI. The model can also simulate the actual EMI source impedance and load impedance. The model provides a model platform for predicting conducted interference. (2) Based on the established high-frequency equivalent circuit model, the transfer functions of CM interference and DM interference at resonance frequency points exceeding the standard limit are established to predict the conducted EMI of high-voltage power lines and determine the main component parameters responsible for the formation of EMI. (3) According to the interference paths and the main components responsible for EMC, an effective method for suppressing wide-band conducted interference at external high voltage port is proposed for high-power motor drive system powered by HV DC of EV. The method can reduce the conducted EMI in 150 kHz–108 MHz wide band to meet the standard requirements. (4) According to the problem of conducted emission exceeding the standard limit due to resonance, a filter optimization design method based on suppressing resonance is proposed. It can be realized in the controller with small size, low cost and high efficiency. It can be realized in different stages of product development. (5) The design method of high-voltage DC power line EMI filter using magnetic ring and the design method of high-voltage DC power EMI filter using air core inductor are proposed. (6) The EMI measurement method and EMI characteristic test research of the motor drive system are described.

3.2 EMI Mechanism of Motor Drive System 3.2.1 Conducted Emission Test of Motor Drive System (1) Structure of PMSM drive system PMSM drive system is an important power component of EV, which mainly consists of a high voltage DC power supply, two high voltage DC power lines, an inverter, three-phase AC lines and a PMSM, as shown in Fig. 3.1. The three-phase fullbridge topology is usually used in the inverter of PMSM, which consists of six fully controlled power devices IGBT (T1–T6). Each IGBT is connected in reverse

3.2 EMI Mechanism of Motor Drive System Fig. 3.1 Schematic diagram of the PMSM drive system

67

High voltage DC cable

High voltage DC power supply

Control Controller Resolver signal Three-phase D5 AC cable

Drive circuit

Gate signal

D3

D1

T5

T3

T1

PMSM

Cdc

D4

D2 T2

T4

D6 T6

Inverter

parallel with a freewheeling diode, and the control signals from the gate determines the turn-on and turn-off of the IGBTs. (2) Test arrangement for conducted electromagnetic emissions According to CISPR25-2016 (CISPR 2015), the test arrangement of the conducted emission-voltage method for the high-voltage DC power supply line of the motor inverter is shown in Fig. 3.2. It mainly consists of the high voltage DC power supply, two linear impedance stabilization networks (LISNs), two high voltage DC lines with a standard length of 1.5 m, the motor inverter, three three-phase AC lines with a standard length of 1 m, a PMSM, a load dynamometer for PMSM and EMI receiver. Under the normal working condition of the PMSM system, the EMI receiver can measure the conducted interference voltage of the HV positive and negative power lines in the frequency bands of 150 kHz–108 MHz by the LISNs. The conducted interference voltage is required to meet the limits defined by CISPR25-2016. Figure 3.3 shows the prototype layout of the test platform which is placed on the test table in the shielding room and connected with the grounding copper plate. The parameters of the test PMSM are shown in Table 3.1. Figure 3.4 shows the experimental results of the conducted voltage in the frequency bands of 0.15–0.30 MHz, 0.53–1.8 MHz, 5.9–6.2 MHz, 30–54 MHz, 48.5–72.5 MHz, 68–87 MHz and 76–108 MHz required by the standard CISPR252016. It can be found that in the 0.53–1.8 MHz, 30–54 MHz, 48.5–72.5 MHz, and 68–87 MHz bands, the peak and average values of the conducted voltage of the HV

High voltage DC power supply

EMI receiver +

1 LISN

HV+ Line 2 -

LISN

HV- Line

Grounding plate

Fig. 3.2 Conducted EMI test layout

Inverter

Three-phase Insulated AC Lines connecting U shaft V W

Motor

Motor load (Dynamometer)

68

3 EMI Prediction and Suppression of Motor Drive System

Insulation Shaft Connected with Dynamometer

Test motor

HV AC Lines

HV LISN

HV DC Lines

Controller LV LISN LV Battery



HV DC Power Supply Lines

EMI Receiver

Monitor

Fig. 3.3 Conducted Emission-Voltage Test Platform Table 3.1 Motor parameters Parameter

Value

Rated power

42

DC voltage

336

Rated speed

3820

Unit

Parameter

Value

Unit

kW

Peak power

120

kW

V

Rated torque

105

N·m

rpm

Motor poles

4

Fig. 3.4 Conducted disturbance voltage test results of the high-voltage positive power supply line of the motor inverter

3.2 EMI Mechanism of Motor Drive System

69

positive power supply line exceed the level 5 average limit defined in the standard. Especially, there are couple of obvious conducted voltage peaks at 1, 30 MHz. Therefore, in order to study the formation mechanism of EMI from the PMSM drive system, and to accurately predict the conducted disturbance voltage of the HV DC power line, it is necessary to establish a high frequency circuit model to quantitatively analyze the relationship between the conducted EMI noise source and the conducted disturbance voltage, and to determine dominated parameters responsible for the conducted EMI exceeding the standard limits.

3.2.2 IGBT EMI Source Infineon FS800R07A2E3 IGBT module as the research object is used in the PMSM driver system. The module is composed of 3 bridge arms and 6 IGBTs, each IGBT is anti-parallel connected with a freewheeling diode and IGBT’s switching frequency is 10 kHz. The main technical parameters of the IGBT module are shown in Table 3.2. As shown in Fig. 3.5a, an equivalent circuit model of an IGBT considering parasitic parameters is established. Where D is a parasitic diode. The three junction capacitors of the IGBT are respectively the inter-electrode capacitance C GE between the gate and the emitter, the inter-electrode capacitance C GC between the gate and the collector and the inter-electrode capacitance C CE between the emitter and the collector, where C ies = C GE + C GC , C oes = C CE + C GC , C res = C GC . L G , L E and L C are the lead inductors of the gate, emitter and collector respectively, which can be calculate based on L S and L SCE. REE’ and RCC’ are the resistances of the corresponding leads, which can be calculate based on RCC’+EE’ and RGint . The voltage waveform V CE obtained between the collector C’ and the emitter E’ of IGBT in ideal case and the case of considering the parasitic parameters are respectively shown in Fig. 3.5b, c. Table 3.2 IGBT module (FS800R07A2E3) Parameters

Typical value

Unit

Condition

V CE

750

V

T J = 25 °C

IC

450

A

T Jmax = 175 °C

LS

20

nH

L sCE

8

nH

RCC’+EE’

0.75

m

T J = 25 °C

RGint

0.7



T J = 25 °C

C ies

80

nF

V GE = 0 V, V

CE

= 50 V, f = 1 MHz

C oes

1

nF

V GE = 0 V, V

CE

= 50 V, f = 1 MHz

C res

0.3

nF

V GE = 0 V, V

CE

= 50 V, f = 1 MHz

tr

80

ns

V CE = 400 V, I C = 450 A, Rg = 2.4 

tf

50

ns

V CE = 400 V, I C = 450 A, Rg = 5.1 

70

3 EMI Prediction and Suppression of Motor Drive System

CGC

LC CCE

LG

D CGE

LE

˄a˅Equivalent circuit model˄b˅Voltage between ideal collector C’ and emitter E’ ˄c˅9oltage between actual collector C’ and emitter E’

Fig. 3.5 IGBT equivalent circuit and voltage V CE between collector C’ and emitter E’

Next, the influence of the IGBT’s turn-on time t on , the rising (falling) time t r (t f ), the switching frequency f or the period T, and the ringing at the rising edge on the amplitude-frequency characteristics of the V CE is analyzed. (1) The influence of t on The voltage, current and power of the motor are regulated by the duty cycle Dy = t on /T of the pulse trapezoidal wave. Therefore, under the given switching frequency, the influence of t on on the amplitude-frequency characteristics of the V CE can be analyzed, and the influence of different motor load conditions on the conducted EMI can be obtained. Let the amplitude of the trapezoidal set as 330 V, f = 10 kHz, t r (t f ) = 0.08 μs, the amplitude frequency characteristics of V CE at duty cycle of 0.1, 0.5, and 0.8 are shown in Fig. 3.6a, respectively. It can be seen that the longer the t on is, the lower the first corner frequency (the intersection of 0 dB/dec and −20 dB/dec) is, and the higher the of V CE amplitude in the low frequency band is. (2) The influence of t r Let the trapezoidal wave of V CE have an amplitude of 336 V, a switching frequency of 10 kHz, a duty cycle of 0.5, and t r = t f . The amplitude-frequency characteristics of V CE for different rise (fall) times of 0.01 μs, 0.08 μs, and 0.4 μs are shown in Fig. 3.6b. It can be seen that the connection points of −20 dB/dec and −40 dB/dec are respectively near 32, 4, and 0.8 MHz respectively, which are consistent with the calculated frequencies of 31.8 M, 3.98 M and 0.795 M. It shows that the shorter the t r is, the higher the second corner frequency is, and the higher the amplitude of V CE in the high frequency band is. Therefore, the conducted EMI noise generated by the IGBT power module has a high frequency component. On the premise of ensuring the normal electrical performance of the motor, the conducted EMI in high frequency band can be reduced by adjusting the gate resistance outside the IGBT module to increase the rising (falling) time of the driving gate signals of the IGBT power module.

3.2 EMI Mechanism of Motor Drive System

(a)Influence of ton

71

(b)Influence of tr

(c) Influence of the switching frequency f

(d) Influence of the ringing

Fig. 3.6 Amplitude-frequency characteristics of VCE of IGBT

(3) Influence of f The switching frequency of the IGBT is related to both power loss and EMI. As shown in Fig. 3.6c, by analyzing the amplitude-frequency characteristics of V CE at different switching frequency of 1 kHz, 10 kHz and 100 kHz, it can be seen that the amplitude of V CE increases with the increase of f . Therefore, although the higher switching frequency of the IGBT can reduce the power loss, it also causes more serious EMI problems. (4) Influence of ringing at edge As shown in Fig. 3.6d, the parasitic capacitance and lead inductance of the IGBTs will induce ringing at the rising and falling edges of the pulse of V CE . The ringing will significantly increase the amplitude of V CE in the high frequency region of 18–108 MHz, and cause a resonance peak at near 62 MHz. It can be seen from above analysis that due to the fast switching of the IGBT, the V CE of the IGBT has a high amplitude in the wide frequency range of 150 kHz– 108 MHz. The EMI noise current is easily formed by the system parasitic parameters, and finally caused the conducted EMI from the high-voltage DC power lines of the inverter. For the actual test conditions, the switching frequency and rise time of the IGBT are usually constant. So the conducted EMI voltage is mainly related to the on-time of noise source signal V CE and parasitic parameters of IGBT module. In addition to the signal of the IGBT noise source, the conducted EMI voltage on the HV DC power supply lines of the motor controller is also determined by the disturbance current propagation path.

72

3 EMI Prediction and Suppression of Motor Drive System

3.2.3 EMI Coupling Path (1) High-frequency equivalent circuit model of motor system It can be seen from Fig. 3.6 that the spectrum of the interference source signal V CE is distributed in the frequency range from150 kHz to 108 MHz, so the influence of the parasitic parameters of the PMSM drive system on the propagation path of the interference current must be considered. According to Fig. 3.3, a high frequency equivalent circuit model of the PMSM drive system is established, as shown in Fig. 3.7. Wherein R1 and R2 are connected in series with C 1 and C 2 , and then connected in parallel with C 3 and C 4 to form positive and negative LISNs. The resistances and inductances of the DC positive and negative lines RDC+ , RDC- and L DC+ , L DC- and their capacitances to ground C DC+ , C DC- form the equivalent circuit model of DC lines. C DC is the internal ripple suppression capacitor of the inverter, RDC and L DC are the lead inductance and equivalent resistance of C DC. The resistances and inductances of three-phase lines RCA , RCB , RCC , L CA , L CB , L CC and ground capacitances C AG , C BG , C CG together form the equivalent circuit model of AC shielded lines. RMA , RMB , RMC , L MA , L MB , L MC are the phase resistances and inductances of the threephase winding of the PMSM, and C M is the winding-to-case parasitic capacitance. C P and C N are bus-to-ground parasitic capacitances of the DC positive and negative for inverter. L IGBT is IGBT lead inductance, C S1 –C S6 are the IGBT inter-electrode equivalent capacitances, C A , C B , and C C are the neutral point-to-ground parasitic capacitances of the three-phase bridges of inverter, L A –L C are the inductances of the copper bar connected by the neutral point of the inverter. The noise current generated by the fast switching of the IGBT finally flows into the LISN through the inverter, the AC lines, the motor, the DC lines, and the parasitic capacitances. The values of main parasitic parameters and component parameters in the system are shown in Table 3.3, obtained by measurement or theoretical calculation (Falck et al. 1997). LISN

DC Lines

AC Lines

Inverter

PMSM

L1 C3 C1

RDC+

LDC+ CDC

R1 VDC

LIGBT

LIGBT

CS1

CS2

CS3

LIGBT

LIGBT

LIGBT

LIGBT

LIGBT

LIGBT

CS4

CS5

CS6

LIGBT

LIGBT

LIGBT

CA

CB

CC

LIGBT LA

RCA

LCA

RMA

LMA

LB

RCB

LCB

RMB

LMB

LC

RCC

LCC

RMC

LMC

LDC C4 R2 L2

C2

RDC RDC- LDC-

CM CCG CBG CAG

CDC-

CDC+ CP CN

Fig. 3.7 Equivalent circuit model of high frequency electromagnetic emission of motor drive system

3.2 EMI Mechanism of Motor Drive System Table 3.3 High frequency equivalent model parameter values

73

Parameters

Value

Parameters

Value

R1 , R2 C1, C2

50 

C S1 –C S6

80 nF

0.1 μF

CP, CN

320 pF

C3, C4

0.1 μF

C A –C C

500 pF

RDC+ , RDC−

0.8 μ

RCA –RCC

0.6 μ

L1 , L2

5 μH

L A –L C

120 nH

L DC+ -, L DC-

32 nH

L CA –L CC

26 nH

RDC+ -, RDC-

0.69 m

RCA –RCC

0.52 m

C DC+ , C DC-

900pF

C AG –C CG

672pF

C DC

1000 μF

RMA –RMC

5.9 m

L DC

100nH

L MA –L MC

0.21 mH

RDC

0.1m

CM

1 nF

L IGBT

10nH

(2) Analysis of the conducted EMI paths of the inverter system (1) Frequency band of voltage exceeding the limits It can be seen from Fig. 3.4 that the conducted voltage of the HV positive power supply line exceeding the standard limits mainly distribute in three frequency bands of 0.53–1.8 MHz, 30–60 MHz, and 60–87 MHz: ➀ Band 1 (0.53–1.8 MHz): In the band 0.53–1.8 MHz, the conducted EMI voltage peak and average values exceed the Level 5 limits by 3 dB and 15 dB, respectively. Due to the existence of parasitic parameters of the system, the resonance is generated at 1 MHz. Therefore, it is important to analyze the CM and DM interference path of the noise current generated by the switching of the IGBT at 1 MHz. So the relationship between the interference source and the conducted EMI voltage is needed to be established to determine the dominated parameters responsible for the conducted EMI. ➁ Band 2 (30–60 MHz): In the 30–60 MHz band, the peak and average value of the conducted EMI voltage exceeds the standard limits by 17 dB and 8 dB, respectively. It is mainly due to the resonance at 30 MHz generated by harmonic of trapezoidal pulse train of the IGBT. ➂ Band 3 (60–87 MHz): In the 60–87 MHz band, the peak and average value of the conducted voltage exceed the limits by 18 dB and 6 dB, respectively, mainly due to the resonance at 68 MHz generated by system low-frequency harmonics under the influence of high-frequency parasitic parameters. In summary, in order to reduce the conducted EMI of the HV power line to meet the requirements of the standard limits, it is only necessary to quantitatively analyze the factors responsible for the conducted EMI at two typical frequency points of 1 MHz and 30 MHz. Since the conducted voltage exceeding standard limits at around 68 MHz are mainly caused by high frequency parasitic parameters, and it is difficult to perform accurate numerical quantitative analysis.

74

3 EMI Prediction and Suppression of Motor Drive System

(2) Switching mode When the inverter is working, the three upper arms have eight switching modes (1,0,0), (0,1,0), (0,0,1), (1,1,0), (1,0,1), (0,1,1), (1,1,1) and (0,0,0). These eight working modes can be divided into three groups: (1,0,0), (0,1,0), (0,0,1); (1,1,0), (1,0,1), (0,1,1); (1,1,1), (0,0,0). The interference sources and the interference paths are dual under the modes of same groups. Therefore, the two IGBT switching states of each bridge arm are complementary, only three switching states are (1,0), (0,1), (0,0). The three switching modes of IGBT can be equivalent to a CM interference source and a DM interference source on each arm. Therefore, the disturbance current path generated by the DM interference sources of three bridge arms are completely identical, and thus the total DM interference voltage is three times of the voltage U R1 generated on the resistance R1 on the resistance R1 measured by LISN when IGBT of single bridge arm works. Similarly, the disturbance current path generated by the CM interference sources of the three bridge arms are also symmetrical. For the above reason, in order to facilitate the quantitative analysis, the switching mode (1,0,0) of the upper bridge arms is taken as an example to analyze the propagation path of the CM and DM current generated by the interference source of single bridge arm. (3) Interference current path analysis In order to determine the type of interference, it is necessary to perform a reactance analysis on the high frequency circuit of the motor drive system. From the experimental results in Fig. 4, it can be seen that the first resonance peak occurs at around 1 MHz with an amplitude up to 87 dBμV. The second resonance peak occurs at around 30 MHz with an amplitude up to 75 dBμV. The third resonant peak occurs at around 68 MHz with an amplitude up to 65 dBμV. ➀ Conducted EMI analysis at 1 MHz Firstly, the reactance of each circuit element is calculated at 1 MHz and marked in the circuit diagram. Then, the CM interference current and DM interference current propagation path are analyzed according to the reactance value. Finally, the relationship between the interference source and the conducted EMI voltage measured by the LISN is established. (a) DM interference paths analysis The interference source caused by the IGBT switching is usually equivalent to a constant current source I DM . Since the inductance reactance of winding of the AC motor is more than ten times higher than that of the DC side, the grounding capacitance reactance of the three-phase AC cable is also relatively large, so the influence of the motor and the three-phase cables on the DM interference current can be neglected. It can be considered that the formation of DM interference is only related to the inverter and its DC side circuit. The following five DM current propagation

3.2 EMI Mechanism of Motor Drive System 0.00069 j0.2

j314 L1 C1

-j0.032

RDC+ LDC+ -j0.34 -j0.00016

LIGBT CDC

CS1

R1 50

VDC

IDM1 C4 R2

-j0.032 L2 j314

75

C2

IDM2 j1.32

LDC IDM3

50 -j0.34 0.0001 RDC- LDC-

RDC

IDM

LIGBT IDM4 LIGBT j0.16 CS4 -j1.99 LIGBT

LIGBT

LIGBT CS2

CS3

LIGBT

LIGBT

LIGBT IDM5 CS5

LIGBT

LIGBT

LIGBT

j0.75 0.00052 LA RCA

j0.16 0.0059 j1319.4 LCA RMA LMA

LB

RCB

LCB

RMB

LMB

LC

RCC

LCC

RMC

LMC

CS6

0.00069 j0.2

Fig. 3.8 DM EMI Current Path at 1 MHz

path are formed in high frequency circuit of the system. The DM EMI propagation path is shown in Fig. 3.8. • DM current path 1: I DM1 → 3L IGBT → L DC+ →RDC+ →C1 → R1 → C2 → R2 → RDC- → L DC- → L IGBT • DM current path 2: I DM2 → 3L IGBT → C DC → L DC → RDC → L IGBT • DM current path 3: I DM3 → L IGBT → C S4 → L IGBT • DM current path 4:I DM4 → 3L IGBT → C S2 → 5L IGBT • DM current path 5:I DM5 → 3L IGBT → C S3 → 5L IGBT Only the DM current I DM1 of path 1 flows through R1 to form the positive DM conducted voltage U R1 . In fact, two switching modes are considered simultaneously, U DM= 3U R1= 3 * 50 * I DM1 The relationship between the HV positive conducted voltage U DM and I DM of the motor inverter is analyzed below. Suppose:  Z 1 = 2 jωL DC+ + R DC+ +

1 + R1 jωC1



1 + jωL DC + R DC ωCDC

Z2 = − j

Z 3 = 2 jωL IGBT +  Z 4 = 4 jωL IGBT +

1 jωC S4

   1 1 // 4 jωL IGBT + jωCs2 jωC53

(3.1) (3.2) (3.3) (3.4)

Z 5 = 2 jωL IGBT

(3.5)

Z = (Z 1 //Z 2 //Z 4 + Z 5 )//Z 3

(3.6)

U D M = 3U R1 = 3 × IDM × Z ×

R1 Z 1 //Z 2 //Z 4 × Z 1 //Z 2 //Z 4 + Z 5 Z1

76

3 EMI Prediction and Suppression of Motor Drive System

Z 1 //Z 2 //Z 4 R1 (Z 1 //Z 2 //Z 4 + Z 5 ) × Z 3 × × Z 1 //Z 2 //Z 4 + Z 5 + Z 3 Z 1 //Z 2 //Z 4 + Z 5 Z1 3IDM R1 Z 3 × (Z 1 //Z 2 //Z 4 ) = × (3.7) Z1 Z 1 //Z 2 //Z 4 + Z 5 + Z 3 = 3 × IDM ×

Since R1 Z 3 Z 4 Z 5 in (3.7) is constant and the DM interference source I DM is a constant current source. so the main variables determining U R1 are included in Z 1 and Z 2 . Increasing the amplitude of Z 1 and decreasing of the amplitude Z 2 can effectively reduce U R1 . As shown in Fig. 9, since the imaginary part of Z 1 + Z 2 //Z 4 is: (-j1.99 + 4 * 0.06)/2//(-j1.99 + 2 * 0.06) + j0.56 = −0.02j, it can be seen that C S2 , C S3 , C S4 and several L IGBT are in parallel resonance with L DC . Therefore, U R1 has a resonance peak at near 1 MHz. In fact, reducing the interference voltage U DM by changing the amplitudes of Z 1 and Z 2 is two different methods. Increasing the Z 1 amplitude is used to reduce the impedance value of each element to reduce the peak value of conducted voltage at the resonant point, which result in reducing the U DM . However, since L DC and C DC in Z 2 are the most important elements for resonating with C S2 , C S3 , C S4 and several L IGBT , changing the amplitude of Z 2 is essentially shifting the resonance point from 1 MHz to a high frequency point without specified limits to reduce the amplitude of conducted voltage at 1 MHz (Fig. 3.9).

CDC

j0.56

LDC

RDC IDM

Fig. 3.9 Resonant circuit

LIGB T j0.06

LIGB T

LIGB T

LIGB T j0.06

CS2 -j1.99 LIGB T j0.06

CS3 -j1.99 LIGB T j0.06

LIGB T

LIGB T

LIGB T

CS4 -j1.99 LIGB T

LIGB T

LIGB T

3.2 EMI Mechanism of Motor Drive System L1 -j0.032 C3 C1 R1

VDC

RDC+ LDC+ j0.2

LIGBT

-j0.34+ UR1 50 -

77 LIGBT

LIGBT

LIGBT

-j1.99 CS2 j0.16 LIGBT

j0.16 0.00052 j4.9 LCA LA RCA

0.0059 RMA

j1319.4 LMA

CS3 LIGBT

LB

RCB

LCB

RMB

LMB

LC

RCC

LCC

RMC

LMC

UCM 50

C4 R2 -j0.032

C2

CS4

-j0.34 j0.2 RDC- LDC-

CM

L2 ICM1 CDC- CDC+ -j176

CP

CN -j497

CA

CB -j318

CC

CAG CBG CCG -j237

Fig. 3.10 CM interference path at 1 MHz

(b) CM interference analysis The CM interference source caused by the IGBT switching is equivalent to a constant voltage source U CM . As shown in Fig. 3.10, the following four CM current propagation paths are formed in the high frequency circuit. • CM current path 1: U CM → 3L IGBT → L DC+ →RDC+ →C 1 → R1 → GND • CM current path 2:U CM → L IGBT → C S4 → L IGBT → L DC- → RDC- → C 2 → R2 → GND • CM current path 3:UCM → L IGBT → C S2 → 3L IGBT → L DC- → RDC- → C 2 → R2 → GND • CM current path 4:U CM → L IGBT → C S3 → 3L IGBT → L DC- → RDC- → C 2 → R2 → GND To further analyze the relationship between the conducted voltage on R1 and the path, the path in Fig. 3.10 is simplified, as shown in Fig. 3.11a. To facilitate analysis, the equivalent simplification of each part of the circuit is carried out, as shown in Fig. 3.11b, where Z 1 = 2L IGBT , Z 2 = 2L IGBT + C S4 , Z 3 = L DC(+/-) + RDC(+/-) + C 1 + R1 , Z 4 = (C S2 + 4L IGBT )// (C S3 + 4L IGBT ), then the following relationship is established: ⎧ ⎪ i1 = i2 + i3 ⎪ ⎪ ⎪ ⎪ ⎨ i5 = i3 + i4 (3.8) i 1 Z 1 + i 2 Z 3 = UCM ⎪ ⎪ ⎪ i Z + i Z = U 4 2 5 3 CM ⎪ ⎪ ⎩i Z + i Z + i Z = U 1 1 3 4 5 3 CM By solving (3.8), the following formula can be obtained: UR1 = i 2 R1 =

UCM (Z 1 Z 3 + Z 2 Z 3 + Z 2 Z 4 + Z 3 Z 4 ) (Z 1 + Z 2 + Z 4 )Z 32 + 2Z 1 Z 2 Z 3 + Z 1 Z 2 Z 4 + Z 1 Z 3 Z 4 + Z 2 Z 3 Z 4 (3.9)

78

3 EMI Prediction and Suppression of Motor Drive System

Fig. 3.11 The simplified CM EMI path and equivalent circuit at1MHz

2LIGBT

UCM

LDC+ RDC+

C1

R1

C2

R2

CS2 4LIGBT 4LIGBT LIGBT

CS4

CS3

LIGBT

LDC-

RDC-

(a) Simplified equivalent circuit i1 UCM

i4

Z1

i2 i3 Z4

Z2

Z3

i5 Z3

(b) Impedance circuit

It can be seen from (3.9) that U CM is related to L IGBT , C S2 , C S3 , C S4 , L DC- , RDC- , C 1 , R1 , C 2 , R2 . The way to reduce U CM is to increase the impedance of CM current path. ➁ Conducted EMI analysis at 30 MHz (a) DM interference analysis From Fig. 3.12, the path of current propagation of DM interference at 30 MHz is the same as that at 1 MHz. However, since the components impedance values has changed, U R1 has been changed. L DC and 4L IGBT can not resonate with C S4 at 30 MHz. Increasing the impedance of L DC+ and L DC- or shunting I R1 with an X capacitor can ultimately reduce U R1 . j9425 L1

0.00069 j6.03

C1

-j0.001

R1

IDM1

VDC

C4 R2 -j0.001

L2 j9425

C2

RDC+ -j0.01 50

LDC+

LIGB T

-j0

CDC

IDM2 j41.5

LDC

50 -j0.01 RDC-

0.0001 LDC-

RDC IDM

CS1

IDM3

LIGB T IDM4 LIGB T j4.7 CS4 -j0.07 LIGB T

LIGB T

LIGB T

CS2

CS3

LIGB T

LIGB T

LIGB T IDM5 CS5

LIGB T

LIGB T

LIGB T

j22.6 0.00052 j4.9 0.0059 LCA LA RCA RMA

LMA

LB

RCB

LCB

RMB

LMB

LC

RCC

LCC

RMC

LMC

CS6

0.00069 j6.03

Fig. 3.12 30 MHz frequency differential mode interference current path

3.2 EMI Mechanism of Motor Drive System

(b) CM interference analysis • CM current path 1: UC M → • CM current path 2: UCM → • CM current path 3: UC M → • CM current path 4: UC M →

UC M →



  

79

2L I G BT →L DC+ →R DC+ →C1 →R1 L I G BT →C S4 →L I G BT →L DC− →R DC− →C2 →R2 2L I G BT →C DC+ L I G BT →C S4 →L I G BT →C DC− 2L I G BT →C P L I G BT →C S4 →L I G BT →C N 3L I G BT →Cs2 →L I G BT L I G BT →C S4 →3L I G BT



3L I G BT → Cs3 → L I G BT L I G BT → Cs4 → 3L I G BT







→ GN D

→ GN D

→ GN D

→ CB → G N D  → CC → G N D

UC M → C A → G N D  I G BT →C S2 →L I G BT • CM current path 5: UC M → 3L → RC B → L C B → C BG → L I G BT →C S4 →3L I G BT GN D 

3L I G BT → C S3 → L I G BT UC M → → RCC → L CC → CC G → G N D L I G BT → C S4 → 3L I G BT UC M → RC A → L C A → C AG → G N D Through the above analysis in Fig. 3.13, there are multiple CM interference paths at 30 MHz, which are related with C S4 , C S5 , C S6 , L IGBT , L DC- , RDC- , L DC+ , RDC+ , L A , L B , L C , C P , C N , C DC+ , C DC- , C A , C B , C C , RCA , RCB , RCC , L CA , L CB , L CC , C AG , C BG , C CG , R1 , C 1 , R1 , and C 1 . (3) Radiated emissions The radiated emission test is essentially to test the radiated signal generated by the two equivalent antennas in the motor drive system: j9425 L1

-j0.001 C3 C1

RDC

LDC+ j6.03

LIGB T

LIGB T

-j0.01

R1 LIGB T

50

LIGB T

-j0.07 CS2 j1.88 LIGB T

j22.6 LA

RCA

j4.9 LCA

0.0059 j39584 RMA LMA

LB

RCB

LCB

RMB

LMB

LC

RCC

LCC

RMC

LMC

CS3 LIGB T

VDC C4 R2 -j0.001

C2

50 CS4

-j0.01 j6.03 RDC- LDC-

CM

j9425 L2

CDC- CDC+ -j6

CP CN -j16.6

CA

CB -j10.6

CC

Fig. 3.13 30 MHz frequency common mode interference current path

CCG CBG CAG -j7.9

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3 EMI Prediction and Suppression of Motor Drive System

One is the equivalent loop antenna, caused by the DM current I D in the loop, the electric field strength E at the distance D from the loop is: E = 1.3S I D f 2 /D

(3.10)

where S is the loop area and f is the frequency of the DM current. It can be seen from (3.10) that the electric field strength is related to the conducted DM current I D . The other is the equivalent monopole antenna or symmetric dipole antenna. The cable of the motor drive system can be equivalent to this antenna, which is caused by the CM current I cm on the cable bundle (the length of the cable is L), I cm is the focus of studying radiated emission, and the electric field intensity E C generated at distance D is E C = 6.28 × 10−7

I cm f L D

(3.11)

From (3.11), it can be seen that the electric field strength is related to the conducted CM current I cm . It can be seen from (3.11) and (3.12) that both DM current and CM current can cause radiated emissions, and controlled conducted emissions can control radiated emissions; in addition, it is easier to reduce conducted emissions than to reduce radiated emissions. Because the conducted emission is only propagated by the motor drive system power line or signal line and control line path. The DC HV cables, three-phase HV cables, and LV power lines of the motor drive system may be equivalent to antennas under the action of high-frequency noise sources, to form electromagnetic radiation. Therefore, considering that the radiated emissions below 108 MHz are mainly caused by cables, the filter suppression technology of conducted interference can be used to suppress the radiated emissions.

3.3 EMI Modeling of Motor Drive System 3.3.1 Modeling and Simulation of EMI in Motor Drive System The high-frequency equivalent circuit model of the motor drive system includes: inverter model, power cable model, motor model, and line impedance stabilization network (LISN) model, as shown in Fig. 3.14. In order to simulate the actual operating state of the EV motor drive system and accurately predict the EMI of the motor drive system, it is also necessary to build an electrical and thermal model of the power module and a pulse width modulation (PWM) control algorithm.

3.3 EMI Modeling of Motor Drive System

81

Fig. 3.14 Motor drive system high-frequency equivalent circuit model

(1) IGBT power module (1) IGBT power module electrical and thermal model The IGBT power module is a key component of the motor controller. Its rapid on and off voltage jump du/dt is up to several thousand volts per microsecond, and current jump di/dt is up to several thousand amperes per microsecond, becoming the main interference source of high-voltage system for EVs. The stray inductance and stray capacitance of the IGBT module form the paths of EMI coupling. The IGBT equivalent circuit built using ANSYS/Simplorer software is shown in Fig. 3.15. In addition to the static characteristics, the charging and discharging effect of the junction capacitance are also considered in the equivalent circuit of the IGBT. The tail current is simulated by an equivalent current source, and the influence of temperature and collector current on the tail current is considered. Finally, the

Fig. 3.15 IGBT high-frequency equivalent circuit model

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3 EMI Prediction and Suppression of Motor Drive System

Fig. 3.16 Equivalent model of IGBT thermal network (Source Zhao 2014)

influence of the collector-emitter voltage on the tail current is also considered. The thermal equivalent model is shown in Fig. 3.16. The PEL is equivalent to a current source, representing the actual current flowing through the IGBT and diode, where the junction temperature, chip internal loss, chip to substrate loss, substrate to heat sink loss, environmental loss are also considered. The eight switching components ST1-ST4, SD1-SD4 are all terminated to ground. According to the relevant parameters and characteristic curves in the IGBT data manual, the Simplorer software is used to establish the electrical and thermal model of the IGBT module. The modeling steps are as follows: first, the model information database needs to be established; then the rated operating point, the limit operating point (module breakdown boundary description), half-bridge test status (on-resistance, stray inductance value and capacitance value), IGBT transfer characteristic curve, IGBT and freewheel diode output characteristic curve, IGBT and freewheel Diode thermal effect, dynamic model input are set dynamic parameters are validated; finally, dynamic parameters are extracted. The fitting results obtained from the simulation are shown in Table 3.4 and Table 3.5. As can be seen from the table, the turn-on loss error is slightly larger, and the temperature error at 125 °C Table 3.4 Simulation error of IGBT module parameters at a junction temperature of 25° Parameters

Turn-on loss

Turn-off loss

On-time delay

Manual data

10.5 (mJ)

21 (mJ)

120 (ns)

Off-time delay 510 (ns)

Simulation data

9.97 (mJ)

21.1 (mJ)

118.1 (ns)

514.7 (ns)

Error

5.0%

0.7%

1.6%

0.92%

Table 3.5 Parameter simulation error of IGBT module at 125 °C Parameters

Turn-on loss

Turn-off loss

On-time delay

Off-time delay

Manual data

12.5 (mJ)

26 (mJ)

130 (ns)

550 (ns)

Simulation data

13.4 (mJ)

25.7 (mJ)

115.6 (ns)

550.2 (ns)

Error

7.2%

1.2%

3.5%

0.036%

3.3 EMI Modeling of Motor Drive System

83

is the largest, 7.2%. The errors of other parameters are all below 5%, it shows that simulation modeling has high accuracy. ➀ Model verification Using the established IGBT model, first the IGBT turn-off and turn-on process are simulated, and then the output characteristic curve is simulated to verify the accuracy of the model. The simulation results of IGBT turn-off and turn-on model circuit and switching characteristics are shown in Fig. 3.17. The values of resistors R1 and R2 are 1.8, and the driving voltage E 2 is a square of amplitude ± 15 V DC and frequency 10 kHz. The voltage E 1 across the IGBT is DC 360 V, and the parasitic inductances of the gate, collector, and emitter wires are 0.1 nH. Figure 3.17b, d is the simulation

Fig. 3.17 IGBT turn-on and turn-off circuit model and voltage and current simulation waveforms

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3 EMI Prediction and Suppression of Motor Drive System

results with a simulation step size of 5 ps, and (c) and (e) are the simulation results with a simulation step size of 500 ps. The simulation found that when smaller steps are used, the waveform at the instant of turn-on and turn-off is oscillating, which is related to the high-frequency parasitic parameters of the IGBT equivalent circuit. In the voltage waveform of (c), there is an obvious peak voltage, which is caused by external inductance, since this peak does not appear when the parasitic inductance of the gate, collector, and emitter wires is not added. The accuracy of the IGBT model is very important for the accuracy of extracting the interference source signal in the later stage. The comparison of the output characteristic curve from IGBT data sheet and the simulation results are shown in Figs. 3.18 and 3.19. Among them, Fig. 3.18 is the output characteristic curve of different gate voltage V ge at the junction temperature of 125 °C. The abscissa is the voltage between collector and emitter V ce , and the ordinate is the collector current I c . Figure 3.19 shows the transfer characteristic curves under different junction temperatures T vj when V ge is 15 V. It can be seen that the simulation results are basically consistent with the parameter curves given in the IGBT data sheet, indicating that the established IGBT electrical and thermal models have high accuracy. ➁ The influence of inductive load The motor can be regarded as an inductive load, and a single-arm simulation circuit of the IGBT power module is established, as shown in Fig. 3.20. The IGBT turn-on and turn-off characteristics under different inductive loads are studied. The driving voltage of the upper and lower bridge IGBTs is a square wave signal with an amplitude of 15 V and a phase difference of 180°, the value of the voltage stabilizing capacitor C 1 is 1100 uF, and load is an inductive load. The simulation results are shown in Figs. 3.21 and 3.22.

Fig. 3.18 IGBT output characteristic curve

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85

Fig. 3.19 IGBT output characteristic curve at different temperature

Fig. 3.20 Simulation circuit of the effect of different inductive loads on the turn-on waveform

Figure 3.21 is the time-domain waveform of the voltage across the IGBT power module and the inductive load of the upper arm. At the moment when the IGBT drive signal of the upper arm is turned off, the voltage across the IGBT power module will have a spike, as shown in Fig. 3.21a. Regardless of whether the IGBT is turned on or off, the voltage across the inductive load will have a spike, as shown in Fig. 3.21b. The voltage pulse spike at the two ends of the upper arm IGBT is 50 V and the oscillation period is 0.3 ns; the voltage pulse spike at the two ends of the inductive

86

3 EMI Prediction and Suppression of Motor Drive System 2mH 0.2mH 20uH 2uH 0.2uH

(a) Voltage waveform across the upper arm 2mH 0.2mH 20uH 2uH 0.2uH

(b) Voltage waveform across inductive load 2mH 0.2mH 20uH 2uH 0.2uH

(c) Local amplification of the voltage waveform across the upper arm (d)Local amplification of voltage across inductive loads

Fig. 3.21 Turn-on and turn-off voltage waveforms of different inductive loads

2mH 0.2mH 20uH 2uH 0.2uH

(a) Positive line current

(b) Local amplification

Fig. 3.22 The turn-on and turn-off current waveform under different inductive loads

3.3 EMI Modeling of Motor Drive System

87

load is 30 V and the ringing period is 0.3 ns. Because there are other stray inductances in the circuit, the voltage pulse spikes across the inductive load are lower than the voltage across the upper arm bridge, but both have the same oscillation period. At the moment when the upper bridge IGBT drive signal is turned off, the positive current waveform will also have over-pulse phenomenon, as shown in Fig. 3.22, the peak current amplitude is 3A, and the period is 0.08 ns. It is found that the period of current ringing is inconsistent with the period of voltage ringing. (2) IGBT module packaging model At low frequencies, the bonding wire and copper conductors inside the IGBT module can be regarded as good conductors. However at higher frequencies (switching frequency), the stray parameters such as distributed inductance and capacitance of the IGBT module will become the coupled paths of high-frequency noise, resulting in the EMI. The EMI generated by the high-voltage part of the IGBT module will be coupled to the control board and drive board of the low-voltage circuit through these paths, affecting the operation of the motor drive system. Therefore, the extraction of the stray parameters of the internal structure of the IGBT module is an important part of the EMI modeling and simulation of the motor drive system. Figure 3.23 shows the internal structure of the IGBT module of the power switching device. As shown in Fig. 3.23b, the structure of the IGBT is chip-solder-DCB boardsolder-substrate-thermally conductive silicone-heat sink in order from top to bottom, where the DCB board is ceramic-based copper clad laminate, adopting a special process method in which copper foil is directly bonded to the surface (single side or double side) of aluminum oxide (Al2 O3 ) or aluminum nitride (AlN) ceramic substrate at high temperature. The chip is soldered to the first copper plate with solder, using ceramic (Al2 O3 /AlN) as the insulating layer, and the copper plate is soldered to the substrate by solder, and the substrate is connected to the heat sink as a whole through thermal silica gel. Q3D software of ANSYS can extract spurious parameters. The IGBT module model is established in the Q3D software, without considering the solder layer, thermally conductive silicone and substrate. The thickness of the copper plate in the DCB

Chip

ChipCase

Solder Copper Ceramics Copper Solder Substrate

Case-Heat sink Heat sinkEnvironment

(a)IGBT physical

Fig. 3.23 IGBT physical and internal structure

Thermal silica Heat sink

(b)IGBT internal structure

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3 EMI Prediction and Suppression of Motor Drive System

Fig. 3.24 The IGBT 3D model established in Q3D

board is 2 mm, the insulating material in the DCB board is 5 mm thick aluminum oxide (Al2 O3 ), and the material of the heat sink is silver. In practice, silver is sprayed on the surface of the iron block, and the size is 216 mm * 100 mm * 50 mm. A 3D model of the IGBT power module is established as shown in Fig. 3.24, in which the wire bonding technology is generally used as electrical connections between the IGBT chip and the electrode terminal of the chip, between the IGBT chip electrode terminal and the diode chip, and between the chip electrode terminal and the insulating liner. The interconnection between the chips is formed by bonding wires to form a loop. Wire bonding is one of the main ways to realize electrical interconnection inside IGBT power devices. One of the most widely used chip interconnection technologies in the industry-aluminum wire bonding are used in the model, with a cross-sectional size of 2 mm * 2 mm cube. After the IGBT 3D model is established, the positive network, negative network, and three-phase output network need to be allocated, as shown in Fig. 3.25. Then, the power is allocated for grid parameter calculation, and finally the scan parameter setting is performed for simulation. The parasitic parameters of the IGBT package at a frequency of 150 kHz are obtained as the RLC matrix through the simulation calculation, as shown in Fig. 3.26. Using Q3D software to extract the stray parameters of the IGBT power module simplifies the tedious work, and the inductance, capacitance, resistance and conductance of each grid and between the two grids can be calculated. The extraction of IGBT power module stray parameters laid the foundation for the EMI simulation of the motor drive system.

Fig. 3.25 Negative grid, positive grid and W phase grid

3.3 EMI Modeling of Motor Drive System

89

Fig. 3.26 The simulation results of IGBT spurious parameters

The existence of spurious parameters provides more coupling paths for highfrequency interference signals, making the electromagnetic environment more complex and more difficult to analyze. The distributed inductance and capacitance of the copper layers and bonding wires can no longer be ignored. When the IGBT is only provided with a current source for the positive electrode, the size and density distribution of the DC current and AC current in the body (surface) of the IGBT packaging model are shown in Fig. 3.27. It can be seen from Fig. 3.27 that under AC excitation, the distributed inductors and capacitors of the IGBT module become the EMI coupling paths, and the current no longer flows just in the positive network like DC, and part of the current forms a loop through the stray capacitors. This makes the analysis of EMC more difficult, and the greater the frequency, the more severe the coupling. Therefore, the establishment

90

3 EMI Prediction and Suppression of Motor Drive System

Fig. 3.27 the DC and AC Current distribution of IGBT

of a high-frequency equivalent model that reflects the true physical structure of the IGBT power module and the extraction of its stray parameters are crucial to the analysis of the mechanism of EMI generated by the EV motor drive system. (2) Modeling and simulation of high-voltage power cables (1) Three-phase AC shielded power cable high-frequency equivalent circuit model The motor inverter uses a three-phase AC shielded power cable to connect with the motor. One end of the cable shield is connected to the inverter shell and the other end is connected to the motor shell. Both the inverter shell and the motor shell are connected to the vehicle. The capacitance between the shielded cable and the vehicle body can be regarded as the capacitance between the inner conductor of the cable and the shield layer. The lumped parameter equivalent circuit model of the threephase AC shielded power cable is shown in Fig. 3.28, where R is the resistance of the inner conductor of the cable, L is the self-inductance of the inner conductor, M Fig. 3.28 The high frequency equivalent circuit model of shielded cables

3.3 EMI Modeling of Motor Drive System

91

Fig. 3.29 Cross-section structure of shielded cable

Outer layer(PVC)

Sheath

Outer conductor(shielding)

Shielding layer Insulation layer

Inner conductor

Conductor

Insulation layer(PVC)

is the mutual inductance between the inner conductors of different phase lines, C is the equivalent capacitance between the inner conductor and the shielding layer. (2) High-voltage power cable model The cross-section of the high-voltage power cable for EVs is shown in Fig. 3.29. From inside to outside, the conductor, insulation layer, shielding layer, and sheath are in order. The conductor is generally bare copper. The insulation layer material and sheath material are generally PE (polyethylene), PVC (polyvinyl chloride), XLPE (cross-linked polyethylene), etc. In order to prevent electromagnetic field leakage, the HV cables needs a shielding layer. And the output port of the inverter is connected to the motor through a high-voltage cable. The longer the cable length, the longer the transmission time of the motor inverter output pulse wave in the HV cable. When the propagation time of the pulse wave is simulative compared to the rise time or fall time of the output pulse wave, if the characteristic impedance of the cable does not match the characteristic impedance of the motor, reflection will occur. Due to the existence of distributed parameters, the cable becomes a coupling path for conducted and radiated EMI. The distribution parameters of the cable mainly depend on the material, geometry, size, physical properties of the insulating material, and the distance between the conductors. The shielding layer is generally a tinned copper mesh with a braided structure. Taking a PVC high-voltage power cable as an example, the cable size and structure are shown in Fig. 3.29. Cable layout is as follows: the cable length is 1500 mm, the AC shielded cable is distributed in parallel, the axial distance is 25 mm, the shielded cable parameters are shown in Table 3.6, and the cable layout is shown in Fig. 3.30. Multi-conductor transmission line theory is used to establish the cable model. Shielded cables are also used as three-phase AC cables. The inner conductor is composed of 64 small copper wires. The cable shielding layer is connected to the ground with the heat sink of the inverter, the motor shell, etc. to form a common mode current circulation circuit. Table 3.6 Shielded cable parameters

Parameters

Values

Parameters

Values

r1

4.0 mm

r2

5.6 mm

r3

6.1 mm

r4

7.5 mm

μ0

1.26 * 10−6 H/m

μr

1

10−12

ε0

8.85 *

F/m

εr

4.3

ρ

1.75 * 10−8 /m

L

1500 mm

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3 EMI Prediction and Suppression of Motor Drive System

Fig. 3.30 Layout of shielded cables

(a) 2D

(b) 3D

➀ Resistance calculation When calculating the resistance of the inner conductor of the shielded cable in low frequency, the copper resistivity of the wire at 20 °C is 1.75 * 10−8 ( * m), the calculation formula is as follows: R=

ρ∗L S

(3.12)

The resistance value of a 1500 mm long cable can be calculated as 0.52 m. ➁ Capacitance calculation The three-dimensional model of the three-phase AC shielded power cable is established in the software ANSYS/Maxwell, the electrostatic field is chosen when calculating the capacitance, and the extracted capacitance of the shielded cable to ground by simulation are shown in Table 3.7. During the capacitor extraction process, it can be clearly seen that after the shielded cable is energized, the voltage and electric field distribution gradually decreases from the outer surface of the inner conductor to the inner surface of the shielding layer, as shown in Figs. 3.31 and 3.32. Table 3.7 Shielded cable capacitance values Parameter

Value

Unit

Parameter

Value

Unit

V-phase-ground

672

pF

U-phase-ground

672

pF

W-phase-ground

672

pF

Fig. 3.31 Shielded cable voltage distribution

3.3 EMI Modeling of Motor Drive System

93

Fig. 3.32 Electric field distribution of shielded cable

➂ Calculation of inductance The transient field is chosen when calculating the inductance. In the process of inductance extraction, the distribution of the magnetic field strength H and magnetic induction strength B of the three-phase cable at different times can be clearly seen. When the rated current of the motor: 258 A, 127 Hz and 120° phase difference is injected to the three-phase UVW cables, the distribution of H and B is shown in Fig. 3.33 a, b. The inductance of the shielded cable extracted by simulation is shown in Table 3.8. (3) Model verification of shielded cable According to the resistance, inductance and capacitance parameters of the threephase shielded cable, the high-frequency equivalent circuit model is established in

Fig. 3.33 H and B distribution

Table 3.8 Inductance value of shielded cable Parameters

Value

Unit

Parameters

Value

Unit

U self-inductance

26

nH

V self-inductance

26

nH

W self-inductance

26

nH

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3 EMI Prediction and Suppression of Motor Drive System

Fig. 3.34 Shielded cable impedance

the electromagnetic simulation software CST, and the impedance parameter Z and the energy scattering coefficient S are analyzed in the 0.15–108 MHz frequency band. With the vector network analyzer, the Z parameter and S parameter of the three-phase AC shielded cable are measured. The simulation results are compared with the test results, as shown in Fig. 3.34 and Fig. 3.35, respectively. It can be found from Fig. 3.34 that the measured values of the shielded cable impedance in the 0.15–20 MHz frequency band agrees well with the simulated values; while they doesn’t agree so well in high frequency bands. The four curves of shielded cables S 11 , S 12 , S 21 , and S 22 in the 0.15–108 MHz frequency band are shown in Fig. 3.35, and the test values have same trends with the simulated values, and have a good correspondence. Simulation and measurement S 11 , S 12 , S 21 , S 22 all have resonance points at 55 MHz. It shows that the cable concentrated equivalent circuit model has high accuracy. (4) Modeling of motor controller DC and AC cable models The motor controller is connected to a DC power source such as a power battery through DC power cables, and connected to the drive motor through three-phase AC cables. Similar to the three-phase AC cable, the DC cable model established in Maxwell is shown in Fig. 3.36. Finite element of the established DC cable model is divided, as shown in Fig. 3.36b, and then through extracting the parameters in Maxwell, the cable resistance, inductance and capacitance and other parameters can be obtained. Combined with the three-phase AC cables model established above, the cable layout of the motor controller can be obtained, as shown in Fig. 3.37. Three-phase AC cables and two DC cables are distributed side by side, and five power cables are distributed on the side of the controller. It can be seen that due to the short distance of the cables, when the motor drive system is working, the large current in the cable will generate a large magnetic field, which will have a certain impact on the adjacent equipment and the human body. When high-frequency interference current flows through the cables, electromagnetic radiation is formed.

Fig. 3.35 Comparison of S-parameter test results and simulation results of shielded cable

(c)S21

(a)S11

(d)S22

(b)S12

3.3 EMI Modeling of Motor Drive System 95

96

3 EMI Prediction and Suppression of Motor Drive System

Fig. 3.36 3D model of DC cables

Fig. 3.37 Motor controller and cable model

(5) Cable electromagnetic field simulation ➀ AC cable Figure 3.38 shows the average values of the frequency domain conducted disturbance current of the three-phase AC lines measured under the 1900 rpm and loading motor conditions. As can be seen from Fig. 3.38, there are couple of obvious conducted current values exceeding the standard limits in the 0.53–1.8 MHz band, and the peak point is at 1 MHz, the dark blue curve is over 45 dBμA, the red and yellow curves are over 35 dBμA, and the sky blue curve is over 25 dBμA. In the 5.9–6.2 MHz frequency band, the dark blue curve, red curve, yellow curve and sky blue curve

Fig. 3.38 Comparison of the average frequency current of AC cables

3.3 EMI Modeling of Motor Drive System

97

each exceed the standard 22, 15, 13 and 20 dBμA; the last excess frequency band appears at 35–48 MHz, and there are couple of conducted current values exceeding the standard limits only in the dark blue curve and sky blue curve, and their peaks appear at 40 and 43 MHz respectively, and the over-scalars are 4 dBμA. According to Fig. 3.38, the frequency points where three spikes appear are selected, which are 0.1, 1 and 5 MHz respectively. It can be seen that the amplitude of the frequency domain current is larger at these three frequency points, and there is a risk of a conducted current exceeding the standard limit. At the same time conducted current will also cause near filed coupling and radiated emission problems. By injecting the currents at three frequency points into the three-phase AC cable model, the electromagnetic field distribution relationship of the cable at these frequency points can be obtained, as shown in Figs. 3.39 and 3.40. ➁ DC cable Under the motor conditions of 3800, 1900, 1000 rpm and 53 N·m, the average value of conducted current on DC positive line were measured at a distance of 50 mm from

(a) 0.1MHz

(b)1MHz

(c)5MHz Fig. 3.39 Three-phase AC cable cross-section and magnetic field distribution diagram

Fig. 3.40 Electric field distribution of three-phase AC cable

(a) Magnetic field distribution

(b) Electric field distribution

98 3 EMI Prediction and Suppression of Motor Drive System

3.3 EMI Modeling of Motor Drive System

(a) Average

99

(b)Peak

Fig. 3.41 Frequency spectrum distribution of mean value of conducted disturbance current of positive line and negative line under different speed conditions

the motor controller, as shown in Fig. 3.41a. Under the motor condition of 1900 rpm and 53 N·m, the average frequency spectrum of the conducted disturbance current of the DC positive and negative lines are shown in Fig. 3.41b. It can be seen from the figure that the measured value of the 3800 rpm speed condition is slightly higher than the other two speed conditions. In the three frequency bands of 0.58–1.8 MHz, 5.9–6.2 MHz and 70–108 MHz, respectively, there are obvious exceedances. At 1 MHz, the over standard is most obviously, about 20 dBμA, and the amount of over standard in the second and third over-frequency bands is basically the same, which is 10 dBμA and 4 dBμA respectively. The green line is the standard mean value limits, the red curve is the measurement result of the conducted disturbance current of the DC positive line, and the blue curve is the measurement result of the conduction disturbance current of the DC negative line. The difference between the positive and negative line measurement results is small. According to the frequency domain current curve of the DC positive and negative cables under various working conditions, three peak frequencies with large current amplitudes are selected, which are 0.1, 1 and 5 MHz, respectively. By injecting currents at three frequency points into the DC cable model, the electromagnetic field distribution relationship of the DC cable can be obtained, as shown in Figs. 3.42 and 3.43. Figure 3.43 is the electromagnetic field distribution diagram obtained by the simulation of the DC cable at a frequency of 1 MHz. According to the equipotential surface, the distribution trends of the electric field and the magnetic field are basically the same. Using this model, the radiated emission of the power line can be predicted. (3) Motor high-frequency model 1) Motor model ➀ Analysis of formation mechanism of CM current The motor parameters that affect the formation of the CM EMI of the motor include: r 0 and L 0 are the equivalent CM resistance and CM inductance of the stator winding, C sf , C sr, and C rf are the equivalent capacitance between stator core and casing, the

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3 EMI Prediction and Suppression of Motor Drive System

(a) 0.1MHz

(b) 1MHz

(c) 5MHz Fig. 3.42 DC cable cross section and magnetic field distribution diagram

(a) Magnetic field distribution (b) Electric field distribution

Fig. 3.43 DC cable electromagnetic field distribution

stator core and rotor, and the rotor and casing. C b and C ins are the bearing oil equivalent capacitance and insulation equivalent capacitance, respectively. Based on these motor parameters, a CM equivalent circuit model of a permanent magnet synchronous motor is established, as shown in Fig. 3.44a, and CM current propagation paths are analyzed, as shown in Fig. 3.44b. uCM is the common-mode interference voltage, which can be obtained through measurement and modeling simulation. The equivalent circuit model is established according to Fig. 3.44, the effect of motor parameters on CM EMI are simulated and analyzed.

3.3 EMI Modeling of Motor Drive System

101

Motor shell

(a) Equivalent circuit diagram of motor CM interference

(b)CM interference current path

Fig. 3.44 Motor CM interference equivalent circuit and CM interference current path

➁ High-frequency equivalent circuit model of three-phase AC motor The EV motor drive system usually adopts SVPWM (space vector pulse width modulation) control. The three-phase PWM pulse voltage output by the inverter power tube IGBT contains a large number of high-frequency components. The three-phase stator windings of the motor are symmetrically distributed along the stator core in the stator slot. The capacitance C ws between the stator winding and the stator core, the selfcapacitance C self of the stator winding, the capacitance C ww of the different phases of the stator winding, and the bearing equivalent capacitance C bearing will has an effect on motor high frequency impedance. According to the distribution of the parasitic capacitance of the PMSM (as shown in Fig. 3.45), the high-frequency equivalent circuit model of the motor shown in Fig. 3.46 is established. In Fig. 3.46, Ru , Rv , Rw are the three-phase stator winding resistance, L u , L v , L w are the effective values of the winding self-inductance, C n represents the winding neutral point and earth capacitance, C wr represents the winding-rotor capacitance, M represents the mutual inductance between different windings. ➂ Three-dimensional model of PMSM PMSM for EV motor drive system is mainly composed of stator core, winding, rotor core, permanent magnet, rotor shaft and bearing. The three-dimensional structure of Fig. 3.45 Parasitic capacitance distribution of PMSM

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3 EMI Prediction and Suppression of Motor Drive System

Fig. 3.46 PMSM high-frequency equivalent circuit model

Fig. 3.47 Schematic diagram of the three-dimensional structure of the motor

Stator core Winding Rotor core Permanent magnets Rotor shaft

a passenger car motor is shown in Fig. 3.47. The motor characteristics parameters are shown in Table 3.9. In the software ANSYS/Maxwell, a three-phase PMSM model is established. First, the electrical and mechanical characteristics of the motor are simulated and analyzed. The input characteristic parameters of the motor are shown in Table 3.10, Table 3.9 Motor characteristics Parameters

Value

Rated power

42

DC voltage

336

Rated speed

3820

Unit

Parameters

Value

Unit

kW

Peak power

120

kW

V

Rated torque

105

N·m

rpm

Motor poles

4

Table 3.10 Input characteristic parameters of PMSM Parameters

Value

Peak voltage

194

Unit

Parameters

Value

Unit

V

Frequency

127

Hz

U-phase voltage phase angle

0

°

V-phase voltage phase angle

120

°

W-phase voltage phase angle

240

°

Rated speed

3820

rpm

3.3 EMI Modeling of Motor Drive System

103

and the motor output rated torque characteristics, three-phase current and threephase induced electromotive force are obtained by simulation, as shown in Fig. 3.48, Fig. 3.49, and Fig. 3.50, respectively. It can be seen from Fig. 3.48 that the motor torque output response includes transient and steady state. After 10 ms, the motor torque stabilized at 105 N·m. Before 10 ms, the motor torque is in the transient response process, the maximum starting torque T st is 230 N·m, the ratio of maximum starting torque to the steady state torque is defined as the coefficient k t , by the formula kt =

T st Te

Fig. 3.48 Motor output torque waveform

Fig. 3.49 Stator three-phase winding current waveform

(3.13)

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3 EMI Prediction and Suppression of Motor Drive System

Fig. 3.50 Three-phase winding input voltage and induced electromotive force waveform

The coefficient k t can be calculated as 2.2. From the torque response curve within 0–10 ms, it can be seen that the rise time is 1 ms and the number of oscillations is 1, indicating that the established motor model has small torque ripple and good starting performance. The curve of the three-phase winding current of the motor with time is shown in Fig. 3.49. The current shows two processes of transient and steady state at startup. After 10 ms, the peak current of the three-phase winding of the motor stabilized at 249A. Before 10 ms, the maximum amplitude I st of the three-phase winding current transient response of the motor is 365 A, and the ratio of the maximum peak current to the steady state current is defined as the coefficient k i , which is determined by the formula ki =

I st Ie

(3.14)

It can be calculated that the coefficient k i is 1.5, and it can be seen from the starting current curve within 0–10 ms that the established motor model has small starting current ripple and good starting performance. The input voltage and induced electromotive force of the three-phase winding of the motor both exhibit sinusoidal periodic changes, as shown in Fig. 3.50, which is based on Faraday’s law of electromagnetic induction eUU = −

dψ UU dt

(3.15)

According to Kirchhoff’s second law, the voltage equation of winding U can be written as uU = RU i U − eUU

(3.16)

3.3 EMI Modeling of Motor Drive System Table 3.11 Motor input and output characteristics

Parameters Torque

105 Rated value

Simulation value

Unit

105

105

Rotating speed

3820

3820

N·m

Winding peak current

258

249

A

Winding peak voltage

194

188

V

rpm

In formula (3.15) and formula (3.16), eUU represents the induced electromotive force of winding U, ψ UU represents the flux linkage of winding U, uU represents the input voltage of winding U, RU represents the resistance of winding U, and iU represents the input current of the winding U. It can be seen from Table 3.11 that the error between the simulation results of the motor characteristic parameters and the actual parameters is small, indicating that the three-dimensional model of the motor can simulate the actual operation of the motor. ➃ High-frequency equivalent circuit parameter extraction of threedimensional model of PMSM (1) Resistance calculation Considering the complicated connection of the motor winding and the influence of the end winding, the resistance value of the motor phase winding is directly extracted by the RMxprt module in the finite element electromagnetic analysis software Ansys Maxwell, and its value is 5.9 m. (2) Calculation of distributed capacitance Using the three-dimensional model of PMSM established in the software ANSYS/Maxwell, the electrostatic field was selected to extract the capacitance value, and the grid division is shown in Fig. 3.51. When the U, V, W three-phase windings of the motor are injected with the peak excitation voltage of 194 V, during the calculation of the distributed capacitance of the motor, the voltage and electric field distribution after the excitation voltage is injected into the motor can be clearly seen, Fig. 3.51 Meshing of 3D motor model

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3 EMI Prediction and Suppression of Motor Drive System

Fig. 3.52 Motor voltage and electric field changes

Table 3.12 Motor capacitance Parameters

Value

Unit

Parameters

Value

Unit

U phase-stator

360

pF

V phase-stator

360

pF

W phase-stator

360

pF

U phase itself

360

pF

V phase itself

360

pF

W phase itself

360

pF

97

pF

U phase-W phase

97

pF

W phase-V phase

97

pF

Stator-rotor

605

pF

U/V/W phase-rotor

12

pF

U phase-V phase

as shown in Fig. 3.52. Table 3.12 is the motor parasitic capacitance extracted by simulation. (3) Inductance calculation The transient field is selected to calculate the motor winding inductance. When the U, V, W three-phase windings of the motor are injected with 194 V, 127.33 Hz, and 120° phase difference voltage which is excitation as the peak phase voltage under the motor rated operating conditions. During the simulation calculation of the motor inductance, after the excitation voltage is injected into the motor, the distribution of the magnetic field strength H and magnetic induction B can be clearly seen in Fig. 3.53. Figure 3.54 is the time-varying curve of motor winding inductance, and Table 3.13 is the motor winding inductance extracted by simulation. (2) Motor model verification ➀ Motor electrical parameters The electrical parameters of PMSM for EVs are measured using resistance testers and LC meters. The comparison between measurement results and simulation results of motor resistance and inductance are shown in Tables 3.14 and 3.15. According to Tables 3.13, 3.14, and 3.15, it can be seen that the deviation between the test results of the motor and the parameters extracted by the modeling simulation is small, which

3.3 EMI Modeling of Motor Drive System

Fig. 3.53 Magnetic field distribution at 0.001 s

Fig. 3.54 Motor winding inductance curve

107

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3 EMI Prediction and Suppression of Motor Drive System

Table 3.13 Motor winding inductance Parameters

Value

Unit

Parameters

Value

Unit

U-phase self-inductance

0.44

mH

V-phase self-inductance

0.44

mH

W-phase self-inductance

0.44

mH

U-V mutual inductance

0.23

mH

U-V mutual inductance

0.23

mH

U-V mutual inductance

0.23

mH

U-phase effective inductance

0.21

mH

V-phase effective inductance

0.21

mH

U-phase effective inductance

0.21

mH

Table 3.14 Motor resistance test results and simulation results Measurement parameters

Value

Unit

Simulation parameters

Value

Unit

U-V phase winding

12.2

m

U phase winding

5.9

m

V-W phase winding

12.2

m

V phase winding

5.9

m

U-W phase winding

W phase winding

5.9

m

12.2

m

U phase winding

6.1

m

V phase winding

6.1

m

W phase winding

6.1

m

Table 3.15 Motor inductance measurement results and simulation results Measurement parameters

Value Unit Simulation parameters

U-V phase winding

0.272 mH

U-phase winding effective value 0.21

mH

V-W phase winding

0.316 mH

V-phase winding effective value

0.21

mH

U-W phase winding

0.498 mH

W-phase winding effective value 0.21

mH

U-phase winding effective value 0.2

mH

V-phase winding effective value

0.2

mH

W-phase winding effective value 0.2

mH

Value Unit

indicates that the accuracy of the established three-dimensional model of the motor is high. ➁ Verification of high-frequency equivalent model of motor According to the resistance, inductance and capacitance parameters of the threephase PMSM, its high-frequency equivalent circuit model is established in the electromagnetic simulation software CST to analyze the impedance characteristics of the three-phase winding of the motor. A vector network analyzer is used to measure the impedance characteristics of the three-phase winding of the motor. The test layout is shown in Fig. 3.55. The motor impedance simulation results with the test results are compared in Fig. 3.56. It can be seen from Fig. 3.56 that in the 0.15–108 MHz frequency band, the change trend of the impedance test results and the simulation results of the PMSM is

3.3 EMI Modeling of Motor Drive System

109

Fig. 3.55 PMSM impedance test layout

Fig. 3.56 Comparison of impedance test results and simulation results of PMSM

consistent. The impedance simulation and measurement results have four identical resonant frequency points, respectively 5.8, 22.7, 47, 71 MHz, indicating that the simulation and measurement have good consistency, and that the established motor equivalent model has high accuracy. (4) Modeling of high-frequency circuits in motor systems ➀ Equivalent circuit modeling of CM interference Based on the parameters of the resistance, inductance and parasitic capacitance of the three-phase shielded cable and the PMSM, an equivalent circuit model of the PMSM drive system CM interference is established in CST software, as shown in Fig. 3.57. The S-parameters and Z-parameters are simulated and analyzed in the frequency band 0.15–108 MHz. The simulation results are shown in Fig. 3.58 and Fig. 3.59.

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3 EMI Prediction and Suppression of Motor Drive System

Fig. 3.57 Equivalent circuit of CM interference of motor drive system

Fig. 3.58 S-parameter simulation results of CM interference equivalent circuit

Fig. 3.59 Z parameter simulation result of CM interference equivalent circuit

3.3 EMI Modeling of Motor Drive System

111

Fig. 3.60 High-frequency DM equivalent circuit of the motor drive system

Fig. 3.61 S-parameter simulation results of the high-frequency DM equivalent circuit of the motor drive system

➁ Equivalent circuit modeling of DM interference The equivalent circuit model of the PMSM drive system DM interference is established in CST, as shown in Fig. 3.60, the S and Z parameters are simulated and analyzed in the frequency range of 0.15–108 MHz, and the simulation results are shown in Figs. 3.61 and 3.62.

3.3.2 Co-simulation of Motor Drive System (1) Simulation based on physical prototype The established IGBT electrical and thermal models, high-voltage cables, IGBT packaging models, and PMSM are connected to form a complete motor drive system model. As shown in Fig. 3.63, the conducted EMI of the system is simulated and analyzed. Software Simplorer can call the SVPWM control algorithm built in

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3 EMI Prediction and Suppression of Motor Drive System

Fig. 3.62 The Z parameter simulation results of the high-frequency DM equivalent circuit of the motor drive system AC Cables LISN A

DC Cables

A A

V

M motor V

IGBT electrical model and thermal model

Control algorithm

IGBT package

Fig. 3.63 Co-simulation of the motor drive system

Simulink, IGBT package module and cable model in Q3D at the same time, and can be connected to the motor model established by Maxwell, that is, the four software Simplorer, Simulink, Q3D and Maxwell can simulation together. The impedance matching network (LISN) is connected to the output end of the high- voltage DC power supply of the inverter to simulate the receiver, and receive the conducted EMI signal transmitted by the DC cable, as shown in Fig. 3.64. As can be seen from Fig. 3.64, the voltage across the 50 resistor in LISN oscillates between ± 50 V, and

3.3 EMI Modeling of Motor Drive System

113

(a)Time-domain voltage curve

(b)Frequency-domain voltage curve Fig. 3.64 Characteristics of voltage signals across 50  in LISN

there are sharp peaks at 5, 10, 15, 20, 25 kHz and other harmonic frequency points, which are related to the IGBT switching frequency. (2) Modeling and simulation of high-frequency equivalent circuit The SVPWM control model of the motor drive system is established in MATLAB/Simulink, as shown in Fig. 3.65a, to obtain the IGBT gate input signal. the high-frequency equivalent circuit model of the motor drive system is established in ANSYS/Simplorer, and by co-simulation with MATLAB/Simulink, the system high-voltage positive bus conducted interference voltage is obtained, as shown in Fig. 3.65b. The comparison of simulation and test results is shown in Fig. 3.65c. Consistent with the changes of the test results, the simulated conducted voltage forms a resonance around 1 and 30 MHz, producing a higher resonance peak voltage. Therefore, the model can be used to design the conducted interference filtering device.

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3 EMI Prediction and Suppression of Motor Drive System

(a)SVPWM control model

(b)High-frequency equivalent circuit model of motor drive system

(c) Comparison of conducted voltage simulation results and test results Fig. 3.65 System conducted EMI modeling and simulation

3.4 Spice Equivalent Circuit Modeling of Electromagnetic Emission …

115

3.4 Spice Equivalent Circuit Modeling of Electromagnetic Emission of Electric Drive System Based on Measurement 3.4.1 Overview of Modeling Methods High dv/dt and di/dt components will be generated when high-power semiconductor devices are turned on and off quickly. These components will produce conducted and/ or radiated emissions through the parasitic parameters of the system, which are easy to form electromagnetic interference (EMI) (Zhong and Lipo 1995; Ran and Gokani 1998; Skibinski et al. 1997). EMI is a major challenge for the design of high-power converters such as drive motor controller and steering motor controller in EV. In order to improve the efficiency and reduce the loss of the insulated gate bipolar transistor (IGBT) in motor inverter, the switching speed of IGBT must be fast enough. Therefore, the values of dv/dt and di/dt are high, which is difficult to reduce. However, the electromagnetic emission can be reduced by improving the electrical architecture design of the inverter, such as the design of the circuit, heatsink, wiring harness, grounding and other structures, adding filters and shielding, etc. usually, these EMI suppression technologies are evaluated through the accurate model of the system in the early product design process. Usually, these EMI suppression techniques could be evaluated through accurate models of the system in the early design process of a product for EV. The full wave model is usually used in electromagnetic simulation. Although the full wave model is accurate, the structure of the motor inverter is complex, and the application of the full wave model requires a lot of calculation time and memory (Chen et al. 2003; Lai et al. 2006). More importantly, this full wave model is often a “black box”, which cannot directly show which parts of the system are responsible for a particular EMI problem. In addition, the nonlinear components of the inverter model cannot be easily modeled by a full wave solver, so circuit analysis should be used to solve the problem. The SPICE-based equivalent model with parasitic parameters is a better method because it can directly give the corresponding relationship between the geometry of the system and the components of the parasitic circuit and the generated common mode (CM) currents or differential mode (DM) currents. A SPICE-based model can be obtained from schematics, harness information, and system layout (e.g., the IGBT module, heatsink, and enclosure geometry). Several methods are available for extracting parasitic SPICE parameters from a complex geometry (Falck et al. 1997; Xing et al. 1998; Schanen et al. 1996). Many of these methods are based on 3-D finite-element analysis (Chen et al. 2003; Lai et al. 2006) or the partial element equivalent circuit (PEEC) method (Ardon et al. 2009, 2010; Ruehli 1974; Lionet et al. 2007). The SPICE-based model, such as the S parameter between two ports, is obtained by the black box measurement of the circuit characteristics by finite element analysis. The spice model of parasitic parameters can be obtained by using PEEC method according to RLGC matrix, but hundreds or thousands of elements may be needed to represent a simple geometry. Although

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3 EMI Prediction and Suppression of Motor Drive System

the model reduction (MOR) technology can be used to represent the equivalent circuit with S parameter, which can not understand the important physical characteristics of the internal components of the equipment (Wittig et al. 2006). Although the equivalent SPICE circuit model can be determined by the Z parameters of the three-dimensional full wave model (Traub et al. 2012), it is not only difficult to model a complete power inverter with complex geometry, but also requires a lot of time and effort. The parasitic parameters can also be obtained by the combination of measurement and full wave simulation. Time domain reflectometry (TDR) and transmission line theory are used to extract parasitic parameters (Zhu et al. 1999) to extract precise parasitic parameters. The equivalent circuit of inverter module can be determined by impedance measurement method (Yang and Odendaal 2007). Due to the impedance mismatch sometimes, this method has some limitations. In reference (Reuter et al. 2013), the S parameter is converted into equivalent common mode and differential mode impedance to model the inverter. Although the model is based on measurement, it is still the “black box” model of the inverter. The model of low frequency induction motor is established by using DM and cm impedance measurement (Sun and Xing 2013). A simple and accurate equivalent model of motor power inverter is proposed. The parasitic circuit elements of the model have obvious correlation with the geometry of the system (Bondarenko et al. 2015; Zhai et al. 2015). This method is suitable for frequencies up to 100 MHz, covering the frequency range of conducted emission of vehicle parts. The motor drive system consists of DC power cables subsystem, DC link capacitor block with DC bus bar, IGBT module, AC bus bar, AC power cables, and motor/ load. Firstly, a simple equivalent model of each subsystem is established and verified by measurement. Then, the subsystem models were assembled to create an equivalent model of the complete motor drive system. Using this models, the most important system components responsible for resonance can be found out, and then the suppression methods mitigating these resonances are proposed.

3.4.2 Construction of the Equivalent Circuit The structure of three-phase AC motor drive system is shown in Fig. 3.66. The equivalent circuit should include high voltage power cables model (both DC and AC side), lead parasitic inductance of Y capacitors, parasitic inductance of DC link capacitor, inductance of neutral point lead end of IGBT module, capacitance between IGBT and chassis, and high frequency impedance model of motor. Since the structure of PWM inverter is symmetrical, it is necessary to model the switching behavior of only one inverter leg. The inverter is packaged in a metal enclosure with good shielding effectiveness. The high and low voltage power cables entering the inverter are shielded, and the cable shielding layer forms a good 360° connection with the shell connector. The simple model of IGBT noise source is shown in Fig. 3.67. For each IGBT, there

3.4 Spice Equivalent Circuit Modeling of Electromagnetic Emission …

117

Fig. 3.66 Power inverter/motor system

Fig. 3.67 Location of the main “noise” source

is parasitic capacitance from emitter to chassis, from collector to chassis, and from phase mid-node to chassis. When IGBT is turned on or off, CM voltage will be generated at the phase mid-node, which can be used as the direct driving between the central conductor and the shielding layer of AC cable connected to the motor. Therefore, many electromagnetic emissions are caused by the voltage from phase mid-node to chassis. The equivalent antenna which forms electromagnetic emission is mainly composed of cable shielding layer, shell of the motor and inverter. Since the conducted disturbance measurement standard specifies the location of these components, the antenna characteristics will not change significantly during the test. Since the antenna is mainly driven by the voltage at the AC cable end, which is mainly generated by the voltage between the IGBT phase node and the chassis, the value of S21 between the phase-node-to-chassis voltage and the center-conductorto-shield voltage of the AC cable connecting the motor to the shielding layer is very important for the study of the methods to suppress the conducted and radiated emission of the motor drive system for EV. ➀ AC and DC Cables The dc and ac cables were modeled as transmission lines. This model requires information like the characteristic impedance, dielectric constant of the insulation, and loss tangent. The datasheet provided only geometrical information, so measurements were made with a TDR and a vector network analyzer and parameters were determined from the measurements. The 1.86 m long cable with Z 0 = 8.56 , ε r =

118

3 EMI Prediction and Suppression of Motor Drive System

Fig. 3.68 Validation of the transmission line model for the open-ended dc cable. Measured values are shown with a solid line and simulated values with a dashed the IGBT module

2.65, and loss tangent of 0.108 is modeled and simulated for transmission line. The comparison results of simulation value and measured value of cable impedance Z11 are shown in Fig. 3.68. L probe is a 4 nH parasitic inductor to simulate the SMA connector connected in series with the cable for measurement. ➁ DC Link The DC link capacitor module consists of DC link capacitor (C Link = 1028 μF) and two Y capacitors (C Y-cap = 0.98 μF), as shown in Fig. 3.69. The inductors L DC-bus bar and M DC-bus bar are caused by the bus bar of capacitor, L 2 and L Link are parasitic inductance related to C Link , L Y-cap and M Y-cap are related to Y capacitor, L 3 is inductance caused by capacitor output (it is connected to IGBT module). For the measurement of DC link filter capacitor module, the capacitor module should be measured separately, and the capacitor module connected with DC cable should be measured to determine the parasitic inductance caused by Y capacitor and DC bus metal bar. The values of Z11 and Z22 are measured when the other ports are made open or short. By measuring the impedance between one of the Y capacitors and the chassis, The value of Z11 is obtained and the value of Y capacitor is calculated. The value of parasitic inductance can not be determined by a single measurement, but can be determined from a set of measured values. The measured Fig. 3.69 Schematic of dc link

3.4 Spice Equivalent Circuit Modeling of Electromagnetic Emission …

119

Table 3.16 Equations describing DC-Link capacitor block inductances Measurement

Equation

DC cable, dc-link block, Y-caps

L @1MHz ≈ L DC cable + L DC−bus bar +

DC cable, dc-link block, DC link, looking into Y-caps

L @2MHz ≈ L DC cable + L DC bus bar + L 2 + L Link

L @10MHz ≈ 2 L Y −cap − MY −cap + L 2 + L Link

DC-link input shorted, Y-caps open

L @14MHz ≈ L 3 + L Link ||(L DC−bus bar + L 2 )

DC link, Y-caps open

L @11MHz ≈ L 3 + L Link

L Y −cap +MY −cap 2

values and the related equations are shown in Table 3.16, and the parameters used in the equation are shown in Fig. 3.69. ➂ IGBT module For the modeling of IGBT module, the value of Z11 is measured twice on the output of the module while the input is open or short circuited in order to determine the IGBT junction capacitance and the loop inductance caused by IGBT bus. As shown in Fig. 3.70, when DC− and DC+ terminals are short circuited, the measured junction capacitance of pull-up and pull-down IGBTs is about 13nF. The predicted value of the loop inductance associated with the single-phase branch of the IGBT module is about 26 nH. The parasitic capacitance from the phase node to the case is about 850pf measured by LCR. According to the measured total capacitance and the conductive area of collector, emitter and IGBT phase node on the IGBT substrate, each parasitic capacitance is estimated to be 412, 380 and 89 pF, as shown in Fig. 3.70 (Yang and Odendaal 2007). The parasitic total capacitance, 850 pF, is very important to the resonance mechanism analysis and resonance suppression. ➃ AC-Bus Bars In order to predict the loop inductance associated with the AC-bus bar, which transmits current from the output of the IGBT to the AC cable. The value of Z11 is measured looking into the output of the IGBT module with the input open circuit and the AC bus bar end short circuited to the chassis. This setting creates two current

Fig. 3.70 Equivalent circuit model of single phase IGBT module

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3 EMI Prediction and Suppression of Motor Drive System

loops: one from the input port through the AC bus bar and the chassis, and the other from the input port through IGBT connection, and the parasitic capacitance flows to the chassis, as shown in Fig. 3.70. The self-induction of IGBT output connector and the AC bus bar is about 25 and 130 nH respectively from Z11. ➄ Complete System The whole circuit model is assembled according to the models of individual subsystems, as shown in Fig. 3.71. The model is suitable for single IGBT bridge arm and one AC cable. Because the impedance value of the motor is large in the frequency band specified in the standard, the establishment of the high frequency model of the motor is ignored. The parameters of the whole circuit are shown in Table 3.17.

Fig. 3.71 Complete equivalent circuit model. Port one: between phase node and chassis. Port 2: between inner and outer conductors at the end of the ac cable

Table 3.17 Values of components within equivalent circuit

Component

Value

Component

Value

L DC-bus bar

50 nH

M DC-bus bar

40 nH

C Y-cap

700 nF

LY-cap

150 nH

M Y -cap

108 nH

L2

2 nH

L3

12 nH

CLink

1028 μF

L Link

10 nH

C Junction

13 nF

L IGBT

26 nH

C phase-to-chassis

412 pF

C collector-to-chassis

89 pF

C emitter-to-chassis

380 pF

L IGBT con

25 nH

L AC-bus bar

130 nH

M AC-bus bar

12 nH

3.4 Spice Equivalent Circuit Modeling of Electromagnetic Emission …

121

Fig. 3.72 Measurement setup used to validate overall inverter model

3.4.3 Validation of the Complete Model Due to the small coupling between subsystems, it must be verified by measuring the complete assembled system. The model was validated by comparing the input impedance and/or S-parameters at ports of both the individual subsystems as well as the overall model. The complete circuit model was validated using the experimental setup shown in Fig. 3.72. Port 1 was placed between the phase node and chassis (at the source of switching) and Port 2 was placed between the inner and outer conductors at the end of the ac cable (where the noise is most likely to drive radiated emissions). Figure 3.73a shows a comparison of simulated and measured transfer characteristics between these ports, i.e. S21 ranging from 100 kHz to 1 GHz. The measured and simulated values of Z11 and Z22 are shown in Fig. 3.73b. The error of the system is less than 4 dB in the frequency range of 100 kHz to 100 MHz. It should be noted that for frequencies above 100 MHz, there are some differences in S21 level, which may be caused by parasitic coupling between subsystems. Here we only consider the frequency band below 100 MHz required by the conducted emission standard.

3.4.4 System Resonance Suppression Method The resonance in the inverter leads to the peak value of transmission characteristics from the phase node to the output of AC cable, which may also lead to the appearance of radiation emission peak. If we can determine which components or current paths are related to these resonances, their influence can be eliminated or suppressed. (1) Two port network of power circuit in motor inverter system The topological structure of the cascade circuit of the motor drive system is established, and the spice equivalent circuit model of the electromagnetic emission of the three-phase power converter of the permanent magnet synchronous motor based on the measurement system behavior level is constructed. The transmission characteristics of DM interference source −DC side, CM interference source −DC side, DM interference source −AC side and CM interference source −AC side are studied, as shown in Fig. 3.74. The impedance characteristics and interference current path of

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3 EMI Prediction and Suppression of Motor Drive System

(a) Magnitude and phase of S21 between the phase node and the output of the ac cable.

(b)Magnitude of Z11 looking into the phase node and Z22 looking into the ac cable output. Fig. 3.73 Comparison of measured and simulated values

resonant frequency point are studied to determine the circuit and components that generate resonance. (2) Path analysis of resonance point At present, the existing EMI suppression methods are mostly carried out outside the motor controller or power port, and the key components causing interference in the system are rarely analyzed. This method may lead to a new resonance point in the EMI suppressed system, resulting in new interference. Therefore, by determining the internal components responsible for EMI, EMI control and EMI suppression

3.4 Spice Equivalent Circuit Modeling of Electromagnetic Emission …

(a) DM interference source-DC side

(c) DM interference source - AC side

123

(b) CM interference source-DC side

(d) CM interference source-AC side

Fig. 3.74 Various forms of two port equivalent circuit network model of the motor drive system

in the motor controller can fundamentally solve the problem of electromagnetic interference. As the main interference source, IGBT produces broadband interference, which forms CM interference and DM interference through system circuit and distributed parameters. The interference is mainly transmitted by DC cables and AC cables. Taking the influence of IGBT interference on AC cable port as an example, the main components of resonance are analyzed. When the parallel resonance occurs in the system circuit, the energy at the resonance point is the highest, and the interference is amplified here. Therefore, it is necessary to study the components responsible for the resonance to suppress EMI. Taking 100 kW motor as the research object, the vector network analyzer is used to measure the internal parameters of the motor inverter system. The parameters of the motor system are shown in Table 3.18. The equivalent circuit model of the two port network of the motor inverter system as shown in Fig. 3.71 is established. According to the equivalent circuit model, the PSICE model of the system is established in ADS software, as shown in Fig. 3.75. In the PSPICE model of the two port network, port1 connects the phase nodes of two IGBTs with the ground, representing the CM interference source generated by IGBT, and port2 connecting the conductor of the single-phase cable and ground of AC motor, representing the interference generated on the single-phase AC cable of the motor. The transmission characteristics of the two ports are shown in Fig. 3.76. The deviation between the simulation results and the test results is less than 6 dB, which indicates that the model has high accuracy. It can be seen from Fig. 3.76 that S21 and Z11 have the same change, so S21 can be analyzed by analyzing the characteristics of Z11. It can be seen from Z11 that parallel

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3 EMI Prediction and Suppression of Motor Drive System

Table 3.18 Values of components within equivalent circuit Elements

Meaning

Value

L DC bus bar

Inductance of DC bus bar

50 nH

MDC bus bar

Mutual inductance of L DC bus bar

40 nH

CY−cap

Capacitance of Y capacitor(Y-cap)

700 nF

L Y−cap

Parasitic inductance of the Y-cap

150 nH

MY−cap

Mutual inductance of L Y−cap

108 nH

L1

Inductance between Y-cap and DC link capacitor

12 nH

L2

Inductance between DC link capacitor and IGBT

2 nH

Clink

Capacitance of the DC link capacitor

1028 uF

L link

Parasitic inductance of the DC link capacitor

10 nH

L IGBT

Lead inductance of IGBT

26 nH

Cphase to chassis

Capacitance of the IGBT phase node to the chassis

412 pF

CJunct

Capacitance between IGBT emitter and collector

13 nF

Ccollector to chassis

Capacitance of the IGBT collector to the chassis

89 pF

Cemitter to chassis

Capacitance of the IGBT emitter to the chassis

280 pF

L IGBT con

Inductance between IGBT phase node and AC bus bar

25 nH

L AC bus bar

Inductance of AC bus bar

130 nH

MAC bus bar

Mutual inductance of L AC bus bar

12 nH

Fig. 3.75 PSPICE model of two port network of motor inverter system in ADS

resonance and common mode interference can be produced at 6, 11 and 26 MHz. Therefore, it is very important to find out the high frequency devices which cause. (1) Analysis of Current Path for 6 MHz The parallel resonance of three frequency points. Next, the interference current source is applied to port1, and the interference current paths of 6, 11 and 26 MHz are analyzed respectively. Through equivalent circuit modeling, the main capacitance

3.4 Spice Equivalent Circuit Modeling of Electromagnetic Emission … Mesurement

S21[dB]

0

125

Simulation

-20 -40 -60 5 10

6

7

10

10 Frequency [Hz] 6MHz 11MHz

8

10

26MHz

60 Z11[dB]

40 20 0 -20 5 10

6

7

10

8

10

10

Frequency [Hz]

Fig. 3.76 Magnitude of S21 and Z11

and inductance responsible for parallel resonance and corresponding devices are determined, so as to propose methods and specific measures to reduce interference. The impedance of each circuit element at 6 MHz was calculated as shown in Fig. 3.77 to find the elements responsible for the resonance at 6 MHz. The Port 1 between the phase node and chassis is supposed to be a CM current source and the CM current paths for 6 MHz are shown in Fig. 3.77. The current paths are mainly composed of a dc side path and an ac side path. However, only the dc side current flowing path is analyzed due to the impedance of the ac side circuit is about –j25 and much higher than that of the dc side, as shown in Fig. 3.78. The CM current flows from the plus of the Port 1 to the plus and minus of the dc cables symmetrically through junction capacitance across both the pull-up and pull-down IGBT C junct and results

-j15 MDC bus bar

-j15

j0.1 j1.9 LDC bus bar L1 . LY-cap j5.7 . CY-cap MY-cap CY-cap . L . Y-cap j5.7 LDC bus bar j1.9

j0.5 L2

j1 LIGBT

LLink

CJunct -j2

CLink Cphase

j1 . LIGBT

to chassis

CJunct -j2

LIGBT j1

Cemitter

Ccollector

to chassis

to chassis

Fig. 3.77 Current following path at 6 MHz

con

MAC bus bar j5 . LAC

-j25

bus bar

Port1

Port2

126

3 EMI Prediction and Suppression of Motor Drive System -j15

j3.4

CDC cable (L+M)DC bus bar . L2 -j0.04

0.08

.

j10

CY-cap (L+M)Y-cap

L3

j0.1 j0.5

MDC bus bar MY-cap CY-cap (L+M)Y-cap .

-j0.04

.

j10

CDC cable (L+M)DC bus bar -j15

j3.4

j1

LIGBT CJunct

-j2 (L-M)IGBT con

CJunct

-j2

LIGBT

j0.45 Port1

j1

Fig. 3.78 Equivalent circuit at 6 MHz

in two current flowing paths. Each path is divided into two parallel paths through L IGBT . One path is through the dc bus bar and the dc cable to the chassis, the other one is through Y-cap to the chassis. Finally the CM current flows back to the minus of the Port 1. The impedance of the DC cable is about −j15 and mainly capacitive at 6 MHz for its impedance characteristic (CISPR 2009). The impedance of the dc bus bars is about j2π f (L DC bus bar +MDC bus bar ) = 2π 6e6 Hz·(50 nH+40 nH) ≈ j3.4 , so the effective impedance for the path is −j11.6 and −j5.8 with the two paths in parallel. The capacitance of the Y-cap is very small and can be negligible, and the inductance of Y-cap is about j2π f (L + M) = 2π 6e6 Hz · (150 nH + 108 nH) ≈ j10 , so the impedance of the two branches in parallel of Y-caps is inductive as j5 . Finally, the capacitive reactance of the dc bus bars path (−j5.8 ) resonates with the inductive reactance of Y-cap path (j5 ). It is obviously that the capacitances of the two parallel branches due to the dc cable and the dc bus bar could generate a parallel resonance with the inductances of the two parallel branches due to the Y-cap. Therefore, the elements responsible for the resonance at 6 MHz mainly are the capacitances of the dc cables, the inductances of the dc bus bar, and the inductances of Y-cap. (2) Analysis of Current Path for 11 MHz The current flowing paths at 11 MHz are mainly composed of a dc side path and an ac side path, which is similar with that at 6 MHz, as shown in Fig. 3.79. However, the impedance of capacitance and inductance of the dc cables at 11 MHz are about −j8.2  and j8.8  respectively. The CM current of the ac side flows from the plus of the Port 1 to the ac bus bar and the ac cable and back to the minus of the Port 1. The impedances of the ac side circuit and the dc side circuit are about − j5  and j4.5  shown in Fig. 3.80, which are close. Therefore, the ac side current path cannot be neglected. The effective inductance of the dc side circuit generates a parallel resonance with the effective capacitance of the ac side. Therefore, the elements responsible for the resonance at 11 MHz mainly are the capacitances of the dc cables, the inductances of the dc bus bar, the inductances of the Y-cap, the inductances of the ac bus bar, and the capacitances of the ac cable.

3.4 Spice Equivalent Circuit Modeling of Electromagnetic Emission …

LY-cap

-j8.2

CY-cap

MDC bus bar

-j8.2 j8.8

j0.14 L1

j3.5 L.DC bus bar

j8.8

.

j0.83 j1.8 L2 LIGBT

j10.4

LLink

MY-cap

CLink

CY-cap . j10.4 . LY-cap LDC bus bar j3.5

127

CJunct -j1.1

j1.73 . LIGBT

Cphase CJunct -j1.1

to chassis

LIGBT j1.8

con

Cemitter

Ccollector

to chassis

to chassis

MAC bus bar . j8.98 LAC

-j13.2

bus bar

Port2

Port1

Fig. 3.79 Current following path at 11 MHz -j8.2

j8.8

j6.2

-j0.02

j17.8

CDC cable LDC cable (L+M)DC bus bar . L1 .

CY-cap (L+M)Y-cap

LIGBT

L2

j0.14 j0.83

MDC bus bar MY-cap CY-cap (L+M)Y-cap .

-j0.02

.

j17.8

CDC cable LDC cable (L+M)DC bus bar -j8.2

j8.8

j6.2

j1.8 -j1.1

CJunct

-j1.1

CJunct

LIGBT

(L-M)IGBT con . j0.9

(L-M)

.

AC bus bar

j8.16

Port1

CAC cable

j1.8

-j13.2

Fig. 3.80 Equivalent circuit at 11 MHz

(3) Analysis of Current Path for 26 MHz The CM current flowing paths at 26 MHz are also composed of a dc side path and an ac side path shown in Fig. 3.81. However, the dc side path at 26 MHz is different from that at 6 and 11 MHz due to the CM current flowing to the chassis not only from the dc bus bar, the dc cables and the Y-cap, but also the distributed capacitance from j8.2 j0.3 j2 j21 LDC bus bar L L 2 1 . LY-cap j24.5 -j3.5 LLink . CY-cap MDC MY-cap CLink Cphase bus bar CY-cap . to chassis -j3.5 . LY-cap j24.5 -j14.86 LIGBT L DC bus bar j21 j8.2 j4

j4 LIGBT CJunct -j0.47 CJunct -j0.47

LIGBT con

MAC bus bar j21.24 . LAC -j5.72 bus bar

Cemitter

Ccollector

to chassis

to chassis

-j16.11

Fig. 3.81 Current following path at 26 MHz

j4.08 .

Port1

Port2

128

3 EMI Prediction and Suppression of Motor Drive System

.

LIGBT j4

LIGBT j4 . (L+M)

(L+M)

.

DC bus bar

j14

(L+M)

j21

j40.5

LDC cable

Y-cap

.

(L+M) Y-cap

j40.5

CDC cable -j3.5

.

(L-M) (L-M)

IGBT con

j2.12

j19.28 Port1

DC bus bar

j14

Cphase

j21

to chassis

LDC cable CDC cable -j3.5

-j14.86 -j16.11

.

AC bus bar

CJunct

CAC cable

-j0.47

-j5.72

Ccollector to chassis

Fig. 3.82 Equivalent circuit at 26 MHz

the phase node to the chassis (C phase to chassis , Ccollector to chassis and Cemmiter to chassis ) to the chassis. Equivalent circuit for 26 MHz is shown in Fig. 3.82, where the effective impedance of inductance for the three loops in parallel is about j6  and the effective impedance of capacitance for the forth loop is about—j5.7 . Therefore the elements responsible for the resonance at 26 MHz mainly are the capacitance of the dc cables, the inductance of the dc bus bar, the inductances of the Y-cap, the capacitance of the IGBT phase node to the chassis, the capacitance from IGBT emitter to the chassis, the capacitance of the ac cable and inductance of the ac bus bar. The elements responsible for critical resonances about 6,11, and 26 MHz are listed in Table 3.19. (3) Mitigation of system resonances According to the analysis of current paths at 6, 11, and 26 MHz, the CM emission from the phase node of two IGBTs of one phase bridge leg of the IGBT module due to fast IGBTs switching can be equivalent to a current source as shown in Fig. 3.83. CM noise current I P1 is mainly composed of two current flowing paths, one is through the dc side elements, another one is through the ac side elements. As shown in Fig. 11, Table 3.19 Elements responsible for resonances Frequency

Responsible elements

6 MHz resonance

Capacitances of DC cables, inductances of DC bus bar and inductances of Y-cap

11 MHz resonance

Capacitances of DC cables, inductances of DC bus bar, inductances of Y-cap, inductances of AC bus bar and capacitances of AC cables

26 MHz resonance

Capacitances of DC cables, inductances of DC bus bar, inductances of Y-cap, capacitance from the IGBT phase node to the chassis, capacitance from IGBT emitter to the chassis, capacitances of AC cable and inductances of AC bus bar

Fig. 3.83 Current source equivalent circuit

IDC ZDC Port1

IAC IP1 ZAC

3.4 Spice Equivalent Circuit Modeling of Electromagnetic Emission …

129

it is expressed as IP1 =IDC +IAC and IDC · Z DC = IAC · Z AC . where I DC is the current of the dc side, Z DC is the equivalent impedance of the dc side, I AC is the current of the ac side, and Z AC is the equivalent impedance of the ac side. Z11 is the effective impedance of the Port 1, as well as the parallel impedance of Z DC and Z AC , as shown in (3.17). The relationship between I P1 and the CM noise current of the ac side I AC is expressed as (3.18). Mitigation elements are added in the dc side circuit, and the impedance of the ac side elements composed of the ac bus bar and the ac cables is supposed to be constant. Therefore, it is shown that the CM current flowing on the ac cable is proportional to the magnitude of Z11 according to (3.19). Z 11 =

Z DC · Z AC Z DC + Z AC

(3.17)

IAC Z DC Z 11 = = IP1 Z DC + Z AC Z AC

(3.18)

IAC Z 11 = IAC− Z 11−

(3.19)

where Z 11 and Z 11- are the effective impedance magnitude of the Port 1 without and with mitigation elements, IAC and IAC- are the effective CM current of the ac side circuit without and with mitigation elements. (1) Mitigation Circuit Design at 6 MHz 1. Design of CM ferrite choke From Fig. 3.76, since the magnitude of Z11 is low at 11 MHz, therefore mitigation circuits are designed at resonance frequencies 6 and 26 MHz. According to Table 3.18 and Fig. 3.77, the capacitances between the dc cables and the shield C DC cable or the inductance of the dc bus bar L DC bus bar and the inductance of the Y-cap L Y-cap can be changed to decrease the magnitude of Z11 and to reduce the value of the resonance at 6 MHz. A CM ferrite choke through the Y-cap is added to change the effective inductance of the branch to reduce the magnitude of Z11 at 6 MHz, and to reduce the CM current and radiated emission through the ac cable. If the value of Z11 is expected to reduce by K times, then: 6 Z 11 =K 6 Z 11−1 6 6 Z 11 (d B) − Z 11−1 (d B) = 20 ∗ log K

(3.20)

130

3 EMI Prediction and Suppression of Motor Drive System

6 6 where Z 11 and Z 11−1 are the magnitude of Z11 at 6 MHz without and with a CM 6 6 (dB) = 30dB, if K = 5, then Z 11−1 (dB) = 16dB. ferrite choke. From Fig. 3.4, Z 11 The ferrite choke is modeled as a simple parallel RLC circuit, L C1 is the inductance 6 6 of the CM ferrite choke, Z 11 and Z 11−1 will be expressed as 6 Z 11 =

6 = Z 11−1

Z DC · Z Y + Z IGBT · (Z DC + Z Y ) + R + Z IGBTcon 2 · (Z DC + Z Y )

(3.21)

Z DC · (Z Y + Z L C1 ) + Z IGBT · (Z DC + Z Y + Z L C1 ) + R + Z IGBTcon 2 · (Z DC + Z Y + Z L C1 ) (3.22)

where Z DC is the impedance of C DC cable and (L + M)DC bus bar, Z Y is the impedance of CY-cap and (L + M)Y-cap in series, Z IGBT is the impedance of L IGBT and C Junct in series, L 2 and L 3 are negligible, R = 0.3, Z IGBTcon is the impedance of LIGBT con, and Z L C1 is the impedance of L C1 . From (3.8) and the impedance of circuit elements in Fig. 3.6, we can achieve     Z DC · (Z Y + Z L C1 ) + (Z IGBT + Z Junct ) · (Z DC + Z Y + Z L C1 ) 6 Z 11−1 =  + R + Z IGBTcon  2 · (Z DC + Z Y + Z L C1 )    − j11.6 · ( j10 + Z L C1 ) − j · (− j11.6 + j10 + Z L C1 )  =  + 0.3 + j0.45 = 6.3 2 · (− j11.6 + j10 + Z L C1 ) Z L C1 = 154.5 L C1 =

Z LC = 4μH 2π 6e − 6

(3.23)

From (3.23), a CM ferrite choke (L C1 = 4μH, RC1 = 10, CC1 = 20pF) simplified as a parallel RLC circuit is chosen as shown in Fig. 3.84. After adding the equivalent j0.1 L1

j1.9 LDC bus bar . -j15 MDC bus bar

-j15

j0.5 L2

j1 LIGBT

LLink

CJunct -j2

LC1 j154.5 LY-cap CY-cap

.

j5.7

CY-cap . j5.7 LY-cap

to chassis

con

CJunct -j2

LC1 j154.5

. LDC bus bar j1.9

Cphase

j1 . LIGBT

MY-cap CLink

LIGBT j1

Cemitter

Ccollector

to chassis

to chassis

MAC bus bar j5 . LAC

-j25

bus bar

Port1

Port2

Fig. 3.84 Current following path at 6 MHz after adding a CM ferrite choke through the Y-caps

3.4 Spice Equivalent Circuit Modeling of Electromagnetic Emission … Without change

131

With ferrite chokes on Y-caps

S21[dB]

0

-20

-40

-60 5 10

6

7

10

10

8

10

Frequency [Hz] 60

Z11[dB]

40 20 0 -20 5 10

6

7

10

10

8

10

Frequency [Hz]

Fig. 3.85 Change in Z11 and S21 after adding a CM ferrite choke on Y-caps

circuit model of the CM ferrite choke in the ADS PSPICE model, the CM current could no longer flow through the Y-cap to the chassis due to the large impedance of L C1 at 6 MHz, as shown in Fig. 3.84. From the simulation results shown in Fig. 3.85, the value of Z11 and IAC are reduced by 14 dB and meet the design requirement, and the value of S21 between the Port 1 and Port 2 was reduced by 7.2 dB, which shows that the CM current and radiated emission through the ac cable generated by fast IGBTs switching can be reduced by adding an appropriate CM ferrite choke through the Y-caps. 2. Design of RC filter A RC filter in front of DC cables is added to change C DC cable and to reduce the magnitude of Z11 at 6 MHz and the quality factor of the resonance Q, it can be expressed as 6 Z 11 =K 6 Z 11−2 6 6 Z 11 (d B) − Z 11−2 (d B) = 20 ∗ log K

(3.24)

6 6 where Z 11 and Z 11−2 - are the magnitude of Z11 at 6 MHz without and with a RC 6 6 and Z 11−2 can be expressed as filter. If K = 5, Z 11 6 Z 11 =

Z DC · Z Y + Z IGBT · (Z DC + Z Y ) + R + Z IGBTcon 2 · (Z DC + Z Y )

(3.25)

132

3 EMI Prediction and Suppression of Motor Drive System 6 Z 11−2 =

where: Z DC =

  · Z Y + Z IGBT · (Z DC + ZY) Z DC + R + Z IGBTcon  2 · (Z DC + Z Y )

Z DC cable ·Z C F Z DC cable +Z C F

(3.26)

+ Z DC bus bar .

6 (dB) Z 11−2

= 16dB and the impedance values in Fig. 3.78 are subtituted When in, one can obtained       Z · Z Y + Z IGBT · (Z DC + ZY) 6  = 6.3 + R + Z =  DC Z 11−2 IGBTcon   2 · (Z DC + Z Y ) Z C F = 13.3 CF =

1 = 2nF 2π 6e − 6 · Z C F

(3.27)

The values of RC were chosen as CF = 2nF, a RC filter with RF = 8 shown in Fig. 3.86. The CM current can flow to the chassis through the RC filter. From the simulation results shown in Fig. 3.87, the values of Z11 and I AC were reduced by 14 dB and the value of S21 was reduced by 8 dB, which indicates that adding an appropriate RC filter in front of the dc cables can reduce the emission and resonance peak at 6 MHz. However, the working current on the dc cables is generally higher than 100A, the voltage and consuming energy of RF is high using the resistance (R F = 8). Therefore, RF should be low enough to guarantee the system work normally. From simulation results shown in Fig. 3.87, although the magnitude of Z11 was decreased at 6 MHz, the magnitude of Z11 at 11 and 26 MHz were both increased using the RC filer (C F = 2nF and R F = 0.2). As a result, it is not recommended to add a RC filter in front of the dc cables to reduce the CM current and radiated emission through the ac cable. (2) Mitigation Circuit Design at 26 MHz A CM ferrite choke was added to the ac bus bar due to the impedance of L AC bus bar larger than other elements in the equivalent circuit at 26 MHz resonance. So the CM ferrite choke was designed by calculating ZDC and ZAC , as shown in Eq. (3.28). RF 10 -j15

CF -j1.3 MDC bus bar

-j15

CF -j1.3 RF 10

LDC bus bar .

L1

j5.7 LY-cap . CY-cap MY-cap CY-cap . j5.7 LY-cap

. LDC bus bar

L2

LIGBT

LLink

CJunct

CLink Cphase CJunct to chassis

MAC bus bar . LIGBT con

LIGBT Cemitter

Ccollector

to chassis

to chassis

. LAC bus bar

Port1

Port2

Fig. 3.86 Current following path at 6 MHz after adding a RC filter in front of the dc cables

3.4 Spice Equivalent Circuit Modeling of Electromagnetic Emission …

133

0 -10 S21[dB]

-20 -30 -40 -50 -60 5 10

Without change With RC filters of R=8ohm and C=2nF on DC cables With RC filters of R=2ohm and C=2nF on DC cables With RC filters of R=0.2ohm and C=2nF on DC cables 6

10

7

Frequency [Hz]

60

10

8

10

Z11[dB]

40 20 0 -20 5 10

6

7

10

10

8

10

Frequency [Hz]

Fig. 3.87 Change in Z11 and S21 after adding a RC filter on DC cables

IAC Z DC = IP1 Z DC + Z AC

(3.28)

The impact of resonances within the power inverter can be reduced by transferring the resonance frequencies to unimportant range through changing the capacitances and inductances generating resonances, or in some cases by reducing their quality factor, for example, reducing the resistance of Z11 at the 26 MHz resonance. The inductance of the CM choke is very high at 26 MHz due to the high frequency CM current and the resistance of the CM is more critical than the inductance at the time, so the CM choke was designed by the expected resistance RC2 . If a CM choke is added in the ac side circuit, and IAC is expected to reduce by K times, then: 26 IAC 26 IAC -1

=K

26 26 (d B) − IAC IAC - 1 (d B) = 20 ∗ log K

(3.29)

26 26 and IAC where IAC - 1 - are the amplitude of IAC at 26 MHz without and with a CM ferrite choke. As shown in Fig. 3.83, the relationship of the noise current and the impedance before and after the mitigation is: 26 IAC 26 IAC -1

=

26 26 + Z AC−1 Z DC−1 26 26 Z DC + Z AC

(3.30)

134

3 EMI Prediction and Suppression of Motor Drive System

26 26 where Z DC and Z DC - 1 are the magnitude of ZDC at 26 MHz without and with a 26 26 and Z AC CM ferrite choke respectively.Z AC - 1 are the magnitude of ZAC at 26 MHz without and with a CM ferrite choke respectively. From (3.28) and the impedance of circuit elements in Fig. 3.82, we can achieve 26 26 26 26 Z DC−1 + Z AC−1 + Z AC + RC2 Z DC = 26 26 26 26 26 I Z DC + Z AC Z DC + Z AC   AC−1  −j12 + j13.56 + RC2  =K =5    −j12 + j13.56 26 IAC

=

RC2 = 7.6 

(3.31)

In order to prevent the CM current flowing through the inductance of the choke, the impedance of the inductance should be ten times more than that of the resistance of the choke, so Z L C2 = 2π · 26 · e6 · L C2 ≥ 10 · RC2 = 76  and L C2 ≥ 0.47 μH. Furthermore, the inductance of the choke should be far lower than that of winding of the motor, finally the inductance of the choke is determined as 0.5 μH. A CM ferrite choke (L C2 = 0.5 μH, RC2 = 7.6 , CC2 = 20 pF) simplified as a parallel RLC circuit is chosen as shown in Fig. 3.88. From the simulation results of Z11 and S21 shown in Fig. 3.89, the magnitude of S21 is reduced by 5 dB, which shows that adding an appropriate CM ferrite choke through the ac bus bar is a better technology to reduce the CM current and radiated emission through the ac cable. The impact of a combination mitigation of adding CM ferrite chokes on the Ycaps and the ac bus bar is shown in Fig. 3.90. The values of Z11 and S21 at critical three resonances frequencies about 6, 11, and 26 MHz are reduced well as shown in Fig. 3.91. The values of Z11 are decreased respectively 15 dB at 6 MHz, 0.4 dB at 11 MHz, and 11.5 dB at 26 MHz, and the values of S11 is decreased respectively 8.6 dB at 6 MHz, 7 dB at 11 MHz, and 6.3 dB at 26 MHz. It is shown that the combination mitigation is a reasonable technology to reduce the CM current and radiated emission through the ac cables at the critical three resonances frequencies. (4) Varification of mitagtion method To validate the mitigation technology, a PSPICE model of the power inverter system for real-time simulation at time domain is built in the CST designer as shown in j8.2 j0.3 j2 j21 LDC bus bar L2 L1 . LY-cap j24.5 LLink -j3.5 . CY-cap MDC MY-cap CLink Cphase bus bar CY-cap . to chassis L -j3.5 -j14.86 . Y-cap j24.5 LIGBT L DC bus bar j21 j8.2 j4

j4 LIGBT CJunct -j0.47 CJunct -j0.47

j4.08 . LIGBT con

Cemitter

Ccollector

to chassis

to chassis

MAC bus bar RC2 7.6 LC2 j76 Port1

j21.24 . LAC -j5.72 bus bar

Port2

-j16.11

Fig. 3.88 Current following path after adding a CM ferrite choke through the ac bus bar

3.4 Spice Equivalent Circuit Modeling of Electromagnetic Emission … Without change

135

With ferrite choke on AC bus bar

S21[dB]

0 -20 -40 -60 5 10

6

10

7

Frequency [Hz]

8

10

10

60

Z11[dB]

40 20 0 -20 5 10

6

7

10

8

10

10

Frequency [Hz]

Fig. 3.89 Change in Z11 and S21 after adding a CM ferrite choke on the ac bus bar

MDC bus bar

LDC bus bar .

L1

LC1 4μH RC110

20pF

LY-cap CY-cap

L2

CC1

.

LLink

RC110 . LDC bus bar

CJunct MAC bus bar

MY-cap CLink

CY-cap . LY-cap LC1 4μH

LIGBT

. Cphase

LIGBT

to chassis

con

CJunct

CC1

CC2 20pF

20pF LIGBT

. LAC RC2 7.6 LC2 0.5μH bus bar

Cemitter

Ccollector

to chassis

to chassis

Port1

Port2

Fig. 3.90 Current following path after adding CM ferrite chokes through the Y-caps and the ac bus bar

Fig. 3.92. A trapezoidal CM interference source is added between the two IGBTs of a single bridge leg as “Source 1”. The cycle time of the trapezoidal wave is 20 μs, and the rise time is 0.032 μs. The probe P1 can obtain the CM current flowing through the ac cable, as shown in Fig. 3.93. The original CM current without any mitigation method is shown as the blue line, there are three problematic frequencies resonance at 7, 11, and 26 MHz similar with that of the model in ADS software shown in Fig. 3.76.

136

3 EMI Prediction and Suppression of Motor Drive System

S21[dB]

0

Without change With ferrite chokes both on Y-caps and AC bus bar

-20 -40 -60 5 10 60

6

7

6

7

10 Frequency [Hz] 10

8

10

Z11[dB]

40 20 0 -20 5 10

10 Frequency [Hz] 10

8

10

Fig. 3.91 Change in Z11 and S21 after adding CM ferrite chokes through the Y-caps and the ac bus bar

Source

Fig. 3.92 SPICE model in time domain simulation added CM ferrite chokes through the Y-caps and the ac bus bar

The value of the CM current flowing through the ac cable at 7 MHz is decreased with the CM ferrite choke through the Y-caps as shown in Fig. 3.93a, and the CM current at 26 MHz decreased with a CM ferrite choke through the ac bus bar as shown in Fig. 3.93b, and the CM current at three problematic resonances are all decreased

3.4 Spice Equivalent Circuit Modeling of Electromagnetic Emission …

I[dBuA]

150

Original current Current with CM ferrite chockes on Y-caps

100 50 0 6 10 150

I[dBuA]

137

7

(a)

10 Frequency [Hz]

8

10

Original current Current with a CM ferrite chocke on AC bus bar

100 50 0 6 10

I[dBuA]

150

7

10 (b) Frequency [Hz]

8

10

Original current Current with CM ferrite chockes both on Y-caps and AC bus bar

100 50 0 6 10

7

10

(c) Frequency [Hz]

8

10

Fig. 3.93 Change in CM current flowing one ac cable from mitigation

with the CM ferrite chokes through the Y-caps and the ac bus bar as shown in Fig. 3.93c. The mitigation method is validated effectively by the real-time simulation results at time domain and divided into the following steps: At first, the power inverter system based on measurement is modeled. Secondly, the responsible components for problematic resonances are determined through the transmission characteristics. Finally, appropriate mitigation method to reduce emission is designed.

3.4.5 Conclusion The development and analysis of a model for a complex power system were illustrated in this paper. While the methodology was applied to a power inverter system, it can be applied to other systems as well, where frequencies of interest are below

138

3 EMI Prediction and Suppression of Motor Drive System

100 MHz. The equivalent model contains only the most important parasitic elements of the system. Each element can be clearly correlated with real system geometry. The model developed here was able to describe the impedance of a real inverter system well from 100 kHz to 100 MHz. Simulations of S21 for this system showed resonances at frequencies similar to those at which peak radiation has been observed from the real system. A substantial advantage of a simple circuit representation is that it allows analytic determination of the possible causes of and mitigations strategies for emissions. This process is possible because the circuit is simple and because circuit elements are directly correlated with physical structures within the system.

3.5 EMI Power Filter In the initial stage of motor controller design, according to EMC standards and filter insertion loss design requirements, a universal DC power filter is designed to suppress EMI noise at the HV DC input port of the DC-AC motor inverter. Considering the radio disturbance characteristics of road vehicles in the wide frequency range of 150 kHz–108 MHz, as well as the motor inverter characteristics, such as the high voltage, high current and variable load, two design methods of high voltage DC power supply filter for the motor drive system of electric drive vehicles are proposed to meet CISPR25- 2016Rd4 standard limit requirements. Method 1: A design method of high-voltage DC power filter for electric drive vehicle motor controller with insertion loss of 60 dB is proposed. This method can reduce conducted EMI in 150 kHz–108 MHz and meet the standard requirements. The filter designed by this method is large in size, heavy in weight and high in cost. It is more convenient to install outside the motor controller. However, due to the large volume occupied in the motor controller, it is not conducive to the engineering realization of electric drive vehicles. Method 2: An EMI filter optimization design method based on resonance suppression is proposed. Using the established high-frequency equivalent circuit model of the motor inverter system, according to the prediction results obtained by the CISPR25-2016 standard conducted emission-voltage method test, the CM and DM current propagation paths and main component parameters that affect EMI is analyzed at the two key frequency points 1 and 30 MHz where resonance occurs. Thus the components that cause conducted disturbance voltage to exceed the standard limit are found, this method can be implemented inside the controller, small in size, low in cost, high in efficiency, and can be used at different stages of product development. Compared with the first method, the second method has a smaller filter module and is only 1/6 the volume of the filter module designed for the first method.

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139

3.5.1 Design of High-Voltage Input Port Filter (1) Topology of the filter The source impedance and load impedance should be considered in the selection of the filter topology. From the test platform shown in Fig. 3.3, it can be seen that the filter will be installed near the input port of HV DC power supply of the motor controller. The input of filter is connected to the input port of HV DC power supply, which is connected to a large capacitor (usually 500uF–3000uF) installed inside the motor controller for absorbing voltage ripple. And the large capacitor is low impedance. The output side of the filter is connected to the LISNs with two resistances of 50. Therefore, the LC-type topology is initially selected as the basic topology of the CM and DM filters. (2) Calculation of LC filter series and corner frequency The number of stages n of the LC filter and the corner frequency f 0 of each stage are primarily related to the frequency range and insertion loss requirements. Considering the effects of source impedance and parasitic parameters, the filter insertion loss is required to be 60 dB, and the conducted EMI can be suppressed in the full frequency range of 150 k–108 MHz. In general, the slope of the insertion loss curve of singlestage LC filter will basically follow 40 dB/10dec between the corner frequency and the 10 times corner frequency. However, due to the influence of parasitic parameters, the insertion loss curve will gradually deviate from the theoretical curve between 10 times and 100 times the corner frequency, but can still maintain the insertion loss of 40 dB/10dec. Beyond 100 times the frequency, the filter suppression effect will gradually deteriorate until there is no filtering effect. Due to the wide frequency range of 150 k–108 MHz, single-stage LC filters is hardly to meet the insertion loss requirements of 60 dB. Usually as the number of filter stages is increased, the insertion loss effect is also increased, however the filter volume is multiplied. Therefore, the filter is initially designed as a two-stage LC filter, as shown in Fig. 3.94. The CM filter circuit consists of two magnetic rings L 1 and L 2 with the same inductance value L, and two pairs of Y capacitors (two C y1 and two C y2 ) to suppress CM interference current. Considering the large capacitance of DM filter inside the motor controller, the self-inductance L D1 and L D2 of the positive and negative copper bars, the two X capacitors C DM1 and C DM2 and 2 pairs of Y capacitors (2 C y1 series and 2 C y2 series) together form a DM filter circuit to suppress DM interference Fig. 3.94 Topology of the filter

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3 EMI Prediction and Suppression of Motor Drive System

current. Where, the capacitance values of C y1 and C y2 are usually much smaller than those of C DM1 and C DM2 , and can be ignored. The classic power filter circuit design method is to obtain the CM interference and DM interference voltages through testing first, and then determine the respective attenuation of CM interference and DM interference, where Attenuation = Actual amplitude-Limit + 6 dB (remaining the amount). Then, the corner frequency f o is calculated according to the formula (3.32). Where, f h_att is the target attenuation frequency, the starting frequency is 150 kHz, Areq_att is the required attenuation amount of 60 dB, and f ilt_att is the attenuation slope of 40 dB/dec. f0 =

f h_att 10

(3.32)

Areq_att f ilt_att

According to (3.32), the corner frequency f 0 of the CM and DM filter is 3.35 kHz. (3) Analysis of the influence of topology on insertion loss Three kinds of filter topologies, LCLC without Y capacitor, LCLC + 2C Y1 and LCLC + 2C Y1 + 2C Y2 are used successively to calculate and determine the parameters of filter elements, and to analyze the insertion loss under constant source impedance and load impedance. Using the high frequency equivalent model of motor drive system, three kinds of topological filters are added to the motor drive system to predict the effective insertion loss of the filter in the actual system. Step1: First, LCLC topology without Y capacitor is adopted. The filter consists of two CM magnetic rings L 1 , L 2 and two DM capacitors C DM1 , C DM2 , as shown in Fig. 3.95a. The DM and CM interference source are U DM and U CM respectively. Their equivalent circuits are shown in Fig. 3.95b, c. The DM filter circuit consists of inductor L D1 and L D2 and X capacitor C DM1 and C DM2 . The CM filter circuit consists of L 1 and L 2 . The value of L D1 and L D2 are calculated according to the size of copper bars. We can get L D1 +=L D1 - = L D2 +=L D2 - = 56nF.  CDM1 =

1 2π f 0

2 ×

1 2L D+

(3.33)

The value of C DM1 and C DM2 are calculated by (3.33) to be 10 μF. The 3D diagram of the LCLC topology filter prototype is shown in Fig. 3.95d. The insertion loss S21 of CM and DM filter are obtained by modeling and simulated in the software ADS, as shown in Fig. 3.95e, f. The LCLC topology without Y capacitor is added to the high-frequency equivalent circuit of the motor drive system, and the conducted disturbance voltage is obtained by simulation, as shown in Fig. 3.95(g). It can be seen that after adding the LCLC topology, the conducted disturbance voltage decreases with increasing frequency below 30 MHz, and rises up to 32 dBμV at 30 MHz. The conducted disturbance

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141

LD1+

LD2+

P L1

N

P’

CDM1 L2

UDM LD2-

LD1-

LD1-

LD2+

CDM1 LD2-

CDM2 50

N’

(a)Topology

L1

LD1+

CDM2

(b)DM equivalent circuit

L2

UCM 25

(c) CM equivalent circuit

(e) S21 of DM

(d) 3D model of principle prototype

(f) S21 of CM

valueinterference voltage

(g) Comparison of conducted disturbance voltages

Fig. 3.95 LCLC filter without Y capacitor characteristics

voltage at the 1 MHz resonance point decreases from 83 dBμV to 23 dBμV. However, the amplitude of conducted disturbance voltage at 30 MHz resonant point decreases slightly, and it is still large at high frequencies. Therefore, the conducted disturbance voltage can be effectively suppressed at 150 kHz–2 MHz by using LCLC topology. Step 2: LCLC + 2C Y1 topology is adopted. A pair of Y capacitors C Y1 is added to the output terminal of the filter based on the LCLC topology without Y-capacitor, as shown in Fig. 3.96a. The DM equivalent circuit is the same as that of Fig. 3.95b. The CM equivalent circuit is shown in Fig. 3.96b. According to the calculated common-mode corner frequency f 0 = 150 kHz, the inductance L 1 of the commonmode magnetic ring and the common-mode Y capacitance C y1 are calculated, and the expression is

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3 EMI Prediction and Suppression of Motor Drive System /2

/1 UCM

(a)Topology

25

(b)CM equivalent circuit

(c) Characteristics of Ultramicrocrystalline magnetic ring

(e)S21 of CM

2Cy1

(d)3D model of principle prototype

(f) Comparison of conducted disturbance voltages

Fig. 3.96 LCLC + 2C Y1 filter

 L1 =

1 2π f 0

2 ×

1 2Cy1

(3.34)

According to the limit value of the leakage current, by formula (3.34) and the inductance L 1 L 2 = 32 μH, C Y1 = 0.034 μF can be calculated. The 3D diagram of the LCLC + 2C Y1 topology filter prototype is shown in Fig. 3.96d. According to Fig. 3.96a and the filter insertion loss standard, the common mode insertion loss of the filter is obtained by modeling and simulation in the software ADS, as shown in Fig. 3.96e. The LCLC + 2C Y1 topology is added to the high-frequency equivalent circuit of motor drive system, and the conducted disturbance voltage is obtained by simulation, as shown in Fig. 3.96(f). Compared with the LCLC without Y capacitor, there is no obvious change. Conducted disturbance voltage has good suppression effect at low-frequency, and its high-frequency disturbance is still larger. In addition, a more obvious resonance is generated near 60 MHz. Therefore, adding a pair of Y capacitors on the output side of the filter has almost no filtering effect, and also makes the conducted disturbance voltage in the high frequency band more serious. Step 3, LCLC + 2C Y1 + 2C Y2 is adopted. Based on the LCLC + 2C Y1 topology, a pair of Y capacitors C Y2 are respectively added to the positive and negative bus bars of the HV input terminal of the filter, and the topology is shown in Fig. 3.94. where, C Y2 = 0.1uF due to the consideration of leakage current. The DM equivalent circuit topology is the same as that of Fig. 3.95b, and the CM equivalent circuit is shown in Fig. 3.97a. The 3D diagram of the filter prototype is

3.5 EMI Power Filter

/1 UCM

143

/2 2Cy2

2Cy1

(a)Topology

(c)Insertion loss S21

25

(b)3D model of principle prototype

(d)Comparison of conducted disturbance voltages

Fig. 3.97 LCLC + 2C Y1 + 2C Y2 filter

shown in Fig. 3.97b. The CM insertion loss of this filter is shown in Fig. 3.97c. This filter circuit is added to the high-frequency equivalent circuit, and the conducted disturbance voltage is obtained by simulation, as shown in Fig. 3.97(d). After adding two additional C Y2 to the filter input port, the conducted voltage at 30 MHz decreases from 40 dBμV to 25 dBμV compared with LCLC + 2C Y1 , and the effective insertion loss is 15 dB, less than 60 dB. Although the high-frequency conducted disturbance voltage is reduced, the attenuated conduction disturbance voltage is still high. Equations (3.33) and (3.34) are indeterminate solution equations for inductance and capacitance. When specific inductance and capacitance parameters are determined, the actual insertion loss requirements should be considered, on the other hand, and the maximum volume of the filter should be considered. Especially, the larger the inductance is, the larger the size of the filter will be. Therefore, it is very important to select the appropriate inductor in consideration of the allowable volume of the filter. From the above analysis, it can be seen that the insertion loss of the filter basically meets the requirement of 60 dB when the source impedance and load impedance are 50, while the effective insertion loss of the filter after adding the filter to the system cannot meet the requirement. In order to verify the actual suppression effect of the filter, it is necessary to add the three topologies to the motor inverter system for experimental verification. (4) Effective insertion loss test of the filter in the system Using the established test platform in Fig. 3.3, the EMI filter is installed at the HV DC port of the motor controller. According to CISPR25-2016, the measurement of the conducted emission of the HV positive power line of the PMSM drive system is carried out.

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3 EMI Prediction and Suppression of Motor Drive System

Case 1: After adding the LCLC filter without Y capacitor, the test result of the conducted interference voltage is shown in Fig. 3.98. It can be seen that the conducted interference voltage in the frequency band below 30 MHz are significantly reduced, and both below the standard limits. The peak value at 1 MHz resonant point decreases from 91dBuV to 29dBuV, and the average value decreases from 80 dBuV to 20 dBuV. However, the peak value and average value of 36– 41 MHz still exceed the limits level. The peak value of 36.8 MHz reaches 63 dBμV, which exceeds the standard limit by 4 dB. The average value of 39.6 MHz reaches 50 dBμV, which exceeds the standard limit by 11 dB. Case 2: LCLCL + 2C Y1 filter is added, and the conducted interference voltage is shown in Fig. 3.99. It can be seen that there is no obvious change in peak and Fig. 3.98 Conducted EMI voltage test results after adding LCLC filter without Y capacitor

(a) 150kHz-108MHz

(b) 35MHz-42MHz

Fig. 3.99 Conducted voltage test results after adding LCLCL + 2C Y1 filter

(a)150kHz-108MHz

(b)36.6MHz-60MHz

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Fig. 3.100 Conducted voltage test results after adding LCLC + 2CY1 + 2C Y2 filter

(a)150kHz-108MHz

(b)36.6MHz-60MHz

average value below 30 MHz band. the maximum peak value of conducted voltage in 36–41 MHz band decreases from 63 dBμV to 53.7 dBμV, which is below the standard limit. The maximum average conduction voltage of 36–41 MHz band decreased from 50 dBμV to 44 dBμV, which exceeded the standard limit of 5 dB. In addition, the average value exceeded the standard limit of 0.4 dB at 57 MHz. Case 3: LCLCL + 2C Y1 + 2C Y2 filter is added, and the conducted interference voltage is shown in Fig. 3.100. It can be seen that there is no significant change in peak and average interference below 30 MHz. The peak and mean values of 36–41 MHz band are reduced and meet the standard limits. However, a new resonance peak appeared near 46 MHz, with the average value of 46 MHz reaching 42.6 dBμV, which exceeded the standard limit by 3.6 dB. The peak value of 47.4 MHz reached 59.8 dBμV, which exceeded the standard by 0.8 dB. In addition, the peak value was still slightly exceeded the standard at about 48 MHz. The experimental results show that after adding 2C Y1 + 2C Y2 does not have a good suppression effect. Although the three suppression methods in the simulation and prediction stage can largely reduce the conducted EMI emission, they still can’t meet the standard limits in the actual test. Since the addition of the Y capacitor may cause a new resonance while suppressing the interference, several C Y2 with different capacitance values are replaced to test obtain the influence of different capacitance values on conducted EMI voltage. In the actual test, a pair of 100nF C Y2 and a pair of 1000nF C Y2 are used respectively. The test results are shown in Fig. 3.101a, c. The comparison of the average values with the three capacitors is shown in Fig. 3.101 (e). From Fig. 3.101a, it can be seen that after adding a 100nF Y capacitor, the conducted interference voltage decreases significantly. Only the average curve reaches 38.5 dBμV at 55 MHz, which exceeds the standard limit. As shown in Fig. 3.101c, after adding a 1000nF Y capacitor, the peak value of the conducted

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3 EMI Prediction and Suppression of Motor Drive System

(a)100nF,150kHz-108MHz

(b)100nF, 35MHz-70MHz

(c) 1000nF, 150 kHz-108MHz

(d) 1000nF,35MHz-70MHz

(e) Comparison of three ways Fig. 3.101 Conducted voltage test results with different Y capacitors

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147

interference voltage and the average curve significantly decrease obviously, which meet the requirement below the standard limits. Therefore, a 1000nF Y capacitor is finally used.

3.5.2 Design of Filter Circuit Based on Resonance Suppression It can be seen from the test results in Fig. 3.4 that the conducted voltage exceeding the standard limits are mainly at around 1 and 30 MHz. According to the path analysis of interference currents at 1 and 30 MHz, it can be seen that the conducted EMI noise at 1 MHz is mainly the DM component, and the conducted EMI noise at 30 MHz is mainly the CM component. (1) DM filter circuit based on 1 MHz resonance suppression There are two main methods to reduce the conducted voltage of the resonant at 1 MHz: ➀ To reduce the equivalent parasitic inductance Ls of the DC filter capacitor C DC , and shift the resonance point from 1 MHz to the other frequency band where the standard has no specified limits. So the solution can be connecting X capacitor in parallel with the C DC . ➁ Increasing the parasitic inductances L P1 and L P2 of DC bus can increase the amplitude of Z 1 , thereby reducing the 1 MHz DM interference resonance peak and conducted voltage. The self-inductances of the positive and negative copper bars of the filter power supply is used as the DM inductance. It can be seen from Fig. 3.4 that the average value exceeds the standard limits by15dBμV, so the DM attenuation is set to be 15 + 6=21 dB, and the DM cutoff frequency is calculated to be 95 kHz using (3.32). Two X capacitors C X1 , C X2 and two DM inductors L D+ , L D- are used to form a one-stage CLC DM filtering topology, as shown in Fig. 3.102. The self-inductance of the positive and negative copper bars of the filter power source is used as a DM inductance, which is 30 nH. Calculated by (3.33), the DM capacitance is 10 μF. The DM filter is placed in the high frequency equivalent circuit of the motor drive system, located between the DC input port of the inverter and the LISNs, as shown in Fig. 3.103. Fig. 3.102 Topology of DM Filter

P

N

Cx1

LD+

LD-

Cx2

P

N

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3 EMI Prediction and Suppression of Motor Drive System

LISN DC Lines Filter

AC Lines

Inverter

PMSM

L1

C3 C1 R1

RDC+ LDC

LX1

+

CX2 CX1

VDC C4 R2 L2 C2 RDC LDC-

LX2

-

CDC-

CDC+

CDC LDC

LIGBT

LIGBT

LIGBT

CS1

CS2

CS3

LIGBT

LIGBT

LIGBT

LIGBT

LIGBT

LIGBT

RD

CS4

CS5

CS6

C

LIGBT

LIGBT

LIGBT

CP CN CA

CB

RMA LMA

RC LCA

LA

A

RC LCB

LB

RMB LMB

B

RC LCC

LC

RMC LMC

C

CM CCG CBG

CC

CAG

Fig. 3.103 Adding the DM filter to high-frequency equivalent circuit of the system

It can be seen from Fig. 3.103 that after adding the DM filter circuit, part of the DM interference current flows back to the interference source via the DM capacitor C x1 and the DM filter inductors L D+ , L D- and the DM capacitor C X2 . Due to the shunt capacitor C x1 , the impedance value of the DC filter branch is reduced, and the DM current through the branch is increased, so that the DM current through the HV DC bus and the conducted voltage measured on the LISNs are reduced. The conducted noise current flowing into the LISN is reduced by increasing the value of L D+ , L D- , and C X2 . (2) CM filter circuit based on 30 MHz resonance suppression Through the analysis of the CM interference current propagation path at 30 MHz, it can be seen that the resonance formation at 30 MHz is related to the inter-pole capacitance of IGBT, the neutral-to-ground capacitance of inverter, the inductance of DC and AC lines, and the inductance of copper bars of inverter. It is difficult to characterize the relationship between the conducted disturbance voltage and the interference source by mathematical equations. Here, the CM filter circuit is designed using the circuit network port theory. Two pairs of CM capacitors (C y1 , C y2 ; C y3 , C y4 ) and a CM magnetic ring L C form a one-stage CLC CM filter topology, as shown in Fig. 3.104. According to the limits of the leakage current, the value of the Y capacitor is determined. The Y capacitor C Y1 at the first corner frequency f 1 = 150 kHz is 0.068 μF. Then according to (3.34) the value of the required inductance of the magnetic ring was calculated to be L c = 30 μH. Fig. 3.104 Topology of CM filter

P

Cy2

N

Cy3 P

Cy1 LC

Cy4

N

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149

(3) Optimization of the filter topology The EMI source impedance of the HV power line of the motor inverter system for EV changes with the PWM control mode and the electrical load in real time. It is difficult to obtain an accurate source impedance through theoretical calculation and numerical solution. Currently, relatively accurate source impedance can be obtained through experimental measurements and system modeling. The high-frequency equivalent model of the motor inverter system established is used to analyze and determine the optimal topology. In order to obtain an optimal solution, the DM and CM filters of four configurations of CL, LC, CLC and LCL considering source impedance in the motor drive system are analyzed below. (1) Optimization of DM filtering topology ➀ CL topology The DM CL filter circuit is added to the DC input side of the high frequency circuit model of the motor drive system. The simulation result of the conducted disturbance voltage obtained on the LISN is shown in Fig. 3.105a. There is a large amplitude attenuation near the 1 MHz, which is reduced by 46 dB at 900 kHz. The resonant peak point is shifted from 900 kHz to 960 kHz and its amplitude is reduced to 54 dBμV.

(a) CL topology

(b) LC topology

(b) CLC topology

(d) LCL topology

(e) Comparison of the four topologies

Fig. 3.105 Conducted interference simulation results after adding DM filter

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3 EMI Prediction and Suppression of Motor Drive System

➁ LC topology Form Fig. 3.105b, the conducted voltage decreases substantially near 1 MHz and 38 dB at 900 kHz. However, the disturbance voltage at 150–220 kHz, 320 kHz, 340 kHz, 560 kHz, 2 MHz to 20 MHz exceed the initial test value without a filter. In particular, the conduced voltage rises more at 150 kHz, and it is easy to exceed the standard limit. The original resonant peak at 900 kHz was shifted from 900 to 560 kHz and the amplitude was also decreased to 64 dBμV. ➂ CLC topology From Fig. 3.105c, the conducted voltage decreases substantially near 1 MHz and is attenuated by 43 dB at 900 kHz. The resonant peak point is shifted from 900 kHz to 1 MHz and its amplitude is reduced to 60 dBμV. However, the conducted voltage at 320, 340, 560 kHz, 2 to 20 MHz exceeds the original test value with no filter. ➃ LCL topology From Fig. 3.105d, the conducted voltage is attenuated in the frequency range of 150 kHz to 2 MHz. The original resonant peak point is shifted from 900 kHz to 1 MHz, and the amplitude is attenuated to 66 dBμV. The conducted disturbance voltage is attenuated by 36 dB at 900 kHz. However, at 1, 2, 3 MHz, etc., some highorder harmonics of 1 MHz appear, and the voltage amplitude exceeds the original value. The simulation results of conducted voltage of the four DM filter topologies are compared, as shown in Fig. 3.105e. It can be seen that each topology can effectively reduce the conducted voltage in the 150 kHz–2 MHz frequency band around 1 MHz, and the LCL topology has the best suppression effect. Compared with LCL topology, the effective insertion loss of CL topology is about 8 dB less than that of LCL between 150 kHz and 250 kHz, and the high frequency is almost the same. The effective insertion loss of the LC topology is about10dB less than the LCL in the 150 kHz–250 kHz band, and about 15 dB less than the LCL in the frequency band above 2 MHz. The effective insertion loss of the CLC topology is about 12 dB smaller than the LCL in the frequency band above 2 MHz. Although LCL has the best performance in terms of effective insertion loss, the structure of two inductors will greatly increase the size of the filter, which is limited by installation space in controller. Therefore, the first-order CL topology is adopted by the final DM filter circuit. (2) Optimization of CM filter topology ➀ CL topology As shown in Fig. 3.106a, the original resonant peak point is shifted from 900 to 440 kHz, and the amplitude is decreased to 75 dBμV. Although the amplitude of this point is high, the limits are not specified in this band. It is decreased by 62 dB at 900 kHz and 57 dB at 30 MHz.

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(a) CL topology

(b) LC topology

(c)CLC topology

(d) LCL topology

(e) Comparison (150kHz-108MHz;150kHz-500kHz;26MHz-108MHz˅

Fig. 3.106 Conducted EMI simulation results after adding CM filter

➁ LC topology As shown in Fig. 3.106b, the original resonant peak is shifted from 900 to 860 kHz, and the amplitude is decreased to 54 dBμV, decreased by 60 dB at 900 kHz, and decreased by 62 dB at 30 MHz. ➂ CLC topology As shown in Fig. 3.106c, the resonance peak point is shifted from 900 to 440 kHz, and the amplitude is 80 dBμV. Although the amplitude of this point is high, the

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3 EMI Prediction and Suppression of Motor Drive System

standard does not specify limits in this band. The conducted voltage is attenuated by 66 dB at 900 kHz and 50 dB at 30 MHz. ➃ LCL topology As shown in Fig. 3.106d, the resonant peak is shifted from 900 kHz to 1 MHz and the amplitude is attenuated to 56 dBμV. The conducted voltage is attenuated by 58 dB, 43 dB, and 25 dB at 900 kHz, 30 MHz and 78 MHz, respectively. The simulation results of conducted voltage of four CM filter topologies in high frequency circuit model of motor drive system for electromagnetic emission are compared, and the results are shown in Fig. 3.106e. The simulation results show that each topology can effectively reduce the CM conducted voltage in the frequency range of 150 kHz–108 MHz, and the LCL topology suppression effect is the best. Compared with the LCL topology, the effective insertion loss of the CLC and the CL topology is about 17 dB less than that of the LCL between 17 kHz and 500 kHz. Especially, a new resonance at the 440 kHz is formed with amplitudes of up to 75 dBμV and 80 dBμV, respectively. The high frequency performance of CL and CLC topologies is basically the same as that of LCL topology. The effective insertion loss of LC topology is about 10 dB smaller than that of LCL between 150 kHz and 500 kHz, and the high frequency performance is basically the same as that of LCL. Likewise, because of the volume, the LC structure is used instead of the LCL topology. (4) Simulation of conducted EMI suppression by CM&DM filter According to the above analysis, CL topology is used for DM filter, and LC topology is used for CM filter. The CM topology structure of the EMI filter based on the 1 MHz and 30 MHz resonance suppression is designed, as shown in Fig. 3.107a. The DM and CM insertion loss with 50  source impedance and load impedance are shown in Fig. 3.107b, c. It can be seen that the filter can effectively decreased the CM and DM LD+ L I S N

LC

Cy1 Cy2

CX1 LD-

Motor inverter

LC

(a) Topology

(b)CM insertion loss

(c) DM insertion loss

Fig. 3.107 Topology and insertion loss of optimized filter network

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Fig. 3.108 Comparison of conducted disturbance voltages

conducted disturbance voltage. The insertion loss of CM filter at 1 MHz is 28 dB, and the insertion loss of DM filter at 1 MHz is 62 dB. The insertion loss can meet the requirement of more than 21 dB at 1 MHz. The Optimized filter is added into the high-frequency circuit for simulation. The contrast results of the conduced voltage before and after adding the optimized filter are shown in Fig. 3.108. The conducted disturbance voltage between 150 kHz and 108 MHz has been effectively suppressed. The conducted emission voltage at the 900 kHz resonance point is reduced from 83 dBμV to 33 dBμV, which is attenuated by 50 dB. The interference value at the 30 MHz resonance point is reduced from 40 dBμV to −2 dBμV, which is attenuated by 42 dB. (5) Experimental verification (1) Principle prototype The capacitance, inductance and shell of the filter are modeled in 3D by CATIA software. The prototype of the filter is shown in Fig. 3.109. (2) Insertion loss test The insertion loss of the optimized filter is tested, and the difference between the actual filtering effect and the simulation predictive filtering effect is compared. The insertion loss test platform consists of a signal generator, a spectrum analyzer, a Fig. 3.109 Filter principle prototype

(a) 3D model

(c)Filter prototype

(b) Comparison of two filter sizes

(d) Filter installed in the motor controller

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3 EMI Prediction and Suppression of Motor Drive System

filter, and an insertion loss test tool housing. During the test, the signal generator transmits 0.5 V swept sine wave with the swept frequency band of 9 kHz–108 MHz. The resolution bandwidth and video bandwidth of the spectrum analyzer are set to 10 kHz between 9 kHz and 30 MHz, 100 kHz between 30 and 108 MHz, and the attenuation is 9 dB. The filter is placed in the test fixture, and the input side and the output side of the filter are separated to prevent the coupling between them. The actual insertion loss test layout is shown in Fig. 3.110. The DM insertion loss of the filter in the 9 kHz–30 MHz band is shown in Fig. 3.111a. It can be seen from the test results that the insertion loss of DM filter is small at 9 kHz and gradually increases with increasing frequency. The insertion loss is 36.8 dB at 1 MHz, and then decreases gradually with the increase of frequency after the fluctuation occurs at 3.5 MHz. The DM insertion loss of the filter in the 30–108 MHz band is shown in Fig. 3.111b The insertion loss is 35.73 dB at 33 MHz. It can be seen that the insertion loss is relatively stable in the frequency range of 30–108 MHz, and is maintained at about 33 dB. The DM insertion loss of the filter in the 9 kHz–30 MHz band is shown in Fig. 3.112a. It can be seen that the CM insertion loss of filter is small at low frequencies, and the insertion loss is 11.12 dB at 159 kHz. It gradually increases with increasing frequency, and its insertion loss is 30 dB at 1 MHz. The resonance point is formed at 6 MHz, and then gradually decreases as the frequency increases. The CM insertion loss of the filter in the 30–108 MHz band is relatively stable and remains at about 40 dB, as shown in Fig. 3.112b. Signal generator

Spectrum analyze

Test fixture

Fig. 3.110 Insertion loss test platform 1.06MHz

32.73MHz

-36 88dBm

-35 73dBm

(a) 9kHz~30MHz

Fig. 3.111 DM insertion loss test results

(b) 30MHz~108MHz

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45.60MHz 36.55dBm -

158.96kHz -11.12dBm

(a) 9kHz~30MHz

(b) 30MHz~108MHz

Fig. 3.112 CM insertion loss test results

Fig. 3.113 Conducted disturbance voltage after adding an optimized filter

(3) Conducted emission experiment Using the test platform shown in Fig. 3.3, conduction EMI test of motor controller with filter is carried out. The test results are shown in Fig. 3.113. It can be seen that the peak and mean values of the conducted disturbance voltage in the frequency range of 150 kHz to 108 MHz meet the limit requirements of the standard CISPR25-2016 HV component level five.

3.5.3 Design of EMI Filter for HV DC Power Supply of Motor Controller 3.5.3.1

EMI HV DC Power Supply Filter- Scheme 1

(1) Filter design The CM filter circuit is preliminarily selected as a two-stage LC structure. Inductors L 1 and L 2 are both selected to use two ultra-microcrystalline magnetic rings, and their inductance is 45 μH at 10 kHz, and the two-stage Y capacitors C y1 -C y4 are

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3 EMI Prediction and Suppression of Motor Drive System

both 68nF safety capacitance. The DM filter circuit chooses to use additional X capacitors and the leakage inductance of the ultra-microcrystalline magnetic ring to form a two-level CL circuit. The capacitance of the two X capacitors is 10uF, and the leakage inductance of the ultra-microcrystalline is about 2% of its self-inductance, 0.9 μH. Figure 3.114a is the filter topology, (b) is the 3D structure diagram of the prototype, (c) is the physical diagram of the filter prototype. (2) Insertion loss modeling and simulation According to Fig. 3.114a, a simulation circuit for filter DM and CM insertion loss is built in the software ADS. The CM insertion loss simulation circuit and simulation results are shown in Figure 3.115a, b, and the DM insertion loss simulation circuit and simulation results are shown in Fig. 3.115c, d. It can be seen that the CM insertion loss is 34.6 dB at 150 kHz, and the DM insertion loss is 76.2 dB at 150 kHz. (3) Insertion loss measurement With a vector network analyzer, the DM S parameters and CM S parameters were measured respectively. The test layout is shown in Fig. 3.116, and the CM and DM insertion loss test results are shown in Fig. 3.117a, b. Whether DM insertion loss or CM insertion loss of the first filter scheme is less than 60 dB in the full frequency band of 150 k–108 MHz. Insertion loss is significantly reduced before 200 kHz and after 10 MHz. In addition, CM and DM S21 both resonate at 30 MHz, and DM S21 also has a resonance point near 1 MHz. The best frequency range of CM insertion loss is 300 kHz–2 MHz, which can achieve 50 dB insertion loss, and DM insertion loss can achieve 35 dB insertion loss in the range of 300 kHz–10 MHz. P CX1 INPUT

Pÿ

R

L1

CX2

L2 CY2

N

CY3

CY1

OUTPUT CY4 Nÿ

(a) Scheme 1 filter topology

(b) Scheme 1 filter 3D model

Fig. 3.114 Scheme 1 filter

(c) Scheme 1 filter installation physical map

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(a)CM simulation circuit

(c)DM simulation circuit

˄b˅CM insertion loss simulation results

(d)DM insertion loss simulation results

Fig. 3.115 Scheme 1 filter insertion loss simulation Fig. 3.116 Scheme 1 filter test layout diagram

(a)Scheme 1 filter CM test result

Fig. 3.117 Scheme 1 filter test result

(b) Scheme 1 filter DM test result

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3 EMI Prediction and Suppression of Motor Drive System P

INPUT

CX1

R1

L1

CY2

CX2

Pÿ

CY5

CY3

CY1

L2

CX3

CY4

OUTPUT

CY6

N

Nÿ

CX1=CX2=CX3=10uH,L1=L2=47uH,CY1=CY2=470pF,CY3=CY4=0.1uF,CY5=CY6=0.22uF

Fig. 3.118 Scheme 2 filter topology diagram

3.5.3.2

EMI High Voltage DC Power Filter- Scheme 2

(1) Filter design In view of the problem that the insertion loss of the filter scheme 1 becomes smaller in the low and high frequency bands, improvements are made on the basis of scheme 1. For the small low-frequency DM insertion loss, another 10 uF X capacitor is added to the output port of the filter; for the small low-frequency CM insertion loss, a pair of Y capacitors are added to the input of the filter and readjust the capacitance of the each level Y capacitors. In view of the small high-frequency insertion loss of the filter, one of the CM magnetic rings is replaced with a ferrite magnetic ring with better high-frequency performance, and the leads of the capacitors are shortened as much as possible to reduce the parasitic inductance caused by the lead of the Y capacitor. The improved filter circuit topology diagram is shown in Fig. 3.118, the filter 3D layout diagram is shown in Fig. 3.119a, and the prototype picture is shown in Fig. 3.119b. The simulation results of CM and DM insertion loss are shown in Fig. 3.119c, d. The insertion loss in 150 kHz–108 MHz is greater than 60 dB. (2) Insertion loss measurement With a vector network analyzer, the filter DM and CM S21 parameters are measured. The test layout is shown in Fig. 3.120, and the CM and DM insertion loss test results are shown in Fig. 3.121a, b. It can be seen that after the improvement, the CM insertion loss and the DM insertion loss have been greatly improved, and the insertion loss in 300 kHz–40 MHz is greater than 60 dB.

3.5.3.3

EMI High Voltage DC Power Supply Filter-Scheme 3

Aiming at the problem that the high frequency insertion loss of filter-scheme 2 does not meet 60 dB in some areas, it is improved by adding high frequency inductors and a pair of 4.7nF Y capacitors to the filter output port. The physical map of the improved filter is shown in Fig. 3.122. A vector network analyzer is used to measure the insertion loss of the filter scheme 3. The test results of CM and DM insertion loss are shown in Fig. 3.123a, b.

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(a) Scheme 2 filter 3D model

159

(b) Scheme 2 filter Installation physical map

˄c˅Filter CM insertion loss simulation result

(d) Filter DM insertion loss simulation result

Fig. 3.119 Scheme 2 filter insertion loss simulation results

Fig. 3.120 Scheme 2 filter S parameter test layout diagram

Through measurement, it is found that the common mode insertion loss of the scheme 3 filter is greater than 60 dB in 266 kHz–108 MHz, and greater than 41 dB in 150KHz–266 kHz; the DM insertion loss is greater than 60 dB in 232 kHz–108 MHz, and greater than 45 dB in 150 kHz–232 kHz. Compared with the first and second scheme, the insertion loss of filter scheme 3 is the best in the entire frequency band, but in actual engineering applications, a magnetic ring needs to be added, which increases the size and cost of the filter.

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3 EMI Prediction and Suppression of Motor Drive System

(a)CM S parameter

(b)DM S parameter Fig. 3.121 Scheme 2 filter S parameter test results

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Fig. 3.122 Scheme 3 filter physical installation diagram

Fig. 3.123 Scheme 3 filter S parameter test results

(a)CM S parameter

˄b˅DM S parameter

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3.5.3.4

3 EMI Prediction and Suppression of Motor Drive System

Insertion Loss Motor System Load Test

(1) Conducted emission The conducted disturbance voltage of the high-voltage positive power line of the motor controller before and after the filter scheme 2 was measured under the working condition of 12Nm and 1900 rpm. Figure 3.124 shows the conducted emission test layout with the filter module. Figure 3.125 is the comparison of the conducted voltage before and after filtering. It can be seen that the conducted voltage of the high-voltage positive power line of the motor controller with filter module meets the CISPR252016 level 3 limit. Figure 3.126 shows the insertion loss of the filter module in the system. It can be seen that the peak and average values of the conducted voltage of the high-voltage positive power line of the motor controller have dropped significantly. Especially at 5 MHz, the average insertion loss of the filter module reaches 48 dB, the peak insertion loss reaches 47 dB. In addition, due to the noise floor of the test

Fig. 3.124 Conducted emission layout with filter module

Fig. 3.125 Comparison of conducted voltage test results (10 kHz–108 MHz)

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Fig. 3.126 Insertion loss of filter module (10 kHz–108 MHz)

environment, the actual insertion loss of the filter module is higher than the test result in Fig. 3.126. (2) Radiated emission In order to verify the suppression effect of the filter module on the radiated emission, the radiation emission of the motor drive system with and without the filter module was measured at 53Nm and 1900 rpm according to the standardCISPR25-2016. The radiated emission test results are shown in Fig. 3.127. Only the vertical polarization test results of the biconical antenna, logarithmic antenna and horn antenna are given, because the vertical polarization test result of the antenna is higher than the horizontal polarization test result. Under load conditions, although the radiated emission of the motor drive system meets the requirements of CISPR25-2016 radiation disturbance level 3 limit, the average value and peak value increase in some frequency bands.

3.5.4 Design of Air-Core Inductor EMI Filter for High-Voltage DC Power Supply (1) Selection of filter topology The topology selection of the filter mainly considers the source impedance and load impedance. The filter is applied to the HV DC port of the motor controller, and the output side of the filter is connected to the HV power battery. The battery is resistive and capacitive and exhibits low impedance. Therefore, the basic topology of the filter is selected as the LC type, and in order to make the filter take into account the suppression of the DM signal and the CM signal, the inductors are selected as DM inductors and the capacitors are selected as Y capacitors, so that the inductor and Y capacitor can be used to suppress CM signals, a DM inductor and the equivalent X capacitor of a pair of Y capacitors in series is used to suppress DM signals. At the same time, the combination of DM inductors and CM capacitors can also make the corner frequencies of the DM and CM filter circuits consistent. There is no need to select the number of stages and corner frequencies for the CM circuit and the DM circuit separately, which can simplify the design process.

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3 EMI Prediction and Suppression of Motor Drive System

(a) Comparison of test results of pole antenna before and after the filter module

(b) Comparison of the biconical antenna vertical polarization test results before and after the filter module

(c) Comparison of the log-period antenna vertical polarization test results before and after the filter module

(d) Comparison of the dual-ridged horn antennas vertical polarization test results before and after filter module

Fig. 3.127 Comparison of the radiated emission before and after the filter module

(2) Determination of the number of filter stages and corner frequencies The number of filter stages and the corner frequency of each stage are mainly related to the frequency range and insertion loss requirements. The filter in this book is required to work in the full range of 150 kHz–108 MHz, and the insertion loss is 60 dB. According to engineering experience, the insertion loss curve of a single-stage filter between the corner frequency and 10 times the corner frequency will basically follow 20n dB/10dec (where n is the number of components in each stage, which is

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165

related to the basic topology of the filter), and the insertion loss curve will gradually deviate from the theoretical curve between 10 times-100 times the corner frequency, but the insertion loss of 20n dB can still be maintained. After 100 times the frequency, the suppression effect will gradually become worse until there is no effect. Due to the wide frequency range of 150 kHz–108 MHz, a single-stage filter cannot meet the requirements. Although the insertion loss effect will become better as the number of filter stages increases, the size of the filter will also increase exponentially, and it will also bring about complex problems such as component resonance. Considering that the conducted interference noise of the motor controller is quite large, the filter is designed as a three-stage filter. Since the basic circuit is an LC circuit, the number of components is n = 2, so the insertion loss below 10 times the frequency meets 40 dB/10dec. If only the first-stage filter circuit is used to meet the 60 dB insertion loss requirement at 150 kHz, the first corner frequency of the stage will be very low, causing the corresponding capacitance and inductance parameters to be too large, affecting the volume of the entire filter. Therefore, the first stage and the second stage filter circuit are used together to meet this requirement. The corner frequency point of the first-stage filter is set to be 20 kHz, and its insertion loss at 150 kHz can reach 35 dB, so the second-stage filter needs to have an insertion loss of 20 dB at 150 kHz, and the second-stage filter’s corner frequency can be calculated through the formula (3.25): f0 =

f1 I L1

10 20n

(3.35)

where, f 0 is the corner frequency, f 1 is the frequency point needed to be suppressed, here is 150 kHz, IL 1 is the target insertion loss that needs to be achieved at the frequency point f 1 , here is 20 dB, and n is the order of the filter, here is 2. Substituting the above data into (3.35), f 20 = 70 kHz can be obtained. The filter corner frequency of the third-stage filter is selected as 130 kHz, and the effective insertion loss range is pulled above 10 MHz. For the filter, it is difficult to predict the insertion loss curve above 10 MHz, and it is more to make some subtle adjustments based on experience, so the corner frequency should not be too high. (3) Determination of filter parameters After determining the corner frequency of each stage of the filter, it is necessary to determine the capacitance and inductance parameters according to the corner frequency. Here, L 1 L 2 L 3 are chosen as three identical inductors. Different corner frequencies are realized by adjusting the size of each level of capacitance. Since these inductors are all DM inductors, the current flowing through each inductor is the entire operating current. In EVs, the operating current of the motor drive system is as high as 300–500A. Under such a large current, the general magnetic core material will be saturated, resulting in extremely small inductance.

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3 EMI Prediction and Suppression of Motor Drive System

Therefore, the air-core inductor is used as the DM inductor. Its main advantage is that there is no saturation problem at high current, and the inductance can be better maintained at high frequency. In order to ensure that the air-core inductor can withstand the current when the motor drive system is working, a solid copper wire with a diameter of 8 mm is selected, and the surface of the copper wire is tinned to prevent its oxidation. The inductance value of the air-core inductor can be calculated as follows: L=

0.01 × D 2 × N 2 l + 0.44 × D

(3.36)

where, L is the inductance of the coil [μH], l is the length of the coil [cm], N is the number of turns of the coil, and D is the cross-sectional diameter of the coil [cm]. Considering that the overall volume of the filter should not be too large when used in vehicles, the average coil diameter is selected to be 4.2 cm, the coil length is 4.1 cm, and the number of turns is 4. The final calculated inductance of air-core inductor is 0.74uH. The air core inductor model and physical diagram are shown in Fig. 3.128a, b respectively. The vector network analyzer is used to measure the inductance of the air-core inductor, and the inductance is 0.64uH, which is consistent with the theoretical calculation. f =

1 √ 2π L 1 CY

(3.37)

The relationship between filter corner frequency and capacitance, inductance is shown in formula (3.37). The corner frequency of the first stage is f 10 = 20 kHz, and C y1 = C y2 = 98.9uF, which is taken as 90uF; the second stage corner frequency is f 20 = 70 kHz, and C y3 = C y4 = 8.1uF, which is taken as 8uF; The third-stage corner frequency is f 30 = 130 kHz, and C y5 = C y6 = 2.3uF, which is taken as 2.2uF. At the same time, because the capacitance of the selected capacitors is large, the energy stored in the capacitors needs to be discharged through an additional channel, so a resistor with the resistance value of 1 M is connected in parallel with each capacitor.

(a) 3d model

Fig. 3.128 Three air-core inductors in series

(b) Physical map

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167

Therefore, the final filter topology and parameters are shown in Fig. 3.129: (4) Design of filter principle prototype The 3D model of the air-core inductor filter is shown in Fig. 3.130, and the physical diagram of the filter is shown in Fig. 3.131. As can be seen from the Fig. 3.131, in order to reduce the coupling of the noise signal between each stage of the filter, the three-stage filters are distributed in a straight line, the Y capacitors are directly arranged under the inductors, and the pin is connected to the inductor and the filter chassis nearby. At the same time, in order to ensure the insulation problem of the filter, the air-core inductor is wrapped with the special insulating green shell paper for the filter and potted with epoxy resin. (5) Simulation of filter insertion loss In order to verify the performance of the filter, referring to the filter measurement standards CISPR 17-2011 and CISPR 25-2016, the insertion loss of the filter is analyzed through ADS software when the source impedance and load impedance are both 50. ➀ Filter CM insertion loss The simulation model and simulation results of the filter CM insertion loss are shown in Fig. 3.132. It can be seen that the filter CM insertion loss is greater than 60 dB at 150 kHz–108 MHz.

L1

L1'

L1

L2

L2

L3

CY1

R

CY3

R

L5 CY5

R

CY2

R L CY4 4

R

CY6 L6

R L2'

L1~L6=0.64uH,CY1=CY2=90uF,CY3=CY4=8uF,CY5=CY6=2.2uF,R=1MΩ Fig. 3.129 Air-core inductor high-voltage DC power supply EMI filter topology 0.64µH air core inductors Chassis

Insulation pad 2.2µF Y capacitor

8µF Y capacitor 90µF Y capacitor

Fig. 3.130 3D model diagram of air core inductor filter

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3 EMI Prediction and Suppression of Motor Drive System

Fig. 3.131 Principle prototype of air-core inductor filter

Fig. 3.132 Modeling and simulation of CM insertion loss of air core inductor filter

(a) Simulation model

(b) Insertion loss simulation results

➁ Simulation of filter DM insertion loss The simulation model and simulation results of the filter DM insertion loss are shown in Fig. 3.133. It can be seen that the CM insertion loss of the filter is greater than 60 dB at 150 kHz–108 MHz. (3) Filter insertion loss test According to the standard CISPR 17-2011, the motor controller filter module is connected with the vector network analyzer, and the two-port CM S parameter S 21-CM and DM S parameter S 21-DM of the filter module are measured and obtained. The insertion loss (ae ) of the filter module can be calculated by the formula (3.38): ae = −20lg|S21 |

(3.38)

➀ S21-CM measurement S 21-CM measurement layout and S 21-CM amplitude measurement results are shown in Fig. 3.134. The S 21-CM amplitude measurement data result is saved, and the MATLAB software is used to calculate according to the formula (3.28) to obtain the CM insertion loss ae-CM curve of the filter module in 150 kHz–108 MHz, as shown in Fig. 3.135,

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(a) Simulation model

(b) Insertion loss simulation results

Fig. 3.133 Modeling and simulation of DM insertion loss of air core inductor filter

(a) S21-CM measurement layout

(b) S parameter amplitude measurement results (9kHz-108MHz)

Fig. 3.134 Measurement of air core inductance filter model S 21-CM Fig. 3.135 Air core inductor filter ae-CM measurement results (150 kHz–108 MHz)

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3 EMI Prediction and Suppression of Motor Drive System

(a) S21-DM measurement layout

(b) S parameter amplitude measurement results (9kHz-108MHz)

Fig. 3.136 Air-core inductor filter model S 21-DM measurement

it can be seen that the insertion loss of ae-CM in 150 kHz–108 MHz is greater than 60 dB. ➁ S21-DM measurement S 21-DM measurement layout and S 21-DM amplitude measurement results are shown in Fig. 3.136. The result of the S 21-DM amplitude measurement data is saved, and the MATLAB software is used to calculate according to the formula (3.38) to obtain the DM insertion loss ae-DM curve of the motor drive system filter module in 150 kHz–108 MHz, as shown in Fig. 3.137, it can be seen that the insertion loss of ae-DM at 150 kHz–108 MHz is greater than 60 dB. (3) Load insertion loss test of air core inductor filter Since the simulation and test of the filter insertion loss are obtained when the source impedance and load impedance are both 50 , and the source impedance of the actual motor drive system is not constant, the air core inductor filter actual insertion loss is tested under the motor rated speed. Figure 3.138 is the test layout of the conducted Fig. 3.137 Air-core inductor filter ae-DM measurement results (150 kHz–108 MHz)

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Fig. 3.138 Conducted emission layout with air-core inductor filter

emission-voltage method of the motor system with an air-core inductor high-voltage DC power supply EMI filter module. The test results of the conducted disturbance voltage of the high-voltage DC positive power line are shown in Fig. 3.139. It can be seen from Fig. 3.139 that the peak and average values of the conducted disturbance voltage of the motor drive system high-voltage positive power line have been greatly reduced with an air-core inductor filter, which meets the CISPR 25-2016 level 3 limit requirements. Especially at 60 kHz, the average insertion loss of the air-core inductor filter reached 55 dB, and the peak insertion loss reached 52 dB. In addition, due to the influence of the noise floor of the test environment, the actual insertion loss of the air-core inductor filter is higher than the test result in Fig. 3.140.

Fig. 3.140 Air core inductor filter insertion loss (10 kHz–108 MHz)

Insertion Loss(dB)

Fig. 3.139 Comparison of conducted disturbance voltage test results (10 kHz–108 MHz)

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3.6 Electromagnetic Disturbance Test 3.6.1 Electromagnetic Disturbance Test Platform At present, the latest Chinese standard for electromagnetic disturbance test of electric vehicle components is GB/T 18655-2018. Appendix 1 “Test Methods for Shielded High Voltage Power Supply Systems for Electric and Hybrid Vehicles” clearly stipulates the test methods, test procedures and limits of motor drive systems, Fig. 3.141 is an example of the test layout of conducted emission and radiated emission, which clearly specifies the length and position of the test harness, the position of the DUT, the position of the antenna, and the grounding position. The motor drive system test platform built in the semi-anechoic chamber is shown in Fig. 3.142. Test items can be divided into five aspects: conducted emission (CE), radiated emission (RE), CS (conducted susceptibility), RS (radiated susceptibility), and ESD (electrostatic discharge).

(a)Conducted emission-voltage method

(b) Conducted emission- current probe method

(c)Radiated emission

Fig. 3.141 Example diagram of conducted emission and radiated emission test layout in Chinese standard GB/T 18655-2018

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173

Fig. 3.142 Motor drive system test platform

3.6.2 Conducted Emission-Voltage Method The methods of conducted emission measurement include voltage method and current method. Voltage method mainly uses LISN and receiver or spectrum analyzer to measure conducted disturbance voltage, and current probe method mainly uses current probe and receiver or spectrum analyzer to measure conducted disturbance current. (1) Conducted emission test of HV power lines The voltage method measurements include the measurements of the conducted disturbance voltage of the HV part and the LV part. The output port of the HV DC power supply is connected to the DC shielded cable through the LISNs to provide DC power to the controller. The high-voltage LISNs and the controller are placed on a grounded copper plate with a height of 1 m from the test ground. It should be noted that there is a 50 mm thick plate between the controller and the grounded copper plate, which is made of non-conductive and low relative dielectric constant materials; LISN is well grounded, the controller and the motor are connected with 1500 mm long shielded cables, and the motor output shaft is connected to the motor load through an insulated connecting shaft. The comparison of conducted disturbance voltage test results between the positive and negative ports of the high-voltage DC bus are shown in Fig. 3.143. The red Fig. 3.143 Comparison of conducted disturbance voltage of HV positive and negative cables

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and pink lines respectively represent the peak limit and average limit of the Chinese standard GB/T 18655-2018 level 3. The blue curve and the green curve respectively indicate the peak and average values of the positive and negative conducted disturbance voltages under 25% motor rated conditions, 1910 rpm rotating speed and 53 N · m torque. It can be seen from the Fig. 3.143 that the positive and negative conducted disturbance voltage curves basically coincide. In the frequency band above 10 MHz, the conducted disturbance voltage of the positive port is slightly larger than that of the negative port by about 3 dB. Neither the positive or negative conducted disturbance voltage can meet the limit requirements of Chinese standard GB/T 18655-2018 Level 3, and the average value exceeds the standard more seriously than the peak value. The peak value of the conducted disturbance voltage exceeds the standard at 5.9–6.2 MHz by about 3 dB, and basically exceeds the standard in the 30–90 MHz frequency band. The average value of conducted interference is more serious in the low frequency band below 30 MHz, like 700 kHz–1.8 MHz, 5.9–6.2 MHz, 35–50 MHz and the maximum over-standard values are 11 dB, 15 dB, 8 dB respectively. The conducted disturbance voltage of the high-voltage cable was measured under no-load conditions 400 rpm, 0 N · m and rated conditions 1900 rpm, 53 N · m. The comparison of the test results is shown in Fig. 3.144. It can be seen from the Fig. 3.144 that the amplitude of the conducted disturbance voltage under the rated operating condition than that during idling. (2) Conducted emission test of LV power line-voltage method Under the rated condition, the conducted interference test results of the positive and negative ports of the LV DC power lines are shown in Fig. 3.145, which shows that the requirements of Chinese standard GB/T 18655-2018 level 3 limit cannot be met. The conducted disturbance voltage curves of the LV positive and negative power lines basically coincide in the frequency band below 7 MHz, but the conducted disturbance voltage of the positive port in the 10–20 MHz frequency band is significantly higher than that of the negative port. Whether it is the positive port or the negative port, the average value exceeds the standard more seriously. Except for the conducted emissions of 0.15 MHz–0.30 MHz, the conducted emissions of other frequency bands are all exceeding the standard level 3 limit. And the exceeding is very serious, especially at 1, 5 MHz, and other frequency points, the exceedance is greater than 20 dB. As shown in Fig. 3.145b, it can also be seen that the conducted disturbance Fig. 3.144 Comparison of conducted interference test results under two different working conditions

175

µ

3.6 Electromagnetic Disturbance Test

(a) 150kHz-108MHz

(b) 30MHz-108MHz

Fig. 3.145 Comparison of the conducted disturbance voltage between the positive and negative ports of LV cables Fig. 3.146 Comparison of conducted interference of HV and LV power lines

voltage has a lot of spikes with an interval of about 30 kHz. This is mainly caused by the harmonics of the pulse signal generated after the IGBT is turned on. Therefore, the EMC problem of the drive system requires attention not only to the HV part, but also to the LV part. (3) Comparison of conducted emission test results of high and low voltage power lines The comparison of conducted interference test results of HV and LV power lines is shown in Fig. 3.146. The overall conducted interference amplitude of HV power lines is much larger than that of LV power lines, but there are spikes in 5–6 MHz, and a valley at 620 kHz, indicating that there is still a certain correlation between them.

3.6.3 Conducted Emission-Current Probe Method (1) Conduction disturbance current test of HV power line The actual test layout of the conducted disturbance current on the HV power line of the motor controller for current probe testing is shown in Fig. 3.147. The current probe in Fig. 3.147a is 750 mm away from the controller, and the current probe in Fig. 3.147b is 50 mm away from the controller.

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3 EMI Prediction and Suppression of Motor Drive System

(a)750mm

(b) 50mm

Fig. 3.147 Test layout of HV power line conducted emission-current method

Fig. 3.148 Comparison of a single HV power cable conducted disturbance current and CM conducted disturbance current

As shown in Fig. 3.148, it is found that the conducted disturbance current of a single positive power line is 2–5 dB larger than the CM current below 10 MHz; above 14 MHz, the CM current is larger than the conducted disturbance current of a single power line, except for a small frequency band near 50 MHz, where the CM current is smaller than the conducted disturbance current of a single power line. The comparison of the CM current test results of the positive and negative power lines at a distance of 50 mm and 750 mm from the controller is shown in Fig. 3.149. It can be seen that in the frequency band 40–60 MHz, the conducted interference at 50 mm is more serious than that at 750 mm, especially at 50 MHz, the CM current of 50 mm is 20 dB larger than that of 750 mm. It can be seen that the CM conducted disturbance currents at different locations are inconsistent at high Fig. 3.149 Comparison of CM conducted disturbance currents of HV power lines at different positions

3.6 Electromagnetic Disturbance Test

(a) 50mm from the controller

177

(b) 750mm from the controller

Fig. 3.150 LV power line conducted interference test-current method actual layout

frequencies, indicating that even if the HV cable has a shielding layer, due to the transfer impedance of the cable, the power line can be equivalent to an antenna at high frequencies and form radiated emission. (2) Conducted disturbance current test of LV power lines The conducted interference test on the LV power lines with current probe is shown in Fig. 3.150. The current probe is 50 mm away from the controller in Figure a, and the current probe is 750 mm away from the controller in Figure b. The test results are shown in Fig. 3.151. Below the 30 MHz, the test results at the two places basically overlap. The conducted interference of the high frequency part at 50 mm is more serious. Above 40 MHz, except for a small band near the 87 and 100 MHz frequencies. The conducted interference at the 50 mm position is larger than that at the 750 mm position, and the maximum value appears at the 63 MHz frequency point, which is 16 dB larger. (3) CM disturbance current test of the three-phase power line of the motor The CM current of the three-phase HV cable connecting the motor controller and the motor is measured at two positions 50 mm and 750 mm away from the controller. The comparison of the test results is shown in Fig. 3.152. Below the frequency of

(a) 150kHz-108MHz

(b) 30MHz-108MHz

Fig. 3.151 Comparison of CM conducted disturbance current of LV lines at different positions

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Fig. 3.152 Comparison of CM currents of motor three-phase power lines

30 MHz, the CM conducted curves at the two positions still basically overlap, and the curves are quite different in the high frequency. The comparison of the CM current test results of the DC cable and the AC cable at the position 50 mm away from the controller is shown in Fig. 3.153. The CM current of the three-phase AC cable at 600 kHz–108 MHz is greater than the CM current of the DC bus, except for the frequency band near 50 MHz. At 150 kHz–600 kHz, the CM current of the DC bus is greater. (4) Comparison of CM conducted disturbance current of HV and LV power lines As shown in Fig. 3.154, the CM current of the LV lines is larger than that of the HV power lines in the entire frequency band, and there are large peaks near 60 MHz. It means that the CM interference on the LV lines, including the LV power lines, CAN bus (communication), and enable line is greater. The electric field probe was used to scan the LV control board of the controller, and there is a large peak amplitude near Fig. 3.153 Comparison of CM current of the HV DC power lines and the motor three-phase AC lines

Fig. 3.154 Comparison of CM conducted disturbance current between HV power lines and LV lines

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179

60 MHz, indicating that the 60 MHz CM current spike is caused by the LV control board and coupled to the HV cable.

3.6.4 Coupling of LV Power Lines to HV Power Lines When only 12 V LV power is provided to the controller, but not high voltage, the drive circuit of the power module works normally, and the motor does not work. In this case, the CM current of the HV power line is measured at the distance of 50 mm from the controller. The test result is shown in Fig. 3.155. The test is mainly to analyze the 30 M-108 MHz interference signal coupling from the LV part to the HV part. Figure 3.155a shows the test results of HV CM current without any methods. It can be seen that the control circuit and drive circuit of the LV part will generate large high-frequency interference signals when they are working. And these interference signals are coupled to the HV part and make the conducted interference of the HV power line when the motor is not running no longer meet the requirements of level 3. Figure 3.155b–g is the test result after adding improvement methods inside the controller, the purpose is to suppress the interference and study the coupling mechanism. Figure b is the test result after adding a 47nF Y capacitor to the HV positive and negative bars inside the controller. The CM current of the HV power line does not decrease but increases, indicating that the Y capacitor increases the coupling of LV part to HV part; Figure c is the test result after adding a magnetic ring to the

(a) Coupling of LV part to HV part (b) Add 47nF ground capacitance to HV positive and negative ports (c) Add magnetic ring to LV power lines (d) Remove the controller grounding lines

˄e˅Add a magnetic ring to the LV power lines˄f˅Remove the 47nF capacitance to ground ˄g˅Add a 3-turn magnetic ring to the LV power lines˄h˅LV conducted interference

Fig. 3.155 EMI coupling of LV part to HV part

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LV power line. It can be seen that the CM current at 50 and 62 MHz has decreased, indicating that the inductance can reduce the coupling of LV part to HV part; Figure (d) is the test result after the controller chassis grounding wires are removed. It can be seen that the CM current of the HV power line drops more than 10 dB near 63 MHz, indicating that grounding wires have a certain effect on the coupling of LV part to HV part. In the frequency band below 30 MHz, the electromagnetic coupling of LV part to HV part is not obvious. Figure (e) is the test result after adding a magnetic ring to the LV power line. It can be seen that the CM current at 62 MHz has no obvious change. The CM current at 30–50 MHz has more spikes, but the amplitude only increases slightly. Figure (f) is the test result after removing the 47nF Y-capacitor. It can be seen that the CM current of 69 MHz drops slightly, but a new spike appears at 50 MHz, but does not exceed the standard. This shows that when there are no grounding wires, the capacitance to ground slightly strengthens the CM current of 69 MHz, but attenuates the CM current spike of 50 MHz. Figure (g) is the test result after a three-turn magnetic ring is connected in series on the LV lines. It is found that the CM current near 45 MHz is greatly enhanced, forming a new peak, and the peak curve and average curve are both exceeding the standard. The CM current at 69 MHz also rises slightly, indicating that after removing the ground wire, increasing the series inductance will increase the coupling of LV part to HV part. Figure (h) shows the CM current test results of the LV power lines after the methods in Figures b–g are implemented. It can be seen that the CM current spikes of the LV lines and the HV lines correspond, which further proves that the conducted interference of HV lines is the result of coupling of LV lines. At the same time, the CM current amplitude of the LV lines is lower than that of HV lines, which shows that after the methods in Figures b–g, the impedance of the CM current propagation path of the LV lines is higher than that of the HV lines, so that more CM current flows through the HV power lines.

3.6.5 Conducted EMI Suppression The conducted disturbance current of either the LV part or the HV part of the motor drive system does not meet the limit requirements of the Chinese standard GB/T 18655-2018 level 3, so measures are taken to suppress the conducted interference and finally the level 3 limit requirements are met. (1) LV power line conducted interference suppression The comparison of the conducted interference of the LV lines when the motor is working normally and when the control circuit is only supplied with 12 V voltage and the drive circuit is not working is shown in Fig. 3.156. The upper two curves are the peak and average values of the conducted disturbance current when the motor is working normally (the green curve is the peak value and the pink curve is the average

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181

Fig. 3.156 Comparison of conducted interference between when the motor is working and when only the control board is working Fig. 3.157 LV lines conducted disturbance current at 30 M-108 MHz when only the control board is working

(a) Only low voltage power lines (b) CAN line (c) Resolver line

value). Figure 3.157 shows the conducted disturbance current of the LV lines at 30– 108 MHz when only 12 V is provided to the control circuit and the drive circuit is not working (the blue curve is the peak value, the green curve is the average value, the red line is the peak limit, and the purple line is the mean limit). It can be seen from Fig. 3.156 and Fig. 3.157 that when only the control circuit is working, large high-frequency interference signals are generated in the LV cables in the 30–108 MHz, and the current peak and average values have over-standard points. When only 12 V voltage is provided to the controller and the control circuit and drive circuit are working, the LV conducted interference are shown in Fig. 3.158. Compared with Fig. 3.157, the driving circuit produces large low-frequency interference signals when it is working. In order to suppress the conducted interference of the LV lines, a 500 μH DM inductor is connected in series with the positive and negative ports of the LV power line near the controller, and a pair of 100uF Y capacitors are connected between the LV power bars and the chassis; a ferrite magnetic ring is placed on the LV power lines and resolver lines near the controller, and the CAN line grounding is removed. The test results are shown in Fig. 3.159. The conducted interference of the LV wiring harness meets the conducted disturbance current limit of the standard level 3 in 150 kHz–108 MHz.

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3 EMI Prediction and Suppression of Motor Drive System

Fig. 3.158 Conducted disturbance current of LV lines when only the control circuit and drive circuit are working

Fig. 3.159 The rectified LV lines conducted disturbance current

(2) HV power lines conducted interference suppression On the basis of rectifying the LV part, the current probe method is used to measure the conducted disturbance current of HV positive power line at 50 mm away from the controller. The test results are shown in Fig. 3.160 (The blue curve is the peak value, the green curve is the average value, and the red line is the peak limit and the purple line is the average limit). It can be seen from the Fig. 3.160 that the requirements of level 3 are met in the whole frequency band except for the peak over-standard points near 46 MHz.

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Fig. 3.160 HV positive power line conducted disturbance current at a distance of 50 mm from the controller

Three pairs of capacitors to ground with the capacitance of 1 μF, 100nF, 47nF are connected to the positive and negative ports of the HV DC bus at the motor controller end. The conducted disturbance voltage of the HV DC positive bus is measured. The test results are shown in Fig. 3.161, which meets the requirements of the standard level 3 limit.

Fig. 3.161 DC bus positive port conducted disturbance voltage

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3.6.6 Voltage and Current of Motor Drive System DC and AC Power Lines Figure 3.162 shows the time-domain waveforms of voltage and current collected by oscilloscope. The green curve is the positive voltage of the DC bus to ground, the yellow curve is the positive current of the DC bus, and the orange curve is the W phase-to-ground voltage at the output port of IGBT power module. The RMS value of the DC bus positive to ground voltage is 170.5 V, and the RMS value of the DC bus current is 34.73A. The voltage and current on the DC bus appear spikes simultaneously, and they are both in the form of ringing. These ringing all appear at the time when the six-channel drive signal SVPWM wave is switched, indicating that each IGBT of the power module will generate voltage and current spikes in the DC bus at the moment of turn-on and turn-off. The partial amplification of voltage and current is shown in Fig. 3.163, where Fig. 3.163a, b are the partial enlarged diagrams of the three-phase output W of the IGBT power module relative to the ground voltage

Fig. 3.162 Voltage and current waveform of motor drive system

(a) The moment when W phase-to-ground voltage drops (b) The moment when W phase-to-ground voltage rises (c) DC bus current (d) DC bus voltage

Fig. 3.163 An enlarged view of the voltage and current waveform of the motor drive system

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185

rise and fall, and (c) is the partial enlarged diagrams of DC bus current when the W phase line relative to ground voltage rises. It can be seen from Fig. 3.163a, b that it takes a period of time to turn on and turn off the IGBT power module. The falling time of W to ground voltage is 0.38 μs, the rising time is 0.32 μs, and the falling peak voltage is 20 V, the rising peak voltage is 40 V. The voltage change rate du/dt is 1218 V/μs in 0.32 μs, then a 1.28A transient electromagnetic pulse can be generated through a 1nF capacitor, which can cause serious electromagnetic interference to the system. From Fig. 3.163a, b, it is also known that the ringing period of the W-phase output voltage to ground is around 0.20 μs, and its frequency is 5 MHz; at the same time, it can be seen from Fig. 3.163(d) that the amplitude of the DC bus voltage oscillation is as high as 34 V, and the ringing period is also around 0.20 μs, which is as large as the ringing period of the W phase output to ground voltage, and the frequency is 5 MHz, which corresponds to the conducted disturbance voltage peak of the motor drive system in Fig. 3.148 near 5 MHz. At the same time, it can be seen from Fig. 3.163c that the current spike oscillation of the DC bus is more serious than the voltage. The current spike of the DC bus can reach 95A. The current changes regularly in a certain period, and the amplitude changes sharply at the moment of switching, the period don’t change. And the current change frequency is about 1 MHz, which corresponds to the conducted disturbance voltage peak of the motor drive system in Fig. 3.148 near 1 MHz. The shorter the peak oscillation time or more severe voltage and current waveforms is, there are more or larger high-frequency spectral components. In the datasheet of the IGBT power module FS800R07A2E3, under the inductive load test with an internal temperature of 25 °C, the voltage rise time and fall time are respectively 0.1 μs and 0.04 μs, while the rise time and fall time of the three-phase output voltage are 0.32 μs and 0.38 μs respectively when the motor drive system is working, which are obviously much larger than those of the individual IGBT chip test in the datasheet, indicating that the stray parameters of the internal structure of the motor controller have a significant delay effect on the turn-on and turn-off time of the actual output voltage of the power module.

3.6.7 The Influence of Speed and Torque on Conducted Disturbance Voltage and Current In order to verify the relationship between the conducted EMI of the drive system and the speed and torque of the motor, the conducted disturbance voltage of the motor drive system was tested at different speeds and torques. The test results are shown in Fig. 3.164. Figure 3.164a is a comparison diagram of the peak and average values of the conducted disturbance voltage for different torques of 25Nm, 53Nm and 100Nm at a constant speed of 1910 rpm. Figure 3.164b is a comparison diagram of the peak and average values of the conducted disturbance voltage for different speeds of 400 rpm, 1910 rpm and 6000 rpm at a constant torque of 53Nm. From the figure,

3 EMI Prediction and Suppression of Motor Drive System 140

120

120

100 µ

µ

186

100

80

80

60

60

40

40 0.1

1

10

30

20 0.1

108

1

10

30

108

10

30

108

(a) When the speed is constant and the torque changes 120

140

100 µ

µ

120 100

80 60

80

40

60 40 0.1

20 1

10

30

108

0 0.1

1

(b) When the torque is constant and the speed changes

Fig. 3.164 The influence of torque and speed on the conducted disturbance voltage

it can be seen that the peak and average values of the conducted disturbance voltage change little with the speed and torque. The spectrum curves basically overlap. When the speed is constant, the larger the torque the overall peak and average value of the conducted disturbance voltage are slightly larger. Similarly, when the torque is constant, the higher the speed, the overall peak and average value of the conducted disturbance voltage are slightly larger, as shown in Fig. 3.164b. When the motor speed is 3800 rpm, 1900 rpm, 1000 rpm and the torque is 53 N · m, the peak value and average value of the conducted current of the DC positive power line measured at a distance of 50 mm from the motor controller are shown in Fig. 3.165a, b, it can be seen that under the same torque, the change of speed has little effect on the conducted current, and the peak value of the 3800 rpm speed condition is slightly higher than the other two speed conditions. When the speed is 1900 rpm, the torque is 25Nm and 53Nm, the peak and average values of the conducted current of the positive power line are shown in Fig. 3.166a,

(a) Peak

(b) Average

Fig. 3.165 The effect of speed on the conducted current of the positive power line

3.6 Electromagnetic Disturbance Test

(a) Peak

187

(b) Average

Fig. 3.166 The effect of torque on the conducted current of the positive power line

(a) Peak

(b) Average

Fig. 3.167 The influence of torque on AC conduction current

b. It can be seen that the change of torque has little effect on the conducted current. At a certain speed, the greater the torque, the overall peak and average values of the conducted disturbance current are slightly larger. Figure 3.167 shows the frequency domain conducted current peak value of threephase AC lines measured at 1900 rpm. It can be seen that under the first condition when the IGBT is already working and the motor torque and speed are still zero, the peak conducted current on a single shielded AC wire exceeds the standard obviously in 0.53 MHz–1.8 MHz. Under the condition of 1900 rpm and 25Nm, the peak conducted current on a single shielded AC line is slightly larger than the first condition. Under the condition of 1900 rpm and 25N · m, the peak conducted current on a single AC line without shielding layer increased by about 10 dBμA overall. The maximum over-standard values in 35–48 MHz and 48.5–55 MHz are 13 dBμA and 11 dBμA respectively. The sky blue curve shows the peak CM conducted current of the three three-phase AC lines at 1900 rpm and 53 N · m. It is about 20 dBμA smaller than the dark blue curve before 2 MHz. There are peak points at 2.3, 4.2 and 6.2 MHz. And the sky blue curve basically coincide with the blue curve between 30 and 108 MHz. The average value of the conducted current in the frequency domain of the three-phase AC lines measured at 1900 rpm are shown in Fig. 3.167b. It can be seen that the change trend is similar to the peak value.

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3 EMI Prediction and Suppression of Motor Drive System

3.6.8 Electromagnetic Radiation Test (1) Radiated emission According to the Chinese standard GB/T 18655-2018, the radiated emission of the motor drive system was measured in the frequency bands of 150 kHz–30 MHz, 30– 200 MHz, 200–1000 MHz and 1 GHz–2.5 GHz under the motor rated operating conditions of 53Nm and 1900 rpm. The test results are shown in Fig. 3.168 (the blue curve is the peak value, the green curve is the average value, the red line is the peak value limit, and the purple line is the average value limit). It can be seen that the radiated emissions in the 150 kHz–30 MHz and 30–200 MHz frequency bands all have over-standard points and cannot meet the requirements of the standard level 3 limit. The vertical polarization of the biconical antenna is higher than the horizontal polarization test value. It is necessary to adopt EMI suppression methods to reduce the radiated emission of the motor drive system to meet the standard limit requirements. (2) Near-field electromagnetic radiation In order to predict the characteristics of the radiated emission generated by the motor cable, the near-field magnetic field near the DC cable and AC cable of the motor unit was tested. (1) Magnetic field test of high voltage positive DC cable The HV DC positive line near-field electromagnetic field test layout is shown in Fig. 3.169, the magnetic field strength of the cable in three directions are measured. The comparison results of the magnetic field peak and average intensity of the DC line in the three directions are shown in Fig. 3.170. As can be seen: ➀ The frequencies of the magnetic field intensity peaks in the three directions are basically the same. There are peaks around 5, 10, 15 and 60 MHz in all of the three results. And there is a certain correspondence with the radiation emission shown in Fig. 3.168a. ➁ The result in the first direction has one more peaks around 1 MHz than the results in the second and third directions, and two more peaks above 30 MHz. ➂ The magnitude of the magnetic field in the three directions differs greatly. The first direction is the largest, followed by the third direction, and the second direction is the smallest. (2) Magnetic field strength test of AC lines The AC line magnetic field strength test layout is shown in Fig. 3.171. The peak and average results of the shielded wire and unshielded wire magnetic field strength are compared, as shown in Fig. 3.172. The following conclusions can be drawn: ➀ The magnetic field results of shielded and unshielded AC lines both have a peak around 12 MHz below 30 MHz, and the magnetic field strength is relatively large around 1 MHz–2 MHz.

3.6 Electromagnetic Disturbance Test

189

(a)Pole antenna

(b) Biconical antenna horizontal polarization

(c) Biconical antenna vertical polarization

(d) Log periodic antenna horizontal polarization (e) Log periodic antenna vertical polarization

(f) Double-ridged horn antenna horizontal polarization (g) Double-ridged horn antenna vertical polarization

Fig. 3.168 Radiated emission test result of motor drive system

➁ Below 30 MHz, the change trend of the magnetic field intensity curve of shielded wire and unshielded wire is basically the same, but the amplitude of shielded wire is about 7 dB lower than that of unshielded wire. ➂ Above 30 MHz, the magnetic field strength of the unshielded wire has a large peak around 60 M, and the magnetic field strength of the shielded wire has no peak.

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3 EMI Prediction and Suppression of Motor Drive System

(a)The first direction (b)The second direction (c)The third direction

Fig. 3.169 Layout of DC line magnetic field strength test

˄a˅Peak

(b) Average

Fig. 3.170 Magnetic field strength in three directions of DC line

(a) Shielded line

(b) Unshielded line

Fig. 3.171 Layout of AC line magnetic field strength test

˄a˅Peak

(b) Average

Fig. 3.172 Magnetic field strength in three directions of AC line

3.6 Electromagnetic Disturbance Test

(a)Peak

191

(b)Average

Fig. 3.173 Conducted disturbance CM and DM components

3.6.9 Conducted Disturbance Voltage DM and CM Separation Test A DM and CM separator is used to separate the conducted disturbance of the HV DC positive power line for DM and CM components. It can be seen from Fig. 3.173 that when the DC operating current is 70A, the peak and the average results of the conducted disturbance voltage of the HV DC positive power line both have overstandard areas; the total peak value and average value of the conducted disturbance voltage almost overlap with the peak value and average value of the CM conducted disturbance voltage. It can be seen that the conducted interference and its overstandard point are mainly caused by CM interference. It provides a test basis for the design of the motor controller HV DC power filter.

References Ardon V, Aime J, Chadebec O, et al (2009) MoM and PEEC method to reach a complete equivalent circuit of a static converter. In: International Zurich symposium on electromagnetic compatibility. IEEE, pp 273–276 Ardon V, Aime J, Chadebec O et al (2010) EMC modeling of an industrial variable speed drive with an adapted PEEC method. IEEE Trans Magn 46(8):2892–2898 Bondarenko N, Zhai L, Xu B et al (2015) A measurement-based model of the electromagnetic emissions from a power inverter. IEEE Trans Power Electron 30(10):5522–5531 Chen S, Nehl TW, Lai JS et al (2003) Towards EMI prediction of a PM motor drive for automotive applications. In: IEEE applied power electronics conference & exposition CISPR (2009) Vehicles, boats and internal combustion engines—Radio disturbance characteristics—Limits and methods of measurement for the protection of off-board receivers: CISPR 12-2009 Ed. 6. 1. IEC, Switzerland CISPR (2015) Vehicles boats and internal combustion engines—Radio disturbance characteristics— Limits and methods of measurement for the protection of on-board receivers: CISPR 25:2015 Ed. 4.0. IEC, Switzerland Falck E, Stoisiek M, Wachutka G (1997) Modeling of parasitic inductive effects in power modules. In: IEEE international symposium on power semiconductor devices & Ics, IEEE, pp 129–132 Lai JS, Huang X, Pepa E et al (2006) Inverter EMI modeling and simulation methodologies. IEEE Trans Ind Electron 53(3):736–744

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Lionet M, Prades R, Floch YL et al (2007) Improving conducted EMI forecasting with accurate layout modeling. In: IEEE symposium on embedded EMC. IEEE Ran L, Gokani S (1998) Conducted electromagnetic emissions in induction motor drive systems. I. Time domain analysis and identification of dominant modes. IEEE Trans Power Electron 13(4):757–767 Reuter M, Friedl T, Tenbohlen S et al (2013) Emulation of conducted emissions of an automotive inverter for filter development in HV networks. In: IEEE International symposium on electromagnetic compatibility. IEEE, pp 236–241 Ruehli AE (1974) Equivalent circuit models for three-dimensional multiconductor systems. IEEE Trans Microw Theory Tech 22(3):216–221 Schanen JL, Clavel E, Roudet J (1996) Modeling of low inductive busbar connections[J]. IEEE Ind Appl Mag 2(5):39–43 Traub F, Hansen J, Ackermann W et al (2012) Generation of physical equivalent circuits using 3D simulations. In: IEEE International symposium on electromagnetic compatibility. IEEE, pp 486–491 Skibinski G, Pankau J, Sladky R et al (1997) Generation, control and regulation of EMI from AC drives. In: Conference Record of the 1997 IEEE industry applications conference thirty-second IAS annual meeting. IEEE, 1997, pp 1571–1583 Sun J, Xing L (2013) Parameterization of three-phase electric machine models for EMI simulation. IEEE Trans Power Electron 29(1):36–41 Wittig T, Schuhmann R, Weiland T (2006) Model order reduction for large systems in computational electromagnetics. Linear Algebra Appl 415(2–3):499–530 Xing K, Lee F C, Boroyevich D (1998) Extraction of parasitics within wire-bond IGBT modules. In: Applied power electronics conference & exposition. IEEE, pp 487–503 Yang L, Odendaal WGH (2007) Measurement-based method to characterize parasitic parameters of the integrated power electronics modules. IEEE Trans Power Electron 22(1):54–62 Zhao L (2014) Co-simulation based on Simplorer field-circuit coupling multi-physics domain. China Water & Power Press, Beijing Zhong E, Lipo TA (1995) Improvements in EMC performance of inverter-fed motor drives. IEEE Trans Ind Appl 31(6):1247–1256 Zhu H, Hefner ARJ, Lai JS (1999) Characterization of power electronics system interconnect parasitics using time domain reflectometry. IEEE Trans Power Electron 14(4):1937–1943

Chapter 4

EMI Prediction and Suppression of DC-DC Converter

4.1 Overview At present, in the field of EVs, most of EMI research is carried out for the DC-DC device on low-voltage power PCB board. However, the HV/LV DC-DC converters of electrically driven vehicles are different from the LV DC-DC modules of PCB boards, there are the following differences between them: First, the input DC voltage range of the DC-DC converter is 200–900 V, and the output voltage is DC12V or 24 V. However, the input voltage range of the DC-DC device on the PCB is DC12V–24 V. Secondly, the DC-DC converter supplies power to all the low voltage components of the EV, so it has the characteristics of high input voltage, high output current and high power. Third, EMI from the HV/LV DC-DC converter may affect other high-voltage components by high-voltage power lines. Fourth, the electrical load of the HV/LV DC-DC converter changes in real time, and the switching technology uses the closed loop control to realize real-time regulation of the output voltage in (Yang et al. 2015; Laour et al. 2017). Finally, the HV/LV DC-DC converter has more and longer low voltage output power supply harness, so the conducted emission is easy to form a radiation emission. Therefore, it is very important to predict and suppress conducted EMI of the DC-DC converters. At present, the classical industrial filter design method is usually used to suppress the conduction EMI of DC-DC converter in electrically driven vehicles (Zhai et al. 2018). There is a lack of design theory and method filter for high voltage system of EVs in the frequency band of 150 kHz–108 MHz. Generally, an EMI filter is added to the high-voltage input port of the DC-DC converter, and several tests are required to optimize the filter topology. This kind of filter can only be installed in the late stage of product design, has a high cost, long design cycle, and it is not easy to realize engineering. In view of the existing problems, this chapter focuses on the followings: (1) A high-frequency equivalent circuit model of the zero voltage switching (ZVS) DC-DC converter, considering the parasitic parameters of the power semiconductor is established to predict conducted EMI. The model can also simulate © China Machine Press 2021 L. Zhai, Electromagnetic Compatibility of Electric Vehicle, Key Technologies on New Energy Vehicles, https://doi.org/10.1007/978-981-33-6165-2_4

193

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4 EMI Prediction and Suppression of DC-DC Converter

the actual EMI source and load impedance, and provide a model platform for the conducted emission prediction. (2) Using the established high-frequency equivalent circuit model, the transfer functions of CM and DM interference at key frequency points are established, and conducted EMI and radiated EMI of high-voltage power lines is predicted, and the main component parameters that affect the formation of EMI are determined. (3) According to the interference path and the main components that affect EMC, aiming at the three typical interferences generated by HV/LV DC-DC converter, switching frequency harmonics, low frequency DM and CM interference and high frequency CM interference, an effective high-voltage port wide-band conducted interference suppression method is proposed to reduce the electromagnetic emission in 150 kHz–108 MHz, to meet the standard limit requirements. (4) Based on the harmonic and resonance peaks exceeding-standard limits, a PCBlevel filter circuit design method is proposed, which can be realized inside the controller. It is characterized by small size, low cost and high efficiency, which can be realized in different development stages of products.

4.2 Conducted Emission Test of DC-DC Converter 4.2.1 Isolated Full-Bridge HV/LV Voltage DC-DC Converter Structure A HV/LV DC-DC converter for commercial EV usually uses isolated full bridge topology. As shown in Fig. 4.1, the HV/LV DC-DC converts DC high voltage of power batteries into DC low voltage to supply power for on-board low voltage electrical components. When the HV/LV DC-DC converter is controlled by PWM mode, the MOSFET operates in the hard switching mode, which causes a large switching

Q1

+ Vin -

D1

C1

Q2

D2

C2

+ D5

Lr 400-900V

ip

CDC Q3

Lf

D3

C3

Q4

D4

C4

Fig. 4.1 Topology of ZVS isolated full-bridge DC-DC converter

Cout

RL

Vo

D6

4.2 Conducted Emission Test of DC-DC Converter

195

loss. The phase-shifted PWM control mode utilizes the resonance between transformer leakage inductance and MOSFET junction capacitance to realize zero voltage switching (ZVS) of the MOSFET, which is beneficial to improve efficiency and switching frequency, while reducing EMI caused by the switching-on and-off of the MOSFET. The input voltage range of the DC-DC converter is DC 400–900 V, the rated output voltage is DC 27 V, the maximum output power can reach 3.6 kW, the maximum input current can reach 10A, and the maximum output current can reach 119A. Four MOSFETs with 100 kHz switching frequency are used in the isolated full bridge topology of the DC-DC converter, as shown in Fig. 4.1. Where, C DC is the filter capacitor on the high-voltage input side, and D1 –D4 are the anti-parallel diodes of the MOSFETs Q1 –Q4 , C 1 –C 4 is the resonant capacitor corresponding to each MOSFET, L r is the resonant inductor, L p is the leakage inductance of the transformer, L f and C out are the filtering inductor and filtering capacitor on the low voltage side respectively, RL is the equivalent load resistance on the low voltage side, and D5 and D6 are rectifier diodes of the secondary side of transformer T r .

4.2.2 Test Setup for the Conducted EMI According to CISPR25-2016 (GB/T 18,655–2018), the test arrangement of the conducted disturbances-voltage method for the HV/LV DC-DC converter is shown in Fig. 4.2. It mainly consists of a high voltage DC power supply, two line impedance stabilization networks (LISNs), the high voltage DC power supply lines with a standard length of 1.5 m, the HV/LV DC-DC converter, two low voltage power supply lines with a standard length of 1 m, a low voltage resistive load and a EMI receiver. Under the normal operation of the HV/LV DC-DC converter, the EMI receiver can measure the conducted voltage of the high-voltage positive bus and the negative bus in the frequency bands of 150 kHz–108 MHz by the LISNs. The conducted voltage

Fig. 4.2 HV/LV DC-DC converter conducted EMI test layout

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4 EMI Prediction and Suppression of DC-DC Converter

Fig. 4.3 Conducted disturbances-voltage method test platform

should meet the limit level defined by CISPR25-2016. Figure 4.3 shows the physical layout of the test platform. Figure 4.4 shows the experimental results of the conducted voltage in the frequency bands of 0.15–0.30 MHz, 0.53–1.8MH z, 5.9–6.2 MHz, 30–54 MHz, 48.5–72.5 MHz, 68–87 MHz and 76–108 MHz required by the standard CISPR252016. It can be found that the conducted voltages of the high voltage positive power supply line exceed the limits value of the standard average level 5. There are couple of obvious conducted voltage values exceeding the standard limits at 800, 900 kHz, 1.2, 1.3, 1.4, 1.5, 1.6, 1.7, 1.8, 5.9–6.2 MHz, and 72 MHz, etc. Therefore, in order to study the formation mechanism of EMI from the high voltage power supply line and to accurately predict the conducted disturbance voltage, it is necessary to establish

Fig. 4.4 Experimental results of the conducted voltage

4.2 Conducted Emission Test of DC-DC Converter

197

a high frequency circuit model to quantitatively analyze the relationship between conducted EMI source and the conducted voltage on high voltage positive power supply line to determine dominated parameters responsible for the conducted EMI.

4.3 EMI Prediction of DC-DC Converter 4.3.1 EMI Source SiHG17N80E MOSFET for automobile is used in the HV/LV DC-DC converter, and its switching frequency is 100 kHz. The main technical parameters of the MOSFET are shown in Table 4.1. As shown in Fig. 4.5a, a equivalent circuit model of MOSFET considering parasitic parameters was established. Where V D is a parasitic diode; the three junction capacitors of the MOSFET are respectively the inter-electrode capacitance C GS between the gate and the source, the inter-electrode capacitance C GD between the gate and the drain, and the inter-electrode capacitance C DS between the source and the drain;and C iss = C GS + C GD , C oss = C DS + C GD , C rss = C GD ; L S and L D are the lead inductors of the source and drain; RS and RD are the resistances of the corresponding leads.The voltage waveform obtained between the drain D’ and the source S’ of MOSFET in ideal case and considering the parasitic parameters are respectively shown in Fig. 4.5b, c. Figure 4.6 shows the spectrum of the voltage waveform V DS in Fig. 4.5b, from which the effect of turn-on time t on and rise time t r of MOSFET, and ringing at the rising edge on V DS spectrum can be seen. It can be seen from Fig. 4.6 that the longer the t on is, the lower the first corner frequency is, and the higher the amplitude of V DS in the low frequency band is. While the shorter the t r is, and the higher the second corner frequency is, the higher the amplitude of V DS in the high frequency band is. The ringing at the rising edge of the pulse of V DS causes a resonance point in the spectrum of V DS around 11 MHz, and then the amplitude of V DS decreases as the frequency increases. Due to the fast switching of the MOSFET in the HV/LV DC-DC converter, the amplitude of V DS is high in the wider frequency Table. 4.1 The main technical parameters of MOSFET (SiHG17N80E) Parameters

Typical value

Unit

Condition

V DS

850

V

T J = 25 °C

RDS(on)

0.25



T J = 25 °C

C iss

2408

pF

V GS = 0 V, V DS = 100 V, f = 1 MHz

C oss

81

pF

V GS = 0 V, V DS = 100 V, f = 1 MHz

C rss

9

pF

V GS = 0 V, V DS = 100 V, f = 1 MHz

tr

24

ns

tf

26

ns

V DD = 480 V, I D = 8.5 A, V GS = 10 V, Rg = 9.1 

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4 EMI Prediction and Suppression of DC-DC Converter D RD

LD

CGD

D G

G CGS

VD

CDS

S LS RS S

a

Equivalent circuit model of MOSFET

VDS(V)

VDS(V)

t(μs)

b

Ideal voltage waveform of VDS

t(μs)

c

Actual voltage waveform of VDS

Fig. 4.5 MOSFET equivalent circuit and drain-source voltage VDS

range of 150 kHz–108 MHz. So the disturbance current is easily formed by parasitic parameters of the HV/LV DC-DC system, and finally causes conducted EMI on the high voltage power supply cables. For bench test conditions, the switching frequency, t on and t r of the MOSFET are usually constant, so the conducted EMI voltage is mainly related to the disturbance current propagation paths.

4.3.2 The Conducted Emission High-Frequency Equivalent Circuit of DC-DC Converter System By analyzing the spectrum of the drain-source voltage of the MOSFET in Fig. 4.6, it can be found that the interference source signal is distributed in the 150 kHz– 108 MHz frequency band, so the influence of the parasitic parameters of the DC-DC converter system circuit must be considered. As shown in Fig. 4.7, the high frequency equivalent circuit model of the HV/LV DC-DC converter system is established. Where the parasitic parameters of Q1 and Q4 are shown in Table 4.1. Other major high frequency parasitic parameters are as follows: L s and Rs , the high frequency parasitic parameters of C DC ; C p1 and C p2 , the

4.3 EMI Prediction of DC-DC Converter

199

(a) Influence of on time ton

(b) Influence of rise time tr

(c) Influence of ring. Fig. 4.6 Spectrum of MOSFET drain-source voltage VDS

distributed capacitance between the chassis and the front arm and rear arm; C ps1 and C ps2 , the distributed capacitance of the transformer; C L ,the distributed capacitance between the load and the chassis; L p1 and L p2 , the equivalent inductance of the highvoltage DC power lines. The values of the main parasitic parameters of the system are shown in Table 4.2, obtained by measurement or theoretical calculation (Bondarenko et al. 2015; Grobler1 et al. 2017; Wang et al. 2013).

200

4 EMI Prediction and Suppression of DC-DC Converter

Fig. 4.7 High-frequency equivalent circuit model

Table. 4.2 The main parasitic parameters of the system Parameters

Description

Value

C DC

HV side filter capacitor

0.4 μF

C Q1 ,C Q2 ,C Q3 ,C Q4

Resonant capacitors

8 nF

Lr

Resonant inductor

30 μH

Lf

LV output filter inductor

50 μH

RL

Load resistance

0.48 

C out

LV output filter capacitor

200 μF

R1 , R2

Standard resistances of LISN

50 

L1 , L2

Standard inductors of LISN

50 μH

C1, C2

Input side capacitors of LISN

5 μF

C3, C4

Output side capacitors of LISN

470 nF

C p1 , C p2

Bridge arm midpoint to chassis distribution capacitors

26 pF

C ps1 , C ps2

Transformer parasitic capacitance

60 pF

CL

Load-to-chassis distributed capacitance

80 pF

Ls

HV side filter capacitor parasitic inductance

0.4 μH

Rs

HV side filter capacitor parasitic resistance

0.1 

L p1 , L p2

HV DC cable inductance

10 nH

4.3.3 EMI Coupling Path It can be seen from Fig. 4.4 that the conducted voltage of the high-voltage positive power supply line exceeding the standard limits mainly distributes in three frequency bands: ➀ Region 1 (0.53–1.8 MHz): In the frequency bands 0.15–0.30 MHz, 0.53– 1.8 MHz, the peaks of the conducted voltage are distributed at intervals of 100 kHz,

4.3 EMI Prediction of DC-DC Converter

201

such as 200, 300, ….. 1.8 MHz. This is due to the harmonics generated by the trapezoidal pulse trains of the MOSFETs operating at a switching frequency of 100 kHz. Theoretically, the harmonic amplitude at 200 kHz is the largest, and the amplitude of the harmonics decreases as the frequency increases, and the conducted voltage in the 0.53–1.8 MHz band would not exceed the standard limits. However, at 800 kHz, 900 kHz, 1.2, 1.3, 1.4, 1.5, 1.6, 1.7, 1.8 MHz, the harmonic voltage amplitude increases and exceeds the limits. Due to the existence of parasitic parameters of the system, the resonance voltage is generated at 2 MHz, resulting in the increase of the harmonic peak voltage at the above-mentioned resonance generated at 2 MHz. Thus, it is important to analyze frequency. Therefore, the conducted voltage exceeding the standard limits in this region is related to the harmonics of the switching frequency 100 kHz of the MOSFET and the resonance generated at 2 MHz. Thus, it is important to analyze the common mode (CM) interference paths and differential mode (DM) interference paths of the disturbance current at two frequencies of 200 kHz and 2 MHz, and determine the circuit parameters responsible for the conducted voltage exceeding the standard limits. ➁ Region 2: In the frequency band 5.9–6.2 MHz, the conducted voltage exceeds the limits by 6 dB, mainly due to the harmonics generated by the switching frequency 100 kHz of the MOSFET and the resonance at 2 MHz. ➂ Region 3: The conducted voltage exceeded the standard limits at 72 MHz, mainly due to the resonance caused by the low frequency harmonics and resonance of the system under the influence of high frequency parasitic parameters. In summary, in order to reduce the conducted EMI of the high-voltage positive power line to meet the standard limits, it is only necessary to quantitatively analyze the factors responsible for the conducted EMI at two typical frequency points of 200 kHz and 2 MHz. Since the exceeding standard limits around 72 MHz are mainly caused by high frequency parasitic parameters, it is difficult to perform accurate quantitative analysis. The DC-DC converter operates with four switching modes:(1,0), (0,1), (0,0) and (1,1). Among them,the EMI source and the EMI propagation path formed under the two working modes (1,0) and (0,1) are dual. Therefore, the switch of MOSFET can be equivalent to a CM EMI source and a DM EMI source on each bridge arm. Since the disturbance current paths generated by the two bridge arm DM EMI sources are completely identical, the actual DM EMI voltage is twice the voltage U R1 of the resistance R1 measured by LISN. However, the disturbance current paths generated by the two CM EMI sources are symmetrically complementary. In order to simplify the analysis, the (1,0) mode is taken as an example to analyze the propagation paths of CM and DM disturbance currents.

202

4 EMI Prediction and Suppression of DC-DC Converter L1 j62.88 C1

R1

-j0.16

50

+ UR1

-

C3

IDM1 -j1.69

+ Vin -

IDM2

C4 -j1.69 C2 -j0.16

R2 50

LP1 j0.013 Q1 CDC

DQ1

CQ1

LS

RS

Q2

-j99.5

DQ2

CQ2

-j13326

IDM Q3

0.1

D1

IDM3 LR -j99.5 I

LP j16.34

DQ3

Q4

DQ4

Cout

+ RL V o -

D2

DM5

CQ3

Lf

CPS1

j31.42

-j1.99 j0.5

IDM4

CQ4 CPS2 -j13326

L2 j62.88

LP2 j0.013

Fig. 4.8 DM EMI current path at 200 kHz

(1) DM EMI analysis at 200 kHz The impedance values of the equivalent circuit parameters at 200 kHz are calculated, as shown in Fig. 4.8. The EMI source caused by the switching of the MOSFET is equivalent to a constant current source I DM. Then the following five DM current paths are formed in the high frequency circuit of the system.     C2 → L 2 L 1 → C1 → → L p2 ➀ DM current path 1: IDM1 → L pl → R1 + C 1 C 4 → R2 ➁ DM current path 2: I DM2 → L p1 → C DC → L s → Rs → L p2 ➂ DM current path 3: I DM3 → L R → L P ➃ DM current path 4: I DM4 → C Q2 ➄ DM current path 5: I DM5 → C Q3 Among them, only the DM current I DM1 of path 1 flows through R1 to form the DM conducted voltage U R1 of the positive high voltage power line. In fact, two switching modes are considered simultaneously, U DM = 2U R1 = 2*50*I DM1 . The relationship between U DM and I DM of the DC-DC converter is analyzed below. Suppose Z 1 = jω(L P1 + L P2 ) 1 + jωL s + Rs ωCDG     1 1 // L 1 − j Z 3 = 2 R1 − j ωC3 ωC1 Z2 = − j

(4.1) (4.2) (4.3)

1  Z4 = − j  ω C Q3 + C Q4

(4.4)

Z 5 = jω(L R + L P )

(4.5)

4.3 EMI Prediction of DC-DC Converter

203

Z C3 = − j

1 ωC3

(4.6)

Z = (Z 1 + Z 2 //Z 3 )//Z 4 //Z 5

(4.7)

UDM = 2UR1 R1 Z 2 //Z 3 1 × Z 1 + Z 2 //Z 3 2 Z C3 + R1 R1 Z 2 //Z 3 1 (Z 1 + Z 2 //Z 3) × Z 4 × Z 5 = 2 × IDM × × × Z 1 + Z 2 //Z 3 2 Z C3 + R1 (Z 1 + Z 2 //Z 3) + Z 4 + Z 5 IDM R1 Z 4 × Z 5 × Z 2 //Z 3 = × Z C3 + R1 (Z 1 + Z 2 //Z 3) + Z 4 + Z 5 IDM R1 Z 3 Z 4 Z 5 1   = × (4.8) Z C3 + R1 (Z 1 + Z 4 + Z 5 ) 1 + ZZ 23 + Z3 = 2 × IDM × (Z 1 + Z 2 //Z 3)//Z 4 //Z 5 ×

It can be seen from (4.8) that both of U DM and I DM1 are related to multiple parameters. The method of reducing U DM and I DM1 is to increase the branch impedance of the DM current I DM1 or reduce the impedance of the other four DM current paths. (2) CM EMI analysis at 200 kHz The CM EMI source caused by the switching of the MOSFET can be equivalent to a constant voltage source U CM . The following four CM current propagation paths are formed in the high frequency circuit as shown in Fig. 4.9. ➀ CM current I CM1 path 1: U CM1 → Cp1 → U CM1   C 3 → R1 → UCM1 ➁ CM current I CM2 path 2: UCM1 → L p1 → C1 → L 1 L1 j62.83 C1

-j0.16

+ Vin -

R1 50

+ UR1

C3

-j1.69

C4

-j1.69

C2 -j0.16

R2 50

L2 j62.83

LP1 j0.013 ICM2 Q1

CDC UCM1

DQ1

CQ1

Q2

DQ2

CQ2

-j99.5 ICM3

DQ3

LP2 j0.013

Fig. 4.9 CM EMI current path at 200 kHz

CQ3

j16.34

ICM1

Q4

ICM4 L j62.83 f

D1

j31.42

Q3

CPS1

LR

LS RS

-j13326

DQ4

CP1 -j30622

Cout

+ RL Vo -

D2

CQ4 CPS2 CP2

CL

-j9952

204

4 EMI Prediction and Suppression of DC-DC Converter

 C2 → L 2 → UCM1 ➂ CM current I CM3 path 3: UCM1 → CQ3 → L p2 → C 4 → R2   Cout → CL → ➃ CM current I CM4 path 4: UCM1 → L r → Cpsl → L f → Rt UCM1 

Among them, only the CM current I CM2 of the path 2 flows through R1 . In fact, considering the two switching modes at the same time, the CM conducted voltage U CM of the positive high voltage power line is formed by the composed of the conducted voltage generated by the CM EMI sources of the two bridge arms, so they need to be considered simultaneously during calculation. The current path of the conducted voltage U R1 at R1 generated by the CM EMI source of the other bridge arm is:   C 3 → R1 → UCM2 . UCM2 → CQ2 → L pl → C1 → L 1 It can be seen from (4.9) that both of U CM and U CM1 are related to the parameters L p1 , C 3 , R1 , C 1 , L 1 , C Q2 in path 2. The method of reducing the value of U CM and I CM1 is to increase the impedance of CM current I CM2 path 2.  UCM = URI + UR1



   1 1 R1 + jωC // jωL 1 + jωC R1 3 1    = UC M1 ×  × 1 1 1 R1 + jωC R1 + jωC // jωL 1 + jωC + jωL P1 3 3 1     ⎞ 1 1 R1 + jωC // jωL 1 + jωC R1 3 1 ⎠.    + UC M2 ×  × 1 1 1 1 R1 + jωC R1 + jωC // jωL 1 + jωC + jωL P1 + jωC 3 3

1

Q2

(4.9) (3) DM EMI analysis at 2 MHz The impedance values of the equivalent circuit parameters at 2 MHz are calculated, as shown in Fig. 4.10. The EMI source caused by the MOSFET switching is equivalent to a current source I DM . The following four DM current propagation paths are formed in the high frequency circuit. ➀ DM current path 1: I DM1 → L p1 → R1 → C 3 → C 4 → R2 → L p2 ➁ DM current path 2: I DM2 → L p1 → C DC → L s → Rs → L p2 ➂ DM current path 3: I DM3 → C Q2 ➃ DM current path 4: I DM4 → C Q3 Among them, only the DM current I DM1 of the path 1 flows through R1 to form U R1 , and the relationship between U DM and the I DM of the DC-DC converter is analyzed below.

4.3 EMI Prediction of DC-DC Converter L1 j628.8 IDM1

C1 -j0.016

+ Vin -

R1

50

+ UR1

-

C3

-j0.169 I

DM2

-j0.169

R2

-j0.016

LP1 j0.13 Q1 CDC

DQ1

CQ1

50

L2 j628.8

LS

IDM4

RS

-j10 IDM Q3 CQ3 DQ3

j5

0.1

-j10

IDM3

Q2

DQ2

CQ2

D1

LR Q4

Lf

CPS1

j314.2

-j0.20

C4

C2

205

j163.4

DQ4

Cout

+ RL V o -

D2

CQ4 CPS2

LP2 j0.13

Fig. 4.10 DM EMI current path at 2 MHz

Suppose UDM = 2UR1 R1 Z 2 //Z 3 1 × Z 1 + Z 2 //Z 3 2 Z C3 + R1 R1 Z 2 //Z 3 1 (Z 1 + Z 2 //Z 3) × Z4 = 2 × IDM × × × Z 1 + Z 2 //Z 3 2 Z C3 + R1 (Z 1 + Z 2 //Z 3) + Z4 IDM R1 Z4 × Z 2 //Z 3 = × . (4.10) Z C3 + R1 (Z 1 + Z 2 //Z 3) + Z4

= 2 × IDM × (Z 1 + Z 2 //Z 3)//Z4 ×

In (4.10), since R1 ,Z 3 , and Z C3 are fixed values and the DM EMI source I DM is a constant current source, so the main variables determining U R1 are Z 1 and Z 2 , and increasing the amplitude of Z 1 and decreasing Z 2 can effectively reduce U R1 . Since Z1 + Z2 //Z3 + Z4 = 0.3267 + 5j + (-5j) = 0.3267, it can be seen that Ls, C DC , C Q3 , C Q4 produce parallel resonance, and the imaginary part of Z1 + Z2 //Z3 + Z4 is 0, so U R1 appears a peak near 2 M. In fact, there are two different ways to reduce U DM by changing the amplitudes of Z 1 and Z 2 . Changing the amplitude of Z 1 is to reduce the impedance value of the path1 in the circuit, that is, the value of the real part of Z 1 + Z 2 //Z 3 + Z 4 , so that the peak value of U DM at the resonance point is decreased, thereby lowering U DM . However, since L s and C DC in Z 2 are the most important components that resonate with C Q3 and C Q4 in Z 3 , changing the amplitude of Z 2 is essentially shifting the resonance point from 2 MHz to a high-frequency point where there is no standard limit requirement, thereby reducing the amplitude of U DM at the 2 MHz.

206

4 EMI Prediction and Suppression of DC-DC Converter LP1 j0.13

L1 j628 R1

C1

-j0.016

50

-

C3

+ Vin -

-j0.17

C4 -j0.17

R2

C2

Q1

+ UR1

CDC

L2 j628

CQ1

UCM1

RS

Q2

DQ2

-j1332

CQ3

Q4

DQ4

Cout

-j3062

+ RL V o -

D2

CQ4

CP1

LP2 j0.13

Lf

j163.4

-j10

DQ3

j628

CPS1

D1

j314.2

Q3

CQ2

LR

LS

50

-j0.016

DQ1

CP2

CPS2

-j3062

CL -j996

Fig. 4.11 CM EMI current path at 2 MHz

(4) CM EMI analysis at 2 MHz The CM EMI source caused by the MOSFET switching is equivalent to a voltage source U CM1 . The following three CM current paths are formed in the high frequency circuit as shown in Fig. 4.11. ➀ CM current I CM1 path 1: U CM1 → C p1 → U CM1 ➁ CM current I CM2 path 2: U CM1 → L p1 → R1 → C 3 → U CM1 ➂ CM current I CM3 path 3: U CM1 → C Q3 → L p2 → R2 → C 4 → U CM1 Among them, only the CM current I CM2 of path 2 flows through R1 . Similarly, The current path through R1 to form the conducted voltage U R1  generated by the CM EMI source of the other bridge arm is as follow: U CM2 → C Q2 → L p1 → R1 → C 3 → U CM2 . So we can obtain:  UCM = UR1 + UR1

= UCM1 ×

R1 +

R1 + UCM2 × + jωL P1 R1 +

1 jωC3

1 jωC3

R1 + jωL P1 +

1 jωC Q2

(4.11) It can be seen from (4.11) that U CM is related to the parameters L p1 , C 3 , R1 , and C Q2 in path 2. The method of reducing U CM and I CM1 is to increase the impedance of I CM2 in path 2.

4.3 EMI Prediction of DC-DC Converter

207

4.3.4 Conducted EMI Prediction Through Simulation According to the Fig. 4.7, the high-frequency circuit model of the HV/LV DC-DC converter system is established by MATLAB/Simulink, as shown in Fig. 4.12a, to obtain the conducted voltage. The simulation and experimental results are compared, as shown in 4.12(b). Similar to the experimental results, the simulation results of the conducted voltage appear couple of points exceeding-standard limits in band 1 (0.53– 1.8 MHz). In addition, a resonance is also generated near 2 MHz, resulting in a high resonant peak voltage. Therefore, the model can be used to design a conducted EMI filter.

4.4 EMI Filter of High Voltage Power Supply Two methods of EMI filter design are described in this section.

(a) High-frequency circuit model of HV/LV DCDC converter

(b) Comparison of simulation and experimental results of the conducted voltage Fig. 4.12 Modeling and simulation of the DC-DC converters for EMI analysis

208

4 EMI Prediction and Suppression of DC-DC Converter

Method 1: the filter design of high-voltage input port of the DC-DC converter is a filtering method that is adopted in the case of unclear knowing the EMI source and propagation path. This method is suitable for EMC rectification in the later stage of product design. However, the research of filters for industrial applications in the early stage is mainly aimed at frequencies below 30 MHz, while the research on filters for EMI suppression in a wide frequency range of 150 kHz to 108 MHz for automobiles are relatively few. Method 2: According to the experimental results obtained by the measurement of the conducted voltage based on CISPR25-2016, the CM and DM current propagation path, and mainly the dominated element parameters responsible for EMI noise at the two key frequencies of 200 kHz and 2 MHz are analyzed. The filter circuit is designed inside the DC-DC converter.

4.4.1 Design of High Voltage Input Port EMI Filter (1) Selection of EMI filter topology The source impedance and load impedance should be considered in the topology design of the filter. The EMI filter is installed between the high-voltage power battery power cable and the high-voltage input port of the on-board HV/LV voltage DC-DC converter. The input of the filter is connected to the high voltage DC power supply port of the DC-DC converter, where the high voltage port has been equipped with a large capacitor for absorbing voltage ripple, its impedance is low. The output of the filter is connected to the high-voltage power battery, which shows capacitance and low impedance. Since the source impedance and the load impedance of the filter are both low impedance, a T-type circuit was selected as the basic circuit of the DM EMI filter. Based on the T-type circuit, a CM EMI filter is designed by adding a CM choke coil and Y capacitors. (2) Determination of filter series and corner frequency The series n of filters and the corner frequency f0 of each stage are primarily related to the frequency range and insertion loss requirements. The filter is required to suppress EMI in the full frequency range of 150 kHz–108 MHz, and requires an insertion loss of 60 dB. In general, the slope of insertion loss curve of a single-stage filter will basically follow 20n dB/10dec between the corner frequency and the 10 times the corner frequency. However, the insertion loss curve will gradually deviate from the theoretical curve between 10–100 times the corner frequency, but it can still maintain 20n dB insertion loss. After 100 times the frequency, the suppression effect will gradually deteriorate until there is no filtering effect. Due to the wide frequency range of 150 k-108 MHz, single-stage filters cannot meet the requirements of insertion loss 60 dB. Although the insertion loss will be improved as the number of filter series increases, it will cause the filter volume to multiply, so the filter is

4.4 EMI Filter of High Voltage Power Supply Fig. 4.13 Filter topology

209

L1

L3

L2

L

N

L CY1

CY2

CY1

CY2

N L1 L2 L3 L1=L2=L3=1.3mH,CY1=0.1uF,CY2=3300pF

initially designed as a two stages LC filter, as shown in Fig. 4.13.The CM filter circuit consists of three same CM chokes with inductance L 1 , L 2 , L 3 respectively and two pairs of Y capacitors (2 Cy1 and 2 Cy2 ) to suppress the CM EMI current. The leakage inductance of the CM choke and the two connected Y capacitors are connected in series to form a DM filter circuit to suppress the DM EMI current. Since the basic circuit is a T-shaped circuit, only when the number of components is 3, the slope of the insertion loss curve below 10 times the corner frequency can satisfy 60 dB/10 decade. Therefore, the corner frequency of the first-stage LCL filter circuit needs to be lower than 15 kHz, so that the filter circuit can meet the 60 dB insertion loss requirement at 150 kHz, and finally the first corner frequency is selected to be 10 kHz. The second stage filter is designed to supplement the first stage filter when the suppression effect become worse after 100 times the first corner frequency. Therefore, the second corner frequency should be below 1 MHz, and the final determination is 400 kHz. As the frequency increases, the influence of the parasitic parameters of the filter elements will cause the filtering effect to decrease. After 10 MHz ie.10 times of the second frequency, the CM choke L 3 is used to suppress the high frequency EMI. (3) Determination of filter parameter Based on the corner frequency of each stage of the filter, the value of the Y capacitor was determined according to the requirement of the leakage current limit, and then the value of the inductor was calculated according to (4.12). The Y capacitor C Y1 at the first corner frequency f 1 = 10 kHz is 0.1 μF. Similarly, the Y capacitor C Y2 at the second corner frequency f 2 = 400 kHz is 3300 pF. f0 =

2π

√

1 L × 2 × CY

(4.12)

According to (4.11) and the values of C y1 and C y2 , the value L of L 1, L 2 ,L 3 was chosen to be 1.3mH. According to (4.13), the CM choke core is selected as Mn-Zn ferrite. Its magnetic permeability is low at low frequencies, but as the frequency increases, its magnetic permeability changes little, so it has excellent performance in high frequency. L=

kμ0 μr N 2 S l

(4.13)

210

4 EMI Prediction and Suppression of DC-DC Converter

where, L is the inductance of the coil, taken as 1.3mH, l is the length of the coil along the magnetic ring, taken as 30 mm, N is the number of turns of the coil, S is the cross-sectional area of the coil, taken as 60mm2 , and u0 is the air permeability, taken as 4π*10 −7 H/m, ur is the relative magnetic permeability of the core, taken as 5000, k is the long-haze coefficient, determined by the ratio of the radius of the winding to the length, here is 0.7.



N=

Ll = kμ0 μr S

1.3 × 10−3 × 30 × 10−3 = 12.12 0.7 × 5000 × 4π × 10−7 × 60 × 10−6

(4.14)

Based on (4.14), N = 12 can be obtained. The leakage inductance of a CM chock coil is 13 uH, which is 1% of the CM inductance. (4) Physical development of the filter Since the operating voltage of the high-voltage DC port filter is 400–1000 V. The withstand AC voltage of the general safety capacitor is 275 V, equivalent to a DC voltage of about 600 V. Therefore, each of C y1 and C y2 needs two capacitors in series. The actual 3D layout and prototype of the filter are shown in Fig. 4.14. (5) Insertion loss measurement of the filter According to Fig. 4.13 and the standard CISPR17-2011 for filter insertion loss measurement, the CM insertion loss and DM insertion loss of the filter are obtained by modeling and simulation by the software ADS, as shown in Fig. 4.15.From Fig. 4.15, it can be seen that the insertion loss of the filter with the source impedance and the load impedance of 50  meets the requirement of 60 dB in the full frequency range of 150 kHz–108 MHz. The actual insertion loss of the filter is obtained with a vector network analyzer, as shown in Fig. 4.16. The measurement layout of S 21-CM is shown in Fig. 4.16a, and the amplitude measurement results of S 21-CM and S 21-DM are shown in Fig. 4.16b, c. As can be seen from Fig. 4.16: the insertion loss ae-CM is greater than 60 dB in the 10 kHz–97 MHz band, and greater than 50 dB between 97 and 108 MHz. The

(a) Filter model Fig. 4.14 High-voltage port EMI filter

(b) Filter actual layout

4.4 EMI Filter of High Voltage Power Supply

211

(a) DM insertion loss

(b) CM insertion loss

Fig. 4.15 Filter insertion loss modeling and simulation results

insertion loss ae-DM is greater than 60 dB at 190 kHz–108 MHz, and greater than 58 dB between 150 and 190 kHz. In order to verify the actual suppression effect of the filter, the filter needs to be added into a DC-DC converter system for verification. (6) Insertion loss and test measurement of the filter in the system According toCISPR 25–2016, after adding the high-voltage port filter to the DC-DC converter, the modeling simulation and measurement of the conducted emission of the high-voltage system are performed. The simulation and test results are shown in Figs. 4.17 and 4.18. It can be seen that after the filter is added to the high-voltage port of the DC-DC converter, the conducted voltage drops significantly, meeting the requirements of the standard limits. The insertion loss at 2 MHz is greater than 60 dB.

4.4.2 Filter Circuit Design Based on Resonance Point Suppression (1) Filter circuit design on PCB board for DC-DC converter According to Fig. 4.4, it can be seen that the conducted voltage of the positive high voltage power supply line exceed the standard limits at 800, 900, 1.2, 1.3, 1.4, 1.5, 1.6, 1.7, 1.8, 5.9–6.2 MHz, which is mainly caused by the generated resonance at

212

4 EMI Prediction and Suppression of DC-DC Converter

(a) Layout drawing

(b) DC-DC converter filter module S21-CM parameter amplitude measurement results (9kHz-108MHz)

(c) DC-DC converter filter module S21-DM parameter amplitude measurement results (9kHz-108MHz)

Fig. 4.16 DC-DC converter filter module insertion loss measurement

4.4 EMI Filter of High Voltage Power Supply

213

Fig. 4.17 DC-DC high-voltage system conducted emission simulation model and conducted disturbance voltage simulation results

Fig. 4.18 DC-DC high-voltage system conducted emission test results

2 MHz. Therefore, eliminating the resonance at the 2 MHz or reducing the amplitude of the 2 MHz conducted current can effectively reduce the amplitude of the conducted voltage below limits. According to the dominated parameters of the DM and the CM EMI current propagation paths at 2 MHz, the filter circuit can be directly designed on the PCB board inside the DC-DC converter. There are two main methods, as described below: ➀ The equivalent parasitic inductance L s of the DC filter capacitor C DC is reduced, so that the resonance point is shifted from 2 MHz to the high frequency band around 10 MHz,because there are no specified limits around 10 MHz. In this paper, we choose to connect 1uF X capacitor C x in parallel with C DC , and its parasitic inductance L cx is 10nF, as shown in Figs. 4.18 and 4.19. In addition, since the capacitance C x is connected in parallel, the impedance value Z 2 of the path 2 is reduced, and the DM current through the branch of the path2 is increased, thereby reducing the DM current flowing on the positive high voltage power line and the conducted voltage measured on the LISN.

214

4 EMI Prediction and Suppression of DC-DC Converter

Fig. 4.19 Filter circuit to reduce the peak of the resonance point at 2 MHz

➁ According to (4.10), increasing the parasitic inductance L P1 and L P2 of the DC bus can increase the amplitude of Z 1 , thereby reducing the resonance peak of the conducted voltage at 2 MHz. In this paper, we added the DM inductors L x1 and L x2 to the positive and negative high voltage traces on the PCB of the DC-DC converter. The inductance value is finally set to1μH, as shown in Fig. 4.19. By this way, the values of L P1 and L P2 can be effective increased. For the conducted voltage exceeding limits near the high frequency of 72 MHz, a pair of Y capacitors C y0 can be added between the high voltage positive and negative bus traces and the ground on the PCB, and the capacitance value of C y0 is 3300pF. Designing the filter circuit on the PCB board inside the DC-DC converter has the following advantages: smaller size, only 1/5 of the high voltage filter (as shown in Fig. 4.20); it can be implemented in the early stage of product design; low cost; easy engineering; and eliminating the electromagnetic coupling between the input and output cables of the filter.

(a) Filter components

Fig. 4.20 Filter circuit on the PCB

(b) Comparison of two filter sizes

4.4 EMI Filter of High Voltage Power Supply

215

(2) Simulation of insertion loss in DC-DC converter system According to Fig. 4.19, the high-frequency circuit model of the DC-DC converter HV system with the filter circuit is established in MATLAB/Simulink to obtain the conducted EMI. The simulation results of the conducted voltage of the positive high-voltage power supply line are shown in Fig. 4.21. Figure 4.21a shows the simulation results after adding 1uF C x and 10 nH L cx . It can be seen that the resonance peak at the 2 MHz disappears and the resonance moves to 10 MHz with no requirement of the standard limit. It can also be found that the conducted voltage at 800 kHz, 1.0, 1.2, 1.4, 1.6, and 1.8 MHz decreases significantly, meeting the standard limits requirements. In addition, due to the shifting of the resonance point, the conducted voltage at 5.9–6.2 MHz is also reduced, which is complied with the limits requirement. Figure 4.21b shows the simulation results after adding L x1 and L x2 . It can be seen that the amplitude of the conducted voltage near the resonance point at 2 MHz is reduced to meet the standard limit requirement, as well as the conducted voltage at 5.9–6.2 MHz. However, there are still same points exceeding standard limits at 800 kHz, 1.0, 1.2, and 1.4 MHz. Figure 4.21c shows the simulation results after adding 1uF,10nH X capacitor in parallel, DM inductors L x1 and L x2 in series. It can be seen that the filter circuit has a good EMI suppression effect, and the conducted voltage between 150 and 15 MHz satisfies the standard requirements. Figure 4.21d shows the comparison of the three methods. Figure 4.21e shows the simulation results after adding C x and C DC in parallel, L x1 and L x2 in series, and a pair of C y0 . It can be seen that after adding the filter circuit on the PCB, the conducted voltage comply with the limit requirements in the 150 kHz–108 MHz frequency band. (3) Conducted emission test Figure 4.22 shows the measurement results of the conducted voltage after adding C x and C DC in parallel, L x1 and L x2 in series, and a pair of C y0 . It can be seen that the conducted voltage exceeding standard limits drops significantly and comply with the standard limit requirements. Comparing the measurement results of effective insertion loss of the filter circuit on the PCB with that of the high-voltage port filter, it can also be seen that although the insertion loss in some frequency bands is lower than that of the high-voltage port filter in some bands, it still meets the standard requirements, which can simplify the structure of filter circuit and reduce size and cost.

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4 EMI Prediction and Suppression of DC-DC Converter

(a) Adding 1uF and 10nH X capacitors in parallel

(b) Adding LX1 and LX2 in series

(c) Adding 1uF and 10nH X capacitors in parallel, and LX1 and LX2 in series. Fig. 4.21 Simulation results of conduction voltage after adding filter circuit on PCB in the DC-DC converter

4.4 EMI Filter of High Voltage Power Supply

217

(d) Comparison of the three methods.

(e) Simulation results for the full frequency band Fig. 4.21 (continued)

Fig. 4.22 Measurement results of the conducted voltage after adding a filter circuit on the PCB

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4 EMI Prediction and Suppression of DC-DC Converter

4.5 Radiated Emission 4.5.1 Theoretical Analysis The radiated emission test is essentially to test the radiated signal generated by the two equivalent antennas in the DC-DC system: One is the equivalent loop antenna, caused by the DM current I D in the loop, the electric field strength E at the distance D from the loop is: E = 1.3S ID f 2 /D

(4.15)

where S is the loop area and f is the frequency of the DM current. It can be seen from (4.15) that the electric field strength is related to the conducted DM current I D . The other is the equivalent monopole antenna or symmetric dipole antenna. The cable of the DC-DC converter can be equivalent to this antenna, which is caused by the CM current I cm on the cable bundle (the length of the cable is L), I cm is the focus of studying radiated emission, and the electric field intensity E C generated at distance D is E C = 6.28 × 10−7

I cm f L D

(4.16)

From (4.16), it can be seen that the electric field strength is related to the conducted CM current I cm . It can be seen from (4.15) and (4.16) that both DM current and CM current can cause radiated emissions, and controlled conducted emissions can control radiated emissions; in addition, it is easier to reduce conducted emissions than to reduce radiated emissions Because the conducted emission is only propagated by the DC-DC converter power line or signal line and control line path. Therefore, considering that the radiated emission below 108 MHz is mainly caused by the cable, the filter suppression technology of conducted interference is used to suppress the radiated emission.

4.5.2 Radiated Emission Measurement According to the radiated emission measurement methods of high-voltage parts and modules of vehicles specified in the test standards CISPR 25–2016, radiated disturbance is measured in the frequency range of 150 kHz–2.5 GHz, and the radiated emission measurement arrangement is shown in Fig. 4.23. The measurement photos are shown in Fig. 4.24, and the measurement results are shown in Fig. 4.25. The

4.5 Radiated Emission

219

HV LISN

50

HV LISN

50

LISN

50

LISN

50

DC-DC

Fig. 4.23 Radiation emission inspection layout

(a) Rod antenna

(d) Logical period antenna (horizontal polarization)

(b) Bi-cone antenna (horizontal polarization)

(e) Log period antenna (vertical polarization)

(c) Bi-cone antenna (vertical polarization)

(f) Horn antenna (horizontal polarization)

(g) Horn antenna (vertical polarization)

Fig. 4.24 DC-DC converter radiation emission inspection photos

radiated emissions after adding the filter all meet the requirements of standard limit level 3. It can be seen that the filter can suppress not only conducted EMI but also radiated EMI.

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4 EMI Prediction and Suppression of DC-DC Converter

Peak limit Average limit Peak curve Average curve

(a)150KHz-30MHz 1m monopole antenna radiated emission peak and average value

Peak limit Average limit Peak curve Average curve

(b)30MHz-200MHz double-cone antenna vertical polarization radiation emission peak value and average value

Fig. 4.25 Radiation emission test results

4.6 Summary (1) A high-frequency equivalent circuit model of the zero voltage switching (ZVS) DC-DC converter, considering the parasitic parameters of the power semiconductor is established to predict conducted EMI. The model can also simulate the actual EMI source and load impedance, and provide a model platform for the conducted emission prediction. (2) Using the established high-frequency equivalent circuit model, the transfer functions of CM and DM interference at key frequency points are established, and conducted EMI of high-voltage power lines is predicted, and the main component parameters that affect the formation of EMI are determined. (3) According to the interference path and the main components that affect EMC, aiming at the three typical interferences generated by HV/LV DC-DC

4.6 Summary

221

Peak limit Average limit Peak curve Average curve

(c) 200MHz-1GHz log period antenna vertical polarization radiation emission peak value and average value Peak limit Average limit Peak curve Average curve

(d) 1GHz-2.5GHz dual-ridge waveguide horn antenna vertical polarization radiation emission peak value and average value

Fig. 4.25 (continued)

converter, switching frequency harmonics, low frequency DM and CM interference and high frequency CM interference, an effective high-voltage port wide-band conducted interference suppression method is proposed to reduce the electromagnetic emission in 150 kHz–108 MHz, to meet the standard limit requirements. (4) Based on the harmonic and resonance peaks exceeding-standard limits, a PCBlevel filter circuit design method is proposed, which can be realized inside the controller. It is characterized by small size, low cost and high efficiency, which can be realized in different development stages of products.

222

4 EMI Prediction and Suppression of DC-DC Converter

References Bondarenko N, Zhai L, Xu B, et al (2015) A Measurement-Based Model of the Electromagnetic Emissions From a Power Inverter. IEEE Transactions on Power Electronics 30(10):5522–5531 Grobler1 I, Gitau M N (2017) Modelling and measurement of high frequency conducted electromagnetic interference in DC–DC converters IET Science, Measurement & Technology 11(4):495–503 Laour M, Tahmi R, Vollaire C (2017) Modeling and analysis of conducted and radiated emissions due to common mode current of a buck converter. IEEE Trans Electromagn Compat 59(4):1260–1267 Wang Q, An Z, Zheng Y, et al (2013) Parameter extraction of conducted electromagnetic interference prediction model and optimization design for a DC–DC converter system. IET Power Electronics 6(7):1449–1461 Yang G, Dubus P, Sadarnac D (2015) Double-phase high-effificiency, wide load range highvoltage/low-voltage LLC DC/DC converter for electric/hybrid vehicles. IEEE Trans Power Electron 30(4):1876–1886 Zhai L, Zhang T, Cao Y, et al (2018) Conducted EMI prediction and mitigation strategy based on transfer function for a high-low voltage DC–DC converter in electric vehicle. Energies 11(5): 1028–1044

Chapter 5

Wireless Charging System Electromagnetic Safety and Electromagnetic Compatibility

5.1 Overview Wireless charging is the development trend of electric vehicle charging technology in the future, and it will promote the development of intelligent networked electric vehicles and driverless vehicles. The efficiency and power of wireless charging systems have always been people’s concerns. In addition, the wireless charging system also needs to focus on two aspects: electromagnetic safety issues and electromagnetic compatibility issues. Technical standards and regulations for wireless charging of electric vehicles at home and abroad, such as SAE J254 and IEC 61980, define the limits of relevant leakage electromagnetic fields and electromagnetic disturbances. In order to improve the safety and electromagnetic compatibility of electric vehicles, the analysis and prediction of the electromagnetic field distribution of the coupling coil of the electric vehicle wireless charging system and the suppression of electromagnetic interference are very important. This chapter describes the magnetic field distribution of the circular coupling coil during alignment and offset, studies the power, efficiency, and electromagnetic field changes during the offset, and also describes the relationship between the power changes during the offset and the electromagnetic field distribution. In addition, according to the standard, a detailed electromagnetic field measurement method is proposed to measure the magnetic field in different areas of the vehicle. First, a bilateral LCC topology circular coupling coil wireless charging system model was established, and the power and efficiency of the coupling device during the offset, and the anti-offset characteristics of the bilateral LCC topology wireless charging device were analyzed; then, through modeling and measurement, the electromagnetic field distribution when the coupling coil is aligned and offset is described. Finally, it shows the modeling and suppression of conducted electromagnetic interference of the DC power line of the wireless charging system, harmonics and suppression methods of the public power line, and the electromagnetic radiation of the vehicle secondary side circuit of the wireless charging system.

© China Machine Press 2021 L. Zhai, Electromagnetic Compatibility of Electric Vehicle, Key Technologies on New Energy Vehicles, https://doi.org/10.1007/978-981-33-6165-2_5

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5 Wireless Charging System Electromagnetic Safety …

5.2 Wireless Charging System Structure 5.2.1 System Structure According to the SAE J2954 (Wireless Power Transfer for Light-Duty PlugIn/Electric Vehicles and Alignment Methodology) standard, the electric vehicle wireless charging. system is divided into four power levels, namely 3.7, 7.7, 11.1, and 22 kW. In this paper, the designs and analyzes the round coupling coil device and its compensation topology of the wireless charging system with the maximum output power of 3.7 KW, which is commonly used in family cars, are described. The system design parameters are shown in Table 5.1. The most classical and widely used system solution is adopted, as shown in Fig. 5.1. The input of the system is the AC power from the power grid, which is boosted by the rectifier and power factor correction (PFC) circuit to output high-voltage DC power, and then converts the high-voltage DC power to 85 kHz AC power by using an inverter. Subsequently, the energy is coupled to the secondary side through the primary and secondary LCC compensation circuits and the coupling coil device, and after rectification and filtering, the output power is used to charge the power battery. According to the state of charge of the vehicle power battery, the primary-side inverter is controlled by wireless communication to adjust the output DC voltage of the secondary-side. As shown in Fig. 5.1, a primary power module (PPM) is arranged on the ground and the charging device outside the vehicle. A secondary power module (SEM) is arranged on the vehicle chassis. Table 5.1 Design parameters of wireless charging system

Design parameter

Symbol

Value

Input grid voltage

U ac

220 V

Input grid voltage frequency

f1

50 Hz

PFC output DC voltage

U in

260–425 V

Output charging voltage

Ub

300–400 V

Working frequency

f

81.38–90 kHz

Maximum power

P

3.7 kW

Fig. 5.1 Schematic diagram of a typical wireless charging system for EV

5.2 Wireless Charging System Structure

225

5.2.2 System Model According to the Fig. 5.1, the LCC wireless charging system is designed. The circuit of the system consists of 8 modules, as shown in Fig. 5.2. The following is a brief description of each module: (a) (b) (c) (d) (e) (f)

Grid input: 220 V, 50 Hz AC; Rectification, and PFC: Full-bridge rectification structure, and boost PFC; Inverter module: Full-bridge inverter structure, power device using MOSFET; Primary topology: LCC compound compensation topology; Coupler: Circular coupling coil and spoke ferrite arrangement; Secondary topology: The same LCC compensation topology as the primary side; (g) Secondary rectification: Full-bridge rectification structure; (h) Secondary filtering: LC filtering structure. The design steps of the coupling device and LCC topology are mainly introduced (1) Structural design of coupling device Coupling device is the main structure of energy transmission in the whole wireless charging system. A reasonable and effective design can improve the efficiency of the system, improve the anti-offset characteristics of the system, and reduce the leakage of magnetic fields. The main structure of the coupling device includes a circular coupling coil and some ferrites. (a) Circular coupling coil The coupling coils can be divided into polar coils (PP) and non-polar coils (NPP) according to their polarities. Polar coil composed of more than two or more coils (including DDP, DDQP, and BP structures), which can generate parallel and vertical flux. Non-polar coils are a pair of coupled coils (including circular coils and rectangular coils), which can only produce vertical flux. Among them, the circular coupling coil is not only simple and reliable in structure, but also has the same anti-offset performance in all directions, so it is currently the most widely used structure in static wireless charging of EVs.

Fig. 5.2 Structure circuit of wireless charging system with bilateral LCC topology

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5 Wireless Charging System Electromagnetic Safety …

As shown in Fig. 5.3, the circular coupling coil has four main design parameters, which are the coil’s inner diameter Di , outer diameter DO , wire diameter DW , and coil turns N. The ratio between the transmission distance d of the coupling coil and the outer diameter DO of the coil needs to be controlled within a certain range Where, DO /d = 4 is used (Suh 2018). The design parameters of the circular coil are shown in Table 5.2. The coil is made of l litz wires twisted by multiple independently insulated enameled wires, which can effectively reduce the skin effect and proximity effect caused by the high-frequency excitation current of the coupling coil and reduce the equivalent series resistance (ESR) of the wire, so as to reduce the AC loss of the coupling coil. (b) Ferrite distribution Because the magnetic resistance of ferrite is much smaller than that of air, more magnetic field can be confined in the coupling area between the primary coil and the secondary coil, so that more magnetic induction lines pass through the secondary coil. According to the shape of the circular coupling coil, a spoke-type ferrite structure is selected. This ferrite structure can be conveniently spliced with rectangular ferrite monomer, which can improve the magnetic field coupling ability of the circular coil. As shown in Fig. 5.4, there are three layout schemes of the spoke-type ferrite structure, namely, a long ferrite arrangement, a short ferrite arrangement, and a long and short ferrite stagger arrangement. Considering the effect and weight cost of the Fig. 5.3 Top view and cross-section view of a circular coupling coil

Table 5.2 Design parameters of round coil

Project selection

Project selection

Winding selection

800 litz strands

Winding material

II (16AWG4 * 5X24/36)

Winding diameter

3.9 mm

Outer radius of coil

300 mm

Inner radius of coil

150 mm

Number of turns

16

5.2 Wireless Charging System Structure

(a) Long ferrite arrangement

227

(b) Short ferrite arrangement

(c) Long and short ferrite arrangement

Fig. 5.4 Ferrite structure layout

three layout schemes, the long and short ferrites are arranged alternately. The specific parameters are shown in Table 5.3. According to the design parameters of coils and ferrites in Tables 5.2 and 5.3, the three-dimensional model of the coils and ferrites and the assembly structure of the coupling device are established by using ANSYS Maxwell software, respectively, as shown in Fig. 5.5. The self-inductance and coupling coefficient of the coupled coil are obtained by simulation, as shown in Table 5.4. By applying an 85 Hz 10 A AC sinusoidal excitation signal to the coil, the magnetic field distribution of the coupling device can be obtained, as shown in Fig. 5.6. It can be seen that the magnetic field is distributed more tightly, the leakage magnetic field is reduced, and the coupling ability of the coupling device is enhanced by using a pair of coupling coils and distributed ferrites. Table 5.3 Ferrite design parameters

Project

Project selection 60 mm × 15 mm × 9 mm

Unit ferrite size Long ferrite size

240 mm × 15 mm × 9 mm

Short ferrite size

180 mm × 15 mm × 9 mm

Number of long ferrites

18

Number of short ferrites

18

Ferrite alignment angle

(a) Fig. 5.5 Circular coil and coupling device

10°

(b)

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5 Wireless Charging System Electromagnetic Safety …

Table 5.4 Parameters of compensation topology and coupling coil

Content

Device

Device name

Coupling Coefficient

k

0.3525

Self-inductance of transmitting coil

Lp

226.44 μH

Self-inductance of receiving coil

Ls

226.44 μH

(a) Magnetic field distribution

(b) Magnetic field vector diagram

Fig. 5.6 magnetic field distribution of the coupling device

(2) Bilateral LCC topology Based on the S-S series topology, a compensation inductance (L 1, L 2 ) and a compensation capacitor (C1, C2) are added to the two sides of the compensation topology to form a T-shaped network, as shown in Fig. 5.7. LCC topology and coil inductance work together to form two. resonant cavities in the circuit. The parameters of L p, C p, L 1, L p L1 , C p, L 1 in the bilateral LCC topology and the switching angular frequency ω of the MOSFET of the inverter ment the following relationship: ω= 

1 1 =√ L s Cs L p Cp

Fig. 5.7 Mutual inductance coupling circuit model of Bilateral LCC topology

(5.1)

5.2 Wireless Charging System Structure

229

Table 5.5 Compensation topology parameters Device name

Parameter

Design value

Device name

Parameter

Design value

Primary compensation inductance

L1

72.1 μH

Primary compensation capacitor 2

Cp

17.5 nF

Secondary compensation inductor

L2

72.1 μH

Secondary compensation capacitor 1

C2

48.9 nF

Primary compensation capacitor 1

C1

48.9 nF

Secondary compensation capacitor 2

Cs

20.7 nF

where, U in is the excitation source voltage, ω is the angular frequency of the excitation source, M is the mutual inductance between the two coils, L p is the transmit coil inductance, C p is the primary-side compensation capacitor, Rp is the transmit coil resistance, and I p is the primary-side loop current; L s is receiving coil inductance, C s is the secondary-side compensation capacitor, Rs is the receiving coil resistance, I s is the secondary-side loop current, and Ro is a purely resistive load. When calculating the parameters of the electrical components of the topology, the internal resistance of both sides is ignored. Assuming that the parameters of the primary and secondary compensation circuits are the same, the parameters of each component in the bilateral LCC topology can be determined according to the following formula (Esteban and Sid 2015):  L1 = L2 =

kU inR0 Lp ωP

(5.2)

where, P is the transmission power. The relationship of compensation capacitance can be obtained by (5.1) (Suh 2018): C1 = C2 = Cp = Cs =

1 ω2 L1 1 − L1)

ω2 (Lp

(5.3) (5.4)

According to the coupler power requirements and ZVS (Zero Voltage Switching) control in the system (Suh 2018), the LCC topology parameters are listed in Table 5.5.

5.2.3 System Performance Simulation In order to verify the feasibility and correctness of the design of bilateral LCC wireless charging system, according to Fig. 5.2, the simulation software ANSYS/Simplorer

230 Table 5.6 Circuit element parameters

5 Wireless Charging System Electromagnetic Safety … Name

Parameter or selection

Rectifier diode

VS-80APS12-M3

PFC boost inductor L PFC

500 uH

PFC capacitor C PFC

1500 uF/600 V

PFC diode

IDW40G65C5SKSA1

Inverter power switch MOSFET

IPW65R048CFDA

Inverter freewheeling diode

IDW10G120C5BFKSA7

Secondary side filter capacitor C 0

66 uF/650 V

Secondary side filter inductor L 0

158 uH

was used to model the wireless charging system. According to the design parameters in Table 5.5 and Table 5.6, the circuit model is established in ANSYS/Simplorer, and the coupler model established by ANSYS/Maxwell. Finally, the collaborative joint simulation is carried out, as shown in Fig. 5.8. Because the coupled device model uses a 3D physical prototype level model, not only the accuracy of the model is improved, but also the simulation results of the electrical performance of the system can be more realistic and reliable. The inverter module uses PWM control. The output voltage is a rectangular square wave, the voltage is 425 V, and the duty cycle is 0.46. The simulation results are shown in Fig. 5.9, which are consistent with the analysis results. It can be seen from Fig. 5.9 that after the system works stably, the current in the receiving coil is larger than that in the transmitting coil, and the current phase in the receiving coil advances the

Fig. 5.8 Co-simulation of system circuits based on Simplorer

Fig. 5.9 (a) Inverter output voltage; (b) primary secondary coil current

5.2 Wireless Charging System Structure

(a) Output voltage

231

(b) Output current

Fig. 5.10 Output voltage and output current curve of load resistor

(a) Time=3.3e-005s

(b) Time=3.4e-005s

(c) Time=4e-005s

Fig. 5.11 Dynamic magnetic field distribution of coupling device

current of the transmitting coil by 90 degrees, which conforms to the principle of energy coupling. It can be known from Fig. 5.10 that the system reaches a stable state after 4 ms, and outputs stable DC voltage and current, and output power of 3.7 kW, meeting the design requirements of the wireless charging system. The distribution of the magnetic field of the coupling device with the current phase is shown in Fig. 5.11.

5.3 Principle of the Wireless Charging Wireless charging technology uses high frequency alternating current of primary coil to generate high frequency alternating magnetic field., By magnetic field coupling, induction electromotive force is generated at the receiving end, which is used for the power supply of load, so as to complete the wireless power transmission. At present, there are two types of theoretical research methods for WPT technology: coupling mode theory and circuit coupling theory. The coupled mode theory studies the coupling coils on both sides as a unified whole, while the circuit coupling theory uses the circuit model to study the primary side and the secondary side separately. In this paper, the circuit coupling theory is used to study the influence of coil offset on the output power, efficiency and EMF by building a mutual inductance coupling model.

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5 Wireless Charging System Electromagnetic Safety …

5.3.1 Transmission Power The impedance analysis method is used to analyze the secondary side circuit of the bilateral LCC topology. The impedance conversion relationship is shown in Fig. 5.12. The open circuit voltage generated by the receiving coil is recorded as U oc , and the overall impedance from the left side of the secondary side, that is, the input terminal to the load terminal, is recorded as Z s , so the relationship of Z s can be obtained as Z s = (Ro + Z L2 )//Z C2 + Z Cs + Z LS + Rs

(5.5)

Taking  Z L2 = jωL√2 and capacitor Z C2 = 1/jωC 2 as well as the expression ω = 1/ L p Cp = 1/ L s L s into the above relationship, the expression of overall impedance can be obtained: Z s = Rs +

Ro (1 − ω2 L 2 C2 ) + Ro ω2 L 2 C2 (1 − ω2 L 2 C2 )ωL 2 j − ωC2 Ro2 j + (1 − ω2 L 2 C2 )2 + Ro2 ω2 C22 (1 − ω2 L 2 C2 )2 + Ro2 ω2 C22 (5.6)

Let the overall impedance of the secondary side be purely resistive, then the imaginary part is taken as 0, so L 2 and C 2 can be expressed as: (1 − ω2 L 2 C2 )L 2 = Ro2 C2

(5.7)

Let ω2 L 2 C2 = 1−1/λ2 , then λ ≥ 1, be brought into the real part of the impedance expression, and we get: Z s = λ2 R o + R s

(5.8)

From the derivation of the above formula, when the overall impedance of the secondary side is purely resistive, compared with the S-S compensation, the equivalent load of the system can be increased to λ2 times by using LCC compensation, and λ is determined by the secondary-side compensation capacitor C 2 and the compensation inductance L 2 (Fig. 5.13). Ls

Fig. 5.12 Secondary side impedance conversion diagram Uoc

Cs

L2

Is

Io C2

Zs

Rs

Ro

5.3 Principle of the Wireless Charging

233

Fig. 5.13 Overall equivalent circuit diagram of the system

Z in = [(Rp + Z r ) + Z Lp + Z Cp ]//Z C1 + Z L1

(5.9)

The reflected impedance Z r is expressed as: Zr =

ω2 M 2 R s + λ2 R o

(5.10)

Combined with (9), the input current of the system can be expressed as: • Iin

•

•

Uin Uin (Rr + Z r + Z C1 ) = = Z in (Rp + Z r )(Z C1 + Z L1 ) + Z L1 Z C1

(5.11)

where Z L1 = jωL 1 , and Z C1 = 1/jωC 1 . According to Kirchhoff’s current law, the expressions of the current I p in the coil and the load output current I o can be summarized as: • Ip

•

Uin = (Rp + Zr )(1−ω2 L 1 C1 ) + Z L1

(5.12)

•

ωM Uin 1 1 · 2 · I = 2 2 o (1 − ω L 1 C1 )(Rp + Z r ) + jωL 1 λ Ro + Rs (1 − ω L 2 C2 ) + j Ro ωC (5.13) •

According to ω2 L 2 C2 = 1 − 1/λ2 , we get •

•

I =

o

ωM Uin λ2 · √ 1 + j λ2 − 1 (1 − ω2 L 1 C1 )[Rp (λ2 Ro + Rs ) + ω2 M 2 ] + jωL 1 (λ2 Ro + Rs )

(5.14) The output power of the system of the bilateral LCC topology is thus obtained:

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5 Wireless Charging System Electromagnetic Safety …

PLCC-LCC =

ω2 M 2 Uin2 λ2 Ro (1 − ω2 L 1 C1 )[Rp (λ2 Ro + Rs ) + ω2 M 2 ]2 + ω2 L 21 (λ2 Ro + Rs )2 (5.15)

It can be seen from the expression (15) that the output power of the system is related to the parameter λ. If λ = 1, the system output power in the S-S topology can be obtained: PS-S =

ω2 M 2 Uin2 Ro [Rp (Ro + Rs ) + ω2 M 2 ]2

(5.16)

The power expression of the system of the two topologies can be plotted, as shown in Fig. 5.14. Using Fig. 5.8 for system simulation, we can get the curve of output power of LCC and S-S topology system with frequency, as shown in Fig. 5.15. The primary power curve is slightly larger than the secondary power curve. In the same case, the S-S topology has more high-power frequency points, while the bilateral LCC topology has a wider power frequency band.

Fig. 5.14 Change curve of system output power with frequency

(a) S-S

Fig. 5.15 Output power versus frequency

(b) Bilateral LCC

5.3 Principle of the Wireless Charging

235

5.3.2 Efficiency In order to reduce energy consumption, it is also necessary to ensure the efficiency of the wireless charging system during power transmission. Combined with the power expression obtained above, the system efficiency can be analyzed. If the system’s primary-side efficiency is ηp and the secondary-side efficiency is ηs , then ⎧ ⎪ ⎪ ⎨

ηp =

⎪ ⎪ ⎩ ηs =

Zr Rp + Z r

λ2 R o λ2 R o + R s

(5.17)

Due to η = ηp ηs , the relationship of the system efficiency of the bilateral LCC topology can be obtained according to the expression of the reflecting impedance: ηLCC-LCC =

ω 2 k 2 L p L s λ2 R o (λ2 Ro + Rs )[Rp (λ2 Ro + Rs ) + ω2 k 2 L p L s ]

(5.18)

By replacing λ2 Ro with Ro in the system efficiency relationship, that is, when λ is taken as 1, the efficiency expression of the system in the S-S topology can be obtained. ηS-S =

ω 2 k 2 L p L s Ro (Ro + Rs )[Rp (Ro + Rs ) + ω2 k 2 L p L s ]

(5.19)

The trend of efficiency curve of the two typologies in Fig. 5.16 is basically the same, however, the range of system efficiency above 80% of the bilateral LCC topology is larger than that of the S-S topology, which can make the system’s resonance frequency have a wider range.

Fig. 5.16 Efficiency Curves of two topologies with frequency

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5 Wireless Charging System Electromagnetic Safety …

(a) S-S

(b) Bilateral LCC

Fig. 5.17 System efficiency versus frequency

Using Simplorer software, the system efficiency curve with frequency is obtained by simulation, as shown in Fig. 5.17. From the efficiency curve, it can be seen that the efficiency of the two topologies is close to 1 at the operating frequency of 85 kHz. The resonance frequency band of the bilateral LCC topology is wider than that of the S-S topology, which can generate resonance at 50 kHz to 110 kHz, but its efficiency fluctuates to some extent.

5.3.3 Offset Characteristics The offset of coupling device in the actual situation usually includes multiple directions, including both lateral offset and longitudinal offset. In this paper, the coupling coil is of a circular symmetrical structure, and only the lateral offset will be used for illustration. When the coupler is shifted, the mutual inductance parameters of the coupling coil will change, resulting in changes in the transmission efficiency and magnetic field distribution of the coupling device. Therefore, it is necessary to analyze the offset characteristics of the coupler and understand the change law of the related parameters of the coupler. The mutual inductance of the coupling coil in a wireless charging system is defined in classic electromagnetics follows: M=

Nϕ I

(5.20)

where ϕ is the magnetic flux of the two coils, which is usually only a part of the total magnetic flux generated by the loosely coupled coils; N is the number of turns of the coil; I is the excitation current of the primary coil. ϕ can be obtained by linear integration of the magnetic vector potential A around the closed path of the coil.  ϕ=

 B · dS =

(∇ A) · d S =

A · dl

(5.21)

5.3 Principle of the Wireless Charging

237

Fig. 5.18 Schematic diagram of coupling coil offset

Figure 5.18 shows a schematic diagram of the coupling coil offset, only considering the case when the coil is one turn. The horizontal deviation of the coil is y. The magnetic vector potential at the point P on the secondary coil is as follows: A=

μ0 I1 dl 4π R

(5.22)

where μ0 is the transmission efficiency. According to the geometric relationship, the relationship between the lateral distance x and the angle α can be obtained:

x=

a2 −

y 2 4

(5.23)

y ) 2a

(5.24)

α = 2 cos−1 (

The crossing portion of the coil is defined as the effective area Beff of the magnetic flux. From the relationship between y and radius a, the overlapping area of the primary and secondary coils can be obtained:

Be f f

⎡ ⎤ 

 2 y (y) 1 y ⎣ ⎦a 3 πa 2 − 2 =2 cos−1 180◦ 2a 2a 2 (2a 2 )2 + 1

(5.25)

Then we can get the mutual inductance relationship: M = (πa 2 )(Be f f )

μ0 √ π( a 2 + h 2 )3

(5.26)

For the primary and secondary coils with N turns, the total magnetic flux can be calculated by the superposition of the individual magnetic flux contributions of each.

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5 Wireless Charging System Electromagnetic Safety …

M=

N  N  μ0 (πai2 )(Be f f, j )  2 2 i=1 j=1 π( a j + h )

(5.27)

It can be seen from formula (27) that when the offset y increases, M will decrease accordingly, which will reduce the energy transmission efficiency of the entire wireless charging system. M is mainly related to ϕ and geometric quantities such as, the shape and size of the coil, and the relative position of the coil. The relationship between the coupling coefficient k and the self-inductance and mutual inductance of the coil is as follows: k=√

M L1 L2

(5.28)

According to the above formula, when the change in coil self-inductance caused by the offset is ignored, that is, when the coil self-inductance L 1 L 2 is constant, the coupling coefficient k is proportional to the mutual inductance of the coil. Then when the offset increases, the ability to transmit electromagnetic fields will also decrease. According to the working characteristics of the wireless charging system and the definition of transmission power, the expression of Ptran can be obtained:  2  2 ω 2 k 2 L p L s Ptran =  Ip  Z r =  Ip  Zs

(5.29)

When the coupling coil is offset, because the system’s resonance frequency ω, primary coil impedance parameters, compensation topology, and load are constant, assuming that the secondary-side equivalent impedance does not change, the system transmission power Ptran is only related to the primary coil current I p and the coupling coefficient k. When the coupling coil is shifted, the coupling coefficient k will change accordingly, and the reflected impedance Zr on the secondary side will also change accordingly, and the transmission power Ptran of the system will fluctuate greatly. Therefore, the primary coil current I p of the compensation topology is required to be adjusted in reverse with equal amplitude according to the change of the coupling coefficient k, which is use to offset the effect of the change in Z r , so as to maintain the relative stability of the system transmission power Ptran . This automatic adjustment characteristic of the primary coil current I p can improve the anti-offset capability of the system. According to the power expressions under the two topologies, the curve of power varying with coupling coefficient can be obtained. Considering that when the coupling coil is offset, the size of the coupling coefficient k will change, and the anti-offset performance can be reflected by the change of efficiency in a certain range. According to the power expression under the two topological structures obtained above, the power variation curve with k can be obtained. When the coupling coil is offset, the mutual inductance between the primary coil and the secondary coil

5.3 Principle of the Wireless Charging

239

Fig. 5.19 Curves of system efficiency with coupling coefficients

will decrease, and the coupling coefficient will change. According to the change curve of the system efficiency with the coupling coefficient, it can be known that the efficiency change curves of the two topologies are basically the same. When the coupling coefficient is greater than 0.2, the system has higher efficiency, and can reach more than 80% (Fig. 5.19).

5.3.4 Effect of Offset on Efficiency (1) S parameter The vector network analyzer (VNA) E5061B is equipped with impedance analysis module and wireless charging module, which can measure the wireless power transmission efficiency between the coupling devices in real time with any load impedance setting. The experimental arrangement of the S-S topology is shown in Fig. 5.20 a. Where a 10 nF capacitor is used in parallel with a 5 nF capacitor to obtain the required capacitance. The S-parameter measurement results are shown in Fig. 5.20b. It can be seen from Fig. 5.20b that energy can be transferred in the frequency bands of 70 kHz to 100 kHz. The experimental layout of the LCC topology is shown in Fig. 5.21a. Where a 40 nF capacitor is used in parallel with a 10 nF capacitor to obtain the required capacitance C 1 . The measured value of C 1 is 50 nF, and two 10 nF capacitors are connected in parallel to obtain the required capacitance C p . The measurement results of C p and S parameters are shown in Fig. 5.21a. As can be seen from Fig. 5.21b, energy can be transferred in the frequency bands of 60 kHz to 110 kHz. Compared with S-S structure, power transmission has a wider frequency range. (2) Efficiency Using the wireless charging module of VNA E5061B, the efficiency of the S-S topology is tested. The equivalent circuit tested is shown in Fig. 5.22. The power transmission efficiency curve of the system is shown in Fig. 5.21. A sharp peak appears at 85 kHz and the peak efficiency can reach 95% (Fig. 5.23).

240

5 Wireless Charging System Electromagnetic Safety …

(a) Compensation capacitance and test platform

(b) S-parameters Fig. 5.20 S-parameter measurement platform for S-S topology

By the above method, the efficiency curves of the coupling device system can be obtained under different resistance values R and reactance values X. A map of the efficiency of the system with R and X in the range of 0 to 100 can be obtained. It can be seen from Fig. 5.24 that the maximum efficiency of the coupling device is close to 95% at X = 0 and R = 70 . Similarly, the efficiency of the bilateral LCC topology is tested. The equivalent circuit tested is shown in Fig. 5.25. From Fig. 5.26, unlike the S-S topology, there are three peaks in the efficiency curve, and the maximum efficiency appears in the second peak. The peak frequency is still 85 kHz, and the peak efficiency is 95%. The three peaks make the coils have high efficiency in a wide frequency range, i.e. from 50 kHz to 110 kHz, with an average value of more than 80%. It can be seen that from Fig. 5.27, the maximum efficiency of the coupling device is close to 95.7% at X = 0 and R = 30O .

5.3 Principle of the Wireless Charging

(a) Compensation capacitance and test platform

(b) S-parameters Fig. 5.21 S-parameter test planform for bilateral LCC topology

Fig. 5.22 S parameters measurement equivalent circuit of S-S topology

241

242

5 Wireless Charging System Electromagnetic Safety …

Fig. 5.23 S-S topology efficiency versus frequency

Fig. 5.24 S-S topology efficiency map

Fig. 5.25 Measurement equivalent circuit for S parameters of bilateral LCC topology

5.3 Principle of the Wireless Charging

243

Fig. 5.26 Curve of efficiency of two-sided LCC topology with frequency

Fig. 5.27 Efficiency map of bilateral LCC topology

(3) The effect of offset on efficiency When the LCC topology is used, the coupling characteristics parameters of the coil system are tested for changes in eight cases, i.e. alignment, coil lateral offset of 50 mm, 100 mm, 150 mm, 200 mm, 250 mm, 300 mm, 350 mm. The effect of offset on the impedance and efficiency of the coupled device system is analyzed. As can be seen from Fig. 5.28, it can be seen that when the coils are aligned, the amplitude of S11 is less than −15 dB in the frequency band of 70–100 kHz, indicating that almost all power is transmitted from the primary side to the secondary side. With the increase of coil offset, the resonance peak of S11 gradually decreases. When the offset is 100 mm, the resonance peak becomes one, and the peak frequency is about 82 kHz, indicating that the frequency band of power coupling is becomes narrow. Finally, when the offset reached half of the coil width, i.e. 350 mm, the peak of the curve disappeared, indicating that almost all power is reflected at this time, and there is almost no power coupling. The same method was used to measure the efficiency of the S-S topology. Figures 5.29 and 5.30 are the comparison curves of the coil efficiency measured under two different topologies at eight different offsets.

244

5 Wireless Charging System Electromagnetic Safety …

Fig. 5.28 S11 curve at different offsets

It can be seen from Fig. 5.29 that the coil efficiency is close to 0 in the frequency band of 0–30 kHz. At around 30 kHz, the efficiency curve starts to rise rapidly, and reaches a maximum at a frequency of 85 kHz, and then the curve begins to decrease. The efficiency curve is basically consistent when the coils are aligned and offset by 50 mm, and the maximum efficiency can reach 95%. However, the efficiency is

5.3 Principle of the Wireless Charging

245

Fig. 5.29 Efficiency comparison of S-S topology with different offsets

Fig. 5.30 Efficiency comparison of bilateral LCC topology with different offsets

significantly reduced when the offset is 100 mm, and the maximum efficiency is reduced to about 75%; It can also be found that the peak frequency has nothing to do with the offset, all of which are 85 kHz. As can be seen from Fig. 5.30, the efficiency changes of the bilateral LCC coupling device system is similar to that of the S-S topology in the frequency band of 0 to 30 kHz. The difference is that at this time, the efficiency curve presents three resonance peaks, and reaches the peak efficiency at the second peak, namely 85 kHz.

246

5 Wireless Charging System Electromagnetic Safety …

After the third spike, the curve begins to fall. The four efficiency curves with offsets of 0, 50 mm, 100 mm and 150 mm are almost coincident, which shows that when the offsets are within 150 mm, the efficiency curve has little effect, however When the offset exceeds 200 mm, it drops from 95 to 90%. Efficiency will be greatly reduced when the offset exceeds 200 mm. When the offset exceeds 250 mm, the highest efficiency is transferred to 60 kHz frequency point. From the S-parameters and efficiency of the coupled device system under offset, compared to the S-S topology, the bilateral LCC topology has better anti-offset characteristics.

5.4 Magnetic Field Distribution Due to the large transmission power, large transmission distance, and easy dislocation of the coupler in the wireless charging system of EVs, a large magnetic field may be generated around the coupling coil, and a magnetic field leakage may occur, which will aggravate the magnetic field strength. Exposure to a large magnetic field for a long time will increase the electric field strength in the human body. When the electric field strength exceeds a certain limit, it will affect the nerve cells and other electrically sensitive cells in the human body.

5.4.1 Vehicle Model The whole vehicle model is established in the 3D modeling software CATIA according to the relevant dimensions of the vehicle, as shown in Fig. 5.31a. The basic dimensions of the car body are 4000 mm * 1700 mm * 1500 mm. According to SAE J2954, in order to simulate the shielding effect of the vehicle body on magnetic fields, the chassis of the body also includes aluminum plates with an area of 1500 mm * 1500 mm and a thickness of 1 mm. The three-dimensional model of the whole

(a) Vehicle model Fig. 5.31 Coupling coil layout

(b) Layout

5.4 Magnetic Field Distribution

247

vehicle and the coupling device are imported into software Maxwell, the coupling coil is arranged in the proper position of the body chassis. The coupling coil is arranged at the right lower part of the driver’s seat and close to the front axle of the vehicle. Considering the positioning function of the front wheel of the vehicle, the relative position of the coupling coil and the vehicle body is determined as shown in Fig. 5.31b.

5.4.2 Distribution of Magnetic Field Test Points According to the standard J2954, the vehicle model is also divided into four corresponding regions. Considering that the area 1 below the body chassis is the main area where the coupling coil carries out power transmission, a large magnetic field will be generated between the primary coil and the secondary coil during normal operation. However, the magnetic field strength of the area 1 is not specified here. For the other three areas, the human body is likely to be in these areas for a long time, so a reasonable and effective test plan needs to be developed to measure the magnetic field at the three areas separately. For area 2a, considering that the area is closest to the working area of the coupling device, the possibility of the magnetic field exceeding the standard is the greatest. Here, the area is mainly tested, and multiple measurement points are selected. Because the coupler is of symmetrical structure, it can be seen from the distribution diagram of the magnetic induction line of the coupler that the maximum value of the magnetic field appears in the cross section of the coupler center, so the section of axis X1 is selected as the measurement plane, which is 75 mm from the ground, as shown in Fig. 5.32. Measurement points P1 (front), P2 (right), P3 (left), and P4 (rear) are selected in an area 100 mm away from the vehicle body. In addition, three groups of measuring points P7–9 , P10–12 , P13–14 are added in the front left and right three directions of the body side according to the size of the coupler. The distribution of each test point is shown in Fig. 5.32a. the measurement points P1–4 are used to describe the magnetic field distribution in the four directions of the

(P5)

X 7HVWDUHD

P2

P10

P11

P12

Z YHKLFOHERG\

P7

P8

Z

P4

2b

2b

&RXSOHU P9

'ULYHUÿVVHDW

P13

P14

P15

700mm

P1

Y

P6

P5 X2

P3

P2 X1

2a

2a

P3 (P6)

Fig. 5.32 Arrangement of magnetic field measurement points

Y

X

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5 Wireless Charging System Electromagnetic Safety …

Fig. 5.33 Seat test point layout

Z1 P16 300mm

P16

P17 320mm

P17

P18 280mm

P18

P19

P19

vehicle body, and measurement points P7–14 are used to describe the magnetic field distribution around the coupler. For the 2b area 700 mm from the ground, select the plane on which the axis X2 is located as the measurement plane, and define the point that is 100 mm from the vehicle body and intersects X2 as the measurement point P5 (right) and the measurement point P6 (left). P5 and P6 are located directly above the measurement points P2 and P3 , respectively. For area 3, this area is the interior area of the vehicle body. Considering the area where the driver is located, only the driver’s seat is measured here. The axis plane in the driver’s seat, that is, the plane on which the axis Z1 is located, is selected as the measurement plane. The magnetic field strength of the head (P16 ), waist (P17 ), seat cushion (P18 ), and foot (P19 ) at the seat position are measured. The relative positions of the four measurement points are shown in Fig. 5.33, and each measurement point is perpendicular to the seat surface, with a relative height of 10 mm.

5.4.3 Coupling Coil Mutual Inductance at Offset As shown in Fig. 5.34, the distance between the central axis of the primary coil and the central axis of the secondary coil is defined as an offset amount Δy. z

x y

Δy

Fig. 5.34 Three-dimensional model of coupling device with offset

5.4 Magnetic Field Distribution Table 5.7 Relationship between offset mutual inductance and coupling coefficient

249 Offset (mm)

Mutual inductance (uH)

Coupling coefficient

0

95.5

0.392

50

83.2

0.341

100

70.4

0.286

150

55.8

0.254

200

41.7

0.179

250

28.3

0.157

300

16.3

0.076

350

3.7

0.021

The width of the coupling device is 700 mm. Where, 50 mm is selected as one offset to study the coupling device from alignment to 50% offset. That is to say, seven cases with offsets of 50 mm, 100 mm, 150 mm, 200 mm, 250 mm, 300 mm and350 mm are selected to analyze the distribution of magnetic field. The mutual inductance and coupling coefficient of coupling device can be calculated by using Maxwell. The simulation results are shown in Table 5.7. After the coupling coil is fixed at the specified offset position, the inductance meter is used to measure the mutual inductance of the coupler under the offset. As shown in Fig. 5.35, the offset test platform of coupling coil is in full alignment, offset of 150 mm and offset of 300 mm (Fig. 5.36).

(a) Alignment

(b) Offset 150mm

(c)Offset 300mm

Fig. 5.35 Coupling coil position with offset

(a)

(b)

Fig. 5.36 (a) Mutual inductance varies with offset; (b) Coupling coefficient varies with offset

250

5 Wireless Charging System Electromagnetic Safety …

5.4.4 Magnetic Field Distribution According to Kirchhoff’s law and the superposition principle of current, we get: I˙Lp =

U˙ jωLp

(5.30)

When resonance occurs, the current of the primary coil is only related to the input voltage, so when the coupling coil is offset, the magnitude of the current on the primary coil remains unchanged, and the strength of the magnetic field generated by it remains unchanged. However, the current of the secondary coil changes with the mutual inductance between the coils, which will lead to the change of electromagnetic field around the coupling device. As shown in Fig. 5.37, within the 150 mm offset, the change of secondary current is small, and the current decrease is larger as the offset continues to increase. The magnetic field distribution when the coupling device is aligned is shown in Fig. 5.38. The red space is the area where the magnetic field strength is higher than the standard magnetic field limit when the coupling device is working. From the front and side views of the magnetic field distribution, it can be seen that the red area is the area with larger magnetic field strength Because the body has a certain shielding effect on the magnetic field, the red area is mainly concentrated below the car chassis. It can be seen from the top view that the magnetic field gradually diffuses outward with the coupler as the center. The measurement points P11 and P14 are in the green area, and the other measurement points are in the blue area or the transition area. The specific magnetic induction values are shown in Table 5.8. It can be seen that the magnetic field strengths at the measurement points P11 and P14 are the largest, 12.80 uT and 12.10 uT, respectively. The magnetic field strengths at the measurement points P2 , P3 , P10 , P12 , P13 , and P15 at the chassis are also at a high level. The magnetic field strength at the measurement points P16–19 in the car is less than 1 uT, which Fig. 5.37 Secondary coil current changes with offset

5.4 Magnetic Field Distribution

251

(a) Magnetic field coupling region 3 3

3

3

3

3 3

3

3

3

3

3

3

3

3

3

3 3

3

3

3

(b) Front view

(c) Top view

(d) Side view

Fig. 5.38 Magnetic field distribution of coupler alignment

Table 5.8 Measurement points and magnetic induction Measuring point

Coordinate

Magnetic induction

Measuring point

Coordinate

Magnetic induction

P1

(0,1600,75)

0.77

P11

(850,0,75)

12.80

P2

(950,0,75)

5.99

P12

(850,−350,75)

7.33

P3

(−950,0,75)

6.30

P13

(−850,350,75)

8.93

P4

(0,−2600,75)

0.09

P14

(−850,0,75)

12.10

P5

(950,0,700)

0.65

P15

(−850,−350,75)

7.59

P6

(−950, 0,700)

0.62

P16

(300,−400,1140)

0.02

P7

(350,1500,75)

0.97

P17

(300,−300,840)

0.02

P8

(0,1500,75)

1.20

P18

(300,−150,520)

0.08

P9

(−350,1500,75)

0.90

P19

(300,300,240)

0.48

(850,350,75)

8.65

P10

is obviously smaller than the outside of the car, indicating that the body has a good shielding effect on the magnetic field. As shown in Figs. 5.39 and 5.40, the front view and side view of the magnetic field distribution are respectively for coil alignment and offset of 150 mm and 300 mm. It can be seen that as the offset increases, the magnetic field on the axial surface of the coil will moves to the left, the distribution of the magnetic field will become irregular, and the area of the red area will also decrease. It can be seen from Fig. 5.40 that as the coil shifts, the working area of the magnetic field will move toward the axial plane of the seat, and the influence on the magnetic field at the seat may increase. Focus on the points where the measured value of magnetic induction intensity changes greatly

252

5 Wireless Charging System Electromagnetic Safety …

(a) 0mm

(b) 150mm

(c) 300mm

Fig. 5.39 Front view of magnetic field distribution of coupler with offset

(a) 0mm

(b) 150mm

(c) 300mm

Fig. 5.40 Side view of magnetic field distribution of coupler offset

with deviation, as shown in Table 5.9. It can be seen that P11 reaches the maximum when the offset is 200 mm and 250 mm, which are 32.86 ut and 46.06 ut respectively, exceeding the standard limits. Table 5.9 Magnetic induction intensity (uT) at the measurement point when the coil is offset Measuring point 0 mm 50 mm 100 mm 150 mm 200 mm 250 mm 300 mm 350 mm P2

5.99

7.19

8.20

9.60

11.23

15.39

9.65

P3

6.30

5.33

4.71

3.55

2.96

2.14

1.42

0.40

P10

8.65

8.46

8.46

11.59

11.07

10.76

7.97

3.09

P11

18.46

20.98

32.86

46.06

28.57

23.93

P12

7.33

9.37

10.67

10.67

9.98

11.35

7.91

3.66

P13

8.93

6.45

5.77

4.11

3.12

3.62

1.77

0.51

9.53

7.25

5.65

4.09

2.39

0.71

5.65

4.34

3.47

2.22

1.35

0.46

P14 P15

12.80 16.76

7.98

12.10 11.2 7.59

6.48

5.4 Magnetic Field Distribution

253

5.4.5 Magnetic Induction Strength Measurement (1) Magnetic field measurement at operating frequency of 85 kHz The coupling device is installed under the chassis of EV, and the test platform is constructed, as shown in Fig. 5.41. When the wireless charging system is stable, magnetic field detectors are used to measure the magnetic induction intensity at 15 measurement points outside the vehicle and 4 measurement points at the driver’s seat inside the vehicle, as shown in Fig. 5.42. The measurement results of the magnetic induction intensity at the 19 measurement points are shown in Table 5.10. Since the measurement points P11 and P14 outside the car are closer to the chassis, the magnetic field strengths are the magnetic field strengths at the measurement points P10 , P12, P13 , and P15 are also at a high level, all near 10 uT. The EV equipped with wireless charging system has so as to ensure that the secondary coil located at the vehicle chassis has an offset of 50 mm, 100 mm, 150 mm, 200 mm, 250 mm, 300 mm and 350 mm respectively with respect to the primary coil fixed on the ground. The magnetic induction intensity of 19 measuring points with different offsets are measured. The value of P11 point with the greatest EMF safety risks is measured. The results are shown in Fig. 5.43. When the offset is 200 mm, the magnetic field strength at P11 exceeds the EMF limit. Therefore, wireless charging systems should try to avoid offsets of 200 mm or more as far as possible. The large offset will not only greatly reduce the system efficiency, but also increase the magnetic field strength in some areas, and even exceed the EMF limit, causing harm to the human body (2) Magnetic field measurement at operating frequency of 85 kHz Using the spectrum analyzer NF-5035, you can test the electric field strength and the magnetic field strength and magnetic induction strength in the XY direction, YZ direction and XZ direction with frequency. Using this device, the low-frequency magnetic field emission was measured, and the magnetic induction intensity spectra in different directions at 7.5 cm and 70 cm outside the vehicle were obtained, as shown in Fig. 5.44. The unit of the electric field in the figure is pT, and the red curve indicates the maximum electric field Value, the green curve is the transient value, and the yellow curve is the average value.

Primary control box

(a) System composition Fig. 5.41 Test platform

Coupler

Secondary control box

(b) Experimental vehicles equipped with wireless charging systems.

254

5 Wireless Charging System Electromagnetic Safety …

(a) Points outside of the car(P2ǃP5ǃP11)

(b) points inside the car(P16ǃP17ǃP18ǃP19) Fig. 5.42 Magnetic induction measurement Table 5.10 Simulation and experimental results of magnetic field strength (uT) at each measuring point Measuring Simulation Experiment Error point

Measuring Simulation Experiment Error point

P1

0.77

0.14

−0.63 P11

12.80

14.63

1.83

P2

5.99

5.21

−0.78 P12

7.33

8.51

1.19

P3

6.30

5.37

−0.93 P13

8.93

8.73

−0.20

P4

0.09

0.06

−0.03 P14

12.10

15.28

3.18

P5

0.65

0.09

−0.56 P15

7.59

9.14

1.55

P6

0.62

0.02

−0.60 P16

0.02

0.01

−0.01

P7

0.97

0.32

−0.65 P17

0.02

0.01

−0.01

P8

1.20

0.48

−0.72 P18

0.08

0.05

−0.03

P9

0.90

0.27

−0.63 P19

0.48

0.23

−0.25

P10

8.65

8.39

−0.16

5.4 Magnetic Field Distribution

Fig. 5.43 Comparison of magnetic field strength at P11

(a) Measurement and the result at 70cm from the ground outside the vehicle body

(b) Measurement and the result at 7.5cm from the ground outside the vehicle body Fig. 5.44 Magnetic induction test chart at different positions

255

256

5 Wireless Charging System Electromagnetic Safety …

It can be seen from the figure that the magnetic field spikes appear near the frequency points of 85 kHz, 170 kHz, 255 kHz, and 340 kHz, respectively. These frequency points are the multiples of the system operating frequency of 85 kHz, and the magnetic induction intensity is maximum at 85 kHz. Therefore, when analyzing the magnetic field during the operation of the wireless charging system, studying the magnetic field at the frequency of operation and frequency doubling can better represent the low-frequency emission characteristics of the system.

5.5 Modeling and Suppression of Power Line Conducted Electromagnetic Interference The problem of electromagnetic interference in electric vehicle wireless charging systems involves conducted emissions and radiated emissions. It focuses on the problem of conducted electromagnetic emissions on DC power cables of DCpowered wireless charging systems. In addition, radiated emissions caused by cables can also cause electromagnetic interference to sensitive equipment inside and outside the vehicle.

5.5.1 Interference Source The high-speed switching of the MOSFET of the power switching device of the inverter in the wireless charging system is the main cause of conducted interference. Therefore, the establishment of a high-frequency equivalent model of the power switching device to analyze the interference source signal is very meaningful for predicting conducted electromagnetic interference. Taking the NW-channel MOSFET of the IPW65R099C6 selected by the inverter as an example, the basic characteristic parameters of the MOSFET are shown in Table 5.11. Use the dynamic MOSFET module of Simploler to set the parameters, and fit the MOSFET transfer characteristic curve, output characteristic curve and drain-source diode characteristic curve according to the characteristics in the datasheet. Finally, you need to set the MOSFET and diode thermal effect curve, as shown in Fig. 5.45. In order to Table 5.11 MOSFET basic characteristic parameters

Parameters

Values

VDS@Tjmax

700

RDS (on), max

0.099

Units V

Qg , typ

127

nC

ID, pulse

115

A

Eoss@400 V

10

μJ

Body diode di/dt

300

A/μs

5.5 Modeling and Suppression of Power Line …

257

Fig. 5.45 Transfer characteristic curve output characteristic curve and drain-diode characteristic curve

compare the difference between the ideal MOSFET model and the dynamic MOSFET model, the corresponding clamping inductance switching circuits are established respectively, as shown in Fig. 5.46. The dynamic characteristics of the two MOSFET models are obtained from the simulation of the clamp inductor switch circuit above as shown in Fig. 5.47. The red curve is the drain-source voltage when the ideal MOSFET model is used, and the green curve is the drain-source voltage when the dynamic MOSFET model is used. It can be seen that with the dynamic MOSFET, the rising edge of the square wave is significantly smoother than the ideal model, which is more in line with reality. The switching characteristics of the dynamic MOSFET model are verified, and the circuit model of the wireless charging system is built in Simplolor, as shown in Fig. 5.48. In addition, considering the parasitic parameters of the MOSFET, the parasitic capacitance is paralleled between the source and the drain, and the pin

Fig. 5.46 Ideal MOSFET (left) and dynamic MOSFET (right) clamping inductance switching circuit

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5 Wireless Charging System Electromagnetic Safety …

Fig. 5.47 Comparison of dynamic characteristics of two MOSFET models

Fig. 5.48 Wireless charging system circuit model using dynamic MOSFET model

Drain-source voltage /V

Drain-source voltage /V

inductance caused by the installation process is connected in parallel. The selection of parasitic parameters can refer to the input capacitance and output capacitance of the datasheet, calculate CDS = 200 pF, and the pin inductance takes 40 nH. Set the DC power supply to 400 V, and the simulation results of the voltage across the source and drain of the MOSFET are shown in Fig. 5.49. It can be seen that the voltage waveform rings at both the rising and falling points. The highest ringing voltage is 440 V, which exceeds the normal operating voltage by 40 V. The ringing gradually stabilizes after 0.4 μs.

Fig. 5.49 MOSFET interference voltage waveform

5.5 Modeling and Suppression of Power Line …

259

5.5.2 Electromagnetic Interference Mechanism (1) Parasitic parameter extraction The parasitic parameters of the system are the main elements that form the transmission of electromagnetic interference. Parasitic parameters can be extracted through a combination of modeling and measurement. The measurement method is usually adopted, and the S-parameter and Z-parameter parasitic parameters of the wireless charging system are measured by the time-domain reflectometry TDR and vector network analyzer VNA, as shown in Table 5.12. Among them, L p1 , L p2 are the lead inductance values; Lp, Rs are the parasitic inductance and equivalent resistance of the filter capacitor C 7 ; RL0 is the high-frequency parasitic resistance of the filter capacitor L 0 ; L C0 is the parasitic inductance of the filter capacitor C 0 ; C 3 –C 6 are the inductance between poles of MOSFETs; RL1 , RL2 are the equivalent resistance of the coupled coil pair; Rlf1 , Rlf2 are the equivalent resistance of L f1 and L f2 ; L Cf1 , L Cf2 are the parasitic inductance of C f1 and C f2 ; L C1 , L C2 are the parasitic inductance of C 1 and C 2 , respectively; L cable1 , L cable2 , L cable3 and Rcable1 , Rcable2 , Rcable3 are the parasitic inductance and equivalent resistance of the connection cable; C 11 , C 12 , C 13 are the primary coil, secondary coil and battery parasitic capacitance; C 10 is the mutual parasitic capacitance between the pair of coupling coils; C 14 and C 15 are the parasitic capacitance to ground at the midpoint of the two bridge arms of the inverter. Based on the parasitic parameters obtained in Table 5.12, build a system highfrequency equivalent circuit model of conducted interference, as shown in Fig. 5.50. The blue devices are all parasitic parameters, where C 10 is the parasitic capacitance generated between the primary coil and the secondary coil. In addition, the connecting cable will also have parasitic parameters at high frequencies, as shown by C 10 and Rcable in the figure. (2) Noise interference source signal Using Matlab/simulink to build a high-frequency equivalent circuit of conducted interference as shown in Fig. 5.50, the electromagnetic interference source signal Table 5.12 Parasitic parameters Parasitic parameters

Values

Parasitic parameters

Values

L p1 /L p2

10 nH

Rlf1 /Rlf2

0.03

L C0

50 nF

RLf3

1 μ

Lp

0.2 μH

RL1 /RL2

115 μ

L cf1 /L cf2

10 nH

C p1 /C p2

380 pF

L c1 /L c2

8 nH

C 3 –C 6

142 pF

L cable1 /L cable2 /L cable3

1.76 μH

C 10

21.4 nF

Rcable1 /Rcable2 /Rcable3

0.0524

C 11 /C 12

80 pF

RS

0.1

C 13

130 pF

260

5 Wireless Charging System Electromagnetic Safety …

Fig. 5.50 Modeling of a high-frequency equivalent circuit of conducted interference in a wireless charging system

at the midpoint of the single bridge arm of the inverter is simulated. Figure 5.51a is the common mode interference voltage, and Fig. 5.51b is the differential mode interference current. (3) Conducted interference propagation path (1) Differential mode interference path Due to the parasitic capacitance between the primary coil and the secondary coil, there may be differential mode interference current in the secondary circuit. The parasitic capacitance is relatively large in the 150 kHz–108 MHz band. Therefore, we assume that differential mode interference is transmitted in the primary side circuit. In differential mode path analysis, the differential mode interference source is equivalent to a current source, and the path of the interference current is shown in Fig. 5.52. Conducted voltage is an important parameter that characterizes conducted electromagnetic interference. The figure consists of R1 , R2 , C 8 and C 9 is a linear network impedance analyzer LISN (line impedance stability network), used to measure the conducted voltage on the system DC power line. The differential mode interference current is mainly composed of two propagation paths of 1 and 2, both of which flow through LISN. The sequence of the differential mode interference current I DM flowing through the components is as follows:

Fig. 5.51 Common mode interference voltage and differential mode interference current

5.5 Modeling and Suppression of Power Line …

261

Fig. 5.52 Differential mode interference current path

Path I 1 :   9 →C 8 →R1 I D M → C5 → L P2 → RR2S→C → L P1 → I D M ; →L P →C9 Path I 2 : I D M → L f 1 → RL f 1 →



L C1 →C1 →L cable1 →Rcable1 →L 1 →R L1 R S →L P →C9



→ L P2 →



R2 →C9 →C8 →R1 R S →L P →C7



→ L P1 → I D M

In order to analyze the input and output changes of electromagnetic interference noise, an impedance equivalent circuit is established according to Fig. 5.52, as shown in Fig. 5.53. Among them, the impedances Z1, Z2, Z3, Z4 and Z5 of each part are expressed by formulas (3.1)–(3.5). Finally, the transfer function is used to express the relationship between the system’s electromagnetic interference noise output and input, that is, the functional relationship between the conducted voltage measured on the LISN and the interference source signal, which is expressed by formula (3.6). In the 150 kHz–108 MHz frequency band, the amplitude-frequency and phase-frequency characteristics of the transfer function are shown in Fig. 5.53. Differential mode interference has the largest amplitude ratio at 150 kHz. As the frequency increases, the amplitude ratio decreases significantly. Therefore, differential mode interference is more pronounced in the low frequency band than in the high frequency band (Fig. 5.54).

Fig. 5.53 Equivalent circuit of differential mode interference path

262

5 Wireless Charging System Electromagnetic Safety …

Fig. 5.54 Amplitude-frequency characteristics of conducted differential mode interference voltage transfer function

Z 1 = R1 + Z2 =

1 1 + + R2 sC8 sC9

1 + s L P + RS sC7

Z 3 = s L f 1 + RL f 1 + Z4 = s Lc f 1 + Z5 =

(5.32)

1 sC1

(5.33)

1 sC f 1

(5.34)

1 + s L cable1 + Rcable1 + s L 1 + R L1 sC1 4 

G(S1 ) =

(5.31)

Z R1 VR D Z1 Z2 i=0   = = 9 VD M (Z 1 Z 2 ) + Z 3 + (Z 4 Z 5 ) C3 + Z L P1 + Z L P2 Z 1  i=2

(5.35)

ai s i bi s i

a0 = 1.25 × 10−9 , a1 = 2.5 × 10−15 , a2 = 3.875 × 1020 , a3 = 7.5 × 10−26 , a4 = 3.75 × 10−32 b2 = 1.575 × 10−19 , b3 = 4.525 × 10−25 , b4 = 2.7 × 10−30 , b5 = 6.9 × 10−36 , b6 = 6.75 × 10−42 b7 = 2.255 × 10−48 , b8 = 6.75 × 10−57 , b9 = 2.25 × 10−63

(5.36)

(2) Common mode interference path Common mode interference is mainly formed by the distribution parameters of the system to the ground. It is necessary to consider the secondary side circuit part when analyzing common mode interference. The common mode interference path of the wireless charging system is shown in Fig. 5.55, where the common mode interference source is added between the midpoint of the first bridge arm and ground, which is equivalent to a voltage source. The common-mode interference current includes four paths of 1, 2, 3, and 4 in the figure. The common-mode interference current I 1 will pass through the LISN to form a conducted disturbance voltage. I 2 and I 4 are a pair of interference currents that flow from the interference source through the primary coil and finally into the ground. However, when I 3 flows through the resonant coil

5.5 Modeling and Suppression of Power Line …

263

Fig. 5.55 Common mode interference current path diagram

L 1 , a part of the interference current will flow through the coupling coil to the parasitic capacitance L 1 to the secondary coil on the vehicle side. Circuits and batteries eventually flow into the ground. Therefore, this part of the interference current may affect the power supply performance of the vehicle’s low-voltage equipment. I 4 flows through the parasitic capacitance C 14 and from the midpoint of the first leg into the ground. The sequence of each current flowing through the component is as follows:   P2 →R2 →C 9 → gr ound → IC M ; Path I 1 : IC M → CL 5P1→L →R1 →C8   E 1 →R L1 → C11 → gr ound → IC M ; Path I 2 : IC M → C5   3 →Vbat →C 13 → gr ound → IC M ; Path I 3 : IC M → E 1 → C10 → ERl22 →E →C12 Path I 4 : IC M → C P1 → gr ound → IC M . Among them, E 1 represents the current flowing through the route: L f 1 → R L f 1 → L C1 → C1 → L cable1 → Rcable1 → L 1 ; E 2 represents the current flowing through the route: L 2 → L cable2 → Rcable2 → C2 → L C2 → L f 2 → R L f 2 ; E 3 represents the current flowing through the route: R L f 3 → L f 3 → L cable3 → Rcable3 . As shown in Fig. 5.56, the impedance of the equivalent circuit of the commonmode interference path consists of Z 1, Z 2, Z 3, Z 4, Z 5 , and Z 6 respectively represent part of the circuit impedance value, and the impedance of each part is expressed by formulas (3.7)–(3.12). The relationship between the output and input of common

Fig. 5.56 Equivalent circuit of common mode interference path

264

5 Wireless Charging System Electromagnetic Safety …

mode electromagnetic interference noise is expressed by the formula (3.13). Z 1 = (R1 +

1 1 1 + s L P1 )//(s L P2 + + R2 + ) sC8 sC9 sC5 Z 2 = s L 2−2 + R2−2 + Rbat

(5.37) (5.38)

1 Z 3 = s L 2−1 + R2−1 + s L cable2 + Rcable2 + s L f 2 + R L f 2 + s L 0 + R L0 + + s L C2 + s L cable3 + Rcable3 sC2

(5.39)

Z 4 = s L c f 1 + R L f 1 + sL C1 +

1 + s L cable1 + Rcable1 + s L C1 + R L1 sC1

Z 5 = (Z 2 //Z 3 ) + Z C13

(5.41)

Z 6 = [(Z 5 + Z 4 )//Z C5 ] + Z C11

(5.42)

G(S2 ) = =

(5.40)

V RC (Z 6 //Z C P1 )R1 = VC M (Z 1 //Z 6 //Z C P1 )2 (Z C P1 + Z C8 + R1 )

7.32 × 10−3 s 4 + 2.446 × 105 s 3 + 2.078 × 1010 s 2 + 4.4418 × 1014 s 5 7.294 × 104 s 7 + 1.23 × 1012 s 6 + 1.56 × 1017 s 5 + 6.631 × 1021 s 4 + 9.4 × 1025 s 3

(5.43)

The amplitude-frequency and phase-frequency characteristics of the commonmode interference voltage transfer function are shown in Fig. 5.57. It can be seen that the common mode interference is not much different in the 150 kHz to 108 MHz frequency band, and the whole frequency band varies from −5 dB to 0 dB. Compared with differential mode interference, common mode interference is more significant in the entire frequency band. Fig. 5.57 Amplitude-frequency characteristics of common-mode interference voltage transfer function

5.5 Modeling and Suppression of Power Line …

265

5.5.3 Conducted Interference Prediction (1) Differential mode conducted interference The establishment of a differential mode interference circuit model is shown in Fig. 5.58a. Inject the differential mode interference current, and the conduction voltage is simulated on the LISN. As shown in Fig. 5.58b, the red curve represents the

Fig. 5.58 Differential mode conduction voltage simulation prediction

266

5 Wireless Charging System Electromagnetic Safety …

differential mode interference negative conduction voltage, and the green line represents the limit required by Chinese standard GB18655. It can be seen that within the frequency range studied from 150 kHz to 108 MHz, the differential mode interference has the largest conducted voltage at 170 kHz (second frequency doubling), width an amplitude of 78.46 dBμV, exceeding the standard limit of 8.46 dBμV. The difference between adjacent harmonics is 85 kHz (working frequency). The superscalar at eighth frequency doubling at 680 kHz is the largest in the entire frequency band, exceeding the standard limit of 16.37 dBμV. The resonance occurs at 30 MHz, and the superscalar is 12.46 dBμV. (2) Common mode conducted interference The common mode interference circuit model established is shown in Fig. 5.59a. Inject the common-mode interference voltage, and the conducted voltage is simulated on LISN, as shown in Fig. 5.59b. Due to the symmetry of the circuit, the common-mode interference sources S1 and S2 have the same characteristics, so the common-mode interference is analyzed only for the case where the common-mode interference source S1 is injected. The common-mode interference has the largest conducted voltage at 170 kHz, with an amplitude of 81.96 dBμV, exceeding the limit of 11.96 dBμV. The difference between adjacent harmonics is 85 kHz, which also exceeds the maximum when the eighth frequency multiplier is 680 kHz, exceeding the limit of 19.89 dBμV. Resonance occurs at 30 MHz, when the standard limit of 15.17 dBμV is exceeded. (3) Differential common mode conducted disturbance voltage In order to better predict the actual situation, common mode interference sources and differential mode interference sources are injected into the simulation model established by CST software, as shown in Fig. 5.60, where yellow port 1 is the injection port of the common mode interference source Port 2 is the injection port for the source of differential mode interference. The conduction voltage simulation comparison of the three modes is shown in Fig. 5.61. Among them, the black curve represents the conduction voltage simulation result when the common mode and the differential mode act simultaneously, the blue curve represents the conduction voltage simulation result under the common mode only, and the red curve represents the conduction voltage simulation result under the differential mode only. According to the results shown in Fig. 5.61, it can be seen that at 170 kHz, 430 kHz, 1.6, and 30 MHz, the three curves all exhibit resonance phenomena. At these four frequencies selected, the conduction voltage result in common mode simulation is closer to the conduction voltage in common mode simulation. However, the simulation results of the common differential mode together in the full frequency band are larger than the differential mode simulation results. The difference is more obvious at high frequencies, especially in the 30–108 MHz band. Therefore, from the perspective of the simulation results of the conduction voltage, the conduction voltage under the common mode interference is closer to the actual

5.5 Modeling and Suppression of Power Line …

Fig. 5.59 Simulation prediction of mode conduction voltage

Fig. 5.60 Common mode and differential mode simulation circuit model

267

268

5 Wireless Charging System Electromagnetic Safety … ——Common difference mode — —Common mode — —Differential mode

Fig. 5.61 Comparison of three simulation results

conduction voltage that the common mode and the differential mode both acting. It can be seen from the high frequency band (30–108 MHz) that the common mode conduction voltage is larger than the differential mode conduction voltage. This is consistent with the calculation result of the transfer function. Therefore, common mode interference is the main mode of conducted interference in wireless charging systems. In addition, when suppressing the conducted electromagnetic interference of the wireless charging system, priority may be given to suppressing the commonmode conducted voltage, and then to suppressing the differential mode interference voltage.

5.5.4 EMI Filter (1) Filter topology Although the important parameters of the filter are the same, its topology is also related to the effect of the filter. In order to select the best filter topology, the suppression effects of filters with different topologies are compared through simulation analysis to determine the best filter structure. The filter is designed between the LISN and the EUT under test, as shown in Fig. 5.62. When designing a filter, you can give priority to designing a common-mode filter, and then consider designing a differential-mode filter and a common-mode differential-mode hybrid filter. If the common mode filter can meet the design requirements, the differential mode filter is no longer designed. In order to select the common-mode filter topology with the best suppression effect, taking into account the size and weight of the filter, three filters with LC, CL, and CLC topologies were designed, and equivalent circuits were established for simulation analysis. See Fig. 5.63. Assuming that CY is 500 nF, the value of LC is 265 uH. The common mode filter simulation results of LC, CL and CLC topologies are shown in Figure 5.64. Some

5.5 Modeling and Suppression of Power Line … /3

C8

C9

C7

Filter

R1

269

S1

LCable3

S2 Lf1

C4

C3

RCable1

RCable2

C5

&3

RL1

C6

RLf2 D D 1 2

C10

D3 D4

RL2

&3

C11

/3

RCable3

L2

L1 S4

S3

R2

RL0 L0

RLf1LC1 C1 LCable1 LCable2 C2 LC2 Lf2

9EDW

C13

C12

Fig. 5.62 Filter circuit model

/,61

(87

/,61

(a) CL

(87

(b) LC

/,61

(87

(c) CLC

Fig. 5.63 Common mode filter topology

Fig. 5.64 Comparison of the suppression effects of common mode filters

harmonics appear in the low frequency band. The nth harmonic (2th = 170 kHz, 3th = 255 kHz, 4th = 340 kHz, 5th = 425 kHz) and 1 MHz conducted voltage values are shown in Table 5.13 Listed. Obviously, the minimum conducted voltage and harmonic content are present in the circuit with the CL structure filter. The use of LC structure filters caused an undesirable new resonance phenomenon and still exceeded the standard limit at 1 MHz. After using the CLC structure filter, the conducted voltage is better suppressed in the research frequency band and meets the

270

5 Wireless Charging System Electromagnetic Safety …

Table 5.13 Comparison of three topological filters in wireless charging system Filter topology

Conducted voltage (dBμV)

Feature

170 kHz

255 kHz

340 kHz

425 kHz

1 MHz

LC type

61.12

43.91

43.91

47.14

55.18

Fewer components, but new resonance points are generated and exceed the standard

CLCtype

56.48

36.86

37.83

36.32

24.70

More components, large volume, good suppression effect

CLtype

38.57

18.26

18.03

14.43

6.36

Fewer components, best suppression effect

requirements of the limit, but the cost and volume of the CLC structure filter are larger than the other two filters. Therefore, compared with CL and CLC topologies, CL topologies are the best choice for common mode filters because they have the best suppression, minimum number of components, minimum volume, and weight. (2) Verification of priority filter design methodology In order to verify the effectiveness of the design of the priority common-mode interference suppression filter again, this section will design a hybrid filter and compare the suppression effect of the hybrid filter with the priority filter. (1) Differential filter topology selection The components of a differential mode filter are generally composed of a differential mode capacitor C x and an inductor Lx. As shown in Fig. 5.65, there are three different topologies of LC, CL, and LCLC. Figure 5.66 shows the conducted disturbance voltage on the DC input cable after the three topological filters are added to the system. It can be seen from the figure that the conducted voltage simulation results of the LC structure filter have dropped significantly in the whole frequency band, but a new resonance point has appeared in the low frequency band and the peak value of the conducted voltage is large. The conduction voltage of the CL structure filter has

(87

/,61

(87

/,61

(a) LC Fig. 5.65 Differential mode filter topology

(b) CL

(87

/,61

(c)LCLC

5.5 Modeling and Suppression of Power Line …

271

Fig. 5.66 Comparison of EMI suppression effects of differential mode filters

dropped in the whole frequency band, but it still exceeds the standard at the 30 MHz resonance point, and a new resonance point has appeared at high frequency. The conducted interference under the action of the LCLC filter is significantly reduced in the whole frequency band, no resonance point appears and the maximum value does not exceed 60 dBuV. By comparing the simulation results of the conducted voltage and differential mode interference after using LC, CL, and LCLC topology filters, the blue curve indicates that the LCLC structure filter has the best suppression effect, no resonance point is generated, and the electromagnetic interference voltage is significantly reduced. In summary, the best topology for differential mode filters is LCLC, and the best topology for common mode filters is CL. However, the position and order of the two filters in the same circuit also affect the overall EMI filtering effect of the wireless charging system. (2) Hybrid filter topology selection The structure of the three differential common mode hybrid filters is shown in Fig. 5.67. A circuit model with a common-mode differential mode filter is established in the CST simulation software. The common-differential mode hybrid filter

LY

LX1

LY

Cx1

CY1

Cx2

EUT

/,61

LX2

(a) Structure 1

Fig. 5.67 Differential filter topology

EUT Cx1 Cx2 LX2

CY1

(b) Structure 2

LX1

LY

CY1

CY1

/,61

LX1

/,61

Cx1

Cx2

LX2

CY1

EUT CY1

(c) Structure 3

272

5 Wireless Charging System Electromagnetic Safety …

is arranged between LISN and EUT, as shown in Fig. 5.68. It can be seen from the figure that the conducted disturbance voltage of the wireless charging system with a hybrid filter is suppressed and the effect is obvious, and the device with the third filter structure has the best filtering effect. (3) Comparison between hybrid filter and priority filter Compare the conducted voltage simulation results of the priority filter and the hybrid filter, see Fig. 5.69. a. Priority filter components are composed of a capacitor and an inductor, and hybrid filter components are composed of three capacitors and three inductors. The hybrid filter is three times the priority filter in terms of the number of components. The volume, weight and cost of the device will increase. b. Although the number of components of the hybrid filter increases, the suppression effect is better in the frequency range of 150 kHz–108 MHz, but the suppression amount is almost all lower than the priority filter by 15 dB. Only at 170 kHz the conducted voltage is about 5 dB below the priority. Fig. 5.68 Comparison of the suppression effect of the hybrid filter

—Hybrid filter —Priority filter

Fig. 5.69 Conducted disturbance voltage of DC power cable after adding filter

5.5 Modeling and Suppression of Power Line …

273

5.5.5 Conducted Disturbance Voltage Measurement

(1) Electrical function test Before measuring the electromagnetic interference of the wireless charging system, you need to test the charging function of the system. Figure 5.70a shows the electric vehicle with a 3.7 kW wireless charger installed. When the system works stably, read the charging information. As shown in Fig. 5.70b, the image information on the dashboard when the electric vehicle is being wirelessly charged. It can be seen that the vehicle speed is 0 km/h at this time, and the fast charging voltage and current are 400 V and 9 A. The cruising range that can be achieved by electricity is 148 km. Therefore, the established wireless charging system can realize the function of charging the vehicle battery. (2) Power line conducted electromagnetic interference test (1) Conducting launch platform construction A high-frequency conducted interference experiment platform for the DC-powered wireless charging system of electric vehicles was built in the EMC laboratory. The layout relationship of the experiment platforms is shown in Fig. 5.71. It can be seen from the figure that the experimental bench is mainly composed of a DC power supply, two 2 m DC shielded power cords, two LISNs, a wireless charging controller, a pair of circular coupling coils, an EMI receiver, and a host computer. Two 1.5-m shielded cables are used between the wireless charging controller and the coupling coil. The wireless charging system is operated at a working frequency of 85 kHz and a working gap of 150 mm, and the conducted disturbance voltage on the DC power line is collected on the EMI receiver.

(a) 3.7kW wireless charging vehicle

Fig. 5.70 Wireless charging experiment vehicle

(b) dashboard information during wireless charging

274

5 Wireless Charging System Electromagnetic Safety …

EMI Receiver 

DC voltage source



LISN

DC power cable

LISN

Wireless charging Controller

Coupler

Battery load

Ground

Fig. 5.71 Test structure diagram of conducted electromagnetic interference for wireless charging system

(2) Conducted emission test According to Fig. 5.71, build a DC-powered wireless charging system conducted emission test platform under the dark room, as shown in Fig. 5.72. According to the test frequency band 150 kHz–108 MH required by Chinese standard GB/T 18655, under the normal working condition of the wireless charging system to charge the battery, the conducted emission test results measured by the EMI receiver are shown in Fig. 5.73. It can be seen that the conducted disturbance voltage exceeds the limit specified by the standard in the frequency range of 500 kHz–2 MHz and 30–108 MHz. The resonance phenomenon occurs when the conduction voltage is 30 MHz, and the high frequency band exceeds the standard is more obvious. The comparison between the conducted disturbance voltage test results and the simulation results is shown in Fig. 5.74. The conducted voltage test results are basically consistent with the simulation results, the second harmonic appears at 170 kHz, and the resonance phenomenon occurs around 30 MHz. The two also exceeded the standard in the 500 kHz–2 MHz and 25–30 MHz bands. However, the difference between the conducted voltages obtained by the experimental method and the simulation method in the 30–108 MHz frequency band is mainly due to the ignorance of some high-frequency parasitic parameters, which needs further research and improvement.

DC source &RQQHFWZLWK/,61

LISN

Load simulator

&RQQHFWZLWKUHFHLYHU

EMI receiver

Wireless charging controller

Power cable

Testing software

Coupler

Fig. 5.72 Conducted emission test platform for wireless charging system

5.5 Modeling and Suppression of Power Line …

275

——Average value of conducted disturbance voltage ——Peak value of conducted disturbance voltage

Fig. 5.73 Conducted emission experiment results

Simulation results

Simulation results

Simulation results

2nd harmonic Resonant frequency

Fig. 5.74 Comparison of conducted emission experiment and simulation results

(3) EMI filter insertion loss test After the CL priority filter is added to the system, the results of the conducted voltage test are shown in Fig. 5.75. The peak and average values of the conducted voltage over the entire frequency band are lower than the limit of Chinese standard GB/T18655, which meets the standard requirements. The suppression frequency of the filter is 30 MHz, and the conducted voltage attenuation is designed to be 21.17 dBμV. As shown in Fig. 5.76, the attenuation of the conducted voltage on the DC power cable of the wireless charging system before and after adding the filter is 21.4 dBμV. The actual attenuation in the experiment is basically the same as the attenuation designed in the simulation, which shows that the proposed CL-first common-mode filter design method is effective.

276

5 Wireless Charging System Electromagnetic Safety … ——Average value of conducted disturbance voltage ——Peak value of conducted disturbance voltage

Fig. 5.75 Conducted voltage test result after adding CL priority filter

The resonant frequency is 30MHz

Before experimental filtering After experimental filtering Standard limit

Fig. 5.76 Comparison of conducted voltage simulation and experimental results

5.5.6 Harmonics of Public Power Supply Lines When the wireless charging system is working, it will generate a lot of harmonics on the public power line, polluting the power grid and affecting the normal operation of the networked electrical equipment. The SAE J2954 standard clearly defines the harmonics of the public power line of the wireless charging system, as shown in Table 5.14.

5.5.7 Harmonic Analysis Because there is a switch-type inverter in the wireless charging system, when the uncontrollable rectifier circuit works, it will turn on and off in turn. In addition, due to the large capacitance existing in the back end, the instantaneous voltage of

5.5 Modeling and Suppression of Power Line …

277

Table 5.14 SAE J2954 Standard harmonic regulations for connecting power grid power lines Odd harmonic n

Maximum allowable harmonic current (A)

Even harmonic n

Maximum allowable harmonic current (A)

3

2.30

2

1.08

5

1.14

4

0.43

7

0.77

6

0.30

9

0.40

8 ≤ n ≤ 40

0.23 * 8/n

11

0.33

13

0.21

15 ≤ n ≤ 39

0.15 * 15/n

the back end large capacitor voltage will also appear When the input voltage of the front-end power grid is higher than that, the uncontrollable rectifier circuit is completely disconnected, which causes severe distortion of the 50 Hz sinusoidal operating current input by the power grid, resulting in distortion and a large amount of current harmonics. The effective value of the total current is I rms , I H1 is the effective current of the fundamental wave, and I Hk is the effective current of the k-th harmonic, then the current in the circuit has the following relationship: I rms =



I H12 + I H22 + I H32 + · · · + I Hk2 + · · ·

(5.44)

I(t) is set to the 50 Hz time-domain change current input by the power grid, then I(t) can be Fourier transformed to obtain the following harmonic relationship with the fundamental frequency of 50 Hz as the following: I (t) =

∝  √ 2I Hn sin(100π nt)

(5.45)

n=1

5.5.8 Harmonics of Traditional Rectifier Filter Booster Circuit System The traditional switching power supply generally uses boost circuit and rectifier large capacitor filtering. The boost circuit does not control the closed-loop power factor correction (PFC) of the switching tube. The working power factor of this circuit is low and the harmonic component is large. It is inefficient and has great pollution to the power grid. The circuit model of the wireless charging system shown in Fig. 5.77 is established in Matlab/Simulink software, that is, the power switch of the booster circuit works under a fixed duty cycle and frequency pulse. Through simulation, the output charging current as shown in Fig. 5.78 and the output charging voltage

278

5 Wireless Charging System Electromagnetic Safety … Inverter control circuit

Determine the duty cycle pulse

C1 LPFC Grid

D5

D6

D7

D8

C2

Lƒ2

L0

Lƒ1

D5 CPFC4

Cƒ1

S5

D1 L1

L2 Cƒ2

D2 C0

D3

D4

Fig. 5.77 Circuit model of traditional PFC-free control circuit system

Fig. 5.78 Charging current charging simulation results Current/(A)

—— current

Time/(s)

as shown in Fig. 5.79 can be obtained, which meet the requirements of the output power of 3.68 kW and the charging voltage of 400 V. The current and voltage of the system input from the power grid are shown in Figs. 5.80 and 5.81. It can be seen that the input voltage of the grid is still a sinusoidal waveform that works normally, but due to the large capacitance of CPFC4, the front-end the full-bridge rectifier diode is not turned on in certain periods, and the sinusoidal current distortion of the grid input appears, resulting in a reduction in power factor. In the case where the output side also needs to reach 3.7 kW, the grid side needs to input a larger current. Due to current distortion, a large number of harmonics are generated. Take the first 15 harmonics for analysis. The simulation results of the 2nd to 15th harmonic currents are shown in Fig. 5.82. The maximum value of each harmonic current is shown in Table 5.15. It can be seen that the even-numbered harmonic currents do not exceed Fig. 5.79 Charging voltage simulation results Voltage/(V)

—— voltage

Time/(s)

5.5 Modeling and Suppression of Power Line …

—Current —Voltage

Current/Voltage(A/V)

Fig. 5.80 Grid input current and voltage

279

Time/(s)

Current/(A)

Fig. 5.81 Grid input current zoom

Time/(s)

the standard limits, but all the odd-numbered harmonic currents of 3, 5, 7, 9, 11, 13, and 15 exceed the standard limits. From the simulation results, it can be seen that the harmonic current of the traditional rectifier filter booster circuit system is large, and the power factor is low, which not only leads to low efficiency, but also pollutes the public power grid.

5.5.9 Harmonics Using PFC Circuit System From the above analysis, we can know that the system using the traditional rectifier large capacitor filter circuit has low power factor, large harmonic component, low efficiency, and great pollution to the power grid, so we use high-frequency PFC closed-loop control circuit for harmonics Suppression, power factor correction and boost. The high-frequency PFC closed-loop control system model circuit is shown in Fig. 5.83. The closed-loop control strategy of the PFC control circuit is shown in Fig. 5.84. The voltage follower PFC control is used, which belongs to the high-frequency active power factor correction technology, mainly based on the PFC module boost target voltage, PFC module actual output voltage, grid input voltage and input current for dual PI control, the control input current follows the voltage sinusoidally, and the phase also follows the voltage change. Using the circuit model of the high-frequency PFC closed-loop control system, the simulation obtained the output charging current and charging voltage shown in

280

5 Wireless Charging System Electromagnetic Safety … —Harmonic current

Current/(s)

Current/(s)

—Harmonic current

Time/(s)

Time/(s)

(a) 2nd harmonic

(b) 3rd harmonic —Harmonic current

Current/(s)

Current/(s)

—Harmonic current

Time/(s)

Time/(s)

(c)4th harmonic

(d)5th harmonic —Harmonic current

Current/(s)

Current/(s)

—Harmonic current

Time/(s)

Time/(s)

(e)6th harmonic

(f)7th harmonic —Harmonic current

Current/(s)

Current/(s)

—Harmonic current

Time/(s)

Time/(s)

(g)8th harmonic

(h)9th harmonic —Harmonic current Current/(s)

Current/(s)

—Harmonic current

Time/(s)

Time/(s)

(i)10th harmonic

(j)11th harmonic —Harmonic current

Current/(s)

Current/(s)

—Harmonic current

Time/(s)

Time/(s)

(k)12th harmonic

(l)13th harmonic —Harmonic current

Current/(s)

Current/(s)

—Harmonic current

Time/(s)

(m)14th harmonic

Fig. 5.82 Simulation results of each harmonic current

Time/(s)

(n)15th harmonic

5.5 Modeling and Suppression of Power Line …

281

Table 5.15 Harmonic current simulation results using traditional rectifier filter boost circuit system Odd harmonic

Actual harmonic current

Maximum allowable harmonic current

Even harmonic

Actual harmonic current

Maximum allowable harmonic current

3

19.40

2.30

2

0.035

1.08

5

13.35

1.14

4

0.024

0.43

7

7.31

0.77

6

0.012

0.30

9

2.92

0.40

8

0.006

0.23

11

1.63

0.33

10

0.005

0.18

13

1.61

0.21

12

0.005

0.15

15

1.04

0.17

14

0.004

0.13

PFC control circuit

Inverter control circuit

C1 LPFC Grid

D5

D6

D7

D8

C2

Lƒ2

L0

Lƒ1

D5 CPFC4

D1

Cƒ1

S5

L1

L2 Cƒ2

D2 C0

D3

D4

Fig. 5.83 High-frequency PFC closed-loop control system circuit model

Vref

Boost target voltage

Gain ÷

V0

Gain

PI

Input target current

+ +

Output voltage

1/311

PI

PWM control signal

Vi

Input voltage

Triangle wave Ii

Input Current

Fig. 5.84 Voltage following control in high-frequency active power factor correction technology

Figs. 5.85 and 5.86, which can be seen to meet the design requirements. At the same time, as shown in the simulation results of the current and voltage input to the system from the public grid as shown in Figs. 5.87 and 5.88, the phase consistency of the current and voltage realizes the correction of the power factor, and the power factor reaches a high power close to 1 Factors to achieve efficiency improvement, and of course harmonic suppression. The harmonic current waveforms from 2nd to 15th are shown in Fig. 5.89. As shown in Table 5.16, it is the maximum value of each harmonic current from 2nd to

282

5 Wireless Charging System Electromagnetic Safety …

Fig. 5.85 Charging current charging simulation results Current/(A)

—Current

Time/(s)

Fig. 5.86 Charging voltage simulation results Voltage/(V)

—Voltage

Time/(s)

Fig. 5.87 Input voltage and current of public grid Current/Voltage(A/V)

—Current —Voltage

—Current —Voltage

Current/Voltage(A/V)

Fig. 5.88 Enlarged view of input voltage and current of public grid

5.5 Modeling and Suppression of Power Line …

283 —Harmonic current

Current/(A)

Current/(A)

—Harmonic current

Time/(s)

Time/(s)

(a)2nd harmonic

(b)3rd harmonic —Harmonic current

Current/(A)

Current/(A)

—Harmonic current

Time/(s)

Time/(s)

(c)4th harmonic

(d)5th harmonic —Harmonic current Current/(A)

Current/(A)

—Harmonic current

Time/(s)

Time/(s)

(e)6th harmonic

(f)7th harmonic

—Harmonic current Current/(A)

Current/(A)

—Harmonic current

Time/(s)

Time/(s)

(g)8th harmonic

(h)9th harmonic

—Harmonic current Current/(A)

Current/(A)

—Harmonic current

Time/(s)

Time/(s)

(i)10th harmonic

(j)11th harmonic —Harmonic current Current/(A)

Current/(A)

—Harmonic current

Time/(s)

Time/(s)

(k)12th harmonic

(l)13th harmonic —Harmonic current Current/(A)

Current/(A)

—Harmonic current

Time/(s)

(m)14th harmonic

Time/(s)

(n)15th harmonic

Fig. 5.89 Simulation results of using PFC closed loop to control each harmonic current

284

5 Wireless Charging System Electromagnetic Safety …

Table 5.16 Harmonic current simulation results using PFC closed-loop control circuit system Odd harmonic

Actual harmonic current

Maximum allowable harmonic current

3 (Exceed standard)

2.50

2.30

5

0.85

7 (Exceed standard)

Even harmonic

Actual harmonic current

Maximum allowable harmonic current

2

0.063

1.08

1.14

4

0.055

0.43

0.78

0.77

6

0.055

0.30

9

0.19

0.40

8

0.042

0.23

11

0.06

0.33

10

0.042

0.18

13 (Exceed standard)

0.25

0.21

12

0.039

0.15

15

0.164

0.17

14

0.034

0.13

15th. It can be seen that compared with the traditional rectifier filter boost circuit, all odd-numbered harmonics have been greatly improved. All odd harmonic currents are significantly reduced. Among them, the third harmonic current is reduced from 19.40 to 2.5 A, the fifth harmonic current is reduced from 13.35 to 0.94 A, and the seventh harmonic current is reduced from 7.31 to 0.78 A. The 9th harmonic current was reduced from 2.92 to 0.19 A, the 11th harmonic current was reduced from 1.61 to 0.06 A, the 13th harmonic current was reduced from 1.61 to 0.25 A, and the 15th harmonic current was reduced from 1.04 to 0.164 A. Although each harmonic has been significantly improved, the currents of the 3rd, 7th and 13th harmonics still slightly exceed the standard limits, and the system needs to be further optimized.

5.5.10 System Harmonic Suppression Method The frequency of the 3th harmonic current is 150 Hz, the current exceeds the standard by 0.2 A (46 dBμA), the frequency of the 7th harmonic current is 350 Hz, the current exceeds the standard by 0.02 A (86 dBμA), the frequency of the 13th harmonic current is 650 Hz, the current exceeds the standard by 0.04 A (92 dBμA). We chose the frequency of the 13th harmonic as an example to design a harmonic suppression filter. Attenuation = actual amplitude-standard value + 6 dBμA (margin). The 13th harmonic is taken as the target attenuation frequency, and the 13th harmonic target is attenuated by 9.1 dBμA with a margin of 6 dBμA, that is, attenuation = 9.1 + 6 = 15.1 dBμA. According to the filter calculation formula, the value of the harmonic LC differential mode filter is obtained, L D1 = L D2 = 340 μH, C X = 500 μF, the structure of the harmonic LC differential mode filter is shown in Fig. 5.90. Add the above-designed filter to the system simulation model to obtain the harmonic optimization of the power supply line connected to the public power grid.

5.5 Modeling and Suppression of Power Line … Fig. 5.90 Structure of a harmonic LC differential mode filter

285

LD1 Public grid

Rectifier module

CX LD2

Other modules of wireless charging system

LC filter

The harmonic simulation results before and after the 2nd to 15th filtering are shown in Fig. 5.91. It can be seen that after the filter device is added, the 3rd, 7th and 13th harmonics that were originally exceeded are significantly suppressed, and the standard limit requirements are met. In addition, the remaining harmonics have been significantly suppressed. According to the simulation results of the above harmonics, the maximum value of each harmonic current as shown in Table 5.17 can be obtained, and it can be seen that the 3rd, 7th and 13th harmonics without the filter suppression measures are exceeded. After adding the designed LC filter, it has been significantly improved. The 3rd harmonic current is reduced from 2.50 A before filtering to 1.50 A, the 7th harmonic is reduced from the original 0.79 A to 0.38 A, and the 13th harmonic is also from 0.25 A before adding the filtering device to 0.056 A, all in line with the limits specified in the standard. The remaining harmonics of the 2nd to 15th orders have been significantly improved compared to before the filter device is added. The most obvious reduction of even harmonics is the 2nd harmonic, which is reduced from 0.065 A before filtering to 0.016 A. The even harmonics of the system also have a very good suppression effect, which reduces the pollution to the public power grid. In addition, the most obvious suppression of odd harmonics is the third harmonic, which is reduced from 2.50 A before filtering to 1.50 A. Greatly suppress the interference of harmonics. It can also be clearly seen from the results summarized in Table 5.17 that all harmonics comply with the limits specified in the standard, which suppresses harmonic interference, improves system performance, and also increases system efficiency and reduces Pollution and interference to the public power grid.

5.6 Electromagnetic Radiation of Secondary Side Circuit of Wireless Charging System 5.6.1 Radiation Emission System Modeling Set up a three-dimensional model of the entire vehicle and a wire harness model of the wireless charging system in the software CST cable studio, as shown in Fig. 5.92a, b is the secondary side wiring harness model established according to the vehicle component radiation emission test requirements, 1.5 m shielded wire, and the electric field probes are placed 1 m on both sides of the harness center point. Co-simulate

286

5 Wireless Charging System Electromagnetic Safety … —Harmonic current before filtering

—Harmonic current before filtering

—Harmonic current after filtering

Current/(A)

Current/(A)

—Harmonic current after filtering

Time/(s)

Time/(s)

(a)2nd harmonic

(b)3rd harmonic —Harmonic current before filtering

—Harmonic current after filtering

—Harmonic current after filtering

Current/(A)

Current/(A)

—Harmonic current before filtering

Time/(s)

Time/(s)

(c)4th harmonic

(d)5th harmonic —Harmonic current before filtering

—Harmonic current after filtering

—Harmonic current after filtering

Current/(A)

Current/(A)

—Harmonic current before filtering

Time/(s)

Time/(s)

(e)6th harmonic

(f)7th harmonic —Harmonic current before filtering

—Harmonic current after filtering

—Harmonic current after filtering

Current/(A)

Current/(A)

—Harmonic current before filtering

Time/(s)

Time/(s)

(g)8th harmonic

(h)9th harmonic —Harmonic current before filtering

—Harmonic current after filtering

—Harmonic current after filtering

Current/(A)

Current/(A)

—Harmonic current before filtering

Time/(s)

Time/(s)

(i)10th harmonic

(j)11th harmonic —Harmonic current before filtering

—Harmonic current after filtering

—Harmonic current after filtering

Current/(A)

Current/(A)

—Harmonic current before filtering

Time/(s)

Time/(s)

(k)12th harmonic

(l)13th harmonic —Harmonic current before filtering

—Harmonic current after filtering

—Harmonic current after filtering

Current/(A)

Current/(A)

—Harmonic current before filtering

Time/(s)

(m)14th harmonic

Fig. 5.91 Harmonics before and after filtering

Time/(s)

(n)15th harmonic

5.6 Electromagnetic Radiation of Secondary Side …

287

Table 5.17 Simulation results of harmonic current before and after filtering using PFC closed-loop control circuit system Harmonic order

Harmonic current before filtering

Whether it exceeds the standard

Harmonic current after filtering

Maximum allowable harmonic current

Whether it exceeds the standard

3

2.50

Y

1.50

2.30

N

5

0.94

N

0.56

1.14

N

7

0.79

Y

0.38

0.77

N

9

0.19

N

0.018

0.40

N

11

0.06

N

0.18

0.33

N

13

0.25

Y

0.056

0.21

N

15

0.17

N

0.08

0.17

N

2

0.065

N

0.016

1.08

N

4

0.05

N

0.018

0.43

N

6

0.06

N

0.01

0.30

N

8

0.05

N

0.008

0.23

N

10

0.05

N

0.006

0.18

N

12

0.041

N

0.006

0.15

N

14

0.028

N

0.005

0.13

N

Fig. 5.92 Wireless charging system model and wiring harness simulation model

the above model with CST design circuit design studio, as shown in Fig. 5.93. The secondary positive and negative cable conduction voltages extracted by the conductive emission simulation are injected into the port 1 and the port 2 of the circuit respectively to simulate and obtain the secondary side harness radiation emission of the vehicle system.

288

5 Wireless Charging System Electromagnetic Safety …

Fig. 5.93 Wireless charging system co-simulation signal excitation circuit

5.6.2 Radiated Emission from Secondary Charging Cable Figure 5.94 shows the frequency domain distribution of the electric field 1 m away from the simulated harness, including the X, Y, Z and composite vector electric field values. The electric field in the X direction>the electric field in the Y direction>the electric field in the Z direction, and the amplitude of the electric field intensity in the X direction is closest to the composite electric field intensity amplitude. The electric field strength reached a peak value of 48.27 dBμV/m at 29.7 MHz, exceeding the standard limit of 44 dBμV/m. After that, the electric field strength showed a downward trend. Therefore, it is necessary to suppress the radiated emission at the frequency of 29.7 MHz. As shown in Fig. 5.95, the field strength distribution at 1, 10, 30 and 100 MHz is analyzed from the electric field distribution cloud diagram. At low frequencies, the electric field is mainly concentrated at the wire harness port, and the electric field at 1 MHz and 10 MHz does not change significantly. When the frequency is up to 30 MHz, the amplitude of the electric field strength increases significantly. The electric field is not only distributed in the harness port, but also the space around the harness has increased. But at 100 MHz, the electric field strength is significantly

Fig. 5.94 Frequency domain distribution of electric field

5.6 Electromagnetic Radiation of Secondary Side …

289

Fig. 5.95 Cloud diagram of electric field distribution

reduced. The change law of the electric field distribution cloud diagram is basically consistent with the electric field distribution characteristics shown by the frequency domain curve. Cable space radiation can be explained by the gain and directivity of the antenna. Figure 5.96 shows the far-field graph of radiated emissions at frequencies of 1, 10, 30, and 100 MHz monitored by the far-field detector. It can be seen that at 1 and 10 MHz, the antenna distribution is approximately spherical, indicating that the radiation emissions generated by the cable are distributed in all directions, and the distribution density is basically the same. At 30 MHz, the antenna gain is maximum.

5.6.3 Radiated EMI Suppression For the problem of radiated EMI caused by the conducted disturbance voltage on the secondary wiring harness, the method of suppressing radiated EMI can be to shield the radiated cable, and the fundamentally effective method is to suppress the signal of the radiated interference source. Therefore, this section adopts the designed common-mode filter for radiated EMI suppression, and uses the filter to reduce the conducted disturbance voltage of the secondary cable, thereby reducing EMI caused by radiated emissions.

290

5 Wireless Charging System Electromagnetic Safety …

Fig. 5.96 Far-field characteristics of radiated emissions

(1) Interference source signal Build a high-frequency circuit model with a filter circuit in the CST design studio. The green element represents the LISN circuit added to the secondary coil to collect the conducted disturbance voltage signal. The CL filter indicated by the red symbol. The simulation model of the high-frequency circuit of the wireless charging system with the added filter is shown in Fig. 5.97. Figure 5.98 shows the positive conducted disturbance voltage and the negative conducted disturbance voltage of the secondary side power line, respectively. It can be seen from the figure that after adding the filter, the resonance disturbance of the frequency domain curve of the conducted disturbance voltage no longer occurs. The maximum value of the positive and negative conduction voltages is significantly reduced from 80 dBμV/m without suppression to 40 dBμV/m.

5.6 Electromagnetic Radiation of Secondary Side … Lf1 RLf1 LC1 C1 LCable1

UCM

LCable2

L1

L2

Cƒ1

RL1

RL2

RLf3 Lf3

D1 D2LCable3

P1

R1

C8

C10 LCf1

C2 LC2 Lf2

RCable2

RCable1

IDM

291

RLf2 Cƒ2

RCable3

C9 R2

LCf2

D3 D4

P2

C11

C12

C13

Fig. 5.97 High-frequency circuit model of WPT system with filter

Fig. 5.98 Conducted disturbance voltage on the charging line on the secondary side of the wireless charging system

292

5 Wireless Charging System Electromagnetic Safety …

Fig. 5.99 Electric field frequency domain curve

(2) Radiated emission from the charging line on the secondary side of the filter As shown in Fig. 5.99, the electric field spectrum distribution at 1 m of the secondary cable after the filter is added to the system. The peak strength of the electric field at 29.2 MHz is 42.3 dBμV/m, which meets the standard limit. The electric field strength drops almost 6 dBμV/m over the entire frequency band. From the analysis of frequency domain characteristics, the addition of conducted interference filters can effectively suppress the cable radiated emissions. The electric field distribution cloud diagram of the 30 MHz frequency point is shown in Fig. 5.100. The maximum value of the electric field strength is 58.2 dBμV/m, which is 3.7 dBμV/m lower than the amplitude of the electric field strength without filter of 61.9 dBμV/m. After adding the filter, the far-field diagram of the radiated emission at the frequency of 30 MHz monitored by the far-field detector is shown in Fig. 5.101. It can be seen that the directivity of the antenna is basically unchanged, and the maximum value of the antenna gain is 6.54 dBi, which is 0.95 dBi less than before the filter is added. Fig. 5.100 Far-field characteristics of 30 MHz radiated emissions

5.6 Electromagnetic Radiation of Secondary Side …

293

Fig. 5.101 Far-field characteristics of 30 MHz radiated emissions

In summary, from the radiated emission frequency domain distribution curve, 30 MHz frequency field electric field distribution cloud diagram and far-field spatial radiation, it can be seen that the addition of the filter has a good suppression effect on the radiated emission of the power line of the secondary side of the wireless charging system.

References Esteban B, Sid A (2015) A comparative study of power supply architectures in wireless EV charging systems. IEEE Trans Power Electron 30(11):6408–6422 Suh I-S (2018) Wireless charging technology and the future of electric transportation (trans Zhai L). Beijing Institute of Technology Press, Beijing, pp 45–49

Chapter 6

Signal Integrity and Electromagnetic Compatibility of Vehicle Controller

6.1 Overview The vehicle controller is a key component of the electric vehicle, which manages and controls various high and low voltage electronic control equipment on the vehicle. Signal integrity and electromagnetic compatibility are important performance indicators that ensure the safe and reliable operation of the vehicle controller. With the application of Ethernet and Internet of Vehicles in intelligent connected electric vehicles, it has brought about an increase in communication speed, and it has put forward new requirements for the design of the electronic and electrical architecture, power integrity, signal integrity and electromagnetic compatibility of the vehicle controller. This section mainly describes the function and structure of the vehicle controller, hardware electromagnetic compatibility design, including electromagnetic emission and electromagnetic sensitivity design; power distribution network (PDN) equivalent circuit modeling, and decoupling capacitor optimization method; Ethernetbased PCB board signal integrity modeling and analysis; PCB electromagnetic emission, and power line conducted disturbance suppression method for Ethernet vehicle controller.

6.2 Function and Structure of Vehicle Controller Specific features include: (1) Vehicle status and driving information collection, system information display Collect vehicle status and driving information to determine the working status of the vehicle controller. (2) Network management The vehicle integrated controller is a node of the vehicle network. It realizes the communication between the sub-controllers through the CAN bus or vehicle © China Machine Press 2021 L. Zhai, Electromagnetic Compatibility of Electric Vehicle, Key Technologies on New Energy Vehicles, https://doi.org/10.1007/978-981-33-6165-2_6

295

296

6 Signal Integrity and Electromagnetic Compatibility …

Ethernet. It is responsible for organizing and transmitting information, network status monitoring, fault diagnosis and processing. (3) Driver’s driving intention information collection According to the collected accelerator pedal information, brake pedal information, steering wheel information, gear switch information, etc., the vehicle controller can combine the state of charge of the power battery to determine the driving torque and power, braking torque and recovered power, whether the power battery needs to be charged, etc. (4) System coordination control and energy management The vehicle controller obtains the highest energy utilization rate through the coordinated management of the motor drive system, battery management system, and other vehicle energy consumption systems such as automobile instruments, headlights, and air conditioners. (5) Fault diagnosis and treatment The vehicle controller can monitor its own faults, CAN bus or Ethernet status, and faults of various connected units on the bus. At the protocol layer, the fault content is classified and given different priorities. The system functional structure of the vehicle controller is shown in Fig. 6.1. The main input of the vehicle controller should include: Ethernet signal, switch signal, analog signal, logic pulse signal, CAN bus signal, etc. The switching signals include parking gear signal (P), reverse gear signal (R), forward gear signal (D), low speed gear signal (L), high speed gear signal (H), charge switch signal, brake Pedal

Hardware reset circuit

Clock circuit

MCU external storage EEPROM Air conditioning relay Water pump relay

Relay drive

Solenoid valve drive

PWM output

Host computer Motor Controller Battery Management System ABS/ESP Other ECU

Power management circuit

MCU XC2268I136FxxL

Solenoid valve drive

CAN transceiver TJA1042T

Ethernet interface circuit

Fig. 6.1 Integrated controller circuit module structure

Signal following conditioning circuit

acceleration pedal Brake pedal Steering wheel angle

Digital signal conditioning circuit

Remote control mode switch Remote control signal Key signal Air conditioner switch

CAN transceiver TJA1042T

Air conditioning instrument

6.2 Function and Structure of Vehicle Controller

297

switch signal, accelerator pedal switch signal, etc. The analog signal is mainly the signal of each sensor, mainly the accelerator pedal sensor, brake pedal sensor and steering wheel angle sensor. These three sensors output continuous voltage signals to the vehicle controller. The vehicle controller calculates the pedal opening according to the signal level, judges the driver’s intention to accelerate or brake, and controls the motor through the motor driver. Logic pulse signals include: key signal, high-grade switching signal, low-gear switching signal, etc. The vehicle output signals of the vehicle controller include: Ethernet signal, CAN bus signal, and execution command output.

6.3 Hardware EMC Design of Vehicle Controller The vehicle controller (VCU), as the core component of the vehicle, is not only affected by the conduction and radiation interference generated by the on-board electronic equipment, but also the high-speed digital chip, DC-DC module, crystal oscillator and the PWM module on the PCB inside the VCU are also sources of electromagnetic interference. The electromagnetic noise generated by the internal interference source of the VCU not only affects the power supply integrity and signal integrity of the PCB, such as CAN bus signals, Ethernet bus signals, highspeed data bus signals, power supply signals, etc., but also forms conduction and radiation electromagnetic interference through the space and wiring harness. It has an impact on on-board equipment, neighboring vehicles and off-vehicle receivers.

6.3.1 Power Line Electromagnetic Immunity (1) Immunity to electrical transient disturbances along the power line The positive pole of the power supply of the vehicle low-voltage equipment is connected together, and the negative pole is connected together with a ground. Transient pulses will be generated along the power line when the vehicle low-voltage equipment is working, and can be transmitted to the vehicle controller through the power cable. GB/T21437.2 (ISO7637.2) “Road Vehicles-Electrical Disturbance Caused by Conduction and Coupling Part 2: Electrical Transient Conduction along Power Lines” specifies the test methods for the immunity of conducted interference of road vehicle components, Fig. 6.2 shows the transient pulse signal specified in GB/T21437.2. (2) Electrostatic discharge immunity along the power line Static electricity is formed due to the accumulation of airborne charge or the contact of the body with the tip of the conductor in the car. A large amount of charge enters the PCB circuit board through the power connector terminal of the VCU.

298

6 Signal Integrity and Electromagnetic Compatibility …

Test pulse P1

Test pulse P2 (a)

Test pulse P2(b)

Test pulse P3 (a)

Test pulse P4

Test pulse P3 (b)

Test pulse P5 (a)

Test pulse P5 (b)

Fig. 6.2 The pulse signal specified in GB/T21437.2

Electrostatic discharge belongs to high-voltage rapid pulse interference. The standard GB/T 19951-2005 (ISO 10605:2001) “Test Method for Electrical Disturbance Generated by Electrostatic Discharge of Road Vehicles” specifies the voltage levels that vehicle electronic modules should withstand when energized and not energized.. As shown in Table 6.1. The electrostatic discharge simulation waveform is shown in Fig. 6.3. The rising edge of the discharge pulse is on the order of several ns. (3) Surge impact along the power line and immunity of electrical fast pulse groups Unipolar surges (shocks) caused by transient overvoltages caused by lightning and switching on and off and electrical fast pulse group interference generated by Table 6.1 Electrostatic discharge voltage level Discharge type

Module status

Severity level (kV) I

II

III

IV

Contact discharge

Power

±4

±6

±7

±8

Air discharge

Power

±4

±8

±14

±15

Contact discharge

No Power

±4

±6

±8

Air discharge

No Power

±4

±15

±25

6.3 Hardware EMC Design of Vehicle Controller

299

Fig. 6.3 Electrostatic discharge waveform

the mechanical switch during the switching process (cutting off the inductive load, relay contact bounce, etc.) will cause serious interference to the VCU. (4) Design of power line interference suppression circuit In summary, the electromagnetic interference along the VCU power line can be divided into three categories: positive high voltage pulse, reverse pulse, and positive high current. Corresponding to the interference caused by the starting system, alternator, load switching, switching jitter and load dump, anti-reverse circuit, pulse interference discharge circuit, and electrostatic discharge protection circuit are designed. (a) Electrostatic capacitance: Use two electrostatic capacitors of 1 μF/200 V in series on the power line to prevent short circuit after one capacitor fails; use a cross layout method on the layout to prevent short circuit damage caused by stress. (b) Transient Voltage Suppressor (TVS): A transient voltage suppressor (TVS) is installed between the anode and cathode of the power input port to suppress all kinds of transient interference pulses, clamping the high voltage transient pulse interference to the acceptable range of the PCB power system. (c) Anti-reverse diode: Sudden disconnection of the inductive load connected in parallel on the VCU power line will produce a large reverse pulse voltage. In addition to the bidirectional TVS absorption, a rectifier diode is used as an antireverse circuit. For example, the voltage stabilizer diode 1N4007 is selected, and its maximum reverse withstand voltage is 1000 V. (d) Common mode choke: The common-mode current on the PCB is easy to form conducted and radiated emissions. In order to suppress the common-mode current on the power line, a common-mode choke is used, and its withstand voltage is 500 V. (e) Filter capacitor: Install an absorption capacitor between the positive and negative poles of the PCB power line to suppress transient pulse groups, using an electrolytic capacitor and two decoupling capacitors. In summary, the power line interference suppression circuit is shown in Fig. 6.4.

300

6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.4 Power line conducted interference suppression circuit

6.3.2 PCB Power Integrity PCB power integrity mainly includes power supply division and decoupling capacitor design. (1) Power supply division The internal power supply of the VCU includes four parts: +5 V logic power supply, +3.3 V logic power supply, +5 V CAN bus chip power supply, and +5 V analog power supply. The power supply and ground of each part must be arranged separately, otherwise, the noise on one power supply area will easily propagate to the other power supply area, and will eventually be grounded in a single-point ground. Use the isolated DC-DC chip to convert the +5 V logic power/ground to the isolated analog chip and CAN bus chip power/ground, as shown in Figs. 6.5 and 6.6, and use an electrolytic capacitor of 10 μF and a chip capacitor of 0.1 μF on the power output pin of the DC-DC chip for basic power network decoupling. The PCB four-layer board uses a power plane and a ground plane to power the chip. Therefore, in addition to using an isolated DC-DC chip to provide separate power/ground for the CAN bus chip and analog chip. It also divides the power plane of logic 5 V, CAN bus chip 5 V, analog signal 5 V, and 3.3 V. The division of the ground plane and the power plane is similar. When the plane is divided, the corner of the edge adopts 45° to avoid the formation of impedance discontinuity at the tip of the plane, which will aggravate the electromagnetic radiation. Figure 6.7 shows the power supply and ground plane segmentation.

(a) Analog power/ground isolation circuit Fig. 6.5 Power isolation circuit

(b) CAN bus power/ground isolation circuit

6.3 Hardware EMC Design of Vehicle Controller

(a) Analog power/ground

301

(b) CAN bus chip power/ground

Fig. 6.6 CAN bus chip power/ground isolation circuit

(a) Stratigraphic division and routing

(b) Power layer division and routing

Fig. 6.7 PCB power/ground plane segmentation

(2) Decoupling capacitor PCB board-level synchronous switching noise and excessive power channel impedance will cause power integrity problems (such as excessive voltage drop, electromagnetic radiation, heat generation, and large energy loss), which will seriously affect the stability of the device and system operation. In the early stage of design, decoupling capacitor design is required for each logic chip. The +5 V logic chip power supply supplies power to the main control chip, the EEPROM external storage chip, and the digital signal acquisition chip. Decoupling capacitors are designed for

302

6 Signal Integrity and Electromagnetic Compatibility …

each power pin of each chip, as shown in Figs. 6.8, 6.9 and 6.10. Each capacitor is installed as close as possible to the chip power pin. Similarly, the decoupling capacitors are designed for the 3.3 V-powered Ethernet interface chip, CAN bus transceiver chip, CAN isolation chip, and the signal following chip in the analog signal processing circuit, as shown in Figs. 6.11, 6.12, 6.13 and 6.14.

Fig. 6.8 Decoupling capacitor of main control chip

Fig. 6.9 Decoupling capacitor of EEPROM external memory chip

6.3 Hardware EMC Design of Vehicle Controller

Fig. 6.10 Decoupling capacitor of digital signal acquisition chip Fig. 6.11 CAN bus transceiver chip decoupling

Fig. 6.12 Decoupling capacitor of CAN bus capacitor isolation chip

303

304

6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.13 Decoupling capacitor of Ethernet

Fig. 6.14 Decoupling capacitor of Analog voltage interface chip follows chip

6.3 Hardware EMC Design of Vehicle Controller

305

6.3.3 Signal Integrity Design When the PCB traces are not continuous, the impedance changes, causing signal reflection and delay. When the signal trace spacing and trace direction are not properly controlled, both will cause crosstalk coupling between adjacent signal lines or adjacent networks. Signal integrity design mainly includes signal integrity problems brought by Ethernet and CAN bus communication. (1) Signal integrity of Ethernet For the power supply of the Ethernet chip, a linear voltage regulator chip is used to step down the 5 V power supply to obtain a 3.3 V power supply, and the noise is suppressed by ferrite beads, as shown in Fig. 6.15. The differential impedance of Ethernet is 100 , and the impedance of the differential line can be controlled by adjusting the wiring width, line spacing, and the thickness of each layer of the board. In order to resist the interference of high voltage pulses such as static electricity, the TVS array is used to protect the signal lines, as shown in Fig. 6.16. Ethernet differential pair traces are designed with equal lengths and equal spacing, as shown in Fig. 6.17.

Fig. 6.15 Using ferrite beads to suppress power noise of Ethernet

Fig. 6.16 Anti-static measures for Ethernet signal lines

306

6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.17 Design of equal length and equal spacing of Ethernet differential lines

(2) CAN bus electromagnetic compatibility design In the transmission range of 1–2 km, the characteristic impedance of the shielded twisted pair with a standard 1.5 square millimeter wire diameter is 120 , that is, the characteristic impedance of the CAN bus cable is 120 . In order to prevent the differential signal from encountering abrupt impedance changes and reflection at the end of the trace, connect a 120  resistor at the farthest end of the bus, as shown in Fig. 6.18a. Install the resistor as close to the CAN bus transceiver as possible, as shown in Fig. 6.19. In addition to impedance matching, the differential pair of the CAN bus may carry a common-mode current, forming a radiated or conductive emission source, and using a common-mode inductance to suppress the common-mode current, as shown in Fig. 6.18(b). In order to reduce the asymmetry of the CAN bus differential signal line and reduce the common mode noise, the form of a serpentine trace is used to ensure that the differential pair is equal in length, as shown in Fig. 6.20. In order to avoid crosstalk between adjacent signal lines, the minimum line spacing between digital

(a) Termination resistance of CAN bus

(b) Common mode inductance of CAN bus

Fig. 6.18 Terminal matching and common mode inductance of CAN bus

6.3 Hardware EMC Design of Vehicle Controller

Fig. 6.19 PCB diagram of CAN bus interface circuit

Fig. 6.20 CAN bus differential pair serpentine routing

307

308

6 Signal Integrity and Electromagnetic Compatibility …

(a) PCB top trace

(b) PCB bottom trace

Fig. 6.21 The top and bottom PCB traces are perpendicular to each other

signal traces is limited to 3 times the line width, and the minimum line spacing between digital and analog signal traces is 5 times the line width. In order to avoid the noise coupling caused by the planar return path shared by the top and bottom signals, the top layer is mainly horizontal traces, and the bottom layer is mainly vertical traces. As shown in Fig. 6.21.

6.4 Power Distribution Network Equivalent Circuit In order to study the influence of the optimization of the decoupling capacitor on the impedance of the power distribution network (PDN), an equivalent circuit model of the PDN is established to analyze the circuit parameters. PCB power distribution network (PDN) frequency band is divided into several parts. In the low frequency band, the power supply noise is mainly filtered by the power conversion chip VRM. In the frequency band of several MHz to several hundred MHz, the power supply noise is mainly filtered by the board-level discrete capacitor and the power ground plane pair of the PCB. In the high-frequency part, the power supply noise is mainly filtered by the PCB power ground plane pair and the high-frequency capacitor inside the chip. The simulation accuracy of the low-frequency and high-frequency parts is still inaccurate. The really meaningful frequency band is mainly in the frequency band of a few MHz to a few hundred MHz.

6.4.1 Equivalent Circuit of Chip Power Distribution Network A typical power distribution network is shown in Fig. 6.22. The PDN consists of VRM, decoupling capacitors (including body capacitance and discrete surface mount

6.4 Power Distribution Network Equivalent Circuit

309

Fig. 6.22 Typical power distribution network

capacitors), power/ground plane, package pad, package power/ground plane, on-chip capacitance, diffusion capacitance and diffusion inductance. The function of PDN is to maintain the low impedance of the power supply channel in each frequency band. In the frequency domain, the PDN action interval is usually divided into 5 parts: DC-10 kHz, 10 KHz–100 kHz, 100 kHz–100 MHz, 100 MHz–1 GHz, 1 GHz and above. Taking the 3.3 V power supply network of the Ethernet interface circuit shown in Fig. 6.23 as the research object, the equivalent circuit modeling of the power distribution network is carried out. The DC-10 kHz range is the VRM action range, VRM determines the impedance value seen from the chip to the PDN, and the VRM equivalent series resistance determines the lower limit of the entire PDN impedance. In the frequency range of 10 kHz–100 kHz, the VRM cannot maintain a rapidly changing current. It is necessary to add a body capacitor to maintain the rapid response of the power supply. The use of electrolytic capacitors or tantalum capacitors in this range can keep the PDN low impedance, but due to the large parasitic inductance of the body capacitor itself, The self-resonance frequency of the body capacitor is limited, therefore, the decoupling range of the body capacitor is 10 kHz–100 kHz. Figure 6.24 shows the impedance frequency distribution of a electrolytic capacitor of 470 μF. The self-resonant frequency is about 76 kHz. The capacitance of the capacitor can be maintained below 76 kHz to achieve the decoupling effect. Capacitors above 76 kHz begin to show inductive, affecting the decoupling effect. Power/ground planes and multilayer ceramic chip capacitors have a frequency range of 100 kHz–100 MHz. Like the body capacitor, the decoupling range of the chip capacitor is also affected by its parasitic inductance and installation inductance. In the 100 MHz–1 GHz frequency range, the impedance of the PDN is mainly affected by the power/ground plane and the parasitic capacitance of the chip package. The minimum impedance of PDN above 1 GHz is determined by the package parasitic parameters and the on-chip capacitance and parasitic parameters of the chip. The parasitic inductance at the package level is usually inductive at high frequencies and becomes the high impedance path of the PDN. Therefore, the frequency of the package determines the upper frequency limit of the PDN design, which is usually 1 GHz. The frequency range of 10 kHz–1 GH is the board-level PDN design range,

310

6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.23 Ethernet interface chip and 3.3 V power network

Fig. 6.24 Impedance curve of a 470 μF electrolytic capacitor

6.4 Power Distribution Network Equivalent Circuit

311

Fig. 6.25 Lumped circuit model of 3.3 V power distribution network

that is, the frequency design range of the body capacitor, chip decoupling capacitor, power supply/ground plane. Through the above analysis, combined with the PCB power distribution network structure of the PCB drawn in Cadence software, the lumped circuit model of the VCU Ethernet 3.3 V power distribution network is established as shown in Fig. 6.25. Next, each part of the model is modeled, and some of the parameters are directly extracted using Q3d Extractor software. (1) VRM model The VRM used by VCU is usually a DC-DC converter. The input voltage is 12 V or 24 V, and the output voltage is usually 5 V and 3.3 V. VRM power supply bandwidth is 1 kHz to several hundred kHz. When the current frequency is higher than this bandwidth frequency, VRM presents high impedance to the current component of this frequency. The equivalent circuit model is shown in Fig. 6.26. There is a voltage adjustment inductor inside the VRM to cope with the impact of large current changes. The installation inductance will occur during installation. The VRM series inductance Lvrm is composed of the above two parts. The internal voltage adjustment inductance Usually it is tens of nH, here the internal voltage regulation inductance is 30 nH, and the installation inductance is 2 nH. The internal resistance Rvrm of VRM is about 1 m. Fig. 6.26 VRM equivalent circuit model Lvrm

VRM Rvrm

Load

312

6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.27 VRM equivalent circuit model frequency domain impedance

According to Fig. 6.26, modeling and simulation in the CST software, the frequency domain distribution of the impedance curve of the VRM model is obtained, as shown in Fig. 6.27. The minimum impedance of VRM is internal resistance 1. Due to the presence of inductance, the impedance of VRM in the frequency band above 1 MHz has exceeded 0.2. With the increase of frequency, VRM can no longer maintain the low impedance of PDN. Therefore, VRM cannot be used for PDN decoupling alone. (2) Via model Vias in the PCB can provide more convenience for layer change and wiring, but the vias themselves contain parasitic parameters, which may cause resonance when high-frequency signals or high-frequency noise pass through the vias. A via can be equivalent to a series connection of an inductor and two capacitors, as shown in Fig. 6.28. Each via has a parasitic capacitance to ground. If the diameter of the via is known as d, the diameter of the via pad is D, the length of the via is h, and the relative dielectric constant of the board substrate is εr, then the size of the via parasitic capacitance C is: Lvia

Fig. 6.28 Via equivalent circuit model

Cvia1

Cvia2

6.4 Power Distribution Network Equivalent Circuit

C=

1.41 · εr hd D−d

313

(6.1)

The unit of length is inch and the unit of capacitance is pF. The parasitic capacitance of the via has a certain decoupling effect, and the operating frequency is within a few hundred MHz. As the diameter of the pad and the diameter D/d of the via decrease, the equivalent parasitic capacitance of the via will increase;As the length of the via increases, the parasitic capacitance will increase. The parasitic inductance of the via is approximated to:   4h L = 5.08h lg( ) + 1 d

(6.2)

The unit of inductance is nH. The outer diameter of the via pad has no effect on the parasitic inductance, and as the via hole diameter increases, the parasitic inductance will decrease; as the via depth h increases, the parasitic inductance will increase. Figure 6.29 shows the frequency domain impedance curve of a 10 mil diameter via (parasitic capacitance of 0.35 pF, parasitic inductance of 0.77 nH). The via’s parasitic capacitance has little effect on suppressing the impedance of the via itself. The impedance of the via has reached close to 1  at 100 MHz. As the frequency increases, the impedance of the via is higher. The way to suppress this effect is not to use vias, or to use large-diameter vias, as shown in Fig. 6.30. (3) Decoupling capacitor model There are many types of decoupling capacitors. VCUs usually use ceramic chip capacitors, aluminum electrolytic capacitors, and tantalum electrolytic capacitors for PDN decoupling. The materials of decoupling capacitors are different. Unlike ideal capacitors, the actual equivalent circuit model of the capacitor is a first-order model

Fig. 6.29 Frequency domain impedance curve of 10 mil vias

314

6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.30 Impedance comparison of large-aperturevias

Fig. 6.31 RLC equivalent model of capacitance

ESLdecap

Cdecap

ESRdecap

composed of a capacitor C, an equivalent series inductance ESL, and an equivalent series resistance ESR, as shown in Fig. 6.31. An LC resonance circuit is formed, and its impedance characteristic can be expressed by Eq. 6.3. In the case of the same capacitance at the same frequency, as the ESL increases, the impedance of the capacitor increases and cannot provide a low impedance path for PDN; when the ESL and capacitance are fixed, and the inductance and capacitance are equal, the impedance decreases until the minimum value of ESR, the frequency f at this time is the self-resonant frequency fSRF. Z = E S R + j2π f · E S L +

1 1 = E S R + j (2π f · E S L − ) j2π f · C 2π f · C (6.3)

fSRF =

2π

√

1

(6.4)

E SL · C

The existence of the ESR of the decoupling capacitor is due to the parasitic resistance of the capacitor itself, and the size is generally between a few milliohms Fig. 6.33 Approximate equivalent circuit model of power/ground plane

L(f) R

R(f) C(f)

6.4 Power Distribution Network Equivalent Circuit

315

and hundreds of milliohms. The ESR of a capacitor is related to the multilayer structure of the capacitor. For ceramic capacitors, ESR can be simply estimated by Eq. 6.5. ESR ≈

180mΩ 2.5lg C

(6.5)

In the formula, C represents the capacitance of the capacitor (unit is nF), and ESR is the equivalent series resistance of the capacitor (unit is m). When the capacitor is actually used, the inductance in series with the ideal capacitor includes the parasitic inductance Lcap and the mounting inductance Lmount. Therefore, the total series inductance of the capacitor is E S L = L cap + L mount

(6.6)

For the mounting structure of the BGA packaged chip decoupling capacitor, it consists of the following four parts: 1. 2. 3. 4.

Surface trace and loop inductance on top of planar cavity; Loop inductance of several vias from the capacitor pad to the cavity plane; Diffusion inductance from capacitor via to BGA via; Loop inductance from the planar cavity under the package to the package pins or solder balls.

However, for SMD ICs, the pads are SMD pads, and BGA is not used, so the mounting inductance Lmount only includes capacitor traces, via inductance, and the loop inductance between the ground plane and the power plane between the capacitor and the load chip. (4) Magnetic bead modeling The magnetic bead can be used as a low-pass filter to filter out high-frequency noise on the power traces at the output port of the VRM, and electromagnetic noise is dissipated in the form of thermal energy. The equivalent circuit model of the magnetic bead consists of four parasitic parameters (see Fig. 6.32). (5) Power/ground plane modeling The power/ground plane is a necessary intermediate layer in the PCB with more than four layers. It is represented as a high-frequency capacitor in the PDN and Fig. 6.32 Magnetic bead equivalent circuit model

Rp Cp

316

6 Signal Integrity and Electromagnetic Compatibility …

provides a low-impedance path in the middle and high frequency bands. The packaged power/ground plane mainly affects the mid-frequency and high-frequency noise. Although the plane can greatly improve the performance of the PDN, when an electromagnetic resonance cavity appears on the plane, a new source of interference will be generated under the excitation. Planar modeling methods are divided into lumped modeling and distributed modeling methods, of which lumped modeling methods mainly include PEEC method and local inductance resistance method. Distributed modeling includes finite element method, finite difference method, finite time domain difference method, transmission matrix method, Green’s function method and other modeling methods. Here, a simplified lumped model is used. The model equivalent plane capacitance shown in Fig. 6.33 is usually used within a few hundred MHz. Where the plane capacitance and plane resistance are A h W R p = 0.0026 × l

C p = 0.225 × εr ×

(6.7)

where εr is the dielectric constant of the dielectric material, A is the area of the plane, the unit is inch2, h is the thickness of the medium between the planes, the unit is mil, W is the width of the plane, the unit is mil, and l is the length of the plane, the unit is mil. (6) PDN lumped circuit modeling Based on the above analysis and calculation, the parameters of the PDN lumped circuit model shown in Fig. 6.24 are shown in Table 6.2. The equivalent circuit model of the lumped parameter model of the Ethernet chip power distribution network is established in the ADS simulation software, which is used for impedance simulation analysis and optimization of decoupling capacitors.

6.4.2 Lumped Parameter Model of Power Distribution Network for Ethernet Chip Through the electric network analysis method, the lumped parameter model of the power distribution network of the Ethernet chip is analyzed to obtain the scattering parameter (Scatter Parameter, S parameter) and the impedance parameter (Z Parameter, Z parameter) of the two-port network model to realize the impedance analysis demand. At present, the method of adding decoupling capacitors based on the peak frequency of impedance is often used to optimize the PDN impedance. This method will introduce new parasitic parameters, generate anti-resonance loops, and form new impedance spikes. Therefore, when designing the PDN impedance optimization method, not only the installation position of the decoupling capacitor but also

6.4 Power Distribution Network Equivalent Circuit

317

Table 6.2 Parameter values in the equivalent circuit model Parameter name Parameter meaning

Parameter value Unit

Rvrm

VRM output resistance

1

m

Lvrm

VRM output inductance

32

nH

ESL Al1

Equivalent series inductance of aluminum electrolytic 1.79 capacitor 1

nH

ESL Al2

Equivalent series inductance of aluminum electrolytic 0.786 capacitor 2

nH

C Al1

Capacitance value of aluminum electrolytic capacitor 100 1

nF

C Al2

Capacitance value of aluminum electrolytic capacitor 10 2

μF

ESRAl1

Equivalent series resistance of aluminum electrolytic capacitor 1

0.06

m

ESRAl2

Equivalent series resistance of aluminum electrolytic capacitor 2

0.004

m

Rbead

Fixed resistance of magnetic beads

0.045



Rbead (f )

Frequency domain resistance of magnetic beads (100 MHz)

121



L bead (f )

Frequency domain inductance of magnetic beads (100 MHz)

133

nH

C bead (f )

Frequency domain capacitance of magnetic beads (100 MHz)

10

pF

L via

Via parasitic inductance

0.59

nH

C via1

Via parasitic capacitance 1

0.53

pF

C via2

Via parasitic capacitance 2

0.53

pF

Rplane

Plane resistance

1

m

C plane

Planar capacitance

3

nF

ESL decap

Decoupling capacitor equivalent series inductance

1.84

nH

C decap

Decoupling capacitance

100

nF

ESRdecap

Decoupling capacitor equivalent series resistance

0.1

m

Rspread

Loop diffusion resistance

1

m

L spread

Loop diffusion inductance

1

nH

the influence of the parasitic parameters of the installation decoupling capacitor must be considered. As shown in Fig. 6.34, a two-port network model of the Ethernet lumped power distribution network is established in the ADS simulation software. S-parameter and Z-parameter simulation results are shown in Fig. 6.35. Z21 has an anti-resonance point at 1 MHz frequency, reaching the lowest impedance value in the range of 100 kHz–1 GHz; above 1 MHz, as the frequency increases, there is a small resonance peak between 3-4 MHz;then the impedance increases with increasing frequency,

318

6 Signal Integrity and Electromagnetic Compatibility …

Z21 magnitude /Ω

S21 magnitude/dB

Fig. 6.34 The 3.3 V power distribution network model established in the ADS software

Frequency/Hz

Frequency/GHz

Fig. 6.35 Z21 and S21 simulation results of the Ethernet chip two-port network model

which is caused by the inductive components connected in series in the circuit. As the frequency increases, S21 increases, that is, the insertion loss decreases, and the value of S21 is higher above 400 MHz, and the influence of parasitic parameters on the S parameter is small.

6.5 PCB Decoupling Capacitor Optimization Method Multiple transistors inside the chip turn on and off at the same time, and the resulting current changes will cause voltage fluctuations in the power distribution network (PDN), which is synchronous switching noise. Synchronous switching noise forms a resonant cavity on the power and ground planes, causing radiated emissions. Therefore, to reduce the radiated emission of the PCB board, the first step is to reduce the synchronous switching noise of the PCB board. It is necessary to use measures such as decoupling capacitors and plane capacitors to reduce the impedance and resonance of the PDN. The crystal frequency of the vehicle controller and the main frequency of the chip are not high. Usually, a decoupling capacitor of 0.1 μF is added to each power pin,

6.5 PCB Decoupling Capacitor Optimization Method

319

but the communication rate of the vehicle Ethernet vehicle controller increases. The voltage tolerance of the Ethernet chip is lower, and the synchronous switching noise is more serious.Current decoupling capacitor design methods for vehicle controllers cannot effectively ensure the power integrity of the chip. Therefore, in order to ensure the power integrity of the Ethernet vehicle controller PCB, a design method for reducing the PDN impedance of the vehicle controller PCB is proposed.

6.5.1 The Impedance of the PCB Power Distribution Network of the Vehicle Controller (1) Target impedance of power distribution network Target impedance design is an important part of power integrity design, which requires independent design of power distribution networks with different voltages. Due to the presence of synchronous switching noise, the current drawn by the chip contains a lot of high-frequency noise, so the frequency spectrum of the power supply current should be considered when designing the target impedance. In the simple PDN model shown in Fig. 6.36, ZPDN (f ) is the complex impedance seen from the die pad to the PDN, including resistive impedance, capacitive reactance, and inductive reactance. When the current I(f) with a certain bandwidth passes through the PDN, a voltage drop ΔU(f ) will be generated on the pad and connection line of the PDN, which needs to be satisfied: I ( f ) × Z P D N ( f ) = V ( f ) < Vri pple Vri pple Z P D N ( f ) < Z target( f ) = I( f )

(6.8)

In the formula, V ripple is the voltage noise tolerance and Z target (f ) is the target impedance. Itransient < Imax Fig. 6.36 Relationship between PDN impedance and voltage drop

(6.9)

320

6 Signal Integrity and Electromagnetic Compatibility …

I transient is the worst-case transient current, and I max is the maximum current drawn by the chip. According to the rule of thumb, we get I transient ≈

1 · Imax 2

(6.10)

Then, the calculation formula of the target impedance is Z t arg et ≤

Vri pple Itransient

≈

Vdd × kri pple × 100% Vdd × kri pple × 100% =2 × (6.11) Itransient Imax

In the formula, V dd is the working voltage of the chip, k ripple is the ripple coefficient. The calculation formula of the target impedance can be expressed by the maximum power Ppeak of the chip: Z t arg et ≤ 2 ×

2 × kri pple Vdd × 100% Ppeak

(6.12)

The target impedance of all the PCB chips can be obtained by formula (6.12), as shown in Table 6.3. It can be seen that the Ethernet vehicle controller has higher requirements on the target impedance, such as Ethernet communication interface chip, MCU chip, CAN transceiver chip and EEPROM chip. Considering the signal characteristics of the Ethernet chip, stricter requirements should be imposed on its target impedance, limiting the ripple requirement to 1%, then the stricter target impedance is 0.2 . (2) PCB impedance modeling and simulation (1) Impedance simulation of main control chip The power pins of the main control chip XC2268I are distributed on the 2, 3, 14, 25, 27, 50, 52, 75, 77, 100 pins of the chip, as shown in Figs. 6.37 and 6.38. Table 6.3 Target impedance of each chip of the vehicle controller Chips

V dd (V)

Micro Control Unit (MCU)

5

Ethernet interface chip

3.3

CAN transceiver

5

CAN digital isolator

5

Digital switch detection chip

5

k ripple 3%

I max (A)

Z target ()

0.14

2.14

10% (1%)

0.132

5.00 (0.2)

10%

0.07

14.29

10%

0.007

142.86

5%

0.005

100.00

–

–

–

5

10%

0.02

50.00

5

10%

0.05

20.00

High-side driver chip

12

Low-side driver chip EEPROM

6.5 PCB Decoupling Capacitor Optimization Method

321

Fig. 6.37 +5 V power distribution network structure and key chip layout

Fig. 6.38 XC2268I chip power pin distribution

The impedance simulation results from each pin pad to PDN are shown in Fig. 6.39. The power pins 2/3/100, 25/27, 50/52 and 75/77 are four pairs of power return paths that are close to each other. The closer the power path is, the closer the parasitic parameters of the power distribution network are, so the closer the impedance curve is. The target impedance of the MCU chip is 2.14 . It can be seen that all 10 power channels below 100 MHz meet the target impedance requirements, but the impedance

322

6 Signal Integrity and Electromagnetic Compatibility …

(a) Impedance of power pin 75

(b) Impedance of power pin 77

(c) Impedance of power pin 50

(d) Impedance of power pin 52

(e) Impedance of power pin 25

(f) Impedance of power pin 27

(h) Power pin 3 impedance

(j) Power pin 14 impedance

(g) Power pin 2 impedance

Fig. 6.39 MCU chip XC2268I impedance simulation results of each power pin

of all power pins begins to be greater than the target impedance around 100 MHz. The maximum impedance at 1 GHz is 20.1 . PND cannot meet the design requirements. (2) Ethernet interface chip impedance The Ethernet interface chip has six 3.3 V power channels, as shown in Fig. 6.40. The allowable power supply range of each channel is 2.97–3.63 V, and the target impedance of the chip is 5. Because only one chip in the entire system uses 3.3 V power supply, so the design does not use the power plane power supply, directly using the etching line power supply, 3.3 V ground is not connected to the ground plane. Figure 6.41 is the impedance simulation result of the 6 pins of the 3.3 V power supply. It meets the target impedance requirements at low frequencies. In the frequency range higher than 1 MHz, the impedance increases with increasing frequency, and exceeds the target impedance at a frequency greater than 260 MHz. In addition, the impedance curves of pins 4, 8, 11, 15, 17 are basically the same, only the impedance of pin 21 exceeds the target impedance at 3 MHz, and resonance occurs.

6.5 PCB Decoupling Capacitor Optimization Method

(a) 3.3V power network structure

323

(b) Chip power pin distribution

Fig. 6.40 3.3 V power network structure in PCB

(a) Impedance of power pin 4

(b) Impedance of power pin 8

(c) Impedance of power pin 11

(d) Impedance of power pin 15

(e) Impedance of power pin 17

(f) Impedance of power pin 21

Fig. 6.41 Impedance simulation results of each power pin of the Ethernet interface chip

(3) Impedance of CAN bus transceiver chip The power supply of the CAN bus transceiver chip is provided by a 5 V power supply through a 5–5 V DCDC chip with a voltage stabilization function to form a +5 V_CAN power supply network, which is isolated from the previous 5 V power supply, and a body capacitor and a decoupling capacitor are installed. The structure of 3 CAN bus transceiver chips and +5 V_CAN is shown in Fig. 6.42. CAN

324

6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.42 Structure of the power distribution network (+5V_CAN) where the CAN bus transceiver chip is located

(a) Power impedance of each CAN bus transceiver chip

(b) Power impedance of each CAN bus isolation chip

Fig. 6.43 Power supply impedance of each chip in the CAN bus circuit

bus transceiver impedance simulation results are shown in Fig. 6.43 (a). The target impedance is 14.29 . The CAN bus interface circuit uses power/ground plane power supply. It can be seen that the impedance is less than the target impedance in the 1 GHz frequency range. The impedance simulation result of the CAN bus digital isolation chip is shown in Fig. 6.43(b). The highest impedance is 17  at 1 GHz. There are two resonance spikes in the DC-1 MHz range, but the impedance of the entire frequency band meets the target impedance of 142 . (4) Impedance simulation of other chips The target impedance of the EEPROM memory chip external to the MCU chip is 20 . Its power supply impedance simulation results are shown in Fig. 6.44. The maximum impedance is 10  within 1 GHz, which meets the target impedance requirements. The power supply impedance simulation results of the digital switch detection chip and the low-side driver chip are shown in Fig. 6.45, and the impedance is less than the target impedance.

6.5 PCB Decoupling Capacitor Optimization Method

325

Fig. 6.44 EEPROM chip power supply impedance

(a) Digital switch chip

(b) Low - side driver chip

Fig. 6.45 Power impedance of digital switch detection chip and low-side driver chip

Through the above simulation analysis, it can be seen that, based on the original PCB decoupling capacitor and power supply plane design, the power supply impedance of the MCU and the Ethernet chip are beyond the standard. The specific results are shown in Table 6.4.

326

6 Signal Integrity and Electromagnetic Compatibility …

Table 6.4 The impedance of power distribution network exceeds the standard Chips Micro Control Unit (MCU) Ethernet interface chip

Target impedance () 2.14 5.00 (0.2 )

Maximum impedance ()

Whether it exceeds the standard

Superscalar

15.6

Yes

629%

20.1

Yes

302%

13

No

0

CAN transceiver

14.29

CAN digital isolator

142.86

19.8

No

0

Digital switch detection chip

100.00

98

No

0

Low-side driver chip

50.00

16.0

No

0

EEPROM

20.00

11.0

No

0

6.5.2 PDN Decoupling Capacitor Optimization (1) Common decoupling capacitor design methods There are two main ideas for optimizing decoupling capacitors. The first is to optimize the equivalent series inductance ESL, including traces, packaging, vias, and the parasitic inductance of the decoupling capacitor itself. This is very important for widening the frequency band of decoupling capacitors. Important; the second is to use optimization algorithms to design the decoupling capacitor values and quantity to reduce the transmission impedance. Commonly used decoupling capacitor design methods include the traditional BIGV method, Multi-Pole (MP) method, and decoupling capacitor design methods that use various mathematical optimization methods that have gradually emerged in recent years. In recent years, decoupling capacitor design methods have developed tremendously, and optimization algorithms have been used to design decoupling capacitors. For example, genetic algorithms and Nature-Inspired algorithms have been applied to decoupling capacitor design. Currently, optimization algorithms for decoupling capacitors have been embedded in various PDN simulation software. For example, genetic algorithms or Nealder-Mead algorithms can be used in CST software to optimize the capacitance of decoupling capacitors. (2) Influence of capacitor parasitic parameters on capacitance impedance The actual capacitor can be equivalent to an RLC series model (Fig. 6.46), where ESR is the parasitic parameter of the capacitor itself, and ESL includes two parts: the installation inductance and the parasitic inductance of the capacitor itself. When designing decoupling capacitors, it is mainly necessary to consider the capacitance

6.5 PCB Decoupling Capacitor Optimization Method Fig. 6.46 Capacitor equivalent circuit model

327

ESLdecap

Cdecap

ESRdecap

value, parasitic inductance, and the number of capacitors. First, study the impact of ESR, ESL, and C on the transmission impedance of the capacitor in the frequency domain. (1) The effect of capacitance on the transmission characteristics of the capacitor First consider an equivalent circuit model of a capacitor of 0.1 μF (Fig. 6.47). First, analyze the effect of the change in capacitance C on the transfer characteristics of the decoupling capacitor when ESR and ESL are unchanged. The transmission characteristics of the capacitor are shown in Fig. 6.48. It can be seen that as C increases, the self-resonance frequency of the capacitor gradually increases, the self-resonance frequency of the 1000 μF “large capacitor” is as low as 160 kHz, and the self-resonance frequency of the 1 pF “small capacitor” The resonant frequency is as high as 5.2121 GHz, it can be seen that as C increases, the frequency band that the capacitor can decouple continues to increase, but this does not mean that using a capacitor with a high self-resonant frequency will get better PDN. As C decreases, The average impedance of the capacitor in a wide frequency band is increasing. For example, a capacitor of 100 pF has a self-resonant frequency of around 530 MHz. At the self-resonant frequency point, the total impedance of the capacitor is about 0.33 , and it is close to self-resonant before the self-resonant frequency. At the point frequency, the impedance of the capacitor of 100 pF has been as high as a few  or even tens of . However, it can still play a decoupling role in the low impedance frequency range. Assuming that the impedance of the capacitor is less

Fig. 6.47 Equivalent model of capacitance in CST software

328

6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.48 ESR and ESL unchanged, the impedance change of the capacitor when C changes

than 1  and is capacitive, it can play a better decoupling role (0.01 m equivalent series resistance, 1 nH equivalent series inductance Under the condition), it can also be seen from Fig. 6.48 that when the frequency is close to 500 MHz, the capacitor of 100 pF can no longer play its capacitive role, while other capacitors with larger capacitances are already in an inductive state. After the self-resonance frequency is 500 MHz, the decoupling effect of the capacitor below 100 pF will be less obvious. Therefore, it is necessary to conservatively estimate the design. The cut-off frequency of the decoupling capacitor is 200 MHz, and the self-resonance frequency is around 1 nF capacitor.. At low frequencies, it is better for large capacitors to maintain low impedance. (2) The effect of ESR on the transmission characteristics of capacitors The size of the ESR is generally between a few milliohms and hundreds of milliohms, and the ten-octave ESR in the range of 1 m to 1  is compared, as shown in Fig. 6.49. It can be seen that as the ESR increases, the impedance curve of the capacitor tends to be flat, and the impedance at the self-resonance frequency continues to increase. When the ESR is very small, the self-resonance spike of the capacitor is obvious, and it is easy to produce parallel resonance with other capacitors with different capacitances. As shown in Fig. 6.50(a), It is a simulation circuit that generates parallel resonance between a capacitor of 0.1 μF and a capacitor of 10 μF with different ESR. The simulation result is shown in Fig. 6.50(b). It can be seen that when the ESR of the capacitor becomes smaller, the resonance generated by the parallel connection of the two capacitors is significantly strengthened. When choosing a capacitor, the ESR can be between 10 m and tens of m. (3) The effect of ESL on the transmission characteristics of capacitors The equivalent series inductance ESL consists of two parts: parasitic inductance and installation inductance. The parasitic inductance of chip capacitors packaged in

6.5 PCB Decoupling Capacitor Optimization Method

329

Fig. 6.49 C and ESL unchanged, the effect of ESR change on the transmission impedance of the capacitor

(a) Simulation model

(b) Simulation results

Fig. 6.50 Parallel resonance simulation of different ESR capacitors

0603 and 0805 is usually in the range of 0.5–2 nH, of which about 1 nH is mostly capacitance. The installation inductance is usually in the order of hundreds of pH, so it can be estimated that the range of an ESL is 0.5–2.5 nH. As can be seen in Fig. 6.51, as the ESL increases, the self-resonant frequency of the capacitor tends to decrease. In addition, as the ESL decreases, the impedance curve of the capacitor shows a smooth trend, which is beneficial to suppress the parallel resonance spike. When the ESL changes from 0.5 to 2 nH, the self-resonance frequency of the capacitor increases from around 10 MHz to around 20 MHz. It can be seen that ESL has a great influence on the self-resonance frequency of the capacitor. If the inductance changes by 0.1 nH in practical applications, it may cause the capacitor self-resonance The frequency varies by several MHz. Therefore, in the design, the capacitor with small parasitic inductance should be selected as much as possible, and the influence

330

6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.51 The effect of ESL changes on the transmission characteristics of capacitors when ESR and C remain unchanged

of the installation inductance should be minimized when installing the decoupling capacitor (3) The practical effect of decoupling capacitance on reducing impedance. The influence of capacitance, ESR and ESL on the impedance characteristics of capacitors in frequency domain has been analyzed, and the practical effect of decoupling capacitors on impedance reduction has been analyzed. The capacitance values of nominal porcelain capacitors include: 1, 1.5, 1.8, 2, 2.2, 2.7, 3.3, 4.7, 5.6, 6.8, 8.2 series. The nominal values of electrolytic capacitors usually include: 1, 2.2, 3.3, 4.7, 6.8 series. If we want to design a fast design method for decoupling capacitors, we need to develop the types of capacitors. Figure 6.52 shows the impedance curves of capacitors of 1, 1.5, 1.8… 27 nf, 33, 47, 100 nF. The self resonance frequency of small capacitors below 10 nf is very close. Therefore, 11

Fig. 6.52 Impedance curve of 1–100 nF nominal capacitor

6.5 PCB Decoupling Capacitor Optimization Method

331

Fig. 6.53 Impedance curve of 100 nF–1 μF nominal capacitor

kinds of capacitors below 10 nf can be replaced by three kinds of capacitance with average distribution. The same is true in the range of 10–100 nf. Figure 6.53 is the impedance curve of all nominal capacitance in the 100 nF–1 μF interval. The larger the capacitance, the smoother the impedance curve. The self resonance frequency of capacitors of 470 and 560 nf is very close, but the impedance curve of the capacitance with twice the capacitance value is quite different. For example, the self resonant frequency of capacitors of 220, 470 nF and 1 μF varies by several MHz respectively, so as to cover the higher low impedance frequency with fewer capacitor types in the range of 100 nF–1 F, priority is given to capacitors with 100, 220 and 470 nF. Figure 6.54 shows impedance curves of nominal capacitors above 1 μF. It can be seen that when ESR and ESL are the same, the larger the capacitance value is, the wider the low impedance range can be covered. Therefore,

Fig. 6.54 Impedance curves of some nominal capacitors above 1 μF

332

6 Signal Integrity and Electromagnetic Compatibility …

in the range below 3 MHz, the capacitor with larger capacitance should be used as decoupling. From the large impedance simulation results of the designed PCB, the most representative impedance curve is shown in Fig. 6.55, where the target impedance is 0.5. The impedance below 750 kHz and above 162 MHz is larger than the target impedance. In addition, there is a resonance point at 11 MHz and the impedance at the resonant point is 0.19 . The influence of decoupling capacitors with different capacitance values is shown in Figs. 6.56 and 6.57. In the range below 10 MHz, the capacitance above 1 μF is very obvious. The effect of capacitors of 1 μF, 10 μF, 100 μF and 330 μF on reducing impedance is enhanced in turn. The capacitance of 330 μF makes impedance reach the target impedance requirement in 1 kHz–10 MHz range, but the effect on the resonant peak at 11 MHz is not obvious. According to the impedance curve analysis of the 330 μF capacitance, the impedance above 10 MHz increases, so the

Fig. 6.55 Impedance curve of a power supply in PCB

Original Parallel 10uF Parallel 10 0uF Parallel 33 0uF Parallel 4.7 uF Parallel 680n F Parallel 330n F Parallel 220n F Parallel 68n F

Fig. 6.56 Actual effect of 68 nF–330 μF decoupling capacitor

6.5 PCB Decoupling Capacitor Optimization Method

333 Original Parallel 10uF Parallel 100uF Parallel 330uF Parallel 4.7uF Parallel 680nF Parallel 330nF Parallel 220nF Parallel 68nF

Fig. 6.57 Actual effect of 68 nF–330 μF decoupling capacitor

decoupling requirement of higher frequency can not be realized. For the analysis of the resonant points above 11 MHz, as shown in the Fig. 6.57, the best decoupling effect is 220 nF capacitance, followed by capacitors of 330, 680, 68, 220 nF, and their self resonant frequency is slightly higher than 11 MHz, so 220 nF capacitance shows low capacitance impedance at 11 MHz.The capacitance of 330 and 680 nF shows low inductive impedance, and the capacitance of 68 nF is higher. Above 20 MHz, the effect of the above capacitors is similar. Therefore, when the decoupling capacitor is used to reduce the PDN impedance, the self resonance frequency of the capacitor should be slightly larger than the frequency to be suppressed, so as to ensure the capacitance is capacitive under low impedance. (4) Decoupling capacitor design method A decoupling capacitor design method suitable for engineering application is proposed according to the type of decoupling capacitor and the decoupling characteristics of capacitors at low frequency and high frequency. The first step: Determine the target impedance of the power supply channel through calculation. The second step: Modeling and simulating the circuit without decoupling capacitor, finding out the impedance exceeding the standard frequency band, setting the initial frequency of n segment exceeding the standard is f (n) a , and terminating frequency is f (n) b . If there is a resonant spike in this frequency band, the frequency of the resonant peak is f (n) c . The third step: Design the types of decoupling capacitance. If the cut-off frequency of the decoupling capacitor is 200 MHz, considering the influence of ESL, the capacitance below 1 nF is not considered. Taking into account that the impedance characteristics of electrolytic capacitors at low frequency range is better, so we should use ceramic capacitors below 1 μF and use electrolytic capacitors above1 μF. Based on the Decades method, the types of decoupling capacitors have been improved. The frequency domain impedance curves of all capacitors have been plotted as shown in Fig. 6.58.

334

6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.58 Frequency domain impedance curve of capacitance below 1 μF

The fourth step: Calculate the self resonant frequency of each capacitor sequentially, and sequentially rank C (m) from low to high capacitor according to the self resonant frequency, and m as the total number of decoupling capacitors, corresponding to the self resonant frequency of f SRF (m). The fifth step: Designe coupling capacitor at low frequency. The self resonant frequency of capacitor of 1 μF is about 5 MHz, and 100 kHz–5 MHz is the decoupling range of low frequency band. If the interval f (n)a< f (n)b is less than 5 MHz, then, in the frequency band between f (n)a and f (n)b , One per decade method is used to design the electrolytic capacitor. The difference is that the self resonance frequency of the capacitor with the largest capacitance is close to f (n)a . The capacitor with the minimum capacitance is 1 μF, and a capacitor is selected every 10 times. The sixth step: Find the frequency of the impedance exceeding the limit within 200 MHz above 5 MHz. According to the impedance diagram of the capacitor, the frequency range of all capacitors whose impedance value is lower than Ztarget is found in the frequency range exceeding the standard. The capacitance of the capacitor is smaller than the target impedance in the frequency range of f decouple (m)a– f decouple (m)a. With the increase of frequency, when the first capacitor is sensible, Select second capacitive capacitors, and so on. The seventh step: Calculate the number of decoupling capacitors in each frequency band. The influence of ESL on the impedance of the capacitor has been analyzed. In order to get the ideal low impedance, ESL must be controlled to a certain range. The average value of L mount = 0.65 nH is obtained here, and the total equivalent series inductance must satisfy 6.4.

6.5 PCB Decoupling Capacitor Optimization Method

335

Z target E SL ≤ N 2π f E S L · 2π f N≥ Z target

E S L max =

(6.13)

where,E S L max The total equivalent series inductance for parallel capacitors, unit: nH; E S L is the equivalent series inductance of a single capacitor, unit: nH; N is the number of capacitors of the same type. f is the highest frequency of the operating frequency band of this type of decoupling capacitor. The eighth step: Repeat the fifth step to the seventh step until the PDN impedance is completely below the target impedance.

6.5.3 Impedance Verification in Frequency Domain The PSPICE model of the Ethernet 3.3 V power PDN in the integrated controller is used to verify the validity of the method. The Ethernet interface circuit did not consider the integrity of the power source in the initial design, only a capacitor of 0.1 μF was connected in parallel at each power pin. A electrolytic capacitor of 100 nF and a electrolytic capacitor of 10 μF were installed at the output end of the VRM power supply. The decoupling capacitor and capacitance are redesigned using the proposed method. The object is the 21 power pin of the Ethernet interface chip. The impedance is shown in Fig. 6.58. After the first step to the ninth step, the following capacitor combinations are finally obtained, as shown in Table 6.5. Install the above decoupling capacitor at the nearest position on the circuit board diagram, as shown in Fig. 6.59, decoupling effect is shown in Fig. 6.60, and the impedance within 200 MHz is reduced to below 0.2 .

6.6 Modeling and Analysis of PCB Board Signal Integrity Ethernet for future vehicular domain controllers is an Ethernet protocol based on TSN (Time-Sensitive Networking) standard. TSN based Ethernet has strict requirements for delay and delay jitter, and it is necessary to analyze and verify its integrity. Table 6.5 Capacitance selection of design method

Capacitance value

Number

Capacitance value

Number

1 nF

3

100 nF

1

3.3 nF

2

470 nF

1

4.7 nF

2

100 μF

1

33 nF

1

336

6 Signal Integrity and Electromagnetic Compatibility …

Modified Original

electrolytic

electrolytic

capacitor

capacitor

(a) Original electrolytic capacitor (b) Modified electrolytic capacitor Original decoupling capacitor

(c) Original decoupling capacitor (small capacitance) Modified decoupling capacitor

(d) Modified decoupling capacitor

Fig. 6.59 Comparison of decoupling capacitance before and after optimization

The transmission line modeling of VCU Ethernet differential line is carried out. The reflection and delay of Ethernet differential transmission line are analyzed, and the crosstalk of Ethernet signal to adjacent signal line signal is analyzed.

6.6 Modeling and Analysis of PCB Board …

337

1

0.2Ohm Impedance (Ohm)

0.1

0.01

0.001

1E-4

Original 7 capacitors of 0.1uF in parallel The method of this parper

1E-5 1E-6

1E-5

1E-4

0.001

0.01

0.1

1

Frequency (GHz)

Fig. 6.60 Impedance suppression effect of decoupling capacitor optimization method

6.6.1 Vehicular Ethernet Signal and CAN Bus Signal Spectrum At present, the vehicle bus is mainly CAN, LIN, FlexRay, MOST, etc. CAN bus is the most widely used. The rate of general CAN bus is 2 Mbps, the CAN bus that meets CAN FD standard can reach the rate of 8 Mbps at the highest rate, and the rate of Ethernet transmission can reach the physical layer of the FD. The Ethernet bus frequency is much higher than that of the bus. As shown in Fig. 6.61, the waveform of 1 MHz and 1 GHz trapezoidal wave signal is approximately equivalent to CAN bus signal and Ethernet signal respectively. For an trapezoidal waves, the amplitude is A, the rise time is tr ,the fall time is t f , the pulse width is τ , the cycle is T , and tr = t f .The Fourier coefficients of trapezoidal

Fig. 6.61 1 MHz and 1 GHz trapezoidal wave signals

338

6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.62 Amplitude frequency characteristics of 1 MHz and 1 GHz trapezoidal waves

waves are Fn =

   sin nω20 τ − j nω0 (tr +τ ) sin nω20 tr cn(2) τ 2 e A = nω0 τ nω0 tr T ( jnω0 )2 2 2

(6.14)

where, n represents the first n harmonic. cn is the harmonic component at n sub harmonic, unit is dBμV. ω0 Represents harmonic frequency. τ represents the time span between the rising edge and the falling edge of a half amplitude point (A/2) in a single cycle. τr represents pulse rise time. The envelope of the spectral amplitude can be obtained from the formula of Fourier coefficients.    sin nω20 τ − j nω0 (tr +τ ) sin nω0 tr 2τ 2 e A nω0 t2r (6.15) E( f ) = |cn | = 2|Fn | = nω0 τ T 2 2 As shown in Fig. 6.62, the amplitude frequency characteristics of 1 MHz and 1 GHz trapezoidal waves in 50 GHz are shown respectively. It can be seen that in the range of low frequency (10 MHz), the amplitude of frequency components of 1 MHz is similar to that of 1 GHz. With the increase of frequency, the frequency components of 1 MHz trapezoidal waves decrease rapidly, while the amplitude of 1 GHz trapezoidal waves decreases slowly. At 1 GHz, 1.5 GHz, 2 GHz, 2.5 GHz and other high frequency points, there are large amplitude components. Generally speaking, the bandwidth of 1 GHz trapezoidal wave greater than that of 1 MHz trapezoidal wave, and the high frequency component energy of 1 GHz trapezoidal wave is much greater than that of 1 MHz trapezoidal wave. Compared with CAN bus, the Ethernet signal has more abundant high-frequency components, and is prone to form high-frequency electromagnetic noise.

6.6 Modeling and Analysis of PCB Board …

339

6.6.2 Reflection Modeling and Simulation (1) Reflection formation mechanism The signal is transmitted along the transmission line in the form of electromagnetic waves, and a transient impedance is felt at all times. This impedance may be the transmission line itself, or it may be other components in the middle or the end. As shown in Fig. 6.63, the transient impedance in the first area is Z 1 , The transient impedance of the second regions is Z 2 , at the interface, part of the signal is propagating forward, part of the signal is transmitted back, and the incident voltage of the signal is Vincident , the reflected voltage is Vr e f lected , the transmission voltage across the interface is Vtransmitted , the ratio of the reflected signal to the amplitude of the incident signal is Vr e f lected Z2 − Z1 = =ρ Vincident Z2 + Z1

(6.16)

where, ρ is the reflection coefficient for voltage. The greater the difference in impedance, the greater the reflected voltage. If the reflected voltage is too large, it may cause undesirable results such as signal tick back or secondary sampling, one of which is to generate ringing. The reason of ringing is that the signal reflects between the driving end and the remote load many times. The output impedance of most chips is very low. If the output impedance is less than the characteristic impedance of PCB wiring, signal ringing is inevitable without source termination, as shown in Fig. 6.64. Larger ringing will lead to signal misjudgment or multiple triggering, which will affect the normal operation of the circuit. (2) Modeling of vehicle differential interconnection Because the stability of differential interconnection is higher than that of single terminal interconnection, the current high-speed signals almost use differential interconnection. However, there is also an obvious defect in the differential interconnection structure. In case of mismatching of interconnection impedance, unexpected common mode current will be generated, which will cause electromagnetic interference on the conductor. The differential interconnection structure is shown in Fig. 6.65. Differential signals and common mode signals can be expressed as Vdi f f = V+ − V−

Z2

Z1 Vincident

Fig. 6.63 Reflection formation mechanism

Vreflected Vtransmitted

340

6 Signal Integrity and Electromagnetic Compatibility …

Ringing at the receiving end Crystal source Receiving end

Fig. 6.64 Signal ringing waveform

Fig. 6.65 Schematic diagram of differential interconnection structure

Vcomm =

TX+

RX+

TX-

RX-

V+ + V− 2

(6.17)

where, V+ is positive signal output for differential driver. V− is negative signals output for differential drivers, the two signals have opposite polarity. There are two transmission modes of differential signals: odd mode and even mode. The transmission mode of two differential lines with opposite voltage is called odd mode, and the same transmission voltage is called even mode. The lumped parameter model of microstrip line is shown in Fig. 6.66. It can be seen that the model is composed of several equivalent models of unit length microstrip line as shown in Fig. 6.67. Differential transmission lines are modeled according to the models in Figs. 6.66 and 6.67, and the characteristic impedance of the microstrip lines can be calculated by the analytic approximation method. The general approximation of characteristic impedance is expressed as Fig. 6.66 Equivalent circuit model of microstrip line

Signal path

Reference path

6.6 Modeling and Analysis of PCB Board …

341

Fig. 6.67 Equivalent model of microstrip line with unit length

  87Ω 5.98h Z0 = √ ln 0.8w + t 1.41 + εr

(6.18)

where, Z 0 represents characteristic impedance. h represents the thickness of the dielectric layer between the signal line and the plane (unit: mil), w represents line width (unit: mil), t represents the thickness of the microstrip line (unit: mil), εr represents dielectric constant. The transmission speed of the signal on the PCB board is c vp = √ μr εr

(6.19)

where, v p represents the propagation speed of signals, c represents the speed of light, μr represents permeability, if the medium is not ferromagnetic material, then, μr = 1 , εr represents dielectric constant, the dielectric constant of FR4 plate is 4. According to this formula, it can be calculated, v p ≈ 6inch/ns.  Z0 =

LL CL

1 vp = √ CL L L

(6.20) (6.21)

The formula 6.19–formula 6.21 obtains the unit length inductance L L and the unit length capacitance C L expressed by the characteristic impedance and transmission speed. √ Z0 Z0 √ = εr = 0.083Z 0 εr n H/in vp c 1 1 √ 83 √ CL = = εr p F/in εr = vp Z0 c · Z0 Z0 LL =

(6.22)

Total inductance of lumped model of microstrip line L total and total capacitance Ctotal can be directly obtained. Ctotal = C L · L en L total = L L · L en where, L en is the equivalent length of the line.

(6.23)

342

6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.68 Ethernet differential line layout diagram

The Ethernet of the vehicle controller is communicated with other controllers through the vehicle connector and is four wire Ethernet. The differential line layout in PCB is shown in Fig. 6.68. The four signal lines are RD−, RD+ , TD− and TD+ from top to bottom, that is two pairs of differential lines. In order to ensure the shielding effectiveness of the vehicle controller shell and the harsh environment tolerance (salt spray environment, high humidity environment), no RJ45 interface was used. The characteristic impedance of the Ethernet differential pair is 100 , the characteristic impedance of the single ended signal line is 50 , and the through-hole characteristic impedance is 40 . Based on the above analysis, the equivalent circuit model of Ethernet differential line as shown in Fig. 6.69 (a) and (b) is established. (3) Ethernet signal reflection and delay The simulation of Ethernet differential line equivalent circuit model analyzes the reflection and delay characteristics of Ethernet two pairs of coupled differential lines. As shown in Fig. 6.70, the reflection simulation results of TD differential pair can be seen. It can be seen that the reflection of TD+ and TD− signals is serious when the termination is not carried out, and the overshoot/ undershoot of more than 0.8 V exists in the signal jump edge of both signals, which forms ringing when the signal rises and falls. Because of the equal length design of Ethernet differential pair wiring, the delay of the differential pair is close, the delay of TD− is 0.163 ns, the delay of TD+ is 0.169 ns, so the time difference between the two lines of TD differential pair is 0.006 ns, which is within the acceptable range. As shown in Fig. 6.71, the results of the RD differential pair reflect that RD− haS larger overshoot and undershoot on the rising and falling edges of the signal than the TD-. The maximum overshoot voltage is close to 1 V, which has more than doubled the differential signal level, and the RD− signal also has ringing. Comparing the waveforms of RD+/TD+ and RD-/TD− signal receivers, as shown in Fig. 6.72, it can be seen that the waveforms of RD+ and TD+ are very close, the

6.6 Modeling and Analysis of PCB Board …

Fig. 6.69 Differential circuit equivalent circuit model of Ethernet

Fig. 6.70 Reflection simulation results of TD+ and TD−

Fig. 6.71 Reflection simulation results of RD+ and RD-

343

344

6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.72 Comparison of waveform between receiver terminals of RD+/TD+ and RD-/TD- signals

undershoot of RD+ is more serious, and the overshoot of RD− is more serious than that of TD−, so take Rd +/ RD− differential pair as the research object to suppress the reflection. (4) Transmission line Z parameter analysis and reflection improvement method The reflection is mainly caused by impedance mismatch, including three types of impedance mismatch: the mismatch between the signal drive end (output) and the transmission line, the mismatch between the signal receiving end and the transmission line, and the impedance mutation in the transmission line. Besides the differential signal, the differential mode will also reflect the common mode signal flowing through it. Therefore, it is necessary to divide the signal into the differential signal and the common mode signal when analyzing the reflection problem of the differential pair. In the odd mode transmission mode, the impedance of the differential signal is all odd mode impedance. As shown in Fig. 6.73, the difference between two single ended signals in the course of transmission is reversed, and the current passed through each single end signal line is I0 =

V0 Z0

(6.24)

V0 is the voltage applied to single ended signals. Z 0 is single ended impedance, the unit is . I0 is the current transmitted on the single ended signal, the current of the two differential lines is opposite, so the return path of the differential signal is the clockwise loop shown in Fig. 6.73. Idi f f = I0 , then, for differential signals, the impedance they encounter is Fig. 6.73 Differential current and differential mode impedance

V0 TX+

Idiff

RX+ Zdiff

TX-

Idiff

V0 RX-

6.6 Modeling and Analysis of PCB Board …

Z di f f =

345

Vdi f f 2 × V0 = = 2 × Z odd Idi f f I0

(6.25)

where, Z odd is odd mode impedance, in odd mode transmission mode, Z odd is equivalent to the characteristic impedance of the single ended transmission line. When the differential line is coupled, for the microstrip line of FR4 material, the differential impedance is approximately   s

Z di f f = 2 × Z 0 1 − 0.48 exp −0.96 h

(6.26)

where, Z 0 is single ended impedance, S is the edge spacing of the signal line, the unit is mil, and h is the dielectric thickness between the signal line and the return plane. The common mode signal is similar to the difference signal, and the common mode components of the two single terminal lines are the same. As shown in Fig. 6.74 is the return path of the common mode signal, and the common mode impedance is the parallel connection of the characteristic impedance of each line. For a common mode signal, the impedance encountered is Z comm =

Z even × Z even Z even = Z even + Z even 2

(6.27)

where, Z comm is common mode impedance, Z even is the impedance of a single terminal transmission line when the differential pair is in the even mode state (the driving voltage of the two lines is the same) Z even =

Vcomm Icomm

(6.28)

When the transmission line is made up of two uncoupled 50 transmission lines, the odd mode impedance and the differential mode impedance are the same. When the pair pair is tightly coupled, the odd mode impedance is about 50  and the even mode impedance is about 55 . Fig. 6.74 Common mode current and common mode impedance

V0 TX+

Icomm Zeven

RX+ V0

TX-

Icomm

RXZeven

346

6 Signal Integrity and Electromagnetic Compatibility …

To solve the problem of signal reflection, the most direct way is to make impedance matching in the transmission path of the signal. The most common method is terminal matching and adjusting the transmission line topology at the end of the transmission line. There are four main termination methods for the differential pair, all of which are at the end of the transmission line, as shown in Fig. 6.75. (a) The first termination method is shown in Fig. 6.75a. At the end of two differential transmission lines, a resistance is connected in parallel, and the resistance values of two termination resistances are Z 0 , that is, the characteristic impedance of a single line. When the differential signal reaches the termination position, the impedance felt is 2Z odd = Z di f f , so that there is no reflection at the end of the transmission line. This structure only terminates the differential signal. (b) The termination method shown in Fig. 6.75c is called T terminal, which can connect the differential signal to the common mode signal simultaneously. Where, R1 = Z odd Z even − Z odd R2 = 2

(6.29)

For differential signals, it flows through two R1 , it is equivalent to the first and second kinds of termination methods. For common mode signals, it flows through RX+

TX+

RX+

TX+

Zodd Zdiff

Zodd TX-

RX-

TX-

RX-

a

b RX+

TX+

R1

RX+

TX+

R2

R2 R1

R1 TX-

RX-

c Fig. 6.75 Four differential terminations

R2 TX-

RX-

d

6.6 Modeling and Analysis of PCB Board …

347

two parallel connections R1 and a series of R2 The common mode termination is completed. (c) The termination method shown in Fig. 6.75 (d) is called π termination R1 =

2Z even Z odd Z even − Z odd R2 = Z even

(6.30)

Differential signal flow through R1 paralleling two R2 , the impedance of the circuit is exactly Z di f f , the differential impedance matching is completed. The common mode signal flows through two R2 in parallel, the impedance felt is R2 /2=Z even /2, similarly, impedance matching is performed for common mode signals. (5) Reflection improvement measures In this paper, vehicular Ethernet is tightly coupled differential pair, odd mode impedance Z odd = 50Ω, even mode impedance Z even = 55Ω, Use π termination, R1 = 1100Ω , R2 = 55Ω. RD +/RD− and TD +/TD− two pairs of differential lines are terminated by π termination, the comparison of the reflection simulation before and after the termination is shown in Fig. 6.76. The signal level after termination is returned to the normal range of 0–1 v or 0 to −1 v, and the overshoot and undershoot of the signal are also improved. The waveform is close to the ideal transmission signal, and the maximum reflection voltage is less than 15% of the signal voltage.

Fig. 6.76 Contrast between RD +/RD-and TD +/TD- two before and after differential terminal connection

348

6 Signal Integrity and Electromagnetic Compatibility …

6.6.3 Crosstalk Modeling and Simulation Crosstalk is the coupling between two signal lines and the mutual inductance and mutual capacitance between the signal lines which cause the noise on the line. The crosstalk noise will pull up or down the level of the signal and affect the quality of the signal. This section mainly analyzes the crosstalk between the high frequency signal of the vehicle Ethernet and the original signal of the vehicle controller after the vehicle controller is added to the vehicle Ethernet. (1) Formation mechanism of crosstalk In the process of signal transmission, the loop between the signal line and its return path will form an electromagnetic field, which will affect the circuit of the signal line and the circuit around the return path. It is called the edge field. The edge field will produce noise on other signal lines in the field, and the noise will be coupled to the victim line through mutual inductance and mutual capacitance between the signal lines. As shown in Fig. 6.77, the attack line crosstalk is caused by capacitive or perceptual coupling on the victim line, and the same as the attack line signal is the far end crosstalk. The opposite direction is the near end crosstalk with the attack line signal. When crosstalk noise is superimposed on the edge of the signal, the edge will shake, which will affect the timing, edge or amplitude of the signal. The signal noise tolerance is usually 15% of the signal voltage amplitude, 5% of which is the noise tolerance of crosstalk. For 1 V signal, the crosstalk that can be tolerated is 50 mV, so it is very necessary to estimate the crosstalk amplitude and reduce the crosstalk when designing the signal routing. (2) Crosstalk simulation of vehicle can bus signal line (1) Signal selection There are three routes in PCB, which need to consider crosstalk, as shown in Figs. 6.78 and 6.79. Where, the parallel 2 lines and the line 3 are the parallel lines of different can bus differential pairs. The 1 line is the TTL level signal received by the master MCU and the can transceiver. In the three part of the crosstalk, the signal line frequency of the 1 line area is the highest, and the parallel length of the

Fig. 6.77 Formation mechanism of crosstalk

6.6 Modeling and Analysis of PCB Board …

349

Fig. 6.78 Crosstalk traces to be considered

Fig. 6.79 Consider the specific signal of the line

adjacent line is the longest. Therefore, the crosstalk in the three part is the most likely to occur crosstalk. And because the resistance of the differential pair to the adjacent crosstalk is stronger than that of the single ended signal, the crosstalk noise of the area 1 signal line may be the largest. Therefore, four parallel lines CANTX1, CANRX1, CANTX0, CANRX0 in zone 1 are selected for crosstalk analysis. (2) Modeling and Simulation of crosstalk in CAN bus Area 1 is four TTL level signal lines, as shown in Fig. 6.80, from top to bottom are CANTX1, CANRX1, CANTX0, CANRX0. The distance between two adjacent signal lines is 3 times the line width. Cantx1 and cantx0 use MCU as transmitter and can transceiver as receiver. Canrx1 and canrx0 take MCU as receiver and can transceiver as transmitter.

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6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.80 Distribution of signal line in line area 1

Fig. 6.81 Cross section of crosstalk model for four lines

Crosstalk mainly takes into account the parallel part marked by the black dotted box in the figure. TTL signal is transmitted in all four signal lines, and its topology is shown in Fig. 6.81. Among them, 1, 2, 3 and 4 are cross-sections of four signal lines that are coupled with each other. The specific crosstalk coupling simulation model is shown in Fig. 6.82. In the dotted box at the left end is the driver/ receiver at MCU end, the transmission line model of microstrip line and coupling part is in the middle, and the driver/ receiver of four can transceivers is at the right end. Each signal is sent out through the driver, through a section of uncoupled line, then through a section of coupled crosstalk line, then through a section of uncoupled crosstalk line, and finally to the receiving end. (3) Crosstalk simulation results and analysis Crosstalk voltage amplitude of signal line CANTX0 and CANRX0 when CANRX1 signal is jumping. The crosstalk between the rising edge signal and the stable high level signal is set as follows: the CANRX1 signal is the square wave pulse rising to 5 ns, and the CANTX0 and CANRX0 signals are 5 V high level signals; the simulation results of CANTX0 and CANRX0 maintaining high level are shown in Fig. 6.83. The maximum crosstalk of signal line CANRX0 is 7 mV. It can be seen that the interference of crosstalk to high

6.6 Modeling and Analysis of PCB Board …

351

Fig. 6.82 Crosstalk coupling model

Fig. 6.83 Terminal voltage of crosstalk cantx0 and canrx0 for high level signal stabilized by rising pulse

352

6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.84 Receiver voltage diagram of cantx0 and canrx0

level signal is very small, and the overlapped crosstalk signal satisfies the threshold of 3.5 V high level. The crosstalk between the rising edge signal and the stable low level signal is set as follows: the CANRX1 signal is the square wave pulse rising to 5 ns, the CANTX0 and CANRX0 signals are the signals maintaining 0 V low level; the simulation results of CANTX0 and CANRX0 keep low level are shown in Fig. 6.84. It can be seen that the maximum crosstalk of signal line CANTX0 is 30.6 mV; the maximum crosstalk of signal line CANRX0 is 6.9 mV. • When CANTX1, CANRX1, CANRX0 three signal lines jump, the crosstalk voltage amplitude of signal line CANTX0. The three signal line rising pulses keep the low level crosstalk on the signal line. The simulation parameters are set as: CANTX1, CANRX1. The three signal of CANRX0 is the square wave pulse with 5 ns rise time. The CANTX0 signal is to maintain 0 V low level signal; CANTX0 remains low level simulation result is shown in Fig. 6.85. It can be seen that when there are three attack lines and crosstalk at the same time, the maximum crosstalk of signal line CANTX0 is 57.2 mV. (3) suppression of crosstalk and simulation verification The CAN signal line is microstrip line, and the crosstalk simulation result is shown in Fig. 6.86. Red: attack line. Blue: victim line. After widening the line spacing, a crosstalk simulation result on a damaged line is shown in Fig. 6.87, for a pulse signal with a frequency of 10 MHz and a fixed duty cycle. Using a stripline, a crosstalk simulation result on a second line with a frequency of 10 MHz and a fixed duty cycle is shown in Fig. 6.88. From Figs. 6.87 and 6.88, it can be seen that under the same parameters and the same line spacing (10 mil), the maximum crosstalk voltage of stripline coupling

6.6 Modeling and Analysis of PCB Board …

353

Fig. 6.85 cantx0 signal line terminal voltage

Fig. 6.86 Two crosstalk simulation results of adjacent microstrip lines: left: simulation result right: victim line crosstalk voltage

Fig. 6.87 Crosstalk simulation results after widened line spacing

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6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.88 Two crosstalk simulation of stripline

is 84.247 mv, while the maximum crosstalk voltage of microstrip line coupling is 231.216 mv. It can be seen that stripline has better crosstalk resistance characteristics than microstrip line. (3) simulation of crosstalk between vehicular Ethernet and adjacent signal lines For example, there are only two switch signal input lines near the differential pair of Ethernet in the vehicle controller, as shown in Fig. 6.89. SP2 and SP3 are switch signal input lines. The main factors affecting crosstalk voltage are coupling distance and attack signal frequency. With the decrease of line spacing, the mutual Fig. 6.89 Crosstalk coupled signal line

Ethernet chip

Digital signal acquisition chip

6.6 Modeling and Analysis of PCB Board …

355

Table 6.6 Minimum spacing between differential pairs and switching signal lines Setting: mm

RD_N

RD_P

TD_N

TD_P

SP3

2.1

1.7

1.15

0.75

SP2

2.5

2.1

1.55

1.15

Fig. 6.90 Crosstalk simulation model SP3

SP2

RD_N RD_P TD_N TD_P

SP2

SP3

capacitance and mutual inductance between transmission lines increase, and the coupling crosstalk noise also increases. The minimum distance between the two Ethernet differential pairs and the signal line is shown in Table 6.6. As shown in Fig. 6.90 is the crosstalk model built in CST. The black box model can simulate the electromagnetic field coupling of crosstalk. The coupling lines are two pairs of differential lines and two switch signal input lines SP2 and SP3. Through this model, the crosstalk effect of single or multiple Ethernet transmission data on switch signal lines is studied. The signal waveform of Ethernet is trapezoidal wave with 1 GHz frequency and 1 V amplitude. The crosstalk of each differential line of Ethernet to SP2 and SP3 signal lines is shown in Fig. 6.91. When a single Ethernet signal line works alone, the crosstalk voltage generated is shown in Table 6.7. The crosstalk voltage of TD_ P signal line closest to the victim line is 2-9 times of that of other Ethernet signal lines. Two differential lines with opposite voltage variations TD_N and TD_P simultaneously produces superimposed crosstalk on the SP2 and SP3 signal lines, ss shown in Fig. 6.92. The maximum crosstalk voltage on SP2 is 5.2 mV, and the maximum crosstalk voltage on SP3 is 6.9 mV, which is about TD_N and TD_P difference of crosstalk voltage generated on SP2 and SP3 respectively. When the four differential lines RD_N, RD_P, TD_N and TD_P work at the same time, the crosstalk generated on the signal lines SP2 and SP3 is shown in Fig. 6.93. It is similar to TD_N and TD_P when they work at the same time, but the voltage amplitude is larger. The maximum crosstalk voltage on SP2 is 5.5 mv, and the maximum crosstalk voltage on SP3 is 7.2 mv.

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6 Signal Integrity and Electromagnetic Compatibility …

Fig. 6.91 Crosstalk of Ethernet differential lines on SP2 and SP3 signal lines separately

Table 6.7 crosstalk generated by separate signal lines on SP2 and SP3 Setting: mV

RD_N

RD_P

TD_N

TD_P

SP2

0.9

1.2

3.2

8.8

SP3

1.5

2

4.1

10.2

Fig. 6.92 Crosstalk caused by simultaneous operation of TD_N and TD_P

Fig. 6.93 Crosstalk caused by simultaneous operation of two pairs of differential lines in Ethernet

In the actual operation of the vehicle controller, a pair of differential lines of Ethernet work at the same time, and the crosstalk of the differential pair to the affected lines is mutually offset. Therefore, it is estimated that the crosstalk voltage of Ethernet to the signal line in the range of 1–2 mm line spacing is about 5.5 mv in the actual operation—Between 12.4 mv, the signal level of SP2 and SP3 is 12 V, and the crosstalk voltage amplitude accounts for 0.1% of the signal level, which will not

6.6 Modeling and Analysis of PCB Board …

357

affect the 12 V signal. In the actual wiring, the rule of ensuring that the Ethernet and the surrounding wiring maintain 3 times of the line width can ensure the crosstalk voltage does not interfere with the adjacent signal lines.

6.7 PCB Plate Electromagnetic Radiation 6.7.1 Electromagnetic Radiation Mechanism Ethernet adopts differential two-wire transmission system to form common mode current and differential mode current on transmission line. Both the differential mode current and the common mode current will form electromagnetic field, which will produce electromagnetic radiation to the adjacent wires or devices. The difference is that the electromagnetic radiation produced by common mode current is far greater than that of differential mode current. Imagine a pair of parallel conductors, as shown in Fig. 6.94. The line flows through current I1 and I2, and the two currents can be decomposed into the superposition of common mode current IC and differential mode current I D . I 1 = IC + I D I 2 = IC − I D

(6.31)

From the formula (6.31), we can get, 1 (I1 − I2 ) 2 1 IC = (I1 + I2 ) 2

ID =

(6.32)

It can be seen that the value of common mode current IC is greater than that of differential mode current ID. When studying the electromagnetic field of the differential transmission line, the electric field Eˆ generated by each current is superposed to obtain the total radiated electric field, as shown in Fig. 6.95. The electric field direction generated by two differential mode currents is opposite to each other, and the electric field direction generated by common mode current is the same and overlapped with each other, so a small common mode current can generate the same electric field strength with a large differential mode current. Differential signal is Fig. 6.94 Common mode current and differential mode current in differential transmission line

I1

IC

ID

I1

IC

ID

358

6 Signal Integrity and Electromagnetic Compatibility … ID

IC

ID

IC

( (

E1

(

(

E1

E2

E2

( (

Enet

Enet

Fig. 6.95 Difference of radiation electric field between common mode current and differential mode current

a useful signal for Ethernet, so the existence of common mode current should be minimized. When the common mode current of PCB differential signal is large, the corresponding electromagnetic radiation of the whole PCB will be large. In addition, in the absence of good suppression measures, PCB common mode current will also form conduction emission or radiation emission through the cable.

6.7.2 PCB Electromagnetic Radiation The transmission of electromagnetic disturbance through space is essentially that the electromagnetic energy of the disturbance source propagates to the surrounding space in the form of field. Induction field refers to a part of electromagnetic field energy flows back and forth periodically between the space around the radiation source and the radiation source, and does not radiate outward. Radiation field refers to another part of electromagnetic field energy separated from the radiator and emitted in the form of electromagnetic wave. Generally speaking, the area with the field source as the center and within the three wavelength range is called near field or inductive field; the space with the field source as the center and the radius beyond the three wavelength range is called far field or radiation field. Electromagnetic simulation software can be used to simulate and predict the electromagnetic radiation characteristics of Ethernet vehicle controller PCB. ANSYS/ SIwave software can simulate the three-dimensional electromagnetic field of PCB and automatically calculate the parasitic parameters, material characteristics and spatial characteristics of PCB. Through siwave software, a voltage source with signal amplitude of 1 V is applied at the head and end of each Ethernet signal line as the signal excitation source to study the electromagnetic field distribution of Ethernet line under different signal frequency and different number of lines.

6.7 PCB Plate Electromagnetic Radiation

359

(1) Electric field Figure 6.96 shows the electric field distribution of PCB on-board Ethernet under the action of excitation source at the frequencies of 66, 100, 600 MHz and 1 GHz. Among them, 66 and 600 MHz are the twisted pair frequencies specified in 100 baset1 and 1000 base-t1 physical layer protocols of vehicle Ethernet. It can be seen that the electric field at both ends of the Ethernet cable is large, and the electric field strength at the connection point between the signal line and the connector is the largest. This is because the signal line encounters the sudden change of the through-hole impedance, resulting in the generation of common mode current, thus forming a large electric field. In addition, with the increase of frequency, the electric field intensity decreases gradually, and the maximum electric field intensity is 14.6 V/m, 14.0 V/m, 12.9 V/m and 12.3 V/m, respectively. The electric field intensity at 66 MHz is the largest. (2) Magnetic field Figure 6.97 shows the distribution of magnetic field of PCB on-board Ethernet under the excitation of 66, 100, 600 MHz and 1 GHz. It can be seen that the magnetic field intensity decreases with the increase of frequency, and the radiation is larger at 66 and 100 MHz. The maximum magnetic field intensity is about 9.8 a/m at 66 MHz and 8.2 a/m at 100 MHz.

˄a˅66MHz

˄b˅100MHz

˄c˅600MHz Fig. 6.96 Electric field distribution of PCB Ethernet wire (RD-)

˄d˅1GHz

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6 Signal Integrity and Electromagnetic Compatibility …

˄a˅66MHz

˄c˅600MHz

˄b˅100MHz

˄d˅1GHz

Fig. 6.97 Magnetic field distribution of PCB single Ethernet wire (RD-)

6.8 Conducted EMI Suppression of Power Line of Ethernet Vehicle Controller In order to reduce the transmission line EMI of the vehicle controller of the Ethernet network, the method of restraining PCB common mode current is mainly adopted. Taking the suppression of 100 MHz common mode current as an example, the following two measures are adopted. (a) The impedance matching method is used to recalculate the differential matching resistance. (b) Design high frequency common mode filters. In view of the common mode current generated by the above PCB Ethernet lines, the LC common mode filter can be used. The common mode inductor LC of the common mode filter is 6.8 uH, and the common mode capacitance CY is 3300 pF. According to the national standard GBBT 18655-2018《Vehicle, boats and internal combustion engines-Radio disturbance characyeristics-Limits and methods of measurement for the protection of on-board receivers》 . The data transmission and reception of 100 Mbps transmission rate between the vehicle controller and the

6.8 Conducted EMI Suppression of Power Line of Ethernet Vehicle Controller

361

MPC5748 development board are carried out through Ethernet bus. As shown in Fig. 6.98, from the test results shown in Fig. 6.99, the peak and average values of conducted emission after the suppression of common mode current are all up to the standard level 3 limit requirement. As shown in Figs. 6.100 and 6.101, in the range of 30–85 MHz, the peak value and average value of conduction voltage after common mode current suppression are reduced by 9 dBμv as a whole. In the range of 85–108 MHz, the maximum attenuation of conduction voltage is about 23 dB μv.

Ethernet Ethernet

Vehicle controller this paper

in

Mpc5748 development board

Power Supply Upper computer

Measurement of conducted interference

Fig. 6.98 Layout of conducted emission test

100 90 80 70

dBµV

60 50

GBT 18655- 2010 Level ㄝ㑻3ዄؐ䰤ؐ 3 peak limit

3 average limit GBT 18655- 2010 Level ㄝ㑻3ᑇഛؐ䰤ؐ

40 30 20 10 0

—— ——

-10

Peak value Average value

-20 1

Frequency (MHz)

10

Fig. 6.99 Test results of conducted emission after suppression of common mode current

100

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6 Signal Integrity and Electromagnetic Compatibility …

100 90 80 70

Emission reduction 100MHz 100MHatz໘থᇘ‫ޣ‬ᇣ

dBµV

60 50 40 30 20 10 0

—— ——

-10

Peak value before inhibition Peak value after inhibition

-20 1

Frequency (MHz)

10

100

Fig. 6.100 Comparison of conducted emission peaks before and after common mode current suppression 100 90 80 70

dBµV

60

The average valueᑇഛؐ‫ޣ‬ᇣ decrease

50 40 30 20 10 0 -10

——

Average value before inhibition

——

Average value after inhibition

-20 1

Frequency (MHz)

10

100

Fig. 6.101 Comparison of conducted emission averages before and after common mode current suppression

Chapter 7

Electromagnetic Compatibility of Battery Management System

7.1 Summary The battery management system (BMS) of electric vehicle is a control system to protect the use safety of power cell. It can monitor the usage status of battery at any time, and guarantee the use safety of new energy vehicles. BMS hardware uses a large number of high-speed digital chips and high-speed buses. The integrity of PCB Power Supply and the electromagnetic compatibility of BMS are important for the normal operation of BMS. In this chapter, firstly, the impedance characteristics of the PCB power distribution network of BMS hardware are analyzed, a decoupling capacitor selection method is proposed and then optimized according to the target impedance. Then, the main interference sources and coupling paths of BMS are described. The clock signal produced by crystal oscillator is one of the main interference sources of electric vehicle BMS. By analyzing the spectrum characteristics of the clock signal and the coupling path of electromagnetic interference, a method to suppress the clock electromagnetic interference is proposed. The switching noise produced by DC-DC is another main interference source of BMS. The spectral characteristics and coupling path of the switching noise produced by MOSFET are analyzed, and the switching noise filter is designed. The mechanism of the switching noise produced by MOSFET is described by the modeling and simulation of the DC-DC buck equivalent circuit. This paper analyzes the interference signal characteristics of CAN bus circuit in BMS, simulates the signal integrity in the Cadence software, analyzes the reflection characteristics of CAN bus, and proposes an anti-interference design method for CAN bus circuit. Finally, the electromagnetic radiation of BMS PCB of electric vehicle is simulated and predicted using the SIwave software. The main contents described in this chapter are shown in Fig. 7.1.

© China Machine Press 2021 L. Zhai, Electromagnetic Compatibility of Electric Vehicle, Key Technologies on New Energy Vehicles, https://doi.org/10.1007/978-981-33-6165-2_7

363

364

7 Electromagnetic Compatibility of Battery Management System

Decoupling capacitor optimization method

•

•

Prediction of differential and common mode interference EMI filter design

Decoupling analysis of PCB Power Distribution Network • Optimization of decoupling capacitor

Filter Design

Electromagnetic noise of clock signal • Spectrum analysis of clock signal • Clock circuit modeling and simulation

Interference source 1

DC-DC Switching Noise DC-DC circuit analysis Interference coupling path analysis • Design of interference suppression circuit • DC-DC circuit modeling and simulation

Interference source 2

• •

Anti-interference analysis of CAN bus • CAN bus circuit interference • CAN bus circuit antiinterference optimization circuit design

Antiinterference design

Electromagnetic emission of PCB • Electromagnetic field distribution simulation of PCB • Far field electric field simulation of PCB

Simulation

Fig. 7.1 BMS electromagnetic contents

7.2 BMS System Function and Structure

365

7.2 BMS System Function and Structure (1) Function The functions of BMS mainly include: acquisition functions (such as single voltage, total voltage, current, temperature acquisition, etc.), charging port detection (CC and CC2) and charge wake-up (CP and A +), relay control and status diagnosis, insulation detection, high-voltage interlock, collision detection, CAN communication and data storage, etc. (2) Structure The BMS hardware architecture is divided into distributed and centralized architectures, as shown in Fig. 7.2. The distributed BMS includes the main control board and the slave control board. Generally, a battery module is equipped with a slave control board. The main control board completes SOC calculation and external bus communication of vehicle controller. The slave control board completes single voltage acquisition, equalization and temperature measurement. Centralized BMS concentrates all electrical components on a large board. The sampling chip channel has the highest usage and the daisy chain communication can be used between the sampling chip and the main chip. The circuit design is relatively simple and the product cost is reduced. However, all the acquisition harnesses will be connected to the main board, posing a greater challenge to the safety and compatibility of BMS, and there could be some stability issues from the daisy chain communication. (3) Communication method There are two methods of information transmission between the sampling chip and the main chip: CAN communication and daisy chain communication, of which the CAN communication is the most stable. High voltage source

Current sensor

High voltage source (415V)

Current Sensor

BMU_1

BMU_2 Power supply (12V)

BMS

Battery Module

Power supply (12V)

Battery Module

HBCU BMU_3

P-CAN P-CAN

Computer monitoring terminal Computer monitoring terminal

(a) Centralized architecture

Fig. 7.2 BMS structure

(b) Distributed architecture

BMU_4

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7 Electromagnetic Compatibility of Battery Management System

Battery equalization circuit

Electrical circuit

EEPROM

Battery voltage acquisition circuit

CAN communication module

CPU

Current acquisition circuit

Temperature acquisition circuit

Power drive circuit

Short circuit protection circuit

Fig. 7.3 BMS Hardware Structure

(4) Structure BMS is mainly divided into two parts, the first part is the front-end analog measurement circuit, including battery voltage conversion and measurement circuit, battery balance drive circuit, switch drive circuit, current measurement circuit, communication circuit; the second part is the back-end data processing module, which is based on the front-end calculation of voltage, current, temperature, etc., and sends the necessary information back to the system through the communication interface for control purposes. As shown in Fig. 7.4, BMS hardware includes power integrated circuit (IC), central processing unit (CPU), sampling IC, driving IC, short circuit protection IC, EEPROM, CAN module, etc. The CPU is the core component,and the sampling IC includes the acquisition of single voltage, module temperature and peripheral configuration of equalization circuit, etc (Fig. 7.3).

7.3 PCB Decoupling Capacitor Power Distribution Network The power integrity of the PCB of the BMS affects the electrical characteristics and electromagnetic compatibility of the controller. The high impedance of PCB power supply channel and the excessive noise of synchronous switch will lead to serious power integrity problems (such as voltage drop, ground bounce, electromagnetic radiation, heat generation and energy loss), which will have noticeable impact on devices and systems. The impedance and power integrity of all 5 V power supply networks of PCB chips are analyzed, the impedance characteristics of each 5 V port (100 KHz–1 GHz) are also analyzed. An optimization method for decoupling capacitor of PCB is proposed for ports exceeding the target impedance.

7.3 PCB Decoupling Capacitor Power Distribution Network

367

7.3.1 Optimal Design Method of Decoupling Capacitor (1) Impedance analysis of power distribution network on controller PCB The main chips on the PCB of BMS include: DC converter DCP010505BP, digital isolator ADUM1201, microprocessor chip XC2267, serial read-only memory AT24C1024 and multi-channel switch detection interface MC33975. The impedance simulation results of the 5 V power port of each chip are shown in Fig. 7.4. (2) Decoupling capacitor optimal design method Using DCP010505BP as an example, the optimization method of decoupling capacitor is introduced. Before the design of decoupling capacitor, the target impedance of the chip needs to be calculated. The maximum current flowing through DCP010505BP chip is not more than 200 mA, the power supply voltage is in between 4.5 and 5.5 V, and the chip’s requirement for voltage ripple is not more than 10%. A decoupling capacitor selection method based on the geometric average method of resonance points is adopted, which is based on the following principle: the self resonance frequency of decoupling capacitor is equal to the average frequency of two adjacent antiresonance points. Figure 7.5 is the impedance curve obtained by paralleling two different capacitors. It can be seen that there are two antiresonance points (M1 and M3) and one resonance point M2. It can be seen from Fig. 7.6 that the minimum impedance peak is achieved by selecting decoupling capacitor with the geometric average method of resonance points. (3) Design procedure of decoupling capacitor Figure 7.7 shows the impedance curve of 5 V power supply of DCP010505BP chip, which is roughly divided into three frequency bands, namely 1–410 MHz, 410–810 MHz and 810 MHz–1 GHz.

Fig. 7.4 Power supply network 5 V PDN frequency domain impedance curve

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7 Electromagnetic Compatibility of Battery Management System

Fig. 7.5 Typical parallel resonance impedance curve

Fig. 7.6 Comparison of 3 decoupling capacitor selection method

Fig. 7.7 DCP010505BP chip’s impedance curve

• 1–410 MHz frequency range The upper limit of parasitic inductance can be expressed as L max ≤

Z target 2π f

(7.1)

7.3 PCB Decoupling Capacitor Power Distribution Network

369

where f =410 MHz, Z target = 2.5 , therefore the upper limit of parasitic inductance is 0.97 nH. The series equivalent inductance of decoupling capacitor is 0.968 nH, and the installation inductance is about 0.3 nH. Thus, the total inductance of decoupling capacitor is 0.968 nH. In this frequency band, the resonant point is 390 MHz, the resonant points on both sides are 140 and 410 MHz respectively; the mean of these two values is 275 MHz. Therefore, the self resonance frequency of the decoupling capacitor to be added should be equal to 275 MHz. Then, the value of decoupling capacitance is calculated according to formula (7.3). f = C=

2π

1 √

LC

1 (2π f )2 L

(7.2) (7.3)

where f = 275 MHz, L = 9.68 × 10−10 H, the capacitance value obtained as C ≈ 346 p F. The value of decoupling capacitance in the frequency band of 1–410 MHz is obtained as 346 pF, the impedance curve is shown in Fig. 7.8a. The impedance curve of the 5 V port network after installing the decoupling capacitor is shown in Fig. 7.8b. • 410–810 MHz frequency range It is known that at f = 810 MHz, It can be calculated by formula (7.1) that the upper limit of parasitic inductance is 0.491 nH, so it is necessary to add several capacitors with same capacitance value and reduce the total inductance using the parallel effect of inductance. The number of decoupling capacitors is given by Eq. (7.4). The parasitic lead inductance is L = 0.968 nH, the upper limit of inductance is L MAX = 0.491 nH, therefore, it is calculated that N = 2. The total parasitic inductance of the final circuit is obtained as 4.84 × 10−10 H .

Fig. 7.8 a The impedance curve of 623 pF decoupling capacitor with 275 MHz self resonant frequency; b Impedance curve of 5 V port network with decoupling capacitor installed

370

7 Electromagnetic Compatibility of Battery Management System

Fig. 7.9 a Impedance curve of 116 pF decoupling capacitor with 670 MHz self resonance frequency; b Impedance cruve of 5 V port network after installing decoupling capacitors

 N=

L



L MAX

(7.4)

Using the geometric average method of resonance point, it is found that the resonance frequency of decoupling capacitor to be selected is 670 MHz, from formula 7.2 it is obtained that f = 670 MHz, L = 4.84 × 10−10 H, and the capacitance C ≈ 116 p F. Therefore, two 116 pF decoupling capacitors with self resonance frequency of 670 MHz are needed in the frequency band of 410–810 MHz. The impedance curve of 116pF decoupling capacitor with self resonance frequency of 670 MHz and the impedance curve of 5 V port network after installing decoupling capacitors are shown in Fig. 7.9a and b respectively. • 810 MHz–1 GHz frequency range It is known that f = 1 GHz, it can be calculated by formula (7.1) that the upper limit of parasitic inductance is 0.398 nH, number of deoupling capacitors is N = 3, and the total parasitic inductance is L = 3.23 × 10−10 H. The resonance frequency of decoupling capacitor is 905 MHz, and the capacitance C ≈ 96 p F. Finally, in the 810 MHz–1 GHz band, three 96 pF decoupling capacitors with self resonance frequency of 905 MHz are needed. The impedance curve of 96 pF decoupling capacitor with self resonance frequency of 905 MHz and the impedance curve of 5 V port network after installing decoupling capacitors are shown in Fig. 7.10a and b respectively.

7.3.2 Decoupling Capacitor Placement The placement of decoupling capacitor relative to the position of chip is also important. The distance between the decoupling capacitor and chip power-supply pin

7.3 PCB Decoupling Capacitor Power Distribution Network

371

Fig. 7.10 a Impedance curve of 96 pF decoupling capacitor with 905 MHz self resonance frequency; b Impedance cruve of 5 V port network after installing decoupling capacitors

shall not be greater than the decoupling radius of decoupling capacitor. The decoupling radius is related to the self resonance frequency of decoupling capacitor. The decoupling radius is given by Eq. 7.5. c

√ 1 λ εr · = lIC−capoptim (i) = 10 4 40 f sel f

(7.5)

where λ is the wavelength, c is the speed of light in vacuum, εr is the dielectric constant. Figure 7.11a shows the layout of decoupling capacitors and vias. Decoupling capacitor with high resonance frequency is placed near the chip power-supply pin in priority, other decoupling capacitors are placed around the chip power-supply pin according to the self-resonance frequency rank.

Fig. 7.11 Layout diagram of decoupling capacitor and optimized 5 V port impedance curve

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7 Electromagnetic Compatibility of Battery Management System

At the same time, in order to minimize the installation inductance, the via can be placed as close as possible to the decoupling capacitor pad. As the PCB is a fourlayer board, for reasonable use of space, three decoupling capacitors are placed on the top layer of the PCB, and the other three decoupling capacitors are placed on the bottom layer of the PCB, as shown in Fig. 7.11a. Figure 7.11b shows the optimized 5 V port impedance curve. It can be seen that the impedance of 5 V power port of DCP010505BP chip are well within the target impedance in full frequency band of 1 MHz–1 GHz after filtering with six decoupling capacitors. From the figure, it can be seen that the geometric average method of resonance point has better decoupling effect compared to the SRF = ARF method. Figure 7.11c shows the installation of decoupling capacitors on the PCB.

7.3.3 Analysis of Power Plane Resonance In a multi-layer board structure, PCB board has power plane and ground plane, due to impedance incompatibility on the ground plane, it is easy to form resonance point on the power plane. The simulation results of resonance frequency point and harmonic distribution of BMS PCB are shown in Figs. 7.12, 7.13, 7.14 and 7.15a. The more complicated four-layer board structure of PCB is made up by complete power plane and ground plane. On the circuit board, due to the existance of metal plane on PCB, electric wall structure is formed. This results in the inability of electromagnetic field energy to spread out effectively and thus, constantly reflecting within the board, leading to the occurance of resonance phenomenon. Resonance is mainly concentrated on the edge of PCB and some relatively isolated plane structures. Frequencies that could cause resonance easily are 859, 756, 691 and 642 MHz. In order to protect the data line and control signal line, methods of installing decoupling capacitor and constructing effective return path to ground, etc. are adopted. Figures 7.12, 7.13, 7.14 and 7.15b shows the simulation results after adding decoupling capacitor in the place with serious resonance, it can be seen that the resonance phenomenon is significantly reduced.

(a) Before installing decoupling capacitor

(b) After installing decoupling capacitor

Fig. 7.12 Simulation results of lower resonance distribution at 859 MHz

7.4 Clock Signal

(a) Before installing decoupling capacitor

373

(b) After installing decoupling capacitor

Fig. 7.13 Simulation results of lower resonance distribution at 756 MHz

(a) Before installing decoupling capacitor

(b) After installing decoupling capacitor

Fig. 7.14 Simulation results of lower resonance distribution at 691 MHz

(a) Before installing decoupling capacitor

(b) After installing decoupling capacitor

Fig. 7.15 Simulation results of lower resonance distribution at 642 MHz

7.4 Clock Signal The clock signal produced by crystal oscillator on PCB is one of the main interference sources. Good clock circuit design is the key to ensure PCB electromagnetic compatibility. Figure 7.16 is the result of testing the radiation emission of a bus

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7 Electromagnetic Compatibility of Battery Management System

Overshoot caused by crystal oscillator

Electric Field Strength (dBμV/m)

Radiate emission result ECE R10 standard

Frequency (MHz) Fig. 7.16 Radiation emission curve with standard exceeded by a bus due to crystal oscillator

according to the ECE R10 standard. It can be seen from the figure that some peak points generated by crystal oscillator are distributed at certain frequency intervals, resulting in the electric field strength of the bus exceeding the standard.

7.4.1 Electromagnetic Interference Mechanism (1) Clock signal spectrum Clock signal frequency is usually the working frequency required by a single chip microcomputer, which can be directly generated by a crystal oscillator, or generated by multiplying or dividing the crystal oscillator frequency with PLL circuit. Taking the Infineon 32-bit microcontroller TC1782F-320F-180 h applied on BMS as an example, the characteristics of EMI signal generated by clock circuit are analyzed. The clock period between the microcontroller and serial flash (SST25VF020B) is 5.5 ns, the magnitude is 3.3 V, the rise time/fall time is 0.33 ns, and the duty cycle is 50%. The clock signal can be regarded as a periodic trapezoidal wave pulse train, the frequency spectrum distribution is shown in Fig. 7.17, and the bandwidth is greater than 1 GHz. (2) Clock signal transmission path Due to the compact layout space, the line between serial flash and microcontroller is not connected in a straight line, but two corners are added. These two corners introduce an impedance mutation, which leads to the whole transmission line impedance being not continuous, causing signal reflection and signal distortion.

375

Magnitude (dBV)

7.4 Clock Signal

Frequency (MHz) Fig. 7.17 Single side magnitude spectrum of clock signal

As shown in Fig. 7.18 a T-type lumped parameter circuit, in which L represents the total inductance of microstrip line and C represents the total capacitance of microstrip line. The equivalent lumped parameter model of clock circuit between microcontroller and serial flash is shown in Fig. 7.19, in which the extra capacitive mutation at the corner is represented by C conner . Figure 7.20 shows the simulation results of input signal and output signal. It can be seen from the figure that ringing is most obviously observed at the end of rising edge of the clock output signal with Fig. 7.18 Transmission line equivalent lumped T-type circuit

L

L

2

2

C

Clock source

Load

(a) Without corner or capacit

Clock source

Load

(b) With corner or capacit

Fig. 7.19 Lumped clock circuit model between microcontroller and serial flash

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7 Electromagnetic Compatibility of Battery Management System

Time(s) Fig. 7.20 Simulation results of clock signal microstrip circuit

corner capacitor and the ringing improved significantly after the influence of corner capacitor is eliminated. In order to avoid the sudden change of impedance, the corner transition is usually introduced into the corner of PCB, as shown in Fig. 7.21, to reduce the capacitive sudden change at the corner. Figure 7.22 shows a 45◦ corner design in actual wiring. The discontinuous impedance structure in the clock circuit increases the possibility of clock interference coupling to other circuits, especially the high frequency component of the clock signal is most sensitive to the parasitic inductance and capacitance in the circuit, making it easy to be coupled to other circuits. The interference current generated by clock signal is also divided into differential mode current and common mode current. Figure 7.23 is a differential mode current coupling path and equivalent circuit, where lz and cz are the unit length inductance and unit length capacitance in the circuit model of microstrip transmission line distribution parameter in PCB. When the differential mode current encounters the impedance mismatch structure, the propagation path will change. It will be coupled to the power plane or ground plane Fig. 7.21 Schematic diagram of 45° corner transition on PCB

Main chip Pin

Main chip Routing

90° corner

45° corner

Off chip memory

Off chip memory

7.4 Clock Signal Fig. 7.22 45° corner design in actual wiring

377

Main chip

Routing

Memory chip

z is the length of the transmission line, c is the capacitance per unit length, and l is the inductance per unit length

Fig. 7.23 Differential mode current coupling path and equivalent circuit

through the parasitic capacitance on the PCB, forming a common mode current. It will then form an equivalent antenna through the PCB, chassis and harness, generating radiation interference. Figure 7.24 shows the propagation path and equivalent circuit of a common mode circuit.

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7 Electromagnetic Compatibility of Battery Management System

Fig. 7.24 Common mode current coupling path circuit diagram

7.4.2 Near Field Scanning Prediction The radiation interference generated by the clock or crystal oscillator high frequency signal inside the chip on the PCB can be analyzed through test. The test is divided into near-field electric field probe test and EMC scan test. Near field probe test: use the near field probe to test the electric field value of PCB circuit. Probe test location selection, location 1: ADUM5402 chip power output pin; location 2: PCB connector; location 3: PCB edge; location 4: microprocessor main chip power-supply pin. The test results are as shown in Fig. 7.25. The peak value of electric field intensity appears at 180 and 360 MHz at multiple test locations of PCB, among which the electric field intensity of location 1 has the highest peak magnitude, the highest peak point, and exhibits frequency doubling phenomenon.

Electric field intensity (dBμV/m)

7.4 Clock Signal

379

ADUM5402-Vdd

Frequency (MHz) Fig. 7.25 Near fied probe test result

EMC Scan is electromagnetic interference scanner, which can display the real time spatial distribution of PCB electromagnetic field, find out the interference source chip on PCB and determine the location of the interference source. Figure 7.26 shows the EMC Scan PCB test layout, Fig. 7.27 shows the spectrum scanning result. It can be seen that the interference frequency points on the PCB are concentrated around 175 and 360 MHz. This is consistent with the results of the near-field probe test. After determining the frequency point of the interference, in order to locate the interference source quickly, the spatial scanning test is carried out for the specific frequency point. The results of spatial scanning are shown in Figs. 7.28 and 7.29. From the results of spatial scanning of 175 MHz frequency points, it can be seen that the area with the largest electric field value corresponds to the area around the chip ADUM5402 on the PCB. Fig. 7.26 EMC scan test

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7 Electromagnetic Compatibility of Battery Management System

Fig. 7.27 EMC scan frequency spectrum result

Fig. 7.28 175 MHz frequency point EMC spatial scan result

ADUM5402 chip is an isolated DC-DC converter with combined iCoupler magnetic isolation technology and isoPower™ technolAogy. It demands the oscillator circuit to control the current flowing into the transformer at a switching frequency between 175 and 300 MHz. The switching frequency signal needs the crystal oscillator signal to provide 90 MHz through frequency doubling, and the rectifier on the secondary side will double the switching frequency in the rectification process. As shown in Fig. 7.30 is the functional block diagram of ADUM5402.

7.4.3 Clock Interference Signal Suppression Magnetic bead is a better choice for the circuit design of high frequency and low power clock signal EMI suppression on PCB. The bead attenuates the high frequency current in the circuit through heating. The overall filtering effect of the bead is related

7.4 Clock Signal

381

Fig. 7.29 360 MHz frequency point EMC spatial scan result

Fig. 7.30 ADUM5402 chip functional block diagram

to the material, frequency, current and bead size. The equivalent circuit of magnetic bead is a parallel circuit of resistor, inductor and capacitor, whose inductive reactance, capacitive reactance and impedance will change with the change of frequency. The principle of selecting magnetic beads for EMI is that the impedance of magnetic beads is the largest at EMI noise frequency. According to the characteristics of

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7 Electromagnetic Compatibility of Battery Management System

Fig. 7.31 Impedance characteristic curve of CBW201209U221T type magnetic bead

DC-DC isolation circuit with DC current of 1A and interference frequency of 100– 800 MHz, CBW201209U221T type bead 111 is selected to suppress high frequency noise. The impedance characteristic curve of magnetic bead is shown in Fig. 7.31. According to the characteristic curve in Fig. 7.31, the equivalent parallel inductance and parallel capacitance of the magnetic bead can be obtained by using formula (7.6). 1

= 150 Ω 2π f 90M H z C 1 2π f 280M H z L − = 0Ω 2π f 280M H z C

2π f 90M H z L −

(7.6)

It is calculated that C ≈ 11 pF, L ≈ 96 nH, R ≈ 200 Ω (100 MHz − 800 MHz). Figure 7.32 shows the circuit diagram of common mode current path added with magnetic bead, and the two ports network model of this model is established on the ADS software, as shown in Fig. 7.33. The simulation results of insertion loss are shown in Fig. 7.34. It can be seen that the insertion loss is significantly reduced at 175 MHz after adding magnetic beads, indicating that the magnetic beads can restrain the interference near this frequency band.

7.4.4 BMS System Electromagnetic Radiation Emission Test Figure 7.35 is a BMS system radiation emission test platform setup in accordance with GB18655-2018. The test results are shown in Fig. 7.36. It can be seen that the electric field intensity test results of BMS system with EMI filter do not exceed the level 3 radiation emission limit of GB 18655, but radiation emission peak values can

7.4 Clock Signal

383

Fig. 7.32 Equivalent circuit of common mode interference with magnetic beads

Fig. 7.33 Circuit model built on ADS software

S21(dB)

Relatively high insertion loss near 175MHz

Frequency (MHz) Fig. 7.34 Comparison of S21 before and after adding EMI filter

With EMI filter Without EMI filter

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7 Electromagnetic Compatibility of Battery Management System

(a) 150k~30MHz monopole antenna

(b) 30M~200MHz biconical antenna

(c) 200M~1000MHz log periodic antenna

(d) 1000M~2500MHz horn antenna

Fig. 7.35 BMS radiation emission test

Frequency (MHz)

PK AV 18655 PK 18655 AV

Magnitude (dBμV/m)

Magnitude (dBμV/m)

PK AV 18655 PK 18655 AV

Frequency (MHz)

PK is the peak value, AV is the average value, 18655PK is the GB18655 peak limit, 18655AV is the GB18655 average limit (a) Vertical direction

(b) Horizontal Direction

Fig. 7.36 Radiation emission test result

be seen at 175, 360, 540 and 720 MHz. It can be seen from the above analysis that these spikes are caused by the clock signal and its related circuits.

7.5 DC-DC Chip Switching Noise

385

7.5 DC-DC Chip Switching Noise The DC-DC power conversion chip on PCB contains MOSFET (metal oxide semiconductor field effect transistor) switching devices, which will generate voltage and current jump and high frequency noise during the opening and closing process of MOSFET.

7.5.1 Switching Noise Mechanism (1) Electromagnetic interference source The DC-DC power conversion chip of BMS converts 24 V of vehicle battery into 5 V, providing power for various chips on PCB. Figure 7.37 shows a circuit structure diagram of commonly used DC-DC chip. This is a simple Buck switch converter, which is widely used in Buck DC-DC circuit. The MOSFET of a Buck converter is the main device of EMI. Figures 7.38 and 7.39 show the drive signal and output signal of MOSFET device and their spectrum distribution, respectively. It can be seen that the magnitude of the high frequency part of the output voltage signal of MOSFET increases a lot compared with the driving signal. In addition, the parasitic parameters of MOSFET will not only affect the switching characteristics of the actual devices, but also lead to the generation of ringing. There are many kinds of parasitic inductors in the return path of MOSFET, such as the excitation winding of isolation transformer, PCB lead inductor and so on. They will form a resonance circuit with the interelectrode capacitance of MOSFET and produce high frequency oscillation. (2) DC-DC electromagnetic interference propagation path (1) Differential mode interference coupling path The differential mode current is a circuit that starts from the positive pole of DC-DC chip and flows back to the negative pole along the normal electrical circuit Fig. 7.37 Circuit schematic diagram of DC-DC chip

Vcc

Vref Vfb

EA

Comp

L

PWM driver

Gnd

Vout

R1 R2 Gnd Gnd

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7 Electromagnetic Compatibility of Battery Management System

Drain Ld

Gate R g

Cgd

Cds

Rds

Cgs

Drive signal

Switch oscillation

Drain output voltage

Ls Source Fig. 7.38 MOSFET circuit model

Magnitude (dBμV)

Drive signal Drive voltage

Frequency (MHz) Fig. 7.39 Spectrum comparison between MOSFET driving signal waveform and drain voltage waveform

through the load. The differential mode current includes normal working current as well as unexpected high-frequency interference current. The propagation path of a typical differential mode interference current is shown in Fig. 7.40. There are two ways to reduce the differential mode interference current: one is to reduce the value of differential mode interference current, the other is to reduce the loop area. The first method can be realized by increasing the rise/fall time of MOSFET driving signal and reducing the switching frequency. The second method

7.5 DC-DC Chip Switching Noise

387

Shell

Equivalent loop antenna

Load

Idm

Interference source

Radiation

Bypass capacitor DC-DC

PCB

GND

Marginal radiation

VCC

Fig. 7.40 DC-DC propagation path of differential mode interference

is to reduce the loop area by placing the positive and negative power lines of DCDC chip, other positive and negative power lines of PCB, or external positive and negative power lines as close as possible, or by adding bypass capacitors. (2) Common mode interference coupling path The common mode current is usually transmitted through the parasitic capacitance between the conductor or interconnector and the ground plane or vehicle shell, Fig. 7.41 shows a typical common mode interference current coupling path. There are two ways to reduce the common mode interference current: one is to reduce the interference current value, the other is to reduce the wire length. For the first

Shell

Load Interference source

DC-DC

Icm1

PCB GND VCC

Icm2

Equivalent loop antenna

Radiation

Fig. 7.41 DC-DC propagation path of common mode interference

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7 Electromagnetic Compatibility of Battery Management System

method, in addition to increasing the rise/fall time of the MOSFET driving signal and reducing the switching frequency, a magnetic ring or bead can also be selected to suppress the common mode current. For the second method, minimize the wire length during routing. (3) Electromagnetic radiation In the PCB of BMS, unintentional antenna is an important part of DC-DC interference propagation path. One part of these antennas is composed of common wires, the other part may be the shell, ground plane and its extension, PCB board and other conductive bodies. Due to the low input impedance of these antennas, the external electric field radiation is generated under the driving of sometimes differential mode or common mode current interference source.

7.5.2 Electromagnetic Interference Model Prediction (1) EMI modeling of DC-DC converter On Matlab/Simulink, build the Buck circuit model as shown in Fig. 7.42. The model includes power circuit, drive circuit, feedback circuit and measurement circuit. The working current waveform, gate drive signal waveform, common mode current

Fig. 7.42 DC-DC converter model

7.5 DC-DC Chip Switching Noise

389

waveform and gate-source voltage waveform of MOSFET are obtained through simulation, as shown in Fig. 7.43. It can be seen from the voltage and current waveforms that when MOSFET is turned on and off, there will be violent voltage and current jumps, forming EMI interference source. Figure 7.44a and b show the positive and negative voltage waveforms of DC-DC chip power supply output respectively. It can be seen that there will be an interference peak in the voltage waveform of DC-DC output for each interval, which is caused by the opening/closing of MOSFET. The spectrum distribution of the voltage waveform output by DC-DC is shown in Fig. 7.45.

Fig. 7.43 Simulation results of MOSFET in DC-DC converter

Negative voltage of DC-DC converter

Voltage (V)

Voltage (V)

Positive voltage of DC-DC converter

Time (s) ˄a˅Positive voltage waveform

Fig. 7.44 DC-DC converter output voltage waveform

Time (s) ˄b˅Negative voltage waveform

7 Electromagnetic Compatibility of Battery Management System

Magnitude (dBμV)

390

Frequency (MHz) Fig. 7.45 DC-DC converter output voltage waveform

7.5.3 DC-DC Interference Suppression Method The power of DC-DC chip on PCB is lower than 6 W. In view of the relatively small power of DC-DC chip but strict space requirements, the LC EMI filter is designed as shown in Fig. 7.46. Among them, the package size of chip beads and chip capacitors is small, which is suitable for circuit board level design. The interference of DCDC chip is mainly concentrated in the frequency band below 100 MHz. Therefore, CBW322513U601T magnetic beads with impedance characteristic curve covering this frequency band is selected. The impedance characteristic curve is shown in Fig. 7.47, and the impedance between 20 and 200 MHz can reach over 300 ohms. Therefore, through calculations it is obtained that the equivalent parallel inductance L ≈ 2.09 µH, the equivalent parallel capacitance C ≈ 4.85 pF and the equivalent parallel resistance of the magnetic beads R ≈ 300 Ω. DC-DC

EMI filter Vcc

Vref Vea

L Comp

Vfb

PWM driver

Ferrite bead

Ramp

Gnd

Lf

R1

Rf

Load Cbypass

Cf

R2

Vs

Cb

VL

Gnd Gnd

Fig. 7.46 Schematic diagram of DC-DC circuit interference suppression circuit and equivalent circuit of EMI filter

7.5 DC-DC Chip Switching Noise

391

X

Z is the impedance curve, R is the DC resistance curve, X is the reactance curve Fig. 7.47 CBW322513U601T bead impedance characteristic curve

The typical characteristic index of filter is insertion loss (IL), which is usually expressed in dB. If VL ,w/o is used for the load voltage before inserting the filter and VL ,w/ is used for the load voltage after inserting the filter, then the insertion loss of EMI filter is defined as: ⎛ 2 ⎞ VL ,w/o     PL ,w/o VL ,w/o R I L d B = 10 log10 = 10 log10 ⎝ V 2 L ⎠ = 20 log10 (7.7) L ,w/ PL ,w/ VL ,w/ RL

where VL ,w/o = VL ,w/ =

RL VS RL + RS

RL 1 jωCb + Rs + R + L

(7.8) VS

1

(7.9)

1 1 1 Rf + jωCf + jωL f

Then the insertion loss of the EMI filter can be expressed as:

I Ld B

⎞ ⎛     RL ⎟ ⎜ V ⎟ ⎜ R L +R S S = 20 log10 ⎜ ⎟ RL ⎠ ⎝ V 1 1 S  jωCb + Rs + R +  1 L   1 + jωC + 1 f jωL R f

(7.10)

f

where in a typical configuration, R L = R S = 50 ,C ≈ 4.85 pF, L ≈ 2.09 µH, R f = 300 Ω, ω = 2π f, f = 30 MHz are taken from a bypass capacitor with self resonance frequency of 30 MHz, then according to the selection method of decoupling capacitance Cb = 40nF, it is obtained that I L d B = 17.5473 dB.

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7 Electromagnetic Compatibility of Battery Management System

7.6 Anti-interference Analysis of CAN Bus CAN (Controller Area Network) is one of the most widely used open field buses in the world. As a kind of remote network communication control mode with strong anti-electromagnetic interference ability, high reliability, fast communication speed, perfect function, simple structure, flexible networking, supporting tree structure and reasonable cost, CAN bus has been widely used in the field of automobile control. The electromagnetic environment of electric vehicle is complex, which is easy to interfere with CAN bus, the issue of wrong frame occurs (several hundred frames per second when the motor is working), and the BMS could crash under the limit condition, all of which brings new challenges to the design of CAN bus. CAN bus circuit as an important information transmission path on the vehicle could affect the driving experience if the key signal is interfered lightly, and if interfered seriously, could lead to catastrophic consequences, for example, sudden trigger of airbag due to interference, which is why the design of CAN bus anti-interference circuit is crucial.

7.6.1 Electromagnetic Interference of CAN Bus Circuit Conducted disturbance generated by high voltage components such as motor drive system and vehicle high-low voltage DC-DC converter could be coupled to the CAN bus through cables, which causes interference to CAN bus signals. Interference source equipment, such as automotive power failure simulator, subminiature automotive transient simulator, voltage change simulator and so on, can produce the common interference waveforms on the simulated automobile. By injecting the interference signals generated by these equipments into the BMS CAN bus, the anti-interference ability of the CAN bus could be tested. As shown in Fig. 7.48 is a test where the CAN bus is interfered while the driving motor is running under the rated working condition. The signal on the CAN bus can

Fig. 7.48 CAN bus high current injection test

7.6 Anti-interference Analysis of CAN Bus

393

be observed using an oscilloscope. It is seen that during the process of injecting large current into CAN harness, at d = 150 mm, 450 mm and 750 mm, the fluctuation of speed and torque is within ± 10%. (d is the distance between the high current injection probe and the motor controller)

7.6.2 Design of Anti-interference Circuit of CAN Bus As shown in Fig. 7.49 is the design schematic diagram of a CAN bus hardware circuit. In the figure, the CAN bus transceiver PCA82C250 produced by Philips is used as an example to design the anti-interference circuit. The function of common mode choke in the anti-interference circuit is to suppress common mode current, while the function of TVS (transient voltage suppression) diode is to prevent surge pulse. Using the SigXplorer simulation design tool provided by Cadence, the transmission-line model is established to simulate the reflection characteristics of CANL and CANH signals. Figure 7.50 shows the SigXplorer simulation model of CAN bus network. The direct cause of reflection of CANL and CANH signal lines is the impedance mismatch between the two ends, which causes the end terminal to feedback part of the voltage to the source terminal. The simulation results of CANL and CANH signal waveforms are shown in Fig. 7.51b. It can be seen that the impedance mismatch of CAN bus circuit causes the CANH signal waveform to not reach the normal level

Fig. 7.49 CAN bus hardware circuit design

Fig. 7.50 SigXplorer CAN bus network simulation model

394

7 Electromagnetic Compatibility of Battery Management System CANH source CANH receiver CANL source CANL receiver

Vcc

TXD

Slope/ standby

RXD Vref

Driver CANH Receiver

CANL

Voltage (V)

Rs

Protection

Reference voltage GND

Time (μs)

(a)

(b)

Fig. 7.51 a CAN transceiver PCA82C250T chip functional structure diagram; b CANL and CANH signal waveform

value. After adjusting the matching resistance, the simulation results as shown in Fig. 7.52 are obtained. The TVS diode verification circuit model as shown in Fig. 7.53 is established on the ADS software. The VTPulse in the figure imitates the load rejection interference waveform on the vehicle. The TVS diode has the function of voltage clamping. When Fig. 7.52 Signal waveform of CANL and CANH under impedance mismatch

Voltage (V)

CAN receiver R=62ohm CAN receiver R=30ohm CAN receiver R=5ohm

Time (μs)

Fig. 7.53 Transient suppression diode circuit model

7.6 Anti-interference Analysis of CAN Bus

395

the load rejection interference voltage exceeds the clamp voltage of the TVS diode, the TVS diode will conduct and maintain the voltage at the clamp voltage to protect the circuit from overvoltage, as shown in Fig. 7.54 (Figs. 7.55 and 7.56). The following CAN bus circuit anti-interference design method is adopted, and the rectification measures on PCB are shown in Fig. 7.57. • Opto-isolator: In order to enhance the anti-interference ability of each node of CAN bus, CAN interface card with photoelectric isolation, CAN/485 conversion module and CAN repeater are selected to electrically isolate the terminal equipment of the network, prevent the influence of different reference ground potential of terminal equipment on other terminal equipment, prevent interference from entering the control circuit and ensure the safety of the control system. The photoelectric coupling has relatively high common mode rejection and strong antiinterference ability. The use of photoelectric coupling can also prevent the strong interference pulse induced by the transmission line from entering the terminal equipment, such as lightning strike. It should be noted that if DC-DC isolated power supply is used, different power supply must be used for the isolation side and the isolated side, otherwise, the isolation would not work.

Without TVS diode With TVS diode

Voltage (V)

Fig. 7.54 Simulation result of voltage clamping of transient voltage suppression diode

Time (ms) Fig. 7.55 Anti-interference optimization design of CAN bus circuit

TVS diode

DC-DC isolator Common mode inductor

Opto isolator

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7 Electromagnetic Compatibility of Battery Management System

(a) 100KHz

(b) 5.6MHz

(c) 13MHz

(d) 64MHz

Fig. 7.57 Simulation results of far field electric field of 5 V power supply network

Magnitude (dBμV/m)

Fig. 7.56 Near field magnetic field distribution under different frequency excitation signals of 5 V power supply network

Frequency (MHz)

• Shielded twisted pair: The twisted pair has the characteristics of mutual winding, which makes the induced magnetic field of the two wires cancel each other. This method is very effective for electromagnetic interference shielding with frequency less than 30 MHz; an aluminium foil shielding layer has better shielding effect for electromagnetic interference with frequency higher than 30 MHz. It must be emphasized that all shielding layers must be well grounded, because the electromagnetic field formed by the large current and arc spark generated by the power bus and motor can cause strong static electricity on the shielding layer through induction or radiation, which will result in potential instability on the shielding wire and thus affects the shielding effect of the shielding wire.

7.6 Anti-interference Analysis of CAN Bus

•

•

•

•

•

397

Therefore, the shielding layer of twisted pair needs to be 360° grounded at both ends. Filter capacitor: A small filter capacitor can be connected in parallel between the CANL terminal, CANH terminal and the ground respectively. The capacitance value is generally several tens of pF to filter out the high-frequency interference on the bus and prevent electromagnetic radiation. Transient voltage suppression diode: A TVS (transient voltage suppression) diode is connected in parallel between the CANL and CANH terminals, which can effectively avoid the impact of load rejection and various surge pulses on the circuit. When there is a high voltage on the CAN bus, TVS is immediately activated to clamp the voltage on the bus at a predetermined value, and discharge the pulse current on the CAN bus to the ground plane, effectively protecting the precision components in the circuit from the damage of various surge pulses. Common mode choke and bead: Common mode choke is a high common mode noise suppression element. As the common mode choke wraps the conductor around the core in a special way, the flux generated by the high frequency differential mode current will cancel each other in the core and will not saturate it. Therefore, the common mode choke can effectively filter the high frequency common mode electromagnetic interference signal on the CAN bus. The magnetic bead has the characteristics of low inductance in low frequency and high impedance in high frequency. It is very effective to use the magnetic bead as a low-pass filter to reduce electromagnetic interference. Matched resistor: Adding matched resistors at the end of CAN bus can match the impedance at both ends of the circuit, prevent wave reflection, reduce interference and improve line communication quality. With the development of intelligent vehicles, more and more control units are added to the vehicle CAN bus, which leads to the increase of bus branches and the complexity of the structure. By adjusting the number of matched resistors in the bus, the impedance of the bus can be adjusted, interference can be reduced, and signal quality can be improved. However, the number of matching resistors should not be too much, as the increase of the number of terminal resistors will result in dropping of bus differential voltage and significant drop in the anti-interference ability. Reduce the baud rate reasonably: The decrease of baud rate can lengthen the time of each CAN bit, lengthen the sampling time of CAN waveform, and miss some electromagnetic interference signals. It is worth noting that the baud rate reduction must be implemented on the premise of satisfying the rapid response of the system.

7.7 PCB Electromagnetic Radiation By importing the PCB of BMS into the SIwave software, considering the factors of PCB geometry, material properties, and spatial position, the simulation can get a realistic electromagnetic field distribution. The voltage source is set at the output

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7 Electromagnetic Compatibility of Battery Management System

pin of the DC-DC chip power supply and the two ends of the clock circuit as the excitation signal. The near-field magnetic field distribution and the far-field electric field frequency distribution under the influence of these two interference signals are simulated and predicted. Figure 7.56 is the simulation result of near field magnetic field distribution under different frequency excitation signals of 5 V power supply network, Fig. 7.57 is the simulation results of far field electric field of 5 V power supply network. It can be seen from the simulation results that the place with the largest magnitude of electric field and magnetic field is located at the corner of the routing line, the signal reflection is caused by the sudden change of impedance at the corner, resulting in the increase of the magnitude of electric field and magnetic field. The simulation results of near field magnetic field distribution of clock circuit under different frequency current excitation signals are shown in Fig. 7.58. The simulation results of far field electric field intensity of PCB clock circuit electromagnetic radiation are shown in Fig. 7.59.

(a) 50MHz

(b) 160MHz

(d) 426MHz

(c) 232MHz

(e) 578MHz

(f) 960MHz

Fig. 7.59 Simulation results of far field electric field of clock circuit

Magnitude (dBµV/m)

Fig. 7.58 Near field magnetic field distribution under different frequency excitation signals of clock circuit

Far field radiation

Frequency (MHz)

Chapter 8

Measurement, Diagnosis and Suppression of Vehicle Electromagnetic Radiation

8.1 Overview Electric vehicles use a lot of electronic equipment different from traditional internal combustion engine driven vehicles, which puts forward new requirements for electromagnetic radiation of the vehicle. As the electromagnetic environment of electric vehicles becomes increasingly complex, the international EMC regulations, as well as the EMC performance index requirements of the vehicle and parts are getting more and more stringent. The national standard GB/T 18387-2008 “Limits and Test Method of Electric and magnetic field intensity from Electric Vehicles, Broadband, 9 kHz to 30 MHz” and GB/T 18387-2017 “Limits and test method of magnetic and electric field intensity from electric vehicles (150 kHz–30 MHz)” stipulated the electric field intensity and magnetic field intensity of electric vehicles under typical operating conditions. GB 14023—2011/CISPR12 “Vehicle Radio disturbance characteristics—Limits and methods of measurement for the protection of off-board receivers” specified the radiation emission limits for vehicles driven by internal combustion engines, electric drives or hybrids electric vehicles. The GB/T 18387-2017 and GB 14023-2011 specified test items are compulsory testing items for China’s new energy vehicles. If the EMC of a vehicle cannot meet the requirements of relevant laws and regulations, it will not be available on the market. Therefore, manufacturers are required to invest a lot of money and manpower to carry out EMC rectification and product re-development. In addition, international standards, such as SAE J551-5 and ECE R10 also stipulate limit requirements on the vehicle emission. Standard limit exceeding of electric vehicle radiation emission will affect the normal operation of the receiving equipment of surrounding vehicles, infrastructure and living environment. With the development of intelligent networking and autonomous vehicles, the electromagnetic radiation of electric vehicles faces new challenges. For example, standard limit exceeding vehicle radiation may interfere the wireless sensor operation of the vehicle, affecting the expected run safety of the vehicle, and even causing safety accidents.

© China Machine Press 2021 L. Zhai, Electromagnetic Compatibility of Electric Vehicle, Key Technologies on New Energy Vehicles, https://doi.org/10.1007/978-981-33-6165-2_8

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The EMC of vehicle components is the basis and premise of vehicle EMC, especially the EMC performance of HV and high-power conversion equipment of electric vehicles, they are not only related to their own working reliability, but also affect the safe operation ability and working reliability of the whole vehicle (Lin 2019). The standard GB/T 18655-2018 (CISPR 25-2016) “Vehicles Radio disturbance characteristics—Limits and methods of measurement for the protection of on-board receivers” specifies the test methods, test procedures and limits of HV power supply system with internal shielding for electric and hybrid electric vehicles, which act as preventive control of vehicle radiation emission. However, the component parts testing cannot represent the vehicle testing, in fact, the exact relationship between the two depends on the installation position of the parts, the length and layout of the wiring harness, the grounding position and the antenna position (Yang 2017). By using measuring instruments and equipment according to the characteristics of electromagnetic emission signal of components, it is possible to diagnose the components and elements that may cause potential vehicle radiation emission exceeding the standard limit in advance. This chapter focuses on the electromagnetic radiation test methods of passenger cars and commercial vehicles; combined with the whole vehicle radiation emission test of commercial vehicles, the whole vehicle radiated electromagnetic interference suppression method is proposed; and using the measurement method, the diagnosis method of electromagnetic interference that causes the electromagnetic radiation emission of the whole vehicle exceeds the standard is proposed, which provides the test basis for the EMC design and rectification of subsequent vehicles.

8.2 Electromagnetic Radiation of Passenger Car 8.2.1 Vehicle Radiation Emission Test GB/T18387 specifies the limits and measurement methods of electromagnetic field emission strength in the range of 9 kHz–30 MHz for electric vehicles. A rod antenna and a loop antenna are both placed at a distance of 3 m or 10 m in the front, rear, left and right side of the electric vehicle to test the electric field intensity and magnetic field intensity generated by the electric vehicle respectively. Using the loop antenna to test the magnetic field intensity as an example, the test arrangement and results are shown in Fig. 8.1.

8.2 Electromagnetic Radiation of Passenger Car

Rear side-X direction

Rear side-Y direction

Rear side-Z direction

Left side-X direction

Fig. 8.1 Magnetic field test results

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8 Measurement, Diagnosis and Suppression of Vehicle …

Left side-Y direction

Left side-Z direction

Right side-X direction

Right side-Y direction

Fig. 8.1 (continued)

8.2 Electromagnetic Radiation of Passenger Car

403

Right side-Z direction

Front side-X direction

Front side-Y direction

Fig. 8.1 (continued)

8.2.2 Vehicle Radiation Emission Suppression It can be seen from Fig. 8.1 that the measured value of magnetic field intensity in X direction on the left side of the vehicle exceeds the standard most, the exceeding frequency points are 59, 78.6, 104 kHz and 2.1 MHz, as shown in Fig. 8.2. By disconnecting the vehicle electronic components one by one, it is found that the amplitude of magnetic field intensity reduced significantly when the motor drive system stops functioning, indicating that the over-standard frequency points in Fig. 8.2 are caused by the work of the motor drive system. The exceeding standard points of magnetic field intensity in low frequency band are 59, 78.6 and 104 kHz, which are mainly

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8 Measurement, Diagnosis and Suppression of Vehicle …

Fig. 8.2 Test results of radiated emission magnetic field intensity of EV—left side X direction of the vehicle

distributed between 10 and 150 kHz. Due to the low frequency, they cannot be caused by crystal oscillator and high-speed chip, but mainly caused by the highspeed switching of the motor inverter power module IGBT. For the high frequency band (2.1 MHz), the EMI is mainly caused by the MOSFET switch in the DC-DC. (1) Installation of class-Y capacitor in motor controller If the interference at 59 kHz frequency point is reduced below the limit value, it is necessary to insert two class-Y capacitors to ground at the input port of the HV positive and negative power lines of the motor controller, as shown in Fig. 8.3a. This is so that the insertion loss I L d B is greater than 11 dB, which meets the standard limit, the calculated capacitance value is 0.14µF. (2) Installation of class-Y capacitor and common mode choke on LV power board The magnetic field intensity of the vehicle exceeds the standard at 2.1 MHz. The common mode choke is mainly caused by the high-speed switching of MOSFET of

(a) HV DC power line with Class-Y capacitor (b) LV power supply board with Class-X capacitor and common mode choke

Fig. 8.3 Installation of suppression link at DC input of motor drive control unit

8.2 Electromagnetic Radiation of Passenger Car

(a) Cable connector

(b) Box gap

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(c) Cable shield grounded to car body

Fig. 8.4 Shielding and grounding measures of motor controller

DC-DC module of LV power board, so it is necessary to insert class-X capacitors into the power line input port of LV power board to suppress EMI. If the interference amplitude at 2.1 MHz is reduced below the limit, the insertion loss is greater than 1.9 dB, and calculated capacitance value is 0.37nF. In order to effectively suppress the common mode interference, a common mode choke is added between the two class-x capacitors of the power line of the LV power board, and the calculated inductance value is 14.4µH. Class-y capacitor and common mode choke are installed on the LV power board inside the motor controller, as shown in Fig. 8.3b. (3) Grounding of the motor controller shield In order to reduce the electromagnetic leakage of the motor controller, shielding and grounding measures are taken at the plug connector, box gap, cable shielding layer and vehicle body, as shown in Fig. 8.4. By adopting the above-mentioned filtering, shielding and grounding measures, the measured value of electromagnetic emission magnetic field intensity of the vehicle is reduced in the frequency band of 9 kHz–30 MHz, no standard exceeding point was observed, all test results meet the standard limit requirements, as shown in Fig. 8.5. In particular, the first figure shows the magnetic field intensity measured in the leftside X direction of the vehicle, the amplitude of the magnetic field intensity greatly reduced after radiation emission rectification.

8.3 Electromagnetic Radiation of Commercial Vehicles 8.3.1 Vehicle Electromagnetic Radiation Test (1) 9 kHz–30 MHz electromagnetic radiation Standard GB/T 18387 is used to measure the electromagnetic field intensity of the whole vehicle in the frequency band 9 kHz–30 MHz, the layout of electric field and magnetic field test is shown in Fig. 8.6. When the vehicle speed is 16 and 64 km/h,

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8 Measurement, Diagnosis and Suppression of Vehicle …

(a) Left side - X, Y, Z directions

(b) Right side – X, Y, Z directions

(c) Rear side – X, Y, Z directions

Fig. 8.5 Test results of magnetic field intensity after vehicle radiation emission rectification

Fig. 8.6 Photo of electric field test (left) and photo of magnetic field test (right)

the electric field intensity test results of the whole vehicle are shown in Fig. 8.7. The over-standard points are mainly concentrated in the vicinity of 13 MHz and the 20–30 MHz frequency band, the amplitude of electric field intensity does not change obviously with vehicle speed. The results of the vehicle magnetic field intensity test are shown in Fig. 8.8. It can be seen that the magnetic field intensity value exceeds

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Electric field intensity (dBμV/m/kHz)

Electric field intensity (dBμV/m/kHz)

8.3 Electromagnetic Radiation of Commercial Vehicles

Frequency(Hz) Peak value

GB/T18387 Limits

Frequency(Hz)

Peak value

(a) Vehicle speed 16km/h

GB/T18387 Limits

(b) Vehicle speed 64km/h

Magnetic field intensity (dBµA/m/kHz)

Magnetic fieldfield intensity (dBµA/m/kHz) Magnetic intensity (dBµV/m)

Fig. 8.7 Electric field intensity test results of the whole vehicle

Frequency(Hz)

Frequency(Hz) Peak value

GB/T18387 Limits

Peak value

GB/T18387 Limits

(a) The loop antenna is arranged in the X direction, the vehicle speed is 16km/h (b) The loop antenna is arranged in the X direction, the vehicle speed is 64km/h

Fig. 8.8 Test results of the magnetic field intensity of the vehicle

the standard at 64 km/h, and the magnetic field intensity amplitude increases with the increase of vehicle speed. (2) 30 MHz–1 GHz electromagnetic radiation The standard GB 14023-2011 is used to measure the electric field intensity of the whole vehicle in the frequency range of 30–1000 MHz. The vehicle operating state during the broadband radiation emission test is: The vehicle speed is set to 40 km/h to run at a constant speed; low beam, hazard warning lights, and front fog lights work; front wipers work at maximum speed; radio works; air conditioning works (cooling, low temperature, maximum wind speed, internal circulation, wind direction Head and feet); the power window is in the middle position; the driver’s seat position is fixed. The measurement results are shown in Fig. 8.9. When the drive motor system is working in the 30–40 MHz frequency band, the vehicle has relatively large external radiation, which exceeds the quasi-peak limit but does not exceed the peak limit. Compared with the limit value between the measured value of the electric field intensity in the frequency band around 100 MHz and around 900 MHz and the limit

8 Measurement, Diagnosis and Suppression of Vehicle …

Amplitude(dB

)

Electric field intensity (dBµV/m)

Electric field intensity (dBµV/m)

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Frequency(Hz) Peak value-3M method

Peak value Quasi peak value-3M method

Frequency(Hz) Peak value

Peak value-3M method

Quasi peak value-3M method

(a) Electric field (left side of the vehicle, antenna horizontal polarization) (b) Electric field (left side of the vehicle, antenna vertical polarization)

Fig. 8.9 Vehicle electric field intensity test −30 MHz–1000 MHz

value between the quasi-peak value, the margin is smaller, but the margin between the peak limit value is larger.

8.3.2 Diagnosis of Electromagnetic Disturbance Source Aiming at the problem of excessive electric field radiation, a spectrum analyzer and near-field probe are used to locate and diagnose the disturbance source. The troubleshooting ideas are as follows: First determine the main troubleshooting order: Drive motor, high voltage components and wiring harness, low voltage components and wiring harness. (1) The impact of the operation of the drive motor on the radiation emission of the vehicle The antenna position is facing the rear of the vehicle. Under the following two conditions, test the electric field intensity of 9 kHz–30 MHz. Situation 1: When the vehicle is powered on with high voltage and the drive motor does not run, the launch test result is the blue curve shown in Fig. 8.10; Situation 2: The vehicle is powered on with high voltage and the motor runs at low speed. The launch test result is the orange curve shown in Fig. 8.10; From Fig. 8.10, it can be seen that the electric field intensity of the motor is significantly increased at low frequencies below 150 kHz, especially in the 15, 30, and 45 kHz frequency bands, which are mainly affected by the IGBT carrier frequency of the motor controller. The actual carrier frequency is 7 kHz. In the 10– 15 MHz frequency band, the electric field intensity increases significantly when the motor is running, and exceeds the standard limit. The electric field intensity value of 15–30 MHz frequency band has little change, which indicates that the excessive

Fig. 8.10 Radiation emission when the HV power supply is stationary and the vehicle is running

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Electricfield fieldintensity intensity(dBμV/m/kHz) (dBµV/m) Electric

8.3 Electromagnetic Radiation of Commercial Vehicles

25.16MHz,29.7 dBµV/m 20.1MHz , 29.2 dBµV/m

No speed

GB/T18387

With speed

Frequency(Hz)

frequency band of 20–30 MHz has little to do with whether the driving motor operates or not Frequency(Hz) (2) The impact of HV system power-on on vehicle radiation emission

Fig. 8.11 Radiation emission of the whole vehicle when the HV power failure occurs

Electric Electricfield fieldintensity intensity(dBμV/m/kHz) (dBµV/m)

Unplug the fast circuit breaker and cut off the HV power supply. Other test arrangements remain unchanged. The radiation emission test results are shown in Fig. 8.11. Compared with Fig. 8.10, the overall electric field intensity amplitude decreases significantly after the HV power supply is cut off. The frequency band 22–26 MHz still exceeds the standard. It shows that HV components have a greater impact on radiation exceeding the standard. And 22–26 MHz still has the phenomenon of exceeding the standard, indicating that this is mainly caused by the work of low-voltage components.

24.87MHz, 26.4dBµV/m/kHz

Peak value

GB/T18387

Frequency(Hz)

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(3) Diagnosis of main disturbance sources in high voltage system In order to further query and locate the main harassment source components, a combination of a portable spectrum analyzer and a near-field probe is used for diagnosis and investigation. (1) First, disconnect the HV fast circuit breaker, turn on only low-voltage power, and turn the vehicle to the ON position to measure the background noise of the rear compartment. Then, scan the following positions with the electric field probe in turn: Vehicle controller peripherals and wiring harness ports; motor controller and signal wiring harness ports; HV fast circuit breaker input and output terminals; HV DC bus positive and negative output lines; wiring harness near DC-DC. The electric field intensity test results are shown in Fig. 8.12. Among them, the measured result near DC-DC is not much different from the background noise. The signal wiring harness of the motor controller and the input end of the fast circuit breaker have the largest near-field radiation, and the other places are relatively small. (2) Close the fast HV circuit breaker, turn the vehicle on to the ON position, and use the electric field probe to scan the position basically the same as the previous step. The measurement position of the HV DC bus positive output line and the wiring harness beside the left rear column are added, and the electric field intensity test results are shown in Fig. 8.13. It can be seen that the near-field electric field intensity of the input and output terminals of the fast circuit breaker is the highest after the HV power on state, the measured value (relative value) of 20–30 MHz is above 40 dBµV, and the highest value is 50 dBµV at 24 MHz.

(a) Vehicle controller and connecting wire harness (b) Motor controller signal wire harness port

(c) HV fast circuit breaker input and output (d) HV DC bus positive output line Fig. 8.12 Near-field scanning result of electric field intensity-no high voltage

8.3 Electromagnetic Radiation of Commercial Vehicles

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(a) Vehicle controller and wiring harness (b) Motor controller and wiring harness diagram (c) Input and output of HV fast circuit breaker

(d) High voltage DC bus positive output line (e) Harness near DC-DC (f) Harness near left rear column

Fig. 8.13 Near-field scanning result of electric field intensity-with HV

(4) Low-voltage equipment interference source diagnosis Disconnect the fast breaker of the HV loop, use a spectrum analyzer and a near-field probe to test the electric field intensity. (1) The quick breaker is disconnected, the key is ON, and the measurement position: the low-voltage harness end of the motor controller, the measured value is shown in Fig. 8.14. (2) The quick breaker is disconnected, the low-voltage wiring harness of the motor controller is fully disconnected, and the switch is turned to ON. Measure the position of the motor controller low-voltage port, the measured value is very small, close to the background noise, the measurement results are shown in Fig. 8.15, it can be seen that the low voltage harness of motor controller will lead to vehicle radiation emission.

Fig. 8.14 Near-field scanning and electric field intensity test results of the control line of the HV power controller

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Fig. 8.15 Near-field scanning and electric field intensity test results after disconnecting the LV line

(3) In the state of step (2), connect the motor resolver signal harness, the measurement position remains unchanged, and the measurement result is close to the background noise. All other positions are measured very small. The measurement situation is shown in Fig. 8.16. It can be seen that the resolver line has no effect on the radiation emission. (4) On the basis of step (2), connect the low-voltage control wiring harness (signal control line) of the motor controller, the measurement position remains unchanged, and the measurement result rises obviously, indicating that the source of the disturbance comes from the low-voltage control module inside the motor controller, as shown in Fig. 8.17. (5) After all the wiring is restored, only disconnect the control harness of the motor controller. First, the near-field probe test, all test positions are very small, equivalent to background noise. Then use the pole antenna to test the electric field intensity of the whole vehicle outside the vehicle with a distance of 3 m. The test results are shown in Fig. 8.18. It can be seen that the electric field intensity is significantly reduced. Based on the above steps 1 to 5, it can be concluded that the source of disturbance mainly comes from the inside of the motor controller, and the propagation path is the low-voltage control wiring harness of the motor controller.

Fig. 8.16 Near-field scanning and electric field intensity test results of the resolver connecting line

8.3 Electromagnetic Radiation of Commercial Vehicles

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Fig. 8.18 Comparison of electric field intensity between on and off motor controller signal harness under power on state

ElectricAmplitude(dBµV/m) field intensity (dBμV/m/kHz)

Fig. 8.17 Near-field scanning and electric field intensity test results after disconnecting the control line

Connection

No Connection

GB/T18387 Limits

Frequency(Hz)

8.3.3 Vehicle Radiated Electromagnetic Interference Suppression (1) EMC design of the vehicle The design method to improve the EMC performance of the vehicle is as follows: (1)

(2)

(3) (4)

The HV wiring harness of the whole vehicle adopts shielded wiring harness, and the shielding layer needs to be well grounded. It is recommended to use single-ended grounding; The motor controller, two-in-one and five-in-one, and the HV connector and locking device of the HV control box also use shielding components and are well grounded; The shielding layer of the HV line should be well grounded, and the singleended or double-ended grounding should be done. The shielding layer of the CAN wire of the whole vehicle should be well grounded, and the single-ended or double-ended ground should be done.

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(5)

The resolver wire shielding layer of the motor controller should be well grounded, and single-ended or double-ended grounding should be done. (6) The motor controller and the five-in-one HV controller box need to be well grounded, and the gaps and reserved HV ports need to be electromagnetically shielded and sealed. (7) Optimize the layout of the whole vehicle wiring harness, reduce the electromagnetic interference coupling path, reduce the unnecessary wiring harness for actual work, and minimize the wiring harness length and loop area on the premise of meeting the electrical performance. (8) Require component suppliers to carry out EMC design of HV and low-voltage components, and do the corresponding filtering, shielding and grounding inside the controller, for example, PCB optimization design, circuit topology optimization design, adding magnetic loops and filter capacitors inside the controller. (9) Keep low-voltage wiring harnesses as far away as possible from HV wiring harnesses to avoid mutual coupling and interference between wires. (10) To ensure that the ground impedance of each HV component is as small as possible, it is recommended to use a short and wide wire or a woven bag for the grounding. (11) The area of the HV loop of each HV component (motor inverter, DC-DC, etc.) (whether external wiring harness or internal wiring of the product) is as small as possible. (2) Electromagnetic radiation suppression methods for electric buses In terms of the installation of the whole vehicle system, the EMC design and rectification of the layout and installation of the HV electrical system were first carried out. First, the original controller was replaced with a HV power integrated controller using a new filter, and the HV connecting harness was replaced with a HV shielded harness. For the wiring harness interface, a locking head with 360-degree shielding is used, and other HV components are also added with independent reinforced grounding. Replace the grounding wire with copper braided tape, as shown in Fig. 8.19 [12].

Replace with die casting parts

DC accessories

360 ° shielding is adopted at both ends of shielding harness

Integrated motor controller

Drive motor

Shielding is used for high voltage wiring harness AC accessories

Strengthen grounding

Fig. 8.19 Schematic diagram of rectification items for HV electrical system

8.3 Electromagnetic Radiation of Commercial Vehicles

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Fig. 8.20 Schematic diagram of 360° shield locking head

(a) Controller implement 360 shielding

(b) Strengthen controller grounding

Fig. 8.21 Rectification diagram of component installation

Replacing the die-casting controller can effectively control the external emission of the electromagnetic field generated by the internal circuit of the controller, but the HV wiring harness connected to it needs to be shielded. At the same time, a 360 degree locking head is used to achieve the integrity of the shield, and the shield is extended to the motor and AC and DC electrical accessories. The structure of the locking head is shown in Fig. 8.20. Implement integrated shielding for the drive motor system and other electrical accessories, and perform reliable grounding to reduce the loop area and reduce external emissions. The main implementation measures are shown in Fig. 8.21. (3) Experimental verification After the above-mentioned EMC rectification, the HV power integrated controller has basically reached the level 3 limit of the standard CISPR 25. On the basis of improving the electromagnetic compatibility of the components, the GB/T18387 standard was adopted to test the electromagnetic field radiation of the vehicle. Before the early rectification of the car, the electric field and magnetic field intensity were tested, and the exceeding condition was serious. The limit was exceeded at 30–150, 200, 250, 300 kHz, 4, 8 and 26 MHz. The HV power integrated controller equipped with a new filter is replaced and tested. The results are shown in Fig. 8.22. As shown in Fig. 8.22a, purple-red is the result of the magnetic field intensity test on the left front of the vehicle before the rectification, and blue is the test result after the rectification. The vehicle is a front-mounted vehicle, and the HV power integrated controller and motor are all arranged on the front side, so the field intensity test results of the

8 Measurement, Diagnosis and Suppression of Vehicle … Electric field intensity (dBµV/m/kHz)

Magnetic field intensity (dBµA/m/kHz)

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Frequency After rectification

Frequency

GB/T18387 Limits After rectification

Before rectification

GB/T18387 Limits

(a) Vehicle front left, magnetic field antenna X direction (b) Vehicle front position, electric field strength

Fig. 8.22 Test results of electromagnetic field intensity before and after the rectification of the test model

front left, front right, and front of the vehicle are higher than those at the rear. After shielding and grounding the high and low voltage wiring harness of the high voltage power integrated controller, it can be found that the external radiation of the electric field and magnetic field of the vehicle in the range of 150 kHz to 30 MHz is greatly reduced. Among them, the electric field and magnetic field intensity caused by the IGBT switch of the HV power integrated controller are significantly reduced, and the amplitude attenuation is nearly 20 dB; the resonance point in the high-frequency area has been greatly improved after the shielded wire and the 360-degree shielded locking head are used, which is below the limit and the margin is more than 6 dB. The margin statistics of test results are shown in Table 8.1. From the test results in Table 8.1, it can be concluded that through the improvement of the main interference source of the HV power integrated controller, combined with the improvement of the layout, installation, and connection of the whole vehicle, the electromagnetic compatibility performance of the whole vehicle has been greatly improved, and the rectification plan effectively solves the problem of low-frequency magnetic field and high-frequency electric and magnetic field Table 8.1 GB/T18387-2008 Test Margin Statistics Table Test situation

Frequency (MHz) Limit

Measured

Front left magnetic field X 7.17

−13.11 dBµA/m/kHz −21.59 dBµA/m/kHz

Front left magnetic field Y 0.139

24.63 dBµA/m/kHz

Front left magnetic field Z

7.17

−13.11 dBµA/m/kHz −25.69 dBµA/m/kHz

Front left electric field

7.17

38.29 dBµV/m/kHz

17.74 dBµA/m/kHz 28.52 dBµV/m/kHz

8.4 EMI Diagnosis Method of Whole Vehicle

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8.4 EMI Diagnosis Method of Whole Vehicle Taking the electromagnetic radiation emission of an electric bus as an example, through the following diagnostic steps, the EMI source equipment responding for radiation emission exceeding the standard limit is found and located, which provides the test basis for the next rectification and design for EMI. Step 1 Vehicle broadband radiation emission test—10 m -left rear side of vehicle According to standard GB/T14023 (ECE R10), the broadband radiation emission of the left rear side of the electric vehicle in the frequency range of 30–1000 MHz is tested in the 10 m anechoic chamber. Since the motor drive system is located at the rear of the vehicle, the radiation emission of the rear part of the vehicle is mainly tested. The picture of the test site is shown in Fig. 8.23a. This step is used to diagnose the broadband radiation emission of the whole vehicle under two working conditions: all the electrical equipment in the vehicle are switched on or off. Test condition: vehicle speed 40 km/h. Antenna position: the logarithmic antenna is aligned to the left rear side of the vehicle and vertically polarized. Detection mode: peak detection mode is used first, and then quasi-peak detection mode is used for points exceeding the standard limit. The test results are shown in Fig. 8.23b, c, and the radiation emission comparison under the two conditions is shown in Fig. 8.23d. It can be seen from Fig. 8.23b that there is a point exceeding the standard limit at 38.9 MHz for the peak value of electric field intensity, and there are several points exceeding the standard limits for the quasi-peak value in the frequency band of 30– 110 MHz. It can be concluded that the peak value point exceeding the standard limit is caused by the operation of electrical equipment in the vehicle. Step 2 Vehicle narrowband radiation emission test—10 m—left rear side of vehicle Then, the narrowband radiation emission of the left rear side of the electric bus is tested in the frequency band of 30–1000 MHz. This step is used to diagnose the narrowband radiation emission of the whole vehicle under the working and no working conditions of high and low voltage DC-DC. Test condition: the vehicle is stationary (the motor is not running), and other electrical equipment are switched on. Detection mode: average detection mode Antenna position: the logarithmic antenna is aligned to the left rear side of the vehicle and vertically polarized The test results are shown in Fig. 8.24a, b, and the radiation emission comparison under the two conditions of high and low voltage DC-DC is shown in Fig. 8.24c. It can be seen from the test results that when DC-DC works, the affected frequency band is 30–157 MHz, and there are some points obviously exceeding the standard

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Field intensity (dBμV/m) Amplitude(dBµV/m)

Amplitude(dBµV/m) Field intensity (dBμV/m)

(a) The picture of the test site-left rear side of vehicle

Frequency(Hz)

Frequency(Hz)

(b) All electrical equipment switched on

(c) All electrical equipment switched off

Switch on Switch off

Frequency(Hz) (d) Comparison results

Fig. 8.23 Broadband radiation emission test-10 m-left rear side of electric bus

limits at 38.7, 44, 50, 64.8, 72 and 106 MHz. The scanning results of portable spectrum analyzer and near-field probe are in good agreement with those measured by antenna, especially the peak distribution trend of 36.7, 44, 64.8, 72, 106, 172, 208, 307 MHz is consistent. During the test, filter capacitance is added between the positive and negative poles of DC-DC output terminal, and the DC-DC harness is shielded and grounded. It can be seen from Fig. 8.24d that the radiation emission suppression effect is not obvious. It can be concluded that the average values exceeding the standard limits are mainly caused by the operation of high and low voltage DC-DC. The filter capacitor is added between the positive and negative poles of the DC-DC output terminal, and the DC-DC harness is shielded and grounded. It can be seen from Fig. 8.24d that the radiation emission suppression effect is not obvious. It can be concluded that the average values exceeding the standard limits are mainly caused by the operation of high and low voltage DC-DC.

8.4 EMI Diagnosis Method of Whole Vehicle

(a) DC-DC switched on

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(b) DC-DC switched off

With DC-DC No DC-DC Near field test

(c) Comparison results With capacitance grounding No capacitance grounding Near field test

(d) Adding capacitance and grounding

Fig. 8.24 Narrowband radiation emission test-10 m- left rear side of electric bus

Step 3 Vehicle narrowband radiation emission test—10 m -left front side of vehicle The narrowband radiation emission of the left rear side of the electric bus is tested in the frequency band of 30–1000 MHz, as shown in Fig. 8.25a. In addition to the DC-DC converter, this step is used to diagnose the vehicle narrowband radiation emission exceeding the standard limits caused by the operation of other electronic equipment. Test condition: the vehicle is stationary (the motor is not running), and other electrical equipment are switched on. Detection mode: average detection mode

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Field intensity (dBμV/m)

Field intensity (dBμV/m)

(a) The picture of the test site- left front of vehicle

(a) horizontal polarization

(c) vertical polarization

Fig. 8.25 Narrowband radiation emission test-10 m-left front of electric bus

Antenna position: the logarithmic antenna is aligned to the left rear side of the vehicle, horizontal polarization and vertical polarization are used. Step 3.1: the antenna vertical polarization and horizontal polarization average detection method is used to test the electric field intensity in the front left of the vehicle. When the DC-DC output cable is pulled out, the battery is used to supply power to other low-voltage electrical equipment. The test results of electric field intensity when other electrical equipment is turned on are shown in Fig. 8.23b, c. It can be seen that the average values have multiple points exceeding the standard limits, and the test result of antenna vertical polarization is more serious. Step 3.2: when the vehicle is stationary (the motor is not running), the antenna vertical polarization average value detection method is used to test the electric field intensity in the front left side of the vehicle, the DC-DC output cables and BMS cables are disconnected, the instrument and centralized lubrication electronic control equipment are turned on, and the vehicle radiation emission when backup camera system is working or not working. The test results of average values of electric field intensity are shown in Fig. 8.26. It can be seen that there are some points exceeding the standard limits at 211.6, 231.8 and 297.1 MHz, when the backup camera system is working. Step 3.3: when the vehicle is stationary (the motor is not running), the antenna vertical polarization average value detection method is used to test the electric field intensity in the front left of the vehicle, the DC-DC output cable and BMS cable are disconnected, the instrument, centralized lubrication equipment and road sign electronic control equipment are closed, and the vehicle radiation emission when the

8.4 EMI Diagnosis Method of Whole Vehicle

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With backup camera No backup camera

Fig. 8.26 Comparison results of radiation emission, when backup camera system is working or not working—left front of electric bus

monitoring host computer is working or not working is tested. The test results are shown in Fig. 8.27. It can be seen that the average value of electric field intensity exceeds the standard limit at 65.4 MHz, when the monitoring host computer is working. Step 3.4: when the vehicle is stationary (the motor is not running), the antenna vertical polarization average value detection method is used to test the electric field intensity in the front left of the vehicle, the DC-DC output cable and BMS cable are disconnected, the instrument, centralized lubrication equipment and road sign electronic control equipment are closed, and the vehicle radiation emission when the stop screen is working or not working is tested. The test results are shown in Fig. 8.28. It can be seen that there are three points of the average values of electric field intensity exceeding the standard limits at 37, 46 and 65 MHz respectively, when the stop screen is working.

With host computer No host computer

Fig. 8.27 Comparison results of radiation emission, when monitoring host computer is switched on or off—left front of electric bus

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With stop screen No stop screen

Fig. 8.28 Comparison results of radiation emission, when stop screen is switched on or off—left front of electric bus

Step 3.5: when the vehicle is stationary (the motor is not running), the electric field intensity in the front left side of the vehicle is tested by the average value detection method with antenna vertical polarization, and the DC-DC converter, instrument, centralized lubrication equipment, road sign electric control equipment and backup camera equipment are turned off, and the vehicle radiation emission when the BMS is working or not working is tested. From the test results as shown in Fig. 8.29, It can be seen that the radiation emission of the whole vehicle is not significantly affected by BMS after EMC rectification. Step 3.6: when the vehicle is stationary (the motor is not running), the electric field intensity in the front left side of the vehicle is tested by the average value detection method with antenna vertical polarization. When the key is on, DC-DC is turned off, the instrument, centralized lubrication equipment, driving recorder and backup camera are turned on, and the radiation emission of the whole vehicle is tested when the station announcer is working or not. The test results are shown in Fig. 8.30. It can

With BMS No BMS Near field test

Fig. 8.29 Comparison results of radiation emission before and after removing BMS output cable— left front of vehicle

8.4 EMI Diagnosis Method of Whole Vehicle

423 With station announcer No station announcer Near field test

Fig. 8.30 Comparison results of radiation emission, when station announcer is switched on or off—left front of electric bus

be seen that the average electric field intensity exceeds the standard limit at 126 MHz when the station announcer is working. Step 3.7: when the vehicle is stationary (the motor is not running), the electric field intensity in the front left side of the vehicle is tested by the average value detection method with antenna vertical polarization. When the key is on, DC-DC is turned off, and the instrument, centralized lubrication equipment, backup camera and station announcer are turned on, then the radiation emission of the whole vehicle is tested when the driving recorder is working or not. The test results are shown in Fig. 8.31. It can be seen that the radiation emission of the whole vehicle is not significantly affected by the operation of driving recorder. Step 3.8: when the vehicle is stationary (the motor is not running), the electric field intensity in the front left side of the vehicle is tested by the average value detection method with antenna vertical polarization. When the key is on, DC-DC is turned off, and the instrument, backup camera and station announcer are turned

With driving recorder No driving recorder

Fig. 8.31 Comparison results of radiation emission, when driving recorder is switched on or off— left front of electric bus

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With centralized lubrication No centralized lubrication

Fig. 8.32 Comparison results of radiation emission, when centralized lubrication is switched on or off—left front of electric bus

on, then the radiation emission of the whole vehicle is tested when the centralized lubrication equipment is working or not. The test results are shown in Fig. 8.32. It can be seen that the average value of electric field intensity decreases by 10 dB and 6 dB respectively at 72.9 and 102.8 MHz when the centralized lubricating electronic equipment is working. Step 4 Vehicle narrowband radiation emission test—10 m -right of vehicle The narrowband radiation emission of the right side of the electric bus is tested in the frequency band of 30–1000 MHz. This step is used to diagnose the vehicle narrowband radiation emission exceeding the standard limits caused by the operation of the electronic equipment located on the right side of the vehicle. Test condition: the vehicle is stationary (the motor is not running), and other electrical equipment are switched on. Detection mode: average detection mode Antenna position: the logarithmic antenna is aligned to the right rear side of the vehicle, vertical polarization is used. Step 4.1: the antenna is used to test the electric field intensity at the right rear of the vehicle. The test results are shown in Fig. 8.33. It can be seen that there are points of the average values of electric field intensity exceeding the standard limits at 34, 42, 46, 54, 66, 89, 105.7, 112.8, 266 and 308 MHz. Step 4.1: the electric field intensity at right front of the vehicle is tested. The test results are shown in Fig. 8.34. It can be seen that there are points of the average values of electric field intensity exceeding the standard limits at 94, 112, 126, 226 and 826 MHz. It can be seen from the above radiation emission test results of the right side of the vehicle that the test results are the same as those of the left side, only a few points exceeding the standard limits are different.

8.4 EMI Diagnosis Method of Whole Vehicle

425 Right rear

Fig. 8.33 Broadband radiated emission test results of electric bus—right rear side of vehicle Right front

Fig. 8.34 Broadband radiated emission test results of electric bus—right front side of vehicle

Step 5 Determine the interference source equipment According to the above diagnosis steps, the interference source equipment causing the vehicle electric field intensity exceeding the standard is obtained, as shown in Table 8.2.

8.5 EMI Measurement on Real Vehicle 8.5.1 Frequency Domain Characteristics of EMI (1) Frequency domain characteristics of conduction current in motor drive system A portable spectrum analyzer is used to test the conducted disturbance current of the motor drive system in frequency domain. The layout diagram is shown in Fig. 8.35.

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Table 8.2 Interference source equipment Left front (MHz)

Interference source

Right front (MHz)

Interference source

Left rear (MHz)

Interference source

Right rear (MHz)

Interference source

37

Stop screen

94

Backup camera system or 24 V power supply for motor controller

38.7

DC-DC

34

DC-DC

46.8

Stop screen

226

Backup camera system or 24 V power supply for motor controller

44

DC-DC

42

DC-DC

65

Monitoring host computer stop screen

50

DC-DC

46

DC-DC

72.9

Centralized lubrication equipment

64.8

DC-DC

54

DC-DC

102.8

Centralized lubrication equipment

72

DC-DC

66

DC-DC

126

Station announcer

106

DC-DC

89

DC-DC

211

Backup camera system

105.7

DC-DC

230.5

Backup camera system

112.8

DC-DC

297.1

Backup camera system

266

DC-DC

308

DC-DC

The current probe is used to clamp the HV DC power lines and AC three-phase power lines respectively to measure the amplitude frequency characteristic curve of conducted disturbance current. The field measurement layout of the real vehicle is shown in Fig. 8.36. The test results of conducted current of HV DC power lines are shown in Fig. 8.37. It can be seen from Fig. 8.37 that under three conditions of acceleration, deceleration and uniform speed, the spectrum waveform distribution

8.5 EMI Measurement on Real Vehicle

427

Fig. 8.35 Test setup of conducted current in motor drive system

(a)

Positive power line

(b) Negative power line

(c) AC power line

Current (dBμA)

Current (dBμA)

Fig. 8.36 Measurement of conducted current of DC bus and AC three-phase line with current probe

(a) Positive power line

(b) Negative power line

Fig. 8.37 Conducted current spectrum of DC positive power line

of conducted current flowing on positive and negative HV DC power lines is basically the same, and the conduction current amplitude under deceleration condition is slightly reduced. The comparison of conducted current between positive and negative power lines of DC HV bus is shown in Fig. 8.38. It is found that the amplitude of current on DC positive power line in the whole frequency band (150 kHz–30 MHz) is 3–5 dB larger than that on negative DC power line. In this frequency band, the maximum conducted current appears on the positive DC bus cable with an amplitude of 70 dbµA and a frequency of 1 MHz. This measurement is consistent with the conducted current results of the motor drive system tested in the laboratory as described in Sect. 3.6.

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Fig. 8.38 The comparison of conducted current between positive and negative power lines of DC HV bus

Fig. 8.39 Comparison of conducted current of AC three-phase power line under acceleration condition

The disturbance current on three-phase AC power line is measured on the real vehicle, the test results are shown in Fig. 8.39. It can be seen from Fig. 8.39 that under the three working conditions of acceleration, deceleration and uniform speed, the current spectrum waveforms of the three AC three-phase power lines are basically the same, and the peaks appear at the frequency points of 2.4, 10.2 and 20.3 MHz, and the maximum peak appears at 2.4 MHz, with the value of 57 dBµA. (2) Frequency domain characteristics of conducted disturbance current of DCDC converter The conducted current of DC-DC converter is tested in frequency domain by using spectrum analyzer. The layout diagram is shown in Fig. 8.40a. Figure 8.40b, c are the layout pictures of real vehicle measurement. The conducted disturbance current of low-voltage positive power line and high-voltage positive and negative power lines are tested respectively. The test results are shown in Figs. 8.41 and 8.42. It can be seen that under the three working conditions of acceleration, deceleration and

8.5 EMI Measurement on Real Vehicle

a

b

429

Test setup of conducted current

LV positive power line

c

HV negative power lines

Fig. 8.40 Test setup of conducted current in DC-DC converter system

Fig. 8.41 Test results of frequency domain conducted current on low voltage power line

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Fig. 8.42 Test results of frequency domain conducted current on HV power line

uniform speed, the current spectrum waveform on the low-voltage positive power line is basically consistent with that on the high-voltage positive power line. The peak values of the current on the low voltage positive power line appear at 700 kHz, 1.1, 1.8, 2.2, 6, 8 and 11 MHz respectively, and the peak values of the current on the high-voltage positive and negative power line appear at 1.9, 4, 11 and 21 MHz, and the current value on the high-voltage positive pole power line is significantly greater than that on the low-voltage positive power line, and the maximum peak value appears at about 1.9 MHz, with the value of 61dBuA. (3) Frequency domain characteristics of common mode current on CAN bus VCU, BMS and motor control system are connected by CAN bus for communication. EMI of CAN bus may cause electromagnetic radiation of the whole vehicle. The test arrangement and test results of common mode current on CAN bus are shown in Fig. 8.43. There are four curves in the diagram, which are the current spectrum of the bottom noise current and three working conditions of acceleration, deceleration and uniform speed. It can be found that the interference current on the CAN bus is affected by the working conditions. The current amplitude under acceleration condition is

Fig. 8.43 The test arrangement and test results of common mode current on CAN bus

8.5 EMI Measurement on Real Vehicle

431

slightly higher in some frequency bands, but the current spectrum changes of CAN bus under acceleration, deceleration and uniform speed are consistent. The common mode current on CAN bus is larger in the range of 0.6–2 MHz, the peak values appear at 0.7 and 1.7 MHz, and the peak value is 63dBuA at 1.7 MHz. (4) Frequency domain characteristics of common mode current in low voltage power line The DC-DC converter converts the high-voltage direct current of power battery into low-voltage direct current, and supplies power to VCU, BMS and other LV equipment through LV lines. In its working process, EMI generated by DC-DC converter and low-voltage equipment is easily coupled to the LV lines, which may cause electromagnetic radiation of the vehicle. The CM current on LV lines is usually measured to estimate its electromagnetic emission. The CM current test layout and test results of LV lines are shown in Fig. 8.44. The current value on the LV lines is relatively large before 2 MHz, and there is a peak value about 0.7 MHz, and its amplitude is about 69dBuA. (5) Frequency domain characteristics of common mode current on Resolver line The resolver line is used to feed back the speed and position information of the motor to the motor controller to realize the closed-loop control of rotor position and speed. There are abundant pulse waves on the resolver line. In addition, the interference signals on the three-phase power line of the motor are easily coupled to the line, which may cause electromagnetic radiation of the whole vehicle. The test arrangement and test results of CM current on resolver line are shown in Fig. 8.45. The current amplitude of motor resolver line is smaller than that of low voltage power line and CAN bus. In addition, the bottom noise is large and discrete distribution, which has an impact on the measurement results.

Fig. 8.44 The CM current test layout and test results of LV lines

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Fig. 8.45 The test arrangement and test results of CM current on resolver line

(6) Frequency domain characteristics of electromagnetic field ➀ Motor drive system The electromagnetic field intensity on the surface of DC HV and LV power lines of the motor system is measured. The measurement results are shown in Figs. 8.46 and 8.47. The measured frequency range is from 30 MHz to 2.5 GHz. With peak detection method, it can be seen that the electromagnetic field intensity on HV line is significantly greater than that on LV line, and the electromagnetic field intensity in X, y and Z directions is slightly different. In addition, the electromagnetic field on the surface of DC bus and three-phase power line in the motor system is measured, and the X, y and Z directions of each line are measured. The test results are shown in Figs. 8.48 and 8.49. The magnetic field intensity of DC bus and three-phase line in X direction is compared, as shown in Fig. 8.50.

Fig. 8.46 Peak electromagnetic field intensity of HV positive and negative lines

8.5 EMI Measurement on Real Vehicle

Fig. 8.47 Peak electromagnetic field intensity of LV positive and negative lines

Fig. 8.48 Electric field and magnetic field intensity of HV DC lines

Fig. 8.49 Electric and magnetic field intensity of three-phase AC line

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Fig. 8.50 Comparison of magnetic field intensity in X direction between DC bus and three-phase AC line

➁ DC-DC converter system Figure 8.51 shows the test layout of the magnetic field on the surface of DC-DC high and low voltage power lines. Among them, the magnetic field intensity of the low-voltage power line in the X-direction and Z-direction is significantly larger than that in the Y-direction, and the magnetic field intensity of the high-voltage power line is the largest in the X-direction. The comparison results of the three are shown in Fig. 8.52. It can be seen that the magnetic field intensity of high voltage power line is greater than that of low voltage power line in the whole test frequency band, and the difference is about 40 dBmW. Figure 8.53 shows the field layout of electric field test, and the corresponding test results are shown in Fig. 8.54. It can be seen from the test results that the electric field intensity of the same high-voltage power line is greater than that of the low-voltage power line in the whole test frequency band, and the difference is also about 40 dBmW.

(a) X direction of LV power line (b) Z direction of LV power line (c) X direction of HV power line

Fig. 8.51 Test setup of magnetic field of power lines of DC-DC

8.5 EMI Measurement on Real Vehicle

435

Fig. 8.52 Comparison of magnetic field intensity of DC-DC power line

(a) Test setup of electric field of LV power line

(b) Test setup of electric field of HV power line

Fig. 8.53 Test setup of electric field of power lines of DC-DC

Fig. 8.54 Comparison of electric field intensity of DC-DC power line

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Fig. 8.55 Schematic diagram of test layout of time domain current and voltage for DC-DC converter

From the above test results, it can be seen that the peak electromagnetic field intensity of low-voltage positive power line will appear at 700 kHz, 1.1, 1.8, 2.2, 6, 8 and 11 MHz, and the peak electromagnetic field intensity of high-voltage positive and negative power lines will appear at 1.9, 4, 11 and 21 MHz. The frequency-domain electromagnetic field intensity of high-voltage positive and negative power lines is significantly greater than that of low-voltage positive power lines. Therefore, the electromagnetic environment of DC-DC high-voltage side is more severe than that of low-voltage side.

8.5.2 Time Domain Characteristics of EMI (1) Influence of high voltage on low voltage side The time domain voltage and current of DC-DC converter are tested by oscilloscope, current probe and voltage probe. The actual layout and test results are shown in Figs. 8.55 and 8.56 respectively. During the high voltage is turned on, the lowvoltage output current, differential voltage, positive to ground voltage and negative to ground voltage measured by the low-voltage power line are shown in Fig. 8.57. (2) Influence of driving conditions on high pressure side Figure 8.58 shows the voltage of the high-voltage positive power line to the ground measured at the high-voltage side. In the process of high voltage turning on and vehicle running. It can be seen from Figs. 8.57 and 8.58 that there is a rapid transient change in the voltage and current of the low-voltage power line and the voltage of the high-voltage power line to the ground during the high-voltage turning on process, and there are obvious voltage changes during the start-up and acceleration and deceleration.

8.5 EMI Measurement on Real Vehicle

437

(a) Current test of LV positive supply line

(b) Differential voltage of LV power line and positive to ground voltage test

(c) Differential voltage test of HV power line

(d) Voltage test between HV positive power line and ground

Fig. 8.56 Test setup of time domain current and voltage test for DC-DC converter

Fig. 8.57 Test results of LV power line

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(a) Voltage between HV positive and ground - starting acceleration

(b) voltage between HV positive and ground - variable working condition

Fig. 8.58 Voltage test results of HV side

References Lin C (2019) Handbook of electric—vehicle design of pure electric vehicle. China Machine Press, Beijing Yang S (2017) Research on electromagnetic interference and suppression method of high voltage power integrated controller for pure electric bus. Beijing Institute of Technology, Beijing